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This is not an exhaustive collection of circuits, but a compendium of preferred ones. Where appropriate, suggested
part numbers and component values are given. Where added
components may be needed for stability, they are shown.
Experienced designers may elect to omit these components
in some applications, but less seasoned practitioners will be
able to put together a working circuit free from the frustration of how to make it stable.
This application guide is intended as a source book for the
design and application of:
Current sources
Current sinks
Floating current sources
Voltage-to-current converters
(transconductance amplifiers)
Current-to-current converters (current mirrors)
Current-to-voltage converters
(transimpedance amplifiers)
The applications shown are intended to inspire the imagination of designers who will move beyond the scope of this
R. Mark Stitt (602) 746-7445
BIDIRECTIONAL CURRENT SOURCES .................................... 16
BIDIRECTIONAL CURRENT SOURCES .................................... 16
Current Source or Sink With Compliance to
Power Supply Rail and Current Out >100µA .......................... 4
BIDIRECTIONAL CURRENT SOURCES .................................... 17
Current Source or Sink With Any Current Out ...................... 5
Floating Current Source With Current Out >100µA ............... 5
SLEW RATE LIMITER ................................................................ 18
Current Sources made with Voltage References ................... 5
Current Sources and Sinks and Current Mirrors
Using an Amplifier and a Series Pass Element .................... 6
Floating Current Source With
Current Out >100µA and No Separate Power Supply ........... 6
CURRENT SOURCE USING INA105 .......................................... 18
Cascoding with FETs ................................................................ 8
200µA Floating Cascoded Current Source Using REF200 ... 9
300µA Floating Cascoded Current Source Using REF200 . 10
400µA Floating Cascoded Current Source Using REF200 . 10
NOISE REDUCTION OF CURRENT SOURCES ........................ 10
MONITORING USING THE INA105 OR THE INA117 ............... 22
ADJUSTMENT USING 5mV REFERENCE ................................. 12
WITH 0 to 5V OUTPUT USES THE RCV420 ............................. 23
FLOATING VOLTAGE REFERENCE .......................................... 13
TO VOLTAGE CONVERTER ....................................................... 23
PHOTODIODE AMPLIFIERS ....................................................... 23
GLOSSARY ............................................................. 27
1989 Burr-Brown Corporation
Printed in U.S.A. March, 1990
Current mirror Q1– Q2 forces equal currents to flow in 8/1
emitter ratioed devices Q7 and Q8. The proportional to
absolute temperature (PTAT) voltage difference between
the emitters—(k • t/q) • ln(8)—is forced across the 4kΩ
resistor resulting in a PTAT current of about 13µA. Because
Q10 matches Q7, and Q3 matches Q4, equal PTAT currents
flow in each of the four Q1 – Q4 legs. The current in the Q4
leg biases a Vbe/12kΩ current generator formed by Q11 and
Q12. The negative TC current from Q11 sums at the output.
The 4kΩ and 12kΩ resistors are actively laser trimmed over
temperature at wafer level to give an accurate zero TC
output. NPN transistors Q5, Q6, and Q9 cascode Q7 and Q8 for
improved accuracy and output impedance. Likewise, J1 and
J2 cascode Q3 and Q4. Using FET cascodes rather than PNPs
eliminates noise due to base current. The capacitor provides
loop compensation.
The REF200 dual current source has two current sources
plus a current mirror in an 8-pin plastic DIP (Figure 1).
Because the circuit is fabricated with the Burr-Brown dielectrically isolated Difet ® Burr-Brown process, the three circuit blocks are completely independent. No power supply
connections are needed to the chip. Just apply 2.5V or more
to a current source for a constant 100µA output. Typical drift
is less than 25ppm/°C and output impedance exceeds 500MΩ.
100µA was chosen as a practical value for the majority of
applications. It is high enough to be used directly for sensor
excitation in many instances, while it is low enough to be
used in low power and battery powered applications where
a higher current might be excessive. Also at higher output
currents, thermal feedback on the chip and self heating
would reduce the output impedance.
FIGURE 1. The REF200 Dual Current Source contains three
completely independent circuit blocks—two
100µA current sources, and a current mirror.
The current mirror is useful in many applications. It uses a
“full Wilson” type architecture as shown in Figure 2, with
laser-trimming to ensure high accuracy.
FIGURE 2. The REF200 Current Mirror uses a “full Wilson”
architecture for high accuracy.
FIGURE 3. The REF200 Current Source cell is powered
from its input terminals. It achieves zero TC by
summing a positive TC current from a bandgap
cell with a negative TC current.
Each of the two current sources are designed as shown in
Figure 3. Zero temperature coefficient (TC) is achieved by
combining positive TC currents with a negative TC current.
The positive TC currents are generated by a bandgap cell.
Difet ® Burr-Brown Corp.
If compliance closer to the negative rail is needed for either
the 50µA sink or source, use the circuit shown in Figure 6,
or Figure 7. Here the mirror input is referenced to the
negative rail with either a resistor and current source, or a
resistor biased zener.
With a 100µA current source as a reference, it is simple to
construct a current source of any value. The REF200 can be
pin strapped for 50µA, 200µA, 300µA, or 400µA, in addition to 100µA.
For a 50µA current sink, use the circuit shown in Figure 4.
A 100µA current source is tied to the mirror common. Since
a current mirror output must equal its input, 50µA flows in
the input to ground, and the output is a 50µA current sink.
FIGURE 6. Compliance of the 50µA Current Sink (Figure 5)
can be extended to –VS+5V by referencing its
bias point to the negative power supply rail using
the other 100µA current and a resistor.
FIGURE 4. A 50µA Current Sink with compliance to ground
can be made using one of the 100µA current
sources and the mirror from the REF200.
50µA Current Source
For a 50µA current source, use the circuit shown in Figure
5. In this circuit a current sink subtracts 50µA from a second
100µA source leaving a 50µA source. Compliance is from
below ground to within 2.5V of the positive rail.
FIGURE 7. If you don’t have a current source to spare, the
50µA Current Sink with compliance to –VS+5V
can be biased using a zener diode.
A 200µA floating current source is formed by simply paralleling the two current sources as shown in Figure 8. For
compliance nearer to the negative rail, use the mirror as
shown in Figure 9. The output of the mirror can swing about
a volt closer to the negative rail than the current source
FIGURE 5. A 50µA Current Source with compliance from
ground to +VS –2.5V can be made using both
100µA current sources and the mirror from the
FIGURE 8. For a 200µA Floating Current Source simply
parallel the two 100µA current sources from the
FIGURE 10. The two 100µA Current Sources and Mirror in
the REF200 can be connected to form a 300µA
floating current source.
FIGURE 9. You can mirror the 100µA or 200µA Current
Sources from the REF200 for a 100µA or
200µA current sink with improved compliance.
FIGURE 11. The two 100µA Current Sources and Mirror in
the REF200 can be connected to form a 400µA
floating current source.
300µA Floating Current Source
A 300µA floating current source can be strapped together as
shown in Figure 10. It is formed by paralleling a 200µA
current source, made with one 100µA source and the mirror,
with the other 100µA current source. The 200µA current
source is made by connecting a 100µA current source to the
mirror input so 100µA flows in the mirror output, and
200µA flows in the mirror common.
Current Source or Sink With Compliance
to Power Supply Rail and Current Out >100µA
You can build a programmable current source of virtually
any value using two resistors, an op amp, and a 100µA
current source as a reference.
The current source shown in Figure 12 can be programmed
to any value above the 100µA reference current. It has
compliance all the way to the negative power supply rail.
The 100µA reference forces a voltage of 100µA • R1 at the
non-inverting input of the op amp. When using a Difet ® op
amp as shown, input bias currents are negligible. The op
amp forces the same voltage across R2. If R1 is N • R2, the
output current is (N+1) • 100µA. So long as the op amps
input common mode range and its output can swing to the
negative rail within the voltage drop across R1, the current
source can swing all the way to the negative rail. If the
voltage drop across R1 is large enough, any op amp can
400µA Floating Current Source
A 400µA floating current source can be strapped together as
shown in Figure 11. It is basically the same as the 200µA
current source of Figure 10, except that 200µA is fed into the
mirror input. This 200µA is summed with the 200µA that
flows in the mirror output for a total of 400µA.
satisfy this requirement. Figure 13 shows the same circuit
turned around to act as a current sink. It has compliance to
the positive rail.
FIGURE 14. Current Source.
FIGURE 12. Current Source.
FIGURE 15. Current Sink.
NOTE: (1) Burr-Brown® OPA602 or OPA128.
NOTE: (1) Burr-Brown OPA602.
FIGURE 13. Current Sink.
FIGURES 12 and 13. For a programmable current source
with any output current greater than
100µA and compliance to +VS or –
VS, use a 100µA current source as a
reference along with an external op
amp and two programming resistors.
➜ Use OPA128
FIGURES 14 and 15. If you don’t need compliance to the
power supply rail, this circuit using a
100µA current source as a reference
along with an external op amp and
two programming resistors can provide virtually any output current.
Current Source or Sink With Any Current Out
For currents less than 100µA, use the circuits shown in
Figures 14 and 15. They can be programmed for virtually
any current (either above or below 100µA). In this case the
100µA current source forms a reference across R1 at the
inverting input of the op amp. Since the reference is not
connected to the output, its current does not add to the
current output signal. So, if R1 is N • R2, then output current
is N • 100µA. Because compliance of the 100µA current
source is 2.5V, the current source, Figure 14, can only
comply within 2.5V of the negative rail—even if the op amp
can go further. Likewise the current sink, Figure 15, has a
2.5V compliance to the positive power supply rail.
Current Sources and Sinks
Using Voltage References
To make a current source with the best possible accuracy use
a zener-based voltage reference. The REF200 uses a bandgap type reference to allow low voltage two-terminal operation. Although this makes a more flexible general-purpose
part with excellent performance, its ultimate temperature
drift and stability cannot compare to the REF102 precision
10.0V buried zener voltage reference.
Make a current source from a voltage reference using the
circuit shown in Figure 15A. The voltage follower connected
op amp forces the voltage reference ground connection to be
equal to the load voltage. The reference output then forces an
accurate 10.0V across R1 so that the current source output is
Negative output compliance for the current sink is limited by
the op amp and further reduced by the 10V drop across R1.
When using the OPA111 on ±15V power supplies, negative
compliance is only guaranteed to ground. When using the
single-supply OPA1013 op amp, negative compliance is
approximately –5V.
Negative output compliance for the current source is limited
by the op amp input common-mode range or output range
(whichever is worse). When using the OPA111 on ±15V
power supplies, the negative compliance is –10V. For compliance almost to the negative power supply rail, use a singlesupply op amp such as the OPA1013.
Positive compliance is limited by the REF102, but is improved by the 10V across R1. For a REF102 operating on
±15V power supplies, the positive compliance is +10V
(limited by the op amp common mode input range).
Positive output compliance is limited by the voltage reference minimum +VS requirement. When using the REF102 on
±15V power supplies, positive compliance is +3.5V.
Make a current sink with a voltage reference using the
circuit shown in Figure 15B. The op amp drives both the
voltage reference ground connection and the current scaling register, R1, so that the voltage reference output is equal
to the load voltage. This forces –10.0V across R 1 so that the
current sink output is –10V/R 1. The R2, C1 network provides local feedback around the op amp to assure loop
stability. It also provides noise filtering. With the values
shown, the reference noise is filtered by a single pole with
f–3dB = 1/(2 • π • R2 • C1).
Keep in mind that the ultimate accuracy of a voltage reference based current source depends on the absolute accuracy
of the current-scaling registor. The absolute TCR and stability of the resistor directly affect the current source temperature drift and accuracy. This is in contrast to circuits using
current source references, as shown in Figures 12 to 15,
where accuracy depends only on the ratio accuracy of the
resistors. It is much easier to get good resistor ratio accuracy
than to get good absolute accuracy especially when using
resistor networks.
Although these current sources made with voltage references
do not have the compliance range of the previous circuits,
they may be the best choice where the utmost in accuracy is
FIGURE 15A. Current Source using Voltage Reference and Op Amp.
FIGURE 15B. Current Sink using Voltage Reference and Op Amp.
Floating Current Source
With Current Out > 100µA
If a completely floating programable current source is needed,
use the circuit shown in Figure 16. It is basically the same
as the current source shown in Figure 12 except that R2 is
driven by a MOSFET. Since no current flows in the gate of
the MOSFET or the inputs of the op amp, all current that
enters the resistors (and no more) leaves. Therefore the
current source is completely floating.
Notice that since a current source is used as a reference, the
circuit can also be used as a precision current mirror. Unlike
mirrors which use matched transistors, this mirror remains
highly accurate no matter what the mirror ratio.
The power supplies of the op amp in this circuit, as in the
other circuits, must be connected to ±VS. Also, the input and
output common mode limitations of the op amp must be
In many cases bipolar devices are adequate and may be
preferred due to their low cost and availability. With a
bipolar device, the base current will add error to the output
signal as discussed in the cascoding section. Using a darlington-connected bipolar device feeds the error current back
into the signal path and reduces the error by the forward
current gain (beta) of the input transistor.
The pass element can bipolar, JFET, MOSFET, or a combination. The examples recommend MOSFETS because their
low gate current minimizes output error. Also, MOSFETs
with very high current ratings are available, and require no
additional drivers.
In some high temperature applications, darlington-connected
bipolar transistors may have lower error than FETs. As a
rule of thumb, the gate current of a FET or MOSFET
doubles for every 8°C increase in temperature, whereas the
beta of a bipolar device increases approximately 0.5%/°C.
Therefore, when operating at 125°C, the gate current of a
FET will be about 6000 times higher than at 25°C, while the
base current of the bipolar will be 1.5 times lower.
When selecting the op amp for this application, pay particular attention to input bias current, input common mode
range, and output range.
The bias current of the op amp adds to the input current, and
subtracts from the output current. For a 1:1 mirror application, the error is only the mismatch of bias currents or IOS of
the amplifier. For other ratios, assume that the error is equal
to the full amplifier bias current. For most applications, the
error will be negligible if a low bias current Difet ® amplifier
such as the Burr-Brown OPA602 is used. Its IB is 1pA max.
Be sure to observe the input common mode range limit of
the op amp. For example, when using the OPA602 in a
current sink application, the voltage between the op amp
negative supply and its input must be at least 4V. In a split
power supply application, R1 and R2 can be connected to
ground, and the op amp negative supply can be connected to
–5V or –15V and there is no problem. In a single supply
application, or when R1, R2, and the op amp’s –VS are all
connected to the negative power supply, a drop of at least 4V
must be maintained across R1.
NOTE: (1) Burr-Brown Difet ® OPA602.
FIGURE 16. Use a 100µA Current Source as a reference, an
external op amp, two programming resistors,
and a series pass element for a programmable
floating curent source.
Current Sources and Sinks and Current Mirrors
Using an Amplifier and a Series Pass Element
In some applications it may be desirable to make a current
source or sink using a series pass element in addition to an
op amp. This approach provides the benefits of cascoding
and also allows arbitrarily high current outputs.
Using a single supply op amp allows the input common
mode range to go to 0V. Especially in single supply current
mirror applications, it is often desirable for the input and
output to go to zero. The OPA1013 has an input commonmode range which extends to its negative power supply, and
its output will swing within a few mV of the negative supply.
Although the OPA1013 has bipolar inputs, its bias current is
low enough for most applications.
The circuit used is the same as for the programmable
floating current source shown in Figure 16. The difference
is that the op amp power supply connection and reference
input are both returned to a fixed potential. The result is
either a current source or sink, but not a floating current
source. The advantage is that the output can be any value,
either more than, less than, or equal to the input reference.
Also, a voltage source or even a variable voltage input can
be used as a reference. The examples shown in Figures 17
through 20 show 100µA current sources used as references.
Components R3, R4, and C1 form a compensation network to
assure amplifier stability when driving the highly capacitive
inputs of some MOSFETs. In many applications they can be
N channel enhancement MOSFET
Supertex, Siliconix, Motorola, etc.
2 x 2N2222
NOTE: Can be connected to –VS with 4V min across R1.
FIGURE 17. Programmable Current Sink using series pass device.
N channel enhancement MOSFET
Supertex, Siliconix, Motorola, etc.
2 x 2N2222
FIGURE 18. Sinking Current Mirror using series pass device.
Supertex, Siliconix, Motorola, etc.
P channel enhancement MOSFET
2 x 2N3906
NOTE: Can be connected to
+VS with 2V min across R1.
FIGURE 19. Programmable Current Source using series pass device.
NOTE: The LF155 input common-mode range typically includes +VS,
but this is not a guaranteed specification.
Supertex, Siliconix, Motorola, etc.
P channel enhancement MOSFET
2 x 2N3906
FIGURE 20. Sourcing Current Mirror using series pass device.
quiescent current flowing from its negative supply pin sums
into the current flowing into R2. The op amp outputs drive
the additional current needed through R2 so the voltage drop
across it matches the voltage drop across R1. If R1 is N • R2,
the output current is (N+1) • 100µA. With the values shown,
the output current is 25mA.
The op amp outputs are connected to R2 through 100Ω
resistors. The current delivered by A1 produces an approximate 0.5V voltage drop across R3. The other three op amps
are connected as voltage followers so that the same voltage
is dropped across the other three 100Ω resistors. The output
current from each op amp is therefore equal and the load is
shared equally. This technique allows any number of 10mA
output op amps to be paralleled for high output current.
Floating Current Source With Current
Out >100µA and No Separate Power Supply
If a programmable current source is needed, and no separate
power supply is available, consider the floating current
source shown in Figure 21. Here the op amp power supplies
are connected to the current source input terminals. The op
amp quiescent current is part of the output current.
Cascoding With FETs
The output impedance and high frequency performance of
any current source can be improved by cascoding. Starting
with a precision current source like the REF200 or any of the
variations previously discussed, it is relatively easy to build
a current source to satisfy just about any need.
Cascoding can also be used to increase high voltage compliance. High voltage compliance of a cascoded current source
is limited solely by the voltage rating of the cascoding
device. High voltage compliance of hundreds or even thousands of volts is possible.
Cascoding is the buffering of the current source from the
load by a series pass device as shown in Figure 22. Here an
N channel JFET cascodes the current source from the output.
The gate of the JFET is tied to ground, its source to the
current source, and its drain to the load. Variations in the
load voltage are taken up by the drain of the JFET while the
source voltage remains relatively constant. In this way, the
voltage drop across the current source remains constant
regardless of voltage changes across the load. With no
changes in the voltage across the current source, and with no
current lost through the JFET drain approaches infinity. AC
performance of the cascoded current sink approaches that of
the JFET.
NOTE: Each amplifier 1/4 LM324.
FIGURE 21. 25mA Floating Current Source using a quad
single-supply op amp needs no external power
There are two special requirements. First, a single supply op
amp must be used (an op amp with an input common mode
range that includes the negative supply rail). Also the output
current must be greater than the op amp quiescent current.
The circuit is basically the same as Figure 12. The 100µA
current flowing through R1 produces a floating voltage
reference at the non-inverting input of A1. The op amp
FIGURE 22. Cascoded Current Sink with compliance to
FIGURE 25. Cascoded Current Source with compliance to
Since the gate of the JFET is tied to ground, the output
compliance is limited to near ground. If greater compliance
is required for the current sink, the gate of the JFET can be
referenced a few volts above the negative rail as shown in
Figures 23 and 24. In Figure 23 the gate reference is derived
from a resistor biased from the second current source. If a
current source is not available, use a resistor biased zener as
shown in Figure 24.
FIGURE 26. Cascoded Current Source with compliance to –
VS+5V (using current source and resistor for
FIGURE 23. Cascoded Current Sink with compliance to –
VS+5V (using zener diode for bias).
FIGURE 27. Cascoded Current Source with compliance to –
VS+5V (using zener diode for bias).
In most applications, JFETs make the best cascoding devices,
but bipolar transistors and MOSFETs can also be used.
MOSFETs can provide equivalent AC and DC performance
to JFETs. Bipolar devices may offer better high frequency
performance, but have a limited DC output impedance. The
output impedance of a bipolar cascoded current source is
limited by changes in base current with changes in collector
voltage. The maximum output impedance of a bipolar cascoded current source is b • RO, where b is the current gain of
the bipolar device, and RO is its output impedance
FIGURE 24. Cascoded Current Sink with compliance to –
VS+5V (using zener diode for bias).
To implement current sources, turn the circuits around and
use P channel JFETs as shown in Figures 25 through 27.
200µA Floating Cascoded
Current Source Using REF200
Floating cascoded current sources with typical output impedances exceeding 10GΩ can be easily implemented. Using the REF200 and a few external devices, sources of
200µA, 300µA, and 400µA can be strapped together.
The 200µA floating cascoded current source is shown in
Figure 28. It is made using a cascoded current source and a
cascoded current sink each biasing the other. Low voltage
compliance is limited to about 8V by the sum of the gate
reference voltages. High voltage compliance is limited by
the lower voltage rated FET.
Regulation (15V–30V) = 0.00003%/V (10GΩ)
FIGURE 29. Cascoded 300µA Floating Current Source.
Regulation (15V–30V) = 0.00005%/V (10GΩ)
FIGURE 28. Cascoded 200µA Floating Current Source.
300µA Floating Cascoded
Current Source Using REF200
The 300µA floating current source is shown in Figure 29. It
is similar to the 200µA current source shown in Figure 28,
except the current source in the cascoded sink section is
derived from the mirror. The gate reference for the sink
cascode is derived from the series combination of the mirror
input and a 27kΩ resistor. The extra 100µA is obtained by
summing the other 100µA current source into the sink
cascode device. Compliance limits are the same as for the
200µA cascoded source.
Regulation (15V–30V) = 0.00025%/V (10GΩ)
400µA Floating Cascoded
Current Source Using REF200
The 400µA floating cascoded current source is shown in
Figure 30. It is the similar to the 300µA cascoded current
source, except that the mirror is driven by a cascoded 200µA
current source derived by the parallel combination of the
two current sources in the REF200. The low voltage compliance of this circuit is about 1V better than the previous two
circuits because the mirror compliance is about 1V better.
High voltage compliance is still limited only by the breakdown of the lower rated FET.
FIGURE 30. Cascoded 400µA Floating Current Source.
In many modern systems, noise is the ultimate limit to
accuracy. And in some systems, performance can be improved with a lower noise current source. Current source
noise can be reduced by filtering, using the same basic
principals used for noise reduction of voltage references.
Reducing the noise bandwidth by filtering can reduce the
total noise by the square root of the bandwidth reduction.
One current source noise reduction circuit is shown in
Figure 31. It is basically a FET cascode circuit with the
addition of an RC noise filtering circuit. The FET, as biased
by the 100µA current source, forces an accurate DC voltage
across the circuit.
In addition to noise reduction, these circuits have the other
advantages of a FET cascoded current sink; output impedance in the GΩ region, improved AC performance, and high
voltage compliance limited only by the FET.
Without the capacitor, noise from the current source would
feed directly through to the output. The capacitor filters the
noise at a –3dB frequency of 1/(2 • π • R• C), or about 30Hz
in this example. Filtering below this frequency will not
reduce noise further, since the 30Hz pole is already below
the 1/f corner of the current source, and noise can not be
reduced by filtering in the 1/f region. Also, the noise of the
FET and resistor are not filtered. Still, using this circuit, the
noise is reduced from the typical 20pA/√Hz to less than
FIGURE 32. Current Noise Filtering Circuit with compliance below ground.
Many design problems can be easily solved with inexpensive, easy-to-use current sources like the REF200. Although
applications are endless, the collection of circuits that follows is intended to stimulate your thinking in several broad
categories: fixed voltage references, floating voltage references, current excitation, fixed current references, steered
current references, and biasing.
FIGURE 31. Current noise from a current source can be
filtered using this circuit.
The value of the resistor used in the noise reduction circuit
determines its ultimate performance. Although the noise of
the resistor increases with the square root of its value, its
noise degenerating effect reduces noise linearly. Therefore,
the noise is reduced by the square root of the resistor
increase. The practical limit for the noise reduction is the
voltage drop which can be placed across the resistor.
Current sources are a versatile means of forming voltage
references. Why not just use a voltage reference? With a
current source, a single resistor provides a programmable
voltage source of any value. Low voltage references are
often needed, and with this approach, it’s as easy to get a
1mV reference as it is to get a 10V reference. Also, the
voltage can be referenced anywhere—to the positive rail, the
negative rail, or floating anywhere in between.
Mathematically, current noise due to the resistor is the
resistor thermal noise divided by the resistor value.
(1.3 •10 –10 )
= 1.3 •10 –10 / R
When impedances driven by the voltage reference are high,
the voltage output from the resistor derived voltage reference can be used directly. The 100µV reference shown in
Figure 33 can be used directly in voltage-to-frequency
converter (VFC) auto-zero applications where an off-zero
reference is needed (since zero frequency would take forever
to measure, off-zero techniques are often used for calibrating VFCs). Where a lower output impedance is needed, a
simple buffer can be added as shown in Figure 34.
With a 50kΩ resistor, the minimum theoretical noise is
.6pA/√Hz, with 10kΩ, it is 1.3pA/√Hz. Noise measurements
of the circuit using both 10kΩ and 50kΩ resistors and the
Siliconix J109 FET agree with these theoretical numbers
within 20%.
The noise reduction circuit in Figure 31 has a low voltage
compliance limit near ground. For compliance below ground,
use the circuit shown in Figure 32.
connected to the output. High-side compliance is limited
only by the op amp. Another advantage of this circuit is its
high input inpedance. The disavantage is limited low side
compliance. Current source compliance limits swing to the
negative rail to 2.5V — regardless of the op amp input
common mode range.
FIGURE 33. 100µV Reference for VFC off-zeroing.
FIGURE 34A. Floating Voltage Reference.
FIGURE 34. Buffered Voltage Reference.
For a floating voltage reference, simply drive the reference
low side (grounded side of the voltage-setting resistor) as
shown in Figure 34A. Notice that in addition to the swing
limitations imposed by the op amp input common-mode
range and output range, the reference high side swing is
limited to 2.5V from the positive rail by the REF200's
minimum compliance voltage. The low side swing is limited
only by the op amp. If the reference voltage is more than
about 3V this limitation can be eliminated by adding gain as
shown in Figure 34B. In this example, the 1V across the
reference setting-registor is amplified to 5V at the output.
Since there is always 4V between the output and the op amp
inputs, the high-side swing is not limited by the current
source compliance or the op amp input common mode
range. It is limited only by the op amp out put swing
FIGURE 34B. Floating Voltage Reference with high-side
compliance limited only by op amp output
swing capability.
Where the voltage reference is lower than about 3V, the high
side compliance will still be limited by the current source
compliance. In these situations, consider the circuit shown in
Figure 34C. In this case, the op amp noninverting input is
driven while the current source connects to the other op amp
input and a voltage setting resistor with its other terminal
FIGURE 34C. Floating Low Voltage Reference with high
impedance input drive and high-side output
compliance limited only by op amp output
swing capability.
Op amp offset adjustment circuits are another application for
millivolt level references. Many op amps, especially duals
and quads, have no built-in provision for offset adjustment.
Even when offset adjustment pins are provided, using them
can degrade offset voltage drift and stability (e.g. the drift of
a typical bipolar input op amp is increased 3µV/°C for each
millivolt of offset adjustment). External offset adjustment
circuits are commonly used to solve these problems.
The circuits shown in Figures 35 and 36 solve these problems. REF200 current sources provide regulated ±5mV
references for stable offset adjustment. This approach provides a truly precision offset adjustment free from problems
associated with power supply variations, noise, and drift.
Conventional external offset adjustment circuits can add
problems of their own. Many of these circuits use the op amp
power supplies as references. Power supply variation feeds
directly into the op amp input. This error appears as poor
power supply rejection. Likewise, noise from the power
supplies appears as op amp input referred noise.
The second circuit, Figure 36, uses a special potentiometer
manufactured by Bourns. It is especially designed for op
amp offset adjustment. It has a tap at the center of the
element which can be connected to ground. Using this
connection eliminates the 51Ω resistors needed in the first
The circuit shown in Figure 35 uses a pair of 51Ω resistors
connected to ground to establish the ±5mV reference. A pot
connected across this reference allows a ±5mV offset adjustment range. Additional pots can be connected, but be sure to
maintain a parallel resistance >50Ω to get >±5mV range.
FIGURE 35. Op Amp Offset Adjustment Circuit uses the two 100µA current sources from a REF200 to provide accurate ±5mV
FIGURE 36. Op amp offset adjustment circuit using Bourns Trimpot®.
excessive errors. The usual solution to this problem is to use
four wire Kelvin connections. Two wires are used to carry
the current excitation signal to the RTD. The other two wires
sense the voltage across the RTD. With no current flowing
in the sense connection, there is no error due to wire
The window comparator circuit, Figure 37, is an example of
a floating reference application. Here, a pair of current
sources is used to provide a floating bipolar window voltage
driven by the VCENTER input. VO is low when VI is either
above VCENTER + 100µA • R, or below VCENTER – 100µA • R.
Otherwise, VO is high. By using different values for the
programming resistors, the threshold can be set asymmetrically around VCENTER if desired.
One problem is that the additional wiring can be very
expensive. The three wire circuit shown in Figure 38 saves
one wire. 200µA is used for excitation of a 1kΩ RTD, and
a matching current from the current mirror is forced in the
ground connection. The voltage drops through the two wires
cancel thereby eliminating error.
Current sources are often used for excitation of resistor type
sensors such as RTDs. If the RTD is located remotely, as it
often is, voltage drops in the interconnecting wire can cause
Notice also, that one common wire (shown as a shield) can
serve multiple sensors. Each additional RTD only needs one
additional pair of wires.
FIGURE 37. Window comparator with voltage programmable window center, and resistor programmable window width.
FIGURE 38. RTD excitation with three-wire lead resistance compensation.
Servo control systems frequently use dead-band and limiting
circuits. The precision dead-band circuit shown in Figures
39 and 40 demonstrates the use of a current source as a fixed
reference. To understand how it works, notice that, without
the current reference, the circuit is an inverting half wave
rectifier. Positive inputs drive the op amp output negative
and feedback is through forward biased D1. No current flows
through reverse biased D2, and the output is held at virtual
ground by R2.
unity gain inverter (VO = –VI ). (Adding the current reference
pre-biases to the input so that the output remains at virtual
ground until the input current through R1 exceeds 100µA.)
The output is zero (dead) until VI < ±100µA • R1. An
alternate approach would be to pre-bias the input through a
precision resistor connected to a voltage reference, but that
would add noise gain increasing offset, drift, and noise at the
For a negative dead-band use the circuit shown in Figure 40.
It’s the same circuit with the diodes reversed.
Negative inputs forward bias D2 and drive the output positive. Feedback to the output through R2 eliminates error due
to the diode drop, and the circuit functions as a precision
For a double dead-band, use the circuit shown in Figure 41.
It uses both the positive and negative dead-band circuits
summed together by a third amplifier.
FIGURE 39. Precision positive dead-band circuit.
FIGURE 40. Precision negative dead-band circuit.
FIGURE 41. Precision double dead-band circuit.
One of the advantages in working with currents is that they
can be steered by diodes or switches without error. As long
as no current is lost through leakage, voltage drops in series
with current signals do not diminish their accuracy.
When R1 is added, the circuit still functions as a follower for
small signals where the current through R1 is less than
100µA. When the current reaches 100µA, the current source
becomes active and limits the output voltage. With the
bidirectional current source, the circuit limits symmetrically
in both directions.
The bidirectional current source shown in Figure 42 is a
versatile circuit building block and an excellent example of
diode steering. This two-terminal element is basically a fullwave bridge rectifier circuit with a current source connected
between its DC terminals. A positive signal on the left
terminal with respect to the right reverse biases D3 and D4 and
accurately steers current through D1 and D2. A negative
signal reverses the situation and accurately steers the same
current in the opposite direction.
For one diode drop better compliance, use the bidirectional
current source shown in Figure 43. The disadvantages of this
circuit are that two current sources are required, and the
inherent current matching of the previous circuit is lost.
FIGURE 42. Bidirectional current source.
FIGURE 44. Precision double limiting circuit.
FIGURE 43. Bidirectional current source with improved
The precision double limiting circuit shown in Figure 44
puts the bidirectional current source to work. To understand
how this circuit works, notice that without R1, it functions as
a precision unity-gain amplifier. The input signal is connected to the non-inverting terminal of A1. Feedback to the
inverting terminal is through the bidirectional current source,
voltage-follower connected A2, and the 1kΩ resistor. When
less than 100µA is demanded from the current source, it
saturates and the total voltage drop across the bidirectional
current source is less than about 2V plus two diode drops.
Since no current flows in the 1kΩ resistor, the circuit output
voltage must equal the input voltage and errors due to A2 and
the drop across the current source are eliminated. The 1k100pF network provides compensation for the extra phase
shift in the feedback loop.
FIGURE 45. Precision limiting circuit.
If a limit in just one direction is required, replace the
bidirectional current source with a current source and single
diode as shown in Figure 45. For limiting in the opposite
direction, reverse the polarity of the current source and
The precision duty cycle modulator shown in Figure 47 is a
variation of the triangle generator. Here the integrating
capacitor is replaced by a true integrator formed by A1 and
C. This allows the summation of a ground referenced signal
through the 100kΩ input resistor. With no input signal, the
output is a square wave with 50% duty cycle. Input signals
sum into the integrator through the 100kΩ resistor. The
integrator then slews faster in one direction, and slower in
the other. The result is a linear duty cycle modulation of the
output signal. The modulator is said to be duty cycle rather
than pulse width because the output frequency varies somewhat with input signal. For a constant frequency duty cycle
modulator add a resistor in series with the inverting input of
A2 and drive that input with a resistor coupled clock signal.
The precision triangle waveform generator shown in Figure
46 makes use of two bidirectional current sources. One
steers a precision current signal into the integrating capacitor
connected to the inverting input of the op amp. The other
steers a precision current into the 10kΩ resistor connected to
the positive op amp terminal to provide ±1V hysteresis. The
result is a relaxation oscillator with precision triangle and
square wave outputs of ±1V.
FIGURE 46. Precision triangle waveform generator.
FIGURE 47. Precision duty-cycle modulation circuit.
Notice that the duty-cycle modulator has a true integrating
input. This is in contrast to conventional modulators which
simply use a comparator connected between the input signal
and a precision triangle wave. With the conventional approach, at crossing, input noise feeds through at the comparator bandwidth resulting in jitter. Not only does the
integrating input filter out input noise, it can be synchronized to input noise (such as 60Hz), completely notching out
its effect. If integration takes place over one or more complete cycles of the noise signal, the undulations of the noise
signal are exactly averaged out.
In some applications, especially when driving inductors, it is
necessary to limit the signal slew rate. The rate limiting
circuit shown in Figure 48 uses a diode bridge for current
steering in a different way. Here two current sources are
connected, one to the positive terminal and one to the
negative terminal, of the bridge. Without the capacitor, the
circuit would act as a unity gain inverting amplifier. Feedback through the 10kΩ – 10kΩ resistor network drives the
left side of the bridge. The right side of the bridge follows,
driving the op amp inverting input. Voltage offset due to
diode mismatch can be mitigated by using a monolithic
bridge such as the one specified. When the integrator capacitor is added, charging and discharging current must flow to
maintain the virtual ground. But when that current exceeds
100µA, the bridge reverse biases limiting the output slew
rate to 100µA/°C regardless of input signal rate.
Single power supply systems are common and the need for
instrumentation amplifiers (IAs) to operate in this environ-
ment is critical. While single supply op amps have been
available for many years, single supply IAs have not. What’s
more, single supply IAs can not be made by simply using
single supply op amps in the traditional manner. In a conventional IA topology the outputs as well as the inputs
would need to swing to the negative rail. Although some op
amps come close, no amplifier output can swing all the way
to its power supply rail, especially when driving a load.
The single supply IA circuit shown in Figure 49 solves this
problem by simply level shifting the input signal up by a
Vbe with a matched pair of matched PNP input transistors.
The transistors are biased as emitter followers by the 100µA
current sources in a REF200. The ensuing circuit is a
traditional three op amp IA. OPA1013s are used for input
amplifiers because they are designed for single supply operation and their output can also swing near the negative rail.
The Burr-Brown INA105 is used as a difference amplifier.
All critical resistor matching is taken care of by the INA105.
The common mode range of the single supply IA typically
extends to 0.5V below the negative rail with a typical CMR
better than 86dB.
The modified Howland current pump (Figures 50-52) is an
extremely versatile voltage controlled current source. Since
it has differential inputs, you can ground one input and drive
the other to get either an inverting or noninverting transfer
function. If you drive both inputs, the output current will be
proportional to the voltage difference between the inputs.
What’s more, unlike current sources made with a series pass
element, which can either sink or source current, this current
source has a bipolar output. It can both sink and source
FIGURE 48. Rate limiting circuit.
FIGURE 49. Single-supply instrumentation amplifier.
Use of this circuit was limited in the past due to the critical
resistor matching and resistor TCR tracking requirements.
By using the INA105 difference amplifier, the circuit can be
easily implemented with the addition of two 1% resistors.
Matching of the external resistors is important, but since
they add to the internal 25kΩ resistors, the matching requirement is divided down by the ratio of resistance.
Output impedance of the current source is proportional to
the common mode rejection (CMR) of the difference amplifier. Mismatch of feedback resistors in the difference ampli-
FIGURE 50. Voltage-controlled current source with differential inputs and bipolar output.
fier caused by the external resistors will degrade CMR and
lower the current source output impedance. Resistor match
of 0.002% is required for 100dB CMR in a unity gain
difference amplifier. Depending on the value of the external
resistor and output impedance requirement, it may be necessary to trim the external resistor.
When the value of the external resister becomes large
consider the alternate circuit shown in Figure. 51.
FIGURE 51. Voltage-controlled current source with differential inputs and bipolar output and circuit to
eliminate feedback resistor error.
You need an extra amplifier to drive the feedback resistor in
the difference amplifier, but only one external resistor is
required, and no matching or trimming is needed.
In any case the output impedance of the current source can
be approximated by the following relationship:
ZO = RX • 10[CMRR/20]
ZO = equivalent output impedance of current source [Ω]
CMRR = difference amp common mode rejection ratio
(for Figure 50)
RX = parallel combination of external resistor and 25kΩ
R • 25kΩ
RX = ————————
R + 25kΩ
(for Figure 51)
RX = external resistor [Ω]
The INA105 can source 20mA and sink 5mA. If higher
output current is required, add a current buffer as shown in
Figure 52. The OPA633 shown allows output currents up to
±100mA. Since the buffer is within the feedback loop, its
DC errors have no effect on the accuracy of the current
source. When using other buffers make sure that their
bandwidth is large enough not to degrade circuit stability.
If you want voltage gain in the voltage to current converter,
use the INA106 for a gain-of-ten difference amplifier.
Don’t forget that source impedance adds directly to the input
resistors of the difference amplifier which can degrade its
performance. A source impedance mismatch of 5Ω will
degrade the CMRR of the INA105 to 80dB. If you are
driving the circuit from an amplifier or other low impedance
source, this should not be a problem. If you have higher
source impedances, buffer the driven input(s) of the difference amplifier, or use an instrumentation amplifier such as
the INA110 instead of a difference amplifier.
The XTR101 is a floating current source designed for two
wire 4-20mA current loop applications. It is a voltage
controlled current source with a precision instrumentation
amplifier input. It also contains two matched 1mA current
sources which makes it suited for remote signal conditioning
of a variety of transducers such as thermocouples, RTDs
thermistors, and strain gauge bridges.
Figure 53 shows the XTR101 connected as a temperature
controlled current source. The temperature sensing element
is a thermocouple, and cold junction compensation is provided by the diode.
The product data sheet for the XTR101, (PDS-627) gives
operating details for the device and shows several other
The XTR110 is a precision single supply voltage to current
converter. Although it is designed specifically for three wire
4-20mA current transmission it can also be used in more
general voltage to current source applications. As shown in
Figure 54, it contains: a precision 10.0V reference and input
resistor network for span offsetting (0V In = 4mA Out), a
voltage to current converter for converting a ground referenced input signal to an output current sink, and a current
mirror for turning the output of the current sink into a current
The current mirror has a gain ratio of 10:1 and uses an
external pass transistor to minimize internal thermal feedback and improve accuracy. Since the mirror transistor is
external, an external mirror ratio setting resistor can be
added for an arbitrarily high output current.
Both the voltage to current converter, and the current mirror
use single supply op amps so that the input and output
signals can go to zero. In the case of the mirror op amp, the
common mode range goes to the positive power supply rail
rather than common.
The following table shows a range of input-output spans that
is available simply by pin strapping the XTR110.
For more details and applications, request product data sheet PDS-555.
FIGURE 52. Voltage-controlled current source with differential inputs and current boosted bipolar output.
FIGURE 53. Temperature-controlled current source using XTR101.
FIGURE 54. Precision single-supply voltage-to-current source transmitter—the XTR110.
Matching resistor, RM, preserves the resistance match of the
INA105 and maintains its high common-mode rejection
(CMR). Because 100Ω is small compared to the 25kΩ
difference resistors, a 1% tolerance is sufficient to maintain
86dB CMR.
Measuring current signals can be as simple as feeding the
current into a precision resistor (V = I • R). If needed, the
voltage developed across the resistor can be buffered or
amplified with an operational amplifier.
If common-mode signals are present on the current return
end of the sense resistor, an instrumentation amplifier (IA)
can be used to reject the common mode signal and reference
the output signal to ground. However, a limitation of conventional IAs is that their common mode input range is
limited to less than 10V.
When you need to reference the current return of the sense
resistor to a higher common mode voltage, consider one of
the following difference amplifiers.
Input Common
Mode Range
± 20V
± 40V
The INA105 references the output signal to ground with a
gain of one. For a 0-10mA input the output is 0 to –1V (a
transfer function of –100V/A). If a positive transfer function
is desired, interchange the input pins of the difference
amplifier. To reference current signals to the positive rail,
simply connect the sense resistor and the other difference
amplifier input to that point.
The 25kΩ input impedance of the difference amplifier
causes a slight error by shunting a portion of the input
current signal. In the noninverting configuration, the matching resistor lowers the difference amplifier gain, but since
the shunting input impedance of the noninverting input is
50kΩ, for a unity gain difference amp, the error turns out to
be the same. For a 100Ω sense resistor, the error is a
approximately 0.4%. For better accuracy, select a slightly
higher value sense resistor to compensate for the error
according to the following equation.
25k • X
RS = RM = ———————
25k – X
NOTES: (1) Common-mode input range specified for operation on standard
±15V power supplies. (2) Also contains a precision reference and offsetting
circuitry to get 0-5V outputs with 4-20mA inputs.
X = desired transfer function [V/A]
Figure 55 shows a current receiver using the INA105. The
input current signal is sensed across 100Ω resistor, RS,
connected to the negative power supply rail. Connecting the
sense resistor to a power supply rail instead of ground
maximizes the voltage drop available across the current
For example:
For 1V/10mA (100V/A) from Figure 55:
RS = RM = 100.4Ω
Voltage divider action of the feedback resistors within the
INA105 divide the common-mode input by two. Therefore,
common mode input signals of up to ±20V are attenuated to
an acceptable level of no more than ±10V at the op amp
The INA117 is a difference amplifier similar to the INA105
except that it has a 20/1 input divider allowing a ±200V
common mode input range. It also has an internal gain of 20
providing an overall gain of one. The penalty is that amplifier DC errors, and resistor and amplifier noise are amplified
by 20. Still for 200V applications that do not require galvanic isolation its has better performance than isolation
amplifiers, and it does not require an isolated power supply.
Figure 56 shows a circuit for measuring load current in a
bridge amplifier application using the INA117. At low
frequencies, a sense resistor could be inserted in series with
the load, and an instrumentation amplifier used to directly
monitor the load current. However, under high frequency or
transient conditions, CMR errors would limit accuracy. This
approach eliminates these problems by gleaning the load
current from measurements of amplifier supply current.
The power supply current of one of the bridge op amps is
measured using INA117s and 0.2Ω sense resistors connected to the power supplies. Because the 0.2Ω sense resistor adds negligible resistance error to the 380kΩ input
resistors of the INA117s, no matching resistors are required.
FIGURE 55. Current-to-voltage converter referenced to the
negative power supply rail.
FIGURE 56. Bridge amplifier load current monitor using the INA117.
To understand how the circuit works, notice that since no
current flows into the inputs of A1:
The RCV420 is a current-to-voltage converter designed
specifically for conversion of 4-20mA input currents into 05V outputs. A pair of precision 75Ω sense resistors are
provided internally allowing both inverting and noninverting
transfer functions. Input common mode signals up to ±40V
can be accommodated due to the internal 4/1 input attenuator. Also, the precision 10.0V reference used for span offseting
is available to the user.
ILOAD = I1 – I2
R1 = R2 = R
e1 = I1 • R,
e2 = –I2 • R,
e1 + e2 = ILOAD • R
Figure 57 shows a typical application. For more details and
applications, request product data sheet PDS-837.
The INA105 is connected as a noninverting summing amplifier with a gain of 5 (the accurate matching of the two 25kΩ
input resistors makes a very accurate summing amplifier).
eO = 5(e1 + e2) = 5(ILOAD • R),
R = 0.2Ω,
eO = ILOAD [1V/A]
When current-to-voltage conversion with no voltage burden
is needed, used the transimpedance amplifier Figures 58—
61. In this circuit, an op amp drives the current input node to
virtual ground by forcing a current equal to IIN through the
feedback resistor, RFB. Notice that the transfer function is
NOTE: (1) May Connect to Gnd or up to ±40V.
FIGURE 57. 4-20mA current loop receiver using the RCV420.
FIGURE 58. Virtual ground current-to-voltage converter.
FIGURE 59. Virtual ground current-to-voltage converter
redrawn to Illustrate phase delay due to input
capacitance and feedback resistor.
The feedback capacitor, CFB, may be needed for circuit
stability. To see why, consider the redrawn circuit, Figure
59. CIN represents the input capacitance of the circuit and
includes input source capacitance, and op amp input capacitance. Notice that RFB and CIN form a single pole filter in the
feedback path to the op amp input. Phase delay through this
circuit subtracts from the op amp phase margin which may
result in instability, especially with the large values of RFB
often used in these circuits. If CFB ≥ CIN, the phase delay will
be less than 20° assuring stability with most unity-gainstable amplifiers.
minimizes the peaking and improves stability as discussed
previously. Capacitors with the small values often required
may be difficult to obtain. By using a capacitor divider
circuit shown in Figure 60A, a larger value capacitor can be
used. In this example, the 10pF capacitor, C1, is reduced to
an effective value of 1pF by the R4, R5 10/1 divider. The
100 pF capacitor, C3, keeps the Impedance of the divider low
beyond the C1, R4 || R5 zero to maintain C1's effect. It also
produces a second-order (40dB/decade) roll-off approximately one decade beyond the C1/10, R2 pole.
The addition of two passive components to the standard
configuration as shown in Figure 61 introduces a second
pole that significantly reduces noise. The modification also
has other advantages.
The added pole of the improved circuit is formed with R3
and C2. Because the pole is placed within the feedback loop,
the amplifier maintains its low output impedance. If the pole
were placed outside the feedback loop, an additional buffer
would be required. The extra buffer would add additional
noise and DC error.
The photodiode amplifier shown in Figure 60 is a common
application of the transimpedance (current-to-voltage) amplifier. In this application, the shunt capacitance of the
photodiode reacting with the relatively large feedback resistor creates excess noise gain. The 1pF feedback capacitor
Photodiode Equivalent Circuit
R1 = 100MΩ
CO = 25pF
FIGURE 60. Standard transimpedance photodiode amplifier.
exhibits two pole response. With C1 • R2 = 2 • C2 • R3, the
transfer function is two pole Butterworth (maximally flat in
the passband). Figure 62 shows the transimpedance frequency response of the two circuits. At DC, the gain is
140dB or 10V/µA. The frequency response of both circuits
is 3dB down at 16kHz. The conventional circuit rolls off at
20dB/decade, while the improved circuit rolls off at 40dB/
FIGURE 60A. Standard Transimpedance Amplifier with
capacitor divider and added feedback pole.
FIGURE 62. Transimpedance signal response of standard
and improved photodiode amplifier.
Figure 63 shows the noise gain of both circuits. The noise
problem is due to the noise gain zero formed by the relatively
high photodiode shunt capacitance, CO, reacting with the
high 10MΩ feedback resistor. The noise zero occurs at:
FIGURE 61. Improved transimpedance photodiode amplifier.
The signal bandwidth of both circuits is 16kHz:
(R1 + R3)
fZ = ——————————— ~ 673Hz in this example.
2 • π • R1 • R3 • (C0 + C1)
f–3dB = ————————— [Hz] ➝ standard circuit
2 • π • R2 • C1
Both curves show peaking in the noise gain at about 673Hz
due to the zero formed by the photodiode shunt capacitance.
The added pole of the improved circuit rolls off the noise
gain at a lower frequency, which reduces the noise above
20kHz. Since the signal bandwidth is 16kHz, the region of
the spectrum above 20kHz contains only noise, not signal.
With the values show, the improved circuit has 3 times less
noise. With the OPA602 (voltage noise = 12nV/√Hz), and
including resistor noise, the improved circuit has 1Hz to
100MHz noise of 68µVrms vs 205µVrms for the standard
f-–3dB = ——————————— [Hz]➝ improved circuit
2 • π • (R2 • C1 • R3 • C2)1/2
Where, for the improved circuit:
C1 • R2 = 2(C2 • R3 )
and R2 >> R3
In the standard circuit, a single 16kHz pole is formed by the
1pF capacitance in the feedback loop. The improved circuit
FIGURE 63. Noise gain of standard and improved
photodiode amplifier.
Another advantage of the improved circuit is its ability to
drive capacitive loads. Since the output of the circuit is
connected to a large capacitor, C2, driving a little extra
capacitance presents no stability problems. Although the
circuit has low DC impedance, the AC transfer function is
affected by load. With reasonable loads, the effect is minimal. With the values shown, a load of 10kΩ in parallel with
100pF has little effect on circuit response.
For applications where the photodiode can be floated consider the noninverting I/V converter shown in Figures 64
and 65. Notice that the buffer amplifier forces zero volts
across the photodiode as in the conventional transimpedance
amplifier configuration.
FIGURES 64 and 65. Photodiode amplifier using floating
virtual ground current-to-voltage
FET input amplifiers are commonly used for photodiode
amplifier applications because of their low input bias currents. However, FET amplifier bias currents increase dramatically at high temperatures (doubling approximately every 8 to 10°C). Seemingly small input bias currents at 25°C
can become intolerable at high temperature. An amplifier
with only 1pA bias current at 25°C could have nearly 6nA
bias current at 125°C.
The difference between input bias currents, offset curent, is
often much better than the absolute bias current. The typical
bias current of an OPA156, for example, is 30pA, while its
offset current is 3pA.
If amplifier bias current is a problem consider the circuit
shown in Figure 66. The added bias current cancellation
resistor R2 cancels the effect of matching op amp input bias
currents. This can provide a ten-to-one or better improvement in performance since voltage offset is due only to IOS
(offset current) reacting with 5MΩ.
FIGURE 66. Differential Photodiode Transimpedance
Amplifier gives bias current cancellation.
A word of caution. Many amplifiers, especially bipolar input
amplifiers, achieve low bias current with internal bias current cancellation circuitry. There may be little or no difference between their IB and IOS. In this case external bias
current cancellation will not improve performance.
Bidirectional Current Source—A floating current source
that provides a constant current independent of the polarity
of applied voltage bias.
Current Source—This may be a general term for any
current source, current sink, or floating current source. In this
text it usually refers to a current generating device referenced
to a positive fixed potential such as +VS of a power supply.
The load must be connected between the current source and
a more negative potential.
a fixed gain while rejecting the common mode signal. An op
amp amplifies the signal at its inputs by its open loop gain
(ideally infinity). An op amp therefore requires feedback
components to make a useful amplifier. It normally takes
three op amps and seven precision resistors to make an IA.
IB—Bias current. The DC current that flows into or out of the
input terminals of an amplifier.
Current Sink—Current generating device referenced to a
negative fixed potential such as –VS of a power supply. The
load must be connected between the current sink and a more
positive potential.
IOS—Offset current. The difference in IB of the two inputs of
an amplifier.
IC—Integrated circuit. Often implies monolithic integrated
circuit, which is a single-chip electronic circuit.
Op Amp—Operational amplifier. An operational amplifier
is a very high gain direct current amplifier with differential
inputs. It is intended for applications where the transfer
function is determined by external feedback components.
RTD—Resistor Temperature Device. A precision temperature transducer using platinum as the active element. Values
at 0°C of 100Ω, 500Ω, and 1000Ω are standard. Due to the
high cost of platinum 1kΩ RTDs are becoming more popular.
Difet ®—Burr-Brown’s trademark for an integrated circuit
process which uses dielectric (DI) instead of reverse biased
junctions (JI) to isolate devices. This technique eliminates
the substrate leakage inherent in JI processes. The result is
lower input bias currents for FET input amplifiers, and
potential for higher temperature operation and radiation
Floating Current Source—A current generating device
with both ends uncommitted. The load may be connected to
either end, or a floating current source may be connected
arbitrary between two loads. The current sources in the
REF200 are floating current sources. A floating current
source may require external power supplies. The floating
current sources in the REF200 are self powered, and require
no external power supply.
TCR—Temperature coefficient of resistance. The change of
DC resistance with temperature of a resistor. Usually expressed in parts per million per °C [ppm/°C].
TCR Tracking—The match or tracking over temperature of
the TCR of two or more resistors.
Transconductance Amplifier—A voltage to current converter.
Transimpedance Amplifier—A current to voltage converter. (Sometimes called transadmittance).
IA—Instrumentation Amplifier. An IA is not an op amp.
Unlike an op amp, an IA amplifies the signal at its inputs by
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