High Linearity HBT Amplifiers for CATV Systems

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RF2312/RF2317: High Linearity HBT Amplifiers for CATV Systems
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The need for high linearity amplifiers arises from stress
placed on communications channels by the addition of
more data and the requirement to handle digitally modulated signals with high fidelity. As the amount of data
increases and the necessity to reduce spurious interference increases, the channel linearity must be
improved over current implementations. A particularly
difficult problem is obtaining the linearity and bandwidth needed for the amplification of multicarrier signals such as those found in the TV distribution industry.
Cable TV, for example, is expanding rapidly in terms of
both the number of channels and additional subscribers. Hence, more bandwidth and power output capacity (without degrading linearity) are necessary. These
systems are also incorporating more technology into
their systems including two-way channels for voice and
data, and digital video modulation. As this additional
capacity is added, the bandwidth and signal handling
capability of the various links in the distribution chain
must be improved. In addition to the need for more
capacity, the linearity of the channel must further
improve because digital modulation schemes are less
tolerant of IM distortion products than the analog systems currently in use. The infrastructure needed to
accommodate these technology innovations must be in
place throughout the distribution chain before it can
become available to the consumer. Hence, there is
considerable pressure to improve channel linearity,
bandwidth, and power handling capability in the many
links of the various distribution channels.
RF Micro Devices is developing a line of high linearity,
wide bandwidth products to address these requirements. The first in the series is the RF2312 amplifier
which has been released as a standard product by
RFMD. Other higher power units are the RF2317 (a
higher power version of the RF2312) and the RF2316,
which is a balanced device having improved second
order distortion over the unbalanced designs. The latter two devices will be in full production in January
1997.
The RF2312 and RF2317 are essentially a low cost,
traditional Darlington “Gain Block” circuits with the
exception that performance far outstrips previously
available units. The gain is essentially flat over the
bandwidth from 1MHz to well over 1GHz and has an
IP3 of over +40dBm. The secret to the performance of
the unit (as in all the units mentioned) is in the use of
GaAs Heterojunction technology in the fabrication of
the device. The unit will replace more expensive and
less reliable discrete amplifiers and permit much better
distortion levels for a given amount of DC power consumption.
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A multicarrier channel consists of many independent
RF carriers each with its own frequency and modulation. It is important that each of these signals remain
“uncontaminated” by all of the other signals in the
channel. For simplicity, consider that if n CW carriers
are present, have the same amplitude, and are coherent with each other, the resulting frequency spectrum
is a “picket fence” as shown in figure 1. The time
waveform resembles an impulse function as shown in
figure 2. (Note that because the frequency spectrum
does not start at zero and because some phase distortion is introduced by filters and amplifiers in the chain,
the time function is generally not, strictly speaking, an
impulse.) The amplitude of the voltage peak is
20log(n)dB above the amplitude peak of a single tone.
This occurs when all of the signals have a peak amplitude occurring at the same point in time. If, however,
each carrier is independent and has a random phase
with respect the other carriers, the time waveform is
similar to that shown in figure 3 while the frequency
spectrum remains the same as in figure 1. In some
newer systems, this spectrum can extend from a few
MHz to about 1GHz. The result is a noise like signal
which has the same average power as in the coherent
case, but has much smaller peak excursions. This is an
important fact, since the amplifier has to be designed
to handle peak excursions in a linear fashion and
designing the part for voltage excursions of 20log(n)
dB above the single tone amplitude would be very difficult and be an “overkill”.
Copyright 1997-2002 RF Micro Devices, Inc.
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TECHNICAL NOTES
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Monolithic Amplifiers using GaAs HBT technology
have been developed. HBT based amplifiers offer
extremely flat frequency response with high dynamic
range and use less DC power than Silicon or GaAs
MESFET based circuits. These inexpensive amplifiers
are packaged in a standard SOIC plastic packages and
feature very flat frequency response to beyond 1GHz
with OIP3 numbers in excess of +40dBm. They are
capable of amplifying multicarrier CATV signals with
very high fidelity.
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Because the signal is noise like, the peak excursions of
the voltage waveform are a statistical function and
therefore the maximum level is indeterminate except to
say that it is smaller than A*20 log(n), where A is the
amplitude of each carrier and n is the number of signals. In fact, successive measurements may yield different answers. Because of the nature and quantity of
the signals present, however, it is possible to estimate
the maximum likely excursion of the waveforms. Many
measurements have shown that as the number of signals increase, the actual peak excursion is always
reduced markedly from the theoretical maximum given
above. The curve of figure 4 shows the probability of
measuring a number of peak to average power ratios
with a typical CATV composite signal. Note that, in
general, 14dB is the practical (but not theoretical) maximum ratio of peak to average power in the composite
signal. Therefore, the design of an amplifier which is to
handle such a signal must have a linear signal handling ability which is 14dB above the average power
and 14+10*Log(n)dB power handling ability over the
power in a single channel.
1.2
1
0.8
0.6
0.4
0.2
0
Chan n
Channel # (Frequency)
Chan 2
Figure 1. CATV Spectrum
1
Time
As an example, consider a 110 channel system where
each carrier is at a level of 50dBmV (0.316VRMS).
Figure 2. Time waveform of signal with zero phase
Channel Power=(0.316)²/75=1.33mW=+1.24dBm.
CompositeAveragePower=110*1.33mW
=146mW=+21.6dBm
Peak Power if each carrier phase is 0° =+43.8dBm
ExpectedNominalPeakPower=10.5+21.6=+32.1dBm
Maximum Peak Power = 14 + 21.6 = 35.6dBm (3.6W)
Time
Figure 3. Time waveform with random carrier phases
Therefore, an amplifier designed to handle this signal
should have the ability to produce +35.6dBm in a linear
fashion - i.e. be able to swing 32.8VPP without clipping.
1
n=77
Relative Number of Occurances
TECHNICAL NOTES
AND ARTICLES
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Maximum Output Voltage Peak = 16.4V (or 32.8VPP)
0.9
0.8
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Four different sources of distortion in an amplifier are
listed below. These issues must be addressed either in
the design of the amplifier or in the choice of technology used to implement the circuit.
0.7
0.6
0.5
0.4
0.3
•
0.2
0.1
0
8
9
10
11
12
13
14
Peak to Average Power Ratio (dB)
Figure 4. Peak to average power ratio for CATV signal
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Voltage compliance: As noted above, an amplifier
which has to handle 110 signals must be capable of
an output peak to peak voltage swing of about 31dB
higher than the peak to peak voltage of each carrier
signal and 40dB higher than the RMS voltage of
each carrier signal. So, if the output is set to
Copyright 1997-2002 RF Micro Devices, Inc.
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•
•
technology. It is a proven technology which uses a
GaAs/AlGaAs heterostructure for producing bipolar
devices which have very high fT, very high Early Voltage, high BVCEO and a capacitance vs. Voltage curve
that is nearly flat. This HBT process is the most reliable
commercially available HBT process in the world and
has been qualified for class S space applications. The
level of ruggedness is absolutely needed for space
applications, but it is also demanded by commercial
applications such as TV distribution systems and cellular base stations which have to be extremely rugged
and reliable.
Drive Current: Although the exponential I-V characteristic of bipolar devices is thought to make the
device nonlinear, experimental results and analysis
prove otherwise. The use of emitter degeneration,
feedback and ample DC collector current permit the
designed to obtain arbitrarily good linearity of GM
(Transconductance). It is necessary to bias the
device at a high enough current level to achieve
suitably linear operation. In general, devices used in
this application have more current drive than would
theoretically be necessary to drive the load at the
required power level. This excess current is used to
provide superior linearity over amplifiers designed
for single carrier applications.
least 107 hours at 125 degrees C.
Distortion due to non-linearity of base-collector
capacitance: Because devices used for multicarrier
systems are, of necessity, fairly large devices
(because of power density), the base-collector
capacitance may be larger than other amplifiers of
the same power level. Ideally, the transistor’s
capacitance is constant, and therefore may cause
roll off in frequency response, but not contribute distortion. Real world devices, however, exhibit a nonlinear capacitance vs. voltage curve which contributes significantly to distortion in the output signal,
particularly when the voltage swing is quite large.
Non-linear ROUT of the transistor: Non-linear
loading of the output by the amplifying device itself
also causes considerable distortion in the output
waveform. Ideally, Rout is high and constant. If it is
not, the loading of the output changes with signal
voltage level, in effect modulating the output signal
and causing distortion.
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The subject amplifiers are fabricated with a GaAs HBT
RF Micro Devices has supplied high volumes of Power
Amplifiers utilizing this process with excellent results.
The HBT process and products built with this process
have been tested to determine that failure rates are at
The main reason that this technology was selected for
this product was that it provides superior linearity and
bandwidth compared to other technology choices. The
desired performance requires amplifying devices which
have a very high transition frequency (fT). The minimum fT is about 15GHz and 20GHz or more is desirable in order to obtain a really flat response with good
return loss through 1GHz. Silicon devices could not be
used for this application because, as the fT of Silicon
Bipolar monolithic transistors increases, a heavy price
is paid in terms of breakdown voltage. The best Si BJT
devices (for ICs) which have an fT in excess of 15GHz,
have a breakdown voltage of 12V (BVCBO) and higher
fT devices result in even less BVCBO. So, there is a limit
to the voltage swing on the output which is too low for
any multicarrier applications. GaAs devices (both HBT
and MESFET) can have Breakdown voltages which
are much higher (18V for MESFET and over 22V for
the HBT) even though they have transition frequencies
well over 20GHz.
For the HBT, whose fT is over 25GHz, the DC collector
voltage can be as high as 10V and the resultant voltage swing is then 20VPP. With a push pull arrangement, the voltage swing doubles to 40VPP which covers
most CATV distribution requirements. This makes the
HBT the best candidate to provide the best voltage
compliance for a wideband amplifier. The second very
real advantage of the HBT is the fact that base - collector capacitance is small and much more constant than
either the Si BJT or the MESFET. Figure 5 shows the
capacitance variation vs. voltage for a Si BJT and an
HBT. Note that the HBT capacitance is nearly constant
with collector voltage. This has a significant effect on
Copyright 1997-2002 RF Micro Devices, Inc.
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TECHNICAL NOTES
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+35dBmV (the RMS voltage of a single signal is
35dB above 1mV=56.2mVRMS), the output swing
must be at least 66dBmV which is a 5.64VPP voltage swing (+/- 2.82V). If the amplifier clips during
this voltage swing, severe distortion results. In practical circuits, there is an emitter degeneration resistor which raises emitter voltage causing early
saturation. It is also necessary to guard band the
saturation of the output device since approaching
VSAT will cause distortion as well. So the DC VCE (or
VDS for FETs) used for the design must be at least
6.5V for this application (or about 3.8V if the collector feed is a choke instead of a resistor). In general,
high transistor breakdown voltages are needed for
these applications in order to prevent voltage compliance distortion.
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57dBmV. Since we are only interested in comparative
data, the signal is left at +60.5dBmV for the comparisons even though all but the RF2317 will not have
good enough performance at the +60.5dBmV level.
The RF2317 obviously has the capability to produce
significantly more than +60.5dBmV per channel since
its IM products are down about 65dB. In fact, it should
meet DIN requirements up to a level of +63dBmV.
linearity as will be shown later.
160
140
Silicon
120
Ccb (fF)
100
80
HBT
60
10
0
40
-10
20
dBm
-20
0
0
1
2
3
4
5
6
-30
-50
-60
Figure 5. Ccb for HBT and Silicon Bipolar
-70
720
TECHNICAL NOTES
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RF2312
-40
Collector Base Voltage
730
740
750
760
770
780
Frequency (MHz)
Figure 6. IM distortion with Three-Tone Input
10
0
-10
dBm
RF Micro Devices has approached the marketplace for
RF products using a philosophy of Optimum Technology Matching (OTM) which uses the best technology at
our disposal (GaAs MESFET, Silicon Bipolar, CMOS,
and GaAs HBT) to address the needs of the market. In
other words, we don’t restrict the selection of the
technology utilized, a priori, but instead, seek that
which will yield the most effective solution for our customers. In this case, the overriding concern is to maximize the linearity of the amplifier with minimum power
consumption, and of course, do so at a reasonable
price. Over the years, our experience is that the most
linear amplifiers we can build are invariably GaAs HBT
devices. In order to illustrate the performance differences and show why we selected HBT for ultra linear
amplifier products, three different parts have been simulated for comparison - two HBT parts, a Si BJT part,
and a GaAs MESFET part. The modeling used to simulate these devices is very accurate and has been
tested in practice for many types of circuit. One of the
HBT parts is the RF2312, and the other is the RF2317.
We have laboratory confirmation of the accuracy of the
simulation since each of these parts have been tested
-20
-30
-40
-50
-60
-70
720
730
740
750
760
770
780
Frequency (MHz)
Figure 7. IM Distortion with Three-Tone Input, Silicon device
10
0
-10
-20
dBm
15
RF2317
-30
-40
-50
Figure 6 shows distortion in the RF2312 and RF2317
which use HBT devices. The test tones are those used
for DIN4500B testing which is a standard test used for
European TV systems. The large tone is at a frequency
of 736MHz. The smaller tones are each 6dB below the
large tone, and the frequency offsets are 18 and
24MHz, respectively. Normally, the DIN rating is the
highest level for the large signal which produces distortion products which are -60dBc. As seen in figure 6,
the RF2312 distortion is -53dBc for the +60.5dBmV
level used for the test. In order to obtain -60dBc, the
level of the large signal must be decreased to about
15-94
-60
-70
720
730
740
750
760
770
780
Frequency (MHz)
Figure 8. IM Distortion with Three-Tone Input, GaAs
MESFET device
The spectrum of figure 7 is the output of the same circuit as the HBT parts except that Silicon Bipolar transistors are used. The transistors are actually
hypothetical transistors, in that they have identical
Copyright 1997-2002 RF Micro Devices, Inc.
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&KLS'HVLJQ
The RF2312 and RF2317 are a simple Darlington
“Cascadable Gain Block” circuit design as shown in
figure 9. This is a broadband feedback amplifier which
has its inputs and outputs matched to 75Ohms by virtue of the feedback resistor. Gain is determined by the
feedback resistor (RFB) and the emitter resistor (RE2).
These resistors along with resistor RB also set the DC
bias conditions on the part. The output is connected to
the power supply through an inductor (to provide an RF
open circuit) and a series resistor which sets the current through the device. The value of the resistor is
determined by calculating the necessary voltage drop
at the desired current level. The voltage drop is the difference between the power supply voltage and the
device voltage as given in figure 10. For example, if the
Output
/Power
RFB
Input
Q1
Q2
RB
RE1
RE2
Figure 9. Darlington gain block
5.6
RF2312
5.5
5.4
5.3
5.2
5.1
5
4.9
4.8
40
50
60
70
80
90
100
110
120
Device Current (mA)
Figure 10. Device voltage versus device current
The inductor value is set such that the reactance at the
lowest frequency of operation is at least 200 Ohms.
The input and output coupling capacitors are set such
that their reactance at the lowest frequency of operation is less than 10Ohms.
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A summary of the measured performance of the
RF2312 and RF2317 is shown in table 1. Note that the
table reflects measurements done in a 50Ohm system.
This is because we were equipped for only 50 Ohm
measurements for some of the parameters. So for consistency, all of the performance measures are listed as
50 Ohm measurements. To perform all of the measurements with laboratory test equipment, a 50 to 75Ohm
transformer must be connected to both the input and
output ports. Unfortunately, the transformers have
Copyright 1997-2002 RF Micro Devices, Inc.
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TECHNICAL NOTES
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Figure 8 shows the same circuit using GaAs MESFET
devices for the active elements of the design. Again,
the bias conditions are identical. In addition, the sizing
(Gate width) and pinch off voltage for these FETs were
optimized for best performance. This circuit is not
hypothetical in the sense used above, because it is
possible to physically realize the circuit. The linearity of
this circuit suffers because of varying capacitance like
the Silicon device, and also because of the strongly
non-linear output conductance of MESFETs. In addition, because of the lower GM for the MESFET, no
source degeneration is possible. In fact, it proved
impossible to obtain the same gain as is possible with
the HBT and Si BJT implementations, so the input signals had to be increased to get the same output power.
The results are clearly very inferior to both the Si Bipolar design and the HBT. So, while it is possible to obtain
MESFET devices which have sufficient breakdown and
fT, they are not a good choice for high linearity amplifiers because gain is lower and distortion is higher.
power supply voltage is to be 10V and the device current is desired to be 100mA, consulting figure 10, we
see that the device voltage is 5.4V and the drop across
the external resistor must be 10-5.4=4.6V. The current
is 100mA, so R=4.6/.1 = 46Ohms. In this manner, the
desired bias conditions may be set up for any power
supply voltage from about 7V to 12V. Voltages outside
of that range are not recommended.
Device Voltage
properties as the HBT circuit except for the base - collector capacitance which is set to have Capacitance vs.
Voltage variations like the Silicon transistor shown in
figure 5. The bias conditions, both current and voltage
are identical to those used for the RF2312 simulation.
This Silicon circuit is a very optimistic case since no
Silicon BJT IC process exists which has the combination of Early Voltage, fT, and BVCEO which is associated with the HBT device. But, if one could find a
Silicon process with these attributes the result would
be as shown. Note that the distortion products
increase by approximately 6dB, because, and only
because, the base - collector capacitance varies with
voltage. The degradation in performance shows clearly
the advantages of the HBT over any process which has
a significant variation in Capacitance vs. Voltage.
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some loss and don’t do a good job over the full bandwidth of the part - hence the 50Ohm measurements.
Because the imaginary parts of ZIN and ZOUT are very
low, the VSWR is pretty good in both 50Ohm and
75Ohm systems, so 50Ohm measurements were performed for the table.
RF2312
RF2317
(expected)
Gain
Gain Flatness
Bandwidth (3dB)
Noise Figure
PMAX @ 500MHz
OIP3 @ 500MHz
OIP2
Input VSWR
Output VSWR
ICC
VD
15.5dB
±0.3dB
2.5GHz
3.8dB
+21dBm
+38dBm
+57dBm
1.4:1
1.6:1
100mA
5.4V
14.5dB
±0.5dB
1.8GHz
5dB
+26dBm
+44dBm
+63dBm
1.2:1
1.3:1
200mA
8.0V
OIP3 (dBm)
35
34
33
32
31
600
700
800
900
Frequency (MHz)
TECHNICAL NOTES
AND ARTICLES
500
600
700
800
900
compression point (P1dB) of the RF2312 vs. Frequency. Note that power handling ability does
decrease as frequency increases. This is inherent in
the Darlington configuration. It is the result of Beta
decreasing in the second transistor (Q2) which forces
the emitter follower stage (Q1) to provide more current
at higher frequencies. A lower fT, as may be found in a
Si device, aggravates that situation - i.e. the higher the
fT of the devices, the better. Distortion resulting from
this effect can be decreased by increasing the collector
current of the first stage (lower the value of RE1). This,
of course, increases overall power consumption. The
current level set in the RF2312 is set to achieve the
performance goals for the part, and yield good multicarrier performance up to 1 GHz.
36
Figure 11. OIP3 versus frequency
A concern with any wideband amplifier of this sort is its
stability - particularly when driving filters and severely
unmatched loads. The HBT parts have been tested
under a variety of conditions using various bandpass
and lowpass filters without evidence of any oscillations.
The parts has also been driven and loaded simultaneously with 10:1 tuners and showed no unstable
properties.
Figures 11 and 12 show the Output Third Order Intercept Point (OIP3) and Power Output at the 1 dB gain
15-96
400
Figure 12. P1dB versus frequency
37
500
300
Frequency (MHz)
38
400
20
18.5
200
39
300
20.5
19
40
15
21
19.5
Table 1. Chip performance
30
200
21.5
P(1dB) dBm
Typical
Performance
22
Figure 15 shows the frequency response of the
RF2312 mounted on a PCB with approximately 3 inch
microstrip traces. Note that while the amplifier has a
very flat frequency response, parasitic elements associated with the construction of the circuit board roll the
response off slightly at the higher frequencies. Generally, this roll off is of the order of a few tenths of a dB if
care is taken in the layout of the PCB. One frequently
overlooked aspect of the layout is the ground traces. To
obtain a flat frequency response, a ground plane
should be placed directly under the amplifier and the
ground pins soldered directly to it. If the opposite side
of the PCB is used for ground plane, multiple vias
should be placed beneath each ground pin. Input and
output traces should be 50 or 75Ohm microstrip and
high quality, high frequency coupling capacitors and
inductors should be used.
Copyright 1997-2002 RF Micro Devices, Inc.
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frequency response. Essentially, a series L-R is added
to the output circuit in parallel with the VCC feed L-R
network. This loads the output at the lower frequencies
hence reducing the gain. As frequency goes up, however, the loading is reduced because the reactance of
the inductor goes up with frequency. This circuit may
be used with any Darlington amplifier which has adequate bandwidth, including the RF2317 and the
RF2316. The output return loss is also improved with
the addition of this circuit.
OUT1
Q3
RF1
600 Ω
IN1
Q1
RE1
2.0
VB2
RE2
2.0
IN2
Q2
R4
600 Ω
Q4
OUT2
Figure 13. Cascade push-pull
VB1
L1
1 µH
C2
1 nF
IN
VCC
L2
1 µH
C3
1 nF
L6
1 µH
L5
1 µH
IN1 OUT1
RF2316
IN2
C1
1 nF
OUT2
C4
1 nF
VCC
VCC
OUT
Figure 14. External connections for the RF2316
16
Without Upslope Compensation
15
13
With compensation
12
11
10
100
200
300
400
500
600
700
800
900
1000
Frequency (MHz)
Figure 15. Gain versus frequency
VCC
R1
150
C2
1 nF
IN
RF2312
RF2317
RCS
30
L1
L2
15 nH 1 µ H
C1
1 nF
OUT
Figure 16. Circuit for high frequency up-slope
compensation
In some applications, however, even a few tenths of a
dB can be a problem. In that case, the circuit of figure
16 can be used to provide a positive “upslope” to the
The RF2316 is a balanced amplifier. It uses external
baluns which are easy to wind and have a minimal
impact on frequency response and power. The connections external to the chip are shown in figure 14. The
unit will produce an output of +44dBmV per channel
with acceptable distortion products.
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The HBT amplifiers offer the best frequency response
and the highest linearity (for a given DC power consumption) of any commercially available integrated circuit. They offer the small size and the inherent
reliability and repeatability of a monolithic design. OIP3
numbers up to +48dBm have been achieved and the
frequency response is extremely flat across a full Gigahertz of bandwidth for all the mentioned parts. They
operate from a single power supply, which can range
from 7V to 12V. Pricing is less than $4.00 for the
RF2312 and RF2317.
Copyright 1997-2002 RF Micro Devices, Inc.
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TECHNICAL NOTES
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Gain (dB)
14
3XVK3XOO&RQILJXUDWLRQ
Figure 13 shows a partial schematic of a Push Pull
Cascode circuit using HBT devices. This architecture is
typically used for high power distribution amplifiers
which produce high output voltages. It has an inherent
advantage over singe ended designs in that the output
voltage swing can be double that of the single ended
design (providing 4 times the power) and, importantly,
the even order distortion products are reduced from a
single ended design. That is, for a given collector current and voltage, the balanced design produces the
same odd order results (same IP3), but even order distortion is reduced typically 10 dB (better IP2) - depending on the balance achieved. The disadvantage of this
circuit configuration is that it requires transformers or
baluns in order to convert the single ended input and
output signals to differential balanced signals at the
ports of the active device. These passive devices have
power loss which must be made up by the amplifier.
The frequency response is also adversely affected.
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AND ARTICLES
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Copyright 1997-2002 RF Micro Devices, Inc.