19-0160; Rev 2; 4/97 KIT ATION EVALU LUDED C IN TION A M R O INF Dual-Output Power-Supply Controller for Notebook Computers ________________________________Features The MAX786 is a system-engineered power-supply controller for notebook computers or similar batterypowered equipment. It provides two high-performance step-down (buck) pulse-width modulators (PWMs) for +3.3V and +5V. Other features include dual, low-dropout, micropower linear regulators for CMOS/RTC back-up, and two precision low-batterydetection comparators. High efficiency (95% at 2A; greater than 80% at loads from 5mA to 3A) is achieved through synchronous rectification and PWM operation at heavy loads, and Idle ModeTM operation at light loads. The MAX786 uses physically small components, thanks to high operating frequencies (300kHz/200kHz) and a new current-mode PWM architecture that allows for output filter capacitors as small as 30µF per ampere of load. Line- and loadtransient responses are terrific, with a high 60kHz unitygain crossover frequency allowing output transients to be corrected within four or five clock cycles. Low system cost is achieved through a high level of integration and the use of low-cost, external N-channel MOSFETs. ♦ Dual PWM Buck Controllers (+3.3V and +5V) Other features include low-noise, fixed-frequency PWM operation at moderate to heavy loads, and a synchronizable oscillator for noise-sensitive applications such as electromagnetic pen-based systems and communicating computers. The MAX786 is a monolithic, BiCMOS IC available in fine-pitch, surface-mount SSOP packages. ___________________________Applications Notebook Computers Portable Data Terminals Communicating Computers Pen-Entry Systems ♦ Two Precision Comparators or Level Translators ♦ 95% Efficiency ♦ 420µA Quiescent Current, 70µA in Standby (linear regulators alive) ♦ 25µA Shutdown Current (+5V linear alive) ♦ 5.5V to 30V Input Range ♦ Small SSOP Package ♦ Fixed Output Voltages: 3.3V (standard) 3.45V (High-Speed Pentium™) 3.6V (PowerPC™) _________________Ordering Information PART TEMP. RANGE VOUT 0°C to +70°C 28 SSOP 3.3V MAX786RCAI 0°C to +70°C 28 SSOP 3.45V Ordering Information continued at end of data sheet. _____________________Pin Configuration TOP VIEW CS3 1 28 FB3 SS3 2 27 DH3 ON3 3 26 LX3 D1 4 D2 5 ________Typical Application Diagram PIN-PACKAGE MAX786CAI 25 BST3 MAX786 24 DL3 VH 6 23 V+ Q2 7 22 VL Q1 8 21 FB5 GND 9 20 PGND REF 10 19 DL5 +3.3V 5.5V TO 30V SHUTDOWN 5V ON/OFF 3.3V ON/OFF SYNC POWER SECTION µP MEMORY +5V PERIPHERALS MAX786 POWER-GOOD LOW-BATTERY WARNING SUSPEND POWER SYNC 11 18 BST5 SHDN 12 17 LX5 ON5 13 16 DH5 SS5 14 15 CS5 SSOP Idle Mode is a trademark of Maxim Integrated Products. Pentium is a trademark of Intel Corp. PowerPC is a trademark of IBM Corp. _______________________________________________________________________ Maxim Integrated Products 1 For free samples & the latest literature: http://www.maxim-ic.com, or phone 1-800-998-8800. For small orders, phone 1-800-835-8769. MAX786 __________________General Description MAX786 Dual-Output Power-Supply Controller for Notebook Computers ABSOLUTE MAXIMUM RATINGS V+ to GND................................................................-0.3V to 36V PGND to GND .......................................................................±2V VL to GND ..................................................................-0.3V to 7V BST3, BST5 to GND .................................................-0.3V to 36V LX3 to BST3 ...............................................................-7V to 0.3V LX5 to BST5 ...............................................................-7V to 0.3V Inputs/Outputs to GND (D1, D2, SHDN, ON5, REF, SS5, CS5, FB5, SYNC, CS3,FB3, SS3, ON3) ............-0.3V to (VL + 0.3V) VH to GND ...............................................................-0.3V to 20V Q1, Q2 to GND ............................................-0.3V to (VH + 0.3V) DL3, DL5 to PGND .......................................-0.3V to (VL + 0.3V) DH3 to LX3 ..............................................-0.3V to (BST3 + 0.3V) DH5 to LX5 ..............................................-0.3V to (BST5 + 0.3V) REF, VL Short to GND................................................Momentary REF Current ........................................................................20mA VL Current ..........................................................................50mA Continuous Power Dissipation (TA = +70°C) SSOP (derate 9.52mW/°C above +70°C)....................762mW Operating Temperature Ranges MAX786CAI/MAX786_CAI .................................0°C to +70°C MAX786EAI/MAX786_EAI ...............................-40°C to +85°C Lead Temperature (soldering, 10sec) ............................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (V+ = 15V, GND = PGND = 0V, IVL = IREF = 0mA, SHDN = ON3 = ON5 = 5V, other digital input levels are 0V or +5V, TA = TMIN to TMAX, unless otherwise noted.) PARAMETER 3.3V AND 5V STEP-DOWN CONTROLLERS CONDITIONS MIN Input Supply Range TYP 5.5 FB5 Output Voltage 0mV < (CS5-FB5) < 70mV, 6V < V + < 30V (includes load and line regulation) FB3 Output Voltage 0mV < (CS3-FB3) < 70mV, 6V < V + < 30V (includes load and line regulation) MAX786 MAX786R MAX786S Load Regulation Either controller (CS_ -FB_ = 0mV to 70mV) Line Regulation Either controller (V+ = 6V to 30V) Current-Limit Voltage CS3-FB3 or CS5-FB5 SS3/SS5 Source Current SS3/SS5 Fault Sink Current INTERNAL REGULATOR AND REFERENCE VL Output Voltage ON5 = ON3 = 0V, 5.5V < V+ < 30V, 0mA < IL < 25mA VL Fault Lockout Voltage Falling edge, hysteresis = 1% VL/FB5 Switchover Voltage Rising edge of FB5, hysteresis = 1% REF Output Voltage No external load (Note 1) REF Fault Lockout Voltage Falling edge REF Load Regulation 0mA < IL < 5mA (Note 2) –———– SHDN = D1 = D2 = ON3 = ON5 = 0V, V+ = 30V V+ Shutdown Current V+ Standby Current D1 = D2 = ON3 = ON5 = 0V, V+ = 30V MAX UNITS 30 V 4.80 5.08 5.20 V 3.17 3.32 3.46 3.35 3.50 3.65 2.5 0.03 100 4.0 3.46 3.60 3.75 V 80 2.5 2 4.5 3.6 4.2 3.24 2.4 120 6.5 % %/V mV µA mA 30 25 70 5.5 4.2 4.7 3.36 3.2 75 40 120 V V V V V mV µA µA Quiescent Power Consumption (both PWM controllers on) D1 = D2 = 0V, FB5 = CS5 = 5.25V, FB3 = CS3 = 3.5V 5.5 8.6 mW V+ Off Current COMPARATORS D1, D2 Trip Voltage D1, D2 Input Current FB5 = CS5 = 5.25V, VL switched over to FB5 30 60 µA 1.69 ±100 V nA 2 Falling edge, hysteresis = 1% D1 = D2 = 0V, 5V 1.61 ________________________________________________________________________________________________ Dual-Output Power-Supply Controller for Notebook Computers (V+ = 15V, GND = PGND = 0V, IVL = IREF = 0mA, SHDN = ON3 = ON5 = 5V, other digital input levels are 0V or +5V, TA = TMIN to TMAX, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS 12 20 30 µA VH = 15V, VOUT = 2.5V 200 500 1000 µA ISOURCE = 5µA, VH = 3V VH - 0 .5 Q1, Q2 Source Current VH = 15V, VOUT = 2.5V Q1, Q2 Sink Current Q1, Q2 Output High Voltage Q1, Q2 Output Low Voltage ISINK = 20µA, VH = 3V Quiescent VH Current VH = 18V, D1 = D2 = 5V, no external load V 0.4 V 4 10 µA OSCILLATOR AND INPUTS/OUTPUTS Oscillator Frequency SYNC = 3.3V 270 300 330 SYNC = 0V, 5V 170 200 230 kHz SYNC High Pulse Width 200 ns SYNC Low Pulse Width 200 ns SYNC Rise/Fall Time Not tested Oscillator SYNC Range Maximum Duty Cycle Input Low Voltage 240 SYNC = 3.3V 89 92 SYNC = 0V or 5V 92 95 SHDN, ON3, ON5, SYNC SHDN, ON3, ON5 Input High Voltage SYNC 200 ns 350 kHz % 0.8 2.4 V V VL - 0 .5 Input Current SHDN, ON3, ON5 VIN = 0V, 5V DL3/DL5 Sink/Source Current VOUT = 2V 1 ±1 µA A DH3/DH5 Sink/Source Current BST3-LX3 = BST5-LX5 = 4.5V, VOUT = 2V 1 A DL3/DL5 On-Resistance High or low 7 Ω DH3/DH5 On-Resistance High or low, BST3-LX3 = BST5-LX5 = 4.5V 7 Ω Note 1: Since the reference uses VL as its supply, its V+ line regulation error is insignificant. Note 2: The main switching outputs track the reference voltage. Loading the reference reduces the main outputs slightly according to the closed-loop gain (AVCL) and the reference voltage load-regulation error. AVCL for the +3.3V supply is unity gain. AVCL for the +5V supply is 1.54. _________________________________________________________________________________________________ 3 MAX786 ELECTRICAL CHARACTERISTICS (continued) ________________________________________________Typical Operating Characteristics (Circuit of Figure 1, TA = +25°C, unless otherwise noted.) EFFICIENCY vs. +5V OUTPUT CURRENT, 200kHz EFFICIENCY vs. +5V OUTPUT CURRENT, 300kHz VIN = 30V 80 70 80 VIN = 30V 70 +3.3V OFF SYNC = 0V, +3.3V OFF 80 50 10m 100m 1 +5V OUTPUT CURRENT (A) 10 50 1m 10m 100m 1 +5V OUTPUT CURRENT (A) 100 VIN = 6V VIN = 30V 70 +5V ON 60 50 100m 1 18 ON3 = ON5 = HIGH 2 1 6 12 18 24 SUPPLY VOLTAGE (V) SHUTDOWN SUPPLY CURRENT vs. SUPPLY VOLTAGE MINIMUM VIN TO VOUT DIFFERENTIAL vs. +5V OUTPUT CURRENT SHDN = 0V 200 100 0 6 12 18 24 SUPPLY VOLTAGE (V) 30 6 12 18 SUPPLY VOLTAGE (V) 24 30 1000 0.8 300kHz 0.6 200kHz 0.4 +5V OUTPUT STILL REGULATING 0.2 0 0 0.5 SWITCHING FREQUENCY vs. LOAD CURRENT SWITCHING FREQUENCY (kHz) MINIMUM VIN TO VOUT DIFFERENTIAL (V) 300 ON3 = ON5 = 0V 1.0 0 1.0 400 1.5 30 +3.3V OUTPUT CURRENT (A) 500 2.0 0 0 10 10 2.5 0 10m 1 STANDBY SUPPLY CURRENT vs. SUPPLY VOLTAGE STANDBY SUPPLY CURRENT (mA) QUIESCENT SUPPLY CURRENT (mA) 90 100m +3.3V OUTPUT CURRENT (A) 19 VIN = 15V 1m 10m 1m 10 QUIESCENT SUPPLY CURRENT vs. SUPPLY VOLTAGE EFFICIENCY vs. +3.3V OUTPUT CURRENT, 300kHz 80 SYNC = 0V, +5V ON 60 50 1m VIN = 30V 70 60 60 EFFICIENCY (%) VIN = 6V VIN = 15V EFFICIENCY (%) VIN = 15V 90 VIN = 6V 90 EFFICIENCY (%) EFFICIENCY (%) 100 VIN = 15V VIN = 6V 90 4 EFFICIENCY vs. +3.3V OUTPUT CURRENT, 200kHz 100 100 SHUTDOWN SUPPLY CURRENT (µA) MAX786 Dual-Output Power-Supply Controller for Notebook Computers 1m 10m 100m 1 +5V OUTPUT CURRENT (A) 10 SYNC = REF (300kHz) ON3 = ON5 = 5V 100 10 +5V, VIN = 7.5V +5V, VIN = 30V +3.3V, VIN = 7.5V 1 0.1 100µ 1m 10m 100m LOAD CURRENT (A) _______________________________________________________________________________________ 1 Dual-Output Power-Supply Controller for Notebook Computers PULSE-WIDTH MODULATION MODE WAVEFORMS IDLE MODE WAVEFORMS +5V OUTPUT 50mV/div LX 10V/div 2V/div +5V OUTPUT 50mV/div 500ns/div +5V OUTPUT CURRENT = 1A VIN= 16V 200µs/div ILOAD = 100mA VIN = 10V +5V LOAD-TRANSIENT RESPONSE +3.3V LOAD-TRANSIENT RESPONSE 3A 0A 3A LOAD CURRENT 0A +5V OUTPUT 50mV/div 200µs/div VIN = 15V LOAD CURRENT +3.3V OUTPUT 50mV/div 200µs/div VIN = 15V _______________________________________________________________________________________ 5 MAX786 _________________________________Typical Operating Characteristics (continued) (Circuit of Figure 1, TA = +25°C, unless otherwise noted.) MAX786 Dual-Output Power-Supply Controller for Notebook Computers _________________________________Typical Operating Characteristics (continued) (Circuit of Figure 1, TA = +25°C, unless otherwise noted.) +5V LINE-TRANSIENT RESPONSE, FALLING +5V LINE-TRANSIENT RESPONSE, RISING +5V OUTPUT 50mV/div +5V OUTPUT 50mV/div VIN, 10V TO 16V 2V/div VIN, 16V TO 10V 2V/div 20µs/div ILOAD = 2A 20µs/div ILOAD = 2A +3.3V LINE-TRANSIENT RESPONSE, RISING +3.3V LINE-TRANSIENT RESPONSE, FALLING +3.3V OUTPUT 50mV/div +3.3V OUTPUT 50mV/div VIN, 10V TO 16V 2V/div VIN, 16V TO 10V 2V/div 20µs/div ILOAD = 2A 6 20µs/div ILOAD = 2A _______________________________________________________________________________________ Dual-Output Power-Supply Controller for Notebook Computers PIN NAME 1 CS3 Current-sense input for +3.3V; +100mV = current limit level referenced to FB3. FUNCTION 2 SS3 Soft-start input for +3.3V. Ramp time to full current limit is 1ms/nF of capacitance to GND. 3 ON3 ON/OFF control input disables the +3.3V PWM. Tie directly to VL for automatic start-up. 4 D1 #1 level-translator/comparator noninverting input, threshold = +1.650V. Controls Q1. Tie to GND if unused. 5 D2 #2 level-translator/comparator noninverting input (see D1) 6 VH External positive supply-voltage input for the level translators/comparators 7 Q2 #2 level-translator/comparator output. Sources 20µA from VH when D2 is high. Sinks 500µA to GND when D2 is low, even with VH = 0V. 8 Q1 #1 level translator/comparator output (see Q2) 9 GND Low-current analog ground 10 REF 3.3V reference output. Sources up to 5mA for external loads. Bypass to GND with 1µF/mA of load or 0.22µF minimum. 11 SYNC 12 SHDN Shutdown control input, active low. Tie to VL for automatic start-up. The 5V VL supply stays active in shutdown, but all other circuitry is disabled. Do not force SHDN higher than VL + 0.3V. 13 ON5 ON/OFF control input disables the +5V PWM supply. Tie to VL for automatic start-up. 14 SS5 Soft-start control input for +5V. Ramp time to full current limit is 1ms/nF of capacitance to GND. Oscillator control/synchronization input. Connect to VL or GND for 200kHz; connect to REF for 300kHz. For external clock synchronization in the 240kHz to 350kHz range, a high-to-low transition causes a new cycle to start. 15 CS5 Current-sense input for +5V; +100mV = current-limit level referenced to FB5. 16 DH5 Gate-drive output for the +5V high-side MOSFET 17 LX5 Inductor connection for the +5V supply 18 BST5 Boost capacitor connection for the +5V supply (0.1µF) 19 DL5 Gate-drive output for the +5V low-side MOSFET 20 PGND 21 FB5 Power ground Feedback and current-sense input for the +5V PWM 22 VL 5V logic supply voltage for internal circuitry. VL is always on and can source 5mA for external loads. 23 V+ Supply voltage input from battery, 5.5V to 30V 24 DL3 Gate-drive output for the +3.3V low-side MOSFET 25 BST3 Boost capacitor connection for the +3.3V supply (0.1µF) 26 LX3 27 DH3 Gate-drive output for the +3.3V high-side MOSFET 28 FB3 Feedback and current-sense input for the +5V PWM Inductor connection for the +3.3V supply _______________________________________________________________________________________ 7 MAX786 _______________________________________________________________________Pin Description MAX786 Dual-Output Power-Supply Controller for Notebook Computers _________________Detailed Description The MAX786 has two close relatives: the MAX782 and the MAX783. The MAX782 and MAX783 each include an extra flyback winding regulator and linear regulators for dual, +12V/programmable PCMCIA VPP outputs. The MAX782/MAX783 data sheet contains extra applications information on the MAX786 not found in this data sheet. The MAX786 converts a 5.5V to 30V input to four outputs (Figure 1). It produces two high-power, PWM, switchmode supplies, one at +5V and the other at +3.3V. The two supplies operate at either 300kHz or 200kHz, allowing for small external components. Output current capability depends on external components, and can exceed 6A on each supply. An internal 5V, 5mA supply (VL) and a 3.3V, 5mA reference voltage are also generated via linear regulators, as shown in Figure 2. Fault protection circuitry shuts off the PWMs when the internal supplies lose regulation. Two precision voltage comparators are also included. Their output stages permit them to be used as level translators for driving external N-channel MOSFETs in load-switching applications, or for more conventional logic signals. INPUT 5.5V TO 30V (NOTE 1) C10 33µF C1 33µF 23 C5 0.1µF R1 25mΩ L1 10µH D1 1N5819 N1 22 V+ D2A 1N4148 +3.3V AT 3A +3.3V Switch-Mode Supply The +3.3V supply is generated by a current-mode, PWM step-down regulator using two N-channel MOSFETs, a rectifier, and an LC output filter (Figure 1). The gate-drive signal to the high-side MOSFET, which must exceed the battery voltage, is provided by a boost circuit that uses a 100nF capacitor connected to BST3. 25 27 VL BST3 BST5 DH3 DH5 LX3 LX5 26 N3 C7 C12 150µF 150µF (NOTE 2) C9 0.01µF +3.3V ON/OFF +5V ON/OFF SHUTDOWN OSC SYNC NOTE 1: INPUT VOLTAGE RANGE 6.5V TO 30V AS SHOWN. SEE LOW-VOLTAGE (6-CELL) OPERATION SECTION FOR DETAILS. NOTE 2: USE SHORT, KELVIN-CONNECTED PC BOARD TRACES PLACED VERY CLOSE TO ONE ANOTHER. 24 DL3 1 CS3 28 FB3 2 3 13 12 11 SS3 ON3 ON5 SHDN SYNC GND 9 DL5 MAX786 +5V AT 5mA 18 16 D2B 1N4148 C4 0.1µF 4.7µF N2 17 19 CS5 15 21 FB5 14 SS5 6 VH 4 D1 8 Q1 5 D2 7 Q2 L2 10µH R2 25m D3 1N5819 N4 (NOTE 2) C8 0.01µF COMPARATOR SUPPLY INPUT IN COMPARATOR 1 OUT IN COMPARATOR 2 OUT REF PGND 10 20 +3.3V AT 5mA C3 1µF Figure 1. MAX786 Application Circuit 8 _______________________________________________________________________________________ +5V AT 3A C6 330µF Dual-Output Power-Supply Controller for Notebook Computers +5V LDO LINEAR REGULATOR V+ VL Programmable soft-start is set by an optional external capacitor; this reduces in-rush surge currents upon start-up and provides adjustable power-up times for power-supply sequencing purposes. P FB3 3.3V PWM CONTROLLER (SEE FIG. 3) 5V 3.3V 4.5V +3.3V REFERENCE ON REF CS3 BST3 DH3 LX3 DL3 ON SS3 SHDN PGND GND 4V FAULT ON3 FB5 CS5 2.8V SYNC 300kHz/200kHz OSCILLATOR ON STANDBY 5V PWM CONTROLLER (SEE FIG. 3) BST5 DH5 LX5 DL5 ON SS5 ON5 VH D1 Q1 1.65V D2 Q2 1.65V Figure 2. MAX786 Block Diagram _______________________________________________________________________________________ 9 MAX786 A synchronous rectifier at LX3 keeps efficiency high by clamping the voltage across the rectifier diode. Maximum current limit is set by an external low-value sense resistor, which prevents excessive inductor current during start-up or under short-circuit conditions. MAX786 Dual-Output Power-Supply Controller for Notebook Computers +5V Switch-Mode Supply The +5V output is produced by a current-mode, PWM step-down regulator, which is nearly identical to the +3.3V supply. The +5V supply’s dropout voltage, as configured in Figure 1, is typically 400mV at 2A. As V+ approaches 5V, the +5V output gracefully falls with V+ until the VL regulator output hits its undervoltagelockout threshold at 4V. At this point, the +5V supply turns off. The default frequency for both PWM controllers is 300kHz (with SYNC connected to REF), but 200kHz may be used by connecting SYNC to GND or VL. +3.3V and +5V PWM Buck Controllers The two current-mode PWM controllers are identical except for different preset output voltages (Figure 3). Each PWM is independent except for being synchronized to a master oscillator and sharing a common reference (REF) and logic supply (VL). Each PWM can be turned on and off separately via ON3 and ON5. The PWMs are a direct-summing type, lacking a traditional integrator error amplifier and the phase shift associated with it. They therefore do not require any external feedback compensation components if the filter capacitor ESR guidelines given in the Design Procedure are followed. The main gain block is an open-loop comparator that sums four input signals: an output voltage error signal, current-sense signal, slope-compensation ramp, and precision voltage reference. This direct-summing method approaches the ideal of cycle-by-cycle control of the output voltage. Under heavy loads, the controller operates in full PWM mode. Every pulse from the oscillator sets the output latch and turns on the high-side switch for a period determined by the duty cycle (approximately VOUT/VIN). As the high-side switch turns off, the synchronous rectifier latch is set and, 60ns later, the low-side switch turns on (and stays on until the beginning of the next clock cycle, in continuous mode, or until the inductor current crosses through zero, in discontinuous mode). Under fault conditions where the inductor current exceeds the 100mV current-limit threshold, the high-side latch is reset and the high-side switch is turned off. At light loads, the inductor current fails to exceed the 25mV threshold set by the minimum current comparator. When this occurs, the PWM goes into idle mode, skipping most of the oscillator pulses in order to reduce the switching frequency and cut back switching losses. The oscillator is effectively gated off at light loads because the minimum current comparator immediately resets the high-side latch at the beginning of each cycle, unless the FB_ signal falls below the reference voltage level. 10 Soft-Start/SS_ Inputs Connecting capacitors to SS3 and SS5 allows gradual build-up of the +3.3V and +5V supplies after ON3 and ON5 are driven high. When ON3 or ON5 is low, the appropriate SS capacitors are discharged to GND. When ON3 or ON5 is driven high, a 4µA constant current source charges these capacitors up to 4V. The resulting ramp voltage on the SS_ pins linearly increases the current-limit comparator setpoint so as to increase the duty cycle to the external power MOSFETs up to the maximum output. With no SS capacitors, the circuit will reach maximum current limit within 10µs. Soft-start greatly reduces initial in-rush current peaks and allows start-up time to be programmed externally. Synchronous Rectifiers Synchronous rectification allows for high efficiency by reducing the losses associated with the Schottky rectifiers. When the external power MOSFET N1 (or N2) turns off, energy stored in the inductor causes its terminal voltage to reverse instantly. Current flows in the loop formed by the inductor, Schottky diode, and load — an action that charges up the filter capacitor. The Schottky diode has a forward voltage of about 0.5V which, although small, represents a significant power loss, degrading efficiency. A synchronous rectifier, N3 (or N4), parallels the diode and is turned on by DL3 (or DL5) shortly after the diode conducts. Since the on resistance (rDS(ON)) of the synchronous rectifier is very low, the losses are reduced. The synchronous rectifier MOSFET is turned off when the inductor current falls to zero. Cross conduction (or “shoot-through”) occurs if the high-side switch turns on at the same time as the synchronous rectifier. The MAX786’s internal break-beforemake timing ensures that shoot-through does not occur. The Schottky rectifier conducts during the time that neither MOSFET is on, which improves efficiency by preventing the synchronous-rectifier MOSFET’s lossy body diode from conducting. The synchronous rectifier works under all operating conditions, including discontinuous-conduction and idle mode. Boost Gate-Driver Supply Gate-drive voltage for the high-side N-channel switch is generated with a flying-capacitor boost circuit as shown in Figure 4. The capacitor is alternately charged from the VL supply via the diode and placed in parallel with the high-side MOSFET’s gate-source terminals. On startup, the synchronous rectifier (low-side) MOSFET forces LX_ to 0V and charges the BST_ capacitor to 5V. On the ______________________________________________________________________________________ Dual-Output Power-Supply Controller for Notebook Computers Ringing seen at the high-side MOSFET gates (DH3 and DH5) in discontinuous-conduction mode (light loads) is a natural operating condition caused by the residual energy in the tank circuit formed by the inductor and stray capacitance at the LX_ nodes. The gate driver negative rail is referred to LX_, so any ringing there is directly coupled to the gate-drive supply. CS_ 1X 60kHz LPF FB_ MAIN PWM COMPARATOR REF, 3.3V (OR INTERNAL 5V REFERENCE) Σ BST_ R LEVEL SHIFT Q S DH_ LX_ SLOPE COMP OSC MINIMUM CURRENT (IDLE MODE) 25mV VL 4µA CURRENT LIMIT SHOOTTHROUGH CONTROL 0mV TO 100mV SS_ 30R 3.3V ON_ N 1R SYNCHRONOUS RECTIFIER CONTROL VL R Q S LEVEL SHIFT DL_ PGND Figure 3. PWM Controller Block Diagram ______________________________________________________________________________________ 11 MAX786 second half-cycle, the PWM turns on the high-side MOSFET by connecting the capacitor to the MOSFET gate by closing an internal switch between BST_ and DH_. This provides the necessary enhancement voltage to turn on the high-side switch, an action that “boosts” the 5V gate-drive signal above the battery voltage. MAX786 Dual-Output Power-Supply Controller for Notebook Computers Modes of Operation PWM Mode Under heavy loads—over approximately 25% of full load — the +3.3V and +5V supplies operate as continuouscurrent PWM supplies (see Typical Operating Characteristics). The duty cycle (%ON) is approximately: %ON = VOUT/VIN Current flows continuously in the inductor: First, it ramps up when the power MOSFET conducts; then, it ramps down during the flyback portion of each cycle as energy is put into the inductor and then discharged into the load. Note that the current flowing into the inductor when it is being charged is also flowing into the load, so the load is continuously receiving current from the inductor. This minimizes output ripple and maximizes inductor use, allowing very small physical and electrical sizes. Output ripple is primarily a function of the filter capacitor (C7 or C6) effective series resistance (ESR) and is typically under 50mV (see the Design Procedure section). Output ripple is worst at light load and maximum input voltage. Idle Mode Under light loads (<25% of full load), efficiency is further enhanced by turning the drive voltage on and off for only a single clock period, skipping most of the clock pulses entirely. Asynchronous switching, seen as “ghosting” on an oscilloscope, is thus a normal operating condition whenever the load current is less than approximately 25% of full load. At certain input voltage and load conditions, a transition region exists where the controller can pass back and forth from idle mode to PWM mode. In this situation, short bursts of pulses occur that make the current waveform look erratic, but do not materially affect the output ripple. Efficiency remains high. can also be driven with an external 240kHz to 350kHz CMOS/TTL source to synchronize the internal oscillator. Normally, 300kHz is used to minimize the inductor and filter capacitor sizes, but 200kHz may be necessary for low input voltages (see Low-Voltage (6-Cell) Operation). Comparators Two noninverting comparators can be used as precision voltage comparators or high-side drivers. The supply for these comparators (VH) is brought out and may be connected to any voltage between +3V and +19V irrespective of V+. The noninverting inputs (D1-D2) are high impedance, and the inverting input is internally connected to a 1.650V reference. Each output (Q1-Q2) sources 20µA from VH when its input is above 1.650V, and sinks 500µA to GND when its input is below 1.650V. The Q1-Q2 outputs can be fixed together in wired-OR configuration since the pull-up current is only 20µA. Connecting VH to a logic supply (5V or 3V) allows the comparators to be used as low-battery detectors. For driving N-channel power MOSFETs to turn external loads on and off, VH should be 6V to 12V higher than the load voltage. This enables the MOSFETs to be fully turned on and results in low rDS(ON). The comparators are always active when V+ is above +4V, even when VH is 0V. Thus, Q1-Q2 will sink current to GND even when VH is 0V, but they will only source current from VH when VH is above approximately 1.5V. If Q1 or Q2 is externally pulled above VH, an internal diode conducts, pulling VH a diode drop below the output and powering anything connected to VH. This voltage will also power the other comparator outputs. BATTERY INPUT VL Current Limiting The voltage between CS3 (CS5) and FB3 (FB5) is continuously monitored. An external, low-value shunt resistor is connected between these pins, in series with the inductor, allowing the inductor current to be continuously measured throughout the switching cycle. Whenever this voltage exceeds 100mV, the drive voltage to the external high-side MOSFET is cut off. This protects the MOSFET, the load, and the battery in case of short circuits or temporary load surges. The current-limiting resistors R1 and R2 are typically 25mΩ for 3A load current. Oscillator Frequency; SYNC Input The SYNC input controls the oscillator frequency. Connecting SYNC to GND or to VL selects 200kHz operation; connecting to REF selects 300kHz operation. SYNC 12 VL BST_ DH_ LEVEL TRANSLATOR LX_ PWM VL DL_ Figure 4. Boost Supply for Gate Drivers ______________________________________________________________________________________ Dual-Output Power-Supply Controller for Notebook Computers MAX786 Table 1. Surface-Mount Components (See Figure 1 for Standard Application Circuit.) COMPONENT 33µF, 35V tantalum capacitors AVX Sprague PART NO. TPSE226M035R0100 595D336X0035R C2 4.7µF, 6V tantalum capacitor AVX Sprague TAJB475M016 595D475X0016A C3 1µF, 20V tantalum capacitor AVX Sprague TAJA105M025 595D105X0020A2B C4, C5 0.1µF, 16V ceramic capacitors Murata-Erie GRM42-6X7R104K50V C6 330µF, 10V tantalum capacitor Sprague 595D337X0010R C7, C12 150µF, 10V tantalum capacitors Sprague 595D157X0010D C8, C9 0.01µF, 16V ceramic capacitors Murata-Erie GRM42-6X7R103K50V CMPD2836 C1, C10 D2A, D2B SPECIFICATION MANUFACTURER 1N4148-type dual diodes Central Semiconductor D1, D3 1N5819 SMT diodes Nihon EC10QS04 L1, L2 10µH, 2.65A inductors Sumida CDR125-100 N1–N4 N-channel MOSFETs (SO-8) Siliconix Si9410DY R1, R2 0.025Ω, 1% (SMT) resistors IRC LR2010-01-R025-F Table 2. Component Suppliers FACTORY FAX [COUNTRY CODE] USA PHONE AVX  (803) 626-3123 (803) 946-0690 (800) 282-4975 Central Semiconductor  (516) 435-1824 (516) 435-1110 IRC  (512) 992-3377 (512) 992-7900 Murata-Erie  (814) 238-0490 (814) 237-1431 Nihon  3-3494-7414 (805) 867-2555 Siliconix  (408) 970-3950 (408) 988-8000 Sprague  (603) 224-1430 (603) 224-1961 Sumida  3-3607-5144 (847) 956-0666 COMPANY Internal VL and REF Supplies An internal linear regulator produces the 5V used by the internal control circuits. This regulator’s output is available on pin VL and can source 5mA for external loads. Bypass VL to GND with 4.7µF. To save power, when the +5V switch-mode supply is above 4.5V, the internal linear regulator is turned off and the high-efficiency +5V switch-mode supply output is connected to VL. The internal 3.3V bandgap reference (REF) is powered by the internal 5V VL supply. It can furnish up to 5mA. Bypass REF to GND with 0.22µF, plus 1µF/mA of load current. The main switching outputs track the reference voltage. Loading the reference will reduce the main outputs slightly, according to the reference load-regulation error. Both the VL and REF outputs remain active, even when the switching regulators are turned off, to supply memory keep-alive power (see Shutdown Mode section). These linear-regulator outputs can be directly connected to the corresponding step-down regulator outputs (i.e., REF to +3.3V, VL to +5V) to keep the main supplies alive in standby mode. However, to ensure start-up, standby load currents must not exceed 5mA on each supply. Fault Protection The +3.3V and +5V PWM supplies and the comparators are disabled when either of two faults is present: VL < +4.0V or REF < +2.8V (85% of its nominal value). __________________Design Procedure Figure 1’s schematic and Table 2’s component list show values suitable for a 3A, +5V supply and a 3A, +3.3V supply. This circuit operates with input voltages from 6.5V to 30V, and maintains high efficiency with output currents between 5mA and 3A (see the Typical Operating Characteristics). This circuit’s components may be changed if the design guidelines described in this section are used — but before beginning the design, the following information should be firmly established: ______________________________________________________________________________________ 13 MAX786 Dual-Output Power-Supply Controller for Notebook Computers VIN(MAX), the maximum input (battery) voltage. This value should include the worst-case conditions under which the power supply is expected to function, such as no-load (standby) operation when a battery charger is connected but no battery is installed. VIN(MAX) cannot exceed 30V. VIN(MIN), the minimum input (battery) voltage. This value should be taken at the full-load operating current under the lowest battery conditions. If VIN(MIN) is below about 6.5V, the filter capacitance required to maintain good AC load regulation increases, and the current limit for the +5V supply has to be increased for the same load level. Inductor (L1, L2) Three inductor parameters are required: the inductance value (L), the peak inductor current (ILPEAK), and the coil resistance (RL). The inductance is: (VOUT) (VIN(MAX) - VOUT) L = ———————————— (VIN(MAX)) (f) (IOUT) (LIR) where: VOUT = output voltage (3.3V or 5V); VIN(MAX) = maximum input voltage (V); f = switching frequency, normally 300kHz; IOUT = maximum DC load current (A); LIR = ratio of inductor peak-to-peak AC current to average DC load current, typically 0.3. A higher value of LIR allows smaller inductance, but results in higher losses and higher ripple. The highest peak inductor current (ILPEAK) equals the DC load current (IOUT) plus half the peak-to-peak AC inductor current (ILPP). The peak-to-peak AC inductor current is typically chosen as 30% of the maximum DC load current, so the peak inductor current is 1.15 times IOUT. The peak inductor current at full load is given by: (VOUT) (VIN(MAX) - VOUT) ILPEAK = IOUT + —————————————. (2) (f) (L) (VIN(MAX)) The coil resistance should be as low as possible, preferably in the low milliohms. The coil is effectively in series with the load at all times, so the wire losses alone are approximately: Power loss = (IOUT2) (RL). In general, select a standard inductor that meets the L, ILPEAK, and RL requirements (see Tables 1 and 2). If a standard inductor is unavailable, choose a core with an LI2 parameter greater than (L) (ILPEAK2), and use the largest wire that will fit the core. 14 Current-Sense Resistors (R1, R2) The sense resistors must carry the peak current in the inductor, which exceeds the full DC load current. The internal current limiting starts when the voltage across the sense resistors exceeds 100mV nominally, 80mV minimum. Use the minimum value to ensure adequate output current capability: For the +3.3V supply, R1 = 80mV / (1.15 x I OUT ); for the +5V supply, R2 = 80mV/(1.15 x IOUT), assuming that LIR = 0.3. Since the sense resistance values (e.g., R1 = 25mΩ for IOUT = 3A) are similar to a few centimeters of narrow traces on a printed circuit board, trace resistance can contribute significant errors. To prevent this, Kelvin connect the CS_ and FB_ pins to the sense resistors; i.e., use separate traces not carrying any of the inductor or load current, as shown in Figure 5. Run these traces parallel at minimum spacing from one another. The wiring layout for these traces is critical for stable, low-ripple outputs (see the Layout and Grounding section). MOSFET Switches (N1-N4) The four N-channel power MOSFETs are usually identical and must be “logic-level” FETs; that is, they must be fully on (have low r DS(ON) ) with only 4V gatesource drive voltage. The MOSFET r DS(ON) should ideally be about twice the value of the sense resistor. MOSFETs with even lower r DS(ON) have higher gate capacitance, which increases switching time and transition losses. MOSFETs with low gate-threshold voltage specifications (i.e., maximum VGS(TH) = 2V rather than 3V) are preferred, especially for high-current (5A) applications. Output Filter Capacitors (C6, C7, C12) The output filter capacitors determine the loop stability and output ripple voltage. To ensure stability, the minimum capacitance and maximum ESR values are: VREF CF > ————————————— (VOUT) (RCS) (2) (π) (GBWP) and, (VOUT) (RCS) ESRCF < —————— VREF where: CF = output filter capacitance (F); VREF = reference voltage, 3.3V; VOUT = output voltage, 3.3V or 5V; RCS = sense resistor (Ω); GBWP = gain-bandwidth product, 60kHz; ESRCF = output filter capacitor ESR (Ω). ______________________________________________________________________________________ Dual-Output Power-Supply Controller for Notebook Computers In idle-mode, the ripple has a capacitive and resistive component: (4) (10-4) (L) VOUT(RPL)(C) = ——————— x (RCS2) (CF) 1 1 ——— + ————— Volts VOUT VIN - VOUT ( ) (0.02) (ESRCF) VOUT(RPL)(R) = ———————Volts RCS The total ripple, VOUT(RPL), can be approximated as follows: if VOUT(RPL)(R) < 0.5 VOUT(RPL)(C), then VOUT(RPL) = VOUT(RPL)(C), otherwise, VOUT(RPL) = 0.5 VOUT(RPL)(C) + VOUT(RPL)(R). Diodes D1 and D3 Use 1N5819s or similar Schottky diodes. D1 and D3 conduct only about 3% of the time, so the 1N5819’s 1A current rating is conservative. The voltage rating of D1 and D3 must exceed the maximum input supply voltage from the battery. These diodes must be Schottky diodes to prevent the lossy MOSFET body diodes from turning on, and they must be placed physically close to their associated synchronous rectifier MOSFETs. Soft-Start Capacitors (C8, C9) A capacitor connected from GND to either SS pin causes that supply to ramp up slowly. The ramp time to full current limit, tSS, is approximately 1ms for every nF of capacitance on SS_, with a minimum value of 10µs. Typical capacitor values are in the 10nF to 100nF range; a 5V rating is sufficient. Because this ramp is applied to the current-limit circuit, the actual time for the output voltage to ramp up depends on the load current and output capacitor value. Using Figure 1’s circuit with a 2A load and no SS capacitor, full output voltage is reached about 600µs after ON_ is driven high. Boost Capacitors (C4, C5) Capacitors C4 and C5 store the boost voltage and provide the supply for the DH3 and DH5 drivers. Use 0.1µF and place each within 10mm of the BST_ and LX_ pins. Boost Diodes (D1A, D1B) Use high-speed signal diodes; e.g., 1N4148 or equivalent. Bypass Capacitors Input Filter Capacitors (C1, C10) Use at least 3µF/W of output power for the input filter capacitors, C1 and C10. They should have less than 150mΩ ESR, and should be located no further than 10mm from N1 and N2 to prevent ringing. Connect the negative terminals directly to PGND. Do not exceed the surge current ratings of input bypass capacitors. Shutdown Mode Shutdown (SHDN = low) forces both PWMs off and disables the REF output and both comparators (Q1 = Q2 = 0V). Supply current in shutdown mode is typically 25µA. The VL supply remains active and can source 25mA for external loads. Note that the VL load capability is higher in shutdown and standby modes than when the PWMs are operating (25mA vs. 5mA). Standby mode is achieved by holding ON3 and ON5 low while SHDN is high. This disables both PWMs, but keeps VL, REF, and the precision comparators alive. Supply current in standby mode is typically 70µA. FAT, HIGH-CURRENT TRACES MAIN CURRENT PATH KELVIN SENSE TRACES SENSE RESISTOR MAX786 Figure 5. Kelvin Connections for the Current-Sense Resistors ______________________________________________________________________________________ 15 MAX786 Be sure to select output capacitors that satisfy both the minimum capacitance and maximum ESR requirements. To achieve the low ESR required, it may be appropriate to use a capacitance value 2 or 3 times larger than the calculated minimum. The output ripple in continuous-current mode is: VOUT(RPL) = ILPP(MAX) x (ESRCF + 1/(2 x π x f x CF) ). MAX786 Dual-Output Power-Supply Controller for Notebook Computers Other ways to shut down the MAX786 are suggested in the applications section of the MAX782/MAX783 data sheet. Table 3. EV Kit Power-Supply Controls (SW1) ON OFF SETTING SETTING SWITCH NAME FUNCTION 1 SHDN Enable shutdown mode Operate Shutdown 2 ON3 Enable 3.3V power supply 3.3V ON 3.3V OFF 3 ON5 Enable 5.0V power supply 5V ON 5V OFF 4 SYNC Oscillator 200kHz 300kHz __________Applications Information Low-Voltage (6-Cell) Operation The standard application circuit can be configured to accept input voltages from 5.5V to 12V by changing the oscillator frequency to 200kHz and increasing the +5V filter capacitor to 660µF. This allows stable operation at 5V loads up to 2A (the 3.3V side requires no changes and still delivers 3A). V+ C10 33µF 35V C1 33µF 35V VL (5V) 23 R9 1k 6 V+ C2 4.7µF 22 VL VH D2 D2 C5 0.1µF R10 OPEN 3.3V OUT R1 0.025Ω N1 L1 10µH D1 1N5819 27 26 N3 C7 C12 150µF 150µF 10V 10V 24 1 28 SW1A BST3 BST5 DH3 DH5 LX3 LX5 DL3 DL5 MAX786 CS3 CS5 FB3 FB5 REF SW1C 3 ON3 ON5 25 SW1B 13 12 D1 4 D2 5 C9 0.01µF 18 16 N2 17 19 N4 SHDN Q1 D1 Q2 D2 SS5 SS3 GND PGND 2 9 20 R2 0.025Ω C6 330µF 10V 21 VREF (3.3V) C3 1µF 20V N1 – N4 = Si9410DY D2 = BAW56L OR TWO 1N4148s 11 SW1D SYNC 8 Q1 7 Q2 14 C8 0.01µF R6 R5 R8 R7 R4 1M 1M 1M 1M 1M Figure 6. MAX786 EV Kit Schematic 16 5V OUT D3 1N5819 10 R3 1M ON5 L2 10µH 15 ON3 SYNC SHDN C4 0.1µF ______________________________________________________________________________________ Dual-Output Power-Supply Controller for Notebook Computers The MAX786 evaluation kit (EV kit) embodies the standard application circuit, with some extra pullup and pull-down resistors needed to set default logic signal levels. The board comes configured to accept battery input voltages between 6.5V and 30V, and provides up to 25W of output power. All functions are con- 1.0" trolled by standard CMOS/TTL logic levels or DIP switches. The kit can be reconfigured for lower battery voltages by setting the oscillator to 200kHz and increasing the 5V output filter capacitor value. The D1 and D2 comparators can be used as precision voltage detectors by installing resistor dividers at each input. 1.0" Figure 7. MAX786 EV Kit Top Component Layout and Silk Screen, Top View Figure 8. MAX786 EV Kit Ground Plane (Layers 2 and 3), Top View 1.0" Figure 9. MAX786 EV Kit Top Layer (Layer 1), Top View ______________________________________________________________________________________ 17 MAX786 _________________EV Kit Information MAX786 Dual-Output Power-Supply Controller for Notebook Computers 1.0" Figure 10. MAX786 EV Kit, Bottom Component Layout and Silk Screen, Bottom View 1.0" Figure 11. MAX786 EV Kit, Bottom Layer (Layer 4), Top View 18 ______________________________________________________________________________________ Dual-Output Power-Supply Controller for Notebook Computers MAX786 ______________________Chip Topography SS3 CS3 FB3 DH3 ON3 D1 D2 LX3 BST3 DL3 VH Q2 Q1 V+ 0.181" VL (4.597mm) FB5 PGND GND DL5 REF BST5 SYNC LX5 SHDN ON5 SS5 CS5 DH5 0.109" (2.769mm) TRANSISTOR COUNT: 1294 SUBSTRATE CONNECTED TO GND ______________________________________________________________________________________ 19 __Ordering Information (continued) PART TEMP. RANGE PIN-PACKAGE VOUT MAX786SCAI 0°C to +70°C 28 SSOP MAX786C/D 0°C to +70°C Dice* MAX786EAI -40°C to +85°C 28 SSOP 3.3V MAX786REAI -40°C to +85°C 28 SSOP 3.45V MAX786SEAI -40°C to +85°C 28 SSOP 3.6V 3.6V — EV KIT TEMP. RANGE BOARD TYPE MAX786EVKIT-SO 0°C to +70°C Surface Mount *Contact factory for dice specifications. ________________________________________________________Package Information SSOP.EPS MAX786 Dual-Output Power-Supply Controller for Notebook Computers Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 1997 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.