LM5037 Dual-Mode PWM Controller With

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LM5037
SNVS578D – NOVEMBER 2008 – REVISED MAY 2015
LM5037 Dual-Mode PWM Controller With Alternating Outputs
1 Features
3 Description
•
•
•
The LM5037 PWM controller contains all the features
necessary to implement balanced double-ended
power converter topologies, such as push-pull, halfbridge and full-bridge. These double-ended
topologies allow for higher efficiencies and greater
power densities compared to common single-ended
topologies such as the flyback and forward. The
device can be configured for either voltage mode or
current mode control with minimum external
components. Two alternating gate drive outputs are
provided, each capable of 1.2-A peak output current.
The device can be configured to operate directly from
the input voltage rail over a wide range of 13 V to
100 V.
1
•
•
•
•
•
•
•
•
•
•
High-Voltage (100-V) Start-up Regulator
Alternating Outputs for Double-Ended Topologies
Current-mode or Feed-forward Voltage-mode
Control
Programmable Maximum Duty Cycle Limit
2% Feedback Reference Accuracy
High Gain-bandwidth Error Amplifier
Programmable Line Undervoltage Lockout (UVLO)
with Adjustable Hysteresis
Versatile Dual Mode Overcurrent Protection with
Hiccup Delay Timer
Programmable Soft-start Time
Precision 5-V Reference Output
Current Sense Leading Edge Blanking
Resistor Programmed 2-MHz Capable Oscillator
Oscillator Synchronization Capability with LowFrequency Lockout Protection
Additional features include programmable maximum
duty cycle limit, line undervoltage lockout, cycle-bycycle current limit and a hiccup mode fault protection
with adjustable timeout delay, soft-start and a 2 MHz
capable oscillator with synchronization capability,
precision reference and thermal shutdown.
Device Information(1)
2 Applications
•
•
PART NUMBER
Telecom Power Converters
Industrial Power Converters
LM5037
PACKAGE
TSSOP (16)
BODY SIZE (NOM)
5.00 mm × 4.4 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Application
VIN
VOUT
LM5037
VIN
VCC
UVLO
OUTA
REF
OUTB
RT1/SYNC
RAMP
RT2
CS
RES
COMP
SS
PGND
FB
Isolated
Feedback
AGND
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM5037
SNVS578D – NOVEMBER 2008 – REVISED MAY 2015
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Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
8
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 10
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 11
7.3 Feature Description................................................. 12
7.4 Device Functional Modes........................................ 17
8
Application and Implementation ........................ 21
8.1 Application Information............................................ 21
8.2 Typical Application ................................................. 26
9 Power Supply Recommendations...................... 32
10 Layout................................................................... 33
10.1 Layout Guidelines ................................................. 33
10.2 Layout Example .................................................... 33
11 Device and Documentation Support ................. 34
11.1
11.2
11.3
11.4
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
34
34
34
34
12 Mechanical, Packaging, and Orderable
Information ........................................................... 34
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision C (March 2013) to Revision D
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section .................................................................................................. 1
Changes from Revision B (March 2013) to Revision C
•
2
Page
Page
Changed layout of National Data Sheet to TI format ........................................................................................................... 26
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5 Pin Configuration and Functions
PW PACKAGE
16 PIN TSSOP
(TOP VIEW)
RAMP
1
16
VIN
UVLO
2
15
REF
COMP
3
14
VCC
FB
4
13
OUTA
RT2
5
12
OUTB
AGND
6
11
PGND
RT1/SYNC
7
10
SS
CS
8
9
RES
Pin Functions
PIN
I/O (1)
DESCRIPTION
NAME
NO.
AGND
6
—
Analog ground. Connect directly to power ground
COMP
3
I/O
Input to the pulse width modulator. Output of the error amplifier and input to the PWM comparator.
CS
8
I
Current sense input. If the CS pin exceeds 250 mV the output pulse terminates, entering cycle-bycycle current limit. An internal switch holds the CS pin low for a period of 65 ns after either output
switches high to blank leading edge transients.
FB
4
I
Feedback. Connected to inverting input of the error amplifier. An internal 1.25-V reference is
connected to the non-inverting input of the error amplifier. In isolated applications using an external
error amplifier, this pin should be connected to the analog ground pin (AGND).
OUTA
13
OUTB
12
O
Output driver. Alternating gate-drive output of the pulse width modulator. These pins are capable of
1.2-A peak source and sink current.
PGND
11
—
Power ground. Connect directly to the analog ground pin (AGND).
RAMP
1
I
Pulse width modulator ramp. Modulation ramp for the PWM comparator. This ramp can be a
representative of the primary current (current mode) or proportional to input voltage (feed-forward
voltage mode). This pin is reset to ground at the conclusion of every cycle by an internal FET.
REF
15
O
Output of a 5V reference. Locally decouple with a capacitor with a value of 0.1-µF or greater.
Maximum output current is 10 mA (typ).
RES
9
I
Restart timer. If cycle-by-cycle current limit is reached during any cycle, the device sources 18 µA of
current to the external RES pin capacitor. If the RES capacitor voltage reaches 2.0 V, the soft-start
capacitor is discharged and then released with a pull-up current of 1 µA. After the first output pulse
(when SS = 1 V), the SS pin charging current increases to the normal level of 100 µA.
RT1/SYNC
7
I
Oscillator dead-time control. The resistance connected between this pin (RT1/SYNC) and the AGND
pin sets the oscillator maximum on-time. The sum of this maximum on-time and the forced dead-time
(set by the RT2 pin) sets the oscillator period.
RT2
5
I
Oscillator dead-time control. The resistance connected between this pin (RT2) and the AGND pin
sets the forced dead-time between switching periods of the alternating outputs.
SS
10
I
Soft-start. An external capacitor and an internal 100-µA current source set the soft-start ramp. The
SS current source is reduced to 1 µA following a restart event (RES pin high).
UVLO
2
I
Line undervoltage lockout. An external voltage divider from the power source sets the shutdown and
standby comparator threshold levels. When the UVLO pin exceeds the 0.45-V shutdown threshold,
the VCC pin and REF pin regulators are enabled. When the UVLO pin exceeds the 1.25-V standby
threshold, the SS pin is released and the device enters the active mode.
VCC
14
I/O
Output of the high voltage start-up regulator. The VCC pin voltage is regulated to 7.7 V. If an auxiliary
winding raises the voltage on this pin above the regulation set point, the internal start-up regulator
shuts down thus reducing the power dissipation of the device. Locally decouple the VCC pin with a
capacitor with a value of 0.47 µF or greater.
VIN
16
I
Input voltage source. Input to the VCC start-up regulator. Operating input range is 13 V to 100 V. For
power sources outside of this range, the LM5037device can be biased directly at the VCC pin by an
external regulator.
(1)
I = Input, O = Output, G = Ground
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
Input voltage (2)
(1)
MIN
MAX
VIN
–0.3
105
VCC, OUTA, OUTB
–0.3
16
CS
–0.3
1.0
UVLO, FB, RT2, RT1/SYNC, RAMP, SS, REF
–0.3
7
UNIT
V
Output voltage
–0.3
7
V
Storage temperature, Tstg
−65
50
°C
(1)
(2)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All input voltage ratings apply w/r/t GND.
6.2 ESD Ratings
VALUE
Electrostatic
discharge
V(ESD)
(1)
(2)
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
VIN
Supply input voltage range
VCC
External voltage applied to VCC
TJ
Operating junction temperature
MAX
UNIT
13
NOM
100
V
8
15
V
–40
125
°C
6.4 Thermal Information
LM5037
THERMAL METRIC
(1)
PW (TSSOP)
UNIT
16 PINS
RθJA
Junction-to-ambient thermal resistance
99.9
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
32.7
°C/W
RθJB
Junction-to-board thermal resistance
45.8
°C/W
ψJT
Junction-to-top characterization parameter
2.0
°C/W
ψJB
Junction-to-board characterization parameter
45.1
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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6.5 Electrical Characteristics
over operating free-air temperature range, VVIN = 48 V, VVCC = 10 V, RRT1/SYNC = 30.1 kΩ, RRT2 = 30.1 kΩ, VUVLO = 3 V (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
STARTUP REGULATOR (VCC PIN)
VVCC
VCC voltage
IVCC(lim)
VCC current limit
IVCC = 10 mA
VVCC = 7 V
VCC undervoltage threshold
8
7.7
45
VVCC = 7 V, TA = 25°C
VVIN = VVCC
VVCC(UV)
7.4
IVCC = 10 mA, TA = 25°C
VCC Reg0.2
VCC Reg0.1
VVIN = VVCC, TA = 25°C
430
VVIN = 100 V, VUVLO = 0 V,
TA = 25°C
350
VVIN = 48 V, VUVLO = 0 V
Supply current into VCC from
external source
µA
370
VVIN = 48 V, VUVLO = 0 V, TA = 25°C
IVCC
V
1.5
VVIN = 100 V, VUVLO = 0 V
Startup regulator current
mA
60
Hysteresis
IVIN
V
325
Output pins and COMP = Open
5.5
Output pins and COMP = Open,
TA = 25°C
mA
3
VOLTAGE REFERENCE REGULATOR (REF PIN)
REF pin voltage
VREF
REF voltage regulation
IREF(lim)
REF current limit
VREF undervoltage threshold
VREF(UV)
IREF = 0 mA
4.75
IREF = 0 mA, TA = 25°C
5.15
5
0 A ≤ IREF ≤ 2.5 mA
25
0 A ≤ IREF ≤ 2.5 mA, TA = 25°C
VREF = 4.5 V
7
5
VREF = 4.5 V, TA = 25°C
3.7
4.3
4
Hysteresis
mV
mA
10
TA = 25°C
V
0.35
V
V
UNDERVOLTAGE LOCKOUT AND SHUTDOWN (UVLO PIN)
1.20
VUVLO
Undervoltage lockout threshold
TA = 25°C
UVLO voltage rising
0.37
UVLO voltage rising, TA = 25°C
Hysteresis current
0.47
0.1
UVLO pin sinking
V
0.42
Hysteresis voltage
IUVLO
1.295
1.25
18
UVLO pin sinking, TA = 25°C
V
25
22
µA
CURRENT SENSE INPUT (CS PIN)
VCS
Current limit threshold
tDLY(CS)
CS delay to output (1)
tBLK(CS)
Leading edge blanking time at CS
RCS(sink)
(1)
(2)
CS sink impedance (clocked) (2)
0.22
TA = 25°C
VCS rising from zero to 1 V. No load.
0.29
0.25
27
ns
66
ns
45
TA = 25°C
V
21
Ω
Time for OUTA pin and OUTB pin to fall to 90% of VCC.
Internal FET sink impedance.
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Electrical Characteristics (continued)
over operating free-air temperature range, VVIN = 48 V, VVCC = 10 V, RRT1/SYNC = 30.1 kΩ, RRT2 = 30.1 kΩ, VUVLO = 3 V (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
CURRENT LIMIT RESTART (RES PIN)
VRES
RES voltage threshold
ICHG(src)
Charge source current
IDSCHG(snk)
Discharge sink current
1.9
TA = 25°C
VRES = 1.5 V
14
VRES = 1.5 V, TA = 25°C
VRES = 1 V
2.2
2
22
18
5
VRES = 1 V, TA = 25°C
11
8
V
µA
µA
SOFT-START (SS PIN)
VSS = 0, Normal operation
ISS
70
VSS = 0, Normal operation, TA =
25°C
Charging current
VSS = 0, Hiccup mode restart
100
0.6
VSS = 0, Hiccup mode restart, TA =
25°C
ISS(sink)
VSS = 2.0 V
Soft-stop current sink
130
1.5
µA
1
70
VSS = 2.0 V, TA = 25°C
130
100
µA
OSCILLATOR (RT1/SYNC AND RT2 PINS)
RRT2 = 15 kΩ
tDEAD
Dead time
40
RRT2 = 15 kΩ, TA = 25°C
75
RRT2 = 75 kΩ
RRT1/SYNC = 30.1 kΩ, RRT2 = 30.1
kΩ
fSW1
Switching frequency 1
(3)
176
223
kHz
200
441
RRT1/SYNC = 11 kΩ, RRT2 = 30.1 kΩ,
TA = 25°C
571
kHz
508
DC voltage level
Input synchronization threshold
voltage
ns
250
RRT1/SYNC = 30.1 kΩ, RRT2 = 30.1
kΩ,
TA = 25°C
RRT1/SYNC = 11 kΩ, RRT2 = 30.1 kΩ
Switching frequency 2 (3)
fSW2
105
2
2.5
TA = 25°C
V
3.4
3
V
PWM CONTROLLER (COMP PIN)
tDLY(pwm)
Delay-to-output time
VPWM-OS
SS to RAMP offset voltage
DMIN
Mimumum duty cycle
COMP open circuit voltage
COMP short circuit current
65
0.7
TA = 25°C
1
VSS = 0 V
VFB = 0 V
VFB = 0 V, VCOMP = 0 V, TA = 25°C
V
0%
4.5
VFB = 0 V, TA = 25°C
VFB = 0 V, VCOMP = 0 V
ns
1.2
5
4.75
0.5
1.5
1
V
mA
VOLTAGE FEED-FORWARD (RAMP PIN)
RRAMP(sink)
(3)
6
RAMP sink impedance (clocked)
20
TA = 25°C
5
Ω
Measured at OUTA, half oscillator frequency
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Electrical Characteristics (continued)
over operating free-air temperature range, VVIN = 48 V, VVCC = 10 V, RRT1/SYNC = 30.1 kΩ, RRT2 = 30.1 kΩ, VUVLO = 3 V (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ERROR AMPLIFIER
GBW
Gain bandwidth
4
MHz
DC gain
Input voltage
COMP pin sink capability
VFB = VCOMP
1.22
VFB = VCOMP, TA = 25°C
VFB = 1.5 V, VCOMP = 1 V
1.27
1.245
5
FB pin bias current
V
13
mA
10
nA
MAIN OUTPUT DRIVERS (OUTA AND OUTB PINS)
VOH
High-level output voltage
VOL
Low-level output voltage
tRISE
tFALL
IOUT = 50 mA, (Source)
IOUT = 50 mA, (Source), TA = 25°C
VVCC–0.5
V
VVCC–0.25
IOUT = 100 mA (Sink)
0.5
V
IOUT = 100 mA (Sink), TA = 25°C
0.2
Rise time
CLOAD = 1 nF
15
ns
Fall time
CLOAD = 1 nF
13
ns
IPEAK(src)
Peak source current
VVCC = 10 V
1.2
A
IPEAK(snk)
Peak source current
VVCC = 10V
1.2
A
Thermal shutdown threshold
165
°C
Thermal shutdown hysteresis
25
°C
THERMAL SHUTDOWN
TSD
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6.6 Typical Characteristics
VUVLO = 0 V
Figure 1. Reference Voltage vs. Input Voltage
Figure 2. Start-Up Regulator Current vs Input Voltage
Figure 3. Input Voltage vs Input Current
Figure 4. Reference Voltage vs. Reference Current
180
50
150
40
120
30
90
20
60
10
30
0
-10
0
-30
-20
-60
-30
-90
-40
-120
-50
-150
-60
10k
100k
1M
FREQUENCY
PHASE (o)
GAIN (dB)
60
-180
10M
Figure 5. Feedback Amplifier Bode Plot
8
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Figure 6. Oscillator Frequency vs
Timing Resistance (RRT1/SYNC)
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Typical Characteristics (continued)
Figure 7. Dead Time vs. Timing Resistance (RRT2)
Figure 8. Feedback Voltage vs. Temperature
Figure 9. Oscillator Frequency vs. Temperature
Figure 10. Dead-Time vs. Temperature
Figure 11. Soft-Start and Restart Current vs. Temperature
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7 Detailed Description
7.1 Overview
The LM5037 PWM controller contains all the features necessary to implement double-ended power converter
topologies such as push-pull, half-bridge and full-bridge. The unique architecture allows the modulator to be
configured for either voltage-mode or current-mode control. The device provides two alternating gate driver
outputs to drive the primary-side power MOSFETs with programmable forced dead-time. The device can be
configured to operate with bias voltages ranging from 13 V to 100 V. Additional features include line undervoltage
lockout, cycle-by-cycle current limit, voltage feed-forward compensation, hiccup mode fault protection with
adjustable delays, soft-start, a 2-MHz capable oscillator with synchronization capability, precision reference and
thermal shutdown. These features simplify the design of double ended topologies. The Functional Block Diagram
section shows the functional block diagram.
10
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7.2 Functional Block Diagram
7.7V Series
Regulator
VCC
VIN
UVLO
0.45V
SHUTDOWN
1.25V
STANDBY
5V
Reference
MODE
CONTROL
LOGIC
REF
VCC/REF
UVLO
THERMAL LIMIT
( 165°C)
UVLO HYSTERESIS (20 PA)
VCC
RT1/SYNC
CLK
J
SET
K
CLR
OUTA
Q
DRIVER
OSCILLATOR
RT2
S
Q
Q
VCC
R
OUTB
RAMP
1.25V
DRIVER
ERROR
AMP
FB
+5V
PGND
5k
PWM
COMP
AGND
PWM
LOGIC
1V
SS
SS Buffer
2.0V
+5V
CS
0.25V
Hiccup
CLK + LEB
SS
+5V
SOFT- START
100 PA
SS
CLK
Restart
Current
Source
Logic
18 PA
RES
+5V
RESTART DELAY
8 PA
1 PA
Shutdown
100 PA
Standby
SOFT- STOP
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7.3 Feature Description
7.3.1 High-Voltage Start-Up Regulator
The LM5037 device contains an internal high voltage, start-up regulator that allows the input pin (VIN) to be
connected directly to the supply voltage over a range of 13V to a maximum of 100 V. The regulator input can
withstand transients up to 105 V. The regulator output at VCC (7.7 V) is internally current limited with a minimum
of 45 mA. When the UVLO pin potential is greater than 0.45 V, the VCC regulator is enabled to charge an
external capacitor connected to the VCC pin. The VCC regulator provides power to the voltage reference (REF)
and the gate drivers (OUTA and OUTB). When the voltage on the VCC pin exceeds its undervoltage (VCC UV)
threshold, the internal voltage reference (REF) reaches its regulation set point of 5 V and the UVLO voltage is
greater than 1.25 V, the controller outputs are enabled. The value selected for the VCC capacitor depends on the
total system design, and its start-up characteristics. The recommended range of values for the VCC capacitor
between 0.47 µF and 10 µF.
Powering VCC from an external supply can reduce the internal power dissipation of the device. In typical
applications, an auxiliary transformer winding is connected through a diode to the VCC pin. This winding must
raise the VCC voltage above 8.1 V to shut off the internal start-up regulator. Powering VCC from an auxiliary
winding improves efficiency while reducing the controller’s power dissipation. The VCC UV circuit function
remains in this mode, requiring that VCC never falls below its nominal threshold during the start-up sequence.
The VCC regulator series pass transistor includes a diode between VCC and VIN that should not be forward
biased in normal operation. Therefore the auxiliary VCC voltage should never exceed the VIN voltage.
An external DC bias voltage can be used instead of the internal regulator by connecting the external bias voltage
to both the VCC and the VIN pins. In this particular case, the external bias must be greater than maximum VCC
pin regulation of 8 V and less than the VCC maximum operating voltage rating (15 V).
7.3.2 Reference
The REF pin is the output of a 5-V linear regulator that can be used to bias an opto-coupler transistor and
external housekeeping circuits. The regulator output is internally current limited to 10 mA (typical).
7.3.3 Error Amplifier
An internal high gain error amplifier is provided within the LM5037. The non-inverting amplifier reference is tied to
a 1.25 V reference. In non-isolated applications the power converter output is connected to the FB pin via the
voltage setting resistors and loop compensation is connected between the COMP and FB pins. A typical
gain/phase plot is shown in the Typical Characteristics section.
For most isolated applications the error amplifier function is implemented on the secondary side. Since the
internal error amplifier is configured as an open drain output, it can be disabled by connecting FB to ground. The
internal, 5-kΩ pull-up resistor connected between the COMP pin and the 5-V reference can be used as the pullup for an opto-coupler or other isolation device .
7.3.4 Cycle-By-Cycle Current Limit
The CS pin is to be driven by a signal representative of the transformer primary current. The current sense signal
can be generated by using a sense resistor or a current sense transformer. If the voltage sensed at the CS pin
exceeds 0.25 V, the current sense comparator terminates the output driver pulse. If the high current condition
persists, the controller operates in a cycle-by-cycle current limit mode with duty cycle determined by the current
sense comparator instead of the PWM comparator. Cycle-by-cycle current limiting may eventually trigger the
hiccup mode restart cycle; depending on the configuration of the RES pin (see Overload Protection Timer
section). To suppress noise, a small R-C filter connected to the CS pin and located near the controller is
recommended. An internal, 21-Ω MOSFET discharges the external current sense filter capacitor at the
conclusion of every cycle. The discharge MOSFET remains on for an additional 65 ns after either OUTA or
OUTB driver switches high to blank leading edge transients in the current sensing circuit. Discharging the CS pin
filter each cycle and blanking leading edge spikes reduces the filtering requirements and improves the current
sense response time. The current sense comparator is very fast and may respond to short duration noise pulses.
Layout considerations are critical for the current sense filter and sense resistor. The capacitor associated with the
12
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Feature Description (continued)
CS filter must be placed very close to the device and connected directly to the CS and AGND pins. If a sense
resistor located in the source of the main MOSFET switch is used for current sensing, a low inductance type of
resistor is required. When designing with a current sense resistor, all the noise sensitive, low power ground
connections should be connected together near the AGND pin, and a single connection should be made to the
power ground (sense resistor ground point).
7.3.5 Soft-Start Sequence
The soft-start circuit allows the regulator to gradually reach a steady state operating point, thereby reducing startup stresses and current surges. When bias is supplied to the LM5037, the SS pin capacitor is discharged by an
internal MOSFET. When the UVLO, VCC and REF pins reach their operating thresholds, the SS capacitor is
released and charged with a 100 µA current source. The PWM comparator control voltage at the COMP pin is
clamped to the SS pin voltage by an internal amplifier. When the PWM comparator input reaches 1 V, output
pulses commence with slowly increasing duty cycle. The voltage at the SS pin eventually increases to 5 V, while
the voltage at the PWM comparator increases to the value required for regulation as determined by the voltage
feedback loop.
One method to disable the regulator is to ground the SS pin. This forces the internal PWM control signal to
ground, reducing the output duty cycle quickly to zero. Releasing the SS pin initiates a soft-start sequence and
normal operation resumes. A second shutdown method is discussed in the Thermal Protection section.
7.3.6 PWM Comparator
The pulse width modulation (PWM) comparator compares the voltage ramp signal at the RAMP pin to the loop
error signal. The loop error signal is derived from the internal error amplifier (COMP pin). The resulting control
voltage passes through a 1-V level shift before being applied to the PWM comparator. This comparator is
optimized for speed in order to achieve minimum controllable duty cycles. The common mode input voltage
range of the PWM comparator is from 0 V to 4.3 V.
7.3.7 Modulation Ramp
The voltage at the RAMP pin provides the modulation ramp for the PWM comparator. The PWM comparator
compares the modulation ramp signal at the RAMP pin to the loop error signal to control the output duty cycle.
The modulation ramp can be implemented either as a ramp proportional to input voltage, known as feed-forward
voltage mode control, or as a ramp proportional to the primary current, known as current mode control. The
RAMP pin is reset by an internal FET with an RDS(on) of 5 Ω (typical) at the end of every cycle. The ability to
configure the RAMP pin for either voltage mode or current mode allows the controller to be implemented for the
optimum control method for the selected power stage topology. Configuring RAMP pin is explained below and
the differences between voltage mode control and current mode control in various double-ended topologies is
explained in Application Information section.
7.3.8 Feed-Forward Voltage Mode
An external resistor (RFF) and capacitor (CFF) connected to VIN, AGND, and the RAMP pins is required to create
the PWM ramp signal as shown in Figure 12. It can be seen that the slope of the signal at RAMP varies in
proportion to the input line voltage. This varying slope provides line feed-forward information necessary to
improve line transient response with voltage mode control. The RAMP signal is compared to the error signal by
the pulse width modulator comparator to control the duty cycle of the outputs. With a constant error signal, the
on-time (tON) varies inversely with the input voltage (VIN) to stabilize the Volt-Second product of the transformer
primary. At the end of clock period, an internal FET engages to reset the CFF capacitor. The formulae for RFF and
CFF and component selection criteria are explained in Application and Implementation section. The amplitude of
the signal driving RAMP pin must not exceed the common mode input voltage range of the PWM comparator
(3.3 V) while in normal operation.
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Feature Description (continued)
SLOPE
PROPORTIONAL
TO Vin
Vin
R FF
VIN
COMP
1V
Gate
Drive
RAMP
CLK
C FF
LM5037
Figure 12. Feed-Forward Voltage Mode Configuration
7.3.9 Current Mode
The LM5037 device can be configured for current mode control by injecting a signal representative of primary
current into the RAMP pin. One way to achieve this is shown in Figure 13. Filter components Rfilter and Cfilter are
used to filter leading edge noise spikes. The signal at the CS pin is thus a ramp on a pedestal. The pedestal
corresponds to the continuous conduction current in the transformer at the beginning of an OUTA or OUTB
conduction cycle. The R-C circuit (RSlope and CSlope), shown in Figure 13, tied to VREF adds an additional ramp to
the current sense signal. This additional ramp signal, known as slope compensation, is required to avoid
instabilities at duty cycles above 50% (25% per phase). The compensated RAMP signal consists of two parts,
the primary current signal and the slope compensation. The compensated RAMP signal is compared to the error
signal by the PWM comparator to control the duty cycle of the outputs. The RAMP capacitor and CS capacitor
are reset through internal discharge FETs. The on-resistance (RDS(on)) of RAMP discharge FET is 5 Ω (typical);
this ensures fast discharge of the RAMP reset capacitor. Any dc voltage source can be used in place of VREF to
generate the slope compensation ramp.
The timing diagram shown in Figure 14 depicts the current mode waveforms and relative timing. When OUTA or
OUTB is enabled, the signal at the RAMP pin consists of the CS pin signal (current ramp on a pedestal) plus the
slope compensation ramp (dotted lines). When OUTA or OUTB is turned off, the primary current component is
absent but the voltage at the RAMP pin continues to rise due to slope compensation component until the end of
the clock period, after which it is reset by the RAMP discharge FET. A component selection example is explained
in detail in the Application and Implementation section.
14
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Feature Description (continued)
VREF (5V)
Rslope
RAMP
CLK
Cslope
Current
Sense
LM5037
Rfilter
cs
CLK + LEB
RCS
Cfilter
Figure 13. Current Mode Configuration with Slope Compensation
CLK
CS
OUTA
OUTB
COMP
RAMP
Slope
Compensation
OUTPUT
OUTA
OUTB
Figure 14. Timing Diagram for Current Mode Configuration
7.3.10 Oscillator
The LM5037 device oscillator frequency and the maximum duty cycle are set by two external resistors connected
between the RT1/SYNC and RT2 pins to AGND. The minimum dead-time between OUTA and OUTB pulses is
proportional to the RT2 resistor value and the overall oscillator frequency is inversely proportional to RRT1/SYNC
and RRT2 resistor values. Each output switches at half the oscillator frequency. Use Equation 1 to calculate a
value of RRT2 that supports a required dead-time. Use Equation 2 to calculate a value of RRT2 that supports a
maximum duty cycle (DMAX).
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Feature Description (continued)
æ t
RRT2 = ç DEAD
è 5 ´ 10-12
50 ns £ tDEAD £
ö
÷
ø
250 ns
(1)
(2)
æ æ (1 - DMAX ) ö ö
çç
÷÷ ÷
ç çè fOSC
ø÷
RRT2 = ç
÷
-12
ç 5 ´ 10
÷
ç
÷
è
ø
(3)
The recommended dead-time range is between 50 ns and 250 ns. Beyond 250 ns, the RRT2 resistance becomes
excessively large, and is prone to noise pickup. Fixed internal delays limit the dead-time to greater than 50 ns.
After the dead-time has been programmed by RT2, the overall oscillator frequency can be set by selecting the
resistor RRT1/SYNC using Equation 4.
ææ 1 ö
ö
çç
÷ - tDEAD ÷
ç fOSC ø
÷
RRT1/SYNC = ç è
-9 ÷
ç 0.162 ´ 10
÷
ç
÷
è
ø
(4)
For example, if the desired oscillator frequency is 400 kHz (OUTA and OUTB each switching at 200 kHz) and
desired dead-time is 100 ns, the maximum duty cycle for each output is 96%. The value of RRT1/SYNC is 15 kΩ
and RRT2 is 20 kΩ.
CLK
tON(max)
OUTA
OUTB
tOSC
tDEAD
tDEAD
Time
Figure 15. Timing Diagram of OUTA, OUTB and Dead-Time Set by RT2
As shown in Figure 15, the internal clock pulse width is the same as the dead-time set by the RT2 pin. This
dead-time pulse is used to limit the maximum duty cycle for each of the outputs. Also, the discharge FET
connected to the RAMP pin is enabled during the dead-time every clock period. The voltages at both the
RT1/SYNC and RT2 pins are internally regulated to a nominal 2 V. Both the RRT1/SYNC and RRT2 resistors should
be located as close as possible to the device, and connected directly to the pins. Consider the tolerance of the
external resistors and the frequency tolerance indicated in the Electrical Characteristics table when determining
the worst case frequency range.
7.3.11 Synchronization Capability
The LM5037 device can be synchronized to an external clock by applying a narrow ac pulse to the RT1/SYNC
pin. The external clock must be at least 10% higher than the free-running oscillator frequency set by the
RRT1/SYNC and RRT2 resistors. If the external clock frequency is less than the programmed frequency, the device
ignores the synchronizing pulses. The synchronization pulse width at the RT1/SYNC pin must be a minimum of
15 ns wide. The synchronization signal should be coupled into the RT1/SYNC pin through a 100 pF capacitor or
16
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Feature Description (continued)
another value small enough to ensure the sync pulse width at RT1/SYNC is less than 60% of the clock period
under all conditions. When the synchronizing pulse transitions from low-to-high (rising edge), the voltage at the
RT1/SYNC pin must be driven to exceed 3.0 V from its nominal 2.0 Vdc level. During the synchronization clock
signal low time, the voltage at the RT1/SYNC pin clamps at 2 V by an internal regulator. The RRT1/SYNC and RRT2
resistors are always required, whether the oscillator is free running or externally synchronized.
7.3.12 Gate Driver Outputs (OUTA and OUTB Pins)
The LM5037 device provides two alternating gate driver outputs, OUTA and OUTB. The internal gate drivers can
each source and sink 1.2-A peak each. The maximum duty cycle is inherently limited to less than 50% and is
based on the value of RRT2 resistor. As an example, if the COMP pin is in a high state, RRT1/SYNC = 15 kΩ and
RRT2 = 20 kΩ then the outputs operate at a maximum duty cycle of 96%.
7.3.13 Thermal Protection
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event the maximum rated
junction temperature is exceeded. When activated, typically at 165°C, the controller is forced into a low power
standby state with the output drivers (OUTA and OUTB) and the bias regulators (VCC and REF) disabled. This
helps to prevent catastrophic failures from accidental device overheating. During thermal shutdown, the soft-start
capacitor is fully discharged and the controller follows a normal start-up sequence after the junction temperature
falls to the operating level (140°C).
7.4 Device Functional Modes
7.4.1 Overload Protection Timer
The LM5037 device provides a current limit restart timer to disable the outputs and force a delayed restart
(hiccup mode) if a current limit condition is repeatedly sensed. The number of cycle-by-cycle current limit events
required to trigger the restart is programmed by the external capacitor at the RES pin. During each PWM cycle,
the device either sources to or sinks current from the RES pin capacitor. If no current limit is detected during a
cycle, a 8-µA discharge current sink is enabled to pull the RES pin towards ground. If a current limit is detected,
the 8-µA sink current is disabled and a 18-µA current source causes the voltage at the RES pin to gradually
increase. The device protects the converter with cycle-by-cycle current limiting while the voltage at RES pin
increases. If the RES voltage reaches the 2.0 V threshold, the following restart sequence occurs (also see
Figure 16):
• The RES capacitor and SS capacitors are fully discharged.
• The soft-start current source is reduced from 100 µA to 1 µA.
• The SS capacitor voltage slowly increases. When the SS voltage reaches approximately 1 V, the PWM
comparator produces the first narrow output pulse. After the first pulse occurs, the SS source current reverts
to the normal 100 µA level. The SS voltage increases at its normal rate, gradually increasing the duty cycle of
the output drivers.
• If the overload condition persists after restart, cycle-by-cycle current limiting begins to increase the voltage on
the RES capacitor again, repeating the hiccup mode sequence.
• If the overload condition no longer exists after restart, the RES pin remains at ground by the 8-µA current sink
and normal operation resumes.
7.4.1.1 Overload Timer Function
This section lists the modes of protection available by configuring the overload timer function.
7.4.1.1.1 Cycle-by-cycle Only
The hiccup mode can be completely disabled by connecting a zero to 50 kΩ resistor from the RES pin to AGND.
In this configuration, the cycle-by-cycle protection limits the output current indefinitely and no hiccup sequences
occurs.
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Device Functional Modes (continued)
7.4.1.1.2 Hiccup Only
The timer can be configured for immediate activation of a hiccup sequence upon detection of an overload by
leaving the RES pin open circuit. In this configuration, the first detection of current limit condition by the CS pin
comparator initiates a hiccup cycle with SS capacitor fully discharged and a delayed restart.
7.4.1.1.3 Delayed Hiccup
Connecting a capacitor to the RES pin provides a programmed interval of cycle-by-cycle limiting before initiating
a hiccup mode restart, as previously described. The dual advantages of this configuration are that a short term
overload does not cause a hiccup mode restart but during extended overload conditions, the average dissipation
of the power converter remains very low.
7.4.1.2 Externally Controlled Hiccup
The RES pin can also be used as an input. The RES pin forces the device into a delayed restart sequence when
the pin rises to a level greater than the 2.0 V hiccup threshold. For example, an external trigger for a delayed
restart sequence may come from an over-temperature protection circuit or an output over-voltage sensor.
Current
Limit
CS
Current
Sense Circuit
5V
Restart
Current
Source Logic
0.25V
CLK
18 PA
RES
C RES
8 PA
SS
Voltage
Feedback
COMP
2.0V
To Output
Drivers
PWM
S
R Q
Restart
Latch
Drivers Off
+5V
Restart
Comparator
+5V
100 PA
SS
1PA
SS
Logic
100 mV
C SS
Drivers Off
Soft-start
LM5037
Figure 16. Current Limit Restart Circuit
Current Limit Detected
at CS
Current Limit Persists
2.0V
RES
0V
100 PA
5V
#1V
SS
1PA
OUTA
OUTB
t1
t3
t2
Figure 17. Current Limit Restart Timing
18
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Device Functional Modes (continued)
7.4.2 Topology and Control Algorithm Choice
The LM5037 device has all the features required to implement double-ended power converter topologies such as
push-pull, half-bridge and full-bridge with minimum external components. One key feature is the flexibility in
control algorithm selection. For example, the device can be used to implement either voltage mode control or
current mode control. Designers familiar with these topologies recognize that conventionally, current mode
control is used for push-pull and full-bridge topologies while voltage mode control is required for the half-bridge
topology. In limited applications, voltage mode control can be used for push-pull and full-bridge topologies as
well, with special care to maintain flux balance, such as using a dc-blocking capacitor in the primary (full-bridge).
The goal of this section is to illustrate implementation of both current mode control and voltage mode control
using the LM5037 device and aid the designer in the design process.
7.4.3 Voltage Mode Control
An external resistor (RFF) and capacitor (CFF) connected to VIN, AGND, and the RAMP pins is required to create
a saw-tooth modulation ramp signal shown in Figure 18. The slope of the signal at RAMP varies in proportion to
the input line voltage. The varying slope provides line feed-forward information necessary to improve line
transient response with voltage mode control. With a constant error signal, the on-time (tON) varies inversely with
the input voltage (VIN) to stabilize the Volt • Second product of the transformer primary. Using a line feed-forward
ramp for PWM control requires very little change in the voltage regulation loop to compensate for changes in
input voltage, as compared to a fixed slope oscillator ramp. Furthermore, voltage mode control is less susceptible
to noise and does not require leading edge filtering, and is therefore a good choice for wide input range power
converters. Voltage mode control requires a more complicated compensation network, due to the complexconjugate poles of the L-C output filter.
In push-pull and full-bridge topologies, any asymmetry in the volt-second product applied to primary in one phase
may not be cancelled by subsequent phase, possibly resulting in a dc current build-up in the transformer, which
pushes the transformer core towards saturation. Special care in the transformer design, such as gapping the
core, or adding ballasting resistance in the primary is required to rectify this imbalance when using voltage mode
control with these topologies. Current mode control naturally corrects for any volt-second asymmetry in the
primary.
The recommended capacitor value range for CFF is 100 pF to 1500 pF. Referring to Figure 18, it can be seen
that value CFF must be small enough such that the capacitor can be discharged within the clock (CLK) pulse
width each cycle. The CLK pulse width is same as the dead-time set by RT2. The minimum possible dead-time
for the device is 50 ns and the internal discharge FET RDS(on) is 5 Ω (typical),
The value of RFF required can be calculated from
-1
RFF =
æ
ö
V
fOSC ´ CFF ´ ln ç 1 - RAMP ÷
ç
VIN(min ) ÷
è
ø
(5)
For example, assuming a VRAMP of 1 V at VIN(min) (a good compromise of signal range and noise immunity),
oscillator frequency, fOSC of 250 kHz, VIN(min) of 24 V, and CFF = 270 pF results in a value for RFF of 348 kΩ.
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Device Functional Modes (continued)
SLOPE
PROPORTIONAL
TO Vin
Vin
R FF
VIN
1V
COMP
Gate
Drive
RAMP
CLK
C FF
LM5037
Figure 18. Feed-Forward Voltage Mode Configuration
7.4.4 Current Mode Control
The LM5037 device can be configured in current mode control by applying the primary current signal into the
RAMP pin. One way to achieve this is shown in Figure 19, which depicts a simplified push-pull converter. The
primary current is sensed using a sense resistor and the current information is then filtered and applied to the
RAMP pin through capacitor Cslope, for use as the modulation ramp. It can be seen that the signal applied to the
RAMP pin consists of the primary current information from the CS pin plus an additional ramp for slope
compensation, added by Rslope and Cslope.
VREF
Rslope
RAMP
+
Vin
Q1
CLK
Q2
Cslope
LM5037
CS
Rfilter
CLK + LEB
Current
Sense
RCS
Cfilter
Figure 19. Current Mode Configuration
Current mode control inherently provides line voltage feed-forward, cycle-by-cycle current limiting and ease of
loop compensation as it removes the additional pole due to output inductor. Also, in push-pull and full-bridge
converters, current mode control inherently balances volt-second product in both the phases by varying the duty
cycle as needed to terminate the cycle at the same peak current for each output phase. For duty cycles greater
than 50% (25% for each phase), peak current mode controlled circuits are subject to sub-harmonic oscillation.
20
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Device Functional Modes (continued)
Sub-harmonic oscillation is normally characterized by observing alternating wide and narrow duty cycles at the
controller output. Adding an artificial ramp (slope compensation) to the current sense signal eliminates this
potential oscillation. Current mode control is also susceptible to noise and layout considerations. It is
recommended that CFilter and Cslope be placed as close to the IC as possible to avoid any noise pickup and trace
inductance. When the converter is operating at low duty cycles and light load, the primary current amplitude is
small and is susceptible to noise. The artificial ramp, added to avoid sub-harmonic oscillations, provides
additional benefits by improving the noise immunity of the converter.
8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Input Supply Voltage (VIN and VCC pins)
The voltage applied to the VIN pin, which may be the same as the system voltage applied to the power
transformer’s primary (VPWR), can vary from 8 V to 100 V. The current into the VIN pin depends primarily on the
gate charge provided by the output drivers, the switching frequency, and any external loads on the VCC and
REF pins. This design uses the filter shown in Figure 20 to suppress transients that may occur at the input
supply. A filter is particularly important when VIN is operated close to the maximum operating rating of the
LM5037.
When power is applied to VIN and the UVLO pin voltage is greater than 0.45 V, the VCC regulator is enabled
and supplies current into an external capacitor connected to the VCC pin. When the voltage on the VCC pin
reaches the regulation point of 7.7 V, the voltage reference (REF) enables. The reference regulation set point is
5 V. The outputs (OUTA and OUTB) enable when the two bias regulators reach their set point and the UVLO pin
potential is greater than 1.25 V. In typical applications, an auxiliary transformer winding connects through a diode
to the VCC pin. In order to shut off the internal start-up regulator, this winding must raise the VCC voltage above
8.1 V.
After the outputs are enabled and the external VCC supply voltage has begun supplying power to the device, the
current into the VIN pin drops below 1 mA. VIN should remain at a voltage equal to or above the VCC voltage to
avoid reverse current through protection diodes.
VPWR
50
VIN
LM5037
0.1 PF
Figure 20. Input Transient Protection
8.1.2 100-V (or Higher) Input Voltage Applications
For applications where the system input voltage exceeds 100 V or the device power dissipation is of concern, the
LM5037 device can be powered from an external start-up regulator as shown in Figure 21. This configuration
shows the VIN and the VCC pins connected together. The voltage at the VCC and VIN pins must be greater than
8.1 V (> VCCMAX reference voltage) and not exceed 15 V. Use an auxiliary winding to reduce the power
dissipation in the external regulator after the power converter activates. The N-P-N base-emitter reverses
breakdown voltage, which can be as low as 5 V for some transistors. Consider this breakdown voltage when
selecting the transistor.
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Application Information (continued)
8.1V to 15V
V PWR
VIN
from aux winding
VCC
LM5037
9V
Figure 21. Start-up Regulator for VPWR >100 V
8.1.3 Current Sense
The CS pin receives an input signal representative of the transformer primary current, either from a current sense
transformer or from a resistor in series with the source of the OUTA and OUTB MOSFET switches. In both
cases, the sensed current creates a voltage ramp across R1, and the RFCF filter suppresses noise and transients
as shown in Figure 22 and Figure 23. Locate components R1, RF and CF as close to the device as possible. Use
a dedicated track from the current sense transformer (R1) to the ground connection (AGND pin). Ensure that the
current sense components provide greater than 220 mV at the CS pin when an over-current condition exists.
V PWR
Current Sense
Power Transformer
Q1
VIN
CS
RF
CF
LM5037
R1
AGND
OUTA
Level
Shift
Q2
OUTB
Figure 22. Current Sense Using Transformer
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Application Information (continued)
Power Transformer
VPWR
Vin
OUTA
Q1
Q2
OUTB
RF
CS
LM5037
CF
R1
Figure 23. Current Sense Using Current Sense Resistor (R1)
Configuration and component selection for current mode control is recommended as follows: The current sense
resistor is selected such that during over current condition, the voltage across the current sense resistor is above
the minimum CS threshold of 220 mV. It is recommended to set the impedances of RFilter and CFilter as seen from
Cslope at relatively low values, so that the slope compensation is primarily dictated by Rslope and Cslope
components. For example, if the filtering time (RFilter and CFilter) for leading edge noise is selected for 50 ns and if
the value selected for RFilter = 25Ω, then
CFilter =
50 x 10-9
3 x 25:
(6)
Equation 6 results in a value of CFilter = 680 pF (approximated to a standard value). In general, the amount of
slope compensation required to avoid sub-harmonic oscillation is equal to at least one-half the down-slope of the
output inductor current, transformed to the primary. To mitigate sub-harmonic oscillation after one switching
period, the slope compensation has to be equal to one times the down slope of the filter inductor current
transformed to primary. This is known as deadbeat control. For circuits where primary current is sensed using a
resistor, the amount of slope compensation for dead-beat control required can be calculated from:
Slope-Comp =
Turns-Ratio x Vout x RCS
FOSC x Lfilter
where
•
turns-ratio is referred with respect to the primary
(7)
For example, for a 5-V output converter with a turns ratio between secondary and primary of 1:2, an oscillator
frequency (fOSC) of 250 kHz, a filter inductance of 4 µH (LFilter) and a current sense resistor (RCS) of 32 mΩ, slope
compensation of 80 mV suffices. The slope compensation "volts" that results from the above expression is the
maximum voltage of the artificial ramp added linearly to the RAMP pin till the end of maximum switching period.
For circuits where a current sense transformer is used for primary current sensing, the turns-ratio of the current
sense transformer has to be taken into account.
Cslope should be selected such that it can be fully discharged by the internal RAMP discharge FET. Capacitor
values ranging from 100 pF to 1500 pF are recommended. The value must be small enough such that the
capacitor can be discharged within the clock (CLK) pulse width each cycle.
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Application Information (continued)
Rslope can be selected from the following formula:
-1
Rslope =
- Rfilter
FOSC x Cslope x In 1 -
Slope-Comp
VREF
(8)
For example, with a Cslope of 1500 pF, FOSC of 250 kHz, reference voltage of 5V (VREF), slope compensation of
80 mV and Rfilter = 25Ω results in Rslope value of 165 kΩ.
8.1.4 UVLO Divider Selection
A dedicated comparator connected to the UVLO pin detects an input under-voltage condition. When the UVLO
pin voltage is below 0.45 V, the LM5037 controller is in a low current shutdown mode. For a UVLO pin voltage
greater than 0.45 V but less than 1.25 V, the controller is in standby mode with VCC and REF regulators active
but no switching. Once the UVLO pin voltage is greater than 1.25 V, the controller is fully enabled. When the
UVLO pin voltage rises above the 1.25-V threshold, an internal 22-µA current source as shown in Figure 24, is
activated thus providing threshold hysteresis. The 22-µA current source is deactivated when the voltage at the
UVLO pin falls below 1.25 V. Use Equation 9 to calculate resistance values for R1 and R2.
R1 = VHYS -
20 x 10-3 x VPWR
1.25
22 PA
R2 =
1.25 x R1
VPWR - 1.25
where
•
•
VPWR is the desired turn-on voltage
VHYS is the desired UVLO hysteresis at VPWR
LM5037
5.0V
VIN
(9)
22 PA
R1
UVLO
R2
1.25V
STANDBY
0.45V
SHUTDOWN
Figure 24. Basic UVLO Configuration
For example, if the device is to be enabled when VPWR reaches 33 V, and disabled when VPWR decreases to 30
V, R1 is 113 kΩ, and R2 is 4.42 kΩ.
CAUTION
Do not allow the voltage at the UVLO pin to exceed 7 V at any time.
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Application Information (continued)
Ensure that both the power and voltage ratings are (0603 resistors can be rated as low as 50 V) for the selected
R1 resistor. Maintain the UVLO threshold accuracy, by using a resistor tolerance of 1% or better.
Remote control of the LM5037 operational modes can be accomplished with open drain device(s) connected to
the UVLO pin as shown in Figure 25.
LM5037
5.0V
VIN
22 PA
R1
1.25V
UVLO
STANDBY
OFF
STANDBY
R2
0.45V
SHUTDOWN
Figure 25. Remote Standby and Disable Control
8.1.5 Hiccup Mode Current Limit Restart (RES Pin)
The basic operation of the hiccup mode current limit is described in the functional description. The delay time to
the initiation of a hiccup cycle is programmed by the selection of the RES pin capacitor CRES as illustrated in
Figure 26.
Current Limit Detected
at CS
Current Limit Persists
2.0V
RES
0V
100 PA
5V
#1V
SS
1PA
OUTA
OUTB
t1
t3
t2
Figure 26. Hiccup Over-Load Restart Timing
In the case of continuous cycle-by-cycle current limit detection at the CS pin, the time required for CRES to reach
the 2.0 V hiccup mode threshold is:
t1 =
CRES x 2.0V
= 111K x CRES
18 PA
(10)
For example, if CRES = 0.01 µF the time t1 is approximately 2.0 ms. The cool down time, t2 is set by the soft-start
capacitor (CSS) and the internal 1 µA SS current source, and is equal to:
t2 =
CSS x 1V
= 1M x CSS
1 PA
(11)
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Application Information (continued)
If CSS = 0.01 µF, t2 is ≊10 ms.
The soft-start time t3 is set by the internal 100 µA current source, and is equal to:
t3 =
CSS x 4V
= 40K x CSS
100 PA
(12)
If CSS = 0.01 µF, t3 is ≊ 400 µs.
The time t2 provides a periodic cool-down time for the power converter in the event of a sustained overload or
short circuit. This off time results in lower average input current and lower power dissipation within the power
components. It is recommended that the ratio of t2 / (t1 + t3) be in the range of 5 to 10 to take advantage of this
feature.
If the application requires no delay from the first detection of a current limit condition to the onset of the hiccup
mode (t1 = 0), the RES pin can be left open (no external capacitor). To disable the hiccup mode entirely, connect
the RES pin to ground (AGND).
8.2 Typical Application
Figure 27 shows an example of an LM5037-controlled 50-W half-bridge converter. The converter provides a
single regulated 5-V output at 10 A, from a standard Telecoms 36-V to 72-V input. The converter is configured
for feed-forward voltage-mode control. An auxiliary winding on the power transformer is used to supply the VCC
voltage externally, to reduce the power dissipation in the device.
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Figure 27. Typical Application Schematic, LM5037
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8.2.1 Design Requirements
• Operating input voltage range: 36 V to 72 V
• Output voltage: 5 V
• Output current: 10 A
• UVLO On Level: 34 V On (rising)
• UVLO Off Level: 30 V Off (falling)
• Output ripple voltage, VRIPPLE(OUT): < 2% (960 mVP-P)
• Oscillator frequency ( 2× fSW per phase): 300 kHz
• Switching frequency ( fSW per phase): 150 kHz
8.2.2 Detailed Design Procedure
8.2.2.1 Oscillator Frequency and Maximum Duty Cycle
The LM5037 oscillator frequency is twice the switching frequency of each switch in the half-bridge power stage.
fOSC = 2 × fSW
(13)
Calculate the dead-time resistor value for RRT2. The recommended of dead-time range is between 50 ns and 250
ns. A value of 175 ns is chosen, which sets a maximum duty cycle of approximately 95%. Equation 14 calculates
the RRT2 resistor value, (R9 in Figure 27) .
æ t
ö 175 ´ 10-9
= 35 kW
RRT2 = ç DEAD ÷ =
5 ´ 10-12
è 5 ´ 10-12 ø
(14)
Use the nearest standard E96 value of 34.8 kΩ.
Use the resistor value on the RT2 pin to calculate the required resistor value for the RT1 pin (R8 in Figure 27)
using Equation 15.
ææ 1 ö
ö ææ
ö
ö
1
çç
- 175 ´ 10-9 ÷
÷ - tDEAD ÷ ç ç
3 ÷
f
ç OSC ø
÷ ç è 300 ´ 10 ø
÷ = 19.5 kW
RRT1 = ç è
=
-9 ÷ ç
-9
÷
0.162 ´ 10
ç 0.162 ´ 10
÷ ç
÷÷
ç
ç
÷ è
ø
è
ø
(15)
Use the nearest E96 value of 20 kΩ.
8.2.2.2 Power Stage Design
As
•
•
•
•
•
shown in the schematic in Figure 27, the primary components of the half-bridge power stage are:
Half-bridge splitter capacitors (C1, C2, C3, C4, C5 and C6)
Power MOSFETs (Q1 and Q2)
Power transformer (T1)
Output rectifier (D3)
Output filter (L2 and C18, C19, and C20)
The half-bridge stage DC input voltage divides evenly across the splitter capacitors, so that one end of the
primary-side of the power transformer connects to a DC level of approximately VIN/2. The other end of the
transformer primary is alternately connected by Q1 to the VIN pin, and then by Q2 to GND, with appropriate
dead-time in between. The device modulates the dead-time or duty cycle in order to regulate the output voltage
to the required level. Thus the primary winding is subjected to a bi-polar voltage swing of ±VIN/2. This voltage is
then scaled by the secondary to primary turns ratio (NS/NP). The secondary winding is center-tapped, so that the
double-diode, D3 can then full-wave rectify the bi-polar secondary waveform. A uni-polar pulse train occurs at the
full fOSC frequency of 300 kHz at the cathode of D3. Output filter inductor L2 and output capacitors C18, C19,
C20 then filter this pulse train to a DC output voltage plus AC ripple at the fOSC frequency.
The PWM controller adjusts the duty cycle with input line voltage in order to regulate the output voltage. The
maximum duty cycle occurs at minimum operating input voltage, which is approximately 30 V (UVLO turn-off
point).
28
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æ 2 ´ VOUT ´ NP ö æ 2 ´ 5 ´ 2 ö
D=ç
÷=ç
÷ = 67%
è VIN ´ NS
ø è 30 ´ 1 ø
(16)
Equation 16 shows that there is sufficient tolerance to the oscillator 95% DMAX setting (set by the RT2 resistor
value). Use Equation 17 to estimate the peak-to-peak ripple current (IP-P) in continuous conduction mode (CCM)
once the duty cycle (D), fOSC frequency and turns ratio, and output inductance (LOUT) are calculated or chosen.
DIP _ P
ææ V
ö
N ö
çç ç IN ´ S ÷ - VOUT ÷÷
è 2 NP ø
ø´ D =
=è
LOUT
fOSC
2
æ
æ 2´ V
öö
OUT ´ NP ÷ ÷
ç VOUT - ç
ç
ç
VIN
NS ÷ø ÷
è
è
ø = 1.24 A
P _P
LOUT ´ fOSC
(17)
8.2.2.3 Half-Bridge MOSFET Driver
Because this application uses half-bridge power stage MOSFETs connected in series between the VIN pin and
GND, the Q1 device is high-side or floating. No high-side floating driver or bootstrap circuit which are necessary
to drive Q1 exists in this device. Both of the outputs (OUTA and OUTB) are low-side or ground referenced.
This design uses an external high-side and low-side half-bridge driver device (U2, LM5100) to interface between
the gate drive outputs and the actual gates of Q1 and Q2. The design requires a bootstrap capacitor (C16) to
generate the necessary high-side or floating bias supply for the high-side driver section of the LM5100.
8.2.2.4 UVLO Setting
To ensure start-up at the required minimum system input voltage of 34 V, with the 4 V of hysteresis to the
desired turn-off level, calculate the UVLO divider resistors R5 and R6 using Equation 18 and Equation 19.
æ
æ 20 ´ 10-3 ´ V ö ö æ
æ 20 ´ 10-3 ´ 34 ö ö
IN ÷ ÷ ç
ç VHYS - ç
4-ç
÷÷
ç
÷÷ ç
ç
÷÷
1.25
1.25
ç
è
ø
è
ø = 157.1 kW
R5 = ç
÷=ç
÷
m
m
22
A
22
A
ç
÷ ç
÷
çç
÷÷ çç
÷÷
ø
è
ø è
(18)
Round this calculated value to the more convenient value of 150 kΩ.
æ 1.25 ´ RR5
R6 = ç
è VIN - 1.25
ö 1.25 ´ 150 kW
= 5.73 kW
÷=
34 - 1.25
ø
(19)
Use the nearest standard E96 value of 5.76 kΩ.
8.2.2.5 VIN, VCC, Start-Up
To reduce the power dissipation in the internal start-up regulator on the VIN pin, use a separate external VCC
supply. Usually, it is the auxiliary winding on the transformer that derives this external VCC supply. The auxiliary
to secondary turns ratio is 2:1, so when the output voltage regulates at 5 V, the auxiliary VCC voltage
approximately 10 V. This is sufficiently greater than the maximum internal VCC regulator level of 8 V to back-bias
the internal regulator after start-up.
8.2.2.6 Voltage-Mode Ramp Input
Because this design uses voltage-mode control, connect an R-C network from the system input voltage to the
RAMP pin, R4 and C9 as shown in Figure 27. Use Equation 5 to calculate the required R-C values. Using a
value of 1 nF for C9, and targeting a ramp amplitude of 850 mV at VIN(min) (a good compromise between signal
range and noise immunity), Equation 20 calculates the required value R4.
æ
ö
ç
÷ æ
ö
ç
÷ ç
÷
-1
-1
÷=ç
÷ = 139.5 kW
R4 = ç
ç
÷ ç
÷
0.85
æ
ö
æ
ö
V
300 kW ´ 1 nF ´ ln ç 1 çf
´ C9 ´ ln ç 1 - RAMP ÷ ÷ ç
36 ÷ø ÷ø
è
ç OSC
ç
VIN(min ) ÷ ÷ è
è
øø
è
(20)
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Use the nearest E96 value of 140 kΩ.
8.2.2.7 Soft-Start Delay
Use Equation 21 to calculate the time period from soft-start delay to commencement of first PWM switching.
æ 1.0 V ´ C12 ö æ 1.0 V ´ 0.1 mF ö
tSS(dly ) = ç
÷=ç
÷ = 1.0 ms
100 mA
è 100 mA ø è
ø
(21)
After the soft-start delay period, the soft-start ramp time depends on the power stage design and the operating
conditions (input voltage and output load).
8.2.2.8 Overload Timer
With a timing capacitance of 10 nF on the RES pin (C13), calculate hiccup-mode timing and duty cycle for a
sustained over-current condition using Equation 22 and Equation 23. Time period t1 describes the hiccup-mode
current-limit persist time and time period t2 describes hiccup-mode cool-down off-time.
æ 2.0 V ´ C13 ö æ 2.0 V ´ 10 nF ö
t1 = ç
÷=ç
÷ = 1.11 ms
18 mA
è 18 mA ø è
ø
(22)
æ 1.0 V ´ C12 ö æ 1.0 V ´ ´100 nF ö
t2 = ç
÷=ç
÷ = 100 ms
1 mA
1 mA
è
ø è
ø
(23)
Calculate the hiccup-mode duty cycle using Equation 24.
t1
1.11
DHICCUP =
=
= 1.09%
t1 + t2 + tSS 1.11 + 100 + 1
(24)
8.2.2.9 Current Sense
In order to improve the efficiency, a current sense transformer (T2) is used. This transformer uses a 1:100 stepdown ratio. This ratio reduces the power dissipation in the current-sense resistor, R11.
Set the current-limit point after calculating the
• output inductor peak-to-peak ripple current when the device operates in CCM
• the turns ratios of the main transformer (NS/NP)
• the turns ratios of the current-sense transformer (CSR)
Using the full load output current of 10 A, and the current limit target of 150%, or 15 A, calculate the required
value for R11. The R11 resistance must generate a voltage at the CS pin to equal the internal cycle-by-cycle limit
of nominally 0.25 V at the current limit level at the output.
0.25 V
0.25 V
=
= 3.2 W
R11 =
DI ö N S
1
1.24 ö 1
1
æ
æ
ç IILIM + 2 ÷ ´ N ´ CSR ç 15 + 2 ÷ ´ 2 ´ 100
è
ø
è
ø
P
(25)
Round the calculated R11 resistance to a standard 3 Ω. If the power stage operates heavily in CCM, the output
inductor ripple component can be ignored to yield a good first-order approximation to the required value.
8.2.2.10 Output Voltage Feedback
Because the output of the DC-DC converter is isolated from the input, this design uses a secondary-side
reference (U5) and error amplifier (U3). The output of the error amplifier, U3 represents the required demand
level to maintain regulation as a function of output load and input line. The design couples the demand signal
across the isolation barrier to the primary through opto-coupler, U6.
Because the error amplifier and reference reside on the secondary side, this design disregards the internal
reference and error amplifier features of the LM5037 device. The FB pin of the LM5037 device is connected to
GND, which forces the COMP pin to pull up to approximately 5 V through the internal 5-kΩ pull-up resistance.
The opto-coupler, U6 then externally pulls down on the COMP pin to set the required level to achieve the
required duty cycle at any given load or line level.
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8.2.3 Application Curves
VIN = 48 Vdc
IOUT = 10 A
Figure 28. Output Voltage During Soft-Start Period
VIN = 36 Vdc
IOUT = 10 A
VIN = 48 Vdc
1 V/div
Time = 1 ms/div
V/div = 1 V
Time = 1 µs/div
Figure 29. Output Voltage During Soft-Stop Period
V/div = 10 V
Time = 2 µs/div
Figure 30. Drain Waveform of Q2
VIN = 72 Vdc
IOUT = 10 A
V/div = 20 V
Time = 2 µs/div
Figure 31. Drain Waveform of Q2
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VIN = 48 Vdc
IOUT = 10 A
Bandwidth Limit = 20 MHz
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VOUT(P-P), 50 mV/div
Time = 5 µs/div
VIN = 48 Vdc
IOUT Step, 5 A to 10 A
Time = 200 µs/div
Figure 32. Output Ripple
Top Trace: IOUT Step, 5 A to 10 A
Bottom Trace: VOUT, 100 mV/div
Bandwidth Limit = 20 MHz
Figure 33. Transient Response
fSW = 250 kHz
Figure 34. Efficiency vs. Load Current
9 Power Supply Recommendations
The VCC pin requires a local decoupling capacitor that is connected to GND. This capacitor ensures stability of
the internal regulator from the VIN pin. The decoupling capacitor also provides the current pulses to drive the
gates of the external MOSFETs through the driver output pins. Place the decoupling capacitor close to the VCC
and PGND pins and track it directly to those pins.
The two ground pins (PGND and AGND) must be connected together with a short, direct PCB connection.
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10 Layout
10.1 Layout Guidelines
The LM5037 device current sense and PWM comparators are very fast, and respond to short duration noise
pulses. Place components for the CS, COMP, SS, UVLO, RT2 and the RT1/SYNC pins as physically close as
possible to the device. This placement minimizes noise pickup on the PC board trace inductances.
Layout considerations are critical for the current sense filter. If a current sense transformer is used, both leads of
the transformer secondary should be routed to the sense filter components and to the device pins. The ground
side of the transformer should be connected via a dedicated PC board trace to the AGND pin, rather than
through the ground plane.
If the current sense circuit employs a sense resistor in the drive transistor source, low inductance resistors
should be used. In this case, all the noise sensitive, low-current ground trace should be connected in common
near the device, and then a single connection made to the power ground (sense resistor ground point).
While employing current mode control, RAMP pin capacitor and CS pin capacitor must be placed close to the
device. Also, a short direct trace should be employed to connect RAMP capacitor to the CS pin.
The gate drive outputs of the device should have short, direct paths to the power MOSFETs in order to minimize
inductance in the PC board The two ground pins (AGND, PGND) must be connected together with a short, direct
connection, to avoid jitter due to relative ground bounce.
If the internal dissipation of the device produces high junction temperatures during normal operation, the use of
multiple vias under the device to a ground plane can help conduct heat away from the device. Judicious
positioning of the PC board within the end product, along with use of any available air flow (forced or natural
convection) helps reduce the junction temperatures. If using forced air cooling, avoid placing the device in the
airflow shadow of tall components, such as input capacitors.
10.2 Layout Example
From VIN
RAMP
VIN
UVLO
REF
COMP
VCC
FB
From FB
OUTA
To GDA
RT2
OUTB
To GDB
AGND
PGND
RT1
CS
From CS
LM5037
SS
RES
Figure 35. LM5037 Board Layout
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11 Device and Documentation Support
11.1 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.2 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.3 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.4 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
www.ti.com
13-Jan-2015
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
LM5037MT/NOPB
ACTIVE
TSSOP
PW
16
92
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
LM5037
MT
LM5037MTX/NOPB
ACTIVE
TSSOP
PW
16
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
LM5037
MT
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
13-Jan-2015
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2015
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
LM5037MTX/NOPB
Package Package Pins
Type Drawing
TSSOP
PW
16
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
2500
330.0
12.4
Pack Materials-Page 1
6.95
B0
(mm)
K0
(mm)
P1
(mm)
5.6
1.6
8.0
W
Pin1
(mm) Quadrant
12.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2015
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM5037MTX/NOPB
TSSOP
PW
16
2500
367.0
367.0
35.0
Pack Materials-Page 2
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