Part 5

Part 5
High Frequency Design
From January 2004 High Frequency Electronics
Copyright © Summit Technical Media, LLC
RF and Microwave Power
Amplifier and Transmitter
Technologies — Part 5
By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington,
Zoya B. Popovich, Nick Pothecary, John F. Sevic and Nathan O. Sokal
The ever-increasing
demands for more bandwidth, coupled
requirements for both
high linearity and high
efficiency create everincreasing challenges in
the design of power
amplifiers and transmitters. A single W-CDMA
signal, for example, taxes the capabilities of a
Kahn-technique transmitter with a conventional class-S modulator. More acute are the
problems in base-station and satellite transmitters, where multiple carriers must be
amplified simultaneously, resulting in peakto-average ratios of 10 to 13 dB and bandwidths of 30 to 100 MHz.
A number of the previously discussed techniques can be applied to this problem, including
the Kahn EER with class-G modulator or splitband modulator, Chireix outphasing, and
Doherty. This section presents some emerging
technologies that may be applied to wideband,
high efficiency amplification in the near future.
Emerging techniques are
examined in this final
installment of our series on
power amplifier technologies, providing notes on
new modulation methods
and improvements in
linearity and efficiency
RF Pulse-Width Modulation
Variation of the duty ratio (pulse width) of
a class-D RF PA [112] produces an amplitudemodulated carrier (Figure 59). The output
envelope is proportional to the sine of the
pulse width, hence the pulse width is varied in
proportion to the inverse sine of the desired
envelope. This can be accomplished in DSP, or
by comparison of the desired envelope to a
full-wave rectified sinusoid. The pulse timing
Figure 59 · RF pulse-width modulation.
conveys signal phase information as in the
Kahn and other techniques.
Radio-frequency pulse-width modulation
(RF PWM) eliminates the series-pass losses
associated with the class-S modulator in a
Kahn-technique transmitter. More importantly, the spurious products associated with
PWM are located in the vicinity of the harmonics of the carrier [113] and therefore easily removed. Consequently, RF PWM can
accommodate a significant RF bandwidth
with only a simple, low-loss output filter.
Ideally, the efficiency is 100 percent. In
practice, switching losses are increased over
those in a class-D PA with a 50:50 duty ratio
because drain current is nonzero during
This series of articles is an expanded version of the paper, “Power Amplifiers and Transmitters for RF and
Microwave” by the same authors, which appeared in the the 50th anniversary issue of the IEEE Transactions on
Microwave Theory and Techniques, March 2002. © 2002 IEEE. Reprinted with permission.
High Frequency Electronics
High Frequency Design
Figure 60 · Current-switching PA for 1 GHz (courtesy
Figure 61 · Prototype class-D PA for delta-sigma modulation (courtesy UCSD).
Previous applications of RF PWM have been limited to
LF and MF transmitters (e.g., GWEN [114]). However, the
recent development of class-D PAs for UHF and
microwave frequencies (Figure 60) offers some interesting
average of the cycles in the PA. Phase is again conveyed
in pulse timing.
The delta-sigma modulator employs an algorithm
such as that shown in Figure 63. The signal is digitized by
a quantizer (typically a single-bit comparator) whose output is subtracted from the input signal through a digital
feedback loop, which acts as a band-pass filter. Basically,
the output signal in the pass band is forced to track the
desired input signal. The quantizing noise (associated
with the averaging process necessary to obtain the
desired instantaneous output amplitude) is forced outside
of the pass band.
The degree of suppression of the quantization noise
depends on the oversampling ratio; i.e., the ratio of the
digital clock frequency to the RF bandwidth and is relatively independent of the RF center frequency. An example of the resultant spectrum for a single 900-MHz carrier and 3.6-GHz clock is shown in Figure 64. The quantization noise is reduced over a bandwidth of 50 MHz,
which is sufficient for the entire cellular band. Out-ofband noise increases gradually and must be removed by a
band-pass filter with sufficiently steep skirts.
As with RF PWM, the efficiency of a practical deltasigma modulated class-D PA is reduced by switching losses associated with nonzero current at the times of switching. The narrow-band output filter may also introduce
significant loss.
Delta-Sigma Modulation
Delta-sigma modulation is an alternative technique
for directly modulating the carrier produced by a class-D
RF PA (Figure 61) [PA8],[PA9]. In contrast to the basically analog operation of RF PWM, delta-sigma modulation
drives the class-D PA at a fixed clock rate (hence fixed
pulse width) that is generally higher than the carrier frequency (Figure 62). The polarity of the drive is toggled as
necessary to create the desired output envelope from the
Carrier Pulse-Width Modulation
Figure 62 · Delta-sigma modulation.
High Frequency Electronics
Carrier pulse-width modulation was first used in a
UHF rescue radio at Cincinnati Electronics in the early
1970s. Basically, pulse-width modulation as in a class-S
modulator gates the RF drive (hence RF drain current) on
and off in bursts, as shown in Figure 65. The width of each
burst is proportional to the instantaneous envelope of the
Figure 63 · Delta-sigma modulator.
Power Spectral Density (dBm/Hz)
Figure 65 · Carrier pulse-width modulation.
Power Recovery
Frequency (MHz)
Figure 64 · Spectrum of delta-sigma modulation.
desired output. The amplitude-modulated output signal is
recovered by a band-pass filter that removes the sidebands associated with the PWM switching frequency. The
PWM signal can be generated by a comparator as in a
class-S modulator or by delta-sigma techniques.
As with RF PWM and delta-sigma modulation, the
series-pass losses and bandwidth limitations of the highlevel modulator are eliminated. The switching frequency
in carrier PWM is not limited by capabilities of powerswitching devices and can therefore easily be tens of
MHz, allowing large RF bandwidths. A second advantage
is that carrier PWM can be applied to almost any type of
RF PA. A disadvantage is that a narrow-band output filter with steep skirts is required to remove the switchingfrequency sidebands, and such filters tend to have losses
of 1 to 2 dB at microwave frequencies. Nonetheless, the
losses in the filter may be more than offset by the
improvement in efficiency for signals with high peak-toaverage ratios.
A number of RF processes result in significant RF
power dissipated in “dump” resistors. Examples include
power reflected from a mismatched load and dumped by a
circulator and the difference between two inputs of hybrid
combiner dumped to the balancing resistor. The notion of
recovering and reusing wasted RF power was originally
applied to the harmonics (18 percent of the output power)
of an untuned LF class-D PA [117].
More recently, power recovery has been applied to outphasing PAs with hybrid combiners [118, 119]. The
instantaneous efficiency of such a system depends upon
both the efficiency of the PA and that of the recovery system. Since the two PAs operate at full power regardless of
the system output, inefficiency in the PA has a significant
impact upon the system efficiency at the lower outputs.
Nonetheless, a significant improvement over conventional hybrid-coupled outphasing is possible, and the PAs are
presented with resistive loads that allow them to operate
optimally. Typically, 50 percent of the dumped power can
be recovered.
The power-recovery technology can also be used to
implement miniature DC-DC converters. Basically, a
high-efficiency RF-power amplifier (e.g., class-E) converts
DC to RF and a high-efficiency rectifier circuit converts
the RF to DC at the desired voltage. Implementation at
microwave frequencies reduces the size of the tuning and
filtering components, resulting in a very small physical
size and high power density. In a prototype that operates
at C band [120], the class-E PA uses a single MESFET to
produce 120 mW with a PAE of 86 percent. The diode rectifier consists of a directional coupler with two Schottky
January 2004
High Frequency Design
Figure 66 · Switched PAs with quarter-wavelength
transmission line combiner.
Figure 67 · Instantaneous efficiency of switched PAs.
diodes connected at the coupled and through ports and
has a 98-percent conversion efficiency and an overall efficiency (including mismatch loss) of 83 percent. For a typical DC output of 3 V, the DC-DC conversion efficiency is
64 percent.
Switched PAs with Transmission-Line Combiners
RF-power amplifiers cannot simply be connected in
series or parallel and switched on and off to make a transmitter module that adapts to variable peak envelope
power. Attempting to do so generally produces either little effect or erratic variations in load impedance, sometimes leading to unstable operation and destruction of the
transistors. Systems of microwave PAs that are toggled
on and off are therefore connected through networks of
quarter-wavelength transmission lines. The Doherty
transmitter (discussed in part 4 of this series) is a classic
example of this sort of technique.
An alternative topology (Figure 66) uses shorting
switches and quarter-wavelength lines to to decouple offstate PAs [121, 122]. The inactive PA is powered-down (by
switching off its supply voltage), after which its output is
shorted to ground. The quarter-wavelength line produces
an open circuit at the opposite end where the outputs
from multiple PAs are connected together to the load.
This technique may be more easy to implement (especially for multiple PAs) than Doherty because a short is more
readily realized than an open.
If PA #1 is the only PA active, its load is simply R0. If
both #1 and #2 are active, the combination produces an
effective load impedance of 2Ro at the load ends of the
lines. Inversion of this impedance through the lines
places loads of R0/2 on the RF PAs. Consequently, the
peak power output for two active PAs is four times that
with a single PA. As in discrete envelope tracking, the RF
PAs operate as linear amplifiers. The number of PAs that
High Frequency Electronics
are active is the minimum needed to produce the current
output power. The peak power is thus kept relatively close
to the saturated output, eliminating most of the effects of
operating in back-off. The efficiency can therefore reach
PEP efficiency at a number of different output levels, as
shown in Figure 67.
The advantage of this technique is the ease in design
associated with relying on short circuits rather than open
circuits to isolate the off-state PAs. A possible disadvantage is operating individual PAs from multiple load
impedances without retuning and a limited number of
power steps available (e.g., 9/9, 4/9, 1/9 for a three-PA system).
Electronic Tuning
The performance of virtually all power amplifiers is
degraded by load- impedance mismatch. Mismatched
loads not only reduce efficiency, but also create higher
stresses on the transistors. Because high-efficiency PAs
generally require a specific set of harmonic impedances,
their use is often restricted to narrow-band applications
with well-defined loads.
Electronic tuning allows frequency agility, matching of
unknown and variable loads, and amplitude modulation.
Components for electronic tuning include pin-diode
switches, MEMS switches, MEMS capacitors, semiconductor capacitors, ceramic capacitors (e.g., BST), and biascontrolled inductors. To date, electronic tuning has been
applied mainly to small-signal circuits such as voltagecontrolled oscillators. Recently demonstrated, however,
are two electronically tuned power amplifiers. One operates in class E, produces 20 W with an efficiency of 60 to
70 percent, and can be tuned from 19 to 32 MHz (1.7:1
range) through the use of voltage-variable capacitors
High Frequency Design
Figure 68 · Electronically tunable class-D PA (courtesy
[123, 124]. The second (Figure 68) operates in class D, produces 100 W with an efficiency of 60 to 70 percent, and
can be tuned from 5 to 21 MHz (4.25:1 range) through the
use of electronically tunable inductors and capacitors
Load Modulation
The output of a power amplifier can be controlled by
varying the drive, gate bias, DC supply voltage, or load
impedance. “Load modulation” uses an electronically
tuned output filter (Figure 69) to vary load impedance
and thereby the instantaneous amplitude of the output
signal. The modulation bandwidth can be quite wide, as it
is limited only by the bias feeds to the tuning components.
A key aspect of load modulation is a diligent choice of
the impedance locus so that it provides both good dynamic range and good efficiency. For ideal saturated PAs of
classes A, B, C, and F, the optimum locus is the pure resistance line on the Smith chart that runs from the nominal
load to an infinite load. For ideal class-E PAs with series
inductance and shunt susceptance for optimum operation
with the nominal load, the optimum locus is the unityefficiency line running from the nominal load upward and
rightward at an angle of 65° [126]. For real PAs, the opti-
Figure 69 · Load modulation by electronic tuning.
High Frequency Electronics
Figure 70
Load-modulated class-E PA (courtesy
mum locus is found by examination of load-pull contours.
The simple T filter has a single electronically variable
element, but provides an approximately optimum locus
for class E over the top 12 dB of the dynamic range. The
experimental 20-W, 30-MHz [124, 126] shown in Figure
70 achieves a 41-dB range of amplitude variation. The
measured instantaneous-efficiency curve (Figure 71) corresponds to a factor of 2.1 improvement in the average
efficiency for a Rayleigh-envelope signal with a 10-dB
peak-to-average ratio.
A load-modulated PA for communications follows the
electronically tuned filter with a passive filter to remove
the harmonics associated with the nonlinear elements.
Predistortion compensates for the incidental phase modulation inherent in dynamic tuning of the filter. Variation
of the drive level can be used to conserve drive power and
to extend the dynamic range.
Figure 71 · Instantaneous efficiency of load modulation compared to class-B linear amplification.
High Frequency Design
112. Figures 8 and 9 in Part 2 of
Electronics, July 2003.
113. F. H. Raab, “Radio frequency
pulsewidth modulation,” IEEE Trans.
Commun., vol. COM-21, no. 8, pp.
958-966, Aug. 1973.
114. F. G. Tinta, “Direct single
sideband modulation of transmitter
output switcher stages,” Proc. RF
Expo East ’86, Boston, MA, pp. 313398, Nov. 10-12, 1986.
115. A. Jayaraman, P. F. Chen, G.
Hanington, L. Larson, and P. Asbeck,
“Linear high-efficiency microwave
power amplifiers using bandpass
delta-sigma modulators,” IEEE
Microwave Guided Wave Lett., vol. 8,
no. 3, pp. 121-123, March 1998.
116. J. Keyzer et al., “Generation
microwave signals using delta-sigma
modulation,” 2002 Int. Microwave
Symp. Digest, vol. 1, pp. 397-400,
Seattle, WA, June 2-7, 2002.
117. J. D. Rogers and J. J. Wormser,
“Solid-state high-power low frequency
telemetry transmitters,” Proc. NEC,
vol. 22, pp. 171-176, Oct. 1966.
118. R. E. Stengel and S. A. Olson,
“Method and apparatus for efficient
signal power amplification,” U.S.
Patent 5,892,395, Apr. 6, 1999.
119. R. Langridge, T. Thornton, P.
M. Asbeck, and L. E. Larson, “A power
re-use technique for improving efficiency of outphasing microwave
power amplifiers,” IEEE Trans.
Microwave Theory Tech., vol. 47, no. 8,
pp. 1467-1470 , Aug. 1999.
120. S. Djukic, D. Maksimovic, and
Z. Popovic, “A planar 4.5-GHz dc-dc
power converter,” IEEE Trans.
Microwave Theory Tech., vol. 47, no. 8,
pp. 1457-1460, Aug. 1999.
121. A. Shirvani, D. K. Su, and B.
A. Wooley, “A CMOS RF power amplifier with parallel amplification for
efficient power control,” IEEE J.
Solid- State Circuits, vol. 37, no. 6,
pp. 684-693, June 2002.
122. C. Y. Hang, Y. Wang, and T.
Itoh, “A new amplifier power combin54
High Frequency Electronics
ing scheme with optimum efficiency
under variable outputs,” 2002 Int.
Microwave Symp. Digest, vol. 2, pp.
913-916, Seattle, WA, June 2-7, 2002.
123. F. H. Raab, “Electronically
tunable class-E power amplifier,” Int.
Microwave Symp. Digest, Phoenix,
AZ, pp. 1513-1516, May 20-25, 2001.
124. F. H. Raab, “Electronically
tuned power amplifier,” Patent pending.
125. F. H. Raab and D. Ruppe,
“Frequency-agile class-D power
amplifier,” Ninth Int. Conf. on HF
Radio Systems and Techniques, pp.
81-85, University of Bath, UK, June
23-26, 2003.
126. F. H. Raab, “High-efficiency
linear amplification by dynamic load
modulation,” Int. Microwave Symp.
Digest, vol. 3, pp. 1717-1720,
Philadelphia, PA, June 8-13, 2003.
In Part 4 of this series (Novermber
2003 issue), Figures 45, 52, 55, 56, 57
and 58 should have been credited as
“Courtesy Andrew Corporation”
instead of “Courtesy WSI.”
Author Information
The authors of this series of articles are: Frederick H. Raab (lead
author), Green Mountain Radio
Research, e-mail:;
Peter Asbeck, University of California
at San Diego; Steve Cripps, Hywave
Associates; Peter B. Kenington,
Andrew Corporation; Zoya B. Popovic,
Colorado; Nick
Pothecary, Consultant; John F. Sevic,
California Eastern Laboratories; and
Nathan O. Sokal, Design Automation.
Readers desiring more information
should contact the lead author.
Acronyms Used in Part 5
Barium Strontium
Ground Wave
Emergency Network
Pulse-Width Modulation
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