Engineering Specification
Engineering Specification
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EMC Design Guide for Printed Circuit Boards
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Released: ES-3U5T-1B257-AA
Initial Release
AE00-I-11448433-000
20021001
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REFERENCE
PREPARED/APPROVED BY
J. A. Tracz, 32-31919
CHECKED BY
DETAILED BY
CONCURRENCE/APPROVAL
SIGNATURES
Design Engineering Supervisor
T. Hermann
Design Engineering Management
C. De Biasi
Manufacturing Engrg.
Quality Control
Purchasing
Supplier Quality Assistance
FRAME 1
PD
May 1988
OF
3947a1e
78
A
Initial Release Date: OCTOBER 01 2002
This document embodies information originated and owned by Ford Motor Company. Reprint by permission.
Printed copies are uncontrolled
Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
TABLE OF CONTENTS
PART I: PREFACE ................................................................................................................ 8
1.
INTRODUCTION .................................................................................................................. 8
PART II: GENERAL EMC ................................................................................................ 9
2. EMC OVERVIEW.................................................................................................................. 9
2.1. The Elements .................................................................................................................. 9
2.2. The Environment .......................................................................................................... 10
2.3.
Regulations and Standards............................................................................................ 11
2.4.
Elements of EMI ........................................................................................................... 12
PART III: DESIGN APPROACH .................................................................................... 14
3. OVERVIEW ......................................................................................................................... 14
3.1. Design Approach for Immunity (Susceptibility) .......................................................... 14
3.1.1.
Design Approach for Radiated Immunity ............................................................. 14
3.1.2.
Design Approach for ESD..................................................................................... 14
3.2. Design Approach for Controlling Radiated and Conducted Emissions........................ 14
3.3. Ground System.............................................................................................................. 15
3.4. Wavelength and Frequency........................................................................................... 19
3.5.
Frequency Domain of Digital Signals........................................................................... 22
3.6.
Radiated Emissions Predictions.................................................................................... 24
3.7.
Crosstalk ....................................................................................................................... 28
3.7.1.
Common Impedance Coupling.............................................................................. 29
3.7.2.
Capacitive and inductive coupling ........................................................................ 30
3.7.3.
Capacitive coupling ............................................................................................... 30
3.7.4.
Inductive coupling ................................................................................................. 33
3.8.
Twisted Pair .................................................................................................................. 36
3.9.
Shielding ....................................................................................................................... 37
3.10. Resistance..................................................................................................................... 40
3.11. Inductance .................................................................................................................... 41
PART IV: IC RE MEASUREMENT PROCEDURE .................................................. 46
4. SCOPE .................................................................................................................................. 46
4.1. Applicable Documents.................................................................................................. 46
4.2.
EMC Test Recommendations ....................................................................................... 47
4.3. Test Procedure Applicability ........................................................................................ 47
4.4. IC Emissions Reference Levels .................................................................................... 48
4.4.1.
Level 1 ................................................................................................................... 49
4.4.2.
Level 2 ................................................................................................................... 49
4.4.3.
Level 3 ................................................................................................................... 49
4.4.4.
Level 4 ................................................................................................................... 50
4.4.5.
Level NR................................................................................................................ 50
4.5.
Data Submission ........................................................................................................... 50
4.6.
Radiated and Conducted Immunity .............................................................................. 50
PART V: EMC DESIGN GUIDELINES FOR PCB .................................................. 51
5. GENERAL............................................................................................................................ 51
5.1. Board Structure/Ground Systems ................................................................................. 52
5.2. Power Systems .............................................................................................................. 57
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5.3.
5.4.
5.5.
5.6.
5.7.
Digital Circuits.............................................................................................................. 61
Analog Circuits ............................................................................................................. 64
Communication Protocols............................................................................................. 65
Shielding ....................................................................................................................... 65
Miscellaneous ............................................................................................................... 67
PART VI: REQUIREMENTS ........................................................................................... 69
6. MANAGEMENT OF CHANGE FOR EMC ....................................................................... 69
6.1.
Radiated Immunity: ...................................................................................................... 69
6.1.1.
For safety critical systems (containing one or more Class C functions) ............... 69
6.1.2
For non-safety critical systems .............................................................................. 70
6.2.
Conducted immunity:.................................................................................................... 70
6.3. Electrostatic Discharge ................................................................................................. 70
6.4.
Conducted Emissions:................................................................................................... 71
6.4.1
CE420 Frequency domain ..................................................................................... 71
6.4.2
CE410 Time Domain............................................................................................. 71
PART VII: CHECKOFF LIST ........................................................................................ 72
7. CHECKOFF LIST – EMC DESIGN GUIDE FOR PCB(S) ................................................ 72
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TABLE OF FIGURES
Figure 2–1. Elements of EMI....................................................................................................... 12
Figure 3–1. Ground Grid.............................................................................................................. 15
Figure 3–2. Inductance of Grounds ............................................................................................. 16
Figure 3–3. Single-Point Ground................................................................................................. 18
Figure 3–4. Multi-Point Ground .................................................................................................. 18
Figure 3–5. Hybrid Ground.......................................................................................................... 18
Figure 3–6. Wavelength of an Electrical Signal .......................................................................... 19
Figure 3–7. Elements of Digital Signal........................................................................................ 22
Figure 3–8. Digital Signal Spectrum ........................................................................................... 22
Figure 3–9. Setup for Measuring CM Currents ........................................................................... 27
Figure 3–10. Elements of Common Impedance .......................................................................... 29
Figure 3–11. Inductive and Capacitive Coupling Between Two Circuits ................................... 30
Figure 3–12. Capacitive Coupling ............................................................................................... 31
Figure 3–13. Inductive Coupling ................................................................................................. 34
Figure 3–14. Mutual Inductance Between Two Wires ................................................................ 35
Figure 3–15. Magnetic Field Coupling into Circuit..................................................................... 36
Figure 3–16. Magnetic Field Coupling into Twisted Wire Pair .................................................. 36
Figure 3–17. Effectiveness of Shielding...................................................................................... 37
Figure 3–18. Inductance in Parallel Wires................................................................................... 42
Figure 3–19. Inductance in Wires over Ground Plane................................................................. 43
Figure 3–20. Inductance of Ground Plane vs. Wire Inductance .................................................. 44
Figure 4–1. IC Radiated Emissions Acceptance Levels .............................................................. 48
Figure 5–1. Relative Costs of EMC vs. NO EMC Design Strategy ............................................ 51
Figure 5–2. Arrangement of Functional Groups on PCB ............................................................ 52
Figure 5–3. Maximizing Ground on PCB.................................................................................... 52
Figure 5–4. Ground Grid Technique............................................................................................ 53
Figure 5–5. Creating 'Faraday's Cage' ......................................................................................... 53
Figure 5–6. Layer Stack-up.......................................................................................................... 54
Figure 5–7. IC Ground................................................................................................................. 54
Figure 5–8. Eliminating Floating Ground.................................................................................... 55
Figure 5–9. Establishing Ground Plane Boundary ...................................................................... 56
Figure 5–10. Power System's Star Point ...................................................................................... 57
Figure 5–11. Power and Ground Routing .................................................................................... 58
Figure 5–12. Primary Loop Area ................................................................................................. 59
Figure 5–13. Secondary Loop Area ............................................................................................. 60
Figure 5–14. Minimizing Digital Bus Length.............................................................................. 61
Figure 5–15. Resistance and Inductance as Functions of Frequency .......................................... 61
Figure 5–16. Crystal/Oscillator placement .................................................................................. 62
Figure 5–17. Transistor Circuit Routing...................................................................................... 64
Figure 5–18. Shielding of Low-Frequency Signals ..................................................................... 66
Figure 5–19. Shielding of High-Frequency Signals .................................................................... 66
Figure 5–20. Packaging Considerations Affecting RE and CE ................................................... 67
Figure 5–21. Use of Interspersed Grounds .................................................................................. 68
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TABLE OF TABLES
Table 2–1. FCC and Ford RE Limits ........................................................................................... 12
Table 3–1. Frequency and Impedance ......................................................................................... 17
Table 3–2. Wavelength as Function of Frequency ...................................................................... 20
Table 3–3. Frequency Allocation and Usage Designation........................................................... 21
Table 3–4. Sample RE Data......................................................................................................... 26
Table 3–5. Ford RE Limit vs. Sample Data................................................................................. 28
Table 3–6. Mutual Capacitance in Two Wires ............................................................................ 32
Table 3–7. Relative Permeability of Common Metals................................................................. 39
Table 3–8. Resistance in Wires.................................................................................................... 40
Table 3–9. Resistance in Grounding Straps ................................................................................. 41
Table 3–10. Inductive Reactance vs. Frequency ......................................................................... 42
Table 3–11. Impedance in Solid Copper Wires ........................................................................... 43
Table 3–12. Self-Inductance in Wires ......................................................................................... 45
Table 4–1. Rating Levels for IC's ................................................................................................ 49
Table 6–1. Analysis of EMC Testing .......................................................................................... 71
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TABLE OF EQUATIONS
Equation 3–1. Wavelength........................................................................................................... 19
Equation 3–2. Duty Cycle............................................................................................................ 23
Equation 3–3. Bandwidth............................................................................................................. 23
Equation 3–4. Current in Square Waves...................................................................................... 24
Equation 3–5. Far-Field Radiated Emissions............................................................................... 24
Equation 3–6. Radiated Emissions from a loop........................................................................... 25
Equation 3–7. Far Field strength.................................................................................................. 25
Equation 3–8. Common Mode current......................................................................................... 27
Equation 3–9. E-field strength due to CM current....................................................................... 27
Equation 3–10. CM current.......................................................................................................... 28
Equation 3–11. Mutual Capacitance in wires .............................................................................. 31
Equation 3–12. Mutual Capacitance ............................................................................................ 32
Equation 3–13. Voltage Noise due to capacitive coupling .......................................................... 33
Equation 3–14. Mutual Inductance .............................................................................................. 34
Equation 3–15. Noise voltage due to inductive coupling ............................................................ 35
Equation 3–16. Noise voltage due to inductive coupling ............................................................ 35
Equation 3–17. Inductive Coupling in twisted-wire pair............................................................. 36
Equation 3–18. Shielding effectiveness ....................................................................................... 38
Equation 3–19. Absorption loss................................................................................................... 38
Equation 3–20. Resistance in Copper .......................................................................................... 40
Equation 3–21. Inductive Reactance............................................................................................ 41
Equation 3–22. Inductance in rectangular conductor .................................................................. 41
Equation 3–23. Inductance in parallel wires................................................................................ 42
Equation 3–24. Self-Inductance................................................................................................... 44
Equation 3–25. Inductance in air-core inductors ......................................................................... 45
Equation 3–26. Inductance in toroids .......................................................................................... 45
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ACRONYMS AND ABREVIATIONS
AC
AWG
BCI
BW
CE
CI
CM
CMOS
CPU
CS
dB
DC
DM
EC
E/E
EMC
EME
EMI
EPDS
ESC
ESD
EU
FCC
FET
FMC
HSIC
IC
IEC
I/O
ISO
LSI
MCU
MOV
NB
PCB
PWB
PWM
RE
RF
RI
RS
SAE
TEM
Alternating Current
American Wire Gauge
Bulk Cable Injection
Bandwidth
Conducted Emissions
Conducted Immunity
Common Mode
Complementary Metal-Oxide Semiconductor
Central Processing Unit
Conducted Susceptibility
decibel
Direct Current
Differential Mode
European Community
Electrical/Electronic
Electromagnetic Compatibility
Electromagnetic Effect
Electromagnetic Interference
Electrical Power Distribution System
Electronic Subsystem Component (Device Under Test)
Electro Static Discharge
European Union
Federal Communication Commission
Field Effect Transistor
Ford Motor Company
High Speed Integrated Circuit
Integrated Circuit
International Electrotechnical Commission
Input/Output
International Standards Organization
Large State Integration
Micro-Controller Unit
Metal-Oxide Varister
Narrowband
Printed Circuit Board
Printed Wiring Board
Pulse Width Modulation
Radiated Emissions
Radio Frequency
Radiated Immunity
Radiated Susceptibility
Society of Automotive Engineers
Transverse Electromagnetic (Cell)
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PART I: PREFACE
1. INTRODUCTION
Due to the tremendous increase in the use of electronic devices, ensuring Electromagnetic
Compatibility (EMC) of a full system in its early design phase is becoming one of the major
technical issues, especially for automotive manufacturers. Safe and reliable operation must
be guaranteed and legal requirements have to be satisfied. From both car-makers and
suppliers sides, the electromagnetic problems occur either when integrating electronic
devices in their operating environment (cross-coupling, interference) or when dealing with
the related EMC regulations (simulation of radiating phenomena due to Common-Mode
currents induced on attached cables). As digital devices become smaller and perform at
greater speeds, their emissions increase, making a thorough understanding of
Electromagnetic Interference (EMI) essential for everyone in electrical engineering and
design today.
This document contains design guidelines to aid in achieving EMC (Electromagnetic
Compatibility) in automotive electrical/electronic components and systems. None of the
material presented herein is new. On the contrary, it is based on well-established EMC
measures and techniques, and on specific automotive EMC experience accumulated over the
years within Ford Motor Company. The "EMC design guide for PCB" simply attempts to
collect that wisdom together.
It should be pointed out that Parts 1 through 6 of this document are meant to be strictly
informative. For example, the various design techniques presented in Section 5 are derived
from a set of fundamental principles, and although the techniques aid each other in achieving
electromagnetic compatibility, they don't guarantee it. Suppliers are ultimately responsible
for assuring full Ford EMC compliance of their products.
Completion of Part 7 is mandatory.
The reader is encouraged to forward any comments, questions or suggestions regarding this
document to the following e-mail address:
mailto:[email protected]
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PART II: GENERAL EMC
2. EMC OVERVIEW
The application of electronic components and devices is increasing in all area of consumer
products as well as within the industrial production environment. This provides an
electromagnetic environment with an increasing overall noise floor due to digital control
applications in virtually any niche of daily life and an ever increasing demand on mobile
telecommunications facilities.
The noise margin – observations in the early 90's have revealed an increase of approximately
3 dB per year – poses an increasing threat onto the immunity margins of the electronic
components. In contrast to the aggression, the immunity margin is falling due to the drastic
increase in the complexity of the components, calling for a reduction in power consumption
in order to control thermal effects for instance. The attempt of controlling costs also leads
towards a trend of replacing solid metal housings with plastics or composites, which decrease
shielding capabilities.
In summary, the trend in Electronics' applications, a raise of a harsh electromagnetic ambient
has to be noticed with a loss of safety margins, making applications more susceptible to
electromagnetic interference and calling for regulations to keep the problems arising under
control.
2.1. The Elements
Electromagnetic radiation due to the operation of electrical or electronic devices may be
grouped into two types:
•
•
Intentional Emissions
Unintentional Emissions
Examples of the first type are television and radio broadcasting systems, communication
and radar systems, and transmitters for navigational purposes. However, even when
performing properly such equipment may also generate undesired electromagnetic
emissions of the second type. This might interfere with the system itself or the overall
emissions might affect other sensitive equipment nearby. In order to control these kind of
effects frequency management is necessary in the first place due to the fact that a certain
part of the emission profile contains valuable information and is intended to be there.
Electronic components provide a frequency band and due to non-linearities in active
devices unintentional harmonics may be created, and modulations might occur. In
general, sources of coherent electromagnetic emission at a given frequency or within a
specified frequency band are intentional transmitters, but both coherent as well as noncoherent emission bear the potential for electromagnetic interference problems.
Electromagnetic emissions may thus be divided into:
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•
•
Radiation due to radio transmitters and similar nearby electrical or electronic
equipment
Transient environment caused by electrical switching operations, electrostatic
discharge and lightning
2.2. The Environment
There are two fundamental classes of transfer types:
•
•
Analogue
Digital
The difference is not only due to the information coding but with regard to EMC the main
difference is due to the quality and vulnerability.
Analogue circuitry reacts immediately on perturbations but the effects remain within
relatively small limits, they might cause a rectification and possibly a drifting of the
operation point. Typically, analogue circuitry recovers from the perturbation by turning
back towards the regular operation. The operational safety margin corresponds to the
signal to noise ratio.
In contrast to the above, digital circuits provide a 'large' safety margin because of the
switching thresholds for the different states. Hence a digital application appears more
robust than an analogue one. However, the move towards low voltage logic, 3V and even
less, will reduce these margins. Another difference lies in the quality of failure which
might be quite unpredictable for digital application – a bit might switch and cause a
system to malfunction in the case of switching to a defined state or to hang because of
turning into an undefined state.
Most problems associated with digital circuits are due to the high bandwidth inherited
from the high-speed clocks and edge rates. Rise times in the realm of a few nanoseconds
are equivalent with bandwidths well above 300 MHz range and the increase of the clock
rates will drive this into the microwave range. In other words, higher bandwidths increase
both emissions and the susceptibility of the circuitry.
This issue is fundamental to the functioning of the designed circuitry and comprises
mainly the aspects of internal or intra-system EMC and Signal Integrity. Intra-system
EMI includes problems due to mixed technologies, e.g. analogue and digital, or
electromechanical and digital. In the former case, the noise created by the digital circuitry
due to the impulsive nature of the power demands might cause some jamming of the
analogue circuits. In the latter case, the noise due to motors and switching relays typically
causes jamming of the digital circuits. In the case of high speed digital application the
digital circuitry might also cause some malfunctions due to crosstalk between such high
speed applications and reflections on the interconnects. A particular characteristic of
analogue components is that they typically operate at low frequencies and low levels and
in addition show very high input impedances.
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2.3. Regulations and Standards
E/E devices on Ford products must comply with a variety of requirements mandated by:
•
•
•
Federal Communications Commission (FCC) regulations
Ford EMC specifications
European Community (EC) EMC Directive
Within the United States, the FCC is responsible for radio spectrum allocations and
assignments outside the federal government sector. FCC 15J is the FCC document that
controls the interference potential of electronic computing devices. A computing device
is defined as any electronic device or system that uses digital techniques. This
encompasses any device that intentionally generates and utilizes frequencies in excess of
10 kHz. The FCC regulates transmitters and receivers under a different rule.
Vehicle radios and remote controlled transceivers must comply with FCC regulations.
Most other E/E modules designed exclusively for a vehicle use, have Section 1
exemption from FCC regulations. However, Ford has a policy of voluntary compliance
for all modules.
The FCC regulates the amount of radiated EMI from an E/E device. Table 2–1 compares
the maximum radiated electric fields allowed by Ford and the FCC. The table shows the
electric field strength in dBµV/m and µV/m. FCC Class B limits are for consumer-type
computing devices.
Ford EMC requirement limits are derived from a variety of SAE, International Standards
Organization (ISO), and EC standards. The latest version of the primary Ford component
level EMC specification is available on line at www.fordemc.com.
By inspection of Table 2–1, it is evident that the Ford limits are much more stringent than
those of FCC; especially when one considers that FCC limits are measured 3 meters away
from the radiating device while Ford measurements are taken at 1 meter. The Ford limits
are more restrictive for two primary reasons:
•
•
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Radiating devices on the vehicle are closer to radio transceivers on board the
vehicle
A vehicle contains many radiating devices
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Table 2–1. FCC and Ford RE Limits
Ford RE limits (ES-XW7T-1A278-AB)
(1 meter)
Frequency
µV/m
dBµV/m
(MHz)
0.15 – 25
31.6
30
25 – 200
3.2
10
200 – 1000
10 – 24
30 – 88
3.2 – 15.8
FCC Class B RE limits
(3 meters)
100
88 – 216
150
43.5
216 – 1000
200
46
40
Outside the United States, the European Community (EC) has recently passed a uniform
EMC directive. Any electronic product and vehicle manufactured in the United States or
elsewhere must fulfill the EC regulations before they can be marketed in Europe. The
main EC EMC Directive for automotive is 95/54/EC (and UN-ECE R10.02).
2.4. Elements of EMI
Figure 2–1 shows the elements of an EMI situation. A generator is an E/E device that
produces EMI. A receptor is an E/E device that receives or couples in EMI. A coupling
path allows EMI from the generator to produce an undesired response in the receptor.
Generator
Receptor
Coupling Path
Figure 2–1. Elements of EMI
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There are 3 ways in which to minimize the EMI:
•
•
•
Reduce the noise from the generator
Alter the coupling path
Improve the immunity of the receptor
Reducing the noise from the generator may include reducing the generator circuit area,
using a slower rise and fall time of a switching signal, using slower digital logic, reducing
the circuit current, re-orienting the generator circuit on PCB, filtering, and/or shielding.
Often the designer must contend with the generator noise. This leaves either reducing the
coupling path or hardening the receptor. Minimizing the coupling path may include
moving the receptor away from the generator, re-orienting the receptor and/or generator
on PCB, or shielding the receptor or generator.
Increasing the immunity of the receptor may include reducing the receptor circuit area,
re-orienting the receptor circuit on PCB, using less susceptible electronic components,
changing the impedance of the circuit, filtering, and/or shielding.
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PART III: DESIGN APPROACH
3. OVERVIEW
Each unit of the overall system must be designed to meet the EMC Specified Limits. These
EMC requirements can be divided into two primary categories: immunity (susceptibility)
requirements and emission requirements.
3.1. Design Approach for Immunity (Susceptibility)
The different susceptibility requirements differ in threat energy and frequency content,
although there is considerable overlap of these parameters among the test requirements.
Of these requirements, the two susceptibility requirements that impact the design the
most are the Radiated Immunity (RI) requirements, and the effects of Electrostatic
Discharge (ESD) requirements.
3.1.1. Design Approach for Radiated Immunity
The Radiated Immunity threat is characterized by moderately high energy and very
high frequency that can propagate in unexpected ways to circuit components,
causing unexpected effects. The frequency range of radiated immunity can be found
in the latest version of the component level spec at www.fordemc.com. Historic
component level RI data shows that the band of frequencies most likely to cause
radiated immunity-related problems for electronic controllers in an automotive
environment is 10 MHz to ~ 900 MHz. The overall design approach for Radiated
Immunity is to keep RI interference contamination out of the ESC (Electronic
Subsystem Component) by removing it from signal and power lines entering the
ESC (at ESC entry point).
3.1.2. Design Approach for ESD
The ESD threat is primarily characterized by short duration, high energy pulses that
can damage I/O components or cause circuit upset. The frequency range of ESD is
from DC to ~300 MHz (DC - 10MHz for lightning). The design approach for
Radiated Immunity will also help prevent circuit upset caused by the high frequency
components of ESD. However, ESD protection requires additional measures to
prevent damage from the low frequency, high energy content of the induced ESD
threats. Therefore, the overall design approach for ESD protection is to maintain
high impedance, with respect to chassis ground, on signal and power lines to the
ESC to minimize the effect of damaging ESD current and the corresponding energy
from entering the ESC.
3.2. Design Approach for Controlling Radiated and Conducted Emissions
The purpose of emission requirements is to provide a level of assurance that the
equipment will not produce electromagnetic emissions great enough to adversely affect
the performance of other equipment in the vehicle (primarily communication, audio and
electronic vehicle control systems). Fortunately, many of the design techniques used to
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prevent Radiated Immunity and ESD interference energy from entering the ESC also
prevent internally generated emissions from leaving the ESC. However, there are several
additional design techniques used to minimize electromagnetic emissions from the ESC.
The overall design approach for reducing emissions from the ESC is to prevent or
minimize the generation of high frequency interference voltage and currents as close to
the interference source as possible. The objective is to stop emissions at their source,
before they can cross couple to other circuits and signal lines that cannot be easily
filtered.
3.3. Ground System
A low inductance ground system is the most important element in designing a PCB for
EMC. There are three types of ground structures on PCB:
•
•
•
Minimal
Ground Grid
Ground Plane
The minimal ground structure connects the ground points on PCB with random, high
inductance connections to other ground points. This is the least desirable method from an
EMC standpoint.
Maximizing the ground area on a PCB minimizes the inductance of the ground system,
which in turn minimizes radiated emissions. In addition, the maximized ground area
provides shielding to improve radiated immunity of the PCB.
Figure 3–1 shows the ground grid structure. A ground grid provides many lowerinductance paths for current to return to its source.
= via
Figure 3–1. Ground Grid
Ground grid is achieved by connecting vertical and horizontal lines on opposite sides of
PCB with vias. A via is a plated through hole that interconnects two or more PCB layers.
Via connections also allow a signal track to 'jump over' a ground grid trace. All two layer
PCBs should use a ground grid. In addition, multi-layer PCBs should use ground grids
even if they employ one or more ground planes.
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A ground plane is the lowest impedance conductor that serves as a current return and a
signal reference. A ground plane is the ideal ground system. It offers the lowest possible
inductance for current to return to its source. A properly designed ground grid is the next
best ground system.
Figure 3–2 compares the inductance of a ground grid and ground plane. The graph
displays the inductance in nanohenries (nH) versus grid spacing in millimeters. To
effectively lower the inductance of a ground grid the grid spacing must be less than 0.5
inches (~13 mm). Figure 3–2 shows that when the grid spacing equals 0.5 inches, the
ground grid inductance has significantly decreased. Reducing the grid spacing further
lowers the inductance.
70
Inductance
(nH)
60
50
40
30
25
Ground grid
Ground
2
3
13
6
Grid spacing in millimeters
Figure 3–2. Inductance of Grounds
The inductance of a ground grid approaches but can never equal the inductance of a
ground plane. They have equal inductances only when the grid spacing equals zero (when
the ground grid becomes a ground plane).
Table 3–1 shows the impedance versus frequency for PCB tracks and ground planes. The
ground plane is square and made of one-ounce copper (0.035 mm thick). The track is also
1 oz. copper with width of 1.0 mm and length 10.0 mm. These impedance calculations do
not take into account the mutual inductance between a signal track and ground plane
which would decrease the total impedance.
Table 3–1 indicates that the track has significantly more impedance than the ground plane
due to a larger inductance. Ground inductance generates ground noise. This is the reason
why ground planes generate less noise than a minimal ground system.
When high frequencies flow through the larger inductances of a minimal ground system
they generate voltage drops within that system. Often this is the source of radiated
emissions from a PCB. Ground noise voltages force EMI currents to flow out onto the
wire harness that connects to the PCB. Moreover, the wires harness is usually much
longer than the PCB, and thus it radiates more efficiently.
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Table 3–1. Frequency and Impedance
100 Hz
Ground plane
impedance
574 µΩ
Track
impedance
5.74 mΩ
1 kHz
574 µΩ
5.74 mΩ
10kHz
574 µΩ
5.76 mΩ
20 kHz
574 µΩ
5.81 mΩ
50 kHz
574 µΩ
6.14 mΩ
100 kHz
574 µΩ
7.21 mΩ
500 kHz
576 µΩ
22.5 mΩ
1 MHz
582 µΩ
44 mΩ
2 MHz
604 µΩ
87.5 mΩ
5 MHz
736 µΩ
218 mΩ
10 MHz
1.04 mΩ
437 mΩ
20 MHz
1.61 mΩ
874 mΩ
50 MHz
2.62 mΩ
2.18 Ω
100 MHz
3.69 mΩ
4.37 Ω
200 MHz
5.22 mΩ
8.74 Ω
500 MHz
8.26 mΩ
21.8 Ω
1 GHz
11.6 mΩ
43.7 Ω
Frequency
Signal grounds usually fall into one of the three categories:
•
•
•
Single-point ground
Multi-point ground
Hybrid ground
Single point grounds, (Figure 3–3), with regards to noise, are very undesirable because of
the series connection of all the individual circuit grounds. At high-frequencies the
inductances of the ground conductors increase the ground impedance. A single-point
ground is preferable below 1 MHz. Between 1 and 10 MHz a single point-ground can
usually be used, provided the length of the longest ground conductor is less than onetwentieth of a wave-length to prevent emissions and to maintain a low impedance.
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Figure 3–3. Single-Point Ground
Multi-point grounds, (Figure 3–4), have very low ground impedance and should be used
at high frequencies and in digital circuitry. The low impedance is due primarily to the
lower inductance of the ground plane. The connection between each circuit and the
ground plane should be kept as short as possible to minimize their impedance. Multipoint grounds should be avoided at low frequencies since ground currents from all
circuits flow through a common ground impedance – the ground plane.
Figure 3–4. Multi-Point Ground
A hybrid ground, (Figure 3–5), is one in which the system grounding configuration
appears differently at different frequencies – a single-point ground at low frequencies,
and a multi-point ground at high frequencies. When different types of circuits (low-level
analog, digital, noisy, etc.) are used in the same system or on the same PCB, then each
must be grounded in a manner appropriate for that type of circuit. The different ground
circuits should be tied together, usually at a single point.
Figure 3–5. Hybrid Ground
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3.4. Wavelength and Frequency
All electrical signals travel as waves with a finite velocity. Figure 3–6 shows the
amplitude plot of a wave as a function of time. Its wavelength is the distance between
any two equivalent points on the waveform.
Amplitude
Wavelength
Time
Figure 3–6. Wavelength of an Electrical Signal
The propagation medium determines the wave's velocity. In space a wave travels at the
speed of light c (c = 3x108 meters/second). However, the wave travels more slowly
through wires or printed circuit board tracks (approx. 0.6 the speed of light).
Equation 3–1 relates wavelength (λ) to frequency in free space or in air. Table 3–2 shows
that the wavelength and frequency are inversely proportional. Consequently, as the
frequency increases, the wavelength will decrease.
λ=
300
[meters]
f (MHz)
Equation 3–1. Wavelength
The atmosphere contains many electromagnetic fields generated by transmitters, which
operate in a range from a few kilohertz to many gigahertz. Table 3–3 shows some of the
frequency spectrum allocations in the United States (US), Europe (E), and Japan (J). All
undesignated sources are in the United States.
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Table 3–2. Wavelength as Function of Frequency
Frequency
λ
10 Hz
30,000 km
60 Hz
5,000 km
100 Hz
3,000 km
1 kHz
300 km
50 kHz
6 km
100 kHz
3 km
500 kHz
600 m
1 MHz
300 m
10 MHz
30 m
100 MHz
3m
1 GHz
30 cm
10 GHz
3 cm
It is crucial that the E/E devices installed in a vehicle are immune to the fields produced
by transmitters such as those listed in Table 3–3. In addition, the vehicle's E/E devices
must not generate emissions that interfere with the intended receivers of these
transmitters.
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Table 3–3. Frequency Allocation and Usage Designation
AM (E)
Frequency
(MHz)
0.15 – 0.285
Typical radiated power
(kW)
320
AM (E & J)
0.525 – 1.605
600 & 500
AM (US)
0.53 – 1.71
50
Amateur
1.8 – 29.7
0.16 (mobile)
Citizens band
26.9 – 27.4
0.004
Amateur
28 – 30
0.2 (mobile)
Land mobile
29 – 54
0.1
Amateur
50 – 54
0.2 (mobile)
TV low VHF
54 – 88
100
Land mobile (E)
65 – 85
0.1
FM (J)
76 – 90
44
FM (US & E)
88 – 108
105
Aircraft
108 – 136
1
Land mobile (E)
120 – 160
0.1
Land mobile
132 – 174
18 – 100
Land mobile (J)
142 – 170
Amateur
144 – 148
0.2 (mobile)
TV high VHF
174 – 216
316
Land mobile
216 – 222
0.2
Amateur
222 – 225
0.1 (mobile)
Land mobile (J)
335 – 384
Land mobile
406 – 512
Land mobile (J)
450 – 470
Amateur
430 – 450
0.1 (mobile)
TV UHF
470 – 806
5000
Land mobile
806 – 947
0.035
Cellular (AMPS)
806 – 947
0.003
Amateur, LM, GPS
1200 – 1600
Cellular (PCS)
1700 – 2000
Bluetooth
2300 - 2500
Source
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3.5. Frequency Domain of Digital Signals
A typical square wave is shown in Figure 3–7.
AC
DC
AC
A
t
A/2
tf
tr
T
Figure 3–7. Elements of Digital Signal
A square wave has an AC component during the transition times and a DC component
during the steady state. The AC current contains all of the frequency components of the
square wave. In addition to the fundamental frequency, a digital signal also contains
harmonic frequencies which are integer multiples of the fundamental frequency. For
example, a digital signal with a fundamental frequency of 10 MHz has harmonic
frequency components at 20, 30, 40, … MHz. Therefore, digital signal current flows at
DC and at 10, 20, 30, 40, … MHz.
The signal spectrum is a plot of the fundamental and harmonic frequencies. Figure 3–8
shows the signal spectrum for a square wave in amplitude versus frequency graph.
Amplitude
Fundamental frequency
2 A × DC
Spectral envelop
-20 dB/decade
-40 dB/decade
Frequency
Bandwidth
f1 =
1
π ×t
f2 =
1
π × tr
Figure 3–8. Digital Signal Spectrum
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The spectrum envelope is a mathematical amplitude limitation of the spectral
components of a digital signal. The maximum amplitude of the spectrum envelope equals
2AxDC, where A is the peak amplitude of the square wave, and DC is a duty cycle (often
denoted as δ), then:
δ =
t
T
Equation 3–2. Duty Cycle
Where, t is the time that a square wave stays above one-half the maximum amplitude.
The spectrum envelope falls off at 20 dB per decade at frequencies above f1 = 1/πt.
The signal rise time (tr) is the time that a digital signal takes to rise from 10% to 90% of
its value (refer to Figure 3–7). Equation 3–1 states that the rise time determines the
bandwidth (BW) of the signal. Use the signal fall time (tf) if it is faster than the rise
time, which it usually is. The spectrum envelope falls of at 40 dB per decade at
frequencies above f2 = 1/πtr.
The magnitude (M) in dB at > f is computed as follows:
M ( f1 < f < f 2 )dB = M 1 = 20 log( 2 Aδ ) − 20 log(
M ( f > f 2 )dB = M dB = ( f = f 2 ) − 40 log(
BW =
f
)
f1
f
)
f2
1
π × tr
Equation 3–3. Bandwidth
In Figure 3–8, the bandwidth contains 99% of the spectral energy of the signal.
The spectrum of the square wave in Figure 3–8 is also its Fourier series. Fourier theory
states that a periodic signal can be expressed in terms of weighted sum of harmonically
related sinusoids.
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Equation 3–4 gives the amplitude for the fundamental and harmonic currents in the
Fourier series of a square wave.
t 

sin nπ r 
sin( nπδ )
T
I n = 2 I maxδ
× 
nπδ
t 
nπ  r 
T 
[Amperes]
Equation 3–4. Current in Square Waves
Where,
1 ≤ n ≤ ∞, Imax is the maximum current.
It is assumed that the rise and the fall times of a square wave are equal. A square wave
with δ = 0.5 has only odd numbered harmonic with the first current harmonic, I1 =
0.641Imax
3.6. Radiated Emissions Predictions
For intentional transmitters (e.g. broadcast towers), the electromagnetic field next to a
transmitting antenna is very complex. This field is called the near field. However, the
field becomes an uniform plane wave some distance from the antenna. This field is called
the far field. The near to far field transition (equation below) occurs at about one-sixth of
a wavelength from the transmitting antenna.
Near to far field transition: ≈
λ
2π
[meters]
This next equation, Equation 3–5, shows how to calculate the far-field radiated emissions
from any RF transmitter:
E=
30 × Pt
r
[Volts/meter]
Equation 3–5. Far-Field Radiated Emissions
For example, the far field for a FM transmitter at 100 MHz occurs at about one-half
meter. The transmitter electric field strength 100 meters (r) away from the transmitter
(Pt=250 kW) equals 27.4 volts per meter.
For unintentional noise sources, E/E designers should consider circuit loops.
Equation 3–6 gives the maximum radiated emission in dBµV per meter from a small
loop. Differential mode (DM) current, In, is the normal signal or power current that
flows in a loop.
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
A × f n2 × I n
E DM = 20 log 10  2.63 × 10 − 8
r





[dBµV/m]
Equation 3–6. Radiated Emissions from a loop
Where,
A is the area of a small loop
fn is the spectral signal frequency
In is the spectral signal current
r is the distance from the small loop to the measurement antenna or the
distance between a radiating generator loop and a receptor circuit.
Equation 3–7 predicts the maximum electric field in the far field from a small loop. It is
accurate when the loop perimeter is less than one-quarter wavelength, and approximate
for larger loops. In the near field multiply Equation 3–6 by Equation 3–7.
2
r
 λ 
E far _ field = 1 + 

 2π × r 
Equation 3–7. Far Field strength
Table 3–4 shows the radiated emissions at 1 meter from a PCB circuit with the following
values:
Area = 5.0x10-4 meter2 (5 cmx1 cm)
Fundamental frequency = 10 MHz
Imax = 10 mA
Rise time = fall time = 5 ns (typical high-speed CMOS)
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Table 3–4. Sample RE Data
Frequency
(MHz)
10
6.34 mA
Electric field
(µV/m)
40.7
Electric field
(dBµV/m)
32.2
30
2.04 mA
45.5
33.2
50
1.15 mA
52.1
34.3
70
0.737 mA
47.5
33.5
90
0.494 mA
52.6
34.4
110
0.331 mA
52.7
34.4
130
0.214 mA
47.6
33.5
150
0.127 mA
37.6
31.5
170
63.7 µA
24.2
27.7
190
17.6 µA
8.4
18.4
Current
Recall from Table 2–1 in Section 2.3 that the Ford RE limit from 1.8 MHz to 200 MHz is
10 dBµV per meter. Table 3–4 shows this PCB circuit would fail the RE limit set by ESXW7T-1A278-AB spec at all of the spectral frequencies.
The predominant contribution to radiated emissions is due to the so-called CommonMode (CM) current flowing in cables attached to an electronic device, and acting as
efficient antennas in the frequency range which is considered (up to 2.5 GHz). The CM
current is simply the net current in the cable. Ideally, this net current should vanish,
because each current that enters the electronic device through the cable, also leaves it
through the cable. Due to parasitic effects, this balance is disturbed and a CM current
results. This CM current determines the amount of radiation because in the balanced case,
the radiated field of each of the different wires in the cable almost cancel each other.
Since only the net current in the cable is important, the cable may be considered as one
single wire carrying this CM current. In automotive electronic devices several hundred
different signals contribute to the overall CM current on attached cables. In order to
estimate the contribution of the different nets, the basic CM current generation principle
has to be understood and two basic mechanisms, i.e. current-driven and voltage-driven
current excitation, need to be considered.
The current-driven mechanism is due to the partial inductance of the return currents in
the ground plane, which produces a voltage drop across the ground plane, and injects the
CM current into the attached cable.
The second mechanism is voltage-driven, because the signal voltage directly drives the
CM current through the parasitic antenna capacitance.
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Figure 3–9 shows the setup for measuring CM currents from an electronic device. The
cable (wire harness) connects the electronic device to a load box that contains all of the
input circuitry and loads the device drives. A two-meter (2 m) harness is the standard
length used for measuring radiated emissions from an electronic device at Ford. The
electronic device, harness, and load box are placed over a ground plane.
Current
probe
ESC
Load
Box
Harness
Ground Plane
Spectrum
analyzer
Figure 3–9. Setup for Measuring CM Currents
The RF current probe measures the net current that exits on the harness. Equation 3–8
gives the CM current calculation as:
I CM =
VSA
Zt
[Amperes]
Equation 3–8. Common Mode current
Where,
VSA is the voltage that the spectrum analyzer measures
Zt is the probe transfer impedance in ohms
Equation 3–9 gives the electric field in dBµV per meter for a short wire (relative to
wavelength) in free space due to the spectral amplitude of current In. Use this equation to
estimate the electric field emissions due to CM current.
I Lf 

ECM = 20 log 10  1.26 n n 
r 

[dBµV/m]
Equation 3–9. E-field strength due to CM current
Where,
Frame 27 of 78
In is the spectral signal current
L is the length of the cable
fn is the spectral signal frequency
r is the distance from the wire to a measuring antenna or the distance between
a radiating generator wire and a receptor circuit.
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Solving Equation 3–9 for the current gives Equation 3–10:
In =
0.4 × E
fn
[µAmperes]
Equation 3–10. CM current
Where,
E is in µV/m
fn is in MHz
This equation includes the factors from ES-XW7T-1A278-AB specification, where the
wire is two (2) meters long and the antenna distance from the wire harness is one (1)
meter. It takes much less CM current than DM to result in the same Radiated Emissions.
Table 3–5 shows the maximum CM current that can flow on a single wire to just meet the
Ford limit for radiated emissions. To find and measure the maximum CM current move
the current probe along the harness length while monitoring the current with a spectrum
analyzer.
Table 3–5. Ford RE Limit vs. Sample Data
Frequency
(MHz)
10
Ford RE limit
(dBµV/m)*
30
Ford RE limit
(µV/m)
31.6
I
(µA)
1.3
30
10
3.16
0.04
100
10
3.16
0.013
200
10
3.16
0.006
Note: Data obtained using Fischer F33-1 current probe.
* Limits per ES-XW7T-1A278-AB specification
3.7. Crosstalk
Vehicles contain many conductors such as wires, vehicle sheet metal, PCB tracks, and
PCB ground planes. Wires can become a dominant factor since they may couple
electromagnetic energy to other wires in the same bundle, and hence into an electronic
device (module). Crosstalk is the coupling of signals between conductors. Crosstalk can
occur through the following mechanisms:
•
•
•
Frame 28 of 78
Common impedance coupling
Capacitive coupling
Inductive coupling
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3.7.1. Common Impedance Coupling
Common Impedance Coupling exists when two or more circuits share a common
conductor to source or sink current. The common impedance is a form of
communication between the two circuits. Current passing through the common
impedance develops a voltage, which appears directly in the receptor circuit. This
shared impedance can occur in the automotive battery feed and ground distributions
and shared signal voltage feed and signal returns. Common impedance coupling can
cause many problems in PCBs and integrated circuits.
Figure 3–10 shows a common impedance in the positive and negative sides of a
battery distribution circuit for two devices, A and B. Current flowing from circuit A
raises the ground potential under circuit A and circuit B. Likewise, current flowing
from circuit B has the same effect on circuit A. The voltage drop caused by current
flow from either circuit changes the ground potential of the other (receptor) circuit.
This is a form of communication between devices A and B, which may cause a
problem, depending on the sensitivity of the other circuit. The same mechanism of
common impedance occurs on the positive side of the battery.
Common
Impedance
Battery
Vn1
Device A
Common
Impedance
IA + IB
Device B
Vn2
IA + IB
Common impedance = Resistance + Inductance
Figure 3–10. Elements of Common Impedance
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3.7.2. Capacitive and inductive coupling
Figure 3–11 shows capacitive and inductive coupling between two circuits.
Cg
Vg
Rg
Cgr
Lgr
Cr
R1
R2
Figure 3–11. Inductive and Capacitive Coupling Between Two Circuits
Where,
Vg is generator voltage
Cg is the generator capacitance
Rg is the generator circuit load
R1 and R2 are terminating resistances of the receptor circuit
Cr is the capacitance of the receptor circuit
Cgr is the mutual capacitance from the generator to the receptor circuit
Lgr is the mutual inductance from the generator to the receptor circuit
3.7.3. Capacitive coupling
A signal voltage creates an electric field from wires and PCB traces. Capacitive
coupling results from the interaction of a time-varying electric field between a
generator and receptor circuit. Figure 3–11 illustrates that capacitive coupling
results from a mutual capacitance Cgr. The mutual capacitance provides a path for
EMI current to flow from the generator circuit to the receptor circuit.
Figure 3–12 shows the equivalent circuit for the capacitive coupling shown in
Figure 3–11. Rr is the parallel equivalent circuit for R1 and R2. Whenever the
generator signal changes, it induces a noise voltage in the receptor circuit. By
inspecting Figure 3–12 one can see that capacitive coupling is essentially a
differentiator circuit. The presence of Cgr differentiates the square wave to produce
the receptor noise voltage.
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Square Wave
Receptor Noise
Cgr
Vg
Cg
Cr
Rr
Figure 3–12. Capacitive Coupling
The amount of noise voltage that the generator circuit induces into the receptor
circuit depends on the generator frequency and Cgr, which is largely a function of:
•
•
Parallel length of the two circuits, and
Separation between the two circuits
Equation 3–11 gives the mutual capacitance in picofarads per inch between two long
conductors:
C
=
l
0.7 ε r ( eff )
2
d
2r  

ln  + 1 −   
r
 d  

[pF/in]
Equation 3–11. Mutual Capacitance in wires
Where,
d is the distance between the center lines of the wires
r is the wire radius
εr is the permittivity of the wire insulation material.
The effective permittivity, εr(eff) depends on the separation distance. It varies from
1<εr(eff)<3.2
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Equation 3–12 gives the capacitance in picofarads per inch between two wires over
a ground plane.
C
=
l
  2h  2 
0.7 ε r ( eff ) ln 1 +   
  d  
  2 h 
ln r 
  
2
[pF/in], for
2h
>> 1
r
Equation 3–12. Mutual Capacitance
Where,
d is the distance between the center lines of the wires
r is the wire radius
h is the distance between the center lines of the wires and the ground plane
(hight)
Table 3–6 shows how mutual capacitance varies between two 18 gauge wires
(radius = 0.024 inch) with and without a ground plane -- the ground plane returns
the currents of both wires. The ground plane reduces the mutual capacitance
between the wires by increasing the self-capacitance – the capacitance to its ground
reference – on each wire. Butting wires show an increase in capacitance due to the
dielectric constant of the wire insulation (PTFE with an Er equal to 2.1).
Table 3–6. Mutual Capacitance in Two Wires
Separation Distance
(inches)
No ground plane
(pF)
Wires butting
18.77
Ground plane
h = 02 in
(pF)
6.28
0.1
6.16
3.01
0.2
3.99
1.71
0.5
2.77
0.53
1.0
2.25
0.16
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Equation 3–13 gives the noise voltage, Vn due to capacitive coupling. Rf is the
parallel equivalence of R1 and R2, which equals 2πxf, where f is the frequency or
frequencies of Vg.
Vn = jω × Rr C grV
whenever
Rr <<
1
jω (C gr + C r )
and
 C gr
Vn = 
C +C
r
 gr

V g


whenever
Rr >>
1
jω (C gr + C r )
Equation 3–13. Voltage Noise due to capacitive coupling
To reduce capacitive coupling:
•
•
•
•
•
•
•
•
Decrease the generator frequency
Decrease the parallel length between the circuits
Increase the separation between the circuits
Orient the receptor circuit to the generator circuit at 90û
Increase Cr
Decrease Rr
Shield the generator and/or the receptor circuit
Place conductors over a ground plane
3.7.4. Inductive coupling
Inductive coupling results from the interaction of a time-varying magnetic field
between a generator and receptor circuit. Inductive coupling can occur at low or
high frequencies. Crosstalk from inductive coupling is more prevalent when highlevel and fast-rising currents transients are conducted in a low-impedance circuits.
Signal current creates a magnetic field that surrounds the conductor. Figure 3–11
illustrates that conductive coupling results from a mutual inductance Lgr. The mutual
inductance provides a path for magnetic flux to couple from the generator circuit to
the receptor circuit.
Figure 3–13 shows that inductive coupling is essentially a simple magnetic
transformer. The generator circuit is the primary and the receptor circuit is the
secondary of the transformer. The figure also illustrates that when Vg is a sine wave
then Vnoise is a sine wave as well but with a reduced amplitude. When Vg is a square
wave then Vnoise shows noise spikes when the square wave changes.
Frame 33 of 78
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Lgr
+
Vg
Ig
Rr
Vnoise
-
Figure 3–13. Inductive Coupling
The amount of noise voltage that the generator circuit induces into the receptor
circuit depends on the generator frequency and Lgr, which is a function of:
•
•
•
Receptor and generator area
Parallel length of the two circuits
Separation between the two circuits
Equation 3–14 gives the mutual inductance in microhenries per inch between two
long circular conductors over a ground plane. The ground plane returns the currents
of both circuits.
  2h  2 
= 0.00254 ln 1 +   
l
  d  
Lgr
[µH/in]
Equation 3–14. Mutual Inductance
Where,
h is the distance between the conductor centers and the ground plane
d is the distance between the center lines of the conductors
Figure 3–14 shows mutual inductance in microhenries per foot between two wires
over a ground plane, versus the ratio of wire height to wire separation. The figure
illustrates that mutual inductance increases as the areas of the generator and receptor
circuits, increase.
Frame 34 of 78
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0.5
Lgr (microhenrys/foot)
0.4
0.3
0.2
0.1
0
0.1
1
10
100
h/d
Figure 3–14. Mutual Inductance Between Two Wires
Equation 3–15 gives induced noise voltage from inductive coupling:
N noise = L gr ×
di g
dt
Equation 3–15. Noise voltage due to inductive coupling
Where,
di g
is the rate of change of the generator current
dt
Therefore, fast changing generator currents will induce larger noise voltages in
receptor circuits.
Equation 3–16 also gives the noise voltage due to inductive coupling:
Vn = jω × BA cos Θ
Equation 3–16. Noise voltage due to inductive coupling
Where,
Frame 35 of 78
B is the magnetic flux density (weber/cm2)
A is the receptor circuit area (cm2)
Θ is the angle between the generator and receptor circuit
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3.8. Twisted Pair
A twisted pair of wires reduces inductive coupling by canceling induced magnetic field
voltages. Figure 3–15 shows magnetic field (B) coupling into a circuit. Vs represents an
input signal to an electronic device on a vehicle. Rin represents the input impedance of the
module. The figure shows that the device input voltage, Vin, is the sum of Vs and the
noise voltage Vn, which the magnetic field induces.
Vn
Vin = Vs + Vn
Rin
Vin
Vs
B
Figure 3–15. Magnetic Field Coupling into Circuit
Vn/4
Vn/4
Vn/4
Vn/4
Rin
Vin
Vs
B
Figure 3–16. Magnetic Field Coupling into Twisted Wire Pair
Figure 3–16 shows the circuit in Figure 3–15 that uses a twisted pair. The twisting
produces four equal loop areas with equal noise voltages. By summing all the voltages
around the circuit the noise voltages cancel due to the twisting. This is why twisted pairs
work best to reduce inductive coupling into a receptor circuit.
Vin = −
Vn Vn
V V
−
+ Vs + n + n = Vs
4
4
4
4
Equation 3–17. Inductive Coupling in twisted-wire pair
Frame 36 of 78
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To reduce inductive coupling:
•
•
•
•
•
•
•
•
•
•
•
•
Reduce the receptor circuit area
Increase the separation between the generator and receptor circuit
Reduce the parallel length between the generator and receptor circuit
Twist the receptor wires if the receptor current returns back through a wire
Orient the receptor circuit to the generator circuit at 90û
Twist the generator wires if the generator current returns back through a wire
Reduce the operating frequency of the generator circuit
Reduce the rate of change of the generator current
Reduce the generator circuit area
Shield the receptor circuit with a shield grounded at both ends
Use a shield of magnetic material
Place the conductors over a ground plane. The ground plane must return the
conductor currents
3.9. Shielding
EMI control must originate in the initial design of an E/E device. Some E/E devices
require shielding to keep radiated energy away from module circuitry, or to keep EM
energy from radiating from the module circuitry. Using a shield as a post-design fix to
provide additional EMI protection adds cost and development time.
Shielding places a conductive partition between two regions in space. Shielding reflects
and absorbs radiated EM energy, as shown in Figure 3–17. The noise source side of the
shield reflects most of the incident energy, and the remaining energy enters the shield. As
the field propagates through the shield, it absorbs some of the energy. When the field
encounters the other surface of the shield some of it reflects back into the shield. The
remaining electromagnetic energy enters the protected region.
E field
Incident
wave
Shield
Transmitted
Wave
Attenuated
incident
wave
H field
Reflected
wave
Internal
reflected
wave
Figure 3–17. Effectiveness of Shielding
Frame 37 of 78
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A shield presents two losses to electromagnetic energy:
•
•
The reflection loss (R) is the air-to-shield and shield-to-air loss. Shield
reflection varies with the type of field.
The absorption loss (A) is the energy lost due to absorption as the field
propagates through the shield. Shield absorption does not vary with the type
of field. However, absorption loss varies with the type of shielding material.
Equation 3–18 shows that the shielding effectiveness (SE) is the sum of the reflection
and absorption losses. The decibel is the unit of SE:
SE = R + A [dB]
Equation 3–18. Shielding effectiveness
A SE of 40 dB indicates that the shield reflects and absorbs 99.99% of electromagnetic
energy. Therefore, only 0.01% of EM energy penetrates the shielding system.
Magnetic material has a relative permeability (µr) greater than 1. Where µr is the ratio
of the material's magnetic field conduction ability to air (µr varies with frequency). At
ratios greater than 1, the magnetic field would rather conduct through the magnetic
material than through the air. Table 3–7 shows the relative permeability of some common
materials. The table also gives the relative conductivity of the material. The relative
conductivity (δr) of the material is the ability to conduct current relative to copper. It is
the inverse of resistivity (ρ).
Absorption loss (A)
1
A = 3.34t ( f × µ r × δ r )2
[dB]
Equation 3–19. Absorption loss
Where,
t is the thickness in inches
Shield absorption does not vary with the type of field. However, absorption loss varies
with the type of shield material.
Reflection loss (R)
A shield can protect against the following:
•
•
•
Frame 38 of 78
Electromagnetic (EM) field
Electric (E) field
Magnetic (H) field
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Electromagnetic, electric, and magnetic fields require different shield design. An
electromagnetic filed has both, an electric and magnetic fields oriented 90 degrees to each
other. These fields travel together as the electromagnetic wave propagates through space.
The electromagnetic field is usually referred to as a far field plane wave.
Any metallic shield will reflect electromagnetic and electric fields. Here, shielding is a
function of:
•
•
•
•
•
Frequency
Shield thickness
Shield's relative conductivity
Shield's relative permeability
Any openings (apertures) in the shield
In general, the field close to an E/E device is either primarily an electric or a magnetic
field. For example, digital circuits on PCB generate a dominant electric field, whereas a
motor generates a dominant magnetic field (at one-sixth of a wavelength distance from
the generator circuit these separate fields do not dominate, and any radiated emissions
become an electromagnetic wave).
In the near field, shielding is a function of the previously mentioned, and these additional
factors:
•
•
The impedance of the field generator
The distance from the field generator
As previously mentioned, any metallic shield will reflect an electric field. However, only
shields constructed from magnetic material are effective reflectors of magnetic field.
Table 3–7. Relative Permeability of Common Metals
Metal material
Silver
Copper-annealed
Gold
Aluminum
Brass
Tin
Beryllium
Nickel
Stainless steel (430)
Iron
Mu-metal (at 1 kHz)
Permalloy (at 1 kHz)
Frame 39 of 78
µr
1
1
1
1
1
1
1
100
500
1000
20,000
80,000
Rev. A
dr
1.05
1.0
0.70
0.61
0.26
0.15
0.28
0.20
0.02
0.17
0.03
0.03
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3.10.
Resistance
Table 3–8 shows some wire characteristics (AWG is the American Wire Gauge).
Table 3–8. Resistance in Wires
AWG
Number of
strands
1
19
Strand
diameter
(inches)
0.0664
Cable
diameter
(inches)
0.332
MΩ
per foot
@ 20ûC
0.1288
2
7
0.0974
0.292
0.1644
4
7
0.0772
0.232
0.2582
6
7
0.0612
0.184
0.4105
8
7
0.0486
0.147
0.6528
10
7
0.0385
0.116
1.028
12
7
0.0305
0.096
1.650
14
7
0.0242
0.073
2.624
16
7
0.0192
0.060
4.172
18
7
0.0152
0.048
6.636
20
7
0.0121
0.035
10.54
22
7
0.010
0.030
14.74
Equation 3–20 gives the resistance (R) of a solid copper slab:
R=
1.724 × 10 −6 × l
wt
[Ω]
Equation 3–20. Resistance in Copper
Where,
Frame 40 of 78
1.724x10-6 Ω-cm equals the resistivity (ρ) of copper
t is the thickness in cm
w is the width in cm
l is the length of the slab in cm
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Table 3–9 shows the resistance of grounding straps in milliohms per foot. The strap size
gives the cross-sectional dimensions.
Table 3–9. Resistance in Grounding Straps
Strap size
(cm)
0.05x0.5
0.05x1.0
0.05x2.0
0.1x0.5
0.1x1.0
0.1x2.0
3.11.
Copper
2.11
1.06
0.53
1.06
0.53
0.26
Resistance mΩ/foot
Steel
Aluminum
11.85
3.27
5.93
1.64
2.96
0.82
5.93
1.63
2.96
0.82
1.49
0.41
Inductance
The inductance L [self inductance] of a circuit or device depends on geometry and the
magnetic properties of the medium in and around the circuit or device. Larger circuits
have more inductance. Devices such as solenoids that use a core of magnetic material
have more inductance.
Inductive reactance is an inductor's resistance to a change in circuit current. Equation 3–
21 gives the inductive reactance (impedance) versus frequency of an inductor:
Z L = 2π × frequency × L
[Ω]
Equation 3–21. Inductive Reactance
Where,
L is the inductance of the conductor
Equation 3–22 gives the inductance of a rectangular conductor or ground strap:


 2l 
w+t
L = 0.002l 2.303 log 
 + 0.2235
 + 0.5 
w+t
 l 


[µH]
Equation 3–22. Inductance in rectangular conductor
Where,
l is the strap length in cm
w is the strap width in cm
t is the strap thickness in cm
Table 3–10 shows the inductive reactance for three ground strap widths. Each strap has a
length of one foot (30.48 cm) and thickness of 0.05 cm. Note that doubling the strap
Frame 41 of 78
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width does not reduce the impedance by one-half due to the logarithmic dependency of
inductance in Equation 3–22.
Table 3–10. Inductive Reactance vs. Frequency
w = 0.5 cm
Inductive Reactance
Ω/foot
w = 1.0 cm
w = 2.0 cm
100 Hz
200x10-6
175x10-6
150x10-6
1 kHz
2.00x10-3
1.75x10-3
1.50x10-3
10 kHz
20.0x10-3
17.5x10-3
15.0x10-3
100 kHz
200.0x10-3
175x10-3
150x10-3
1 MHz
1.997
1.750
1.497
100 MHz
199.7
175.0
149.7
500 MHz
998.3
875.2
748.4
1 GHz
1,996.7
1,750.4
1,496.9
Frequency
Figure 3–18 shows two wires where one wire carries signal or power current and the
other wire carries the signal or power return current. Both wires have the same radius, r.
I
I
Radius = r
d
Figure 3–18. Inductance in Parallel Wires
Equation 3–23 gives the self-inductance in micro henrys per foot for two circular
conductors of equal radii:
L
d 
= 0.12 ln 
l
r
[µH/foot]
Equation 3–23. Inductance in parallel wires
Frame 42 of 78
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Where,
d is the distance between the center lines of the two conductors in inches
r is the wire radii in inches
Table 3–11 below, shows the impedance of various solid copper wire pairs versus
frequency. Each wire has a length of one (1) foot and wire separation of one (1) inch.
Note: wires with smaller diameters will have higher inductances.
Table 3–11. Impedance in Solid Copper Wires
4 AWG
Inductive Reactance
Ω/foot
16 AWG
22 AWG
100 Hz
172x10-6
277x10-6
329x10-6
1 kHz
1.72x10-3
2.77x10-3
3.29x10-3
10 kHz
17.2x10-3
27.7x10-3
32.9x10-3
100 kHz
172x10-3
277x10-3
329x10-3
1 MHz
1.72
2.77
3.29
100 MHz
172
276.9
329.3
500 MHz
860
1.38x10+3
1.65x10+3
1 GHz
1.72x10+3
2.77x10+3
3.29x10+3
Frequency
Figure 3–19 shows a circular conductor of radius, r, located at height, h, over a ground
plane. The ground plane returns the current that flows through the circular conductor.
I
I
h
Ground plane
Figure 3–19. Inductance in Wires over Ground Plane
Equation 3–24 gives the self-inductance in micro henrys per foot for a circular wire over
a ground plane, with ground plane returning the current. Note that the inductance is about
one-half the inductance in Equation 3–23.
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L
 2h 
= 0.06 ln 
l
 r 
[µH/foot]
Equation 3–24. Self-Inductance
Figure 3–20 shows the self-inductance, L(d) in micro henrys per foot of an 18 gauge
solid-copper wire pair versus the ratio of the separation distance (d) to the wire radius (r).
The wire separation varies from 0.1 to 10 inches and wire radius is 0.020 inches.
Figure 3–20 also shows the inductance, L(h) in micro Henrys per foot of an 18 gauge
solid-copper wire over a ground plane versus the ratio of the wire height (h) to the wire
radius (r). The wire height varies from 0.1 to 10 inches. Note the significant difference
when a ground plane returns the current instead of a wire. The ground plane provides
more mutual inductance to the wire to reduce the overall inductance circuit.
0.08
0.06
L(d)
0.04
L(h)
0.02
0
0
2
4
6
8
10
d/r & h/r
Figure 3–20. Inductance of Ground Plane vs. Wire Inductance
Frame 44 of 78
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Table 3–12 also compares the self-inductances of the wires in Figure 3–20.
Table 3–12. Self-Inductance in Wires
0.1
L(h)
Wire over
ground plane
µH/foot
0.132
0.2
0.168
0.252
0.5
0.228
0.360
1
0.264
0.444
2
0.312
0.528
5
0.360
0.636
10
0.408
0.720
Wire height (h) or
wire separation (d)
(inches)
L(d)
Wire pair
µH/foot
0.168
Equation 3–25 gives the inductance of an air-core inductor as:
L=
r2 × N 2
9 r + 10l
[µH]
Equation 3–25. Inductance in air-core inductors
Where,
r is the core radius in inches
l is the core length in inches
N is the number of turns
Equation 3–26 gives the inductance of a toroid as:
d 
L = 0.00508 N 2 × b × µ r × ln 2 
 d1 
[µH]
Equation 3–26. Inductance in toroids
Where,
Frame 45 of 78
N is the number of turns
b is the core length in inches
d1 is the inside diameter in inches
d2 is the outside diameter in inches
µr is the permeability of the toroid material
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PART IV: EMISSIONS FROM INTEGRATED CIRCUITS
4. SCOPE
With today's advancements in semiconductor technology and push towards faster MCUs and
peripherals, the main concern is the amount of emissions generated by the microcontroller.
At present, no single specification on the pass or fail criteria for integrated circuits exists, and
limits in general depend upon the application, functional requirements, and local or mandated
emission requirements.
In order to promote international co-operation concerning standardization in the E/E fields
several procedures have been developed to facilitate verification of the electromagnetic
behavior of integrated circuits. These test methods guarantee a high degree of repeatability
and correlation of RE measurements. Ford recommends that integrated circuits, prior to ESC
installation, be subjected to emissions profiling in order to quantify their EMI contribution to
the overall system performance.
Rather than imposing a single limit on all integrated circuits, they ought to be classified
according to their radiated emissions levels.
4.1. Applicable Documents
IEC Documents:
IEC-61967*
"Integrated Circuits, Measurement of Electromagnetic Emissions,
150 kHz to 1 GHz", IEC SC47A/WG9, 2001 or later edition
SAE Documents:
SAE-J1752/3**
"Electromagnetic Compatibility Measurement Procedures for
Integrated Circuits - Integrated Circuit Radiated Emissions
Measurement Procedure 150 kHz to 1000 MHz, TEM Cell", SAE
1999 or later edition
Ford Documents:
XW7T-1A278-AB
"Electronic Component EMC Requirements and Test
Procedures", FMC 1999 or later edition
* IEC-61967, as part of International Standard IEC SC47A/WG9, is a project number for
emissions test methods. This specification recommends test procedures to use and provides
reference levels required in order to comply with automotive requirements
** SAE-J1752/3 procedure - for further details refer to this Society of Automotive Engineers
manual, 1999 or later edition
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4.2. EMC Test Recommendations
It is recommended that all applicable integrated circuits be subjected to radiated
emissions testing according to procedures specified in International Standard IEC-61967*
- "Integrated Circuits, Measurement of Electromagnetic Emissions, 150 kHz to 1 GHz".
The suppliers are encouraged to obtain test reports from IC manufacturers following the
testing to establish a quantitative measure of RF emissions from ICs. The test reports
should contain all salient information and parameters of the tests and test results. The
reports should include test data, test set up description, photograph or sketch of the set up,
software operating modes, and a summary of test results.
The recommended frequency range is 150 kHz – 1 GHz, but this may be extended if the
specific procedure is usable over an extended frequency range.
4.3. Test Procedure Applicability
The following criteria may be used to determine if a part is a candidate for RE testing:
•
•
Digital technology, LSI, products with oscillators or any technology which
has the potential of producing radiated emissions capable of interfering with
communication receiver devices. Examples include microprocessors, high
speed digital ICs, FETs incorporating charge pumps, devices with watchdogs,
switch mode regulator control and driver IC's
For all new, re-qualified or existing IC's that have undergone revisions from
previous versions
Examples of factors that would be expected to affect emissions are changes in the
components:
•
•
•
•
•
•
Frame 47 of 78
Clock drive (internal or external)
I/O drive
Manufacturing process or material composition that reduces rise/fall time
Minimum feature size (die shrink)
Package or pinout configuration
Lead-frame material
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4.4. IC Emissions Reference Levels
The IC emissions acceptance levels proposed by Ford differ from those proposed by SAE
J1752/3** document, and are shown in Figure 4–1. These reference levels apply to
measurements over the frequency range of 150 KHz to 1000 MHz in units of dBµV.
60
50
40
Ref
Ref
Ref
Ref
dBuV
30
20
10
Level 4
Level 3
Level 2
Level 1
0
-10
-20
0.1
1
10
100
1000
MHz
Figure 4–1. IC Radiated Emissions Acceptance Levels
Note: Limits for the TEM Cell are derived from the Magnetic Dipole Moment Reference
Levels in SAE J1752/2 using the equation in J1752/3 Appendix D.
Frame 48 of 78
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The following rating system was developed to help classify IC emissions:
Table 4–1. Rating Levels for IC's
Classification
Description
1
RE ≤ Reference Level 1
2
Reference Level 1 ≤ RE ≤ Reference Level 2
3
Reference Level 2 ≤ RE ≤ Reference Level 3
4
Reference level 3 ≤ RE ≤ Reference Level 4
NR – Not
Recommended
RE ≥ Reference Level 4
4.4.1. Level 1
No EMC RE risk. Full EMC RE compliance may be achieved with minimum effort.
Greatest flexibility in EMC design application. Potential cost reduction opportunity
through reduction of:
•
•
•
•
•
Number of components on PCB
Material cost (single-sided PCB opportunity)
PCB design iterations
Testing cycles
PCB assembly time
4.4.2. Level 2
Minor EMC RE risk. Less flexibility in EMC design than Level 1 but full
compliance may still be achieved with little effort. Additional components and/or
stricter adherence to industry's fundamental EMC design practices may be required.
4.4.3. Level 3
Moderate EMC RE risk. Close follow up of EMC fundamental design principles is
strongly recommended. Additional preventive measures may be required, and may
include the introduction of T-filters, Pi-filters, ferrites, inductors, etc. Extra PCB
layers, components, and/or component shielding may also be necessary.
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4.4.4. Level 4
High EMC RE risk. Strict adherence to EMC fundamental design principles is
strongly recommended. Additional preventive measures, such as PCB shielding
(metal housing), harness and/or component shielding, extra layers, T-filters, Pifilters, ferrites, chokes, etc., may also be required at a substantial additional expense.
Supplier(s) choosing Level 4 IC(s) should keep in mind that in order to implement
and validate these additional preventive techniques an extra design time and/or test
time may be necessary.
4.4.5. Level NR
IC(s) classified as category NR may not be able, even with extra preventive
measures, to comply with Ford RE requirements and therefore should not be
considered. ESC(s) with NR category parts found to exceed Ford RE requirements
may be denied EMC approval.
4.5. Data Submission
It is recommended that suppliers request from IC manufacturers emissions plots (both
orientations) along with setup information (ref. SAE-J1752/3**, figure 4) for all
applicable integrated circuits prior to PCB layout. Such data should be available for
review by Ford EMC personnel upon request.
4.6. Radiated and Conducted Immunity
If the supplier is aware of any radiated or conducted immunity concerns which have the
potential of jeopardizing product robustness or timely delivery, Ford EMC representative
should be notified immediately.
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PART V: EMC DESIGN GUIDELINES FOR PCB
5. GENERAL
Electromagnetic compatibility MUST be considered early in the design stage of any E/E
device. If it is ignored early in the design cycle, as problems are encountered during testing
or in the field, fixes become very expensive – primarily because the design is less flexible.
A noise problem on a printed circuit board can be solved at the layout stage for a relatively
small cost. However, if it is dealt with after the design has been completed the cost can
increase 10 or even 100 fold. Figure 5–1 illustrates that EMC achieved at the design stage is
a one-time cost, while EMI problems and fixes in the field can cost an enormous amount of
time and money if engineering changes are necessary.
Cost ($)
Equipment Designed WITHOUT
EMC Consideration
Equipment Designed with
EMC Consideration
Early Design
& Functional
Specs.
Prototype
Development
Testing
Production
Time
(Equipment life)
Figure 5–1. Relative Costs of EMC vs. NO EMC Design Strategy
Attention to good practices in circuit (hardware) design can provide inherent
electromagnetic compatibility. Examples of such practices are decoupling power and I/O
lines to ground, providing selectivity and voltage limiting, maintaining linearity, limiting
signal bandwidth to only that needed by the circuit, using negative feedback, wave
shaping, signal interlocks, synchronization, refresh cycling, and fault tolerant software.
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EMC Design Guide for Printed Circuit Boards
5.1. Board Structure/Ground Systems
1. When creating a schematic each component should have an appropriate reference
designator identifying it as a member of a specific functional group. This will assure
correct placement of components on PCB during layout.
A recommended arrangement of functional groups on PCB is shown in Figure 5–2.
All components should be placed with an appropriate functional group and their
tracks routed within their designated PCB area.
Digital
Group 4
Analog
Group 3
Power Supply
Group 2
I/O (Connector)
Noisy
Group 5
Group 1
Figure 5–2. Arrangement of Functional Groups on PCB
2. Place ground plane(s) under all components and all their associated tracks - a
continuous ground plane with no avoidance in IC package or I/O connector area is
recommended. (Figure 5–3).
3. Maximize copper areas to provide low impedance for power supply decoupling careful arrangement of components and connections (traces) may allow large areas of
PCB to be filled with ground. Figure 5–3.
YES
NO
Figure 5–3. Maximizing Ground on PCB
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EMC Design Guide for Printed Circuit Boards
4. All two-layer PCBs, where ground plane is not feasible, should utilize a ground grid
system.
The top and bottom side of PCB should look like a ground plane with horizontal
ground tracks on one side, and vertical ground tracks on the other side. Ground grid
traces should be as wide as possible and be placed apart as close as possible. A
ground grid is achieved by connecting vertical and horizontal lines on opposite sides
of PCB with vias. A via is a plated through hole that interconnects two or more PCB
layers. In addition, multi-layer PCBs should use ground grids even if they employ one
or more ground planes. A properly designed ground grid is the next best ground
system (Figure 5–4)
= via
Figure 5–4. Ground Grid Technique
5. Install ground vias around the perimeter of the PCB every 0.5 inches or less as shown
in Figure 5–5. Connect these vias together with 15 mil (0.4 mm) minimum trace
thickness on all layers.
This should help to contain frequencies up to 5 GHz on the PCB board by forming a
"Faraday's cage". Routing of traces outside the ground vias should not be permitted
except for connections with the 'outside world'.
Vias 0.5” max. apart
connect Gnd planes
together
PCB
Figure 5–5. Creating 'Faraday's Cage'
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EMC Design Guide for Printed Circuit Boards
6. For multi-layer boards the recommended layer stack-up is shown in Figure 5–6.
Component Side - horizontal routing
Ground plane
Power plane
High speed - vertical routing
High speed – horizontal routing
Bottom side – vertical routing
Figure 5–6. Layer Stack-up
7. On boards without a ground plane, i.e. two-sided boards, power and ground traces
should be routed adjacent to or on top of one another on different layers to reduce
loop area (Figure 5–11).
8. A solid ground 'island' should exist underneath all High Speed Integrated Circuits
(HSICs) on surface layers. Figure 5–7.
9. Whenever possible, place a ground via next to all IC's ground pins as shown in Figure
5–7. Frequent use of vias interconnecting grounds on both sides of PCB or on
different layers of the board may help lower RF impedance in the ground structure.
Via to GND
High Speed
Integrated
Circuit
C
Via to VCC
d
Figure 5–7. IC Ground
10. All ground planes, belonging to the same net, should be conductively tied together
with low impedance connections at each component's ground pin.
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EMC Design Guide for Printed Circuit Boards
11. Ground returns from high-frequency digital circuits and low-level analog circuits
should never be mixed. Assure that ground return paths for analog, digital, or power
signals don't flow through each other's circuits.
12. Keeping ground leads shorter than one-twentieth (1/20) of a wavelength may prevent
excessive radiated emissions and may help to maintain low impedance.
13. Single-point grounding scheme should only be used for low level and low frequency
circuits (below 1 MHz). Figure 3–3.
14. Multi-point grounding scheme should be used for high frequency circuits (above
1MHz) to keep ground impedance low. Figure 3–4.
15. Assure even distribution of ground pins across all connector pin fields (including
ribbon cables or custom device packages) to prevent local ground upset due to
transient currents. The number of connector ground pins required should be
determined prior to start of layout (Figure 5–21).
16. There should be no floating metal of any kind near any PCB. All ground segments
with length-to-width ratio greater than 10:1 should have, at the minimum, one GND
via at each end tying them to rest of PCB ground structure. Figure 5–8.
Ground
Via
Tracks
NO
NO
YES
Figure 5–8. Eliminating Floating Ground
17. For PCBs without a ground plane, a minimum of one ground-return track should be
routed adjacent to each eight lines of address and data lines to minimize the loop area.
Keep the lines as short as possible. For the address lines, route the ground return next
to the least significant bit (LSB) since this line is likely to be the most active.
18. Avoid ground loops. They can be the source of radiated emissions. A ground plane or
ground grid are helpful in preventing ground loops from forming. Breaking a loop
with small gap may work at DC but gap capacitance may effectively close the loop at
higher frequencies, creating a large loop antenna. Apart from the RE problems, large
ground loops are known to cause the system to be susceptible to malfunction when
subjected to external EMI sources.
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EMC Design Guide for Printed Circuit Boards
19. Extend ground planes as far as possible beyond the boundaries of components and
their tracks and power planes – ground planes should extend beyond power planes
and any tracks by at least 20 times their layer spacing (Figure 5–9).
PCB
Tracks,
Components
Layer
Spacing
Ground
Power
Figure 5–9. Establishing Ground Plane Boundary
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EMC Design Guide for Printed Circuit Boards
5.2. Power Systems
20. Power supplies should be located close to power entry point to PCB, and as close as
possible to powered circuitry. Closely routed tracks (to minimize the area between
conductors, and hence the inductance) should be used to connect the power source to
the local power distribution system.
21. Power feeds should always be decoupled at their entry points onto the PCB.
22. Bulk capacitors should always be parallel decoupled with one or more smaller highfrequency capacitors with low ESL (equivalent series inductance). Place the smallest
value decoupling capacitor closest to a device to be decoupled.
23. Power should be distributed with a 'star', or grid, or power plane configuration but
never with point-to-point wiring (daisy-chaining). Use the positive side of the bulk
capacitor on the output of the voltage regulator as the "star" point (Figure 5–10).
Power Supply
D1
IN
OUT
C3
C4
VCC
'Star' Point
BATT
IN
C1
C2
+
+
C5
Figure 5–10. Power System's Star Point
24. The value of the bulk capacitor should be at least ten (10) times greater than the sum
of all the values of decoupling capacitors.
25. High-frequency, low-inductance ceramic capacitors should be used for integrated
circuit (IC) decoupling at each power pin – use 0.1 µF for up to 15 MHz, and 0.01 µF
over 15 MHz. The decoupling capacitor should be located as close as physically
possible from the IC's power pin. Figure 5–7.
Power distribution system must provide sufficient current, in time, for the device to
function properly. This includes high-peak current requirements during output
switching. Local discrete capacitors, when placed next to the device and attached to
power and ground with low inductance connections, will provide this current.
26. Printed circuit board traces which carry high switching current with fast rise/fall times
(5 – 10 ns) should maintain at least 3 mm spacing from other signal traces which run
parallel to them, and/or a ground guard traces should be placed between them.
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EMC Design Guide for Printed Circuit Boards
27. Corresponding power and ground signals should always be routed in parallel (sideby-side) or on top of each other (on adjacent layers) to minimize loop area thus
reducing loop impedance (Figure 5–11).
GND
NO
VCC
YES
Figure 5–11. Power and Ground Routing
28. VCC (clean power) traces should never be routed parallel to unfiltered (dirty) traces
that carry battery, ignition, high-current, or fast switching signals.
29. Use the lowest power, slowest logic that satisfies circuit requirements.
30. Power, ground, and signal traces on the board should be kept short and as wide as
possible. The traces should be shorter than the diagonal dimension of the board, and
ideally their length-to-width ratio should be kept at 10:1.
31. Placing ferrite beads on power tracks may provide attenuation of unwanted signals
above 1 MHz. When properly sized, these beads can be very effective in damping
high-frequency switching transients or parasitic ringing due to line reflections without
causing a DC loss. CAUTION: using ferrites may impede AC current flow.
32. Devices sensing battery or ignition, such as sensing resistors, should be placed at the
power entry point to PCB (close to I/O connector).
33. Devices, such as Zener diodes, MOV's or transzorbs should be placed at the power
entry point to PCB as their function is to limit/clip transients and spikes. Assure lowimpedance connection to ground.
34. Provide enough current storage (capacitor) on the incoming battery line when
designing switching power supplies and/or other circuits drawing discontinuous
currents from the battery, so that these currents do not appear on the wiring harness
where they can be radiated or conducted to other circuits.
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EMC Design Guide for Printed Circuit Boards
35. Closely grouped Power Switching and High Current circuits should be kept separate
from digital, low level analog, and relay circuits.
36. All switching mode power supply (SMPS) traces should be routed on one layer of
PCB with the SMPS reference plane placed directly on adjacent layer to minimize the
loop area.
37. Heatsink of the power switching transistor should be connected to the same potential
as the transistor's tab, either power or ground. Sometimes the heatsink is not directly
connected to the power switching transistor, but is insulated from it by a dielectric
material. This produces a parasitic capacitance between the power transistor and the
heatsink. Attaching the heatsink to a reference plane other than the power or ground
used by the power transistor may provide a path for common-mode currents.
38. The loop area of Switching Mode Power Supply (SMPS) snubber circuit should be as
small as physically possible (Figure 5–12).
39. The primary loop area of SMPS that uses a transformer should be minimized as much
as possible (Figure 5–12). The loop includes the positive lead of the bulk capacitor,
the primary windings of the transformer, collector or drain of the switching transistor,
current sense resistor; the ground lead of the current sense resistor, and the ground
lead of the bulk capacitor.
BATT
IN
TR
QSWITCH
RGATE
PWM
CBULK
RSNUBBER
RCURRENT_SENSE
CSNUBBER
Primary Loop Area
Snubber Loop Area
Figure 5–12. Primary Loop Area
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40. The secondary loop area of SMPS that uses a transformer should be minimized as
much as possible (Figure 5–13). The loop includes the positive side of the secondary
windings of the transformer, the series diode, the bulk capacitor; the ground lead of
the bulk capacitor, and the ground side of the secondary winding of the transformer.
DFLYBACK
TR
V1OUT
CBULK
V2OUT
Secondary Loop Area
Figure 5–13. Secondary Loop Area
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5.3. Digital Circuits
41. Digital clock connections (being very aggressive signals) should be the first 'nets' to
be routed, and they should be run on a single PCB layer adjacent to a ground plane.
42. All clock/address/data bus connections should be as short and direct as possible with
adjacent ground guard tracks or ground plane (Figure 5–14). Avoid using wires, stubs
or ribbon cables to distribute clock signals.
Bus
Microprocessor
Latch
RAM
Clock
RAM
Bus
RAM
Clock
RAM
RAM
RAM
Latch
Microprocessor
RAM
RAM
NO
YES
Figure 5–14. Minimizing Digital Bus Length
43. High-speed digital signals, such as data, address, and control lines of
microprocessors, should be grouped together and located as far from the I/O
connector as possible.
44. Always route signal tracks and their associated ground returns as close to one another
as possible to minimize the area (loop) enclosed by current flow (Figure 5–15).
Load
Circuit
trace
Driving
gate
Circuit
trace
Driving
gate
a) At low frequencies current
follows the path of least
resistance
Load
High-speed
return
current stays
close to
signal trace
b) At high-frequencies current
follows the path of least
inductance
Figure 5–15. Resistance and Inductance as Functions of Frequency
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EMC Design Guide for Printed Circuit Boards
45. Avoid running any traces other than ground next to/under crystals or any other
inherently noisy circuits.
46. Keep oscillators and clock generating IC's away from I/O connectors and close to the
chips they service, to keep the loop area small.
47. Always choose the lowest clock frequency and slowest rise and fall time for digital
signals that meets system requirements.
48. All critical nets, such as clocks, data strobes, etc., should be routed manually adjacent
to ground tracks or ground plane.
49. Place crystals or resonators as close as physically possible to the device they service,
and ideally on the same side of PCB. Minimize the track length between the oscillator
and the IC (Figure 5–16).
Microprocessor
S
G
Crystal/
Oscillator
Figure 5–16. Crystal/Oscillator placement
50. Placing RF filters ahead of such components as diodes, transistors, or integrated
circuits may prevent the RF from being converted to a DC or low frequency
disturbance signals.
51. Using terminators for traces which length exceeds twice the signal rise time may
prevent reflections at either end of a transmission line.
52. For long busses, keep high-speed traces separated from low speed signals by adding
extra spacing between the high-speed and low-speed signals, and by running high
frequency signals next to a ground trace.
53. All differential signal lines should be routed adjacent to one another to take full
advantage of magnetic field cancellation. Place ground guard traces on both sides of
the entire length of the differential-pair signals.
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EMC Design Guide for Printed Circuit Boards
54. Routing signal tracks perpendicular (90û) to each other on adjacent layers of a printed
circuit board may help to minimize cross-talk.
55. Controlling the rise and fall times, the duty cycle, and the fundamental frequency of
switched signals may help to minimize harmonic generation.
56. All unused IC inputs should be terminated to prevent unintentional random switching
and noise generation – i.e. unterminated CMOS inputs tend to self-bias into a linear
region of operation, significantly increasing the DC current drawn. Consult IC
manufacturer for recommendations.
57. Provide good ground imaging for long traces, high speed signals.
58. Keep high-speed traces away from the edge of a PCB.
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EMC Design Guide for Printed Circuit Boards
5.4. Analog Circuits
59. Analog or peripheral circuitry should be located as close to the I/O connector as
possible, and be kept away from high speed digital, high current, or power switching
circuits. Figure 5–2.
60. Routing of low level analog signals should be confined to analog section of PCB
only.
61. Low pass filtering should always be used on all analog inputs.
62. Printed circuit board traces which terminate at the device connector should be
decoupled of RF at the connector.
63. Ground guard tracks should always be routed adjacent to analog signals. Attach the
guard tracks at both ends with vias to sending and receiving circuits' ground.
64. If using suppression device across coils of relays and/or solenoids the suppressor
should be placed as close to the coil terminals as possible.
65. If a PWM signal is used to drive a solenoid, resistor suppression may be used. This
will prevent high rate of change of current (di/dt), which can cause excessive
magnetic field radiation.
66. Biasing resistors when placed as close as physically possible to the base of transistors
may prevent RF from coupling in and turning the transistor on or off. (Figure 5–17)
67. Base and emitter bypass capacitors should be located very close to transistors (Figure
5–17). They should be connected to ground with low impedance connection to
minimize inductance and loop area.
RB
Q1
CB
CE
Loop Area
Figure 5–17. Transistor Circuit Routing
68. Treat every trace carrying sensitive signals (especially into high input impedance
loads i.e. higher than 10 kΩ) as a receiving antenna when considering its routing.
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EMC Design Guide for Printed Circuit Boards
5.5. Communication Protocols
69. SCP physical layer EMC layout guidelines should always be considered and
followed. Use the latest revision available. For details refer to the following web site:
http://www_eese.ford.com/mux
70. CAN physical layer EMC layout guidelines should always be considered and
followed. Use the latest revision available. For details refer to the following web site:
http://www_eese.ford.com/mux
71. UBP physical layer EMC layout guidelines should always be considered and
followed. Use the latest revision available. For details refer to the following web site:
http://www_eese.ford.com/mux
72. FORD ISO-9141 physical layer EMC layout guidelines should always be considered
and followed. Use the latest revision available. For details refer to the following web
site: http://www_eese.ford.com/mux
73. ACP physical layer EMC layout guidelines should always be considered and
followed. Use the latest revision available. For details refer to the following web site:
http://www_eese.ford.com/mux
74. Always consult Ford EMC representative prior to PCB layout when/if a
communication protocol not listed here is considered for use in the design.
5.6. Shielding
75. All metallic shields of a system should be interconnected and grounded. Each shield
ought to have a low-impedance contact to ground in at least two places in order to
prevent its noise potential from coupling to the enclosed object. An ungrounded
shield's potential will vary with conditions and location, and therefore the noise
coupled to the object inside will vary also.
76. Placing a shield over the whole harness may limit radio frequency (RF) emissions
from it. To reduce susceptibility and cross-talk between high impedance lines within
a wiring harness use individual shields. Since shielding of harness is generally less
cost-effective and more labor intensive than other EMI suppression measures such as
filtering, it should not be the first choice in EMI suppression.
77. In order to realize shielding effectiveness, the shield ought to completely enclose the
electronics eliminating any penetrations such as holes, seams, slots, or cables. Any
penetrations in the shield unless properly treated, may drastically reduce the
effectiveness of the shield.
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EMC Design Guide for Printed Circuit Boards
78. Shields for low frequency signals (below 10 MHz) should be terminated and
grounded only at the source, thus preventing undesirable ground loops (Figure 5–18).
Conductor
Shield
Ground
Figure 5–18. Shielding of Low-Frequency Signals
79. Shields for frequency signals (above 10 MHz) should be terminated and grounded at
both ends (Figure 5–19).
Conductor
Shield
Ground
Figure 5–19. Shielding of High-Frequency Signals
80. Using twisted-pair wiring to the load should avoid creation of loop antennas that can
radiate magnetic fields.
81. When routing a wire harness along sheet metal keep it away from any openings as
much as possible. Openings may act as slot antennas.
82. Keep wire harness at least 5 inches away from electric field sources such as
distributors and magnetic field sources such as alternators and solenoids.
83. The exposed (unshielded) end of a shielded cable near a connector or a terminal
should not exceed 10 mm in length.
84. Always try to minimize the length of wire harness to reduce coupling and pickup.
85. Use twisted pair for sensitive low-frequency signals (below 1.0 MHz) and for circuits
with impedances less than 1.0 kΩ to provide accurate reference voltages.
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EMC Design Guide for Printed Circuit Boards
86. Coaxial cable should be used for transmission of RF (above 10 MHz) and where
impedance match over a broad frequency range is important (such as video
applications).
87. Circuits generating large, abrupt current variations should be provided with a separate
return lead to the ground in order to reduce transient pickup in other circuits.
5.7. Miscellaneous
88. All unused multipurpose Integrated Circuit (IC) ports should be configured as outputs
to prevent unintentional random state switching and noise generation – i.e.
unterminated CMOS inputs tend to self-bias into the linear region of operation, thus
significantly increasing DC current draw. Use appropriate pull-up or pull-down
discrete components. Consult IC manufacturer for recommendation.
89. Software may be used to disable (turn off) all unused clock outputs from an IC.
Consult IC manufacturer for recommendation.
90. Reducing output buffer drive from IC's may reduce radiated emissions. Consult IC
manufacturer for recommendations.
91. All output drivers should be protected against flyback transient from inductive loads.
92. ESD sensitive devices should never be located close to I/O connectors or any other
accessible openings where they may be damaged by an ESD event.
93. Keep ribbon cables and jumper strips away from IC's and oscillator circuits. Routing
over or near IC's should be avoided at all cost (Figure 5–20).
Ribbon Cable
Printed Circuit
Board
Osc
uP
Connector
NO
Figure 5–20. Packaging Considerations Affecting RE and CE
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94. When attaching ribbon cables to PCB's always provide multiple ground returns to
minimize loop area (Figure 5–21).
Ground
Signal
Critical
Signal
Figure 5–21. Use of Interspersed Grounds
95. Critical signals should never be placed on the outside conductors of shielded ribbon
cables (Figure 5–21).
.
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EMC Design Guide for Printed Circuit Boards
PART VI: REQUIREMENTS
6. MANAGEMENT OF CHANGE FOR EMC
Electronic modules are often changed after their initial release for many reasons and each
change must be evaluated for its potential impact on the EMC performance. It is a regular
dilemma as to which EMC tests to repeat if at all. Repeating all tests is the safest answer but
not cost effective. So it is necessary to define a logical process for considering the
performance of original module and the potential impact of the change.
Evaluating the exact impact of a change on the final EMC performance is not very easy and
requires in depth understanding of both electronic and electromagnetic characteristic of the
module. The product designers and the EMC experts of the module's supplier are best placed
to analyze the change and decide which EMC tests to repeat. Here at Ford, we expect to be
advised by the supplier as to what is changed, the analysis of the expected impact on EMC
characteristics and what tests shall be repeated. We will review suppliers' analysis and may
request additional tests as we see fit based on the information provided.
It is proposed that the following process is used for establishing what tests are to be run:
(Only tests that were deemed to be applicable in the initial tests should be considered)
Repeat RE310 tests if one or all of the following statements are true:
•
•
•
•
•
The change involved a software modification
The change resulted in a modified printed circuit board
Changes in any clock or PWM frequency, duty rate or other relevant
parameter
If the module is in a metallic housing and it has been modified in some way
Any other change that can reasonably be expected to influence radiated
emissions profile
6.1. Radiated Immunity:
6.1.1. For safety critical systems (containing one or more Class C functions)
Repeat RI 11X tests if change influences items listed for RE310 or any other change
that can reasonably be expected to influence radiated immunity profile.
Repeat RI120 and RI130 if the change impacts any aspect of input / output interface
(changes to hardware or software)
Repeat RI140 if the change involves any items that are sensitive to magnetic fields
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6.1.2 For non-safety critical systems
Repeat RI120 and RI130 if the change impacts any aspect of input / output interface
(changes to hardware or software)
Repeat RI140 if the change involves any items that are sensitive to magnetic fields
If the change involved any of the modifications listed for radiated missions listed
above, run radiated emissions first. If the results are different (+/- 3 dB in amplitude
and +/- 1 MHz in frequency) compared to the original data then repeat the RI11x
series of Radiated Immunity tests.
6.2. Conducted immunity:
Repeat CI210, CI220, CI230, CI240, CI260 and CI270 tests if the change influences any
of the circuit interfaces directly or indirectly connected to the vehicle supply network.
•
•
•
Changes to the component specification such as package size, value or rating
The placement or routing changes influencing component specifically
intended for EMC mitigation
Change involved a voltage regulator or any associated circuitry such as
capacitors, resistors, indictors or active components such as diodes, transistors
or supply voltage watchdogs.
In addition consider repeating CI260 if there has been software change that results in a
different operating loop time or influences power supply or reset management.
6.3. Electrostatic Discharge
Repeat CI280 A if the change involved any modification to parts or PCB associated or
within close proximity (within 25mm) of external connector pins.
Repeat CI280B and/or CI280C tests if there has been a change to module packaging such
as changing case material type, addition of metallic labels, and changing aperture sizes.
It is also necessary to consider the impact of PCB changes if the PCB is part of the man
machine interface such as boards that contain displays or push buttons.
Any modifications to module's software which influence the core operating routine, power supply
management, communications, input output management or similar parts which can reasonable be
expected to result in different radiated emissions profile.
Any modifications to printed circuit board (PCB), which result in a change of the substrate material or
on the copper area resulting from actions such as moving components, rerouting tracks, redefining
earth fills, movement, deletion or addition of vias (layer to layer interconnections).
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6.4. Conducted Emissions:
6.4.1
CE420 Frequency domain
Repeat if RE310 is run and the results are different ((+/- 3 dB in amplitude and +/- 1
MHz in frequency) in the band of 0.15 – 108 MHz,
or
If the change involves any parts that are used for controlling conducted emissions
such as suppression components,
or
Any other change that can reasonably be expected to influence conducted emissions
profile.
6.4.2
CE410 Time Domain
Repeat if conditions for Conducted Immunity listed above are met.
Table 6–1. Analysis of EMC Testing
Radiated
Immunity
Conducted
Immunity
Test
RI 110
RI 120
RI 130
RI 140
CI 210
CI 220
CI 230
CI 240
CI 250
CI 260
CI 270
CI 280 (A)
CI 280 (B)
CI 280 (C)
Radiated
Emissions
Conducted
Emissions
Reasons for NOT testing
RE 310
CE 410
CE 420
Note: Use this table to record your own analysis
Frame 71 of 78
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Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
PART VII: CHECKOFF LIST
7. CHECKOFF LIST – EMC DESIGN GUIDE FOR PCB(S)
The following list shall be completed and submitted to the approving activity, together with
the EMC test plan, at least 60 days prior to commencement of component level EMC testing.
7.1.
General description
ESC Name:
Manufacturer:
Ford P/N(s):
Model Year:
7.2.
7.3.
7.4.
PWB #:
Vehicle Application:
Physical segregation of circuits has been employed right from the beginning of PCB
design
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
All relevant components/tracks have been contained within their designated PCB area
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
Which type of Power distribution technique best describes the PCB?
c
d
e
f
g
N/A
c
d
e
f
g
Minimal (daisy chain) distribution
c
d
e
f
g
Single-point (star) distribution
c
d
e
f
g
Power plane
c
d
e
f
g
Other, explain: __________________________________________
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Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
7.5.
7.6.
7.7.
Which type of Ground distribution technique best describes the PCB?
c
d
e
f
g
Minimal (daisy chain)
c
d
e
f
g
Single-Point (star)
c
d
e
f
g
Multi-point
c
d
e
f
g
Hybrid
c
d
e
f
g
Ground grid
c
d
e
f
g
Ground plane
c
d
e
f
g
Other, explain: __________________________________________
List all Power (voltage) levels present on PCB, and specify which ground they reference
Power Level
(Volts)
Reference Ground
(Example: 3.3 VDC)
(Example: Logic ground)
List PCB layer stack-up and content, indicating the location of all crystal(s)/resonator(s)
Layer number
Layer content
Layer 1
Layer 2
Layer 3
Layer 4
Layer 5
Layer 6
Frame 73 of 78
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Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
7.8.
7.9.
7.10.
All possible surface areas of the PCB have been filled with ground
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
All ground segments have been tied together with multiple vias and/or with many short
thick tracks, and there is no 'floating' copper of any kind on the PCB
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
List names and provide actual track lengths as well as rise/fall time of all periodic clock
signals (e.g. ECLK, MCLK, ALE, CLKOUT, etc.)
Signal name
7.11.
Signal rise
time
(tr)
Signal length
(mm)
Signal fall
time
(tf)
All clock signals have adjacent ground-return tracks
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
Frame 74 of 78
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Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
7.12.
7.13.
7.14.
All unused connector pins have been terminated to PCB ground
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
The PCB contains the following communications protocols:
c
d
e
f
g
N/A
c
d
e
f
g
ACP
c
d
e
f
g
CAN
c
d
e
f
g
ISO-9141
c
d
e
f
g
SCP
c
d
e
f
g
UBP
c
d
e
f
g
Other, explain: __________________________________________
Discrete components supporting the individual communication protocols, also known as
physical layers, have been placed and routed according to their unique EMC design
requirements as specified by Ford -- list the applicable specification used*
c
d
e
f
g
N/A
c
d
e
f
g
ACP . . . . . . . Spec.#: ___________________________________
c
d
e
f
g
CAN . . . . . . Spec.#: ___________________________________
c
d
e
f
g
ISO-9141 . . . Spec.#: ___________________________________
c
d
e
f
g
SCP . . . . . . . Spec.#: ___________________________________
c
d
e
f
g
UBP . . . . . . . Spec.#: ___________________________________
c
d
e
f
g
Other, explain: __________________________________________
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Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
7.15.
7.16.
Potential radiated emissions from CPU(s) have been considered prior to PCB layout
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
List and classify each CPU according to its highest radiated emissions level**
CPU Name
7.17.
7.18.
IC RE Reference Level
Has the PCB been grounded to its enclosure (housing)? Specify type of connection
c
d
e
f
g
N/A (non-metallic housing)
c
d
e
f
g
Yes =>
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
c
d
e
f
g
DC connection
c
d
e
f
g
AC connection
Unused clock outputs from CPU(s) have been addressed to minimize radiated emissions
(i.e. proper termination, disabled in software)
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
Frame 76 of 78
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Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
7.19.
7.20.
EMC support has been available throughout the PCB layout phase
c
d
e
f
g
N/A
c
d
e
f
g
Yes
c
d
e
f
g
No
c
d
e
f
g
If No, explain: __________________________________________
Please provide the name, company, job title, and contact information of the person(s)
responsible for EMC signoff of this product
Name:
Company:
Job Title:
Phone #:
E-mail:
Note:
*
Recommended EMC design guidelines for each physical layer are available form Ford's
external web site at – http://www_eese.ford.com/mux.
** For IC RE reference levels refer to Ford's General IC Specification Limits for Radiated
Emissions, part number ES-3U5T-1B257-AA, 10/01/2002 or latest revision.
Frame 77 of 78
Rev. A
10/01/2002
Printed copies are uncontrolled
Engineering Specification
ES-3U5T-1B257-AA
EMC Design Guide for Printed Circuit Boards
Collected References:
XW7T-1A278-AB, Component Specification Electromagnetic Compatibility, Ford Motor
Company, April 1999
C.R. Paul, Introduction to Electromagnetic Compatibility, John Wiley Interscience, 1992
A.R. Macko, Electromagnetic Compatibility and Electromagnetic Interference Control in the
Automotive Electrical Environment, Ford/Visteon/EMCARM Co., 1995
A.R. Macko, A. Nielsen, P. Bator, Electromagnetic Compatibility for Printed Circuits
Boards, Ford Motor Company/Visteon, December, 1994
A. Gunsaya, Management of Change for EMC, Ford Motor Company, March 2002
H.W. Ott, Noise Reduction Techniques in Electronic Systems, Second Edition, John Wiley
Interscience, 1998
Howard W. Johnson, Martin Graham, High-Speed Digital Design, Prentice Hall, 1993
Michel Mardiguian, Interference Control in Computers and Microprocessor-Based
Equipment, First Edition, Don White Consultants, 1987
Mike Catherwood, Designing for Electromagnetic Compatibility (EMC) with CMOS
Microcontrollers, Document Ref. No. AN1050, Motorola, Inc.
Imad Kobeissi, Noise Reduction Techniques for Microcontroller-Based Systems, Document
Ref. No. AN1705, Motorola, Inc.
Hewlett-Packard, Designing for Electromagnetic Compatibility, Student Workbook, Course
No. HP 11949A, Hewlett-Packard Company, 1989
Darryl Lindsey, The Design & Drafting of Printed Circuits, Revised Edition, Bishop
Graphics, Inc. 1984
Keith Armstrong, PCB Design Techniques for Lowest-Cost EMC Compliance, Part 1,
Electronics and Communication Engineering Journal, August 1999
Jean-Claude Kedzia, Giuseppe Guglielmetti, Stefan Dickmann, Improving EMC Compliance
of Electronic Devices through Numerical Simulation, EC project, BRPR-CT97-0592
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