NEWS BRIEFS IN-DEPTH ARTICLES

NEWS BRIEFS IN-DEPTH ARTICLES
Volume Forty-Four
NEWS BRIEFS
2
IN-DEPTH ARTICLES
DESIGN SHOWCASE
Reference voltage for multiple ADCs
3
Designing compact telecom power supplies
6
Small, high-voltage boost converters
15
Compact, inductorless boost circuit regulates white-LED bias current
22
Analog switch lowers relay power consumption
23
LOW-NOISE REFERENCE CIRCUIT
3V
3V
0.1µF
0.1µF
2.048V
N.C.
29
31
1
32
2
16.2kΩ
3
IC1
1µF
3
10Hz LOWPASS
FILTER
4 -
0.1µF
5
+
IC2
2
1
162Ω
2
1
REFOUT
REFIN
REFP
REFN
IC3
ADC 1
COM
100µF
0.1µF 0.1µF 0.1µF
10Hz LOWPASS
FILTER
0.1µF
NOTE: ONE FRONT-END REFERENCE CIRCUIT DESIGN
MAY BE USED WITH UP TO 1000 ADCs.
N.C.
29
31
32
1
0.1µF
2
REFOUT
REFIN
REFP
REFN
COM
IC4
ADC N
0.1µF 0.1µF 0.1µF
For ultrasound applications, a single, low-noise reference circuit drives up to 1000 ADCs. (See article inside, page 3.)
2.2µF
10V
News Briefs
MAXIM REPORTS REVENUES AND EARNINGS
FOR THE SECOND QUARTER OF FISCAL 2002
Maxim Integrated Products, Inc., (Nasdaq: MXIM) reported net revenues of $247.1 million for its fiscal second
quarter ending December 29, 2001, down from the $438.3 million reported for the second quarter of fiscal 2001 and up from
the $239.4 million reported for the first quarter of fiscal 2002. Net income for the quarter was $62.6 million, a decrease from
the $122.2 million reported last year and an increase over the $61.3 million reported for the previous quarter. Diluted
earnings per share were $0.18 for the second quarter, down from the $0.34 reported for the same period a year ago and up
from the $0.17 reported for the first quarter of fiscal 2002.
During the quarter, cash and short-term investments increased $22.1 million after the Company repurchased
2.0 million shares of its common stock for $67.8 million and acquired a total of $18.7 million of capital equipment. The
stock repurchase was enabled by temporary suspension by the SEC of regulations relating to stock repurchases after
pooling-of-interests transactions. Accounts receivable decreased by $4.7 million in the second quarter to $94.8 million, and
inventories decreased $4.9 million to $154.4 million.
Gross margin for the second quarter increased slightly to 70.1%, after increasing inventory reserves $3.0 million,
compared to 70.0% reported for the first quarter. Research and development expense increased from the $66.0 million
reported in the first quarter or 27.6% of net revenues to $68.6 million or 27.7% of net revenues, primarily the result of hiring
additional engineers and investing in new product development. Selling, general and administrative expenses decreased from
$25.1 million in the first quarter to $22.9 million in the second quarter, primarily due to control of discretionary spending.
Second quarter bookings were approximately $230 million, a 9% increase over the first quarter’s level of $211 million.
Turns orders received during the quarter were $125 million, a 23% increase over the $102 million received in the prior quarter
(turns orders are customer orders that are for delivery within the same quarter and may result in revenue within the same
quarter if the Company has available inventory that matches those orders). Order cancellations were down 39% from the first
quarter and down 81% from the year ago level. Bookings increased in all geographic regions except the U.S., where bookings
decreased slightly compared to the prior quarter. Bookings increased in most of the Company’s 14 business units, with
particular strength in those business units with products for portable equipment. Bookings were down from the first quarter for
those business units with products for the test equipment, fiber, and telecommunication equipment markets.
Second quarter ending backlog shippable within the next 12 months was approximately $187 million, including
$170 million requested for shipment in the third quarter of fiscal 2002.
Jack Gifford, Chairman, President, and Chief Executive Officer, commented on the quarter: “We are encouraged by
the continued increase in bookings during our second quarter. Although visibility remains limited, we currently expect
bookings to increase again in our third quarter.”
Mr. Gifford continued: “We are confident that inventory levels for much of the end equipment that our products address
are either significantly reduced or have moderated. We remain cautious about the short-term outlook for the telecom and ATE
markets, where our customers are still working through inventories. Although we believe that our customers’ visibility is better
now than it has been in the past several quarters, turns orders remain high because of our short lead times. As lead times
increase, we expect customers to place orders beyond their immediate needs.”
Mr. Gifford concluded: “Unlike some companies in our industry, Maxim has continued to increase its engineering
headcount and its investment in research and development activities. In our second quarter, R&D expenses were nearly 28% of
net revenues, a historically high level for the Company and a remarkably high level for any mature company in our industry.”
Certain statements in this press release are forward-looking statements within the meaning of the Private Securities
Litigation Reform Act of 1995. These statements involve risk and uncertainty. All forward-looking statements included in this
news release are made as of the date hereof, based on the information available to the Company as of the date hereof, and the
Company assumes no obligation to update any forward-looking statement.
Maxim Integrated Products is a leading international supplier of quality analog and mixed-signal products for applications
that require real world signal processing.
ADC accuracy
Reference voltage
for multiple ADCs
For many applications, gain and noise level have a major
effect on ADC accuracy. The gain of an ADC is
represented by the slope of its transfer function, which
relates analog inputs to the allowable range of digital
output codes. One way to quantify gain is to measure the
full-scale (FS) input range, which is directly controlled by
the reference-voltage level. For medical ultrasound
imaging systems, variation in the full-scale range of the
ADCs can cause errors in beam formation. It also varies
the ADCs’ clipping point—an effect that may be important
in certain signal-demodulation schemes.
Ultrasound-imaging systems usually host a large array of
analog-to-digital converters (ADCs). Not only do these
systems require precise channel-to-channel matching, but
also a careful design of the voltage reference system in
order to maintain sufficient dynamic performance.
These reference systems typically provide a precise,
low-noise reference to a large bank of converters. To
further ensure accuracy and stability, the ADCs’ reference
inputs should be buffered to minimize the loading effects
on the reference system. In response to these requirements,
high-speed ADC manufacturers have begun supplying
converters with reference inputs that can support the use of
both the internal precision reference and an external,
precision, reference-voltage source.
An ADC’s noise level determines its useable dynamic
range; this dynamic range should generally be as large as
possible. The reference-noise component of ADC noise
can be additive or multiplicative. Additive noise is easily
filtered by local bypass capacitors on the individual ADCs,
which in most designs are already present to optimize the
ADC’s dynamic performance.
Multiplicative noise, on the other hand, is more insidious.
For ultrasound applications, reference noise in the audio
frequency spectrum can modulate large “stationary”
signals in the RF spectrum. Such signals are produced by
stationary tissue in the ultrasound target. Audio modulation produces sidebands on the RF signal that can be
demodulated by a Doppler detector, producing audio tones
in the detected Doppler output signal.
Medical ultrasound-imaging systems, for example,
commonly use a large number of ADCs in the receiver’s
beamforming electronics, usually organized in groups of
16, 24, 32, etc. Maximum beam accuracy requires the
absolute minimum beamforming path errors. A major
source of these errors can be traced to poor accuracy in
terms of noise and the actual reference voltage present at
each of the individual ADCs. Other error sources include
variances in the distributed load seen by the reference’s
source. In any particular system architecture, this load can
consist of the many individual resistive and capacitive
loads seen by the reference. Several approaches can
provide the reference voltage for such ADC arrays:
To estimate the amount of audio noise tolerable in an ultrasound application, assume a nearly full-scale RF signal
applied to a 10-bit ADC like the MAX1448. The dynamic
range of that device (almost 60dB) implies a noise floor of
-60dBFS. That noise level can be normalized to a 1Hz
bandwidth. The Nyquist bandwidth for an 80MHz
sampling rate is 40MHz. The correction factor is √40MHz
= 76dB, which places the ADC’s noise floor at -60dBFS 76dBFS = -136dBFS. Because a conservative design
requires the reference-voltage noise to be at least 20dB
lower (-156dBFS), a 2.0V reference requires an extremely
low noise level of 33nVP-P (approximately 8nVRMS/√Hz).
• Individual on-chip references. Though it offers a
convenient connection locally to each ADC, this option
features relatively poor matching between the converters.
• A single, external reference voltage applied to all
reference inputs of the ADC array. Such a configuration allows the user to engineer an external reference
voltage of arbitrary accuracy, but incurs error due to
small variations among the resistor ladders (one ladder
internal to each ADC).
A multi-ADC array may require a reference voltage more
accurate than the one internal to each converter (the
reference internal to MAX144x converters, for example, is
±1% accurate). The following two circuits are submitted as
reference designs for such arrays. They feature a single,
common low-frequency noise filter, and they achieve
high-frequency noise suppression with local decoupling
capacitors on individual ADCs.
• An external reference driving the ADCs’ referenceladder taps directly. This option delivers maximum gain
accuracy by directly controlling the reference voltage
applied to each ADC ladder. However, it requires driving
the (relatively) low resistance of the ladders, and some
ADCs do not allow access to that internal bias point.
3
building blocks in the signal path of an ultrasound receiver.
Note that proper power-up/down sequencing is ensured
because all active parts are driven from the same supply
voltage. That approach yields excellent gain matching and a
very low noise level with minimal circuitry and should
suffice in many applications that require multiple gainmatched ADCs.
Single external reference
Multiple-converter systems based on the MAX144x (IC3,
IC4) family are well-suited for use with a common reference
voltage. The REFIN pin of those converters can be connected
directly to an external reference source, eliminating the need
for any circuit modification. Moreover, the high input
impedance of REFIN (even of multiple REFIN terminals
connected in parallel) draws only a small load current.
Generating a precision external reference
A precision source like the MAX6062 (IC1) generates an
external DC level of 2.048V (Figure 1), and exhibits a noisevoltage density of 150nV/√Hz. Its output passes through a 1pole lowpass filter (with 10Hz cutoff frequency) to an op
amp (IC2) like the MAX4250, which buffers the reference
before its output is applied to a second 10Hz lowpass filter.
IC2 provides a low offset voltage (for high-gain accuracy)
and a low noise level. The passive 10Hz filter following the
buffer attenuates noise produced in the voltage-reference and
buffer stages. This filtered-noise density, which decreases for
higher frequencies, meets the noise levels specified for
precision ADC operation.
For applications requiring more stringent gain matching
(Figure 2), the MAX144x family again is well suited.
Connecting each REFIN to analog ground disables the
internal reference of each device, allowing the internal
reference ladders to be driven directly by a set of external
reference sources. These voltages can have an arbitrarily tight
tolerance, and the ADCs track them within 0.1% (typ).
ADCs of this family also feature a 4kΩ resistance across the
ladder reference connection, which allows the load to be
easily driven even with many ADCs in parallel.
A DC level of 2.500V can be generated by a precision source
(IC1) such as the MAX6066, followed by a 10Hz lowpass
filter and precision voltage divider. The buffered outputs of
this divider are set to 2.0V, 1.5V, and 1.0V, with an accuracy
that depends on the tolerance of the divider resistors.
Converters of the MAX144x family are specified for a
typical gain error of ±4.4% (better than ±0.5dB). This
performance is better than the gain tolerance of all other
3V
3V
0.1µF
0.1µF
2.048V
N.C.
29
31
1
32
2
16.2kΩ
3
IC1
1µF
3
10Hz LOWPASS
FILTER
4 -
0.1µF
5
+
IC2
2
1
162Ω
2
1
REFOUT
REFIN
REFP
REFN
IC3
ADC 1
COM
100µF
0.1µF 0.1µF 0.1µF
10Hz LOWPASS
FILTER
0.1µF
NOTE: ONE FRONT-END REFERENCE CIRCUIT DESIGN
MAY BE USED WITH UP TO 1000 ADCs.
N.C.
29
31
32
1
0.1µF
2
0.1µF 0.1µF 0.1µF
Figure 1. For ultrasound applications, a single, low-noise reference circuit drives up to 1000 ADCs.
4
REFOUT
REFIN
REFP
REFN
COM
IC4
ADC N
2.2µF
10V
Those three voltages are buffered by the quad op amp IC2,
which is selected for its low noise and DC offset. The
individual voltage followers are connected to 10Hz lowpass
filters which filter both the reference-voltage noise and
buffer-amplifier noise to a level of 3nV/√Hz. The 2.0V and
1.0V reference voltages set the differential full-scale range of
the associated ADCs at 2VP-P. The 2.0V and 1.0V buffers
drive the ADCs’ internal-ladder resistances between them:
4kΩ divided by the number of ADCs in the circuit. As an
example, 32 ADCs will draw 8mA from those supplies—a
load current well within the capability of IC2.
that require more than 32 matched ADCs, a voltage
reference and divider string common to all converters are
highly recommended.
Conclusion
Systems requiring large numbers of data converters with
good channel-to-channel matching require careful design of
the voltage-reference system. The use of a common, highprecision, low-noise reference driving all the ADCs is a
valuable approach for achieving high-accuracy matching.
The flexible reference inputs and exceptional dynamic
performance of 10-bit ADCs in the MAX144x family make
them compelling candidates for such applications.
This configuration’s gain accuracy can be good, depending
on the accuracy grade of IC1 and the tolerance of resistors in
the voltage divider. The gain matching of each ADC in this
configuration is typically 0.1%. With the noise level below
3nV/√Hz at 100Hz, this circuit provides exemplary performance. As in Figure 1, the common power supply for all
active components removes any concern about power-supply
sequencing when powering up or down.
(A similar idea appeared in the 1/24/02 issue of EDN.)
References:
1. Maxim MAX1444 data sheet, Rev. 0, 10/00.
2. Maxim MAX1448EVKIT data sheet, Rev. 1, 12/01.
3. Maxim MAX4249–MAX4252 data sheet, Rev. 4, 1/02.
4. Maxim MAX6061–MAX6068 data sheet, Rev. 1, 5/01.
With the outputs of the op amp buffers matching better than
0.1%, the buffers and subsequent lowpass filters can be
replicated to support as many as 32 ADCs. For applications
3V
0.1µF
N.C.
29
31
1
2 21.5k Ω
3
IC1
2
3
+
-
4
IC2A
11
1
1.5V
5
0.1µF
+
21.5kΩ
21.5kΩ
9
2
0.1µF
REFP
REFN
COM
IC3
ADC 1
0.1µF 0.1µF
1.5V AT 0
4
7
-
4
IC2C
11
47Ω
10µF
6V
1.47kΩ
21.5kΩ
1.0V
10 +
MAX4254 POWER-SUPPLY BYPASSING.
PLACE CAPACITOR AS CLOSE AS
POSSIBLE TO THE OP AMP.
1
REFIN
3V
IC2B
6
11
3V
2.0V AT 8mA
47Ω
10µF
6V
330µF
1.47kΩ 6V
21.5kΩ
1µF
32
3V
2V
REFOUT
3V
8
330µF
6V
0.1µF
1.0V AT -8mA
47Ω
10µF
6V
330µF
6V
N.C.
1.47kΩ
29
31
REFOUT
REFIN
32
REFP
1
REFN
2
COM
NOTE: ONE FRONT-END REFERENCE CIRCUIT DESIGN MAY BE USED WITH UP TO 32 ADCs.
0.1µF 0.1µF 0.1µF
Figure 2. Also for ultrasound applications, a precision, low-noise reference circuit drives up to 32 ADCs.
5
IC4
ADC 2
2.2µF
10V
Designing compact
telecom power
supplies
telecom power supplies. It enables the design of small,
efficient, power-converter circuits. A fixed-switching
frequency of 262kHz controls switching losses while
allowing moderately small power components. The IC
includes undervoltage lockout capability with large
hysteresis and a low startup current. This results in lowloss designs for power supplies that feature a wide inputvoltage range and low output power. Cycle-by-cycle
current limiting (achieved with a fast internal
comparator) reduces overdesign in the MOSFET and
transformer. Other features include maximum-dutycycle limiting and high-peak capability for the
source and sink-drive currents. A reference design
(Figure 1) illustrates the 5W flyback converter with an
input-voltage range of 36V to 72V.
Telecom power supplies are specified for operation over
a wide input-voltage range (36V to 75V) but with circuit
performance optimized at 48V. Such circuit designs
should be compact, efficient, and have a low profile to
comply with the tight spacing between cards. This
article discusses a 5W flyback converter for telecom
applications, based on the MAX5021—a universal
offline power-supply controller.
Telecom systems include numerous line cards. Connected
in parallel to the high-power backplane, each has its own
input-filter capacitor and low-voltage power converter.
The large number of input-filter capacitors in parallel
limits the value of each to a few microfarads, making the
power-supply design fairly difficult.
Power-stage design
The first step in designing a power supply is deciding on
a conversion topology. Topology selection criteria
include the input-voltage range, output voltage, peak
currents in the primary and secondary circuits,
efficiency, form factor, and cost.
The MAX5021 IC is a high-frequency, current-mode
PWM controller suitable for wide-input-range, isolated
48VIN
EPC13-CTX03-15498 COOPER
1
T1
J1
R11
7.5kΩ
C1
1µF,
100V
C2
1µF,
100V
D2 MBRS340
1
C10
560pF
R1
82Ω
J3
C5
330µF,
10V
NS_A
D3
BAS21LT1
J2
GND
L1
2kΩ AT 100MHz
D1
C4
0.22µF
5
6
VCC
U1
NDRV
MAX5021
VIN
OPTO
1
Q1
IRF5802
R10
51Ω
2
R5
249kΩ
COM
C7
10nF
7
5
3
J4
R8
510Ω
MOC207 U3 1
6
C8
0.22µF
4
C9
10µF
C11
2.2µF, 250VAC
MMBD914
1
5V, 1A
+VOUT
L2
120Ω AT 100MHz
C6
220µF
3
R12
47kΩ
4
TLV431AC 5
GND 2
CS
1
R9
1.3Ω
C3
1nF
U2
R7
8.06kΩ
R3
100Ω
R4
1.3Ω
Figure 1. Based on the MAX5021 PWM controller, this flyback converter for telecom applications delivers 5W at 5V.
6
R6
24.9kΩ
• Calculate the AL value of the core.
• Calculate the RMS current in the primary, and
estimate the secondary RMS current.
• Consider the proper winding sequence and transformer
construction for low leakage.
1) Use the following equation to estimate the minimum
area product required:
The best choice for a 5W output with a 1:2 input-voltage
range and small form factor is a flyback topology whose
minimum component count reduces the cost and form
factor. The flyback transformer can be designed to
operate either in continuous or discontinuous mode.
Discontinuous mode causes the transformer core to
complete its energy transfer during the off cycle, and
continuous mode allows the next cycle to begin before
the energy transfer is complete. In the present case,
discontinuous mode is chosen for the following reasons:
it maximizes energy storage in the magnetic component
(thereby reducing the component’s size); it simplifies
compensation (no right-half-plane zero); and it yields a
higher unity-gain bandwidth.
AP ≥
(1.1 × POUT × DMAX)
...( m 4 )
η × KP × KU × J × KT × BMAX × ƒ SW ( MIN)
AP ≥
(2 × POUT ) × 10 −12
...( m 4 )
η × BMAX
where:
η = expected efficiency of the converter;
KP = area assigned to the primary (usually 0.5);
KT = ratio of RMS to average current in the primary (0.55
to 0.65 for discontinuous flyback);
KU = window utilization factor (0.4 to 0.5);
J = current density (9.862x106A/m2 for winding temperature
rise less than 40°C); and
BMAX = maximum operating flux density in Teslas (use
between 0.12T to 0.15T).
A disadvantage of the discontinuous operating mode is
the higher ratio of peak-to-average current in the primary
and secondary circuits. A higher ratio means higher RMS
current, which leads to higher loss and lower efficiency.
For low-power conversion, the advantages of discontinuous mode easily surpass the disadvantages. Moreover,
the IC’s drive capability is sufficient for driving the large
switching MOSFET necessary to carry these peak
currents. A telecom application using the MAX5021 in
this topology easily achieves power outputs to 15W using
standard MOSFETs.
Select a core with area product (AP) equal to or greater
than the figure calculated above, and note its core cross
section area. Refer to the following table for output power
vs. core size, AP, and core cross-section area (Ae):
Transformer design
Output Power vs. Core Size
The key to low loss and high efficiency in the transformer
is a proper core. The core and the winding-area product
determine the amount of power the transformer can
handle with an acceptable rise in temperature. Also
considered in the core selection are the topology (ratio of
average to RMS current in the winding), output current,
efficiency, and form factor. The design of a discontinuous-mode transformer is explained below, step by step.
Note that the first equation is a general one, and the
second is specific to the MAX5021 power supply with a
40°C temperature rise.
Output
Power (W)
Core Size
Ap (mm4)
Ae (mm2)
Up to 2
EPC-10
30
9.4
3 to 4
EEM-12.7
90
12
5 to 8
EPC-13
145
12.5
9 to 12
EPD-15
216
13.5
(Refer to the Appendix on page 13 for an example.)
2) As discussed earlier, discontinuous operation
requires that the core be discharged during the
off-cycle. Secondary inductance determines the time
required to discharge the core. Use the following
equations to calculate secondary inductance:
• Estimate the minimum area-product requirement, and
select a core and bobbin with suitable form factor.
• Calculate the secondary-winding inductance for
guaranteed core discharge within the minimum
off-time.
• Calculate the primary-winding inductance for sufficient
energy to support maximum load.
• Calculate the number of turns in the primary.
• Calculate the number of turns in the secondary and
bias windings.
7
LS ≤
(VO + VD ) × (DOFF( MIN) )2
...(H)
2 × IOUT × ƒSW (MAX )
LS ≤
430 × 10 −9 × (VO + VD )
...(H)
IOUT
where:
VD = secondary-diode forward-voltage drop in volts.
IOUT = maximum-rated output current in amperes.
AL =
7) The transformer manufacturer must know the RMS
currents in the primary, secondary, and bias windings
to decide the thickness of the wire. To keep skin-effect
loss under control, only wires thinner than 28AWG are
recommended. Multiple wires in parallel can be used
to achieve the required copper thickness. Multifilar
windings are very common in high-frequency
converters. Maximum RMS current in the primary and
secondary windings occurs at 50% duty cycles
(minimum input voltage) and maximum output power.
Use the following equations to calculate primary and
secondary RMS currents:
POUT
DMAX
IPRMS =
×
...( A )
0.5 × DMAX × η × VIN ( MIN )
3
3) Rising current in the primary builds energy in the core
during the on-cycle, which is then released to deliver
output power during the off-cycle. The primary inductance must hold enough energy during the on-time to
support the maximum output power.
LP =
LP =
VIN ( MIN ) 2 × DMAX 2 × η
2 × POUT × ƒS( MAX )
0.4 × 10 −6 × VIN ( MIN )2 × η
...(H)
POUT
4) Next, calculate the primary number of turns necessary
to keep the maximum flux density within limits at the
maximum V-s product across the primary. The
maximum-operating peak current occurs at the
maximum duty cycle.
NP =
NP =
IPRMS =
VIN ( MIN ) × DMAX
Ae × BMAX × ƒS( MIN )
ISRMS =
2.1 × 10 −6 × VIN ( MIN )
Ae × BMAX
1.63 × POUT
...( A )
η × VIN ( MIN )
IOUT
DOFF ( MAX )
...( A )
0.5 × DOFF ( MAX )
3
ISRMS = 1.63 × IOUT.....( A )
Bias current is usually less than 10mA, so the selection
of wire thickness depends more on the convenience of
winding the wire than its current capacity.
where:
Ae = core cross section area in square meters.
5) Round off the primary number of turns to the closest
integer, and calculate the number of turns for the
secondary and bias windings using the rounded-off
primary number of turns. Refer to the following
equation:
NS = NP ×
nH
LP
× 10 9...
2
2
NP
8) The winding technique and sequencing is important in
achieving a lower leakage-inductance spike at switch
turn-off. As an example, interleave the secondary
between two primary halves and keep the bias winding
close to the secondary, so the bias voltage follows the
output voltage.
LS
LP
MOSFET selection
11.7
NBIAS =
VOUT + 0.2
Selection criteria for the MOSFET include maximum
drain voltage, peak/RMS current in the primary, and
maximum-allowable power dissipation for the package
(without exceeding the junction temperature limits).
Voltage at the MOSFET drain is the sum of the input
voltage, the secondary voltage reflected through the transformer turns ratio, and the leakage-inductance spike.
(Figure 2 illustrates the relationship between drain
voltage and primary current.) The MOSFET’s absolute
maximum VDS rating must be higher than the worst-case
drain voltage (maximum input voltage and output load).
The forward-bias drops of secondary- and biasrectifier diodes are assumed to be 0.2V and 0.7V,
respectively. Refer to the diode manufacturer’s data
sheet to verify these numbers. Again, round off the
number of turns for secondary and bias windings to
the closest integers.
6) The core’s AL value depends on the air gap in the
magnetic path length. Most of the energy is stored in
the air gap during the MOSFET’s on time. To reduce
electromagnetic radiation, insert the air gap in the
center leg of the core.
VDS (MAX ) = VIN ( MAX ) +
8
NP
× (VOUT + VD ) + VSPIKE ...(V)
NS
RCD snubber-network design
To avoid an excessive VDS requirement for the MOSFET,
we recommend using an RCD snubber across the primary
to suppress the spike caused by energy in the leakage
inductance. The snubber dissipates energy that would
otherwise dissipate in the MOSFET itself. The snubber
capacitor’s value should be high enough to absorb the
leakage-inductance energy without allowing the MOSFET
drain voltage to rise beyond an acceptable limit. Use the
following equation to calculate this capacitance:
DRAIN
VOLTAGE
50V/div
PRIMARY
CURRENT
0.65A/div
C=
where:
LL = leakage inductance, which should be specified by
the transformer vendor. (Values of 1µH to 3µH are
common for the transformer under discussion.) VSPIKE =
spike voltage, typically 30V to 50V. IPK = peak primary
current, which in this case (for a worst-case spike) equals
the current-limit threshold divided by RSENSE.
1µs/div
Figure 2. This scope photo shows Figure 1’s circuit operating at
VIN = 36V, VOUT = 5V, and IOUT = 1A. The switching
MOSFET (Q1) exhibits drain voltage (upper trace) at
50V/div, and primary current (lower trace) at 0.65A/div.
The diode must be a fast-switching type, with reverseblocking voltage at least equal to the VDS(MAX) rating of
the MOSFET. The resistor is selected for an RC time
constant 2 to 3 times the switching period. Power dissipation in the resistor is the sum of the leakage-inductance
energy times frequency, plus the power loss caused by
DC bias across the capacitor. The following equation lets
you estimate power dissipation in the resistor:
1
PR = ( × CSNUBBER × VSPIKE 2 ×) ƒS ( MAX)
2
N
[(VO + VD) × P ]2 × (1 − DMIN )
NS
+
RSNUBBER
where:
D MIN = minimum duty cycle = D MAX /2. (A 50%
derating is recommended for chip resistors.)
A lower VDS Absolute Maximum Rating means a shorter
channel, lower RDS(ON), lower gate charge, and a smaller
package. Thus, it is advisable to keep VDS(MAX) low by
choosing a lower NP/NS ratio and keeping the leakageinductance spike under control. A resistor/capacitor/diode
(RCD) snubber network can suppress such spikes.
The RMS current in the primary can be used to calculate
DC loss in the MOSFET. Switching loss in the MOSFET
depends on the operating frequency, the total gate
charge, and the cross-conduction loss during turn-off.
Cross-conduction loss during turn-on is negligible,
because in discontinuous-conduction mode the primary
current starts from zero. To avoid damage during poweron and during fault conditions, it may be necessary to
derate the MOSFET. Use the following equation to
estimate the MOSFET’s power dissipation:
Input filter design
The input filter reduces the amplitude of AC components
in the converter’s current pulses, thereby making the
converter appear to the source as a DC load. Design
parameters for this filter are the RMS ripple-current
capability, input voltage, and the allowable level for AC
components reflected back to the source.
PMOS = (1.4 × RDS (ON ) × IPRMS) +
(Q G × VCC × ƒS (MAX ) ) +
+
(VIN ( MAX ) × IPK × toff × ƒS (MAX ) )
4
(CDS × VDS 2 × ƒS (MAX ) )
2
L L × IPK 2
...( F )
VSPIKE 2
...( W )
Because discontinuous-mode flyback converters draw
peak triangular currents through the capacitor ESR
during each cycle, large aluminum electrolytic capacitors
are needed for their low ESR and high ripple-current
ratings. Unfortunately for a distributed power system, the
input-filter capacitances of parallel converters add
where:
QG = total gate charge of the MOSFET in coulombs;
VCC = bias voltage in volts;
tOFF = turn-off time in seconds;
CDS = drain-to-source capacitance in farads.
9
together and may produce an unacceptable inrush current
at startup. As an alternative, you can use ceramic capacitors to achieve low ESR and a high ripple-current rating
while keeping the total capacitance low.
secondary replenishes those lost charges by
discharging the core during the off-cycle, and simultaneously supplies the load current. Again, output ripple
is the sum of the voltage drop due to the output
capacitor’s ESR (∆VESR), and the charge loss (∆VC)
during the switch on-time. High switching frequency in
the MAX5021 reduces the capacitance requirement.
Use low-ESR tantalum capacitors for their favorable
combination of capacitance and ESR, and use the
following equations to calculate the capacitance and
ESR:
4 × (1 − DOFF ) × IO × 10 −6
C OUT =
...( F )
∆VC
And,
∆VESR
ESR O =
...(Ω)
IO
where:
DOFF is the discharge duty cycle, calculated using the
following equation:
The input peak-to-peak ripple voltage is a combination
of the voltage drop due to capacitor ESR (∆VESR) and
the loss of charge from the capacitor (∆VC). For lowESR ceramic capacitors, use a 3:1 contribution from
charge loss and ESR ripple respectively, and use the
following equation to estimate capacitance and ESR for
the capacitor:
4 × POUT × 10 −6
CIN =
...( F )
η × VIN ( MIN ) × ∆VC
And,
ESRIN =
∆VESR
....(Ω)
IPK
Choose a capacitor that can handle the necessary RMS
ripple without increasing its internal temperature
(Figure 3). Use the following equation to estimate RMS
ripple in the input capacitor:
ICRMS =
DOFF =
1.63 × POUT
...( A )
η × VIN ( MIN )
IO × LS
2 × 10 −6 × (VO + VD )
Additional noise spikes ride on the output ripple, caused
by the di/dt of secondary current flowing through the
output capacitor’s ESL. A small LC filter can suppress
these low-energy spikes, and it helps in attenuating
switching-frequency ripple as well. To minimize the
filter’s effect on phase loss and to ensure that it does not
interfere with compensation, you should design its corner
frequency more than one decade away from the estimated
closed-loop bandwidth. Figure 4 shows the peak-to-peak
ripple waveforms with and without the LC filter. Use a
low-ESR ceramic capacitor of 1µF to 10µF, and calculate
the inductance using the following equation:
Output filter design
The output capacitance required depends on the level
of peak-to-peak ripple acceptable at the load end. The
output capacitor for flyback converters supports load
current during the switch on-time. The transformer
L≤
DRAIN
VOLTAGE
50V/div
1
...(H)
4 × 10 3 × ƒC 2 × C
where:
fC = estimated closed-loop bandwidth.
Power loss consideration
INPUTVOLTAGE
RIPPLE
500mV/div
High-frequency switching converters can be very lossy,
since switching loss simply adds to the DC loss. Careful
component selection is necessary to keep switching loss at
a minimum. The MAX5021 is designed to operate at a
sufficiently high frequency to reduce the size of passive
components while minimizing switching losses. The
MAX5021’s low startup current and low quiescent
operating current minimize power loss in the control
circuitry. To reduce switching loss even further and
achieve higher converter efficiency, use a MOSFET with
1µs/div
Figure 3. This scope photo shows Figure 1’s circuit operating at
VIN = 72V, VOUT = 5V, and IOUT = 1A. The circuit exhibits
input-voltage ripple at 500mV/div (lower trace) and drain
voltage at 50V/div (upper trace).
10
tF = turn-off time in seconds;
VD = drain voltage at turn-off in volts;
fSW = switching frequency (262kHz);
IPK = primary peak current in amperes.
OUTPUT
RIPPLE
WITHOUT
LC FILTER
200mV/div
Use a Schottky diode in the secondary to achieve low
VDS and low reverse-recovery loss. Use the following
equation to calculate DC loss in the secondary diode
while neglecting reverse-recovery loss due to switching:
OUTPUT
RIPPLE
WITH
LC FILTER
100mV/div
PD = VFB × IO
where:
VFB = forward drop for the secondary diode at ISPK/2,
in volts.
2µs/div
You can reduce the transformer-leakage inductance
between primary and secondary by sandwiching the
secondary between two halves of the primary. Use a
multifilar winding structure to reduce skin-effect loss.
Figure 4. This scope photo shows Figure 1’s circuit operating at
VIN = 72V, VOUT = 5V, and IOUT = 1A. The circuit has
less output-voltage ripple with an LC filter (lower trace
at 100mV/div) than without (upper trace at 200mV/div).
Frequency compensation
The absence of a right-half-plane (RHP) zero in the
discontinuous flyback converter simplifies the closedloop frequency compensation to a single pole-zero pair.
No bandwidth limit arises from the location of the RHP
zero. The loop is closed through the shunt regulator,
optocoupler, and PWM comparator internal to the
MAX5021. The location of the error amplifier’s pole
and zero is determined from the existing PWM gain, the
output-filter pole, and the ESR zero frequency. Use the
following equations to calculate the current-mode
converter’s PWM gain, the output-capacitor filter pole,
and the output capacitor’s ESR zero:
low gate charge and low gate-to-drain capacitance, and
balance the MOSFET’s DC and switching-power losses.
See graph of Converter Efficiency vs. Output Current
(Figure 5), and use the following equation to calculate
DC and switching losses in the MOSFET:
PMOS = (1.4 R DS (ON ) × IPRMS)
+ (Q G × ƒSW × VCC ) +
( IPK × VD 6× tF × ƒSW )
where:
QG = total gate charge for the MOSFET in nanocoulombs;
VCC = voltage at VCC (pin 4 of MAX5021);
A PWM =
EFFICIENCY vs. OUTPUT CURRENT
RL × LP × ƒSW × η
6.2 × 10 3
2
×
× CTR
RS
RLED
85
48VIN
80
EFFICIENCY (%)
75
A PWM =
36VIN
70
5 × 61 × 10 −6 × 262 × 10 3 × 0.8
6.2 × 10 3
2
×
×1
0.65
510
65
72VIN
60
APWM = 105
55
50
Capacitor filter pole (fP):
45
40
0
0.2
0.4
0.6
0.8
1.0
ƒP =
1.2
IOUT (A)
ƒP =
Figure 5. This graph demonstrates Efficiency vs. Output Current curves
for the circuit in Figure 1.
1
...(Hz)
2 × π × RL × CO
1
...(Hz)
2 × π × 5 × 330 × 10 −6
ƒP = 96...(Hz)
11
response time, which may also cause an output
overshoot during turn-on. Figure 8 depicts the result of
an optimally compensated loop.
Capacitor ESR zero (fz):
1
ƒz =
...(Hz)
2 × π × CO × ESR
1
....(Hz)
2 × π × 330 × 10 −6 × 0.06
CLOSED-LOOP RESPONSE
ƒz = 8038....(Hz)
80
where:
RL = load resistance;
CTR = current-transfer ratio of optocoupler;
RS = current-sense resistor in the primary path; and
CO = output-filter capacitor.
180
135
60
90
PHASE
GAIN (dB)
40
45
0
20
-45
0
Total loop gain equals the PWM gain (APWM) times the
gain of the voltage divider and the error amplifier (shunt
regulator). The available worst-case phase margin (PM)
occurs at full load.
-90
GAIN
-20
PHASE (DEGREES)
ƒz =
-135
-180
-40
100
1k
10k
100k
FREQUENCY (Hz)
The response of the combined-error amplifier, optocoupler,
and PWM is too complicated to estimate analytically. You
should therefore use the existing compensation network to
plot a Bode diagram of the closed-loop transfer function
from control to output. Then place the zero and pole at
appropriate locations for maximum “phase bump” at the
crossover frequency. To maintain a gain slope of -1 to well
beyond the crossover frequency, place the error-amplifier
pole at the ESR zero location. Use the following equations
to calculate the zero (fZE) and poles (fPE) of the error
amplifier:
1
ƒZE =
...(Hz)
2 × π × Rƒ × Cƒ
ƒZE =
Figure 6. This Bode plot illustrates stability for Figure 1’s circuit
operating with component values as shown.
OUTPUT
CURRENT
1A/div
1
...(Hz)
2 × π × 47 × 10 3 × 10 × 10 −9
OUTPUT
VOLTAGE
100mV/div
ƒZE = 338...(Hz)
ƒPE =
ƒPE =
1
....(Hz)
2 × π × Rƒ × Cƒƒ
50µs/div
1
....(Hz)
2 × π × 47 × 10 3 × 220 × 10 −12
Figure 7. The transient response for Figure 1’s circuit: IOUT at
1A/div (lower trace), and VOUT at 100mV/div (upper trace).
ƒPE = 15, 392....(Hz)
Optimization on the board produces a closed-loop
bandwidth of 8kHz with 44° of phase margin. The Bode
plot of Figure 6 is based on the circuit of Figure 1, with
values as shown for the compensation components.
Layout and safety guidelines
High-frequency switching converters produce current
and voltage waveforms with high slew rates. To
minimize voltage spikes and electromagnetic radiation,
you should minimize inductance in the current loops and
PC traces. Component placement is critical in keeping
the high-frequency traces short. Follow the steps below
for good layout:
We can verify the load-transient response for a smalldeviation, fast-settling perturbation in the output voltage
by switching the load from 100mA to 1A in 20µs
(Figure 7). An overcompensated converter increases the
12
Step 1. Area product (AP):
OUTPUT
VOLTAGE
2V/div
AP ≥
(2 × POUT ) × 10 −12
...( m 4 )
η × BMAX
AP ≥
(2 × 5.61) × 10 −12
...( m 4 )
0.8 × 0.12
AP ≥ 117 × 10 −12....( m 4 )
INPUT
VOLTAGE
20V/div
Select EPC13 (TDK Part Number – PC44EPC13-Z)
Core Ap and Ae:
A P = 145 × 10 −12....( m 4 )
2ms/div
A e = 12.5 × 10 −6....( m 2 )
Figure 8. These startup waveforms appear in Figure 1’s circuit with an
optimally compensated loop. The circuit has a 48V input
voltage at 20V/div (lower trace), and a 5V output voltage
with 1A load at 2V/div (upper trace).
Step 2. Secondary inductance (LS):
• Minimize the loop formed by the input capacitor
positive terminal, transformer primary, MOSFET
switch, current-sense resistor, and input-capacitor
negative terminal.
LS ≤
430 × 10 −9 × (VO + VD )
...(H)
I OUT
S≤
430 × 10 −9 × (5.1 + 0.4)
...( )
1.1
• Keep the gate-drive trace from the MAX5021 to the
switching MOSFET short.
L S ≤ 2.15 × 10 −6...(H)
• Place the RCD snubber components close to the
input capacitor and MOSFET switch.
Step 3. Primary inductance (LP):
• Place the ceramic capacitors connected to the
MAX5021 VCC, VIN, and CS pins close to the IC.
LP =
• Minimize the loop formed by the transformer
secondary, secondary diode, and output capacitor.
• For effective heatsinking on the PC board, connect a
large copper area to the MOSFET drain, transformer
secondary, and secondary diode.
0.4 × 10 −6 × VIN ( MIN )2 × η
...(H)
POUT
P
=
0.4 × 10 −6 × 34 2 × 0.8
...( )
5.6
L P = 65 × 10 −6....(H)
The type of circuit (SELV, TNV-1, TNV-2, or TNV-3)
and its degree of pollution (determined by the circuit
surroundings) determine the requirements for clearance
and creepage between the primary and secondary
circuits. For the minimum clearance and creepage
distances between different circuit components, contact
your safety engineer or refer to Underwriters Laboratory
standard UL60950.
Step 4. Primary turns (NP):
NP =
P=
Appendix—Transformer Design
Given the specifications VIN = 36V to 72V, VOUT =
5.1V, and IOUT = 1.1A, proceed as follows:
2.1 × 10 −6 × VIN ( MIN )
A e × BMAX
2.1 × 10 −6 × 36
12.5 × 10 −6 × 0.12
P
= 47.6
Round off the primary turns, NP = 48.
13
Step 5. Secondary- and bias-winding turns (NS and Nbias):
NS = NP ×
S
LS
LP
= 8.7
Round off the secondary turns, NS = 9:
Nbias =
11.7
× NS
VOUT + 0.2
Nbias = 19.8
Round off the bias-winding turns, Nbias = 20.
Step 6. Value of the core:
AL =
LP
nH
× 10 9...
2
NP 2
AL = 26...
nH
2
Step 7. Primary and secondary RMS currents (IPRMS and
ISRMS):
IPRMS =
IPRMS =
1.63 × POUT
....( A )
η × VIN MIN
1.63 × 5.1 × 1.1
....(A )
0.8 × 34
IPRMS = 0.33....( A )
ISRMS = 1.63 × IOUT....( A )
ISRMS = 1.79....( A )
14
Small, high-voltage
boost converters
reduced, allowing the use of the MAX1605, which has a
28V internal MOSFET. The complete layout (smaller
than an 8-pin DIP) fits on a 6mm x 8.5mm double-sided
board (Figure 2).
Theory of operation
The biasing of avalanche photodiodes (APDs), piezoelectric transducers (PZTs), vacuum fluorescent displays
(VFDs), and microelectromechanical systems (MEMS)
require high-voltage power supplies. This article presents
three topologies (Figures 1a, 1b, and 1c) for generating
a high output voltage from a low input voltage. The
advantages and disadvantages of each are discussed
with focus given to power density and circuit size. The
end of the article presents experimental data to contrast
transformer-based and inductor-based solutions.
The high-voltage bias required in many APD applications
(75V) is derived from a 3V supply. That requirement
presents the following challenges:
Figure 2. This 6mm x 8.5mm DC-DC converter converts 2.5V to 75V
using the MAX1605. The front and back layout of the circuit
is shown.
• High-voltage MOSFETs generally do not operate with
a low 3V gate drive.
• The larger drain-source capacitance of high-voltage
MOSFETs requires energy in the inductor to slew the
drain to the output voltage. The resulting losses are as
large as 1/2 fswitch x CDSVOUT2.
Standard boost and flyback DC-DC converters can be
merged to form the hybrid topology of Figure 1c. The
resulting merged topology stacks secondary-winding
flyback voltage on top of the input voltage and the
primary-winding flyback voltage (a standard flyback
converter only takes advantage of flyback voltage
produced at the secondary side). Compared with a
standard boost converter, this topology produces higher
output voltages from a lower voltage MOSFET by
limiting the voltage seen at LX.
• High-voltage MOSFETs are larger and more expensive
than their lower voltage counterparts. High-voltage
power MOSFETs are seldom found within switchingcontroller ICs.
• Extreme duty cycles impose inefficiently small
off-times or low switching frequencies. Lower
switching frequency causes higher ripple and
requires larger magnetics.
A transformer provides the following advantages:
• Higher attainable output voltages;
• Lower operating duty cycle;
• Lower voltage on the MOSFET.
The circuit of Figure 1c solves these challenges by using
an autotransformer. The peak voltage on the MOSFET is
Figure 1a
Figure 1b
Figure 1c
NODE 1
VIN
VOUT
ILIM VCC
LX
BOOST
FB
CONVERTER
VOUT
VIN
ILIM VCC
VIN
2.5V TO 5.5V
D1
T1
NS
NP
LX
FLYBACK
CONVERTER FB
GND
6-PIN
SOT23
VOUT = 75V AT MORE
THAN 6mA
ILIM VCC
CONTROL
LOGIC
LX
MAX1605
GND
N
COUT
FB
A SMALL 6-PIN SOT23 WITH INTERNAL FET CONVERTS
2.5V TO 130V OR MORE THAN 6mA AT 75V.
Figures 1a, 1b, and 1c. These high-voltage DC-DC converters in three topologies are used to create high output voltage from low input voltage.
15
The following advantages also accrue when operating the
transformer in discontinuous mode, with a constant peak
current in the MOSFET:
IPK
• Higher switching frequency produces lower output ripple
• Higher frequency ripple
• Smaller magnetics
IL
0
T
DxT
STEADY-STATE CURRENT RIPPLE FOR VIN = 2.5V AND VOUT = 50V
Figure 3. This analysis of the inductor current for the circuit of
Figure 1a can be used to determine its duty cycle.
Standard boost
The circuit of Figure 1b could be made equivalent to the
circuit of Figure 1a by connecting the left side of the
transformer’s secondary winding to VIN and setting the
turns ratio to 1. The secondary-side diode can be reflected
to the primary side, making the relationship between a
flyback converter and a boost converter easier to perceive.
The standard boost converter is shown in Figure 1a.
When the MOSFET turns on, the inductor current ramps
up. When the MOSFET turns off, LX flies up to VOUT +
VD, and the inductor current ramps down. Intuitively, if
the inductor spends 1/nth of its time delivering energy to
the output, the output voltage (VOUT) is n times the input
voltage (V IN ), thus the following equation can be
generated:
Turns ratios larger than 1:1 provide leverage by allowing
higher output voltage with less extreme duty cycles.
Alternatively, node 1 of the transformer can be connected
to any power supply, enabling leverage with respect to
that supply. Because LX flies up during the off-cycle, an
additional voltage step can be obtained by connecting
node 1 to LX, as shown in Figure 1c. That connection
also helps catch some of the leakage energy that would
otherwise be dumped from the transformer’s primary side
to the MOSFET, producing short, high-voltage spikes at
the MOSFET’s drain. If the voltage spikes are higher than
the MOSFET’s voltage tolerance, a snubber circuit will
be necessary to dissipate the leakage energy.
VOUT
1
=
VIN 1 − D
where D is the duty cycle. A more analytical proof can
be found by using Figure 3. The key to this proof lies in
steady-state operation, for which the current must ramp
down the same amount that it ramps up:
∆I UP = ∆I DOWN
Thus, the final inductor current equals the initial
inductor current.
∆IDOWN =
LX is shorted to ground in Figure 1b, allowing the
primary-side current to ramp up as for an inductor. No
current flows through the secondary side of the
transformer and the diode is reverse-biased because
VIN
V
tON = IN D × T
LBST
LBST
VOUT − VIN
V
− VIN
tOFF = OUT
(1 − D) × T
LBST
LBST
VPRIMARY = − VIN
VSECONDARY = NS / NP × VPRIMARY
Because they are equal,
Primary-side current must cease when the switch at LX
turns off, but the N x I product must remain constant:
− VIN
VIN
V
(1 − D) × T
D × T = OUT
LBST
LBST
IP_ initial × NP + IS_ initial × NS = IP_ final × NP + IS_ final × NS
where subscript “P” indicates primary side, subscript “S”
indicates secondary side, “initial” indicates current at the
moment before the MOSFET is switched off, and “final”
indicates current at the moment after the MOSFET is
switched off.
VIN × D = VOUT − VIN × (1 − D)
VOUT
1
=
VIN
(1 − D)
dI = VOUT - VIN
dt
LBST
IOUTMAX
The MAX1605 and many other boost converters can be
used in this topology. Maximum output voltage is limited
by the transformer turns ratio, the transformer and diode
voltage ratings, the MOSFET’s voltage rating and drain
capacitance, and the diode’s reverse-recovery time.
∆I UP =
IINMAX
dI = VIN
dt
LBST
and
D = 1−
VIN
VOUT
16
Because IS_initial = IP_final = 0,
I S_ final =
ISAT = IPK / N
NP
IP_ initial
NS
LTOT = LBST × N
LP = LTOT / N 2 = LBST / N
The circuit of Figure 1c is similar, except that IP_final =
IS_final, thus
NP
I S_ final =
I P_ initial
NP + NS
For the steady state, it is clear from Figure 4 that:
∆IUP = IPK
To simplify, the turns ratio ‘N’ is expressed as:
N=
∆IUP =
IL
∆IDOWN =
so
dI VOUT - VIN
=
LTOT
dt
VOUT N × D
=
+1
VIN
1− D
and
D=
VOUT − VIN
VIN ( N − 1) + VOUT
Figures 3 and 4 are both drawn to scale and have the
same off-time (set to some optimal minimum). The
shaded regions in Figures 3 and 4 represent energy
delivered to the load, and the energy per pulse is
proportional to the area of those regions. That energy
can also be calculated from the expression 1/2 L x I2 (note
that L in Figure 4 is N times larger, and I is N times
smaller). Because the circuit of Figure 1c delivers less
energy per pulse, the ripple is N times smaller. Thus, a
transformer not only leverages the output voltage up; it
also leverages the output ripple down.
IINMAX
IOUTMAX
STEADY-STATE CURRENT RIPPLE FOR VIN = 2.5V and VOUT = 50V
The topology of Figure 1c delivers less energy per pulse,
but compensates by delivering more pulses, as clearly
indicated in Figure 4. Figure 1c requires an N times larger
inductance, but the saturation current can be N times
smaller because the primary and secondary side see only
IPK/N as much current simultaneously. With an ISAT
smaller by a factor of N and an inductance N times larger,
the transformer’s energy-storage capability may also be N
times smaller. Transformer size is a function of its energystorage capability, so in theory you can use a transformer
that is physically smaller by a factor of N. (In practice, the
attainable size is determined by market limitations.)
Figure 4. This analysis of the primary-side transformer current for the
circuit of Figure 1c can be used to determine its duty cycle.
The primary-side inductance is:
LP =
VOUT − VIN
V
− VIN
tOFF = OUT
(1 − D) × T
LTOT
LBST × N
Solving for VOUT /VIN yields:
IPK
IPK
N
VIN
VIN
tON =
D×T
LP
LBST / N
VOUT − VIN
VIN
(1 − D) × T =
D×T
LBST × N
LBST × N / N
tOFF = (1 - D) x T
MOSFET SWITCHES t = D X T
ON
OFF
IPK ∆IUP
=
N
N
and
Figure 4 shows the primary-side current waveform for
Figure 1c. For step-up converters whose output is more
than twice the input voltage, off-time has a greater effect
on the efficiency than on-time. Assume (for a comparable
boost converter) that off-time has been minimized by
minimizing inductance (LBST), which also minimizes
component size to the point that further reduction would
lead to undesired efficiency loss. Then, select the Figure
4 transformer’s total inductance to be N times as large.
Because the primary-side current ramps down from
IPK/N instead of IPK, the primary inductance must be N
times as large to maintain the same off-time.
t
∆IDOWN =
where ∆IUP is the upward step in the primary-side current
and ∆IDOWN is the downward step. ∆IUP and ∆IDOWN can
be calculated as follows:
NP + NS
NP
because the secondary side of Figure 1c never acts
independently. Though unconventional, this definition
for N is more appropriate for Figure 1c.
2
dI = VIN x N
LTOT
dt
and
LTOT
N2
where LTOT is the total inductance of the autotransformer
since LP is N2 smaller than LTOT, and LTOT is N times
larger than LBST, LP is LBST/N. As a result, the primary
side ramps faster than a simple boost converter.
17
Output ripple
Efficiency considerations
For discontinuous conduction, either converter’s output
ripple can be calculated by equating energy change in the
inductor or transformer with energy change in the output
capacitor during the off-cycle. Because the inductor/transformer has zero energy at the end of the cycle, no-load ripple
can be calculated as:
There are three main efficiency losses to consider in the
transformer topology. Transformer/inductor DC resistance combined with switch resistance produces losses
proportional to the square of the peak current.
Transformer-leakage inductance produces loss because
transformer energy is not completely coupled to the
output. When the diode is quickly and heavily reversebiased (when the MOSFET turns on), any delay in the
diode (the reverse-recovery time, tRR) can also cause
significant loss.
1
2
1
2
LI 2 = 12 COUTVfinal 2 − 12 COUTVinitial2
LI 2 = 12 COUT(Vfinal − Vinitial ) × (Vfinal + Vinitial )
1
2
LI 2 = COUT ∆VOUT × VOUT
∆VOUT =
1
2
The percent efficiency loss due to DC resistance in the
switch and in the primary side of the transformer is
independent of load, and can be approximated as:
LI 2
COUT VOUT
ER _ LOSS
×100%
ER _ LOSS + Edelivered
For the boost converter L = LBST and I = IPK. For the circuit
of Figure 1c, L = LBST x N and I = IPK/N, so:
∆VOUTA =
∆VOUT C =
1
2
1
2
where ER_LOSS is the energy loss due to resistance per
pulse, and Edelivered is the energy delivered per pulse. By
taking the integral of the primary-side resistive power
dissipation, the resistive efficiency loss for large duty
cycles can be approximated as:
LBST( IPK )2
CVOUT
IPK 2
)
N = ∆VOUTA
CVOUT
N
N × LBST (
1I R
3 PK
1 I R + 1V
3 PK
2 IN
where ∆VOUTA is the output ripple of the boost configuration, and ∆VOUTC is the output ripple for the Figure 1c
circuit. Ripple in Figure 1c is 1/Nth as large for the boost
configuration and the switching frequency is N times higher.
where D is the duty cycle expressed as a percentage, and
R is the sum of the switch resistance and the primaryside resistance. For operation in discontinuous mode, the
same equation applies for the circuit of Figure 1a or 1c.
The efficiency loss due to leakage inductance can be
approximated as:
L leakage
×100%
L leakage + L primary
Figure 5 offers a comparison of ripple for the circuits of
Figures 1a and 1c, when both circuits are designed for the
same off-time. Because the duty cycle is normalized in the
transformer circuit (made closer to 50%), the controller can
operate at a frequency N times higher for the same off-time.
ESR RIPPLE
× 100%
where Lleakage is the total leakage inductance seen at the
primary side. Transformers with higher turns ratios
provide larger leakage inductance, higher frequency,
and deliver lower energy per pulse, so the source of
inefficiency becomes more significant.
BOOST WITH TRANSFORMER LEVERAGE
CAPACITIVE RIPPLE
Transformer selection
Since the selection of off-the-shelf transformers is much
narrower than that of equivalent inductors, transformers
generally cost more than inductors of equivalent energy
and energy density. The customer base for transformers
is smaller, yet the set of possible transformer
configurations is much larger than that of corresponding
inductor configurations. As a result, magnetic designs
based on custom transformers are often necessary.
STANDARD BOOST
Figure 5. This illustration contrasts the ripple produced by the
circuit of Figure 1a vs. the circuit of Figure 1c assuming
both are optimized for a minimum acceptable off-time.
18
Figure 7 shows the MAX1605 maximum output current
(the measured typical, for which the controller falls out
of regulation by 5%) as a function of output voltage and
input voltage.
When specifying an autotransformer, consider an equivalent inductor. The following inductor, for example, is
available from Toko: D32FU 680µH, 74mA, 20Ω, 3.5mm
x 3.5mm x 2.2mm. It is reasonable to ask for an autotransformer whose end-to-end characteristics are similar. For
such a transformer with a 1:9 turns ratio, the primary-side
rating would be 6.8µH, 740mA, and 2Ω. That inductance
rating is based on a value of N2 (where N is the total
number of turns divided by the number of primary-side
turns). For a 1:9 turns ratio there must be some multiple of
10 turns total. N = 10 must be used for the previous calculation. The saturation current is inversely proportional to
N and resistance is proportional to N.
MAXIMUM LOAD vs. OUTPU
10
VIN = 5V
MAXIMUM LOAD (mA)
8
6
VIN = 3.3V
4
2
Occasionally, thermal limitations do not allow the
maximum current rating to scale by N. Also, a limited
product selection may bar you from that ideal starting
point. This analysis provides a starting point and a decent
edge when discussing possibilities with a custom transformer vendor. When wound as an autotransformer, an
equivalent magnetic component should require less space
(a smaller wiring cavity) because the lower currents allow
thinner wire on the secondary side. (Extra manufacturing
costs usually prohibit this approach, however.)
VIN = 2.5V
0
30 40 50 60 70 80 90 100
OUTPUT VOLTAGE (V)
Figure 7. This Maximum Load vs. Output Voltage graph illustrates the
maximum load deliverable by the circuit of Figure 6.
Output ripple for the circuit of Figure 1c can be calculated
as:
1L ×I 2
P
PK
VRIPPLE = 2
COUT × VOUT
Application
where LP is the primary-side inductance, IPK is the peak
primary-side current (500mA), C OUT is the output
capacitance (0.47µF), and VOUT is the output voltage.
For a 75V output, the ripple is 16mVP-P. The low inductance that produces such a low ripple is typically not
efficient in a straight boost configuration like Figure 1a.
The circuit of Figure 6 produces an APD bias of 75V.
Because the transformer reduces voltage stress on the
switch, you can use a small 6-pin SOT23 device such as
the MAX1605. The 28V, 500mA MOSFET in that IC is
more than adequate since it only sees peak voltages of
VIN + (VOUT - VIN) / N = 17V. With a higher turns ratio,
the circuit can handle higher voltages.
SUMIDA CMD4D13 4365-T020,
NP:NS IS 1:5, LP = 4.53µH
IP_SAT = 562mA, RP = 0.852Ω
SIZE IS 4.1mm x 4.3mm x 1.45mm
VIN
2.5V TO 5.5V
0.47µF
CENTRAL SEMICONDUCTOR
CMOD4448 BK SOD-523
D1
T1
NP
Even 16mV P-P ripple is not small enough for many
applications. To bias an APD, high ripple is unacceptable
because it couples directly into the signal. Such applications can use an RC or LC filter following the power
supply, but the resistor in an RC filter produces a loadregulation error. (Typical load currents are small, but the
ripple filter may require large resistors.)
NS
SHDN LIM VCC
CONTROL
LOGIC
LX
MAX1605
GND
N
750kΩ
FB
13.7kΩ
At 100V, high-capacitance values require board space, so
the filter is composed largely of resistance. For the same
cutoff frequency (using the same resistor and capacitor),
you can reduce the load-regulation error by a factor of β
using the circuit of Figure 8. Although a fixed VBE drop
is introduced instead, that approach dramatically reduces
the dependence of VOUT on load. To achieve greater
ripple reduction for the same level of load regulation, a
β-times larger filter resistor can be used.
VOUT = 75V AT MORE
THAN 6mA
470pF
COUT
0.47µF 100V
CERAMIC CAPACITOR
IN 1210 CASE
(TDK C322X7R2A474K)
Figure 6. This circuit, whose layout is shown in Figure 2, is used to
produce 75V from 2.5V.
19
components were used in conjunction with the MAX668
Evaluation Kit:
OUT
RFILTER
• Inductor: Coilcraft DO1813P-472HC 4.7µH,
2.6A 0.054Ω inductor
• Ultra-fast diode: ES1D 200V 15ns
reverse-recovery time
• MOSFET: IRF640NS 200V 0.15Ω QG = 67nC,
COSS = 185pF, and provides over 2A with 5.5V
gate drive
• Sense resistor: 50mΩ sense resistor
CFILTER
FILTERED OUT
Another resistor connected between the FB terminal and a
voltage source allows that source to adjust the output
voltage by sinking or sourcing current to the FB terminal.
You can then adjust the output voltage to 150V and the
input voltage to 6V.
Figure 8. A filter further reduces ripple.
Experimental comparison: inductor and
transformer approaches
For a fair comparison of inductors versus transformers in a
high-voltage converter, a switching converter with the
following features was selected:
•
•
•
•
For the inductor-based solution, the maximum load current
is 18mA at 150V (2.7W). Peak efficiency (65%) occurs at
maximum load and the quiescent (no-load) current is 91mA
with a 6V supply. The quiescent-current loss in the inductor
circuit is due to the diode’s reverse-recovery time and the
MOSFET’s drain capacitance. Those effects are illustrated
in Figure 10.
External MOSFETs
Adjustable switching frequency
Adjustable current limit
Evaluation kit available
The transformer-based approach (Figure 11) was
implemented by exchanging the inductor for the following
transformer, using the topology of Figure 1c:
The MAX668 current-mode controller fulfills these criteria
while eliminating the need for a feedforward capacitor. The
circuit of Figure 9 allows you to compare performance by
swapping the transformer for an inductor and by swapping
MOSFETs.
Sumida CMD-8LN 6313-T036,
LP = 5.6µH, IP = 2.3A, NP:NS = 1:9,
RP = 0.5Ω
The MAX668 includes a MOSFET driver that can
efficiently drive the 48nC gate charge of an IRF7401
MOSFET. With the following components, it forms a
150V inductor-based boost converter. The following
CEE98 6343-T361,
NP:NS is 1:9, LP = 2.4µH,
IP_SAT = 7A, RP = 0.055Ω
SIZE IS 8.2mm x 9.5mm x 6.0mm
D1
NP
SYNC/
SHDN
VCC
EXT
CS
MAX668
FREQ
R4
121kΩ
ES1D ULTRA-FAST, 200V,
15ns DIODE
T1
VIN = 5V
68µF
LX
Using a transformer with a 1:9 turns ratio requires only a
22V MOSFET, but an actual application would use a 30V
MOSFET (instead of a 200V MOSFET) for the transformer circuit. Yet, the efficiency is 77% while delivering
25.5mA at 150V (3.8W). Peak efficiency is 88% at 15mA,
and the no-load supply current is only 1.8mA total.
NS
Using the 200V MOSFET with a transformer allows the
possibility of much higher voltage. A 200V MOSFET and
1:9 transformer can approach output voltages as high as
2kV in theory, but in practice the transformer windings
cannot sustain such high voltage. A greater problem,
however, is obtaining >1kV diodes with fast reverserecovery times. For slow reverse-recovery times, the
switching speed must be reduced.
VOUT
R2
10MΩ
N1
IRF7811W
R1
50mΩ
COUT
FOUR 0.1µF 250V
CERAMIC
CAPACITORS
PGND
GND
FB
VADJ
R3
75kΩ
220pF
Substituting a 400V CMR1U-04 diode from Central
Semiconductor (50ns t RR ) for the ES1D diode and
changing the output capacitors allows the circuit to
produce outputs as high as 400V. The ES1D diode
cannot reliably produce outputs greater than 346V,
because its anode goes to -9 x VIN when the MOSFET
Figure 9. This circuit can be used to deliver higher power and
higher output voltage.
20
ZOOM OF NO-LOAD SWITCHING WAVEFORM
NO-LOAD SWITCHING WAVEFORM FOR AN INDUCTOR SOLUTION
INDUCTOR CURRENT
500mA/div
CHARGE INTO
OUTPUT
0
CHARGE OUT OF OUTPUT
0.8A REVERSE-INDUCTOR
CURRENT
INDUCTOR CURRENT
SLEWS CLX UP
VLX
50V/div
0
INDUCTOR CURRENT
SLEWS CLX DPWN
MOSFET BODY DIODE
CLAMPS NEGATIVE
CURRENT
DIODE IS
CONDUCTING
MOSFET IS ON
400ns/div
20ns/div
Figure 10. This scope photo (left) demonstrates the effect of the ES1D diode’s 15ns reverse-recovery time. The zoom of the waveform (right)
reveals that the inductor current actually ramps negative since the diode fails to turn off in time.
turns on. Configured for VOUT = 330V, the circuit was
able to produce 9.6mA (3.1W) at 60% efficiency and the
peak efficiency was 66% at 4mA.
MOSFET. The dominant loss is associated with the
transformer’s primary-side resistance, which is about
0.5Ω. You can scale this system to provide higher
power. For example, another custom transformer
specified for IPSAT = 5A and LP = 1.7µH should deliver
more than twice as much power.
As mentioned earlier, a 30V MOSFET is more logical for
the 150V output. The IRF640NS was replaced with a
logic-level IRF7811W (30V, 0.012Ω, QG = 18nC, CDSS =
500pF). Resistance drops dramatically (from 0.15Ω to
0.012Ω), but the efficiency improvement is minor. When
delivering a maximum load of 25.7mA at 150V, the
efficiency is 82.3% (vs. 77%). Peak efficiency at 15.5mA
is 88%. Efficiency results are summarized in Figure 12.
Thus, besides use of the smaller, cheaper, and more
efficient IRF7811W MOSFET, a transformer boost
converter can operate with lower input voltages.
Transformer leverage can improve power density and
efficiency, reduce ripple, and allow the use of smaller,
cheaper, and sometimes internal MOSFETS. The cost of
applying transformer leverage is mostly due to market
limitations. When size and power density are a high
priority, consider using transformers.
The insignificant improvement in efficiency implies two
things. Primary losses are in the transformer (resistive
loss and leakage energy), and capacitive losses are in the
EFFICIENCY vs. LOAD
100
TRANSFORMER-BASED, 30V MOSFET, 150V OUTPUT
90
ILPRIM
0.5A/div
EFFICIENCY (%)
80
O
VLXI
20V/div
TRANSFORMER-BASED, 200V MOSFET,
150V OUTPUT
70
60
50
40
TRANSFORMER-BASED,
200V MOSFET, 330V OUTPUT
30
O
20
OUTPUT RIPPLE
AC-COUPLED
200mV/div
10
INDUCTOR-BASED, 200V MOSFET, 150V OUTPUT
0
0
5
10
15
20
25
LOAD (mA)
Figure 12. This Efficiency vs. Load graph compares and contrasts
transformer-based, step-up DC-DC converters vs.
inductor-based, step-up DC-DC converters. The
maximum load, quiescent current, and efficiency are
greatly improved using a transformer.
Figure 11. The switching waveform shows the transformer-based
approach for the MAX668 circuit, a 150V step-up
DC-DC converter shown in Figure 9.
21
DESIGN SHOWCASE
Compact, inductorless boost circuit
regulates white-LED bias current
The increasing use of color LCDs in hand-held
equipment is creating a need for smaller and cheaper
sources of white backlight. Cold-cathode fluorescent
lamps (CCFLs) and electro-luminescent (EL) panels
have been used in the past, but those circuits are
excessively large, expensive, and complex for today’s
hand-held, consumer-electronics devices. Fortunately,
recent advances in LED technology have produced an
LED that emits white light. White LEDs have several
advantages over conventional backlights, including
small size, low cost, low complexity, and high reliability.
0.22µF
7
CXN
1Ω
IN
1µF
3,4
8
CXP
OUT
IN
1µF
9
WHITE LED
0.22µF
U1
FB
10
MAX1759
2
ON
OFF
SHDN
1
To µC
16Ω
PGND
6
The typical forward-bias voltage for white LEDs is
about 3.5V ±10%. To obtain white light, you simply
forward-bias the device, but a boost circuit is required
if the white LED’s forward voltage can be greater
than the battery voltage. The conventional approach
to this problem—a boost regulator that biases the
LEDs through a ballast resistor—has two drawbacks.
First, the wide variation of forward voltage in white
LEDs causes a large variation in bias current and the
resulting light output. Second, the conventional boost
converter has a DC path between input and output
(even in shutdown) that allows an inactive LED to
drain the battery.
POK
GND
5
Figure 1. Unorthodox connections enable this regulated
charge-pump IC to directly regulate the white
LED’s bias current.
secondary concern in this application and allows the
use of a small (0.22µF) output capacitor. Even so, the
output ripple is only 400mVP-P.
(A similar idea appeared in the 3/5/01 issue of
Electronic Design.)
Figure 1’s compact circuit overcomes these
problems. The regulated buck/boost charge pump in a
small µMAX package (U1) has a 100mA outputcurrent capability. Configured as shown, the circuit
directly regulates bias current flowing through the
white LED. By biasing multiple white LEDs in
parallel, it provides good light distribution. The U1’s
design eliminates the troublesome input-output path
in shutdown, and its SHDN input lets the user turn
the backlight on and off. The circuit also includes a
power-OK (POK) output for signaling a microprocessor when the backlight is available.
Though not necessary in this case, the input RC “π”
filter limits voltage ripple reflected back to the input
to just 40mVP-P (for VIN = 3.6V). Because outputvoltage ripple is not visible to the eye, it is of
22
DESIGN SHOWCASE
Analog switch lowers relay
power consumption
Table 1. Power dissipated by relay
Relays are often used as electrically controlled
switches. Unlike transistors, their switch contacts are
electrically isolated from the control input. On the
other hand, the power dissipation in a relay coil may
be unattractive for battery-operated applications.
Adding an analog switch lowers the dissipation,
allowing the relay to operate at a lower voltage
(Figure 1).
Voltage
(V)
5 (Normal
OperatingVoltage)
3.5
(Pickup Voltage)
2.5 (Circuit
of Figure 1)
Power consumed by the relay coil equals V2/RCOIL.
The circuit lowers this dissipation (after actuation) by
applying less than the normal operating voltage of 5V.
Note that the voltage required to turn a relay on
(pickup voltage) is greater than that required to keep it
on (dropout voltage). The relay shown has a 3.5V
pickup voltage and a 1.5V dropout voltage, yet the
circuit allows it to operate from an intermediate
supply voltage of 2.5V. Table 1 compares the relay’s
power dissipation with fixed-operating voltages across
it, and with the Figure 1 circuit in place.
SW1
D1
+
2
6
C2
0.15µF
U1
MAX4624
63
221
45
250
The values for R2 and C2 are selected to allow C1 to
charge almost completely before C2’s voltage reaches
the logic threshold of the analog switch. In this case, the
time constant C2R2 is about seven times C1(R1 +
RON). Larger C2R2 values increase the delay between
switch closure and relay activation.
D2
4
1
450
Component values for this circuit depend on the relay
characteristics and the supply voltage. The value of R1,
which protects the analog switch from the initial current
surge through C1, should be sufficiently small to allow
C1 to charge rapidly, but large enough to prevent the
surge current from exceeding the specified peak current
for the analog switch. The switch’s peak current (U1) is
400mA, and the peak surge current is IPEAK = (VIN VD1)/(R1 + RON), where RON is the on-resistance of
the analog switch (typically 1.2Ω). C1’s value depends
on the relay characteristics and on the difference
between VIN and the relay’s pickup voltage. Relays that
need more turn-on energy require larger C1 values.
VIN = 2.5V
5
90
When C2 reaches that threshold, the analog switch
connects C1 in series with the 2.5V supply and relay
coil. This action turns on the relay by boosting the
voltage across its coil to 5V (twice the supply voltage).
As C1 discharges through the coil, the coil voltage
drops back to 2.5V minus the drop across D1, but the
relay remains on because that voltage is above the
relay’s dropout voltage (1.5V).
By closing SW1, current flows in the relay coil, and
C1 and C2 begin to charge. The relay remains inactive
because the supply voltage is less than its pickup
voltage. The RC time constants are such that C1
charges almost completely before the voltage across
C2 reaches the logic threshold of the analog switch.
C1
100µF
R2
27kΩ
Current
Total Power
(mA)
Dissipation (mW)
R1
4.7Ω
3
(A similar idea appeared in the 12/20/01 issue of EDN.)
Figure 1. An analog switch lowers relay power dissipation.
23
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