Miniature Wireless Magnetoelastic Resonant Motor with Frequency Selectable Bi

Miniature Wireless Magnetoelastic Resonant Motor with Frequency Selectable Bi
Miniature Wireless Magnetoelastic Resonant
Motor With Frequency Selectable
Bidirectional Rotation
Jun Tang, Scott R. Green, and Yogesh B. Gianchandani, Fellow, IEEE
Abstract—This paper presents the analysis, fabrication, and experimental results of wirelessly actuated chip-scale rotary motors.
Two designs are described. Design M is actuated by a φ8-mm magnetoelastic stator lithographically micromachined from Metglas
2826MB-bulk-foil with 25 μm thickness. It operates at a resonant
frequency of 11.35 kHz while 3-Oe dc and 2-Oe amplitude ac
magnetic fields are applied. The measured rotation speed, start
torque, calculated driving step size, and payload are 44 r/min,
2 nN · m, ≈23 mdeg, and 9 mg, respectively. Design S uses a
stator that is a sandwich of Si (φ8 mm diameter and 65 μm
thickness) and magnetoelastic foil (φ8 mm diameter and 25 μm
thickness) to tailor the stiffness. The typical resonant frequencies
of clockwise (CW) mode and counterclockwise (CCW) mode are
6.1 and 7.9 kHz, respectively. The CCW mode provides a rotation
rate of about 100 r/min, start torque of 30 nN · m, and driving step
size of 74 mdeg while 8-Oe dc and 6-Oe amplitude ac magnetic
fields are applied. Bidirectional rotation is realized by switching
the applied frequency, thereby exciting the stator in a slightly
different mode shape. Design S shows at least 43-mg payload
Index Terms—Magnetostrictivity, Metglas, photochemical machining (PCM), transient liquid phase bonding.
HIP-SCALE rotary motors have been of interest since the
1980s. There are a number of microsystem applications
that would benefit from rotary micromotors, including defense,
robotics, aerospace, laboratory automation, medical, and so on.
One interesting example application involves the integration
of a rotary micromotor with a MEMS gyro. The controlled
rotational input can be used for in situ recalibration of signal
drift from the gyro, rather than requiring the gyro to be taken
out of service for calibration. Another possible application for
a wirelessly actuated miniature motor is to provide propulsive
force for microbots in cavities and ducts. Similar robotic technology can potentially conduct in situ monitoring and repair of
large-scale complex machines, such as engines, eliminating the
need for disassembly.
Manuscript received July 18, 2012; revised December 3, 2012; accepted
January 14, 2013. Date of publication February 22, 2013; date of current
version May 29, 2013. Subject Editor D.-I. Cho.
The authors are with the Center for Wireless Integrated MicroSensing and
Systems (WIMS2 ), University of Michigan, Ann Arbor, MI 48109-2125 USA
(e-mail: [email protected]; [email protected]; [email protected]).
Color versions of one or more of the figures in this paper are available online
Digital Object Identifier 10.1109/JMEMS.2013.2242253
Rotary motor actuation methods have included electrostatic,
electromagnetic, piezoelectric, thermal, and so on. Electrostatic
motors [1], [2] are considered as one of the milestones in
MEMS history. These motors usually can operate at very
high rotation rate and are compatible with IC processes, but
packaging requirements, output torque, and payload capacity
limit applicability. In recent years, there have been improvements in torque and payload capacity by utilizing miniature
ball bearings [3] and liquid bearing [4] systems. Macroscale
electromagnetic motors are very mature, but efforts toward their
miniaturization have resulted in landmark accomplishments for
the microsystem research community [5]–[8]. Relative to other
technologies, these motors typically offer high rotation rate and
torque, but integration of coils requires careful consideration.
Another successful and well-studied technology is piezoelectric
micromotors [9]–[14]. Piezoelectric ultrasonic motors usually
have a piezoelectric stator resonating at a high frequency. In
each cycle, the rotor is pushed by a small tangential step. These
motors usually have high payload, high stall torque, and high
angular resolution. Thermally actuated rotary motors [15] can
be used for applications requiring large forces. In this paper,
another option for chip-scale rotary motors—magnetoelastic
actuation—is investigated.
Magnetoelastic coupling represents the interaction between
the material conditions of strain, stress, and magnetization.
Magnetoelastic materials exhibit strains under an external magnetic field due to the field-directed rotation and alignment of
tightly coupled, elongated structural, and magnetic domains in
the materials. The strain induces stress in the material, which, in
turn, alters the magnetization. When excited with an oscillatory
field, these two simultaneously existing effects make magnetoelastic materials attractive for wireless resonant sensing.
Various magnetoelastic sensing systems have been studied for
pressure, temperature, liquid viscosity and density, fluid flow
rate, pH, glucose, Young’s modulus, stent occlusion, etc. [16].
Utilizing magnetoelastic material for actuation in miniature and
microscale systems remains an open challenge. Magnetoelastic
actuators can operate by mechanically resonating in response
to an alternating magnetic field. The resonant motion is used
to drive movement of other parts of the system. The use of
magnetoelastic transduction enables wireless actuation, which
simplifies device architecture and allows for remote operation.
The transduction method exhibits relatively high payload carrying capacity. It also offers noise immunity due to resonant
1057-7157/$31.00 © 2013 IEEE
Fig. 2. (a) Schematic of design M geometry. (b) Customized teeth design. In
design M, the stator is fabricated from magnetoelastic metal foil alone.
Fig. 1. Wireless magnetoelastic resonant rotary motor operation concept.
(a) Standing wave in stator. (b) Bidirectional rotation operation method.
This paper describes two types of magnetoelastic rotary
motors: in design M,1 the motor is actuated by a magnetoelastic
bulk foil stator; in design S, the motor is driven by a stator that
uses a sandwich of Si and magnetoelastic foil. Both designs
successfully demonstrate the wireless magnetoelastic rotary
actuation method. Furthermore, bidirectional rotation of the
design S motor is achieved by exciting the stator vibration in
slightly different mode shapes. Section II provides a theory
of motor operation, simulation of the magnetoelastic rotary
actuation method, and geometrical design of the two prototypes. Section III details the fabrication process of the motors.
Section IV describes the experimental setup and preliminary results. Section V discusses the advantages of the wireless rotary
actuation method, performance comparison of the two designs,
and performance improvement plans. Section VI provides the
Singular magnetoelastic material, which can either be the
vibrating stator itself or the actuator for the stator, can be wirelessly driven by magnetic fields. The deformation is substantially enhanced if a proper dc magnetic field is superimposed
on the ac magnetic field. Thus, both dc and ac magnetic fields
are required for rotary motor wireless actuation [18].
B. Modeling
A custom magnetomechanical harmonic finite-element technique [18] is used to estimate modal displacements, shapes, and
frequencies for the magnetoelastic material. Although magnetoelastic materials are generally nonlinear, it is appropriate to
use linearized constitutive equations describing the coupling
between flux, field strength, stress, and strain in a magnetostrictive material
[C][d]T B
μ0 μr
= − [d][C] ε + 1 B
μ0 μr
μ0 μr
A. Theory
The rotary actuation mechanism in this effort is similar to
that for piezoelectric ultrasonic rotary motors [9]. A vibratory
wave, which could be either standing wave or travelling wave, is
generated in the stator. In this case, the vibratory micromotion
is a standing flexural resonant wave and is generated magnetoelastically. The resulting vibratory mode shape of the stator
has antinodes at which maximal out-of-plane deflection occurs.
Teeth are located on the stator such that they are slightly offset
from the antinodes, resulting in elliptical motion of the teeth
tip and a contact force with a tangential component that causes
the rotor to rotate (Fig. 1). By selecting resonant frequency
of different mode shapes of the stator, bidirectional rotation
can be achieved. A rotor, which can contain other microsystem
components, is then stacked above the stator.
1 Portions
of this paper have appeared in conference abstract form in [17].
σ = [C]ε −
where σ is the stress vector, C is the stiffness matrix, ε is the
strain, d is the magnetostrictivity matrix, B is the magnetic
flux density vector, H is the field strength vector, μ0 is the
permeability of free space, and μr is the relative permeability.
Equations (1) and (2) are implemented in this paper by utilizing
COMSOL Multiphysics and coupled time-harmonic induction
current and stress–strain frequency response modes. A detailed
look at a finite-element analysis (FEA) implementation for
magnetostrictive materials is presented in [19]; the approach
used in this work is modified for application to resonant actuators. In this paper, two types of magnetoelastic rotary motor,
namely, design M and design S, are modeled and investigated.
C. Design M
The design M motor consists of a magnetoelastic material—
Metglas 2826MB stator—and two stainless steel bases
Fig. 4. Schematic of design S geometry. In design S, the stator is fabricated
from a layer of magnetoelastic metal foil bonded to Si.
Fig. 3. FEA simulation results of the frequency response of the bulkmagnetoelastic-foil stator of design M.
[Fig. 2(a)]. The stacked architecture is modular, which significantly simplifies the fabrication process. Each layer can be
easily fabricated by utilizing photochemical machining (PCM)
process, as described in the fabrication section. Alignment
pins ensure reasonable assembly accuracy. Two stainless steel
bases provide a recess allowing stator vibration and initial
positions for alignment pins and a hub. The ring-shaped stator
is suspended with four crab-leg springs, which are stiff in the
rotational direction but flexible in the out-of-plane direction—
preventing stator rotation during rotor actuation and allowing
large vertical deformation.
For a stator fabricated solely from Metglas 2826MB, careful
consideration is required for the placement of the driving teeth.
The number of driving teeth is usually the same as the number
of antinodes in the desired mode shape for a resonant rotary
motor [9]. The driving teeth are typically offset from the
location of the antinodes in order to ensure elliptical motion of
the tooth tip. However, the mass loading added by the driving
teeth causes the antinodes to shift to the same location as the
teeth. Consequently, the teeth no longer move elliptically, and
no rotation is generated. Therefore, as shown in Fig. 2(b), a
design employing eight teeth is proposed to address the issue.
There are still four sets of teeth, but one set of teeth is a
combination of two teeth: an auxiliary tooth and a driving tooth,
each separated slightly from each other. The auxiliary teeth and
driving teeth have the same mass so that the mass center and
antinodes of the vibratory mode shape will be in the center
between the two different teeth. However, the driving tooth is
taller, so only it will contact the rotor. As a result, the driving
teeth can move in an elliptical manner and can drive the rotor
The FEA results shown in Fig. 3 predict that a stator (4 mm
inner diameter, 8 mm outer diameter, and 25 μm thickness)
has the desired mode shape at a resonant frequency of about
11.43 kHz, with 0.1-μm out-of-plane deformation under harmonic excitation with an amplitude of 2 Oe. Due to the orientation and size of the teeth and spring suspension, a slightly
different performance is predicted for different directions of
alternating magnetic excitation. Simulation results suggest that
applying magnetic field between the suspension springs gives
the largest out-of-plane displacement amplitude. As desired,
the vibration mode shape demonstrates that the antinodes are
located in the center between a driving tooth and an auxiliary
Fig. 5. FEA simulation results of the frequency response of the design M
stator with two mode shapes.
D. Design S
As shown in Fig. 4, design S has a stacked structure similar
to that of design M, but the stator is made of a silicon–Metglas
sandwich instead of solely Metglas 2826MB. The benefit of
silicon is that it has better surface uniformity compared to magnetoelastic foil. Furthermore, the higher resonant frequency and
quality factor of the thicker silicon can potentially increase the
rotation rate and can improve the positioning resolution. Using
standard micromachining processes, silicon should facilitate
the direct integration of the driving teeth on the stator. The
thicker silicon also means that no auxiliary teeth are required
in this design, and the intended bidirectional rotation can be
realized by switching the driving frequency and mode shapes.
A ring-shaped Metglas 2826MB disc is attached to the design
S stator so that the motor is still wirelessly driven by magnetic
fields. The rotor is patterned to have six radial segments so that
the rotation rate can be measured visually or optically.
The FEA simulation results show that the design of the
teeth can be simplified because the stator is thick and robust
compared to the teeth. Additionally, bidirectional rotation can
be achieved by switching the mode shapes of the stator (Fig. 5).
The intended clockwise (CW) vibration mode shape of a design
S stator with 4 mm inner diameter, 8 mm outer diameter, and
65 μm thickness is simulated to be at 8 kHz with 3.2-μm
amplitude; the counterclockwise (CCW) vibration mode shape
exists at 11 kHz with 2.7-μm out-of-plane amplitude. A 6-Oe
magnitude ac magnetic field is applied in the simulation.
Fig. 7. Fabrication process flow of design M.
Fig. 6. (a) Fabrication process flow of design M. (b) Optical image of the
assembled device.
E. Metglas 2826MB and Driving Methods
Metglas alloys provide excellent magnetostrictive properties
as well as adequate mechanical properties. In this design,
Metglas 2826MB, an amorphous NiFeMoB alloy, is used. Its
saturation magnetostriction is 12 ppm, and its dc permeability is
larger than 50 000 [20]. These materials are readily available in
foils (≈ 25 μm thick) and are easy to pattern by utilizing PCM.
Relative to other Metglas alloys and magnetoelastic materials,
2826MB also requires a small dc bias (less than 10 Oe) and can
be stimulated with a relatively small alternating field.
In principle, an onboard coil and permanent magnets can
be implemented to provide the driving ac and dc magnetic
fields. For example, inductive coils for generating ac magnetic
fields can be patterned on an underlying silicon substrate using
standard micromachining techniques. On the same substrate,
permanent magnet materials such as permalloy or samarium
cobalt can be deposited and used to provide the dc fields
required to bias the magnetoelastic material. However, for the
motors presented in this paper, external coils are used.
The fabrication process flow of a design M motor is illustrated in Fig. 6(a). First, the magnetoelastic stator is batchpatterned using PCM [21] from the Metglas 2826MB foil. The
ring-shaped stator is patterned with an inner diameter of 4 mm,
an outer diameter of 8 mm, and a thickness of 25 μm. Other
base layers are also fabricated using PCM from 0.5-mm-thick
stainless steel foils. The layers are stacked and aligned with
pins, and bonded to each other with epoxy. The stainless steel
auxiliary teeth (500 μm wide and 300 μm tall) and driving teeth
(300 μm wide and 500 μm tall) are manually placed on the
stator and fixed with epoxy. The rotor is microelectrodischarge
machined (μEDM) from Metglas 2826MB foil; it has a total
mass of about 9 mg. A hub with a diameter of 2 mm is used
to constrain the rotor. The overall size of the chip is 2 × 2 cm2
[Fig. 6(b)].
For design S, the stator and rotor are fabricated side by side
using a two-mask deep reactive-ion etching (DRIE) process,
as shown in Fig. 7(a)–(d). The fabrication process starts with
coating a layer of KMPR 1010 on the backside of a silicon
wafer, to serve as an etch stop and support layer for throughwafer etching. Silicon oxide of 4 μm thickness is then deposited
on the front side of the wafer by PECVD and is patterned using
mask 1. The silicon oxide layer is wet etched in 5 : 1 buffered
hydrofluoric acid solution. Photoresist is used to make the first
DRIE step in which the ring-shaped stator is defined. Then, the
PR is removed, and the remaining silicon oxide layer is used
to etch through the remainder of the wafer. At the end of this
process, the ring-shaped stator is fully etched out along with
the teeth that are located on the stator. After removing KMPR
1010 by immersing the wafer in Remover PG (MicroChem
Corporation, Newton, MA, USA) at 80 ◦ C for 1 h, the silicon
rotor is ready for assembling. The silicon stator must be bonded
to Metglas 2826MB as described in the following steps.
Silicon–Metglas 2826MB bonding is an important step because it affects the actuation of the silicon stator. An Au–In
liquid transient phase bonding (TLP) is used because it can be
performed at relatively low temperature, ensuring that magnetic
properties of Metglas 2826MB will not change. Additionally,
gold and indium can be easily coated using standard microfabrication steps. The possible stages of TLP bonding for indium
rich are shown in Fig. 8 [22], [23]. Liquid–solid interdiffusion,
stage 1, occurs when two substrates are brought into intimate
contact while the temperature is above 157 ◦ C, and the appropriate pressure is applied. Liquid indium dissolves gold layers
on both the silicon substrate and the Metglas 2826MB substrate.
Simultaneously, dissolved gold diffuses into the indium layer
to form AuIn2 . Stage 2 occurs upon cooling; the final bonding
Fig. 8. Composition of Au–In TLP bonding and possible stages of the bonding process: stage 1, melting of In and liquid–solid interdiffusion at 157 ◦ C–
200 ◦ C; stage 2, solidification of the mixture below 157 ◦ C [22], [23].
layer is a mixture of solid indium and AuIn2 because the
quantity of gold is insufficient to react with all indium.
The use of this bonding step with the process sequence
of the design S motors is shown in Fig. 7(d)–(g). Initially,
chrome (0.1 μm) and gold (0.5 μm) are evaporated on both the
Metglas 2826MB ring-shaped structure and the backside of the
silicon stator. The step is followed by electroplating of ≈8-μm
indium on the backside of the silicon stator. The silicon stator
and the Metglas 2826MB component are aligned and bonded.
The bonding process is performed in a vacuum oven so that
oxidation is prevented. The temperature is held at 200 ◦ C for 1 h
during this step. After bonding, the connections between ringshaped disc and frame in the Metglas 2826MB layer are cut
using μEDM. An SEM image of the silicon stator with bonded
ring-shaped Metglas 2826MB is shown in Fig. 9(a).
As the final step before testing, the design S stator, stainless
steel bases, and a hub are aligned, stacked, and fixed to each
other with epoxy. The silicon rotor located directly above the
stator is constrained by hub and touches the stator only at the
teeth. The optical image of the assembled design S motor is
shown in Fig. 9(b).
A. Experimental Methods
The motors were actuated wirelessly using two sets of coils.
Two coils with a diameter of 12.5 cm, shown in Fig. 10, were
connected to a dc power supply providing constant current to
provide a dc magnetic field. This arrangement was used to bias
the magnetoelastic material into an operating region in which
the strain is sensitive to magnetic field. For design M, based
on the relationship between the magnetostriction coefficient and
the applied field for Metglas 2826MB [24], the dc bias field
was set to 3 Oe, with ±2-Oe ac field (Fig. 11). However, the
effective magnetostriction curve appears to be shifted for design
S, as suggested by the magnetic bias at which the rotation rate
is highest. The cause of bias shift for design S may be related to
residual stresses from the bonding process; the reference curve
Fig. 9. (a) SEM image of the design S stator. (b) Optical image of the
assembled device (b).
Fig. 10. Experimental set up for motor actuation and frequency response
was measured on a free–free sample. Thus, the dc bias field of
design S was set to 8 Oe to achieve a high rotation rate.
Another two coils with a diameter of 11.5 cm were placed
inside the dc coils; these smaller coils generated an alternating
magnetic field with amplitude from 2 to 6 Oe. A dc current,
equal to the measured ac current amplitude, was first applied on
the ac coils. Subsequently, the dc magnetic field, equivalent to
the ac magnetic field amplitude, was experimentally measured
by an FW Bell 5180 Hall Effect Gaussmeter (Pacific Scientific
OECO, Milwaukie, OR, USA). The device under test was
placed between the coils.
To measure the frequency response of the stator, a laser
vibrometer and a network analyzer were used in conjunction
(Fig. 10). The vibration mode shape can be determined by
Fig. 11. Magnetostriction versus applied field for Metglas 2826MB (reproduced from [24]) and rotation rate versus applied dc bias at fixed ac field. The
derivative of the reference curve at a bias point leads to the magnetostrictivity
(“d”) in the constitutive equations in Section II-B at that bias point. The
effective magnetostriction curve appears to be shifted for design S, as suggested
by the magnetic bias at which the rotation rate is highest.
Fig. 12. Experimentally measured frequency response of bulkmagnetoelastic-foil stator of design M, with 3-Oe dc and 2-Oe amplitude ac
magnetic fields.
measuring a number of points along the rim of the stator;
antinodes of a given mode shape exist at locations of strong
response at a given frequency.
A video camera was used to record the motors from above,
and the video is analyzed frame by frame to derive the rotation
rate and initial acceleration. In addition to this method, a laser
displacement sensor was used to monitor the patterned surface
of the rotor. The frequency of the resulting square wave was
used to calculate the rotation rate.
Fig. 13. Optical images of design M rotating with 3-Oe dc and 2-Oe amplitude
ac magnetic fields at a frequency of 11.35 kHz and 9-mg payload.
Fig. 14. Experimentally measured angular velocity of design M as a function
of time with 3-Oe dc and 2-Oe amplitude ac magnetic fields at a frequency
of 11.35 kHz. Linear fits of different trials were used to calculate initial
is ≈2 nN · m. This is calculated by τ = αI, where τ is the
start torque and α is the angular acceleration. The angular
acceleration, before the angular velocity plateaus, was derived
from the average of linear fit slopes from different trials. The
moment of inertia I is further given by I = (1/2)m(r12 + r22 ),
where m is the mass of the rotor and r1 and r2 are the inner and
outer radii of the rotor, respectively.
B. Design M Results
C. Design S Results
The desired mode shape with four antinodes was confirmed
to exist at 11.35 kHz, with 0.2-μm unloaded out-of-plane
deflection at the antinodes. As shown in Fig. 12, the frequency
response is stable over at least 5 min of operation time. In preliminary tests, a typical rotation rate of approximately 44 r/min
(4.6 rad/s) was obtained (Fig. 13). An angular driving step
size of about 23 mdeg is calculated from the measured angular
velocity and resonant frequency. The angular velocities from
different trials were calculated from the recordings and are
shown in Fig. 14.
The associated start torque (i.e., the motor driving torque
minus friction torque when rotation rate is approximately zero)
The frequency response of design S was also measured, and
it was confirmed that two mode shapes exist that can provide
bidirectional operation. The CW mode shape was confirmed
to exist at 6.08 kHz, with about 2-μm unloaded out-of-plane
deflection at the antinodes [Fig. 15(a)], whereas the CCW mode
shape was confirmed to exist at 7.85 kHz, with about 1.8-μm
unloaded out-of-plane deflection at the antinodes [Fig. 15(b)].
The frequencies are lower than the FEA results. Further study
is required to understand the reason for the lower-than-expected
resonant frequency.
The bidirectional rotation was demonstrated at corresponding resonant frequencies. The typical rotation rates of CW and
Fig. 16. Rotation rate measurement of design S with 43-mg payload and 8-Oe
dc and 6-Oe amplitude ac magnetic fields at a frequency of 7.85 kHz.
Fig. 15. Experimentally measured frequency response of design S stator with
no load.
CCW directions are about 30 and 100 r/min when an 22-mg
rotor was used and an ac magnetic field with about 6-Oe
amplitude and a dc magnetic field with 8-Oe amplitude were
applied. However, the CW rotation rate is relatively slow, and
rotor wobbling was observed. This is possibly due to the tooth
locations that are too close to the antinodes of CW vibration
mode shape. Consequently, vertical movement, instead of tangential movement, dominates the motion of the tooth tip so that
the rotor is mainly pushed up and down rather than tangentially.
This problem can be solved by placing the tooth equally far
from the antinodes of both CW and CCW mode shapes in future
designs. More characterization was performed on the CCW
The effect of payload on CCW rotation rate was characterized by using rotors with different weights. For example, for
the 43-mg rotor with six patterned segments used in this test,
a square wave was obtained to calculate the rotation rate of
about 60 r/min (Fig. 16). As shown in Fig. 17, the rotation
rate decreases when the payload increases in an approximate
linear relationship. Design S showed at least 43-mg payload
capability. The rotation rate was also closely related to applied
power or applied ac magnetic field strength. The rotation rates
were measured when the magnitude of the ac magnetic field
increases from 3 to 6 Oe. As shown in Fig. 18, rotation
rate increases with the increasing ac magnetic field in an ap-
Fig. 17. Experimentally obtained effect of payload on rotation rate of design S,
with 8-Oe dc and 6-Oe amplitude ac magnetic fields at a frequency of 7.85 kHz.
Fig. 18. Experimentally obtained effect of applied ac magnetic field on
rotation rate of design S, with 8-Oe dc, with 22-mg payload, and at a frequency
of 7.85 kHz.
proximately linear manner. This indicates that magnetoelastic
material still operates in the intended linear region of magnetostriction versus applied field curve.
The transient response of design S was experimentally evaluated. The results obtained with a 22-mg payload are shown in
Fig. 19. These data were obtained by analyzing a slow-motion
video. The typical performance achieved a rotation rate of about
100 r/min, start torque of 30 nN · m, and step size of 74 mdeg.
Fig. 19. Experimentally measured angular velocity of design S as a function
of time with 8-Oe dc and 6-Oe amplitude ac magnetic fields, a frequency
of 7.85 kHz, and a 22-mg payload. Linear fit was used to calculate initial
The start torque is calculated using the same method for
design M.
The performance of the two designs is summarized and
compared in Table I.
The magnetoelastic wireless resonant motors demonstrate
that the magnetoelastic material has significant potential for
actuation in miniature and microscale devices. The main
advantage—the wireless aspect of the actuation approach—
allows miniaturization and eliminates lead transfer actuator,
which is especially preferred for implantable applications.
Along with being wireless, magnetoelastic resonant motors
offer the advantages of ultrasonic piezoelectric motors, such as
high precision. Design S was able to move at least a 43-mg
payload. These capabilities will enable integration of inertial
sensors on the rotor in the future.
Design M demonstrated the concept of wireless magnetoelastic actuation, but the performance was compromised by two
factors. First, the positioning and assembling of the teeth, which
was performed manually, contributed to a loss of performance.
Second, an uneven magnetoelastic stator surface resulted in
unstable rotation rate. Furthermore, because the customized
teeth design limits vibration mode shapes of the stator, only
CW rotation was achieved. Design S was designed based on
the results of design M. The thick silicon stator has better
surface uniformity and simplifies the tooth design. This resulted
in stable rotation rate and bidirectional operation capability.
Design M also exhibited a larger payload capability. The Au–In
TLP bonding process was customized to attach magnetoelastic
material to silicon. This bonding process has potential applications in other magnetoelastic actuation and sensing systems.
Although integration of the actuation coil with the stator chip
is beyond the scope of the present effort, it is worthwhile to
consider what options exist for generating the magnetic field
on-chip. Utilizing FEA simulations similar to those described
in Section II, the power consumption for design S can be
estimated. The power required to achieve the necessary magnetic field with small coils placed in close proximity to the
motor is about 74 mW. A rectangular copper sheet located
50 μm beneath the magnetoelastic stator carrying 150 mA,
together with a permanent magnetic providing the dc field,
should be sufficient to achieve actuation. In contrast, longdistance wireless control can be realized by compromising the
performance or power efficiency. For example, if utilizing an
experimental setup similar to that in this paper, an ac current
of 5-A amplitude provides a 6-Oe amplitude ac magnetic field
with a 50-turns Helmholtz coil of 40-cm radius (which is equivalent to the wireless distance). A power amplifier is necessary in
this case. In general, the wireless control distance is determined
by performance requirement, power budget, and other factors,
on a case-by-case basis.
Compared to conventional electromagnetic motors, magnetoelastic micromotors have very simple structures and only require a single ac excitation signal. The performance of rotation
rate and step size can be potentially improved by increasing the
resonant frequency of the silicon stator. This can be achieved
by either increasing the thickness of the stator or decreasing the
size of the stator. Additionally, a method for braking the motion
of the rotor can be provided by exploiting bidirectional rotation
capability. For example, CCW rotation can be stopped abruptly
by switching the exciting frequency from CCW to CW with
a calibrated ac magnetic field. Finally, as mentioned before,
external coils can be potentially replaced by on-chip driving
coils for short-range wireless operation.
This paper has presented the analysis, fabrication, and experimental results for two types of magnetoelastically actuated chip-scale rotary motors. Both designs were successfully
wirelessly actuated using external coils. For design M, a new
approach that employs auxiliary teeth and driving teeth was
used to customize the resonant behavior of the stator. A stator of
φ8 mm diameter, fabricated from 25-μm-thick magnetoelastic
foil typically achieved a rotation rate of about 44 r/min, angular
driving step size of about 23 mdeg, and payload of 9 mg. Design
S used a stator that is a sandwich of Si (65 μm thick) and magnetoelastic foil (25 μm thick). An Au–In TLP bonding process
was used to bond bulk-magnetoelastic-foil to the silicon stator.
Bidirectional rotation was realized by switching the applied
frequency and exciting at a slightly different mode shape. The
typical performance achieved by an 8-mm-diameter stator with
a 22-mg payload was a rotation rate of about 100 r/min, start
torque of 30 nN · m, and driving step size of 74 mdeg. Design S
also showed at least 43-mg payload capability. Future efforts
will be directed at increasing the rotation rate, improving the
angular resolution, and integrating driving and control circuits.
In the long term, such motors can be potentially used in calibrating microgyroscopes or for robotic navigation in tubes or pipes.
The authors would like to thank Prof. K. Grosh from the Department of Mechanical Engineering, University of Michigan,
Ann Arbor, MI, USA, for providing access to a laser vibrometer
used in testing, and Metglas, Inc., for the samples provided for
this paper.
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Jun Tang received the B.S. degree in electronic
science and technology from Huazhong University
of Science and Technology, Wuhan, China, in 2007
and the M.S. degree in microelectronics and solidstate electronics from Shanghai Jiao Tong University,
Shanghai, China, in 2010. He is currently working
toward the Ph.D. degree in the Department of Mechanical Engineering, University of Michigan, Ann
Arbor, MI, USA, with a focus on microelectromechanical systems.
He is currently a Graduate Student Research
Assistant with the Department of Mechanical Engineering, University of
Michigan. His research interests include wireless magnetoelastic sensors and
Scott R. Green received the B.S. degree in mechanical engineering from Rose–Hulman Institute
of Technology, Terre Haute, IN, USA, in 2003 and
the M.S. and Ph.D. degrees in mechanical engineering from the University of Michigan, Ann Arbor,
MI, USA, with a focus on microelectromechanical
He was a Senior Design Engineer with the Instruments Division, Stryker Corporation, from 2003
to 2005, developing minimally invasive surgical devices for vertebroplasty and discectomy procedures.
He is currently an Assistant Research Scientist with the University of Michigan.
His research interests include wireless magnetoelastic sensors and actuators,
miniature implantable medical systems, and miniature and microfabricated
high-vacuum sputter ion pumps.
Yogesh B. Gianchandani (S’83–M’85–SM’05–
F’10) received the B.S. degree in electrical engineering from the University of California, Irvine, CA,
USA, in 1984, the M.S. degree in electrical engineering from the University of California, Los Angeles,
CA, USA, in 1986, and the Ph.D. degree in electrical
engineering from the University of Michigan, Ann
Arbor, MI, USA, in 1994.
He is currently a Professor with the University of
Michigan with a primary appointment in the Department of Electrical Engineering and Computer Science and a courtesy appointment in the Department of Mechanical Engineering.
He also serves as the Director of the Center for Wireless Integrated MicroSensing and Systems (WIMS2 ). He has published more than 250 papers in journals
and conference proceedings and has about 35 U.S. patents issued or pending.
He was a Chief Coeditor of Comprehensive Microsystems: Fundamentals,
Technology, and Applications, published in 2008. He serves several journals
as an Editor or a Member of the Editorial Board. His research interests include
all aspects of design, fabrication, and packaging of micromachined sensors and
actuators and their interface circuits.
Dr. Gianchandani served as the General Cochair for the IEEE/ASME International Conference on Microelectromechanical Systems in 2002. From 2007
to 2009, he also served at the National Science Foundation as the Program
Director for Micro and Nano Systems within the Electrical, Communication,
and Cyber Systems Division.
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