ADP2105
1 Amp/1.5 Amp/2 Amp Synchronous,
Step-Down DC-to-DC Converters
ADP2105/ADP2106/ADP2107
FEATURES
GENERAL DESCRIPTION
Extremely high 97% efficiency
Ultralow quiescent current: 20 μA
1.2 MHz switching frequency
0.1 μA shutdown supply current
Maximum load current
ADP2105: 1 A
ADP2106: 1.5 A
ADP2107: 2 A
Input voltage: 2.7 V to 5.5 V
Output voltage: 0.8 V to VIN
Maximum duty cycle: 100%
Smoothly transitions into low dropout (LDO) mode
Internal synchronous rectifier
Small 16-lead 4 mm × 4 mm LFCSP_VQ package
Optimized for small ceramic output capacitors
Enable/shutdown logic input
Undervoltage lockout
Soft start
The ADP2105/ADP2106/ADP2107 are low quiescent current,
synchronous, step-down dc-to-dc converters in a compact 4 mm ×
4 mm LFCSP_VQ package. At medium to high load currents,
these devices use a current mode, constant frequency pulsewidth modulation (PWM) control scheme for excellent stability
and transient response. To ensure the longest battery life in portable
applications, the ADP2105/ADP2106/ADP2107 use a pulse
frequency modulation (PFM) control scheme under light load
conditions that reduces switching frequency to save power.
The ADP2105/ADP2106/ADP2107 run from input voltages of
2.7 V to 5.5 V, allowing single Li+/Li− polymer cell, multiple
alkaline/NiMH cells, PCMCIA, and other standard power sources.
The output voltage of ADP2105/ADP2106/ADP2107 is adjustable
from 0.8 V to the input voltage (indicated by ADJ), whereas the
ADP2105/ADP2106/ADP2107 are available in preset output
voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V (indicated by x.x V).
Each of these variations is available in three maximum current
levels: 1 A (ADP2105), 1.5 A (ADP2106), and 2 A (ADP2107). The
power switch and synchronous rectifier are integrated for minimal
external part count and high efficiency. During logic controlled
shutdown, the input is disconnected from the output, and it
draws less than 0.1 μA from the input source. Other key features
include undervoltage lockout to prevent deep battery discharge
and programmable soft start to limit inrush current at startup.
APPLICATIONS
Mobile handsets
PDAs and palmtop computers
Telecommunication/networking equipment
Set top boxes
Audio/video consumer electronics
TYPICAL OPERATING CIRCUIT
0.1μF
VIN
10Ω
100
VIN = 3.3V
INPUT VOLTAGE = 2.7V TO 5.5V
VOUT = 2.5V
VIN = 3.6V
95
10μF
OFF
16
15
14
13
FB
GND
IN
PWIN1
OUTPUT VOLTAGE = 2.5V
LX2 12
1 EN
2μH
2 GND
PGND 11
ADP2107-ADJ
85kΩ
LX1 10
3 GND
PWIN2 9
4 GND
5
70kΩ
SS
40kΩ
AGND NC
6
7
8
80
FB
VIN
VIN = 5V
85
4.7μF
10μF
LOAD
0A TO 2A
75
1nF
0
200
400
600
800
1000 1200 1400 1600 1800 2000
LOAD CURRENT (mA)
120pF
NC = NO CONNECT
06079-002
COMP
10μF
90
06079-001
ON
EFFICIENCY (%)
FB
Figure 2. Efficiency vs. Load Current for the ADP2107 with VOUT = 2.5 V
Figure 1. Circuit Configuration of ADP2107 with VOUT = 2.5 V
Rev. C
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rights of third parties that may result from its use. Specifications subject to change without notice. No
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Fax: 781.461.3113 ©2006–2008 Analog Devices, Inc. All rights reserved.
ADP2105/ADP2106/ADP2107
TABLE OF CONTENTS
Features .............................................................................................. 1
External Component Selection ................................................ 16
Applications ....................................................................................... 1
Setting the Output Voltage ........................................................ 16
General Description ......................................................................... 1
Inductor Selection ...................................................................... 17
Typical Operating Circuit ................................................................ 1
Output Capacitor Selection....................................................... 18
Revision History ............................................................................... 2
Input Capacitor Selection .......................................................... 18
Functional Block Diagram .............................................................. 3
Input Filter................................................................................... 19
Specifications..................................................................................... 4
Soft Start Period.......................................................................... 19
Absolute Maximum Ratings............................................................ 6
Loop Compensation .................................................................. 19
Thermal Resistance ...................................................................... 6
Bode Plots .................................................................................... 20
Boundary Condition .................................................................... 6
Load Transient Response .......................................................... 21
ESD Caution .................................................................................. 6
Efficiency Considerations ......................................................... 22
Pin Configuration and Function Descriptions ............................. 7
Thermal Considerations............................................................ 22
Typical Performance Characteristics ............................................. 8
Design Example .............................................................................. 24
Theory of Operation ...................................................................... 14
External Component Recommendations .................................... 25
Control Scheme .......................................................................... 14
Circuit Board Layout Recommendations ................................... 27
PWM Mode Operation.............................................................. 14
Evaluation Board ............................................................................ 28
PFM Mode Operation................................................................ 14
Evaluation Board Schematic for ADP2107 (1.8 V) ............... 28
Pulse-Skipping Threshold ......................................................... 14
100% Duty Cycle Operation (LDO Mode) ............................. 14
Recommended PCB Board Layout (Evaluation Board Layout)
....................................................................................................... 28
Slope Compensation .................................................................. 15
Application Circuits ....................................................................... 30
Design Features ........................................................................... 15
Outline Dimensions ....................................................................... 33
Applications Information .............................................................. 16
Ordering Guide .......................................................................... 33
REVISION HISTORY
9/08—Rev. B to Rev. C
Changes to Table Summary Statement .......................................... 4
Changes to LX Minimum On-Time Parameter, Table 1 ............. 5
7/08—Rev. A to Rev. B
Changes to General Description Section ...................................... 1
Changes to Figure 3 .......................................................................... 3
Changes to Table 1 ............................................................................ 4
Changes to Table 2 ............................................................................ 6
Changes to Figure 4 .......................................................................... 7
Changes to Table 4 ............................................................................ 7
Changes to Figure 26 ...................................................................... 11
Changes to Figure 31 Through Figure 34 .................................... 12
Changes to Figure 35 ...................................................................... 13
Changes to PMW Mode Operation Section and Pulse Skipping
Threshold Section ........................................................................... 14
Changes to Slope Compensation Section .................................... 15
Changes to Setting the Output Voltage Section ........................ 16
Changes to Figure 37 ...................................................................... 16
Changes to Inductor Selection Section........................................ 17
Changes to Input Capacitor Selection Section ........................... 18
Changes to Figure 47 through Figure 52 ..................................... 21
Changes to Transition Losses Section and Thermal
Considerations Section .................................................................. 22
Changes to Table 11 ....................................................................... 25
Changes to Circuit Board Layout Recommendations Section..27
Changes to Table 12 ....................................................................... 26
Changes to Figure 53...................................................................... 28
Changes to Figure 56 Through Figure 57.................................... 30
Changes to Figure 58 Through Figure 59.................................... 31
Changes to Outline Dimensions .................................................. 33
3/07—Rev. 0 to Rev. A
Updated Format .................................................................. Universal
Changes to Output Characteristics and
LX (Switch Node) Characteristics Sections ...................................3
Changes to Typical Performance Characteristics Section ...........7
Changes to Load Transient Response Section ............................ 21
7/06—Revision 0: Initial Version
Rev. C | Page 2 of 36
ADP2105/ADP2106/ADP2107
FUNCTIONAL BLOCK DIAGRAM
COMP 5
SS 6
14 IN
SOFT
START
9 PWIN2
REFERENCE
0.8V
CURRENT SENSE
AMPLIFIER
13 PWIN1
FB 16
FB 16
GM ERROR
AMP
PWM/
PFM
CONTROL
AGND 7
GND 2
FOR PRESET
VOLTAGE
OPTIONS ONLY
DRIVER
AND
ANTISHOOT
THROUGH
GND 3
GND 4
CURRENT
LIMIT
10 LX1
12 LX2
SLOPE
COMPENSATION
NC 8
GND 15
OSCILLATOR
ZERO CROSS
COMPARATOR
THERMAL
SHUTDOWN
11 PGND
06079-037
EN 1
Figure 3.
Rev. C | Page 3 of 36
ADP2105/ADP2106/ADP2107
SPECIFICATIONS
VIN = 3.6 V @ TA = 25°C, unless otherwise noted. 1
Table 1.
Parameter
INPUT CHARACTERISTICS
Input Voltage Range
Undervoltage Lockout Threshold
Min
Typ
2.7
Max
Unit
Conditions
5.5
V
V
V
V
V
mV
−40°C ≤ TJ ≤ +125°C
VIN rising
VIN rising, −40°C ≤ TJ ≤ +125°C
VIN falling
VIN falling, −40°C ≤ TJ ≤ +125°C
VIN falling
V
V
V
2.4
2.2
2.6
2.2
2.0
Undervoltage Lockout Hysteresis 2
OUTPUT CHARACTERISTICS
Output Regulation Voltage
2.5
200
3.267
3.3
3.3
3.201
1.782
1.8
1.8
1.5
1.5
1.2
1.2
0.4
0.5
0.6
0.1
0.1
Line Regulation 3
Output Voltage Range
FEEDBACK CHARACTERISTICS
FB Regulation Voltage
FB Bias Current
0.33
0.3
VIN
%/A
%/A
%/A
%/V
%/V
V
V
V
μA
μA
μA
μA
μA
μA
μA
μA
μA
ADJ
ADJ, −40°C ≤ TJ ≤ +125°C
ADJ, −40°C ≤ TJ ≤ +125°C
1.2 V output voltage
1.2 V output voltage, −40°C ≤ TJ ≤ +125°C
1.5 V output voltage
1.5 V output voltage, −40°C ≤ TJ ≤ +125°C
1.8 V output voltage
1.8 V output voltage, −40°C ≤ TJ ≤ +125°C
3.3 V output voltage
3.3 V output voltage, −40°C ≤ TJ ≤ +125°C
1.515
1.545
1.164
Load Regulation
1.236
V
V
V
1.818
1.854
1.455
1.188
3.3 V, load = 10 mA
3.3 V, VIN = 3.6 V to 5.5 V, no load to full load
3.3 V, VIN = 3.6 V to 5.5 V, no load to full load,
−40°C ≤ TJ ≤ +125°C
1.8 V, load = 10 mA
1.8 V, VIN = 2.7 V to 5.5 V, no load to full load
1.8 V, VIN = 2.7 V to 5.5 V, no load to full load,
−40°C ≤ TJ ≤ +125°C
1.5, load = 10 mA
ADP210x-1.5 V, VIN = 2.7 V to 5.5 V, no load to full load
ADP210x-1.5 V, VIN = 2.7 V to 5.5 V, no load to full load,
−40°C ≤ TJ ≤ +125°C
1.2 V, load = 10 mA
1.2 V, VIN = 2.7 V to 5.5 V, no load to full load
1.2 V, VIN = 2.7 V to 5.5 V, no load to full load,
−40°C ≤ TJ ≤ +125°C
ADP2105
ADP2106
ADP2107
ADP2105, measured in servo loop
ADP2106 and ADP2107, measured in servo loop
ADJ
3.399
1.746
1.485
3.333
0.8
1.212
0.8
0.784
−0.1
0.816
+0.1
3
6
4
8
5
10
10
20
V
V
V
V
V
V
Rev. C | Page 4 of 36
ADP2105/ADP2106/ADP2107
Parameter
INPUT CURRENT CHARACTERISTICS
IN Operating Current
Min
Typ
Max
Unit
Conditions
30
μA
μA
μA
30
μA
1
μA
ADP210x(ADJ), VFB = 0.9 V
ADP210x(ADJ), VFB = 0.9 V, −40°C ≤ TJ ≤ +125°C
ADP210x(x.x V) output voltage 10% above regulation
voltage
ADP210x(x.x V) output voltage 10% above regulation
voltage, −40°C ≤ TJ ≤ +125°C
VEN = 0 V
20
20
IN Shutdown Current4
LX (SWITCH) NODE CHARACTERISTICS
LX On Resistance 4
0.1
190
165
mΩ
mΩ
mΩ
mΩ
230
mΩ
mΩ
140
mΩ
mΩ
270
100
160
90
LX Leakage Current4, 5
LX Peak Current Limit5
0.1
2.9
2.6
1
3.3
2.25
2.0
2.6
1.5
1.3
LX Minimum On-Time
ENABLE CHARACTERISTICS
EN Input High Voltage
EN Input Low Voltage
EN Input Leakage Current
1.8
110
2
0.4
−0.1
−1
OSCILLATOR FREQUENCY
SOFT START PERIOD
THERMAL CHARACTERISTICS
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
COMPENSATOR
TRANSCONDUCTANCE (gm)
CURRENT SENSE AMPLIFIER GAIN (GCS)2
+1
1.2
1
750
1000
1.4
1200
μA
A
A
A
A
A
A
ns
P-channel switch, ADP2105
P-channel switch, ADP2105, −40°C ≤ TJ ≤ +125°C
P-channel switch, ADP2106 and ADP2107
P-channel switch, ADP2106 and ADP2107,
−40°C ≤ TJ ≤ +125°C
N-channel synchronous rectifier, ADP2105
N-channel synchronous rectifier, ADP2105,
−40°C ≤ TJ ≤ +125°C
N-channel synchronous rectifier, ADP2106 and ADP2107
N-channel synchronous rectifier, ADP2106 and ADP2107,
−40°C ≤ TJ ≤ +125°C
VIN = 5.5 V, VLX = 0 V, 5.5 V
P-channel switch, ADP2107
P-channel switch, ADP2107, −40°C ≤ TJ ≤ +125°C
P-channel switch, ADP2106
P-channel switch, ADP2106, −40°C ≤ TJ ≤ +125°C
P-channel switch, ADP2105
P-channel switch, ADP2105, −40°C ≤ TJ ≤ +125°C
In PWM mode of operation, −40°C ≤ TJ ≤ +125°C
V
V
μA
μA
MHz
MHz
μs
VIN = 2.7 V to 5.5 V, −40°C ≤ TJ ≤ +125°C
VIN = 2.7 V to 5.5 V, −40°C ≤ TJ ≤ +125°C
VIN = 5.5 V, VEN = 0 V, 5.5 V
VIN = 5.5 V, VEN = 0 V, 5.5 V, −40°C ≤ TJ ≤ +125°C
VIN = 2.7 V to 5.5 V
VIN = 2.7 V to 5.5 V, −40°C ≤ TJ ≤ +125°C
CSS = 1 nF
50
°C
°C
μA/V
1.875
2.8125
3.625
A/V
A/V
A/V
140
40
1
ADP2105
ADP2106
ADP2107
All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). Typical values are at TA = 25°C.
Guaranteed by design.
The ADP2105/ADP2106/ADP2107 line regulation was measured in a servo loop on the automated test equipment that adjusts the feedback voltage to achieve a
specific COMP voltage.
4
All LX (switch) node characteristics are guaranteed only when the LX1 pin and LX2 pin are tied together.
5
These specifications are guaranteed from −40°C to +85°C.
2
3
Rev. C | Page 5 of 36
ADP2105/ADP2106/ADP2107
ABSOLUTE MAXIMUM RATINGS
THERMAL RESISTANCE
Table 2.
Parameter
IN, EN, SS, COMP, FB to AGND
LX1, LX2 to PGND
PWIN1, PWIN2 to PGND
PGND to AGND
GND to AGND
PWIN1, PWIN2 to IN
Operating Junction Temperature Range
Storage Temperature Range
Soldering Conditions
Rating
−0.3 V to +6 V
−0.3 V to (VIN + 0.3 V)
−0.3 V to +6 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
−40°C to +125°C
−65°C to +150°C
JEDEC J-STD-020
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
θJA is specified for the worst-case conditions, that is, a device
soldered in a circuit board for surface-mount packages.
Table 3. Thermal Resistance
Package Type
16-Lead LFCSP_VQ/QFN
Maximum Power Dissipation
θJA
40
1
Unit
°C/W
W
BOUNDARY CONDITION
Natural convection, 4-layer board, exposed pad soldered to the PCB.
ESD CAUTION
Rev. C | Page 6 of 36
ADP2105/ADP2106/ADP2107
13 PWIN1
14 IN
15 GND
16 FB
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
12 LX2
EN 1
11 PGND
10 LX1
9
PWIN2
NC = NO CONNECT
06079-003
NC 8
COMP 5
AGND 7
GND 4
TOP VIEW
(Not to Scale)
GND 2
SS 6
GND 3
ADP2105/
ADP2106/
ADP2107
Figure 4. Pin Configuration
Table 4. Pin Function Descriptions
Pin No.
1
Mnemonic
EN
2, 3, 4, 15
GND
5
COMP
6
SS
7
AGND
8
9, 13
NC
PWIN2,
PWIN1
10, 12
LX1, LX2
11
PGND
14
IN
16
FB
Description
Enable Input. Drive EN high to turn on the device. Drive EN low to turn off the device and reduce the input
current to 0.1 μA.
Test Pins. These pins are used for internal testing and are not ground return pins. These pins are to be tied to the
AGND plane as close as possible to the ADP2105/ADP2106/ADP2107.
Feedback Loop Compensation Node. COMP is the output of the internal transconductance error amplifier. Place
a series RC network from COMP to AGND to compensate the converter. See the Loop Compensation section.
Soft Start Input. Place a capacitor from SS to AGND to set the soft start period. A 1 nF capacitor sets a 1 ms soft
start period.
Analog Ground. Connect the ground of the compensation components, the soft start capacitor, and the voltage
divider on the FB pin to the AGND pin as close as possible to the ADP2105/ ADP2106/ADP2107. The AGND is
also to be connected to the exposed pad of ADP2105/ADP2106/ADP2107.
No Connect. This is not internally connected and can be connected to other pins or left unconnected.
Power Source Inputs. The source of the PFET high-side switch. Bypass each PWIN pin to the nearest PGND plane with a
4.7 μF or greater capacitor as close as possible to the ADP2105/ADP2106/ ADP2107. See the Input Capacitor
Selection section.
Switch Outputs. The drain of the P-channel power switch and N-channel synchronous rectifier. These pins are to
be tied together and connected to the output LC filter between LX and the output voltage.
Power Ground. Connect the ground return of all input and output capacitors to the PGND pin using a power
ground plane as close as possible to the ADP2105/ADP2106/ADP2107. The PGND is then to be connected to the
exposed pad of the ADP2105/ADP2106/ADP2107.
Power Input. The power source for the ADP2105/ADP2106/ADP2107 internal circuitry. Connect IN and PWIN1
with a 10 Ω resistor as close as possible to the ADP2105/ADP2106/ADP2107. Bypass IN to AGND with a 0.1 μF or
greater capacitor. See the Input Filter section.
Output Voltage Sense or Feedback Input. For fixed output versions, connect to the output voltage. For
adjustable versions, FB is the input to the error amplifier. Drive FB through a resistive voltage divider to set the
output voltage. The FB regulation voltage is 0.8 V.
Rev. C | Page 7 of 36
ADP2105/ADP2106/ADP2107
TYPICAL PERFORMANCE CHARACTERISTICS
100
100
95
95
VIN = 3.6V
EFFICIENCY (%)
80
VIN = 4.2V
75
VIN = 5.5V
INDUCTOR: SD14, 2.5µH
DCR: 60mΩ
TA = 25°C
1
VIN = 4.2V
VIN = 5.5V
80
75
65
60
85
10
70
65
1000
100
INDUCTOR: SD3814, 3.3µH
DCR: 93mΩ
TA = 25°C
1
10
LOAD CURRENT (mA)
1000
100
LOAD CURRENT (mA)
Figure 5. Efficiency—ADP2105 (1.2 V Output)
Figure 8. Efficiency—ADP2105 (1.8 V Output)
100
100
VIN = 3.6V
95
95
90
90
VIN = 5.5V
VIN = 4.2V
VIN = 2.7V
VIN = 3.6V
85
EFFICIENCY (%)
EFFICIENCY (%)
06079-086
70
VIN = 2.7V
VIN = 3.6V
90
VIN = 2.7V
85
06079-084
EFFICIENCY (%)
90
85
80
75
80
VIN = 4.2V
75
70
VIN = 5.5V
65
70
1
10
1
10
Figure 6. Efficiency—ADP2105 (3.3 V Output)
10k
Figure 9. Efficiency—ADP2106 (1.2 V Output)
100
100
VIN = 3.6V
95
90
95
90
VIN = 2.7V
85
EFFICIENCY (%)
85
VIN = 4.2V
80
75
VIN = 5.5V
70
65
VIN = 5.5V
80
VIN = 4.2V
75
70
65
60
60
INDUCTOR: D62LCB, 2µH
DCR: 28mΩ
TA = 25°C
55
1
10
100
1k
06079-062
EFFICIENCY (%)
1k
LOAD CURRENT (mA)
LOAD CURRENT (mA)
50
100
06079-008
50
1000
100
INDUCTOR: D62LCB, 2µH
DCR: 28mΩ
TA = 25°C
55
VIN = 3.6V
INDUCTOR: D62LCB, 3.3µH
DCR: 47mΩ
TA = 25°C
55
50
10k
LOAD CURRENT (mA)
1
10
100
1k
LOAD CURRENT (mA)
Figure 7. Efficiency—ADP2106 (1.8 V Output)
Figure 10. Efficiency—ADP2106 (3.3 V Output)
Rev. C | Page 8 of 36
06079-053
60
06079-085
60
INDUCTOR: CDRH5D18, 4.1μH
DCR: 43mΩ
TA = 25°C
65
10k
ADP2105/ADP2106/ADP2107
100
100
95
95
90
VIN = 2.7V
85
EFFICIENCY (%)
80
VIN = 4.2V
75
70
VIN = 5.5V
65
VIN = 4.2V
80
VIN = 5.5V
75
70
65
60
60
55
50
1
10
100
06079-010
INDUCTOR: SD12, 1.2µH
DCR: 37mΩ
TA = 25°C
50
10k
1k
INDUCTOR: D62LCB, 1.5µH
DCR: 21mΩ
TA = 25°C
55
1
10
LOAD CURRENT (mA)
Figure 14. Efficiency—ADP2107 (1.8 V)
1.23
100
95
1.22
VIN = 5.5V
VIN = 4.2V
75
70
65
2.7V, +25°C
3.6V, +25°C
5.5V, +25°C
2.7V, +125°C
3.6V, +125°C
5.5V, +125°C
1.21
1.20
1.19
INDUCTOR: CDRH5D28, 2.5µH
DCR: 13mΩ
TA = 25°C
55
1
10
100
1.18
1.17
0.01
10k
1k
06079-082
60
50
2.7V, –40°C
3.6V, –40°C
5.5V, –40°C
VIN = 3.6V
06079-054
EFFICIENCY (%)
OUTPUT VOLTAGE (V)
90
80
10k
1k
LOAD CURRENT (mA)
Figure 11. Efficiency—ADP2107 (1.2 V)
85
100
06079-063
EFFICIENCY (%)
VIN = 2.7V
90
VIN = 3.6V
85
VIN = 3.6V
0.1
1
10
100
1k
10k
LOAD CURRENT (mA)
LOAD CURRENT (mA)
Figure 12. Efficiency—ADP2107 (3.3 V)
Figure 15. Output Voltage Accuracy—ADP2107 (1.2 V)
1.85
3.38
3.36
3.6V, –40°C
5.5V, –40°C
3.6V, +25°C
5.5V, +25°C
3.6V, +125°C
5.5V, +125°C
OUTPUT VOLTAGE (V)
1.81
1.79
3.34
3.32
3.30
3.28
3.26
1.77
1.75
0.1
1
2.7V, +25°C
3.6V, +25°C
5.5V, +25°C
10
2.7V, +125°C
3.6V, +125°C
5.5V, +125°C
100
1k
3.24
3.22
0.01
10k
LOAD CURRENT (mA)
06079-081
2.7V, –40°C
3.6V, –40°C
5.5V, –40°C
06079-064
OUTPUT VOLTAGE (V)
1.83
0.1
1
10
100
1k
LOAD CURRENT (mA)
Figure 13. Output Voltage Accuracy—ADP2107 (1.8 V)
Figure 16. Output Voltage Accuracy—ADP2107 (3.3 V)
Rev. C | Page 9 of 36
10k
ADP2105/ADP2106/ADP2107
10k
190
SWITCH ON RESISTANCE (mΩ)
1k
+25°C
–40°C
10
1
0.8
1.2
1.6
2.0
2.4
2.8
3.2
3.6
4.0
4.4
4.8
PMOS POWER SWITCH
150
140
130
120
NMOS SYNCHRONOUS RECTIFIER
110
06079-016
+125°C
160
100
2.7
5.2
06079-093
100
170
3.0
3.3
INPUT VOLTAGE (V)
Figure 17. Quiescent Current vs. Input Voltage
SWITCH ON RESISTANCE (mΩ)
FEEDBACK VOLTAGE (V)
5.4
PMOS POWER SWITCH
0.800
0.799
0.798
0.797
06079-017
0.796
–20
0
20
40
60
80
100
100
80
40
20
0
2.7
120 125
NMOS SYNCHRONOUS RECTIFIER
60
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
INPUT VOLTAGE (V)
TEMPERATURE (°C)
Figure 21. Switch On Resistance vs. Input Voltage—ADP2106 and ADP2107
Figure 18. Feedback Voltage vs. Temperature
1260
1.75
1.70
1250
1.60
1.55
ADP2105 (1A)
1.50
1.45
1.40
1.30
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
06079-073
1.35
5.7
1240
1230
+125°C
1220
+25°C
–40°C
1210
1200
1190
2.7
06079-021
SWITCHING FREQUENCY (kHz)
1.65
PEAK CURRENT LIMIT (A)
5.1
120
0.801
1.25
2.7
4.8
Figure 20. Switch On Resistance vs. Input Voltage—ADP2105
0.802
0.795
–40
3.6
3.9
4.2
4.5
INPUT VOLTAGE (V)
06079-018
QUIESCENT CURRENT (µA)
180
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 22. Switching Frequency vs. Input Voltage
Figure 19. Peak Current Limit of ADP2105
Rev. C | Page 10 of 36
5.4
ADP2105/ADP2106/ADP2107
2.35
LX (SWITCH) NODE
2.30
3
ADP2106 (1.5A)
2.20
2.15
INDUCTOR CURRENT
2.05
2.00
1
1.95
OUTPUT VOLTAGE
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
06079-074
TA = 25°C
3.0
CH1 1V
CH3 5V
5.7
INPUT VOLTAGE (V)
Figure 23. Peak Current Limit of ADP2106
2.85
ADP2107 (2A)
2.80
2.75
2.70
2.65
2.55
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
06079-071
2.60
PULSE-SKIPPING THRESHOLD CURRENT (mA)
PEAK CURRENT LIMIT (A)
2.90
120
105
90
VOUT = 1.2V
75
60
45
VOUT = 2.5V
15
0
2.7
5.7
VOUT = 1.8V
30
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
5.7
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 27. Pulse-Skipping Threshold vs. Input Voltage for ADP2105
Figure 24. Peak Current Limit of ADP2107
195
135
120
105
VOUT = 1.2V
90
75
60
VOUT = 2.5V
VOUT = 1.8V
15
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
06079-067
30
5.7
PULSE-SKIPPING THRESHOLD CURRENT (mA)
150
0
2.7
1.78V
135
2.95
45
A CH1
Figure 26. Short -Circuit Response at Output
3.00
2.50
2.7
M 10µs
T 45.8%
CH4 1AΩ
06079-066
1.85
2.7
4
06079-072
1.90
PULSE-SKIPPING THRESHOLD CURRENT (mA)
Δ: 260mV
@: 3.26V
2.10
180
VOUT = 1.2V
165
150
135
VOUT = 1.8V
120
105
90
VOUT = 2.5V
75
60
45
30
15
0
2.7
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
06079-068
PEAK CURRENT LIMIT (A)
2.25
5.7
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 28. Pulse-Skipping Threshold vs. Input Voltage for ADP2107
Figure 25. Pulse-Skipping Threshold vs. Input Voltage for ADP2106
Rev. C | Page 11 of 36
ADP2105/ADP2106/ADP2107
190
170
160
3
PMOS POWER SWITCH
150
LX (SWITCH) NODE
140
1
130
120
OUTPUT VOLTAGE (AC-COUPLED)
NMOS SYNCHRONOUS RECTIFIER
4
06079-093
110
100
2.7
3.0
3.3
3.6
3.9
4.2
4.5
INPUT VOLTAGE (V)
4.8
5.1
06079-033
SWITCH ON RESISTANCE (mΩ)
180
INDUCTOR CURRENT
CH1 50mV
CH3 2V
5.4
CH4 200mAΩ
M 400ns
T 17.4%
A CH3
3.88V
Figure 32. DCM Mode of Operation at Light Load (100 mA)
Figure 29. Switch On Resistance vs. Temperature—ADP2105
140
LX (SWITCH) NODE
SWITCH ON RESISTANCE (mΩ)
120
PMOS POWER SWITCH
100
3
80
NMOS SYNCHRONOUS RECTIFIER
60
1
0
–40
06079-083
20
–20
0
20
40
60
80
100
06079-034
OUTPUT VOLTAGE (AC-COUPLED)
40
INDUCTOR CURRENT
4
CH1 20mV
CH3 2V
120
JUNCTION TEMPERATURE (°C)
Figure 30. Switch On Resistance vs. Temperature—ADP2106 and ADP2107
CH4 1AΩ
M 2µs
T 13.4%
A CH3
1.84V
Figure 33. Minimum Off Time Control at Dropout
LX (SWITCH) NODE
LX (SWITCH)
NODE
3
3
1
1
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
CH4 200mAΩ
M 2µs
T 6%
A CH3
06079-031
INDUCTOR CURRENT
CH1 50mV
CH3 2V
INDUCTOR CURRENT
06079-030
4
4
3.88V
CH1 20mV
CH3 2V
Figure 31. PFM Mode of Operation at Very Light Load (10 mA)
CH4 1AΩ
M 1µs
T 17.4%
A CH3
3.88V
Figure 34. PWM Mode of Operation at Medium/Heavy Load (1.5 A)
Rev. C | Page 12 of 36
ADP2105/ADP2106/ADP2107
LX (SWITCH) NODE
ENABLE VOLTAGE
3
3
OUTPUT VOLTAGE
CHANNEL 3
FREQUENCY
= 336.6kHz
Δ: 2.86A
@: 2.86A
1
INDUCTOR CURRENT
INDUCTOR CURRENT
OUTPUT VOLTAGE
1
CH1 1V
CH3 5V
CH4 1AΩ
M 4µs
T 45%
A CH3
1.8V
Figure 35. Current Limit Behavior of ADP2107 (Frequency Foldback)
06079-035
4
06079-032
4
CH1 1V
CH3 5V
CH4 500mAΩ
M 400µs
T 20.2%
A CH1
1.84V
Figure 36. Startup and Shutdown Waveform (CSS = 1 nF → SS Time = 1 ms)
Rev. C | Page 13 of 36
ADP2105/ADP2106/ADP2107
THEORY OF OPERATION
The ADP2105/ADP2106/ADP2107 are step-down, dc-to-dc
converters that use a fixed frequency, peak current mode architecture with an integrated high-side switch and low-side synchronous rectifier. The high 1.2 MHz switching frequency and tiny
16-lead, 4 mm × 4 mm LFCSP_VQ package allow for a small stepdown dc-to-dc converter solution. The integrated high-side switch
(P-channel MOSFET) and synchronous rectifier (N-channel
MOSFET) yield high efficiency at medium to heavy loads. Light
load efficiency is improved by smoothly transitioning to variable
frequency PFM mode.
The ADP2105/ADP2106/ADP2107 (ADJ) operate with an input
voltage from 2.7 V to 5.5 V and regulate an output voltage down
to 0.8 V. The ADP2105/ADP2106/ADP2107 are also available with
preset output voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V.
CONTROL SCHEME
The ADP2105/ADP2106/ADP2107 operate with a fixed frequency,
peak current mode PWM control architecture at medium to high
loads for high efficiency, but shift to a variable frequency PFM
control scheme at light loads for lower quiescent current. When
operating in fixed frequency PWM mode, the duty cycle of the
integrated switches is adjusted to regulate the output voltage, but
when operating in PFM mode at light loads, the switching
frequency is adjusted to regulate the output voltage.
The ADP2105/ADP2106/ADP2107 operate in the PWM mode
only when the load current is greater than the pulse-skipping
threshold current. At load currents below this value, the converter
smoothly transitions to the PFM mode of operation.
PWM MODE OPERATION
In PWM mode, the ADP2105/ADP2106/ADP2107 operate at a
fixed frequency of 1.2 MHz set by an internal oscillator. At the
start of each oscillator cycle, the P-channel MOSFET switch is
turned on, putting a positive voltage across the inductor. Current
in the inductor increases until the current sense signal crosses
the peak inductor current level that turns off the P-channel
MOSFET switch and turns on the N-channel MOSFET synchronous rectifier. This puts a negative voltage across the inductor,
causing the inductor current to decrease. The synchronous
rectifier stays on for the remainder of the cycle, unless the
inductor current reaches zero, which causes the zero-crossing
comparator to turn off the N-channel MOSFET. The peak
inductor current is set by the voltage on the COMP pin. The
COMP pin is the output of a transconductance error amplifier that
compares the feedback voltage with an internal 0.8 V reference.
PFM MODE OPERATION
The ADP2105/ADP2106/ADP2107 smoothly transition to the
variable frequency PFM mode of operation when the load current
decreases below the pulse skipping threshold current, switching
only as necessary to maintain the output voltage within regulation.
When the output voltage dips below regulation, the ADP2105/
ADP2106/ADP2107 enter PWM mode for a few oscillator cycles
to increase the output voltage back to regulation. During the wait
time between bursts, both power switches are off, and the output
capacitor supplies all the load current. Because the output voltage
dips and recovers occasionally, the output voltage ripple in this
mode is larger than the ripple in the PWM mode of operation.
PULSE-SKIPPING THRESHOLD
The output current at which the ADP2105/ADP2106/ADP2107
transition from variable frequency PFM control to fixed frequency
PWM control is called the pulse-skipping threshold. The pulseskipping threshold is optimized for excellent efficiency over all
load currents. The variation of pulse-skipping threshold with
input voltage and output voltage is shown in Figure 25, Figure 27,
and Figure 28.
100% DUTY CYCLE OPERATION (LDO MODE)
As the input voltage drops, approaching the output voltage, the
ADP2105/ADP2106/ADP2107 smoothly transition to 100%
duty cycle, maintaining the P-channel MOSFET switch-on continuously. This allows the ADP2105/ADP2106/ADP2107 to regulate
the output voltage until the drop in input voltage forces the Pchannel MOSFET switch to enter dropout, as shown in the
following equation:
VIN(MIN) = IOUT × (RDS(ON) − P + DCRIND) + VOUT(NOM)
The ADP2105/ADP2106/ADP2107 achieve 100% duty cycle
operation by stretching the P-channel MOSFET switch-on time
if the inductor current does not reach the peak inductor current
level by the end of the clock cycle. When this happens, the oscillator remains off until the inductor current reaches the peak
inductor current level, at which time the switch is turned off and
the synchronous rectifier is turned on for a fixed off time. At
the end of the fixed off time, another cycle is initiated. As the
ADP2105/ADP2106/ADP2107 approach dropout, the switching
frequency decreases gradually to smoothly transition to 100%
duty cycle operation.
Rev. C | Page 14 of 36
ADP2105/ADP2106/ADP2107
SLOPE COMPENSATION
Short-Circuit Protection
Slope compensation stabilizes the internal current control loop
of the ADP2105/ADP2106/ADP2107 when operating beyond
50% duty cycle to prevent subharmonic oscillations. It is implemented by summing a fixed, scaled voltage ramp to the current
sense signal during the on-time of the P-channel MOSFET switch.
The ADP2105/ADP2106/ADP2107 include frequency foldback
to prevent output current runaway on a hard short. When the
voltage at the feedback pin falls below 0.3 V, indicating the possibility of a hard short at the output, the switching frequency is
reduced to 1/4 of the internal oscillator frequency. The reduction
in the switching frequency results in more time for the inductor to
discharge, preventing a runaway of output current.
The slope compensation ramp value determines the minimum
inductor that can be used to prevent subharmonic oscillations
at a given output voltage. For slope compensation ramp values,
see Table 5. For more information see the Inductor Selection
section.
Table 5. Slope Compensation Ramp Values
Part
ADP2105
ADP2106
ADP2107
Slope Compensation Ramp Values
0.72 A/μs
1.07 A/μs
1.38 A/μs
Undervoltage Lockout (UVLO)
To protect against deep battery discharge, UVLO circuitry is
integrated on the ADP2105/ADP2106/ADP2107. If the
input voltage drops below the 2.2 V UVLO threshold, the
ADP2105/ADP2106/ADP2107 shut down, and both the power
switch and synchronous rectifier turn off. When the voltage
again rises above the UVLO threshold, the soft start period is
initiated, and the part is enabled.
Thermal Protection
DESIGN FEATURES
Drive EN high to turn on the ADP2105/ADP2106/ADP2107.
Drive EN low to turn off the ADP2105/ADP2106/ADP2107,
reducing the input current below 0.1 μA. To force the
ADP2105/ADP2106/ADP2107 to automatically start when
input power is applied, connect EN to IN. When shut down, the
ADP2105/ADP2106/ADP2107 discharge the soft start capacitor,
causing a new soft start cycle every time they are re-enabled.
In the event that the ADP2105/ADP2106/ADP2107 junction
temperatures rise above 140°C, the thermal shutdown circuit turns
off the converter. Extreme junction temperatures can be the
result of high current operation, poor circuit board design, and/or
high ambient temperature. A 40°C hysteresis is included so that
when thermal shutdown occurs, the ADP2105/ADP2106/
ADP2107 do not return to operation until the on-chip temperature drops below 100°C. When coming out of thermal shutdown,
soft start is initiated.
Synchronous Rectification
Soft Start
In addition to the P-channel MOSFET switch, the ADP2105/
ADP2106/ADP2107 include an integrated N-channel MOSFET
synchronous rectifier. The synchronous rectifier improves efficiency, especially at low output voltage, and reduces cost and
board space by eliminating the need for an external rectifier.
The ADP2105/ADP2106/ADP2107 include soft start circuitry
to limit the output voltage rise time to reduce inrush current at
startup. To set the soft start period, connect the soft start capacitor
(CSS) from SS to AGND. When the ADP2105/ADP2106/ADP2107
are disabled, or if the input voltage is below the undervoltage
lockout threshold, CSS is internally discharged. When the
ADP2105/ADP2106/ADP2107 are enabled, CSS is charged through
an internal 0.8 μA current source, causing the voltage at SS to rise
linearly. The output voltage rises linearly with the voltage at SS.
Enable/Shutdown
Current Limit
The ADP2105/ADP2106/ADP2107 have protection circuitry to
limit the direction and amount of current flowing through the
power switch and synchronous rectifier. The positive current
limit on the power switch limits the amount of current that can
flow from the input to the output, and the negative current limit
on the synchronous rectifier prevents the inductor current from
reversing direction and flowing out of the load.
Rev. C | Page 15 of 36
ADP2105/ADP2106/ADP2107
APPLICATIONS INFORMATION
into account when calculating resistor values. The FB bias
current can be ignored for a higher divider string current, but
this degrades efficiency at very light loads.
EXTERNAL COMPONENT SELECTION
The external component selection for the ADP2105/ADP2106/
ADP2107 application circuits shown in Figure 37 and Figure 38
depend on input voltage, output voltage, and load current
requirements. Additionally, trade-offs between performance
parameters like efficiency and transient response can be made
by varying the choice of external components.
To limit output voltage accuracy degradation due to FB bias
current to less than 0.05% (0.5% maximum), ensure that the
divider string current is greater than 20 μA. To calculate the
desired resistor values, first determine the value of the bottom
divider string resistor (RBOT) using the following equation:
SETTING THE OUTPUT VOLTAGE
RBOT =
The output voltage of ADP2105/ADP2106/ADP2107(ADJ) is
externally set by a resistive voltage divider from the output
voltage to FB. The ratio of the resistive voltage divider sets the
output voltage, and the absolute value of those resistors sets the
divider string current. For lower divider string currents, the
small 10 nA (0.1 μA maximum) FB bias current is to be taken
0.1μF
where:
VFB = 0.8 V, the internal reference.
ISTRING is the resistor divider string current.
VIN
10Ω
VFB
I STRING
INPUT VOLTAGE = 2.7V TO 5.5V
CIN1
VOUT
16
15
FB
ON
OFF
1
14
13
GND IN
PWIN1
LX2 12
EN
OUTPUT VOLTAGE = 1.2V, 1.5V, 1.8V, 3.3V
L
2
GND
3
GND
4
GND
ADP2105/
ADP2106/
ADP2107
COUT
LX1 10
PWIN2 9
COMP
SS
5
6
VOUT
PGND 11
VIN
CIN2
AGND NC
7
LOAD
8
CSS
RCOMP
06079-065
CCOMP
NC = NO CONNECT
Figure 37. Typical Applications Circuit for Fixed Output Voltage Options of ADP2105/ADP2106/ADP2107(x.x V)
0.1μF
VIN
10Ω
INPUT VOLTAGE = 2.7V TO 5.5V
CIN1
FB
OFF
16
15
14
13
FB
GND
IN
PWIN1
LX2 12
1
EN
2
GND
3
GND
4
GND
OUTPUT VOLTAGE
= 0.8V TO VIN
L
ADP2105/
ADP2106/
ADP2107
SS
5
6
RTOP
LX1 10
PWIN2 9
COMP
RCOMP
PGND 11
VIN
AGND NC
7
COUT
LOAD
FB
CIN2
RBOT
8
CSS
CCOMP
NC = NO CONNECT
06079-038
ON
Figure 38. Typical Applications Circuit for Adjustable Output Voltage Option of ADP2105/ADP2106/ADP2107(ADJ)
Rev. C | Page 16 of 36
ADP2105/ADP2106/ADP2107
When RBOT is determined, calculate the value of the top resistor
(RTOP) by using the following equation:
Ensure that the maximum rms current of the inductor is greater
than the maximum load current and that the saturation current
of the inductor is greater than the peak current limit of the
converter used in the application.
⎡V − VFB ⎤
RTOP = RBOT ⎢ OUT
⎥
⎣ VFB
⎦
The ADP2105/ADP2106/ADP2107(x.x V) include the resistive
voltage divider internally, reducing the external circuitry required.
For improved load regulation, connect the FB to the output
voltage as close as possible to the load.
INDUCTOR SELECTION
The high switching frequency of ADP2105/ADP2106/ADP2107
allows for minimal output voltage ripple even with small inductors.
The sizing of the inductor is a trade-off between efficiency and
transient response. A small inductor leads to larger inductor
current ripple that provides excellent transient response but
degrades efficiency. Due to the high switching frequency of
ADP2105/ADP2106/ADP2107, shielded ferrite core inductors
are recommended for their low core losses and low electromagnetic
interference (EMI).
As a guideline, the inductor peak-to-peak current ripple (ΔIL) is
typically set to 1/3 of the maximum load current for optimal
transient response and efficiency, as shown in the following
equations:
V
× (V IN − VOUT ) I LOAD (MAX )
ΔI L = OUT
≈
V IN × f SW × L
3
⇒ LIDEAL =
2.5 × VOUT × (VIN − VOUT )
μH
VIN × I LOAD ( MAX )
where fSW is the switching frequency (1.2 MHz).
The ADP2105/ADP2106/ADP2107 use slope compensation in
the current control loop to prevent subharmonic oscillations
when operating beyond 50% duty cycle. The fixed slope compensation limits the minimum inductor value as a function of
output voltage.
For the ADP2105
L > (1.12 μH/V) × VOUT
Table 6. Minimum Inductor Value for Common Output
Voltage Options for the ADP2105 (1 A)
VOUT
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
2.7 V
1.67 μH
1.68 μH
2.02 μH
2.80 μH
3.70 μH
3.6 V
2.00 μH
2.19 μH
2.25 μH
2.80 μH
3.70 μH
VIN
4.2 V
2.14 μH
2.41 μH
2.57 μH
2.80 μH
3.70 μH
5.5 V
2.35 μH
2.73 μH
3.03 μH
3.41 μH
3.70 μH
Table 7. Minimum Inductor Value for Common Output
Voltage Options for the ADP2106 (1.5 A)
VOUT
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
2.7 V
1.11 μH
1.25 μH
1.49 μH
2.08 μH
2.74 μH
3.6 V
2.33 μH
1.46 μH
1.50 μH
2.08 μH
2.74 μH
VIN
4.2 V
2.43 μH
1.61 μH
1.71 μH
2.08 μH
2.74 μH
5.5 V
1.56 μH
1.82 μH
2.02 μH
2.27 μH
2.74 μH
Table 8. Minimum Inductor Value for Common Output
Voltage Options for the ADP2107 (2 A)
VOUT
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
2.7 V
0.83 μH
0.99 μH
1.19 μH
1.65 μH
2.18 μH
3.6 V
1.00 μH
1.09 μH
1.19 μH
1.65 μH
2.18 μH
VIN
4.2 V
1.07 μH
1.21 μH
1.29 μH
1.65 μH
2.18 μH
5.5 V
1.17 μH
1.36 μH
1.51 μH
1.70 μH
2.18 μH
Table 9. Inductor Recommendations for the ADP2105/
ADP2106/ADP2107
For the ADP2106
L > (0.83 μH/V) × VOUT
Vendor
Sumida
For the ADP2107
L > (0.66 μH/V) × VOUT
Toko
Inductors 4.7 μH or larger are not recommended because they
may cause instability in discontinuous conduction mode under
light load conditions. It is also important that the inductor be
capable of handling the maximum peak inductor current (IPK)
determined by the following equation:
Coilcraft
Cooper
Bussmann
⎛ ΔI ⎞
I PK = I LOAD( MAX ) + ⎜ L ⎟
⎝ 2 ⎠
Rev. C | Page 17 of 36
Small-Sized Inductors
(< 5 mm × 5 mm)
CDRH2D14, 3D16,
3D28
1069AS-DB3018,
1098AS-DE2812,
1070AS-DB3020
LPS3015, LPS4012,
DO3314
SD3110, SD3112,
SD3114, SD3118,
SD3812, SD3814
Large-Sized Inductors
(> 5 mm × 5 mm)
CDRH4D18, 4D22,
4D28, 5D18, 6D12
D52LC, D518LC,
D62LCB
DO1605T
SD10, SD12, SD14, SD52
ADP2105/ADP2106/ADP2107
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
15
20
0
–20
1
–40
2
3
–60
–80
0
2
4
06079-060
–100
14.7µF 0805 X5R MURATA GRM21BR61A475K
210µF 0805 X5R MURATA GRM21BR61A106K
322µF 0805 X5R MURATA GRM21BR60J226M
6
VOLTAGE (VDC)
Figure 40. Percentage Drop-In Capacitance vs. DC Bias for Ceramic
Capacitors (Information Provided by Murata Corporation)
For example, to get 20 μF output capacitance at an output voltage
of 2.5 V, based on Figure 40, as well as to give some margin for
temperature variance, a 22 μF and a 10 μF capacitor are to be
used in parallel to ensure that the output capacitance is sufficient
under all conditions for stable behavior.
Table 10. Recommended Input and Output Capacitor
Selection for the ADP2105/ADP2106/ADP2107
06079-070
OVERSHOOT OF OUTPUT VOLTAGE (%)
The output capacitor selection affects both the output voltage ripple
and the loop dynamics of the converter. For a given loop crossover
frequency (the frequency at which the loop gain drops to 0 dB), the
maximum voltage transient excursion (overshoot) is inversely
proportional to the value of the output capacitor. Therefore, larger
output capacitors result in improved load transient response. To
minimize the effects of the dc-to-dc converter switching, the crossover frequency of the compensation loop should be less than 1/10
of the switching frequency. Higher crossover frequency leads to
faster settling time for a load transient response, but it can also
cause ringing due to poor phase margin. Lower crossover
frequency helps to provide stable operation but needs large output
capacitors to achieve competitive overshoot specifications.
Therefore, the optimal crossover frequency for the control loop of
ADP2105/ADP2106/ADP2107 is 80 kHz, 1/15 of the switching
frequency. For a crossover frequency of 80 kHz, Figure 39 shows
the maximum output voltage excursion during a 1 A load transient,
as the product of the output voltage and the output capacitor is
varied. Choose the output capacitor based on the desired load
transient response and target output voltage.
When choosing output capacitors, it is also important to
account for the loss of capacitance due to output voltage dc bias.
Figure 40 shows the loss of capacitance due to output voltage dc
bias for three X5R MLCC capacitors from Murata.
CAPACITANCE CHANGE (%)
OUTPUT CAPACITOR SELECTION
20
25
30
35
40
45
50
55
60
65
70
OUTPUT CAPACITOR × OUTPUT VOLTAGE (μC)
Figure 39. Percentage Overshoot for a 1 A Load Transient Response vs.
Output Capacitor × Output Voltage
For example, if the desired 1 A load transient response (overshoot)
is 5% for an output voltage of 2.5 V, then from Figure 39
Output Capacitor × Output Voltage = 50 μC
⇒ Output Capacitor =
50 μ C
2 .5
≈ 20 μ F
The ADP2105/ADP2106/ADP2107 have been designed for
operation with small ceramic output capacitors that have low
ESR and ESL. Therefore, they are comfortably able to meet tight
output voltage ripple specifications. X5R or X7R dielectrics are
recommended with a voltage rating of 6.3 V or 10 V. Y5V and Z5U
dielectrics are not recommended, due to their poor temperature
and dc bias characteristics. Table 10 shows a list of recommended
MLCC capacitors from Murata and Taiyo Yuden.
Capacitor
4.7 μF, 10 V
X5R 0805
10 μF, 10 V
X5R 0805
22 μF, 6.3 V
X5R 0805
Vendor
Murata
Taiyo Yuden
GRM21BR61A475K
LMK212BJ475KG
GRM21BR61A106K
LMK212BJ106KG
GRM21BR60J226M
JMK212BJ226MG
INPUT CAPACITOR SELECTION
The input capacitor reduces input voltage ripple caused by the
switch currents on the PWIN pins. Place the input capacitors as
close as possible to the PWIN pins. Select an input capacitor
capable of withstanding the rms input current for the maximum
load current in your application.
For the ADP2105, it is recommended that each PWIN pin be
bypassed with a 4.7 μF or larger input capacitor. For the ADP2106,
bypass each PWIN pin with a 10 μF and a 4.7 μF capacitor, and
for the ADP2107, bypass each PWIN pin with a 10 μF capacitor.
As with the output capacitor, a low ESR ceramic capacitor is
recommended to minimize input voltage ripple. X5R or X7R
dielectrics are recommended, with a voltage rating of 6.3 V or
10 V. Y5V and Z5U dielectrics are not recommended due to
their poor temperature and dc bias characteristics. Refer to
Table 10 for input capacitor recommendations.
Rev. C | Page 18 of 36
ADP2105/ADP2106/ADP2107
INPUT FILTER
The IN pin is the power source for the ADP2105/ADP2106/
ADP2107 internal circuitry, including the voltage reference and
current sense amplifier that are sensitive to power supply noise.
To prevent high frequency switching noise on the PWIN pins from
corrupting the internal circuitry of the ADP2105/ADP2106/
ADP2107, a low-pass RC filter should be placed between the IN
pin and the PWIN1 pin. The suggested input filter consists of
a small 0.1 μF ceramic capacitor placed between IN and AGND
and a 10 Ω resistor placed between IN and PWIN1. This forms a
150 kHz low-pass filter between PWIN1 and IN that prevents any
high frequency noise on PWIN1 from coupling into the IN pin.
SOFT START PERIOD
To set the soft start period, connect a soft start capacitor (CSS) from
SS to AGND. The soft start period varies linearly with the size
of the soft start capacitor, as shown in the following equation:
TSS = CSS × 109 ms
The transconductance error amplifier drives the compensation
network that consists of a resistor (RCOMP) and capacitor (CCOMP)
connected in series to form a pole and a zero, as shown in the
following equation:
⎛
1
ZCOMP (s) = ⎜⎜ RCOMP +
sC
COMP
⎝
⎞
⎟
⎟
⎠
At the crossover frequency, the gain of the open loop transfer
function is unity. For the compensation network impedance at
the crossover frequency, this yields the following equation:
⎛ (2π )FCROSS ⎞⎛ COUTVOUT
⎟⎜
ZCOMP (FCROSS ) = ⎜
⎜ G G
⎟⎜ V
m CS
REF
⎝
⎠⎝
⎞
⎟
⎟
⎠
where:
FCROSS = 80 kHz, the crossover frequency of the loop.
COUTVOUT is determined from the Output Capacitor Selection
section.
To ensure that there is sufficient phase margin at the crossover
frequency, place the compensator zero at 1/4 of the crossover
frequency, as shown in the following equation:
For a soft start period of 1 ms, a 1 nF capacitor must be
connected between SS and AGND.
LOOP COMPENSATION
The ADP2105/ADP2106/ADP2107 utilize a transconductance
error amplifier to compensate the external voltage loop. The
open loop transfer function at angular frequency (s) is given by
⎛Z
(s) ⎞⎛ V
H (s) = GmGCS ⎜⎜ COMP ⎟⎟⎜⎜ REF
⎝ sCOUT ⎠⎝ VOUT
⎞ ⎛ 1 + sRCOMP CCOMP
⎟=⎜
⎟ ⎜
sCCOMP
⎠ ⎝
⎞
⎟
⎟
⎠
F
(2 π)⎛⎜ CROSS
⎝ 4
⎞R
⎟ COMP CCOMP = 1
⎠
Solving the three equations in this section simultaneously yields
the value for the compensation resistor and compensation
capacitor, as shown in the following equation:
where:
VREF is the internal reference voltage (0.8 V).
VOUT is the nominal output voltage.
ZCOMP(s) is the impedance of the compensation network at the
angular frequency.
COUT is the output capacitor.
gm is the transconductance of the error amplifier (50 μA/V
nominal).
GCS is the effective transconductance of the current loop.
GCS = 1.875 A/V for the ADP2105.
GCS = 2.8125 A/V for the ADP2106.
GCS = 3.625 A/V for the ADP2107.
Rev. C | Page 19 of 36
⎛ (2 π)FCROSS
RCOMP = 0.8 ⎜⎜
⎝ GmGCS
CCOMP =
2
πFCROSS RCOMP
⎞⎛ COUT VOUT
⎟⎜
⎟⎜ V
REF
⎠⎝
⎞
⎟
⎟
⎠
ADP2105/ADP2106/ADP2107
BODE PLOTS
60
60
ADP2106
ADP2105
50
50
180
0
CROSSOVER
OUTPUT VOLTAGE = 1.8V
FREQUENCY = 87kHz
–10 INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A
–20 INDUCTOR = 2.2µH (LPS4012)
OUTPUT CAPACITOR = 22µF + 22µF
–30 COMPENSATION RESISTOR = 180kΩ
COMPENSATION CAPACITOR = 56pF
–40
10
100
1
FREQUENCY (kHz)
20
10
135
180
CROSSOVER
OUTPUT VOLTAGE = 1.2V
–10
FREQUENCY = 79kHz
INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A
–20 INDUCTOR = 3.3µH (SD3814)
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF
–30 COMPENSATION RESISTOR = 267kΩ
COMPENSATION CAPACITOR = 39pF
–40
1
10
100
FREQUENCY (kHz)
300
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
Figure 44. ADP2105 Bode Plot at VIN = 5.5 V, VOUT = 1.2 V and Load = 1 A
Figure 41. ADP2106 Bode Plot at VIN = 5.5 V, VOUT = 1.8 V and Load = 1 A
60
60
90
LOOP PHASE
0
300
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
45
PHASE
MARGIN = 49°
06079-058
135
LOOP PHASE
LOOP GAIN (dB)
10
90
06079-055
LOOP GAIN (dB)
PHASE
MARGIN = 48°
30
LOOP PHASE (Degrees)
45
30
20
0
LOOP GAIN
0
LOOP GAIN
LOOP PHASE (Degrees)
40
40
ADP2106
ADP2107
50
50
0
180
CROSSOVER
OUTPUT VOLTAGE = 1.8V
–10 INPUT VOLTAGE = 3.6V
FREQUENCY = 83kHz
LOAD CURRENT = 1A
–20 INDUCTOR = 2.2µH (LPS4012)
OUTPUT CAPACITOR = 22µF + 22µF
–30 COMPENSATION RESISTOR = 180kΩ
COMPENSATION CAPACITOR = 56pF
–40
10
100
1
FREQUENCY (kHz)
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
10
Figure 45. ADP2107 Bode Plot at VIN = 5 V, VOUT = 2.5 V and Load = 1 A
60
45
0
90
135
180
CROSSOVER
OUTPUT VOLTAGE = 1.2V
FREQUENCY = 71kHz
–10
INPUT VOLTAGE = 3.6V
LOAD CURRENT = 1A
–20 INDUCTOR = 3.3µH (SD3814)
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF
–30 COMPENSATION RESISTOR = 267kΩ
COMPENSATION CAPACITOR = 39pF
–40
1
10
100
FREQUENCY (kHz)
300
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
Figure 43. ADP2105 Bode Plot at VIN = 3.6 V, VOUT = 1.2 V, and Load = 1 A
0
LOOP GAIN
30
LOOP GAIN (dB)
30
40
LOOP PHASE (Degrees)
0
06079-057
LOOP GAIN (dB)
LOOP GAIN
LOOP PHASE
ADP2107
50
40
PHASE
MARGIN = 51°
300
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LOAD TRANSIENT.
50
10
180
CROSSOVER
OUTPUT VOLTAGE = 2.5V
–10 INPUT VOLTAGE = 5V
FREQUENCY = 76kHz
LOAD CURRENT = 1A
–20 INDUCTOR = 2µH (D62LCB)
OUTPUT CAPACITOR = 10µF + 4.7µF
–30 COMPENSATION RESISTOR = 70kΩ
COMPENSATION CAPACITOR = 120pF
–40
1
10
100
FREQUENCY (kHz)
ADP2105
20
135
0
300
Figure 42. ADP2106 Bode Plot at VIN = 3.6 V, VOUT = 1.8 V, and Load = 1 A
60
90
LOOP PHASE
45
PHASE
MARGIN = 70°
20
10
LOOP PHASE
0
90
135
180
CROSSOVER
OUTPUT VOLTAGE = 3.3V
–10 INPUT VOLTAGE = 5V
FREQUENCY = 67kHz
LOAD CURRENT = 1A
–20 INDUCTOR = 2.5µH (CDRH5D28)
OUTPUT CAPACITOR = 10µF + 4.7µF
–30 COMPENSATION RESISTOR = 70kΩ
COMPENSATION CAPACITOR = 120pF
–40
1
10
100
FREQUENCY (kHz)
LOOP PHASE (Degrees)
0
20
06079-059
135
LOOP PHASE
45
PHASE
MARGIN = 65°
300
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LOAD TRANSIENT.
Figure 46. ADP2107 Bode Plot at VIN = 5 V, VOUT = 3.3 V, and Load = 1 A
Rev. C | Page 20 of 36
06079-069
10
90
06079-056
LOOP GAIN (dB)
PHASE
MARGIN = 52°
20
LOOP GAIN (dB)
45
30
0
LOOP GAIN
30
LOOP PHASE (Degrees)
LOOP GAIN
LOOP PHASE (Degrees)
40
40
ADP2105/ADP2106/ADP2107
LOAD TRANSIENT RESPONSE
T
T
OUTPUT CURRENT
OUTPUT CURRENT
3
3
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
2
2
1
1
LX NODE (SWITCH NODE)
LX NODE (SWITCH NODE)
M 20.0µs
T 10.00%
A
CH3
700mA
CH3
700mA
Figure 50. 1 A Load Transient Response for ADP2105-1.2
with External Components Chosen for 10% Overshoot
Figure 47. 1 A Load Transient Response for ADP2105-1.2
with External Components Chosen for 5% Overshoot
T
A
OUTPUT CAPACITOR: 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESISTOR: 135kΩ
COMPENSATION CAPACITOR: 82pF
06079-087
OUTPUT CAPACITOR: 22µF + 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESISTOR: 270kΩ
COMPENSATION CAPACITOR: 39pF
M 20.0µs
T 10.00%
06079-090
CH1 2.00V
CH2 100mV~
CH3 1.00A Ω
CH1 2.00V
CH2 100mV~
CH3 1.00A Ω
T
OUTPUT CURRENT
OUTPUT CURRENT
3
3
OUTPUT VOLTAGE (AC-COUPLED)
2
2
OUTPUT VOLTAGE (AC-COUPLED)
1
1
LX NODE (SWITCH NODE)
LX NODE (SWITCH NODE)
M 20.0µs
T 10.00%
A
CH3
700mA
CH3
700mA
Figure 51. 1 A Load Transient Response for ADP2105-1.8
with External Components Chosen for 10% Overshoot
Figure 48. 1 A Load Transient Response for ADP2105-1.8
with External Components Chosen for 5% Overshoot
T
A
OUTPUT CAPACITOR: 10µF + 10µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESISTOR: 135kΩ
COMPENSATION CAPACITOR: 82pF
06079-088
OUTPUT CAPACITOR: 22µF + 22µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESISTOR: 270kΩ
COMPENSATION CAPACITOR: 39pF
M 20.0µs
T 10.00%
06079-091
CH1 2.00V
CH2 100mV~
CH3 1.00A Ω
CH1 2.00V
CH2 100mV~
CH3 1.00A Ω
T
OUTPUT CURRENT
OUTPUT CURRENT
3
3
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
2
2
1
1
LX NODE (SWITCH NODE)
M 20.0µs
T 10.00%
A
CH3
700mA
OUTPUT CAPACITOR: 22µF + 4.7µF
INDUCTOR: CDRH5D18, 4.1µH
COMPENSATION RESISTOR: 270kΩ
COMPENSATION CAPACITOR: 39pF
M 20.0µs
T 10.00%
A
CH3
700mA
OUTPUT CAPACITOR: 10µF + 4.7µF
INDUCTOR: CDRH5D18, 4.1µH
COMPENSATION RESISTOR: 135kΩ
COMPENSATION CAPACITOR: 82pF
06079-089
CH1 2.00V
CH2 200mV~
CH3 1.00A Ω
CH1 2.00V
CH2 200mV~
CH3 1.00A Ω
Figure 52. 1 A Load Transient Response for ADP2105-3.3
with External Components Chosen for 10% Overshoot
Figure 49. 1 A Load Transient Response for ADP2105-3.3
with External Components Chosen for 5% Overshoot
Rev. C | Page 21 of 36
06079-092
LX NODE (SWITCH NODE)
ADP2105/ADP2106/ADP2107
EFFICIENCY CONSIDERATIONS
Transition Losses
Efficiency is the ratio of output power to input power. The high
efficiency of the ADP2105/ADP2106/ADP2107 has two distinct
advantages. First, only a small amount of power is lost in the dcto-dc converter package that reduces thermal constraints. Second,
the high efficiency delivers the maximum output power for the
given input power, extending battery life in portable applications.
Transition losses occur because the P-channel MOSFET power
switch cannot turn on or turn off instantaneously. At the middle of
an LX (switch) node transition, the power switch is providing all
the inductor current, while the source to drain voltage of the
power switch is half the input voltage, resulting in power loss.
Transition losses increase with load current and input voltage
and occur twice for each switching cycle.
There are four major sources of power loss in dc-to-dc
converters like the ADP2105/ADP2106/ADP2107:
•
•
•
•
The amount of power loss can be calculated by
PTRAN =
Power switch conduction losses
Inductor losses
Switching losses
Transition losses
where tON and tOFF are the rise time and fall time of the LX
(switch) node, and are both approximately 3 ns.
THERMAL CONSIDERATIONS
Power Switch Conduction Losses
Power switch conduction losses are caused by the flow of output
current through the P-channel power switch and the N-channel
synchronous rectifier, which have internal resistances (RDS(ON))
associated with them. The amount of power loss can be approximated by
PSW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT2
where D = VOUT/VIN.
The internal resistance of the power switches increases with
temperature but decreases with higher input voltage. Figure 20
and Figure 21 show the change in RDS(ON) vs. input voltage,
whereas Figure 29 and Figure 30 show the change in RDS(ON) vs.
temperature for both power devices.
Inductor Losses
Inductor conduction losses are caused by the flow of current
through the inductor, which has an internal resistance (DCR)
associated with it. Larger sized inductors have smaller DCR,
which can improve inductor conduction losses.
Inductor core losses are related to the magnetic permeability of
the core material. Because the ADP2105/ADP2106/ADP2107
are high switching frequency dc-to-dc converters, shielded ferrite
core material is recommended for its low core losses and low EMI.
The total amount of inductor power loss can be calculated by
PL = DCR × IOUT2 + Core Losses
Switching Losses
Switching losses are associated with the current drawn by the
driver to turn on and turn off the power devices at the
switching frequency. Each time a power device gate is turned on
and turned off, the driver transfers a charge ΔQ from the input
supply to the gate and then from the gate to ground.
The amount of power loss can by calculated by
PSW = (CGATE − P + CGATE − N) × VIN2 × fSW
V IN
× I OUT × (t ON + t OFF ) × f SW
2
In most applications, the ADP2105/ADP2106/ADP2107 do not
dissipate a lot of heat due to their high efficiency. However, in
applications with high ambient temperature, low supply voltage,
and high duty cycle, the heat dissipated in the package is large
enough that it can cause the junction temperature of the die to
exceed the maximum junction temperature of 125°C. Once the
junction temperature exceeds 140°C, the converter goes into
thermal shutdown. To prevent any permanent damage it recovers
only after the junction temperature has decreased below 100°C.
Therefore, thermal analysis for the chosen application solution
is very important to guarantee reliable performance over all
conditions.
The junction temperature of the die is the sum of the ambient
temperature of the environment and the temperature rise of the
package due to the power dissipation, as shown in the following
equation:
TJ = TA + TR
where:
TJ is the junction temperature.
TA is the ambient temperature.
TR is the rise in temperature of the package due to the power
dissipation in the package.
The rise in temperature of the package is directly proportional
to the power dissipation in the package. The proportionality
constant for this relationship is defined as the thermal resistance
from the junction of the die to the ambient temperature, as
shown in the following equation:
TR = θJA × PD
where:
TR is the rise in temperature of the package.
PD is the power dissipation in the package.
θJA is the thermal resistance from the junction of the die to the
ambient temperature of the package.
where:
(CGATE − P + CGATE − N) ≈ 600 pF.
fSW = 1.2 MHz, the switching frequency.
Rev. C | Page 22 of 36
ADP2105/ADP2106/ADP2107
For example, in an application where the ADP2107(1.8 V) is
used with an input voltage of 3.6 V, a load current of 2 A, and a
maximum ambient temperature of 85°C, at a load current of 2 A,
the most significant contributor of power dissipation in the dc-todc converter package is the conduction loss of the power switches.
Using the graph of switch on resistance vs. temperature (see
Figure 30), as well as the equation of power loss given in the
Power Switch Conduction Losses section, the power dissipation
in the package can be calculated by the following:
PSW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT2 =
The θJA for the LFCSP_VQ package is 40°C/W, as shown in
Table 3. Therefore, the rise in temperature of the package due to
power dissipation is
TR = θJA × PD = 40°C/W × 0.40 W = 16°C
The junction temperature of the converter is
TJ = TA + TR = 85°C + 16°C = 101°C
Because the junction temperature of the converter is below the
maximum junction temperature of 125°C, this application operates
reliably from a thermal point of view.
[109 mΩ × 0.5 + 90 mΩ × 0.5] × (2 A)2 ≈ 400 mW
Rev. C | Page 23 of 36
ADP2105/ADP2106/ADP2107
DESIGN EXAMPLE
Consider an application with the following specifications:
4.
Input Voltage = 3.6 V to 4.2 V.
Output Voltage = 2 V.
Typical Output Current = 600 mA.
Maximum Output Current = 1.2 A.
Soft Start Time = 2 ms.
Overshoot ≤ 100 mV under all load transient conditions.
1.
2.
3.
5.
Choose the dc-to-dc converter that satisfies the maximum
output current requirement. Because the maximum output
current for this application is 1.2 A, the ADP2106 with a
maximum output current of 1.5 A is ideal for this
application.
See whether the output voltage desired is available as a
fixed output voltage option. Because 2 V is not one of the
fixed output voltage options available, choose the adjustable
version of ADP2106.
The first step in external component selection for an
adjustable version converter is to calculate the resistance of
the resistive voltage divider that sets the output voltage.
R BOT =
VFB
I STRING
=
0.8 V
20 μA
⇒ Output Capacitor =
6.
For the ADP2106:
L > (0.83 μH/V) × VOUT
⇒ L > 0.83 μH/V × 2 V
⇒ L > 1.66 μH
Next, calculate the ideal inductor value that sets the
inductor peak-to-peak current ripple (ΔIL) to 1/3 of the
maximum load current at the maximum input voltage as
follows:
L IDEAL =
Output Capacitor × Output Voltage = 60 μC
= 40 kΩ
⎡ 2 V − 0.8 V ⎤
⎡V
− V FB ⎤
RTOP = R BOT ⎢ OUT
⎥ = 60 kΩ
⎥ = 40 kΩ × ⎢
VFB
⎣
⎦
⎣⎢ 0.8 V ⎦⎥
Calculate the minimum inductor value as follows:
The closest standard inductor value is 2.2 μH. The maximum
rms current of the inductor is to be greater than 1.2 A, and
the saturation current of the inductor is to be greater than
2 A. One inductor that meets these criteria is the LPS40122.2 μH from Coilcraft.
Choose the output capacitor based on the transient response
requirements. The worst-case load transient is 1.2 A, for
which the overshoot must be less than 100 mV, which is 5%
of the output voltage. For a 1 A load transient, the overshoot
must be less than 4% of the output voltage, then from
Figure 39:
7.
8.
9.
2.5 × VOUT × (V IN − VOUT )
μH =
V IN × I LOAD ( MAX )
Rev. C | Page 24 of 36
2. 0 V
≈ 30 μF
Taking into account the loss of capacitance due to dc bias, as
shown in Figure 40, two 22 μF X5R MLCC capacitors from
Murata (GRM21BR60J226M) are sufficient for this
application.
Because the ADP2106 is being used in this application, the
input capacitors are 10 μF and 4.7 μF X5R Murata capacitors
(GRM21BR61A106K and GRM21BR61A475K).
The input filter consists of a small 0.1 μF ceramic capacitor
placed between IN and AGND and a 10 Ω resistor placed
between IN and PWIN1.
Choose a soft start capacitor of 2 nF to achieve a soft start
time of 2 ms.
Calculate the compensation resistor and capacitor as
follows:
⎛ (2 π)FCROSS
RCOMP = 0.8 ⎜⎜
⎝ GmGCS
⎞⎛ COUT VOUT
⎟⎜
⎟⎜ V
REF
⎠⎝
⎞
⎟=
⎟
⎠
⎛
⎞⎛ 30 μF × 2 V ⎞
(2 π) × 80 kHz
⎟⎜
⎟ = 215 kΩ
0.8 ⎜
⎜ 50 μA / V × 2.8125 A / V ⎟⎜ 0.8 V ⎟
⎝
⎠⎝
⎠
C COMP =
2.5 × 2 × (4.2 − 2)
μH = 2.18 μH
4 . 2 × 1 .2
60 μC
2
2
=
= 39 pF
πFCROSS RCOMP π × 80 kHz × 215 kΩ
ADP2105/ADP2106/ADP2107
EXTERNAL COMPONENT RECOMMENDATIONS
For popular output voltage options at 80 kHz crossover frequency with 10% overshoot for a 1 A load transient (refer to Figure 37 and
Figure 38).
Table 11. Recommended External Components
Part
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2105-1.2
ADP2105-1.5
ADP2105-1.8
ADP2105-3.3
ADP2106-1.2
ADP2106-1.5
ADP2106-1.8
ADP2106-3.3
ADP2107-1.2
ADP2107-1.5
ADP2107-1.8
ADP2107-3.3
VOUT (V)
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
CIN1 1 (μF)
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
CIN21 (μF)
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
10
10
10
10
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
COUT 2 (μF)
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
L (μH)
2.0
2.5
3.0
3.3
3.6
4.1
1.5
1.8
2.0
2.2
2.5
3.0
1.2
1.5
1.5
1.8
1.8
2.5
2.5
3.0
3.3
4.1
1.8
2.0
2.2
3.0
1.5
1.5
1.8
2.5
1
RCOMP (kΩ)
135
135
135
135
135
135
90
90
90
90
90
90
70
70
70
70
70
70
135
135
135
135
90
90
90
90
70
70
70
70
CCOMP (pF)
82
82
82
82
82
82
100
100
100
100
100
100
120
120
120
120
120
120
82
82
82
82
100
100
100
100
120
120
120
120
RTOP 3 (kΩ)
5
20
35
50
85
125
5
20
35
50
85
125
5
20
35
50
85
125
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
RBOT3 (kΩ)
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
4.7 μF 0805 X5R 10 V Murata—GRM21BR61A475KA73L. 10 μF 0805 X5R 10 V Murata—GRM21BR61A106KE19L.
4.7 μF 0805 X5R 10 V Murata—GRM21BR61A475KA73L. 10 μF 0805 X5R 10 V Murata—GRM21BR61A106KE19L. 22 μF 0805 X5R 6.3 V Murata—GRM21BR60J226ME39L.
3
0.5% accuracy resistor.
2
Rev. C | Page 25 of 36
ADP2105/ADP2106/ADP2107
For popular output voltage options at 80 kHz crossover frequency with 5% overshoot for a 1 A load transient (refer to Figure 37 and
Figure 38).
Table 12. Recommended External Components
Part
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2105(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2106(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2107(ADJ)
ADP2105-1.2
ADP2105-1.5
ADP2105-1.8
ADP2105-3.3
ADP2106-1.2
ADP2106-1.5
ADP2106-1.8
ADP2106-3.3
ADP2107-1.2
ADP2107-1.5
ADP2107-1.8
ADP2107-3.3
VOUT (V)
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
CIN1 1 (μF)
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
CIN21 (μF)
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
10
10
10
10
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
COUT 2 (μF)
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
22 + 4.7
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
22 + 4.7
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
22 + 4.7
L (μH)
2.0
2.5
3.0
3.3
3.6
4.1
1.5
1.8
2.0
2.2
2.5
3.0
1.2
1.5
1.5
1.8
1.8
2.5
2.5
3.0
3.3
4.1
1.8
2.0
2.2
3.0
1.5
1.5
1.8
2.5
1
RCOMP (kΩ)
270
270
270
270
270
270
180
180
180
180
180
180
140
140
140
140
140
140
270
270
270
270
180
180
180
180
140
140
140
140
CCOMP (pF)
39
39
39
39
39
39
56
56
56
56
56
56
68
68
68
68
68
68
39
39
39
39
56
56
56
56
68
68
68
68
RTOP 3 (kΩ)
5
20
35
50
85
125
5
20
35
50
85
125
5
20
35
50
85
125
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
RBOT 3(kΩ)
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
4.7μF 0805 X5R 10V Murata—GRM21BR61A475KA73L. 10μF 0805 X5R 10V Murata—GRM21BR61A106KE19L.
4.7μF 0805 X5R 10V Murata—GRM21BR61A475KA73L. 10μF 0805 X5R 10V Murata—GRM21BR61A106KE19L. 22μF 0805 X5R 6.3V Murata—GRM21BR60J226ME39L.
3
0.5% accuracy resistor.
2
Rev. C | Page 26 of 36
ADP2105/ADP2106/ADP2107
CIRCUIT BOARD LAYOUT RECOMMENDATIONS
Good circuit board layout is essential to obtaining the best
performance from the ADP2105/ADP2106/ADP2107. Poor
circuit layout degrades the output ripple, as well as the
electromagnetic interference (EMI) and electromagnetic
compatibility (EMC) performance.
•
Make the high current path from the PGND pin through L
and COUT back to the PGND plane as short as possible. To
accomplish this, ensure that the PGND pin is tied to the
PGND plane as close as possible to the input and output
capacitors.
Figure 54 and Figure 55 show the ideal circuit board layout for
the ADP2105/ADP2106/ADP2107 to achieve the highest
performance. Refer to the following guidelines if adjustments to
the suggested layout are needed:
•
The feedback resistor divider network is to be placed as
close as possible to the FB pin to prevent noise pickup. The
length of trace connecting the top of the feedback resistor
divider to the output is to be as short as possible while
keeping away from the high current traces and the LX
(switch) node that can lead to noise pickup. An analog
ground plane is to be placed on either side of the FB trace
to reduce noise pickup. For the low fixed voltage options
(1.2 V and 1.5 V), poor routing of the OUT_SENSE trace
can lead to noise pickup, adversely affecting load regulation.
This can be fixed by placing a 1 nF bypass capacitor close to
the FB pin.
•
The placement and routing of the compensation components
are critical for proper behavior of the ADP2105/ADP2106/
ADP2107. The compensation components are to be placed
as close to the COMP pin as possible. It is advisable to use
0402-sized compensation components for closer placement,
leading to smaller parasitics. Surround the compensation
components with an analog ground plane to prevent noise
pickup. The metal layer under the compensation components
is to be the analog ground plane.
•
Use separate analog and power ground planes. Connect
the ground reference of sensitive analog circuitry (such as
compensation and output voltage divider components) to
analog ground; connect the ground reference of power
components (such as input and output capacitors) to power
ground. In addition, connect both the ground planes to the
exposed pad of the ADP2105/ADP2106/ADP2107.
•
For each PWIN pin, place an input capacitor as close to the
PWIN pin as possible and connect the other end to the closest
power ground plane.
•
Place the 0.1 μF, 10 Ω low-pass input filter between the IN
pin and the PWIN1 pin, as close to the IN pin as possible.
•
Ensure that the high current loops are as short and as wide
as possible. Make the high current path from CIN through
L, COUT, and the PGND plane back to CIN as short as possible.
To accomplish this, ensure that the input and output
capacitors share a common PGND plane.
Rev. C | Page 27 of 36
ADP2105/ADP2106/ADP2107
EVALUATION BOARD
EVALUATION BOARD SCHEMATIC FOR ADP2107 (1.8 V)
C7
0.1µF
VCC
R3
10Ω
C1
10µF1
OUT
J1
U1
GND
16
15
FB
GND IN
1
EN
2
GND
INPUT VOLTAGE = 2.7V TO 5.5V
VIN
VCC
14
13
PWIN1
LX2 12
EN
PGND 11
ADP2107-1.8
1
LX1 10
GND
3
L12
2µH
PWIN2 9
GND
4
OUT
6
7
8
C4
22µF1
C3
22µF1
GND
R5
NS
R1
140kΩ
C6
68pF
R4
0Ω
C2
10µF1
COMP SS AGND PADDLE NC
5
OUTPUT VOLTAGE = 1.8V, 2A
VOUT
2
VCC
1 MURATA
X5R 0805
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
2 2μH INDUCTOR D62LCB TOKO
C5
1nF
NC = NO CONNECT
06079-044
R2
100kΩ
Figure 53. Evaluation Board Schematic of the ADP2107-1.8 (Bold Traces are High Current Paths)
RECOMMENDED PCB LAYOUT (EVALUATION BOARD LAYOUT)
JUMPER TO ENABLE
ENABLE
GROUND
VIN
100kΩ PULL-DOWN
GROUND
INPUT
INPUT CAPACITOR
POWER GROUND
PLANE
PLACE THE FEEDBACK RESISTORS AS
CLOSE TO THE FB PIN AS POSSIBLE.
RTOP RBOT
CONNECT THE GROUND RETURN OF
ALL POWER COMPONENTS SUCH AS
INPUT AND OUTPUT CAPACITORS TO
THE POWER GROUND PLANE.
OUTPUT CAPACITOR
CIN
COUT
LX
OUTPUT
PGND
ADP2105/ADP2106/ADP2107
VOUT
LX
RCOMP
CIN
CCOMP
PLACE THE COMPENSATION
COMPONENTS AS CLOSE TO
THE COMP PIN AS POSSIBLE.
INDUCTOR (L)
COUT
OUTPUT CAPACITOR
CSS
ANALOG GROUND PLANE
POWER GROUND
INPUT CAPACITOR
06079-045
CONNECT THE GROUND RETURN OF ALL
SENSITIVE ANALOG CIRCUITRY SUCH AS
COMPENSATION AND OUTPUT VOLTAGE
DIVIDER TO THE ANALOG GROUND PLANE.
Figure 54. Recommended Layout of Top Layer of ADP2105/ADP2106/ADP2107
Rev. C | Page 28 of 36
ADP2105/ADP2106/ADP2107
ENABLE
VIN
GND
GND
ANALOG GROUND PLANE
POWER GROUND PLANE
INPUT VOLTAGE PLANE
CONNECTING THE TWO
PWIN PINS AS CLOSE
AS POSSIBLE.
VIN
VOUT
CONNECT THE PGND PIN
TO THE POWER GROUND
PLANE AS CLOSE TO THE
ADP2105/ADP2106/ADP2107
AS POSSIBLE.
FEEDBACK TRACE: THIS TRACE CONNECTS THE TOP OF THE
RESISTIVE VOLTAGE DIVIDER ON THE FB PIN TO THE OUTPUT.
PLACE THIS TRACE AS FAR AWAY FROM THE LX NODE AND HIGH
CURRENT TRACES AS POSSIBLE TO PREVENT NOISE PICKUP.
Figure 55. Recommended Layout of Bottom Layer of ADP2105/ADP2106/ADP2107
Rev. C | Page 29 of 36
06079-046
CONNECT THE EXPOSED PAD OF
THE ADP2105/ADP2106/ADP2107
TO A LARGE GROUND PLANE TO
AID POWER DISSIPATION.
ADP2105/ADP2106/ADP2107
APPLICATION CIRCUITS
0.1μF
VIN
10Ω
INPUT VOLTAGE = 5V
10μF1
VOUT
OFF
16
15
FB
GND IN
14
13
PWIN1
LX2 12
1
EN
2
GND
2.5μH2
PGND 11
ADP2107-3.3
3
GND
LX1 10
4
GND
PWIN2 9
COMP
SS
5
6
10μF1
8
1nF
70kΩ
OUTPUT VOLTAGE = 3.3V
4.7μF1
LOAD
0A TO 2A
VIN
AGND NC
7
VOUT
10μF1
1 MURATA
X5R 0805
10μF: GRM21BR61A106KE19L
4.7μF: GRM21BR61A475KA73L
2 SUMIDA CDRH5D28: 2.5μH
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 10% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
120pF
06079-047
ON
Figure 56. Application Circuit—VIN = 5 V, VOUT = 3.3 V, Load = 0 A to 2 A
0.1μF
VIN
10Ω
INPUT VOLTAGE = 3.6V
10μF1
VOUT
OFF
16
15
FB
GND IN
14
13
PWIN1
LX2 12
1 EN
1.5μH2
PGND 11
2 GND
22μF1
ADP2107-1.5
3 GND
LX1 10
4 GND
PWIN2 9
COMP
SS
5
6
140kΩ
68pF
AGND NC
1nF
7
8
VOUT
OUTPUT VOLTAGE = 1.5V
22μF1
LOAD
0A TO 2A
VIN
10μF1
1 MURATA X5R 0805
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
2 TOKO D62LCB OR COILCRAFT LPS4012
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 57. Application Circuit—VIN = 3.6 V, VOUT = 1.5 V, Load = 0 A to 2 A
Rev. C | Page 30 of 36
06079-048
ON
ADP2105/ADP2106/ADP2107
0.1μF
VIN
10Ω
INPUT VOLTAGE = 2.7V TO 4.2V
4.7μF1
VOUT
16
15
FB
GND IN
14
13
PWIN1
LX2 12
1 EN
OFF
2.7μH2
PGND 11
2 GND
ADP2105-1.8
3 GND
LX1 10
4 GND
PWIN2 9
COMP
SS
5
6
4.7μF1
LOAD
0A TO 1A
1 MURATA X5R 0805
4.7μF: GRM21BR61A475KA73L
22μF: GRM21BR60J226ME39L
8
2 TOKO 1098AS-DE2812: 2.7μH
1nF
270kΩ
OUTPUT VOLTAGE = 1.8V
22μF1
VIN
AGND NC
7
VOUT
22μF1
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
39pF
06079-049
ON
Figure 58. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.8 V, Load = 0 A to 1 A
0.1μF
VIN
10Ω
INPUT VOLTAGE = 2.7V TO 4.2V
4.7μF1
VOUT
OFF
16
15
FB
GND IN
14
13
PWIN1
LX2 12
1
EN
2
GND
2.4μH2
PGND 11
ADP2105-1.2
3
GND
LX1 10
4
GND
PWIN2 9
COMP
SS
5
6
135kΩ
82pF
AGND NC
1nF
7
VOUT
22μF1
8
OUTPUT VOLTAGE = 1.2V
4.7μF1
LOAD
0A TO 1A
VIN
4.7μF1
1 MURATA
X5R 0805
4.7μF: GRM21BR61A475KA73L
22μF: GRM21BR60J226ME39L
2 TOKO 1069AS-DB3018HCT OR
TOKO 1070AS-DB3020HCT
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 10% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 59. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.2 V, Load = 0 A to 1 A
Rev. C | Page 31 of 36
06079-050
ON
ADP2105/ADP2106/ADP2107
0.1μF
VIN
10Ω
INPUT VOLTAGE = 5V
10μF1
FB
OFF
16
15
14
13
FB
GND
IN
PWIN1
LX2 12
1
EN
2
GND
2.5μH2
ADP2106-ADJ
85kΩ
3
GND
LX1 10
4
GND
PWIN2 9
COMP
SS
5
6
180kΩ
56pF
OUTPUT VOLTAGE = 2.5V
PGND 11
AGND NC
1nF
7
10μF1
22μF1
LOAD
0A TO 1.5A
FB
VIN
40kΩ
4.7μF1
8
1 MURATA
X5R 0805
4.7μF: GRM21BR61A475KA73L
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
2 COILTRONICS SD14: 2.5μH
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 60. Application Circuit—VIN = 5 V, VOUT = 2.5 V, Load = 0 A to 1.5 A
Rev. C | Page 32 of 36
06079-051
ON
ADP2105/ADP2106/ADP2107
OUTLINE DIMENSIONS
4.00
BSC SQ
0.60 MAX
0.60 MAX
13
12° MAX
1.00
0.85
0.80
0.65 BSC
TOP
VIEW
12
0.50
0.40
0.30
0.80 MAX
0.65 TYP
(BOTTOM VIEW)
9
8
5
4
0.25 MIN
1.95 BSC
0.05 MAX
0.02 NOM
SEATING
PLANE
2.50
2.35 SQ
2.20
EXPOSED
PAD
3.75
BSC SQ
0.35
0.30
0.25
PIN 1
INDICATOR
1
0.20 REF
COPLANARITY
0.08
THE EXPOSED PAD ON THE
BOTTOM OF THE LFCSP PACKAGE
MUST BE SOLDERED TO PCB GROUND
FOR PROPER HEAT DISSIPATION AND
ALSO FOR NOISE AND MECHANICAL
STRENGTH BENEFITS.
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC
010606-0
PIN 1
INDICATOR
16
Figure 61. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad
(CP-16-10)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADP2105ACPZ-1.2-R7 1
ADP2105ACPZ-1.5-R71
ADP2105ACPZ-1.8-R71
ADP2105ACPZ-3.3-R71
ADP2105ACPZ-R71
ADP2106ACPZ-1.2-R71
ADP2106ACPZ-1.5-R71
ADP2106ACPZ-1.8-R71
ADP2106ACPZ-3.3-R71
ADP2106ACPZ-R71
ADP2107ACPZ-1.2-R71
ADP2107ACPZ-1.5-R71
ADP2107ACPZ-1.8-R71
ADP2107ACPZ-3.3-R71
ADP2107ACPZ-R71
ADP2105-1.8-EVALZ1
ADP2105-EVALZ1
ADP2106-1.8-EVALZ1
ADP2106-EVALZ1
ADP2107-1.8-EVALZ1
ADP2107-EVALZ1
1
Output
Current
1A
1A
1A
1A
1A
1.5 A
1.5 A
1.5 A
1.5 A
1.5 A
2A
2A
2A
2A
2A
Temperature
Range
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
Output Voltage
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.8 V
Adjustable, but set to 2.5 V
1.8 V
Adjustable, but set to 2.5 V
1.8 V
Adjustable, but set to 2.5 V
Z = RoHS Compliant Part.
Rev. C | Page 33 of 36
Package Description
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Package Option
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
CP-16-10
ADP2105/ADP2106/ADP2107
NOTES
Rev. C | Page 34 of 36
ADP2105/ADP2106/ADP2107
NOTES
Rev. C | Page 35 of 36
ADP2105/ADP2106/ADP2107
NOTES
©2006–2008 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06079-0-9/08(C)
Rev. C | Page 36 of 36
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