Vorperian 1 [562,16 KiB]
INTRODUCTION
Simplified Analysis of
PWM Converters Using
Model of PWM Switch
Part I: Continuous Conduction
Mode
VATCHEVORPERIAN
Virginia Polytechnic Institute and State University
A circuit-oriented approach to the analysis of pulsewidth
modulation (PWM)converters is presented. This method relies
on the identification of a three-terminal nonlinear device, called
the PWM switch, which consists of only the active and passive
switches in a PWM converter. Once the invariant properties of
the PWM switch are determined, an average equivalent circuit
model for it can be derived. This model is versatile enough that
it can easily account for storage-time modulation of BJTs. The
dc and small-signal characteristics of a large class of PWM
converters can then be obtained by a simple substitution of the
P W M switch with its equivalent circuit model. The methodology
presented is very similar to linear amplifier circuit analysis
.whereby the transistor is replaced by its equivalent circuit model.
Consequently, for the novice, this method should serve as a very
smooth introduction to the analysis of PWM converters.
Manuscript received February 16, 1989; revised June 16, 1989.
IEEE Log No. 33463A.
Author’s address: Virginia Power Electronics Center, Dep’t. of
Electrical Engineering, Virginia Polytechnic Institute and State
University, Blacksburg, VA 24061.
0018-9251/90/0500-0490$1.00 @ 1990 IEEE
490
In this article the concept of the pulsewidth
modulation (PWM) switch is introduced which leads
to considerable simplification in the analysis (linear
and nonlinear) and generation of dc-to-dc converters.
To show the simplicity and elegance of the Pu.7M switch
model the dc and small-signal analysis of basic dc-to-dc
converters is given. Of course, the results of the dc
and small-signal analysis of dc-to-dc PWM converters
are well known today from the systematic method of
state-space averaging and its canonical circuit model [l,
21. Hence, before going any further, a brief description
of the two models mentioned above is given and the
diflerence in the analytical approach between them
is explained in order to justify the purpose of this
article. First, the canonical circuit model completely
represents the dc and small-signal characteristics of a
PWM converter and is obtained after a considerable
amount of matrix manipulations in order to single out
the desirable input and output characteristics of the
converter (input impedance, line-to-output transfer
function, and output impedance) in addition to its
control-to-output characteristics. The P W switch
model, on the other hand, represents only the dc and
small-signal characteristics of the nonlinear part of
the converter, which consists of the active and passive
switches (the PWM switch), and is obtained after a few
lines of very simple algebra. The dc and small-signal
characteristics of a PWM converter are then obtained
by replacing the PWh4 switch with its equivalent
circuit model in a manner similar to obtaining the
small-signal characteristics of linear amplifiers whereby
the transistor is replaced by its equivalent circuit
model. There are two advantages in using the model
of the PWM switch. First, the PWM switch model
allows many PWM converters to be analyzed using
simple linear electronic circuit analysis programs
(P-SPICE, MICRO-CAP to name a few), which allow
for user-defined models (macros), without recourse to
special-purpose programs which manipulate state-space
equations. The second and more important advantage
is a pedagogical one. More and more universities are
beginning to teach power electronics at the senior level
in their undergraduate program. Although students
at this level are familiar with some matrix algebra, it
would be far easier for them to learn all about the dc
and small-signal properties of PWM converters using
the method of equivalent circuit model which they
learned in their electronics courses earlier. Students
spend quite a bit of time learning about the nonlinear
characteristics of the transistor and its small-signal
equivalent circuit model. They later use this model
to analyze the small-signal characteristics of linear
amplifiers simply by replacing the transistor with
its equivalent circuit model. Why not d o the same
in power electronics? Ultimately, whether or not
the PWM switch model presented here is a useful
IEEE TRANSACTIONS ON AEROSPACE AND ELECTRONIC SYSTEMS VOL. 24, NO. 3 MAY 19N
educational tool depends on how well it is received
by the educators in the field of power electronics.
The model of the PWM switch along with the
model of the resonant switch were developed while
work on the small-signal analysis of quasi-resonant
With
. a model
converters was being conducted [MI
of the PWM and resonant switches, the small-signal
characteristics of quasi-resonant and PWM converters
could be compared easily. The relationship of the
PWM switch to the converter cell approach to the
generation of converters has been discussed in [7],
while a similar but more limited approach was given
earlier in [ l l , 161. Subsequently, the nonlinear analysis
of the PWM switch was performed [9, 101 which
resulted in considerable simplification over previous
attempts namely, the perturbation series method [SI,
and the Volterra functional series method [14].
C
PWM SWITCH A N D ITS INVARIANT PROPERTIES
The four familiar PWM converters are shown
in Fig. 1 where the active and passive switches are
lumped together in a single functional block called
the PWM switch. This functional block represents the
total nonlinearity in these converters and is shown
as a three-terminal nonlinear device in Fig. 2. For
obvious reasons, the terminal designations a, p , c
refer to active, passive, and common, respectively. The
voltage and current port designations of this device
are important if we are to think of the PWM switch
as the basic building block of the converters in Fig.
1. It can be seen that in each converter the external
circuit elements are connected to the switch in such a
way as to provide the proper port conditions given in
Fig. 2. Hence, we see that the converters of Fig. 1 are
obtained by a simple permutation of the PWM switch.
A similar cyclic permutation of the PWM switch to
generate some basic converters was discussed in [ l l ]
but its significance on the analysis of converters was
never realized.
Next we investigate the invariant relationships
between the terminal currents and voltages of the
PWM switch which are shown in Fig. 3. It can be
easily seen that the instantaneous current in the active
terminal is always the same as the current in the
common terminal during the on-interval DT, no matter
which configuration the switch is implemented in.
Also, the instantaneous port voltages Gcp(t)and G a p ( t )
are always coincident during DT,. Hence, the invariant
relations in the instantaneous terminal quantities are
given by
VORPmIAN: CONTINUOUS CONDUCTION MODE
Fig. 1. Four basic PWM converters
Fig. 2. PWM switch.
Since in dc-to-dc converters the behavior of the
average quantities is of greater interest in determining
the dc and small-signal characteristics of these
converters, we seek to determine the invariant
relationships between the average terminal quantities.
Hence, for the average terminal currents i, and i, we
have from very simple considerations
i, = di,
(2)
The instantaneous and average voltages across port
a - p require special attention. Since this port is a
voltage port, it is connected either across a voltage
source or a capacitor which in general has an ESR.
Hence, the voltage waveform across this port consists
in general of a small square wave (with a tilted top)
riding on top of a large average value as shown in
Fig. 3. The source of the square wave is the ESR of
the capacitor which absorbs a pulsating current of
peak-to-peak amplitude equal to the maximum value
of the current in the common terminal. Hence, if
the ESR of this capacitor is equal to zero, then the
instantaneous voltage Gap(t)is continuous and consists
only of the capacitive ripple (which in the averaging
process is neglected). Furthermore, this capacitor
may be connected directly across terminals a - p as
in the case of the Cuk, boost, and buck with input
filter converters, or indirectly as in the case of the
491
7;
+
i --
I
DT,+
*
L-
’
n-
DT$-u
I
7; -4
U,
,
I
I8
I
8
1
1
I
FDT,?
(1
,
I1
,1
vv- 3 5- - - - tGP(O
0
,
I
I
,
I
I
I
I
‘
0’7; A,
,
I
’
I
Fig. 4. Terminal voltage C a P ( t ) in presence of ESR of capacitor
which absorbs pulsating current in converter.
DD‘rC
I1
Fig. 3. Relation between terminal voItages and currents of PWM
switch.
buck-boost converter. If we neglect the ripple in the
current and consider Only average
value i,, then the peak-to-peak ripple voltage due to
the ESR shown in Fig. 4 can be expressed as
v, = icre
(3)
. .
where re is in general a function of the ESR of the
capacitor and the load resistor R. For example, in
the boost and buck-boost converters the pulsating
current of amplitude i, is absorbed by the outpucfilter
capacitor which is in parallel with the output load
resistor so that re is given by
re = rc,IIR,
boost, buck-boost.
(4a)
On the other hand, for the Cuk converter, we can see
that
re = rc,,
Cuk
(4b)
+
because the peak-to-peak pulsating current (i, = ii,
io) is absorbed only by the energy transfer capacitor
and the square ripple is only due to its ESR. Hence,
depending upon the converter, one can figure out
easily re. Referring to Fig. 4 we can now easily see that
the relation between the average port voltages is given
bY
v,, = d(v,, - &red’), d’ = 1 - d .
(5)
The desired invariant relations of the PWM switch are
given by (1) and (5) and are summarized below
i, = di,
v,, = d(v,, - icred’)
of the PWM switch are perturbed because of some
perturbation in either thk input voltage or the load of
the converter. Perturbing (6a)and (6b) for a fixed duty
ratio, we get basically the same thing
2,
(6b)
(7a)
2,
(8b)
,D,
+ 1,;
= D(P,, + I c r e i - icreD’)+ d(V,,
= D;,
- ZcreD’)
(9a)
which can be rearranged as
Yap
ci
- + iCreD’- [V,, + 1,(D - D’)re]-.
D
Qc,
=
(9b)
These equations correspond to the dc and small-signal
model of the PWM switch shown in Fig. 6. Notice
that if we neglect re, the model simplifies to the
model shown in Fig. 7, which can also be obtained by
perturbing (7a) and (7b).Equations (sa) and (9b) are
summarized as follows:
+ Zd,
= 0%
,D,
DC AND SMALL-SIGNAL MODEL OF PWM SWITCH
492
Pa)
These equations correspond to the model of the
PWM switch for a fixed duty ratio as shown in Fig. 5
which is also valid all the way down to dc. Suppose
we wanted to compute the open-loop line-to-output
transfer function of a converter. The model to use
then would be the one shown in Fig. 5 as is explained
in the next section. If on the other hand, we would
like to determine the response of the converter to
perturbations in the duty ratio; we perturb (6a) and
(6b) as follows
v,, = dv,,.
Let us first assume that the duty ratio is fixed at
d = D and that the terminal currents and voltages
= Di,
9,, = D(D,, - ;,reD’).
A
(6a)
If the ESR of the capacitor, which absorbs the
pulsating current, can be neglected then (6) reduce
to
i, = di,
Fig. 5. Equivalent average circuit model of P W M switch for fixed
duty ratio D .
where
=
D
2
+ ;,re D’ - VD D
(lob)
+
( W
V, = Vap Zc(D- D’)re.
IEEE TRANSACTIONS ON AEROSPACE AND ELECTRONIC SYSTEMS VOL. 26, NO. 3 MAY 1990
~
Fig. 6. Equivalent dc and small-signal model of PWM switch.
Fig. 10. Boost converter to be analyzed for line-to-output and
input impedance functions using model of PWM switch shown in
Fig. 5.
Open-Loop Line-to-Ouqut Danger Function:
6 P
Fig. 7. Simplified dc and small-signal model of PWM switch.
Under open-loop considerations the model in Fig. 5
still applies. Hence, to determine the line-to-output
transfer the dc input voltage is replaced with a
signal source 9, as shown in Fig. 10 whence we can
immediately determine
where
Fig. 8. Boost converter.
s,1
1
=‘Cf
=
Q
Cf
rLf
+ r,D‘
WO
1
,
Fig. 9. Boost converter to be analyzed for dc characteristics using
model of PWM switch shown in Fig. 5.
ANALYSIS OF PWM CONVERTERS USING MODEL
OF PWM SWITCH
We now show how the model of the PWM
switch is used to determine the dc as well as all the
small-signal characteristics of PWM converters. The
boost converter with all the parasitics shown in Fig. 8
is used as an example.
DC [email protected]: Point-by-point substitution of the
dc model of the PWM switch (Fig. 5 ) in the boost
converter gives the circuit in Fig. 9 from which we can
immediately determine the dc conversion ratio
where re = rCf IIR as discussed earlier in (4a).
Open-Loop Input Impedance: From Fig. 10 we can
continue to determine the input impedance
Zi, = R;, 1+ S/WOQ
+ s’/w,”
1 s/sp
+
where
R;, = rLf i-r,DD’
+ D”R
Pa)
The denominator is (of course) the same as in (13) and
and Q are given by (14b) and (14c).
Open-Loop Output Impedance: In this case we
short the input to ground and connect a test voltage
source at the output as shown in Fig. 11 and obtain
WO
where re = r C f IIR as explained in (4a). The two other
dc quantities which determine the operating point of
the switch are I, and Vap.These are easily determined
from Fig. 9
where
Hence VDcan be calculated from (1Oc).
VORPkRIAN: CONTINUOUS CONDUCTION MODE
493
~
;*i..
-
Fig. 11. Boost converter to be analyzed for output impedance
function using model of PWM switch shown in Fig. 5.
The quadratic in the denominator is the same as in
(13) and szl is the same as in (14a).
Control-to-&put Danger Function: In this case
the input voltage source V, is shorted to ground
and the P W M switch is replaced by its equivalent
circuit model of Fig. 6 as shown in Fig. 12. Although
one can directly determine the control-to-output
transfer function in a straightforward manner from
Fig. 12, we can take some short cuts and use some
tricks of network analysis [15]. Hence, we expect the
denominator of the control-to-output transfer function
to be the same as that of the line-to-output transfer
function as given in (13) so that we have
where W O and Q are given by (14b) and (14c). The low
frequency asymptote K d is simply given by
as can be easily verified from (11). All that remains
to be determined is the numerator in (19) which
corresponds to the zeros or the nulls of the output
voltage. To determine these zeros we simply consider
the circuit of Fig. 12 under null conditions of the
output, i.e., v0(sz2) = 0 and vo(-sZ1) = 0. The
transform circuit under null conditions of the output
is shown in Fig. 13. The first null is determined by
the zero of the impedance of the output filter which
is simply given by
The second null in the output voltage is given by the
null in the passive terminal current, i.e., i,(s,z) = 0.
ForAthesecond null we see that the control generator
VDd/D appears across the transformer whose
secondary voltage appears across r ~ , r,DD'
,
and L f .
Hence we have
1
Fig. 12. Boost converter to be analyzed for control-to-output
transfer function using model of PWM switch shown in Fig. 6 .
Fig. 13. Determination of zeros, or nulls, of control-toautput
transfer function.
Solving (22) and (23) we get
Finally, use of the operating point, (L'D,&),
in (1%) and (12b) in (24) gives
determined
Hence, the control-to-output transfer function has been
completely determined.
MODEL OF PWM SWITCH INCLUDING
STORAGE-TIME M0DULATlON
The model of the PWM switch is versatile enough
to include parasitic effects such as storage-time
modulation which Middlebrook has described in [12].
The effect of storage-time modulation in BJTs can be
accounted for by a simple change in the perturbation
of the duty ratio as follows
where I,, is a modulation parameter which depends
on the type of base drive (proportional, direct, etc.)
as discussed in [13]. Substitution of the above in
(1Oa)-(1Oc) gives
(27)
2VD
-D
D
= ic(rLf
+ r,DD' + s z 2 L f )
(22)
Yap = -
where re = rc, IIR as discussed earlier in (4a). Also, at
the active terminal junction we have
zc = I,d + i,D.
*
494
*
(23)
D
where
VD
r,,, = modulation resistance.
(29)
Zm e
IEEE TRANSACTIONS ON AEROSPACE AND ELECTRONIC SYSTEMS VOL. 26, NO. 3 MAY 1990
’1
D
a - + f l C
Dr,
C
i
+ D’r,
rd
P
Fig. 14. Equivalent circuit model of PWM switch including effects
of storage-time modulation and resistances of diode and transistor
branches. Under d c conditions, nondissipative modulation resistance
rm disappears.
Fig. 15. PWM switch showing resistances of transistor and diode
branches.
control-to-output transfer function was not observed,
it was immediately suspected that the culprit was the
numerator in the line-to-output transfer function. The
According to the approximations explained in [12
converter was first modeled including all the parasitic
and 131, D >> &/Ime
so that (27) reduces to its
elements of the passive components assuming ideal
original form. With this approximation, the above
switches, but no anomalous behavior in the phase
equations correspond to a simple modification in the
response was seen. Finally, when the modulation
model of the P W M switch shown in Fig. 14 where
resistance and the resistances of the diode and the
a nondissipative modulation resistance r, has been
transistor branches were included, a new zero was
added in series with the common terminal.
discovered which could move anywhere from the
Other parasitic elements such as the lead
left-half plane to the right-half plane depending
resistances in the active and passive switches shown
upon the values of rm (which can be made positive
in Fig. 15, can be easily included in the switch model
without much difficulty. Intuitively, it can be seen from or negative), rd, and r,. In order to appreciate the
model of the P W M switch, one should go through
Fig. 15 that the common terminal spends D-percent of
the method of state-space averaging and its canonical
its time in series with r, and D’-percent of its time in
circuit model employed in [13] and compare it with
series with rd so that the total effective resistance in
the method given below using the model of the P W M
the common terminal must be Dr, + D’rd as shown
switch which yields the same desired results almost
in Fig. 14. This can be verified formally if we write
by inspection. Thus, the model of the PWM switch in
the correct relationship between the average terminal
Fig. 14, without the control sources, is substituted in
voltages (as we did in (5)) in the presence of r, and rd
the Cuk converter as shown in Fig. 16 to determine
as
vCp= d(vap- icred’ - ecrt)- d’icrd.
(30) the open-loop line-to-output transfer function. Since
we are interested here only in the numerator, we
If this equation is perturbed, our intuitive derivation is
study this circuit in the transform domain under null
confirmed with a minor adjustment in VD
conditions of the output. The first null is clearly given
VD = V a p + (D - D’)lcrc zc(Td - r,).
(31) by the null of the impedance of the output ffilter as
in (21). The second null is given by the null in the
Almost always one can use the approximation VD N
s )0. For this second null we
inductor current ? ~ ~ ( =
Vapas can be seen from the above. The equations
can immediately see from Fig. 16 that
corresponding to the model of the PWM switch in Fig.
14 including all the parasitic effects, are summarized
below
+
ia 21 D;, + Ice?
(3h)
which can be immediately solved for s to give the
second zero
1
(34)
where
Recalling r, from (32c) and re = rc, from (4b) we get
from (34)
nn r
UY
s22 =
As a final example we consider the line-to-output
transfer function of the Cuk converter including
all the parasitic elements as discussed in [13].
Researchers in the early days of this converter were
puzzled with the peculiar behavior in the phase
response of the line-to-output transfer function
which at higher frequencies showed a variation from
-90 to 90 degrees. Since a similar behavior in the
VORP6RIAN: CONTINUOUS CONDUCTION MODE
Cc(Dr,
+ D‘rd 4- r,)
(35)
where
-
VD
-
VQP = (VO+%) - - Vo
(36)
D I me
where we have made use of the fact that for the Cuk
converter Vo/V, = D/D’. Hence, the numerator of the
rm
Zme
Ime
I me
495
~
Fig. 16. Cuk converter, including all parasitic elements, to be
analyzed for audio. Null condition corresponding to second zero of
audio transfer function is shown.
line-toautput transfer function can now be written as
where s1, and sZ2 are determined above in (21)
and (35), respectively, and D ( s ) is a fourth-order
polynomial. It is this second zero s,2 which is entirely
dependent on the parasitic resistances of the switches
that causes variations in the high end of the phase
response. It has been shown in [13] that rm can be
easily made sufficiently negative that the zero can
migrate from the right-half plane to the left-half plane
resulting in variations from -90 to 90 degrees at the
high end of the phase response.
CONCLUSIONS
A new, simple, and circuit-oriented method of
analysis of PWM converters, which uses the model of
the PWM switch developed in this article, is presented.
In this method, the model of the PWM switch is used
in the same way as the model of the transistor is used
in the analysis of electronic amplifier circuits. The
PWM switch is presented as a three-terminal nonlinear
device which represents the total nonlinearity in a
PWM converter just as the transistor represents the
total nonlinearity in an electronic amplifier. Hence,
as one does not linearize the entire equations of an
amplifier along with the Ebers-Moll equations of
the transistor to determine its dc and small-signal
characteristics, one does not need to linearize the
entire equations of a PWM converter as is done in
the case of state-space averaging, circuit averaging or
hybrid modeling.
Finally, with a model of the PWh4 switch
PWM converters can be easily analyzed for dc and
small-signal characteristics on standard electronic
circuit analysis programs in closed-loop operation
without the need for special-purpose programs.
REFERENCES
[l]
4%
Cuk, S. (1976)
Modelling, analysis and design of switching converters.
Ph.D. dissertation, California Institute of Technology,
Pasadena, Nov. 1976.
Middlebmk, R. D., and Cuk, S. (1976)
A general unified approach to modelling
switching-converter power stages.
In IEEE Power Electronics Specialists Conference
Proceedings, 1976.
VorptSrian, V. (1986)
Average mcdelof the single-pole double throw switch in
PWM converters.
Internal correspondence, Feb. 11, 1986.
VorptSrian, V., 'Qmerski, R., Liu, K., and Lee, E C. (1986)
Generalized cesonant switches: part I.
In 1986 W E C Conference Proceedings.
[email protected], V., 'Qmerski, R., Liu, K., and Lee, E C. (1986)
Generalized resonant switches: part 11.
In 1986 W E C Conference Proceedings.
VorptSrian, V. (1986)
Equivalent circuit models for PWM and resonant switches.
In Proceedings of 1986 Conference Internatwnal Symposium
or Circuits and S y s t m , 3.
qmerski, R., and Vorperian, V. (1986)
Generation, classification and analysis of switched-mode
dc-todc converters by the useof converter cells.
INTELEC Conference Proceedings, 1986.
Erickson, R. W. (1982)
Large signals in switching converters.
Ph.D. dissertation, California Institute of Technology,
Pasadena, Nov. 1982.
'Qmerski, R., Vorperian, V., Baumann, W., and Lee, E C.
Nonlinear modeling of the PWM switch.
Accepted for publication, to appear in the IEEE Journal of
Power Electronics.
'Qmerski, R. (1988)
Topology and analysis in power conversion and inversion
circuits.
Ph.D. dissertation, Virginia Polytechnic Institute and State
University, Blacksburg, VA, May 1988.
Cuk, S. (1977)
A new optimum topology switching dc-todc converter.
In IEEE Power Electronics Specialists Conference
Proceedings, 1977.
Middlebmk, R. D. (1975)
A continuous model of the tapped boost converter.
In IEEE Power Electronics Specialists Conference
Proceedings, 1975.
Polivka, W., Chetty, P., and Middlebrook, R. (1980)
State-space average modelling of converters with parasitics
and storage-time modulation.
In IEEE Power Electronics Specklists Conference
Proceedings, 1980.
'Qmerski, R. (1985)
The large signal forced response of switching amplifiers
using the Volterra functional series expansion.
Technical note T173, Power Electronics Group, Caltech,
Pasadena, CA, Apr. 1985.
VorptSrian, V.
Circuitaiented analysis of PWM converters using the
model of PWM switch.
These are examples worked out in detail using many
tricks of network analysis. Available from the author upon
request.
Landsman, E. E. (1979)
A unifying derivation of switching dcdc converter
topologies.
In IEEE Power Electronics Specialists Conference
Proceedings, 1979,239-243.
lor's photograph and biography will be found on
505 of this issue following Part I1 of this article.
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