POWER FACTOR CORRECTION (PFC) HANDBOOK, REV. 4
Power Factor Correction (PFC) Handbook
Choosing the Right Power Factor Controller Solution
HBD853/D
Rev. 4, Feb−2011
© SCILLC, 2011
Previous Edition E 2007
All Rights Reserved”
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Table of Contents
Page
Foreward . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Preface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Chapter 1: Overview of Power Factor Correction Approaches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Chapter 2: Methodology for Comparison of Active PFC Approaches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Chapter 3: Critical Conduction Mode (CrM) PFC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Chapter 4: Frequency Clamped Critical Conduction Mode (FCCrM) PFC . . . . . . . . . . . . . . . . . . . . . . . . . . 45
Chapter 5: Continuous Conduction Mode (CCM) PFC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59
Chapter 6: Interleaved PFC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71
Chapter 7: Bridgeless PFC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85
Chapter 8: Single Stage, Isolated Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97
Chapter 9: Detailed Analyses and Comparisons . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 107
For additional information on Power Factor Correction, contact the Technical Information Center at 800−282−9855 (from the
USA and Canada) or www.onsemi.com/tech−support.
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FOREWORD
Designing power supplies in a global energy efficiency context
Designing power supplies has always been a challenging task. But just as many of the traditional
problems have been solved, emerging regulatory standards governing efficiency levels are about to start
the cycle over again.
The first phase of this cycle is already well underway and has focused on improving standby power
consumption levels (passive mode). The next phase is tackling the tougher problem of improving active
mode efficiency levels. Government agencies around the world, driven by the US Environmental
Protection Agency (EPA) and its ENERGY STAR® program and by the China National Institute of
Standardization (CNIS), are announcing new performance standards for active mode efficiency for
power supplies.
The standards are aggressive and it will take the joint efforts of manufacturers and their suppliers
(including semiconductor suppliers) to provide solutions that meet the new challenges.
Amidst these trends, power factor correction (PFC) or harmonic reduction requirements as mandated
by IEC 61000−3−2 stands out as the biggest inflection point in power supply architectures in recent years.
With increasing power levels for all equipment and widening applicability of the harmonic reduction
standards, more and more power supply designs are incorporating PFC capability. Designers are faced
with the difficult tasks of incorporating the appropriate PFC stage while meeting the other regulatory
requirements such as standby power reduction, active mode efficiency and EMI limits.
ON Semiconductor is committed to providing optimal solutions for any given power supply
requirement. Our commitment is reflected in providing design guidance in choosing between many
options for topology and components. In this handbook we have attempted to provide a detailed
comparison between various options for PFC implementation while keeping it in the context of total
system requirements. As new technologies and components are developed, the balance of choice may
shift from one approach to the other, but the methodology used in this handbook will remain applicable
and provide a means for the power supply designer to arrive at the best choice for a given application.
We at ON Semiconductor sincerely hope this book will help you to design efficient, economical PFC
circuits for your products. Please see our Web site, www.onsemi.com, for up−to−date information on this
subject.
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Preface
Choices for the power factor correction solutions range from passive circuits to a variety of active circuits. Depending on
the power level and other specifics of the application, the appropriate solution will differ. The advances in the discrete
semiconductors in recent years, coupled with availability of lower priced control ICs have made the active PFC solutions more
appropriate in a wider range of applications. When evaluating the PFC solutions, it is important to look at them in the context
of full system implementation cost and performance.
This handbook is an updated version of the first PFC handbook published by ON Semiconductor in 2003. This current
version was updated with the assistance of Dhaval Dalal from Acptek ([email protected]).
In this handbook, a number of different PFC approaches are evaluated for a 300 W (400 V, 0.75 A) application. An overview
of select single−stage isolated converters including an LED driver is also presented. By providing step-by-step design
guidelines and system level comparisons, it is hoped that this effort will help the power electronics designers select the right
approach for their application.
Chapter 1 provides a comprehensive overview of PFC circuits and details of operation and design considerations for
commonly used PFC circuits.
Chapter 2 describes the methodology used for comparing different active PFC approaches for a given application (400 V, 0.75
A output). It also briefly introduces the proposed approaches.
Chapter 3 contains the design guidelines, discussion and salient operational results for the two variations of the critical
conduction mode topologies (voltage mode and current mode)
Chapter 4 contains the design guidelines, discussion and salient operational results for the Frequency Clamped CrM topology
Chapter 5 contains the design guidelines, discussion and salient operational results for the continuous conduction mode
topology
Chapter 6 contains the information on the interleaved PFC topology and salient operational results
Chapter 7 contains the information on the bridgeless PFC approach and salient operational results
Chapter 8 contains information on the single stage PFC operation and salient operational results
Chapter 9 provides a detailed analysis of the results obtained from the three different implementations (CrM, FCCrM and
CCM) for the same applications. Comparative analyses and rankings are provided for the topologies for given
criteria. It also includes guidelines for the designers based on the results described in the previous chapters.
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CHAPTER 1
Overview of Power Factor Correction Approaches
ABSTRACT
Designing power factor correction (PFC) into modern switched-mode power supplies (SMPS) has evolved over the past few
years due to the introduction of many new controller integrated circuits (ICs). Today, it is possible to design a variety of PFC
circuits with different modes of operation, each with its own set of challenges. As the number of choices has increased, so has
the complexity of making the choice and then executing the design. In this chapter, the design considerations and details of
operation for the most popular approaches are provided.
Introduction
Power factor correction shapes the input current of off-line power supplies to maximize the real power available from the
mains. Ideally, the electrical appliance should present a load that emulates a pure resistor, in which case the reactive power
drawn by the device is zero. Inherent in this scenario is the absence of input current harmonics---the current is a perfect replica
of the input voltage (usually a sine wave) and is exactly in phase with it. In this case the current drawn from the mains is at
a minimum for the real power required to perform the needed work, and this minimizes losses and costs associated not only
with the distribution of the power, but also with the generation of the power and the capital equipment involved in the process.
The freedom from harmonics also minimizes interference with other devices being powered from the same source.
Another reason to employ PFC in many of today’s power supplies is to comply with regulatory requirements. Today,
electrical equipment in Europe and Japan must comply with the IEC61000-3-2. This requirement applies to most electrical
appliances with input power of 75 W (Class D equipment) or greater, and it specifies the maximum amplitude of line-frequency
harmonics up to and including the 39th harmonic. Additionally, many energy efficiency requirements also carry a PFC
requirement such as the Energy Star 5.0 for Computers and Energy Star 2.0 for External Power Supplies, and for TV effective
November 2008.
Definition
Power factor correction is simply defined as the ratio of real power to apparent power, or:
PF +
(expressed in Watts)
Real Power
Apparent Power (expressed in VA)
where the real power is the average, over a cycle, of the instantaneous product of current and voltage, and the apparent power
is the product of the rms value of current times the rms value of voltage. If both current and voltage are sinusoidal and in phase,
the power factor is 1.0. If both are sinusoidal but not in phase, the power factor is the cosine of the phase angle. In elementary
courses in electricity this is sometimes taught as the definition of power factor, but it applies only in the special case, where
both the current and voltage are pure sine waves. This occurs when the load is composed of resistive, capacitive and inductive
elements and all are linear (invariant with current and voltage).
Switched-mode power supplies present nonlinear impedance to the mains, as a result of the input circuitry. The input circuit
usually consists of a half-wave or full-wave rectifier followed by a storage capacitor capable of maintaining a voltage of
approximately the peak voltage of the input sine wave until the next peak comes along to recharge the capacitor. In this case
current is drawn from the input only at the peaks of the input waveform, and this pulse of current must contain enough energy
to sustain the load until the next peak. It does this by dumping a large charge into the capacitor during a short time, after which
the capacitor slowly discharges the energy into the load until the cycle repeats. It is not unusual for the current pulse to be 10%
to 20% of the cycle, meaning that the current during the pulse must be 5 to 10 times the average current. Figure 1−1 illustrates
this situation.
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PFC Handbook
Top: Input Voltage
Bottom: Input Current
Figure 1−1. Input Characteristics of a Typical
Switched−Mode Power Supply without PFC
Note that the current and voltage are perfectly in phase, in spite of the severe distortion of the current waveform. Applying
the “cosine of the phase angle” definition would lead to the erroneous conclusion that this power supply has a power factor
of 1.0.
Figure 1−2 shows the harmonic content of the current waveform in Figure 1−1. The fundamental (in this case 60 Hz) is shown
with reference amplitude of 100%, and the higher harmonics are then given with their amplitudes shown as percentages of the
fundamental amplitude. Note that the even harmonics are barely visible; this is a result of the symmetry of the waveform.
Since only the fundamental component produces real power, while the other harmonics contribute to the apparent power,
the actual power factor is well below 1.0. This deviation is represented by a term called distortion factor and is primarily
responsible for the non-unity power factor in SMPS. The general equation governing the relationship between the real power
and apparent power is given by:
Real power expressed in W
Apparent power expressed in VA
ǀPinǁ + Vin(rms) · Iin(rms) · cos ö. · cos q
Where cosϕ is the displacement factor coming from the phase angle ϕ between the voltage and current waveforms and cosθ
is the distortion factor. Incidentally, the power factor of the power supply with the waveform in Figure 1−2 is approximately
0.6.
100%
80%
60%
40%
20%
0%
1
3
5
7
9
11
13
15
17
19
21
Harmonic Number
Figure 1−2. Harmonic Content of the Current Waveform in Figure 1−1
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For reference, Figure 1−3 shows the input of a power supply with perfect power factor correction. It has a current waveform
that mimics the voltage waveform, both in shape and in phase. Note that its input current harmonics are nearly zero.
Top: Input Voltage
Bottom: Input Current
100%
80%
60%
40%
20%
0%
1
3
5
7
9
11
13
15
17
19
21
Harmonic Number
Figure 1−3. Input Characteristics of a Power Supply with Near−Perfect PFC
Power Factor Correction vs. Harmonic Reduction
It is clear from the previous illustrations that high power factor and low harmonics go hand-in-hand. It is generally thought
that specifying limits for each of the harmonics will do the better job of controlling the “pollution” of the input current, both
from the standpoint of minimizing the current and reducing interference with other equipment. So, while the process of shaping
this input current is commonly called “power factor correction,” the measure of its success in the case of the international
regulations is the harmonic content. In the case of SMPS, usually the displacement factor is close to unity, so the following
relationships between the harmonic distortion and power factor apply.
Ǹȍ
R
THD(%) + 100 @
Ip2
p+2 I12
cos q + PF +
1
Ǹ1 ) THD
2
Here, THD is the Total Harmonic Distortion which is quadratic sum of the unwanted harmonics over the fundamental that
gives the relative weight of the harmonic content with respect to the fundamental. The second equation uses the absolute value
of THD (not percentage) and demonstrates that THD has to be zero for PF to be unity.
Types of Power Factor Correction
The input characteristics shown in Figure 1−3 were obtained with “active” power factor correction, using a switched-mode
boost converter placed between the input rectifier and the storage capacitor, with the converter controlled by a PFC IC
(Integrated Circuit) and its attendant circuitry in a manner to shape the input current to match the input voltage waveform. This
is the most popular type of PFC used in today’s power supplies, as shown in Figure 1−4. It isn’t the only type, however. There
are no rules demanding that the PFC task be accomplished by active circuits (transistors, ICs, etc.). Any method of maintaining
the harmonics below the regulatory limits is fair game. It turns out that one inductor, placed in the same location as the active
circuit, can do the job. An adequate inductor will reduce the peaks of the current and spread the current out in time well enough
to reduce the harmonics enough to meet the regulations. This method has been used in some power supplies where the large
size of the inductor and its weight (due to its iron core and copper winding) are not objectionable. At higher power levels, the
size and weight of the passive approach become unpopular. Figure 1−5 shows the input characteristics of three different 250-W
PC power supplies, all with the current waveforms at the same scale factor. As shown, the peak current levels in passive PFC
circuit are still 33% higher than the peak currents in active circuits. In addition, while the harmonic levels of the second levels
may meet the IEC61000-3-2, it will fail the more stringent 0.9 PF requirement being imposed by some recent regulations.
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PFC Handbook
Rectifiers
PFC Preconverter
AC
Line
+
High
Frequency
Bypass
Capacitor
Active
Power Factor
Controller
Converter
+
Bulk
Storage
Capacitor
Load
Figure 1−4. PFC Preconverter Stage
In recent years, market trends (rising cost of copper and magnetic core material and falling costs of semiconductors) have
tilted the balance decidedly in favor of active PFC even in the most cost-sensitive consumer applications. Coupled with the
additional system benefits afforded by the active PFC circuits [1], this seems to be a trend that is likely to continue in the future
and lead to more advanced active PFC solutions becoming available to the designers.
Waveforms:
1. Input current with no PFC
2. Input current with passive PFC
3. Input current with active PFC
4. Input voltage
Figure 1−5. Input Characteristics of PC Power Supplies with Different PFC Types (None, Passive, and Active)
Input Line Harmonics Compared to IEC61000-3-2
Figure 1−6 shows the input harmonics of three 250-W PC power supplies, along with the limits according to IEC61000-3-2.
These limits are for Class D devices, which include personal computers, televisions and monitors. The harmonic amplitudes
are proportioned to the input power of these devices. For lighting products, class C limits are applied, which are also
proportional to input power and even more stringent. In the case of other products not used in such high volume, the limits are
fixed at the values corresponding to 600 W input. The performance of the passive PFC, as shown in this graph, just barely
complies with the limit for the third harmonic (harmonic number 3).
10.000
1.000
IEC61000−3−2 Limit
Unit A, Without PFC
0.100
Unit B, Passive PFC
Unit C, Active PFC
0.010
0.001
3
13
23
33
Harmonic Number
Figure 1−6. Input Harmonics of Three PC Power Supplies Relative to IEC61000−3−2 Limits
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Passive PFC
Figure 1−7 shows the input circuitry of the PC power supply with passive PFC. Note the line-voltage range switch connected
to the center tap of the PFC inductor. In the 230-V position (switch open) both halves of the inductor winding are used and the
rectifier functions as a full-wave bridge. In the 115-V (switch closed) position only the left half of the inductor and the left half
of the rectifier bridge are used, placing the circuit in the half-wave doubler mode. As in the case of the full-wave rectifier with
230 Vac input, this produces 325 Vdc (230 S √2) at the output of the rectifier. This 325 Vdc bus is, of course, unregulated and
moves up and down with the input line voltage.
Inrush Current
Limiter
(Thermistor)
0.01
0.0047
PFC Inductor
+
230 Vac
0.1
0.022
470
325 Vdc to
Forward
Converter
0.22
115 Vac
1M
Differential
Mode
Inductor
(L2)
Common
Mode
Inductor
(L3)
0.022
470
−
0.0047
Figure 1−7. Passive PFC in a 250 W PC Power Supply
The passive PFC circuit suffers from a few disadvantages despite its inherent simplicity. First, the bulkiness of the inductor
restricts its usability in many applications. Second, as mentioned above, for worldwide operation, a line-voltage range switch
is required. Incorporation of the switch makes the appliance/system prone to operator errors if the switch selection is not
properly made. Finally, the voltage rail not being regulated leads to a cost and efficiency penalty on the dc-dc converter that
follows the PFC stage.
Critical Conduction Mode (CrM) Controllers
Critical Conduction Mode or Transitional Mode (also known as Borderline Conduction Mode BCM) controllers are very
popular for lighting and other lower power applications. These controllers are simple to use as well as inexpensive. A typical
application circuit is shown in Figure 1−8.
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PFC Handbook
Output
Vin
+
−
2.5 V
Error
Amp
Zero
Current
RDC1
Rac1
Current
Shaping
Network
Reference
Multiplier
AC Input
Cout
Control
Logic
Out
Rac2
RDC2
Rshunt
Out Gnd
Figure 1−8. Basic Schematic for a Critical Conduction Mode Converter
The basic CrM PFC converter uses a control scheme (current mode control) similar to that shown above. An error amplifier
with a low frequency pole provides an error signal into the reference multiplier. The other input to the multiplier is a scaled
version of the input rectified ac line voltage. The multiplier output is the product of the near dc signal from the error amplifier
and the full-wave rectified sine waveform at the ac input.
The signal out of the multiplier is also a full-wave rectified sine wave that is scaled by a gain factor (error signal), and is used
as the reference for the input voltage. The amplitude of this signal is adjusted to maintain the proper average power to cause
the output voltage to remain at its regulated value.
The current shaping network forces the current to follow the waveform out of the multiplier, although the line frequency
current signal (after filtering) will be half of the amplitude of this reference. The current shaping network functions as follows:
Vref
Iinductor
Iavg
V
t
Figure 1−9. CRM Waveforms
In the waveforms of Figure 1−9, Vref is the signal out of the multiplier. This signal is fed into one input of a comparator, with
the other input connected to the current waveform.
When the power switch turns on, the inductor current ramps up until the signal across the shunt reaches the level of Vref.
At this point the comparator changes states and turns off the power switch. With the switch off, the current ramps down until
it reaches zero. The zero current sense circuit measures the voltage across the inductor, which will fall to zero when the current
reaches zero. At this point the switch is turned on and the current again ramps up.
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As the name implies, this control scheme keeps the inductor current at the borderline limit between continuous and
discontinuous conduction, or critical conduction. This is important, because the wave shape is always known, and therefore,
the relationship between the average and peak current is also known. For a triangular waveform, the average is exactly one
half of the peak. This means that the average current signal (Inductor current S Rsense) will be at a level of one half of the
reference voltage.
The frequency of this type of regulator varies with line and load. At high line and light load, the frequency is at a maximum,
but also varies throughout the line cycle (high frequency near zero crossing and low frequency near the peak).
Critical Conduction Mode without a Multiplier (Voltage Mode)
A novel approach to the critical conduction mode controller is available in some ON Semiconductor ICs, most recent
example being NCP1607. These chips provide the same input-output function as the controllers described above; however they
accomplish this without the use of a multiplier [2].
As was explained in the previous section, the current waveform for a CrM controller ramps from zero to the reference signal
and back to zero. The reference signal is a scaled version of the rectified input voltage, and as such can be referred to as k S
Vin, where k is a scaling constant from the ac voltage divider, error amplifier and multiplier in a classic circuit. Given this, and
knowing the relation of the slope of the inductor with the input voltage, the following are true:
k Vin
Iinductor
Ipk + k · Vin(t) and Ipk +D I +
Vin(t)
· ton
L
Figure 1−10. CRM Current Envelope
Equating the peak current for these two equations gives:
k · Vin(t) +
Vin(t)
t
L on
Therefore, ton + k · L
This equation shows that ton is a constant for a given reference signal (k S Vin). Toff will vary throughout the cycle, which
is the cause of the variable frequency that is necessary for critical conduction. The fact that the on time is constant for a given
line and load condition is the basis for this control circuit.
In the circuit of Figure 1−11, the programmable one-shot timer determines the on time for the power switch. When the on
period is over, the PWM will switch states and turn off the power switch. The zero current detector senses the inductor current,
and when it reaches zero, the switch is turned on again. This creates the same dc output as with the classic scheme, without
the use of the multiplier. The benefit of the voltage mode CrM control is that the multiplier is not needed and the input voltage
sensing network is eliminated. In addition, the current sensing is needed only for protection purpose.
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PFC Handbook
Vbulk
Error
Amplifier
FB
2.5 V
Vcontrol
Cl
PWM
Disable
DRV
Logic
Control
DRV
CS
QCL
DRV
Figure 1−11. Simplified Schematic of CrM Controller Without Multiplier
Since a given value of on time is only valid for a given load and line condition, a low frequency error amplifier for the dc
loop is connected to the one-shot. The error signal modifies the charging current and therefore, the on time of the control circuit
so that regulation over a wide range of load and line conditions can be maintained.
One of the voltage mode CrM controllers, the MC33260 contains a number of other features including a circuit that will allow
the output voltage to follow the input voltage. This is called follower boost operation (shown in Figure 1−12). In the follower
boost mode, the output voltage is regulated at a programmed level above the peak of the input voltage. In most cases, the output
of the PFC converter is connected to a dc-dc converter. Many dc-dc converter topologies (e.g. flyback converters) are capable
of regulating over a wide range of input voltages, so a constant input voltage is not necessary. On the other hand, if a topology
can not function well over a wide input range, the follower boost output range needs to be narrowed (if it is used).
Vout
Vac
Figure 1−12. Follower Boost
Follower boost operation offers the advantages of a smaller and therefore, less expensive inductor, and reduced on-time
losses for the power FET [3]. This is normally used in systems where the lowest possible system cost is the main objective.
Frequency Clamped Critical Conduction Mode (FCCrM)
Although the Critical Conduction Mode is widely used in the industry, it has some known limitations. The primary limitation
being the variable switching frequency which reaches peak at light loads and also near the zero crossing of the sinusoid. Some
solutions which clamped the frequency excursion by putting a maximum frequency clamp resulted in the distortion of current
(since the Ton was not adjusted for this) and lower power factor as the inductor entered the discontinuous mode of operation.
This is illustrated in Figure 1−13.
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Figure 1−13. CrM Line Current Distortion Due to Frequency Clamping
Recently, a new technique has been introduced which allows true power factor correction even in discontinuous mode
(DCM). This technique is summarized in Figure 1−14 and following equations.
tcycle = ton + tdemag
ICOIL
ton
tdemag
tDT
Time
tcycle
TSW
Figure 1−14. DCM Operating Waveforms
With reference to Figure 1−14, the coil peak current is given by:
Icoil, pk +
vin(t)
L
· ton
The average coil current over a switching cycle (which is also taken as the instantaneous line current for that switching cycle,
since the switching frequency is much higher compared to the line frequency at which the line voltage varies) is given by:
Icoil, pk tcycle
ǀIcoilǁ
+
·
+ Iin(t)
TSW
2
TSW
Combining these equations and simplifying leads to:
Iin(t) +
vin(t)
2·L
ǒ
· ton ·
tcycle
TSW
Ǔ
From this equation, we can deduce that, if we devise an algorithm that keeps ton S tcycle /Tsw constant for a given load and
line condition, we can achieve a sinusoidal line current and unity power factor even in the discontinuous mode. ON
Semiconductor has introduced NCP1601 and NCP1605 which incorporate this principle.
Despite its benefit of fixed frequency, the DCM is not the ideal mode in all situations as it leads to highest peak current levels.
A comparison of the three different modes of operation is shown in Figure 1−15.
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PFC Handbook
Symbol
Rating
Unit
IL
Continuous
Conduction Mode
(CCM)
S Always hard−switching
S Inductor value is largest
S Minimized rms current
Discontinuous
Conduction Mode
(DCM)
S Highest rms current
S Reduce coil inductance
S Best stability
Critical
Conduction Mode
(CrM)
S Largest rms current
S Switching frequency is not fixed
IL
IL
Figure 1−15. Comparison of PFC Operating Modes
A more judicious choice would be to allow the PFC to slide between the DCM and CrM modes seamlessly and extract the
best of both worlds. So, at light loads, when CrM can go to high switching frequency, it is preferable to go into DCM. Similarly,
when the load current is higher, it is desirable to stay in CrM to avoid the high peak currents. This optimization is best depicted
by Figure 1−16. NCP1601 and NCP1605 offer programmable frequency clamp that enables selection of appropriate mode
boundary. As shown in Figure 1−16, the option 3 is the ideal solution as it combines the best of both worlds (low frequency
variation and contained peak currents). More details and results of this mode of operation are provided in Chapter 4.
Figure 1−16. PFC Mode Selection Through Frequency Clamping
Continuous Conduction Mode (CCM) Control
The Continuous conduction mode control has been widely used in a broad range of applications because it offers several
benefits. First, the peak current stress is low and that leads to lower losses in the switches and other components. Also, input
ripple current is low and at constant frequency, making the filtering task much easier. The following attributes of the CCM
operation need further consideration.
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Vrms2 Control
As is the case with many of the PFC controllers on the market, one essential element is a reference signal that is a scaled
replica of the rectified input voltage, which is used as a reference for the circuit that shapes the current waveform. These chips
all use a multiplier to accomplish this function; however, the multiplier system is more complex than a conventional two-input
multiplier.
Figure 1−17 shows the classic approach to continuous-mode PFC. The boost converter is driven by an average current-mode
pulse width modulator (PWM) that shapes the inductor current (the converter’s input current) according to the current
command signal, Vi. This signal, Vi, is a replica of the input voltage, Vin, scaled in magnitude by VDIV. VDIV results from
dividing the voltage error signal by the square of the input voltage (filtered by Cf, so that it is simply a scaling factor proportional
to the input amplitude).
IIN
VIN
Load
PWM
Control
Ac in
Vi
KIN
Kf
Cf
VSIN
Kf VIN
Mult.
KM
IIN
VDIV
Square
KS
Div.
KD
-
E/A
VERR
+
VREF
KS Kf2 VIN2
Vi + VDIV · KM · VSIN +
VERR · KM · VSIN
KD · KS · Kf2 · VIN2
Figure 1−17. Block Diagram of the Classic PFC Circuit
It may seem unusual that the error signal is divided by the square of the input voltage magnitude. The purpose is to make
the loop gain (and hence the transient response) independent of the input voltage. The voltage squared function in the
denominator cancels with the magnitude of VSIN and the transfer function of the PWM control (current slope in the inductor
is proportional to the input voltage). The disadvantage of this scheme lies in the production variability of the multiplier. This
makes it necessary to overdesign the power-handling components, to account for the worst-case power dissipation.
Average Current Mode Control
The ac reference signal output from the multiplier (Vi) represents the wave shape, phase and scaling factor for the input
current of the PFC converter in Figure 1−17. The job of the PWM control block is to make the average input current match
the reference. To do this, a control system called average current mode control is implemented in these controllers [4], [5]. This
scheme is illustrated in Figure 1−18.
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Output
L
Vs
Oscillator
Vin
Vca
+
Cout
PWM
Rshunt
−
Rcp
+
Out Gnd
−
+
Icp
+
Current
Amplifier
Figure 1−18. Diagram for Average Current Mode Control Circuit
Average current mode control employs a control circuit that regulates the average current (input or output) based on a control
signal Icp. For a PFC controller, Icp is generated by the low frequency dc loop error amplifier (and it is simply the current
equivalent of the signal Vi as depicted in Figure 1−17. The current amplifier is both an integrator of the current signal and an
error amplifier. It controls the wave shape regulation, while the Icp signal controls the dc output voltage. The current Icp
develops a voltage across Rcp. For the current amplifier to remain in its linear state, its inputs must be equal. Therefore, the
voltage dropped across Rshunt must equal the voltage across Rcp, since there can be no dc current in the input resistor to the
non-inverting input of the current amplifier. The output of the current amplifier is a “low frequency” error signal based on the
average current in the shunt, and the Icp signal.
This signal is compared to a sawtooth waveform from an oscillator, as is the case with a voltage mode control circuit. The
PWM comparator generates a duty cycle based on these two input signals.
ON Semiconductor NCP1650 Family
ON Semiconductor offers a line of highly integrated PFC controllers, with a novel control scheme [6]. This chip’s control
circuit uses elements from the critical conduction mode units, as well as an averaging circuit not used before in a power factor
correction chip. The basic regulator circuit includes a variable ac reference, low frequency voltage regulation error amplifier
and current shaping network.
This chip incorporates solutions to several problems that are associated with PFC controllers, including transient response,
and multiplier accuracy. It also includes other features that reduce total parts count for the power converter [7]. The simplified
block diagram of this approach is shown in Figure 1−19. More details of this approach can be found in the references provided
at the end of this chapter.
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L
Vin
Rect −
+
FB/SD
Loop
Comp
−
4V
1.08 Vref
Error
Amp
Output
−
DS
+
Overshoot
Comparator
RDC1
Rac1
Current
Shaping
Network
Reference
Multiplier
R7
AC Input
Oscillator
C7
Rac2
Cout
Control
Logic
Current
Sense
Amplifier
Q1
Out
RDC2
+
Rshunt
−
IS
−
Out Gnd
Rect −
Figure 1−19. Simplified Block Diagram of the NCP1650 PFC Controller
In addition to the NCP1650, which works in a traditional boost PFC topology, the NCP165x family also consists of NCP1651
and NCP1652. The NCP1651 and NCP1652 allow a single-stage, isolated step-down power conversion with PFC for many
low-mid power applications where the output voltage is not very low and can handle some ripple. As shown in Figure 1−20,
the NCP1651 based flyback converter provides a uniquely simple alternative to two-stage approaches commonly used. The
NCP1651 includes all the relevant significant feature improvements of the NCP1650 and also includes a high-voltage start-up
circuitry.
TO VCC
VIN
VO
ACIN
STARTUP
OUT
VCC GND CT
FB
Secondary FB
& Protection
IS
Figure 1−20. Single Stage PFC Using the NCP1651
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Predictive Control of CCM PFC
The preceding section outlined some of the inherent complexities of CCM PFC control and how NCP1650 helps overcome
some of those complexities. In recent years, a newer control technique has been introduced which greatly simplifies the control
algorithm of the CCM PFC controllers. As incorporated in the NCP1653 and NCP1654 from ON semiconductor, this technique
is known as predictive control since it uses the sensed current to determine (predict) the required duty cycle instead of
generating the reference signal based on input voltage sensing.
The average inductor current in a system with good PF must be proportional to the input voltage:
V
ǀIcoil(t)ǁ
+ Iin(t) T Vin(t), and doff + in
TSW
Vout
for CCM operation, where doff is the duty cycle of power switch off time.
ǀ
ǁ
Hence doff T Icoil(t) TSW
The way the predictive CCM PFC controller works is to control the power switch on time by summing a ramp signal with
a signal proportional to the coil current. As a result, the higher the sensed coil current, lower the on time and higher the off time,
satisfying the relationship above. Figure 1−21 shows the current shaping scheme. Some ramp is summed with a signal
proportional to the coil current.
vsum + Vsense )
Iref · ton
Cramp
The power switch stops conducting when Vsum exceeds the current reference. Hence, one can deduct ton and consequently
doff .
t
doff + 1 * on
TSW
doff + 1 *
(Iref · TSW) * (Cramp · Vref) Cramp · Vsense
)
Iref · TSW
Iref · TSW
If Iref TSW = Cramp Vref, i.e. Iref , Cramp , and Vref also act as the oscillator to control the operating frequency, one can obtain
V
doff + sense + k · Icoil(t)
Vref
which leads to near-unity power factor. More details on this approach are provided in a later chapter.
Vref
(Current Reference)
Ramps
Sensed Coil Current
Figure 1−21. Predictive CCM Control Waveforms
Advanced Approaches for PFC
The major control algorithms (CrM, CCM and DCM) and their combinations allow many options for the designers. In
addition to these, the search for higher efficiency and modularity has lead to advanced architectures being utilized for the
leading edge applications. These approaches are getting into the mainstream applications only now. However, given their
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highly advanced nature, the designer has to be careful about staying clear of any intellectual property (IP) implications when
considering these approaches. In this handbook, two such advanced approaches are presented in later chapters.
The bridgeless PFC solutions arose from the recognition that the diode bridge at the front-end of any PFC typically
contributes 2% power losses at full load. If the bridge can either be eliminated or combined with other functions, these losses
can be averted. With this in mind, many topologies have been presented in the industry publications and also have been use
in some of the higher end applications (UPS being one of them) in the past few years. The bridgeless solutions involve distinctly
more complex control and also require acute awareness of the grounding loops when implementing them. Most known
implementations involve moving the boost inductor to the ac side of the bridge and replacing the lower diodes of the rectifier
bridge with switches in order to replicate boost converters for each leg.
Another recent trend is to apply the interleaving concept to the PFC circuits. In interleaving operation, a single converter
is replaced by 2 or more paralleled converters each operating out of phase so that the ripple current when summed at the output
or input has a cancelling effect and results in lower filtering requirements. Other benefits of interleaving are modularization,
heat dispersion and ability to optimize cost/performance of a smaller module which is much easier due to component
availability. Against this, there are potential negatives such as higher component count and a more complex control function.
Component Selection for PFC Circuits
The basic PFC boost converter is one of the simplest converter types (along with buck and buck-boost converters) around.
Consequently, the number of components required for power stage is minimal – one inductor, one power switch, one diode
and one output capacitor. So, when adding an active PFC circuit to an existing power converter, the component requirements
are not very complex. Additional components such as the input bridge and EMI filter are already existent in all ac connected
power converters.
While the power stage is simple, the component selection is by no means trivial and there are many critical choices to be
made while optimizing the design for required performance. Given the recent trend for higher efficiency, the component
selection plays an even more significant role and it has been shown that a proper component selection alone can boost the PFC
efficiency by 2-3% for a given topology.
PFC Inductor
The PFC inductor (also referred to as boost inductor or coil or PFC choke), is very important to the operation of the PFC
circuit. It must be designed to prevent saturation and consistently provide good power factor. The value of inductance is selected
based on commonly available equations. For CCM, the value of inductor tends to be higher than the value for the DCM or CrM
operation. However, that does not mean that the size of the inductor is always higher for the CCM operation. The size depends
on the inductance value and the rms current through the inductor. The key consideration is in selecting the right core material
and winding size for a given inductor. The higher peak-peak current ripple means that the core losses are higher for the CrM
operation. It is often more difficult to use a cheap powder iron core for the CrM operation and achieve the required efficiency.
The other key consideration is the type of core. Toroids are the most popular because they offer low cost, but if the number
of winding turns is high, a bobbin based winding may facilitate easier winding. Some advanced designs use this approach with
Ferrite cores to get better flux control.
PFC Diode
The choice of PFC diode plays a crucial role in the efficiency and EMI performance of a CCM boost converter. At the instance
of the boost switch turn-on, the diode is carrying significant current in the CCM operation. Since this diode is a high voltage
diode, it typically suffers from reverse recovery phenomenon (forced by slower recombination of minority carriers) which adds
to the losses and ringing. It is important to use ultrafast diodes (preferably with soft recovery characteristics) to mitigate this
problem. In recent years, alternative rectifier solutions based on more advanced materials have been proposed, but cost
constraints limit their applicability in mainstream applications. For the CrM or FCCrM operations, the requirement is different
since the diode always turns off at zero current and hence there are no reverse recovery issues to be faced. In these applications,
the important criterion is to optimize the forward drop of the diode to improve efficiency. ON Semiconductor has recently
introduced PFC diodes (MUR550 series) for these applications.
PFC Switch
The choice of PFC switch is based on the cost vs. performance trade-off. The recent advances in MOSFET technology has
helped in moving this trade-off more quickly in the direction of higher performance. With a 500 V or 600 V FET, the important
issue is to select the FET with right level of Rds(on) to get low conduction losses without increasing switching losses
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significantly. Blindly selecting the lowest available Rds(on) FET will not yield the highest efficiency and will actually increase
the cost of implementation. Of equal importance is effective MOSFET drain capacitance. This capacitance must be charged
and discharged every switching cycle. Choosing a MOSFET with low capacitance will reduce switching losses and increase
efficiency.
Current Sense Resistor
The current sense resistor is another important contributor to conduction losses and it is important to minimize the voltage
drop across it at full load. In higher end designs, this is achieved by employing current sense transformers. However, in more
typical designs, the choice of current sense resistor is dictated by the requirements of the PFC controllers. Many controllers
require a fixed 1 V signal and that leads to contribution of about 1% losses. In many of ON Semiconductor’s PFC controllers,
through negative sensing scheme, the current sense signal is user programmable and hence can be optimized further.
Conclusion
The number of choices available to the PFC designer has grown significantly over the past few years. This is due to the
increased interest in complying with IEC61000-3-2 and its derivatives, coupled with an enthusiastic spirit of competition
among the semiconductor suppliers. The end users reap increasing benefits as PFC becomes better and more cost effective.
Power Supply designers benefit from the increasing capability of these IC controllers, with more options available to execute
the designs.
On the other hand, the designer’s job has become more complicated as a result of the plethora of design approaches at his
fingertips. Just surveying them is difficult enough, but understanding each of them well enough to make an informed,
cost-effective choice is a big challenge. It has been an objective of this chapter to increase the designer’s awareness of this trend
and to provide some insight into the details. In the remaining chapters of this handbook, we expand on the individual approaches
and attempt to provide benchmarking that will make this selection easier.
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References
The following references were chosen for their relevance to the material in this paper, and are but a small sample of the vast
library available to the interested reader.
[1] “Boosting Power Supply Efficiency for Desktop Computers”, Dhaval Dalal, Power Electronics Technology
Magazine, February 2005. http://powerelectronics.com/mag/power_boosting_power_supply/
[2] “Design of Power Factor Correction Circuit Using GreenlineTM Power Factor Controller MC33260,” AND8016/D,
Rev. 1, Ming Hain Chew, ON Semiconductor, June 2002.
[3] “An Innovative Controller for Compact and Cost-Effective PFC Solutions,” Joel Turchi, ON Semiconductor,
www.chipcenter.com/analog/tn029.htm.
[4] “High Power Factor Preregulators for Off-Line Power Supplies,” Lloyd H. Dixon, Jr., Unitrode (now Texas Inst.),
Power Supply Design Seminar, SEM-800, 1991.
[5] “Average Current Mode Control of Switching Power Supplies,” Lloyd H. Dixon, Jr., Unitrode (now Texas Inst.),
Application Note U140.
[6] “NCP1650/D Power Factor Controller,” Rev. 1, Alan Ball, ON Semiconductor, March 2002.
[7] “NCP1650 Benchtop Assistance,” AND8084, Rev 0, Alan Ball, ON Semiconductor, May 2002.
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CHAPTER 2
Methodology for Comparison of Active PFC Approaches
There are many different driving factors for designing PFC circuits as outlined in Chapter 1. Depending on end
applications requirements and the prominent driving factors, the choice of a PFC circuit will vary. Until very recently, only
one or two topologies have been widely utilized for PFC implementations. For higher power circuits, the traditional topology
of choice is the boost converter operating in continuous conduction mode (CCM) and with average current mode control
(ACMC). For lower power applications, typically the critical conduction mode (CrM) boost topology is utilized. As the range
of circuits and applications incorporating PFC has expanded, the need for more diversified PFC solutions has grown. Many
of the emerging solutions use variations of the established topologies, while some truly novel techniques have also emerged.
It is often difficult to provide an instantaneous answer to the question: “Which approach is the most suitable for a given
application or power range?” The answer depends in part on the design priorities and various trade-offs. However, the other
part of the answer lies in benchmarking of different approaches for a given application. In this handbook, results of such
a benchmarking effort have been presented with detailed analysis. Previous version of the PFC handbook by ON
Semiconductor (published in 2003) also presented similar benchmarking, but the technology enhancements since then have
shifted the basis of comparison somewhat.
The choice of a correct application is critical in carrying out such a benchmarking study. It is commonly accepted that
at power levels below 200 W, the CrM approach is more appropriate, while for power levels above 300 W, the CCM approach
is admittedly sensible. However, in the market, there is no dearth of power supplies which implement CCM at lower power
and CrM at higher power – ultimately, it is the designer’s comfort factor that counts. The power range of 200-300 W
represents the gray area where either approach could be used. As a result, it is most pertinent to evaluate the performance
of different approaches somewhere within this power range. A 270 W (output) power level was chosen as a target application.
Also, since most applications are required to operate over universal input voltage (88-264 Vac, 50/60 Hz), that was chosen
as the input voltage range. All the systems were designed to a hold-up time (line drop-out) specification of 16 ms (1 line
cycle). The output voltage for the benchmarking is chosen to be 385 Vdc which is commonly used for the universal input
PFC applications.
Choice of Approaches
From the approaches described in Chapter 1 and other available approaches, following were identified as the suitable
candidates for this benchmarking. The accompanying figures for each approach depict the complete system implementation
including input filtering.
1. Critical Conduction Mode (CrM) boost converter with fixed output voltage. As shown in Figure 2−1, this approach
creates a fixed (385 V) output voltage at the PFC output using NCP1607 – a new voltage mode CrM controller
from ON Semiconductor.
85−265 V
50/60 Hz
AC Input
F1
EMI filter for
variable freq
ripple
P1
PFC Boost
Front−end using
NCP1607
385 V
DC out
Figure 2−1. Critical Conduction Mode PFC
2. Frequency Clamped Critical Conduction Mode (FCCrM) boost converter with variable output voltage. As shown in
Figure 2−2, this approach uses the Frequency Clamped CrM (where the DCM and CrM approaches are judiciously
mixed), to generate the 385 V dc output. NCP1605 is used as the PFC controller.
85−265 V
50/60 Hz
AC Input
F2
EMI filter for
clamped freq
ripple
P2
PFC Boost
Front−end using
NCP1605
385 V
DC out
Figure 2−2. Frequency Clamped Critical Conduction Mode PFC
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3. Continuous conduction mode boost converter with fixed output voltage. As shown in Figure 2−3, this approach
creates a fixed (385 V) output voltage using a CCM boost topology. The NCP1654 is used as the PFC controller for
this approach.
85−265 V
50/60 Hz
AC Input
F3
EMI filter for
fixed freq
ripple
P3
PFC Boost
Front−end using
NCP1654
385 V
DC out
Figure 2−3. Continuous Conduction Mode PFC
4. Interleaved converter with fixed output voltage. Since there is an increasing level of interest in interleaving of PFC,
additional benchmarking is provided for the interleaved converter in the same power range as shown in Figure 2−4.
85−265 V
50/60 Hz
AC Input
F4
EMI filter for
clamped freq
ripple
P4
Interleaved PFC Boost
Front−end using
NCP1631
385 V
DC out
Figure 2−4. Interleaved PFC
Test Methodology
All the above PFC approaches (P1-P4) were designed, built and characterized. Each converter went through minor
modifications in order to achieve local optimization without making major component changes. It is recognized that each
approach can be optimized further through a more aggressive design and selection of components. However, the focus of
this work was to compare the different approaches and the design approach for all the circuits was very similar. Each PFC
circuit was tested for the following parameters:
1.
2.
3.
4.
Operation over line and load ranges (Vin = 85 to 265 Vac, zero to full load)
Line and load regulation
Input current total harmonic distortion (THD), individual harmonic contributions, and power factor
Power Conversion Efficiency (Vin = 100, 115, 230 Vac, Pout = 20%, 50% and 100% of full load)
The test set-up is depicted in Figure 2−5 below.
Power
Analyzer
Pin, Vin, Iin
PF, THD
A
Out+
Sense+
AC Line
AC Source
0−300Vac
2 kVA
Unit
Under
Test
V
Sense−
Out−
Figure 2−5. Test Set−Up for Performance Measurements
Equipment used for measurements
AC Source: Chroma AC Source (6520)
Power Analyzer: YOKOGAWA Precision Power Analyzer (WT21)
Load: Chroma Electronic Load (63105) was used
Voltmeter: Agilent Digital Multimeter (34401A)
Current Meter: Current measurement were done by the Agilent Digital Multimeter (34401A)
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ON Semiconductor
The circuits were tested utilizing an isolated ac source with input voltages ranging from 85 to 265 Vac. Input parameters
were measured with the power analyzer. They included input power (Pin), rms input voltage (Vin), rms input current (Iin),
power factor level (PF), and total harmonic distortion (THD). All the measurements were made after the circuit was
thermally stabilized by operating it at full load and 100 Vac input for 30 minutes.
The output voltage was measured directly at the output sense pins using a Kelvin sensing scheme. There was virtually no
current flowing through the sense leads and therefore no voltage drop that can cause an erroneous reading. On the contrary,
measuring output voltage across the resistor load can cause a wrong reading as voltage drops occur between the UUT and
the load, the voltage drop varying with the amount of current flowing. The load current was measured using the current meter
built in the Agilent Digital Multimeter.
The tests were conducted in similar settings for all the boards. However, it should be realized that there is always an
element of inaccuracy in PFC power measurements due to the nature of these measurements. This inaccuracy is more
pronounced at lighter load conditions, so the reader is advised to take this into account whenever interpreting the results
presented in this handbook and also those taken in any lab environment.
Criteria for Comparisons
The comparisons were carried out between the performances of circuits P1−P3. These are summarized in Chapter 9. The
key metrics for comparing power systems are cost, size and performance. It is not possible to provide an absolute cost metric
for this handbook as the cost structures depend on many factors. However, the comparisons take into account relative costs
of different approaches and provide details of the trade-offs involved. The size comparison is based on comparison of the
sizes of major power train components for the different approaches.
Trend Chart/Effects on Variations in Conditions
While all the comparisons are made based on identical input and output conditions to provide a true comparative picture,
in real life, different applications will have varying requirements. In such cases, one approach or topology may be more
suitable for a given application than others. Following variations in operating or applications conditions are explored in
Chapter 9.
1.
2.
3.
4.
Cost/complexity as a function of power
Efficiency as a function of power
Filter cost/complexity as a function of power
Single Line condition (instead of universal line operation)
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CHAPTER 3
Critical Conduction Mode (CrM) PFC
PFC Converter Modes
The boost converter is the most popular topology used in PFC applications. It can operate in various modes such as
continuous conduction mode (CCM), discontinuous conduction mode (DCM), and critical conduction mode (CrM). This
chapter provides the analysis of the CrM operation using the NCP1607. As shown in Chapter 1, in this mode the inductor current
reaches zero before the start of the next cycle and the frequency varies with line and load conditions. One benefit of CrM is
that the current loop is intrinsically stable and there is no need for ramp compensation. In addition, the inductor current reaching
zero every cycle causes the diode to turn off without reverse recovery losses and enables the use of a less expensive boost diode
without performance penalties. Similarly, the MOSFET turn-on can be at a low voltage, which reduces switching losses. This
chapter contains the background, design details, and results of a CrM PFC converter. It should be noted that the critical
conduction mode is also known as boundary conduction mode (BCM), borderline conduction mode (BCM), and transition
conduction mode (TCM). It is also mistakenly referred to as discontinuous conduction mode. The true DCM PFC is different
and only recently introduced by ON Semiconductor and is covered in Chapter 4.
Basics of CrM Operation
The overview of the CrM operation is provided in Chapter 1. This chapter presents a detailed view of the basic operation.
The power switch ON state and OFF state are the two fundamental states of operation for a CrM converter. Figure 3−1 shows
the idealized circuit diagrams and waveforms for each state. A few key equations for understanding the operation of the CrM
converter are derived next. The first relationship is derived from the triangular nature of the inductor (coil) current waveform.
Figure 3−1. States of Operation in Critical Mode Conduction PFC
I in (t ) = I coil
T SW
=
I coil , pk
2
Where Iin(t) is the low (line) frequency instantaneous value of the input current, which is equal to the averaged switching
frequency inductor current <Icoil>TSW. Using this equation and the input voltage and current relationship for unity power factor,
it is determined that the on time of the switch is constant for a particular line voltage and load current condition [1]:
ton =
2 Pin L
Vac 2
As described in chapter 1, this relationship is the basis of the voltage-mode control of the CrM PFC converter. Traditional
CrM controllers use current-mode control and industry contains many pin and function compatible, low-cost devices from
various semiconductor suppliers. The key common attributes of these controllers are:
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• Feedback error processing using a transconductance error amplifier in order to accommodate the overvoltage
•
protection (OVP) function that is critical to PFC operation. Some devices use alternative techniques for OVP
detection and managed to retain the use of traditional error amplifier for feedback processing. Pins 1 and 2 are
used for this function.
Multiplier input (Pin 3) that senses the scaled instantaneous input voltage information to create the reference
signal for the current waveform.
Instantaneous current sense input (Pin 4) that is connected to the PWM comparator and protection circuits.
•
• A zero current detection (ZCD) input (Pin 5) that senses when the inductor current reaches zero and causes the
driver to turn on.
• Vcc (Pin 8), Ground (Pin 6) and Driver (Pin 7).
The reasons for success of this function set and pinout are easy to explain, in a very compact package, these features included
all the necessary control functionality for the current mode CrM PFC. In addition, the layout is made easy by separation of noise
sensitive signals (pins 1-4) and higher noise generation pins (pin 6 and 7). By buffering the driver between Vcc and ground,
optimum noise protection is achieved.
NCP1607 – Industry Pin Compatible Voltage Mode CrM Controller
The voltage mode CrM offers certain benefits compared to the current-mode CrM operation, namely:
1. There is no requirement to sense the input voltage through a resistive divider. In addition to reducing component
count, it also reduces the power consumption at light or no load conditions. This is a critical feature for compliance
with low standby power requirements.
2. The current sense signal is used for protection only and is not used by the PWM comparator. This is beneficial near
the input voltage zero crossing where the current sense signal amplitude is low and the possibility of noise injection
negatively affecting the circuit performance is high.
3. The lack of a multiplier eliminates a common source of inaccuracies in the circuit.
These benefits make it advantageous to convert from current-mode to voltage-mode control. Due to the popularity and
designer familiarity with the traditional CrM controllers, it is important that all the desirable attributes of the current-mode
controllers be retained. ON Semiconductor’s NCP1607 and NCP1608 accomplish this objective by retaining the pin out and
functionality of the current mode controllers while offering the benefits of the voltage-mode control.
As shown in Figure 3−2, the only change required to convert from current-mode controllers to the NCP1607 or NCP1608
is to modify the components connected to pin 3. The input voltage sense resistors are removed and the multiplier filter capacitor
is modified to match the design equation for the timing capacitor for voltage-mode control. All other functionality is retained
in the NCP1607 and NCP1608. In many cases, parametric and protection features are enhanced such as the OVP threshold
accuracy or the floating pin protection. The NCP1607 incorporates a traditional error amplifier with dynamic OVP sensing
based on the feedback path current. The dynamic OVP sensing enables the designer to program the overvoltage threshold. The
NCP1608 incorporates a transconductance error amplifier with fixed OVP sensing from the feedback voltage. The advantage
of the transconductance error amplifier is that the overvoltage fault is sensed independently of the error amplifier dynamic
operation. Both the NCP1607 and NCP1608 offer a low peak current sense threshold (typically 0.5 V) for additional power
savings as desired by designers. Additionally, open loop protection (UVP) is provided by both controllers. Finally, the
controllers enable the shutdown of the output driver if the FB pin is pulled to ground. The NCP1607 enables the designer to
shut down the output driver if the ZCD pin is pulled to ground. More details of the NCP1607 and NCP1608 operation are found
in [2] and [3] respectively.
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ON Semiconductor
LBOOST
VOUT
DBOOST
LOAD
(Ballast,
SMPS, etc.)
R1 RZCD
AC Line
EMI
Filter
+
CIN
ROUT1
1
R2
CCOMP
ROUT2
R3
2
3
4
CT
NCP1607
FB
VCC
Control DRV
Ct
CS
GND
ZCD
VCC
8
7
6
+
CBULK
5
RS
Figure 3−2. Converting from Current−Mode Control to the NCP1607
Design Steps with 270 W Example
The systematic design procedure for CrM PFC using the NCP1607 is performed to illustrate the design of a 270 W CrM PFC
converter used as a basis for comparison (design P1 described in Chapter 2).
Step 1: Define the key specifications
Equipment used for measurements
Minimum Input voltage (VacLL): 88 Vac (this is usually 10-12% below the minimum typical voltage, which can be 100 Vac
in many countries).
Maximum Input voltage (VacHL): 264 Vac (this is usually 10% above the maximum typical voltage, which can be 240 Vac in
many countries)
Line frequency (fLINE): 50 Hz /60 Hz (This is usually specified in a range of 47-63 Hz and for calculations such as hold-up
time, the lowest value specified is used for the calculation)
Output Voltage (Vout): 385V (This value is at least greater than 1.414 S VacHL and is typically between 385 and 400 V for
universal input operation)
Maximum output voltage (Vout(max)) : 415 V (This value is usually 7-10% above the Vout value and is determined by the
accuracy of the OVP threshold of the PFC controller – conversely, the selection of the PFC controller must be made based on
this specification if it is determined by other component stress levels and derating factors).
PFC maximum output power (Pout): 270 W (This is the specified output power for the PFC stage. It is important to factor in
the follow-on stage efficiency when specifying this parameter – it will always be higher than the specified maximum system
output power)
Minimum switching frequency (fsw(min)): 40 kHz (This parameter helps set the value of the boost inductor. Choosing it too
low increases the inductor size and choosing it too high leads to very high frequency operation near zero crossing and at light
loads)
Output voltage ripple (Vripple(p-p)): 20 V (This parameter is often specified in percentage of output voltage, +/- 5% is a very
typical specification)
Hold-up time (thold-up): 16 ms (This parameter specifies the amount of time the output remains within a specified limit during
line drop-out. One line cycle is typically specified. For PFC alone, this spec is not applicable, but the PFC output capacitor is
the single largest determinant of the hold-up time)
Estimated efficiency (η): 93% (This parameter is an initial estimate that is used to size the power stage components. A high
level of accuracy is not needed for the design procedure).
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PFC Handbook
Step 2: Design the Boost Inductor (PFC coil)
The (maximum) peak inductor current is calculated using:
I coil , pk (max) =
2 2 Pout 2 2 270
=
= 9.33 A
h Vac LL
0.93 88
The rms current is calculated using:
I coil ,rms (max) =
I coil , pk (max)
6
=
9.33
= 3.81A
2.45
The boost inductor design is calculated using the equation below (generally low line presents the worst case situation, but
the following equation should also be applied for the high line condition) :
V
385
2
h Vac LL ( out − Vac LL ) 0.93 88 2 (
− 88)
2
2
=
= 225 mH
Lw
2 Vout Pout f sw(min)
2 400 270 40000
For this design a 250 μH inductor is chosen, which results in a minimum frequency of 36 kHz. This design uses a Ferrite
core (PQ3230, PC40 material) in order to reduce the core losses and litz wire (0.1f S 80) to reduce the skin effect (where current
flows only in the periphery of the conductor) and hence the conduction losses.
When designing the inductor, the flux density has to be taken into account. The first point is to keep the flux density low
to prevent core from saturation. The ferrite cores have low saturation flux density, 0.25 to 0.5 T, which is temperature
dependent. The second point is to get a trade-off point between the usage of flux density and the hysteresis power loss. Based
on the lab experience, targeting the peak flux density around 0.25 T seems reasonable:
B=
I coil , pl (max) L
Ae N
10 4 =
9.33 A 250 m
10 4 v0.25T
2
1.67cm N
N w55.86
For this design, a 56-turn winding is implemented using Litz wire as indicated above. The inductor includes a secondary
winding to sense the zero current detection (ZCD). The turns ratio for this winding is calculated in Step 4.
Step 3: Select the timing capacitor
The voltage mode CrM circuit (NCP1607) operates by setting a fixed on time over the line cycle. This on time changes with
the line and load conditions. The on time will be at its maximum value at low line and full load (as that is the time maximum
energy has to be transferred to the output). The timing capacitor connected to pin 3 adjusts the maximum on time. The equation
for the maximum on time is:
TON (max) =
2 L Pout
h Vac LL
2
=
2 225 10 −6 270
= 16.87 ms
0.93 88 2
Next, the value of Ct is calculated using following equation:
Ct >
I ch arg e TON (max)
VEA( diff )
=
297 10 −6 16.87 10 −6
= 1.727 nF
2 .9
Where Icharge and VEA(diff) are the datasheet specified values for the Ct charge current and the error amplifier peak and valley
voltages respectively. To ensure that the desired maximum on time can be delivered, the specified maximum and minimum
values are used for the numerator and denominator respectively. Based on above calculations, a standard value of 1.8 nF is used
in the present design.
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ON Semiconductor
Step 4: ZCD network design
During the switch off condition, a voltage of (Vout-Vin) is across the boost inductor. As the current reaches zero, this voltage
starts collapsing and ZCD winding senses this event. In order to ensure fail-safe operation, the turns ratio of ZCD winding is
selected such that the reflected (Vout-Vin) voltage is above the ZCD threshold under all conditions as shown in the equation:
(V − 2 Vac HL ) (385 − 1.414 264)
NB
< out
=
= 5.06
2 .3
N ZCD
VZCDH
The resistor RZCD (R125) is selected to limit the current into the ZCD pin. If it is selected too low, the current may trigger
the ZCD shutdown feature. So, the value of RZCD is given by:
RZCD w
2 Vac HL N ZCD 1.414 264
=
0.2 = 29.5 kW
I CL _ NEG
NB
2.5 10 −3
In the design, a 56 kΩ resistor is used. The value of RZCD is finalized empirically to get the lowest switching losses. The
resistor and parasitic (or real) capacitance on the ZCD pin create a short delay circuit that allows the drain voltage of the FET
to fall after ZCD condition is reached. If the switch turn-on condition matches the valley of the drain voltage, the switching
losses are minimized.
Step 5: Set FB, OVP, and UVP Voltages
The design of the feedback components RFB1 and RFB2 determine the nominal output voltage, the undervoltage protection
threshold, and the overvoltage protection threshold. RFB1 sets the overvoltage protection by the following equation:
Vout (max) − Vout
RFB1 =
I OVP
=
415 − 385
= 3 MW
1 10 − 5
RFB1 is selected as three 1 MΩ resistors (R130, R131, and R132) due to the high output voltage.
The NCP1607 includes Floating Pin Protection (FPP) that protects the system if the FB pin is floating. FPP is implemented
with a pull−down resistor (RFB shown the NCP1607 datasheet). The inclusion of RFB is compensated by adjusting RFB2. The
parallel combination of RFB and RFB2 form an equivalent resistor REQ that is calculated using:
REQ =
2.5 RFB1 2.5 3 10 6
=
= 19.61 kW
Vout − 2.5 385 − 2.5
RFB 2 =
REQ RFB
RFB − REQ
=
19.61 4700
= 19.7 kW
4700 − 19.61
RFB2 is selected as a standard value of 19.6 kΩ (R134).
Finally, the UVP threshold is given by:
Vout (UV ) =
RFB1 + REQ
REQ
VUVP =
3000 + 19.61
0.3 = 46 V
19.61
The NCP1607 detects an undervoltage fault when the output voltage is less than Vout(UV) and disables the output driver. This
feature also protects against open feedback path conditions (missing RFB1).
Step 6: Power Stage Components (MOSFET, Diode, and Current Sense resistor)
The power stage components are designed based on their current and voltage ratings. The inductor design is covered in step
2. The MOSFET is selected based on peak voltage stress (Vout(max)+margin) and rms current stress:
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PFC Handbook
I M ( rms ) =
2 Pout
3 h Vac LL
1−
8 2 Vac LL
2 270
8 2 88
=
1−
= 3.24 A
3 p Vout
3 p 385
3 0.93 88
Using a 600 V, 0.19 Ω MOSFET and assuming that Rds(on) increases by 80% due to temperature effects, the conduction losses
are:
2
Pcond = I M ( rms ) Rds ( on ) = 3.24 2 0.19 1.8 = 3.6 W
While switching losses are difficult to predict without the details of switching waveforms and diode recovery characteristics,
it is assumed that for a CrM PFC, the switching losses are less than the conduction losses. One component of the switching
losses that is easily predicted is the capacitive turn-on losses:
Psw,cap =
2
Coss 25 25 Von1.5 f = 0.666 780 10 −12 5 (385)1.5 36000 = 0.71 W
3
The nonlinear nature of the capacitance is accounted for in the equation above. As mentioned in step 4, the ZCD network
design in reducing the Von from the nominal voltage of 385 V to a much lower value and reduce the losses even more. For
CrM operation, the conduction losses dominate. The lowest economically feasible Rds(on) MOSFET is chosen. For this design,
the 20 A, 600 V FET with 0.19 ohm Rds(on) was chosen.
The boost diode is a simple selection for CrM since there are no reverse recovery issues. The goal is to choose the correct
voltage rating (Vout(max)+margin) and lowest forward voltage drop available. The average diode current is the same as the
output current (Id = Pout/Vout = 270/385 = 0.7 A). The losses are simply Id*Vf . The peak current of the diode is equal to the
inductor peak current (9.33 A). The chosen diode is the MUR860, which is an ultrafast rectifier with 8 A average and 16 A
repetitive peak current rating and 600 V reverse voltage rating.
The current sense resistor is calculated using the peak current and the current sense voltage threshold.
Rsense =
0.5
VILIM
=
= 0.0536 W
I coil , pk 9.33
In this design, Rsense (R100) is selected as 0.04 Ω, which provides 33% margin over the maximum peak current. The
dissipation in the sense resistor is calculated using:
2
2
PRsesnse = I M
( rms ) Rsense = 3.24 0.04 = 0.42 W
Step 7: Output Capacitor Design
The output capacitor is designed considering three factors: output voltage ripple, output current ripple, and the hold-up time.
The output voltage ripple is calculated using:
Vripple ( p − p ) =
Pout
2 p f line C out Vout
The capacitor rms current is calculated using:
I Cout ( rms )
2
32 2 Pout
P
=
− ( out ) 2
2
Vout
9 p Vac LL Vout h
However, it is common to size the value of the capacitor based on the hold-up time that is calculated using:
t hold −up =
2
2
C out (Vout
− Vmin
)
2 Pout
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ON Semiconductor
In this case, a 220 μF capacitor is selected to satisfy the conditions above. The peak-peak ripple is 10 V, the rms current is
1.87 A, and the hold-up time is 18.6 ms (for Vmin = 320 V).
These major design steps enable the functional prototype to be built and tested. In addition to these steps, other common steps
such as Vcc generation, feedback compensation, and inrush limiting are required to complete the design. These steps are not
covered here as they follow conventional methods.
Circuit Schematic and Bill of Materials
3
BD001
GBU806
D101
1N5406
L101
1
C101
0.47 mF
C103
1 nF
C102
1 mF
R120
301 k
2
FG
C002
1 nF
C001
1 nF
R121
301 k
C104
1 nF
C006
0.33 mF
R122
143 k
FG
PFC
VCC
R125
56 k
R105
22
R003
330 k
R002
330 k
R001
330 k
NCP1607
DRV
ZCD
VCC
ZD101
1N4745A
C122
1 mF
C121
100 mF
R126
39
Q100
SPP20N60C3
R127
39
R130
1 MEG
C124
0.01 mF
R131
1 MEG
Ctrl
CT
FB
GND
CS
C120
2.2 nF
R133
10 k
R128
10.2 k
R129
10
C126
220 pF
R2
TB1
Figure 3−3. Schematic of NCP1607 Based PFC Converter
37
+ C110
220 mF
450 V
ZD100
1N4745A
R134
19.6 k
F001
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C 111
102PF/5T
R132
1 MEG
C125
0.1 mF
L001
LC-TR-25_EE
L1
PFC
OUT
IC100
L002
LC-TR-25_EE
C005
0.47 mF
D100
MUR860
L100
PQ3230
4
R100
0.04
PFC
GND
PFC
VCC
PFC Handbook
Table 3−1. Bill of Materials for CrM Board
Reference
Quanity
Part
BD001
1
GBU806
C001, C002, C103, C104
4
1nF
C005, C101
2
0.47mF
C006
1
0.33mF
C102, C122
2
1mF
C110
1
220mF
C111
1
102PF/5T
C120
1
2.2nF
C121
1
100mF
C124
1
0.01mF
C125
1
0.1mF
C126
1
220pF
D100
1
MUR860
D101
1
1N5406
F001
1
FUSE 5*20 (Axial Lead)
IC100
1
NCP1607
L001, L002
2
CHOKE (EMI Choke)
L100
1
PQ3230
L101
1
CHOKE (Power Choke)
Q100
1
SPP20N60C3
R001, R002, R003
3
330k
R100
1
0.04
R105
1
22
R120, R121
2
30k
R122
1
143k
R125
1
56k
R126, R127
2
39
R128
1
10.2k
R129
1
10
R130, R131, R132
3
1MEG
R133
1
10k
R134
1
19.6k
TB1
1
AC inlet
ZD100
1
1N4733A
ZD101
1
1N4745A
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ON Semiconductor
Results and Analysis
The results for the 270 W design based on the NCP1607 are shown in Table 3−2 below.
Table 3−2. Performance Results for the 270 W PFC Based on NCP1607
Vin
(Vac)
Pin
(W)
Vo
(V)
Io
(A)
Output
Power
(W)
Po
Efficiency
PF
THD
100
28.91
386.97
0.071
27.43
10%
94.88%
0.954
18.2
42.24
387.04
0.104
40.40
15%
95.63%
0.971
14.4
56.93
387.10
0.141
54.59
20%
95.89%
0.970
11.7
141.59
387.24
0.350
135.60
50%
95.77%
0.996
5.9
287.10
387.30
0.704
272.53
100%
94.93%
0.998
3.7
28.57
386.88
0.070
27.06
10%
94.72%
0.927
23.1
41.96
386.97
0.104
40.20
15%
95.80%
0.960
15.8
56.62
387.02
0.141
54.49
20%
96.24%
0.972
14.4
140.57
387.16
0.350
135.56
50%
96.44%
0.993
8.0
283.52
387.28
0.700
271.19
100%
95.65%
0.998
4.0
28.63
387.21
0.071
27.57
10%
96.29%
0.533
100.0
42.18
387.04
0.105
40.82
15%
96.78%
0.642
100.0
55.64
387.07
0.140
54.10
20%
97.23%
0.762
74.1
137.85
386.89
0.349
135.00
50%
97.93%
0.892
30.5
275.48
387.06
0.699
270.63
100%
98.24%
0.970
12.3
115
230
As shown in Table 3−2 the efficiency of the circuit is nearly 95% at 100 Vac and full load. The efficiency is comfortably higher
than 95% for all other line conditions between 15% and 100% load. The wide load range where the efficiency is high is
attributable to low switching losses for the CrM operation. The THD at high line and full load is measured to be 12.3% with
a PF of 0.97. This is within the allowable range of the IEC61000-3-2 harmonics requirements as depicted in Figure 3−4. The
PF and THD performance at light load and high line is not that great for this board. The reasons for this are the distortions
applied by the propagation delays.
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PFC Handbook
3.50
Harmonic Current Percentage (mA/W)
3.00
2.50
2.00
1.50
1.00
0.50
0.00
3
5
7
9
11
13
15
17
19
21
23
25
27
29
31
33
35
37
39
Harmonic #
Class D limitation
270 W result
75 Wresult
Figure 3−4. Harmonic Content of NCP1607 Based PFC Converter at Full Load and 75 W
The no load power consumption of the circuit is given in Table 3−3. As shown, the no load losses are higher at low line. The
no load losses are greater at low line because of the large difference between the input and output voltages. The large difference
between the input and output voltages requires increased switching to boost the input voltage up to the output voltage..
Table 3−3. No Load Power Consumption
Vin
Pin
(W)
100 Vac
1.084
115 Vac
0.974
230 Vac
0.526
The efficiency optimization involves component tuning. For example, the diode and the MOSFET selections are made using
different combinations and the results are compared. For this design, two types of diodes are evaluated. The MUR550, which
is a diode designed for CrM and FCCrM applications and MUR860, which is a standard 600 V ultrafast rectifier. In addition,
the MOSFET was changed from 20 A to 15 A and its effect was observed. The results are shown in Table 3−4.
Table 3−4. Efficiency Change as a Result of Major Component Changes
100 Vac
115 Vac
100% load
15 A FET
20 A FET
100% load
15 A FET
20 A FET
5 A diode (MUR550)
93.5%
94.4%
5 A diode (MUR550)
95.1%
95.6%
94.9%
8 A diode (MUR860)
8 A diode (MUR860)
95.7%
20% load
15 A FET
20 A FET
20% load
15 A FET
20 A FET
5 A diode (MUR550)
95.0%
95.7%
5 A diode (MUR550)
95.7%
96.1%
95.9%
8 A diode (MUR860)
8 A diode (MUR860)
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96.2%
ON Semiconductor
From Table 3−4, it is clear that the combination of the 8A diode and 20 A FET provides the highest efficiency results.
However, at 115 Vac, the difference between performance results is not large. The impact of the FET on efficiency performance
is easy to explain. The conduction losses scale with Rds(on) and increase significantly as the FET Rds(on) increases. By contrast,
the efficiency difference between the MUR550 and the MUR860 is not easy to explain.
The relevant operating waveforms of the circuit at full load and 115 Vac input are shown in Figure 3−5. Here, Ch1 is the input
voltage (the captured waveform is taken at the peak of the input votlage), Ch2 is the inductor current, which shows the CrM
nature of the circuit, Ch3 is the Vds of the main FET, and Ch4 is the Vgs.
Figure 3−5. Operating Waveforms at Full Load, 115 Vac
The switching frequency of the CrM varies over a wide range with line, load, and the phase angle. Figure 3−6 shows these
variations at the peak of the line voltage for low line and high line operations. Also included in the figure are the measured
values at some corners. As shown there, the theoretical low line frequency range is 36 kHz to 360 kHz from full load to 10%
load. The measured full load frequency matches the theoretical value closely. However, the measured frequency at light load
and low line is much lower than the theoretical value. This difference can be ascribed to the internal propagation delays and
forward drops which make the actual values different from idealized values used in calculations. The actual frequency being
much lower than calculated value is good news from the viewpoint of EMI and light load efficiency. However, the propagation
delays also lead to higher distortion in the input current waveform which reduces the PF and increases THD.
Similar results are plotted for the zero crossing instances in Figure 3−7. The difference in the calculated and measured values
is apparent and is attributed to the same factors described above. One interesting difference between the two plots is that for
zero crossing, the low line frequencies are lower compared to the high line values, while the sinusoid peak shows exactly
opposite behavior. At high line, the peak is higher, so the frequency range from zero crossing to sinusoid peak will be wider.
Thus, even starting from a smaller number at the peak, it goes to much higher peak during one-half line cycle.
In general, the zero crossing distortion is increased for CrM operation at high line. This is because the theoretical frequency
is never reached due to inherent circuit propagation delays. The result is that the PF decreases and THD is increases.
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PFC Handbook
Figure 3−6. Frequency Variation at the Peak of the Sinusoid in CrM PFC
Figure 3−7. Frequency Variation at Zero Crossing in CrM PFC
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ON Semiconductor
Following figures provide the graphical representation of the data shown in Table 3−2.
Figure 3−8. Efficiency Performance Over Line and Load
Figure 3−9. Power Factor Performance Over Line and Load
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PFC Handbook
References
[1] “Power Factor Correction Stages Operating in Critical Conduction Mode”, AND8123/D, ON Semiconductor.
http://www.onsemi.com/pub_link/Collateral/AND8123−D.PDF
[2] “Implementing Cost Effective and Robust Power Factor Correction with the NCP1607”, AND8353/D,
ON Semiconductor. http://www.onsemi.com/pub/Collateral/AND8353−D.PDF
[3] “Implementing Power Factor Correction with the NCP1608”, AND8396/D, ON Semiconductor.
http://www.onsemi.com/pub/Collateral/AND8396−D.PDF
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ON Semiconductor
CHAPTER 4
Frequency Clamped Critical Conduction Mode (FCCrM) PFC
Introduction
The Critical Conduction Mode (CrM) described in Chapter 3 offers many advantages, especially for low power applications.
However, the results and the characteristics also showed some limitations of the approach at light load conditions. The
prominent limitation was that the frequency range of the converter varies significantly over line and load conditions. In the
design example used in Chapter 3, at high line and full load, the frequency will go from 14.5 kHz at the peak of the sinusoid
to 480 kHz near the zero crossing. At lighter loads, this frequency range gets even wider. Theoretically, the frequency has to
go to a few MHz to achieve the PFC under light load and high line conditions. However, in practice, the inherent delays within
the controllers and in surrounding circuits limit the switching frequency (also, some CrM controllers such as the MC33260
offer frequency clamp mechanisms to limit the frequency excursion). Limiting of the switching frequency helps limit the EMI
generation and also cuts down the magnetic core losses and filter requirements. However, they come at a price of distortion
in the power factor, especially near the zero crossings, where the frequency tends to get to its highest level. Thus, a designer
always faces a trade-off between keeping the maximum frequency too low and letting it go high.
This difficulty has led to development of a new control approach called the Frequency Clamped Critical Conduction Mode
(FCCrM).
Basics of FCCrM Operation
The overview of the FCCrM operation was already provided in Chapter 1. Here, let us take a more detailed view of the basic
operation. Since the key differentiation between the CrM and FCCrM is when the converter operates in fixed frequency mode
(DCM), we will examine that in detail. The operation in the CrM follows what was described in Chapter 3. There are three
fundamental states of operation in DCM – Boost Switch ON state, OFF state with current conduction and OFF state with no
current conduction. Figure 4−1 provides the idealized circuit waveforms for these states. A few key equations for the
understanding the operation of the DCM mode can be derived next. The first relationship was derived from the nature of the
inductor (coil) current waveform in Chapter 1:
ǒ
Ǔ
t
V (t)
Iin(t) + in · ton · cycle
TSW
2·L
tcycle = ton + tdemag
ICOIL
ton
tdemag
tDT
Time
tcycle
TSW
Figure 4−1. States of Operation in Discontinuous Conduction Mode PFC
This relationship between the input current and input voltage points to a PFC control algorithm that can be implemented to
make the quantity (ton S tcycle/TSW) constant over a line cycle (for a given set of line and load conditions). As described in [1],
this is achieved by including a switched amplifier block as shown in Figure 4−2 that integrates two different values to equal
the error amplifier input voltage. This results in:
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PFC Handbook
Vton =
Tsw Vcontrol
t cycle
Where Vcontrol is the voltage at the error amplifier output which represents the line and load conditions and Vton is the voltage
presented to the timing comparator that sets the on time. Since the timing comparator controls the on time such that it is
proportional to Vton, we get:
t on TVton and
t on t cycle
Tsw
TVcontrol which is constant over a line cycle.
This yields the desired power factor algorithm. This algorithm is patented and incorporated in three of ON Semiconductor’s
offerings – NCP1601, NCP1605, and NCP1631. NCP1601 is an 8-pin simple PFC controller, while the NCP1605 is an
advanced feature set controller. NCP1631 is an interleaved FCCrM PFC controller, which will be depicted in Chapter 6. In this
design, the advance feature set of NCP1605 was used and it is described in the next section.
Vton
Vcontrol
R1
C1
IN1
−> Vton during cycle
−> 0 V during dead−time
−> (Vton S dcycle) in average
Isense
Coil Current
Monitoring for
Dead−Time
Detection
PWM
Comparator
To PWM
Latch
S1
Timing
Capacitor
Saw−Tooth
S2
VDT
High During Dead−Time
Figure 4−2. DCM PFC Algorithm to Modulate on Time
NCP1605 – Advanced Feature Set FCCrM Controller
Figure 4−3 shows the block diagram of the NPC1605 PFC controller. It has many features to ease and optimize the
implementation of PFC in the system.
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ON Semiconductor
Vout Low Detect
95.5% Vref
200 mA
VoutL
+
-
V/I
Iout
Vcontrol
VDD
TSD
UVP
BO_NOK
STDWN
3V
FLAG1
VREGUL
R
VSTBY
300 mV
SKIP
+
Ich=K.Iout.Iout
Vref
Vton
processing
circuitry
VDD
DT
R R
R
Q
R PWM
S Latch
OFF
outON
+
CLK
+
Iref
Regul
outON
Ct_OK
+
100 mV
Output
Buffer
Oscillator /
Synchronization
Block
Drv
OSC /
SYNC
ZCD
Q
R
Dead−time
Detection
Latch
S
OCP
Ics > 250 mA
VCC
VCC
outON
DT
Ics
CSout
50 mV
BO_NOK
Ics
CSin
OFF
SKIP
OCP
1 V / 0.5 V
NC
Vt(on)
PWM
Comparator
OVP
Ct
BO /
Stdwn
FAULT
Management
OVP
VREGUL
pfcOK
VoutL
HVCS_ON
2.R
LSTBY
S
Q
R
+
UVLO (Vcc<VccOFF)
FLAG1
SKIP
+
HV
UVLOs
Latch
Reset
Internal
Thermal
Shutdown
OFF
STBY
HVCS_ON
UVLO
pfcOK
Error Amplifier
VSTBY
+
±20 mA
Vref
FB
All the RS latches are
RESET dominant
ZCD
+
-
REF5V
pfcOK
OVP
12% Vref
Vovp = Vref
+
UVP
S
FLAG1
Lstup Q
R
OFF
+
-
OVP
Vref
+
S
R
Q
stdwn
Vcc<VccRST
STDWN
Figure 4−3. Block Diagram
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pfcOK /
REF5V
GND
PFC Handbook
The block diagram helps identify key functional improvements [1] of the NCP1605 which are described below.
• The high voltage input (pin 16) which allows Vcc to be derived directly from the PFC bulk output by charging
the Vcc capacitor through an internal current source. [Not used in this design]
• A separate OVP/UVP pin (pin 14) that allows independent protection feature desired in some applications.
• A latched shutdown input (pin 13) that can be used to protect against catastrophic faults. [Not used in this
•
•
•
•
design]
A PFCOK/reference output (pin 12) that can be used for sequencing the second stage to start after PFC is
operational. [Not used in this design]
An Oscillator/Sync input that helps set the maximum switching frequency. When the coil current has not
reached zero, the switching period is automatically extended to enable CrM.
The current sense resistor is put in the return path to sense the true inductor current. Hence the ZCD winding is
not required.
The light load operation is improved by including a cycle-skip function driven by input on pin 1. [Not used in
this design]
Window comparators to speed up the transient response when output voltage goes outside specified limits.
•
• Brown-out detection (pin 2) to turn-off the circuit when input voltage is too low.
Design Steps with 270 W Example
The step-by-step design procedure for FCCrM PFC using the NCP1605 is applied next to illustrate the design of a 270 W
FCCrM PFC used as a basis for comparison (design P2 described in Chapter 2).
Step 1: Define the key specifications
Minimum Input voltage (VacLL): 88 Vac (this is usually 10-12% below the minimum typical voltage which could be 100 Vac
in many countries).
Maximum Input voltage (VacHL): 264 Vac (this is usually 10% above the maximum typical voltage which could be 240 Vac
in many countries)
Line frequency (fLINE): 50 Hz /60 Hz (This is often specified in a range of 47-63 Hz and for calculations such as hold-up time,
one has to factor in the lowest value specified)
Output Voltage (Vout): 385 V (This value has to be above 1.414 S VacHL and is typically between 385 and 400 V for universal
input operation)
Maximum output voltage (Vout(max)) : 415 V (This value is usually 7-10% above the Vout value and is determined by the
accuracy of the OVP level of the PFC controller – conversely, the selection of the PFC controller has to be made based on this
specification if it is determined by other component stress levels and derating factors).
PFC maximum output power (Pout): 270 W (This is the specified output power for the PFC stage. It is important to factor in
the follow-on stage efficiency when specifying this parameter – it will always be higher than the specified maximum system
output power)
Minimum switching frequency (fsw(min)): 40 kHz (This parameter helps set the value of the boost inductor. Choosing it too
low increases the inductor size and choosing it too high leads to very high frequency operation near zero crossing and at light
loads)
Maximum switching frequency (fsw(max)): 65 kHz (This parameter helps set the value of the oscillator capacitor and whenever
CrM frequency tries to go above this value, the converter enters the DCM mode.
Output voltage ripple (Vripple(p-p)): 20 V (This parameter is often specified in percentage of output voltage, +/- 5% is a very
typical specification)
Hold-up time (thold-up): 16 ms (This parameter specifies the amount of time the output will remain valid during line drop-out.
One line cycle is typically specified. For PFC alone, this spec is not applicable, but the PFC output capacitor is the single largest
determinant of the hold-up time)
Estimated efficiency (η): 93% (This parameter is an initial estimate that is used to size the power stage components – high level
of accuracy is not needed for the design procedure).
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ON Semiconductor
Step 2: Design the Boost Inductor (PFC coil)
The (maximum) peak inductor current is the same as the CrM, since the converter operates in CrM for the most stressful
condition (low line, full load). This current is given by:
I coil , pk (max) =
2 2 Pout 2 2 270
=
= 9.33 A
0.93 88
h Vac LL
The rms current is given by:
I coil ,rms (max) =
I coil , pk (max)
6
=
9.33
= 3.81A
2.45
The boost inductor design (which is also similar to the CrM design) is given by the equation below (generally low line
presents the worst case situation, but the following equation should also be applied for the high line condition) :
V
385
2
h Vac LL ( out − Vac LL ) 0.93 88 2 (
− 88)
2
2
= 225 mH
Lw
=
2 Vout Pout f sw(min)
2 385 270 40000
In the design under consideration, a 250-mH inductor was chosen, resulting in a minimum frequency of 36 kHz. In order to
facilitate direct comparison with CrM, an identical inductor was chosen and used for both CrM (Chapter 3) and FCCrM
(Chapter 4).
In the absence of a need to make a direct comparison such as in this handbook, it would have been advisable to push the
minimum frequency higher for a FCCrM application. The benefit of this would be smaller inductor requirement as indicated
in the equation above. In a normal CrM operation, this would have created a very high maximum frequency, but because of
frequency clamping, this situation is averted in the FCCrM operation. Other point to note is that with CrM and FCCrM, the
turn-on switching losses are averted, so the switching loss penalty of increasing the switching frequency is not that severe.
However, the inductor still has core losses which are function of switching frequency and these should be kept in mind while
increasing the switching frequency.
Step 3: Select the oscillator capacitor
The oscillator capacitor sets the maximum frequency. In this case, the maximum frequency is chosen to be 65 kHz. Choosing
a maximum frequency very close to the minimum frequency calculated in the above step lets the converter operate in CrM for
only a brief operating range. All other times, the converter will operate in DCM. In this case, this choice was made intentionally
for two reasons. First, it demonstrates the functionality of the FCCrM in the widest range of operating condition and second,
it allows the FCCrM circuit to operate as close in frequency to the CCM solution to make the comparisons between the two
modes more relevant.
The value of Cosc is derived using following equation:
Cosc =
I charge
2 DV f sw(max)
− 20 10 −12 =
100 10 −6
− 20 10 −12 = 750 pF
2 1 65000
Where Icharge and DV are the datasheet specified values for the Cosc charge current and the oscillator voltage swing
respectively. Based on above calculations, 2 capacitors (270 pF and 470 pF) in parallel are used in the present design to have
the required capacitance.
Step 4: Current sense and ZCD design
The current sense signal across the sense resistor is actually a negative voltage. However, insertion of a resistor between
Rsense and pin 5 of the NCP1605 allows that signal to be converted into a current. The value of Rsense is chosen to minimize
its impact on efficiency at low line, full load (here, 0.1 W is chosen). The equation for Rocp is given by:
Rocp =
I coil , pk Rsense
I pin 5(max)
=
9.33 0.1
= 3 .6 k W
250 10 −6
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PFC Handbook
The ZCD design in the NCP1605 is eased because the inductor current is sensed through pin 5. The current into pin 5 is
reflected and sourced out of pin 6 through termination resistor RZCD. The value of RZCD should be about 3 times the value of
Rocp. A natural hysteresis is provided by the offset resistor that is tied to the DRV pin (RDRV). This resistor should be about
3 times the value of RZCD. The final component values are:
RZCD = 12 kW, RDRV = 33 kW
If a further optimized efficiency is preferred, a ZCD circuit that is similar as the one in NCP1607 application schematic would
be useful. This circuit is to add the ZCD winding on PFC choke (NZCD), and insert one diode (e.g. 1N4148) and one resistor
(RZCD1) between the ZCD winding and pin 6. During the switch off condition, a voltage of (Vout-Vin) is across the boost
inductor. As the current reaches zero, this voltage starts collapsing and ZCD winding senses this event. The turns ratio of ZCD
winding is selected such that the reflected (Vout-Vin) voltage is above the ZCD threshold under all conditions as shown in the
equation:
(Vout − 2 Vac HL ) N ZCD
RZCD
−V f > VZCDH
NB
RZCD + RZCD1
Assume RZCD1 has 2 times higher resistance of RZCD.
NB
<
N ZCD
(Vout − 2 Vac HL )
(385 − 1.414 264)
=
= 19.5
RZCD + RZCD1
0 .1 3 + 0 .3
VZCDH +Vf
RZCD
The NZCD would be then above 56 turns / 19.5 = 2.87 turns. Choose 3 turns as NZCD.
The resistance of RZCD1 and parasitic capacitance on the pin 6 create a short delay circuit which allows the drain voltage
of the FET to fall after ZCD condition is reached. If the switch turn-on condition matches the valley of the drain voltage, the
switching losses are minimized. The final component value of RZCD1 is:
RZCD1 = 24 kW
Step 5: Set FB, OVP and UVP levels
The NCP1605 provides 2 separate pins for the feedback and OVP/UVP functions. Using a single feedback chain consisting
of ROUT1, ROUT2 and ROUT3, these functions can be accomplished. The equations are given by:
Vout =
ROUT 1 + ROUT 2 + ROUT 3
+ ROUT 2 + ROUT 3
R
Vref and Vout (max) = OUT 1
Vref
ROUT 2 + ROUT 3
ROUT 2
Since there are only two equations and three unknown quantities, we have the freedom to choose the impedances to achieve
the right trade-off between noise susceptibility when very high impedances are chosen and high standby current when low
impedances are chosen. In this case, ROUT2 is chosen as 24.3 kW and from there, the ROUT1 and ROUT3 are calculated to be
4.0 MW and 1.82 kW respectively.
Finally, the UVP level is given by:
Vout (UV ) = 0.12 Vout (max) = 0.12 415 = 49.8 V
This is the output voltage below which the NCP1605 will enter under-voltage protection and shut down output pulses. This
feature also protects against an open feedback path.
Step 6: Select the on-time capacitor (Ct) and offset circuit
The Ct capacitor on pin 7 can offer multiple functionalities in NCP1605. By dimensioning it correctly, the follower boost
operation can be programmed to take effect at a given power level. However, in this particular case, the follower boost function
is not desired, so the Ct capacitor is sized to provide full power at low line. The equation for required capacitance for no follower
boost is:
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50
ON Semiconductor
Ct =
120 m Vref2 L Pout
2
h Vac LL
120 10 −6 2.5 2 250 10 −6 270
=
= 7.03 nF
0.93 88 2
Where Vref is the NCP1605 internal reference (2.5 V) and 120 m represents the internal constant of NCP1605 when the Ct
charge current is made proportional to the feedback voltage squared.
A known issue with any fixed on-time circuit is that the on-time at high line (especially at light loads) is extremely small
- in the 200-500 ns range. Such small on-times are susceptible to propagation delay changes and circuit variations. Additionally,
it presents a unique problem to the NCP1605, since there is an intervening circuit that is used to create the Vton that corresponds
to the on-time. The accuracy of this circuit suffers when on-time is small. A trick is to add an intentional offset on the on-time
to change its functional range. This offset is externally introduced by inserting a resistor Roffset between Ct pin and ground
and tying a resistor Rdrv2 from the gate drive output to the junction of Ct and R4. To achieve accurate Vout, the resistive divider
of Rdrv2 and Roffset should form an offset of 400-500 mV when the switch is on. On the other hand, since NCP1605
implements the skip mode feature, a lower offset could be trimmed depending on the required accuracy of Vout. In the present
design, with choice of Rdrv2=4.7 kW and Roffset=56 W, the offset is 0.18 V for 15 V Vcc.
Once the offset is applied to the Ct pin, its effective range becomes narrow. Without offset, it is full 1 V, but the offset reduces
it by 0.18 V in this case. As a result, in order to achieve the same on-time, the Ct value has to be revised up by same ratio. Hence,
the new Ct value is given by:
Ct (offset ) =
Ct (nom)
7.03
=
= 8.57 nF
1 − Voffset 1 − 0.18
For ensuring no follower boost operation, a higher value of Ct (10 nF) is used in this application.
Step 7: Power Stage components (MOSFET, Diode and sense resistor)
The power stage components are designed based on their current and voltage ratings. The inductor design is already covered
in step 2. The MOSFET is selected based on peak voltage stress (Vout(max)+margin) and rms current stress:
I M ( rms ) =
2 Pout
3 h Vac LL
1−
8 2 Vac LL
2 270
8 2 88
=
1−
= 3.24 A
3 p Vout
3 p 385
3 0.93 88
Using a 600 V, 0.19 ohm FET, will give conduction losses of (assuming that Rds(on) increases by 80% due to temperature
effects):
2
Pcond = I M ( rms ) Rds ( on ) = 3.24 2 0.19 1.8 = 3.6 W
While switching losses are harder to predict without getting into the details of switching waveforms and diode recovery
characteristics, etc., it can be assumed that for a FCCrM PFC, the switching losses will be lower than the conduction losses.
One component of the switching losses that can be predicted is the capacitive turn-on losses.
Psw,cap =
2
Coss 25 25 Von1.5 f = 0.666 780 10 −12 5 (385)1.5 36000 = 0.71 W
3
Here, the nonlinear nature of the capacitance is taken into account. However, as mentioned in step 4, the ZCD network design
can help in reducing the Von from the nominal voltage of 385 V to a much lower value and reduce the losses even more. For
the CrM, the conduction losses dominate, hence, lowest economically feasible Rds(on) MOSFET is chosen. In this design, the
20 A, 600 V FET with 0.19 ohm Rds(on) was chosen.
The boost diode is a simple selection for FCCrM since there are no reverse recovery issues to worry about. The goal is to
choose the correct voltage rating (Vout(max)+margin) and lowest forward drop available. The average diode current is the same
as the output current (Id = Pout/Vout = 270/385 = 0.7 A). So, the losses are just Id S Vf. The peak current seen by the diode will
be the same as the inductor peak current. The diode chosen here is the MUR550 which offers lower forward drop by trading
off on the reverse recovery characteristics.
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PFC Handbook
Step 8: Output capacitor design
The output capacitor is designed considering 3 factors: output voltage ripple, output current ripple and the hold-up time.
The output voltage ripple is given by:
Vripple ( p − p ) =
Pout 2 p f line C out Vout
The capacitor rms current is given by:
I Cout ( rms ) =
2
P
32 2 Pout
− ( out ) 2
2
Vout
9 p Vac LL Vout h
However, almost always, the size and the value of the capacitor is determined by the hold-up time which is given by:
t hold −up =
2
2
− Vmin
C out (Vout
)
2 Pout
In this case, a 220-mF capacitor was chosen that satisfies the conditions above. The peak-peak ripple is 10 V, the rms current
is 1.87 A and the hold-up time is 18.6 ms (for a 320 V Vmin).
These major design steps allow the functional prototype to be built and tested. In addition to these steps, other common steps
such as Vcc generation, brown-out detection, feedback compensation and inrush limiting are also needed to complete the
design. These steps are not covered here as they follow conventional methods.
Circuit Schematic and Bill of Materials
GBU8J 8A 600V
DB1
L2
150uH
2M
2M
0.47 uF
C2
2
3
Rout1a
D1
C1
0.47 uF
2 x 6.8 mH
4
N
Rout1b
L3
C3
TB2
1
MUR550
L1
250uH
TB1
390 V
2
SPP20N60C3
Q1
C4
1uF
2
Rzcd1
24k
1
L
G
5 A f use
F1
1
3
AC inlet
Dzcd
R4
10 k
1n4148
0.1
Rbo1a
3.6 M
R2
R1
Vcc
Rocp
Rbo1b
3.6 M
12k
91k
3.6 k
1
2
3
4
5
6
7
Ct
8
STBY
HV
IC2
BO
Vctrl
NC
OVP/UVP
FB
STDWN
pf cOK
CSin
CSout/ZCD
Vcc
Ct
DRV
OSC/SY NC
GND
10n
16
Rout2
1.82k
D2
R5
10
15
1N4148
14
13
12
Vcc
11
TB3
1
10
2
9
NCP1605
C8
C9
Rout3
Cbo
220n
180 uF 450 V
Rsense
Rbo2 Ccomp
62k
680n Rzcd
12k
Rof f set Cosc1
56
270p
Cosc2
470p
24.3k
0.1 uF
22 uF
Rdrv 2
4.7k
Rdrv 1
33k
Figure 4−4. Circuit Schematic of NCP1605 Based PFC Converter
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ON Semiconductor
Table 4−1. Bill of Materials for the NCP1605 based PFC Converter
Quantity
Reference
Description
Part Number
Manufacturer
1
C1
0.47 uF / 275 V type X2
F1772-447-2000
VISHAY
1
C2
0.47 uF / 275 V type X2
F1772-447-2000
VISHAY
1
C3
1u, 400V, High Ripple, Polypropylene Cap
ECWF4105JL
Matsushita
1
C4
180 uF 450 V
2222 159 47181
BC Components
1
Cbo
0.22 uF / 50V
K224K20X7RF53H5
VISHAY
1
Ccomp
0.68 uF / 50V
K684K20X7RF53H5
VISHAY
1
Cosc1
270 pF / 50 V
K271K15X7RF53H5
VISHAY
1
Cosc2
470 pF / 50 V
K471K15X7RF53H5
VISHAY
1
Ct
10 nF / 50 V
K103K15X7RF53H5
VISHAY
1
C9
0.1 uF / 50V
K104K15X7RF53H5
VISHAY
1
C8
22 uF / 25 V
2222 013 36229
BC Components
1
DB1
600V, 8.0A bridge diode
GBU8J
VISHAY
1
D1
5.0A, 520V
MUR550PF
ON Semiconductor
2
D2
1N4148
1N4148
VISHAY
Dzcd
1N4148
1N4148
VISHAY
1
F1
5 A fuse, Time Delay Fuse (FST 5x20)
0034.3124
SCHURTER
1
IC1
FCCrM PFC controller
NCP1605
ON Semiconductor
1
L1
250uH (56 turns to 3 turns)
1
L2
4 A, 2 x 6.8 mH, CM choke
B82725-J2402-N20
EPCOS
1
L3
150 uH, 5A, WE-FI series, DM choke
7447055
Wurth Elektronik
1
Q1
20 A 600 V MOSFET
SPP20N60C3
Infineon
2
Rbo1a
Resistor, Axial Lead, 3.6 M, 1/4 W, 1%
CCF503M60FKE36
VISHAY
Rbo1b
Resistor, Axial Lead, 3.6 M, 1/4 W, 1%
CCF503M60FKE36
VISHAY
1
Rbo2
Resistor, Axial Lead, 62 k, 1/4 W, 1%
CCF5062K0FKE36
VISHAY
1
Rdrv1
Resistor, Axial Lead, 33 k, 1/4 W, 1%
CCF5033K0FKE36
VISHAY
1
Rdrv2
Resistor, Axial Lead, 4.7 k, 1/4 W, 1%
CCF504K70FKE36
VISHAY
1
Rocp
Resistor, Axial Lead, 3.6 k, 1/4 W, 1%
CCF503K60FKE36
VISHAY
1
Rdrv2
Resistor, Axial Lead, 56, 1/4 W, 1%
CCF5056R0FKE36
VISHAY
2
Rout1a
Resistor, Axial Lead, 2 M, 1/4 W, 1%
CCF502M00FKE36
VISHAY
Rout1b
Resistor, Axial Lead, 2 M, 1/4 W, 1%
CCF502M00FKE36
VISHAY
1
Rout2
Resistor, Axial Lead, 1.82 k, 1/4 W, 1%
CCF501K82FKE36
VISHAY
1
Rout3
Resistor, Axial Lead, 24.3 k, 1/4 W, 1%
CCF5024K3FKE36
VISHAY
1
Rsense
Resistor, Axial Lead, 0.1, 3 W, 1% LVR3 series
LVR03 R1000 F E12
VISHAY
2
Rzcd
Resistor, Axial Lead, 12 k, 1/4 W, 1%
CCF5012K0FKE36
VISHAY
R2
Resistor, Axial Lead, 12 k, 1/4 W
CCF5012K0FKE36
VISHAY
1
Rzcd1
Resistor, Axial Lead, 24 k, 1/4 W, 1%
CCF5024K0FKE36
VISHAY
1
R1
Resistor, Axial Lead, 91 k, 1/4 W
CCF5091K0FKE36
VISHAY
1
R4
Resistor, Axial Lead, 10 k, 1/4 W
CCF5010K0FKE36
VISHAY
1
R5
Resistor, Axial Lead, 10, 1/4 W
CCF5010R0FKE36
VISHAY
1
TB1
AC inlet connector
GSF1.1201.31
SCHURTER
1
TB2
DC output plug socket
20.101/2 (order code 3044531)
IMO
WEIDMULLER
1
TB3
Vcc connector plug socket
PM5.08/2/90.
(order code 5015571)
1
HS1
Heatsink (2.9 °C/W)
SK481 100mm
Fischer elektronik
2
Q1
Isolator TO220
3223-07FR-43
BERGQUIST
D1
Isolator TO220
3223-07FR-43
BERGQUIST
DB1
Clip for heatsink (TO220)
THFU 1
Fischer elektronik
Q1
Clip for heatsink (TO220)
THFU 1
Fischer elektronik
D1
Clip for heatsink (TO220)
THFU 1
Fischer elektronik
3
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PFC Handbook
Results and Performance Curves
The results of the NCP1605 based design described above are shown below.
Table 4−2. Performance Results of the 270 W NCP1605 PFC
Vin (Vac)
Pin (W)
Vo (V)
Io (A)
Output
Power (W)
Pout
Efficiency
PF
THD
100
28.63
388.24
0.070
27.21 W
10%
95.05%
0.972
4.6
42.52
388.19
0.105
40.77 W
15%
95.89%
0.987
4.3
56.41
388.17
0.140
54.32 W
20%
96.29%
0.992
4.3
140.88
387.96
0.350
135.84 W
50%
96.43%
0.998
2.9
284.04
387.64
0.700
271.40 W
100%
95.55%
0.996
7.5
28.58
388.28
0.070
27.21 W
10%
95.21%
0.956
6.2
42.40
388.23
0.105
40.77 W
15%
96.15%
0.977
3.9
56.25
388.19
0.140
54.32 W
20%
96.56%
0.987
3.4
140.29
388.04
0.350
135.87 W
50%
96.85%
0.997
2.7
281.60
387.77
0.700
271.44 W
100%
96.39%
0.997
6.1
29.26
400.11
0.070
28.06 W
10%
95.90%
0.734
18.6
42.35
389.76
0.105
41.05 W
15%
96.94%
0.825
13.6
115
230
55.77
388.58
0.140
54.37 W
20%
97.49%
0.877
9.6
138.25
388.19
0.350
135.91 W
50%
98.30%
0.969
6.7
276.11
388.07
0.700
271.61 W
100%
98.37%
0.988
7.2
As shown in Table 4−2 the full load efficiency of the circuit at 100 Vac input is comfortably above 95%. The efficiency shows
even better performance at light load (96.29% at 20% load). These are derived from the low switching losses and frequency
clamping associated with FCCrM approach. The PF and THD at high line, full load are 0.988 and 7.2% respectively – which
are quite good. At lighter load, the PF does go down, but not very significantly. Another interesting result to note is that the
high line full load efficiency crosses 98%.
The harmonic analysis shown in Figure 4−5 shows that the converter passes class D limits easily. The data for the 75 W input
condition were captured as shown in Figure 4−6 also show positive result.
Amplitude (mA/W)
4.00
Result
3.50
Class D requirement
3.00
2.50
2.00
1.50
1.00
0.50
0.00
3
5
7
9
11
13
15
17
19
21
23
25
27
29
31
33
Harmonic number
Figure 4−5. Harmonic Spectrum of 270 W FCCrM Converter at Full Load
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35
37
39
ON Semiconductor
Amplitude (mA/W)
4.00
Result
3.50
Class D requirement
3.00
2.50
2.00
1.50
1.00
0.50
0.00
3
5
7
9
11
13
15
17
19
21
23
25
27
29
31
33
35
37
39
Harmonic number
Figure 4−6. Harmonic Spectrum of 270 W FCCrM Converter at 75 W Load
Another key attribute to measure is the no load power drawn from the line at various input conditions. Table 4−3 shows these
measurements. These measurements indicate that the no load input power is quite low for the FCCrM due to the frequency
clamping and is also independent of the input voltage level.
Table 4−3. No Load Power Consumption
Vin
Pin (W)
100 Vac
0.125
115 Vac
0.118
230 Vac
0.131
The other areas of optimization for FCCrM investigated included the choice of optimal boost diode and also the optimization
of the ZCD feature to get valley switching and minimize turn-on losses. It should be remembered that the relatively high rms
current means that the trade-off of Rds on vs. switching loss is not an option and hence the 20 A MOSFET was not modified.
By changing the boost diode from the 8 A, 600 V ultrafast diode (optimized for CCM) to 5 A, 520 V diode (MUR 550, optimized
for CrM/FCCrM), the efficiency improvement at 100 Vac and full load was 0.22%. Next, by fine tuning the ZCD circuit to
achieve precise valley switching, another efficiency improvement of 0.56% at 100 Vac and full load was achieved. The final
results shown in Table 4−2 incorporate these enhancements. Figure 4−7 depicts the effects of the ZCD optimization effect. In
the left figure, the switch turns on prior to Vds going to its valley and hard turn-on results in higher switching losses. By adding
the ZCD delay (NZCD = 3 turns, RZCD1 = 24 kΩ, and DZCD = 1N4148), the Vds can be brought to near zero as shown in the
right waveform and the efficiency is improved. Here, Ch1 is the input voltage, Ch2 is the inductor current, Ch3 is the Vds
waveform and Ch4 is the gate drive waveform.
Figure 4−7. Turn−On Waveforms with (Left) No ZCD Delay and (Right) Optimized ZCD Delay
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PFC Handbook
The Figure 4−8 shows the switching waveforms of the converter at full load and 115 Vac input. Here, Ch1 is the input voltage,
Ch2 is the inductor current, Ch3 is the Vds waveform and Ch4 is the gate drive waveform. As seen here, the switching frequency
is about 47 kHz.
Figure 4−8. Switching Waveforms at Full Load, 115 Vac Input
Another critical contribution of the FCCrM circuit is the frequency clamping. As indicated in step 3, the maximum frequency
chosen is 65 kHz. So, the frequency variation is limited compared to pure CrM operation shown in Chapter 3. As shown in
Table 4−4, the frequency stays clamped at 62 kHz (close to the designed value). As a result, the light load operation is always
in fixed frequency mode. Even at high line, full load and zero crossing, the frequency clamping comes into place. However,
at other full load conditions, the converter operates in the CrM mode, keeping the peak currents in check. The frequency at
peak and zero crossing appears to be the same for low line and full load conditions. This may be attributable to measurement
error margins. Typically, the low line frequency variation between the peak and the zero crossing is not that much, so the
measurement errors make them look the same.
Table 4−4. Frequency Variation in FCCrM Application
Full load (0.7 A)
10 % load (0.07 A)
Peak Vin
(kHz)
Zero Cross
(kHz)
Peak Vin
(kHz)
Zero Cross
(kHz)
88 Vac
35.2
37.6
59.8
61.9
264 Vac
25.3
62.2
618
62.5
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ON Semiconductor
Efficiency
The results from Table 4−2 are plotted in graph form in figures below.
98.5%
98.0%
97.5%
97.0%
96.5%
96.0%
95.5%
95.0%
100 Vac
115 Vac
230 Vac
0%
20%
40%
60%
80%
100%
Output Power
Figure 4−9. Efficiency Plot for FCCrM PFC
Power Factor
100.0%
95.0%
90.0%
85.0%
100 Vac
115 Vac
230 Vac
80.0%
75.0%
70.0%
0%
20%
40%
60%
80%
100%
Output Power
Figure 4−10. Power Factor Plot for FCCrM PFC
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PFC Handbook
References
[1] NCP1601 Datasheet, ON Semiconductor. http://www.onsemi.com/pub_link/Collateral/NCP1601A­D.PDF
[2] “Implementing the NCP1605 to drive the PFC stage of a 19 V/8 A Power Supply ”, AND8281/D,
ON Semiconductor. http://www.onsemi.com/pub_link/Collateral/AND8281­D.PDF
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ON Semiconductor
CHAPTER 5
Continuous Conduction Mode (CCM) PFC
The continuous conduction mode (CCM) PFC has been around for many years and as indicated in [1], average current mode
control is the most convenient way of achieving CCM PFC. The traditional average current mode controllers and their
improvements have been described before [2]. More recently, predictive CCM PFC controllers have been introduced as
described in Chapter 1. This section walks the reader through the design of a continuous conduction mode boost PFC circuit
utilizing the NCP1654 predictive PFC controller, which has 65 kHz, 133 kHz, and 200 kHz operating frequency versions. The
description here is restricted to major design choices and their analyses. More in-detail designs are provided in the products’
datasheets and application notes.
CCM Introduction
The CCM operation is more popular at higher power levels as it has minimal peak and rms currents. In comparison with the
CrM operation, the peak currents can be 50% lower and rms currents can be 25% lower. This reduces the stress in power FET,
diode and inductor. In addition, the filtering is easier as the current through the boost inductor is more continuous. Finally, the
switching frequency remains constant for the CCM operation, so the boost inductor design and EMI filter design are easier.
However, traditional control algorithms and circuitry for CCM PFC have always been more complex and 16-pin controllers
with significant number of external components were common in the CCM applications for a long time. In recent years, the
predictive control method has resulted in a simplified approach being available and NCP1653 and NCP1654 are examples of
such simple methodology for CCM PFC.
NCP1654 – Simplified CCM PFC Controller
Housed in a DIP8 or SO8 package, the NCP1654 minimizes the number of external components and drastically simplifies
the PFC implementation. It also integrates high safety protection features that make the NCP1654 a driver for robust and
compact PFC stages (e.g. input power runaway clamping circuitry). Following features make the NCP1654 a unique solution:
•
•
•
•
•
•
•
•
•
•
Flexibility to operate in traditional or follower boost mode
Low power consumption and shutdown capability
Key safety and protection features including:
Maximum current limit –directly applicable to the inductor current
Feedback network failure protection (Under voltage protection/shutdown)
Brown-out detection and shutdown to prevent thermal overheating
Output overvoltage protection
Overpower protection and shutdown
Thermal shutdown
Improved output voltage regulation accuracy (compared to NCP1653)
The block diagram of NCP1654 is shown in Figure 5−1.
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PFC Handbook
IN
EMI
Filter
AC
Input
Iin Vin
+
L
IL
Cfilter
Cbulk
Output
Voltage
(VOUT)
+
−
IL
RSENSE
OVP
+
105% Vref
UVP
+
RboU
RfbU
BO
Vdd
OPL
8% Vref with 4% Vref
Hysteresis
Vdd
UVLO
200 A
Soft Start
Undervoltage
Lock−Out
S
FB
6
Reference
Block
Q
R
Vref
RfbL
95% Vref
+
Vref
+
Vcontrol
Off
Vout Low Detect
Iref
OTA
Vcc
7
+
Output
Buffer
Vcontrol(min)
CP
Vdd
±28 A
5
RZ
Bias Block
UVP BO
CZ
8
1
GND
Thermal
Shutdown
BO
Fault
4
RboL
BO
+
CBO
VboH / VboL
VboH = 1.3 V, VboL = 0.7 V
OL OVP
Vref
Vdd
Vramp
Ics
RCS
R
S
Iref
Vbo
Vref/10% Vref
Vm
+
Division
C1
Ics*Vbo > 200 VA
OPL
PWM
Latch
+ -
CS
3
R
Vdd
Current Mirror
Ics
Q
+
Vdd
Ics
DRV
S1
2
65/133/200 kHz
Oscillator
OL
Im = (Ics*Vbo) / (4*(Vcontrol − Vcontrol(min))
Ics > 200 A
OCP
Figure 5−1. NCP1654 Block Diagram
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RM
CM
ON Semiconductor
Design Steps with 270 W Example
The step-by-step design procedure for CCM PFC using the NCP1654 is applied next to illustrate the design of a 270 W CCM
PFC used as a basis for comparison (design P3 described in Chapter 2).
Step 1: Define the key specifications
Minimum Input voltage (VacLL): 88 Vac (this is usually 10-12% below the minimum typical voltage which could be 100 Vac
in many countries).
Maximum Input voltage (VacHL): 264 Vac (this is usually 10% above the maximum typical voltage which could be 240 Vac
in many countries)
Line frequency (fLINE): 50 Hz /60 Hz (This is often specified in a range of 47-63 Hz and for calculations such as hold-up time,
one has to factor in the lowest value specified)
Output Voltage (Vout): 385 V (This value has to be above 1.414 S VacHL and is typically between 385 and 400 V for universal
input operation)
Maximum output voltage (Vout(max)) : 415 V (This value is usually 7-10% above the Vout value and is determined by the
accuracy of the OVP level of the PFC controller – conversely, the selection of the PFC controller has to be made based on this
specification if it is determined by other component stress levels and derating factors).
PFC maximum output power (Pout): 270 W (This is the specified output power for the PFC stage. It is important to factor in
the follow-on stage efficiency when specifying this parameter – it will always be higher than the specified maximum system
output power)
Switching frequency (fsw): 65 kHz (This parameter helps set the value of the boost inductor). For the NCP1654, the frequency
is set internally with 3 options, which are 65 kHz, 133 kHz, and 200 kHz. The 65 kHz is the common choice for easier EMI
compliance, while 133 kHz and 200 kHz are chosen for applications needing lower profile or smaller inductor.
Output voltage ripple (Vripple(p-p)): 20 V (This parameter is often specified in percentage of output voltage, +/- 5% is a very
typical specification)
Hold-up time (thold-up): 16 ms (This parameter specifies the amount of time the output will remain valid during line drop-out.
One line cycle is typically specified. For PFC alone, this spec is not applicable, but the PFC output capacitor is the single largest
determinant of the hold-up time)
Estimated efficiency (η): 93% (This parameter is an initial estimate that is used to size the power stage components – high level
of accuracy is not needed for the design procedure).
Step 2: Design the Boost Inductor (PFC coil)
The (maximum) peak inductor current is the about half the value of the corresponding value in CrM and FCCrM. This does
not take into account the peak to peak ripple due to non-infinite inductor value. This current is given by:
I coil , pk (max) =
2 Pout
2 270
=
= 4.67 A
h Vac LL 0.93 88
Similarly, ignoring the contribution of the p-p ripple (which can be shown to contribute less than 10% to the rms value for
most selections), the rms current in the inductor is given by:
I coil ,rms =
Pout
270
=
= 3.3 A
h Vac LL 0.93 88
Unlike the CrM and FCCrM topologies, there is no minimum or maximum inductance value equation for the CCM PFC
(except that to maintain the CCM operation, the ripple should be less than 100% p-p). The inductor value selection is somewhat
iterative and is determined based on the peak current, ripple current, output ripple voltage, components stress and losses, as
well as board space. Typically, higher inductance value will reduce ripple and stress levels, but use up significant board space.
In contrast, lower inductance value will increase the ripple and peak currents, but will have benefits of smaller size and lower
current at diode turn-off – as will be shown in the results section, this lower diode turn-off current results in significant
improvement of diode switching characteristics and efficiency.
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PFC Handbook
A first approximation of the inductor value L can be obtained with the following equation:
L=
2
Vac LL
2 I% f sw P
out
where
h
ƪ ǒ
1 *
Ǹ2 @ Vac
LL
Vout
Ǔƫ
L = inductance value
I% = ratio of allowable pk-pk ripple current to peak current in the inductor (25-45% typical)
Inductance (mH)
The following chart helps in defining a range of inductances based on the allowable ripple current. It is recommended to use
a value of inductance that falls within the 25-45 % range of input current ripple. In this particular design, p-p ripple value applied
was 45% and it resulted in the inductor value of 650 μH. The resultant modified peak current is 22.5% above the calculated
value above (4.67 S 1.225=5.72 A).
Figure 5−2. CCM Inductance Value as a Function of p−p Ripple
Step 3: Design the Power Stage Components
The power stage components are designed based on their current and voltage ratings. The inductor design is already covered
in step 2. The bridge diode should be selected based on peak inductor current rating and will have the power dissipation given
by:
Pbridge =
V
V
P
P
4 2
f out [1.8 f out
p Vac LL h
Vac LL h
The MOSFET is selected based on peak voltage stress (Vout(max)+margin) and rms current stress:
I M ( rms ) =
Pout
8 2 Vac LL
270
8 2 88
1−
=
1−
= 2.81 A
h Vac LL
3 p Vout
0.93 88
3 p 385
Using a 600 V, 0.19 ohm FET, will give conduction losses of (assuming that Rds(on) increases by 80% due to temperature
effects):
2
Pcond = I M ( rms ) Rds ( on ) = 2.812 0.19 1.8 = 2.7 W
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While switching losses are harder to predict without getting into the details of switching waveforms and diode recovery
characteristics, etc., one component of the switching losses that can be predicted is the capacitive turn-on losses.
Psw,cap =
2
Coss 25 25 Von1.5 f = 0.666 780 10 −12 5 (385)1.5 36000 = 0.71 W
3
Here, the nonlinear nature of the capacitance is taken into account. The other switching losses of the FET are dependent on
the choice of boost diode, switching speeds and the choice of snubbers in the circuit. In the CCM circuit, the diode recovery
losses often dominate the switching losses and hence, the choice of FET is less critical in this application.
The choice of boost diode is more critical in the CCM applications. As mentioned before, the diode current at its turn-off
is non-zero unlike the CrM application. As a result, there is significant reverse recovery phenomenon that leads to dissipation
in the diode and the FET. One option is to use advanced rectifiers such as SiC diodes and such. However, they are expensive
and their higher forward drop offsets some of the gains of reduced switching losses. For this application, a soft recovery boost
rectifier (LQA08TC600) was used and it worked well due to its softer recovery characteristics and additional snubbers were
not necessary. The average diode current is the same as the output current (Id = Pout/Vout = 270/385 = 0.7 A). So, the diode
conduction losses are Id S Vf (generally less than 1 W). The peak current seen by the diode will be the same as the inductor peak
current (4.67 A).
Step 4: Output capacitor design
The output capacitor is designed considering 3 factors: output voltage ripple, output current ripple and the hold-up time. The
output voltage ripple is given by:
Vripple ( p − p ) =
Pout 2 p f line C out Vout
The capacitor rms current is given by (assuming a resistive load):
I Cout ( rms ) =
2
32 2 Pout
P
− ( out ) 2
2
Vout
9 p Vac LL Vout h
However, almost always, the size and the value of the capacitor are determined by the hold-up time which is given by:
t hold −up =
2
2
C out (Vout
− Vmin
)
2 Pout
In this case, a 220 μF capacitor was chosen that satisfies the conditions above. The peak-peak ripple is 10 V, the rms current
is 1.83 A and the hold-up time is 20.82 ms (for a 330 V Vmin).
Step 5: Set FB, OVP and UVP levels
The OVP level in the NCP1654 is set at 5% above the nominal Vout given by
Vout ,OVP = 105% Vout ,nom
As the nominal output voltage is set at 385 V. The OVP level is:
Vout ,OVP = 105% 385 = 404 V
The choice of feedback resistors does not play a critical role in setting the OVP level. As a result, the feedback divider values
are selected as follows.
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PFC Handbook
First, choose the value of the lower resistor, RfbL. There is a trade-off between the noise immunity and the power losses when
choosing RfbL. In this application, we select 23.2 kΩ as RfbL that leads a 108 mA feedback current and 42 mW losses. The value
of upper resistor RfbU is then given by:
R fbU =
Vout − VREF
R fbL
VREF
where:
VREF is the internal reference voltage for Vout feedback (2.5 V typical).
RfbU = RfbU1 + RfbU2 is the total feedback resistor placed between Vout and FB pin.
In this case, Vout is 385 V and RfbL is 23.2 k, one must then select the following RfbU resistance:
R fbU =
385 − 2.5
23.2 kW = 3.549 MW
2.5
The feedback string is implemented using two equal resistors RfbU1 = RfbU2 = 1.8 MΩ.
The Under-Voltage Protection (UVP) function has 2 purposes.
• Open Loop Protection - Protect the power stage from damage at feedback loop abnormal, such as Vfb is
shorted to ground, the feedback resistor RfbU is open, or the FB pin is left open.
• Shutdown mode - Disables the PFC stage and forces a low consumption mode. This feature helps to meet
stringent stand-by specifications. Power Factor being not necessary in stand-by, the PFC stage is generally
inhibited to save the pre-converter losses. To further improve the stand-by performance, the PFC controller
should consume minimum current in this mode.
The UVP level in the NCP1654 is set when Vfb is below 8% of VREF, the device is shut down. The device automatically starts
operation when the output voltage goes above 12% of VREF. In normal situation of boost converter configuration, Vfb has to
be greater than 12 % of VREF to enable the NCP1654 to operate. For this case, the UVP level is set at:
Vout ,UVP = 12% Vout ,nom = 12% 385 = 46V
It is around 32 Vac input, which is much lower than the minimum input voltage, i.e. 85 Vac, and is ok for start-up.
Step 6: Input voltage sensing
The NCP1654 monitors the input voltage, Vin, which is the rectified AC line sinusoid for brown-out, over-power limitation
(OPL), and PFC duty cycle modulation. This sensing circuit consists of:
RboU and RboL are dimensioned to adjust the threshold of brown out protection. Because of the safety consideration, it is
recommended to split this upper side brown out resistor into 2 or more resistors.
CBO that forms a low pass filtering together with RboL to get the average value of input signal. A time constant in the range
of around 5 times the Vin period should be targeted to make Vbo substantially constant and proportional to the mean input
voltage as the rule of thumb:
Vbo =
RboL
Vin
RboL + RboU
The NCP1654 starts to operate as Vbo exceeds 1.3 V and keeps operating until Vbo goes below 0.7 V. The 600 mV hysteresis
prevents the system from oscillating.
RboU + RboL should be relatively high impedance to limit the current drawn and the associated losses. Please note however
that given the bias current of the brown-out comparator (0.5 mA maximum), it is recommended to set the current flowing
through RboU and RboL to be in the range or higher than 5 μA at low line. In this application, we use 82.5 kW for RboL, which
leads to a bias current of:
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ON Semiconductor
0.7V
82.5k
= 8.5 mA
Second, select RboU according to Vac,on, the minimum AC input voltage to start PFC, which comes from:
RboU =
2Vac,on − VBOH
RboL
V BOH
In this application, 75 Vac is targeted as Vac,on. Hence,
RboU =
2 75V − 1.3V
82.5 kW [6.65 MW
1.3V
Here RboU is split into 2 parts, RboU1 and RboU2 both equal to 3.3 MW for a total of 6.6 MW resistance.
Third, select CBO to make the time constant be around 5 times TVin, the cycle time of Vin by
5 T
C BO [ Vin
RboL
In this application, TVin is 10 ms since the ac input line is 50 Hz. So . Here 0.47 mF is selected because it is the closest
normalized value.
5 10ms
C BO [
= 0.6 mF
82.5kW
Fourth, check Vacoff, the PFC brown-out off threshold of AC input voltage.
Vac,off =
V BOL
f
2 2 ⎛
K BO ⎢⎢1 − BO
p ⎝ 3 f line
⎞
⎟⎟
⎠
Where KBO is the scaling factor formed by RboU and RboL and fBO is the corner frequency of the BO filter.
In this application,
Vac,off =
0.7
2 2 ⎛
4.2 Hz ⎞
0.0123 ⎢1 −
⎟
p ⎝ 3 50 Hz ⎠
= 64.8 Vac
which seems acceptable. By reducing CBO, one can increase this level and vice versa. Note that this calculation result is just
a reference since it doesn’t include the voltage drop on the current loop, i.e. EMI filter and bridge diode etc.
Step 7: Current Sense Circuit
The current sense signal across the sense resistor is actually a negative voltage. However, insertion of a resistor between
Rsense and pin 3 of the NCP1654 allows that signal to be converted into a current. The value of Rsense is chosen to minimize
its impact on efficiency at low line, full load (here, 0.1 Ω is chosen). The equation for Rcs is given by:
Rcs =
I coil , pk Rsense
I pin 3(min)
=
4.67 0.1
= 2.52 kW
185 10 −6
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PFC Handbook
RM adjusts the maximum power the PFC stage can supply given the chosen output voltage level. By choosing RM high
enough, you can force the “Follower Boost Operation”. Use the following equation to select RM:
2pRCS @ DVCONTROL @ VREF
RM + 70% @ h
@ VacLL + 45.4 kW
Ǹ2 @ R
sense @ KBO @ VoutLL @ Pout, max
In this example, follower boost is not utilized, so the VoutLL value is same as Vout and plugging in the other values results
in a RM value of 45.4 kW - a 47 kW value is used. The value of CM is chosen so that the RM.CM time constant is at least 5 times
the switching period. Based on this, the CM value is calculated to be 1 nF.
Additional design steps such as feedback compensation, Vcc generation follow conventional methods and are not covered
here.
Circuit Schematics and Bill of Materials
The circuit schematic of the CCM PFC converter described in the design steps above is provided in Figure 5−3. The Bill of
Materials follows in Table 5−1.
R2
F1
L
R1
DB1
L1
G
N
C1
D1
C2
TB2
C3
Q1
TB1
C4
+
R4
R6
R9
R7
IC1
R13
R10
C3
R 11
C6
VM
VCC
CS
FB
BO
Vcontrol
R8
C5
D2
R5
NCP1654
DRV
GND
+15 V
C8
R12
C9
C12
R3
C10
Figure 5−3. Circuit Schematic of the CCM PFC Circuit Based on NCP1654
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+
TB3
ON Semiconductor
Table 5−1. Bill of Materials for CCM PFC Circuit Based on the NCP1654
Quantity
Reference
Description
Part Number
Manufacturer
1
C1
0.47 uF / 275 V type X2
F1772-447-2000
VISHAY
1
C2
0.47 uF / 275 V type X2
F1772-447-2000
VISHAY
1
C3
0.1u, 400V, High Ripple, Polypropylene Cap
ECWF4104JL
Matsushita
1
C4
180 uF 450 V
2222 159 47181
BC Components
1
C5
0.22 uF / 50V
K224K20X7RF53H5
VISHAY
1
C7
0.47 uF / 50V
K474K20X7RF53H5
VISHAY
1
C9
0.1 uF / 50V
K104K15X7RF53H5
VISHAY
1
C6
1 nF / 50 V
K102K15X7RF53H5
VISHAY
1
C10
100 pF / 50 V
K101K15X7RF53H5
VISHAY
1
C8
22 uF / 25 V
2222 013 36229
BC Components
1
C12
2.2 uF / 50 V
B32529D5225J
EPCOS
1
DB1
600V, 8.0A bridge diode
GBU8J
VISHAY
1
D1
8.0A, 600V
MSR860G
ON Semiconductor
1
D2
1N4148
1N4148
VISHAY
1
F1
5 A fuse, Time Delay Fuse (FST 5x20)
0034.3124
SCHURTER
1
IC1
CCM PFC controller
NCP1654
ON Semiconductor
1
L1
650uH
GA3199-AL
CoilCraft
2702.0010A
Pulse
1
L2
4 A, 2 x 6.8 mH, CM choke
B82725-J2402-N20
EPCOS
1
L3
150 uH, 5A, WE-FI series, DM choke
7447055
Wurth Elektronik
1
Q1
20 A 600 V MOSFET
SPP20N60C3
Infineon
2
R1
Resistor, Axial Lead, 1.8 M, 1/4 W, 1%
CCF501M80FKE36
VISHAY
R2
Resistor, Axial Lead, 1.8 M, 1/4 W, 1%
CCF501M80FKE36
VISHAY
R9
Resistor, Axial Lead, 3.3 M, 1/4 W, 1%
CCF503M30FKE36
VISHAY
VISHAY
2
R13
Resistor, Axial Lead, 3.3 M, 1/4 W, 1%
CCF503M30FKE36
1
R10
Jumper
Jumper
1
R3
Resistor, Axial Lead, 23.7 k, 1/4 W, 1%
CCF5023K7FKE36
VISHAY
1
R4
Resistor, Axial Lead, 10 k, 1/4 W
CCF5010K0FKE36
VISHAY
1
R5
Resistor, Axial Lead, 10, 1/4 W
CCF5010R0FKE36
VISHAY
1
R6
Resistor, Axial Lead, 0.1, 3 W, 1% LVR3 series
LVR03 R1000 F E12
VISHAY
1
R7
Resistor, Axial Lead, 3.6 k, 1/4 W 1%
CCF503K60FKE36
VISHAY
1
R8
Resistor, Axial Lead, 47 k, 1/4 W
CCF5047K0FKE36
VISHAY
1
R11
Resistor, Axial Lead, 82.5 k, 1/4 W, 1%
CCF5082K5FKE36
VISHAY
1
R12
Resistor, Axial Lead, 12 k, 1/4 W, 1%
CCF5012K0FKE36
VISHAY
1
TB1
AC inlet connector
GSF1.1201.31
SCHURTER
1
TB2
DC output plug socket
20.101/2
(order code 3044531)
IMO
1
TB3
Vcc connector plug socket
PM5.08/2/90.
(order code 5015571)
WEIDMULLER
1
HS1
Heatsink (2.9 °C/W)
SK481 100mm
Fischer elektronik
2
Q1
Isolator TO220
3223-07FR-43
BERGQUIST
D1
Isolator TO220
3223-07FR-43
BERGQUIST
3
DB1
Clip for heatsink (TO220)
THFU 1
Fischer elektronik
Q1
Clip for heatsink (TO220)
THFU 1
Fischer elektronik
D1
Clip for heatsink (TO220)
THFU 1
Fischer elektronik
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PFC Handbook
Results and Performance Curves
The operating results of the NCP1654 based design described above are shown in Table 5−2.
Table 5−2. Results of the 270 W CCM PFC Using NCP1654
Vin (ac)
Pin (W)
Vo (V)
Io (A)
Output Power (W)
Po
Efficiency
PF
100
28.44
382.78
0.070
26.97
10%
94.83%
0.975
41.56
381.95
0.104
39.80
15%
95.76%
0.984
115
230
56.07
381.79
0.141
53.88
20%
96.10%
0.991
139.59
381.40
0.350
133.65
50%
95.75%
0.996
280.91
381.26
0.701
267.10
100%
95.08%
0.998
28.70
383.03
0.072
27.39
10%
95.43%
0.968
42.15
382.18
0.105
40.31
15%
95.64%
0.981
55.52
381.83
0.140
53.42
20%
96.22%
0.985
138.41
381.43
0.349
133.28
50%
96.30%
0.994
278.29
381.30
0.700
266.77
100%
95.86%
0.997
27.67
382.63
0.069
26.55
10%
95.97%
0.757
42.04
382.31
0.106
40.63
15%
96.65%
0.858
55.33
382.13
0.141
53.78
20%
97.21%
0.905
136.59
381.64
0.350
133.71
50%
97.89%
0.978
272.60
381.36
0.701
267.15
100%
98.00%
0.991
THD%
4.6
4.6
7.2
From the information in Table 5−2, we can confirm that the full load efficiency at 100 Vac is above 95%. In addition, even
at 20% load (which is a key performance criterion from regulatory viewpoint), the efficiency is above 96%. These results are
achieved with a combination of optimal component choices and careful layout of the PCB. It can also be observed that the power
factor remains above 0.9 for high line (230 Vac) at loads down to about 50 W (20%). The THD of the circuit at full load and
high line is measured at 7.2% which is quite respectable.
The input harmonic spectrum for 230 Vac and full load is shown in Figure 5−4 against class D limit and it is shown that the
circuit comfortably passes the IEC61000-3-2 requirements. Similar result was observed with the load reduced to 75 W.
Figure 5−4. Harmonic Spectrum of the 270 W CCM Converter
Some of the design trade-offs involved in component choices are illustrated next using changes in two key components and
how they impact the efficiency at critical points. In the above design, the MOSFET used is a 20 A, 600 V (0.19 Ω) and the boost
diode is 8 A, 600 V. By experimenting with 15 A, 600 V FET and 3 A, 600 V diode from the same vendors/component class,
following result variations are obtained at 100 Vac input. Similar performance changes are observed at the 115 Vac input.
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Table 5−3. Impact of Component Variations on Efficiency (100 Vac)
100% load
15 A FET
20 A FET
3 A diode
94.82%
94.99%
8 A diode
94.89%
95.08%
20% load
15 A FET
20 A FET
3 A diode
95.88%
95.91%
8 A diode
96.06%
96.08%
The results of Table 5−3 offer interesting insight into the trade-offs involved in component selection for PFC circuits. First,
it is interesting to note that the efficiency change when going from [15 A FET, 3 A diode] combination to [20 A FET, 8 A diode]
combination results in about 0.25% efficiency improvement at full load (0.2% at 20% load). This implies that the gains in the
conduction losses made by going to bigger FET are offset by the increased switching losses associated with higher Coss.
Depending on the costs of the available devices and efficiency targets, Table 3 offers an interesting set of options for selecting
right components and optimizing the design. In another set of experiments, the boost diode was replaced with a different
category of diode (Q series to X series) resulted in efficiency drop of about 0.5% at low line and 20% load.
Finally, a change in the inductor value from 650 μH to 250 μH was tried. Such a drastic reduction in inductor value takes
the circuit closer to DCM at lighter loads and the power factor gets worse as shown in Table 5−4. However, since the ripple
value gets bigger, the inductor current at the diode turn-off instance is significantly below its peak and average values. The result
of this is that the diode recovery effects are not as bad and the efficiency and EMI improve. For a given design requirement,
this may be a good trade-off to make. Comparing with Table 2, the efficiency improves by 0.2% at full load and by about 0.33%
at 50% load (at 100 Vac). However, the 20% load efficiency reduces by about 0.27% - this is caused by higher ripple in the
inductor causing additional core losses.
Table 5−4. Impact of Reducing the Inductor Value in CCM
Vin (ac)
Pin (W)
Vo (V)
Io (A)
Output Power
(W)
Po
Efficiency
PF
THD%
100
28.55
383.71
0.071
27.09
10%
94.90%
0.966
13.6
41.97
382.58
0.105
40.13
15%
95.62%
0.977
12.2
56.52
382.06
0.142
54.16
20%
95.83%
0.982
12.2
139.08
381.53
0.350
133.62
50%
96.08%
0.992
9.3
280.26
381.31
0.700
267.05
100%
95.29%
0.996
5.9
28.97
383.66
0.072
27.54
10%
95.06%
0.969
12.0
42.29
382.67
0.106
40.54
15%
95.85%
0.972
11.9
55.75
382.16
0.140
53.60
20%
96.14%
0.978
11.8
138.42
381.59
0.350
133.48
50%
96.43%
0.989
11.6
277.90
381.35
0.700
266.93
100%
96.05%
0.996
5.1
27.70
382.70
0.070
26.60
10%
96.02%
0.731
23.0
41.08
382.45
0.104
39.77
15%
96.82%
0.839
15.5
55.43
382.14
0.141
53.88
20%
97.21%
0.893
12.4
136.72
381.61
0.351
133.83
50%
97.89%
0.954
15.7
272.05
380.29
0.701
266.58
100%
97.99%
0.980
13.9
115
230
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PFC Handbook
The results from Table 5−2 are plotted in the following figures for understanding performance over line and load.
Output Power
Figure 5−5. Efficiency Performance of the 270 W CCM Design
Output Power
Figure 5−6. Power Factor Performance of the 270 W CCM Design
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CHAPTER 6
Interleaved PFC
Interleaved PFC is an emerging solution that becomes particularly popular in applications where a strict form factor has to
be met, for instance, in low profile notebook adapters or in LCD TVs. This section will consider the interleaving of two FCCrM
PFC stages that efficiently address the 300−W application of our interest. Interleaving CCM PFC stages is also possible.
However, this option should be devoted to much higher power applications (above 1 kW) and will not be detailed here.
This section will give an overview of the interleaved PFC characteristics and will deal with the main design steps. Practical
results are given as obtained with the NCP1631. More detailed information is available at www.onsemi.com (see references).
Introduction
Interleaving consists in paralleling two “small” stages in lieu of a bigger one, which may be more difficult to design.
Practically, two 150−W PFC stages are combined to form our 300−W PFC pre−regulator. This approach has several merits like
the ease of implementation, the use of smaller components or better heat distribution.
Also, Interleaving extends the power range of Critical Conduction Mode that is an efficient and cost−effective technique (no
need for low trr diodes). Furthermore, if the two stages are operated out−of−phase, the current ripple is significantly reduced.
In particular, the input current looks like a CCM one and the rms current within the bulk capacitor is dramatically reduced.
Interleaving Method
When designing a CrM or FCCrM interleaved PFC, maintaining an out−of phase operation is the main difficulty. This is
because the switching frequency is not fixed and a nominal operation requires the MOSFET turn on to be delayed until the very
moment the valley is detected. Hence, each phase must beat at its own rhythm and at the same time stay synchronized to the
other branch, thus requiring a sophisticated control circuitry.
In the traditional master/slave approach, the master branch operates freely, while the other phase is controlled to follow with
a 180° phase shift. The challenge is to drive the slave branch so that it never enters CCM nor exhibits undesired dead−times
[1].
The NCP1631 utilizes an interactive−phase option where the two branches operate independently, in CrM, inherently
preventing risks of undesired dead−times or continuous conduction mode sequences. Still, the two phases interact with each
other for out−of−phase operation. It is worth noting that the NCP1631’s unique interleaving technique (which will be described
in this chapter) substantially maintains the wished 180° phase shift between branches in all conditions including start−up, fault
or transient sequences.
More generally, the NCP1631 integrates a dual MOSFET driver for interleaved, 2−phase PFC applications. It drives the two
branches in so−called Frequency Clamped Critical conduction Mode (FCCrM) where each phase operates in Critical
conduction Mode (CrM) in the most stressful conditions and in Discontinuous Conduction Mode (DCM) otherwise, acting as
a CrM controller with a frequency clamp (given by the oscillator). According to the conditions, the PFC stage actually
transitions from DCM to CrM (and vice versa) with no discontinuity in operation and without degradation of the current shape.
Traditional CrM control methods fail to offer this integrity of current shape, even though some of them transition to DCM
through use of frequency clamping.
The NCP1631 capitalizes on its FCCrM operation mode to manage the out−of−phase operation by the simple means of an
oscillator. As sketched by Figure 6−1, the oscillator voltage swings at twice the frequency of each branch. When it reaches
its 4−V lower threshold, the circuit generates a clock signal alternatively for phase 1 or phase 2.
There are two possible cases to consider:
1. The cycle time of the inductor current is shorter than two oscillator periods. When the clock signal is generated, the
inductor of the branch corresponding to the clock is demagnetized (the inductor current has already reached zero) and
a new cycle can immediately start. We have a fixed frequency DCM operation mode in each branch as depicted by
the right side of Figure 6−1. Out−of−phase operation is naturally obtained as the result of the interleaved clock
signals.
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PFC Handbook
2. The inductor current is not at zero when the clock signal corresponding to the inductor branch occurs. The circuit
would enter Continuous Conduction Mode (CCM) if the MOSFET turned on in that moment. Instead, the next
conduction phase is delayed until the core is reset and the inductor current reaches zero. The clock signal remains
high for this waiting period and the MOSFET turns on as soon as the coil current has reached zero. In other words,
critical conduction mode (CrM) operation is obtained. The oscillator keeps on discharging for the waiting period,
reducing the oscillator frequency and delaying the occurrence of the next clock signals. It can be analytically shown
that an appropriate selection of the oscillator charge and discharge currents leads to stable out−of−phase operation as
depicted by the left side of Figure 6−1.
5V
Vpin4
4V
Phase 1
CLK1
Wait for
demag of
DRV1 phase 1
CLK2
Wait for
demag of
phase 1
DRV2
CRM Operation
DCM Operation
Figure 6−1. Interleaved Clock Generation
As a voltage mode controller, the NCP1631 forces the MOSFET on−time to be identical in both branches. The
demagnetization time that only depends on the conduction time and on the line and output voltages is then the same in both
⎛
Vin
⎢⎢ tde m ag = ton V
out − Vin
branches as well ⎝
⎞
⎟⎟
⎠ . Hence if we neglect the tolerance in the timing circuitry that adjusts the on−time
in response to the control signal for each circuit, the current cycle duration is the same in the two branches even if their
respective inductors do not have the same inductance.
Finally, the only source of current unbalancing is the inductor tolerance. One can easily show that the current sharing is
governed by the following equation:
Iin ( branch1)
Iin ( branch 2)
=
Lbranch 2
Lbranch1
Where:
Iin(branch1) and Iin(branch2) are the averaged input currents drawn by phase 1 and phase 2 respectively
Lbranch1 and Lbranch2 are the inductance values of phase 1 and phase 2 respectively
Practically, if the inductance dispersion is in the range of ±5%, the possible current imbalance is less than ±10%.
Housed in a SOIC16 package, the NCP1631 also offers many other useful features. In particular, the circuit adapts its
frequency clamp to optimize the PFC stage efficiency over the line/load range: below a programmable load level, the frequency
linearly decays as a function of the power to maintain high efficiency levels even in very light load (frequency fold−back).
A NCP1631−driven PFC stage also eases the design of the downstream converter by providing it with a narrow range dc
voltage. With this goal, the circuit features a “pfcOK” output pin to disable the downstream converter until the bulk capacitor
is charged and no fault is detected. In addition, the NCP1631 dramatically reduces the output voltage deviations during abrupt
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load or line transients (when the low bandwidth of the PFC loop normally causes large overshoot and undershoot problems)
by:
• Dedicating one specific pin for an optimal programming of the Over−Voltage Protection level
• Drastically speeding−up the regulation loop when the output voltage is 4.5 % below its desired level (dynamic
response enhancer).
As a matter of fact, the downstream converter can be optimally designed since the PFC stage provides it with a very narrow
voltage range.
Finally, the NCP1631 protections (maximum current limitation, in−rush current detection, under−voltage protection,
brown−out detection...) protect the system from most possible over−stresses and make the PFC stage extremely robust and
reliable.
Main Merits of Interleaved PFC
The simplified schematic of Figure 6−2 shows the two paralleled channels. The first branch absorbs a current Ibranch1 and
provides the output with ID1. The other branch draws Ibranch2 and generates ID2. These currents are similar to those exhibited
by a classical one−phase FCCrM PFC stage. However, we will see that as a result of the out−of−phase operation, the total
current absorbed by the interleaved PFC (Iin(total) of Figure 6−2) and the total current provided by the boost diodes (ID(total)
of Figure 6−2) have a significantly reduced ripple compared to that obtained with conventional one−phase PFC stages.
AC Line
Vin
Iin(total)
Ibranch1
EMI
Filter
ID1
Vout
DRV1
Ibranch2
ID2
DRV1
ID(total)
NCP1631
LOAD
DRV2
Rsense
Figure 6−2. Simplified Schematic of an Interleaved PFC Stage
1. The input current ripple is minimized:
I in(total)
I branch2
I branch1
time
Figure 6−3. The Total Current Exhibits a Reduced Ripple
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PFC Handbook
The inductor current within each branch exhibits a large ripple (CrM operation) but as illustrated by Figure 6−3 and detailed
in [2], out−of−phase operation results in a very small ripple on the total current drawn by the PFC stage. In fact, there is no
ripple at all when the input voltage is half of the output voltage (the up and down−slopes being the same) and goes to a maximum
value (100 % of the averaged current − peak to peak) when Vin is near zero or close to Vout (see Figure 6−4).
More specifically, assuming a permanent CrM operation (no frequency clamp − (1)) and a 180− phase shift, one can compute
the input current ripple as follows:
DIin ( pk − pk )
Iin
DIin ( pk − pk )
Iin
⎛ Vin ⎞
⎢⎢ 2 V ⎟⎟ − 1
out ⎠
=⎝
Vin
Vout
if
Vin
> 0.5
Vout
⎛ V
1 − ⎢⎢ 2 in
⎝ Vout
=
V
1 − in
Vout
if
Vin
< 0.5
Vout
⎞
⎟⎟
⎠
⎛ Vin
⎢⎢ V
The following figure summarizes the ripple variations as a function of ⎝ out
⎞
⎟⎟
⎠.
120
Pk to pk ripple (%)
100
80
60
40
20
0
0
0.25
0.5
0.75
1
Vin/Vo ut
Figure 6−4. Peak−to−Peak Ripple of the Input Current as a Function of the Line Magnitude
As a matter of fact, the input current looks similar to that of a CCM PFC in the sense that the current ripple is small.
2. The ripple of the current that feeds the bulk and the load is minimized:
As shown by Figure 6-5, still assuming a CrM operation, the 180° phase shift reduces the ac content of the total current delivered
by the 2 branches to the bulk capacitor and the load (ID(total) of Figure 6−2). This is because the two phases exhibit interleaved
refueling sequences. Furthermore, if the input voltage is less than half of the output voltage, there is no overlap at all between the
1 - The current ripple increases if the PFC stage operates in DCM as a result of the dead-times. However, the PFC stage is designed
to operate in CrM in the most severe conditions, the frequency being only clamped when the line current is relatively low. Thus,
FCCrM does not alter the benefits in term of input and output current ripple.
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two-branch refueling intervals. This characteristic leads to a dramatically reduced rms current within the bulk capacitor at low
line conditions in wide mains applications, for an easier, cheaper and more robust design.
In the wide mains case (no overlap between the refueling times of the two phases), the maximum rms value of the global
refueling current can be expressed as follows:
ID( rm s ) =
2
16 2 Pout
9 p Vac LL Vout h 2
The rms current is thus 30% lower (divided by Ǹ2) when compared to what is obtained in traditional CrM or FCCrM
operation.
Similarly, the rms current flowing through the bulk capacitor is also diminished. Assuming a constant load current, we can
obtain a good approximation of this current using the following equation:
ICout ( rm s ) =
2
16 2 Pout
9 p Vac LL Vout h
2
−(
Pout 2
)
Vout
Whereas, in a traditional CrM or FCCrM PFC, we would have:
ICout ( rm s ) =
2
32 2 Pout
9 p Vac LL Vout h 2
−(
Pout 2
)
Vout
ID(total)
time
ID1
time
ID2
time
Figure 6−5. Reduction in the AC content of the Refueling Current as a Result of Interleaving
Design steps with the 300 W example
A 300−W interleaved PFC consists of two 150−W parallel CrM (Chapter 3) or FCCrM (Chapter 4) stages. Hence, the main
task lies in their dimensioning. As the main steps in designing a CrM or FCCrM PFC boost have already been covered in this
handbook, this chapter will mainly focus on the aspects that are specific to interleaving.
Step 1: Define the key specifications
Minimum Input voltage (VacLL): 88 Vac (this is usually 10−12% below the minimum typical voltage which could be 100 Vac
in many countries).
Maximum Input voltage (VacHL): 264 Vac (this is usually 10% above the maximum typical voltage which could be 240 Vac
in many countries)
Line frequency (fLINE): 50 Hz /60 Hz (This is often specified in a range of 47−63 Hz and for calculations such as hold−up time,
one has to factor in the lowest value specified)
Output Voltage (Vout): 390 V (This value has to be above 1.414 S VacHL and is typically between 385 and 400 V for universal
input operation)
Maximum output voltage (Vout(max)) : 415 V (This value is usually 7−10% above the Vout value and is determined by the
accuracy of the OVP level of the PFC controller).
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PFC Handbook
PFC maximum output power (Pout): 300 W (This is the specified output power for the PFC stage. It is important to factor in
the follow−on stage efficiency when specifying this parameter − it will always be higher than the specified maximum system
output power)
Clamp frequency (fsw(min)): 130 kHz per phase leading to a 260−kHz clamp frequency for the interleaved PFC. The clamp
frequency is key when dimensioning the boost inductors that must be large enough to force CrM operation in the most severe
conditions (full load). The higher fsw(min), the lower the inductance can be.
Output voltage ripple (Vripple(p−p)): <27 V (This parameter is often specified in percentage of output voltage, ±3.5% maximum
in our case)
Estimated efficiency (h): 94% (This parameter is an initial estimate that is used to size the power stage components − high level
of accuracy is not needed for the design procedure).
Step 2: Design the Boost Inductor (for each branch)
As already seen in the CrM and FCCrM sections, the (maximum) peak and rms inductor currents within one branch are:
Icoil, pk (ma x)
⎛P ⎞
2 2 ⎢ out ⎟
⎝ 2 ⎠ = 2 2 150 = 5.13 A
=
0.94 88
h Vac LL
And:
Icoil,rm s (ma x) =
Icoil, pk (ma x)
6
5.13
^
^2.09 A
6
As discussed in chapter 4, the boost inductor design is given by the equation below (where fsw(min) is the clamp frequency):
V
390
h Vac LL 2 ( out − Vac LL ) 0.94 88 2 (
− 88)
2
2
Lw
=
^127 μH
⎛P ⎞
2 390 150 130 k
2 Vout ⎢ out ⎟ fs w (min)
⎝ 2 ⎠
Finally, a 150−μH / 6−A pk / 2.5−A rms inductor was selected.
Step 3: Design the Power Stage Components
The power stage components are designed based on their current and voltage ratings. The inductor design is already covered
in step 2. The bridge diode should be selected based on peak inductor current rating and will have the power dissipation given
by:
Pbridge =
P
Vf
V
4 2
300
^6.5 Vf
out ^1.8 f p
h
Vac LL
88 0.94
Assuming a forward voltage of 1 V for each diode (Vf + 1 V), the bridge approximately dissipates 6.5 W.
For each branch, the MOSFET is selected based on peak voltage stress (Vout(max)+margin) and rms current stress:
IM ( rm s ) =
⎛P ⎞
2 ⎢ out ⎟
⎝ 2 ⎠ 1 − 8 2 Vac LL =
3 p Vout
3 h Vac LL
2 150
3 0.94 88
1−
8 2 88
^1.79 A
3 p 390
Using a 550−V, 250−mΩ FET, per branch, each MOSFET will give conduction losses of (assuming that RDS(on) increases by
80% due to temperature effects):
Pcond = IM ( rm s )2 R DS ( on ) = 1.79 2 0.25 180% ^1.44 W
One Infineon IPP50R250 MOSFET per branch was used in this design.
Two IPP50R250 placed in parallel would cause the same conduction losses in a conventional CrM or FCCrM PFC stage.
Switching losses are difficult to predict accurately. They are not computed here.
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One can simply note that whether a traditional or an interleaved PFC is used, the switching event occurs under the same voltage
stress. The only difference is in the current magnitude that is half in one branch of the interleaved PFC compared to that of the
conventional PFC. Thus, assuming that the switching times are the same in both cases and remembering that the interleaved
PFC consists of two branches:
• If the same MOSFETs are used (one in a traditional PFC, 1 per branch in an interleaved one), the global
switching losses would be the same in the two solutions
• In practice, they should be lower in interleaved PFC because of the use of a smaller MOSFET per phase
leading to less parasitic capacitance and higher transition speed.
Interleaved PFC requires two boost diodes (one per branch). As explained in the FCCrM chapter, there are no reverse recovery
issues to worry about. Simply, they must meet the correct voltage rating (Vout(max)+margin) and exhibit a low forward voltage
drop. Supposing a perfect current sharing, the average diode current is the half of the load current
Id ( total )
⎛ Id ( total ) Vf
⎛
⎞
P
= out ^0.38 A ⎟⎟
⎢⎢
⎢⎢ Id 1 = Id 2 =
V
2
2
2
out
⎝
⎠ . So, the losses are just ⎝
⎞
⎟⎟
⎠ per diode. For each phase, the peak current seen
by the diode will be the same as the corresponding inductor peak current.
Step 4: Output capacitor design
The output capacitor is designed considering 3 factors: output voltage ripple, output current ripple and the hold−up time.
The output voltage ripple is given by:
Vripple ( p − p ) =
Pout 2p fline C out Vout
The capacitor rms current is given by (assuming a resistive load):
ICout ( rm s ) =
2
16 2 Pout
9 p Vac LL Vout
⎛P
− ⎢⎢ out
2
h
⎝ Vout
⎞
⎟⎟
⎠
2
However, almost always, the value of the capacitor is determined by the hold−up time which is approximated by:
t hold −up =
2
2
− Vmin
C out (Vout
)
2 Pout
The hold−time being not considered here, a 100−mF capacitor was chosen to satisfy the other above conditions. The peak−peak
ripple peaks to 24 Vpp at full load (50−Hz line) (±3% of Vout) and the rms current is 1.34 A.
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PFC Handbook
Step 5: Current Sense Circuit
AC Line
EMI
Filter
Vaux2
Vin
L2
Vaux1
Cin
D1
Current Mirror
ICS
L1
OCP
M2
IOCP = 210 mA
ICS
(ICS is
proportional
to the coil
current)
CS
9
DRV2
Cbulk
ICS
In−rush
DRV1
QZCD1
QZCD2
IZCD = 21 mA
Negative Clamp
ICS
M1
Load
ICS
Vout
D2
Iin
(from ZCD block)
ROCP
Iin
RCS
Figure 6−6. Current Sense Block
The NCP1631 is designed to monitor a negative voltage proportional to the inductor current. Practically, a current sense resistor
(RCS of Figure 6−6) is inserted in the return path to generate a negative voltage proportional to the total current absorbed by
the two branches. The circuit incorporates an operational amplifier that sources the current necessary to maintain the CS pin
voltage null (refer to Figure 6−6). By inserting a resistor ROCP between the CS pin and RCS , we adjust the pin9 current as
follows:
− (R CS Iin ) + (R OCP ICS ) = V pin 9 ^0
Where Iin is the total current drawn by the two phases of the interleaved PFC stage.
Finally the current ICS absorbed by pin9 is proportional to the inductor current as shown by the following equation:
ICS = Ipin 9 =
R CS
Iin
R OCP
The circuit compares ICS to an internal 210−μA current reference for a cycle by cycle current limitation. Hence, the
Over−Current protection trips when:
Iin ,ma x =
R OCP
210 μA
R CS
Finally, the ratio (ROCP / RCS ) sets the over−current limit in accordance with the following equation:
I
R OCP
= in ,ma x
R CS
210 μA
As we have two external components to set the current limit (ROCP and RCS ), the current sense resistor can be optimized to
have the best trade−off between losses and noise immunity. Following procedure is used to arrive at the trade−off:
Calculate the maximum current drawn by the two branches:
As shown in [2], the following equations give the total current that is absorbed by the interleaved PFC
Iin ,ma x = 2
⎛
Pin ,avg )
(
⎢
ma x
2
1−
Iin ,ma x = 2 2 (Vin,rm s )LL
⎢
⎢
⎝
(Pin,avg )ma x ⎛⎢1 −
(Vin,rm s )LL
⎢
⎝
Vout ,nom
(
4 Vout ,nom −
Vout ,nom
(
4 2 Vin ,rm s
( 2 (V
)LL
)
in ,rm s LL
⎞
⎟
⎟
⎟
⎠
))
⎞
⎟
⎟
⎠
(
if Vin ,rm s
(
if Vin ,rm s
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V
)LL v out2 ,nom
2
V
)LL w out2 ,nom
2
ON Semiconductor
Where:
• (Vin,rms )LL is the lowest level of the line rms voltage.
• (Pin,avg )max is the maximum level of the input power.
• Vout,nom is the nominal level of the output voltage (or the output regulation voltage)
⎛
⎢ Vin ,rm s
In our case, ⎝
(
)LL = 88
V
⎞
390
v out ,nom =
^138 ⎟
2 2
2 2
⎠
.
Hence:
⎛
Pin ,avg )
(
⎢
ma x
1−
2
Iin ,ma x = 2
Assuming
⎢
⎢
⎝
(Vin,rm s )LL
300
^320 W
(Pin,avg )ma x = 94%
Iin ,ma x = 2 2 320
88
Vout ,nom
(
4 Vout ,nom −
( 2 (V
)
in ,rm s LL
⎞
⎟
⎟
⎟
⎠
))
(assuming a 94% minimum efficiency), it comes
⎛
390
⎢
⎢1 −
− 2 88
4
390
⎢
⎝
(
(
⎞
⎟
⎟ ^6.5 A
⎟
⎠
))
Selecting ROCP and RCS :
If we neglect the input current ripple, the RCS losses are given by the following simplified equation:
pR CS = R CS
⎛ Pin ,avg
⎢
⎢ Vin ,rm s
⎝
⎞
⎟
⎟
⎠
2
One can choose RCS as a function of its relative impact on the PFC stage efficiency at low line and full power. If α is the relative
percentage of the power that can be consumed by RCS , this criterion leads to:
(
a Pin ,avg
(
⎛ P
in ,avg
⎢
⎢⎢ V
⎝ in ,rm s
)ma x = R CS (
)ma x ⎞⎟
2
)min ⎟⎟⎠
Finally:
R CS = a (Vin,rm s )min
2
(Pin,avg )ma x
And:
R OCP = R CS Iin ,ma x
210 μA
Generally (a + 0.2%) gives a good trade−off between losses and noise immunity (0.2% of the power is lost in the RCS at low
line). Lower levels can be set when superior efficiency ratios are targeted. In this case, a special attention is to be paid to the
PCB layout.
This criterion leads to the following RCS value:
R CS = 0.2% 88 2
^50 m W
320
This selection results in the following ROCP resistor:
R OCP = 50 m 6.5 A
^1.54 k W
210 μA
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A 1.8−kΩ resistor is selected for ROCP that gives a 15% margin in the over−current level.
When an over−current event is detected, the NCP1631 decreases the conduction time in both branches as a function of the
excess current for a gradual response in case of a moderate over−current and a sharp reaction in a severe situation. This method
detailed in the data−sheet [3] avoids risks of discontinuity in the operation because of “normal” transients that would trigger
the OCP function.
Application Schematic
U1
KBU6K
D16
1N5406
Vout
+
C18
680nF
C1
100nF
IN
I
in
Vin
-
C12
4.7nF
Type = Y1
CM1
C13
4.7nF
Type = Y1
L4
150μH
C6
1μF
Type = X2
Vaux2
R32
1800k
R42
1800k
R39
1800k
R43
1800k
R38
1800k
R44
1800k
R23x
560k
R123
680k
R121
680k
Vout
R41
1800k
OVP
in
R46
120k
R122
680k
C28
220nF
C27
1nF
R40
27k
R31
1800k
R14
22k
I
R25
27k
C15
220pF
C20
150nF
R36
39k
R34
270k
R33
18k
N
Earth
90−265VAC
pfcOK
16
1
2
15
3
14
4
13
5
12
6
11
7
10
8
9
DRV2
C25
1μF
R37
4.7k
OVP
in
L
X1
R15
22k
R18
820k
C22
1nF
I
C30
100nF
R24
50m (3W )
R7
2.2
D21
15V
R2
1k
Vaux2
L1
D15
1N4148
+
C33
100nF
C32x
100 μF/25V
D5
MUR550
X4
IPP50R250
DRV2
Vcc
I
L2
D4
MUR550
R11
10k
Q1
2N2907
C34
10nF
R1
1.8k
D14
1N4148
X7
15V
−
Q2
2N2907
+
390V
−
R17
2.2
X6
IPP50R250
R20
10k
C32
100 μF/450V
in
The circuit latches off if VCC
exceeds 17.5 V
Figure 6−7. Application Schematic
The presented application is that of the NCP1631 demo−board detailed in [4]. It was populated so that it must be fed by a 15−V
external power source. However, it is possible to add the elements necessary to have it self−powered: start−up resistors and
charge pump.
The boards is equipped with two 150−μH, PQ2620 inductors.
The MOSFETS are IPP50R250 from Infineon.
The boost diodes are axial MUR550 from ON Semiconductor. These Ultra−Fast diodes are specifically designed for CrM or
FCCrM PFC.
The latch off capability is highlighted in schematic. Thermal protection could utilize the NCP1631 latch off capability.
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ON Semiconductor
Bill of Materials
Results and Performance Curves
Evaluation Board Manual [4] details the performance of the presented 300−W interleaved PFC stage.
The main results are however reported here.
Figure 6−8 shows the input current at low and high line. As expected, the input current looks like a CCM one. At high line,
the frequency clamp slightly increases the input current ripple.
The bottom waveforms are magnified views. The drive signals are shown to check that the 180° phase shift is properly
obtained. At low line (left), the branches operate in CrM. At high line they operate in DCM (right) as attested by the oscillating
current near the valley since the frequency clamp is set to 130 kHz at nominal load (at a lower level at light load as dictated
by the frequency foldback function to improve the efficiency). It is worth reminding that the NCP1631 like other FCCrM
circuits, automatically modulates the MOSFET conduction time to still exhibit near unity power factor in DCM. PF and THD
performance are detailed in [4].
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PFC Handbook
Figure 6−8. Typical Waveforms at full load
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ON Semiconductor
98.5
98.0
97.5
Efficiency (%)
97.0
96.5
96.0
95.5
95.0
100V
94.5
115V
94.0
230 V
93.5
93.0
0
10
20
30
40
50
60
70
80
90
100
110
Lo ad (%)
Figure 6−9. Efficiency versus Load
As shown by Figure 6−9, the efficiency characteristic over load is particularly flat.
This is because the interleaved PFC limits the weight of the conduction losses at full load by sharing the stress between two
branches. At light−load, the frequency clamp and the frequency fold−back functions mitigate the switching losses and hence,
effectively limit the efficiency decay usually observed at light load.
These efficiency performances were obtained using the typical frequency fold−back characteristic. When superior light−load
efficiency ratios are required, the frequency fold−back can be tweaked. For instance, application note AND8456/D [5] teaches
how to abruptly drop the switching frequency by adding a simple resistor, when the power goes below a programmed level.
Figure 6−10 shows that such a technique can further improve the efficiency at 10% and 20% of the load by forcing a 20−kHz
operation at these power levels. The PFC stage keeps running in CrM at 50% and 100% of the load as necessary for an efficient
operation.
Figure 6−10. Tweaking the frequency fold−back characteristic can help improve the light−load efficiency
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PFC Handbook
References
[1] L. Huber, B. T. Irving and M. M. Jovanovic, “Open−Loop Control Methods for Interleaved DCM/CCM Boundary
Boost PFC Converters”, IEEE trans. Power Electron., vol. 23, no. 4, pp 1649−1657, July 2008
[2] Joel Turchi, “Characteristics of Interleaved PFC Stages”, Application note AND8355/D, www.onsemi.com
[3] NCP1631 data−sheet, www.onsemi.com
[4] Stéphanie Conseil, “Performance of a 300−W Interleaved PFC Driven by the NCP1631”, Evaluation Board
Manual, NCP1631EVB/D, www.onsemi.com
[5] Joel Turchi, “Further Improve the Low−Power Efficiency of Your NCP1631−Driven Interleaved PFC”, Application
note AND8456/D, www.onsemi.com
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ON Semiconductor
CHAPTER 7
Bridgeless PFC
Introduction
The need for higher efficiencies from the PFC stage has led the circuit designers to look closely at all sections of the circuit
and develop possible lower loss alternatives. One section that contributes significantly to the losses is the input bridge rectifier.
As a result, the alternatives to eliminate the diode bridge or convert it into a dual-use circuit have been explored for many years.
This elimination/conversion of diode bridge brings about its own set of challenges. This chapter provides a more in-depth look
at the bridgeless techniques and works through a design example for an 800-W bridgeless PFC converter.
Why remove the bridge?
D4
AC
Line
D3
EMI
Filter
PFC
Stage
D2
D1
Figure 7−1. The Input Current Flows Through Two Diodes
Figure 7−1 portrays the diode bridge that is usually inserted between the EMI filter and the PFC stage. This bridge rectifies
the line voltage to feed the PFC stage with a rectified sinusoid input voltage. It is well known that as a result of this structure,
the input current must flow through two diodes before being processed by the PFC boost stage:
♦
For one line half-wave, D1 and D4 conduct (red arrows of Figure 7−1)
♦
For the other one, D2 and D3 convey the current (blue arrows of Figure 7−1)
As a matter of fact, two diodes of the bridge are permanently inserted in the current path. Unfortunately, these components
exhibit a forward voltage that leads to conduction losses.
The mean value of the current seen by the bridge is the line current averaged over one half−line cycle. Hence we can write
the following equation:
ǂIbridgeǃT
line
+ ǂI in(t)ǃ T
line
+
2 Ǹ2
@ I in(rms)
p
(eq. 1)
2
2
The line rms current can be easily expressed as a function of the power and of the line voltage:
I in(rms) +
P out
h @ V in(rms)
(eq. 2)
Where:
♦
Pout is the output power
♦
h is the efficiency
♦
Vin(rms) is the rms line voltage
Since two diodes permanently see the average input current, the bridge consumes a power that can be computed as follows:
2 Ǹ2 @ P out
P bridge + 2 @ V f @ I bridge T
^ 2 @ Vf @
h @ p @ V in(rms)
line
ǂ
ǃ
(eq. 3)
2
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PFC Handbook
Finally, if we assume a 1-V forward voltage per diode and computing the losses at the usual low line rms voltage (90 V),
it comes:
P bridge ^ 2 @ 1 @
2 Ǹ2 @ P out
h @ p @ 90
^ 2% @
P out
(eq. 4)
h
In other words, an input bridge consumes about 2% of the input power at low line of a wide mains application. Hence, if one
of the series diodes could be suppressed, 1% of the input power could be saved and the efficiency could for instance, rise from
94% to 95%. Also, the major hot spot produced by a traditional diode bridge would be eliminated with the benefit of an
improved reliability of the application.
“Basic” Bridgeless Architecture
Switching Cell When
PH2 is High Q1 is Off
PH1
D1
L
D2
AC
Line
PH2
Q1
Q2
+
Switching Cell When
PH1 is High Q2 is Off
Figure 7−2. “Traditional” Bridgeless Solution
Figure 7−2 portrays a classical option for bridgeless PFC. There are two switching cells. Each of them consists of a power
MOSFET and of a diode:
♦ The first cell (Q1, D1) processes the power for the half-line cycle when the terminal “PH1” of the line is high and is
in idle mode for the rest of the line period.
♦ The second cell (Q2, D2) is active for the other half-wave when “PH1” is low compared to terminal “PH2”.
The line and the PFC inductor are placed in series and the arrangement they form is connected to the switching nodes of the
two switching cells. The input current is processed by the switching cell that is active for the considered line half-wave. The
MOSFET of the inactive cell has a role anyway, since its body diode serves as the current return path. Compared to a
conventional PFC stage, the losses due to the bridge are saved but the body diode of the inactive MOSFET conducts the coil
current. Finally, this structure eliminates the voltage drop of one diode in the line-current path for an improved efficiency.
However, the above architecture presents the following inconveniences, resulting from the fact that the line is floating with
respect to the PFC stage ground (as opposed to the conventional PFC, where the line is connected to PFC ground):
♦ Certain PFC controllers need to sense the input voltage. In this structure, a simple circuitry cannot do the job.
♦ Similarly, the coil current cannot be easily monitored.
Besides these difficulties in the circuit implementation, EMI filtering is the main issue. When “PH1” is high, the negative
terminal “PH2” is attached to ground by the Q2 body diode. Hence, the application ground is connected to ac line as it happens
in a conventional PFC. Now, when “PH2” is high, the MOSFET Q2 switches and the voltage between the line terminals and
the application ground pulses as well. More specifically, the potential of the “PH2” node nearly oscillates between 0 (when
the MOSFET is on) and the PFC output voltage (when the MOSFET is off). This large “dV/dt” leads to an increased
common-mode noise that is difficult to filter. This is probably the major drawback of the above solution ([1] and [2]).
Two-Phase Approach
Figure 7−3 portrays another option for bridgeless PFC. This solution was proposed by Professors Alexandre Ferrari de Souza
and Ivo Barbi [3]. As shown in Figure 7−3, there is no full bridge. Instead, the ground of the PFC circuit is linked to the line
by diodes D1 and D2 and each terminal feeds a PFC stage. Hence, the solution could be viewed as 2-phase PFC where the two
branches operate in parallel:
♦ For the half-wave when the terminal “PH1” of the line is high, diode D1 is off and D2 connects the PFC ground to
the negative line terminal (“PH2”). D2 grounds the input of the “PH2 PFC stage” branch that thus, is inactive and the
“PH1 PFC stage” processes the power.
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ON Semiconductor
♦
For the second half-line cycle (when “PH2” is high), the “PH2 PFC stage” branch is operating and “PH1 PFC stage”
that has no input voltage, is inactive.
PH1 PFC
Stage
PH1
AC
Line
PH2 PFC
Stage
PH2
DRV
D2
+
D1
Figure 7−3. Two−Phase Architecture
Figure 7−4 gives an equivalent schematic for the two half-waves. Similarly to the traditional bridgeless structure presented
in the previous paragraph, this architecture eliminates one diode in the current path and hence improves the efficiency.
One other interesting characteristic of this structure is that the PFC stage that is active, behaves as a conventional PFC boost
would do:
♦ When the “PH1” terminal is positive (see Figure 7−4a), diode D1 opens and D2 offers the return path. The input
voltage for the “PH1” PFC stage is a rectified sinusoid referenced to ground.
♦
For the other half-wave (see Figure 7−4b), when “PH2” is the positive terminal, D1 offers the return path. Diode D2
is off and sees a rectified sinusoid that inputs the “PH2” PFC stage. Again, we have a conventional PFC where the
input voltage and the boost are traditionally referenced to ground.
PH1 PFC Stage
+
−
PH1
PH2 PFC Stage
Ac Line
AC Line
PH2
−
DRV
PH1
PH2
+
+
D1
D2
DRV
+
b) Terminal PH2 is the High One
a) Terminal PH1 is the High One
Figure 7−4. Equivalent Schematic for the Two Half−Waves
It is also worth noting that the two-phase structure does not require any specific control scheme to switch between the phases.
The MOSFETs of the two branches are referenced to ground and they can be permanently driven even when their phase is in
idle mode. The MOSFET of the inactive branch would then be turned on and off with no function, but:
♦ At the benefit of a simplified circuitry since there is no need for detecting the active phase and for directing the drive
signal to the right MOSFET according to the half-line cycle.
♦
At the price of the additional losses due to the inactive MOSFET drive. The loss is not very high anyway since the
voltage across the MOSFET is null when its input voltage is zero. Hence, the gate charge to be provided is
approximately halved compared to that of the active MOSFET.
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PH1
Ac Line
0V
PH2
D2
D1
DRV
+
Rsense
Figure 7−5. Operation for the ”PH2” Half−Wave
One should however note that the current does not necessary return by the D1 and D2 diodes only. Figure 7−5 portrays (in
blue and red) the “expected” current path when “PH2” is high (the same analysis could have been done for “PH1” high):
♦ The blue path is supposed to be the current path when the MOSFET is on
♦
The red one, that of the current when the MOSFET is open.
Actually, a large portion of the current flows as indicated in black irrespective of the MOSFET state. This is because the body
diode of the supposedly inactive MOSFET provides the current with another path. Since the coil presents low impedance at
the line frequency, we have two diodes in parallel and the current is shared between them in inverse proportion of actual
impedance of the paths.
Phase 1 current (5 A/div)
5 ms/div
0A
0A
Part of the active
phase current flows
through the inactive
MOSFET and coil!
Phase 2 current (5 A/div)
Figure 7−6. Part of the Current Flows through the Supposedly Inactive MOSFET and Coil
Figure 7-6 portrays the input current for each branch. One can see on this plot that a negative current takes place through the
body diode during the “inactive” half­wave. The main inconvenience of this behavior is that the input current cannot be sensed
by inserting a RSENSE resistor in the supposed return path (as shown by Figure 7-5) since part of the current takes another route.
A better alternative is to insert current sense transformers as detailed in the “step 6” part of the next section.
Design Procedure with an 800-W Example
The step-by-step procedure of an 800-W bridgeless PFC is described in this section with reference to Figure 7-7, which
highlights the main parts of the design.
Step 1: Key Specifications
Minimum Input voltage (VacLL): 88 Vac
Maximum Input voltage (VacHL): 264 Vac
Line frequency (fLINE): 50 Hz / 60 Hz (47-63 Hz)
Output Voltage (Vout): 385 V
Maximum output voltage (Vout(max)) : 420 V
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PFC maximum output power (Pout): 800 W
Switching frequency (fsw): 100 kHz
Output voltage ripple (Vripple(p-p)): 15 V
Hold-up time (thold-up): 16 ms (330 V being the permissible minimal level at the end of the hold-up time)
Estimated efficiency (h): 95%.
Step 2: Diode Bridge
While this is claimed as a bridgeless circuit, a diode bridge is implemented! However, it does not serve as a traditional input
bridge to rectify the line (as shown by Figure 7−7, the two branches are directly connected to the line terminals). Here, the upper
diodes (D3 and D4) simply provide path for the in-rush current that takes place when the PFC stage is plugged in (Note 1.).
Unless an overload situation occurs, these diodes are off for the rest of time. The bottom diodes (D1 and D2) have the function
that was described in the preceding section. Each of these diodes is a low-speed diode, but has a high peak current rating (given
by the coil peak current or in-rush current) and 600 V voltage rating. GSIB1580 from Vishay (15-A, 800-V) diodes were
chosen.
Diodes Bridge
D4
In−rush current path
D3
Branch 1
PH1
Current sense
transformer
AC Line
+
Bulk
Capacitor
EMI
Filter
Branch 2
PH2
D2
D1
FB
8
2
7
4
NCP1653
1
3
RETURN
VCC
Current sense
transformer
DS
1N5406
6
5
In−rush
current
detection
(optional)
RSENSE
RETURN
Input voltage
sensing
CS
One control circuitry
Figure 7−7. Simplified Application Schematic of Bridgeless Converter
Step 3: Power components for each PFC branch
Each PFC branch (branch 1 and branch 2) is designed to handle full power for half the line cycle. Hence, the design of
components (power FET, boost inductor and boost diode) follows the methodology described in Chapter 5 for CCM operation.
However, the fact that each branch is active for one half-line cycle only, improves the heating distribution. Also, the rms current
being halved in each branch, the power components does not need to be as large as those of a conventional PFC. Resultant
choices for each branch are:
♦ Boost diodes (1 per branch): CSD10060 (10-A, 600-V SiC diode from CREE)
♦ Power MOSFETs (2 per branch): SPP20N60 from Infineon (20-A, 600-V, 0.19-W)
♦ Inductors (1 per branch): 200-mH / 9.7 Arms / 16-Apk / 5 App coil (ferrite core)
Part Name:
PFC−Choke LDU80025
Part Number: 203860 (J−Lasslop)
1.. As in a conventional PFC boost, there is no switch between the line and the bulk capacitor able to prevent the line from directly
charge the bulk capacitor. That is why an in−rush takes place that charges the bulk capacitor to the line peak voltage. This in−
rush current can be huge if not limited by some dedicated circuitry.
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Step 4: Bulk capacitor selection
The PFC bulk capacitor is common to both branches as shown in Figure 7−7. Its selection follows the common methodology
based on the 3 factors: Output voltage ripple, output current ripple and hold-up time. These three factors have to be computed
as it would be computed in a conventional CCM PFC. Two 330­mF, 450­V capacitors are used considering these factors.
Step 5: Input voltage sensing:
The input voltage needs to be sensed by the PFC controller for feed­forward (NCP1653, NCP1654) and for brown-out
protection (NCP1654). As shown by Figure 7−7, two diodes are used that re-construct the rectified line voltage from the two
branches that can then be monitored by the circuit. Our prototype is driven by the NCP1653. As detailed in [4], the input voltage
sensing network is to be designed so that it feeds pin3 with a 15-mA dc current at low line. If the NCP1654 is used, the ripple
of the voltage applied to the brown-out pin must be adjusted to program the desired line brown-out levels as shown in [5].
Step 6: Current Sense Circuit and Inrush detection
As described in the previous section, a simple current sense resistor cannot sense the MOSFET current. Thus, the MOSFETs
current has to be monitored using current sense transformers. Here, two current sense transformers are used (one in series with
each FET branch). The chosen current sense transformer is rated for 20 A and has 50:1 turns ratio (WCM 601-2). The current
delivered by the boost diodes must also be sensed to provide the controller with a signal representative of the total current
flowing through the coils. This can be done with the help of a third current sense transformer.
In our designs, however, another option is chosen in order to detect inrush current. A current sense resistor is inserted in the
bulk capacitor return path that monitors the current that re­fuels the bulk capacitor, i.e.:
♦
in-rush currents during the start-up phase or in over-load situations
♦
the current provided by the boost diodes in normal operation
Due to that, the MOSFET and the input bridge are referenced to the “RETURN” potential instead of ground. The voltage
between the RETURN and ground potentials is the negative voltage developed across the RSENSE resistor. If this voltage
becomes too large (during in-rush sequences for instance), the MOSFETs’ source potential may dramatically drop and some
accidental MOSFET turn on may follow. That is why the voltage across the RSENSE resistor is limited by a diode.
This diode must be able to sustain the in-rush current and its forward voltage must be high enough so that the RSENSE voltage
is not clamped until the current largely exceeds its permissible level in normal operation. Otherwise, the clamping diode would
prevent the RSENSE voltage from becoming high enough to trigger the over-current protection.
The secondary of the two current sense transformers and the RSENSE voltage are combined to provide the circuit with a
current proportional to the input current for over-current limitation and duty-cycle control.
Figure 7−8 summarizes the current sensing circuitry:
PH1
1N4148
Ns/Np=50
15 k
3R3
D1
DRV
RETURN
RETURN
PH2
1N4148
Ns/Np=50
15 k
3R3
D2
DRV
RETURN
33m • I DIODES
3.3
2
S IMOSFET = 33m S IMOSFET
Ns
Np
33m
Figure 7−8. Details of the Current Sense Circuits
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90
2k2
To Current
Sense Pin
2k2
ON Semiconductor
In our configuration, the current sense transformers and the current sense resistor provide a voltage that in both cases is linked
to the sensed current by the same coefficient (33 S 10−3). That is why R4 and R24 of the detailed circuit schematics in Figure
7−10 on page 92 have the same value. R4 and R24 adjust the current sourced by the circuit to maintain the current sense pin
to zero. This current is proportional to the sensed current. If the current sense transformer and the current sense resistor had
different coefficients, R4 and R24 would have to be adapted so that the same current leads to the same current sunk from the
controller current sense pin, this absorbed current having to reach 200 mA when the maximum input current is exceeded.
Step 7: Control circuitry
As already mentioned, the two-phase bridgeless PFC does not require any complex control circuitry. A single NCP1653 PFC
controller directly drives the two branches. The NCP1653 is a compact 8-pin PFC controller that operates in continuous
conduction mode. As it directly adjusts the conduction time as a function of the coil current, there is no inner current loop to
be compensated for an eased design. Alternatively, NCP1654, a derivative of NCP1653 with additional features such as
brown-out protection and better line regulation can be used (see [4] and [5]).
Results and Performance Curves
Figure 7−9 shows the annotated picture of the 800-W prototype.
Figure 7−9. Photograph of the Evaluation Board
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91
92
http://onsemi.com
Figure 7−10. Application Schematic
F1
10 A
Earth
90 to 265 Vrms
50 or 60 Hz line voltage
N
R21x
680k
R22x
680k
RETURN
CS
R24
2. 2k
C1
100 nF /
63 V
C17
1mF
R4
2. 2k
R3
470k
R2
2700k
R1
2200k
D1
1N 4007
28
R6
100k
4
3
2
VCC
Vm
GND
Drv
5
6
7
N C P1653
CS
In
Vctrl
FB
−
+
R7
56k
4
3
2
1
NC
5
6
R20x
10k
R17
DRV1 10
VCC 2
DRV2
PH 2
R19
10k
R18
10
R10
100m / 3W
R9
100m / 3W
R8
100m / 3W
M C 33152
In B Ou tB
GND VCC
7
8
DRV1
In A Ou tA
NC
C28
220nF
VCC 2
C27
100pF
C6
22mF
R11
180k
C24
220nF
C4
1nF
R13
100
C25
22mF
R15
10
R12
680k
RETURN
VCC
R13x
680k
X5
NCP1653
1
8
R14
390k
D5
1N 5406
C5
10nF
R5
D3
390
1N 5817
C2
100nF
C3
220nF
D2x
1N 4007
PH 2
IN
R29
15k
D4
R23
10k
CS
R28
3
C12
330 m/450 V
R21
10k
C11
330 m/450 V +
R20
10
R28
3
CS
D8
CSD10060
R22
10
1N4148
DRV2
X1
SPP20N60
RETURN
L4
0. 2m
RETURN
DRV1
X2
SPP20N60
D2
X3
SPP20N60
LN
C13
1 mF
Type = X2
R23x
680k
C14
1 mF
Type = X2
C18
4. 7 nF
Type = Y2
C16
1mF
PH 1
R25
15k
1N4148
D7
CSD10060
X7
SPP20N60
CM 1
CM 2
C19
4. 7 nF
Type = Y2
C15
1mF
Type = X2
X4
D iode bridge
PH 1
L1
0. 2m
+
Vout
PFC Handbook
Circuit Schematics
ON Semiconductor
The performance of the board is captured in following waveforms and graphs. Figure 7−11 shows the typical waveforms
(line current, output voltage, current sense signal and rectified input voltage) for one branch at two line voltage conditions.
90 Vrms
Iline (10 A/div)
230 Vrms
Iline (10 A/div)
Vout
Vout
CS (negative sensing)
CS (negative sensing)
Vin,1 (input voltage for branch 1)
Vin,1
Figure 7−11. Typical Waveforms at Low Line (Left) and High Line (Right)
Plots of Figure 7−11 portray typical waveforms at full load (Iout = 2.1 A). “CS” is the negative voltage provided by the current
sense transformers. It is representative of the current flowing into the MOSFETs of the two branches (“CS” is the common
output of the two current sense transformers). As expected, the input voltage of the “PH1 PFC stage” (“Vin,1”) is a rectified
sinusoid for one half-line cycle and null for the other one. The line current is properly shaped.
Figure 7−12 provides a magnified view at the top of the line sinusoid. The switching frequency is 100 kHz. The signal
“Vsense” (identical to “CS”) is a negative representation of the MOSFET current. The current sense transformers are wired so
that only the current drawn by the MOSFET drain is monitored (possible current flowing in the opposite direction cannot be
sensed).
The waveforms are similar to those of a traditional CCM PFC.
90 Vrms
rms
Iline (10 A/div)
Iline (10 A/div)
230 Vrms
rms
Vout
Vout
Vin,1
Vsense (negative sensing)
Vsense (negative sensing)
Vin,1 (input voltage for branch 1)
Figure 7−12. Magnified Views of Figure 7−11 Plots
Thermal measurements
The following results were obtained using a thermal camera, after a 1/2 hour operation. The board was operating at a 25°C
ambient temperature, without fan. These data are indicative.
For the bridge, the MOSFETs and diodes, the measurements were actually made on the heat-sink as near as possible to the
components of interest.
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PFC Handbook
Measurement Conditions:
Vin(rms) = 88 V
Pin(avg)) = 814 W
Vout = 381 V
Iout = 2 A
PF = 0.995
THD = 9 %
Devices
Bridge MOSFET1 Diode1
Temperature (5C)
85
95
Coil1
MOSFET2
Diode2
Coil2
Bulk Capacitor
CM EMI coil
47
86
80
48
40
45
77
Efficiency and Total Harmonic Distortion
100
20
230 Vrms
15
90 Vrms
120 Vrms
96
THD (%)
EFFICIENCY (%)
98
90 Vrms
94
10
230 Vrms
5
92
90
100
200
20% Pmax
300
400
500
600
700
OUTPUT POWER (W)
0
100
800
Pmax
120 Vrms
200
20% Pmax
Figure 7−13. Efficiency Performance
300
400
500
600
OUTPUT POWER (W)
700
800
900
Pmax
Figure 7−14. Total Harmonic Distortion Over the
Load Range
Figure 7−13 portrays the efficiency over the line range, from 20% to 100% of the load.
The efficiency was measured under following conditions:
♦ The measurements were made after the board was operated at full load, low line for 30 minutes
♦
All the measurements were made consecutively without interruptions
♦
PF, THD, Iin(rms) were measured by a power meter PM1200
♦
Vin(rms) was measured directly at the input of the board by a HP 34401A multi­meter
♦
Vout was measured by a HP 34401A multi­meter
♦
The input power was computed according to: Pin(avg) + Vin(rms) @ Iin(rms) @ PF
♦
Open frame, ambient temperature, no fan
Reviewing Figure 7−13, it can be noted that:
♦ Like in a conventional PFC, the efficiency is higher at high line.
♦
At low line (90 Vrms), full load, the efficiency is in the range of 94% without a fan. When measured @ 100 Vrms
input, full load efficiency of 95% was recorded.
♦
The light load efficiency is very good. For instance, at 20% of full load, efficiency is in the range of or higher than
96%. One of the reasons for this lies in the fact that a bridgeless PFC requires relatively low Qg MOSFETs
compared to a conventional PFC for the same efficiency target at full load.
Figure 7−14 portrays the THD at 90, 120 and 230 Vrms over the load. One can note that the total harmonic distortion remains
very low even in high line, light load (<15%) where the line current is small and more sensitive to all the sources of distortion
like the system inaccuracies and mainly the EMI filter.
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ON Semiconductor
Conclusion
A bridgeless PFC based on the two-phase architecture has several merits among which one can list the ease of control or the
absence of high frequency noise injected to the line (eased EMI). In this chapter, the basics of the bridgeless architecture along
with a summary design procedure are covered. The designed prototype has been tested at full load (800-W output) without a
fan (open frame, ambient temperature). In these conditions, the full-load efficiency was measured in the range of 94% at 90
Vrms and as high as 95% at 100 Vrms. The THD remains very low. A NCP1653 or NCP1654-driven two-phase bridgeless PFC
is a solution of choice for very efficient, high power applications.
It should be noted that when traditional topologies (CCM/CrM/FCCrM boost) are scaled up to high power levels such as
800-W, they encounter several design challenges related to component size and dissipation. Hence, topologies such as
two-phase bridgeless or interleaved, which spread the heat dissipation and offer other benefits, make a lot of sense. Further
information on bridgeless PFC can be found in AND8392 [7] from ON Semiconductor.
Please note that a 300−W bridgeless PFC has been recently experimented. An application note is available on the ON
Semiconductor website that details this application [9].
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PFC Handbook
References
1. Laszlo Huber, Yungtaek Jang and Milan M. Jovanovic, “Performance Evaluation of Bridgeless PFC Boost
Rectifiers”, APEC 2007
2. Pengju Kong, Shuo Wang, and Fred C. Lee, “Common Mode EMI Noise Suppression for Bridgeless PFC
Converters”
3. Alexandre Ferrari de Souza and Ivo Barbi, “High Power Factor Rectifier with Reduced Conduction and
Commutation Losses”, Intelec, 1999
4. Joel Turchi, “Four Key Steps to Design a Continuous Conduction Mode PFC Stage Using the NCP1653”,
AND81842/D, ON Semiconductor, http://www.onsemi.com/pub/Collateral/AND8184­D.PDF
5. Patrick Wang, “Four Key Steps to Design a Continuous Conduction Mode PFC Stage Using the NCP1654”,
AND8322/D, ON Semiconductor, http://www.onsemi.com/pub/Collateral/AND8322­D.PDF
6. NCP1653 data sheet and application notes, www.onsemi.com
7. NCP1654 data sheet and application notes, www.onsemi.com
8. Joel Turchi, “A 800−W Bridgless PFC Stage”, AND8392/D, ON Semiconductor, http://www.onsemi.com/pub/
Collateral/AND8392−D.PDF
9. Joel Turchi, “A High−Efficiency, 300−W Bridgeless PFC Stage,”AND8481/D, ON Semiconductor,
http://www.onsemi.com/pub/Collateral/AND8481−D.PDF
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CHAPTER 8
Single Stage, Isolated Power Factor Correction
Introduction
Applications that require an isolated, regulated output voltage or current along with input power factor correction typically
involve a two stage conversion process as depicted in Figure 8−1. This scheme is composed of an input boost or bridge−less
power factor corrector stage which converts and pre−regulates the input line into a nominal 400 Vdc bus. This bus then provides
the input voltage for a conventional dc−to−dc converter which can be any appropriate topology. For lower power applications
of 100 W and less, a flyback converter is commonly used.
AC
input
EMI
Filter
Main
Converter
Boost
PFC
DC
output
Figure 8−1. Conventional Two−Stage Conversion
With a few performance compromises, a simpler technique to develop an isolated regulated dc output voltage or current can
be implemented in which the power factor corrector and main converter sections are merged into a single conversion, The
advantage of this approach is reduced parts count and the potential to improve efficiency and power density in lower power
applications. The block diagram is shown in Figure 8−2.
EMI
Filter
AC
input
Main
Converter
DC
output
Figure 8−2. Single Stage Conversion with NCP1652
In this single stage PFC converter, the most useful power circuit is the flyback topology or an equivalent buck−boost
derivation. The flyback stage not only handles the output voltage regulation and input to output isolation functions, but can
provide power factor correction as well. The circuit essentially functions as a conventional boost PFC converter with the output
being derived from an isolated secondary winding on the boost choke rather than using a high voltage diode directly after the
choke as in the conventional high voltage boost mode. The dc input to the converter following the ac bridge rectifier is a
full−wave rectified sine wave, operating at twice the line frequency (100 or 120 Hz) instead of a pure dc voltage. The normal
input “bulk” capacitor following the bridge rectifier must be reduced to a value of 1 μF or less so that the capacitive input filter
does not have any significant effect on the power factor. The value of this capacitor should be sufficient to provide a low
impedance at the converter’s switching frequency, but small enough to offer very high impedance at the ac line frequency.
Single Stage Converter Characteristics
The single stage, isolated PFC converter can be configured from the conventional buck−boost derived flyback topology. The
operational mode can be in discontinuous conduction mode (DCM), critical conduction mode (CrM), or continuous conduction
mode (CCM). The most common operational mode for lower power circuits is CrM because of the overall control simplicity
and ease of implementation of synchronous output rectification for lower voltage outputs (<12 Vdc). The ON Semiconductor
NCL30000 PFC controller is particularly well suited for the lower power applications of 50 W or less which includes LED
lighting applications as well as small power adapters. The CrM operational characteristics will be similar to any conventional
critical conduction mode boost PFC or flyback converter, namely the switching frequency will vary with line and load, and
recovery losses in the output rectifier will be negligible due to the secondary flyback current going to zero prior to re−activation
of the main power switch. This latter characteristic makes it straightforward to implement MOSFET synchronous rectifiers
in designs where low output voltages require minimal conduction losses in the output rectifier. The ON Semiconductor
NCP4302 and NCP4303 are particularly suited for secondary side synchronous rectifier control.
© Semiconductor Components Industries, LLC, 2009
April, 2009 − Rev. 1
97
Publication Order Number:
AND8397/D
PFC Handbook
Continuous conduction mode however, can offer significant advantages for applications that require fixed frequency
operation; this is especially true for output voltages of 15 Vdc or higher where the use of synchronous rectification yields
marginal efficiency improvements. In CCM the peak MOSFET current can be significantly less than in CrM resulting in lower
MOSFET switching losses, particularly at power levels above 75 W. CCM operation also reduces the high frequency output
capacitor ripple current, and the overall conversion efficiency is generally higher. The NCP1652 controller is designed
particularly for CCM operation and also provides a second gate drive output for the implementation of an active clamp snubber
for even higher power applications where voltage spikes caused by the flyback transformer’s leakage inductance energy can
become a significant issue.
The single stage PFC conversion process, regardless of the operational mode, has a few compromises over the traditional
two−stage conversion scheme of Figure 8−1. They are as follows:
1. As with any power factor corrector circuit, the gain bandwidth of the control loop is very low, typically in the
10−30 Hz range. This is necessary, otherwise the control loop would attempt to regulate off the line variations of
the input and this would result in poor power factor. As a consequence of the low bandwidth, transient response to
load step changes will be compromised although dc regulation will be excellent. For adapter applications where
point−of−load regulation (POL) is utilized anyway, the slow transient response is inconsequential.
2. Because the loop cannot regulate away the line ripple, it will appear as a ripple component on the output. Therefore
sufficient output capacitance must be utilized to keep the line ripple component minimized for the specific
application. As mentioned previously, the use of downstream POLs in the system should preclude excessive output
capacitance since the ripple will be attenuated sufficiently by the POL regulators. In constant current LED
applications, the low frequency ripple will even be less of an issue as long as the peak current requirement for the
LED is not violated.
3. Due to the lack of a large input bulk capacitor, the converter has no significant inherent hold−up time other than
that provided by the stored energy in the transformer and the output capacitors.
4. The power factor for the single stage converter tends to degrade with increasing line and decreasing load due to
factors related to the duty cycle [D/(1−D)] transfer function and non−linearity effects in the magnetics, however, for
most typical line and load conditions the PF will be above 0.90.
5. Just like any flyback topology, the effects of leakage inductance in the main power transformer will have a direct
impact on performance, EMI, and overall efficiency. The transformer should be wound with proper wire sizing to
minimize AC losses due to skin effect, and winding layers should be kept to a minimum to avoid proximity effect
and winding self−capacitance. In most cases with off−the−shelf cores, the primary will require 2 layers so it is best
to use the “sandwiched” secondary construction where half the primary is wound first, then the secondary, and then
the second half of the primary over the secondary. The primaries are then connected in series externally. This
construction will help minimize the leakage inductance of the magnetic structure. Under no circumstances should
windings be wound in different layers and then paralleled. This construction will result in high circulating currents.
Windings requiring multiple strands of wire should be wound “multifilar” and constrained to two layers or less.
Moreover, the leakage inductance effects will be more prominent the lower the secondary output voltage due to the
coupling effect of the primary to the secondary turns.
Despite these tradeoffs, the single stage, isolated PFC converter is an efficient and very cost effective solution for offline
LED power supply units, notebook adapters and similar applications needing compact size with minimal parts count. For LED
driver applications, an additional current feedback loop with output current sense resistors can be implemented to provide a
constant current, constant voltage (CCCV) characteristic. The ON Semiconductor NCS1002 dual op amp plus reference is well
suited for such applications.
Design Example using the NCL30000
An implementation of a CrM/DCM single stage PFC converter based on the compact SO−8 packaged NCL30000 is shown
in Figure 8−3. This circuit is optimized for driving high brightness LEDs with constant current control but could be modified
for fixed voltage output by changing feedback to a constant voltage configuration. As shown, this circuit provides up to 15
W at a constant 350 mA which is typical for many high brightness LEDs and is capable of powering 4 to 17 white LEDs from
a source spanning 90 to 305 Vac.
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99
1A
1
Line
1
Neutral
J1-2
F1
47nF
C1
BAW56 D7
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R10
6.2k
2
RT1
R11
100k
4
27mH
1 L1 3
T
4.7k
R14
R2 5K6
L2 2.2mH
R15
100k
RV1
CS
Ct
ZCD
GND
Comp DRV
VCC
NCL30000
C4
100nF
MFP
C9
820 pF
4
3
2
1
C8
10uF
D4
MRA4007
V300LA4
1nF R17
100
C7
L3 2.2mH
R3 5K6
C2
47nF
Q2
MMBTA06
D9
MMBZ5245
Q1
D8
15V
MMBTA06
BZX84C5V1
5.1V
R9 6.2k
Figure 8−3. Schematic of a 15 W High Power Factor Single Stage LED Driver
5
6
7
8
Q3
SPD02N80
R20
0.33 W
R18 100
4
3
T1A
T1E
T1D
C12
470uF
+
+
D10 MURD330
C10 4.7 nF
U2
PS2561L_1
2
T1C
1
5
3
R7
47K T1B
R6
47K
R19 10
R16
47k
D6
BAS21
D5
ES1M
4700 pF
C5
4
R22
1k
BZX84C5V6
1
D11
2
MMBTA06
Q5
MMBTA06
R24
47k
100pF
C14
1k
R25
C15
220nF
U3
8
IN1+
1 VCC IN1−
OUT1
IN2+
OUT2
7
IN2−
GND
LM2904
D12
BZX84C56
R23
1k
D13 BAW56
3
J1-1
3
2
5
6
4
R28
470
R27
200
16k
R26
0.2 W
R29
R30
24k
U4
TL431A
24k
R31
C16
100nF
LED
Cathode
J2-2
1
LED
Anode
J2-1
1
ON Semiconductor
Q4
C13
100nF
PFC Handbook
400
84%
390
83%
380
82%
370
81%
360
80%
350
79%
340
78%
330
77%
LED Current
320
Efficiency
LED Current (mA)
The key to high power factor performance in a CrM single stage converter is maintaining fixed on−time over a half cycle
of input sine wave. The pulse width is determined by average AC input voltage and power delivered to the load. This control
method is explained in detail in the ON Semiconductor NCL30000 datasheet. Switching frequency will vary with input voltage
and output load. The zero current detection feature built in to the NCL30000 controller ensures CrM operation by initiating
a switching cycle only when the current in the flyback transformer is depleted. The primary inductance of the flyback
transformer sets the switching frequency range for a given set of input voltage and output load conditions.
Energy delivered by a single stage converter follows the input sine wave shape starting at zero, reaching a peak, and then
falling to zero every input half cycle. Output ripple amplitude is determined by the output filter capacitor. An open load circuit
protects the output capacitor in the event the load is disconnected by providing feedback to the NCL30000 controller.
The LED driver shown in Figure 8−3 provides tight current regulation and high efficiency driving 12 white LEDs over the
range of 90 to 305 Vac as shown in Figure 8−4. LED current varies less than 1% over the input voltage range. Efficiency is
greater than 81% for 115 − 230 Vac applications at nominal load.
76%
Efficiency
310
75%
300
74%
90
115
140
165
190
215
240
265
290
315
Input Voltage (Vac)
Figure 8−4. Line Regulation and Efficiency Driving 12 LED Load (Vf = 37 Vdc)
Figure 8−5 shows the power factor is maintained above 0.9 and harmonic distortion is less than 13% over the input range
of 90 to 305 Vac with a 12 LED load (Vf = 37 Vdc).
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14
1.00
13
0.99
12
0.98
11
0.97
10
0.96
9
0.95
8
Power Factor
Input Current THD
ON Semiconductor
0.94
THD
Power Factor
7
0.93
6
90
115
140
165
190
215
240
265
290
0.92
315
Input Voltage (Vac)
23
1.00
20
0.99
17
0.98
14
0.97
11
0.96
THD
8
0.95
Power Factor
5
0.94
0
10
20
30
40
50
LED Voltage (V dc)
Figure 8−6. Power Factor and THD vs. Load (115 Vac Input)
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60
Power Factor
Input Current THD
Figure 8−5. THD and Power Factor with 12 LED (Vf = 37 Vdc) Load Over a Wide Input Range
PFC Handbook
Power factor is also a function of loading for a given application. Figure 8−6 shows the effect of load on the LED driver
at 115 Vac input. Note the input current distortion reaches a minimum near 43 V which is about 15 W output. Power factor
reaches a maximum at this same load point. Out of phase line current due to EMI filter capacitors is constant at a given input
line voltage, but is a larger percentage of the total line current at lighter loads and has a greater effect on power factor. Figure
8−6 shows this reduction in power factor at light loads.
IEC 61000−3−2 Class C requirements establish the maximum allowable harmonic content for input current in lighting
applications. Two limit categories are based on input power draw above or below 25 W. This example driver draws less than
25 W input and must meet requirements only at the third and fifth harmonic of input current as a percentage of fundamental
line current draw as shown in Table 8−1 below.
Table 8−1. Line Current Harmonics and IEC 61000−3−2 Limits
Third Harmonic
Fifth Harmonic
Input
(Volts)
Fundamental
(Amperes)
Reading
Limit
Reading
Limit
115
0.13
6.95%
86.0%
0.61%
61.0%
230
0.07
9.91%
86.0%
2.21%
61.0%
Table 8−1 above shows the harmonic content of line current is well below the required levels. Note that requirements for
lighting applications equal to or greater than 25 W must meet more restrictive levels which extend up to the 40th harmonic of
input current fundamental.
The NCL30000 provides a single stage CrM/DCM solution for driving constant current in LEDs. A power factor greater
than 0.9, THD performance of less than 20% across line, high efficiency, and tight regulation over a wide operating range are
demonstrated in a circuit with few external components.
Design Example using the NCP1652A
Figure 8−7 shows the schematic of a 19 V, 5 A single stage PFC converter utilizing the NCP1652A controller. This power
supply design is intended for laptop and notebook adapters requiring up to 100 W output. The flyback transformer was designed
with a high primary inductance (600 μH) to force the circuit to operate in “deep” CCM. This mode reduces the peak MOSFET
current and output capacitor rms ripple current for maximum efficiency. The SO−16 and SO−20 packages provide additional
pins for additional control features including brownout detection, feed−forward compensation, over−current protection, and
a latch input for overvoltage and/or over−temperature protection. The loop gain, multiplier bandwidth and other control
parameters can also be tailored to fit the intended application. Design details for this 19 V adapter supply and a similar 48 V,
2 A output power supply intended for telecom and LED applications can be found in ON Semiconductor application notes
AND8397 and AND8394 respectively at ON Semiconductor’s website.
A version of the NCP1652A intended specifically for LED lighting applications is the NCL30001 [12]. This controller is
identical to the NCP1652A but without the “OUT B” drive output. Applications using this controller in constant voltage,
constant current (CVCC) LED lighting power supplies is shown in application notes AND8427 [13], AND8470 [14], and
design note DN06068 [15].
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F1
2.5A
D1 − D4
1N5406 x 4
0.47
C2 ”X”
MRA4007T
R2
560k
0.5W
D10
R6
365k
R10
R11
365k
R9
30.1k
332k
R8
2k
1/2W
365k
D11
R4
11
10
C8
0.1
100k
Q1
10, 1/2W2.2nF
+
C24 19V,
C23 0.1
D8
10,11,12
5A
_
220uF
25V
MBR30H100CTG
R31
0 ohm
R23
3.3 ohm
9
D9 MMSD
4148T
MMSZ
5245B
R17
C14 39k
10nF
L3
4,700uF 3.3uH
25V x 3
7,8,9
R24 C19 C20 C21 C22
100
SFH615A−4
U2
1
4
R26
2.7k
2
R21
C15 C16
Z3
0.1 0.1
10
R20
0.10 ohm
0.5W
Z4
(deleted)
R27
1k
3
R18
2.2k
Z2
R16
R15
33nF
C13
C12
R14
8.6k
C11
4.7uF
25V
470pF
+
T1
(6:1)
100 MMSD
MURS120T
4148T C6
6
C5
100uF
470uF
D7
1
35V
35V
MURS
R5
160T
R22
10k
49.9k
R25
20k
R13
C29
0.1
C10
1nF
6
7
8
7.32k
680pF
C9
MMBT
2222A Q2
30k
R12
C28
1uF
16
15
14 NC
U1 13
12
68nF 2
400V
5
NCP1652
1
2
3
4
5
D6
C7
SPP11N80C3
C4
22uF
400V
R3
36k
3W
76.8k R19
R7
D5
C18
Z1
C3
0.22uF
400V
C17
1M
0.5W
1nF
0.47
”X”
10nF
L2
MRA4007T
R1
C1
1.5KE440A
L1
AC
In
R29
22k
R28 C26
4.7k 1uF
U3
TL431A
C25
0.1
MMSZ
5248B
R30
3.3k
C27
2.2nF ”Y”
Notes:
1. Crossed schematic lines are not connected.
2. Heavy lines indicate power traces/planes.
3. Z2/D9 is for optional OVP.
4. L1 is Coilcraft BU10−1012R2B or equivalent.
5. L2 is Coilcraft P3221−AL or equivalent.
6. L3 is Coilcraft RFB0807−3R3L or equivalent.
7. Q1 and D8 will require small heatsinks.
NCP1652 90 Watt PFC Adapter Supply
19 Vout, 90−265VAC Input (Rev 6)
Figure 8−7. 90 W Adapter, Single Stage Converter Schematic
The 19 V, 90 W adapter circuit of Figure 8−7 achieved an average efficiency approaching 90% and a power factor greater
than 0.9 for typical operational loads. The data below compares the key parametric results at different line and load conditions.
Power factor, THD, and efficiency measurements were taken at loads of 25%, 50%, 75% and 100% at both mains voltages.
The efficiencies were averaged per Energy Star criteria. The results are shown in the table below.
Table 8−2
Vin = 120 Vac
Vin = 230 Vac
Load
100%
75%
50%
25%
THD =
6.6
4.5
7.2
11.4
PF =
0.995
0.993
0.986
0.948
Eff =
88.0
88.9
89.8
89.1
THD =
7.1
10.3
13.8
14.6
PF =
0.975
0.951
0.901
0.713
Eff =
89.9
89.4
90.4
87.1
Eff avg = 89%
Eff avg = 89.2%
Note that for either AC line value, the efficiency easily exceeded the 87% minimum Energy Star requirement for adapters.
At no load the input power consumption was less than 500 mW for either 120 or 230 Vac line conditions. The light load
efficiency was as follows:
Table 8−3
Output Load
0.5 W
1.0 W
1.7 W
120 Vac in
57%
69%
73%
230 Vac in
47%
59%
69%
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Figure 8−8 below shows the output voltage profile for the 19 V adapter at supply turn−on with no load and with full load
indicating a controlled voltage rise with no overshoot that is sometimes typical with slow control loops.
Figure 8−8. Turn−on Profiles
Figure 8−9 below displays the 120 Hz output ripple that passes through the converter. The ripple amplitude is strictly a
function of the output capacitance and output load on the power supply since the regulation loop bandwidth is necessarily less
than the ripple frequency to assure high power factor.
Figure 8−9. Output Ripple
Some Final Comments
Single stage power factor corrected switch mode topologies provide designers with a new tool to develop power supplies
when the requirements call for high power factor and compact size and the application can allow some amount of line ripple
on the output waveform. In addition to what has been highlighted with these CrM and CCM topologies, it is important to be
aware of the limitation in applying these topologies for lower output voltages. Traditionally when the output voltage is 12 volts
and less, the efficiency can be enhanced by the use of synchronous output rectifiers instead of convention PN or Schottky
diodes. Synchronous output rectifiers are not, however, wholly compatible with continuous conduction mode (CCM)
operation.
This is because CCM or DCM operation will almost always transition into the other depending on the load situation. At light
load CCM will transition to DCM, and with DCM operation, the startup and over−current conditions usually revert to CCM
for fixed frequency converters. As a result of these two different modes of operation, the required gate drive signal to the
synchronous MOSFET must be based on different sensing criteria for each mode which causes additional circuit complexity.
The “problem mode” is CCM because there has to be a delayed timing sequence to the sync MOSFET to prevent simultaneous
conduction overlap with the main primary side MOSFET. Even though the necessary timing sequence can be achieved, one
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critical issue still remains. When the sync MOSFET is turned off just prior to the main primary MOSFET coming on, the
intrinsic body diode of the sync MOSFET must carry the still flowing continuous flyback current. This parasitic body diode
usually has very poor recovery characteristics and when the main MOSFET turns on, the body diode is force commutated off
and significant reverse current will flow in the body diode during the recovery process. This current along with the associated
circuit reactive parasitics generates large voltage spikes and ringing on the sync MOSFET and main MOSFET during this
transition. This usually necessitates the addition of larger snubbers and/or TVS clamping circuits to avoid MOSFET failure.
In many cases, the added circuit cost and dissipative issues are generally not worth the increased cost and circuit complexity.
So, if synchronous rectification is desired, the control technique to use is critical conduction mode (CrM) where all of the
critical switching transitions can take place simultaneously when current in both the primary side and synchronous MOSFETs
are zero. In this case no timing sequencing is required and a simple secondary current detection scheme is all that is necessary
for effective synchronous rectifier control. Unlike DCM or CCM implementations, CrM does not have any load dependent
mode transition, and currents are always zero when switching transitions takes place. For further information on these circuits,
please view the detailed applications notes and data sheets available from ON Semiconductor.
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References − ON Semiconductor website (www.onsemi.com)
[1] NCP1652/D: Device Data Sheet
[2] AND8394/D: A 48 V, 2 A High Efficiency, Single Stage, Isolated Power Factor Corrected Power Supply for LED
Drivers and Telecom Power (NCP1652)
[3] AND8397/D: A 90 Watt High Efficiency Notebook Adapter Power Supply with Inherent Power Factor Correction
(NCP1652)
[4] AND8124/D: 90 W, Universal Input, Single Stage, PFC Converter (NCP1651)
[5] AND8147/D: An Innovative Approach to Achieving Single Stage PFC and Step−Down Conversion for
Distributive Systems (NCP1651)
[6] AND8209/D: 90 W, Single Stage, Notebook Adaptor
[7] TND317/D: 90 W Notebook AC/DC Adapter Reference Design
[8] DN06069/D: NCP1028: Single Stage, Off−line, Isolated 12 V, 800 mA Converter with High Power Factor
[9] AND8470/D: A 25 to 55 V, 0.7 to 1.5 A, Single Stage Power Factor Corrected Constant Current Offline LED
Driver with Flexible Dimming Options
[10] AND8427/D: A Constant Current Adjustable 0.7 A to 1.5 A, Up to 55 Vdc Single Stage Power Factor Corrected
LED Power Supply
[11] NCL30000/D: Power Factor Corrected TRIAC Dimmable LED Driver − Device Data Sheet
[12] NCL300001/D: High Efficiency Single Stage Power Factor Corrected LED Driver − Device Data Sheet
[13] AND8427/D: A Constant Current Adjustable 0.7 A to 1.5 A, Up to 55 Vdc Single Stage Power Factor Corrected
LED Power Supply (NCL30001)
[14] AND8470/D: A 25 to 55 V, 0.7 to 1.5 A, Single Stage Power Factor Corrected Constant Current Offline LED
Driver with Flexible Dimming Options (NCL30001)
[15] DN06068/D: NCL30001, NCS1002: Up to 28 V, 3.3 A Constant Current Offline High PF LED Driver
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CHAPTER 9
Detailed Analyses and Comparisons
This chapter provides a detailed analysis of the results obtained with the three topologies for power factor correction where
direct comparison is applicable. These topologies are Continuous Conduction Mode (CCM), Critical Conduction Mode
(CrM) and Frequency-Clamped Critical Conduction Mode (FCCrM). Comparative analyses and rankings are provided for
the topologies for given criteria.
Comparison Methodology
As discussed in Chapter 2, the comparisons between different topologies are necessary to answer the question of how these
topologies perform for a given application. However, the methodology and basis for comparison have to be kept fully under
control in order to make meaningful comparisons. If that is not done, the results would come out different and could be
misleading.
In order to keep the number of variables as low as possible, the comparisons between the 3 topologies were made with
a single power stage PCB. This PCB shown in Figure 9−1 was originally designed for the NCP1654 (CCM) PFC and hence
the NCP1654 circuitry was on-board. For the FCCrM and CrM topologies, a daughter card using the NCP1605 and the
NCP1607 respectively were made and connected to the main board as shown in Figure 9−2. The flexibility of NCP1605
allows it to be used for pure CrM mode or in the FCCrM mode depending on the choice of the Cosc capacitor that sets the
maximum frequency. For FCCrM, the Cosc choice was set to 730 pF for a maximum frequency of 65 kHz. For CrM, there
was an option to use the NCP1605 in pure CrM mode by changing Cosc to 100 pF (max frequency to 400 kHz), however,
a true CrM controller (NCP1607) was used for more definitive comparisons.
Figure 9−1. Picture of NCP1654 Demo Board Used for Comparisons
The daughter card for FCCrM (NCP1605) shown in Figure 9−2 was modified from the board used in the application note
AND8281 [1]. For CrM (NCP1607), the daughter card was modified from the board used in the application note AND8353
[2].
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Figure 9−2. Picture of Daughter Board Used for FCCrM Application
This methodology ensured that the same power traces and layout are used for all the topologies. In addition, the
consistency of measurements was maintained to make the comparisons more meaningful. The only components that were
changed (as needed) for different topologies were the FET, diode and the power inductor and the control circuit. The
summary of these choices is shown in Table 9−1 with appropriate comments.
Table 9−1. Component Comparisons for PFC Options
Attribute
P1 − CrM
P2 − FCCrM
P3 − CCM
Comments
Inductor
250 mH
9.33 Apk
PQ3230 Core
11.5 cm2
37.7 cm3
250 mH
9.33 Apk
PQ3230 Core
11.5 cm2
37.7 cm3
650 mH
5.42 Apk
RM14 Core
20.4 cm2
61.3 cm3
Despite the higher inductance value, the
CCM inductor size is not much greater than
the other options.
Power
Switch
600 V
99 mW RDS(on)
IPP60R099CP
TO−220 Package
600 V
99 mW RDS(on)
IPP60R099CP
TO−220 Package
600 V
190 mW RDS(on)
SPP20N60C3
TO−220 Package
In CCM, relatively higher RDS(on) (and lower
cost) FET can be used
Power Diode
Ultrafast
600 V, 8 A
MUR550
TO−220 Package
Ultrafast
520 V, 5 A
MUR550
TO−220 Package
Merged PN Schottky
600 V, 8 A
LQA08TC600
TO−220 Package
CrM and FCCrM allow use of Ultrafast
diodes but CCM requires a low Qrr diode for
good efficiency
Frequency
Range
20−400 kHz
22−67 kHz
65 kHz Fixed
Frequency range of FCCrM offers the
advantage of CCM without the diode
recovery issues
Control
V−mode
V−mode
I−mode
Dictated by controller design − performance
can be optimized around each
Inductor (Lp)
Based on the results in Table 9−1, it may appear at first that the CrM and FCCrM represent the best solution, considering
the low inductance value. As mentioned earlier however, this inductor is subjected to larger flux swings as its peak-to-peak
current and frequency vary over the line and extra care has to be given to the selection of the magnetic core. Furthermore,
P1 and P2 are exposed to much larger inductor currents and will typically require larger gage wire to handle the current
capacity. When designing the inductor it is also very important to minimize DCR in order to reduce conduction losses.
Strictly comparing the P1 and P2, the FCCrM is better as it subjects the inductor to lower frequency variations.
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It should also be noted that one major FCCrM merit is the possibility to use smaller inductors compared to those required
by traditional CrM circuits. Since FCCrM clamps the switching frequency, there is no need for a large inductor to pull down
the CrM switching frequency range and smaller inductors can be used without any significant efficiency reduction [3]. This
study is however based on the same inductor for CrM and FCCrM approaches for the sake of consistency.
The CCM boost inductor needs a higher inductance. Thanks to its lower flux density swing, the CCM topology could use
a low loss core material such as KoolMu or MPP. In FCCrM or CrM, because the flux density swing is high, the core loss
dominates. It is therefore necessary to choose Ferrite to reduce the core losses. In this test, we still use Ferrite core for the
CCM topology for a more fair comparison, because the Ferrite’s core losses are minimal.
Power Switch
For CrM and FCCrM, a lower RDS(on) and new generation MOSFET (IPP60R099CP) was used, that allowed conduction
loss reduction without increasing switching losses. However, as was shown in Chapters 3 and 5, the CCM topology is less
sensitive to the MOSFET change to a higher RDS(on) value, so it was implemented using a FET that is twice the RDS(on)
(SPP20N60C3) from older generation. This is understandable because the rms and peak currents are lower in CCM and
hence, when a higher RDS(on) FET is used, the conduction loss increase is not that significant. On the other hand, the lower
Coss of the FET will result in lower switching losses. For the CrM, the conduction loss is a dominant factor at heavy load
while the turn−on losses are low. So, the CrM and FCCrM take benefit from a low RDS(on) power switch.
Power Diode
The diode choices were made based on the appropriate technology and availability of components. For CCM, for example,
the diode choice is more critical – this led to the selection of the LQA08TC600 from QSpeed. Similarly, the FCCrM topology
was able to utilize a lower current, lower voltage diode (MUR550) without impacting the efficiency. That same diode was
tried in the CrM topology.
Table 9−2. PFC Preregulator Stage
Attribute
P1 − CrM
P2 − FCCrM
P3 − CCM
THD
(230 Vac)
12.3%
8.3%
7.2%
Efficiency
(100 Vac)
94.4%
95.4%
95.1%
Hold−Up Capability
(ms)
20
20
20
Results
Table 9−2 summarizes the results for the PFC stages. It can be seen that the three topologies offer a high efficiency. The
CCM solution is however a bit worse at low line, full load as a consequence of higher switching losses. FCCrM provides
a slightly better efficiency under most conditions than the other two. The CrM performance is similar to the FCCrM at low
line, full load but declines at light load and high line as a result of the switching frequency taking off. In terms of power
density, the dominant factor was the inductor, and it is larger for the CCM topology. All other aspects of the designs are
similar. From a cost perspective, the higher cost of the CCM PFC inductor and diode is likely to be offset by the lower cost
of the power MOSFET, leading to a very comparable cost metric at these power levels. Detailed cost analysis is not within
the purview of this study as it depends on many other commercial considerations.
The next three figures have a graphical representation of the efficiency observed at different power levels and different
input voltages.
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Figure 9−3. Efficiency Performance at 100 Vac
Figure 9−4. Efficiency Performance at 115 Vac
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Figure 9−5. Efficiency Performance at 230 Vac
The THD data are plotted in Figure 9−6 and the corresponding PF and THD table is given in Table 9−3. Table 9−4 further
gives the PF performance at mid−load. The THD and PF data indicate that all the three topologies yield low harmonics and
high power factor which will comfortably meet the existing IEC61000−3−2 and Energy Star requirements. However,
amongst the three topologies, the CCM consistently provides better power factor and lower harmonic distortion. This can
be attributed to the fixed frequency, the lower current ripple operation and the total absence of dead−time (1). Also, the EMI
filter that influences the performance was optimized for the CCM topology. The FCCrM topology yields THD and PF
performance which is very comparable to the CCM topology. At low line, the data are slightly worse, but that can be partly
attributed to the daughter board set−up. Finally, the CrM topology seems to perform slightly better at low line, but
deteriorates at high line – this can be attributed to much higher frequency variation at high line compared to low line. Since
the THD performance is typically measured at high line, it is apparent that both CCM and FCCrM topologies give superior
performance for THD compared to the CrM topology.
Table 9−3. Total Harmonic Distortion Comparison
CrM
CCM
FCCrM
Vin
PF
THD (%)
PF
THD (%)
PF
THD (%)
100
0.997
7.2
0.996
7.5
0.998
5.2
115
0.996
7.9
0.997
6.1
0.998
4.9
230
0.980
15.9
0.988
7.2
0.993
6.3
(1) In CrM and FCCrM, the MOSFET does not turn on immediately when the inductor current reaches zero but after the short dead-time
necessary for the MOSFET drain-source voltage to drop to its valley (for reduced turn-on losses). This dead-time and the small resonant
current that takes place are sources of distortion.
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Figure 9−6. Total Harmonic Distortion Comparison
Table 9−4. Measured Power Factor for Different Line and Load Conditions
Load
Input Voltage
(V)
CrM
FCCrM
CCM
50%
100
0.996
0.994
0.996
50%
115
0.993
0.995
0.994
50%
230
0.892
0.945
0.978
100%
100
0.998
0.994
0.998
100%
115
0.998
0.992
0.997
100%
230
0.970
0.967
0.991
Trends and Projections
While the results in these sections have provided a good direct performance comparison between different topologies for
PFC, they are for a specific application and power level. As we look for broader range of power and applications, some
general observations and priorities can be made which are useful guidelines in selecting the appropriate topology for that
power level and application.
Applicability of PFC topologies
In addition to the efficiency, there are many other attributes that can dictate the choice of the most appropriate PFC
topology. Following table (Table 9−5) provides the comparison of the different topologies with respect to the key design
attributes.
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Table 9−5. Ranking of PFC Topologies for Different Attributes
CrM
FCCrM
Low Profile
****
EMI
***
Single
Stage
Follower
Boost
****
**
****
****
***
***
CCM
Interleaved
Bridgeless
****
***
*****
****
****
*****
Isolation
*****
Efficiency
****
****
****
*****
*****
****
****
Compactness
****
****
****
***
****
*****
****
Hold−Up Time
****
****
****
****
****
**
***
Transient Response
****
****
****
****
****
**
*****
Harmonic Performance
****
*****
*****
*****
*****
***
****
Relative Cost
****
****
***
***
**
*****
****
High Power Capability
***
****
*****
****
*****
**
***
Also, depending on end application segments, certain topologies are more preferred than others. Following Table 9−6
highlights the choices of PFC circuits for different topologies.
Table 9−6. Application Based Preferences of PFC Topologies
Power Range (W)
Preferred Topology 1
Preferred Topology 2
Preferred Topology 3
75−150
CrM
FCCrM
Single Stage
ATX Power Supply
200−500
CCM
FCCrM
CrM
Server Power Supply
400−1000
CCM
Bridgeless
Interleaved
Flat TV Power Supply
200−500
Interleaved
FCCrM
CCM
Lighting
20−100
Single Stage
CrM
FCCrM
Telecom Rectifier
1000+
Bridgeless
CCM
Interleaved
On−Line UPS
500+
Bridgeless
CCM
Notebook Adapter >75 W
Tables 9−5 and 9−6 are indicative as specific requirements or particular embodiment of the considered topology can
change the ranking. For instance, the need for a superior efficiency, may lead to prefer the bridgeless or interleaved topology
to the CCM one in server applications. Similarly, the need for compactness may cause the FCCrM or CCM solution to be
adopted in lieu of the CrM option for adapters.
Summary
The above comparisons yield some interesting information and the following observations can be made for the three
different control modes:
CrM
♦
Pros: Good efficiency for power levels below 300 W.
♦
Cons: Wide frequency, bigger PFC choke, a differential mode choke is needed to reduce the input current ripple
and EMI. This differential mode choke introduces more losses that affect the overall efficiency.
FCCrM
♦
Pros: At power levels below 300 W, FCCrM PFC can have the same efficiency as CrM at low AC line and an even
better efficiency at high line. Furthermore, additional benefits include a smaller choke (verified in our testing) and
a substantial EMI and THD improvement thanks to the frequency clamp.
♦
Cons: Its complexity makes it difficult to understand how to control the on time (but it has been done inside the
controller). A differential mode choke is also needed to reduce the input current ripple and EMI, which affects the
overall efficiency.
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CCM
♦
Pros: Fixed frequency, lower EMI filter, and a smaller ripple on the input and output stage which creates little
stress for the input filter and output capacitor. Therefore this control mode is more suitable for high power
applications.
♦
Cons: This control mode exhibits higher switching losses due to the Qrr of the boost diode. In this control mode,
the boost diode therefore becomes a critical component of the PFC stage.
CCM seems to produce a lower efficiency than FCCrM and CrM at 270 W. It is however important to note that the
efficiency was measured with an EMI stage tailored for the CCM solution (keep in mind that the NCP1654 demo−board is
used for the three topologies). The CrM and FCCrM topologies would require a bigger differential mode choke that would
lead to some additional losses.
Based on the results, it can be said that the FCCrM topology offers the best alternative for the given application. However,
the other topologies also yielded very good efficiency and THD performance and can certainly be made to work in the
applications without any problems.
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References - ON Semiconductor website (www.onsemi.com)
[1] AND8281/D- Implementing the NCP1605 to Drive the PFC Stage of a 19 V/8 A Power Supply
[2] AND8353/D – Implementing Cost Effective & Robust Power Factor Correction with NCP1607
[3] “Optimizing Inductor Design for Efficient PFC Stage”, EDN Asia, 2010 November issue,
http://www.ednasia.com/article−27331−optimizinginductordesignforef ficientpfcstage−Asia.html
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NOTES
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NOTES
Special thanks to Laurent Jenck, Dhaval Dalal and Steve West for all of their assistance in completing this revision of the PFC Handbook.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
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ON Semiconductor and the ON logo are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee
regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or
incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for
each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant
into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such
unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or
indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This
literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
Literature Distribution Center for ON Semiconductor
P.O. Box 5163, Denver, Colorado 80217 USA
Phone: 303-675-2175 or 800-344-3860 Toll Free USA/Canada
Fax: 303-675-2176 or 800-344-3867 Toll Free USA/Canada
Email: [email protected]
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N. American Technical Support: 800-282-9855 Toll Free
USA/Canada.
Europe, Middle East and Africa Technical Support:
Phone: 421 33 790 2910
Japan Customer Focus Center
Phone: 81-3-5773-3850
ON Semiconductor Website: www.onsemi.com
Order Literature: http://www.onsemi.com/orderlit
For additional information, please contact your local
Sales Representative
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