RF and Microwave Power Amplifier and Transmitter Technologies

RF and Microwave Power Amplifier and Transmitter Technologies
From May 2003 High Frequency Electronics
Copyright © 2003 Summit Technical Media, LLC
High Frequency Design
RF and Microwave Power
Amplifier and Transmitter
Technologies — Part 1
By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington,
Zoya B. Popovic, Nick Pothecary, John F. Sevic and Nathan O. Sokal
F and microwave
power amplifiers
and transmitters
are used in a wide variety
of applications including
wireless communication,
jamming, imaging, radar,
and RF heating. This
article provides an introduction and historical
background for the subject, and begins the
technical discussion with material on signals,
linearity, efficiency, and RF-power devices. At
the end, there is a convenient summary of the
acronyms used—this will be provided with all
four installments. Author affiliations and contact information are also provided at the end
of each part.
With this issue, we begin a
four-part series of articles
that offer a comprehensive
overview of power amplifier
technologies. Part 1 covers
the key topics of amplifier
linearity, efficiency and
available RF power devices
The generation of significant power at RF
and microwave frequencies is required not
only in wireless communications, but also in
applications such as jamming, imaging, RF
heating, and miniature DC/DC converters.
Each application has its own unique requirements for frequency, bandwidth, load, power,
efficiency, linearity, and cost. RF power can be
generated by a wide variety of techniques
using a wide variety of devices. The basic
techniques for RF power amplification via
classes A, B, C, D, E, and F are reviewed and
illustrated by examples from HF through Ka
band. Power amplifiers can be combined into
transmitters in a similarly wide variety of
architectures, including linear, Kahn, enve-
lope tracking, outphasing, and Doherty.
Linearity can be improved through techniques
such as feedback, feedforward, and predistortion. Also discussed are some recent developments that may find use in the near future.
A power amplifier (PA) is a circuit for converting DC input power into a significant
amount of RF/microwave output power. In
most cases, a PA is not just a small-signal
amplifier driven into saturation. There exists
a great variety of different power amplifiers,
and most employ techniques beyond simple
linear amplification.
A transmitter contains one or more power
amplifiers, as well as ancillary circuits such as
signal generators, frequency converters, modulators, signal processors, linearizers, and
power supplies. The classic architecture
employs progressively larger PAs to boost a
low-level signal to the desired output power.
However, a wide variety of different architectures in essence disassemble and then
reassemble the signal to permit amplification
with higher efficiency and linearity.
Modern applications are highly varied.
Frequencies from VLF through millimeter
wave are used for communication, navigation,
and broadcasting. Output powers vary from 10
mW in short-range unlicensed wireless systems to 1 MW in long-range broadcast transmitters. Almost every conceivable type of modulation is being used in one system or another. PAs and transmitters also find use in systems such as radar, RF heating, plasmas, laser
drivers, magnetic-resonance imaging, and
miniature DC/DC converters.
This series of articles is an expanded version of the paper, “Power Amplifiers and Transmitters for RF and
Microwave” by the same authors, which appeared in the the 50th anniversary issue of the IEEE Transactions on
Microwave Theory and Techniques, March 2002. © 2002 IEEE. Reprinted with permission.
High Frequency Electronics
High Frequency Design
No single technique for power
amplification nor any single transmitter architecture is best for all
applications. Many of the basic techniques that are now coming into use
were devised decades ago, but have
only recently been made practical
because of advances in RF-power
devices and supporting circuitry such
as digital signal processing (DSP).
The development of RF power
amplifiers and transmitters can be
divided into four eras:
Spark, Arc, and Alternator
In the early days of wireless communication (from 1895 to the mid
1920s), RF power was generated by
spark, arc, and alternator techniques.
The original RF-power device, the
spark gap, charges a capacitor to a
high voltage, usually from the AC
mains. A discharge (spark) through
the gap then rings the capacitor, tuning inductor, and antenna, causing
radiation of a damped sinusoid.
Spark-gap transmitters were relatively inexpensive and capable of
generating 500 W to 5 kW from LF to
MF [1].
The arc transmitter, largely
attributed to Poulsen, was a contemporary of the spark transmitter. The
arc exhibits a negative-resistance
characteristic which allows it to operate as a CW oscillator (with some
fuzziness). The arc is actually extinguished and reignited once per RF
cycle, aided by a magnetic field and
hydrogen ions from alcohol dripped
into the arc chamber. Arc transmitters were capable of generating as
much as 1 MW at LF [2].
The alternator is basically an AC
generator with a large number of
poles. Early RF alternators by Tesla
and Fessenden were capable of operation at LF, and a technique developed by Alexanderson extended the
operation to LF [3]. The frequency
was controlled by adjusting the rotation speed and up to 200 kW could be
High Frequency Electronics
generated by a single alternator. One
such transmitter (SAQ) remains
operable at Grimeton, Sweden.
Vacuum Tubes
With the advent of the DeForest
audion in 1907, the thermoionic vacuum tube offered a means of electronically generating and controlling
RF signals. Tubes such as the RCA
UV-204 (1920) allowed the transmission of pure CW signals and facilitated the transition to higher frequencies of operation.
Younger readers may find it convenient to think of a vacuum tube as
a glass-encapsulated high-voltage
FET with heater. Many of the concepts for modern electronics, including class-A, -B, and -C power amplifiers, originated early in the vacuumtube era. PAs of this era were characterized by operation from high voltages into high-impedance loads and
by tuned output networks. The basic
circuits remained relatively unchanged throughout most of the era.
Vacuum tube transmitters were
dominant from the late 1920s
through the mid 1970s. They remain
in use today in some high power
applications, where they offer a relatively inexpensive and rugged means
of generating 10 kW or more of RF
Discrete Transistors
Discrete solid state RF-power
devices began to appear at the end of
the 1960s with the introduction of silicon bipolar transistors such as the
2N6093 (75 W HF SSB) by RCA.
Power MOSFETs for HF and VHF
appeared in 1974 with the VMP-4 by
Siliconix. GaAs MESFETs introduced
in the late 1970s offered solid state
power at the lower microwave frequencies.
The introduction of solid-state
RF-power devices brought the use of
lower voltages, higher currents, and
relatively low load resistances.
Ferrite-loaded transmission line
transformers enabled HF and VHF
PAs to operate over two decades of
bandwidth without tuning. Because
solid-state devices are temperaturesensitive, bias stabilization circuits
were developed for linear PAs. It also
became possible to implement a variety of feedback and control techniques through the variety of opamps and ICs.
Solid-state RF-power devices
were offered in packaged or chip
form. A single package might include
a number of small devices. Power outputs as high as 600 W were available
from a single packaged push-pull
device (MRF157). The designer basically selected the packaged device
that best fit the requirements. How
the transistors were made was
regarded as a bit of sorcery that
occurred in the semiconductor houses
and was not a great concern to the
ordinary circuit designer.
Custom/Integrated Transistors
The late 1980s and 1990s saw a
proliferation variety of new solidstate devices including HEMT,
pHEMT, HFET, and HBT, using a
variety of new materials such as InP,
SiC, and GaN, and offering amplification at frequencies to 100 GHz or
more. Many such devices can be operated only from relatively low voltages.
However, many current applications
need only relatively low power. The
combination of digital signal processing and microprocessor control allows
widespread use of complicated feedback and predistortion techniques to
improve efficiency and linearity.
Many of the newer RF-power
devices are available only on a madeto-order basis. Basically, the designer
selects a semiconductor process and
then specifies the size (e.g., gate
periphery). This facilitates tailoring
the device to a specific power level, as
well as incorporating it into an RFIC
or MMIC.
The need for linearity is one of the
principal drivers in the design of
High Frequency Design
Figure 1 · SRRC data pulses.
modern power amplifiers. Linear
amplification is required when the
signal contains both amplitude and
phase modulation. It can be accomplished either by a chain of linear
PAs or a combination of nonlinear
PAs. Nonlinearities distort the signal
being amplified, resulting in splatter
into adjacent channels and errors in
Signals such as CW, FM, classical
FSK, and GMSK (used in GSM) have
constant envelopes (amplitudes) and
therefore do not require linear amplification. Full-carrier amplitude modulation is best produced by high level
amplitude modulation of the final RF
PA. Classic signals that require linear amplification include single sideband (SSB) and vestigal-sideband
(NTSC) television. Modern signals
that require linear amplification
include shaped-pulse data modulation and multiple carriers.
Shaped Data Pulses
Classic FSK and PSK use abrupt
frequency or phase transitions, or
equivalently rectangular data pulses.
The resultant RF signals have constant amplitude and can therefore be
amplified by nonlinear PAs with good
efficiency. However, the resultant
sinc-function spectrum spreads signal energy over a fairly wide bandwidth. This was satisfactory for rela26
High Frequency Electronics
Figure 2 · RF waveforms for SRRC and multicarrier signals.
tively low data rates and a relatively
uncrowded spectrum.
Modern digital signals such as
QPSK or QAM are typically generated by modulating both I and Q subcarriers. The requirements for both
high data rates and efficient utilization of the increasingly crowded spectrum necessitates the use of shaped
data pulses. The most widely used
method is based upon a raised-cosine
channel spectrum, which has zero
intersymbol interference during
detection and can be made arbitrarily close to rectangular [4]. A raisedcosine channel spectrum is achieved
by using a square-root raised-cosine
(SRRC) filter in both the transmitter
and receiver. The resultant SRRC
data pulses (Figure 1) are shaped
somewhat like sinc functions which
are truncated after several cycles. At
any given time, several different data
pulses contribute to the I and Q modulation waveforms. The resultant
modulated carrier (Figure 2) has
simultaneous amplitude and phase
modulation with a peak-to-average
ratio of 3 to 6 dB.
Depending on the application, the
signals can have different amplitudes, different modulations, and
irregular frequency spacing.
In a number of applications
including HF modems, digital audio
broadcasting, and high-definition
television, it is more convenient to
use a large number of carriers with
low data rates than a single carrier
with a high data rate. The motivations include simplification of the
modulation/demodulation hardware,
equalization, and dealing with multipath propagation. Such Orthogonal
(OFDM) techniques [5] employ carriers with the same amplitude and
modulation, separated in frequency
so that modulation products from one
carrier are zero at the frequencies of
the other carriers.
Regardless of the characteristics
of the individual carriers, the resultant composite signal (Figure 2) has
simultaneous amplitude and phase
modulation. The peak-to-average
ratio is typically in the range of 8 to
13 dB.
Multiple Carriers and OFDM
Applications such as cellular base
stations, satellite repeaters, and
multi-beam “active-phased-array”
transmitters require the simultaneous amplification of multiple signals.
Nonlinearities cause imperfect
reproduction of the amplified signal,
resulting in distortion and splatter.
Amplitude nonlinearity causes the
instantaneous output amplitude or
High Frequency Design
The traditional
measure of linearity is the carrier-toER
(C/I) ratio. The PA
is driven with two
or more carriers
(tones) of equal
cause the producE
tion of intermodulation products at
frequencies correFrequency Offset from Carrier (MHz)
sponding to sums
Figure 3 · ACPR offsets and bandwidths.
Figure 4 · Error vector.
and differences of
multiples of the
carrier frequencies
envelope to differ in shape from the act upon the instantaneous signal [6]. The amplitude of the third-order
corresponding input. Such nonlinear- voltage or envelope. However, memo- or maximum intermodulation distorities are the variable gain or satura- ry effects can also occur in high- tion (IMD) product is compared to
tion in a transistor or amplifier. power PAs because of thermal effects that of the carriers to obtain the C/I.
Amplitude-to-phase conversion is a and charge storage. Thermal effects A typical linear PA has a C/I of 30 dB
phase shift associated with the signal are somewhat more noticeable in III- or better.
amplitude and causes the introduc- V semiconductors because of lower
Noise-Power Ratio (NPR) is a tration of unwanted phase modulation thermal conductivity, while charge- ditional method of measuring the lininto the output signal. Amplitude-to- storage effects are more prevalent in earity of PAs for broadband and
phase conversion is often associated overdriven BJT PAs.
noise-like signals. The PA is driven
with voltage-dependent capacitances
with Gaussian noise with a notch in
in the transistors. While imperfect Measurement of Linearity
one segment of its spectrum.
Linearity is characterized, mea- Nonlinearities cause power to appear
frequency response also distorts a
signal, it is a linear process and sured, and specified by various tech- in the notch. NPR is the ratio of the
therefore does not generate out-of- niques depending upon the specific notch power to the total signal power.
signal and application. The linearity
band signals.
Adjacent Channel Power Ratio
Amplitude nonlinearity and of RF PAs is typically characterized (ACPR) characterizes how nonlinearamplitude-to-phase conversion are by C/I, NPR, ACPR, and EVM ity affects adjacent channels and is
described by transfer functions that (defined below).
widely used with modern shapedpulse digital signals such as NADC
Offset 1
Offset 2
and CDMA. Basically, ACPR is the
ratio of the power in a specified band
NADC [13]
±30 kHz
±60 kHz
32.8 kHz
outside the signal bandwidth to the
–26 dBc
–45 dBc
rms power in the signal (Figure 3).
PHS [14]
±600 kHz
±900 kHz
37.5 kHz
In some cases, the actual power spec–50 dBc
–55 dBc
trum S(f) is weighted by the frequenEDGE [15]
±400 kHz
±600 kHz
30 kHz
cy response H(f) of the pulse-shaping
–58 dBc
–66 dBc
filter; i.e. (eq. 1)
Normalized Magnitude (dB)
TETRA [16]
25 kHz
–60 dBc
50 kHz
–70 dBc
25 kHz
IS-95 CDMA [17]
885 kHz
–45 dBc
1980 kHz
–55 dBc
30 kHz
4.68 MHz
W-CDMA (3G-PP) 5.00 MHz
–33 dB
10.0 MHz
–43 dB
Table 1 · ACPR and EVM requirements of various systems.
High Frequency Electronics
fc − fo + BW / 2
ACPRlower =
fc − fo − BW / 2
H ( f ) S ( f ) df
∫ H ( f ) S ( f ) df
Figure 5 · Envelope PDFs.
Figure 6 · Power-output PDFs.
where fc is the center frequency, B is
the bandwidth, fo is the offset, and fL
and fU are the band edges. The
weighting, frequency offsets, and
required ACPRs vary with application as shown in Table 1. ACPR can
be specified for either upper or lower
sideband. In many cases, two different ACPRs for two different frequency offsets are specified. ACPR2, based
upon the outer band, is sometimes
called “Alternate Channel Power
Error Vector Magnitude (EVM) is
a convenient measure of how nonlinearity interferes with the detection
process. EVM is defined (Figure 4) as
the distance between the desired and
actual signal vectors, normalized to a
fraction of the signal amplitude.
Often, both peak and rms errors are
specified (Table 1).
Efficiency, like linearity, is a critical factor in PA design. Three definitions of efficiency are commonly used.
Drain efficiency is defined as the
ratio of RF output power to DC input
η = Pout/Pin
Power-added efficiency (PAE)
incorporates the RF drive power by
subtracting it from the output power;
i.e. (Pout – PDR)/Pin. PAE gives a reasonable indication of PA performance
when gain is high; however, it can
become negative for low gains. An
overall efficiency such as Pout/(Pin +
PDR) is useable in all situations. This
definition can be varied to include
driver DC input power, the power
consumed by supporting circuits, and
anything else of interest.
Average Efficiency
The instantaneous efficiency is
the efficiency at one specific output
level. For most PAs, the instantaneous efficiency is highest at the
peak output power (PEP) and
decreases as output decreases.
Signals with time-varying amplitudes (amplitude modulation) therefore produce time-varying efficiencies. A useful measure of performance
is then the average efficiency, which
is defined [7] as the ratio of the average output power to the average DCinput power:
This concept can be used with any
of the three definitions of efficiency.
The probability-density function
(PDF) of the envelope gives the relative amount of time an envelope
spends at various amplitudes (Figure
5). Also used is the cumulative distri-
bution function (CDF), which gives
the probability that the envelope
does not exceed a specified amplitude. CW, FM, and GSM signals have
constant envelopes and are therefore
always at peak output. SRRC data
modulation produces PDFs that are
concentrated primarily in the upper
half of the voltage range and have
peak-to-average ratios on the order of
3 to 6 dB. Multiple carriers [8] produce random-phasor sums much like
random noise and therefore have
Rayleigh-distributed envelopes; i.e.,
p(E) = 2 E ξexp (–V2 ξ)
Peak-to-average ratio ξ is typically
between 6 and 13 dB.
The average input and output
powers are found by integrating the
product of their variation with amplitude and the PDF of the envelope.
Two cases are of special interest.
When the DC input current is constant (class-A bias), the DC input
power is also constant. The average
efficiency is then ηPEP/ξ. If the DC
input current (hence power) is proportional to the envelope (as in classB), the average efficiency is (4/π)1/2
ηPEP, for a Rayleigh-distributed signal. Thus for a multicarrier signal
with a 10 dB peak-to-average ratio,
ideal class-A and B PAs with PEP
efficiencies of 50 and 78.5 percent,
May 2003
High Frequency Design
respectively, have average efficiencies
of only 5 and 28 percent, respectively.
The need to conserve battery
power and to avoid interference to
other users operating on the same
frequency necessitates the transmission of signals whose peak amplitudes well below the peak output
power of the transmitter. Since peak
power is needed only in the worstcase links, the “back-off” is typically
in the range of 10 to 20 dB.
For a single-carrier mobile transmitter, back-off rather than envelope
PDF is dominant in determining the
average power consumption and
average efficiency. The PDF of the
transmitting power (Figure 4)
depends not only upon the distance,
but also upon factors such as attenuation by buildings, multipath, and
orientation of the mobile antenna [8],
[9], [10]. To facilitate prediction of the
power consumption, the envelope and
back-off PDFs can be combined [11].
A wide variety of active devices is
currently available for use in RFpower amplifiers, and RF-power
transistors are available in packaged,
die, and grown-to-order forms.
Packaged devices are used at frequencies up to X band, and are dominant for high power and at VHF and
lower frequencies. A given package
can contain one or more die connected in parallel and can also include
internal matching for a specific frequency of operation. Dice (chips) can
be wire-bonded directly into a circuit
to minimize the effects of the package
and are used up to 20 GHz. In
MMICs, the RF-power device is
grown to order, allowing its size and
other characteristics to be optimized
for the particular application. This
form of construction is essential for
upper-microwave and millimeterwave frequencies to minimize the
effects of strays and interconnects.
Virtually all RF power transistors
High Frequency Electronics
are npn or n-channel types because
the greater mobility of electrons (vs.
holes) results in better operation at
higher frequencies.
Bipolar Junction Transistor (BJT)
The Si BJT is the original solidstate RF power device, originating in
the 1960s. Since the BJT is a vertical
device, obtaining a high collectorbreakdown voltage is relatively simple and the power density is very
high. Si BJTs typically operate from
28 V supplies and remain in use at
frequencies up to 5 GHz, especially in
high-power (1 kW) pulsed applications such as radar. While Si RF
power devices have higher gain at
high frequencies, their fundamental
properties are basically those of ordinary bipolar transistors. The positive
temperature coefficient of BJTs tends
to allow current hogging, hot-spotting, and thermal runaway, necessitating carefully regulated base bias.
Since RF power BJTs are generally
composed of multiple, small BJTs
(emitter sites), emitter ballasting
(resistance) is generally employed to
force even division of the current
within a given package.
Metal-Oxide-Silicon Field-Effect
Transistor (MOSFET)
MOSFETs are constructed with
insulated gates. Topologies with both
vertical and later current flow are
used in RF applications, and most are
produced by a double-diffusion process. Because the insulated gate conducts no DC current, MOSFETs are
very easily biased.
The negative temperature coefficient of a MOSFET causes its drain
current to decrease with temperature. This prevents thermal runaway
and allows multiple MOSFETs to be
connected in parallel without ballasting. The absence of base-charge storage time allows fast switching and
also eliminates a mechanism for subharmonic oscillation. An overdriven
(saturated) MOSFET can conduct
drain current in either direction,
which is very useful in switchingmode operation with reactive loads.
Vertical RF power MOSFETs are
useable through VHF and UHF.
Gemini-packaged devices can deliver
up to 1 kW at HF and 100s of watts
at VHF. VMOS devices typically operate from 12, 28, or 50-V supplies,
although some devices are capable of
operation from 100 V or more.
Laterally Diffused MOS (LDMOS)
LDMOS is especially useful at
UHF and lower microwave frequencies because direct grounding of its
source eliminates bond-wire inductance that produces negative feedback and reduces gain at high frequencies. This also eliminates the
need for the BeO insulating layer
commonly used in other RF-power
LDMOS devices typically operate
from 28-V supplies and are currently
available with power outputs of 120
W at 2 GHz. They are relatively low
in cost compared to other devices for
this frequency range and are currently the device of choice for use in highpower transmitters at 900 MHz and 2
Junction FET (JFET)
JFETs for power applications are
Transistors (SITs). Impressive power
and efficiency have been obtained
from RF JFETs based upon Si, SiGe,
and SiC at frequencies through UHF.
However, the JFET has never become
as popular as other RF-power FETs.
GaAs MEtal Semiconductor FET
GaAs MESFETs are JFETs based
upon GaAs and a Schottky gate junction. They have higher mobility than
do Si devices and are therefore capable of operating efficiently at higher
frequencies. GaAs MESFETs are
widely used for the production of
microwave power, with capabilities of
up 200 W at 2 GHz and 40 W at 20
GHz in packaged devices. These
High Frequency Design
devices have relatively low breakdown
MOSFETs or JFETs and are typically operated from supply voltages
(drain biases) of 5 to 10 V. Most
devices and require a negative gate
bias, although some enhance-mode
devices that operate with a positive
bias have been developed. Linearity
is often poor due to input capacitance
variation with voltage; the output
capacitance is also often strongly
bias- and frequency-dependent.
although technically an “HFET” has
a doped channel that provides the
carriers (instead of the heterojunction). The acronyms “HFET and
“HJFET” (HeteroJunction FET)
appear to be used interchangeably.
GaAs HEMTs/HFETs with fT as
high as 158 GHz are reported. PAs
based upon these HEMTs exhibit 15W outputs at 12 GHz with a poweradded efficiency (PAE) of 50 percent.
Outputs of 100 W are available at S
band from packaged devices.
Pseudomorphic HEMT
Heterojunction FET (HFET) / HighElectron-Mobility Transistor (HEMT)
HFETs and HEMTs improve upon
the MESFET geometry by separating
the Schottky and channel functions.
Added to the basic MESFET structure is a heterojunction consisting of
an n-doped AlGaAs Schottky layer,
an undoped AlGaAs spacer, and an
undoped GaAs channel. The discontinuity in the band gaps of AlGaAs and
GaAs causes a thin layer of electrons
(“two-dimensional electron gas or 2DEG”) to form below the gate at the
interface of the AlGaAs and GaAs
layers. Separation of the donors from
the mobile electrons reduces collisions in the channel, improving the
mobility, and hence high-frequency
response, by a factor of about two.
AlGaAs has crystal-lattice properties similar to those of GaAs, and this
makes it possible to produce a potential difference without lattice stress.
The GaAs buffer contributes to a relatively high breakdown voltage.
Their fabrication employs advanced
epitaxial technologies (Molecular
Beam Epitaxy or Metal Organic
Chemical Vapor Deposition) which
tends to increase their cost.
The GaAs HEMT is known in the
literature by a wide variety of different names, including MODFET
(Modulation-Doped FET), TEGFET
FET), and SDFET (Selectively Doped
FET). It is also commonly called an
High Frequency Electronics
(pHEMT) further improves upon the
basic HEMT by employing an
InGaAs channel. The increased
mobility of In with respect to GaAs
increases the bandgap discontinuity
and therefore the number of carriers
in the two-dimensional electron gas.
The lattice mismatch between the
GaInAs channel and GaAs substrate
is also increased, however, and this
limits the In content to about 22 percent.
The efficiency of PAs using
pHEMTs does not begin to drop until
about 45 GHz and pHEMTs are useable to frequencies as high as 80
GHz. Power outputs vary from 40 W
at L band to 100 mW at V band.
While pHEMTs are normally grown
to order, a packaged device pHEMT
has recently become available.
and efficiency than the GaAs
pHEMT, with the PA efficiency beginning to drop at 60 GHz. However, it
has a lower breakdown voltage (typically 7 V) and must therefore be operated from a relatively low drain-voltage supply (e.g., 2 V). This results in
lower output per device and possibly
loss in the combiners required to
achieve a specified output power.
Nonetheless, the InP HEMT generally has a factor-of-two efficiency
advantage over the pHEMT and
InP HEMTs have been fabricated
with fmax as high as 600 GHz (0.1
µm gate length), and amplification
has been demonstrated at frequencies as high as 190 GHz. The efficiency does not begin to drop until about
60 GHz. Power levels range from 100
to 500 mW per chip.
Metamorphic HEMT (mHEMT)
The mHEMT allows channels
with high-In content to be built on
GaAs substrates. The higher electron
mobility and higher peak saturation
velocity result in higher gain than is
possible in a pHEMT. mHEMTs are
generally limited to low-power applications by their relatively low breakdown voltage (<3 V). However, an
mHEMT capable of 6-V operation
and a power output of 0.5 W has been
recently reported.
Heterojunction Bipolar Transistor
The InP HEMT places an
AlInAs/GaInAs heterojunction on an
InP substrate. The lattices are more
closely matched, which allows an In
content of up to about 53 percent.
This results in increased mobility,
which in turn results in increased
electron velocity, increased conduction-band discontinuity, increased
two-dimensional electron gas, and
higher transconductance. The thermal resistance is 40 percent lower
than that of a comparable device
built on a GaAs substrate.
The InP HEMT has higher gain
HBTs are typically based upon
the compound-semiconductor material AlGaAs/GaAs. The AlGaAs emitter
is made as narrow as possible to minimize base resistance. The base is a
thin layer of p GaAs. The barrier is
created by heterojunction (AlGaAs/
GaAs) rather than the doping. The
base can therefore be doped heavily
to minimize its resistance. Base sheet
resistance is typically two orders of
magnitude lower than that of an
ordinary BJT, and the frequency of
operation is accordingly higher. The
current flow is (in contrast to a MES-
High Frequency Design
FET) vertical so that surface imperfections have less effect upon performance. The use of a semi-insulating
substrate and the higher electron
mobility result in reduced parasitics.
The DC curves are somewhat similar
to those of a conventional BJT, but
often contain a saturation resistance
as well as saturation voltage.
Currently available AlGaAs/GaAs
HBTs are capable of producing several watts and are widely used in wireless handsets, GaAs HBTs are also
widely used in MMIC circuits at frequencies up to X band and can operate in PAs at frequencies as high as
20 GHz.
power densities of 10 W/mm, which is
ten times that of a GaAs MESFET.
The high thermal conductivity of the
SiC substrate is particularly useful
in high-power applications. The higher operating voltage and associated
higher load impedance greatly simplify output networks and power
combining. SiC MESFETs typically
operate from a 48-V supply. Devices
with outputs of 10 W are currently
available, and outputs of 60 W or
more have been demonstrated experimentally. The cost of SiC devices is
at presently about ten times that of
MMICs integrate RF power
devices and matching/decoupling elements such as on-chip inductors,
capacitors, resistors, and transmission lines. The proximity of these elements to the RF-power devices is
essential for input, output, and interstage matching at microwave and
millimeter-wave frequencies.
Radio Transmitters and Carrier
Currents, Scranton PA: International
Textbook Company, 1928.
4. J. B. Groe and L. E. Larson,
CDMA Mobile Radio Design,
Norwood, MA: Artech House, 2000.
5. R. van Nee and R. Prasad,
OFDM for Wireless Multimedia
Communications, Norwood, MA:
Artech House, 2000.
6. H. L. Krauss, C. W. Bostian, and
F. H. Raab, Solid State Radio
Engineering, New York: Wiley, 1980.
7. F. H. Raab, “Average efficiency
of power amplifiers,” Proc. RF
Technology Expo '86, Anaheim, CA,
pp. 474-486, Jan. 30-Feb. 1, 1986.
8. N. Pothecary, “Feedforward linear power amplifiers,” in Workshop
Notes WFB, Int’l. Microwave Symp.,
Boston, MA, June 16, 2001.
9. J. F. Sevic, “Statistical characterization of RF power amplifier efficiency for wireless communication
systems,” Proc. Wireless Commun.
Conf., Boulder, CO, pp. 1-4, Aug. 1997.
10. G. Hanington, P.-F. Chen, P. M.
Asbeck, and L. E. Larson, “High-efficiency power amplifier using dynamic power-supply voltage for CDMA
Microwave Theory Tech., vol. 47, no. 8,
pp. 1471-1476, Aug. 1999.
11. I. Kipnis, “Refining CDMA
Microwaves & RF, vol. 39, no. 6, pp.
71-76, June 2000.
12. J. Staudinger, “Applying
switched gain stage concepts to
improve efficiency and linearity for
mobile CDMA power amplification,”
Microwave Journal, vol. 43, no. 9, pp.
152-162, Sept. 2000.
Table 1 References
1. W. J. Bryon, “Arcs and sparks,
Part 1,” Communications Quarterly,
vol. 4, no. 2, pp. 27-43, Spring 1994.
2. W. J. Bryon, “The arc method of
Communications Quarterly, vol. 8, no.
3, pp. 47-65, Summer 1998.
3. K. M. MacIlvain and W. H.
Freedman, Radio Library, Vol. III:
13. “Mobile station – base station
interoperability standard for dualmode cellular system,” ANSI-136
Standard, Telecommun. Industries
Assoc., 2000.
14. “Digital cellular communication systems,” RCR STD-27, Research
and Development Center for Radio
Systems (RCR), April 1991.
The use of SiGe rather than Si in
the base of the HBT both increases
the maximum operating frequency
and decreases the base resistance.
However, they are generally less efficient than GaAs HBTs and can have
lower breakdown voltages. One
experimental SiGe HBT is capable of
delivering over 200 W at L band.
The use of InP in an HBT further
boosts mobility and therefore the
high frequency response. In addition,
InP HBTs have lower turn-on and
knee voltages, resulting in higher
gain and efficiency. InP HBTs for RFpower applications incorporate two
heterojunctions (AlInAs/GaInAs and
GaInAs/InP). The InP in the collector
increases the breakdown voltage,
allowing higher output power. To
date, outputs of about 0.5 W at 20
GHz have been demonstrated, but it
is anticipated that operation to 50 or
60 GHz will be possible.
The wide band gap of SiC results
in both high mobility and high breakdown voltage. An SiC MESFET can
therefore have a frequency response
comparable to that of a GaAs MESFET, but breakdown voltages double
that of Si LDMOS. This results in
High Frequency Electronics
GaN offers the same high breakdown voltage of SiC, but even higher
mobility. Its thermal conductivity is,
however, lower, hence GaN devices
must be built substrate such as SiC
or diamond. While the GaN HEMT
offers the promise of a high-power,
high-voltage device operating at frequencies of 10 GHz or more, it is still
in an experimental state. Power outputs of 8 W at 10 GHz with 30 percent efficiency have been demonstrated.
Monolithic Microwave Integrated
Circuit (MMIC)
High Frequency Design
Acronyms Used in Part 1
Alternating Current
Adjacent-Channel Power Ratio
Bipolar-Junction Transistor
Cumulative Distribution Function
Code-Division Multiple Access
Continuous Wave
Direct Current
Digital Signal Processing
Error-Vector Magnitude
Field-Effect Transistor
Frequency-Shift Keying
Gaussian Minimum Shift Keying
Global System for Mobile communication
Heterojunction bipolar transistor
High Electron-Mobility Transistor
Heterojunction FET (also HJFET)
Integrated Circuit
Junction Field-Effect Transistor
Laterally Diffused MOS (FET)
15. “Digital cellular telecommunications system
(phase 2+), radio transmission and reception,” GSM 5.05
Standard, v. 8.4.1, European Telecommun. Standards
Inst., 1999.
16. “Terrestrial trunked radio (TETRA) voice+data air
interface,” TETRA
Standard, European
Telecommun. Standards Inst., 1999.
17. “Mobile station – base station interoperability
standard for dual-mode wideband spread-spectrum cellular system,” TIA/EIA IS-95 Interim Standard,
Telecommun. Industries Assoc., July 1993.
18. “UE radio transmission and reception (FDD),” TS
25.101, v. 3.4.1, Third Generation Partnership Project,
Technical Specification Group, 1999.
Series Notes
1. The remaining three parts of this series will appear
in successive issues of High Frequency Electronics (July,
September and November 2003 issues).
High Frequency Electronics
MEtal Semiconductor FET
Metamorphic HEMT
Microwave Monolithic Integrated Circuit
Metal-Oxide-Silicon Field-Effect Transistor
North American Digital Cellular
Noise-Power Ratio
National Television Standards Committee
Orthogonal Frequency-Division Multiplex
Power Amplifier
Power-Added Efficiency
Probability-Density Function
Peak-Envelope Power
Pseudomorphic HEMT
Phase-Shift Keying
Quadrature Amplitude Modulation
Quadrature Phase Shift Keying
Radio Frequency
Square-Root Raised Cosine
Single SideBand
2. To maintain continuity, all figures, tables, equations
and references will be numbered sequentially throughout
the entire series.
3. Like all articles in High Frequency Electronics, this
series will be archived and available for downloading (for
personal use by individuals only) online at the magazine
website: www.highfrequencyelectronics.com
Author Information
The authors of this series of articles are: Frederick H.
Raab (lead author), Green Mountain Radio Research, email: f.raab@ieee.org; Peter Asbeck, University of
California at San Diego; Steve Cripps, Hywave
Associates; Peter B. Kenington, Andrew Corporation;
Zoya B. Popovic, University of Colorado; Nick Pothecary,
Consultant; John F. Sevic, California Eastern
Laboratories; and Nathan O. Sokal, Design Automation.
Readers desiring more information should contact the
lead author.
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