LT1228
LT1228
100MHz Current Feedback
Amplifier with DC Gain Control
DESCRIPTIO
U
FEATURES
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The LT®1228 makes it easy to electronically control the gain
of signals from DC to video frequencies. The LT1228
implements gain control with a transconductance amplifier
(voltage to current) whose gain is proportional to an externally controlled current. A resistor is typically used to
convert the output current to a voltage, which is then
amplified with a current feedback amplifier. The LT1228
combines both amplifiers into an 8-pin package, and operates on any supply voltage from 4V (±2V) to 30V (±15V). A
complete differential input, gain controlled amplifier can be
implemented with the LT1228 and just a few resistors.
Very Fast Transconductance Amplifier
Bandwidth: 75MHz
gm = 10 × ISET
Low THD: 0.2% at 30mVRMS Input
Wide ISET Range: 1µA to 1mA
Very Fast Current Feedback Amplifier
Bandwidth: 100MHz
Slew Rate: 1000V/µs
Output Drive Current: 30mA
Differential Gain: 0.04%
Differential Phase: 0.1°
High Input Impedance: 25MΩ, 6pF
Wide Supply Range: ±2V to ±15V
Inputs Common Mode to Within 1.5V of Supplies
Outputs Swing Within 0.8V of Supplies
Supply Current: 7mA
The LT1228 transconductance amplifier has a high impedance differential input and a current source output with wide
output voltage compliance. The transconductance, gm, is
set by the current that flows into Pin 5, ISET. The small signal
gm is equal to ten times the value of ISET and this relationship
holds over several decades of set current. The voltage at Pin
5 is two diode drops above the negative supply, Pin 4.
U
APPLICATIO S
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Video DC Restore (Clamp) Circuits
Video Differential Input Amplifiers
Video Keyer/Fader Amplifiers
AGC Amplifiers
Tunable Filters
Oscillators
The LT1228 current feedback amplifier has very high input
impedance and therefore it is an excellent buffer for the
output of the transconductance amplifier. The current feedback amplifier maintains its wide bandwidth over a wide
range of voltage gains making it easy to interface the
transconductance amplifier output to other circuitry. The
current feedback amplifier is designed to drive low impedance loads, such as cables, with excellent linearity at high
frequencies.
, LTC and LT are registered trademarks of Linear Technology Corporation.
U
TYPICAL APPLICATIO
Frequency Response
6
Differential Input Variable Gain Amp
15V
–
3
+
1
gm
R2A
10k
2
–
4
R3
100Ω
R2
100Ω
+
R4
1.24k
R6
6.19k
6
CFA
8
ISET
4.7µF
+
5
–15V
ISET = 1mA
–3
7
R5
10k
R1
270Ω
–
RG
10Ω
VOUT
GAIN (dB)
+
VIN
0
4.7µF
+
R3A
10k
VS = ±15V
RL = 100Ω
3
–6
–9
–12
ISET = 300µA
–15
RF
470Ω
–18
–21
HIGH INPUT RESISTANCE
EVEN WHEN POWER IS OFF
–18dB < GAIN < 2dB
VIN ≤ 3VRMS
LT1228 • TA01
–24
100k
ISET = 100µA
1M
10M
100M
FREQUENCY (Hz)
LT1228 • TA02
1
LT1228
U
U
RATI GS
W
W W
W
AXI U
U
ABSOLUTE
PACKAGE/ORDER I FOR ATIO
(Note 1)
ORDER PART
NUMBER
LT1228CN8
LT1228CS8
TOP VIEW
Supply Voltage ...................................................... ±18V
Input Current, Pins 1, 2, 3, 5, 8 (Note 8) ............ ±15mA
Output Short Circuit Duration (Note 2) ......... Continuous
Operating Temperature Range
LT1228C ................................................ 0°C to 70°C
LT1228M (OBSOLETE) .............. –55°C to 125°C
Storage Temperature Range ..................–65°C to 150°C
Junction Temperature
Plastic Package .............................................. 150°C
Ceramic Package (OBSOLETE) ................ 175°C
Lead Temperature (Soldering, 10 sec).................. 300°C
IOUT
1
–IN
2
+IN
3
6
VOUT
V–
4
5
ISET
+ –
gm
8
GAIN
7
V+
S8 PART MARKING
N8 PACKAGE
S8 PACKAGE
8-LEAD PLASTIC DIP
8-LEAD PLASTIC SOIC
TJ MAX = 150°C, θJA = 100°C/W (N)
TJ MAX = 150°C, θJA = 150°C/W (S)
1228
ORDER PART
NUMBER
LT1228MJ8
LT1228CJ8
J8 PACKAGE
8-LEAD CERAMIC DIP
TJ MAX = 175°C, θJA = 100°C/W (J)
OBSOLETE PACKAGE
Consider the N8 or S8 Packages for Alternate Source.
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ±5V ≤ VS ≤ ±15V, ISET = 0µA,
VCM = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
TA = 25°C
MIN
TYP
MAX
UNITS
±3
±10
±15
mV
mV
●
Input Offset Voltage Drift
IIN+
Noninverting Input Current
±0.3
±3
±10
µA
µA
±10
±65
±100
µA
µA
●
IIN–
Inverting Input Current
TA = 25°C
●
en
Input Noise Voltage Density
in
Input Noise Current Density
f = 1kHz, RF = 1k, RG = 10Ω, RS = 10k
RIN
Input Resistance
VIN = ±13V, VS = ±15V
VIN = ±3V, VS = ±5V
f = 1kHz, RF = 1k, RG = 10Ω, RS = 0Ω
CIN
Input Capacitance (Note 3)
VS = ±5V
Input Voltage Range
VS = ±15V, TA = 25°C
●
●
●
VS = ±5V, TA = 25°C
●
CMRR
Common Mode Rejection Ratio
Inverting Input Current
Common Mode Rejection
PSRR
2
Power Supply Rejection Ratio
VS = ±15V, VCM = ±13V, TA = 25°C
VS = ±15V, VCM = ±12V
VS = ±5V, VCM = ±3V, TA = 25°C
VS = ±5V, VCM = ±2V
VS = ±15V, VCM = ±13V, TA = 25°C
VS = ±15V, VCM = ±12V
VS = ±5V, VCM = ±3V, TA = 25°C
VS = ±5V, VCM = ±2V
VS = ±2V to ±15V, TA = 25°C
VS = ±3V to ±15V
●
●
6
nV/√Hz
1.4
pV/√Hz
2
2
25
25
MΩ
MΩ
6
pF
±13
±12
±3
±2
±13.5
V
V
V
V
55
55
55
55
69
±3.5
2.5
2.5
●
60
60
dB
dB
dB
dB
69
●
●
µV/°C
10
●
TA = 25°C
80
10
10
10
10
µA/V
µA/V
µA/V
µA/V
dB
dB
LT1228
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ±5V ≤ VS ≤ ±15V, ISET = 0µA,
VCM = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
Noninverting Input Current
Power Supply Rejection
VS = ±2V to ±15V, TA = 25°C
VS = ±3V to ±15V
MIN
●
Inverting Input Current
Power Supply Rejection
VS = ±2V to ±15V, TA = 25°C
VS = ±3V to ±15V
●
AV
Large-Signal Voltage Gain
VS = ±15V, VOUT = ±10V, RLOAD = 1k
VS = ±5V, VOUT = ±2V, RLOAD = 150Ω
●
●
55
55
65
65
dB
dB
ROL
Transresistance, ∆VOUT/∆IIN–
VS = ±15V, VOUT = ±10V, RLOAD = 1k
VS = ±5V, VOUT = ±2V, RLOAD = 150Ω
●
●
100
100
200
200
kΩ
kΩ
VOUT
Maximum Output Voltage Swing
VS = ±15V, RLOAD = 400Ω, TA = 25°C
±13.5
●
±12
±10
±3
±2.5
30
25
65
●
125
125
mA
mA
6
11
mA
●
VS = ±5V, RLOAD = 150Ω, TA = 25°C
IOUT
Maximum Output Current
RLOAD = 0Ω, TA = 25°C
MAX
10
50
50
UNITS
nA/V
nA/V
0.1
5
5
µA/V
µA/V
V
V
V
V
±3.7
Is
Supply Current
SR
Slew Rate (Notes 4 and 6)
TA = 25°C
500
V/µs
SR
Slew Rate
3500
V/µs
tr
Rise Time (Notes 5 and 6)
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 400Ω
TA = 25°C
BW
Small-Signal Bandwidth
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
100
MHz
tr
Small-Signal Rise Time
3.5
ns
Propagation Delay
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
3.5
ns
Small-Signal Overshoot
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
15
%
Settling Time
0.1%, VOUT = 10V, RF =1k, RG= 1k, RL =1k
45
ns
Differential Gain (Note 7)
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k
0.01
%
0.01
DEG
0.04
%
0.1
DEG
ts
Differential Phase (Note 7)
Differential Gain (Note 7)
Differential Phase (Note 7)
VOUT = 0V, ISET = 0V
TYP
●
300
10
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω
20
ns
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ±5V ≤ VS ≤ ±15V, ISET =
100µA, VCM = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
ISET = 1mA, TA = 25°C
MIN
TYP
MAX
UNITS
±0.5
±5
±10
mV
mV
●
Input Offset Voltage Drift
IOS
Input Offset Current
TA = 25°C
40
200
500
nA
nA
0.4
1
5
µA
µA
●
IB
Input Bias Current
µV/°C
10
●
TA = 25°C
●
en
Input Noise Voltage Density
f = 1kHz
20
nV/√Hz
RIN
Input Resistance-Differential Mode
VIN ≈ ±30mV
●
30
200
kΩ
Input Resistance-Common Mode
VS = ±15V, VCM = ±12V
VS = ±5V, VCM = ± 2V
●
●
50
50
1000
1000
MΩ
MΩ
3
pF
CIN
Input Capacitance
3
LT1228
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ±5V ≤ VS ≤ ±15V, ISET =
100µA, VCM = 0V unless otherwise noted.
SYMBOL
CMRR
PSRR
gm
PARAMETER
CONDITIONS
Input Voltage Range
VS = ±15V, TA = 25°C
VS = ±15V
VS = ±5V, TA = 25°C
VS = ±5V
Common Mode Rejection Ratio
Power Supply Rejection Ratio
Transconductance
MIN
VS = ±15V, VCM = ±13V, TA = 25°C
VS = ±15V, VCM = ±12V
VS = ±5V, VCM = ±3V, TA = 25°C
VS = ±5V, VCM = ±2V
VS = ±2V to ±15V, TA = 25°C
VS = ±3V to ±15V
●
●
Maximum Output Current
ISET = 100µA
60
60
100
●
0.75
1.00
IOL
Output Leakage Current
ISET = 0µA (+IIN of CFA), TA = 25°C
●
V
V
V
V
100
dB
dB
dB
dB
100
dB
dB
1.25
– 0.33
70
UNITS
±4
●
●
IOUT
MAX
±14
60
60
60
60
●
ISET = 100µA, IOUT = ±30µA, TA = 25°C
Transconductance Drift
±13
±12
±3
±2
TYP
µA/mV
%/°C
100
130
µA
0.3
3
10
µA
µA
●
VOUT
Maximum Output Voltage Swing
VS = ±15V , R1 = ∞
VS = ±5V , R1 = ∞
●
●
±13
±3
±14
±4
RO
Output Resistance
VS = ±15V, VOUT = ±13V
VS = ±5V, VOUT = ±3V
●
●
2
2
8
8
MΩ
MΩ
Output Capacitance (Note 3)
VS = ±5V
6
pF
IS
Supply Current, Both Amps
ISET = 1mA
THD
Total Harmonic Distortion
VIN = 30mVRMS at 1kHz, R1 = 100k
0.2
%
BW
Small-Signal Bandwidth
R1 = 50Ω, ISET = 500µA
80
MHz
tr
Small-Signal Rise Time
R1 = 50Ω, ISET = 500µA, 10% to 90%
5
ns
Propagation Delay
R1 = 50Ω, ISET = 500µA, 50% to 50%
5
ns
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: A heat sink may be required depending on the power supply
voltage.
Note 3: This is the total capacitance at Pin 1. It includes the input
capacitance of the current feedback amplifier and the output capacitance
of the transconductance amplifier.
Note 4: Slew rate is measured at ±5V on a ±10V output signal while
operating on ±15V supplies with RF = 1k, RG = 110Ω and RL = 400Ω. The
slew rate is much higher when the input is overdriven, see the applications
section.
4
●
9
V
V
15
mA
Note 5: Rise time is measured from 10% to 90% on a ±500mV output
signal while operating on ±15V supplies with RF = 1k, RG = 110Ω and
RL = 100Ω. This condition is not the fastest possible, however, it does
guarantee the internal capacitances are correct and it makes automatic
testing practical.
Note 6: AC parameters are 100% tested on the ceramic and plastic DIP
packaged parts (J and N suffix) and are sample tested on every lot of
the SO packaged parts (S suffix).
Note 7: NTSC composite video with an output level of 2V.
Note 8: Back to back 6V Zener diodes are connected between Pins 2
and 3 for ESD protection.
LT1228
U W
TYPICAL PERFOR A CE CHARACTERISTICS Transconductance Amplifier, Pins 1, 2, 3 & 5
100
VS = ±15V
Small-Signal Transconductance
and Set Current vs Bias Voltage
100
R1 = 100Ω
TRANSCONDUCTANCE (µA/mV)
10
R1 = 10k
1
10000
VS = ±2V TO ±15V
TA = 25°C
2.0
1000
10
100
1
0.1
10
0.01
1.0
R1 = 100k
100
1000
0.001
0.9
1.0
SET CURRENT (µA)
1.1
1.2
0.1
1.5
25°C
0.8
0.6
125°C
0.4
0
–200 –150 –100 –50
0.1
100 150 200
LT1228 • TPC03
Input Common Mode Limit vs
Temperature
V+
VS = ±2V TO ±15V
TA = 25°C
ISET = 1mA
100
ISET = 100µA
V + = 2V TO 15V
–0.5
COMMON MODE RANGE (V)
SPOT NOISE (pA/√Hz)
ISET = 100µA
50
0
INPUT VOLTAGE (mVDC)
ISET = 1mA
–1.0
–1.5
–2.0
2.0
1.5
V – = –2V TO –15V
1.0
0.5
0.01
V–
–50
10
100
10
10
1000
100
INPUT VOLTAGE (mVP–P)
10k
1k
100k
50
75
125
Output Saturation Voltage vs
Temperature
V+
OUTPUT SATURATION VOLTAGE (V)
0.9
CONTROL PATH GAIN (µA/µA)
100
LT1228 • TPC06
1.0
∆IOUT
∆ISET
25
TEMPERATURE (°C)
Small-Signal Control Path
Gain vs Input Voltage
VS = ±2V TO ±15V
VIN = 200mV
(PIN 2 TO 3)
0
LT1228 • TPC05
Small-Signal Control Path
Bandwidth vs Set Current
10
–25
FREQUENCY (Hz)
LT1228 • TPC04
–3dB BANDWIDTH (MHz)
1.0
LT1228 • TPC02
1000
1
100
–55°C
1.2
Spot Output Noise Current vs
Frequency
VS = ±15V
1
1.4
BIAS VOLTAGE, PIN 5 TO 4, (V)
Total Harmonic Distortion vs
Input Voltage
OUTPUT DISTORTION (%)
1.4
1.3
LT1228 • TPC01
10
1.6
0.2
0.1
10
V S = ±2V TO ±15V
ISET = 100µA
1.8
SET CURRENT (µA)
–3dB BANDWIDTH (MHz)
R1 = 1k
Small-Signal Transconductance
vs DC Input Voltage
TRANSCONDUCTANCE (µA/mV)
Small-Signal Bandwidth vs
Set Current
0.8
0.7
0.6
0.5
∆IOUT
∆ISET
0.4
0.3
0.2
–0.5
–1.0
±2V ≤ VS ≤ ±15V
R1 = ∞
+1.0
+0.5
0.1
1
0
10
100
1000
SET CURRENT (µA)
0
40
80
120
160
200
INPUT VOLTAGE, PIN 2 TO 3, (mVDC)
LT1228 • TPC07
LT1228 • TPC08
V–
–50
–25
0
25
50
75
100
125
TEMPERATURE (°C)
LT1228 • TPC09
5
LT1228
U W
TYPICAL PERFOR A CE CHARACTERISTICSCurrent Feedback Amplifier, Pins 1, 6, 8
Voltage Gain and Phase vs
Frequency, Gain = 6dB
160
6
90
140
GAIN
135
4
180
3
225
2
VS = ±15V
RL = 100Ω
RF = 750Ω
RF = 500Ω
120
RF = 750Ω
100
80
RF = 1k
60
40
RF = 2k
20
–1
–2
0.1
160
0
1
10
100
2
4
6
8
10
14
12
0
160
90
140
135
18
180
17
225
16
VS = ±15V
RL = 100Ω
RF = 750Ω
PHASE SHIFT (DEG)
VOLTAGE GAIN (dB)
GAIN
45
19
10
100
RF = 250Ω
80
RF = 500Ω
RF = 750Ω
60
RF = 1k
40
RF = 2k
2
4
6
8
10
14
12
16
0
18
2
4
–3dB BANDWIDTH (MHz)
RF = 500Ω
10
RF = 1k
8
6
RF = 2k
4
0
14
12
16
18
RF = 500Ω
12
RF = 1k
10
8
RF = 2k
6
4
2
0
0
2
4
6
8
10
12
14
16
18
SUPPLY VOLTAGE (±V)
LT1228 • TPC16
10
–3dB Bandwidth vs Supply
Voltage, Gain = 100, RL = 1kΩ
14
12
2
FREQUENCY (MHz)
8
LT1228 • TPC15
14
32
0.1
100
6
SUPPLY VOLTAGE (±V)
16
PHASE SHIFT (DEG)
VOLTAGE GAIN (dB)
RF = 2k
20
90
33
6
RF = 1k
40
18
225
10
RF = 750Ω
60
16
37
VS = ±15V
RL = 100Ω
RF = 750Ω
RF = 500Ω
80
45
180
1
RF = 250Ω
100
18
38
18
120
0
135
16
140
LT1228 • TPC14
39
34
14
12
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
–3dB Bandwidth vs Supply
Voltage, Gain = 100, RL = 100Ω
36
10
0
0
Voltage Gain and Phase vs
Frequency, Gain = 40dB
GAIN
8
–3dB Bandwidth vs Supply
Voltage, Gain = 10, RL = 1kΩ
SUPPLY VOLTAGE (±V)
PHASE
6
LT1228 • TPC12
160
LT1228 • TPC13
35
4
180
FREQUENCY (MHz)
40
2
SUPPLY VOLTAGE (±V)
120
100
42
RF = 2k
RF = 1k
0
0
1
40
18
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
20
13
41
16
180
–3dB BANDWIDTH (MHz)
21
12
0.1
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
60
–3dB Bandwidth vs Supply
Voltage, Gain = 10, RL = 100Ω
PHASE
14
80
LT1228 • TPC11
Voltage Gain and Phase vs
Frequency, Gain = 20dB
15
100
SUPPLY VOLTAGE (±V)
LT1228 • TPC10
20
RF = 750Ω
120
0
0
FREQUENCY (MHz)
22
RF = 500Ω
140
20
–3dB BANDWIDTH (MHz)
0
180
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
–3dB BANDWIDTH (MHz)
1
PHASE SHIFT (DEG)
5
–3dB Bandwidth vs Supply
Voltage, Gain = 2, RL = 1k
–3dB BANDWIDTH (MHz)
180
45
PHASE
–3dB BANDWIDTH (MHz)
0
7
8
VOLTAGE GAIN (dB)
–3dB Bandwidth vs Supply
Voltage, Gain = 2, RL = 100Ω
0
2
4
6
8
10
12
14
16
18
SUPPLY VOLTAGE (±V)
LT1228 • TPC17
LT1228 • TPC18
LT1228
U W
TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6, 8
Maximum Capacitive Load vs
Feedback Resistor
Total Harmonic Distortion vs
Frequency
0.10
VS = ±15V
100
RL = 1k
PEAKING ≤ 5dB
GAIN = 2
10
1
0.01
VO = 7VRMS
1
2
10
100
1k
FEEDBACK RESISTOR (kΩ)
10k
–1.5
–2.0
2.0
V – = –2V TO –15V
1.0
0.5
Output Short-Circuit Current vs
Temperature
25
50
75
100
70
–0.5
V–
0
–1.0
RL = ∞
±2V ≤ VS ≤ ±15V
1.0
0.5
V–
–50 –25
125
TEMPERATURE (°C)
0
25
75
50
100
60
50
40
30
–50 –25
125
0
TEMPERATURE (°C)
75 100 125 150 175
LT1228 • TPC24
Power Supply Rejection vs
Frequency
Output Impedance vs
Frequency
80
100
VS = ±15V
RL = 100Ω
RF = RG = 750Ω
60
POSITIVE
40
NEGATIVE
20
OUTPUT IMPEDANCE (Ω)
POWER SUPPLY REJECTION (dB)
100
en
50
LT1228 • TPC23
Spot Noise Voltage and Current vs
Frequency
10
25
TEMPERATURE (°C)
LT1228 • TPC22
–in
100
LT1228 • TPC21
OUTPUT SHORT-CIRCUIT CURRENT (mA)
V + = 2V TO 15V
–50 –25
10
FREQUENCY (MHz)
V+
OUTPUT SATURATION VOLTAGE (V)
COMMON MODE RANGE (V)
1
Output Saturation Voltage vs
Temperature
V+
SPOT NOISE (nV/√Hz OR pA/√Hz)
–70
100k
LT1228 • TPC20
Input Common Mode Limit vs
Temperature
1.5
3rd
–50
FREQUENCY (Hz)
LT1228 • TPC19
–1.0
–40
–60
3
–0.5
2nd
VO = 1VRMS
0.001
0
VS = ±15V
VO = 2VP–P
RL = 100Ω
RF = 750Ω
AV = 10dB
–30
DISTORTION (dBc)
VS = ±5V
1k
–20
VS = ±15V
RL = 400Ω
RF = RG = 750Ω
TOTAL HARMONIC DISTORTION (%)
CAPACITIVE LOAD (pF)
10k
2nd and 3rd Harmonic
Distortion vs Frequency
VS = ±15V
10
1.0
RF = RG = 2k
RF = RG = 750Ω
0.1
0.01
+in
1
10
100
1k
10k
100k
FREQUENCY (Hz)
0
10k
100k
1M
10M
100M
FREQUENCY (Hz)
LT1228 • TPC25
0.001
10k
100k
1M
10M
100M
FREQUENCY (Hz)
LT1228 • TPC26
LT1228 • TPC27
7
LT1228
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TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6 & 8
Settling Time to 10mV vs
Output Step
Settling Time to 1mV vs
Output Step
INVERTING
4
4
OUTPUT STEP (V)
6
2
NONINVERTING
8
6
VS = ±15V
RF = RG = 1k
0
10
–2
–4
9
8
INVERTING
SUPPLY CURRENT (mA)
NONINVERTING
8
OUTPUT STEP (V)
Supply Current vs Supply Voltage
10
10
2
VS = ±15V
RF = RG = 1k
0
–2
–4
INVERTING
–6
–6
–8
NONINVERTING
0
20
40
60
80
100
5
125°C
4
175°C
3
1
–10
–10
25°C
6
2
NONINVERTING
–8
INVERTING
–55°C
7
0
4
8
12
16
20
SETTLING TIME (µs)
SETTLING TIME (ns)
0
0
2
4
6
8
10
LT1228 • TPC29
LT1228 • TPC28
W
W
7 V+
BIAS
ISET
–IN
3
2
IOUT
8 GAIN
6 VOUT
1
5
4 V–
LT1228 • TA03
8
14
16
18
LT1228 • TPC30
SI PLIFIED SCHE ATIC
+IN
12
SUPPLY VOLTAGE (±V)
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The LT1228 contains two amplifiers, a transconductance
amplifier (voltage-to-current) and a current feedback amplifier (voltage-to-voltage). The gain of the transconductance amplifier is proportional to the current that is externally programmed into Pin 5. Both amplifiers are designed
to operate on almost any available supply voltage from 4V
(±2V) to 30V (±15V). The output of the transconductance
amplifier is connected to the noninverting input of the
current feedback amplifier so that both fit into an eight pin
package.
Resistance Controlled Gain
If the set current is to be set or varied with a resistor or
potentiometer it is possible to use the negative temperature coefficient at Pin 5 (with respect to Pin 4) to compensate for the negative temperature coefficient of the transconductance. The easiest way is to use an LT1004-2.5, a 2.5V
reference diode, as shown below:
Temperature Compensation of gm with a 2.5V Reference
R
ISET
TRANSCONDUCTANCE AMPLIFIER
The LT1228 transconductance amplifier has a high impedance differential input (Pins 2 and 3) and a current source
output (Pin 1) with wide output voltage compliance. The
voltage to current gain or transconductance (gm) is set by
the current that flows into Pin 5, ISET. The voltage at Pin 5
is two forward biased diode drops above the negative
supply, Pin 4. Therefore the voltage at Pin 5 (with
respect to V –) is about 1.2V and changes with the log of
the set current (120mV/decade), see the characteristic
curves. The temperature coefficient of this voltage is
about –4mV/°C (–3300ppm/°C) and the temperature coefficient of the logging characteristic is 3300ppm/°C. It is
important that the current into Pin 5 be limited to less than
15mA. THE LT1228 WILL BE DESTROYED IF PIN 5 IS
SHORTED TO GROUND OR TO THE POSITIVE SUPPLY. A
limiting resistor (2k or so) should be used to prevent more
than 15mA from flowing into Pin 5.
The small-signal transconductance (gm) is equal to ten
times the value of ISET (in mA/mV) and this relationship
holds over many decades of set current (see the characteristic curves). The transconductance is inversely proportional to absolute temperature (–3300ppm/°C). The input
stage of the transconductance amplifier has been designed to operate with much larger signals than is possible
with an ordinary diff-amp. The transconductance of the
input stage varies much less than 1% for differential input
signals over a ±30 mV range (see the characteristic curve
Small-Signal Transconductance vs DC Input Voltage).
gm
Vbe
4
5
R
2.5V
2Eg
ISET
Vbe
LT1004-2.5
V–
LT1228 • TA04
The current flowing into Pin 5 has a positive temperature
coefficient that cancels the negative coefficient of the
transconductance. The following derivation shows why a
2.5V reference results in zero gain change with temperature:
q ISET
×
= 10 • ISET
kT 3.87
 cT n 
akT
= Eg –
where a = In 

q
 Ic 
Since g m =
and V be
(
≈ 19.4 at 27°C c = 0.001, n = 3, Ic = 100µA
)
Eg is about 1.25V so the 2.5V reference is 2Eg. Solving
the loop for the set current gives:
ISET =

akT 
2E g – 2  E g –

q 

R
or ISET =
2akT
Rq
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Substituting into the equation for transconductance gives:
a
10
=
1.94R R
The temperature variation in the term “a” can be ignored
since it is much less than that of the term “T” in the
equation for Vbe. Using a 2.5V source this way will maintain the gain constant within 1% over the full temperature
range of –55°C to 125°C. If the 2.5V source is off by 10%,
the gain will vary only about ±6% over the same temperature range.
gm =
We can also temperature compensate the transconductance without using a 2.5V reference if the negative power
supply is regulated. A Thevenin equivalent of 2.5V is
generated from two resistors to replace the reference. The
two resistors also determine the maximum set current,
approximately 1.1V/RTH. By rearranging the Thevenin
equations to solve for R4 and R6 we get the following
equations in terms of RTH and the negative supply, VEE.
R4 =
R TH
 2.5V 
1 – V 

EE 
and R6 =
R THV EE
2.5V
is two diode drops above the negative supply, a single
resistor from the control voltage source to Pin 5 will
suffice in many applications. The control voltage is referenced to the negative supply and has an offset of about
900mV. The conversion will be monotonic, but the linearity is determined by the change in the voltage at Pin 5
(120mV per decade of current). The characteristic is very
repeatable since the voltage at Pin 5 will vary less than
±5% from part to part. The voltage at Pin 5 also has a
negative temperature coefficient as described in the previous section. When the gain of several LT1228s are to be
varied together, the current can be split equally by using
equal value resistors to each Pin 5.
For more accurate (and linear) control, a voltage-tocurrent converter circuit using one op amp can be used.
The following circuit has several advantages. The input no
longer has to be referenced to the negative supply and the
input can be either polarity (or differential). This circuit
works on both single and split supplies since the input
voltage and the Pin 5 voltage are independent of each
other. The temperature coefficient of the output current is
set by R5.
R3
1M
Temperature Compensation of gm with a Thevenin Voltage
1.03k
R1
1M
R'
V1
ISET
gm
R6
6.19kΩ
R'
Vbe
4
5
V2
+
R5
1k
LT1006
–
ISET
IOUT
TO PIN 5
OF LT1228
R4
1M
VTH = 2.5V
Vbe
50pF
R1 = R2
R3 = R4
R4
1.24kΩ
–15V
LT1228 • TA05
Voltage Controlled Gain
To use a voltage to control the gain of the transconductance amplifier requires converting the voltage into a
current that flows into Pin 5. Because the voltage at Pin 5
10
R2
1M
IOUT =
(V1 – V2) R3
•
= 1mA/V
R5
R1
LT1228 • TA19
Digital control of the transconductance amplifier gain is
done by converting the output of a DAC to a current flowing
into Pin 5. Unfortunately most current output DACs
sink rather than source current and do not have output
LT1228
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compliance compatible with Pin 5 of the LT1228. Therefore, the easiest way to digitally control the set current is
to use a voltage output DAC and a voltage-to-current
circuit. The previous voltage-to-current converter will take
the output of any voltage output DAC and drive Pin 5 with
a proportional current. The R, 2R CMOS multiplying DACs
operating in the voltage switching mode work well on both
single and split supplies with the above circuit.
Transconductance Amp Small-Signal Response
ISET = 500µA, R1 = 50Ω
Logarithmic control is often easier to use than linear
control. A simple circuit that doubles the set current for
each additional volt of input is shown in the voltage
controlled state variable filter application near the end of
this data sheet.
Transconductance Amplifier Frequency Response
The bandwidth of the transconductance amplifier is a
function of the set current as shown in the characteristic
curves. At set currents below 100µA, the bandwidth is
approximately:
–3dB bandwidth = 3 • 1011 ISET
The peak bandwidth is about 80MHz at 500µA. When a
resistor is used to convert the output current to a voltage,
the capacitance at the output forms a pole with the
resistor. The best case output capacitance is about 5pF
with ±15V supplies and 6pF with ±5V supplies. You must
add any PC board or socket capacitance to these values to
get the total output capacitance. When using a 1k resistor
at the output of the transconductance amp, the output
capacitance limits the bandwidth to about 25MHz.
The output slew rate of the transconductance amplifier is
the set current divided by the output capacitance, which is
6pF plus board and socket capacitance. For example with
the set current at 1mA, the slew rate would be over
100V/µs.
CURRENT FEEDBACK AMPLIFIER
The LT1228 current feedback amplifier has very high
noninverting input impedance and is therefore an excellent buffer for the output of the transconductance amplifier. The noninverting input is at Pin 1, the inverting input
at Pin 8 and the output at Pin 6. The current feedback
amplifier maintains its wide bandwidth for almost all
voltage gains making it easy to interface the output levels
of the transconductance amplifier to other circuitry. The
current feedback amplifier is designed to drive low impedance loads such as cables with excellent linearity at high
frequencies.
Feedback Resistor Selection
The small-signal bandwidth of the LT1228 current feedback amplifier is set by the external feedback resistors and
the internal junction capacitors. As a result, the bandwidth
is a function of the supply voltage, the value of the
feedback resistor, the closed-loop gain and load resistor.
The characteristic curves of bandwidth versus supply
voltage are done with a heavy load (100Ω) and a light load
(1k) to show the effect of loading. These graphs also show
11
LT1228
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the family of curves that result from various values of the
feedback resistor. These curves use a solid line when the
response has less than 0.5dB of peaking and a dashed line
for the response with 0.5dB to 5dB of peaking. The curves
stop where the response has more than 5dB of peaking.
Current Feedback Amp Small-Signal Response
VS = ±15V, RF = RG = 750Ω, RL = 100Ω
Capacitance on the Inverting Input
Current feedback amplifiers want resistive feedback from
the output to the inverting input for stable operation. Take
care to minimize the stray capacitance between the output
and the inverting input. Capacitance on the inverting input
to ground will cause peaking in the frequency response
(and overshoot in the transient response), but it does not
degrade the stability of the amplifier. The amount of
capacitance that is necessary to cause peaking is a function of the closed-loop gain taken. The higher the gain, the
more capacitance is required to cause peaking. For example, in a gain of 100 application, the bandwidth can be
increased from 10MHz to 17MHz by adding a 2200pF
capacitor, as shown below. CG must have very low series
resistance, such as silver mica.
1
VIN
+
6
CFA
8
VOUT
–
RF
510Ω
RG
5.1Ω
CG
LT1228 • TA08
Boosting Bandwidth of High Gain Amplifier
with Capacitance On Inverting Input
49
46
CG = 4700pF
43
40
GAIN (dB)
At a gain of two, on ±15V supplies with a 750Ω feedback
resistor, the bandwidth into a light load is over 160MHz
without peaking, but into a heavy load the bandwidth
reduces to 100MHz. The loading has so much effect
because there is a mild resonance in the output stage that
enhances the bandwidth at light loads but has its Q
reduced by the heavy load. This enhancement is only
useful at low gain settings, at a gain of ten it does not boost
the bandwidth. At unity gain, the enhancement is so
effective the value of the feedback resistor has very little
effect on the bandwidth. At very high closed-loop gains,
the bandwidth is limited by the gain-bandwidth product of
about 1GHz. The curves show that the bandwidth at a
closed-loop gain of 100 is 10MHz, only one tenth what it
is at a gain of two.
CG = 2200pF
37
34
CG = 0
31
28
25
22
19
1
10
100
FREQUENCY (MHz)
LT1228 • TA09
12
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Capacitive Loads
The LT1228 current feedback amplifier can drive capacitive loads directly when the proper value of feedback
resistor is used. The graph of Maximum Capacitive Load
vs Feedback Resistor should be used to select the appropriate value. The value shown is for 5dB peaking when
driving a 1k load, at a gain of 2. This is a worst case
condition, the amplifier is more stable at higher gains, and
driving heavier loads. Alternatively, a small resistor (10Ω
to 20Ω) can be put in series with the output to isolate the
capacitive load from the amplifier output. This has the
advantage that the amplifier bandwidth is only reduced
when the capacitive load is present and the disadvantage
that the gain is a function of the load resistance.
The output slew rate is set by the value of the feedback
resistors and the internal capacitance. At a gain of ten with
a 1k feedback resistor and ±15V supplies, the output slew
rate is typically 500V/µs and –850V/µs. There is no input
stage enhancement because of the high gain. Larger
feedback resistors will reduce the slew rate as will lower
supply voltages, similar to the way the bandwidth is
reduced.
Current Feedback Amp Large-Signal Response
VS = ±15V, RF = 1k, RG = 110Ω, RL = 400Ω
Slew Rate
The slew rate of the current feedback amplifier is not
independent of the amplifier gain configuration the way it
is in a traditional op amp. This is because the input stage
and the output stage both have slew rate limitations. The
input stage of the LT1228 current feedback amplifier slews
at about 100V/µs before it becomes nonlinear. Faster
input signals will turn on the normally reverse biased
emitters on the input transistors and enhance the slew rate
significantly. This enhanced slew rate can be as much as
3500V/µs!
Current Feedback Amp Large-Signal Response
VS = ±15V, RF = RG = 750Ω Slew Rate Enhanced
Settling Time
The characteristic curves show that the LT1228 current
feedback amplifier settles to within 10mV of final value in
40ns to 55ns for any output step less than 10V. The curve
of settling to 1mV of final value shows that there is a slower
thermal contribution up to 20µs. The thermal settling
component comes from the output and the input stage.
The output contributes just under 1mV/V of output change
and the input contributes 300µV/V of input change.
Fortunately the input thermal tends to cancel the output
thermal. For this reason the noninverting gain of two
configuration settles faster than the inverting gain of one.
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Power Supplies
The LT1228 amplifiers will operate from single or split
supplies from ±2V (4V total) to ±18V (36V total). It is not
necessary to use equal value split supplies, however the
offset voltage and inverting input bias current of the
current feedback amplifier will degrade. The offset voltage
changes about 350µV/V of supply mismatch, the inverting
bias current changes about 2.5µA/V of supply mismatch.
For example, let’s calculate the worst case power dissipation in a variable gain video cable driver operating on ±12V
supplies that delivers a maximum of 2V into 150Ω. The
maximum set current is 1mA.
(
) (
P D = 2VS ISMAX + 3.5ISET + VS – V OMAX
[
(
)] (
) VOMAX
RL
P D = 2 • 12V • 7mA + 3.5 • 1mA + 12V – 2V
) 1502VΩ
= 0.252 + 0.133 = 0.385W
Power Dissipation
The worst case amplifier power dissipation is the total of
the quiescent current times the total power supply voltage
plus the power in the IC due to the load. The quiescent
supply current of the LT1228 transconductance amplifier
is equal to 3.5 times the set current at all temperatures. The
quiescent supply current of the LT1228 current feedback
amplifier has a strong negative temperature coefficient
and at 150°C is less than 7mA, typically only 4.5mA. The
power in the IC due to the load is a function of the output
voltage, the supply voltage and load resistance. The worst
case occurs when the output voltage is at half supply, if it
can go that far, or its maximum value if it cannot reach half
supply.
The total power dissipation times the thermal resistance of
the package gives the temperature rise of the die above
ambient. The above example in SO-8 surface mount package (thermal resistance is 150°C/W) gives:
Temperature Rise = PDθJA = 0.385W • 150°C/W
= 57.75°C
Therefore the maximum junction temperature is 70°C
+57.75°C or 127.75°C, well under the absolute maximum
junction temperature for plastic packages of 150°C.
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TYPICAL APPLICATI
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Basic Gain Control
The basic gain controlled amplifier is shown on the front
page of the data sheet. The gain is directly proportional to
the set current. The signal passes through three stages
from the input to the output.
First the input signal is attenuated to match the dynamic
range of the transconductance amplifier. The attenuator
should reduce the signal down to less than 100mV peak.
The characteristic curves can be used to estimate how
much distortion there will be at maximum input signal. For
single ended inputs eliminate R2A or R3A.
The signal is then amplified by the transconductance
amplifier (gm) and referred to ground. The voltage gain of
the transconductance amplifier is:
14
g m • R1 = 10 • ISET • R1
Lastly the signal is buffered and amplified by the current
feedback amplifier (CFA). The voltage gain of the current
feedback amplifier is:
1+
RF
RG
The overall gain of the gain controlled amplifier is the
product of all three stages:
 R3 
 RF 
AV = 
•
10
•
I
•
R
1
•
SET

1 + R 
 R3 + R3A 

G
More than one output can be summed into R1 because the
output of the transconductance amplifier is a current. This
is the simplest way to make a video mixer.
LT1228
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Video Fader
Video DC Restore (Clamp) Circuit
NOT NECESSARY IF THE SOURCE RESISTANCE IS LESS THAN 50Ω
VIN1
1k
3
+
1
gm
2
1000pF
LT1223
CFA
–
5
100Ω
+
200Ω
3
VOUT
V+
7
+
–
1k
–
10k
–5V
10k
CFA
8
VOUT
–
RF
V–
5.1k
6
10k
5.1k
VS = ±5V
+
0.01µF
5
4
10k
1
gm
2
RG
5V
1k
VIN2
1k
100Ω
3
gm
2
3k
5
+
VIDEO
INPUT
1
LOGIC
INPUT
–
LT1228 • TA12
The video fader uses the transconductance amplifiers
from two LT1228s in the feedback loop of another current
feedback amplifier, the LT1223. The amount of signal
from each input at the output is set by the ratio of the
set currents of the two LT1228s, not by their absolute
value. The bandwidth of the current feedback amplifier
is inversely proportional to the set current in this
configuration. Therefore, the set currents remain high
over most of the pot’s range, keeping the bandwidth over
15MHz even when the signal is attenuated 20dB. The pot
is set up to completely turn off one LT1228 at each end of
the rotation.
RESTORE
2N3906
3k
LT1228 • TA13
The video restore (clamp) circuit restores the black level of
the composite video to zero volts at the beginning of every
line. This is necessary because AC coupled video changes
DC level as a function of the average brightness of the
picture. DC restoration also rejects low frequency noise
such as hum.
The circuit has two inputs: composite video and a logic
signal. The logic signal is high except during the back porch
time right after the horizontal sync pulse. While the logic is
high, the PNP is off and ISET is zero. With ISET equal to zero
the feedback to Pin 2 has no affect. The video input drives
the noninverting input of the current feedback amplifier
whose gain is set by RF and RG. When the logic signal is
low, the PNP turns on and ISET goes to about 1mA. Then the
transconductance amplifier charges the capacitor to force
the output to match the voltage at Pin 3, in this case zero
volts.
This circuit can be modified so that the video is DC coupled
by operating the amplifier in an inverting configuration.
Just ground the video input shown and connect RG to the
video input instead of to ground.
15
LT1228
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TYPICAL APPLICATI
S
Single Supply Wien Bridge Oscillator
resistor and the transconductance amplifier must be about
11, resulting in a set current of about 600µA at oscillation.
At start-up there is no set current and therefore no attenuation for a net gain of about 11 around the loop. As the
output oscillation builds up it turns on the PNP transistor
which generates the set current to regulate the output
voltage.
100Ω
2N3906
V+
6V TO 30V
V+
10kΩ
470Ω
+
10µF
7
3
+
5
1
gm
10kΩ
2
–
+
8
4
12MHz Negative Resistance LC Oscillator
0.1µF
6
CFA
V+
–
RF
680Ω
51Ω
9.1k
10µF
160Ω
1000pF
8
VO
51Ω
–
V–
50Ω
6
750Ω
1k
160Ω
30pF
50Ω
4.3k
330Ω
2N3906
2N3904
LT1228 • TA14
In this application the LT1228 is biased for operation from
a single supply. An artificial signal ground at half supply
voltage is generated with two 10k resistors and bypassed
with a capacitor. A capacitor is used in series with R G to set
the DC gain of the current feedback amplifier to unity.
The transconductance amplifier is used as a variable
resistor to control gain. A variable resistor is formed by
driving the inverting input and connecting the output back
to it. The equivalent resistor value is the inverse of the g m.
This works with the 1.8k resistor to make a variable
attenuator. The 1MHz oscillation frequency is set by the
Wien bridge network made up of two 1000pF capacitors
and two 160Ω resistors.
For clean sine wave oscillation, the circuit needs a net gain
of one around the loop. The current feedback amplifier has
a gain of 34 to keep the voltage at the transconductance
amplifier input low. The Wien bridge has an attenuation of
3 at resonance; therefore the attenuation of the 1.8k
16
CFA
4
4.7µH
f = 1MHz
VO = 6dBm (450mVRMS)
2nd HARMONIC = –38dBc
3rd HARMONIC = –54 dBc
FOR 5V OPERATION SHORT OUT 100Ω RESISTOR
+
5
–
10µF
1000pF
1
gm
2
+
+
7
+
1k
RG
20Ω
1.8k
3
VO
10k
VO = 10dB
0.1µF
V–
AT VS = ±5V ALL HARMONICS 40dB DOWN
AT VS = ±12V ALL HARMONICS 50dB DOWN
LT1228 • TA15
This oscillator uses the transconductance amplifier as a
negative resistor to cause oscillation. A negative resistor
results when the positive input of the transconductance
amplifier is driven and the output is returned to it. In this
example a voltage divider is used to lower the signal level
at the positive input for less distortion. The negative
resistor will not DC bias correctly unless the output of the
transconductance amplifier drives a very low resistance.
Here it sees an inductor to ground so the gain at DC is zero.
The oscillator needs negative resistance to start and that
is provided by the 4.3k resistor to Pin 5. As the output level
rises it turns on the PNP transistor and in turn the NPN
which steals current from the transconductance amplifier
bias input.
LT1228
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TYPICAL APPLICATI
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Filters
Single Pole Low/High/Allpass Filter
VIN
LOWPASS
INPUT
R3A
1k
3
R3
120Ω
2
+
1
gm
–
C
330pF
5
ISET
+
8
VOUT
–
RF
1k
RG
1k
VIN
HIGHPASS
INPUT
6
CFA
R2A
1k
R2
120Ω
fC =
R2
R +1
10
I
× SET × F
×
RG
R2 + R2A
C
2π
fC = 109 ISET FOR THE VALUES SHOWN
LT1228 • TA16
Allpass Filter Phase Response
PHASE SHIFT (DEGREES)
0
1mA SET CURRENT
–45
–90
–135
100µA SET CURRENT
–180
10k
100k
1M
10M
FREQUENCY (Hz)
LT1228 • TA17
Using the variable transconductance of the LT1228 to
make variable filters is easy and predictable. The most
straight forward way is to make an integrator by putting a
capacitor at the output of the transconductance amp and
buffering it with the current feedback amplifier. Because
the input bias current of the current feedback amplifier
must be supplied by the transconductance amplifier, the
set current should not be operated below 10µA. This limits
the filters to about a 100:1 tuning range.
The Single Pole circuit realizes a single pole filter with a
corner frequency (fC) proportional to the set current. The
values shown give a 100kHz corner frequency for 100µA
set current. The circuit has two inputs, a lowpass filter
input and a highpass filter input. To make a lowpass filter,
ground the highpass input and drive the lowpass input.
Conversely for a highpass filter, ground the lowpass input
and drive the highpass input. If both inputs are driven, the
result is an allpass filter or phase shifter. The allpass has
flat amplitude response and 0° phase shift at low frequencies, going to –180° at high frequencies. The allpass filter
has –90° phase shift at the corner frequency.
17
LT1228
UO
TYPICAL APPLICATI
S
Voltage Controlled State Variable Filter
+
1k
LT1006
10k
–
2N3906
VC
100pF
180Ω
51k
3k
–5V
5V
3.3k
3
VIN
3k
7
+
100Ω
5
1
gm
2
–
+
6
CFA
4
8
18pF
BANDPASS
OUTPUT
–
–5V
1k
3.3k
3.3k
100Ω
5V
3
7
+
5
gm
100Ω
2
–
1
+
6
CFA
4
18pF
–5V
8
LOWPASS
OUTPUT
–
1k
3.3k
fO = 100kHz AT VC = 0V
fO = 200kHz AT VC = 1V
fO = 400kHz AT VC = 2V
fO = 800kHz AT VC = 3V
fO = 1.6MHz AT VC = 4V
The state variable filter has both lowpass and bandpass
outputs. Each LT1228 is configured as a variable integrator whose frequency is set by the attenuators, the capacitors and the set current. Because the integrators have both
positive and negative inputs, the additional op amp normally required is not needed. The input attenuators set the
circuit up to handle 3VP–P signals.
The set current is generated with a simple circuit that gives
logarithmic voltage to current control. The two PNP transistors should be a matched pair in the same package for
18
LT1228 • TA18
best accuracy. If discrete transistors are used, the 51k
resistor should be trimmed to give proper frequency
response with VC equal zero. The circuit generates 100µA
for VC equal zero volts and doubles the current for every
additional volt. The two 3k resistors divide the current
between the two LT1228s. Therefore the set current of
each amplifier goes from 50µA to 800µA for a control
voltage of 0V to 4V. The resulting filter is at 100kHz for VC
equal zero, and changes it one octave/V of control input.
LT1228
U
PACKAGE DESCRIPTIO
J8 Package
8-Lead CERDIP (Narrow .300 Inch, Hermetic)
(Reference LTC DWG # 05-08-1110)
0.300 BSC
(0.762 BSC)
CORNER LEADS OPTION
(4 PLCS)
0.008 – 0.018
(0.203 – 0.457)
0° – 15°
0.015 – 0.060
(0.381 – 1.524)
0.023 – 0.045
(0.584 – 1.143)
HALF LEAD
OPTION
0.045 – 0.068
(1.143 – 1.727)
FULL LEAD
OPTION
0.405
(10.287)
MAX
0.005
(0.127)
MIN
8
NOTE: LEAD DIMENSIONS APPLY TO SOLDER
DIP/PLATE OR TIN PLATE LEADS
0.014 – 0.026
(0.360 – 0.660)
5
0.025
(0.635)
RAD TYP
0.220 – 0.310
(5.588 – 7.874)
1
0.045 – 0.065
(1.143 – 1.651)
6
7
2
3
4
0.125
3.175
MIN
0.100
(2.54)
BSC
0.200
(5.080)
MAX
J8 1298
OBSOLETE PACKAGE
N8 Package
8-Lead PDIP (Narrow .300 Inch)
(Reference LTC DWG # 05-08-1510)
0.300 – 0.325
(7.620 – 8.255)
0.045 – 0.065
(1.143 – 1.651)
0.065
(1.651)
TYP
0.009 – 0.015
(0.229 – 0.381)
(
0.400*
(10.160)
MAX
0.130 ± 0.005
(3.302 ± 0.127)
8
7
6
5
1
2
3
4
0.255 ± 0.015*
(6.477 ± 0.381)
+0.035
0.325 –0.015
+0.889
8.255
–0.381
)
0.125
(3.175) 0.020
MIN
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
0.100
(2.54)
BSC
N8 1098
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
S8 Package
8-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.053 – 0.069
(1.346 – 1.752)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
8
7
6
5
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
SO8 1298
1
2
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
3
4
19
LT1228
UO
TYPICAL APPLICATI
S
RF AGC Amplifier (Leveling Loop)
15V
RF INPUT
0.6VRMS to 1.3VRMS
25MHz
10k
3
7
+
1
gm
100Ω
2
–
+
5
OUTPUT
2VP–P
CFA
300Ω
8
–
4
470Ω
0.01µF
10k
–15V
10k
4pF
10Ω
0.01µF
10k
15V
–
10k
100k
4.7k
–15V
A3
LT1006
AMPLITUDE
ADJUST
+
1N4148’s
COUPLE THERMALLY
LT1004
1.2V
–15V
LT1228 • TA20
Inverting Amplifier with DC Output Less Than 5mV
Amplitude Modulator
5V
2
7
–
3
1
gm
3
5
+
4
4.7µF
+
V+
+
100µF
R5
CFA
8
VO
RF
1k
VS = ±5V, R5 = 3.6k
RG
VS = ±15V, R5 = 13.6k
1k
VOUT MUST BE LESS THAN
200mVP–P FOR LOW OUTPUT OFFSET
VIN
BW = 30Hz TO 20MHz
INCLUDES DC
1
gm
6
–
V–
7
+
+
2
CARRIER
INPUT
30mV
+
5
–
4
CFA
10k
1k
8
–
6
VOUT
0dBm(230mV) AT
MODULATION = 0V
RF
750Ω
4.7µF +
RG
750Ω
–5V
MODULATION
INPUT ≤ 8VP–P
LT1228 • TA21
LT1228 • TA22
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1227
140MHz Current Feedback Amplifier
1100V/µs Slew Rate, 0.01% Differential Gain, 0.03% Differential Phase
LT1251/LT1256
40MHz Video Fader
Accurate Linear Gain Control: ±1% Typ, ±3% Max
LT1399
400MHz Current Feedback Amplifier
800V/µs Slew Rate, 80mA Output Current
20
Linear Technology Corporation
1228fbs sn1228 LT/CP 0801 1.5K REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
 LINEAR TECHNOLOGY CORPORATION 1994
This datasheet has been download from:
www.datasheetcatalog.com
Datasheets for electronics components.
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