LT1228 100MHz Current Feedback Amplifier with DC Gain Control DESCRIPTIO U FEATURES ■ ■ ■ ■ ■ ■ The LT®1228 makes it easy to electronically control the gain of signals from DC to video frequencies. The LT1228 implements gain control with a transconductance amplifier (voltage to current) whose gain is proportional to an externally controlled current. A resistor is typically used to convert the output current to a voltage, which is then amplified with a current feedback amplifier. The LT1228 combines both amplifiers into an 8-pin package, and operates on any supply voltage from 4V (±2V) to 30V (±15V). A complete differential input, gain controlled amplifier can be implemented with the LT1228 and just a few resistors. Very Fast Transconductance Amplifier Bandwidth: 75MHz gm = 10 × ISET Low THD: 0.2% at 30mVRMS Input Wide ISET Range: 1µA to 1mA Very Fast Current Feedback Amplifier Bandwidth: 100MHz Slew Rate: 1000V/µs Output Drive Current: 30mA Differential Gain: 0.04% Differential Phase: 0.1° High Input Impedance: 25MΩ, 6pF Wide Supply Range: ±2V to ±15V Inputs Common Mode to Within 1.5V of Supplies Outputs Swing Within 0.8V of Supplies Supply Current: 7mA The LT1228 transconductance amplifier has a high impedance differential input and a current source output with wide output voltage compliance. The transconductance, gm, is set by the current that flows into Pin 5, ISET. The small signal gm is equal to ten times the value of ISET and this relationship holds over several decades of set current. The voltage at Pin 5 is two diode drops above the negative supply, Pin 4. U APPLICATIO S ■ ■ ■ ■ ■ ■ Video DC Restore (Clamp) Circuits Video Differential Input Amplifiers Video Keyer/Fader Amplifiers AGC Amplifiers Tunable Filters Oscillators The LT1228 current feedback amplifier has very high input impedance and therefore it is an excellent buffer for the output of the transconductance amplifier. The current feedback amplifier maintains its wide bandwidth over a wide range of voltage gains making it easy to interface the transconductance amplifier output to other circuitry. The current feedback amplifier is designed to drive low impedance loads, such as cables, with excellent linearity at high frequencies. , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATIO Frequency Response 6 Differential Input Variable Gain Amp 15V – 3 + 1 gm R2A 10k 2 – 4 R3 100Ω R2 100Ω + R4 1.24k R6 6.19k 6 CFA 8 ISET 4.7µF + 5 –15V ISET = 1mA –3 7 R5 10k R1 270Ω – RG 10Ω VOUT GAIN (dB) + VIN 0 4.7µF + R3A 10k VS = ±15V RL = 100Ω 3 –6 –9 –12 ISET = 300µA –15 RF 470Ω –18 –21 HIGH INPUT RESISTANCE EVEN WHEN POWER IS OFF –18dB < GAIN < 2dB VIN ≤ 3VRMS LT1228 • TA01 –24 100k ISET = 100µA 1M 10M 100M FREQUENCY (Hz) LT1228 • TA02 1 LT1228 U U RATI GS W W W W AXI U U ABSOLUTE PACKAGE/ORDER I FOR ATIO (Note 1) ORDER PART NUMBER LT1228CN8 LT1228CS8 TOP VIEW Supply Voltage ...................................................... ±18V Input Current, Pins 1, 2, 3, 5, 8 (Note 8) ............ ±15mA Output Short Circuit Duration (Note 2) ......... Continuous Operating Temperature Range LT1228C ................................................ 0°C to 70°C LT1228M (OBSOLETE) .............. –55°C to 125°C Storage Temperature Range ..................–65°C to 150°C Junction Temperature Plastic Package .............................................. 150°C Ceramic Package (OBSOLETE) ................ 175°C Lead Temperature (Soldering, 10 sec).................. 300°C IOUT 1 –IN 2 +IN 3 6 VOUT V– 4 5 ISET + – gm 8 GAIN 7 V+ S8 PART MARKING N8 PACKAGE S8 PACKAGE 8-LEAD PLASTIC DIP 8-LEAD PLASTIC SOIC TJ MAX = 150°C, θJA = 100°C/W (N) TJ MAX = 150°C, θJA = 150°C/W (S) 1228 ORDER PART NUMBER LT1228MJ8 LT1228CJ8 J8 PACKAGE 8-LEAD CERAMIC DIP TJ MAX = 175°C, θJA = 100°C/W (J) OBSOLETE PACKAGE Consider the N8 or S8 Packages for Alternate Source. Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ±5V ≤ VS ≤ ±15V, ISET = 0µA, VCM = 0V unless otherwise noted. SYMBOL PARAMETER CONDITIONS VOS Input Offset Voltage TA = 25°C MIN TYP MAX UNITS ±3 ±10 ±15 mV mV ● Input Offset Voltage Drift IIN+ Noninverting Input Current ±0.3 ±3 ±10 µA µA ±10 ±65 ±100 µA µA ● IIN– Inverting Input Current TA = 25°C ● en Input Noise Voltage Density in Input Noise Current Density f = 1kHz, RF = 1k, RG = 10Ω, RS = 10k RIN Input Resistance VIN = ±13V, VS = ±15V VIN = ±3V, VS = ±5V f = 1kHz, RF = 1k, RG = 10Ω, RS = 0Ω CIN Input Capacitance (Note 3) VS = ±5V Input Voltage Range VS = ±15V, TA = 25°C ● ● ● VS = ±5V, TA = 25°C ● CMRR Common Mode Rejection Ratio Inverting Input Current Common Mode Rejection PSRR 2 Power Supply Rejection Ratio VS = ±15V, VCM = ±13V, TA = 25°C VS = ±15V, VCM = ±12V VS = ±5V, VCM = ±3V, TA = 25°C VS = ±5V, VCM = ±2V VS = ±15V, VCM = ±13V, TA = 25°C VS = ±15V, VCM = ±12V VS = ±5V, VCM = ±3V, TA = 25°C VS = ±5V, VCM = ±2V VS = ±2V to ±15V, TA = 25°C VS = ±3V to ±15V ● ● 6 nV/√Hz 1.4 pV/√Hz 2 2 25 25 MΩ MΩ 6 pF ±13 ±12 ±3 ±2 ±13.5 V V V V 55 55 55 55 69 ±3.5 2.5 2.5 ● 60 60 dB dB dB dB 69 ● ● µV/°C 10 ● TA = 25°C 80 10 10 10 10 µA/V µA/V µA/V µA/V dB dB LT1228 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ±5V ≤ VS ≤ ±15V, ISET = 0µA, VCM = 0V unless otherwise noted. SYMBOL PARAMETER CONDITIONS Noninverting Input Current Power Supply Rejection VS = ±2V to ±15V, TA = 25°C VS = ±3V to ±15V MIN ● Inverting Input Current Power Supply Rejection VS = ±2V to ±15V, TA = 25°C VS = ±3V to ±15V ● AV Large-Signal Voltage Gain VS = ±15V, VOUT = ±10V, RLOAD = 1k VS = ±5V, VOUT = ±2V, RLOAD = 150Ω ● ● 55 55 65 65 dB dB ROL Transresistance, ∆VOUT/∆IIN– VS = ±15V, VOUT = ±10V, RLOAD = 1k VS = ±5V, VOUT = ±2V, RLOAD = 150Ω ● ● 100 100 200 200 kΩ kΩ VOUT Maximum Output Voltage Swing VS = ±15V, RLOAD = 400Ω, TA = 25°C ±13.5 ● ±12 ±10 ±3 ±2.5 30 25 65 ● 125 125 mA mA 6 11 mA ● VS = ±5V, RLOAD = 150Ω, TA = 25°C IOUT Maximum Output Current RLOAD = 0Ω, TA = 25°C MAX 10 50 50 UNITS nA/V nA/V 0.1 5 5 µA/V µA/V V V V V ±3.7 Is Supply Current SR Slew Rate (Notes 4 and 6) TA = 25°C 500 V/µs SR Slew Rate 3500 V/µs tr Rise Time (Notes 5 and 6) VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 400Ω TA = 25°C BW Small-Signal Bandwidth VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω 100 MHz tr Small-Signal Rise Time 3.5 ns Propagation Delay VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω 3.5 ns Small-Signal Overshoot VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω 15 % Settling Time 0.1%, VOUT = 10V, RF =1k, RG= 1k, RL =1k 45 ns Differential Gain (Note 7) VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k 0.01 % 0.01 DEG 0.04 % 0.1 DEG ts Differential Phase (Note 7) Differential Gain (Note 7) Differential Phase (Note 7) VOUT = 0V, ISET = 0V TYP ● 300 10 VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω 20 ns ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ±5V ≤ VS ≤ ±15V, ISET = 100µA, VCM = 0V unless otherwise noted. SYMBOL PARAMETER CONDITIONS VOS Input Offset Voltage ISET = 1mA, TA = 25°C MIN TYP MAX UNITS ±0.5 ±5 ±10 mV mV ● Input Offset Voltage Drift IOS Input Offset Current TA = 25°C 40 200 500 nA nA 0.4 1 5 µA µA ● IB Input Bias Current µV/°C 10 ● TA = 25°C ● en Input Noise Voltage Density f = 1kHz 20 nV/√Hz RIN Input Resistance-Differential Mode VIN ≈ ±30mV ● 30 200 kΩ Input Resistance-Common Mode VS = ±15V, VCM = ±12V VS = ±5V, VCM = ± 2V ● ● 50 50 1000 1000 MΩ MΩ 3 pF CIN Input Capacitance 3 LT1228 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ±5V ≤ VS ≤ ±15V, ISET = 100µA, VCM = 0V unless otherwise noted. SYMBOL CMRR PSRR gm PARAMETER CONDITIONS Input Voltage Range VS = ±15V, TA = 25°C VS = ±15V VS = ±5V, TA = 25°C VS = ±5V Common Mode Rejection Ratio Power Supply Rejection Ratio Transconductance MIN VS = ±15V, VCM = ±13V, TA = 25°C VS = ±15V, VCM = ±12V VS = ±5V, VCM = ±3V, TA = 25°C VS = ±5V, VCM = ±2V VS = ±2V to ±15V, TA = 25°C VS = ±3V to ±15V ● ● Maximum Output Current ISET = 100µA 60 60 100 ● 0.75 1.00 IOL Output Leakage Current ISET = 0µA (+IIN of CFA), TA = 25°C ● V V V V 100 dB dB dB dB 100 dB dB 1.25 – 0.33 70 UNITS ±4 ● ● IOUT MAX ±14 60 60 60 60 ● ISET = 100µA, IOUT = ±30µA, TA = 25°C Transconductance Drift ±13 ±12 ±3 ±2 TYP µA/mV %/°C 100 130 µA 0.3 3 10 µA µA ● VOUT Maximum Output Voltage Swing VS = ±15V , R1 = ∞ VS = ±5V , R1 = ∞ ● ● ±13 ±3 ±14 ±4 RO Output Resistance VS = ±15V, VOUT = ±13V VS = ±5V, VOUT = ±3V ● ● 2 2 8 8 MΩ MΩ Output Capacitance (Note 3) VS = ±5V 6 pF IS Supply Current, Both Amps ISET = 1mA THD Total Harmonic Distortion VIN = 30mVRMS at 1kHz, R1 = 100k 0.2 % BW Small-Signal Bandwidth R1 = 50Ω, ISET = 500µA 80 MHz tr Small-Signal Rise Time R1 = 50Ω, ISET = 500µA, 10% to 90% 5 ns Propagation Delay R1 = 50Ω, ISET = 500µA, 50% to 50% 5 ns Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: A heat sink may be required depending on the power supply voltage. Note 3: This is the total capacitance at Pin 1. It includes the input capacitance of the current feedback amplifier and the output capacitance of the transconductance amplifier. Note 4: Slew rate is measured at ±5V on a ±10V output signal while operating on ±15V supplies with RF = 1k, RG = 110Ω and RL = 400Ω. The slew rate is much higher when the input is overdriven, see the applications section. 4 ● 9 V V 15 mA Note 5: Rise time is measured from 10% to 90% on a ±500mV output signal while operating on ±15V supplies with RF = 1k, RG = 110Ω and RL = 100Ω. This condition is not the fastest possible, however, it does guarantee the internal capacitances are correct and it makes automatic testing practical. Note 6: AC parameters are 100% tested on the ceramic and plastic DIP packaged parts (J and N suffix) and are sample tested on every lot of the SO packaged parts (S suffix). Note 7: NTSC composite video with an output level of 2V. Note 8: Back to back 6V Zener diodes are connected between Pins 2 and 3 for ESD protection. LT1228 U W TYPICAL PERFOR A CE CHARACTERISTICS Transconductance Amplifier, Pins 1, 2, 3 & 5 100 VS = ±15V Small-Signal Transconductance and Set Current vs Bias Voltage 100 R1 = 100Ω TRANSCONDUCTANCE (µA/mV) 10 R1 = 10k 1 10000 VS = ±2V TO ±15V TA = 25°C 2.0 1000 10 100 1 0.1 10 0.01 1.0 R1 = 100k 100 1000 0.001 0.9 1.0 SET CURRENT (µA) 1.1 1.2 0.1 1.5 25°C 0.8 0.6 125°C 0.4 0 –200 –150 –100 –50 0.1 100 150 200 LT1228 • TPC03 Input Common Mode Limit vs Temperature V+ VS = ±2V TO ±15V TA = 25°C ISET = 1mA 100 ISET = 100µA V + = 2V TO 15V –0.5 COMMON MODE RANGE (V) SPOT NOISE (pA/√Hz) ISET = 100µA 50 0 INPUT VOLTAGE (mVDC) ISET = 1mA –1.0 –1.5 –2.0 2.0 1.5 V – = –2V TO –15V 1.0 0.5 0.01 V– –50 10 100 10 10 1000 100 INPUT VOLTAGE (mVP–P) 10k 1k 100k 50 75 125 Output Saturation Voltage vs Temperature V+ OUTPUT SATURATION VOLTAGE (V) 0.9 CONTROL PATH GAIN (µA/µA) 100 LT1228 • TPC06 1.0 ∆IOUT ∆ISET 25 TEMPERATURE (°C) Small-Signal Control Path Gain vs Input Voltage VS = ±2V TO ±15V VIN = 200mV (PIN 2 TO 3) 0 LT1228 • TPC05 Small-Signal Control Path Bandwidth vs Set Current 10 –25 FREQUENCY (Hz) LT1228 • TPC04 –3dB BANDWIDTH (MHz) 1.0 LT1228 • TPC02 1000 1 100 –55°C 1.2 Spot Output Noise Current vs Frequency VS = ±15V 1 1.4 BIAS VOLTAGE, PIN 5 TO 4, (V) Total Harmonic Distortion vs Input Voltage OUTPUT DISTORTION (%) 1.4 1.3 LT1228 • TPC01 10 1.6 0.2 0.1 10 V S = ±2V TO ±15V ISET = 100µA 1.8 SET CURRENT (µA) –3dB BANDWIDTH (MHz) R1 = 1k Small-Signal Transconductance vs DC Input Voltage TRANSCONDUCTANCE (µA/mV) Small-Signal Bandwidth vs Set Current 0.8 0.7 0.6 0.5 ∆IOUT ∆ISET 0.4 0.3 0.2 –0.5 –1.0 ±2V ≤ VS ≤ ±15V R1 = ∞ +1.0 +0.5 0.1 1 0 10 100 1000 SET CURRENT (µA) 0 40 80 120 160 200 INPUT VOLTAGE, PIN 2 TO 3, (mVDC) LT1228 • TPC07 LT1228 • TPC08 V– –50 –25 0 25 50 75 100 125 TEMPERATURE (°C) LT1228 • TPC09 5 LT1228 U W TYPICAL PERFOR A CE CHARACTERISTICSCurrent Feedback Amplifier, Pins 1, 6, 8 Voltage Gain and Phase vs Frequency, Gain = 6dB 160 6 90 140 GAIN 135 4 180 3 225 2 VS = ±15V RL = 100Ω RF = 750Ω RF = 500Ω 120 RF = 750Ω 100 80 RF = 1k 60 40 RF = 2k 20 –1 –2 0.1 160 0 1 10 100 2 4 6 8 10 14 12 0 160 90 140 135 18 180 17 225 16 VS = ±15V RL = 100Ω RF = 750Ω PHASE SHIFT (DEG) VOLTAGE GAIN (dB) GAIN 45 19 10 100 RF = 250Ω 80 RF = 500Ω RF = 750Ω 60 RF = 1k 40 RF = 2k 2 4 6 8 10 14 12 16 0 18 2 4 –3dB BANDWIDTH (MHz) RF = 500Ω 10 RF = 1k 8 6 RF = 2k 4 0 14 12 16 18 RF = 500Ω 12 RF = 1k 10 8 RF = 2k 6 4 2 0 0 2 4 6 8 10 12 14 16 18 SUPPLY VOLTAGE (±V) LT1228 • TPC16 10 –3dB Bandwidth vs Supply Voltage, Gain = 100, RL = 1kΩ 14 12 2 FREQUENCY (MHz) 8 LT1228 • TPC15 14 32 0.1 100 6 SUPPLY VOLTAGE (±V) 16 PHASE SHIFT (DEG) VOLTAGE GAIN (dB) RF = 2k 20 90 33 6 RF = 1k 40 18 225 10 RF = 750Ω 60 16 37 VS = ±15V RL = 100Ω RF = 750Ω RF = 500Ω 80 45 180 1 RF = 250Ω 100 18 38 18 120 0 135 16 140 LT1228 • TPC14 39 34 14 12 PEAKING ≤ 0.5dB PEAKING ≤ 5dB –3dB Bandwidth vs Supply Voltage, Gain = 100, RL = 100Ω 36 10 0 0 Voltage Gain and Phase vs Frequency, Gain = 40dB GAIN 8 –3dB Bandwidth vs Supply Voltage, Gain = 10, RL = 1kΩ SUPPLY VOLTAGE (±V) PHASE 6 LT1228 • TPC12 160 LT1228 • TPC13 35 4 180 FREQUENCY (MHz) 40 2 SUPPLY VOLTAGE (±V) 120 100 42 RF = 2k RF = 1k 0 0 1 40 18 PEAKING ≤ 0.5dB PEAKING ≤ 5dB 20 13 41 16 180 –3dB BANDWIDTH (MHz) 21 12 0.1 PEAKING ≤ 0.5dB PEAKING ≤ 5dB 60 –3dB Bandwidth vs Supply Voltage, Gain = 10, RL = 100Ω PHASE 14 80 LT1228 • TPC11 Voltage Gain and Phase vs Frequency, Gain = 20dB 15 100 SUPPLY VOLTAGE (±V) LT1228 • TPC10 20 RF = 750Ω 120 0 0 FREQUENCY (MHz) 22 RF = 500Ω 140 20 –3dB BANDWIDTH (MHz) 0 180 PEAKING ≤ 0.5dB PEAKING ≤ 5dB –3dB BANDWIDTH (MHz) 1 PHASE SHIFT (DEG) 5 –3dB Bandwidth vs Supply Voltage, Gain = 2, RL = 1k –3dB BANDWIDTH (MHz) 180 45 PHASE –3dB BANDWIDTH (MHz) 0 7 8 VOLTAGE GAIN (dB) –3dB Bandwidth vs Supply Voltage, Gain = 2, RL = 100Ω 0 2 4 6 8 10 12 14 16 18 SUPPLY VOLTAGE (±V) LT1228 • TPC17 LT1228 • TPC18 LT1228 U W TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6, 8 Maximum Capacitive Load vs Feedback Resistor Total Harmonic Distortion vs Frequency 0.10 VS = ±15V 100 RL = 1k PEAKING ≤ 5dB GAIN = 2 10 1 0.01 VO = 7VRMS 1 2 10 100 1k FEEDBACK RESISTOR (kΩ) 10k –1.5 –2.0 2.0 V – = –2V TO –15V 1.0 0.5 Output Short-Circuit Current vs Temperature 25 50 75 100 70 –0.5 V– 0 –1.0 RL = ∞ ±2V ≤ VS ≤ ±15V 1.0 0.5 V– –50 –25 125 TEMPERATURE (°C) 0 25 75 50 100 60 50 40 30 –50 –25 125 0 TEMPERATURE (°C) 75 100 125 150 175 LT1228 • TPC24 Power Supply Rejection vs Frequency Output Impedance vs Frequency 80 100 VS = ±15V RL = 100Ω RF = RG = 750Ω 60 POSITIVE 40 NEGATIVE 20 OUTPUT IMPEDANCE (Ω) POWER SUPPLY REJECTION (dB) 100 en 50 LT1228 • TPC23 Spot Noise Voltage and Current vs Frequency 10 25 TEMPERATURE (°C) LT1228 • TPC22 –in 100 LT1228 • TPC21 OUTPUT SHORT-CIRCUIT CURRENT (mA) V + = 2V TO 15V –50 –25 10 FREQUENCY (MHz) V+ OUTPUT SATURATION VOLTAGE (V) COMMON MODE RANGE (V) 1 Output Saturation Voltage vs Temperature V+ SPOT NOISE (nV/√Hz OR pA/√Hz) –70 100k LT1228 • TPC20 Input Common Mode Limit vs Temperature 1.5 3rd –50 FREQUENCY (Hz) LT1228 • TPC19 –1.0 –40 –60 3 –0.5 2nd VO = 1VRMS 0.001 0 VS = ±15V VO = 2VP–P RL = 100Ω RF = 750Ω AV = 10dB –30 DISTORTION (dBc) VS = ±5V 1k –20 VS = ±15V RL = 400Ω RF = RG = 750Ω TOTAL HARMONIC DISTORTION (%) CAPACITIVE LOAD (pF) 10k 2nd and 3rd Harmonic Distortion vs Frequency VS = ±15V 10 1.0 RF = RG = 2k RF = RG = 750Ω 0.1 0.01 +in 1 10 100 1k 10k 100k FREQUENCY (Hz) 0 10k 100k 1M 10M 100M FREQUENCY (Hz) LT1228 • TPC25 0.001 10k 100k 1M 10M 100M FREQUENCY (Hz) LT1228 • TPC26 LT1228 • TPC27 7 LT1228 U W TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6 & 8 Settling Time to 10mV vs Output Step Settling Time to 1mV vs Output Step INVERTING 4 4 OUTPUT STEP (V) 6 2 NONINVERTING 8 6 VS = ±15V RF = RG = 1k 0 10 –2 –4 9 8 INVERTING SUPPLY CURRENT (mA) NONINVERTING 8 OUTPUT STEP (V) Supply Current vs Supply Voltage 10 10 2 VS = ±15V RF = RG = 1k 0 –2 –4 INVERTING –6 –6 –8 NONINVERTING 0 20 40 60 80 100 5 125°C 4 175°C 3 1 –10 –10 25°C 6 2 NONINVERTING –8 INVERTING –55°C 7 0 4 8 12 16 20 SETTLING TIME (µs) SETTLING TIME (ns) 0 0 2 4 6 8 10 LT1228 • TPC29 LT1228 • TPC28 W W 7 V+ BIAS ISET –IN 3 2 IOUT 8 GAIN 6 VOUT 1 5 4 V– LT1228 • TA03 8 14 16 18 LT1228 • TPC30 SI PLIFIED SCHE ATIC +IN 12 SUPPLY VOLTAGE (±V) LT1228 U W U UO APPLICATI S I FOR ATIO The LT1228 contains two amplifiers, a transconductance amplifier (voltage-to-current) and a current feedback amplifier (voltage-to-voltage). The gain of the transconductance amplifier is proportional to the current that is externally programmed into Pin 5. Both amplifiers are designed to operate on almost any available supply voltage from 4V (±2V) to 30V (±15V). The output of the transconductance amplifier is connected to the noninverting input of the current feedback amplifier so that both fit into an eight pin package. Resistance Controlled Gain If the set current is to be set or varied with a resistor or potentiometer it is possible to use the negative temperature coefficient at Pin 5 (with respect to Pin 4) to compensate for the negative temperature coefficient of the transconductance. The easiest way is to use an LT1004-2.5, a 2.5V reference diode, as shown below: Temperature Compensation of gm with a 2.5V Reference R ISET TRANSCONDUCTANCE AMPLIFIER The LT1228 transconductance amplifier has a high impedance differential input (Pins 2 and 3) and a current source output (Pin 1) with wide output voltage compliance. The voltage to current gain or transconductance (gm) is set by the current that flows into Pin 5, ISET. The voltage at Pin 5 is two forward biased diode drops above the negative supply, Pin 4. Therefore the voltage at Pin 5 (with respect to V –) is about 1.2V and changes with the log of the set current (120mV/decade), see the characteristic curves. The temperature coefficient of this voltage is about –4mV/°C (–3300ppm/°C) and the temperature coefficient of the logging characteristic is 3300ppm/°C. It is important that the current into Pin 5 be limited to less than 15mA. THE LT1228 WILL BE DESTROYED IF PIN 5 IS SHORTED TO GROUND OR TO THE POSITIVE SUPPLY. A limiting resistor (2k or so) should be used to prevent more than 15mA from flowing into Pin 5. The small-signal transconductance (gm) is equal to ten times the value of ISET (in mA/mV) and this relationship holds over many decades of set current (see the characteristic curves). The transconductance is inversely proportional to absolute temperature (–3300ppm/°C). The input stage of the transconductance amplifier has been designed to operate with much larger signals than is possible with an ordinary diff-amp. The transconductance of the input stage varies much less than 1% for differential input signals over a ±30 mV range (see the characteristic curve Small-Signal Transconductance vs DC Input Voltage). gm Vbe 4 5 R 2.5V 2Eg ISET Vbe LT1004-2.5 V– LT1228 • TA04 The current flowing into Pin 5 has a positive temperature coefficient that cancels the negative coefficient of the transconductance. The following derivation shows why a 2.5V reference results in zero gain change with temperature: q ISET × = 10 • ISET kT 3.87 cT n akT = Eg – where a = In q Ic Since g m = and V be ( ≈ 19.4 at 27°C c = 0.001, n = 3, Ic = 100µA ) Eg is about 1.25V so the 2.5V reference is 2Eg. Solving the loop for the set current gives: ISET = akT 2E g – 2 E g – q R or ISET = 2akT Rq 9 LT1228 W U U UO APPLICATI S I FOR ATIO Substituting into the equation for transconductance gives: a 10 = 1.94R R The temperature variation in the term “a” can be ignored since it is much less than that of the term “T” in the equation for Vbe. Using a 2.5V source this way will maintain the gain constant within 1% over the full temperature range of –55°C to 125°C. If the 2.5V source is off by 10%, the gain will vary only about ±6% over the same temperature range. gm = We can also temperature compensate the transconductance without using a 2.5V reference if the negative power supply is regulated. A Thevenin equivalent of 2.5V is generated from two resistors to replace the reference. The two resistors also determine the maximum set current, approximately 1.1V/RTH. By rearranging the Thevenin equations to solve for R4 and R6 we get the following equations in terms of RTH and the negative supply, VEE. R4 = R TH 2.5V 1 – V EE and R6 = R THV EE 2.5V is two diode drops above the negative supply, a single resistor from the control voltage source to Pin 5 will suffice in many applications. The control voltage is referenced to the negative supply and has an offset of about 900mV. The conversion will be monotonic, but the linearity is determined by the change in the voltage at Pin 5 (120mV per decade of current). The characteristic is very repeatable since the voltage at Pin 5 will vary less than ±5% from part to part. The voltage at Pin 5 also has a negative temperature coefficient as described in the previous section. When the gain of several LT1228s are to be varied together, the current can be split equally by using equal value resistors to each Pin 5. For more accurate (and linear) control, a voltage-tocurrent converter circuit using one op amp can be used. The following circuit has several advantages. The input no longer has to be referenced to the negative supply and the input can be either polarity (or differential). This circuit works on both single and split supplies since the input voltage and the Pin 5 voltage are independent of each other. The temperature coefficient of the output current is set by R5. R3 1M Temperature Compensation of gm with a Thevenin Voltage 1.03k R1 1M R' V1 ISET gm R6 6.19kΩ R' Vbe 4 5 V2 + R5 1k LT1006 – ISET IOUT TO PIN 5 OF LT1228 R4 1M VTH = 2.5V Vbe 50pF R1 = R2 R3 = R4 R4 1.24kΩ –15V LT1228 • TA05 Voltage Controlled Gain To use a voltage to control the gain of the transconductance amplifier requires converting the voltage into a current that flows into Pin 5. Because the voltage at Pin 5 10 R2 1M IOUT = (V1 – V2) R3 • = 1mA/V R5 R1 LT1228 • TA19 Digital control of the transconductance amplifier gain is done by converting the output of a DAC to a current flowing into Pin 5. Unfortunately most current output DACs sink rather than source current and do not have output LT1228 U W U UO APPLICATI S I FOR ATIO compliance compatible with Pin 5 of the LT1228. Therefore, the easiest way to digitally control the set current is to use a voltage output DAC and a voltage-to-current circuit. The previous voltage-to-current converter will take the output of any voltage output DAC and drive Pin 5 with a proportional current. The R, 2R CMOS multiplying DACs operating in the voltage switching mode work well on both single and split supplies with the above circuit. Transconductance Amp Small-Signal Response ISET = 500µA, R1 = 50Ω Logarithmic control is often easier to use than linear control. A simple circuit that doubles the set current for each additional volt of input is shown in the voltage controlled state variable filter application near the end of this data sheet. Transconductance Amplifier Frequency Response The bandwidth of the transconductance amplifier is a function of the set current as shown in the characteristic curves. At set currents below 100µA, the bandwidth is approximately: –3dB bandwidth = 3 • 1011 ISET The peak bandwidth is about 80MHz at 500µA. When a resistor is used to convert the output current to a voltage, the capacitance at the output forms a pole with the resistor. The best case output capacitance is about 5pF with ±15V supplies and 6pF with ±5V supplies. You must add any PC board or socket capacitance to these values to get the total output capacitance. When using a 1k resistor at the output of the transconductance amp, the output capacitance limits the bandwidth to about 25MHz. The output slew rate of the transconductance amplifier is the set current divided by the output capacitance, which is 6pF plus board and socket capacitance. For example with the set current at 1mA, the slew rate would be over 100V/µs. CURRENT FEEDBACK AMPLIFIER The LT1228 current feedback amplifier has very high noninverting input impedance and is therefore an excellent buffer for the output of the transconductance amplifier. The noninverting input is at Pin 1, the inverting input at Pin 8 and the output at Pin 6. The current feedback amplifier maintains its wide bandwidth for almost all voltage gains making it easy to interface the output levels of the transconductance amplifier to other circuitry. The current feedback amplifier is designed to drive low impedance loads such as cables with excellent linearity at high frequencies. Feedback Resistor Selection The small-signal bandwidth of the LT1228 current feedback amplifier is set by the external feedback resistors and the internal junction capacitors. As a result, the bandwidth is a function of the supply voltage, the value of the feedback resistor, the closed-loop gain and load resistor. The characteristic curves of bandwidth versus supply voltage are done with a heavy load (100Ω) and a light load (1k) to show the effect of loading. These graphs also show 11 LT1228 U W U UO APPLICATI S I FOR ATIO the family of curves that result from various values of the feedback resistor. These curves use a solid line when the response has less than 0.5dB of peaking and a dashed line for the response with 0.5dB to 5dB of peaking. The curves stop where the response has more than 5dB of peaking. Current Feedback Amp Small-Signal Response VS = ±15V, RF = RG = 750Ω, RL = 100Ω Capacitance on the Inverting Input Current feedback amplifiers want resistive feedback from the output to the inverting input for stable operation. Take care to minimize the stray capacitance between the output and the inverting input. Capacitance on the inverting input to ground will cause peaking in the frequency response (and overshoot in the transient response), but it does not degrade the stability of the amplifier. The amount of capacitance that is necessary to cause peaking is a function of the closed-loop gain taken. The higher the gain, the more capacitance is required to cause peaking. For example, in a gain of 100 application, the bandwidth can be increased from 10MHz to 17MHz by adding a 2200pF capacitor, as shown below. CG must have very low series resistance, such as silver mica. 1 VIN + 6 CFA 8 VOUT – RF 510Ω RG 5.1Ω CG LT1228 • TA08 Boosting Bandwidth of High Gain Amplifier with Capacitance On Inverting Input 49 46 CG = 4700pF 43 40 GAIN (dB) At a gain of two, on ±15V supplies with a 750Ω feedback resistor, the bandwidth into a light load is over 160MHz without peaking, but into a heavy load the bandwidth reduces to 100MHz. The loading has so much effect because there is a mild resonance in the output stage that enhances the bandwidth at light loads but has its Q reduced by the heavy load. This enhancement is only useful at low gain settings, at a gain of ten it does not boost the bandwidth. At unity gain, the enhancement is so effective the value of the feedback resistor has very little effect on the bandwidth. At very high closed-loop gains, the bandwidth is limited by the gain-bandwidth product of about 1GHz. The curves show that the bandwidth at a closed-loop gain of 100 is 10MHz, only one tenth what it is at a gain of two. CG = 2200pF 37 34 CG = 0 31 28 25 22 19 1 10 100 FREQUENCY (MHz) LT1228 • TA09 12 LT1228 U W U UO APPLICATI S I FOR ATIO Capacitive Loads The LT1228 current feedback amplifier can drive capacitive loads directly when the proper value of feedback resistor is used. The graph of Maximum Capacitive Load vs Feedback Resistor should be used to select the appropriate value. The value shown is for 5dB peaking when driving a 1k load, at a gain of 2. This is a worst case condition, the amplifier is more stable at higher gains, and driving heavier loads. Alternatively, a small resistor (10Ω to 20Ω) can be put in series with the output to isolate the capacitive load from the amplifier output. This has the advantage that the amplifier bandwidth is only reduced when the capacitive load is present and the disadvantage that the gain is a function of the load resistance. The output slew rate is set by the value of the feedback resistors and the internal capacitance. At a gain of ten with a 1k feedback resistor and ±15V supplies, the output slew rate is typically 500V/µs and –850V/µs. There is no input stage enhancement because of the high gain. Larger feedback resistors will reduce the slew rate as will lower supply voltages, similar to the way the bandwidth is reduced. Current Feedback Amp Large-Signal Response VS = ±15V, RF = 1k, RG = 110Ω, RL = 400Ω Slew Rate The slew rate of the current feedback amplifier is not independent of the amplifier gain configuration the way it is in a traditional op amp. This is because the input stage and the output stage both have slew rate limitations. The input stage of the LT1228 current feedback amplifier slews at about 100V/µs before it becomes nonlinear. Faster input signals will turn on the normally reverse biased emitters on the input transistors and enhance the slew rate significantly. This enhanced slew rate can be as much as 3500V/µs! Current Feedback Amp Large-Signal Response VS = ±15V, RF = RG = 750Ω Slew Rate Enhanced Settling Time The characteristic curves show that the LT1228 current feedback amplifier settles to within 10mV of final value in 40ns to 55ns for any output step less than 10V. The curve of settling to 1mV of final value shows that there is a slower thermal contribution up to 20µs. The thermal settling component comes from the output and the input stage. The output contributes just under 1mV/V of output change and the input contributes 300µV/V of input change. Fortunately the input thermal tends to cancel the output thermal. For this reason the noninverting gain of two configuration settles faster than the inverting gain of one. 13 LT1228 U W U UO APPLICATI S I FOR ATIO Power Supplies The LT1228 amplifiers will operate from single or split supplies from ±2V (4V total) to ±18V (36V total). It is not necessary to use equal value split supplies, however the offset voltage and inverting input bias current of the current feedback amplifier will degrade. The offset voltage changes about 350µV/V of supply mismatch, the inverting bias current changes about 2.5µA/V of supply mismatch. For example, let’s calculate the worst case power dissipation in a variable gain video cable driver operating on ±12V supplies that delivers a maximum of 2V into 150Ω. The maximum set current is 1mA. ( ) ( P D = 2VS ISMAX + 3.5ISET + VS – V OMAX [ ( )] ( ) VOMAX RL P D = 2 • 12V • 7mA + 3.5 • 1mA + 12V – 2V ) 1502VΩ = 0.252 + 0.133 = 0.385W Power Dissipation The worst case amplifier power dissipation is the total of the quiescent current times the total power supply voltage plus the power in the IC due to the load. The quiescent supply current of the LT1228 transconductance amplifier is equal to 3.5 times the set current at all temperatures. The quiescent supply current of the LT1228 current feedback amplifier has a strong negative temperature coefficient and at 150°C is less than 7mA, typically only 4.5mA. The power in the IC due to the load is a function of the output voltage, the supply voltage and load resistance. The worst case occurs when the output voltage is at half supply, if it can go that far, or its maximum value if it cannot reach half supply. The total power dissipation times the thermal resistance of the package gives the temperature rise of the die above ambient. The above example in SO-8 surface mount package (thermal resistance is 150°C/W) gives: Temperature Rise = PDθJA = 0.385W • 150°C/W = 57.75°C Therefore the maximum junction temperature is 70°C +57.75°C or 127.75°C, well under the absolute maximum junction temperature for plastic packages of 150°C. UO TYPICAL APPLICATI S Basic Gain Control The basic gain controlled amplifier is shown on the front page of the data sheet. The gain is directly proportional to the set current. The signal passes through three stages from the input to the output. First the input signal is attenuated to match the dynamic range of the transconductance amplifier. The attenuator should reduce the signal down to less than 100mV peak. The characteristic curves can be used to estimate how much distortion there will be at maximum input signal. For single ended inputs eliminate R2A or R3A. The signal is then amplified by the transconductance amplifier (gm) and referred to ground. The voltage gain of the transconductance amplifier is: 14 g m • R1 = 10 • ISET • R1 Lastly the signal is buffered and amplified by the current feedback amplifier (CFA). The voltage gain of the current feedback amplifier is: 1+ RF RG The overall gain of the gain controlled amplifier is the product of all three stages: R3 RF AV = • 10 • I • R 1 • SET 1 + R R3 + R3A G More than one output can be summed into R1 because the output of the transconductance amplifier is a current. This is the simplest way to make a video mixer. LT1228 UO TYPICAL APPLICATI S Video Fader Video DC Restore (Clamp) Circuit NOT NECESSARY IF THE SOURCE RESISTANCE IS LESS THAN 50Ω VIN1 1k 3 + 1 gm 2 1000pF LT1223 CFA – 5 100Ω + 200Ω 3 VOUT V+ 7 + – 1k – 10k –5V 10k CFA 8 VOUT – RF V– 5.1k 6 10k 5.1k VS = ±5V + 0.01µF 5 4 10k 1 gm 2 RG 5V 1k VIN2 1k 100Ω 3 gm 2 3k 5 + VIDEO INPUT 1 LOGIC INPUT – LT1228 • TA12 The video fader uses the transconductance amplifiers from two LT1228s in the feedback loop of another current feedback amplifier, the LT1223. The amount of signal from each input at the output is set by the ratio of the set currents of the two LT1228s, not by their absolute value. The bandwidth of the current feedback amplifier is inversely proportional to the set current in this configuration. Therefore, the set currents remain high over most of the pot’s range, keeping the bandwidth over 15MHz even when the signal is attenuated 20dB. The pot is set up to completely turn off one LT1228 at each end of the rotation. RESTORE 2N3906 3k LT1228 • TA13 The video restore (clamp) circuit restores the black level of the composite video to zero volts at the beginning of every line. This is necessary because AC coupled video changes DC level as a function of the average brightness of the picture. DC restoration also rejects low frequency noise such as hum. The circuit has two inputs: composite video and a logic signal. The logic signal is high except during the back porch time right after the horizontal sync pulse. While the logic is high, the PNP is off and ISET is zero. With ISET equal to zero the feedback to Pin 2 has no affect. The video input drives the noninverting input of the current feedback amplifier whose gain is set by RF and RG. When the logic signal is low, the PNP turns on and ISET goes to about 1mA. Then the transconductance amplifier charges the capacitor to force the output to match the voltage at Pin 3, in this case zero volts. This circuit can be modified so that the video is DC coupled by operating the amplifier in an inverting configuration. Just ground the video input shown and connect RG to the video input instead of to ground. 15 LT1228 UO TYPICAL APPLICATI S Single Supply Wien Bridge Oscillator resistor and the transconductance amplifier must be about 11, resulting in a set current of about 600µA at oscillation. At start-up there is no set current and therefore no attenuation for a net gain of about 11 around the loop. As the output oscillation builds up it turns on the PNP transistor which generates the set current to regulate the output voltage. 100Ω 2N3906 V+ 6V TO 30V V+ 10kΩ 470Ω + 10µF 7 3 + 5 1 gm 10kΩ 2 – + 8 4 12MHz Negative Resistance LC Oscillator 0.1µF 6 CFA V+ – RF 680Ω 51Ω 9.1k 10µF 160Ω 1000pF 8 VO 51Ω – V– 50Ω 6 750Ω 1k 160Ω 30pF 50Ω 4.3k 330Ω 2N3906 2N3904 LT1228 • TA14 In this application the LT1228 is biased for operation from a single supply. An artificial signal ground at half supply voltage is generated with two 10k resistors and bypassed with a capacitor. A capacitor is used in series with R G to set the DC gain of the current feedback amplifier to unity. The transconductance amplifier is used as a variable resistor to control gain. A variable resistor is formed by driving the inverting input and connecting the output back to it. The equivalent resistor value is the inverse of the g m. This works with the 1.8k resistor to make a variable attenuator. The 1MHz oscillation frequency is set by the Wien bridge network made up of two 1000pF capacitors and two 160Ω resistors. For clean sine wave oscillation, the circuit needs a net gain of one around the loop. The current feedback amplifier has a gain of 34 to keep the voltage at the transconductance amplifier input low. The Wien bridge has an attenuation of 3 at resonance; therefore the attenuation of the 1.8k 16 CFA 4 4.7µH f = 1MHz VO = 6dBm (450mVRMS) 2nd HARMONIC = –38dBc 3rd HARMONIC = –54 dBc FOR 5V OPERATION SHORT OUT 100Ω RESISTOR + 5 – 10µF 1000pF 1 gm 2 + + 7 + 1k RG 20Ω 1.8k 3 VO 10k VO = 10dB 0.1µF V– AT VS = ±5V ALL HARMONICS 40dB DOWN AT VS = ±12V ALL HARMONICS 50dB DOWN LT1228 • TA15 This oscillator uses the transconductance amplifier as a negative resistor to cause oscillation. A negative resistor results when the positive input of the transconductance amplifier is driven and the output is returned to it. In this example a voltage divider is used to lower the signal level at the positive input for less distortion. The negative resistor will not DC bias correctly unless the output of the transconductance amplifier drives a very low resistance. Here it sees an inductor to ground so the gain at DC is zero. The oscillator needs negative resistance to start and that is provided by the 4.3k resistor to Pin 5. As the output level rises it turns on the PNP transistor and in turn the NPN which steals current from the transconductance amplifier bias input. LT1228 UO TYPICAL APPLICATI S Filters Single Pole Low/High/Allpass Filter VIN LOWPASS INPUT R3A 1k 3 R3 120Ω 2 + 1 gm – C 330pF 5 ISET + 8 VOUT – RF 1k RG 1k VIN HIGHPASS INPUT 6 CFA R2A 1k R2 120Ω fC = R2 R +1 10 I × SET × F × RG R2 + R2A C 2π fC = 109 ISET FOR THE VALUES SHOWN LT1228 • TA16 Allpass Filter Phase Response PHASE SHIFT (DEGREES) 0 1mA SET CURRENT –45 –90 –135 100µA SET CURRENT –180 10k 100k 1M 10M FREQUENCY (Hz) LT1228 • TA17 Using the variable transconductance of the LT1228 to make variable filters is easy and predictable. The most straight forward way is to make an integrator by putting a capacitor at the output of the transconductance amp and buffering it with the current feedback amplifier. Because the input bias current of the current feedback amplifier must be supplied by the transconductance amplifier, the set current should not be operated below 10µA. This limits the filters to about a 100:1 tuning range. The Single Pole circuit realizes a single pole filter with a corner frequency (fC) proportional to the set current. The values shown give a 100kHz corner frequency for 100µA set current. The circuit has two inputs, a lowpass filter input and a highpass filter input. To make a lowpass filter, ground the highpass input and drive the lowpass input. Conversely for a highpass filter, ground the lowpass input and drive the highpass input. If both inputs are driven, the result is an allpass filter or phase shifter. The allpass has flat amplitude response and 0° phase shift at low frequencies, going to –180° at high frequencies. The allpass filter has –90° phase shift at the corner frequency. 17 LT1228 UO TYPICAL APPLICATI S Voltage Controlled State Variable Filter + 1k LT1006 10k – 2N3906 VC 100pF 180Ω 51k 3k –5V 5V 3.3k 3 VIN 3k 7 + 100Ω 5 1 gm 2 – + 6 CFA 4 8 18pF BANDPASS OUTPUT – –5V 1k 3.3k 3.3k 100Ω 5V 3 7 + 5 gm 100Ω 2 – 1 + 6 CFA 4 18pF –5V 8 LOWPASS OUTPUT – 1k 3.3k fO = 100kHz AT VC = 0V fO = 200kHz AT VC = 1V fO = 400kHz AT VC = 2V fO = 800kHz AT VC = 3V fO = 1.6MHz AT VC = 4V The state variable filter has both lowpass and bandpass outputs. Each LT1228 is configured as a variable integrator whose frequency is set by the attenuators, the capacitors and the set current. Because the integrators have both positive and negative inputs, the additional op amp normally required is not needed. The input attenuators set the circuit up to handle 3VP–P signals. The set current is generated with a simple circuit that gives logarithmic voltage to current control. The two PNP transistors should be a matched pair in the same package for 18 LT1228 • TA18 best accuracy. If discrete transistors are used, the 51k resistor should be trimmed to give proper frequency response with VC equal zero. The circuit generates 100µA for VC equal zero volts and doubles the current for every additional volt. The two 3k resistors divide the current between the two LT1228s. Therefore the set current of each amplifier goes from 50µA to 800µA for a control voltage of 0V to 4V. The resulting filter is at 100kHz for VC equal zero, and changes it one octave/V of control input. LT1228 U PACKAGE DESCRIPTIO J8 Package 8-Lead CERDIP (Narrow .300 Inch, Hermetic) (Reference LTC DWG # 05-08-1110) 0.300 BSC (0.762 BSC) CORNER LEADS OPTION (4 PLCS) 0.008 – 0.018 (0.203 – 0.457) 0° – 15° 0.015 – 0.060 (0.381 – 1.524) 0.023 – 0.045 (0.584 – 1.143) HALF LEAD OPTION 0.045 – 0.068 (1.143 – 1.727) FULL LEAD OPTION 0.405 (10.287) MAX 0.005 (0.127) MIN 8 NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE OR TIN PLATE LEADS 0.014 – 0.026 (0.360 – 0.660) 5 0.025 (0.635) RAD TYP 0.220 – 0.310 (5.588 – 7.874) 1 0.045 – 0.065 (1.143 – 1.651) 6 7 2 3 4 0.125 3.175 MIN 0.100 (2.54) BSC 0.200 (5.080) MAX J8 1298 OBSOLETE PACKAGE N8 Package 8-Lead PDIP (Narrow .300 Inch) (Reference LTC DWG # 05-08-1510) 0.300 – 0.325 (7.620 – 8.255) 0.045 – 0.065 (1.143 – 1.651) 0.065 (1.651) TYP 0.009 – 0.015 (0.229 – 0.381) ( 0.400* (10.160) MAX 0.130 ± 0.005 (3.302 ± 0.127) 8 7 6 5 1 2 3 4 0.255 ± 0.015* (6.477 ± 0.381) +0.035 0.325 –0.015 +0.889 8.255 –0.381 ) 0.125 (3.175) 0.020 MIN (0.508) MIN 0.018 ± 0.003 (0.457 ± 0.076) 0.100 (2.54) BSC N8 1098 *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm) S8 Package 8-Lead Plastic Small Outline (Narrow .150 Inch) (Reference LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.053 – 0.069 (1.346 – 1.752) 0.008 – 0.010 (0.203 – 0.254) 0°– 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.014 – 0.019 (0.355 – 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 8 7 6 5 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) SO8 1298 1 2 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 3 4 19 LT1228 UO TYPICAL APPLICATI S RF AGC Amplifier (Leveling Loop) 15V RF INPUT 0.6VRMS to 1.3VRMS 25MHz 10k 3 7 + 1 gm 100Ω 2 – + 5 OUTPUT 2VP–P CFA 300Ω 8 – 4 470Ω 0.01µF 10k –15V 10k 4pF 10Ω 0.01µF 10k 15V – 10k 100k 4.7k –15V A3 LT1006 AMPLITUDE ADJUST + 1N4148’s COUPLE THERMALLY LT1004 1.2V –15V LT1228 • TA20 Inverting Amplifier with DC Output Less Than 5mV Amplitude Modulator 5V 2 7 – 3 1 gm 3 5 + 4 4.7µF + V+ + 100µF R5 CFA 8 VO RF 1k VS = ±5V, R5 = 3.6k RG VS = ±15V, R5 = 13.6k 1k VOUT MUST BE LESS THAN 200mVP–P FOR LOW OUTPUT OFFSET VIN BW = 30Hz TO 20MHz INCLUDES DC 1 gm 6 – V– 7 + + 2 CARRIER INPUT 30mV + 5 – 4 CFA 10k 1k 8 – 6 VOUT 0dBm(230mV) AT MODULATION = 0V RF 750Ω 4.7µF + RG 750Ω –5V MODULATION INPUT ≤ 8VP–P LT1228 • TA21 LT1228 • TA22 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1227 140MHz Current Feedback Amplifier 1100V/µs Slew Rate, 0.01% Differential Gain, 0.03% Differential Phase LT1251/LT1256 40MHz Video Fader Accurate Linear Gain Control: ±1% Typ, ±3% Max LT1399 400MHz Current Feedback Amplifier 800V/µs Slew Rate, 80mA Output Current 20 Linear Technology Corporation 1228fbs sn1228 LT/CP 0801 1.5K REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1994 This datasheet has been download from: www.datasheetcatalog.com Datasheets for electronics components.
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