REI Datasheet LMX2305 550 MHz Frequency Synthesizer for RF Personal Communications The LMX2305 is a high performance frequency synthesizer with an integrated prescaler designed for RF operation up to 550 MHz. The LMX2305 contains a dual modulus prescaler which can select either a 64/65 or a 128/129 divide ratio at input frequencies of up to 550 MHz. LMX2305, which employs the digital phase lock loop technique, combined with a high quality reference oscillator and a loop ilter, provides the tuning voltage for the voltage controlled oscillator to generate a very stable, low noise local oscillator signal. Rochester Electronics Manufactured Components Quality Overview Rochester branded components are manufactured using either die/wafers purchased from the original suppliers or Rochester wafers recreated from the original IP. All recreations are done with the approval of the OCM. • ISO-9001 • AS9120 certiication • Qualiied Manufacturers List (QML) MIL-PRF-38535 • Class Q Military • Class V Space Level • Qualiied Suppliers List of Distributors (QSLD) • Rochester is a critical supplier to DLA and meets all industry and DLA standards. Parts are tested using original factory test programs or Rochester developed test solutions to guarantee product meets or exceeds the OCM data sheet. Rochester Electronics, LLC is committed to supplying products that satisfy customer expectations for quality and are equal to those originally supplied by industry manufacturers. The original manufacturer’s datasheet accompanying this document relects the performance and speciications of the Rochester manufactured version of this device. Rochester Electronics guarantees the performance of its semiconductor products to the original OEM speciications. ‘Typical’ values are for reference purposes only. Certain minimum or maximum ratings may be based on product characterization, design, simulation, or sample testing. © 2014 Rochester Electronics, LLC. All Rights Reserved 01162014 To learn more, please visit www.rocelec.com August 1996 LMX2305 PLLatinum TM 550 MHz Frequency Synthesizer for RF Personal Communications General Description Features The LMX2305 is a high performance frequency synthesizer with an integrated prescaler designed for RF operation up to 550 MHz. It is fabricated using National’s ABiC IV BiCMOS process. The LMX2305 contains a dual modulus prescaler which can select either a 64/65 or a 128/129 divide ratio at input frequencies of up to 550 MHz. LMX2305, which employs the digital phase lock loop technique, combined with a high quality reference oscillator and a loop filter, provides the tuning voltage for the voltage controlled oscillator to generate a very stable, low noise local oscillator signal. Y Serial data is transferred into the LMX2305 via a three line MICROWIRETM interface (Data, Enable, Clock). Supply voltage can range from 2.65V to 5.5V. The LMX2305 features very low current consumption, typically 4.0 mA at 2.75V. The LMX2305 is available in a TSSOP 20-pin surface mount plastic package. Y Y Y Y Y RF operation up to 550 MHz 2.65V to 5.5V operation Low current consumption: ICC e 4.0 mA (typ) at VCC e 2.75V Dual modulus prescaler: 64/65 or 128/129 Internal balanced, low leakage charge pump Small-outline, plastic, surface mount TSSOP, 0.173× wide package Applications Y Y Y Y Y Analog Cellular telephone systems (AMPS, ETACS, NMT) Portable wireless communications (PCS/PCN, cordless) Wireless local area networks (WLANs) Other wireless communication systems Pagers Block Diagram TL/W/12459– 1 TRI-STATEÉ is a registered trademark of National Semiconductor Corporation. MICROWIRETM and PLLatinumTM are trademarks of National Semiconductor Corporation. C1996 National Semiconductor Corporation TL/W/12459 RRD-B30M126/Printed in U. S. A. http://www.national.com LMX2305 PLLatinum 550 MHz Frequency Synthesizer for RF Personal Communications PRELIMINARY Connection Diagram LMX2305 TL/W/12459– 2 20-Lead (0.173× Wide) Thin Shrink Small Outline Package (TM) Order Number LMX2305TM or LMX2305TMX See NS Package Number MTC20 Pin Descriptions Pin No. Pin Name I/O Description 1 OSCIN I 3 OSCOUT O 4 VP Power supply for charge pump. Must be t VCC. 5 VCC Power supply voltage input. Input may range from 2.65V to 5.5V. Bypass capacitors should be placed as close as possible to this pin and be connected directly to the ground plane. 6 Do 7 GND 8 LD O Lock detect. Output provided to indicate when the VCO frequency is in ‘‘lock’’. When the loop is locked, the pin’s output is HIGH with narrow low pulses. 10 fIN I Prescaler input. Small signal input from the VCO. 11 CLOCK I High impedance CMOS Clock input. Data is clocked in on the rising edge, into the various counters and registers. 13 DATA I Binary serial data input. Data entered MSB first. LSB is control bit. High impedance CMOS input. 14 LE I Load enable input (with internal pull-up resistor). When LE transitions HIGH, data stored in the shift registers is loaded into the appropriate latch (control bit dependent). Clock must be low when LE toggles high or low. See Serial Data Input Timing Diagram. 15 FC I Phase control select (with internal pull-up resistor). When FC is LOW, the polarity of the phase comparator and charge pump combination is reversed. 16 BISW O Analog switch output. When LE is HIGH, the analog switch is ON, routing the internal charge pump output through BISW (as well as through Do). 17 fOUT O Monitor pin of phase comparator input. CMOS output. 18 wp O Output for external charge pump. wp is an open drain N-channel transistor and requires a pull-up resistor. 19 PWDN I Power Down (with internal pull-up resistor). PWDN e HIGH for normal operation. PWDN e LOW for power saving. Power down function is gated by the return of the charge pump to a TRI-STATE condition. 20 wr O 2,9,12 NC http://www.national.com O Oscillator input. A CMOS inverting gate input intended for connection to a crystal resonator for operation as an oscillator. The input has a VCC/2 input threshold and can be driven from an external CMOS or TTL logic gate. May also be from a reference oscillator. Oscillator output. Internal charge pump output. For connection to a loop filter for driving the input of an external VCO. Ground. Output for external charge pump. wr is a CMOS logic output. No connect. 2 Functional Block Diagram TL/W/12459– 3 Note 1: The power down function is gated by the charge pump to prevent any unwanted frequency jumps. Once the power down pin is brought low the part will go into power down mode when the charge pump reaches a TRI-STATE condition. 3 http://www.national.com Recommended Operating Conditions Absolute Maximum Ratings (Notes 1 and 2) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/Distributors for availability and specifications. Power Supply Voltage b 0.3V to a 6.5V VCC b 0.3V to a 6.5V VP Voltage on Any Pin b 0.3V to VCC a 0.3V with GND e 0V (VI) b 65§ C to a 150§ C Storage Temperature Range (TS) a 260§ C Lead Temperature (TL) (solder, 4 sec.) Power Supply Voltage VCC VP 2.65V to 5.5V VCC to a 5.5V b 40§ C to a 85§ C Operating Temperature (TA) Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics. The guaranteed specifications apply only for the test conditions listed. Note 2: This device is a high performance RF integrated circuit with an ESD rating k 2 keV and is ESD sensitive. Handling and assembly of this device should only be done at ESD workstations. Electrical Characteristics VCC e 2.75V, VP e 2.75V; b40§ C k TA k 85§ C, except as specified Typ Max Units ICC Symbol Power Supply Current Parameter Conditions Min 4 6 mA ICC-PWDN Power Down Current 30 180 mA fIN RF Input Operating Frequency 45 550 MHz fOSC Oscillator Input Operating Frequency 5 fw Phase Detector Frequency MHz MHz a6 dBm PfIN Input Sensitivity VCC e 2.65V to 5.5V VOSC Oscillator Sensitivity OSCIN VIH High-Level Input Voltage * VIL Low-Level Input Voltage * IIH High-Level Input Current (Clock, Data) VIH e VCC e 5.5V b 1.0 1.0 mA IIL Low-Level Input Current (Clock, Data) VIL e 0V, VCC e 5.5V b 1.0 1.0 mA IIH Oscillator Input Current 100 mA IIL b 10 22 10 0.5 VPP 0.7 VCC V 0.3 VCC VIH e VCC e 5.5V VIL e 0V, VCC e 5.5V b 100 V mA IIH High-Level Input Current (LE, FC) VIH e VCC e 5.5V b 1.0 1.0 mA IIL Low-Level Input Current (LE, FC) VIL e 0V, VCC e 5.5V b 100 1.0 mA *Except fIN and OSCIN http://www.national.com 4 Electrical Characteristics VCC e 2.75V, VP e 2.75V; b40§ C k TA k 85§ C, except as specified (Continued) Symbol IDo-source Parameter Charge Pump Output Current Conditions Min VDo e VP/2 Typ Max Units b 2.5 b 1.0 mA IDo-sink VDo e VP/2 IDo-Tri Charge Pump TRI-STATEÉ Current 0.5V s VDo s VP b 0.5V TA e b40§ C k TA k 85§ C VOH High-Level Output Voltage IOH e b1.0 mA** VOL Low-Level Output Voltage IOL e 1.0 mA** VOH High-Level Output Voltage (OSCOUT) IOH e b200 mA VOL Low-Level Output Voltage (OSCOUT) IOL e 200 mA IOL Open Drain Output Current (wp) VOL e 0.4V IOH Open Drain Output Current (wp) VOH e 2.75V tCS Data to Clock Set Up Time See Data Input Timing 50 tCH Data to Clock Hold Time See Data Input Timing 10 ns tCWH Clock Pulse Width High See Data Input Timing 50 ns tCWL Clock Pulse Width Low See Data Input Timing 50 ns tES Clock to Enable Set Up Time See Data Input Timing 50 ns tEW Enable Pulse Width See Data Input Timing 50 ns 1.0 b 5.0 2.5 mA 5.0 VCC b 0.8 nA V 0.4 VCC b 0.8 V V 0.4 1.0 V mA 100 mA ns **Except OSCOUT 5 http://www.national.com Functional Description The simplified block diagram below shows the 19-bit data register, the 14-bit R Counter and the R15 Latch, and the 18-bit N Counter (intermediate latches are not shown). The data stream is clocked (on the rising edge) into the DATA input, MSB first. If the Control Bit (last bit input) is HIGH, the DATA is transferred into the R Counter (programmable reference divider) and the S Latch (prescaler select: 64/65 or 128/129). If the Control Bit (LSB) is LOW, the DATA is transferred into the N Counter (programmable divider). TL/W/12459– 1 PROGRAMMABLE REFERENCE DIVIDER (R COUNTER) AND PRESCALER SELECT (R15 LATCH) If the Control Bit (last bit shifted into the Data Register) is HIGH, data is transferred from the 19-bit shift register into a 14-bit latch (which sets the 14-bit R Counter) and the 1-bit R15 Latch, which sets the prescaler: 64/65 or 128/129. Serial data format is shown below. TL/W/12459– 14 14-BIT PROGRAMMABLE REFERENCE DIVIDER RATIO (R COUNTER) Divide R R R R R R R R R R R R R R Ratio 14 13 12 11 10 9 8 7 6 5 4 3 2 1 R 1-BIT PRESCALER SELECT (R15 LATCH) Prescaler Select P R 15 3 0 0 0 0 0 0 0 0 0 0 0 0 1 1 128/129 0 4 0 0 0 0 0 0 0 0 0 0 0 1 0 0 64/65 1 # # # # # # # # # # # # # # # 16383 1 1 1 1 1 1 1 1 1 1 1 1 1 1 Notes: Divide ratios less than 3 are prohibited. Divide ratio: 3 to 16383 R1 to R14: These bits select the divide ratio of the programmable reference divider. C: Control bit (set to HIGH level to load R counter and R15 Latch) Data is shifted in MSB first. http://www.national.com 6 Functional Description (Continued) PROGRAMMABLE DIVIDER (N COUNTER) The N counter consists of the 7-bit swallow counter (A counter) and the 11-bit programmable counter (B counter). If the Control Bit (last bit shifted into the Data Register) is LOW, data is transferred from the 19-bit shift register into a 7-bit latch, which sets the 7-bit Swallow (A) Counter, and an 11-bit latch, which sets the 11-bit programmable (B) Counter. Serial data format is shown below. TL/W/12459– 15 Note: S8 to S18: Programmable counter divide ratio control bits (3 to 2047) 7-BIT SWALLOW COUNTER DIVIDE RATIO (A COUNTER) 11-BIT PROGRAMMABLE COUNTER DIVIDE RATIO (B COUNTER) Divide Ratio A N 7 N 6 N 5 N 4 N 3 N 2 N 1 Divide Ratio B N 18 N 17 N 16 N 15 N 14 N 13 N 12 N 11 N 10 N 9 N 8 0 0 0 0 0 0 0 0 3 0 0 0 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1 4 0 0 0 0 0 0 0 0 1 0 0 # # # # # # # # # # # # # # # # # # # # 127 1 1 1 1 1 1 1 2047 1 1 1 1 1 1 1 1 1 1 1 Note: Divide ratio: 0 to 127 Note: Divide ratio: 3 to 2047 (Divide ratios less than 3 are prohibited) BtA BtA PULSE SWALLOW FUNCTION fVCO e [(P c B) a A] c fOSC/R fVCO: Output frequency of external voltage controlled oscillator (VCO) B: Preset divide ratio of binary 11-bit programmable counter (3 to 2047) A: Preset divide ratio of binary 7-bit swallow counter (0 s A s 127, A s B) fOSC: Output frequency of the external reference frequency oscillator R: Preset divide ratio of binary 14-bit programmable reference counter (3 to 16383) P: Preset modulus of dual modulus prescaler (64 or 128) 7 http://www.national.com Functional Description (Continued) SERIAL DATA INPUT TIMING TL/W/12459– 16 Notes: Parenthesis data indicates programmable reference divider data. Data shifted into register on clock rising edge. Data is shifted in MSB first. Test Conditions: The Serial Data Input Timing is tested using a symmetrical waveform around VCC/2. The test waveform has an edge rate of 0.6 V/ns with amplitudes of 2.2V @ VCC e 2.7V and 2.6V @ VCC e 5.5V. Phase Characteristics In normal operation, the FC pin is used to reverse the polarity of the phase detector. Both the internal and any external charge pump are affected. Depending upon VCO characteristics, FC pin should be set accordingly: When VCO characteristics are like (1), FC should be set HIGH or OPEN CIRCUIT; When VCO characteristics are like (2), FC should be set LOW. When FC is set HIGH or OPEN CIRCUIT, the monitor pin of the phase comparator input, fout, is set to the reference divider output, fr. When FC is set LOW, fout is set to the programmable divider output, fp. VCO Characteristics TL/W/12459– 17 PHASE COMPARATOR AND INTERNAL CHARGE PUMP CHARACTERISTICS TL/W/12459– 18 Notes: Phase difference detection range: b 2q to a 2q The minimum width pump up and pump down current pulses occur at the Do pin when the loop is locked. FC e HIGH http://www.national.com 8 Analog Switch The analog switch is useful for radio systems that utilize a frequency scanning mode and a narrow band mode. The purpose of the analog switch is to decrease the loop filter time constant, allowing the VCO to adjust to its new frequency in a shorter amount of time. This is achieved by adding another filter stage in parallel. The output of the charge pump is normally through the Do pin, but when LE is set HIGH, the charge pump output also becomes available at BISW. A typical circuit is shown below. The second filter stage (LPF-2) is effective only when the switch is closed (in the scanning mode). TL/W/12459– 19 Typical Crystal Oscillator Circuit Typical Lock Detect Circuit A typical circuit which can be used to implement a crystal oscillator is shown below. A lock detect circuit is needed in order to provide a steady LOW signal when the PLL is in the locked state. A typical circuit is shown below. TL/W/12459– 20 TL/W/12459– 21 9 http://www.national.com Typical Application Example Operational Notes: * ** TL/W/12459– 22 VCO is assumed AC coupled. RIN increases impedance so that VCO output power is provided to the load rather than the PLL. Typical values are 10X to 200X depending on the VCO power level. fIN RF impedance ranges from 40X to 100X. *** 50X termination is often used on test boards to allow use of external reference oscillator. For most typical products a CMOS clock is used and no terminating resistor is required. OSCIN may be AC or DC coupled. AC coupling is recommended because the input circuit provides its own bias. (See Figure below) TL/W/12459– 23 Application Hints: Proper use of grounds and bypass capacitors is essential to achieve a high level of performance. Crosstalk between pins can be reduced by careful board layout. This is an electrostatic sensitive device. It should be handled only at static free work stations. http://www.national.com 10 Application Information LOOP FILTER DESIGN A block diagram of the basic phase locked loop is shown. TL/W/12459– 24 FIGURE 1. Basic Charge Pump Phase Locked Loop An example of a passive loop filter configuration, including the transfer function of the loop filter, is shown in Figure 2 . TL/W/12459– 25 Z(s) e s (C2 # R2) a 1 s2 (C1 # C2 # R2) a sC1 a sC2 FIGURE 2. 2nd Order Passive Filter Define the time constants which determine the pole and zero frequencies of the filter transfer function by letting (1a) T2 e R2 # C2 TL/W/12459– 26 FIGURE 3. Open Loop Transfer Function Thus we can calculate the 3rd order PLL Open Loop Gain in terms of frequency and T1 e R2 # G(s) # H(s)ls e j # 0 e C1 # C2 C1 a C2 b Kw # KVCO (1 a j0 # T2) T1 # 02C1 # N(1 a j0 # T1) T2 (2) From equation 2 we can see that the phase term will be dependent on the single pole and zero such that w(0) e tanb1 (0 # T2) b tanb1 (0 # T1) a 180§ (3) (1b) The PLL linear model control circuit is shown along with the open loop transfer function in Figure 3 . Using the phase detector and VCO gain constants [Kw and KVCO] and the loop filter transfer function [Z(s)], the open loop Bode plot can be calculated. The loop bandwidth is shown on the Bode plot (0p) as the point of unity gain. The phase margin is shown to be the difference between the phase at the unity gain point and b180§ . By setting T2 T1 dw e b e0 d0 1 a (0 # T2)2 1 a (0 # T1)2 (4) we find the frequency point corresponding to the phase inflection point in terms of the filter time constants T1 and T2. This relationship is given in equation 5. 0p e 1/0T2 # T1 (5) For the loop to be stable the unity gain point must occur before the phase reaches b180 degrees. We therefore want the phase margin to be at a maximum when the magnitude of the open loop gain equals 1. Equation 2 then gives TL/W/12459– 27 C1 e Open Loop Gain e ii/ie e H(s) G(s) e Kw Z(s) KVCO/Ns Closed Loop Gain e io/ii e G(s)/[1 a H(s) G(s)] 11 Ó Kw # KVCO # T1 (1 a j0p # T2) 0p2 # N # T2 (1 a j0p # T1) Ó (6) http://www.national.com Application Information (Continued) Therefore, if we specify the loop bandwidth, 0p, and the phase margin, wp, Equations 1 through 6 allow us to calculate the two time constants, T1 and T2, as shown in equations 7 and 8. A common rule of thumb is to begin your design with a 45§ phase margin. secwp b tanwp T1 e 0p (7) T2 e In choosing the loop filter components a trade off must be made between lock time, noise, stability, and reference spurs. The greater the loop bandwidth the faster the lock time will be, but a large loop bandwidth could result in higher reference spurs. Wider loop bandwidths generally improve close in phase noise but may increase integrated phase noise depending on the reference input, VCO and division ratios used. The reference spurs can be reduced by reducing the loop bandwidth or by adding more low pass filter stages but the lock time will increase and stability will decrease as a result. 1 0p2 # T1 (8) From the time constants T1, and T2, and the loop bandwidth, 0p, the values for C1, R2, and C2 are obtained in equations 9 to 11. THIRD ORDER FILTER A low pass filter section may be needed for some applications that require additional rejection of the reference sidebands, or spurs. This configuration is given in Figure 4 . In order to compensate for the added low pass section, the component values are recalculated using the new open loop unity gain frequency. The degradation of phase margin caused by the added low pass is then mitigated by slightly increasing C1 and C2 while slightly decreasing R2. The added attenuation from the low pass filter is: ATTEN e 20 log[(2qfref # R3 # C3)2 a 1] (12) T1 Kw # KVCO 1 a (0p # T2)2 # T2 0p2 # N 1 a (0p # T1)2 (9) T2 b1 C2 e C1 # (10) T1 T2 R2 e C2 (11) KVCO (MHz/V) Voltage Controlled Oscillator (VCO) Tuning Voltage constant. The frequency vs voltage tuning ratio. Kw (mA) Phase detector/charge pump gain constant. The ratio of the current output to the input phase differential. N Main divider ratio. Equal to RFopt/fref 0 C1 e # J RFopt (MHz) Radio Frequency output of the VCO at which the loop filter is optimized. fref (kHz) Frequency of the phase detector inputs. Usually equivalent to the RF channel spacing. T2 e Defining the additional time constant as T3 e R3 # C3 (13) Then in terms of the attenuation of the reference spurs added by the low pass pole we have 0 10ATTEN/20 b 1 (14) (2q # fref)2 We then use the calculated value for loop bandwidth 0c in equation 11, to determine the loop filter component values in equations 15–17. 0c is slightly less than 0p, therefore the frequency jump lock time will increase. T3 e 1 0c2 # (T1 a T3) (15) Ð01 (T1 a T3)2 a T1 # T3 b1 [tanw # (T1 a T3)] 2 0c e tanw # (T1 a T3) # [(T1 a T3)2 a T1 # T3] C1 e (/2 (1 a 0c2 # T22) T1 Kw # KVCO # # T2 0 c2 # N (1 a 0c2 # T12) (1 a 0c2 # T32) http://www.national.com a Ð ( 12 ( (16) (17) Application Information (Continued) EXTERNAL CHARGE PUMP The LMX PLLatinum series of frequency synthesizers are equipped with an internal balanced charge pump as well as outputs for driving an external charge pump. Although the superior performance of NSC’s on board charge pump eliminates the need for an external charge pump in most applications, certain system requirements are more stringent. In these cases, using an external charge pump allows the designer to take direct control of such parameters as charge pump voltage swing, current magnitude, TRI-STATE leakage, and temperature compensation. One possible architecture for an external charge pump current source is shown in Figure 9 . The signals wp and wr in the diagram, correspond to the phase detector outputs of the LMX2305 frequency synthesizer. These logic signals are converted into current pulses, using the circuitry shown in Figure 9 , to enable either charging or discharging of the loop filter components to control the output frequency of the PLL. Referring to Figure 9 , the design goal is to generate a 5 mA current which is relatively constant to within 0.5V of the power supply rail. To accomplish this, it is important to establish as large of a voltage drop across R5, R8 as possible without saturating Q2, Q4. A voltage of approximately 300 mV provides a good compromise. This allows the current source reference being generated to be relatively repeatable in the absence of good Q1, Q2/Q3, Q4 matching. (Matched transistor pairs is recommended.) The wp and wr outputs are rated for a maximum output load current of 1 mA while 5 mA current sources are desired. The voltages developed across R4, 9 will consequently be approximately 258 mV, or 42 mV k R8, 5, due to the current density differences À0.026*1n (5 mA/1 mA)Ó through the Q1, Q2/Q3, Q4 pairs. In order to calculate the value of R7 it is necessary to first estimate the forward base to emitter voltage drop (Vfn,p) of the transistors used, the VOL drop of wp, and the VOH drop of wr’s under 1 mA loads. (wp’s VOL k 0.1V and wr’s VOH k 0.1V.) Knowing these parameters along with the desired current allow us to design a simple external charge pump. Separating the pump up and pump down circuits facilitates the nodal analysis and give the following equations. VR5 b VT # ln R4 e R9 e R8 e R6 e R7 e isource p max Design Parameters bn e 100, bp e 50 VP e 5.0V; Vcntl e 0.5V b 4.5V; Vwp e 0.0V; Vwr e 5.0V ISINK e ISOURCE e 5.0 mA; Vfn e Vfp e 0.8V Irmax e Ipmax e 1 mA VR8 e VR5 e 0.3V VOLwp e VOHwr e 100 mV TL/W/12459– 28 FIGURE 9 Therefore select R4 e R9 e J 0.3V b 0.026 # 1n(5.0 mA/1.0 mA) e 51.6X 5 mA R5 e 0.3V e 300X 1.0 mA R8 e 0.3V e 300X 1.0 mA R6 e R7 e (5V b 0.1V) b (0.3V a 0.8V) e 3.8 kX 1.0 mA isource VR8 b VT # ln R5 e #i EXAMPLE Typical Device Parameters Typical System Parameters #i isink n max J isink VR5 ip max VR8 ir max (Vp b VVOLwp) b (VR5 a Vfp) ip max (VP b VVOHwr) b (VR8 a Vfn) imax 13 http://www.national.com LMX2305 PLLatinum 550 MHz Frequency Synthesizer for RF Personal Communications Physical Dimensions inches (millimeters) unless otherwise noted NS Package Number MTC20 20-Lead (0.173× Wide) Thin Shrink Small Outline Package (TM) Order Number LMX2305TM For Tape and Reel Order Number LMX2305TMX (2500 Units per Reel) LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. National Semiconductor Corporation Americas Tel: 1(800) 272-9959 Fax: 1(800) 737-7018 Email: support @ nsc.com http://www.national.com 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. National Semiconductor Europe Fax: a49 (0) 180-530 85 86 Email: europe.support @ nsc.com Deutsch Tel: a49 (0) 180-530 85 85 English Tel: a49 (0) 180-532 78 32 Fran3ais Tel: a49 (0) 180-532 93 58 Italiano Tel: a49 (0) 180-534 16 80 National Semiconductor Southeast Asia Fax: (852) 2376 3901 Email: sea.support @ nsc.com National Semiconductor Japan Ltd. Tel: 81-3-5620-7561 Fax: 81-3-5620-6179 National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
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