A 1.57-GHz RF Front-End for Triple Conversion GPS Receiver

A 1.57-GHz RF Front-End for Triple Conversion GPS Receiver
A 1.57-GHz RF Front-End for
Triple Conversion GPS Receiver
Francesco Piazza and Qiuting Huang, Senior Member, IEEE
Abstract— A low-power, 1.57-GHz RF front-end for a global
positioning system (GPS) receiver has been designed in a 1.0-m
BiCMOS technology. It consists of a low noise amplifier with
15 dB of gain, a single balanced mixer with 6.3 mS of conversion
gm , a Colpitts LC local oscillator, and an emitter coupled logic
(ECL) divide-by-eight prescaler. This front-end has a singal
sideband (SSB) noise figure of 8.1 dB and is part of a triple
conversion superheterodyne receiver whose IF frequencies are
179, 4.7, and 1.05 MHz. Low power consumption has been
achieved, with 10.5 mA at 3-V supply voltage for the front-end,
while the complete receiver is expected to draw about 12 mA.
Index Terms— CDMA, frequency synthesis, GPS, low noise
amplifier, low power, mixer, MMIC, receiver, RF front end.
HE fast growing wireless communications market has
created urgent demands for low-power, low-cost solutions
to implement both the digital processing and the RF analog
parts of communications receivers. Until a few years ago, RF
design was dominated by discrete circuits or low integration
level monolithic microwave integrated circuits (MMIC’s) in
the low GHz range, where most communication services,
including the global positioning system (GPS), are allocated.
The general trend toward miniaturization and low cost makes
integrated solutions with low external part count highly desirable, and today complete GPS receivers on a single silicon chip
have become commonplace [1]. These single-chip receivers,
although performing very well, are usually targeted toward
relatively big handheld or fixed (e.g., on a car or ship)
applications, where the power consumption or the number and
size of external components is less a problem. If GPS receivers
can be made much less power hungry and much smaller than
those on the market today, however, many applications outside
navigation and surveying, such as those of the consumer
market, can be envisaged. The latter include receivers that
are small enough to be carried in the pocket or worn on the
wrist, for better convenience. In this paper we present a GPS
chip targeted toward these applications. The paper is organized
as follows. Section II provides a general description of the
GPS system, which contains background technical information that has impact on our receiver planning. The planned
receiver architecture is presented next (Section III), before
front-end specifications and design are described (Section IV).
Manuscript received February 11, 1997; revised July 8, 1997. This work
was performed in collaboration with Asulab S.A., Marin-Epagnier.
The authors are with the Integrated Systems Laboratory, Swiss Federal
Institute of Technology (ETH), CH-8092 Zurich, Switzerland.
Publisher Item Identifier S 0018-9200(98)00718-5.
Sections V–VII discuss achieved power consumption, issues
in layout, and measurements respectively.
The GPS is based on 24 satellites located in six orbital
planes at a height of 20 200 km and circle the Earth every
12 h. Each plane is inclined at 55 to the Earth’s equator and
contains four satellites. The GPS positioning is based on oneway time-of-arrival ranging. Each satellite sends the universial
time (UTC) and navigation data using a spread spectrum code
division multiple access (CDMA) technique. A receiver can
calculate its own position and speed by correlating the signal
delays from any four satellites and combining the result with
orbit-correction data sent by the satellites. Two services are
provided by GPS: a precise positioning service (P-code) whose
use is restricted to military and a standard positioning service
(coarse acquisition, C/A-code), less precise than the P-code
but available to everyone.
All 24 satellites send on the same two frequencies: L1
is the primary frequency and carries the C/A-code, and L2
is the secondary frequency and carries the P-code. The two
frequencies are derived from a 10.23-MHz atomic frequency
standard. The frequency of L1 is 1575.42 MHz (154 times
the atomic clock) and that of L2 is 1227.6 MHz (120 times
the atomic clock). Interference between signals of different
satellites is avoided using pseudorandom signals with low
cross-correlation for the CDMA modulation. The C/A-code
uses 1023 chips Gold codes [2], [3].
The integrated circuit reported in this work is a low-power
RF front-end for a GPS receiver for the 1575.42 MHz civilian
L1 band. The immediate application of such an integrated
receiver is to provide GPS time reference—GPS positioning
will be used to set the correct time zone—for small, portable
(wearable) consumer products. Low power consumption is
therefore a primary requirement, and the specified power
source is a small lithium battery (2.4–3.5 V). The number
and size of external components are also important requirements, not only due to cost, but also the space available.
Both requirements must already be addressed during system
To reduce power consumption, the most obvious thing to do
is to reduce the number of components working at the highest
frequency, while the high gain IF amplifiers must work at the
lowest practical frequency. On the other hand, filter size and
selectivity requirements prevent the intermediate frequency
0018–9200/98$10.00  1998 IEEE
Fig. 1. Block diagram of the complete GPS receiver.
from being too low. Surface acoustic wave (SAW) filters for
frequencies lower than roughly 100–150 MHz are typically
too big and incompatible with our application. As a tradeoff
between these requirements, a triple conversion architecture
has therefore been chosen. A block diagram of this receiver,
including the most important external components, is shown
in Fig. 1.
The 1.57-GHz GPS L1 signal is received by an active patch
antenna and filtered with a SAW filter to remove signals at the
image frequency and other strong out-of-band signals which
may overload the front-end. This signal is then amplified
by about 15 dB and mixed down to first IF with a single
balanced mixer. The first IF signal is then routed to an off-chip
179-MHz SAW filter where channel filtering takes place. The
performance of this filter is critical to system performance,
due to the rather low second IF chosen. A relatively highfrequency channel filter has been chosen because of its small
size. The wide 1.8-MHz passband makes this possible, without
having to use highly selective filters. The filtered signal is then
amplified by 12 dB and down-converted to the 4.7 MHz second
IF with a double balanced mixer. Since channel filtering occurs
at first IF, the second IF filter need only remove the higher
frequency mixing products and the local oscillator feedthrough
of the first mixer, both being well above 4.7 MHz. The required
performance is therefore quite moderate, allowing the filter to
be fully integrated on chip. The main benefit of integrating the
second IF filter is the reduction of the receiver complexity to a
level comparable to that of a single superheterodyne receiver.
A fifth-order Butterworth active RC filter has been chosen.
After 68 dB of gain, the second IF signal is amplitude limited
and converted to digital with a 1-b AD converter. Using a 1-b
converter results in slightly degraded performance compared
to that of a multibit converter, but it allows the design of a
simpler, lower power receiver without automatic gain control.
Sampling at 3.6 MHz, the AD converter also down-converts
the second IF signal to the third IF of 1.05 MHz. Signal
detection is then performed digitally on a second chip that
contains all the digital processing and controlling parts of
the receiver. Since GPS uses CDMA, the receiver needs only
to receive one channel. This means only a single frequency
Fig. 2. GPS receiver specifications.
has to be generated by the local oscillator. The frequency
synthesizer is therefore a very simple design, and fixed dividers
with simple division ratios can be used, resulting in a further
reduction in power consumption.
Although a single-chip implementation is envisaged, due to
time constraints, the receiver has been split into two blocks.
The first is the 1.57-GHz RF front-end to the left of the dashed
line in Fig. 1. It consists of a low noise amplifier (LNA), a
single balanced mixer, an LC-oscillator and an emitter coupled
logic (ECL) divide-by-eight prescaler, and has been integrated
in a 1- m BiCMOS technology, whose NPN bipolar transistors
have an
GHz. The IF and baseband components
form the second chip, which has been designed in the same
BiCMOS technology [4]. The two chips will be merged later
into a single-chip receiver and will be reported separately.
The choice of gain for the various blocks of a receiver
is always a tradeoff, especially at the front end. The main
parameters that have to be taken into consideration are intermodulation (IP3), noise figure (NF), and power consumption.
Setting high gain to the active antenna and LNA will help
reduce the noise figure by minimizing noise contribution of
the mixer, but at the expense of higher power consumption in
these blocks and risk of early mixer overloading. Lower LNA
gain may improve linearity and power consumption, but a very
low noise mixer would be required to maintain an acceptable
noise figure. Such a mixer will very likely consume much
power and require a large local oscillator amplitude. In other
Fig. 3. Schematic diagram of the LNA-mixer combination.
words, low gain LNA combined with a low noise mixer may
not offer a significant advantage in total power consumption
over high gain LNA combined with a mixer with higher NF.
For the GPS signal itself, front-end nonlinearity is not much
of a problem. The received signal level is in fact very low
and relatively constant, consisting mainly of thermal (cosmic)
noise. The linearity specification is dictated by the required
system performance in the presence of external interfering
signals. The very small patch antenna required by wearable
applications has necessarily low gain. This tends to relax
the linearity requirement somewhat, but imposes a low noise
figure for the receiver. A front end with relatively high
gain has therefore been chosen. Fig. 2 shows the resulting
receiver specifications with gains, NF, and 1 dB compression
points (CP) given below each block. The total single sideband
(SSB) noise figure, referred to the input of the on-chip LNA,
is 7.43 dB. At the receiver input it is 3.81 dB, which is
adequate considering the application of this receiver. The first
RF amplifier block (active antenna) is not on-chip. It will
be constructed using a discrete transistor and a few passive
components and will not be discussed further in this paper.
A. Low Noise Amplifier
In this application, the gain of the LNA was specified as
15 dB, while the requirements for both the noise figure and
1-dB compression point were quite moderate, 7 and 43 dBm,
respectively. Gain independence from temperature, process
tolerances or
variation is desirable, as is 50- input
impedance, to avoid external matching networks.
To obtain a gain of 15 dB at 1.57 GHz, a two-stage
amplifier is needed. The schematic diagram is shown in Fig. 3.
It is a transconductance amplifier driving a transimpedance
load, similar to [6]. The main difference is the much higher
current levels required by this design, dictated by the higher
operating frequency. This requires a different strategy for
transistor sizing to minimize NF. The noise here is dominated
, while shot noise is negligible due to the high input
. A large
transconductance implemented by transistor
transistor, 24 times the minimum emitter, has been used to
, and thus thermal noise. The replica circuit (
, and
) is used to set the collector current of
3 mA, which is sufficient to achieve the required bandwidth.
The input impedance of the amplifier at high frequency is
mainly defined by
and the reactance of
. A
small, partially capacitive impedance around 12 is therefore
expected. Matching this impedance to 50
could be easily
obtained with an external low LC-network, but such external
components are undesirable in our application and should
be replaced with on-chip components whenever possible. A
common technique to obtain approximate matching is to add
a small inductance at the emitter of
, which will set the
input impedance of the amplifier to approximately
The required value is about 1 nH, which can be realized using
bonding wires. Since this value is quite small, the emitter
connection of
has been brought out on two separate pads,
such that the emitter inductor may be realized as two bonding
wires in parallel, effectively halving the total inductance.
In our test circuit, however, the best results were obtained
with a single, short, bonding wire. Wide bandwidth and low
sensitivity to manufacturing tolerances may be obtained if the
of the matching network is low. In this case, the is about
1.3, thus ensuring good matching over a wide bandwidth that
includes the image frequency. Had there been a mismatch at
the image frequency, the noise figure of the front end would
have been degraded, because no image filter is used before
the mixer. Using this technique, matching to 50 is achieved
with no external component, and a very satisfactory
10.6 dB has been achieved, as shown in Fig. 4. The first
LNA stage, with its bonding wire matching network, has a
voltage gain of 12 dB.
The transimpedance stage is formed by
and is
dc coupled to the first stage. Using dc coupling between stages
helps minimize the parasitic capacitances on sensitive nodes
and allows a flat frequency response (without peaking) to be
achieved. In order to save chip area, external components, and
the power required to go off-chip and drive a 50- load, the
image-reject filter has been omitted. The output of the LNA
has been connected directly to the mixer at the expense of
an increased noise figure due to the LNA noise at the image
frequency of the mixer. As explained later, the use of an image
reject filter could improve the noise figure by about 1.4 dB.
Fig. 5. Temperature compensation of LNA gain.
Fig. 4. Measured
of the LNA-mixer combination.
To reduce power consumption further, a small transistor (four
emitters) with low parasitic capacitances has been used as
To obtain the required gain and bandwidth, a collector current
of only 2 mA was required. At this current, the voltage gain
for the second stage is 4.7 dB, yielding a total LNA gain of
16.7 dB. Linearity performance is decreased at low current,
but the moderate requirement of our receiver is easily fulfilled
at 2 mA. Having a high
, transistor
is the main noise
source of the second stage, but its contribution to LNA noise
is low because it is scaled by the gain of the first stage. In
fact, the noise of the second stage is only about 30% of the
also generates the bias voltage for the mixer.
Since the
of a bipolar transistor is inversely proportional
to the absolute temperature, temperature compensation of the
LNA gain requires that its bias current have a proportional to
absolute temperature (PTAT) characteristic. In this prototype,
no temperature compensation has been implemented, but this
can be easily obtained by regulating the LNA supply voltage
as shown in Fig. 5.
Due to the internal connection to the mixer, no measurement can be performed on the LNA alone, thus performance
has been estimated with calculations and SPICE simulations
only. The voltage gain of the LNA, including its impedance
matching network, is 16.7 dB at 1.57 GHz, while its noise
figure is about 2.5 dB. The voltage gain of the complete front
end is 26.7 dB. The 1 dB CP of the LNA alone is dominated by
the second stage and is calculated to be 22.7 dBm, which is
much better than the specification. The overall front end 1 dB
CP, however, will be dominated by the mixer. Measurements
on the whole front end, as shown later, confirm these estimates.
B. Single Balanced Mixer
In order to minimize power consumption, a single balanced
RF mixer has been chosen. The conversion
of a single
balanced mixer is roughly twice that of a double balanced
mixer for the same bias current, but is less linear and has a high
local oscillator feedthrough to the IF port. This mixer, also
shown in Fig. 3, consists of
, a 12- emitter degeneration
resistor and the chopping differential pair
. The
IF output is high impedance open collector, which will be
connected to the external SAW filter via an RCL matching
network. The nominal source and load impedances for the
SAW filter are specified as 500 . Assuming ideal switching,
the conversion
of a single balanced mixer is
Assuming a 250- total load (500
for the SAW filter, in
parallel with its 500- matching resistor), a
of 12.6 mS is
required to obtain a gain of 10 dB. Taking into consideration
the effect of the emitter degeneration resistor, a bias current
of 2 mA is required for
. The emitter degeneration resistor
was added to improve both linearity and dc bias stability, with
only a slight degradation of noise figure. The latter is in fact
dominated by the noise generated by the switching differential
pair. The large local oscillator feedthrough typical of single
balanced mixers is of no consequence in this application, since
the oscillator signal is removed by the external IF filter.
DC biasing of the mixer is provided by the LNA second
, with which
forms a current mirror. By biasing
the LNA as in Fig. 5, the bias current of
will have a PTAT
characteristic. This means that temperature compensation of
is also obtained. DC bias of the local oscillator
the mixer
(LO) port is provided by a 5-k resistor to
. The resistor
is also part of the local oscillator.
The performance of this mixer, as in the case of the LNA,
can only be estimated, since the internal connection to the
LNA prevents any direct measurement. The noise figure of a
mixer is quite difficult to predict, because it depends on the
precise switching behavior of
and . When one of the two
transistors is completely cut off, the mixer looks like a cascode
amplifier, thus the noise is mostly contributed by
. Around
the zero-crossing of the LO signal, however, both
conduct, during which time they form a differential amplifier
that contributes its own amplified noise to the total noise of the
mixer. Experience has shown that the double sideband (DSB)
noise figure of a mixer of this type is around 16–18 dB, with
a moderate LO drive around 0 dBm. Given the DSB noise
figure of the mixer, and considering the 3-dB loss due to the
unfiltered noise at the image frequency for the LNA, the noise
figure of the whole front end can be calculated with
dB for the LNA, an SSB noise figure
between 7.1 and 7.9 dB will then result. The measured value
was 8.1 dB, which corroborates the estimate. The 1 dB
compression point of the front end will be dominated by that
of the mixer, which is around 5 dB below the compression
point of the LNA. A value of 28 dBm has been measured at
the input of the front end, which corresponds to 11.4 dBm
at the mixer input.
If the relative contribution of the mixer noise is small,
the noise figure of the front-end can be improved by using
an image filter between LNA and mixer, at the expense of
extra external components and higher power consumption. If
the relative mixer noise contribution is high, as is often the
case, the noise figure improvement by image-reject filter is
very inefficient. Assuming an ideal filter that is lossless at
the desired frequency and rejects the noise of the LNA at the
image frequency completely, the total SSB noise figure of the
front end can be calculated with
For the same noise figures of 18 and 2.5 dB for mixer and
LNA, respectively, a noise figure of 6.5 dB can be calculated,
which corresponds to a 1.4 dB improvement over the case
without filter. This is lower than the 3 dB one might have
expected, because the image filter, in fact, cannot suppress the
noise at the image frequency generated by the mixer itself.
A real filter will result in an even smaller improvement, due
to insertion losses and imperfect image rejection. A suitable
image-reject filter is a high- LC or a SAW filter. If a secondorder LC bandpass with, e.g., a of ten is used, it will reject
the 1.217 GHz image frequency by only 15 dB. Despite the
relatively moderate performance, such a filter cannot be easily
integrated, because it will require high inductors, which will
be difficult to implement using on-chip inductors or bonding
wires. More importantly, since the latter’s values cannot be
readily adjusted, tolerances in the resonant frequency will incur
unacceptable loss of gain. An off-chip SAW filter, on the other
hand, typically has a high insertion loss, although its center
frequency is better defined. Both approaches partially defeat
the purpose of the LNA. For these reasons, no filter has been
used in this design.
C. LC Local Oscillator
In our application, the local oscillator must provide a
1.4-GHz signal to the mixer, with an amplitude of about
3 dBm, or 450 mVpp , for best performance. With GPS being
a spread spectrum system, phase noise is not a critical parameter, and even a value around 80–85 dBc/Hz at 100 kHz
offset is still acceptable.
From the point of view of only phase noise, many oscillator architectures such as fully integrated ring or relaxation
oscillators, or LC oscillators with integrated low- inductor,
fulfill these specifications. Integrated oscillators, however, tend
Fig. 6. Schematic diagram of the local oscillator.
to consume much more power—a critical parameter in our
application—than those based on an external high- resonator.
The local oscillator chosen for our application is therefore the
varactor-tuned Colpitts oscillator with external LC tank, whose
schematic diagram appears in Fig. 6. Transistor
forms the
form a 10 : 1
active part of the oscillator, while
biasing current mirror.
The bias current is a function of the required signal amplitude and the of the external resonator and can be calculated
from the formula [5]
is the peak voltage at
is the voltage
divider ratio
, and
is the total parallel
resistive loss of the LC tank. To estimate the of the tank, a
few low-cost 10-nH inductors have been measured, showing
around 30–50 at 1.4 GHz. The corresponding series
resistance is therefore 1.8 to 2.9 . To account for the losses
in the capacitors, connections, and bonding pads, a total series
resistance of 4
has been assumed. Based on this estimate
and knowing that
in this circuit, we can derive that
A is required for 3 dBm amplitude. This current
is less than 9% of the total current consumption of the front
The oscillator is connected to the rest of the circuit directly,
without using any buffer, again in order to save power. The
mixer LO port is connected to the base of
, while the
divide-by-eight prescaler is ac coupled to its emitter. Usually,
a buffer with high reverse isolation is required at the input of
the prescaler to prevent the latter from injecting the switching
transients at submultiples of the oscillator’s frequency back to
the oscillator, thus degrading its performance. In our design,
a symmetrical layout, as explained later, has been used to
reduce these transients, thereby enabling the unbuffered
connection. During the measurements, no spurious signals
originating from the prescaler has been noticed, while
oscillator performance was not affected substantially by
switching the prescaler on and off.
D. Divide-by-Eight Prescaler
Since in a GPS receiver only one local oscillator frequency
has to be synthesized, fixed dividers with simple division ratios
can be used. In order to save power, for the first divider, a
Fig. 7. Schematic diagram of the ECL prescaler.
divide-by-eight prescaler has been chosen which consists of a
cascade of three ECL flip-flops. A schematic diagram of these
flip-flops is shown in Fig. 7. A signal amplitude of 200 mVPP
has been chosen as a compromise between noise immunity,
speed, and signal coupling into the substrate. The first flipflop is biased with 400 A and is ac coupled to the oscillator’s
emitter with a 2-pF capacitor. Its bias voltage is established
with a voltage divider such that no level shifters are needed,
therefore the input buffers
have been omitted.
Since the operating frequency of the second and third flip-flops
are 1/2 and 1/4 of the input frequency, the bias current has been
scaled down proportionally to 200 and 100 A, respectively.
These two flip-flops require input buffers (
) which
are biased with 50 and 25 A, respectively. This results in a
current consumption of 850 A for the whole prescaler. The
output of the prescaler is then buffered with a differential pair
and brought off-chip for measurements. This buffer consumes
400 A and is only required for this prototype. In the final
single-chip version of the receiver, the buffer is no longer
required since the output of the prescaler stays on-chip and is
connected directly to the rest of the frequency synthesizer.
E. Biasing
The local oscillator and the prescaler require a bias current
source of 90 and 40 A, respectively. These currents are
generated by the
current source shown in Fig. 8. A
current source has been used here because of its PTAT
temperature coefficient, which compensates that of the
of the bipolar transistors. The
core is formed by
(4 ) and
(1 ) and is cascoded by
for better
precision. Transistors
, , and related parts form a start-up
circuit that inject some current into . This circuit is then shut
down just before the output current reaches its nominal value.
Due to its high operating frequency, the power consumption
of the complete GPS receiver is dominated, as expected, by
Fig. 8. Schematic diagram of the bias current source.
the front end. The nominal supply current of this front-end
chip, including all the biasing circuitry and the output buffer
for the prescaler, is only 10.3 mA at 3 V, resulting in a
power consumption of 30.9 mW. The remainder of the receiver
is expected to draw approximately 1.7 mA, resulting in an
estimated power consumption for the complete receiver chip
of about 36 mW, which is sufficiently low for our application
and which compares very well to existing solutions to date.
While having a performance similar to our chip, the typical
consumption of existing commercial GPS chips is from 2.5 to
11 times higher [1].
A chip photomicrograph is shown in Fig. 9, the chip size
982 m2 . In a chip working at 1.57 GHz, many
is 1134
layout issues may affect the performance, as, for example,
component and pad placement, coupling through the substrate,
and between parallel lines, etc.
During the placement of the various blocks, attention has
been paid to parasitic coupling which could degrade performance or cause oscillation. The LNA and mixer are the block
Fig. 9. Photomicrograph of the chip.
at the bottom left, the VCO is placed top left, while the pads
of LNA and VCO are on opposite sides of the chip. All signal
pads are placed between ground pads or pads at ac ground
, bias). Large substrate contacts and guard rings separate
all stages. Since the substrate is high ohmic, guard rings are an
effective means to prevent parasitic coupling. More than 30 pF
of on-chip decoupling capacitors has been placed between
the supply lines to prevent coupling through them. The ECL
prescaler has been placed top right, with the low-speed stages
and the output buffer far right, to prevent interaction with the
VCO. Particular attention is paid to layout symmetry, while
the lines carrying true and complement have been routed in
parallel. The bias circuit is located bottom right and has its
own supply decoupling capacitor. Its outputs are connected
to pads and decoupled externally to ground with low ESL
ceramic surface mounted (SMD) capacitors.
Metal layers 1 and 2 have been used in most pads, while the
pads carrying high frequency signals are metal2 only for lower
capacitance against substrate. All pads have ESD protection
diodes, 50 m wide for the high frequency pads and 100 m
wide for the rest. The
pads have an NPN clamp. A total
of six pads are dedicated to ground, two for the LNA input
and four on opposite sides for the rest. Double pads have
been used for the VCO to maintain the stray inductance of the
connections (bonding wires) as low as possible.
Thick metal2 or metal1–metal2 lines have been used for
the ground connections, while metal1 has been used for the
. The RF connections are mainly metal2 and have been
kept short wherever possible, routed away from other sensitive
lines and partially shielded with metal1.
In this application, due to its stringent space constraints,
hybrid construction technique using unpackaged chips will be
used. For the measurements, a similar technique has been used.
The naked chip has been directly bonded to a small printed
circuit board (PCB), which contains all the necessary external
components. The PCB, measuring 25.4 25.4 mm2 , has one
signal plane and one ground plane on opposite sides and is
mounted on a test fixture that holds PCB and connectors in
place. Only SMD components have been used, while the VCO
inductor has been constructed with a loop of copper wire. The
supply voltage was set to its nominal value of 3 V.
The measured voltage gain of the front-end is 26.5 dB,
and driven by
with the mixer output terminated on 250
about 3 dBm of local oscillator amplitude. The acceptable
local oscillator range is 15 dBm to 3 dBm. At the two
extremes, the gain of the front end decreases by 1 dB. The
is 10.6 dB at 1575.42 MHz and 8.4 dB at the image
is plotted in Fig. 4 from 1
frequency of 1217 MHz. The
to 2 GHz. The SSB noise figure is 8.1 dB, slightly higher
than the specified 7.43 dB, while the input-referred 1 dB
compression point is 28 dBm. No IP3 measurement has been
performed, but a value of 18 dBm can be estimated from the
compression point measurement.
Due to the lack of a direct output, the phase noise of the
oscillator has been measured after mixing its signal down to
IF with a clean 1.57 GHz signal using the internal mixer. A
value of 95.1 dBc/Hz at 100 kHz offset has been measured.
Since the local oscillator was not fast enough to measure
the maximum operating frequency of the prescaler, the selfoscillating frequency with the oscillator stopped has been
measured, in addition to the regular divide-by-eight operation
at the local oscillator frequency. Self-oscillation with no input
signal is a normal behavior of any ECL divider. In fact, with
(see Fig. 7)
no input signal, all transistors from
are biased and form, together with the stray capacitances, a
relaxation oscillator. A value of 1728 MHz has been measured.
The maximum operating frequency is expected to be about 10
to 20% higher.
The bias currents generated by the internal current source
were 96.4 and 43 A at 3 V, while the total current consumption of the chip was 10.5 mA at the same supply voltage.
Although no temperature and supply voltage independent
biasing has been implemented in this prototype for the LNA
and mixer, the chip is fully functional over the specified 2.4
to 3.6 V supply voltage range, but some gain variation must
clearly be expected. The front-end gain varies between 22
and 29 dB for a 2.4 to 3.6 V supply voltage variation, while
the total current consumption varies from 8.1 to 12.8 mA.
Using the bias scheme shown in Fig. 5, the supply voltage
dependence will be reduced to approximately that of the bias
current source. For the latter, the bias currents were 94.5 and
42.2 A at a supply voltage of 2.4 V, increasing to 98.2 and
43.8 A for a supply voltage of 3.6 V.
A summary of the most important characteristics of this
GPS front end is shown in Table I.
A low-power RF front end suitable for small portable GPS
receivers has been demonstrated. Power consumption as low
as 31.5 mW, combined with a minimal number of external
components, has been achieved with nearly no compromise
to performance. The front-end gain of 26.5 dB and the noise
figure of 8.1 dB are comparable with most existing chips to
date while consuming much less power.
[1] Datasheets from GEC Plessey (GP1010, GP2010), Sony (CXA1951Q),
NEC (UPB1004GS), Philips (SA1570) and Rockwell.
[2] I. A. Getting, “The global positioning system,” IEEE Spectrum, Dec.
[3] E. D. Kaplan, Understanding GPS, Principles and Applications.
Boston/London: Artech, 1996.
[4] F. Piazza and Q. Huang, “An IF-strip with integrated 2nd IF filter for
a triple conversion GPS receiver,” in ESSCIRC’97, Southampton, UK,
Sept. 16–18, 1997, pp. 148–151.
[5] D. O. Pederson and K. Mayaram, Analog Integrated Circuits for Communication. Boston/Dordrecht/London: Kluwer, 1991.
[6] F. Piazza and Q. Huang, “A 170MHz RF front-end for ERMES pager
applications,” in ISSCC Dig. Tech. Papers, San Francisco, CA, Feb.
1995, pp. 324–325.
[7] R. G. Meyer and W. D. Mack, “A 1-GHz BiCMOS RF front-end IC,”
IEEE J. Solid-State Circuits, vol. 29, pp. 350–355, Mar. 1994.
Francesco Piazza was born in Olivone, Switzerland, on January 5, 1962. He received the Dipl.Ing.
degree in electrical engineering from the Swiss
Federal Institute of Technology (ETH), Zurich, in
Since May 1992, he has been with the Integrated
Systems Laboratory at the same university where he
is working toward the Ph.D. degree in the area of
RF and other high-speed integrated circuits.
Qiuting Huang (S’86–M’87–SM’96) graduated
from the Department of Precision Instruments,
Harbin Institute of Technology in 1982. He received
the Ph.D. degree from Katholieke Universiteit
Leuven, Departement Elektrotechniek, ESAT
Laboratories, Heverlee, Belgium, in 1987.
Between 1987 and 1992, he was a Lecturer at
the University of East Anglia, Norwich, UK. Since
January 1993, he has been Assistant Professor at
the Integrated Systems Laboratory, Swiss Federal
Institute of Technology, Zurich. His general field
of research is in analog and mixed analog-digital integrated circuits and
systems. His current research projects include RF front-end for wireless
communications, interface circuits to sensors and actuators, and low-noise,
low-power IC’s for biomedical applications.
Was this manual useful for you? yes no
Thank you for your participation!

* Your assessment is very important for improving the work of artificial intelligence, which forms the content of this project

Download PDF