Power MOSFET Basics

Power MOSFET Basics
Source
Gate
Power MOSFET Basics
N+
P-body
Table of Contents
1.
2.
3.
4.
5.
6.
7.
8.
9.
10.
11.
12.
13.
14.
15.
N- Epi
Basic Device Structure
Breakdown Voltage
On-State Characteristics
Capacitance
Gate Charge
Gate Resistance
Turn-on and Turn-off
Body Diode Forward Voltage
Body Diode Reverse Recovery
Avalanche capability and ratings
dV/dt ratings
Thermal Resistance Characterization
Power Dissipation
Safe-Operating Area
Current Ratings
N+ Substrate
Drain
Figure 1b: Planar MOSFET Structure
2. Breakdown Voltage
In most power MOSFETs the N+ source and P-body junction
are shorted through source metallization to avoid accidental
turn-on of the parasitic bipolar transistor. When no bias is
applied to the Gate, the Power MOSFET is capable of
supporting a high Drain voltage through the reverse-biased Pbody and N- Epi junction. In high voltage devices, most of the
applied voltage is supported by the lightly doped Epi layer. A
thicker and more lightly doped Epi supports higher breakdown
voltage but with increased on-resistance. In lower voltage
devices, the P-body doping becomes comparable to the N- Epi
layer and supports part of the applied voltage. If the P-body is
not designed thick or heavy enough, the depletion region can
punch-through to the N+ source region and cause lower
breakdown. But if it is over designed, the channel resistance
and threshold voltage will also increase. So careful design of
the body and Epi doping and thickness is needed to optimize
the performance.
1. Basic Device Structure
Power MOSFETs (Metal Oxide Semiconductor Field Effect
Transistor) are the most commonly used power devices due to
their low gate drive power, fast switching speed and superior
paralleling capability. Most power MOSFETs feature a
vertical structure with Source and Drain on opposite sides of
the wafer in order to support higher current and voltage.
Figure 1a and 1b show the basic device structures of Trench
and Planar MOSFETs respectively. Trench MOSFETs are
mainly used for <200V voltage rating due to their higher
channel density and thus lower on-resistance. Planar
MOSFETs are good for higher voltage ratings since onresistance is dominated by epi-layer resistance and high cell
density is not beneficial. The basic MOSFET operation is the
same for both structures. Unless specified, the N-channel
trench MOSFET is discussed in this application note.
In the datasheet, BVDSS is usually defined as the drain to
source voltage when leakage current is 250uA. The leakage
current flowing between source and drain is denoted by IDSS. It
is measured at 100% of the BVDSS rating. As temperature
increases, IDSS increases and BVDSS also increases for power
MOSFETs.
3. On-State Characteristics
We consider here power MOSFET under two different modes
of operations: the first quadrant operation and the third
quadrant operation.
N+
P-body
Gate
First-Quadrant Operation:
For an n-channel MOSFET, the device operates in the first
quadrant when a positive voltage is applied to the drain, as
shown in figure 2. As the gate voltage (VG) increases above
the threshold voltage (VTH), the MOSFET channel begins to
conduct current. The amount of current it conducts depends
on the on-resistance of the MOSFET, as defined by
N- Epi
N+ Substrate
Drain
RDSON = VD / I D
Figure 1a: Trench MOSFET Structure
1
For sufficiently large gate overdrive (VG >> VTH), the ID-VD
curve appears linear because the MOSFET channel is fully
turned on. Under low gate overdrive, the drain current reaches
a saturation point when VD > (VG-VTH) due to a pinch-off
effect of the channel.
MOSFETs benefit from high density scaling to achieve very
low RDSON.
Rs
Rch
Rch
Rs
RJFET
Repi
Rsubs
Figure 3b: RDSON components of a planar MOSFET
The channel resistance (RCH) is highly dependent on the
amount of the gate overdrive. RCH decreases with increasing
VGS. RDSON initially decreases rapidly as VGS increases above
VTH, indicating the turning-on of the MOSFET channel. As
VGS increases further, RDSON drops to a flat region because the
channel is fully turned on and the MOSFET resistance is
limited by the other resistance components.
Figure 2: On-region characteristics (first-quadrant
operation)
For a trench MOSFET, RDSON consists of the following
components:
- RS: source resistance
- RCH: channel resistance
- RACC: resistance from the accumulation region
- REPI: resistance from the top layer of silicon (epitaxial
silicon, also known as epi); epi controls the amount of
blocking voltage the MOSFET can sustain
- RSUBS: resistance from the silicon substrate on which the
epi is grown
RDSON increases with temperature due to the decreasing carrier
mobility. This is an important characteristic for device
paralleling.
Rs
Rch
Racc
Repi
Figure 4: RDSON vs. gate bias and temperature
Rsubs
Threshold Voltage
Threshold voltage, VGS(TH) , is defined as the minimum gate
bias which can form a conducting channel between the source
and drain. For power MOSFETs, it is usually measured at the
drain-source current of 250uA. Gate oxide thickness and
doping concentration of the channel can be used to control the
VGS(TH). Typically, 2~4V is designed for gate drive of 10-15V.
With the scaling down of the CMOS technology, the gate
drive of the power MOSFET drops to 2.5-4.5V. Therefore,
lower threshold voltages of 1-2V are needed for these
applications. The threshold voltage has a negative temperature
Figure 3a: RDSON components of a trench MOSFET
For a planar MOSFET, the RDSON components are similar to
that of a trench MOSFET. The primary difference is the
presence of a JFET component. As devices scale down to
smaller dimensions, RS, RCH, RACC are reduced because more
individual unit cells can be packed in a given silicon area.
RJFET on the other hand suffers from a “JFET”-effect where
current is constrained to flow in a narrow n-region by the
adjacent P-body region. Due to the absence of RJFET, trench
2
coefficient, which means it decreases with increasing
temperature.
Capacitance
The MOSFET’s switching behavior is affected by the parasitic
capacitances between the device’s three terminals, that is,
gate-to-source (CGS), gate-to-drain (CGD) and drain-to-source
(CDS) capacitances as shown in Figure 6. These capacitances’
values are non-linear and a function of device structure,
geometry, and bias voltages.
Transconductance
Transconductance, gfs, which is defined as the gain of the
MOSFETs, can be expressed as the following equation:
ΔI DS
ΔVGS
μ C W
g fs = n⋅ OX
LCH
g fs =
Source
It is usually measured at saturation region with fixed VDS. The
transconductance is influenced by gate width (W), channel
length (LCH), mobility (μn), and gate capacitance (COX) of the
devices. gfs decreases with increasing temperature due to the
decreasing carrier mobility.
N+
Cgs Gate
Cds
Third-Quadrant Operation:
Third-quadrant operation for power MOSFET is common in
DC-DC buck converters, where current conduction occurs
under at VDS (for an n-channel MOSFET). Current flows in
the reverse direction compared to first-quadrant operation.
The same RDSON components apply.
Drain
Figure 6: Illustration of MOSFET parasitic capacitances
During turn on, capacitors CGD and CGS are charged through
the gate, so the gate control circuit design must consider the
variation in this capacitance. The MOSFET parasitic
capacitances are provided in the data sheet parameters as CISS,
COSS, and CRSS:
CGD = CRSS
CGS = CISS − CRSS
CDS = COSS − CRSS
CRSS = small-signal reverse transfer capacitance.
CISS = small-signal input capacitance with the drain and source
terminals are shorted.
COSS = small-signal output capacitance with the gate and
source terminals are shorted.
Differences appear only under sufficient large current, and
therefore sufficient large VDON. When VDON approaches the
forward drop voltage of the body diode, the body diode starts
to conduct. As a result, the current increases and no current
saturation behavior is observed.
V DS (V)
-0.8
-0.6
-0.4
-0.2
N- Epi
N+ Substrate
Under relatively low current, the on-state characteristics for
the third-quadrant operation are symmetric to that of the first
quadrant operation. We may assume the same RDSON for both
types of operation.
-1
Cgd
P-body
0
0
The MOSFET capacitances are non-linear as well as a
function of the dc bias voltage. Figure 7a shows how
capacitances vary with increased VDS voltage. All the
MOSFET capacitances come from a series combination of a
bias independent oxide capacitance and a bias dependent
depletion (Silicon) capacitance. The decrease in capacitances
with VDS comes from the decrease in depletion capacitance as
the voltage increases and the depletion region widens.
-5
V GS=0V
(Body diode)
-15
ID (A)
-10
2.5V
-20
4.5V
10V
Figure 7b shows that the MOSFET gate capacitance also
increases when the VGS voltage increases past the threshold
voltage (for low VDS values) because of the formation of an
inversion layer of electrons in the MOS channel and an
accumulation layer of electrons under the trench bottom. This
why the slope of the gate charge curve increases once the
voltage goes beyond the Qgd phase.
-25
-30
Figure 5: Third-Quadrant Operation
4.
3
this state is QGS. Once the drain current reaches ID the drain
voltage starts to fall. At this period of time, VGS remains
constant at VGP. The gate current is used to charge the CGD
capacitance and Ig= CGD. dVDS/dt . The plateau phase ends
when VDS reaches its on-state value. The gate charge injected
during this plateau phase is QGD and is often used to estimate
voltage transition times and switching loss.
Next, the DUT gate continues charging to its final value, and
the drain-to-source voltage becomes equal to RDSON x ID. The
gate-to-source voltage is free to rise with a slope controlled by
the gate charging current and the CISS (which is higher at
VGS>VTH as shown in figure 7b, leading to a lower slope in the
gate charge curve) until the gate-to-source voltage reaches its
maximum value. This gate charge is the total gate charge QG.
Figure 7a: Typical variation of Capacitances with VDS
5000
Ciss (pF)
4500
4000
Ciss
3500
V DS =0V
3000
0
1
2
3
4
5
V GS (V)
Figure 7b: Typical variation of Ciss with VGS
5. Gate Charge
Figure 8: Gate charge test circuit & waveform
Gate charge parameter can be used to estimate switching times
of the power MOSFET once the gate drive current is known. It
depends only on the device parasitic capacitances. This
parameter is also weakly dependent of the drain current, the
supply voltage, and the temperature.
6. Gate Resistance
The power MOSFET gate presents an impedance like an RC
network to its gate drive. The equivalent R is referred to as the
gate resistance Rg. The gate resistance is caused by the finite
resistance of the Polysilicon gate conductors, and the metal
and contact structures that route the gate signal to the pad for
connection to external package leads. For polysilicon gate
power trench MOSFETS, the resistance of the gate electrode
depends on doping level and type (N type or P-type) of
polysilicon material, gate trench geometry and the device
layout arrangement. N-type trench power MOSFETs usually
have lower gate resistance than that of P-type trench power
MOSFETs for the same device layout due to lower sheet
resistance of N-type in situ doped polysilicon.
A schematic the gate charge test circuit and its waveform is
shown in Figure 8. In this circuit a constant gate current
source Ig charges the gate of the device under test, while drain
current ID is sourced. Measuring VGS vs. gate charging time
provides a direct indication of the energy spent to switch drain
current from 0 to ID as the Drain voltage swings from VDC to
its on-state voltage.
Before the gate current is turned on, the DUT withstands all
the supply voltage VDC, while the voltage VGS and the drain
current are zero. Once the gate current Ig flows, the gate-tosource capacitance CGS and gate-to-drain capacitance CGD start
to charge and the gate-to-source voltage increases. The rate of
charging is given by IG/CISS. Once the voltage VGS reaches
threshold voltage of the power MOSFET, drain current starts
to flow. The gate voltage continues to rise to the plateau
voltage VGP (VGSTH+ID/gFS), while the voltage across the DUT
remains equal to VDC. The charge (Ig*time) needed to reach
Most switching devices are 100% final tested for Rg using
LCR meters
7.
4
Turn-on and Turn-off
Body Diode Forward Voltage
VSD is a measure of the forward voltage drop of the integral
body diode, by applying a set current to the source. The
applied current is typically 1A and is specified in the datasheet
along with the maximum limit of forward voltage drop.
Figure 10 shows typical forward I-V characteristics for the
diode at two temperatures. For AOS SRFET, the typical VSD is
lower than that of a normal MOSFET, with typical value of
0.4V; this low VSD can help to reduce power loss during diode
conduction duration. SRFET is therefore an ideal choice for
low side FETs for DC-DC conversion, and other applications
where a certain period of body diode conduction is needed.
1.0E+02
1.0E+01
IS (A)
125°C
125°C
1.0E+00
SRFET
Figure 9: Resistive switching test circuit & waveforms
25°C
25°C
Regular FET
1.0E-01
Power MOSFET datasheets often contain the resistive
switching characteristics, which depend on Rg, Ciss and Crss.
While practical measurements are influenced by parasitic
inductances and gate drive details, we examine the basic
physics here. Figure 9 shows the power MOSFET resistive
switching test circuit and waveforms.
1.0E-02
0.0
0.2
0.4
0.6
0.8
1.0
V SD (Volts)
td(on) – Turn-on Delay Time
This is the time from when Vgs rises over 10% of the gate
drive voltage to when the drain current rises past 10% of the
specified current. At the moment of td(on), VGS reaches up to
the threshold voltage VTH. This period is controlled by the
time constant Rg.Ciss.
Figure 10: Body-Diode forward Characteristics
9. Body Diode Reverse Recovery
MOSFET parasitic body diode reverse recovery occurs during
diode switching from the on-state to the off-state, because its
stored minority charges must be removed, either actively via
negative current, or passively via recombination inside the
device. There are three dynamic parameters listed in the
datasheet for diode reverse recovery:
tr – Rise Time
This is the time between the drain current rising from 10% to
90% of load current. This depends on the VTH,
transconductance gFS and the time constant Rg.Crss.
trr: body diode reverse recovery time
IRM: body diode reverse peak current
Qrr: Body diode reverse recovery charge, defined by
td(off) – Turn-off Delay Time
It is the time from when Vgs drops below 90% of the gate
drive voltage to when the drain current drops below 90% of
the load current. It is the delay before current starts to
transition in the load, and depends on Rg.Ciss.
Qrr = ∫ isd (t )dt
trr
the area within the negative portion of the diode current
waveform.
tf – Fall Time
It is the time between the drain current falling from 90% to
10% of load current. This depends on the VTH,
transconductance gFS and the time constant Rg.Crss.
The above parameters vary with test condition, such as applied
voltage VDS and di/dt etc. The parameter definitions and test
circuit are shown in Figure 11.
8.
5
Avalanche capability and ratings
Physics of avalanche breakdown
As the voltage of a power MOSFET is increased, the electric
field increases at the body-epi junction. When this field
reaches a critical value EC (about 3E5V/cm in Si), avalanche
multiplication of carriers occurs, leading to an abrupt increase
in current. Avalanche multiplication is not a destructive
process. However, since the current flow path involves hole
current flow IH (=ID) in the path shown in Figure 12, there is
the possibility at high current density of turning on the
parasitic bipolar when VBE=IH*(Rp+Rc)>0.7V. When this
occurs, the gate can no longer turn-off the FET current. Also,
since the BVCEO is typically lower than the MOSFET
breakdown, current filaments into the weakest cell where local
non-uniformities first cause the parasitic bipolar to turn-on.
From this basic description it follows that:
• Failure occurs above a critical current density (even
for short low energy high voltage pulses)
• High values of Rp (body pinch resistance under the
source) and Rc (contact resistance) degrade UIS
capability
• High density cell construction reduces the length of
the current path. This decreases Rp, and increases the
critical current density at which failure occurs.
• Since both Rp and Rc increase with temperature, and
the emitter-base turn on voltage VBE decreases with
temperature, UIS capability decreases with
temperature.
If avalanche capability is tested at lower currents over a long
duration, the energy dissipated Ipk/2*BV*tAV heats the device.
The failing current is therefore determined by the peak
temperature the device reaches during this event. Since large
chips have a greater heat capacity, they have higher UIS
capability in this mode of operation.
Figure 11: Diode reverse recovery test circuit &
waveforms
The gate and source of DUT are shorted to test the body diode.
A control device is subjected to a double pulse. The current
ramps in the lower control device, and freewheels through the
DUT body diode when the control device turns off. When it is
turned on again by the second pulse, the DUT body diode
must recover before the control FET voltage can drop.
During diode reverse recovery, its reverse current also goes to
the lower FET in Figure 11, along with the load current; in
addition, the reverse recovery di/dt can cause large voltage
overshoots (Ldi/dt) due to circuit stray inductance. These
voltage overshoots are minimized if the di/dt during the
second phase of the trr (after crossing IRM) is kept low. Such a
diode is said to have soft recovery. Lower QRR leads to lower
switching loss. This is often the largest single component of
switching loss in a switching converter.
Source
N+
P-body
AOS SDMOS and SRFET have been designed with advanced
technology specifically to improve body diode reverse
recovery performance with low Qrr and good softness
coefficient compare to regular MOSFETs, which can greatly
reduce the voltage overshoot, and improve the overall
efficiency.
Rc
Gate
Rp
N- Epi
N+ Substrate
Drain
Figure 12: Trench MOSFET cell construction. Parasitic
NPN is shown, along with parasitic base resistances Rp
and Rc
10.
6
Avalanche ratings
Repetitive ratings:
Power MOSFETs may be driven to voltages in excess of rated
VDS(MAX) due to inductive spikes during circuit operation.
Therefore, manufacturers commonly specify single and
repetitive ratings, and many perform 100% single pulse testing
on shipped units.
If a power MOSFET is subjected to repetitive UIS pulses, its
junction temperature undergoes an increase in average value,
based on the average power dissipated as well as peaks of
temperature with each pulse. When the current density is high
enough, and the peak temperature is high enough, the device
can fail from the same mechanisms as described for single
pulse avalanche.
Typical single pulse ratings are captured using time in
avalanche curves as shown in figure 13. These are guaranteed
performance data, and the actual point of destruction is above
this level. As expected, avalanche capability decreases with
temperature for the same time duration. The current capability
decreases with longer times in avalanche at a given starting
junction temperature due to heating during the avalanche
event.
No common standard is being used for specifying repetitive
avalanche ratings. Two methods are described here.
Method 1: Select a small inductor, say L=1μH, with duty
cycle of 0.01, f=100kHz. Increase the current until average
temperature reaches TJ =150C to set IAR. Or increase current
until destruction occurs and de-rate to establish the IAR rating.
This method has the disadvantage that it only relates to one
inductor and one frequency. If the frequency is raised, IAR
drops. If the inductor is higher, IAR drops. In fact, if the
frequency is low enough such that the device returns to its
starting TJ (25C) after each pulse, well designed power
MOSFETs will have EAR=EAS, and IAR=IAS.
These curves are generated using the circuit in figure 14,
where the current is ramped up in the device under test
through an inductor. When the device turns off, since the
inductor current cannot be interrupted, the device voltage flies
up to the breakdown voltage of the device. When the device
turns off, the switch in series with the supply Vdd is also turned
off, forcing the current to freewheel though the diode. Now
the voltage across the inductor is –BV, which causes the
current to ramp down to zero. Using different inductors, one
may obtain different times in avalanche. The basic equations
relating Energy, current, time and inductance are listed below:
Ear = ∫ iL × VDS dt =
tav =
i pk × L
Method 2: Do not distinguish EAR, EAS, and IAR, IAS since they
are the same at low enough frequency. The user may use the
time in avalanche curves for short time durations to estimate
the maximum allowable current in avalanche, beginning with
an estimate of starting TJ from the average power dissipation P
and thermal resistance.
1 2
× i pk × L
2
P = 0.5 fLI 2
P
TJ =
RθJA
BV
11.
Figure 13: Time in avalanche vs. Peak current as a
function of temperature
Figure 14: Test circuit used for UIS (avalanche)
measurements
7
dV/dt ratings
Thermal Resistance Characterization
Power MOSFETs fail from excessive drain source dV/dts
under various scenarios. In each case, the failure is caused by
displacement or conduction current flow via Rp+Rc, leading
to turn-on of the parasitic bipolar, and consequent failure of
the device by the same mechanism described before for
avalanche failures.
• If the gate is shorted to source via a resistor, and a
fast dV/dt applied between Drain and source, the
displacement current Coss*dV/dt flows under the
source, and can develop sufficient voltage drop
across Rp+Rc to exceed the VBE (0.7V) of the
parasitic bipolar. Due to the low Coss values of most
modern power MOSFETs, this current is low even
for dV/dts of 10-50V/ns, and is not considered a
major failure mode. If however, the resistance
shorting the gate to source is large, the Crss*dV/dt
current will develop enough voltage drop across it to
turn on the gate, leading to current flow which if
unconstrained, can lead to device failure.
• During body diode reverse recovery, hole current
flows out of the source contact via Rp+Rc. This
current adds to and often far exceeds the Coss*dV/dt
current at also flows as the voltage develops across
the body diode of the FET. Since the diode stored
charge and its removal is non-uniform, the diode
recovery dV/dt failure is seen at lower values of
dV/dt. The failure mechanism is again caused by
turn-on of the parasitic bipolar. If the gate-source
shorting resistor is too large, there is the further
possibility of exacerbating the dV/dt current by
turning on the gate of the MOSFET by developing
sufficient voltage across the resistor as it sinks the
Crss*dV/dt current.
• Both modes of dV/dt failure get worse with
temperature.
Junction Calibration
Before the thermal resistance of any device is to be measured,
a calibration curve must be made. Each silicon device has its
own unique calibration, but once determined, is valid for any
package it may be put into.
The calibration curve is measured by treating the device as a
diode and forcing a 10mA sense current (IS) and measuring the
forward voltage drop (VFSD) at each junction temperature. A
sample calibration curve for a device is shown in Figure 15.
On subsequent thermal resistance tests, the same 10mA sense
current will be forced through the device and the junction
temperature will be calculated from the resulting forward
voltage drop.
Figure 15: Sample Temperature Calibration Curve.
12.
Junction-to-Ambient/Lead/ Case Thermal Resistance
The junction-to-ambient thermal resistance RθJA is defined as
the thermal resistance from the device junction to the ambient
environment. The junction-to-lead thermal resistance RθJL is
the thermal resistance from the device junction to the drain
lead of the device. For larger devices (Ultra SO8 and bigger)
with a back exposed drain pad, the RθJC must also be
measured. RθJC is defined as the thermal resistance from the
device junction to the device case. Both can be calculated
from the following equation:
RΘJX =
TJ − T X
PD
Where TJ is the junction temperature of the device, it can be
read out from junction calibration curve by measuring forward
voltage drop at different junction temperature. TX is the
ambient, lead or case temperature depending on whether RθJA,
RθJL or RθJC is being measured, and PD is the power dissipation
of the device, which is calculated by input voltage and current.
8
Transient Thermal Heating Curve, Junction to Ambient/
Case
Power Dissipation
Power dissipation PD and PDSM are the maximum power that is
allowed for device safe operation. Power dissipation is
calculated using the following formula:
Transient thermal curves can be used to estimate instantaneous
temperatures resulting from power loss on a transient basis.
These curves can be Junction to Ambient or Case based.
Namely they are characterizing transient thermal impedance
form device junction to ambient or case.
PD =
The thermal measurement machine applies a single pulse with
various durations; junction temperature is read again by
measuring forward voltage drop after each single pulse. This
measures the ‘single pulse’ transient thermal curve. Based on
single pulse curve, a 3 or 4 stages RC network is simulated to
generate the rest curves in Transient Thermal Heating curve
group, as show in Figure 16.
Thermal Resistance
RΘJC (max)
PDSM =
TJ (max) − TA
RΘJA(max)
PD is based on junction to case thermal resistance. To achieve
power dissipation of PD, case temperature needs to be
maintained at 25oC.
PDSM is based on junction to ambient thermal resistance. The
device is mounted on a 1 square inch 2 oz. copper PCB, and
PDSM is the power that raises Tj to 150oC
10
Z θ JA Normalized Transient
TJ (max) − TC
1
0.1
14. Safe-Operating Area
0.01
0.001
0.00001
0.0001
0.001
0.01
0.1
1
10
100
SOA (FBSOA) curves define the maximum value of drain to
source voltage and drain current which guarantees safe
operation when the device is in forward bias.
1000
Pulse Width (s)
Figure 16: Sample Transient Thermal Heating Curve
based on Junction-to-Ambient measurements.
13.
Figure 17: Maximum Forward Biased Safe Operating
Area
The right hand vertical boundary is maximum drain to source
voltage (VDS).
The upper horizontal limit is maximum pulsed drain current
(IDM).
The slope on left had side is limited by drain to source on
resistance (RDS(ON))
The paralleled lines in the middle are the maximum drain to
source current for different pulse widths. These currents are
determined by the transient thermal impedance.
15.
9
2.
Current Ratings
Continuous Drain current - ID and IDSM
Reference:
[1] B. J. Baliga, “Fundamentals of Power Semiconductor
Devices”, 2008.
[2] Application notes MOS-006, “Power MOSFET
Continuous Drain current rating and Bonding wire limitation’,
www.aosmd.com.
Excluding package limitations, the continuous Drain Current
ID and IDSM is the maximum drain current corresponding to PD
and PDSM.
ID =
Calculation based on transient thermal resistance
at 260μs pulse duration.
PD
RDS ( on ) max @ TJ (max)
ID will be de-rated with increasing case temperature, as shown
in Figure 18, based on the reduced power dissipation allowed.
Figure 18: Current rating vs. case temperature
Package Limitation
Continuous current rating is limited by two factors:
1. thermal resistance
2. package limitation
Package limitation usually refers to bond wire current
handling capability. The conventional way to rate bond wire
current limit is based on bond wire fusing temperature, which
is not correct because:
1. Wire temperature can not exceed 220oC, or it
will cause the degradation of the plastic molding
compound.
2. In most cases the silicon resistance is ~10 times
higher than wire resistance. Most of the heat is
generated on the silicon surface. The hottest spot
is on silicon.
Silicon maximum junction temperature is lower than 220oC,
that’s why bond wire fusing problem doesn’t exist in most of
the cases. Bond wires fuse only when devices fail. Please refer
to application note [MOS-006] on AOS website.
Pulsed Drain Current - IDM
Pulsed Drain Current is rated for 260μs current pulse. The
value on datasheet is the lower value of the following two:
1. Actual single pulse current measurement with
260μs current pulse.
10
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