A Scanning Thermal Microscopy System with a Temperature Dithering, Servo

A Scanning Thermal Microscopy System with a Temperature Dithering, Servo
A SCANNING THERMAL MICROSCOPY SYSTEM WITH A TEMPERATURE
DITHERING, SERVO-CONTROLLED INTERFACE CIRCUIT
Joohyung Lee1 and Yogesh B. Gianchandani1,2
1
2
ECE Department, University of Wisconsin, Madison, WI 53706, USA
EECS Department, University of Michigan, Ann Arbor, MI 48109, USA
ABSTRACT
This paper describes a thermal imaging system which includes
a customized micromachined thermal probe and circuit interface for
a scanning microscopy instrument. The probe shank is made from
polyimide for mechanical compliance and high thermal isolation,
and has a thin-film metal tip of ≈50 nm in diameter. The circuit
provides closed-loop control of the tip temperature and also permits
it to be dithered, facilitating scanning microcalorimetry applications.
This paper explains system design and optimization including both
electrical and thermal analyses. Sample scans of patterned
photoresist demonstrate noise-limited resolution of 29 pW/K in
thermal conductance. Applications of the thermal imager extend
from ULSI lithography research to biological diagnostics.
can be controlled to operate the scan at a fixed temperature. The
interface also provides electronic dithering of the tip temperature.
Its design includes consideration of thermal and electrical
interactions between the probe and circuit components based on
MatLab™ modeling of the overall system. The functionality of the
system is demonstrated with both microcalorimetric and imaging
applications of patterned photoresist and calibration materials. To
further evaluate the operation of the circuit, nodal measurements
taken during a practical scan (not in a test mode), and are presented
along with the scanned image obtained.
AC
+
PI controller V
integ
I. INTRODUCTION
In the past decade, scanning microscopy using thermallysensitive probes has been applied to a variety of applications,
ranging from ULSI lithography research to cellular diagnostics in
biochemistry [Oc96, Li02]. Thermal probes have also been
employed for data storage and other applications [Ve00, Le00,
Ma99]. These are generally made from dielectric thin films on a
silicon substrate, and use a metal or semiconductor film bolometer
for sensing the tip temperature. Other approaches that use more
involved micromachining methods have also been reported [Gi97].
A commercially available probe uses a narrow gauge wire bent into
a V-shape to form a self-supporting resistor. However, for many
applications, thermal probes must have very low mechanical spring
constants to prevent damage to soft samples. In addition, for many
applications they must have very high thermal isolation to minimize
the thermal load presented to the sample. Both of these needs can
be met by the use of a polymer for the probe shank. Furthermore,
thermal and mechanical design challenges must be considered in
conjunction with the interface circuit for best performance of the
overall system.
In a frequently used microscopy technique, the scanning tip is
mechanically dithered so that the sample spacing is modulated at a
known frequency. This is akin to chopping the signal, which
permits the detection to be phase locked to the dither, improving the
overall signal-to-noise ratio. In the context of thermal microscopy,
this also permits thermal capacitance measurement. However, with
ultracompliant probes, the mechanical spring constant is far too low
to permit physical dithering. The alternative is to dither the bias
current in the bolometer, thereby placing the burden on the interface
circuit. Furthermore, the very high thermal resistance of the probe
shank, designed to minimize thermal loading of the sample, has the
impact of reducing the thermal bandwidth of the probe below 1
KHz. This increases the susceptibility of the pick-off to flicker
noise and further raises the burden on the circuit.
This paper describes a thermal imaging system (Fig. 1) which
uses a customized micromachined bolometer probe and circuit
interface to a commercial scanning microscopy instrument
(TopoMetrix™ SPMLab v.3.06). The bias current in the bolometer
V
VCC
CC
Summing
amplifier
+
Electronic
dithering
source
V5
VE E
LPF
Vout
R1
∆Vout
Frequency
Doubler
DEMODULATOR
+
Voltage
difference
amplifier
R2
V2 V1
Rc
PSPD
Ip
Laser
Mirror
Heat
Transfer
Rp
Mount
Thermal
Probe
Sample
Feedback Loop
Vref
X, Y, Z
Piezo-drive
Rp : Probe
resistance
Rc : control
resistance
Vinteg : integrator
output
Vout : demod.
output
Fig. 1: The overall system configuration of the custom probe and
circuit which interface with a commercial instrument.
II. SYSTEM DESCRIPTION
A. Sensor Element
The scanning thermal probe is fabricated on a Si substrate
using a 7 mask process similar to those described in [Li00, Li01]. A
metal thin film bolometer is sandwiched between two layers of
polyimide that form a cantilever (Fig. 2). At one end of the
cantilever the Ni thin film protrudes through an opening in lower
polyimide layer, where it is molded into a pyramidal tip by a notch
that was anisotropically wet-etched into the substrate. A tip
diameter of ≈50 nm is achieved by sharpening the notch by
anisotropic thermal oxidation. The tip and a portion of the probe
shank are then released from the substrate by etching an underlying
sacrificial layer. The released length is then folded over to extend
past the die edge for clearance, and held in place by a thermocompression bond across a thin film of Au which is deposited as the
final layer on top of the polyimide. This film also serves as a mirror
to permit use of the probe for AFM. The entire fabrication process
is performed below 350°C, and is compatible with post-CMOS
processing to accommodate the possible integration of an interface
circuit. Typical dimensions of the probes after assembly are 250 µm
length, 50 µ m width, and 3 µ m thickness, which result in a
mechanical spring constant of 0.08 N/m, which is upto 100× below
commercial probes. The bolometer, which has Cr/Ni at the tip and
Cr/Au leads, is ≈45 Ω.
Fig. 2: Schematic and optical micrograph of a fabricated probe.
B. Interface Circuit
The bolometer readout is through a Wheatstone bridge, which
is commonly used for piezoresistive pressure sensors, strain gauges,
etc. It is well suited for microfabrication and allows a differential
measurement that offers a higher common-mode noise rejection than
a single-element measurement. Historically, the conversion of
bridge resistance to current or voltage for readout has suffered from
non-linearity and restricted dynamic range [Yo00]. Additionally, in
DC mode the signal is subject not only to thermal noise from the
resistor bridge, but also 1/f flicker noise from the electronics. To
overcome these challenges, many efforts have been made to convert
resistance variation to frequency [Mo95, Hu87, Gi76], to duty
cycle/time [Ci90, Go93], and to both of them [Fe97]. Some require
components such as a pulsed bridge supply current, or an input
amplifier with very low offset and drift [Gi76]. Furthermore, these
approaches are constrained by switching delays causing nonlinearity between frequency (or pulse width) and resistance change,
are expensive to implement, and most importantly cannot be applied
directly to operating the microbolometer or anemometer in a
constant temperature mode.
The system used in this effort (Fig. 1) utilizes two separate
feedback loops: electrical and optomechanical. As the probe (Fig. 2)
scans the sample surface, topography is mapped by detecting the
laser signal reflected off a mirror located near the tip and using this
in a mechanical feedback loop to maintain constant contact force.
Since variations in heat loss through the probe tip cause variations in
the probe resistance, this quantity maps the temperature or thermal
conductance of the sample.
When both a DC and an AC signal (at ω o) are applied to the
bridge (Fig. 1), the bolometer is modulated by the square of
VDC+VACcos(ωot+θ), and its resistance changes proportional to:
∆Rp ∝ VDC2+2VDCVACcos(ωot+θ) + VAC2cos2(ωot+θ)
(1)
Therefore, bolometer resistance is approximately represented as:
(2)
Rp ≈ RpDC + RpACcos(ωot+θ) if VAC2 << 2⋅VDC⋅VAC
making ω o the dominant resistance-modulation frequency. The
output of the bridge voltage difference amplifier is:
∆Vout=0.5⋅IACRpACcos(2ωot+θ)+ IDCRpACcos(ωot+θ)
(3)
If the 2 ω o term (second harmonic) of the voltage-modulation
frequency is selected, the impact of 1/f flicker noise can be reduced.
In addition, better signal-to-noise ratio is expected as IAC becomes
high to a certain extent. In the selected implementation, VDC was 5
V and VAC was 0.8 V.
The interface circuit includes a PI controller (which is
comprised of an integrator and an inverting amplifier), and a simple
homodyne demodulator (Fig. 1), in which the input signal is
multiplied by in-phase local oscillator and then low pass filtered
(Fig. 3). The PI controller has integral gain of 104 and proportional
gain of 1, showing settling time <10 msec. This demodulation
technique (Method A) is applicable when phase change in the input
signal is negligibly small compared to change in its magnitude. The
Method A is simple to implement and does not have mismatch
problems which are faced in quadrature homodyne demodulation
(Method B). In Method B, in-phase (I signal) and quadrature (Q
signal) signals are generated, low pass filtered, and root mean
squared. Problems are caused by mismatches between the
amplitude of I and Q signals and errors in the nominally 90° phase
shift. Method A is consequently preferred. According to our
previous investigations [Li01], the -3 dB frequency of thermal
response of the probe is about 0.5 kHz with an open-loop interface
circuit. It is somewhat higher with a closed loop interface circuit
because external power is used to increase effective thermal
conductance of the probe [Sa93]. Consequently, for this project a 1
kHz dither is selected, and scan speeds are set to provide a measured
data bandwidth <50 Hz. The bridge output voltage is band pass
filtered at the second harmonic 2 kHz, and multiplied by the
frequency doubled output of the dither oscillator (Fig. 3). The phase
of the local oscillator is synchronized with that of input carrier
signal to avoid signal distortion. The final output is obtained by low
pass filtering. The Q factor of band pass filter and -3 dB frequency
of low pass filter are based on the dither frequency and data
bandwidth, but adjusted for low frequency noise near the band edge
of scan data. Bi-quad band pass filters and bi-quad low pass filters
are used because of their excellent tuning features and good
stability. The gain, quality factor, and salient frequencies of the
filters can be independently controlled.
III. SYSTEM MODELING AND SIMULATION
The simulation of the whole sensing subsystem provides an
understanding of the interaction between thermal behavior of the
probe and electrical behavior of the interface circuit. It is
accomplished using electrical parameters of the Simulink tool within
MatLab™. Figure 4 shows the state diagram for the combined
subsystem. Using this, it is demonstrated how the demodulator
achieves noise reduction compared to a non-dithered DC closed loop
interface circuit.
A challenge in modeling the subsystem is how to transform a
thermal probe into electrical parameters. The dotted block in bottom
left of Fig. 4 represents the thermal probe model. The three inputs
shown are used to mimic the time variation of thermal conductance
encountered during a scan of photoresist lines on a Si substrate.
Thermal conductance changes smoothly in a real scan, but the
variation should have a non-zero and finite bandwidth to test the
circuit for signal distortion. The sum of these inputs is multiplied by
the temperature bias of the tip to calculate the power variation in the
probe. This variation, which would otherwise modulate the
bolometer, is instantly compensated by the interface circuit which
keeps the probe temperature constant.
An important optimization parameter for simulations is the
ratio of VDC to VA C in eqn. (1). As VA C increases, modulation of
probe resistance by the second harmonic of applied power cannot be
ignored. Additionally, simulations show that the PI controller loses
its feedback control, even though the signal-to-noise at the output of
demodulator becomes better in a certain range of VAC. The probe
temperature is supposed to be almost constant despite small AC
temperature variations introduced for dither operation by the PI
controller. However, as VAC increases, power supplied by the AC
component becomes comparable to DC power, causing the tip
temperature to fluctuate significantly. Now the PI controller
receives a significant AC signal in addition to the DC signal that is
the differential output from the resistor bridge. The output of PI
controller thus contains not only DC compensation power but also a
significant amount of unnecessary AC power, which derails the PI
feedback control. A low pass filter can be placed between bridge
circuit and PI controller to avoid this problem. However, it is only
useful when the dither frequency is much higher than bandwidth of
the scan signal from the bridge. In the simulated system the
mimicked signal at the input of the system contains frequency
components at higher frequencies than the dithering signal.
Figure 4(a) represents a noiseless input to the system. When
low frequency noise exists at 100 Hz, with a 20% variation in
bolometer resistance the bridge output is deteriorated in the absence
of electrical dithering (Fig. 4(b)). In contrast, the output of
demodulator (Fig. 4(c)), which is used with electrical dithering,
shows a much better signal-to-noise ratio. However, the output of
demodulator can be distorted because high-frequency components of
the input signal can be inadvertently screened by band pass filters
with high quality factor. This motivates the use of the highest dither
frequency (and thus a fast thermal response) to secure the maximum
signal bandwidth.
IV. MEASUREMENT
Insets in Fig. 3 show frequency spectra at various circuit nodes
taken while scanning a photoresist sample at a tip temperature of
45°C. Figure 3(a) shows that at the output of the bridge circuit,
where the second harmonic contains the pursued power-modulated
thermal signal, the amplitude ratio of the first harmonic to the
second is 24.6, which is very close to the theoretical value of 25
obtained from eqn. (3). This demonstrates that the bandwidth of the
thermal probe can be wider than 2 kHz and the 2 kHz-dithered
signal is not distorted by thermal delay. Figure 3(b) shows that the
band pass filtered signal has a dominant second harmonic. Filters
with higher Q-factor can be used to suppress other harmonics, but
could cause signal distortion due to reduced bandwidth. Figure 3(c)
shows the output of the frequency doubling circuit, which serves as
the local oscillator in demodulation. The dominant 2 kHz harmonic
is obtained using a high Q-factor band pass filter. Figure 3(d) shows
the multiplier output, where the DC component contains
demodulated thermal signal. The output of the low pass filter shows
that other harmonics can be effectively removed (Fig. 3(e)).
Figure 5 is a comparison of the thermal image with the
topographic image obtained using closed loop interface circuit
during measurements shown in Fig. 3. The sample was 350 nm
thick, developed Shipley UV6TM photoresist, with a 1 µm pitch. The
similarity between the two images is self-evident. The somewhat
flatter top seen for the ridges in the thermal image is as expected
because of the thermal diffusivity of the sample. According to a line
scan across the photoresist patterns of Fig.5, the noise-limited
minimum detectable thermal conductance change is ≈29 pW/K.
V. CONCLUSION
A scanning thermal imager with micromachined bolometer
type probes and a custom interface circuit was described. Unified
simulation of the transducer and circuit permits the components to
be optimized together. The probe temperature can be precisely
controlled by a PI controller while electrical dithering provides
relative immunity to thermal bridge noise even for sub-µV lowfrequency signals. Scanning thermal images obtained showed a
high signal-to-noise ratio of 6 for 350nm UV photoresist in which
the minimum detectable thermal conductance change was ≈29
pW/K.
ACKNOWLEDGEMENTS
This work was funded in part by the Semiconductor Research
Corporation contract # 98-LP-452.005. Valuable discussions with
Profs. F. Cerrina and P. Nealey, and Drs. M.-H. Li and R. Tate, all
of UW-Madison, and Dr. L. Ocola of Argonne National Laboratory
are gratefully acknowledged.
REFERENCES
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[Fe97] V.Ferrari, C.Ghidini, D.Marioli, A.Taroni, “Conditioning circuit for
resistive sensors combining frequency and duty-cycle modulation on the
same output signal,” Measurement Science & Technol., 8(7), pp. 827-9, ‘97
[Gi76] B.Gilbert, “A versatile monolithic voltage-to-frequency converter,”
IEEE J. Solid-State Circuits, SC-11(6), pp. 852-64, 1976
[Gi97] Y. Gianchandani, K. Najafi, "Scanning Thermal Profilers with
Integrated Sensing and Actuation," Transactions on Electron Devices, 44
(11), Nov. 1997, pp. 1857-68
[Go93] F.M.L. van der Goes, P.C.De Jong, G.C.M.Meijer, “Concepts for
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(Transducers ’93), Yokohama, Japan, 1993, pp. 331-4
[Hu87] J.H.Huijsing, G.A.Van Rossum, M.van der Lee, “Two-wire bridgeto-frequency converter,” J. Solid-State Circuits, SC-22(3), pp. 343-9, 1987
[Le00] J. Lerchner, D. Caspary, G.Wolf, "Calorimetric detection of volatile
organic compounds," Sensors and Actuators B, 70, pp. 57-66, 2000
[Li00] M.-H.Li, Y.B.Gianchandani, "Microcalorimetry applications of a
surface micromachined bolometer-type thermal probe," J. Vac. Sci. Technol.
B, 18(6), pp. 3600-3, 2000
[Li01] M.-H.Li, J.J.Wu, Y.B.Gianchandani, "Surface micromachined
polyimide scanning thermocouple probes," J. Microelectromech. Sys., 10(1),
pp. 3-9, 2001
[Li02] M.-H.Li, J.-H.Lee, F.Cerrina, A.K.Menon, Y.B.Gianchandani,
“Chemical and biological diagnostics using fully insulated ultracompliant
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[Mo95] K.Mochizuki, K.Watanabe, “A linear resistance-to-frequency
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[Oc96] L.E.Ocola, D.Fryer, P.Nealey, J.dePablo, F.Cerrina, S. Kämmer,
“Latent image formation: Nanoscale topography and calorimetric
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14(6), pp. 3974-9, 1996
[Sa93] G.R.Sarma, “Analysis of a constant voltage anemometer circuit,”
Proc., Instrumentation and Measurement Technology Conf. (IMTC),
Waltham, MA, 1993, pp. 731-6.
[Ve00] P.Vettiger, et al., “The “Millipede”-More than one thousand tips for
future AFM data storage,” IBM J. Res. Develop., 44(3), pp. 323-40, 2000
[Yo00] D.J.Yonce, P.P.Bey Jr., T.L.J.Fare, “A DC autonulling bridge for
real-time resistance measurement,” Trans. Circuits & Systems, 47(3), pp.
273-7, 2000
0
DEMODULATOR
R B13
GND
R B23
+
R B12
Vinteg R
B11
-
(a)
OP3
0
-
-
-
OP6
+
AC
2
3
4
Frequency (kHz)
X2
5
C B31
-
C ps
+
R B31
OP 9
R ps3
+
OP 11
+
2
3
4
Frequency (kHz)
R L23
C L21
C L12
R L12
R L14
-
-
R L21
OP15
-40
-60
-100
+
C L22
R L24
-
+
G=12, Q=0.7
f 3dB=200 Hz
V out
OP18
+
(e)
Low Pass Filter 2
0
(c)
-20
RL24
-
R L22
OP16
+
G=3, Q=0.7
f 3dB =200 Hz
(c)
5
OP17
-80
Band Pass Filter 3
1
RL24
Low Pass Filter 1
G=10, Q=50
Frequency Doubler
0
+
0
R B34
R B34
-
-120
R L14
+
+
RB32
5
-
OP13
OP10
R ps1
OP 12
-
-80
(e)
-20
VLPF (V)
1
R B34
RB33
Phase Shifter
R ps2
(d)
Vdoubler (V)
0
C L11
Vmod
OP8
-60
OP14
C B32
-60
2
3
4
Frequency (kHz)
R L13
(b)
R L11
-40
-80
1
R L14
(a)
-20
-100
R B24
+
+
0
R B24
RB22
R B2
OP5
+
-120
(d)
-40
-100
+
C B21
R B14
R B14
-80
-100
-
6
-60
OP7
OP4
C B11
Vbridge (V)
Band Pass Filter 2
CB22G=5, Q=15
R B24
-40
-20
Vmixer (V)
Band Pass Filter 1
CB12 G=5, Q=15
R B14
-Vs
VBPF (V)
+V s
0
(b)
-20
-40
-60
-80
-100
0
1
2
3
4
Frequency (kHz)
5
-120
0
0.5
1
1.5
Frequency (kHz)
2
Fig. 3: Demodulator section of the interface circuit in Fig. 1. Embedded frequency spectra were obtained while scanning a real sample, not in
a test mode. The scan results are present in Fig. 5.
Fig. 4: State diagram for the scanning thermal microscopy system including thermal response of the probe and the electrical behavior of the
circuit. The MatLab™ Simulink tool is used to optimize the circuit and evaluate overall performance, including noise immunity.
Fig. 5:
Topographic (left)
and thermal (right)
images of
developed 350 nm
thick UV6TM
photoresist scanned
at 45°C tip
temperature.
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