"Ka-Band Miniaturized Quasi-Planar High

"Ka-Band Miniaturized Quasi-Planar High
Ka-Band Miniaturized Quasi-Planar
High-Q Resonators
Kenneth J. Vanhille, Student Member, IEEE, Daniel L. Fontaine, Christopher Nichols, Member, IEEE,
Zoya Popović, Fellow, IEEE, and Dejan S. Filipović, Member, IEEE
Abstract—Air-filled copper
-band resonators with heights
between 250–700 m are demonstrated with measured unloaded
factors between 440–829. The low-profile quasi-planar
resonators are fabricated using a sequential metal deposition
process. Miniaturization of up to 70% with respect to a TE101
quasi-planar resonator is accomplished by reducing the footprint
of the resonator while maintaining the height. Finite-element
simulations predict the measured resonant frequency within less
than 1%. Due to fabrication imperfections, un is measured to
be approximately 15% less than predicted. The resonators are
compatible with air-filled rectangular microcoaxial feed lines
fabricated in the same metal-deposition process.
Index Terms—Cavity resonator, coaxial transmission line, photolithography, quality ( ) factor.
HE miniaturization of microwave resonators, including
reentrant cavity designs, has been of interest since the
development of microwave engineering [1]. Work with filters
using evanescent waveguide for miniaturization and suppression of spurious modes with the ability to do wideband filters
is presented in [2]. More recently, interest in microfabricated
microwave and millimeter-wave resonators using various
techniques has been seen. The loading of -band surface
micromachined cavity resonators with barium titanate and alumina is examined in [3]; ’s in the hundreds are demonstrated
with fabrication tolerances being the most important limiting
factor on performance. Silicon micromachining of Ku-band resonators with drain holes used to release the sacrificial material
are fabricated as outlined in [4]; the electrical performance is
decreased by a small amount of leftover silicon in the resonator
after the processing. Silicon micromachined and metal-plated
layered polymer resonators with miniaturization factors up to
-bands are presented in [5]. More recently,
70% at - and
tuning from 5.4 to 10.9 GHz of an evanescent-mode resonator
was demonstrated [6]. The resonators presented in [5] and
[6] use rib-shaped interdigital capacitor loading (among other
Manuscript received September 29, 2006; revised February 3, 2007.
K. J. Vanhille, Z. Popović, and D. S. Filipović are with the Department
of Electrical and Computer Engineering, University of Colorado at Boulder,
Boulder, CO 80309-0425 USA (e-mail: [email protected];
[email protected]; [email protected]).
D. L. Fontaine is with the Advanced Systems and Technology Division, BAE
Systems, Nashua, NH 03060 USA (e-mail: [email protected]).
C. Nichols is with Rohm and Haas Electronic Materials LLC, Blacksburg,
VA 24060 USA (e-mail: [email protected]).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TMTT.2007.895232
techniques), and are an excellent starting point for the designs
- and -band rectangular coaxial transpresented here.
mission line filters with nickel walls using the electrochemical
fabrication (EFAB) process have been shown in [7] and [8],
respectively. In this study, the resonators are not characterized
alone; however, the largest theoretical ’s that can be obtained
from nickel coaxial resonators with the same dimensions as in
[7] and [8] are less than 80. The same technology with gold
plating has recently demonstrated factors near 250 at 60 GHz
[9]. Laser etching of copper has been used to fabricate rect-band filters [10]. 2.5-D resonators using
angular coaxial
U-shaped half-wavelength resonators in a cavity operating
band have been fabricated using a low-temperature
in the
co-fired ceramic (LTCC) implementation [11]. This method
allows compact filter design by including multiple resonators
in the same cavity.
Presented here are loaded-cavity resonators fabricated
using a planarizing sequential-layering process that enables
simultaneous fabrication of integrated rectangular coaxial
transmission lines. Microfabrication of air-filled rectangular
coaxial transmission lines and other components allow high
isolation, dense circuit integration, multilayer topologies,
low-loss performance, and dominant TEM propagation into the
400-GHz range [12]. Rectangular coaxial transmission lines,
as well as other air-loaded surface-micromachined passive
structures, such as high-quality factor inductors, have been
demonstrated in the millimeter-wave frequency range [13].
Transmission lines, multiport couplers, and bandpass filters
have been fabricated using nickel air-filled coax lines [14].
Quasi-planar microfabricated copper branch-line couplers and
transmission-line resonators have been recently presented in
[15] with improved electrical properties up to
Results for several quasi-planar resonators fabricated using
the process presented here are given in [16]. Initial results for a
miniaturized quasi-planar resonator are introduced in [17]. This
paper gives a comprehensive analysis, design, and measurement
study for miniaturized resonators at different frequencies and
with different loadings. Sensitivity studies of the release holes,
new resonator topologies, and behavioral circuit models are presented. Fig. 1(a) shows an example of a miniaturized cavity
resonator with an arbitrary loading mechanism. Two types of
predominately capacitive loading are presented in this paper.
Fig. 1(b) shows what we refer to as the “puck-loading” used
at 26 GHz and Fig. 1(c) shows the 36-GHz “rib-loading” structure, described in detail in Section II.
It is worth clarifying the meaning of “quasi-planar” in the
for a
context of this study. Fig. 2 shows simulated values of
0018-9480/$25.00 © 2007 IEEE
VANHILLE et al.:
Fig. 3. Basic fabrication steps for miniaturized quasi-planar resonators. The
first eight steps (S1–S8) are identical for the five- and ten-layer structures.
Fig. 1. (a) Sketch of a loaded quasi-planar resonator showing the loading, relevant dimensions, and resonator features. (b) Sketch of the metal puck loading
in the 26-GHz resonator. The four support posts are also shown. (c) Half-view
sketch of the ribs in the 36-GHz loaded cavity resonator. The ribs are cut along
the xy -plane to simplify their presentation. The relevant dimensions of the resonators are given in Table I.
allows resonators and microcoaxial feeds to be fabricated in the
same process, while simultaneously providing factors higher
than those of coaxial transmission line resonators of the same
This paper is organized as follows.
• Section II details the design of the resonators. Brief descriptions of the fabrication process, and the resonator
feeding and probing structures are also given.
• Section III provides a portion of the analysis necessary to
understand the various design parameter effects on the performance of the resonators.
• Section IV describes a method for deriving a behavioral
circuit model of the loaded cavity resonators.
• Section V shows the measurement results for the designed
• Section VI discusses a few final points and indicates potential directions for future research.
Fig. 2. Percentage of Q of a resonator with respect to the maximum Q of
metallic resonator as a function of normalized cavity height h =w for
a TE
several metals. The measured Q values for 26- and 36-GHz fabricated cavity
resonators are given.
square cavity with a footprint
on the side, and with
different metal conductivity values. The height of the cavity ( )
is varied from a small value to . The measured unloaded
factors for two unloaded quasi-planar cavity resonators are indicated in Fig. 2 for comparison. The normalized heights of the
, justifying the terminology
two cavities are
“quasi planar.” “Low profile” describes these resonators with
regard to their height equally well. The small electrical height
The PolyStrata process developed by Rohm and Haas Electronic Materials LCC, Blacksburg, VA, consists of sequentially
depositing layers of metal and photoresist with high aspect ratios (2 : 1) [18]. The resonators are fabricated on a low-resistivity silicon substrate using a sequence of standard photolithographic steps, as depicted in Fig. 3. Each copper layer is chemically polished before the next layer is deposited. The final step
involves removing the photoresist, leaving microfabricated airfilled copper structures.
Devices with up to ten layers are demonstrated in this paper.
The fabrication and resulting mechanical parameters dictate the
electrical design. The major fabrication-related factors and their
influence on component design are as follows.
1) Total resonator height is limited by the number of layers
and thickness of each layer. In our case, with ten layers, a
maximal cavity height of 700 m is possible, which will
factor as presented in Fig. 2. The electrical
limit the
height of the tallest resonator presented in this paper is one
twelfth of a free-space (TEM) wavelength.
Fig. 5. Normalized Q for different sized release holes on a cavity resonator.
The surface area of the release holes is kept constant with respect to the top
surface area of the cavity. This is shown for the 250-m-tall cavities at 26 GHz
and 700-m-tall cavities at 36 GHz.
Fig. 4. (a) Puck-loaded cavity resonator taken from [17]. The ports are indi200 m. (b) Two cavity
cated by P1 and P2. The release hole size is 200
resonators fabricated for operation at 36 GHz. The resonator at left (R1) is full
sized, and the resonator at right (R2) uses miniaturization. The release hole size
is 400 400 m. Probing of the structure is done on the ends of the resonator at
ports indicated by P1 and P2. The loading region in each resonator is indicated
by the dashed lines.
2) Photoresist release holes on the top wall and on the sidewalls limit the factor. However, a relatively large number
of holes are required to completely evacuate the lossy dielectric from the cavity. The size and number of release
holes were carefully examined within fabrication parameters and are described in Section II-A.
3) The design of the loading to reduce the size of the resonator is limited to what can be fabricated with this technology. The main limitation was the size of the loading
structure, which needed to be mechanically fixed to the
top wall while providing enough loading. We could not use
previously published designs, as given the mechanical constraints, they would not provide a sufficiently large loading
reactance. This constraint resulted in a 3-D capacitance,
which is unique to this paper and differs from the fins presented in [5].
4) The aspect ratio of a thin cavity is such that the top wall can
sag or bulge. The mechanical support posts, which increase
the footprint of the 26-GHz loaded cavity, are eliminated
in the 36-GHz design because of the miniaturization with
the 3-D loading.
5) The inductive coupling between the input -coaxial cable
and cavity is constrained by the height of the center conductor and the required impedance match; the analysis is
detailed in Section II-C.
Results from two sets of wafers are given in order to show the
fabrication constraints on the resonator factor, i.e., 1) a wafer
with coaxial components fabricated in a five-layer process with
design frequencies near 26 GHz and 2) a wafer with 36-GHz
components made using a ten-layer process. Resonators from
the two wafers are shown in Fig. 4.
A. Design of Release Holes
In order to remove lossy photoresist after the top metal layer
is deposited, release holes in the top layer and on the sidewalls
are necessary. Electrically, these holes perturb the current flow
and, therefore, affect the resonant frequency, as well as the
factor. While no holes or small holes will give the highest
factor, many larger holes are needed for high fabrication yield,
thus an optimization study is required. The release holes on the
sides of these resonators do not have a large effect on the electrical performance because of the cavity heights used so only
the effects of the top holes will be examined. Fig. 5 shows the
results of a 3-D finite-element method (FEM) study, using the
Ansoft High Frequency Structure Simulator (HFSS) version 10,
of a cavity resonator versus the number of rows of holes
). The
on the cavity resonator top surface (total holes
ratio of the surface area of the release holes to the total top-wall
surface area of the resonator is kept constant. For what is fabricated, this ratio is 8.74% for the five-layer 26-GHz designs
(250- m cavity height) and 21.80% for the ten-layer 36-GHz
designs (700- m cavity height). These numbers are dictated
200 m square release holes at 26 GHz
by the use of 200
and 400 400 m square release holes at 36 GHz
. Fig. 5 shows curves obtained numerically for both
ratios at both design frequencies. The resonant frequency of the
resonators changes slightly as the number of holes increases. To
for a fixed frequency, it is necescompare these changes in
sary to normalize the values.
We begin with the formula for due to the conductor losses
of a
rectangular metallic cavity resonator, as can be found
in [19]
where is the wavenumber, is the wave impedance of the
is the surface resistance of the
cavity-filling medium, and
cavity walls. The length, width, and height of the cavity are ,
VANHILLE et al.:
, and , respectively. We assume that and are equal and that
the cavity is air filled as follows:
To compare two resonators with the same size, but different
configurations of release holes, it is necessary to normalize the
calculated unloaded quality factors based on the frequency of
resonance. If the cavity dimensions and materials do not change,
we find the ratio of quality factors to be related to a power of the
ratio of the resonant frequencies
The data shown in Fig. 5 are simulated for two frequencies
and two top plate surface area to release hole surface area ratios
using the normalization in (3). The cavity is 250- m tall for
the 26-GHz resonator and 700- m tall for the 36-GHz resonator.
We see that the resonator performance is minimally affected by
the release hole size when the cavity is 250- m tall; however,
the unloaded of the cavity is large enough in the 700- m-tall
case in that an effect on the is seen for large hole sizes.
B. Mechanical Support
The values of the dimensions labeled in Fig. 1 are given in
Table I. The puck-loaded cavity contains four mechanical support posts that connect the top and bottom cavity walls. The
added mechanical strength comes with a penalty in size, as the
posts behave as inductances in parallel, decreasing the overall
inductance of the cavity and increasing the necessary cavity
footprint for a given frequency [17]. The metal rib loaded cavity
contains two ribs connected to the bottom cavity wall and a
single rib hangs from the top cavity wall. A cross configuration
is used with the ribs to increase the strength of the connection of
the ribs to the top and bottom cavity walls. The ribs have the effect of increasing the capacitance of the cavity, operating much
like an interdigitated capacitor. The rib widths that are not explicitly labeled in Fig. 1(c) are 100 m in size.
C. Coupling
The resonators are measured in a two-port topology with
probing structures in the regions marked P1 and P2 in Fig. 4.
150- m-pitch ground–signal–ground (GSG) probes are placed
in contact with a short section of vertical rectangular coax, as
shown in the figure, where the relatively complex geometry
Fig. 6. (a) Top-view photograph of the probing area for the 36-GHz resonator.
(b) Cross-sectional view of the port area (courtesy of the Mayo Foundation).
(c) 3-D model of the resonator feed.
serves for impedance matching by reactively tuning the discontinuity parasitics. The center conductor of the rectangular
coax in the probing region bends downwards to connect to the
bottom wall of the cavity, resulting in inductive coupling, as
shown in Fig. 1(a). A top view of one of the probe structures
of the 36-GHz resonator is shown in Fig. 6(a). A wafer is diced
down the center line of the rib-loaded resonators to reveal the
cross-sectional view shown in Fig. 6(b), giving the sidewall
profiles. An indication of the detailed analysis involved in
producing high-quality transitions to the on-wafer environment
is shown in Fig. 6(c), which gives the computer-aided design
(CAD) model of the resonator probe port.
Comprehensive numerical studies are conducted to examine
the effects of several of the design parameters that affect the
resonator performance. The more efficient eigenmode analysis
is applied to examine parameter effects and trends, while the
driven analysis provides frequency-dependent -parameters.
The loading effect of the capacitive puck and the more complex
3-D metallic ribs is addressed here.
Fig. 7 shows the results of a numerical study of the size of
the capacitive loading puck of the 26-GHz resonator for puck
heights of 75 m and 175 m, corresponding to the two possible values using the five-layer process. The position and size
of the support posts and the overall dimensions of the resonator
are kept constant. The fabrication process allows three possible
discrete heights (0, 75, and 175 m), as there are three layers
that make up the actual cavity (two more layers exist, one for
the top wall and one for the bottom wall, making a total of five
layers). The tallest height is chosen to maximize the factor,
whose theoretical value is less than 2% lower than the highest
ideal value for a copper resonator without release holes.
Fig. 7. Results of a parametric analysis of a simplified model of the 26-GHz
resonator. The release holes are not included for this analysis. The FEM eigenmode analysis examines how different diameters for the loading puck affect the
Q and f of the resonator. Two puck heights (75 and 175 m) corresponding
to the two possible heights using the five-layer fabrication process are examined.
Values corresponding to the fabricated resonator are indicated on the graph.
Fig. 9. Equivalent circuit is generated for the resonator with feed combination
for the miniaturized resonator using a loading puck and support posts. (a) Circuit
topology used to create the model. (b) Measured input impedance of the resonator/feed combination compared to the input impedance of the circuit model
derived from the measured data.
Fig. 8. Results of a parametric analysis of the 36-GHz resonator. The analysis
examines different spacings of the loading ribs S , as given in the inset. Eigenmode analysis using the FEM is employed, and values corresponding to the
fabricated resonator are indicated on the graph.
Fig. 8 presents the results of a parametric study examining
the effect of the rib spacing on
of the resonator.
The values corresponding to the dimensions used for the fabrication are highlighted on the graph. As the rib separation is
increased, the capacitance between the ribs diminishes; however, the capacitive coupling from the ribs to the opposite wall
of the cavity remains strong. The scaling of the two -axes is
the same relative to the first point of each data set, emphasizing
the frequency changes more rapidly than
over the range of
separation values considered.
Returning to Fig. 4, the puck-loaded resonator, shown at
the top of the figure, has a footprint that is 15% smaller than
a normal cavity resonator and 50% smaller than a resonator
using the four support posts [17]. A photograph of an unloaded
ten-layer 36-GHz resonator is shown on the left in Fig. 4(b).
A miniaturized rib-loaded resonator operating at the same
frequency, but with a 71% reduction in surface area, is shown
on the right-hand side of this figure.
Though full-wave FEM simulations are accurate and flexible
for detailed studies, they are computationally expensive. For
example, on a dual-processor Intel Xeon 3.4 machine, it takes
four minutes per frequency point with a 70-k tetrahedron mesh.
Therefore, behavioral circuit models are developed for the two
loaded resonators and compared with the measured results to
validate this approach.
A circuit model, as shown in Fig. 9(a), is developed for the
two-port resonator. The steps for extracting the model are similar to those outlined in [20]. Once the capacitance of the gap
between the puck and top wall is approximated using quasistatic analysis, the other circuit parameters can be determined.
The other parameters of interest are the series resistance of the
, the turns ratio , which is approximately the coufeed
pling coefficient, the total inductance of the support posts ,
the characteristic impedance of the lines connecting to the resonator , and the nominal capacitance, resistance, and inductance of the cavity ,
, and , respectively. Starting values
for these parameters are found using full-wave analysis, and
then the parameters are fit to correspond to the measured results.
Good agreement can be seen between the measured response
and the derived equivalent circuit [see Fig. 9(b)]. The calculated
values for the circuit parameters are
pH, and
k . A similar exercise using the
simulated -parameter data from the full-wave response yields a
circuit model whose input impedance is indistinguishable from
that calculated in the simulation.
Fig. 10(a) shows the model and Fig. 10(b) shows the input
impedance predicted by the model, as compared with the sim-
VANHILLE et al.:
Fig. 10. Equivalent circuit is generated for the resonator with feed combination
for the miniaturized resonator using three ribs for miniaturization. (a) Circuit
topology used to create the model. (b) Simulated input impedance of the resonator/feed combination and the input impedance of the circuit model derived
from the simulated data.
ulated input impedance for the rib-loaded resonator. A comparison with the simulated results demonstrates that the circuit model predicts the measured response and agrees with the
full-wave simulations. The circuit model has one inductor less
than the model in Fig. 9(a), as there are no support posts in this
, is caldesign. The loading capacitance from the ribs, i.e.,
culated using Ansoft’s Maxwell 3D, as the approximation of
the capacitance is more involved than what was done for the
loading puck. This method calculates the electrostatic capacitance, but this is a good approximation since the dimensions of
at the resonant frequency.
the loading ribs are less than
is 0.068 pF. The nominal capacitance of the resonator is
pF, the nominal inductance is
k , the turns ratio is
the nominal resistance
, and the series resistance of the feed is
An examination of the derived circuit models of the two
miniaturized resonators presented in Figs. 9(a) and 10(a) reveals that quite similar behavioral models describe the two
resonators with seemingly different loading topologies. This
approach can be generalized to other possible quasi-planar
loading topologies.
The resonators are measured using 150- m-pitch probes
from Cascade Microtech on a Cascade Summit 9000 probe
station connected to an HP-8510C network analyzer. An external short-open-load-thru (SOLT) calibration on an alumina
substrate is performed. A comparison of the measured and
simulated results for the two miniaturized resonators is given
for the
in Table II. The simulated and measured values of
puck-loaded resonator are 508 and 442, respectively. Compared
Fig. 11. S -parameter transmission data for the 26-GHz resonator taken from
[17]. A comparison of measured, circuit model, and simulated results is shown.
The simulated resonator is modeled using HFSS. The offset in frequency between the measured and simulated results is less than 0.7%.
Fig. 12. S -parameter transmission data for the 36-GHz resonator. A comparison of measured, circuit model, and simulated results is shown. The simulated
resonator is modeled using HFSS. The offset in frequency between the measured and simulated results is less than 0.4%.
to a cavity resonator with four support posts, but no capacitive
loading, the 50% reduction in footprint results in virtually no
for a cavity resreduction in quality factor. The simulated
onator using this technology is 541; therefore, a 15% footprint
reduction with four supporting posts is realized by sacrificing
. The values of
for the rib-loaded resonator
6% of
are 995 and 829 for the simulated and measured results. This
of 1308 for a full-sized cavity
compares to a simulated
resonator with the same cavity height. A 71% reduction in
footprint is thus achievable for a 25% reduction in
The measurement results of the puck-loaded 26-GHz resfrom the full-wave simulation
onator are shown in Fig. 11.
and the equivalent-circuit model of Fig. 9(a) are also plotted
for comparison. The resonant frequency of the measured resonator differs by less than 0.7% from the predicted value. The
for the rib-loaded 36-GHz resonator
measurement results of
are given in Fig. 12. Again, a comparison is made between the
full-wave simulation and the derived equivalent-circuit model;
the deviation of the measured from predicted resonant frequency
is less than 0.4%.
Results from two miniaturized resonators have been presented
and their performance and size has been compared to that of
full-sized cavity resonators operating at the same frequencies.
The puck-loaded resonator achieves a 15% miniaturization in
the footprint of the resonator, while including mechanical supporting posts (this gives a 50% reduction in footprint compared to
a cavity with just the support posts). The rib-loaded resonator has
a 71% miniaturization in the cavity footprint compared to a stanresonator. The electrical effects of release holes and
their size and placement are studied. The electrical phenomena
resulting in the miniaturization of the resonators are explored
and the quality factors and cavity footprints are compared to
standard cavity resonators. Behavioral equivalent circuits are
derived and their usefulness validated by measurements. From
here, one could develop filters, using the circuit models as the
building blocks for rapid synthesis. Such resonators can also
be integrated with rectangular coaxial lines and active elements
for circuits that require high- factors such as low phase-noise
oscillators and frequency diplexers.
The authors would like to thank G. Potvin, BAE systems,
Nashua, NH, D. Sherrer and the Rohm and Haas Microfabrication Team, Blacksburg, VA, W. Wilkins and her measurement team, Mayo Clinic, Rochester, MN, Dr. J. Evans, Defense
Advanced Research Projects Agency (DARPA), Arlington, VA,
and E. Adler, Army Research Laboratory (ARL), Adelphi, MD,
for their support. The authors would also like to thank M. Lukić,
and Dr. S. Rondineau, both with the University of Colorado at
Boulder, for helpful technical input.
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Kenneth J. Vanhille (S’00) received the B.S. degree
in electrical engineering from Utah State University,
Logan, in 2002, the M.S.E.E. degree from the University of Colorado at Boulder, in 2005, and is currently
working toward the Ph.D. degree at the University of
Colorado at Boulder.
From 2000 to 2003, he was with the Space
Dynamics Laboratory, Logan, UT, where he designed space science instrumentation for sounding
rocket campaigns. In 2002, he was a member of
the National Aeronautics and Space Administration
(NASA) Academy, Goddard Space Flight Center. His current interests include
millimeter-wave components and antenna design.
Daniel L. Fontaine was born in Holyoke, MA, on
February 17, 1966. He received the B.S. and M.S.
degrees in electrical engineering from the University
of Massachusetts at Amherst, in 1988 and 1991,
From 1988 to 1996, he was a Senior Design Engineer with the Raytheon Company, Tewksbury, MA.
Since 1996, he has been a Principal Design Engineer
with the Advanced Systems and Technology Division, BAE Systems, Nashua, NH. His professional
design experience and interests include microwave
and millimeter-wave patch antennas and arrays, quasi-optical feed networks,
transmit/receive (T/R) modules, and frequency-selective surfaces.
VANHILLE et al.:
Christopher Nichols (M’03) received the B.S. degree in physics from Arkansas State University, State
University, in 1990, and the M.S. degree in physics
and Ph.D. degree in applied science from The College of William and Mary, Williamsburg, VA, in 1992
and 1996, respectively. His doctoral dissertation involved the engineering of a novel hyperthermal neutral stream etch process tool for charge-free wafer
Prior to graduation, he was with IBM, Yorktown
Heights, NY, where he was involved with ionized
physical vapor deposition. He is currently a Senior Engineer and Microfabrication Program Manager with Rohm and Haas Electronic Materials LLC,
Blacksburg, VA.
Zoya Popović (S’86–M’90–SM’99–F’02) received
the Dipl.Ing. degree from the University of Belgrade,
Belgrade, Serbia, in 1985, and the Ph.D. degree from
the California Institute of Technology, in 1990.
She is currently the Hudson Moore Jr. Chaired Professor of Electrical and Computer Engineering with
the University of Colorado at Boulder. Her research
interests include high-efficiency and low-noise microwave circuits, quasi-optical millimeter-wave techniques, smart and multibeam antenna arrays, intelligent RF front ends, RF optics, and wireless powering
for batteryless sensors.
Dr. Popović was the recipient of the 1993 and 2006 IEEE Microwave Theory
and Techniques Society (IEEE MTT-S) Microwave Prize for best journal papers. She was also the recipient of the 1996 URSI Issac Koga Gold Medal, a
2000 Humboldt Research Award for Senior U.S. Scientists, and a 1993 National
Science Foundation (NSF) Presidential Faculty Fellow Award.
Dejan S. Filipović (S’97–M’02) received the Dipl.
Eng. degree in electrical engineering from the University of Nis, Nis, Serbia, in 1994, and the M.S.E.E.
and Ph.D. degrees from The University of Michigan
at Ann Arbor, in 1999 and 2002, respectively.
From 1994 to 1997, he was a Research Assistant
with the University of Nis. From 1997 to 2002,
he was a Graduate Student with the University of
Michigan at Ann Arbor. He is currently an Assistant
Professor with the University of Colorado at Boulder.
His research interests are in the development of
millimeter-wave components and systems, multiphysics modeling, antenna
theory and design, as well as in computational and applied electromagnetics.
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