© Copr. 1949-1998 Hewlett-Packard Co.
A New Programmable, Building-Block
Pulse and Digital System
A pulse generator system consisting of a series of plug-ins
that can be combined to provide a wide variety of digital
test signals. Variable rise and fall time pulses and digital
words in a number of formats are among its capabilities.
By Gordon K. Blanz and Ronald L. Knauber
vide a total of 300 watts.
requires compatibility with
Any supply is available to
state-of-the-art develop
any plug-in via connectors
ments and components and
on a motherboard. There
techniques. There is a need
are two 10- volt unregulated
for a variety of pulse genera
supplies, two 25-volt fixed,
tion, timing and shaping
regulated supplies and two
methods for easier digital
20-to-70 volt variable regu
systems design, for remote
lated supplies. To minimize
operation of digital equip
use of power, the variable
ment, and at the same time
supplies were designed with
a need to reduce the effects
switching regulators, Fig. 2.
Fig. 1. Four of the building blocks of the Model 1900
of electrical radiation. To
This circuit is easily pro
Pulse System are shown here. They are the Model 1900A
meet these needs the HP
grammed to supply large
Mainframe, the Model 1905A Rate Generator, the Model
Model 1900 Pulse System,
1908A Delay Generator and the Model 1915A Variable
currents at variable voltages.
Transition Time Output.
Fig. 1 was developed as a
To reduce electromagnetic
versatile plug-in system. At
radiation in the system en
present it consists of two mainframes and seven plug-ins.
vironment, several techniques are used, Fig. 3. A line
Others are planned. It was designed to be electronically
filter is used as well as two inner and two outer top and
programmable as an option, and special care was taken
bottom covers. There are gaskets between plug-ins and
to reduce radiation.
mainframes, and beads between V4 -module plug-ins.
The two mainframes, the key to the system flexibility,
Also die castings are used for plug-in front panels.
are the Models 1900A and 1901 A. The Model 1901 A
omits two high voltage variable power supplies included
in the Model 1900A.
Plug-ins initially available include the Model 1905 A
The mainframes are five-inch high cabinets with four
Rate Generator, Model 1908 A Delay Generator, Model
compartments designed to accept any combination of 141910A Delay Generator, Model 191 5 A Variable Transi
module and V2 -module plug-ins (except two Model
tion Time Output, Model 1917A Variable Transition
19 ISA's). The Model 1900A has six supplies which proTime Output, Model 1920A 350 ps Transition Time
© Copr. 1949-1998 Hewlett-Packard Co.
Output and the Model 1925 A Word Generator. The
Model 1905 A has an internal clock rate of 25 Hz to
25 MHz in six decade ranges with a 10:1 continuous
vernier. It supplies a positive clock pulse of less than
10 ns pulse width. In the external mode it can be driven
from dc to 25 MHz with a 0.5 volt peak-to-peak positive
signal with selectable trigger level and slope. By over
driving, it will provide a 50 MHz clock rate for the
Model 1925 A. The Model 1905 A has a gating feature in
which the clock pulses are synchronous with the gating
signal. Programming allows remote control, computer
interfacing, and phase/frequency locked loops.
The Model 1908 A Delay Generator provides trigger
pulses and drive pulses (each similar to the Model 1905 A
rate output pulse) which are used for the system timing
signals. The Model 1908 A can be operated in any of the
following modes: drive pulse delay, drive pulse advance,
double pulse and programmed. The time interval varies
in six ranges from 15 ns to 10 ms with a 10: 1 continuous
vernier. The double pulse mode can be used to provide a
50 MHz pulse train.
The Model 1910A Delay Generator provides trigger
and drive pulses up to a repetition rate of 125 MHz. The
delay between trigger and drive pulses is available in
twenty ranges from 5 ns to 100 ns in 5 ns increments.
The Model 1910A has low jitter and can be used to ob
tain delays greater than a period at high repetition rates.
Varying Transition Time
Transition times variable from 7 ns to 1 ms with a
100:1 vernier can be obtained with the Model 19 ISA
Variable Transition Time Output. Variable transition
time is especially useful for testing of magnetic memory
devices and MOS integrated circuits. See page 5. The
leading edge and trailing edge transition times are de
termined by a capacitance-current source circuit, Fig. 4.
A synchronous switch alternately connects a charging
current source and a discharging current source to the
selected capacitance C to determine the rise and fall time
of the pulse. With no signal at the base of Ql, Ql and Q4
are off, Q2 and Q3 are on and trailing edge constant cur
rent IT linearly charges C to the baseline voltage deter
mined by Vc. Leading edge constant current IL flows
through Q2. A negative signal at the base of Ql turns on
Ql and Q4, turns off Q2 and Q3, and IL linearly dis
charges C through Q4 to clamp CR3. A positive signal at
the base of Ql reverses the procedure and returns the
output to its quiescent level. Assuming a fixed voltage
Vc, the time rate of change of the transition time output
voltage is:
Fig. 2. The voltage reference from a plug-in changes
the inverse current source through the adjustable bias.
The current controls the duty cycle of the astable multi
vibrator which in turn controls the operation of the
transistor switch. When the switch is closed, the LC
filter charges from the rectifier. When it is open, the
LC filter discharges. The voltage at the output of the LC
filter is variable from 20 to 70 volts. Since power is dis
sipated only during the switching interval and not during
the open and closed interval, power in the transistor
switch (the high-current path) is reduced.
d v I
— = -pr = constant since the charging currents are
(v = peak-to-peak output voltage, t = time required for
peak-to-peak voltage change, I = IL or IT, C = selected
capacitance.) IL and IT can be changed by a ratio of
Variable rise and fall time pulse generators can be
classified as either constant transition time or constant
slope as a function of amplitude change. Unlike the con
stant slope instruments, the Model 1915A provides the
highly useful constant transition time.
If the output pulse amplitude is to change without
changing the pulse width, the rise time or the fall time,
then both leading and trailing edge transition time cur-
Co ver: Two Model 1925 A Word Generators
are cascaded to produce a 32-bit word in a
nonreturn-to-zero format shown on the face of
a Model 143 A Oscilloscope.
In this Issue: A New Programmable, Building
Block Pulse and Digital System ¡page 2. Why
Use Variable Rise and Fall?; page 5. Generat
ing Words for Digital Testing; page 8. Fre
quency Domain Oscilloscope Now Measures
To 1250 MHz; page 14. The Meaning of 'Fre
quency Domain Oscilloscope'; page 16. Be
yond Traditional Spectrum Analyzer Uses;
page 18.
© Copr. 1949-1998 Hewlett-Packard Co.
òo <7Ã- = - - = —r- = constant
(k = /ro«Ã- parte/ amplitude vernier setting)
Fig. 3. Reducing RFI is accomplished
by this combination of techniques.
rents must be modified, as well as the output current, in
accordance with any change in amplitude. This is ac
complished by the amplitude vernier circuitry (see Fig.
5). As the amplitude is changed by the front panel ampli
tude vernier control, the pulse baseline clamp voltage in
the transition time circuit is changed. This results in a
change in waveform 1 amplitude VA. Simultaneously, the
leading and trailing edge currents which charge and dis
charge capacitor C are changed by the amplitude vernier
circuit to keep the transition times constant. From the
basic transition equation given previously:
The amplitude vernier circuit also changes the current in
the output current source I0 so the voltage across re
sistor R, VB, is proportional to the pulse amplitude. In
Fig. 5 on waveform 2, VB = I0RThe Model 1915A can provide 50 mA to 1 ampere
output current (or 50 volts maximum into 50 ohms) in
four ranges at a repetition rate from dc to 25 MHz. This
current is provided by five current sources which supply
four output differential amplifiers. A simplified positive
output amplifier and representative waveforms are shown
in Fig. 5. During the quiescent state Q7 is on and Q14
is off. When a positive pulse (waveform 2 for the positive
output amplifier) is applied to the base of Q7, the se
quence of turning on Q14 and turning off Q7 is started.
At about the 10% level of maximum amplitude of the
positive transition, Q14 turns on and conducts more
heavily as the amplitude increases. At about the 90%
level Q7 turns off completely, clipping the top 10% of
the pulse. As the negative transition of waveform 2 starts,
Q7 is fully off and Q14 is fully on. At about 90% of
maximum amplitude Q7 turns on. As the negative transi
tion falls toward the baseline, Q7 current increases and
Q14 current decreases. When the 10% level is again
reached, Q14 turns off, clip
ping the bottom 10% of the
pulse. This 'window^ framed
by the clipping levels and
represented by voltage VB in
Fig. 5, can be adjusted by
changing the base voltage of
Q 14. Its purpose is to main
tain a clean output pulse.
However, it causes a small
corner shift as amplitude is
Fig. 4. Linear leading edge and
trailing edge transition times
are produced by discharging
capacitor C with constant cur
rent /L and by charging the ca
pacitor with constant current I-,.
© Copr. 1949-1998 Hewlett-Packard Co.
Why Use Variable Rise and Fall?
A variable rise and fall pulse generator is an extremely val
uable tool in electronic circuit design. It can be used to
determine the effect of the speed of an input driving pulse
on a circuit output. For example, magnetic memory devices
are generally tested with pulses with linear rise and fall
times ranging from 10 to 700 ns. Drive currents up to 800 mA
are needed. Testing of cores in a magnetic core plane is
another application of a variable rise and fall pulse system.
To prevent excessive power dissipation in the output
amplifier at low output levels, the high voltage variable
supply level is controlled by a sample and hold circuit,
the peak detector, which samples the peak output volt
age and stores it on a capacitor (see Fig. 5). This peak
voltage controls the reference voltage input to the vari
able supply in the mainframe. It maintains the variable
supply (and thus the output base supply referenced to it)
at minimum for outputs less than 10 volts. Above 10
volts, both supplies change one volt for every one volt
change in the output. These supply changes keep the
output amplifier about five volts from saturation, which
reduces its power dissipation.
The Model 1915A has unique overload protection
circuits for both positive and negative output stages. The
instantaneous power in the output stage is monitored. If
excessive dissipation occurs, the output is disabled. At
A typical setup (below) is used to test the output of the
sense winding of a core.
In these scope photos, the upper waveform is the drive
signal (500 mA/cm) and the lower waveform is the sense
output (200 mV/cm). The horizontal calibration is 500 ns/cm.
Note that as the input drive risetime increases, output
amplitude increases, output delay decreases and reflections
appear and increase in amplitude.
the same time an overload light on the front panel is
lighted. It stays on until the overload is eliminated by
reducing the output amplifier dissipation with the front
panel controls.
Negative Overload Protection
The currents in current sources 2 and 4, Fig. 6, are
proportional to the currents in output current sources 1
and 3. Since V-, is at the same voltage as VB, the current
in Q30A, It, is proportional to Q 14 emitter current and
the current in Q29A, L, is proportional to Q14 basecollector voltage. Q29A, Q29B and Q30A are matched
transistors connected as diodes. The collector current of
Q30B, derived from basic diode voltage-current relation
ships in the lil-, multiplier circuit, is proportional to the
product of Ij and L. Therefore, the output voltage of
© Copr. 1949-1998 Hewlett-Packard Co.
Fig. leading This amplitude vernier circuit maintains constant leading
and trailing edge transition times as output amplitude is changed.
emitter follower Q3 1 will be proportional to the instan
taneous power in Q 14. Due to the 3 ns response time of
the diode multiplier, a small capacitor Cl was included
to prevent very short overload signals from energizing
the disable driver. During a longer overload, the negative
disable driver (a Schmitt trigger circuit) switches, disabling
the output current sources and energizing the overload
light. As I, and L are reduced, the circuit returns to its
quiescent state. The cycle repeats as long as the overload
continues. The repetition rate of the circuit is determined
by the time rate of discharge of C2 through Q3 1 and the
severity of the overload. It varies from 5 kHz to 30 kHz
(see Fig. 7).
Varying Pulse Widths
Internal pulse widths from 10 ns to 40 ms with a con-
Fig. current proportional currents, one proportional to output amplifier current and one proportional
to output amplifier base-collector voltage are multiplied in a diode multiplier circuit.
This product represents the instantaneous output amplifier power. A voltage propor
tional level. the power disables the output when the power exceeds a predetermined level.
© Copr. 1949-1998 Hewlett-Packard Co.
tinuous 10:1 vernier are supplied by the Model 19 15 A.
Duty cycles up to 90% can be obtained with internal
width operation. The Model 1915A also provides an ex
ternal width setting which converts the width circuit to a
pulse amplifier. The external width mode allows the
Model 1915A to accept a variable width pulse train, and
to shape and amplify the pulses but retain the initial
widths. In this mode 100% duty cycles are possible. Also
three of the four basic logic formats can be used with the
external width mode: return- to-zero, non-return-to-zero
and bi-phase. Only bi-polar logic cannot be used. The ex
ternal width mode is extremely useful in conjunction with
the HP Model 1925 A Word Generator (see page 8).
To fill the needs of bipolar transistor and 1C testing
and general purpose work, a low-power output plug-in
was designed. This Model 1917A Variable Transition
Time Output is a low-power, low-cost version of the
Model 1915A with many of the same features. Width
capabilities are identical. Transition times available are
7 ns to 500 fus in five ranges with a 50: 1 vernier. Five
amplitude settings span a range from 0.2 to 10 volts into
an external 50 ohm load. A maximum of 100 m A offset
(or 2.5 volts into an external 50 ohms) in both polarities
is provided.
A fast 350 ps, fixed rise and fall time is available in
the Model 1920A 350 ps Transition Time Output. It
provides an output amplitude of 0.5 to 5 volts in three
ranges, continuously adjustable, at repetition rates to
25 MHz. Pulse width is 0 to 10/is in four ranges, also
continuously adjustable. Outputs are available in either
positive or negative polarity with offsets up to 2 volts
into 50 ohms.
The Model 1925 A is a serial digital word generator
(see accompanying article) which provides variable word
length at clock rates from dc to 50 MHz. It has a pseudo
random bit sequence, and programming that is compat
ible with integrated circuits.
Fig. 7. How the overload protection circuit disables out
put current during excessive power dissipation in the
output amplifier. This output waveform is 50 volts into
50 ohms with a rise and fall time of 1 ms.
mechanical attenuator. Also, the set of range capacitors
are linearly charged and discharged in the rate, delay,
width and transition time range circuits, Fig. 8.
At the present time the two mainframes can be wired
with a program cable assembly used to connect the plugins to the rear panel. Four rear panel connectors, one
for each V* -module plug-in compartment, provide the
interface between the Model 1 900 system and an external
programming system. The required programming cir
cuitry is provided by plug-in boards or other boards
which can be installed at the factory or in the field.
Designed for Programming
Programming is an integral part of the pulse system
design. The circuits are designed so that the programming
response times are as short as possible. With few excep
tions, response times are between 3 and 30 /xs. Circuit
functions are designed with electronic controllability in
mind, that is, with the application of a voltage or a cur
rent, not by a mechanical method. For example, in the
Model 19 15 A, the output current sources are turned on
and off by applying a voltage to transistor bases. This
allows easy programming of the output current and
avoids wasteful dissipation (as much as 50 watts) in a
Fig. 8. The transition time range capacitor CT can be
energized mechanically or electronically with the cir
cuit shown here. When point A is grounded, 12 mA
passes into the base of Q1 causing it to saturate. Thus
CT is connected to the -25 V supply. Then capacitor CT
will charge and discharge in accordance with the tran
sition time switching circuit. During saturation, both the
forward and reverse beta characteristics of the tran
sistor 01 are important, since current will flow in both
directions through the transistor.
© Copr. 1949-1998 Hewlett-Packard Co.
Gordon Blanz (right) has a degree of BEE from the Univer
sity of Minnesota (1960) and a degree of MSEE from the
University of Colorado (1968). He joined Hewlett-Packard
in Palo Alto in 1960 as a development engineer and worked
on the design of the HP Model 140A Oscilloscope. He later
transferred to Colorado Springs in the high-frequency de
sign group and was responsible for the design of the
Model 1755A Dual Trace Amplifier. He is presently part
of the Model 1900 System design group.
Among his hobbies, Gordon plays tennis and climbs
mountains. He has climbed 37 of 53 Colorado peaks over
14,000 feet. Gordon is a member of Eta Kappa Nu.
Ron Knauber (left) attended the University of Nebraska
where he received his BSEE in 1961. He was project
leader on the initial design phase of the Model 1900 Pulse
System. Prior to joining Hewlett-Packard in 1965, Ron
worked on flight control systems.
Ron is an accomplished pianist, and like many of his
associates at HP, is a mountain climber and enjoys hunt
ing. He is a member of Eta Kappa Nu and Pi Mu Epsilon.
The range and the 'not-continuously-adjustable' func
tions are digitally programmed by a contact closure to
ground that also is a 10 mA current sink. An open circuit
deactivates a program line. A continuously adjustable
function requires an analog program current of 0 to
James M. Umphrey. Early project work was done by
Larry Nevin and Ronald J. Huppi. The product design
was done by Albert C. Knack, Spencer M. Ure and Nor
man L. O'Neal. Technical support was contributed by
Paul P Ficek, Charles T. Small and E. Yates Keiter. We
would especially like to acknowledge the valuable work
done on 1900 system multilayer printed circuit boards,
both by Paulette Metcalf and Sheila McCullough for the
printed circuit artwork and by Charles Canfield's proc
essing shop for providing many prototype and production
boards. Blair H. Harrison's encouragement and ideas are
also greatly appreciated. S
The major contributors to the Model 1900 System
circuit design were Gordon K. Blanz, Dee Broadhead,
James D. Dolan, Edward S. Donn, Ronald L. Knauber.
Stanley R. Lang, Robert L. Morrell, Jeffrey H. Smith and
Generating Words for Digital Testing
By Eddie Donn
DIGITAL EQUIPMENT TESTING requires a wide variety of
special test signals. The most practical way to meet these
varied requirements is with the programmable plug-in
system used in the HP Model 1900 Pulse System. A key
element of this system, the HP Model 1925A Word Gen
erator, Fig. 1, is capable of generating a variable length,
serial digital word at clock rates up to 50 MHz, in several
operating modes. Front-panel switching on the plug-in
permits selection of: return-to-zero (RZ) or non-returnto-zero (NRZ) format. Fig. 2(a), complementary output.
Fig. 2(b), command or automatic word recycling, and
electronic programming. In addition a long pseudo-ran
dom sequence (32,767 bits) is provided for testing communcations channels and LSI memories. The internal
registers may be set or cleared from the front panel to
establish reference levels and sequences.
The Word Generator will interface with all other plugins in the 1900 system. It accepts clock signals from the
Rate Generator (HP Model 1905) or the Delay Genera
tor (HP Model 1908A), and it can supply compatible
© Copr. 1949-1998 Hewlett-Packard Co.
trigger pulses to the Output Stages (Models 1915 or
Most pulse generators accept only return-to-zero (RZ)
trigger inputs because they operate only upon the leading
edge of the trigger pulse. A special trigger input on the
1900 output stages, External Width, assures compati
bility with all data formats of the Word Generator. In
this mode the output stage is operated as a pulse amplifier
(the pulse width is determined by the incoming signal).
This special mode is essential when NRZ digital wave
forms or any form of the biphase formats is used. The
usual Internal Width mode on the Output Stage will ac
commodate RZ formats (the pulse width is determined
by the output stage).
Logic power supplies in the Model 1900A mainframe
are capable of driving two Word Generators; power
supplies in the Model 1901 mainframe are capable of
driving four Word Generators. Both positive and negative
voltages are available for powering either saturated or
emitter-coupled types of logic.
Fig. 1. Characters are generated by setting the
appropriate toggle switches on the front panel
of this HP Model 1925A Word Generator plugin. Complement, pseudo-random noise, and RZ
and NRZ formats can be selected.
of the word. The parallel data can be set by the front
panel switches or brought in from the electronic pro
gramming inputs. The word is then shifted out of the
register in synchronism with the clock input. The shift
register output is operated upon to produce the desired
format: non-return-to-zero or return-to-zero, normal or
complemented. A transmission line driver then delivers
the word to its destination.
WORD recycling is accomplished by a flip-flop which
keeps track of the WORD state. When the WORD state
is false the instrument loads the shift register with incom-
Emitter-coupled integrated circuit logic is used in the
Word Generator. Emitter-coupled, non-saturated circuits
are quite at home in the Model 1900 system, since that is
basically the technique used in most of the pulse gen
erator circuits.
The digital word is generated by first loading it in par
allel into an open-ended shift register at the end of each
word, Fig. 3. This is essential for rapid reprogramming
Fig. 2. nonreturn-to-zero traces (left) show a return-to-zero (top traes) and nonreturn-to-zero
format is trace) with END pulses on the bottom trace. At (right), the top trace is
a normal output signal while its complement is the middle trace. The WORD output
is forced to zero at the END signal (bottom trace).
© Copr. 1949-1998 Hewlett-Packard Co.
Fig. 3. For rapid programming,
the input word to the Model
1925A Word Generator is
loaded in parallel into the shift
register. The word is then
shifted out in sync with the
clock. Pseudo-random noise
sequences are generated by a
digital feedback technique.
^ f ~
ing data and presets a counter according to the desired
word length. In the AUTO recycle mode, this occurs for
one period of the clock. In the MANUAL recycle mode
the WORD state remains false until the receipt of a
START signal or a command from the MANUAL push
button, Fig. 4. The END output is the logical comple
ment of the WORD state. It goes true between words in
the AUTO recycle mode. This information may be used
for scope sync or to command a new word from an elec
tronic programmer.
of the desired word length — 0001 for 16, 1000 for 9,
etc. Unused data switches are simply set to zero.
Long Words
Words longer than 16 bits, Fig. 6, are generated by
either continually programming new sequences with the
parallel inputs or by stacking the word generators. Stack
ing is accomplished by connecting the END of the one
generator to the START of the next. The loop is com
pleted by connecting the END of the last generator to
the START of the first.
Variable Word Length
Pseudo Random Noise
Word length is determined by a variable modulus
counter. Words shorter than 16 are generated by reduc
ing the modulus of the counter which controls the word
state, Fig. 5. This is accomplished by four switches inside
the Model 1925 A which are set in the 17's complement
In the pseudo-random-noise (PRN) mode, the WORD
state is forced true, hence END is inoperative. In this
mode a digital feedback circuit is enabled such that the
input to the first register is equal to the Exclusive-Or of
Fig. source pro Synchronous gating of a word from an external source is possible by pro
viding Gen signal (top trace) to both the Rate Generator GATE input and the Word Gen
erator START input. The resulting word (middle trace) and the END pulses (bottom
trace) are shown. Fig. 5. (right) Word lengths from 2 to 16 bits may be constructed.
Shown from top to bottom are word lengths of 16, 11,7 and 3 bits.
© Copr. 1949-1998 Hewlett-Packard Co.
the contents of the 14th and 15th cells in the register.
This results in a maximal length linear shift register se
quence of 215-1 or 32,767 bits. Other sequences can be
provided. The sequence starts with the current contents
of the shift register. This will be the same as the data
input if the machine is in the MANUAL mode. The se
quence will continuously recycle.
The sequence has the following randomness proper
ties: ones and zeros occur equally often; after a run of
ones and zeros there is a 50 % chance the run will end with
the next bit; and it is not possible to predict an entire
sequence from any partial sequence. The runs of ones
and zeros are useful for investigating duty cycle prob
lems. In this sequence there are 4096 runs of length one,
2048 runs of length two, 1024 runs of length three, etc.,
1 run of 14 zeros, and 1 run of 15 ones.
Fig. 6. Long words are constructed by ganging word
generators together. The top trace shows a 48-bit word.
Below are END pulses from the first, second and third
word generators.
Logic is on three multilayer printed circuit boards
separated into logical functions. The shift register flipflops and their loading gates are located on one board
to minimize the transmission of high frequency signals
to other parts of the instrument. The other two circuit
boards contain the control circuitry for the various modes
of operation. Two additional multilayer PC boards con
tain interconnections and programming circuitry. Multi
layer PC boards allow better transmission of the high
speed ECL (emitter-coupled logic) signals and reduce
the need for elaborate wire harnesses.
The plug-in boards permit fast trouble-shooting with
high packaging density. The problem of rigidity is solved
by anchoring the boards with a sheet metal top cover.
This also provides good airflow and increases RFI
Two methods of programming may be used. The fast
est is to reprogram at the start of each word. This allows
the maximum possible time for the programming lines to
settle to their new values. Alternately, the programmed
data may be strobed into the word generator during
END. The data gates to the internal memory are open
during END and the memory will latch on the strobed
data bits. The time constant of the interface network is
about 0. 1 ,ns, so END must have a duration of at least
0.2 /is. The need for buffer storage of the parallel pro
gramming data is eliminated.
CLOCK and START inputs are terminated in 50
ohms and designed to receive 1 volt or 20 mA signals.
The width of the CLOCK signal is important; the word
generator is designed to receive the output of the 1905 A
or 1906 A Rate Generators. The rate generators, how
ever, are designed to receive external signals with arbi
trary waveforms. The leading edge of the START input
is differentiated internally, so it may be of any width
between 10 ns and the period of the word cycle.
WORD and END outputs are from current source line
drivers. The complementary MECL outputs are used to
drive two pairs of emitter-coupled differential amplifiers.
The result is a fast, clean signal suitable for triggering.
The propagation delay between the incoming CLOCK
and the WORD and END outputs is less than 20 ns.
Since the outputs of the Word Generator are intended
to drive 50 ohm trigger inputs of the 1900 system, they
Programming is accomplished by an interface network
which transforms contact-closure or TTL type inputs (0
to +4 V) to MECL levels (-0.7 to -1.5 V). The front
panel switches are disabled by gates during programmed
operation. When they are enabled, the front panel
switches override any information on the programming
When fast programming is desired, the electronic in
puts should be provided via transmission lines (twisted
pairs are most economical). The programming source
should be matched to the impedance of the transmission
line by a series resistor if it is a voltage source (approxi
mately 100 ohms for twisted PVC wire).
© Copr. 1949-1998 Hewlett-Packard Co.
boards via coaxial transmission lines.
Noise between plug-ins is reduced by having separate
power supply regulators in each digital plug-in. This also
provides better voltage regulation and very fast current
limiting in case of a malfunction.
Eddie Dor, n
Ed Donn graduated from the
University of Florida with a
BSEE in 1962. After graduating
he worked on missile guidance,
anti-submarine warfare,
satellite communications, and
ground support systems. Since
joining HP in 1966 he has
worked on the pulse shaping
circuits in the 1915 Variable
Rise and Fall Output Stage and
product planning for the
1900 series digital plug-ins.
Several features of the word generator may be illus
trated by a typical application which requires a returnto-zero format with 11 bits of data of width t,, and a
sync bit of width t,., Fig. 7(a). The equipment needed
includes a rate generator and two word generators. One
of the following combinations of output stages may be
used: A 1901 A low power mainframe with two 1920A's
for fast risetime applications, or a 1 900A with two 1 9 1 5 A
plug-ins modified for two positive, two negative, or posi
tive and negative outputs. The rate generator provides
the clock for both word generators. Word generator
#1 provides timing for the data (which could be pro
grammed), and word generator #2 provides timing for
the sync. Similarly, output stage #1 provides the data
outputs of width t, and output stage #2 provides the sync
output of width t,. The two output stages are summed
together for the desired output. Both word generators
are set for a word length of 12 bits (1 1 data bits plus 1
sync bit).
Using a similar configuration it is possible to generate
bipolar outputs, Fig. 7(b). In that case output #1 would
generate the positive pulses and output #2 would gen
erate the negative pulses of the bipolar pulse train. Using
Model 1915A or the Model 1917A plug-ins, the pulse
widths, rise and fall times, amplitude, current offset
and polarity can be independently controlled and pro
Ed enjoys motorcycling,
mountain and rock climbing.
He has climbed 19 of
Colorado's 14,000-foot peaks
and led several local roped
are unterminated. To drive unterminated or ac termi
nated loads (such as some scope trigger inputs), END and
WORD may be terminated at the plug-in or at the load,
whichever is appropriate.
Similar to other 1900 System plug-ins, the electronic
inputs and outputs of the Word Generator may be
switched to the front panel or to the mainframe con
nector. Complicated configurations are thus patched to
gether inside the mainframe rather than by cables on the
front panel. Any one or all of the inputs and outputs may
be connected in this manner.
Noise induced problems are eliminated by adequate
grounding and power supply filters. Ground loops are
eliminated with balun transformers. For accurate phase
control, all clock signals are transmitted to the logic
Fig. generate (left) word generators and two output stages are used to generate this word (left)
consisting of one wide sync bit and 11 narrow data bits. Bipolar words (right) can be
generated with the same combination.
© Copr. 1949-1998 Hewlett-Packard Co.
Much of the convenience and economy of the Word
Generator — both to the user and in production — is
the result of innovations contributed by Al Knack (prod
uct design) and Chuck Small (electronic technician).
Rodger Earley contributed many ideas which helped
speed the transfer from breadboard to production. 5
1. Solomon W. Golomb, et al. 'Digital Communications
with Space Applications^ Prentice-Hall, Englewood Cliffs,
N.J. 1964.
2. Solomon W. Golomb, 'Shift Register Sequences^ HoldenDay, San Francisco 1967.
3. George C. Anderson, Brian W. Finnic and Gordon T.
Roberts, 'Pseudo-Random and Random Test Signals] Hew
lett-Packard Journal, Sept. 1967.
drive output delay mode; approx 14 ns in drive output ad
vance mode
HP Model 1900
PRICE: HP Model 1908A, S200.00.
Pulse System
The HP Model 1900 Pulse System is all solid-slate with plug-in
capability and can be assembled in a variety of combinations
All major functions can be programmed with an added option.
Model 1900A Mainframe
Mainframe accepts any 1900 series quarter-sue or halt-sue
plug-ins. Plug-ins may be interchanged in any manner within
Either external (with BNC cables) or internally selectable with
switches in the plug-ins
DIMENSIONS: 16* in. wide. 5V« in. high. 21 H In. deep overall
(425 x 133 x 543 mm).
POWER: 115 or 230 volts ±10% 50 to 60 Hz. 300 watts max
PRICE: HP Model 1900A, $790.00.
Model 1910A Delay Generator
RANGE: 5 ns to 100 ns. 5 ns steps
INPUT IMPEDANCE: 50 II, de-coupled.
SENSITIVITY: >+1 volt peak.
CONNECTION: Rate input may be connected internally or ex
ternally from other plug-ins, selected by internal switch.
AMPLITUDE: > + 1 volt into 25 ohms (drives two 1900 series
PRICE: HP Model 1910A, $150.00.
Model 1901A Mainframe
Mainframe accepts low power 1900 series quarter-size or halfsize plug-Ins. Plug-ins may be interchanged in any manner
switches in the plug-ins
:able with
DIMENSIONS: 16V< in wide. 5'< in. high. 21 H in. deep overall
(425 x 133 x 543 mm).
POWER: 115 or 230 volts ±10%. 50 to 60 Hz. 250 watts max
PRICE: HP Model 1901A, $490.00.
Model 1905A Rate Generator
REPETITION RATE: 25 Hz to 25 MHz in 6 decade ranges. 10:f
vernier allows continuous adjustment on any range.
PERIOD JITTER: <0.1% of selected period.
INPUT IMPEDANCE: 50 Q. de-coupled.
SENSITIVITY: 0.5 volts peak-to-peak.
LEVEL: Continuously variable over ±3 volt range.
SLOPE: + or -.selectable
DELAY: Appro». 10 ns between trigger input and rate output
SENSITIVITY: -2 volts or more required to gate pulse train on.
INPUT IMPEDANCE: 50 '..' de-coupled.
DELAY: Approx. 27 ns between gate input and first rate output
MANUAL OPERATION: Pushbutton for single pulse.
AMPLITUDE: >1 volt into 25 0 (drives two 1900 series plugins).
R1SETIME: - 5 ns
WIDTH: <10 n».
PRICE: HP Model 1905A 1200.00.
Model 1908A Delay Generator
DELAY: Drive output delayed with respect to trigger output.
ADVANCE: Trigger output delayed with respect to drive output
DOUBLE PULSE Generated from drive output connector. Spac
ing determined by time interval setting.
RANGE: 15 ns to 10 ms in 8 ranges. 10:1 vernier allows con
tinuous adjustment on any range
JITTER: <0 1% of selected time interval
EXCESSIVE DELAY INDICATOR: Light comes on when se
lected time interval exceeds pulse period.
INPUT IMPEDANCE: 50 0. de-coupled
SENSITIVITY: > +1 volt peck.
WIDTH; Portion of input trigger above 0.8 volts must be <7ns.
AMPLITUDE: > + 1 volt into 25 Q (drives two 1900 series
WIDTH: <10 ns.
RISETIME: <5 ns.
in approx 14 ns in drive output delay mode: appro» 29 ns in
Model 1915A Variable Transition Time Output
50 '.: or high Z; self contained 50 '..' termination
may be connected or disconnected
HIGH 2 OUTPUT: Approx 5 k ohms shunted by 45 pF.
50 '..' OUTPUT: Approx 50 '.! shunted by 45 pF.
50 milliamperes to 1 ampere in 4 ranges: 2.5:1 vernier allows
continuous adjustment on any range. Voltage Into external
50 0 is ±2.5 V to ±50 V with high Z source; ±1.25 V to
-25 V with 50 U source. Maximum amplitude (including
offset) is ±50 V.
WITH 50 12 SOURCE AND 50 Ià LOAD: ±5% for transition
times 7 ns to 10 ns; ±2% for transition times >10 ns.
WITH HIGH Z SOURCE AND 50 II LOAD: ±5% for all transi
tion times.
POLARITY: + or - , selectable.
DUTY CYCLE: 0 to >90% internal width mode; 0 to 100% ex
ternal width mode.
BASELINE OFFSET: -60 milliamperes. Maximum offset into
external 50 O is ±1.5 volts with 50 tÃ- source; ±3 volts with
high Z source.
TRANSITION TIMES: 7 ns (10 ns with high Z source) to 1 ms
in 11 ranges (1,2,5 sequence); two 100:1 verniers allow inde
pendent control of rise and fall times.
RANGES: 10 ns to 40 ms in 7 decade ranges (except for first
range which is 10 to 40 ns): 10:1 vernier allows continuous
adjustment on any range.
WIDTH JITTER: <0-5% of selected pulse width
Provides pulse amplifier operation; output pulse width de
termined by width of drive input.
INPUT IMPEDANCE: 50 0. de-coupled.
SENSITIVITY: > + 1 volt pe»k.
PRICE. HP Model 1915A $1600.00.
Model 1917A Variable Transition Time Output
SOURCE IMPEDANCE: Approx 50 ohms shunted by 45 pF am
plitude (volts into 50 ohms) 02 to 10 volts; 2.5:1 vernier
allows continuous adjustment on any range
PULSE TOP VARIATIONS: ±5% for transition times >7 ns.
POLARITY: + or - , selectable.
DUTY CYCLE; Oto >90% internal width mode
0 to 100% external width mode
BASELINE OFFSET: ±25 volts into external 50 ohms
TRANSITION TIMES: 7 ns to 500 PS in 5 ranges; two 50:1
verniers allow independent control of rise and fall times
RANGES: 10 ns to 40 ms in 7 decade ranges (except for
first range which is 10 to 40 ns); 10:1 vernier allows con
tinuous adjustment on any range.
Provides pulse amplifier operation, output pulse width
determined by width of drive input.
© Copr. 1949-1998 Hewlett-Packard Co.
INPUT IMPEDANCE: 50 ohms, dc coupled.
SENSITIVITY: > + 1 volt peak.
PRICE: HP Model 1917A $525.00.
Model 1920A 350 ps Transition Time Output
SOURCE IMPEDANCE: Approx 50 ohms.
AMPLITUDE: 0.5 to 5 volts in three ranges; 25:1 vernier allows
continuous adjustment on any range.
TRANSITION TIME: Rise and fall times -350 ps at max am
PULSE TOP VARIATIONS: <8% at max amplitude.
POLARITY: + or - . selectable.
BASELINE OFFSET: Max offset into external 50 ohms is -2
WIDTH: 0 to 10 ns in four ranges. 10:1 vernier allows contin
uous adjustment on any range.
INPUT IMPEDANCE: 50 ohms, de-coupled.
SENSITIVITY: >-M volt peak.
PRICE: HP Model 1920A, $1750.00.
Model 1925A Word Generator
WORD LENGTH: 2 to 16 bits, set by internal switches, not pro
WORD CONTENT: Set by front panel switches or rear panel
programming. Loaded into shift register between each word
cycle during END.
NRZ/RZ. selectable from front panel or programmed. RZ
pulse width less than clock period/2. WORD or WORD select
able from front panel switch.
WORD CYCLING: Automatic (continuous with one clock period
delay between words), external start command, or manual
MANUAL/AUTO: Selectable from front panel switch or pro
grammed. In AUTO mode, word continuously recycles with
one clock period delay between words. In program mode.
contenÃ- of each word corresponds to the previous parallel
word input that existed >200 ns before and during END. In
manual mode, a word Starts after receiving an external start
signal or pressing MANUAL push button and stops after 16
clock pulses.
END OUT: Available from front panel BNC corresponding to
SET: Serially loads ones into shift register. Output word bits
are all ones after 16 clock pulses
CLEAR: Resets shift register in parallel. Output word bits are
all zero.
PSEUDO-RANDOM NOISE: Provides a linear shift register se
quence of 32.767 bits. The sequence starts with the last 16
bit word in the shift register.
PROGRAMMING; All data bits. NRZ/RZ. PRN/WORD. and
CLOCK INPUT; (1905A or 1906A)
AMPLITUDE: >0.9 volts. <4 volts
WIDTH: >4 ns. <18 ns at +06 volts
INPUT IMPEDANCE: 50 ohms de-coupled
PERIOD; Word length +15 ns
AMPLITUDE: >0.9 volts. <4 volts
WIDTH: >5 ns
INPUT IMPEDANCE: 50 ohms de-coupled
TRUE; Contact closure, saturated DTL, or voltage source
(TiL) <+0.2V.
FALSE: Open, off DTL or voltage source (T"L) >25 V.
<4.0 V.
TRUE: 40 ±10 ma current source or 20 ±0.5 V into 50
FALSE; < 1 ma
RISE AND FALL TIMES: <4 ns (10% to 90%).
SOURCE IMPEDANCE: Unterminated current source.
PRICE: HP Model 1925A. $(50.00.
1900 Garden of the Gods Road
Colorado Springs, Colorado 80907
Frequency-Domain Oscilloscope Now
Measures to 1250 MHz
With analyzer new RF plug-in, HP's absolutely calibrated RF spectrum analyzer can display
any part of the frequency range from 500 kHz to 1250 MHz— or the whole range at once.
By Siegfried Linkwitz
TION, and other electronic equipment operating in the
frequency range from 500 kHz to 1250 MHz can now
make absolutely calibrated frequency-domain measure
ments with a spectrum analyzer as easily as they have
always made lower-frequency time-domain measure
ments with an oscilloscope. The instrument that makes
this possible consists of a Display Section (Model 140S,
14 IS, or 143S), a plug-in IF section (Model 8552A),
and a plug-in RF section, the new 0.5 to 1250 MHz
Model 8554L (Fig. 1). Al
though it's a spectrum an
alyzer, this instrument has
many of the qualities that
have made the oscilloscope
such a universal instrument.
It has absolute calibration
on both horizontal and ver
tical axes, it's easy to oper
ate, it gives unambiguous
spurious-free displays, and it
has high stability and sensi
tivity. Because it has these
qualities, it is beginning to
acquire the name frequencydomain oscilloscope. *
The new Model 8554L is the second spectrum an
alyzer RF Section to be designed for the same display
sections. The display sections, the IF plug-in, and the
first RF plug-in, a 1 kHz to 1 10 MHz instrument, were
described in these pages in August 1968.|1¡'-i Also de
scribed at that time were many of the frequency-domain
measurements that the new spectrum analyzer can make.
They include such traditional spectrum-analyzer meas
urements as spectrum surveillance and EMI testing,
measuring pulse spectra, and checking multichannel com
munications systems. More important, however, are the
measurements the new analyzer can make in general RF
circuit design. For example, it can measure the flatness,
harmonic content, and spectral purity of oscillators; it
can measure AM and FM modulation indexes; it can
measure gain, frequency response, harmonic and inter-
Fig. 1. A new plug-in RF section makes this a 0.5-to-1 250-MHz
spectrum analyzer— or frequency-domain oscilloscope, if you
prefer. It has absolute amplitude calibration, automatic phase
lock, and simple controls. Its frequency response is flat within
±1 dB from 1 to 1000 MHz. It has a 60 dB spurious free dis
play range, —120 dBm maximum sensitivity, and scan widths
from 20 kHz to 1250 MHz. Variable-persistence and largescreen displays are optional.
* See page 16 for more about frequency-domain oscilloscopes.
© Copr. 1949-1998 Hewlett-Packard Co.
2050 MHz IF
Fig. IF spurious quadruple conversion process, plus low-pass input and IF filters, keep spurious
responses off the display. For wide scan widths the YIG-tuned first LO is swept. For
narrow stable widths the third LO is swept and the first LO is phase-locked to a stable
reference to reduce residual FM to /ess than 300 Hz.
modulation distortion, and parasitic oscillations in am
plifiers; it can measure mixer conversion loss and localoscillator suppression in balanced mixers. The new RF
section extends all these measurements to 1 250 MHz. A
few examples of its use are described on page 18.
Analytical Capabilities
One of the new spectrum analyzer's most powerful
capabilities is its scanning versatility. It can display the
full spectrum from 0 to 1250 MHz, or any part of it.
When set for the 1 250-MHz scan, a marker pip appears
on the display at the frequency to which the analyzer is
tuned. The pip points downwards to distinguish it from a
signal. When the analyzer is switched to any of its ten
narrower scan widths, the scan is symmetrical about the
marker frequency. The narrowest scan width is 20 kHz,
and the widest symmetrical scan width is 1000 MHz.
Alternatively, scanning can be halted and the analyzer
can be operated as a sensitive manually tuned receiver
with variable bandwidth.
The analyzer's amplitude scale is absolutely calibrated
in dBm and ,uV Hence the analyzer doesn't merely dis
play a spectrum; it accurately measures the components
of the spectrum. Input signals can be as small as — 120
dBm (0.3 ,uV) or as large as +10 dBm (0.8 V). Fre
quency response is flat within ± 1 dB from 1 MHz to
1000 MHz, and most amplitude measurements can be
made to within ± 1 .5 dB or better.
Residual responses and harmonic mixing responses
are kept off the display by microcircuit filters. This makes
it easy to identify signals and to read their frequencies
directly on the analyzer's scales. The distortion-free and
spurious-free display range is greater than 60 dB.
The analyzer has seven calibrated bandwidths ranging
from 300 Hz to 300 kHz. The narrowest bandwidths give
high resolution for analyzing signals close together in fre
quency, and the wider bandwidths allow fast scanning of
wide frequency ranges. If the operator selects a band
width too narrow for the scan rate, a red light warns him
that the display is uncalibrated.
To provide the stability needed to make its narrow
bandwidths and narrow scan widths useful the analyzer
has a phase-lock system that automatically stabilizes the
first local oscillator when a narrow scan width is selected.
The phase-lock system reduces residual FM to less than
300 Hz peak-to-peak. The operator doesn't have to ma
nipulate any controls to achieve phase lock.
Advanced technology was needed to get many of the
new RF plug-in's capabilities into such a small package.
Most noteworthy are its YIG-tuned solid-state first local
© Copr. 1949-1998 Hewlett-Packard Co.
oscillator, its microcircuit filters, and the microcircuit
sampler in its phase-lock loop.
The block diagram, Fig. 2, illustrates the operation in
detail. The low-pass filter at the analyzer input has a fre
quency response which is flat to 1 300 MHz, down 3 dB
at 1500 MHz, and down 70 dB at 2050 MHz. Following
the filter, the input mixer and the YIG-tuned solid-state
first local oscillator convert the incoming signal to 2050
MHz, the first intermediate frequency. Even signals as
low as 500 kHz are transformed to this high frequency.
The high first IF and the low-pass input filter eliminate
image responses which might otherwise occur for input
frequencies from 4000 to 5300 MHz. This is a major
What's Inside
The new spectrum analyzer is essentially an electron
ically tuned superheterodyne receiver. The amplitude of
the received signal is displayed on the CRT as vertical
deflection. The frequency of the receiver is changed in
synchronism with the horizontal movement of the elec
tron beam across the CRT. The result is a display of
amplitude vs frequency, that is, a display of the spectrum.
The Meaning of 'Frequency-Domain Oscilloscope'
Excerpts from an informal talk by Roderick Carlson of Hewlett-Packard's Microwave Division
They complement each other. Each has its own place: the
oscilloscope is the instrument for working in the time do
main, and the spectrum analyzer is the instrument for work
ing in the frequency domain. When we call a spectrum an
alyzer a frequency-domain oscilloscope we mean that the
spectrum analyzer is exactly analogous to the oscilloscope,
but is for the frequency domain, and that the spectrum
analyzer has the same general usefulness as the oscilloscope!'
" 'Frequency-domain oscilloscope' is a term thai we at HewlettPackard have adopted to describe a certain type of spectrum
analyzer. As a name, if s not important; in fact, 'spectrum
analyzer' is a much more accurate name than 'oscilloscope'.
What is important is the concept behind the name — a new
way of thinking about spectrum analyzers and a new way
of using them.
"To deserve the name 'frequency-domain oscilloscope', a
spectrum analyzer has to be fully calibrated, like an oscil
loscope, and just as easy to use as an oscilloscope. There
are now two spectrum analyzers that have evolved this far;
they are the 110 MHz instrument described in the August
1968 HP Journal (Model 8552A/8553L), and the new 0.5
to 1250 MHz unit described in the accompanying article
(Model 8554L/8552A).
"These fully calibrated, easy-to-use instruments have an
area ap application that is, quite literally, immense. This ap
plication area is general circuit design — the same area in
which the oscilloscope finds most of its applications. The
spectrum analyzer is a basic measuring tool for designing
oscillators, amplifiers, mixers, modulators, filters and so on.
Like a scope or a dc voltmeter, if s a general-purpose, con
stant-use tool — not a special purpose instrument, but one
that has a broad range of uses'.'
Characteristics of F DO 's
"Our new spectrum analyzers are designed to be well suited
to general circuit-design work, and so are true frequencydomain oscilloscopes. They have several characteristics that
other spectrum analyzers don't have. One is absolute ampli
tude calibration, which allows you to measure signal levels
accurately with the spectrum analyzer. Some day all spectrum
analyzers will have absolute amplitude calibration — users are
going to demand it. Imagine how far you would get with an
oscilloscope these days if it didn't have vertical calibration.
"Another characteristic of these frequency-domain oscil
loscopes is an easy-to-interpret, unambiguous display. This
comes from an input filter which allows the analyzer to have
only a single response, thereby avoiding the confusion of
images and multiple responses due to harmonic mixing.
"A third characteristic of the new analyzers is a phase-lock
system that operates almost automatically, so the operator
doesn't even realize that the local oscillator has been stabil
ized by phase-locking. This is something that in the past has re
quired some skill and a bit of hope on the part of the operator.
"Finally, the new spectrum analyzers are much smaller in
size and lower in cost than are many older analyzers'.'
Usefulness Found by Experience
"We recognized the broad usefulness of the spectrum an
alyzer through experience in our own laboratories with our
microwave spectrum analyzer (Model 851B/855IB). In our
laboratories we do much the same kind of circuit design that
everyone else does. Our microwave spectrum analyzer was
conceived with the classical spectrum-analyzer applications
in mind. These are such things as looking at the spectra of
radar pulses or looking at the signals in microwave carrier
systems. This analyzer was also designed for some new ap
plications, such as spectrum surveillance and radio-frequencyinterference measurement. However, as soon as the analyzer
was put to use in our laboratories, it became apparent that
it was an excellent general-purpose tool for observing the
ever\day signals we were working with. In fact, it was better
than an oscilloscope for most of our purposes. Now we have
several of these analyzers and they are in constant use. We
wonder how we ever got along without them. To lose them
would be like losing one of our senses, like going deaf or
going blind.
"Now, oscilloscopes and spectrum analyzers aren't rivals.
Frequency-Domain Measurements
"Very few of the frequency-domain measurements that can
be made with the spectrum analyzer are new. Nearly all are
very familiar. They just haven't been made with spectrum
analyzers in the past. What engineers need to realize now is
that the spectrum analyzer has evolved to a point where it is
the most convenient and accurate instrument available for
making these measurements. In many cases, with a fully cal
ibrated spectrum analyzer on his bench a design engineer
doesn't need an RF voltmeter, or a power meter, or a wave
analyzer, or a distortion meter, or a swept-frequency indi
cator. What's more, he can make much more eye-opening
measurements with his spectrum analyzer than he can with a
collection of these other instruments"
© Copr. 1949-1998 Hewlett-Packard Co.
factor in keeping the display free of spurious signals.
To get narrow analyzing bandwidths, it is necessary to
convert to a lower intermediate frequency. This is done
in 3 steps, to avoid undesired image responses. A 1500
MHz fixed-frequency transistor oscillator and a second
mixer convert the 2050 MHz first IF signal to a 550 MHz
signal, the second IF. The signal has so far undergone two
conversions and is now amplified for the first time in a
550 MHz amplifier.
After further mixing with a 500 MHz third-localoscillator signal, followed by amplification of the result
ing 50 MHz IF signal and mixing with a 47 MHz fourthlocal-oscillator signal, the final 3 MHz IF is obtained.
It is at this last frequency that the bandwidth is narrowed
to seven calibrated bandwidths between 300 kHz and 300
Hz, one of which is selected by the user. Finally, the 3
MHz signal is amplified by a logarithmic amplifier which
has a dynamic range of 70 dB, and the amplifier output
is detected to produce the video signal which is applied to
the vertical deflection circuits of the CRT.
The horizontal deflection of the electron beam of the
CRT is controlled by a sawtooth generator. The same
sawtooth generator causes the analyzer's frequency to
scan, centered on the value set by the coarse frequency
tuning control and indicated on the slide-rule dial on the
front panel. For scan widths of 1250 MHz to 5 MHz, the
first local oscillator is swept. For narrower scan widths,
the sawtooth voltage is removed from the first LO and
applied to the third LO. The bandwidth of the IF stages
preceding the third mixer is sufficiently wide to allow for
the maximum tuning range of the third LO. When the
third LO is being swept, the first LO is automatically
phase-lock stabilized to a constant reference frequency,
and acts strictly as an up-converter. Narrow-band fre
quency tuning and scanning are accomplished with the
third LO. Stepping the scanning operation between the
first LO and the third LO gives in one instrument the ad
vantages of both very wide scans and very stable narrow
band scans.
YIG-Tuned Solid-State
First LO
First and Second
Converter Section
Microcircuit Low-Pass Filter
(3 dB Cutoff Frequency:
1500 MHz)
Fig. 3. Four hybrid microcircuits are used in the new RF
plug-in. One is the YIG-tuned transistor first local oscil
lator, two are low-pass filters (one is hidden under the
first and second converter section), and the fourth is a
sampler in the phase-lock system (see Fig. 5).
YIG-Tuned Solid-State First LO
The YIG-tuned solid-state first local oscillator made
it possible to get the new RF plug-in into the required
small package. This oscillator is not only smaller, but also
far superior in frequency stability and tuning linearity
to previously used backward-wave-tube oscillators. The
tuning element is a highly polished 0.035 inch diameter
sphere of Yttrium-Iron-Garnet, a ferrite material. Its
electrical equivalent is a parallel tuned circuit of low loss.
The resonance frequency of the YIG is a linear function
Fig. 4. When the analyzer is switched to stabilized oper
ation and the phase-lock system is unlocked, the posi
tive feedback loop oscillates at a rate of about 5 Hz,
tuning the YIG oscillator until its frequency is equal to
a harmonic of 1 MHz. Then, with the sampler acting as
a phase detector, the negative feedback loop takes over
to keep the YIG oscillator phase-locked to the reference
oscillator. The offset voltage to the third LO keeps the
display from shifting when phase-lock occurs.
© Copr. 1949-1998 Hewlett-Packard Co.
of the strength of the magnetic field applied to it.
The YIG sphere is mounted between the pole pieces
of an electromagnet and the magnet current controls the
first LO frequency, to the degree with which the magnetic
field follows this current. By careful selection of the mag
net material, excellent tuning linearity and small hyster
esis were obtained over the range from 2000 to 3300
MHz. The oscillator's center frequency is always within
10 MHz of the front-panel dial setting, and the frequency
error between any two points on the display is less than
10% of the indicated separation.
A loop around the YIG sphere forms the RF coupling
to a single-transistor oscillator. The oscillator is followed
by two stages of power amplification, which act as a
buffer against variations in the load on the oscillator,
which could otherwise pull the frequency.
The magnet/YIG combination has high tuning sensi
tivity (20 kHz//xA), so an extremely low-noise-current
power supply had to be designed for the magnet. Careful
magnetic shielding against extraneous fields was also re
quired to maintain spectral purity, since the frequency
depends on magnetic field strength. In the final design,
residual FM is less than 10 kHz peak to peak without
phase-lock stabilization.
Beyond Traditional Spectrum Analyzer Uses
Absolute calibration, ease of use, free
dom from spurious responses, and
automatic stabilization — characteristics
that qualify the new 1250 MHz Spec
trum Analyzer as a frequency-domain
Signal spectrum observed at Palo Alto, Califor
nia with a single-turn 16" x 24" loop antenna
Vertical scale (amplitude): Logarithmic, -20
dBm to -90 dBm, 10 dB/division, reference
-20 dBm.
oscilloscope — obviously give it sub
stantially greater analytical capabilities
than older spectrum analyzers in the
same applications. But a frequencydomain oscilloscope isn't limited to
such traditional areas as radar, com
munications, and EMI measurements.
It's a general-purpose design tool, use
ful for measurements on oscillators,
modulators, mixers, amplifiers, and fil
ters. Shown here are a few measure
ments made with the new analyzer.
Three main-frame Display Sections
will accept the 1250 MHz Model 8554L
RF Section. Model 140S is the basic
mainframe; it has a 5 inch internalgraticule CRT with a normal-persist
ence P11 phosphor. Model 141S has
the additional advantages of variable
persistence and storage; these features
are useful for measuring intermittent
signals, for comparing signals before
and after adjustments, and for making
high-resolution measurements at low
0 to 200 MHz Band at 20 MHz/division
Note FM band at center screen, TV Channels 2,
4, 5, 7, and 9 all clearly visible
0 to WOO MHz Band at 100 MHz/division
In addition to FM and VHF-TV channels, note
UHF Channel 36 at 600 MHz, and other signals.
Center frequency = 300 MHz
Horizontal scale (frequency) = 20 kHz/division,
thus FM deviation = 30 kHz.
sweep speeds where flicker might
otherwise be a problem. The third main
frame, Model 143S, has a large 8 by 10
inch display, useful in production areas
or classrooms.
Measuring Gain and Frequency Response
of an Amplifier
Horizontal scale (frequency): 0-1000 MHz, 100
Vertical scale (amplitude): Logarithmic, +10
dBm to -60 dBm, 10 dB/division, reference
+ 10 dBm.
Swept Source Directly Into Analyzer
© Copr. 1949-1998 Hewlett-Packard Co.
Amplifier Output
Gain = 20 dB, -3 dB point •
700 MHz
Thin-film microcircuit technology, computer-aided de
sign, and the ability to accurately characterize microwave
devices by scattering parameters were key factors in the
development of the YIG oscillator circuitry. The oscil
lator and amplifier are small enough that they can be
mounted together with the YIG sphere in the 0.10 inch
gap between the pole pieces of the driving magnet. The
entire oscillator, including magnets, is in a package just
2 inches high and 2.5 inches in diameter (Fig. 3).
. Sampled Signal
Microcircuit Filters
RF Input
The input low-pass filter, which helps establish the 60
dB spurious-response-free display range of the new RF
plug-in, and the low-pass filter in the first IF, which
rejects undesired mixing products, were designed using
thin-film circuit elements. These filters operate in the
microwave region and exhibit excellent attenuation char
acteristics in their stop bands, a result of the small phys
ical size of the filter elements in relation to the wavelength
at which they are used. Because of their size, individual
capacitors and inductors in the filter can be characterized
as discrete ralhci than distributed elements. A com
puterized synthesis procedure determines the actual di
mensions and the layout for the plating masks. Element
losses are taken into account so filters with sharp cutoffs
in their attenuation characteristics can be realized.
Both filters are thirteen-element, thirteen-pole Tschebyscheff low-pass filters. The input filter has a 3 dB cutoff
frequency of 1500 MHz and has more than 70 dB rejec
tion in its stop band from 2050 MHz through 1 2 GHz.
It is 2 inches long and 0.25 inch in diameter. The filter
in the first IF has a cutoff frequency of 5.0 GHz and has
70 dB rejection through 20 GHz. Its dimensions are 0.7
by 0.1 by 0.04 inch.
Fig. 5. This balanced, two-diode hybrid-microcircuit
sampler acts as a phase detector in the automatic
phase-lock system of the Model 8554L RF Section. The
reference signal, a 1 MHz square wave, drives the two
step-recovery-diode stages to produce voltage steps
with very fast risetimes. The balanced configuration of
shorted transmission lines then differentiates this wave
form and the resulting narrow pulses switch the two hot
carrier diodes on and off to sample the voltage from the
YIG oscillator.
Coupling Loops to 5.0 GHz
Microcircuit Low-Pass Filter (not shown)
Sampling Phase-Lock System
The resolution which can be obtained in any spectrum
analyzer is limited by the bandwidths and shape factors
of the IF filters and by the stability of the local oscillators
which are used to frequency-translate the input signal. In
practice the limitation is set by oscillator stability, which
in turn determines the narrowest usable filter bandwidths.
One way to improve the stability of an oscillator is to
phase-lock it to a stable reference frequency. This ap
proach has been widely used in spectrum analyzers, but
it has always required some effort from the operator in
setting up the instrument. In the new RF plug-in, phaselocking of the first local oscillator (YIG) occurs auto
matically for narrow scan widths and does not require
operator intervention.
Fig. 6. To get flat frequency response in the first mixer,
the two hot carrier mixer diodes (it's a balanced mixer)
are mounted so their packages and leads are part of the
mixer operation and don't contribute parasitic effects.
The two diodes are inductively coupled to the stripline
from the first LO and to the first IF filter cavity. A similar
technique is used in the second mixer.
The phase-lock system is shown in Fig. 4. An impor
tant part of the system is a balanced, two-diode microcircuit sampler which produces a voltage proportional to
the phase difference between the YIG oscillator output
signal and the reference oscillator output signal. Fig. 5
is a photograph of the sampler.
© Copr. 1949-1998 Hewlett-Packard Co.
HP Model 8554L/85S2A
Spectrum Analyzer
SCAN WIDTH: (on 10 division CRT horizontal axis)
Per Division: 15 calibrated scan widths from 100 MHz/div to
2 kHz/div in 1, 2, 5 sequence.
Preset: 0-1250 MHz
Zero: Analyzer is fixed-tuned receiver.
Center Frequency Accuracy: The dial indicates the display
center frequency within 10 MHz.
Scan Linearity: Frequency error between two points on the
display is less than 10% of the indicated separation.
IF Bandwidth: Bandwidths of 03 to 300 kHz provided in a f,
3 sequence
IF Bandwidth Accuracy: Individual bandwidth»' 3 dB points
calibrated to ±20% (10 kHz bandwidth ±5%).
IF Bandwidth Selectivity: 60 dB/3 dB bandwidth ratio <20:1
for IF bandwidths from 1 kHz to 300 kHz. 60 dB/3 dB
bandwidth ratio <25:1 for 300 Hz IF bandwidth.
Residual FM:
Stabilized: .'300 Hz peak-to-peak.
Unstabilized: <10 kHz peak-to-peak.
Noise Sidebands: More than 60 dB belo
CW signal. 20 kHz
or more away from signal, for 1 kHz b ndwidth.
Calibrator Output: Amplitude -30 dBm. ±0.3 dB
Frequency 30 MHz, ±0.3 MHz
LIN: From 0.1 nWdlv to 100 mV/div in a 1, 2 sequence on an
8 division display.
Average Noise Level: < - 102 dBm with 10 kHz IF bandwidth.
Spurious Responses: For -40 dBm signal level to the input
mixer*, image responses, out of band mixing responses,
harmonic and intermodulation distortion are all more than
60 dB below the input signal level.
Residual Responses: <- 100 dBm.
• Signal level to
level at input - input RF alten
A M P L I T U D E A C C U R A C Y : L O G
Frequency Response (Flatness):
( 1 M H ! t o 1 . 0 G H z ) Â ± 1 d B
(500 kHz to 1.25 GHz) ±2dB
Switching between Bandwidths: ±0.5 dB
(At 20°C)
First and Second Converters
In an absolutely calibrated spectrum analyzer, the fre
quency response of the input mixer is very important.
The responses of the other mixers, amplifiers, and filters
in the signal path have an effect only at fixed frequencies
or over very narrow bandwidths. The input mixer, how
ever, is broadband, so its response largely determines the
instrument's accuracy. The input mixer in the new RF
plug-in is a balanced mixer mounted inside the coaxial
cavity of the 2050 MHz first IF bandpass filter. Two
standard hot carrier diodes in glass packages mounted on
a printed circuit board are inductively coupled to the IF
cavity and to a balanced stripline which carries the firstlocal-oscillator signal (Fig. 6). Thus the diode packages
and their leads are used as part of the mixer operation
to avoid parasitic effects.
A similar technique is used for the second mixer,
which uses a single hot carrier diode. The diode is
mounted in the wall between the IF filter cavity and the
second LO cavity. It couples to the second LO cavity
with one lead and to the IF filter cavity with its other lead.
Input Impedance: 50 I! nominal. Reflection coefficient <0.30
(Return loss >105 dB).
Maximum Input Level: Peak or average power +13 dBm {1.4
Vac peak, ±50 Vdc).
Scan Time: 16 internal scan rates from 0.1 ms/div to 10 s/div
in a 1, 2. 5 sequence.
Scan Time Accuracy:
0.1 ms/div to 20 ms/div ± 10%
50 ms/div to 10 8/div ±20%
d e l
d e l
d e l 1 4
d e l 1 4
5 4 L
S e c t i o n
5 2 A
S e c t i o n
S D i s p l a y S e c t i o n
S D i s p l a y S e c t i o n
3 3 0 0 .
1 9 0 0 .
$ 7 2 5 .
S 1 5 2 5 .
1501 Page Mill Road
Palo Alto, California 94304
who designed the sampler and phase-lock system, Wil
liam Swift and Melvin D. Humpherys, who designed the
third converter, John E. Nidecker and Fred H. Meyers,
who did the product design, and many others, particularly
Roderick Carlson, who guided the project through sev
eral critical phases. S
[1]. Brian D. Unter, 'Fully Calibrated Frequency-Domain
Measurements; Hewlett-Packard Journal, August 1968.
[2]. Thomas L. Grisell, Irving H. Hawley, Jr., Brian D.
Unter, and Paul G. Winninghoff, 'Design of a Third-Gen
eration RF Spectrum Analyzer; Hewlett-Packard Journal,
August 1968.
Siegfried Linkwitz
Curiosity about the possible
course of human development,
which he shares with
many other contemporary
scientists and engineers, has
caused Siegfried Linkwitz
to give much of his leisure time
to social and psychological
studies. One aspect of
this activity has been to lead
encounter groups concerned
with human potential.
Merely to list all those who worked with me on the
design of the Model 8554L RF Section doesn't do justice
to their efforts and dedication. However, even a list of
those who contributed their talents at various stages of
the project would be quite long. It would include Harley
L. Malversan, who was project leader during the early
phases, Richard C. Keiier and James C. Harmon, who
designed the first and second converters, John J. Dupre,
who designed the YIG oscillator, Fendall G. Winston,
Amplitude Display: t 0.25 dB/da ±2.8% of full
but not more 8 dlv deflection
than ±1.5dB
over the full
70 dB display
Siegfried attended the
Technische Hochschule in Darmstadt, Germany, where he
received his Diplom Ingenieur in 1961. He joined
Hewlett-Packard in the same year and worked on the
design of several instruments and systems before
becoming project manager for the 8554L A member of
IEEE, Siegfried has taken some graduate courses
at Stanford and holds one patent.
HEWLETT-PACKARD JOURNAL © APRIL 1969 Volume 20 -Number 8
Editor/a/ Stall: F. J. BUmHARD, R. P. DOLÃA!, L. D. SHCRGALIS. P H. SNYDER Art Director R. A. ERICKSON
© Copr. 1949-1998 Hewlett-Packard Co.
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