teplotni_cidla_prirucka_vyvojare_analog_devices.pdf

teplotni_cidla_prirucka_vyvojare_analog_devices.pdf
TEMPERATURE SENSORS
SECTION 7
TEMPERATURE SENSORS
Walt Kester, James Bryant, Walt Jung
INTRODUCTION
Measurement of temperature is critical in modern electronic devices, especially
expensive laptop computers and other portable devices with densely packed circuits
which dissipate considerable power in the form of heat. Knowledge of system
temperature can also be used to control battery charging as well as prevent damage
to expensive microprocessors.
Compact high power portable equipment often has fan cooling to maintain junction
temperatures at proper levels. In order to conserve battery life, the fan should only
operate when necessary. Accurate control of the fan requires a knowledge of critical
temperatures from the appropriate temperature sensor.
APPLICATIONS OF TEMPERATURE SENSORS
n Monitoring
u Portable Equipment
u CPU Temperature
u Battery Temperature
u Ambient Temperature
n Compensation
u Oscillator Drift in Cellular Phones
u Thermocouple Cold-Junction Compensation
n
Control
u Battery Charging
u Process Control
Figure 7.1
Accurate temperature measurements are required in many other measurement
systems such as process control and instrumentation applications. In most cases,
because of low-level nonlinear outputs, the sensor output must be properly
conditioned and amplified before further processing can occur.
Except for IC sensors, all temperature sensors have nonlinear transfer functions. In
the past, complex analog conditioning circuits were designed to correct for the sensor
nonlinearity. These circuits often required manual calibration and precision
resistors to achieve the desired accuracy. Today, however, sensor outputs may be
7.1
TEMPERATURE SENSORS
digitized directly by high resolution ADCs. Linearization and calibration is then
performed digitally, thereby reducing cost and complexity.
Resistance Temperature Devices (RTDs) are accurate, but require excitation current
and are generally used in bridge circuits. Thermistors have the most sensitivity but
are the most non-linear. However, they are popular in portable applications such as
measurement of battery temperature and other critical temperatures in a system.
Modern semiconductor temperature sensors offer high accuracy and high linearity
over an operating range of about –55ºC to +150ºC. Internal amplifiers can scale the
output to convenient values, such as 10mV/ºC. They are also useful in cold-junctioncompensation circuits for wide temperature range thermocouples. Semiconductor
temperature sensors can be integrated into multi-function ICs which perform a
number of other hardware monitoring functions.
Figure 7.2 lists the most popular types of temperature transducers and their
characteristics.
TYPES OF TEMPERATURE SENSORS
THERMOCOUPLE
RTD
THERMISTOR
SEMICONDUCTOR
Widest Range:
Range:
Range:
Range:
–184ºC to +2300ºC
–200ºC to +850ºC
0ºC to +100ºC
–55ºC to +150ºC
High Accuracy and
Fair Linearity
Poor Linearity
Linearity: 1ºC
Repeatability
Accuracy: 1ºC
Needs Cold Junction
Requires
Requires
Compensation
Excitation
Excitation
Low-Voltage Output
Low Cost
High Sensitivity
Requires Excitation
10mV/K, 20mV/K,
or 1µA/K Typical
Output
Figure 7.2
THERMOCOUPLE PRINCIPLES AND COLD-JUNCTION
COMPENSATION
Thermocouples are small, rugged, relatively inexpensive, and operate over the
widest range of all temperature sensors. They are especially useful for making
measurements at extremely high temperatures (up to +2300°C) in hostile
environments. They produce only millivolts of output, however, and require
precision amplification for further processing. They also require cold-junctioncompensation (CJC) techniques which will be discussed shortly. They are more
linear than many other sensors, and their non-linearity has been well characterized.
Some common thermocouples are shown in Figure 7.3. The most common metals
used are Iron, Platinum, Rhodium, Rhenium, Tungsten, Copper, Alumel (composed
7.2
TEMPERATURE SENSORS
of Nickel and Aluminum), Chromel (composed of Nickel and Chromium) and
Constantan (composed of Copper and Nickel).
COMMON THERMOCOUPLES
JUNCTION MATERIALS
TYPICAL
NOMINAL
ANSI
DESIGNATION
USEFUL
SENSITIVITY
RANGE (ºC)
(µV/ºC)
38 to 1800
7.7
B
0 to 2300
16
C
Chromel - Constantan
0 to 982
76
E
Iron - Constantan
0 to 760
55
J
Chromel - Alumel
–184 to 1260
39
K
Platinum (13%)/Rhodium-
0 to 1593
11.7
R
0 to 1538
10.4
S
–184 to 400
45
T
Platinum (6%)/ RhodiumPlatinum (30%)/Rhodium
Tungsten (5%)/Rhenium Tungsten (26%)/Rhenium
Platinum
Platinum (10%)/RhodiumPlatinum
Copper-Constantan
Figure 7.3
Figure 7.4 shows the voltage-temperature curves of three commonly used
thermocouples, referred to a 0°C fixed-temperature reference junction. Of the
thermocouples shown, Type J thermocouples are the most sensitive, producing the
largest output voltage for a given temperature change. On the other hand, Type S
thermocouples are the least sensitive. These characteristics are very important to
consider when designing signal conditioning circuitry in that the thermocouples'
relatively low output signals require low-noise, low-drift, high-gain amplifiers.
To understand thermocouple behavior, it is necessary to consider the non-linearities
in their response to temperature differences. Figure 7.4 shows the relationships
between sensing junction temperature and voltage output for a number of
thermocouple types (in all cases, the reference cold junction is maintained at 0°C). It
is evident that the responses are not quite linear, but the nature of the non-linearity
is not so obvious.
Figure 7.5 shows how the Seebeck coefficient (the change of output voltage with
change of sensor junction temperature - i.e., the first derivative of output with
respect to temperature) varies with sensor junction temperature (we are still
considering the case where the reference junction is maintained at 0°C).
When selecting a thermocouple for making measurements over a particular range of
temperature, we should choose a thermocouple whose Seebeck coefficient varies as
little as possible over that range.
7.3
TEMPERATURE SENSORS
THERMOCOUPLE OUTPUT VOLTAGES FOR
TYPE J, K, AND S THERMOCOUPLES
THERMOCOUPLE OUTPUT VOLTAGE (mV)
60
50
TYPE K
40
TYPE J
30
20
TYPE S
10
0
-10
-250
0
250
500
750
1000
1250
1500
1750
TEMPERATURE (°C)
Figure 7.4
THERMOCOUPLE SEEBECK COEFFICIENT
VERSUS TEMPERATURE
70
SEEBECK COEFFICIENT - µV/ °C
60
TYPE J
50
TYPE K
40
30
20
TYPE S
10
0
-250
0
250
500
750
1000
TEMPERATURE (°C)
Figure 7.5
7.4
1250
1500
1750
TEMPERATURE SENSORS
For example, a Type J thermocouple has a Seebeck coefficient which varies by less
than 1µV/°C between 200 and 500°C, which makes it ideal for measurements in this
range.
Presenting these data on thermocouples serves two purposes: First, Figure 7.4
illustrates the range and sensitivity of the three thermocouple types so that the
system designer can, at a glance, determine that a Type S thermocouple has the
widest useful temperature range, but a Type J thermocouple is more sensitive.
Second, the Seebeck coefficients provide a quick guide to a thermocouple's linearity.
Using Figure 7.5, the system designer can choose a Type K thermocouple for its
linear Seebeck coefficient over the range of 400°C to 800°C or a Type S over the
range of 900°C to 1700°C. The behavior of a thermocouple's Seebeck coefficient is
important in applications where variations of temperature rather than absolute
magnitude are important. These data also indicate what performance is required of
the associated signal conditioning circuitry.
To use thermocouples successfully we must understand their basic principles.
Consider the diagrams in Figure 7.6.
THERMOCOUPLE BASICS
A. THERMOELECTRIC VOLTAGE
C. THERMOCOUPLE MEASUREMENT
Metal A
Metal A
V1 – V2
Metal A
V1
Thermoelectric
EMF
Metal B
T1
V1
T1
T2
Metal B
D. THERMOCOUPLE MEASUREMENT
B. THERMOCOUPLE
Copper
Metal A
R
Metal A
T1
V
Metal A
T3
T2
Copper
Metal A
I
V1
V2
V2 V1
Metal B
T4
T1
T2
V2
Metal B
R = Total Circuit Resistance
I = (V1 – V2) / R
V = V1 – V2, If T3 = T4
Figure 7.6
If we join two dissimilar metals at any temperature above absolute zero, there will
be a potential difference between them (their "thermoelectric e.m.f." or "contact
potential") which is a function of the temperature of the junction (Figure 7.6A). If we
join the two wires at two places, two junctions are formed (Figure 7.6B). If the two
junctions are at different temperatures, there will be a net e.m.f. in the circuit, and a
current will flow determined by the e.m.f. and the total resistance in the circuit
(Figure 7.6B). If we break one of the wires, the voltage across the break will be
7.5
TEMPERATURE SENSORS
equal to the net thermoelectric e.m.f. of the circuit, and if we measure this voltage,
we can use it to calculate the temperature difference between the two junctions
(Figure 7.6C). We must always remember that a thermocouple measures the
temperature difference between two junctions, not the absolute temperature at one
junction. We can only measure the temperature at the measuring junction if we
know the temperature of the other junction (often called the "reference" junction or
the "cold" junction).
But it is not so easy to measure the voltage generated by a thermocouple. Suppose
that we attach a voltmeter to the circuit in Figure 7.6C (Figure 7.6D). The wires
attached to the voltmeter will form further thermojunctions where they are
attached. If both these additional junctions are at the same temperature (it does not
matter what temperature), then the "Law of Intermediate Metals" states that they
will make no net contribution to the total e.m.f. of the system. If they are at
different temperatures, they will introduce errors. Since every pair of dissimilar
metals in contact generates a thermoelectric e.m.f. (including copper/solder,
kovar/copper [kovar is the alloy used for IC leadframes] and aluminum/kovar [at the
bond inside the IC]), it is obvious that in practical circuits the problem is even more
complex, and it is necessary to take extreme care to ensure that all the junction
pairs in the circuitry around a thermocouple, except the measurement and reference
junctions themselves, are at the same temperature.
Thermocouples generate a voltage, albeit a very small one, and do not require
excitation. As shown in Figure 7.6D, however, two junctions (T1, the measurement
junction and T2, the reference junction) are involved. If T2 = T1, then V2 = V1, and
the output voltage V = 0. Thermocouple output voltages are often defined with a
reference junction temperature of 0ºC (hence the term cold or ice point junction), so
the thermocouple provides an output voltage of 0V at 0ºC. To maintain system
accuracy, the reference junction must therefore be at a well-defined temperature
(but not necessarily 0ºC). A conceptually simple approach to this need is shown in
Figure 7.7. Although an ice/water bath is relatively easy to define, it is quite
inconvenient to maintain.
Today an ice-point reference, and its inconvenient ice/water bath, is generally
replaced by electronics. A temperature sensor of another sort (often a semiconductor
sensor, sometimes a thermistor) measures the temperature of the cold junction and
is used to inject a voltage into the thermocouple circuit which compensates for the
difference between the actual cold junction temperature and its ideal value (usually
0°C) as shown in Figure 7.8. Ideally, the compensation voltage should be an exact
match for the difference voltage required, which is why the diagram gives the
voltage as f(T2) (a function of T2) rather than KT2, where K is a simple constant. In
practice, since the cold junction is rarely more than a few tens of degrees from 0°C,
and generally varies by little more than ±10°C, a linear approximation (V=KT2) to
the more complex reality is sufficiently accurate and is what is often used. (The
expression for the output voltage of a thermocouple with its measuring junction at
T°C and its reference at 0°C is a polynomial of the form V = K1T + K2T2 + K3T3 +
..., but the values of the coefficients K2, K3, etc. are very small for most common
types of thermocouple. References 8 and 9 give the values of these coefficients for a
wide range of thermocouples.)
7.6
TEMPERATURE SENSORS
CLASSICAL COLD-JUNCTION COMPENSATION USING AN
ICE-POINT (0°C) REFERENCE JUNCTION
METAL A
METAL A
V1 – V(0°C)
T1
V1
METAL B
V(0°C)
ICE
BATH
T2
0°C
Figure 7.7
USING A TEMPERATURE SENSOR
FOR COLD-JUNCTION COMPENSATION
V(OUT)
V(COMP)
COPPER
METAL A
T1
TEMPERATURE
COMPENSATION
CIRCUIT
COPPER
SAME
TEMP
METAL A
V(T1)
TEMP
SENSOR
V(T2)
T2
METAL B
V(COMP) = f(T2)
V(OUT)
ISOTHERMAL BLOCK
= V(T1) – V(T2) + V(COMP)
IF V(COMP) = V(T2) – V(0°C), THEN
V(OUT)
= V(T1) – V(0°C)
Figure 7.8
7.7
TEMPERATURE SENSORS
When electronic cold-junction compensation is used, it is common practice to
eliminate the additional thermocouple wire and terminate the thermocouple leads in
the isothermal block in the arrangement shown in Figure 7.9. The Metal A-Copper
and the Metal B-Copper junctions, if at the same temperature, are equivalent to the
Metal A-Metal B thermocouple junction in Figure 7.8.
TERMINATING THERMOCOUPLE LEADS
DIRECTLY TO AN ISOTHERMAL BLOCK
COPPER
V(OUT) = V1 – V(0°C)
T2
METAL A
COPPER
T1
V1
METAL B
TEMPERATURE
COMPENSATION
CIRCUIT
TEMP
SENSOR
COPPER
T2
ISOTHERMAL BLOCK
Figure 7.9
The circuit in Figure 7.10 conditions the output of a Type K thermocouple, while
providing cold-junction compensation, for temperatures between 0ºC and 250ºC. The
circuit operates from single +3.3V to +12V supplies and has been designed to
produce an output voltage transfer characteristic of 10mV/ºC.
A Type K thermocouple exhibits a Seebeck coefficient of approximately 41µV/ºC;
therefore, at the cold junction, the TMP35 voltage output sensor with a temperature
coefficient of 10mV/ºC is used with R1 and R2 to introduce an opposing cold-junction
temperature coefficient of –41µV/ºC. This prevents the isothermal, cold-junction
connection between the circuit's printed circuit board traces and the thermocouple's
wires from introducing an error in the measured temperature. This compensation
works extremely well for circuit ambient temperatures in the range of 20ºC to 50ºC.
Over a 250ºC measurement temperature range, the thermocouple produces an
output voltage change of 10.151mV. Since the required circuit's output full-scale
voltage change is 2.5V, the gain of the circuit is set to 246.3. Choosing R4 equal to
4.99kΩ sets R5 equal to 1.22MΩ. Since the closest 1% value for R5 is 1.21MΩ, a
50kΩ potentiometer is used with R5 for fine trim of the full-scale output voltage.
Although the OP193 is a single-supply op amp, its output stage is not rail-to-rail,
and will only go down to about 0.1V above ground. For this reason, R3 is added to
the circuit to supply an output offset voltage of about 0.1V for a nominal supply
voltage of 5V. This offset (10°C) must be subtracted when making measurements
7.8
TEMPERATURE SENSORS
referenced to the OP193 output. R3 also provides an open thermocouple detection,
forcing the output voltage to greater than 3V should the thermocouple open.
Resistor R7 balances the DC input impedance of the OP193, and the 0.1µF film
capacitor reduces noise coupling into its non-inverting input.
USING A TEMPERATURE SENSOR FOR
COLD-JUNCTION COMPENSATION (TMP35)
3.3V TO 5.5V
0.1µF
TMP35
TYPE K
THERMO
COUPLE
R5*
1.21MΩ
Ω
R4*
4.99kΩ
Ω
R1*
24.9kΩ
Ω
P1
50kΩ
Ω
0 °C < T < 250 °C
–
R3*
1.24MΩ
Ω
CHROMEL
–
OP193
Cu
+
COLD
JUNCTION
+
R7*
4.99kΩ
Ω
Cu
R2*
102Ω
Ω
ALUMEL
VOUT
0.1 - 2.6V
10mV/°C
R6
100kΩ
Ω
0.1µF
FILM
* USE 1% RESISTORS
ISOTHERMAL
BLOCK
Figure 7.10
The AD594/AD595 is a complete instrumentation amplifier and thermocouple cold
junction compensator on a monolithic chip (see Figure 7.11). It combines an ice point
reference with a precalibrated amplifier to provide a high level (10mV/°C) output
directly from the thermocouple signal. Pin-strapping options allow it to be used as a
linear amplifier-compensator or as a switched output set-point controller using
either fixed or remote set-point control. It can be used to amplify its compensation
voltage directly, thereby becoming a stand-alone Celsius transducer with 10mV/°C
output. In such applications it is very important that the IC chip is at the same
temperature as the cold junction of the thermocouple, which is usually achieved by
keeping the two in close proximity and isolated from any heat sources.
The AD594/AD595 includes a thermocouple failure alarm that indicates if one or
both thermocouple leads open. The alarm output has a flexible format which
includes TTL drive capability. The device can be powered from a single-ended supply
(which may be as low as +5V), but by including a negative supply, temperatures
below 0°C can be measured. To minimize self-heating, an unloaded AD594/AD595
will operate with a supply current of 160µA, but is also capable of delivering ±5mA
to a load.
The AD594 is precalibrated by laser wafer trimming to match the characteristics of
type J (iron/constantan) thermocouples, and the AD595 is laser trimmed for type K
7.9
TEMPERATURE SENSORS
(chromel/alumel). The temperature transducer voltages and gain control resistors
are available at the package pins so that the circuit can be recalibrated for other
thermocouple types by the addition of resistors. These terminals also allow more
precise calibration for both thermocouple and thermometer applications. The
AD594/AD595 is available in two performance grades. The C and the A versions
have calibration accuracies of ±1°C and ±3°C, respectively. Both are designed to be
used with cold junctions between 0 to +50°C. The circuit shown in Figure 7.11 will
provide a direct output from a type J thermocouple (AD594) or a type K
thermocouple (AD595) capable of measuring 0 to +300°C.
AD594/AD595 MONOLITHIC THERMOCOUPLE AMPLIFIERS
WITH COLD-JUNCTION COMPENSATION
+5V
0.1µF
BROKEN
THERMOCOUPLE
ALARM
4.7kΩ
Ω
OVERLOAD
DETECT
TYPE J: AD594
TYPE K: AD595
THERMOCOUPLE
VOUT
10mV/°C
AD594/AD595
+A
–
–
G
+
G
+
+
ICE
POINT
COMP
–TC
+TC
Figure 7.11
The AD596/AD597 are monolithic set-point controllers which have been optimized
for use at elevated temperatures as are found in oven control applications. The
device cold-junction compensates and amplifies a type J/K thermocouple to derive an
internal signal proportional to temperature. They can be configured to provide a
voltage output (10mV/°C) directly from type J/K thermocouple signals. The device is
packaged in a 10-pin metal can and is trimmed to operate over an ambient range
from +25°C to +100°C. The AD596 will amplify thermocouple signals covering the
entire –200°C to +760°C temperature range recommended for type J thermocouples
while the AD597 can accommodate –200°C to +1250°C type K inputs. They have a
calibration accuracy of ±4°C at an ambient temperature of 60°C and an ambient
temperature stability specification of 0.05°C/°C from +25°C to +100°C.
None of the thermocouple amplifiers previously described compensate for
thermocouple non-linearity, they only provide conditioning and voltage gain. High
7.10
TEMPERATURE SENSORS
resolution ADCs such as the AD77XX family can be used to digitize the
thermocouple output directly, allowing a microcontroller to perform the transfer
function linearization as shown in Figure 7.12. The two multiplexed inputs to the
ADC are used to digitize the thermocouple voltage and the cold-junction
temperature sensor outputs directly. The input PGA gain is programmable from 1
to 128, and the ADC resolution is between 16 and 22 bits (depending upon the
particular ADC selected). The microcontroller performs both the cold-junction
compensation and the linearization arithmetic.
AD77XX ADC USED WITH
TMP35 TEMPERATURE SENSOR FOR CJC
3V OR 5V
(DEPENDING ON ADC)
0.1µF
AIN1+
CONTROL
REGISTER
TMP35
AIN1–
THERMO
COUPLE
MUX
AIN2+
AIN2–
PGA
Σ∆
ADC
OUTPUT
REGISTER
G=1 TO 128
AD77XX SERIES
(16-22 BITS)
SERIAL
INTERFACE
TO MICROCONTROLLER
Figure 7.12
RESISTANCE TEMPERATURE DETECTORS (RTDS)
The Resistance Temperature Detector, or the RTD, is a sensor whose resistance
changes with temperature. Typically built of a platinum (Pt) wire wrapped around a
ceramic bobbin, the RTD exhibits behavior which is more accurate and more linear
over wide temperature ranges than a thermocouple. Figure 7.13 illustrates the
temperature coefficient of a 100Ω RTD and the Seebeck coefficient of a Type S
thermocouple. Over the entire range (approximately –200°C to +850°C), the RTD is
a more linear device. Hence, linearizing an RTD is less complex.
7.11
TEMPERATURE SENSORS
RESISTANCE TEMPERATURE DETECTORs (RTD)
n Platinum (Pt) the Most Common
n 100Ω,
Ω, 1000Ω
Ω Standard Values
n Typical TC = 0.385% / °C,
0.385Ω
Ω / °C for 100Ω
Ω Pt RTD
n Good Linearity - Better than Thermocouple,
Easily Compensated
11.5
0.400
RTD
RESISTANCE
TC, ∆Ω / °C
100Ω
Ω Pt RTD
10.5
TYPE S
THERMOCOUPLE
0.375
9.50
0.350
TYPE S
THERMOCOUPLE
SEEBECK
COEFFICIENT,
µV / °C
8.50
0.325
7.50
0.300
0.275
6.50
0
400
800
5.50
TEMPERATURE - °C
Figure 7.13
Unlike a thermocouple, however, an RTD is a passive sensor and requires current
excitation to produce an output voltage. The RTD's low temperature coefficient of
0.385%/°C requires similar high-performance signal conditioning circuitry to that
used by a thermocouple; however, the voltage drop across an RTD is much larger
than a thermocouple output voltage. A system designer may opt for large value
RTDs with higher output, but large-valued RTDs exhibit slow response times.
Furthermore, although the cost of RTDs is higher than that of thermocouples, they
use copper leads, and thermoelectric effects from terminating junctions do not affect
their accuracy. And finally, because their resistance is a function of the absolute
temperature, RTDs require no cold-junction compensation.
Caution must be exercised using current excitation because the current through the
RTD causes heating. This self-heating changes the temperature of the RTD and
appears as a measurement error. Hence, careful attention must be paid to the
design of the signal conditioning circuitry so that self-heating is kept below 0.5°C.
Manufacturers specify self-heating errors for various RTD values and sizes in still
and in moving air. To reduce the error due to self-heating, the minimum current
should be used for the required system resolution, and the largest RTD value chosen
that results in acceptable response time.
Another effect that can produce measurement error is voltage drop in RTD lead
wires. This is especially critical with low-value 2-wire RTDs because the
temperature coefficient and the absolute value of the RTD resistance are both small.
If the RTD is located a long distance from the signal conditioning circuitry, then the
lead resistance can be ohms or tens of ohms, and a small amount of lead resistance
can contribute a significant error to the temperature measurement. To illustrate
7.12
TEMPERATURE SENSORS
this point, let us assume that a 100Ω platinum RTD with 30-gauge copper leads is
located about 100 feet from a controller's display console. The resistance of 30-gauge
copper wire is 0.105Ω/ft, and the two leads of the RTD will contribute a total 21Ω to
the network which is shown in Figure 7.14. This additional resistance will produce a
55°C error in the measurement! The leads' temperature coefficient can contribute an
additional, and possibly significant, error to the measurement. To eliminate the
effect of the lead resistance, a 4-wire technique is used.
A 100Ω
Ω Pt RTD WITH 100 FEET
OF 30-GAUGE LEAD WIRES
R = 10.5Ω
Ω
COPPER
100Ω
Ω
Pt RTD
R = 10.5Ω
Ω
COPPER
RESISTANCE TC OF COPPER = 0.40%/°C @ 20°C
RESISTANCE TC OF Pt RTD
= 0.385%/ °C @ 20°C
Figure 7.14
In Figure 7.15, a 4-wire, or Kelvin, connection is made to the RTD. A constant
current is applied though the FORCE leads of the RTD, and the voltage across the
RTD itself is measured remotely via the SENSE leads. The measuring device can be
a DVM or an instrumentation amplifier, and high accuracy can be achieved provided
that the measuring device exhibits high input impedance and/or low input bias
current. Since the SENSE leads do not carry appreciable current, this technique is
insensitive to lead wire length. Sources of errors are the stability of the constant
current source and the input impedance and/or bias currents in the amplifier or
DVM.
RTDs are generally configured in a four-resistor bridge circuit. The bridge output is
amplified by an instrumentation amplifier for further processing. However, high
resolution measurement ADCs such as the AD77XX series allow the RTD output to
be digitized directly. In this manner, linearization can be performed digitally,
thereby easing the analog circuit requirements.
7.13
TEMPERATURE SENSORS
FOUR-WIRE OR KELVIN CONNECTION TO Pt RTD
FOR ACCURATE MEASUREMENTS
FORCE
LEAD
RLEAD
100Ω
Ω
Pt RTD
I
FORCE
LEAD
SENSE
LEAD
RLEAD
TO HIGH - Z
IN-AMP OR ADC
SENSE
LEAD
Figure 7.15
Figure 7.16 shows a 100Ω Pt RTD driven with a 400µA excitation current source.
The output is digitized by one of the AD77XX series ADCs. Note that the RTD
excitation current source also generates the 2.5V reference voltage for the ADC via
the 6.25kΩ resistor. Variations in the excitation current do not affect the circuit
accuracy, since both the input voltage and the reference voltage vary ratiometrically
with the excitation current. However, the 6.25kΩ resistor must have a low
temperature coefficient to avoid errors in the measurement. The high resolution of
the ADC and the input PGA (gain of 1 to 128) eliminates the need for additional
conditioning circuits.
The ADT70 is a complete Pt RTD signal conditioner which provides an output
voltage of 5mV/°C when using a 1kΩ RTD (see Figure 7.17). The Pt RTD and the
1kΩ reference resistor are both excited with 1mA matched current sources. This
allows temperature measurements to be made over a range of approximately –50°C
to +800°C.
The ADT70 contains the two matched current sources, a precision rail-to-rail output
instrumentation amplifier, a 2.5V reference, and an uncommitted rail-to-rail output
op amp. The ADT71 is the same as the ADT70 except the internal voltage reference
is omitted. A shutdown function is included for battery powered equipment that
reduces the quiescent current from 3mA to 10µA. The gain or full-scale range for the
Pt RTD and ADT701 system is set by a precision external resistor connected to the
instrumentation amplifier. The uncommitted op amp may be used for scaling the
internal voltage reference, providing a "Pt RTD open" signal or "over temperature"
warning, providing a heater switching signal, or other external conditioning
determined by the user. The ADT70 is specified for operation from –40°C to +125°C
and is available in 20-pin DIP and SOIC packages.
7.14
TEMPERATURE SENSORS
INTERFACING A Pt RTD TO A HIGH RESOLUTION ADC
3V OR 5V
(DEPENDING ON ADC)
+VREF
RREF
6.25kΩ
Ω
–VREF
+
400µA
100Ω
Ω
Pt RTD
CONTROL
REGISTER
AIN1+
MUX
PGA
–
Σ∆
ADC
OUTPUT
REGISTER
AIN1–
G=1 TO 128
AD77XX SERIES
(16-22 BITS)
SERIAL
INTERFACE
TO MICROCONTROLLER
Figure 7.16
CONDITIONING THE PLATINUM RTD USING THE ADT70
+5V
0.1µF
Ω Pt
1kΩ
RTD
ADT70
+
2.5V
REFERENCE
1kΩ
Ω REF
RES
–
MATCHED
1mA SOURCES
SHUT
DOWN
+
INST
AMP
–
GND
REF
OUT = 5mV/ °C
RG = 50kΩ
Ω
-1V TO -5V
Note: Some Pins Omitted
for Clarity
Figure 7.17
7.15
TEMPERATURE SENSORS
THERMISTORS
Similar in function to the RTD, thermistors are low-cost temperature-sensitive
resistors and are constructed of solid semiconductor materials which exhibit a
positive or negative temperature coefficient. Although positive temperature
coefficient devices are available, the most commonly used thermistors are those with
a negative temperature coefficient. Figure 7.18 shows the resistance-temperature
characteristic of a commonly used NTC (Negative Temperature Coefficient)
thermistor. The thermistor is highly non-linear and, of the three temperature
sensors discussed, is the most sensitive.
RESISTANCE CHARACTERISTICS OF A
10kΩ
Ω NTC THERMISTOR
40
ALPHA THERMISTOR, INCORPORATED
RESISTANCE/TEMPERATURE CURVE 'A'
10 kΩ
Ω THERMISTOR, #13A1002-C3
30
THERMISTOR
RESISTANCE
kΩ
Ω
20
10
Nominal Value @ 25 °C
0
0
20
40
60
80
100
TEMPERATURE - °C
Figure 7.18
The thermistor's high sensitivity (typically, – 44,000ppm/°C at 25°C, as shown in
Figure 7.19), allows it to detect minute variations in temperature which could not be
observed with an RTD or thermocouple. This high sensitivity is a distinct advantage
over the RTD in that 4-wire Kelvin connections to the thermistor are not needed to
compensate for lead wire errors. To illustrate this point, suppose a 10kΩ NTC
thermistor, with a typical 25°C temperature coefficient of –44,000ppm/°C, were
substituted for the 100Ω Pt RTD in the example given earlier, then a total lead wire
resistance of 21Ω would generate less than 0.05°C error in the measurement. This is
roughly a factor of 500 improvement in error over an RTD.
7.16
TEMPERATURE SENSORS
TEMPERATURE COEFFICIENT OF
10kΩ
Ω NTC THERMISTOR
-60000
ALPHA THERMISTOR, INCORPORATED
RESISTANCE/TEMPERATURE CURVE 'A'
10 kΩ
Ω THERMISTOR, #13A1002-C3
-50000
THERMISTOR
TEMPERATURE
COEFFICIENT
ppm/ °C
-40000
-30000
-20000
0
20
40
60
80
100
TEMPERATURE - °C
Figure 7.19
However, the thermistor's high sensitivity to temperature does not come without a
price. As was shown in Figure 7.18, the temperature coefficient of thermistors does
not decrease linearly with increasing temperature as it does with RTDs; therefore,
linearization is required for all but the narrowest of temperature ranges. Thermistor
applications are limited to a few hundred degrees at best because they are more
susceptible to damage at high temperatures. Compared to thermocouples and RTDs,
thermistors are fragile in construction and require careful mounting procedures to
prevent crushing or bond separation. Although a thermistor's response time is short
due to its small size, its small thermal mass makes it very sensitive to self-heating
errors.
Thermistors are very inexpensive, highly sensitive temperature sensors. However,
we have shown that a thermistor's temperature coefficient varies from –44,000
ppm/°C at 25°C to –29,000ppm/°C at 100°C. Not only is this non-linearity the
largest source of error in a temperature measurement, it also limits useful
applications to very narrow temperature ranges if linearization techniques are not
used.
It is possible to use a thermistor over a wide temperature range only if the system
designer can tolerate a lower sensitivity to achieve improved linearity. One approach
to linearizing a thermistor is simply shunting it with a fixed resistor. Paralleling the
thermistor with a fixed resistor increases the linearity significantly. As shown in
Figure 7.20, the parallel combination exhibits a more linear variation with
temperature compared to the thermistor itself. Also, the sensitivity of the
combination still is high compared to a thermocouple or RTD. The primary
7.17
TEMPERATURE SENSORS
disadvantage to this technique is that linearization can only be achieved within a
narrow range.
LINEARIZATION OF NTC THERMISTOR
USING A 5.17kΩ
Ω SHUNT RESISTOR
40
30
RESISTANCE
kΩ
Ω
20
THERMISTOR
PARALLEL COMBINATION
10
0
0
20
40
60
80
100
TEMPERATURE - °C
Figure 7.20
The value of the fixed resistor can be calculated from the following equation:
R=
RT2 ⋅ ( RT1 + RT3 ) − 2 ⋅ RT1 ⋅ RT3
,
RT1 + RT3 − 2 ⋅ RT2
where RT1 is the thermistor resistance at T1, the lowest temperature in the
measurement range, RT3 is the thermistor resistance at T3, the highest
temperature in the range, and RT2 is the thermistor resistance at T2, the midpoint,
T2 = (T1 +T3)/2.
For a typical 10kΩ NTC thermistor, RT1 = 32,650Ω at 0°C, RT2 = 6,532Ω at 35°C,
and RT3 = 1,752Ω at 70°C. This results in a value of 5.17kΩ for R. The accuracy
needed in the signal conditioning circuitry depends on the linearity of the network.
For the example given above, the network shows a non-linearity of – 2.3°C/ + 2.0 °C.
The output of the network can be applied to an ADC to perform further linearization
as shown in Figure 7.21. Note that the output of the thermistor network has a slope
of approximately –10mV/°C, which implies a 12-bit ADC has more than sufficient
resolution.
7.18
TEMPERATURE SENSORS
LINEARIZED THERMISTOR AMPLIFIER
226µA
VOUT ≈ 0.994V @ T = 0°C
VOUT ≈ 0.294V @ T =70°C
∆VOUT/∆
∆T ≈ −10mV/°C
−
10kΩ
Ω NTC
THERMISTOR
AMPLIFIER
OR ADC
5.17kΩ
Ω
LINEARIZATION
RESISTOR
LINEARITY ≈ ± 2°C, 0°C TO +70°C
Figure 7.21
SEMICONDUCTOR TEMPERATURE SENSORS
Modern semiconductor temperature sensors offer high accuracy and high linearity
over an operating range of about –55°C to +150°C. Internal amplifiers can scale the
output to convenient values, such as 10mV/°C. They are also useful in cold-junctioncompensation circuits for wide temperature range thermocouples.
All semiconductor temperature sensors make use of the relationship between a
bipolar junction transistor's (BJT) base-emitter voltage to its collector current:
VBE =
kT  I c 
ln 
q
 Is 
where k is Boltzmann's constant, T is the absolute temperature, q is the charge of
an electron, and Is is a current related to the geometry and the temperature of the
transistors. (The equation assumes a voltage of at least a few hundred mV on the
collector, and ignores Early effects.)
If we take N transistors identical to the first (see Figure 7.22) and allow the total
current Ic to be shared equally among them, we find that the new base-emitter
voltage is given by the equation
VN =
kT  I c 
ln

q
 N ⋅ Is 
7.19
TEMPERATURE SENSORS
BASIC RELATIONSHIPS FOR SEMICONDUCTOR
TEMPERATURE SENSORS
IC
IC
ONE TRANSISTOR
VBE
VBE =
N TRANSISTORS
VN
kT  IC 
ln 
q  IS 
VN =
∆ VBE = VBE − VN =
kT  IC 
ln

q  N ⋅ IS 
kT
ln(N)
q
INDEPENDENT OF IC, IS
Figure 7.22
Neither of these circuits is of much use by itself because of the strongly temperature
dependent current Is, but if we have equal currents in one BJT and N similar BJTs
then the expression for the difference between the two base-emitter voltages is
proportional to absolute temperature and does not contain Is.
∆VBE = VBE − VN =
kT  I c  kT  I c 
ln  −
ln

q
q
 Is 
 N ⋅ Is 
∆VBE = VBE − VN =
 Ic  
kT   I c 

 ln  − ln
q   Is 
 N ⋅ Is  
 I c 

 

kT  Is 
 = kT ln( N )
∆VBE = VBE − VN =
ln
 Ic  

q
q



 N ⋅ Is  

The circuit shown in Figure 7.23 implements the above equation and is known as
the "Brokaw Cell" (see Reference 10). The voltage ∆VBE = VBE – VN appears across
resistor R2. The emitter current in Q2 is therefore ∆VBE/R2. The op amp's servo
loop and the resistors, R, force the same current to flow through Q1. The Q1 and Q2
currents are equal and are summed and flow into resistor R1. The corresponding
voltage developed across R1 is proportional to absolute temperature (PTAT) and
given by:
7.20
TEMPERATURE SENSORS
VPTAT =
2R1( VBE − VN )
R1 kT
=2
ln( N ) .
R2
R2 q
CLASSIC BANDGAP TEMPERATURE SENSOR
+VIN
R
"BROKAW CELL"
R
Q2
NA
Q1
A
VN
kT
ln(N)
∆ VBE = VBE − VN =
q
VBANDGAP = 1.205V
+
I2 ≅ I1
VBE
(Q1)
R2
VPTAT = 2
R1 kT
ln(N)
R2 q
R1
Figure 7.23
The bandgap cell reference voltage, VBANDGAP, appears at the base of Q1 and is
the sum of VBE(Q1) and VPTAT. VBE(Q1) is complementary to absolute
temperature (CTAT), and summing it with VPTAT causes the bandgap voltage to be
constant with respect to temperature (assuming proper choice of R1/R2 ratio and N
to make the bandgap voltage equal to1.205V). This circuit is the basic band-gap
temperature sensor, and is widely used in semiconductor temperature sensors.
Current and Voltage Output Temperature Sensors
The concepts used in the bandgap temperature sensor discussion above can be used
as the basis for a variety of IC temperature sensors to generate either current or
voltage outputs. The AD592 and TMP17 (see Figure 7.24) are current output
sensors which have scale factors of 1µA/K. The sensors do not require external
calibration and are available in several accuracy grades. The AD592 is available in
three accuracy grades. The highest grade version (AD592CN) has a maximum error
@ 25ºC of ±0.5ºC and ±1.0ºC error from –25ºC to +105ºC. Linearity error is ±0.35ºC.
The TMP17 is available in two accuracy grades. The highest grade version
(TMP17F) has a maximum error @ 25ºC of ±2.5ºC and ±3.5ºC error from –40ºC to
+105ºC. Typical linearity error is ±0.5ºC. The AD592 is available in a TO-92 package
and the TMP17 in an SO-8 package.
7.21
TEMPERATURE SENSORS
CURRENT OUTPUT SENSORS: AD592, TMP17
V+
AD592: TO-92 PACKAGE
TMP17: SO-8 PACKAGE
V–
n 1µA/K Scale Factor
n Nominal Output Current @ +25°C: 298.2µA
n Operation from 4V to 30V
n ±0.5°C Max Error @ 25°C, ±1.0°C Error Over Temp,
±0.1°C Typical Nonlinearity (AD592CN)
n ±2.5°C Max Error @ 25°C, ±3.5°C Error Over Temp,
±0.5°C Typical Nonlinearity (TMP17F)
n AD592 Specified from –25°C to +105°C
n TMP17 Specified from –40°C to +105°C
Figure 7.24
RATIOMETRIC VOLTAGE OUTPUT SENSORS
VS = +3.3V
0.1µF
REFERENCE
I(VS)
ADC
+
VOUT
INPUT
–
R(T)
GND
AD22103
VOUT =
VS
28mV


×  0.25 V +
× TA 

°C
3.3 V 
Figure 7.25
7.22
TEMPERATURE SENSORS
In some cases, it is desirable for the output of a temperature sensor to be ratiometric
with its supply voltage. The AD22103 (see Figure 7.25) has an output that is
ratiometric with its supply voltage (nominally 3.3V) according to the equation:
VOUT =
VS 
28mV

×  0.25V +
× TA  .

3.3V 
°C
The circuit shown in Figure 7.25 uses the AD22103 power supply as the reference to
the ADC, thereby eliminating the need for a precision voltage reference. The
AD22103 is specified over a range of 0°C to +100°C and has an accuracy better than
±2.5°C and a linearity better than ±0.5°C.
The TMP35/TMP36/TMP37 are low voltage (2.7V to 5.5V) SOT-23 (5-pin), SO-8, or
TO-92 packaged voltage output temperature sensors with a 10mV/°C (TMP35/36) or
20mV/°C (TMP37) scale factor (see Figure 7.26). Supply current is below 50µA,
providing very low self-heating (less than 0.1°C in still air). A shutdown feature is
provided which reduces the current to 0.5µA.
The TMP35 provides a 250mV output at +25°C and reads temperature from +10°C
to +125°C. The TMP36 is specified from –40°C to +125°C. and provides a 750mV
output at 25°C. Both the TMP35 and TMP36 have an output scale factor of
+10mV/°C. The TMP37 is intended for applications over the range +5°C to +100°C,
and provides an output scale factor of 20mV/°C. The TMP37 provides a 500mV
output at +25°C.
ABSOLUTE VOLTAGE OUTPUT SENSORS
WITH SHUTDOWN
+VS = 2.7V TO 5.5V
SHUTDOWN
0.1µF
TMP35
TMP36
TMP37
ALSO
SO-8
OR TO-92
VOUT
SOT-23-5
n VOUT:
u TMP35, 250mV @ 25°C, 10mV/°C (+10°C to +125°C)
u TMP36, 750mV @ 25°C, 10mV/°C (–40°C to +125°C)
u TMP37, 500mV @ 25°C, 20mV/°C ( +5°C to +100°C)
n ±2°C Error Over Temp (Typical), ±0.5°C Non-Linearity (Typical)
n Specified –40°C to +125°C
n 50µA Quiescent Current, 0.5µA in Shutdown Mode
Figure 7.26
7.23
TEMPERATURE SENSORS
The ADT45/ADT50 are voltage output temperature sensors packaged in a SOT-23-3
package designed for an operating voltage of 2.7V to 12V (see Figure 7.27). The
devices are specified over the range of –40ºC to +125ºC. The output scale factor for
both devices is 10mV/ºC. Typical accuracies are ±1ºC at +25ºC and ±2ºC over the –
40ºC to +125ºC range. The ADT45 provides a 250mV output at +25ºC and is
specified for temperature from 0ºC to +100ºC. The ADT50 provides a 750mV output
at +25ºC and is specified for temperature from –40ºC to +125ºC.
ADT45/ADT50 ABSOLUTE VOLTAGE OUTPUT SENSORS
+VS = 2.7V TO 12V
VOUT
ADT45
ADT50
0.1µF
SOT-23
n VOUT:
u ADT45, 250mV @ 25°C, 10mV/°C Scale Factor
u ADT50, 750mV @ 25°C, 10mV/°C Scale Factor
n ±2°C Error Over Temp (Typical), ±0.5°C Non-Linearity (Typical)
n Specified –40°C to +125°C
n 60µA Quiescent Current
Figure 7.27
If the ADT45/ADT50 sensors are thermally attached and protected, they can be
used in any temperature measurement application where the maximum
temperature range of the medium is between –40°C to +125°C. Properly cemented
or glued to the surface of the medium, these sensors will be within 0.01°C of the
surface temperature. Caution should be exercised, as any wiring to the device can
act as heat pipes, introducing errors if the surrounding air-surface interface is not
isothermal. Avoiding this condition is easily achieved by dabbing the leads of the
sensor and the hookup wires with a bead of thermally conductive epoxy. This will
ensure that the ADT45/ADT50 die temperature is not affected by the surrounding
air temperature.
7.24
TEMPERATURE SENSORS
In the SOT-23-3 package, the thermal resistance junction-to-case, θJC, is 180°C/W.
The thermal resistance case-to-ambient, θCA, is the difference between θJA and
θJC, and is determined by the characteristics of the thermal connection. With no air
flow and the device soldered on a PC board, θJA is 300°C/W. The temperature
sensor's power dissipation, PD, is the product of the total voltage across the device
and its total supply current (including any current delivered to the load). The rise in
die temperature above the medium's ambient temperature is given by:
TJ = PD × (θJC + θCA) + TA.
Thus, the die temperature rise of an unloaded ADT45/ADT50 (SOT-23-3 package)
soldered on a board in still air at 25°C and driven from a +5V supply (quiescent
current = 60µA, PD = 300µW) is less than 0.09°C. In order to prevent further
temperature rise, it is important to minimize the load current, always keeping it less
than 100µA.
The transient response of the ADT45/ADT50 sensors to a step change in
temperature is determined by the thermal resistances and the thermal mass of the
die and the case. The thermal mass of the case varies with the measurement
medium since it includes anything that is in direct contact with the package. In all
practical cases, the thermal mass of the case is the limiting factor in the thermal
response time of the sensor and can be represented by a single-pole RC time
constant. Thermal mass is often considered the thermal equivalent of electrical
capacitance.
The thermal time constant of a temperature sensor is defined to be the time
required for the sensor to reach 63.2% of the final value for a step change in the
temperature. Figure 7.28 shows the thermal time constant of the ADT45/ADT50
series of sensors with the SOT-23-3 package soldered to 0.338" x 0.307" copper PC
board as a function of air flow velocity. Note the rapid drop from 32 seconds to 12
seconds as the air velocity increases from 0 (still air) to 100 LFPM. As a point of
reference, the thermal time constant of the ADT45/ADT50 series in a stirred oil bath
is less than 1 second, which verifies that the major part of the thermal time constant
is determined by the case.
The power supply pin of these sensors should be bypassed to ground with a 0.1µF
ceramic capacitor having very short leads (preferably surface mount) and located as
close to the power supply pin as possible. Since these temperature sensors operate
on very little supply current and could be exposed to very hostile electrical
environments, it is important to minimize the effects of EMI/RFI on these devices.
The effect of RFI on these temperature sensors is manifested as abnormal DC shifts
in the output voltage due to rectification of the high frequency noise by the internal
IC junctions. In those cases where the devices are operated in the presence of high
frequency radiated or conducted noise, a large value tantalum electrolytic capacitor
(>2.2µF) placed across the 0.1µF ceramic may offer additional noise immunity.
7.25
TEMPERATURE SENSORS
THERMAL RESPONSE IN FORCED AIR FOR SOT-23-3
35
SOT-23-3 SOLDERED TO 0.338" x 0.307" Cu PCB
V+ = 2.7V TO 5V
NO LOAD
30
25
TIME
CONSTANTSECONDS
20
15
10
5
0
0
100
200
300
400
500
600
700
AIR VELOCITY - LFPM
Figure 7.28
Digital Output Temperature Sensors
Temperature sensors which have digital outputs have a number of advantages over
those with analog outputs, especially in remote applications. Opto-isolators can also
be used to provide galvanic isolation between the remote sensor and the
measurement system. A voltage-to-frequency converter driven by a voltage output
temperature sensor accomplishes this function, however, more sophisticated ICs are
now available which are more efficient and offer several performance advantages.
The TMP03/TMP04 digital output sensor family includes a voltage reference,
VPTAT generator, sigma-delta ADC, and a clock source (see Figure 7.29). The
sensor output is digitized by a first-order sigma-delta modulator, also known as the
"charge balance" type analog-to-digital converter. This converter utilizes timedomain oversampling and a high accuracy comparator to deliver 12 bits of effective
accuracy in an extremely compact circuit.
The output of the sigma-delta modulator is encoded using a proprietary technique
which results in a serial digital output signal with a mark-space ratio format (see
Figure 7.30) that is easily decoded by any microprocessor into either degrees
centigrade or degrees Fahrenheit, and readily transmitted over a single wire. Most
importantly, this encoding method avoids major error sources common to other
modulation techniques, as it is clock-independent. The nominal output frequency is
35Hz at + 25ºC, and the device operates with a fixed high-level pulse width (T1) of
10ms.
7.26
TEMPERATURE SENSORS
DIGITAL OUTPUT SENSORS: TMP03/04
+VS = 4.5 TO 7V
REFERENCE
VOLTAGE
TEMP
SENSOR
VPTAT
CLOCK
(1MHz)
SIGMA-DELTA
ADC
OUTPUT
(TMP04)
OUTPUT
(TMP03)
TMP03/TMP04
GND
Figure 7.29
TMP03/TMP04 OUTPUT FORMAT
T1
T2
 400 × T1
TEMPERATURE (° C) = 235 − 


T2 
 720 × T1
TEMPERATURE (° F) = 455 − 


T2 
n
n
n
n
n
n
n
T1 Nominal Pulse Width = 10ms
±1.5°C Error Over Temp, ±0.5°C Non-Linearity (Typical)
Specified –40°C to +100°C
Nominal T1/T2 @ 0°C = 60%
Nominal Frequency @ +25°C = 35Hz
6.5mW Power Consumption @ 5V
TO-92, SO-8, or TSSOP Packages
Figure 7.30
7.27
TEMPERATURE SENSORS
The TMP03/TMP04 output is a stream of digital pulses, and the temperature
information is contained in the mark-space ratio per the equations:
 400 × T1 
Temperature ( ° C) = 235 − 

 T2 
 720 × T1 
Temperature ( ° F) = 455 − 
.
 T2 
Popular microcontrollers, such as the 80C51 and 68HC11, have on-chip timers
which can easily decode the mark-space ratio of the TMP03/TMP04. A typical
interface to the 80C51 is shown in Figure 7.31. Two timers, labeled Timer 0 and
Timer 1 are 16 bits in length. The 80C51's system clock, divided by twelve, provides
the source for the timers. The system clock is normally derived from a crystal
oscillator, so timing measurements are quite accurate. Since the sensor's output is
ratiometric, the actual clock frequency is not important. This feature is important
because the microcontroller's clock frequency is often defined by some external
timing constraint, such as the serial baud rate.
INTERFACING TMP04 TO A MICROCONTROLLER
+5V
XTAL
0.1µF
V+
OSCILLATOR
÷12
TIMER 0
TMP04
OUT
CPU
P1.0
GND
TIMER
CONTROL
TIMER 1
80C51 MICROCONTROLLER
NOTE:
ADDITIONAL
PINS OMITTED
FOR CLARITY
Figure 7.31
Software for the sensor interface is straightforward. The microcontroller simply
monitors I/O port P1.0, and starts Timer 0 on the rising edge of the sensor output.
The microcontroller continues to monitor P1.0, stopping Timer 0 and starting Timer
1 when the sensor output goes low. When the output returns high, the sensor's T1
and T2 times are contained in registers Timer 0 and Timer 1, respectively. Further
software routines can then apply the conversion factor shown in the equations above
and calculate the temperature.
7.28
TEMPERATURE SENSORS
The TMP03/TMP04 are ideal for monitoring the thermal environment within
electronic equipment. For example, the surface mounted package will accurately
reflect the thermal conditions which affect nearby integrated circuits. The TO-92
package, on the other hand, can be mounted above the surface of the board to
measure the temperature of the air flowing over the board.
The TMP03 and TMP04 measure and convert the temperature at the surface of
their own semiconductor chip. When they are used to measure the temperature of a
nearby heat source, the thermal impedance between the heat source and the sensor
must be considered. Often, a thermocouple or other temperature sensor is used to
measure the temperature of the source, while the TMP03/TMP04 temperature is
monitored by measuring T1 and T2. Once the thermal impedance is determined, the
temperature of the heat source can be inferred from the TMP03/TMP04 output.
One example of using the TMP04 to monitor a high power dissipation
microprocessor or other IC is shown in Figure 7.32. The TMP04, in a surface mount
package, is mounted directly beneath the microprocessor's pin grid array (PGA)
package. In a typical application, the TMP04's output would be connected to an
ASIC where the mark-space ratio would be measured. The TMP04 pulse output
provides a significant advantage in this application because it produces a linear
temperature output, while needing only one I/O pin and without requiring an ADC.
MONITORING HIGH POWER MICROPROCESSOR
OR DSP WITH TMP04
FAST MICROPROCESSOR, DSP, ETC.,
IN PGA PACKAGE
PGA SOCKET
PC BOARD
TMP04 IN SURFACE
MOUNT PACKAGE
Figure 7.32
Thermostatic Switches and Setpoint Controllers
Temperature sensors used in conjunction with comparators can act as thermostatic
switches. ICs such as the ADT05 accomplish this function at low cost and allow a
single external resistor to program the setpoint to 2ºC accuracy over a range of –
40ºC to +150ºC (see Figure 7.33). The device asserts an open collector output when
the ambient temperature exceeds the user-programmed setpoint temperature. The
ADT05 has approximately 4ºC of hysteresis which prevents rapid thermal on/off
cycling. The ADT05 is designed to operate on a single supply voltage from +2.7V to
7.29
TEMPERATURE SENSORS
+7.0V facilitating operation in battery powered applications as well as industrial
control systems. Because of low power dissipation (200µW @ 3.3V), self-heating
errors are minimized, and battery life is maximized. An optional internal 200kΩ
pull-up resistor is included to facilitate driving light loads such as CMOS inputs.
The setpoint resistor is determined by the equation:
R SET =
39 MΩ° C
− 90.3 kΩ .
TSET ( ° C) + 281.6° C
The setpoint resistor should be connected directly between the RSET pin (Pin 4) and
the GND pin (Pin 5). If a ground plane is used, the resistor may be connected
directly to this plane at the closest available point.
The setpoint resistor can be of nearly any resistor type, but its initial tolerance and
thermal drift will affect the accuracy of the programmed switching temperature. For
most applications, a 1% metal-film resistor will provide the best tradeoff between
cost and accuracy. Once RSET has been calculated, it may be found that the
calculated value does not agree with readily available standard resistors of the
chosen tolerance. In order to achieve a value as close as possible to the calculated
value, a compound resistor can be constructed by connecting two resistors in series
or parallel.
ADT05 THERMOSTATIC SWITCH
+VS = 2.7V TO 7V
ADT05
200kΩ
Ω
RPULL-UP
TEMP
SENSOR
OUT
0.1µF
SETPOINT
SOT-23-5
RSET
n
n
n
n
±2°C Setpoint Accuracy
4°C Preset Hysteresis
Specified Operating Range: –40°C to + 150°C
Power Dissipation: 200µW @ 3.3V
Figure 7.33
7.30
TEMPERATURE SENSORS
The TMP01 is a dual setpoint temperature controller which also generates a PTAT
output voltage (see Figure 7.34 and 7.35). It also generates a control signal from one
of two outputs when the device is either above or below a specific temperature
range. Both the high/low temperature trip points and hysteresis band are
determined by user-selected external resistors.
TMP01 PROGRAMMABLE SETPOINT CONTROLLER
TMP01
VREF
TEMPERATURE
SENSOR AND
VOLTAGE
REFERENCE
2.5V
R1
–
SET
HIGH
V+
OVER
+
R2
WINDOW
COMPARATOR
SET
LOW
+
R3
UNDER
–
GND
HYSTERESIS
GENERATOR
VPTAT
Figure 7.34
The TMP01 consists of a bandgap voltage reference combined with a pair of matched
comparators. The reference provides both a constant 2.5V output and a PTAT
output voltage which has a precise temperature coefficient of 5mV/K and is 1.49V
(nominal) at +25ºC. The comparators compare VPTAT with the externally set
temperature trip points and generate an open-collector output signal when one of
their respective thresholds has been exceeded.
Hysteresis is also programmed by the external resistor chain and is determined by
the total current drawn out of the 2.5V reference. This current is mirrored and used
to generate a hysteresis offset voltage of the appropriate polarity after a comparator
has been tripped. The comparators are connected in parallel, which guarantees that
there is no hysteresis overlap and eliminates erratic transitions between adjacent
trip zones.
7.31
TEMPERATURE SENSORS
The TMP01 utilizes laser trimmed thin-film resistors to maintain a typical
temperature accuracy of ±1ºC over the rated temperature range. The open-collector
outputs are capable of sinking 20mA, enabling the TMP01 to drive control relays
directly. Operating from a +5V supply, quiescent current is only 500µA maximum.
TMP01 SETPOINT CONTROLLER KEY FEATURES
n VC: 4.5 to 13.2V
n Temperature Output: VPTAT, +5mV/K
n Nominal 1.49V Output @ 25°C
n ±1°C Typical Accuracy Over Temperature
n Specified Operating Range: –55°C to + 125°C
n Resistor-Programmable Hysteresis
n Resistor-Programmable Setpoints
n Precision 2.5V ±8mV Reference
n 400µA Quiescent Current, 1µA in Shutdown
n Packages: 8-Pin Dip, 8-Pin SOIC, 8-Pin TO-99
n Other Setpoint Controllers:
u Dual Setpoint Controllers: ADT22/ADT23
(3V Versions of TMP01 with Internal Hysteresis)
u Quad Setpoint Controller: ADT14
Figure 7.35
The ADT22/23-series are similar to the TMP01 but have internal hysteresis and are
designed to operate on a 3V supply. A quad (ADT14) setpoint controller is also
available.
ADCs With On-Chip Temperature Sensors
The AD7816/7817/7818-series digital temperature sensors have on-board
temperature sensors whose outputs are digitized by a 10-bit 9µs conversion time
switched capacitor SAR ADC. The serial interface is compatible with the Intel 8051,
Motorola SPI™ and QSPI™, and National Semiconductor's MICROWIRE™
protocol. The device family offers a variety of input options for further flexibility.
The AD7416/7417/7418 are similar but have standard serial interfaces. Functional
block diagrams of the AD7816, AD7817, and AD7818 are shown in Figures 7.36, 37,
and 38, and key specifications in Figure 7.39
7.32
TEMPERATURE SENSORS
AD7816 10-BIT DIGITAL TEMPERATURE SENSOR
WITH SERIAL INTERFACE
REFIN
+VDD = 2.7V TO 5.5V
AD7816
TEMP
SENSOR
2.5V
REF
OVER TEMP
REGISTER
A>B
DIN/OUT
CLOCK
10-BIT
CHARGE
REDISTRIBUTION
SAR ADC
MUX
AGND
OTI
OUTPUT
REGISTER
SCLK
CONTROL
REGISTER
RD/WR
CONVST
Figure 7.36
AD7817 10-BIT MUXED INPUT ADC WITH TEMP SENSOR
REFIN
+VDD = 2.7V TO 5.5V
AD7817
TEMP
SENSOR
2.5V
REF
OVER TEMP
REGISTER
A>B
DOUT
CLOCK
VIN1
VIN2
VIN3
10-BIT
CHARGE
REDISTRIBUTION
SAR ADC
MUX
VIN4
OTI
OUTPUT
REGISTER
CONTROL
REGISTER
SCLK
RD/WR
DIN
CS
AGND
DGND
BUSY
CONVST
Figure 7.37
7.33
TEMPERATURE SENSORS
AD7818 SINGLE INPUT 10-BIT ADC WITH TEMP SENSOR
+VDD = 2.7V TO 5.5V
AD7818
TEMP
SENSOR
2.5V
REF
OVER TEMP
REGISTER
A>B
DIN/OUT
CLOCK
VIN1
10-BIT
CHARGE
REDISTRIBUTION
SAR ADC
MUX
OTI
OUTPUT
REGISTER
SCLK
CONTROL
REGISTER
RD/WR
CONVST
AGND
Figure 7.38
AD7816/7817/7818 - SERIES TEMP SENSOR
10-BIT ADCs WITH SERIAL INTERFACE
n 10-Bit ADC with 9µs Conversion Time
n Flexible Serial Interface (Intel 8051, Motorola SPI™ and QSPI™,
National MICROWIRE™)
n On-Chip Temperature Sensor: –55°C to +125°C
n Temperature Accuracy: ± 2°C from –40°C to +85°C
n On-Chip Voltage Reference: 2.5V ±1%
n +2.7V to +5.5V Power Supply
n 4µW Power Dissipation at 10Hz Sampling Rate
n Auto Power Down after Conversion
n Over-Temp Interrupt Output
n Four Single-Ended Analog Input Channels: AD7817
n One Single-Ended Analog Input Channel: AD7818
n AD7416/7417/7418: Similar, but have I2C Compatible Interface
Figure 7.39
7.34
TEMPERATURE SENSORS
MICROPROCESSOR TEMPERATURE MONITORING
Today's computers require that hardware as well as software operate properly, in
spite of the many things that can cause a system crash or lockup. The purpose of
hardware monitoring is to monitor the critical items in a computing system and take
corrective action should problems occur.
Microprocessor supply voltage and temperature are two critical parameters. If the
supply voltage drops below a specified minimum level, further operations should be
halted until the voltage returns to acceptable levels. In some cases, it is desirable to
reset the microprocessor under "brownout" conditions. It is also common practice to
reset the microprocessor on power-up or power-down. Switching to a battery backup
may be required if the supply voltage is low.
Under low voltage conditions it is mandatory to inhibit the microprocessor from
writing to external CMOS memory by inhibiting the Chip Enable signal to the
external memory.
Many microprocessors can be programmed to periodically output a "watchdog"
signal. Monitoring this signal gives an indication that the processor and its software
are functioning properly and that the processor is not stuck in an endless loop.
The need for hardware monitoring has resulted in a number of ICs, traditionally
called "microprocessor supervisory products," which perform some or all of the above
functions. These devices range from simple manual reset generators (with
debouncing) to complete microcontroller-based monitoring sub-systems with on-chip
temperature sensors and ADCs. Analog Devices' ADM-family of products is
specifically to perform the various microprocessor supervisory functions required in
different systems.
CPU temperature is critically important in the Pentium II microprocessors. For this
reason, all new Pentium II devices have an on-chip substrate PNP transistor which
is designed to monitor the actual chip temperature. The collector of the substrate
PNP is connected to the substrate, and the base and emitter are brought out on two
separate pins of the Pentium II.
The ADM1021 Microprocessor Temperature Monitor is specifically designed to
process these outputs and convert the voltage into a digital word representing the
chip temperature. The simplified analog signal processing portion of the ADM1021
is shown in Figure 7.40.
The technique used to measure the temperature is identical to the "∆VBE" principle
previously discussed. Two different currents (I and N·I)are applied to the sensing
transistor, and the voltage measured for each. In the ADM1021, the nominal
currents are I = 6µA, (N = 17), N·I = 102µA. The change in the base-emitter voltage,
∆VBE, is a PTAT voltage and given by the equation:
∆VBE =
kT
ln( N ) .
q
7.35
TEMPERATURE SENSORS
Figure 7.40 shows the external sensor as a substrate transistor, provided for
temperature monitoring in the microprocessor, but it could equally well be a discrete
transistor. If a discrete transistor is used, the collector should be connected to the
base and not grounded. To prevent ground noise interfering with the measurement,
the more negative terminal of the sensor is not referenced to ground, but is biased
above ground by an internal diode. If the sensor is operating in a noisy environment,
C may be optionally added as a noise filter. Its value is typically 2200pF, but should
be no more than 3000pF.
ADM1021 MICROPROCESSOR TEMPERATURE MONITOR
INPUT SIGNAL CONDITIONING CIRCUITS
VDD = +3V TO +5.5V
I
µP
REMOTE
SENSING
TRANSISTOR
N×I
IBIAS
OSCILLATOR
D+
C
D–
65kHz
LOWPASS
FILTER
kT
∆VBE = q ln N
SPNP
BIAS
DIODE
GAIN
=G
VOUT
TO ADC
CHOPPER
AMPLIFIER
AND RECTIFIER
kT
VOUT = G • q ln N
Figure 7.40
To measure ∆VBE, the sensing transistor is switched between operating currents of
I and N·I. The resulting waveform is passed through a 65kHz lowpass filter to
remove noise, then to a chopper-stabilized amplifier which performs the function of
amplification and synchronous rectification. The resulting DC voltage is proportional
to ∆VBE and is digitized by an 8-bit ADC. To further reduce the effects of noise,
digital filtering is performed by averaging the results of 16 measurement cycles.
7.36
TEMPERATURE SENSORS
In addition, the ADM1021 contains an on-chip temperature sensor, and its signal
conditioning and measurement is performed in the same manner.
One LSB of the ADC corresponds to 1ºC, so the ADC can theoretically measure from
–128ºC to +127ºC, although the practical lowest value is limited to –65ºC due to
device maximum ratings. The results of the local and remote temperature
measurements are stored in the local and remote temperature value registers, and
are compared with limits programmed into the local and remote high and low limit
registers as shown in Figure 7.41. An ALERT output signals when the on-chip or
remote temperature is out of range. This output can be used as an interrupt, or as
an SMBus alert.
The limit registers can be programmed, and the device controlled and configured, via
the serial System Management Bus (SMBus). The contents of any register can also
be read back by the SMBus. Control and configuration functions consist of:
switching the device between normal operation and standby mode, masking or
enabling the ALERT output, and selecting the conversion rate which can be set
from 0.0625Hz to 8Hz.
ADM1021 SIMPLIFIED BLOCK DIAGRAM
D–
SIGNAL CONDITIONING
AND ANALOG MUX
ADDRESS POINTER
REGISTER
TEMP
SENSOR
ONE-SHOT
REGISTER
CONVERSION RATE
REGISTER
LOCAL TEMPERATURE
VALUE REGISTER
8-BIT
ADC
BUSY
D+
REMOTE TEMPERATURE
VALUE REGISTER
LOCAL TEMPERATURE
LOW LIMIT COMPARATOR
LOCAL TEMPERATURE
LOW LIMIT REGISTER
LOCAL TEMPERATURE
HIGH LIMIT COMPARATOR
LOCAL TEMPERATURE
HIGH LIMIT REGISTER
REMOTE TEMPERATURE
LOW LIMIT COMPARATOR
REMOTE TEMPERATURE
LOW LIMIT REGISTER
REMOTE TEMPERATURE
HIGH LIMIT COMPARATOR
REMOTE TEMPERATURE
HIGH LIMIT REGISTER
RUN/STANDBY
CONFIGURATION
REGISTER
STBY
EXTERNAL DIODE OPEN CIRCUIT
STATUS
REGISTER
INTERRUPT
MASKING
ALERT
SMBUS INTERFACE
TEST VDD NC GND GND NC NC TEST
SDATA
SCLK
ADD0
ADD1
Figure 7.41
7.37
TEMPERATURE SENSORS
ADM1021 KEY SPECIFICATIONS
n On-Chip and Remote Temperature Sensing
n 1°C Accuracy for On-Chip Sensor
n 3°C Accuracy for Remote Sensor
n Programmable Over / Under Temperature Limits
n 2-Wire SMBus Serial Interface
n 70µA Max Operating Current
n 3µA Standby Current
n +3V to +5.5V Supplies
n 16-Pin QSOP Package
Figure 7.42
7.38
TEMPERATURE SENSORS
REFERENCES
1.
Ramon Pallas-Areny and John G. Webster, Sensors and Signal
Conditioning, John Wiley, New York, 1991.
2.
Dan Sheingold, Editor, Transducer Interfacing Handbook, Analog
Devices, Inc., 1980.
3.
Walt Kester, Editor, 1992 Amplifier Applications Guide, Section 2, 3,
Analog Devices, Inc., 1992.
4.
Walt Kester, Editor, System Applications Guide, Section 1, 6, Analog
Devices, Inc., 1993.
5.
Jim Williams, Thermocouple Measurement, Linear Technology
Application Note 28, Linear Technology Corporation.
6.
Dan Sheingold, Nonlinear Circuits Handbook, Analog Devices, Inc.
7.
James Wong, Temperature Measurements Gain from Advances in Highprecision Op Amps, Electronic Design, 15 May 1986.
8.
OMEGA Temperature Measurement Handbook, Omega Instruments, Inc.
9.
Handbook of Chemistry and Physics, Chemical Rubber Co.
10.
Paul Brokaw, A Simple Three-Terminal IC Bandgap Voltage Reference,
IEEE Journal of Solid State Circuits, Vol. SC-9, December, 1974.
7.39
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