TDA2030A

TDA2030A
TDA2030A
18W Hi-Fi AMPLIFIER AND 35W DRIVER
DESCRIPTION
The TDA2030A is a monolithic IC in Pentawatt 
package intended for use as low frequency class
AB amplifier.
With VS max = 44V it is particularly suited for more
reliable applications without regulated supply and
for 35W driver circuits using low-cost complementary pairs.
The TDA2030A provides high output current and
has very low harmonic and cross-over distortion.
Further the device incorporates a short circuit protection system comprising an arrangement for
automatically limiting the dissipated power so as to
keep the working point of the output transistors
within their safe operating area. A conventional
thermal shut-down system is also included.
PENTAWATT
ORDERING NUMBERS : TDA2030AH
TDA2030AV
TYPICAL APPLICATION
March 1995
1/15
TDA2030A
PIN CONNECTION (Top view)
TEST CIRCUIT
THERMAL DATA
Symbol
Rth (j-case)
2/15
Parameter
Thermal Resistance Junction-case
Max
Value
Unit
3
°C/W
TDA2030A
ABSOLUTE MAXIMUM RATINGS
Symbol
Parameter
Value
Unit
± 22
V
Vs
Supply Voltage
Vi
Input Voltage
Vi
Differential Input Voltage
± 15
V
Io
Peak Output Current (internally limited)
3.5
A
Ptot
Total Power Dissipation at Tcase = 90 °C
Tstg, Tj
Vs
Storage and Junction Temperature
20
W
– 40 to + 150
°C
ELECTRICAL CHARACTERISTICS
(Refer to the test circuit, VS = ± 16V, Tamb = 25oC unless otherwise specified)
Symbol
Parameter
Vs
Supply Voltage
Id
Quiescent Drain Current
Ib
Input Bias Current
Vos
Input Offset Voltage
Ios
Input Offset Current
PO
Output Power
Test Conditions
Power Bandwidth
SR
Slew Rate
Typ.
±6
Max.
Unit
± 22
V
50
80
mA
VS = ± 22V
0.2
2
µA
VS = ± 22V
±2
± 20
mV
± 20
± 200
nA
d = 0.5%, Gv = 26dB
f = 40 to 15000Hz
VS = ± 19V
BW
Min.
Po = 15W
W
RL = 4Ω
RL = 8Ω
RL = 8Ω
15
10
13
RL = 4Ω
18
12
16
100
kHz
8
V/µsec
Gv
Open Loop Voltage Gain
f = 1kHz
Gv
Closed Loop Voltage Gain
f = 1kHz
80
d
Total Harmonic Distortion
RL = 4Ω
Po = 0.1 to 14W
f = 40 to 15 000Hz
f = 1kHz
Po = 0.1 to 9W, f = 40 to 15 000Hz
RL = 8Ω
0.08
0.03
%
%
0.5
%
25.5
26
dB
26.5
dB
d2
Second Order CCIF Intermodulation
Distortion
PO = 4W, f2 – f1 = 1kHz, RL = 4Ω
0.03
%
d3
Third Order CCIF Intermodulation
Distortion
f1 = 14kHz, f2 = 15kHz
2f1 – f2 = 13kHz
0.08
%
eN
Input Noise Voltage
B = Curve A
B = 22Hz to 22kHz
2
3
10
µV
µV
B = Curve A
B = 22Hz to 22kHz
50
80
200
pA
pA
R L = 4Ω, Rg = 10kΩ, B = Curve A
PO = 15W
PO = 1W
106
94
dB
dB
iN
S/N
Input Noise Current
Signal to Noise Ratio
Ri
Input Resistance (pin 1)
(open loop) f = 1kHz
SVR
Supply Voltage Rejection
R L = 4Ω, Rg = 22kΩ
Gv = 26dB, f = 100 Hz
Tj
Thermal Shut-down Junction
Temperature
0.5
5
MΩ
54
dB
145
°C
3/15
TDA2030A
Figure 1 : Single Supply Amplifier
Figure 2 :
Open Loop-frequency Response
Figure 4 : Total Harmonic Distortion versus
Output Power (test using rise filters)
4/15
Figure 3 : Output Power versus Supply Voltage
Figure 5 : Two Tone CCIF Intremodulation
Distortion
TDA2030A
Figure 6 :
Large Signal Frequency Response
Figure 7 : Maximum Allowable Power Dissipation
versus Ambient Temperature
Figure 8 :
Output Power versus Supply Voltage
Figure 9 : Total Harmonic Distortion versus
Output Power
Figure 10 : Output Power versus Input Level
Figure 11 : Power Dissipation versus Output
Power
5/15
TDA2030A
Figure 12 : Single Supply High Power Amplifier (TDA2030A + BD907/BD908)
Figure 13 : P.C. Board and Component Layout for the Circuit of Figure 12 (1:1 scale)
6/15
TDA2030A
TYPICAL PERFORMANCE OF THE CIRCUIT OF FIGURE 12
Symbol
Parameter
Test Conditions
Vs
Supply Voltage
Id
Quiescent Drain Current
Vs = 36V
Po
Output Power
d = 0.5%, RL = 4Ω, f = 40 z to 15Hz
Vs = 39V
Vs = 36V
d = 10%, RL = 4Ω, f = 1kHz
Vs = 39V
Vs = 36V
Gv
Voltage Gain
SR
Slew Rate
d
f = 1kHz
S/N
Typ.
Max.
36
44
19.5
Unit
V
50
mA
35
28
W
W
44
35
W
W
20
20.5
dB
8
V/µsec
f = 1kHz
f = 40Hz to 15kHz
0.02
0.05
%
%
Input Sensitivity
Gv = 20dB, f = 1kHz, Po = 20W, RL = 4Ω
890
mV
Signal to Noise Ratio
RL = 4Ω, Rg = 10kΩ, B = Curve A
Po = 25W
Po = 4W
108
100
Total Harmonic Distortion
Po = 20W
Vi
Min.
dB
Figure 14 : Typical Amplifier with Spilt Power Supply
Figure 15 : P.C. Board and Component Layout for the Circuit of Figure 14 (1:1 scale)
7/15
TDA2030A
Figure 16 : Bridge Amplifier with Split Power Supply (PO = 34W, VS = ± 16V)
Figure 17 : P.C. Board and Component Layout for the Circuit of Figure 16 (1:1 scale)
MULTIWAY SPEAKER SYSTEMS AND ACTIVE
BOXES
Multiway loudspeaker systems provide the best
possible acoustic performance since each loudspeaker is specially designed and optimized to
handle a limited range of frequencies. Commonly,
these loudspeaker systems divide the audio spectrum into two or three bands.
To maintain aflat frequencyresponseover the Hi-Fi
audio range the bands covered by each loudspeaker must overlap slightly. Imbalance between
the loudspeakers produces unacceptable results
8/15
therefore it is important to ensure that each unit
generates the correct amount of acoustic energy
for its segmento of the audio spectrum. In this
respect it is also important to know the energy
distribution of the music spectrum to determine the
cutoff frequencies of the crossover filters (see Figure 18). As an example a 100W three-way system
with crossover frequencies of 400Hz and 3kHz
would require 50W for the woofer, 35W for the
midrange unit and 15W for the tweeter.
TDA2030A
Figure 18 : Power Distribution versus Frequency
A more effective solution, named ”Active Power
Filter” by SGS-THOMSON is shown in Figure 19.
Figure 19 : Active Power Filter
Both active and passive filters can be used for
crossovers but today active filters cost significantly
less than a good passive filter using air cored
inductors and non-electrolytic capacitors. In addition, active filters do not suffer from the typical
defects of passive filters:
- power less
- increased impedance seen by the loudspeaker
(lower damping)
- difficulty of precise design due to variable loudspeaker impedance.
Obviously, active crossovers can only be used if a
power amplifier is provided for each drive unit. This
makes it particularly interesting and economically
sound to use monolithic power amplifiers.
In some applications, complex filters are not really
necessary and simple RC low-pass and high-pass
networks (6dB/octave) can be recommended.
The result obtained are excellent because this is
the best type of audio filter and the only one free
from phase and transient distortion.
The rather poor out of band attenuation of single
RC filters means that the loudspeaker must operate linearly well beyond the crossover frequency to
avoid distortion.
The proposed circuit can realize combined power
amplifiers and 12dB/octave or 18dB/octave highpass or low-pass filters.
In practice, at the input pins of the amplifier two
equal and in-phase voltages are available, as required for the active filter operation.
The impedance at the pin (-) is of the order of 100Ω,
while that of the pin (+) is very high, which is also
what was wanted.
The component values calculated for fc = 900Hz
using a Bessek 3rd order Sallen and Key structure
are :
C 1 = C2 = C3
R1
R2
R3
22nF
8.2kΩ
5.6kΩ
33kΩ
Using this type of crossover filter, a complete 3-way
60W active loudspeaker system is shown in Figure 20.
It employs 2nd order Buttherworth filters with the
crossover frequencies equal to 300Hz and 3kHz.
The midrange section consists of two filters, a high
pass circuit followed by a low pass network. With
VS = 36V the output power delivered to the woofer
is 25W at d = 0.06% (30W at d = 0.5%).
The power delivered to the midrange and the
tweeter can be optimized in the design phase
taking in account the loudspeaker efficiency and
impedance (RL = 4Ω to 8Ω).
It is quite common that midrange and tweeter
speakers have an efficiency 3dB higher thanwoofers.
9/15
TDA2030A
Figure 20 : 3 Way 60W Active Loudspeaker System (VS = 36V)
10/15
TDA2030A
MUSICAL INSTRUMENTS AMPLIFIERS
Another important field of application for active
systems is music.
In this area the use of several medium power
amplifiers is more convenient than a single high
power amplifier, and it is also more realiable.
A typical example (see Figure 21) consist of four
amplifiers each driving a low-cost, 12 inch loudspeaker. This application can supply 80 to
160WRMS.
can be used down to the values as low as 0.002%
in high power amplifiers.
Figure 22 : Overshoot Phenomenon in Feedback
Amplifiers
Figure 21 : High Power Active Box
for Musical Instrument
TRANSIENT INTERMODULATION DISTORTION (TIM)
Transient intermodulation distortion is an unfortunate phenomen associated with negative-feedback amplifiers. When a feedback amplifier
receives an input signal which rises very steeply,
i.e. contains high-frequencycomponents, the feedback can arrive too late so that the amplifiers
overloads and a burst of intermodulation distortion
will be produced as in Figure 22. Since transients
occur frequently in music this obviously a problem
for the designer of audio amplifiers. Unfortunately,
heavy negative feedback is frequency used to reduce the total harmonic distortion of an amplifier,
which tends to aggravate the transient intermodulation (TIM situation. The best known method for
the measurement of TIM consists of feeding sine
waves superimposed onto square waves, into the
amplifier under test. The output spectrum is then
examined using a spectrum analyser and compared to the input. This method suffers from serious
disadvantages : the accuracy is limited, the measurement is a rather delicate operation and an expensive spectrum analyser is essential. A new
approach (see Technical Note 143) applied by
SGS-THOMSON to monolithic amplifiers measurement is fast cheap-it requires nothing more sophisticated than an oscilloscope - and sensitive - and it
The ”inverting-sawtooh” method of measurement
is based on the response of an amplifier to a 20kHz
sawtooth waveform. The amplifier has no difficulty
following the slow ramp but it cannot follow the fast
edge. The output will follow the upper line in Figure 23 cutting of the shaded area and thus increasing the mean level. If this output signal is filtered to
remove the sawtooth, direct voltage remains which
indicates the amount of TIM distortion, although it
is difficult to measure because it is indistinguishable from the DC offset of the amplifier. This problem is neatly avoided in the IS-TIM method by
periodically inverting the sawtooth waveform at a
low audio frequency as shown in Figure 24.
Figure 23 : 20kHz Sawtooth Waveform
Figure 24 : Inverting Sawtooth Waveform
11/15
TDA2030A
In the case of the sawtooth in Figure 25 the mean
level was increased by the TIM distortion, for a
sawtooth in the other direction the opposite is true.
The result is an AC signal at the output whole
peak-to-peak value is the TIM voltage, which can
be measured easily with an oscilloscope. If the
peak-to-peak value of the signal and the peak-topeak of the inverting sawtooth are measured, the
TIM can be found very simply from:
VOUT
⋅ 100
TIM =
Vsawtooth
In Figure 25 the experimental results are shown for
the 30W amplifier using the TDA2030A as a driver
and a low-cost complementary pair. A simple RC
filter on the input of the amplifier to limit the maximum signal slope (SS) is an effective way to reduce
TIM.
Figure 26 : TIM Design Diagram (fC = 30kHz)
POWER SUPPLY
Figure 25 : TIM Distortion versus Output Power
Using monolithic audio amplifier with non-regulated
supply voltage it is important to design the power
supply correctly. In any working case it must provide a supply voltage less than the maximum value
fixed by the IC break-down voltage.
It is essential to take into account all the working
conditions,in particular mains fluctuationsand supply voltage variations with and without load. The
TDA2030A(VS max = 44V) is particularly suitable for
substitution of the standard IC power amplifiers
(with VS max = 36V) for more reliable applications.
An example, using a simple full-wave rectifier followed by a capacitor filter, is shown in the table 1
and in the diagram of Figure 27.
Figure 27 : DC Characteristics of
50W Non-regulated Supply
The diagram of Figure 26 originated by SGSTHOMSON can be used to find the Slew-Rate (SR)
required for a given output power or voltage and a
TIM design target.
For example if an anti-TIM filter with a cutoff at
30kHz is used and the max. peak-to-peak output
voltage is 20V then, referring to the diagram, a
Slew-Rate of 6V/µs is necessary for 0.1% TIM.
As shown Slew-Rates of above 10V/µs do not
contribute to a further reduction in TIM.
Slew-Rates of 100/µs are not only useless but also
a disadvantage in Hi-Fi audio amplifiers because
they tend to turn the amplifier into a radio receiver.
12/15
TDA2030A
APPLICATION SUGGESTION
The recommended values of the components are
those shown on application circuit of Figure 14.
Different values can be used. The Table 2 can help
the designer.
Table 1
Mains
(220V)
Secondary
Voltage
+ 20%
DC Outpu t Voltage (Vo)
Io = 0
Io = 0.1A
Io = 1A
28.8V
43.2V
42V
37.5V
+ 15%
27.6V
41.4V
40.3V
35.8V
+ 10%
26.4V
39.6V
38.5V
34.2V
–
24V
36.2V
35V
31V
– 10%
21.6V
32.4V
31.5V
27.8V
– 15%
20.4V
30.6V
29.8V
26V
– 20%
19.2V
28.8V
28V
24.3V
SHORT CIRCUIT PROTECTION
The TDA2030A has an original circuit which limits
the current of the output transistors. This function
can be considered as being peak power limiting
rather than simple current limiting. It reduces the
possibility that the device gets damaged during an
accidental short circuit from AC output to ground.
A regulated supply is not usually used for the power
output stages because of its dimensioning must be
done taking into account the power to supply in the
signal peaks. They are only a small percentage of
the total music signal, with consequently large
overdimensioning of the circuit.
Even if with a regulated supply higher output power
can be obtained (VS is constant in all working conditions), the additional cost and power dissipation do
not usually justify its use. Using non-regulated supplies, there are fewer designe restriction. In fact,
when signal peaks are present, the capacitor filter
acts as a flywheel supplying the required energy.
In average conditions, the continuous power supplied is lower. The music power/continuous power
ratio is greater in this case than for the case of
regulated supplied, with space saving and cost
reduction.
THERMAL SHUT-DOWN
The presence of a thermal limiting circuit offers the
following advantages:
1. An overload on the output (even if it is
permanent), or an above limit ambient
temperature can be easily supported since the
Tj cannot be higher than 150oC.
2. The heatsink can have a smaller factor of
safety compared with that of a conventional
circuit. There is no possibility of device damage
due to high junction temperature. If for any
reason, the junction temperature increases up
to 150oC, the thermal shut-down simply
reduces the power dissipation and the current
consumption.
Table 2
Comp.
Recom.
Value
Purpose
Larger than
Recommended Value
Smaller than
Recommended Value
R1
22kΩ
Closed loop gain setting
Increase of gain
R2
680Ω
Closed loop gain setting
Decrease of gain (*)
Increase of gain
R3
22kΩ
Non inverting input biasing
Increase of input impedance
Decrease of input impedance
R4
1Ω
Frequency Stability
R5
≅ 3 R2
Danger of oscillation at high
frequencies with inductive
loads
Poor High Frequencies
Attenuation
C1
1µF
Input DC Decoupling
C2
22µF
Inverting DC Decoupling
Upper Frequency Cut-off
Decrease of gain
Danger of Oscillation
Increase of low frequencies
cut-off
Increase of low frequencies
cut-off
Danger of Oscillation
C3, C4
0.1µF
Supply Voltage Bypass
C5, C6
100µF
Supply Voltage Bypass
Danger of Oscillation
C7
Frequency Stability
Larger Bandwidth
C8
0.22µF
1
≈
2 πBR1
D1, D2
1N4001
To protect the device against output voltage spikes
Upper Frequency Cut-off
Smaller Bandwidth
Larger Bandwidth
(*) The value of closed loop gain must be higher than 24dB.
13/15
TDA2030A
PENTAWATT PACKAGE MECHANICAL DATA
DIM.
mm
TYP.
MIN.
A
C
D
D1
E
F
F1
G
G1
H2
H3
L
L1
L2
L3
L5
L6
L7
M
M1
Dia
MAX.
4.8
1.37
2.8
1.35
0.55
1.05
1.4
2.4
1.2
0.35
0.8
1
3.4
6.8
10.4
10.4
10.05
MIN.
inch
TYP.
0.094
0.047
0.014
0.031
0.039
0.126
0.260
0.134
0.268
MAX.
0.189
0.054
0.110
0.053
0.022
0.041
0.055
0.142
0.276
0.409
0.409
0.396
17.85
15.75
21.4
22.5
0.703
0.620
0.843
0.886
2.6
15.1
6
3
15.8
6.6
0.102
0.594
0.236
0.118
0.622
0.260
4.5
4
0.177
0.157
3.65
3.85
0.144
0.152
E
L
D1
C
D
M
A
M1
L1
L2
L5
L7
L6
14/15
F
H2
F1
Dia.
G1
G
H3
L3
TDA2030A
Information furnished is believed to be accurate and reliable. However, SGS-THOMSON Microelectronics assumes no responsibility for the
consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No
license is granted by implication or otherwise under any patent or patent rights of SGS-THOMSON Microelectronics. Specifications mentioned
in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied.
SGS-THOMSON Microelectronics products are not authorized for use as critical components inlife support devices or systems without express
written approval of SGS-THOMSON Microelectronics.
 1995 SGS-THOMSON Microelectronics - All Rights Reserved
PENTAWATT is a Registered Trademark of SGS-THOMSON Microelectronics
SGS-THOMSON Microelectronics GROUP OF COMPANIES
Australia - Brazil - France - Germany - Hong Kong - Italy - Japan - Korea - Malaysia - Malta - Morocco - The Netherlands - Singapore Spain - Sweden - Switzerland - Taiwan - Thaliand - United Kingdom - U.S.A.
15/15
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