MAX1858A, MAX1875A, MAX1876A

MAX1858A, MAX1875A, MAX1876A
19-2966; Rev 0; 10/03
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
Features
The MAX1858A/MAX1875A/MAX1876A dual, synchronized, step-down controllers generate two outputs from
input supplies ranging from 4.5V to 23V. Each output is
adjustable from sub-1V to 18V and supports loads of 10A
or higher. Input voltage ripple and total RMS input ripple
current are reduced by synchronized 180° out-of-phase
operation.
♦ 4.5V to 23V Input Supply Range
The switching frequency is adjustable from 100kHz to
600kHz with an external resistor. Alternatively, the controller can be synchronized to an external clock generated by another MAX1858A/MAX1875A/MAX1876A or a
system clock. One MAX1858A/MAX1875A/MAX1876A
can be set to generate an in-phase, or 90° out-ofphase, clock signal for synchronization with additional
controllers. This allows two controllers to operate either
as an interleaved two- or four-phase system with each
output shifted by 90°. The MAX1858A/MAX1875A/
MAX1876A feature soft-start. The MAX1858A also features first-on/last-off power sequencing and soft-stop.
The MAX1858A/MAX1875A/MAX1876A eliminate the
need for current-sense resistors by utilizing the low-side
MOSFET’s on-resistance as a current-sense element.
This protects the DC-DC components from damage during output-overload conditions or output short-circuit
faults without requiring a current-sense resistor.
Adjustable foldback current limit reduces power dissipation during short-circuit conditions. The MAX1858A/
MAX1876A include a power-on reset (POR) output to signal the system when both outputs reach regulation.
♦ Optional Synchronization
The MAX1858A/MAX1875A/MAX1876A ensure that the
output voltage does not swing negative when the input
power is removed or when EN is driven low. The
MAX1875A/MAX1876A also allow prebias startup without discharging the output.
The MAX1858A/MAX1875A/MAX1876A are available in a
24-pin QSOP package. Use the MAX1875 evaluation kit
or the MAX1858 evaluation kit to evaluate the
MAX1858A/MAX1875A/MAX1876A.
Applications
Network Power Supplies
Telecom Power Supplies
DSP, ASIC, and FPGA Power Supplies
Set-Top Boxes
Broadband Routers
Servers
♦ 0 to 18V Output Voltage Range (Up to 10A)
♦ Adjustable Lossless Foldback Current Limit
♦ Adjustable 100kHz to 600kHz Switching
Frequency
♦ Clock Output for Master/Slave Synchronization
♦ 4 x 90° Out-of-Phase Step-Down Converters
(Using Two Controllers, Figure 7)
♦ Prebias Startup (MAX1875A/MAX1876A)
♦ Power Sequencing (MAX1858A)
♦ RST Output with 140ms Minimum Delay
(MAX1858A/MAX1876A)
♦ Fixed-Frequency Pulse-Width Modulation (PWM)
Operation
Ordering Information
PART
TEMP RANGE
PIN-PACKAGE
MAX1858AEEG
-40°C to +85°C
24 QSOP
MAX1875AEEG
-40°C to +85°C
24 QSOP
MAX1876AEEG
-40°C to +85°C
24 QSOP
Pin Configuration
TOP VIEW
COMP2 1
24 EN
FB2 2
23 DH2
ILIM2 3
22 LX2
OSC 4
21 BST2
V+ 5
REF 6
GND 7
20 DL2
MAX1858A
MAX1875A
MAX1876A
19 VL
18 PGND
CKO 8
17 DL1
SYNC 9
16 BST1
ILIM1 10
15 LX1
FB1 11
14 DH1
COMP1 12
13 RST (N.C.)
Desknote Computers
QSOP
() ARE FOR THE MAX1875A ONLY
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1858A/MAX1875A/MAX1876A
General Description
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
ABSOLUTE MAXIMUM RATINGS
V+ to GND ..............................................................-0.3V to +25V
PGND to GND .......................................................-0.3V to +0.3V
VL to GND ..................-0.3V to the lower of +6V and (V+ + 0.3V)
BST1, BST2 to GND ...............................................-0.3V to +30V
LX1 to BST1..............................................................-6V to +0.3V
LX2 to BST2..............................................................-6V to +0.3V
DH1 to LX1 ..............................................-0.3V to (VBST1 + 0.3V)
DH2 to LX2 ..............................................-0.3V to (VBST2 + 0.3V)
DL1, DL2 to PGND........................................-0.3V to (VL + 0.3V)
CKO, REF, OSC, ILIM1, ILIM2,
COMP1, COMP2 to GND ..........................-0.3V to (VL + 0.3V)
FB1, FB2, RST, SYNC, EN to GND...........................-0.3V to +6V
VL to GND Short Circuit .............................................Continuous
REF to GND Short Circuit ...........................................Continuous
Continuous Power Dissipation (TA = +70°C)
24-Pin QSOP (derate 9.4mW/°C above +70°C)...........762mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V+ = 12V, EN = ILIM_ = VL, SYNC = GND, IVL = 0mA, PGND = GND, CREF = 0.22µF, CVL = 4.7µF (ceramic), ROSC = 60kΩ,
compensation components for COMP_ are from Figure 1, TA = -40°C to +85°C (Note 1), unless otherwise noted.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
GENERAL
V+ Operating Range
(Note 2)
4.5
23.0
VL = V+ (Note 2)
4.5
5.5
V
V+ Operating Supply Current
VL unloaded, no MOSFETs connected
3.5
6
mA
V+ Standby Supply Current
EN = LX_ = FB_ = 0V
0.3
0.6
mA
Thermal Shutdown
Rising temperature, typical hysteresis = 10°C
Current-Limit Threshold
PGND - LX_
VL REGULATOR
Output Voltage
ILIM_ = VL
75
100
RILIM_ = 100kΩ
32
50
62
RILIM_ = 600kΩ
225
300
375
5.5V < V+ < 23V, 1mA < ILOAD < 50mA
4.75
5
5.25
V
4.1
4.2
4.3
V
VL Undervoltage Lockout Rising
Trip Level
VL Undervoltage Lockout
Hysteresis
°C
+160
(Note 3)
125
100
mV
mV
REFERENCE
Output Voltage
IREF = 0µA
Reference Load Regulation
0µA < IREF < 50µA
1.98
2.00
2.02
V
0
4
10
mV
SOFT-START
Digital Ramp Period
Internal 6-bit DAC for one converter to ramp from 0V to
full scale (Note 4)
Soft-Start Steps
1024
DC-DC
clocks
64
Steps
FREQUENCY
Low End of Range
ROSC = 60kΩ
High End of Range
ROSC = 10kΩ
DH_ Minimum Off-Time
ROSC = 10kΩ
2
0°C to +85°C
84
100
115
-40°C to +85°C
80
100
120
540
600
660
kHz
250
303
ns
_______________________________________________________________________________________
kHz
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
(V+ = 12V, EN = ILIM_ = VL, SYNC = GND, IVL = 0mA, PGND = GND, CREF = 0.22µF, CVL = 4.7µF (ceramic), ROSC = 60kΩ,
compensation components for COMP_ are from Figure 1, TA = -40°C to +85°C (Note 1), unless otherwise noted.)
PARAMETER
CONDITIONS
MIN
SYNC Range
Switching frequency must be set to half of the SYNC
frequency
200
SYNC Input Pulse Width
(Note 4)
High
100
Low
100
SYNC Rise/Fall Time
(Note 4)
TYP
MAX
UNITS
1200
kHz
ns
100
ns
250
nA
ERROR AMPLIFIER
FB_ Input Bias Current
FB_ Input Voltage Set Point
FB_ to COMP_ Transconductance
0°C to +85°C
0.985
1.00
1.015
-40°C to +85°C
0.98
1.00
1.02
0°C to +85°C
1.25
1.8
2.70
-40°C to +85°C
1.2
1.8
2.9
V
mS
DRIVERS
DL_, DH_ Break-Before-Make Time CLOAD = 5nF
DH_ On-Resistance
DL_ On-Resistance
30
ns
Low
1.5
2.5
High
3
5
Low
0.6
1.5
High
3
5
Ω
Ω
LOGIC INPUTS (EN, SYNC)
Input Low Level
Typical 15% hysteresis, VL = 4.5V
Input High Level
VL = 5.5V
2.4
0.8
Input High/Low Bias Current
VEN = 0 or 5.5V
-1
V
V
+0.1
+1
µA
0.4
V
LOGIC OUTPUTS (CKO)
Output Low Level
VL = 5V, sinking 5mA
Output High Level
VL = 5V, sourcing 5mA
4.0
V
COMP_
Pulldown Resistance During
Shutdown and Current Limit
Ω
17
RST OUTPUT (MAX1858A/MAX1876A ONLY)
Output-Voltage Trip Level
Output Low Level
Both FBs must be over this to allow the reset timer to
start; there is no hysteresis
0.87
0.9
0.93
VL = 5V, sinking 3.2mA
0.4
VL = 1V, sinking 0.4mA
0.3
Output Leakage
V+ = VL = 5V, V RST = 5.5V, VFB = 1V
Reset Timeout Period
VFB_ = 1V
FB_ to Reset Delay
FB_ overdrive from 1V to 0.85V
140
315
4
V
V
1
µA
560
ms
µs
Note 1: Specifications to -40°C are guaranteed by design and not production tested.
Note 2: Operating supply range is guaranteed by VL line regulation test. Connect V+ to VL for 5V operation.
Note 3: When VL falls and UVLO is tripped, the device is latched and VL must be discharged below 2.5V before normal operation
can resume.
Note 4: Guaranteed by design and not production tested.
_______________________________________________________________________________________
3
MAX1858A/MAX1875A/MAX1876A
ELECTRICAL CHARACTERISTICS (continued)
Typical Operating Characteristics
(Circuit of Figure 1, VIN = 12V, TA = +25°C, unless otherwise noted.)
60
50
40
30
20
10
0.8
0.6
0.4
0.2
OUT2
0
-0.2
OUT1
-0.4
0.5
MAX1858A/75A/76A toc03
OUT1
70
1.0
MAX1858A/75A/76A toc02
80
EFFICIENCY (%)
MAX1858A/75A/76A toc01
OUT2
OUTPUT VOLTAGE ACCURACY (%)
100
90
VL VOLTAGE ACCURACY
vs. LOAD CURRENT
OUTPUT VOLTAGE ACCURACY vs. LOAD
0
VL VOLTAGE ACCURACY
EFFICIENCY vs. LOAD
-0.5
-1.0
-1.5
-0.6
-0.8
0
-2.0
-1.0
0.1
1
100
10
0
5
10
15
LOAD (A)
SWITCHING FREQUENCY vs. ROSC
LOAD TRANSIENT RESPONSE (OUTPUT 1)
400
50
100
MAX1858A/75A/76A toc06
VOUT2
50mV/div
AC-COUPLED
VOUT1
50mV/div
AC-COUPLED
VOUT1
50mV/div
AC-COUPLED
VOUT2
50mV/div
AC-COUPLED
10A
IOUT1
10A
IOUT2
0A
0A
300
200
100
0
0
10
20
30
40
50
10µs/div
60
10µs/div
ROSC (kΩ)
SOFT-START AND SOFT-STOP WAVEFORM
(MAX1858A ONLY)
SOFT-START AND SOFT-STOP WAVEFORM
(MAX1858A ONLY)
MAX1858A/75A/76A toc08
MAX1858A/75A/76A toc07
10V
EN
0V
EN PULLED HIGH BEFORE VOUT1 REACHES 0V.
VOUT1
1V/div
IOUT1 = 300mA
0V
2ms/div
5V
EN
0V
VOUT1
1V/div
IOUT1 = 300mA
0V
VOUT2
1V/div
IOUT2 = 300mA
0V
VOUT2
1V/div
IOUT2 = 300mA
0V
4
150
LOAD TRANSIENT RESPONSE (OUTPUT 2)
MAX1858A/75A/76A toc05
MAX1858A/75A/76A toc04
500
0
LOAD CURRENT (mA)
LOAD (A)
600
SWITCHING FREQUENCY (kHz)
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
2ms/div
_______________________________________________________________________________________
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
START AND STOP WAVEFORM
(MAX1875A/MAX1876A ONLY)
RESET TIMEOUT
(MAX1858A/MAX1876A ONLY)
INPUT POWER REMOVAL
MAX1858A/75A/76A toc09
MAX1858A/75A/76A toc10
10V
EN
0V
MAX1858A/75A/76A toc11
VIN
10V/div
0V
VOUT1
1V/div
IOUT1 = 300mA
0V
VOUT1
1V/div
IOUT1 = 300mA
0V
VOUT2
1V/div
IOUT2 = 300mA
VOUT2
1V/div
IOUT2 = 300mA
0V
0V
EN
0
VOUT2
0V
VOUT1
0V
VRST
0V
PREBIAS STARTUP
2ms/div
5ms/div
OUT-OF-PHASE WAVEFORM
EXTERNALLY SYNCHRONIZED
SWITCHING WAVEFORM
100ms/div, 5V/div
MAX1858A/75A/76A toc13
MAX1858A/75A/76A toc12
VOUT1
20mV/div
CKO OUTPUT WAVEFORM
MAX1858A/75A/76A toc14
5V
VSYNC
0V
SYNC = GND
12V
VLX1
5V
VCK0
0V
5V
VCK0
0V
0V
12V
VLX2
10V
VLX1
0V
10V
VLX1
0V
0V
VOUT1
10mV/div
AC-COUPLED
VOUT1
10mV/div
AC-COUPLED
VOUT2
20mV/div
400ns/div
1µs/div
400ns/div
SHORT-CIRCUIT CURRENT FOLDBACK
AND RECOVERY
CKO OUTPUT WAVEFORM
MAX1858A/75A/76A toc15
MAX1858A/75A/76A toc16
SYNC = VL
IOUT1 = 10A (5A/div)
VOUT1 = 1.8V (1V/div)
5V
VCK0
0V
10V
VLX1
0V
SHORT
VOUT2 = 2.5V (1V/div)
VOUT2
IOUT2 = 10A (5A/div)
VOUT1
10mV/div
400ns/div
4ms/div
_______________________________________________________________________________________
5
MAX1858A/MAX1875A/MAX1876A
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 12V, TA = +25°C, unless otherwise noted.)
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
MAX1858A/MAX1875A/MAX1876A
Pin Description
PIN
1
2
3
4
FUNCTION
Compensation Pin for Regulator 2 (REG2). Compensate REG2’s control loop by connecting a series resistor
COMP2 (RCOMP2) and capacitor (CCOMP2A) to GND in parallel with a second compensation capacitor (CCOMP2B) as
shown in Figure 1.
FB2
Feedback Input for Regulator 2 (REG2). Connect FB2 to a resistive divider between REG2’s output and GND to
adjust the output voltage between 1V and 18V. To set the output voltage below 1V, connect FB2 to a resistive
voltage-divider from REF to REG2’s output. See the Setting the Output Voltage section.
ILIM2
Current-Limit Adjustment for Regulator 2 (REG2). The PGND–LX2 current-limit threshold defaults to 100mV if
ILIM2 is connected to VL. Connect a resistor (RILIM2) from ILIM2 to GND to adjust the REG2’s current-limit
threshold (VITH2) from 50mV (RILIM2 = 100kΩ) to 300mV (RILIM2 = 600kΩ). See the Setting the Valley Current
Limit section.
OSC
Oscillator Frequency Set Input. Connect a resistor from OSC to GND (ROSC) to set the switching frequency from
100kHz (ROSC = 60kΩ) to 600kHz (ROSC = 10kΩ). The controller still requires ROSC when an external clock is
connected to SYNC. When using an external clock, select ROSC as described above, and set the external clock
frequency to twice the desired switching frequency.
5
V+
Input Supply Voltage. 4.5V to 23V.
6
REF
2V Reference Output. Bypass to GND with a 0.22µF or greater ceramic capacitor.
7
GND
Analog Ground
CKO
Clock Output. Clock output for external 2- or 4-phase synchronization (see the Clock Synchronization (SYNC,
CKO) section).
9
SYNC
Synchronization Input or Clock Output Selection Input. SYNC has three operating modes. Connect SYNC to a
200kHz to 1200kHz clock for external synchronization. Connect SYNC to GND for 2-phase operation as a master
controller. Connect SYNC to VL for 4-phase operation as a master controller (see the Clock Synchronization
(SYNC, CKO) section).
10
ILIM1
Current-Limit Adjustment for Regulator 1 (REG1). The PGND–LX1 current-limit threshold defaults to 100mV if
ILIM1 is connected to VL. Connect a resistor (RILIM1) from ILIM1 to GND to adjust REG1’s current-limit threshold
(VITH1) from 50mV (RILIM1 = 100kΩ) to 300mV (RILIM1 = 600kΩ). See the Setting the Valley Current Limit section.
11
FB1
8
12
13
6
NAME
Feedback Input for Regulator 1 (REG1). Connect FB1 to a resistive divider between REG1’s output and GND to
adjust the output voltage between 1V and 18V. To set the output voltage below 1V, connect FB1 to a resistive
voltage-divider from REF and REG1’s output. See the Setting the Output Voltage section.
Compensation Pin for Regulator 1 (REG1). Compensate REG1’s control loop by connecting a series resistor
COMP1 (RCOMP1) and capacitor (CCOMP1A) to GND in parallel with a second compensation capacitor (CCOMP1B) as
shown in Figure 1.
RST
Open-Drain Reset Output (MAX1858A/MAX1876A Only). RST is low when either output voltage is more than 10%
below its regulation point. After soft-start is completed and both outputs exceed 90% of their nominal output
voltage (VFB_ > 0.9V), RST becomes high impedance after a 140ms delay and remains high impedance as long
as both outputs maintain regulation. Connect a resistor between RST and the logic supply for logic-level
voltages.
N.C.
Connect to GND or leave unconnected for the MAX1875A.
_______________________________________________________________________________________
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
PIN
NAME
FUNCTION
14
DH1
High-Side Gate-Driver Output for Regulator 1 (REG1). DH1 swings from LX1 to BST1. DH1 is low during UVLO.
15
LX1
External Inductor Connection for Regulator 1 (REG1). Connect LX1 to the switched side of the inductor. LX1
serves as the lower supply rail for the DH1 high-side gate driver.
16
BST1
Boost Flying-Capacitor Connection for Regulator 1 (REG1). Connect BST1 to an external ceramic capacitor and
diode according to Figure 1.
17
DL1
Low-Side Gate-Driver Output for Regulator 1 (REG1). DL1 swings from PGND to VL. DL1 is low during UVLO.
18
PGND
19
VL
20
DL2
Low-Side Gate-Driver Output for Regulator 2 (REG2). DL2 swings from PGND to VL. DL2 is low during UVLO.
21
BST2
Boost Flying-Capacitor Connection for Regulator 2 (REG2). Connect BST2 to an external ceramic capacitor and
diode according to Figure 1.
22
LX2
23
DH2
24
EN
Power Ground
Internal 5V Linear-Regulator Output. Supplies the regulators and powers the low-side gate drivers and external
boost circuitry for the high-side gate drivers.
External Inductor Connection for Regulator 2 (REG2). Connect LX2 to the switched side of the inductor. LX2
serves as the lower supply rail for the DH2 high-side gate driver.
High-Side Gate-Driver Output for Regulator 2 (REG2). DH2 swings from LX2 to BST2. DH2 is low during UVLO.
Active-High Enable Input. A logic low shuts down both controllers. Connect to VL for always-on operation.
Detailed Description
DC-DC PWM Controller
The MAX1858A/MAX1875A/MAX1876A step-down converters use a PWM voltage-mode control scheme (Figure
2) for each out-of-phase controller. The controller generates the clock signal by dividing down the internal oscillator or SYNC input when driven by an external clock, so
each controller’s switching frequency equals half the
oscillator frequency (fSW = fOSC/2). An internal transconductance error amplifier produces an integrated error
voltage at the COMP pin, providing high DC accuracy.
The voltage at COMP sets the duty cycle using a PWM
comparator and a ramp generator. At each rising edge
of the clock, REG1’s high-side N-channel MOSFET turns
on and remains on until either the appropriate duty cycle
or until the maximum duty cycle is reached. REG2 operates out-of-phase, so the second high-side MOSFET
turns on at each falling edge of the clock. During each
high-side MOSFET’s on-time, the associated inductor
current ramps up.
During the second-half of the switching cycle, the highside MOSFET turns off and the low-side N-channel
MOSFET turns on. Now the inductor releases the stored
energy as its current ramps down, providing current to
the output. Under overload conditions, when the induc-
tor current exceeds the selected valley current limit (see
the Current-Limit Circuit (ILIM_) section), the high-side
MOSFET does not turn on at the appropriate clock edge
and the low-side MOSFET remains on to let the inductor
current ramp down.
Synchronized Out-of-Phase Operation
The two independent regulators in the MAX1858A/
MAX1875A/MAX1876A operate 180° out-of-phase to
reduce input filtering requirements, reduce electromagnetic interference (EMI), and improve efficiency. This
effectively lowers component cost and saves board
space, making the MAX1858A/MAX1875A/MAX1876A
ideal for cost-sensitive applications.
Dual-switching regulators typically operate both
controllers in-phase, and turn on both high-side
MOSFETs at the same time. The input capacitor must
then support the instantaneous current requirements of
both controllers simultaneously, resulting in increased
ripple voltage and current when compared to a single
switching regulator. The higher RMS ripple current
lowers efficiency due to power loss associated with the
input capacitor’s effective series resistance (ESR). This
typically requires more low-ESR input capacitors in
parallel to minimize input voltage ripple and ESR-related
losses, or to meet the necessary ripple-current rating.
_______________________________________________________________________________________
7
MAX1858A/MAX1875A/MAX1876A
Pin Description (continued)
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
VIN
6V - 23V
CMPSH-3A
RV+
4.7Ω
0.1µF
V+
CV+
0.22µF
VL
4.7Ω
CIN1
2 × 10µF
BST1
COUT1
4 × 220µF
NH1*
L1
1.1µH
R1A
8.06kΩ
**
NL1*
CIN2
2 × 10µF
BST2
CBST2
0.1µF
CBST1
0.1µF
OUTPUT1
VOUT = 1.8V
CVL
4.7µF
4.7Ω
DH1
DH2
LX1
LX2
DL1
DL2
NH2*
L2
1.1µH
**
NL2*
R2A
15kΩ
OUTPUT2
VOUT = 2.5V
COUT2
4 × 220µF
PGND
FB1
FB2
RCOMP1
5.9kΩ
R1B
10kΩ
RCOMP2
8.2kΩ
COMP1
CCOMP1A
0.01µF
CCOMP1B
100pF
D2
CMSSH-3
R2B
10kΩ
COMP2
CCOMP2B
100pF
10kΩ
OSC
MAX1858A
MAX1875A
MAX1876A
REF
CCOMP2A
6800pF
D3
CMSSH-3
CREF
0.22µF
GND
CLOCK OUTPUT
CKO
RESET OUTPUT
RST (MAX1858A/
MAX1876A ONLY)
ILIM1
EN
ILIM2
SYNC
VL
118kΩ
96.5kΩ
ON
140kΩ
OFF
84.5kΩ
*IRF7811W
**OPTIONAL
Figure 1. Standard 600kHz Application Circuit
With dual, synchronized, out-of-phase operation, the
MAX1858A/MAX1875A/MAX1876As’ high-side MOSFETs
turn on 180° out-of-phase. The instantaneous input current peaks of both regulators no longer overlap, resulting
in reduced RMS ripple current and input voltage ripple.
This reduces the required input capacitor ripple-current
rating, allowing fewer or less expensive capacitors, and
reduces shielding requirements for EMI. The Out-ofPhase Waveforms in the Typical Operating Characteristics demonstrate synchronized 180° out-of-phase
operation.
8
Internal 5V Linear Regulator (VL)
All MAX1858A/MAX1875A/MAX1876A functions are
internally powered from an on-chip, low-dropout 5V
regulator. The maximum regulator input voltage (V+) is
23V. Bypass the regulator’s output (VL) with a 4.7µF
ceramic capacitor to PGND. The VL dropout voltage is
typically 500mV, so when V+ is greater than 5.5V, VL is
typically 5V. The MAX1858A/MAX1875A/MAX1876A
also employs an undervoltage lockout circuit that disables both regulators when V L falls below 4.2V. V L
should also be bypassed to GND with a 0.1µF capacitor. When VL falls and UVLO is tripped, the device is
latched and VL must be discharged below 2.5V before
normal operation can resume.
_______________________________________________________________________________________
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
MAX1858A/MAX1875A/MAX1876A
REF
V+
5V LINEAR
REGULATOR
MAX1858A
MAX1875A
MAX1876A
VREF
2.0V
GND
VL
COMP1
BST1
FB1
CONVERTER 1
DH1
SOFT-START DAC
(SEQUENCING—
MAX1858A ONLY)
LX1
R
Q
S
Q
DL1
PGND
1VP-P
OSC
SYNC
OSCILLATOR
CK0
5µA
RST
(MAX1858A/
MAX1876A ONLY)
ILIM1
RESET
EN
VREF
UVLO
AND
SHUTDOWN
VL - 0.5V
VL
BST2
DH2
CONVERTER 2
LX2
DL2
COMP2
FB2
ILIM2
Figure 2. Functional Diagram
The internal VL linear regulator can source over 50mA to
supply the IC, power the low-side gate driver, charge the
external boost capacitor, and supply small external
loads. When driving large FETs, little or no regulator current may be available for external loads.
For example, when switched at 600kHz, a single large
FET with 18nC total gate charge requires 18nC ✕ 600kHz
= 11mA. To drive larger MOSFETs, or deliver larger
loads, connect VL to an external power supply from 4.5V
to 5.5V.
_______________________________________________________________________________________
9
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
A
UVLO
B
C
D
F
E
G
H
J
I
VL
EN
VOUT_
SS_
DH_
DL_
M
N
O
MAX1875A/MAX1876A POWER-ON-OFF SEQUENCING DEFINITIONS
SYMBOL
UVLO
VL
EN
VOUT_
SS_
DH_
DL_
A
B
C
D
E
F
G
H
I
J
K
L
M
N
O
DEFINITION
Undervoltage lockout trip level is provided in the Electrical Characteristics table.
Internal 5V Linear-Regulator Output
Active-High Enable Input
Output Voltage
Internal Soft-Start Input Signal into Error Amplifier
High-Side Gate-Driver Output
Low-Side Gate-Driver Output
VL rising while below the UVLO threshold. EN is low.
VL is greater than the UVLO threshold. EN is low.
EN is pulled high.
Normal operation
VL enters UVLO.
VL exits UVLO.
Resumes normal operation
EN is pulled low.
EN is pulled high.
Resumes normal operation
VL drops below UVLO threshold while EN is high.
Resumes normal operation
UVLO is activated and DL_ is latched low.
Exiting UVLO: DL_ remains latched low until the first fall of DH_ is detected.
DL_ is low after EN is pulled low.
Figure 3. MAX1875A/MAX1876A Detailed Power-On-Off Sequencing
10
______________________________________________________________________________________
K
L
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
A
B
C
D
E
F
G
H
J
I
K
L
MAX1858A/MAX1875A/MAX1876A
UVLO
M
VL
EN
VOUT1
SS1
VOUT2
SS2
DH1
DL1
DH2
DL2
N
O
P
MAX1858A POWER-ON-OFF SEQUENCING DEFINITIONS
SYMBOL
UVLO
VL
EN
VOUT1
SS1
VOUT2
SS2
DH1
DL1
DH2
DL2
A
B
C
DEFINITION
Undervoltage threshold value is provided in the
Electrical Characteristics table.
Internal 5V Linear-Regulator Output
Active-High Enable Input
Regulator 1 Output Voltage
Regulator 1: Internal Soft-Start Input Signal into Error Amplifier
Regulator 2 Output Voltage
Regulator 2: Internal Soft-Start Input Signal into Error Amplifier
Regulator 1: High-Side Gate-Driver Output
Regulator 1: Low-Side Gate-Driver Output
Regulator 2: High-Side Gate-Driver Output
Regulator 2: Low-Side Gate-Driver Output
VL rising while below the UVLO threshold. EN is low.
VL is greater than the UVLO threshold. EN is low.
EN is pulled high. DH1 and DL1 start switching. DH2 and
DL2 are off.
SYMBOL
D
E
F
G
H
I
J
K
L
M
N
DEFINITION
Normal operation
VL enters UVLO.
VL exits UVLO.
Resumes normal operation. DH1 and DL1 start switching.
DH2 and DL2 are off.
EN is pulled low and then high.
VOUT1 must reach 0V before restarting due to the cycling
of the enable in region H (above).
VOUT1 recovers.
VOUT2 recovers.
VL enters UVLO before VOUT2 fully recovers.
VL exits UVLO.
UVLO latches DL_ low.
O
Exiting UVLO: DL_ remains latched low until the first fall
of DH_ is detected.
P
DL_ is high after EN is pulled low and soft-stop is complete.
Figure 4. MAX1858A Detailed Power-On-Off Sequencing
______________________________________________________________________________________
11
High-Side Gate-Drive Supply (BST_)
Gate-drive voltages for the high-side N-channel switches are generated by the flying-capacitor boost circuits
(Figure 5). A boost capacitor (connected from BST_ to
LX_) provides power to the high-side MOSFET driver.
On startup, the synchronous rectifier (low-side MOSFET)
forces LX_ to ground and charges the boost capacitor to
5V. On the second half-cycle, after the low-side MOSFET
turns off, the high-side MOSFET is turned on by closing
an internal switch between BST_ and DH_. This provides
the necessary gate-to-source voltage to turn on the highside switch, an action that boosts the 5V gate-drive
signal above VIN. The current required to drive the highside MOSFET gates (fSWITCH ✕ QG) is ultimately drawn
from VL.
MOSFET Gate Drivers (DH_, DL_)
The DH and DL drivers are optimized for driving moderate-size N-channel high-side and larger low-side power
MOSFETs. This is consistent with the low duty factor
seen with large VIN - VOUT differential. The DL_ low-side
drive waveform is always the complement of the DH_
high-side drive waveform (with controlled dead time to
prevent cross-conduction or “shoot-through”). An adaptive dead-time circuit monitors the DL_ output and prevents the high-side FET from turning on until DL_ is fully
off. There must be a low-resistance, low-inductance path
from the DL_ driver to the MOSFET gate in order for the
adaptive dead-time circuit to work properly. Otherwise,
the sense circuitry in the MAX1858A/MAX1875A/
MAX1876A interprets the MOSFET gate as “off” while
there is actually charge still left on the gate. Use very
short, wide traces (50mils to 100mils wide if the MOSFET
is 1in from the device). The dead time at the DH-off edge
is determined by a fixed 30ns internal delay.
Current-Limit Circuit (ILIM_)
The current-limit circuit employs a “valley” current-sensing algorithm that uses the on-resistance of the low-side
MOSFET as a current-sensing element. If the currentsense signal is above the current-limit threshold, the
MAX1858A/MAX1875A/MAX1876A do not initiate a new
cycle (Figure 6). Since valley current sensing is
employed, the actual peak current is greater than the
current-limit threshold by an amount equal to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability are a function of
the low-side MOSFET’s on-resistance, current-limit
threshold, inductor value, and input voltage. The reward
for this uncertainty is robust, lossless overcurrent sensing that does not require costly sense resistors.
INPUT
(VIN)
VL
BST_
Synchronous rectification reduces conduction losses in
the rectifier by replacing the normal low-side Schottky
catch diode with a low-resistance MOSFET switch.
Additionally, the MAX1858A/MAX1875A/MAX1876A use
the synchronous rectifier to ensure proper startup of the
boost gate-driver circuit and to provide the current-limit
signal.
The internal pulldown transistor that drives DL_ low is
robust, with a 0.5Ω (typ) on-resistance. This low onresistance helps prevent DL_ from being pulled up during the fast rise time of the LX_ node, due to capacitive
coupling from the drain to the gate of the low-side synchronous-rectifier MOSFET. However, for high-current
applications, some combinations of high- and low-side
FETs can cause excessive gate-drain coupling, leading
to poor efficiency, EMI, and shoot-through currents.
This can be remedied by adding a resistor (typically
less than 5Ω) in series with BST_, which increases the
turn-on time of the high-side FET without degrading the
turn-off time (Figure 5).
-IPEAK
4.7Ω
ILOAD
DH_
LX_
MAX1875A
Figure 5. Reducing the Switching-Node Rise Time
12
INDUCTOR CURRENT
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
ILIMIT
0
TIME
Figure 6. “Valley” Current-Limit Threshold Point
______________________________________________________________________________________
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
Undervoltage Lockout and Startup
IF VL drops below 4.2V, the MAX1858A/MAX1875A/
MAX1876A assume that the input supply and reference
voltages are too low to make valid decisions and activate
the undervoltage lockout (UVLO) circuitry, which latches
DL and DH low to inhibit switching. RST is also forced
low during UVLO. To reset the latch and be ready for the
next VL rise, VL must be pulled below 2.5V.
In addition, to ensure proper startup, the value of the
capacitor at REF to GND must meet the following condition:
CREF > ((8.29 x 10-4) / V+_SLOPE) - (1.97 x 10-1 / fS_MAX)
where V+_SLOPE is the actual input-voltage rise time’s
slew rate.
For example, if the switching frequency is set at
600kHz nominal, which is 660kHz (max), and the inputvoltage rise time’s slew rate is 1.6V/mS, then C REF
should be greater than 0.22µF. Make sure CREF is chosen large enough to cover for worst-case capacitance
tolerances and temperature coefficient.
Enable (EN), Soft-Start, and Soft-Stop
Pull EN high to enable or low to shut down both regulators. See the timing diagrams, Figures 3 and 4, for
more detail.
Output-Voltage Sequencing
After the startup circuitry enables the controller, the
MAX1858A begins the startup sequence. Regulator 1
(OUT1) powers up with soft-start enabled. Once the first
converter’s soft-start sequence ends, regulator 2 (OUT2)
powers up with soft-start enabled. Finally, when both converters complete soft-start and both output voltages
exceed 90% of their nominal values, the reset output
(RST) goes high (see the Reset Output section). Soft-stop
is initiated by pulling EN low. Soft-stop occurs in reverse
order of soft-start, allowing last-on/first-off operation.
Reset Output (RST) (MAX1858A/
MAX1876A Only)
RST is an open-drain output. RST pulls low when either
output falls below 90% of its nominal regulation voltage.
Once both outputs exceed 90% of their nominal regulation
voltages and both soft-start cycles are completed, RST
goes high impedance. To obtain a logic-voltage output,
connect a pullup resistor from RST to the logic supply voltage. A 100kΩ resistor works well for most applications. If
unused, leave RST grounded or unconnected.
Clock Synchronization (SYNC, CKO)
SYNC serves two functions: SYNC selects the clock output (CKO) type used to synchronize slave controllers, or it
serves as a clock input so the MAX1858A/MAX1875A/
MAX1876A can be synchronized with an external clock
signal. This allows the MAX1858A/MAX1875A/MAX1876A
to function as either a master or slave. CKO provides a
clock signal synchronized to the MAX1858A/MAX1875A/
MAX1876As’ switching frequency, allowing either inphase (SYNC = GND) or 90° out-of-phase (SYNC = VL)
synchronization of additional DC-DC controllers (Figure 7).
The MAX1858A/MAX1875A/MAX1876A support the following three operating modes:
• SYNC = GND: The CKO output frequency equals
REG1’s switching frequency (fCKO = fDH1) and the
CKO signal is in phase with REG1’s switching frequency. This provides 2-phase operation when synchronized with a second slave controller.
• SYNC = VL: The CKO output frequency equals two
times REG1’s switching frequency (fCKO = 2fDH1)
and the CKO signal is phase shifted by 90° with
respect to REG1’s switching frequency. This provides 4-phase operation when synchronized with a
second MAX1858A/MAX1875A/MAX1876A (slave
controller).
______________________________________________________________________________________
13
MAX1858A/MAX1875A/MAX1876A
The adjustable current limit accommodates MOSFETs
with a wide range of on-resistance characteristics (see
the Design Procedure section). The current-limit threshold is adjusted with an external resistor at ILIM_ (Figure
1). The adjustment range is from 50mV to 300mV, corresponding to resistor values of 100kΩ to 600kΩ. In
adjustable mode, the current-limit threshold across the
low-side MOSFET is precisely 1/10th the voltage seen
at ILIM_. However, the current-limit threshold defaults
to 100mV when ILIM is tied to VL. The logic threshold
for switchover to this 100mV default value is approximately VL - 0.5V.
Adjustable foldback current limit reduces power dissipation during short-circuit conditions (see the Design
Procedure section).
Carefully observe the PC board layout guidelines to
ensure that noise and DC errors do not corrupt the current-sense signals seen by LX_ and PGND. The IC
must be mounted close to the low-side MOSFET with
short, direct traces making a Kelvin-sense connection
so that trace resistance does not add to the intended
sense resistance of the low-side MOSFET.
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
• SYNC Driven by External Oscillator: The controller
generates the clock signal by dividing down the
SYNC input signal, so that the switching frequency
equals half the synchronization frequency (fSW =
fSYNC/2). REG1’s conversion cycles initiate on the rising edge of the internal clock signal. The CKO output
frequency and phase match REG1’s switching frequency (f CKO = f DH1 ) and the CKO signal is in
phase. Note that the MAX1858A/MAX1875A/
MAX1876A still require ROSC when SYNC is externally clocked and the internal oscillator frequency should
be set to 50% of the synchronization frequency (fSW
= 0.5 fSYNC).
Thermal Overload Protection
Thermal overload protection limits total power dissipation
in the MAX1858A/MAX1875A/MAX1876A. When the
device’s die-junction temperature exceeds TJ = +160°C,
an on-chip thermal sensor shuts down the device, forcing
DL_ and DH_ low, allowing the IC to cool. The thermal
sensor turns the part on again after the junction temperature cools by 10°C. During thermal shutdown, the regulators shut down, RST goes low, and soft-start is reset. If
the VL linear-regulator output is short circuited, thermaloverload protection is triggered.
Design Procedure
Effective Input Voltage Range
Although the MAX1858A/MAX1875A/MAX1876A controllers can operate from input supplies ranging from
4.5V to 23V, the input voltage range can be effectively
limited by the MAX1858A/MAX1875A/MAX1876As’
duty-cycle limitations. The maximum input voltage is
limited by the minimum on-time (tON(MIN)):
VIN(MAX) ≤
⎡V
+ VDROP1 ⎤
VIN(MIN) = ⎢ OUT
⎥ + VDROP2 - VDROP1
⎢⎣ 1- fSW t OFF(MIN) ⎥⎦
where VDROP1 is the sum of the parasitic voltage drops
in the inductor discharge path, including synchronous
rectifier, inductor, and PC board resistances. VDROP2 is
the sum of the resistances in the charging path, including high-side switch, inductor, and PC board resistances.
MAX1858A
MAX1875A
MAX1876A
MAX1858A
MAX1875A
MAX1876A
CK0
SYNC
CK0
OSC
SYNC
SYNC
VL
VL
MASTER
SLAVE
180° PHASE SHIFT
OSC
SLAVE
4-OUTPUT APPLICATION
90° PHASE SHIFT
DH1
MASTER
SYNC
MASTER
3-OUTPUT APPLICATION
DH1
MASTER
DH2
SLAVE
t ON(MIN)fSW
where tON(MIN) is 100ns. The minimum input voltage is
limited by the switching frequency and minimum offtime, which determine the maximum duty cycle
(DMAX = 1 - fSWtOFF(MIN)):
MAX1858A
MAX1875A
MAX1876A
OSC
VOUT
DH
DH2
DH1
SLAVE
DH2
Figure 7. Synchronized Controllers
14
______________________________________________________________________________________
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
⎡⎛ V
⎞ ⎤
R _ A = R _ B ⎢⎜ OUT ⎟ -1⎥
⎢⎣⎝ VSET ⎠ ⎥⎦
where VSET = 1V (see the Electrical Characteristics)
and VOUT can range from VSET to 18V.
For output voltages below 1V, set the MAX1858A/
MAX1875A/MAX1876A output voltage by connecting a
voltage-divider from the output to FB_ to REF (Figure
8). Select R_C (FB to REF resistor) in the 1kΩ to 10kΩ
range. Calculate R_A with the following equation:
⎛V
⎞
V
R _ A = R _ C ⎜ SET- OUT ⎟
V
V
⎝ REF- SET ⎠
where V SET = 1V, V REF = 2V (see the Electrical
Characteristics), and VOUT can range from 0 to VSET.
Setting the Switching Frequency
The controller generates the clock signal by dividing
down the internal oscillator or SYNC input signal when
driven by an external oscillator, so the switching frequency equals half the oscillator frequency (fSW = fOSC/2).
The internal oscillator frequency is set by a resistor
(ROSC) connected from OSC to GND. The relationship
between fSW and ROSC is:
6 × 109 (Ω - Hz)
ROSC =
fSW
where fSW is in Hz and ROSC is in Ω. For example, a
600kHz switching frequency is set with ROSC = 10kΩ.
Higher frequencies allow designs with lower inductor
values and less output capacitance. Consequently,
peak currents and I 2 R losses are lower at higher
switching frequencies, but core losses, gate-charge
currents, and switching losses increase.
A rising clock edge on SYNC is interpreted as a synchronization input. If the SYNC signal is lost, the internal oscillator takes control of the switching rate,
returning the switching frequency to that set by ROSC.
This maintains output regulation even with intermittent
SYNC signals. When an external synchronization signal
is used, ROSC should set the switching frequency to
one-half SYNC rate (fSYNC).
OUT_
REF
R_A
R_C
FB_
FB_
MAX1858A
MAX1875A
MAX1876A
R_B
MAX1858A
MAX1875A
MAX1876A
R_A
OUT_
VOUT_ < 1V
VOUT_ > 1V
Figure 8. Adjustable Output Voltage
Inductor Selection
Three key inductor parameters must be specified for
operation with the MAX1858A/MAX1875A/MAX1876A:
inductance value (L), peak-inductor current (IPEAK), and
DC resistance (RDC). The following equation assumes a
constant ratio of inductor peak-to-peak AC current to DC
average current (LIR). For LIR values too high, the RMS
currents are high, and therefore I2R losses are high.
Large inductances must be used to achieve very low LIR
values. Typically, inductance is proportional to resistance (for a given package type), which again makes I2R
losses high for very low LIR values. A good compromise
between size and loss is a 30% peak-to-peak ripple current to average-current ratio (LIR = 0.3). The switching
frequency, input voltage, output voltage, and selected
LIR determine the inductor value as follows:
V
(V - V
)
L = OUT IN OUT
VINfSWIOUTLIR
where VIN, VOUT, and IOUT are typical values (so that
efficiency is optimum for typical conditions). The switching frequency is set by R OSC (see the Setting the
Switching Frequency section). The exact inductor value
is not critical and can be adjusted in order to make
trade-offs among size, cost, and efficiency. Lower
inductor values minimize size and cost, but also
improve transient response and reduce efficiency due
to higher peak currents. On the other hand, higher
inductance increases efficiency by reducing the RMS
current. However, resistive losses due to extra wire turns
can exceed the benefit gained from lower AC current
levels, especially when the inductance is increased
without also allowing larger inductor dimensions.
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. The
______________________________________________________________________________________
15
MAX1858A/MAX1875A/MAX1876A
Setting the Output Voltage
For 1V or greater output voltages, set the MAX1858A/
MAX1875A/MAX1876A output voltage by connecting a
voltage-divider from the output to FB_ to GND (Figure
8). Select R_B (FB_ to GND resistor) to between 1kΩ
and 10kΩ. Calculate R_A (OUT_ to FB_ resistor) with
the following equation:
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
inductor’s saturation rating must exceed the peakinductor current at the maximum defined load current
(ILOAD(MAX)):
⎛ LIR ⎞
IPEAK = ILOAD(MAX ) + ⎜
⎟I
⎝ 2 ⎠ LOAD(MAX )
RFBI =
PFB × VOUT
5 × 10-6 (1- PFB )
and
RILIM =
10 × VITH (1- PFB ) × RFBI
V
[ OUT -10 × VITH(1- PFB )]
Setting the Valley Current Limit
The minimum current-limit threshold must be high
enough to support the maximum expected load current
with the worst-case low-side MOSFET on-resistance
value since the low-side MOSFET’s on-resistance is
used as the current-sense element. The inductor’s valley
current occurs at ILOAD(MAX) minus half of the ripple
current. The current-sense threshold voltage (V ITH)
should be greater than voltage on the low-side MOSFET
during the ripple-current valley:
⎛ LIR ⎞
VITH > RDS(ONMAX
⎟
,
) × ILOAD(MAX) × ⎜1⎝
2 ⎠
where R DS(ON) is the on-resistance of the low-side
MOSFET (N L). Use the maximum value for R DS(ON)
from the low-side MOSFET’s data sheet, and additional
margin to account for RDS(ON) rise with temperature is
also recommended. A good general rule is to allow
0.5% additional resistance for each °C of the MOSFET
junction temperature rise.
Connect ILIM_ to VL for the default 100mV (typ) current-limit threshold. For an adjustable threshold, connect a resistor (R ILIM _) from ILIM_ to GND. The
relationship between the current-limit threshold (VITH_)
and RILIM_ is:
RILIM _ =
VITH _
0.5µA
where RILIM_ is in Ω and VITH_ is in V.
An RILIM resistance range of 100kΩ to 600kΩ corresponds to a current-limit threshold of 50mV to 300mV.
When adjusting the current limit, 1% tolerance resistors
minimize error in the current-limit threshold.
For foldback current limit, a resistor (RFBI) is added
from ILIM pin to output. The value of RILIM and RFBI
can then be calculated as follows:
First select the percentage of foldback, PFB, from 15%
to 30%, then:
16
If RILIM_ results in a negative number, select a low-side
MOSFET with lower RDS(ON) or increase PFB_ or a combination of both for the best compromise of cost, efficiency, and lower power dissipation during short circuit.
Input Capacitor
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor must meet the ripple current
requirement (IRMS) imposed by the switching currents
as defined by the following equation:
IRMS = ILOAD
VOUT (VIN - VOUT )
VIN
IRMS has a maximum value when the input voltage equals
twice the output voltage (VIN = 2VOUT), so IRMS(MAX) =
ILOAD / 2. For most applications, nontantalum capacitors
(ceramic, aluminum, polymer, or OS-CON) are preferred
at the input due to their robustness with high inrush currents typical of systems that can be powered from very
low impedance sources. Additionally, two (or more)
smaller-value low-ESR capacitors can be connected in
parallel for lower cost. Choose an input capacitor that
exhibits less than +10°C temperature rise at the RMS
input current for optimal long-term reliability.
Output Capacitor
The key selection parameters for the output capacitor
are capacitance value, ESR, and voltage rating. These
parameters affect the overall stability, output ripple voltage, and transient response. The output ripple has two
components: variations in the charge stored in the output capacitor, and the voltage drop across the capacitor’s ESR caused by the current flowing into and out of
the capacitor:
VRIPPLE ≅ VRIPPLE(ESR) + VRIPPLE(C)
______________________________________________________________________________________
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
VRIPPLE(ESR) = IP-PRESR
VRIPPLE(C) =
To determine the loop gain (AL), consider the gain from
FB to COMP (ACOMP/FB), from COMP to LX (ALX/COMP),
and from LX to FB (AFB/LX). The total loop gain is:
AL = A COMP / FB × ALX / COMP × AFB / LX
IP-P
8COUT fSW
where:
⎛ V -V
⎞⎛ V
⎞
IP-P = ⎜ IN OUT ⎟ ⎜ OUT ⎟
⎝ fSWL ⎠ ⎝ VIN ⎠
where IP-P is the peak-to-peak inductor current (see the
Inductor Selection section). These equations are suitable
for initial capacitor selection, but final values should be
verified by testing in a prototype or evaluation circuit.
As a general rule, a smaller inductor ripple current results
in less output ripple voltage. Since inductor ripple current
depends on the inductor value and input voltage, the output ripple voltage decreases with larger inductance and
increases with higher input voltages. However, the inductor ripple current also impacts transient-response performance, especially at low VIN - VOUT differentials. Low
inductor values allow the inductor current to slew faster,
replenishing charge removed from the output filter capacitors by a sudden load step. The amount of output-voltage sag is also a function of the maximum duty factor,
which can be calculated from the minimum off-time and
switching frequency:
gM _ COMP
V
×
ACOMP / FB = COMP ≅
VFB
SCCOMP
1 + sRCOMPCCOMP _ A
1 + sRCOMPCCOMP _ B
assuming an ideal integrator, and assuming that
CCOMP_B is much less than CCOMP_A:
ALX / COMP =
VLX
VIN
=
VCOMP VRAMP
where VRAMP = 1VP-P:
1+ sRESRCOUT
V
V
AFB / LX = FB = SET 2
VLX VOUT S LCOUT + SRESRCOUT + 1
1+ SRESRCOUT
V
≅ SET
VOUT VOUT S2LCOUT + 1
Therefore:
⎡⎛ V
⎤
⎞
L(ILOAD1 - ILOAD2 )2 ⎢⎜ OUT ⎟ + t OFF(MIN) ⎥
⎢⎣⎝ VINfSW ⎠
⎥⎦
VSAG =
⎡⎛ VIN - VOUT ⎞
⎤
2COUT VOUT ⎢⎜
⎟ - t OFF(MIN) ⎥
⎢⎣⎝ VINfSW ⎠
⎥⎦
where t OFF(MIN) is the minimum off-time (see the
Electrical Characteristics), and fSW is set by ROSC (see
the Setting the Switching Frequency section).
Compensation
Each voltage-mode controller section employs a
transconductance error amplifier whose output is the
compensation point of the control loop. The control loop
is shown in Figure 9. For frequencies much lower than
Nyquist, the PWM block can be simplified to a voltage
amplifier. Connect RCOMP_ and CCOMP_A from COMP
to GND to compensate the loop (Figure 9). The inductor,
output capacitor, compensation resistor, and compensation capacitors determine the loop stability. Since the
inductor and output capacitor are chosen based on performance, size, and cost, select the compensation resistor and capacitors to optimize control-loop stability.
AL ≅
gM _ COMP
SCCOMP _ A
×
1 + SRCOMPCCOMP _ A
1 + SRCOMPCCOMP _ B
×
VIN
VRAMP
V
1 + SRESR COUT
× SET ×
VOUT
S2 LC
+1
OUT
For an ideal integrator, this loop gain approaches infinity
at DC. In reality the gM amplifier has a finite output
impedance, which imposes a finite, but large, loop gain.
It is this large loop gain that provides DC load accuracy.
The dominant pole occurs due to the integrator, and for
this analysis, it can be approximated to occur at DC.
RCOMP creates a zero at:
fZ _ COMP _ A =
1
2π × RCOMP _ CCOMP _ A
The inductor and capacitor form a double pole at:
fLC =
1
2π × LCOUT
______________________________________________________________________________________
17
MAX1858A/MAX1875A/MAX1876A
The output voltage ripple as a consequence of the ESR
and output capacitance is:
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
At some higher frequency, the output capacitor’s
impedance becomes insignificant compared to its ESR,
and the LC system becomes more like an LR system,
turning a double pole into a single pole. This zero
occurs at:
fESR =
1
The following loop-gain equation can be found by using
these previous approximations with Figure 9:
AL ≅
Setting the loop gain to 1 and solving for the crossover
frequency yields:
2π × RESR COUT
A final pole is added using CCOMP_B to reduce the
gain and attenuate noise after crossover. This pole
(fCOMP_B) occurs at:
fCOMP _ B =
V
× SET
VRAMP VOUT
gM _ COMP × RCOMP × RESR
×
2π × L
fCO = GBW =
1
2π × RCOMPCCOMP _ B
Figure 10 shows a Bode plot of the poles and zeros in
their relative locations.
Near crossover, the following approximations can be
made to simplify the loop-gain equation:
• RCOMP has much higher impedance than CCOMP.
This is true if, and only if, crossover occurs above
fZ_COMP_A. If this is true, CCOMP_A can be ignored
(as a short to ground).
• RESR is much higher impedance than COUT. This is
true if, and only if, crossover occurs well after the output capacitor’s ESR zero. If this is true, C OUT
becomes an insignificant part of the loop gain and can
be ignored (as a short to ground).
• CCOMP_B is much higher impedance than RCOMP
and can be ignored (as an open circuit). This is true
if, and only if, crossover occurs far below fCOMP_B.
P
DH
DL
To ensure stability, select RCOMP to meet the following
criteria:
• Unity-gain crossover must occur below 1/5th of the
switching frequency.
• For reasonable phase margin using type 1 compensation, fCO must be larger than 5 ✕ fESR.
Choose CCOMP_A so that fZ_COMP_A equals half fLC
using the following equation:
CCOMP _ A =
CCOMP _ B =
1
(
)
2π × 3 × fCO × RCOMP
L
VOUT
LX
N
FB
LX
=
COUT
RESR
FB
COMP_
gM_COMP
gM_COMP
VSET
RCOMP_
RCOMP
Choose CCOMP_B so that fCOMP_B occurs at 3 times
fCO using the following equation:
RESR
COMP_
2 × LCOUT
GAIN = +VIN/VRAMP
L
M
VIN
N
W
VC
gM _ COMP × RCOMP × RESR
VIN
V
× SET ×
VRAMP VOUT
sL
CCOMP_A
CCOMP_A
VSET
RCOMP_
CCOMP_B
CCOMP_B
Figure 9. Fixed-Frequency Voltage-Mode Control Loop
18
______________________________________________________________________________________
COUT
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
All four N-channel MOSFETs must be a logic-level type
with guaranteed on-resistance specifications at VGS ≥
4.5V. For maximum efficiency, choose a high-side
MOSFET (NH_) that has conduction losses equal to the
switching losses at the optimum input voltage. Check to
ensure that the conduction losses at minimum input
voltage do not exceed MOSFET package thermal limits,
or violate the overall thermal budget. Also, check to
ensure that the conduction losses plus switching losses
at the maximum input voltage do not exceed package
ratings or violate the overall thermal budget.
Ensure that the MAX1858A/MAX1875A/MAX1876A DL_
gate drivers can drive NL _. In particular, check that the
dv/dt caused by NH _ turning on does not pull up the NL _
gate through NL _’s drain-to-gate capacitance. This is the
most frequent cause of cross-conduction problems.
Gate-charge losses are dissipated by the driver and do
not heat the MOSFET. All MOSFETs must be selected
so that their total gate charge is low enough that VL can
power all four drivers without overheating the IC:
PVL = VIN × QG _ TOTAL × fSW
MOSFET package power dissipation often becomes a
dominant design factor. I2R power losses are the greatest heat contributor for both high-side and low-side
MOSFETs. I2R losses are distributed between NH_ and
NL _ according to duty factor as shown in the equations
below. Switching losses affect only the high-side
MOSFET, since the low-side MOSFET is a zero-voltage
switched device when used in the buck topology.
Calculate MOSFET temperature rise according to package thermal-resistance specifications to ensure that
both MOSFETs are within their maximum junction temperature at high ambient temperature. The worst-case
dissipation for the high-side MOSFET (PNH) occurs at
both extremes of input voltage, and the worst-case dissipation for the low-side MOSFET (PNL) occurs at maximum input voltage.
50
fLC
40
30
fESR
20
GAIN (dB)
• Total gate charge (Qg)
• Reverse transfer capacitance (CRSS)
• Power dissipation
BODE PLOT FOR VOLTAGEMODE CONTROLLERS
10
fZ-COMP_A
fSWITCH
fCO
0
-10
-20
fCOMP_B
-30
-40
0.001
0.01
0.1
1
FREQUENCY (MHz)
Figure 10. Voltage-Mode Loop Analysis
⎛ Q + QGD ⎞
PNH(SWITCHING) = VINILOADfSW ⎜ GS
⎟
⎝ IGATE ⎠
IGATE is the average DH driver-output current capability
determined by:
IGATE =
VL
(
2 RDS(ON)DH + RGATE + RGMOSFET
)
where RDS(ON)DH is the high-side MOSFET driver’s onresistance (5Ω max), RGATE is any series resistance
between DH and BST (Figure 5), and RGMOSFET is the
internal gate resistance of the external MOSFET:
⎛V
⎞
PNH(CONDUCTION) = ILOAD2RDS(ON)NH ⎜ OUT ⎟
⎝ VIN ⎠
PNH(TOTAL) = PNH(SWITCHING) + PNH(CONDUCTION)
⎛ ⎛V
⎞⎞
PNL = ILOAD2RDS(ON)NL ⎜1- ⎜ OUT ⎟ ⎟
⎝ ⎝ VIN ⎠ ⎠
where PNH(CONDUCTION) is the conduction power loss
in the high-side MOSFET, and PNL is the total low-side
power loss.
To reduce EMI caused by switching noise, add a 0.1µF
ceramic capacitor from the high-side switch drain to
the low-side switch source or add resistors in series
with DL_ and DH_ to increase the MOSFETs’ turn-on
and turn-off times.
______________________________________________________________________________________
19
MAX1858A/MAX1875A/MAX1876A
MOSFET Selection
The MAX1858A/MAX1875A/MAX1876As’ step-down
controller drives two external logic-level N-channel
MOSFETs as the circuit switch elements. The key
selection parameters are:
• On-resistance (RDS(ON))
• Maximum drain-to-source voltage (VDS(MAX))
• Minimum threshold voltage (VTH(MIN))
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
Applications Information
Dropout Performance
When working with low input voltages, the output-voltage adjustable range for continuous-conduction operation is restricted by the minimum off-time (tOFF(MIN)).
For best dropout performance, use the lowest (100kHz)
switching-frequency setting. Manufacturing tolerances
and internal propagation delays introduce an error to
the switching frequency and minimum off-time specifications. This error is more significant at higher frequencies. Also, keep in mind that transient response
performance of buck regulators operated close to
dropout is poor, and bulk output capacitance must
often be added (see the VSAG equation in the Design
Procedure section).
The absolute point of dropout is when the inductor current ramps down during the minimum off-time (∆IDOWN)
as much as it ramps up during the maximum on-time
(∆IUP). The ratio h = ∆IUP/∆IDOWN is an indicator of the
ability to slew the inductor current higher in response to
increased load, and must always be greater than 1. As
h approaches 1, the absolute minimum dropout point,
the inductor current cannot increase as much during
each switching cycle and V SAG greatly increases
unless additional output capacitance is used.
A reasonable minimum value for h is 1.5, but adjusting
this up or down allows tradeoffs between VSAG, output
capacitance, and minimum operating voltage. For a
given value of h, the minimum operating voltage can be
calculated as:
⎡V
+ VDROP1 ⎤
VIN(MIN) = ⎢ OUT
⎥ + VDROP2 - VDROP1
⎢⎣ 1- hfSW t OFF(MIN) ⎥⎦
where VDROP1 is the sum of the parasitic voltage drops
in the inductor discharge path, including synchronous
rectifier, inductor, and PC board resistances; VDROP2 is
the sum of the resistances in the charging path, including high-side switch, inductor, and PC board resistances; and t OFF(MIN) is from the Electrical
Characteristics. The absolute minimum input voltage is
calculated with h = 1.
If the calculated V+(MIN) is greater than the required
minimum input voltage, then reduce the operating frequency or add output capacitance to obtain an acceptable VSAG. If operation near dropout is anticipated,
calculate V SAG to be sure of adequate transient
response.
20
Dropout design example:
VOUT = 5V
fSW = 600kHz
tOFF(MIN) = 250ns
VDROP1 = VDROP2 = 100mV
h = 1.5
⎡
⎤
5V + 100mV
VIN(MIN) = ⎢
⎥
⎣ 1- 1.5(600kHz)(250ns) ⎦
+100mV − 100mV = 6.58V
Calculating again with h = 1 gives the absolute limit of
dropout:
⎡
⎤
5V + 100mV
VIN(MIN) = ⎢
⎥
(
)(
)
1
600
250
kHz
ns
⎣
⎦
+100mV − 100mV = 6V
Therefore, VIN must be greater than 6V, even with very
large output capacitance, and a practical input voltage
with reasonable output capacitance would be 6.58V.
Improving Noise Immunity
Applications where the MAX1858A/MAX1875A/
MAX1876A must operate in noisy environments can
typically adjust their controller’s compensation to
improve the system’s noise immunity. In particular,
high-frequency noise coupled into the feedback loop
causes jittery duty cycles. One solution is to lower the
crossover frequency (see the Compensation section).
PC Board Layout Guidelines
Careful PC board layout is critical to achieve low switching losses and clean, stable operation. This is especially
true for dual converters where one channel can affect
the other. Refer to the MAX1858 EV kit or MAX1875 EV
kit data sheet for specific layout examples.
If possible, mount all the power components on the top
side of the board with their ground terminals flush
against one another. Follow these guidelines for good
PC board layout:
• Isolate the power components on the top side from
the analog components on the bottom side with a
ground shield. Use a separate PGND plane under
the OUT1 and OUT2 sides (referred to as PGND1
and PGND2). Avoid the introduction of AC currents
into the PGND1 and PGND2 ground planes. Run the
power plane ground currents on the top side only.
______________________________________________________________________________________
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable, jitter-free operation.
• Connect GND and PGND together close to the IC.
Do not connect them together anywhere else.
Carefully follow the grounding instructions under
step 4 of the Layout Procedure section.
• Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PC boards (2oz vs. 1oz) to enhance
full-load efficiency by 1% or more.
• LX_ and PGND connections to the synchronous rectifiers for current limiting must be made using Kelvinsense connections to guarantee the current-limit
accuracy. With 8-pin SO MOSFETs, this is best done
by routing power to the MOSFETs from outside
using the top copper layer, while connecting PGND
and LX_ underneath the 8-pin SO package.
• When trade-offs in trace lengths must be made,
allow the inductor-charging path to be made longer
than the discharge path. Since the average input
current is lower than the average output current in
step-down converters, this minimizes the power dissipation and voltage drops caused by board resistance. For example, allow some extra distance
between the input capacitors and the high-side
MOSFET rather than to allow distance between the
inductor and the low-side MOSFET or between the
inductor and the output filter capacitor.
• Ensure that the feedback connection to COUT_ is
short and direct.
• Route high-speed switching nodes (BST_, LX_, DH_,
and DL_) away from the sensitive analog areas
(REF, COMP_, ILIM_, and FB_). Use PGND1 and
PGND2 as EMI shields to keep radiated noise away
from the IC, feedback dividers, and analog bypass
capacitors.
• Make all pin-strap control input connections (ILIM_,
SYNC, and EN) to analog ground (GND) rather than
power ground (PGND).
Layout Procedure
1) Place the power components first, with ground terminals adjacent (NL _ source, CIN_, and COUT_). Make
all these connections on the top layer with wide, copper-filled areas (2oz copper recommended).
2) Mount the controller IC adjacent to the synchronousrectifier MOSFETs (NL _), preferably on the back
side in order to keep LX_, PGND_, and DL_ traces
short and wide. The DL_ gate trace must be short
and wide, measuring 50mils to 100mils wide if the
low-side MOSFET is 1in from the controller IC.
3) Group the gate-drive components (BST_ diodes and
capacitors, and VL bypass capacitor) together near
the controller IC.
4) Make the DC-DC controller ground connections as
follows: create a small analog ground plane near the
IC. Connect this plane to GND and use this plane for
the ground connection for the reference (REF) V+
bypass capacitor, compensation components, feedback dividers, OSC resistor, and ILIM_ resistors (if
any). Connect GND and PGND together under the
IC (this is the only connection between GND and
PGND).
5) On the board’s top side (power planes), make a star
ground to minimize crosstalk between the two sides.
Chip Information
TRANSISTOR COUNT: 6688
PROCESS: BiCMOS
______________________________________________________________________________________
21
MAX1858A/MAX1875A/MAX1876A
• Use a star-ground connection on the power plane to
minimize the crosstalk between OUT1 and OUT2.
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information
go to www.maxim-ic.com/packages.)
QSOP.EPS
MAX1858A/MAX1875A/MAX1876A
Dual 180° Out-of-Phase Buck Controllers with
Sequencing/Prebias Startup and POR
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
21-0055
F
1
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
22 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2003 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
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