MAX796, MAX797, MAX799

MAX796, MAX797, MAX799
19-0221; Rev 4; 9/05
KIT
ATION
EVALU
E
L
B
AVAILA
Step-Down Controllers with
Synchronous Rectifier for CPU Power
The MAX796/MAX797/MAX799 high-performance, stepdown DC-DC converters with single or dual outputs
provide main CPU power in battery-powered systems.
These buck controllers achieve 96% efficiency by using
synchronous rectification and Maxim’s proprietary Idle
Mode™ control scheme to extend battery life at full-load
(up to 10A) and no-load outputs. Excellent dynamic
response corrects output transients caused by the latest
dynamic-clock CPUs within five 300kHz clock cycles.
Unique bootstrap circuitry drives inexpensive N-channel
MOSFETs, reducing system cost and eliminating the
crowbar switching currents found in some PMOS/NMOS
switch designs.
The MAX796/MAX799 are specially equipped with a secondary feedback input (SECFB) for transformer-based
dual-output applications. This secondary feedback path
improves cross-regulation of positive (MAX796) or negative (MAX799) auxiliary outputs.
The MAX797 has a logic-controlled and synchronizable
fixed-frequency pulse-width-modulating (PWM) operating
mode, which reduces noise and RF interference in sensitive mobile-communications and pen-entry applications.
The SKIP override input allows automatic switchover to
idle-mode operation (for high-efficiency pulse skipping) at
light loads, or forces fixed-frequency mode for lowest noise
at all loads.
The MAX796/MAX797/MAX799 are all available in 16pin DIP and narrow SO packages. See the table below
to compare these three converters.
PART
MAIN OUTPUT
SPECIAL FEATURE
MAX796
3.3V/5V or adj
Regulates positive secondary
voltage (such as +12V)
MAX797
3.3V/5V or adj
Logic-controlled low-noise mode
MAX799
3.3V/5V or adj
Regulates negative secondary
voltage (such as -5V)
________________________Applications
____________________________Features
♦ 96% Efficiency
♦ 4.5V to 30V Input Range
♦ 2.5V to 6V Adjustable Output
♦
♦
♦
♦
Preset 3.3V and 5V Outputs (at up to 10A)
Multiple Regulated Outputs
+5V Linear-Regulator Output
Precision 2.505V Reference Output
♦
♦
♦
♦
♦
Automatic Bootstrap Circuit
150kHz/300kHz Fixed-Frequency PWM Operation
Programmable Soft-Start
375μA Typ Quiescent Current (VIN = 12V, VOUT = 5V)
1μA Typ Shutdown Current
_______________Ordering Information
PART†
TEMP RANGE
0°C to +70°C
16 Plastic DIP
MAX796CSE
MAX796C/D
MAX796EPE
0°C to +70°C
0°C to +70°C
-40°C to +85°C
16 Narrow SO
Dice*
16 Plastic DIP
MAX796ESE
MAX796MJE
-40°C to +85°C
-55°C to +125°C
16 Narrow SO
16 CERDIP
Ordering Information continued at end of data sheet.
*Contact factory for dice specifications.
___________________Pin Configuration
TOP VIEW
SS 1
16 DH
(SECFB) SKIP 2
15 LX
Notebook and Subnotebook Computers
REF 3
PDAs and Mobile Communicators
GND 4
Cellular Phones
SYNC 5
14 BST
MAX796
MAX797
MAX799
13 DL
12 PGND
SHDN 6
11 VL
FB 7
10 V+
CSH 8
Idle Mode is a trademark of Maxim Integrated Products.
†U.S. and foreign patents pending.
PIN-PACKAGE
MAX796CPE
9
CSL
DIP/SO
( ) ARE FOR MAX796/ MAX799.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX796/MAX797/MAX799
_______________General Description
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
ABSOLUTE MAXIMUM RATINGS
V+ to GND .................................................................-0.3V, +36V
GND to PGND........................................................................±2V
VL to GND ...................................................................-0.3V, +7V
BST to GND ...............................................................-0.3V, +36V
DH to LX...........................................................-0.3V, BST + 0.3V
LX to BST.....................................................................-7V, +0.3V
SHDN to GND............................................................-0.3V, +36V
SYNC, SS, REF, FB, SECFB, SKIP, DL to GND..-0.3V, VL + 0.3V
CSH, CSL to GND .......................................................-0.3V, +7V
VL Short Circuit to GND..............................................Momentary
REF Short Circuit to GND ...........................................Continuous
VL Output Current ...............................................................50mA
Continuous Power Dissipation (TA = +70°C)
SO (derate 8.70mW/°C above +70°C) ........................696mW
Plastic DIP (derate 10.53mW/°C above +70°C) .........842mW
CERDIP (derate 10.00mW/°C above +70°C) ..............800mW
Operating Temperature Ranges
MAX79_C_ _ ......................................................0°C to +70°C
MAX79_E_ _....................................................-40°C to +85°C
MAX79_MJE .................................................-55°C to +125°C
Storage Temperature Range .............................-65°C to +160°C
Lead Temperature (soldering, 10s) .....................................+300
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V+ = 15V, GND = PGND = 0V, I VL = I REF = 0A, T A = 0°C to +70°C for MAX79_C, T A = 0°C to +85°C for MAX79_E,
TA = -55°C to +125°C for MAX79_M, unless otherwise noted.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
+3.3V AND +5V STEP-DOWN CONTROLLERS
Input Supply Range
MAX79_C
4.5
30
MAX79_E/M
5.0
30
V
5V Output Voltage (CSL)
0mV < (CSH-CSL) < 80mV, FB = VL, 6V < V+ < 30V,
includes line and load regulation
4.85
5.10
5.25
V
3.3V Output Voltage (CSL)
0mV < (CSH-CSL) < 80mV, FB = 0V, 4.5V < V+ < 30V,
includes line and load regulation
3.20
3.35
3.46
V
Nominal Adjustable Output
Voltage Range
External resistor divider
REF
6
V
Feedback Voltage
(CSH-CSL) = 0V
2.43
2.57
V
Load Regulation
Line Regulation
2.505
0mV < (CSH-CSL) < 80mV
2.5
25mV < (CSH-CSL) < 80mV
1.5
6V < V+ < 30V
0.04
%
0.06
CSH-CSL, positive
80
100
120
CSH-CSL, negative
-50
-100
-160
SS Source Current
2.5
4.0
6.5
SS Fault Sink Current
2.0
Current-Limit Voltage
%/V
mV
µA
mA
FLYBACK/PWM CONTROLLER
SECFB Regulation Setpoint
Falling edge, hysteresis = 15mV (MAX796)
2.45
2.505
2.55
Falling edge, hysteresis = 20mV (MAX799)
-0.05
0
0.05
V
INTERNAL REGULATOR AND REFERENCE
VL Output Voltage
SHDN = 2V, 0mA < IVL < 25mA, 5.5V < V+ < 30V
4.7
5.3
V
VL Fault Lockout Voltage
Rising edge, hysteresis = 15mV
3.8
4.1
V
VL/CSL Switchover Voltage
Rising edge, hysteresis = 25mV
4.2
4.7
V
2
_______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
(V+ = 15V, GND = PGND = 0V, I VL = I REF = 0A, T A = 0°C to +70°C for MAX79_C, T A = 0°C to +85°C for MAX79_E,
TA = -55°C to +125°C for MAX79_M, unless otherwise noted.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
MAX79_C
2.46
2.505
2.54
UNITS
MAX79_E/M
2.45
2.55
1.8
2.3
V
50
mV
µA
Reference Output Voltage
No external load (Note 1)
V
Reference Fault Lockout Voltage
Falling edge
Reference Load Regulation
0µA < IREF < 100µA
CSL Shutdown Leakage Current
SHDN = 0V, CSL = 6V, V+ = 0V or 30V, VL = 0V
0.1
1
V+ Shutdown Current
SHDN = 0V, V+ = 30V,
CSL = 0V or 6V
MAX79_C
1
3
MAX79_E/M
1
5
V+ Off-State Leakage Current
FB = CSH = CSL = 6V,
VL switched over to CSL
MAX79_C
1
3
MAX79_E/M
1
5
Dropout Power Consumption
V+ = 4V, CSL = 0V (Note 2)
4
8
mW
Quiescent Power Consumption
CSH = CSL = 6V
4.8
6.6
mW
µA
µA
OSCILLATOR AND INPUTS/OUTPUTS
Oscillator Frequency
SYNC = REF
270
300
330
SYNC = 0V or 5V
125
150
175
SYNC High Pulse Width
200
SYNC Low Pulse Width
200
SYNC Rise/Fall Time
Maximum Duty Cycle
190
89
91
SYNC = 0V or 5V
93
96
SHDN, SKIP
Input Low Voltage
Input Current
ns
SYNC = REF
SYNC
Input High Voltage
ns
Guaranteed by design
Oscillator Sync Range
kHz
200
ns
340
kHz
%
VL - 0.5
V
2.0
SYNC
0.8
SHDN, SKIP
0.5
SHDN, 0V or 30V
2.0
SECFB, 0V or 4V
0.1
SYNC, SKIP
1.0
CSH, CSL, CSH = CSL = 6V, device not shut down
V
µA
50
FB, FB = REF
±100
nA
DL Sink/Source Current
DL forced to 2V
1
A
DH Sink/Source Current
DH forced to 2V, BST-LX = 4.5V
1
DL On-Resistance
High or low
7
Ω
DH On-Resistance
High or low, BST-LX = 4.5V
7
Ω
A
Note 1: Since the reference uses VL as its supply, V+ line-regulation error is insignificant.
Note 2: At very low input voltages, quiescent supply current may increase due to excess PNP base current in the VL linear
regulator. This occurs only if V+ falls below the preset VL regulation point (5V nominal). See the Quiescent Supply Current
vs. Supply Voltage graph in the Typical Operating Characteristics.
_______________________________________________________________________________________
3
MAX796/MAX797/MAX799
ELECTRICAL CHARACTERISTICS (continued)
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
ELECTRICAL CHARACTERISTICS (continued)
(V+ = 15V, GND = PGND = 0V, IVL = IREF = 0A, TA = -40°C to +85°C for MAX79_E, unless otherwise noted.) (Note 3)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
30
V
+3.3V and +5V STEP-DOWN CONTROLLERS
Input Supply Range
5.0
5V Output Voltage (CSL)
0mV < (CSH - CSL) < 80mV, FB = VL, 6V < V+ < 30V,
includes line and load regulation
4.70
5.10
5.40
V
3.3V Output Voltage (CSL)
0mV < (CSH - CSL) < 80mV, FB = VL, 4.5V < V+ < 30V,
includes line and load regulation
3.10
3.35
3.56
V
Nominal Adjustable Output
Voltage Range
External resistor divider
REF
6.0
V
Feedback Voltage
(CSH-CSL) = 0V
2.40
2.60
V
Line Regulation
6V < V+ < 30V
0.06
%/V
Current-Limit Voltage
0.04
CSH - CSL, positive
70
CSH - CSL, negative
-40
130
-100
-160
mV
FLYBACK/PWM CONTROLLER
SECFB Regulation Setpoint
Falling edge, hysteresis = 15mV (MAX796)
2.40
2.60
Falling edge, hysteresis = 20mV (MAX799)
-0.08
0.08
V
INTERNAL REGULATOR AND REFERENCE
VL Output Voltage
SHDN = 2V, 0mA < IVL < 25mA, 5.5V < V+ < 30V
4.7
5.3
V
VL Fault Lockout Voltage
Rising edge, hysteresis = 15mV
3.75
4.05
V
VL/CSL Switchover Voltage
Rising edge, hysteresis = 25mV
4.2
4.7
V
Reference Output Voltage
No external load (Note 1)
2.43
Reference Load Regulation
0µA < IREF < 100µA
V+ Shutdown Current
SHDN = 0V, V+ = 30V, CSL = 0V or 6V
V+ Off-State Leakage Current
FB = CSH = CSL = 6V, VL switched over to CSL
Quiescent Power Consumption
2.505
2.57
V
50
mV
1
10
µA
1
10
µA
4.8
8.4
mW
OSCILLATOR AND INPUTS/OUTPUTS
Oscillator Frequency
SYNC = REF
250
300
350
SYNC = 0V or 5V
120
150
180
kHz
SYNC High Pulse Width
250
ns
SYNC Low Pulse Width
250
ns
Oscillator Sync Range
Maximum Duty Cycle
210
320
SYNC = REF
89
91
SYNC = 0V or 5V
93
96
kHz
%
DL On-Resistance
High or low
7
Ω
DH On-Resistance
High or low, BST - LX = 4.5V
7
Ω
Note 3: All -40°C to +85°C specifications above are guaranteed by design.
4
_______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
INPUT
4.5V TO 30V
V+
VL
SHDN
MAX797
DH
BST
SS
LX
REF
DL
+3.3V
OUTPUT
PGND
SYNC
CSH
GND
CSL
SKIP
FB
INPUT
6V TO 30V
V+
SECFB
SHDN
FB
+12V
OUTPUT
VL
MAX796
DH
BST
+5V
OUTPUT
LX
DL
SS
PGND
REF
CSH
GND
CSL
SYNC
_______________________________________________________________________________________
5
MAX796/MAX797/MAX799
__________________________________________________Typical Operating Circuits
_____________________________________Typical Operating Circuits (continued)
FROM
REF
INPUT 6V TO 30V
V+
SECFB
SHDN
FB
VL
MAX799
–5V
OUTPUT
DH
BST
+5V
OUTPUT
LX
DL
SS
PGND
REF
CSH
GND
CSL
SYNC
__________________________________________Typical Operating Characteristics
(TA = +25°C, unless otherwise noted.)
VIN = 6V
VIN = 5V
EFFICIENCY (%)
VIN = 30V
80
70
STANDARD MAX797 5V/3A
CIRCUIT, FIGURE 1
f = 300kHz
60
VIN = 12V
80
VIN = 30V
70
STANDARD MAX797 3.3V/3A
CIRCUIT, FIGURE 1
f = 300kHz
60
50
0.01
0.1
LOAD CURRENT (A)
1
10
80
SKIP = HIGH
70
60
STANDARD MAX797 3.3V/10A
CIRCUIT, FIGURE 1
f = 300kHz
VIN = 5V
50
40
50
0.001
6
SKIP = LOW
90
90
EFFICIENCY (%)
90
100
MAX796-02
100
MAX796-01
100
EFFICIENCY
vs. LOAD CURRENT, 3.3V/10A CIRCUIT
EFFICIENCY
vs. LOAD CURRENT, 3.3V/3A CIRCUIT
MAX796-03
EFFICIENCY
vs. LOAD CURRENT, 5V/3A CIRCUIT
EFFICIENCY (%)
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
0.001
0.01
0.1
LOAD CURRENT (A)
1
10
0.1
1
LOAD CURRENT (A)
_______________________________________________________________________________________
10
Step-Down Controllers with
Synchronous Rectifier for CPU Power
QUIESCENT SUPPLY CURRENT
vs. SUPPLY VOLTAGE,
3.3V/3A CIRCUIT IN IDLE MODE
600μ
400μ
600
0
0
16
20
24
28
0
32
4
16
20
24
0
28
1.4
VIN - VOUT (mV)
1.0
0.8
DEVICE CURRENT ONLY
SHDN = LOW
f = 300kHz
500
400
300
f = 150kHz
200
0.4
STANDARD MAX797 APPLICATION
CONFIGURED FOR 5V
VOUT > 4.8V
100
0.2
0
0
8
12
16
20
24
28
0.01
32
SWITCHING FREQUENCY (kHz)
300
200
100
0
20
1
10
0.1
100μ
100
1000
MAX796
MAXIMUM SECONDARY CURRENT
vs. SUPPLY VOLTAGE, 5V/15V CIRCUIT
+5V, VIN = 7.5V
+5V, VIN = 30V
+3.3V, VIN = 7.5V
1
32
5
450
10
28
REF LOAD CURRENT (μA)
SYNC = REF (300kHz)
SKIP = LOW
100
24
0
10
1000
MAX796-10
400
20
10
SWITCHING FREQUENCY
vs. LOAD CURRENT
VL LOAD-REGULATION ERROR
vs. LOAD CURRENT
0
1
16
15
LOAD CURRENT (A)
SUPPLY VOLTAGE (V)
500
0.1
12
20
MAXIMUM SECONDARY CURRENT (mA)
4
8
SUPPLY VOLTAGE (V)
MAX796-08
700
600
1.2
0
4
REF LOAD-REGULATION ERROR
vs. LOAD CURRENT
800
MAX796-07
1.6
0.6
0
32
DROPOUT VOLTAGE
vs. LOAD CURRENT
SHUTDOWN SUPPLY CURRENT
vs. SUPPLY VOLTAGE
LOAD REGULATION ΔV (mV)
12
f = 150kHz
10
SUPPLY VOLTAGE (V)
SUPPLY VOLTAGE (V)
SUPPLY CURRENT (μA)
8
f = 300kHz
MAX796-09
12
20
STANDARD MAX797 3.3V/3A
CIRCUIT, FIGURE 1
SKIP = HIGH
STANDARD MAX797 3.3V/3A
CIRCUIT, FIGURE 1
SKIP = LOW
SYNC = REF
LOAD REGULATION ΔV (mV)
8
NOT SWITCHING
(FB FORCED TO 3.5V)
400
200μ
4
MAX796-05
800
200
0
SWITCHING
1000
MAX796-11
STANDARD MAX797 APPLICATION
CONFIGURED FOR 5V
SKIP = LOW
SYNC = REF
SUPPLY CURRENT (mA)
14m
800μ
1200
SUPPLY CURRENT (μA)
SUPPLY CURRENT (A)
15m
30
1400
MAX796-04
16m
QUIESCENT SUPPLY CURRENT
vs. SUPPLY VOLTAGE, LOW-NOISE MODE
MAX796-06
QUIESCENT SUPPLY CURRENT
vs. SUPPLY VOLTAGE,
5V/3A CIRCUIT IN IDLE MODE
IOUT (MAIN) = 0A
400
350
300
IOUT (MAIN) = 3A
250
200
150
CIRCUIT OF FIGURE 11
TRANSFORMER = TTI5870
VSEC > 12.75V
100
50
0
1m
10m
100m
1
28
32
_______________________________________________________________________________________
7
40
60
VL LOAD CURRENT (mA)
80
LOAD CURRENT (A)
0
4
8
12
16
20
24
SUPPLY VOLTAGE (V)
MAX796/MAX797/MAX799
____________________________Typical Operating Characteristics (continued)
(TA = +25°C, unless otherwise noted.)
____________________________Typical Operating Characteristics (continued)
(TA = +25°C, unless otherwise noted.)
MAX796
MAXIMUM SECONDARY CURRENT
vs. SUPPLY VOLTAGE, 3.3V/5V CIRCUIT
800
900
MAXIMUM SECONDARY CURRENT (mA)
IOUT (MAIN) = 2A
750
IOUT (MAIN) = 0A
600
450
300
CIRCUIT OF FIGURE 12
TRANSFORMER = TDK 1.5:1
VSEC ≥ 4.8V
150
0
IOUT (MAIN) = 0A
700
600
500
400
IOUT (MAIN) = 1A
300
CIRCUIT OF FIGURE 13
TRANSFORMER = TTI5926
VSEC ≤ -5.1V
200
100
0
0
3
6
9
12
15
18
21
24
0
4
SUPPLY VOLTAGE (V)
8
12
16
20
24
28
32
SUPPLY VOLTAGE (V)
PULSE-WIDTH-MODULATION MODE WAVEFORMS
IDLE-MODE WAVEFORMS
LX VOLTAGE
10V/div
+5V OUTPUT
50mV/div
+5V OUTPUT
VOLTAGE
50mV/div
2V/div
500ns/div
200μs/div
ILOAD = 1A, VIN = 16V,
CIRCUIT OF FIGURE 1
ILOAD = 100mA, VIN = 10V,
CIRCUIT OF FIGURE 1
+5V LOAD-TRANSIENT RESPONSE
3A
0A
LOAD CURRENT
+5V OUTPUT
50mV/div
200μs/div
VIN = 15V, CIRCUIT OF FIGURE 1
8
MAX796-13
MAX799
MAXIMUM SECONDARY CURRENT
vs. SUPPLY VOLTAGE, ±5V CIRCUIT
MAX796-12
1050
MAXIMUM SECONDARY CURRENT (mA)
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
_______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
PIN
NAME
1
SS
SECFB
(MAX796/
MAX799)
2
SKIP
(MAX797)
FUNCTION
Soft-Start timing capacitor connection. Ramp time to full current limit is approximately 1ms/nF.
Secondary winding Feedback input. Normally connected to a resistor divider from an auxiliary output.
Don’t leave SECFB unconnected.
• MAX796: SECFB regulates at VSECFB = 2.505V. Tie to VL if not used.
• MAX799: SECFB regulates at VSECFB = 0V. Tie to a negative voltage through a high-value current-limiting resistor (IMAX = 100µA) if not used.
Disables pulse-skipping mode when high. Connect to GND for normal use. Don’t leave SKIP unconnected.
With SKIP grounded, the device will automatically change from pulse-skipping operation to full PWM operation when the load current exceeds approximately 30% of maximum. (See Table 3.)
3
REF
Reference voltage output. Bypass to GND with 0.33µF minimum.
4
GND
Low-noise analog Ground and feedback reference point.
5
SYNC
Oscillator Synchronization and frequency select. Tie to GND or VL for 150kHz operation; tie to REF for
300kHz operation. A high-to-low transition begins a new cycle. Drive SYNC with 0V to 5V logic levels (see the
Electrical Characteristics table for VIH and VIL specifications). SYNC capture range is 190kHz to 340kHz
guaranteed.
6
SHDN
Shutdown control input, active low. Logic threshold is set at approximately 1V (VTH of an internal N-channel
MOSFET). Tie SHDN to V+ for automatic start-up.
Feedback input. Regulates at FB = REF (approximately 2.505V) in adjustable mode. FB is a Dual-ModeTM
input that also selects the fixed output voltage settings as follows:
• Connect to GND for 3.3V operation.
• Connect to VL for 5V operation.
• Connect FB to a resistor divider for adjustable mode. FB can be driven with +5V rail-to-rail logic in order to
change the output voltage under system control.
7
FB
8
CSH
Current-Sense input, High side. Current-limit level is 100mV referred to CSL.
9
CSL
Current-Sense input, Low side. Also serves as the feedback input in fixed-output modes.
10
V+
Battery voltage input (4.5V to 30V). Bypass V+ to PGND close to the IC with a 0.1µF capacitor. Connects to a
linear regulator that powers VL.
11
VL
5V Internal linear-regulator output. VL is also the supply voltage rail for the chip. VL is switched to the output
voltage via CSL (VCSL > 4.5V) for automatic bootstrapping. Bypass to GND with 4.7µF. VL can
supply up to 5mA for external loads.
12
PGND
13
DL
Low-side gate-drive output. Normally drives the synchronous-rectifier MOSFET. Swings 0V to VL.
14
BST
Boost capacitor connection for high-side gate drive (0.1µF).
15
LX
Switching node (inductor) connection. Can swing 2V below ground without hazard.
16
DH
High-side gate-drive output. Normally drives the main buck switch. DH is a floating driver output that swings
from LX to BST, riding on the LX switching-node voltage.
Power Ground.
Dual Mode is a trademark of Maxim Integrated Products.
_______________________________________________________________________________________
9
MAX796/MAX797/MAX799
______________________________________________________________Pin Description
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
If the 3A or 5A circuit must be guaranteed to withstand
a continuous output short circuit indefinitely, see the
section MOSFET Switches under Selecting Other
Components. Don’t change the frequency of these circuits without first recalculating component values (particularly inductance value at maximum battery voltage).
______Standard Application Circuit
It is easy to adapt the basic MAX797 single-output 3.3V
buck converter (Figure 1) to meet a wide range of
applications with inputs up to 28V (limited by choice of
external MOSFET). Simply substitute the appropriate
components from Table 1. These circuits represent a
good set of tradeoffs between cost, size, and efficiency
while staying within the worst-case specification limits
for stress-related parameters such as capacitor ripple
current. Each of these circuits is rated for a continuous
load current at TA = +85°C, as shown. The 1A, 2A and
10A applications can withstand a continuous output
short-circuit to ground. The 3A and 5A applications can
withstand a short circuit of many seconds duration, but
the synchronous-rectifier MOSFET overheats, exceeding the manufacturer’s ratings for junction temperature
by 50°C or more.
_______________Detailed Description
The MAX796 is a BiCMOS, switch-mode power-supply
controller designed primarily for buck-topology regulators in battery-powered applications where high efficiency and low quiescent supply current are critical.
The MAX796 also works well in other topologies such
as boost, inverting, and CLK due to the flexibility of its
floating high-speed gate driver. Light-load efficiency is
enhanced by automatic idle-mode operation—a variable-frequency pulse-skipping mode that reduces
INPUT
C1
C7
0.1μF
10
V+
ON/OFF
CONTROL
LOW-NOISE
CONTROL
6
+5V AT
5mA
11
D2
CMPSH-3
VL
DH
SHDN
BST
2
MAX797
SKIP
LX
DL
PGND
1
CSH
SS
C6
0.01μF
(OPTIONAL)
CSL
GND
FB
7
NOTE: KEEP CURRENT-SENSE
LINES SHORT AND CLOSE
TOGETHER. SEE FIG. 10
SYNC
REF
16
C4
4.7μF
Q1
14
15
C3
0.1μF
L1
+3.3V
OUTPUT
R1
C2
13
Q2
D1
12
8
9
4
3
5
C5
0.33μF
J1
150kHz/300kHz
JUMPER
Figure 1. Standard 3.3V Application Circuit
10
GND
OUT
______________________________________________________________________________________
REF OUTPUT
+2.505V AT 100μA
Step-Down Controllers with
Synchronous Rectifier for CPU Power
LOAD CURRENT
COMPONENT
1A
2A
3A
4A
10A
Input Range
4.75V to 18V
4.75V to 18V
4.75V to 28V
4.75V to 24V
4.5V to 6V
Application
PDA
Sub-Notebook
Notebook
High-End Notebook
Desktop 5V-to-3V
Frequency
150kHz
300kHz
300kHz
300kHz
300kHz
Q1 High-Side
MOSFET
Motorola 1/2
International Rectifier
MMDF3N03HD or 1/2
1/2 IRF7101
Si9936
Motorola
MMSF5N03HD or
Si9410
Motorola
MTD20N03HDL
DPAK
Motorola
MTD75N03HDL
D2PAK
Q2 Low-Side
MOSFET
Motorola 1/2
International Rectifier
MMDF3N03HD or 1/2
1/2 IRF7101
Si9936
Motorola
MMSF5N03HD or
Si9410
Motorola
MTD20N03HDL
DPAK
Motorola
MTD75N03HDL
D2PAK
C1 Input
Capacitor
22µF, 35V AVX TPS
or Sprague 595D
2 x 220µF, 10V
2 x 22µF, 35V AVX
4 x 22µF, 35V AVX
Sanyo OS-CON
TPS or Sprague 595D TPS or Sprague 595D
10SA220M
C2 Output
Capacitor
150µF, 10V AVX TPS 150µF, 10V AVX TPS
or Sprague 595D
or Sprague 595D
D1 Rectifier
1N5817 Motorola
MBR0502L SOD-89
1N5817 NIEC
EC10QS02L or
Motorola MBRS130T3
1N5819 NIEC
1N5821 NIEC
1N5820 NIEC
EC10QS03 or
NSQ03A04 or
NSQ03A02, or
Motorola MBRS130T3 Motorola MBRS340T3 Motorola MBRS340T3
R1 Resistor
0.062Ω IRC
LR2010-01-R062
0.039Ω IRC
LR2010-01-R039
0.025Ω IRC
LR2010-01-R025
0.015Ω IRC
LR2010-01-015
L1 Inductor
47µH, 1.2A Ferrite or
33µH, 2.2A Ferrite
Kool-Mu
Dale LPE6562-330MB
Sumida CD75-470
10µH, 3A Ferrite
Sumida CDRH125
1.5µH, 11A, 3.5mΩ
4.7µH, 5.5A Ferrite
Coiltronics
Coilcraft DO3316-472
CTX03-12357-1
2 x 22µF, 35V AVX
TPS or Sprague 595D
220µF, 10V AVX TPS
or Sprague 595D
4 x 220µF, 10V
3 x 220µF, 10V AVX
Sanyo OS-CON
TPS or Sprague 595D
10SA220M
3 x 0.02Ω IRC
LR2010-01-R020
(3 in parallel)
Table 2. Component Suppliers
MANUFACTURER
AVX
Central Semiconductor
Coilcraft
Coiltronics
Dale
International Rectifier
IRC
Kemet
Matsuo
Motorola
USA PHONE
(803) 946-0690
(516) 435-1110
(847) 639-6400
(561) 241-7876
(605) 668-4131
(310) 322-3331
(512) 992-7900
(864) 963-6300
(714) 969-2491
(602) 303-5454
FACTORY FAX
[Country Code]
[1] 803-626-3123
[1] 516-435-1824
[1] 847-639-1469
[1] 561-241-9339
[1] 605-665-1627
[1] 310-322-3332
[1] 512-992-3377
[1] 864-963-6521
[1] 714-960-6492
[1] 602-994-6430
USA PHONE
FACTORY FAX
[Country Code]
Murata-Erie
(814) 237-1431
(800) 831-9172
[1] 814-238-0490
NIEC
Sanyo
(805) 867-2555* [81] 3-3494-7414
(619) 661-6835 [81] 7-2070-1174
Siliconix
(408) 988-8000
(800) 554-5565
[1] 408-970-3950
Sprague
Sumida
TDK
Transpower Technologies
(603) 224-1961
(847) 956-0666
(847) 390-4461
(702) 831-0140
[1] 603-224-1430
[81] 3-3607-5144
{1} 847-390-4405
[1] 702-831-3521
MANUFACTURER
* Distributor
losses due to MOSFET gate charge. The step-down
power-switching circuit consists of two N-channel
MOSFETs, a rectifier, and an LC output filter. The output voltage is the average of the AC voltage at the
switching node, which is adjusted and regulated by
changing the duty cycle of the MOSFET switches. The
gate-drive signal to the N-channel high-side MOSFET
must exceed the battery voltage and is provided by a
flying capacitor boost circuit that uses a 100nF capacitor connected to BST.
The MAX796 contains nine major circuit blocks, which
are shown in Figure 2.
______________________________________________________________________________________
11
MAX796/MAX797/MAX799
Table 1. Component Selection for Standard 3.3V Applications
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
BATTERY VOLTAGE
TO
CSL
V+
+5V LINEAR
REGULATOR
4.5V
OUT
VL
+5V AT 5mA
AUXILIARY
OUTPUT
SHDN
BST
SECFB
DH
PWM
LOGIC
LX
DL
+2.505V
REF
+2.505V
AT 100μA
PGND
PWM
COMPARATOR
CSH
REF
CSL
LPF
60kHz
GND
ON/OFF
3.3V FB
5V FB
SHDN
SS
ADJ FB
FB
4V
MAX796
SYNC
1V
Figure 2. MAX796 Block Diagram
12
______________________________________________________________________________________
MAIN
OUTPUT
Step-Down Controllers with
Synchronous Rectifier for CPU Power
These internal IC blocks aren’t powered directly from
the battery. Instead, a +5V linear regulator steps down
the battery voltage to supply both the IC internal rail (VL
pin) as well as the gate drivers. The synchronousswitch gate driver is directly powered from +5V VL,
while the high-side-switch gate driver is indirectly powered from VL via an external diode-capacitor boost circuit. An automatic bootstrap circuit turns off the +5V
linear regulator and powers the IC from its output voltage if the output is above 4.5V.
PWM Controller Block
The heart of the current-mode PWM controller is a
multi-input open-loop comparator that sums three signals: output voltage error signal with respect to the reference voltage, current-sense signal, and slope
compensation ramp (Figure 3). The PWM controller is a
direct summing type, lacking a traditional error amplifier and the phase shift associated with it. This directsumming configuration approaches the ideal of
cycle-by-cycle control over the output voltage.
Under heavy loads, the controller operates in full PWM
mode. Each pulse from the oscillator sets the main
PWM latch that turns on the high-side switch for a period determined by the duty factor (approximately
VOUT/VIN). As the high-switch turns off, the synchronous rectifier latch is set. 60ns later the low-side switch
turns on, and stays on until the beginning of the next
clock cycle (in continuous mode) or until the inductor
current crosses zero (in discontinuous mode). Under
fault conditions where the inductor current exceeds the
100mV current-limit threshold, the high-side latch
resets and the high-side switch turns off.
At light loads (SKIP = low), the inductor current fails to
exceed the 30mV threshold set by the minimum-current
comparator. When this occurs, the controller goes into
idle mode, skipping most of the oscillator pulses in
order to reduce the switching frequency and cut back
gate-charge losses. The oscillator is effectively gated
off at light loads because the minimum-current comparator immediately resets the high-side latch at the
Table 3. Operating-Mode Truth Table
SHDN
SKIP
LOAD
CURRENT
MODE
NAME
Low
X
X
Shutdown
DESCRIPTION
All circuit blocks
turned off; supply
current = 1µA typ
High
Low
Low,
<10%
Idle
Pulse-skipping;
supply current =
700µA typ at VIN =
10V; discontinuous
inductor current
High
Low
Medium,
<30%
Idle
Pulse-skipping;
continuous inductor
current
High
Low
High,
>30%
PWM
Constant-frequency
PWM; continuous
inductor current
High
High
X
Constant-frequency
PWM regardless of
Low Noise*
load; continuous
(PWM)
inductor current
even at no load
* MAX796/MAX799 have no SKIP pin and therefore can’t go
into low-noise mode.
X = Don’t Care
beginning of each cycle, unless the feedback signal
falls below the reference voltage level.
When in PWM mode, the controller operates as a fixedfrequency current-mode controller where the duty ratio
is set by the input/output voltage ratio. The currentmode feedback system regulates the peak inductor
current as a function of the output voltage error signal.
Since the average inductor current is nearly the same
as the peak current, the circuit acts as a switch-mode
transconductance amplifier and pushes the second
output LC filter pole, normally found in a duty-factorcontrolled (voltage-mode) PWM, to a higher frequency.
To preserve inner-loop stability and eliminate regenerative inductor current “staircasing,” a slope-compensation ramp is summed into the main PWM comparator to
reduce the apparent duty factor to less than 50%.
The relative gains of the voltage- and current-sense
inputs are weighted by the values of current sources
that bias three differential input stages in the main PWM
comparator (Figure 4). The relative gain of the voltage
comparator to the current comparator is internally fixed
at K = 2:1. The resulting loop gain (which is relatively
low) determines the 2.5% typical load regulation error.
The low loop-gain value helps reduce output filter
capacitor size and cost by shifting the unity-gain
crossover to a lower frequency.
______________________________________________________________________________________
13
MAX796/MAX797/MAX799
PWM Controller Blocks:
• Multi-Input PWM Comparator
• Current-Sense Circuit
• PWM Logic Block
• Dual-Mode Internal Feedback Mux
• Gate-Driver Outputs
• Secondary Feedback Comparator
Bias Generator Blocks:
• +5V Linear Regulator
• Automatic Bootstrap Switchover Circuit
• +2.505V Reference
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
CSH
1X
CSL
REF
FROM
FEEDBACK
DIVIDER
MAIN PWM
COMPARATOR
BST
R
LEVEL
SHIFT
Q
S
DH
LX
SLOPE COMP
OSC
30mV
SKIP
(MAX797
ONLY)
VL
4μA
CURRENT
LIMIT
SHOOTTHROUGH
CONTROL
24R
SS
2.5V
N
SHDN
1R
SYNCHRONOUS
RECTIFIER CONTROL
R
–100mV
VL
Q
LEVEL
SHIFT
S
DL
PGND
REF (MAX796)
GND (MAX799)
SECFB
1μs
SINGLE-SHOT
NOTE 1
NOTE 1: COMPARATOR INPUT POLARITIES
ARE REVERSED FOR THE MAX799.
MAX796, MAX799 ONLY
Figure 3. PWM Controller Detailed Block Diagram
14
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
R1
R2
TO PWM
LOGIC
UNCOMPENSATED
HIGH-SPEED
LEVEL TRANSLATOR
AND BUFFER
OUTPUT DRIVER
FB
I1
I2
I3
REF
CSH
CSL
SLOPE COMPENSATION
Figure 4. Main PWM Comparator Block Diagram
The output filter capacitor C2 sets a dominant pole in
the feedback loop. This pole must roll off the loop gain
to unity before the zero introduced by the output
capacitor’s parasitic resistance (ESR) is encountered
(see Design Procedure section). A 60kHz pole-zero
cancellation filter provides additional rolloff above the
unity-gain crossover. This internal 60kHz lowpass compensation filter cancels the zero due to the filter capacitor’s ESR. The 60kHz filter is included in the loop in
both fixed- and adjustable-output modes.
Synchronous-Rectifier Driver (DL Pin)
Synchronous rectification reduces conduction losses in
the rectifier by shunting the normal Schottky diode with
a low-resistance MOSFET switch. The synchronous rectifier also ensures proper start-up of the boost-gate driver circuit. If you must omit the synchronous power
MOSFET for cost or other reasons, replace it with a
small-signal MOSFET such as a 2N7002.
If the circuit is operating in continuous-conduction
mode, the DL drive waveform is simply the complement
of the DH high-side drive waveform (with controlled
dead time to prevent cross-conduction or “shootthrough”). In discontinuous (light-load) mode, the synchronous switch is turned off as the inductor current
falls through zero. The synchronous rectifier works
under all operating conditions, including idle mode.
The synchronous-switch timing is further controlled by
the secondary feedback (SECFB) signal in order to
improve multiple-output cross-regulation (see
Secondary Feedback-Regulation Loop section).
Internal VL and REF Supplies
An internal regulator produces the 5V supply (VL) that
powers the PWM controller, logic, reference, and other
blocks within the MAX796. This +5V low-dropout linear
regulator can supply up to 5mA for external loads, with
a reserve of 20mA for gate-drive power. Bypass VL to
GND with 4.7µF. Important: VL must not be allowed to
exceed 6V. Measure VL with the main output fully
loaded. If VL is being pumped up above 5.5V, the
probable cause is either excessive boost-diode capacitance or excessive ripple at V+. Use only small-signal
diodes for D2 (1N4148 preferred) and bypass V+ to
PGND with 0.1µF directly at the package pins.
The 2.505V reference (REF) is accurate to ±1.6% over
temperature, making REF useful as a precision system
reference. Bypass REF to GND with 0.33µF minimum.
REF can supply up to 1mA for external loads. However,
if tight-accuracy specs for either VOUT or REF are
essential, avoid loading REF with more than 100µA.
Loading REF reduces the main output voltage slightly,
according to the reference-voltage load regulation
error. In MAX799 applications, ensure that the SECFB
divider doesn’t load REF heavily.
When the main output voltage is above 4.5V, an internal Pchannel MOSFET switch connects CSL to VL while simultaneously shutting down the VL linear regulator. This
action bootstraps the IC, powering the internal circuitry
from the output voltage, rather than through a linear regulator from the battery. Bootstrapping reduces power dissipation caused by gate-charge and quiescent losses by
providing that power from a 90%-efficient switch-mode
source, rather than from a 50%-efficient linear regulator.
______________________________________________________________________________________
15
MAX796/MAX797/MAX799
VL
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
It’s often possible to achieve a bootstrap-like effect, even
for circuits that are set to VOUT < 4.5V, by powering VL
from an external-system +5V supply. To achieve this
pseudo-bootstrap, add a Schottky diode between the
external +5V source and VL, with the cathode to the VL
side. This circuit provides a 1% to 2% efficiency boost
and also extends the minimum battery input to less than
4V. The external source must be in the range of 4.8V to
6V. Another way to achieve a pseudo-bootstrap is to add
an extra flyback winding to the main inductor to generate
the +5V bootstrap source, as shown in the +3.3V/+5V
Dual-Output Application (Figure 12).
Boost High-Side
Gate-Driver Supply (BST Pin)
Gate-drive voltage for the high-side N-channel switch is
generated by a flying-capacitor boost circuit as shown
in Figure 5. The capacitor is alternately charged from
the VL supply and placed in parallel with the high-side
MOSFET’s gate-source terminals.
On start-up, the synchronous rectifier (low-side MOSFET) forces LX to 0V and charges the BST capacitor to
5V. On the second half-cycle, the PWM turns on the
high-side MOSFET by closing an internal switch
between BST and DH. This provides the necessary
enhancement voltage to turn on the high-side switch,
an action that “boosts” the 5V gate-drive signal above
the battery voltage.
Ringing seen at the high-side MOSFET gate (DH) in
discontinuous-conduction mode (light loads) is a natural operating condition, and is caused by the residual
energy in the tank circuit formed by the inductor and
stray capacitance at the switching node LX. The gatedriver negative rail is referred to LX, so any ringing
there is directly coupled to the gate-drive output.
Current-Limiting and
Current-Sense Inputs (CSH and CSL)
The current-limit circuit resets the main PWM latch and
turns off the high-side MOSFET switch whenever the
voltage difference between CSH and CSL exceeds
100mV. This limiting is effective for both current flow
directions, putting the threshold limit at ±100mV. The
tolerance on the positive current limit is ±20%, so the
external low-value sense resistor must be sized for
80mV/R1 to guarantee enough load capability, while
components must be designed to withstand continuous
current stresses of 120mV/R1.
For breadboarding purposes or very high-current applications, it may be useful to wire the current-sense inputs
with a twisted pair rather than PC traces. This twisted
pair needn’t be anything special, perhaps two pieces of
wire-wrap wire twisted together.
16
BATTERY
+5V
VL SUPPLY INPUT
VL
VL
MAX796
MAX797
MAX799
BST
DH
LEVEL
TRANSLATOR
PWM
LX
VL
DL
Figure 5. Boost Supply for Gate Drivers
Oscillator Frequency and
Synchronization (SYNC Pin)
The SYNC input controls the oscillator frequency.
Connecting SYNC to GND or to VL selects 150kHz
operation; connecting SYNC to REF selects 300kHz.
SYNC can also be used to synchronize with an external
5V CMOS or TTL clock generator. SYNC has a guaranteed 190kHz to 340kHz capture range.
300kHz operation optimizes the application circuit for
component size and cost. 150kHz operation provides
increased efficiency and improved load-transient
response at low input-output voltage differences (see
Low-Voltage Operation section).
Low-Noise Mode (SKIP Pin)
The low-noise mode (SKIP = high) is useful for minimizing RF and audio interference in noise-sensitive applications such as Soundblaster™ hi-fi audio-equipped
systems, cellular phones, RF communicating computers, and electromagnetic pen-entry systems. See the
summary of operating modes in Table 3. SKIP can be
driven from an external logic signal.
The MAX797 can reduce interference due to switching
noise by ensuring a constant switching frequency
regardless of load and line conditions, thus concentrating the emissions at a known frequency outside the
system audio or IF bands. Choose an oscillator freSoundblaster is a trademark of Creative Labs.
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
Remote sensing of the output voltage, while not possible in fixed-output mode due to the combined nature of
the voltage- and current-sense input (CSL), is easy to
achieve in adjustable mode by using the top of the
external resistor divider as the remote sense point.
Fixed-output accuracy is guaranteed to be ±4% over
all conditions. In special circumstances, it may be necessary to improve upon this output accuracy. The HighAccuracy Adjustable-Output Application (Figure 18)
provides ±2.5% accuracy by adding an integrator-type
error amplifier.
The breakdown voltage rating of the current-sense
inputs (7V absolute maximum) determines the 6V maximum output adjustment range. To extend this output
range, add two matched resistor dividers and speedup capacitors to form a level translator, as shown in
Figure 8. Be sure to set these resistor ratios accurately
(using 0.1% resistors), to avoid adding excessive error
to the 100mV current-limit threshold.
Secondary Feedback-Regulation Loop
(SECFB Pin)
A flyback winding control loop regulates a secondary
winding output (MAX796/MAX799 only), improving
cross-regulation when the primary is lightly loaded or
when there is a low input-output differential voltage. If
SECFB crosses its regulation threshold (VREF for the
Adjustable-Output Feedback
(Dual-Mode FB Pin)
Adjusting the main output voltage with external resistors is easy for any of the devices in the MAX796 family,
via the circuit of Figure 6. The nominal output voltage
(given by the formula in Figure 6) should be set approximately 2% high in order to make up for the MAX796’s
-2.5% typical load-regulation error. For example, if
designing for a 3.0V output, use a resistor ratio that
results in a nominal output voltage of 3.06V. This slight
offsetting gives the best possible accuracy.
Recommended normal values for R5 range from 5kΩ to
100kΩ. To achieve a 2.505V nominal output, simply
connect FB to CSL directly. To achieve output voltages
lower than 2.5V, use an external reference-voltage
source higher than VREF, as shown in Figure 7. For best
accuracy, this second reference voltage should be
much higher than VREF. Alternatively, an external op
amp could be used to gain-up REF in order to create
the second reference source. This scheme requires a
minimum load on the output in order to sink the R3/R4
divider current.
V+
DH
REMOTE
SENSE
LINES
MAIN
OUTPUT
MAX796
MAX797 DL
MAX799
R4
CSH
CSL
FB
GND
R5
R4
VOUT = VREF 1 + –––
R5
WHERE VREF (NOMINAL) = 2.505V
(
)
Figure 6. Adjusting the Main Output Voltage
______________________________________________________________________________________
17
MAX796/MAX797/MAX799
quency where harmonics of the switching frequency
don’t overlap a sensitive frequency band. If necessary,
synchronize the oscillator to a tight-tolerance external
clock generator.
The low-noise mode (SKIP = high) forces two changes
upon the PWM controller. First, it ensures fixed-frequency operation by disabling the minimum-current comparator and ensuring that the PWM latch is set at the
beginning of each cycle, even if the output is in regulation. Second, it ensures continuous inductor current
flow, and thereby suppresses discontinuous-mode
inductor ringing by changing the reverse current-limit
detection threshold from zero to -100mV, allowing the
inductor current to reverse at very light loads.
In most applications, SKIP should be tied to GND in
order to minimize quiescent supply current. Supply current with SKIP high is typically 10mA to 20mA, depending on external MOSFET gate capacitance and
switching losses.
Forced continuous conduction via SKIP can improve
cross regulation of transformer-coupled multiple-output
supplies. This second function of the SKIP pin produces a result that is similar to the method of adding
secondary regulation via the SECFB feedback pin, but
with much higher quiescent supply current. Still,
improving cross regulation by enabling SKIP instead of
building in SECFB feedback can be useful in noisesensitive applications, since SECFB and SKIP are
mutually exclusive pins/functions in the MAX796 family.
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
V+
VREF2 >>VREF
(4.096V)
MAX874
DH
V+
R4
MAIN
OUTPUT
RSENSE
MAX796
MAX797
DL
MAX799
MAX796
MAX797 DL
MAX799
R1
2.43k
0.01μF
R3
2.43k
0.01μF
CSH
CSL
CSH
FB
GND
CSL
FB
GND
OUTPUT
(8V AS
SHOWN)
DH
R5
R4
VOUT = VREF - (VREF2 - VREF) (–––)
R5
R2
1.1k
R4
1.1k
R3
VOUT = VREF (1 + –––)
R4
DIVIDER IMPEDANCE ≤ 5kΩ
(EACH LEG)
Figure 7. Output Voltage Less than 2.5V
Figure 8. Adjusting the Output Voltage to Greater than 6V
MAX796), a 1µs one-shot is triggered that extends the
low-side switch’s on-time beyond the point where the
inductor current crosses zero (in discontinuous mode).
This causes the inductor (primary) current to reverse,
which in turn pulls current out of the output filter capacitor
and causes the flyback transformer to operate in the forward mode. The low impedance presented by the transformer secondary in the forward mode dumps current into
the secondary output, charging up the secondary capacitor and bringing SECFB back into regulation. The SECFB
feedback loop does not improve secondary output accuracy in normal flyback mode, where the main (primary)
output is heavily loaded. In this mode, secondary output
accuracy is determined, as usual, by the secondary rectifier drop, turns ratio, and accuracy of the main output
voltage. So, a linear post-regulator may still be needed in
order to meet tight output accuracy specifications.
The secondary output voltage-regulation point is determined by an external resistor divider at SECFB. For negative output voltages, the SECFB comparator is referenced
to GND (MAX799); for positive output voltages, SECFB
regulates at the 2.505V reference (MAX796). As a result,
output resistor divider connections and design equations
for the two device types differ slightly (Figure 9).
Ordinarily, the secondary regulation point is set 5% to
10% below the voltage normally produced by the flyback
effect. For example, if the output voltage as determined
by the turns ratio is +15V, the feedback resistor ratio
should be set to produce about +13.5V; otherwise, the
SECFB one-shot might be triggered unintentionally, causing an unnecessary increase in supply current and output
noise. In negative-output (MAX799) applications, the
resistor divider acts as a load on the internal reference,
which in turn can cause errors at the main output. Avoid
overloading REF (see the Reference Load-Regulation
Error vs. Load Current graph in the Typical Operating
Characteristics). 100kΩ is a good value for R3 in MAX799
circuits.
18
Soft-Start Circuit (SS)
Soft-start allows a gradual increase of the internal current-limit level at start-up for the purpose of reducing
input surge currents, and perhaps for power-supply
sequencing. In shutdown mode, the soft-start circuit
holds the SS capacitor discharged to ground. When
SHDN goes high, a 4µA current source charges the SS
capacitor up to 3.2V. The resulting linear ramp waveform causes the internal current-limit level to increase
proportionally from 20mV to 100mV. The main output
capacitor thus charges up relatively slowly, depending
on the SS capacitor value. The exact time of the output
rise depends on output capacitance and load current
and is typically 1ms per nanofarad of soft-start capacitance. With no SS capacitor connected, maximum current limit is reached within 10µs.
Shutdown
Shutdown mode (SHDN = 0V) reduces the V+ supply
current to typically 1µA. In this mode, the reference and
VL are inactive. SHDN is a logic-level input, but it can
be safely driven to the full V+ range. Connect SHDN to
V+ for automatic start-up. Do not allow slow transitions
(slower than 0.02V/µs) on SHDN.
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
The five pre-designed standard application circuits
(Figure 1 and Table 1) contain ready-to-use solutions
for common applications. Use the following design procedure to optimize the basic schematic for different
voltage or current requirements. Before beginning a
design, firmly establish the following:
VIN(MAX), the maximum input (battery) voltage. This
value should include the worst-case conditions, such
as no-load operation when a battery charger or AC
adapter is connected but no battery is installed.
VIN(MAX) must not exceed 30V. This 30V upper limit is
determined by the breakdown voltage of the BST floating gate driver to GND (36V absolute maximum).
VIN(MIN), the minimum input (battery) voltage. This
should be taken at full-load under the lowest battery
conditions. If VIN(MIN) is less than 4.5V, a special circuit
must be used to externally hold up VL above 4.8V. If
the minimum input-output difference is less than 1.5V,
the filter capacitance required to maintain good AC
load regulation increases.
Inductor Value
The exact inductor value isn’t critical and can be
adjusted freely in order to make tradeoffs among size,
cost, and efficiency. Although lower inductor values will
minimize size and cost, they will also reduce efficiency
due to higher peak currents. To permit use of the physically smallest inductor, lower the inductance until the
circuit is operating at the border between continuous
and discontinuous modes. Reducing the inductor value
even further, below this crossover point, results in discontinuous-conduction operation even at full load. This
helps reduce output filter capacitance requirements but
causes the core energy storage requirements to
increase again. On the other hand, higher inductor values will increase efficiency, but at some point resistive
losses due to extra turns of wire will exceed the benefit
gained from lower AC current levels. Also, high inductor values can affect load-transient response; see the
VSAG equation in the Low-Voltage Operation section.
The following equations are given for continuous-conduction operation since the MAX796 is mainly intended
for high-efficiency battery-powered applications. See
Appendix A in Maxim’s Battery Management and DCDC Converter Circuit Collection for crossover point and
discontinuous-mode equations. Discontinuous conduction doesn’t affect normal idle-mode operation.
0.33μF
REF
R3
R3
SECFB
SECFB
1-SHOT
TRIG
1-SHOT
TRIG
R2
V+
2.505V REF
R2
POSITIVE
SECONDARY
OUTPUT
NEGATIVE
SECONDARY
OUTPUT
V+
DH
DH
MAIN
OUTPUT
MAX796
MAX799
MAIN
OUTPUT
DL
DL
R2
+VTRIP = VREF 1 + –––
R3
(
)
WHERE VREF (NOMINAL) = 2.505V
R2
-VTRIP = -VREF –––
R3
( )
R3 = 100kΩ (RECOMMENDED)
Figure 9. Secondary-Output Feedback Dividers, MAX796 vs. MAX799
______________________________________________________________________________________
19
MAX796/MAX797/MAX799
_________________Design Procedure
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
Three key inductor parameters must be specified:
inductance value (L), peak current (IPEAK), and DC
resistance (R DC). The following equation includes a
constant LIR, which is the ratio of inductor peak-topeak AC current to DC load current. A higher value of
LIR allows smaller inductance, but results in higher
losses and ripple. A good compromise between size
and losses is found at a 30% ripple current to load current ratio (LIR = 0.3), which corresponds to a peak
inductor current 1.15 times higher than the DC load
current.
VOUT (VIN(MAX) - VOUT)
L = ———————————
VIN(MAX) x f x IOUT x LIR
where:
f = switching frequency, normally 150kHz or
300kHz
IOUT = maximum DC load current
LIR = ratio of AC to DC inductor current,
typically 0.3
The peak inductor current at full load is 1.15 x IOUT if
the above equation is used; otherwise, the peak current
can be calculated by:
VOUT (VIN(MAX) - VOUT)
IPEAK = ILOAD + ———————————
2 x f x L x VIN(MAX)
The inductor’s DC resistance is a key parameter for efficiency performance and must be ruthlessly minimized,
preferably to less than 25mΩ at IOUT = 3A. If a standard off-the-shelf inductor is not available, choose a
core with an LI2 rating greater than L x IPEAK2 and wind
it with the largest diameter wire that fits the winding
area. For 300kHz applications, ferrite core material is
strongly preferred; for 150kHz applications, Kool-mu
(aluminum alloy) and even powdered iron can be
acceptable. If light-load efficiency is unimportant (in
desktop 5V-to-3V applications, for example) then lowpermeability iron-powder cores, such as the
Micrometals type found in Pulse Engineering’s 2.1µH
PE-53680, may be acceptable even at 300kHz. For
high-current applications, shielded core geometries
(such as toroidal or pot core) help keep noise, EMI, and
switching-waveform jitter low.
Current-Sense Resistor Value
The current-sense resistor value is calculated according to the worst-case-low current-limit threshold voltage
(from the Electrical Characteristics table) and the peak
inductor current. The continuous-mode peak inductorcurrent calculations that follow are also useful for sizing
the switches and specifying the inductor-current saturation ratings. In order to simplify the calculation, ILOAD
20
may be used in place of IPEAK if the inductor value has
been set for LIR = 0.3 or less (high inductor values)
and 300kHz operation is selected. Low-inductance
resistors, such as surface-mount metal-film resistors,
are preferred.
80mV
RSENSE = ————
IPEAK
Input Capacitor Value
Place a small ceramic capacitor (0.1µF) between V+
and GND, close to the device. Also, connect a low-ESR
bulk capacitor directly to the drain of the high-side
MOSFET. Select the bulk input filter capacitor according to input ripple-current requirements and voltage rating, rather than capacitor value. Electrolytic capacitors
that have low enough ESR to meet the ripple-current
requirement invariably have more than adequate
capacitance values. Aluminum-electrolytic capacitors
such as Sanyo OS-CON or Nichicon PL are preferred
over tantalum types, which could cause power-up
surge-current failure, especially when connecting to
robust AC adapters or low-impedance batteries. RMS
input ripple current is determined by the input voltage
and load current, with the worst possible case occurring at VIN = 2 x VOUT:
————————
√VOUT (VIN - VOUT)
IRMS = ILOAD x ——————————
VIN
IRMS = ILOAD / 2 when VIN is 2 x VOUT
Output Filter Capacitor Value
The output filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance
requirements for loop stability. In other words, the lowESR electrolytic capacitor that meets the ESR requirement usually has more output capacitance than is
required for AC stability. Use only specialized low-ESR
capacitors intended for switching-regulator applications,
such as AVX TPS, Sprague 595D, Sanyo OS-CON, or
Nichicon PL series. To ensure stability, the capacitor
must meet both minimum capacitance and maximum
ESR values as given in the following equations:
VREF (1 + VOUT / VIN(MIN))
CF > ––––––––––––––––———–––
VOUT x RSENSE x f
RSENSE x VOUT
RESR < ————————
VREF
(can be multiplied by 1.5, see note below)
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
There is no well-defined boundary between stable and
unstable operation. As phase margin is reduced, the
first symptom is a bit of timing jitter, which shows up as
blurred edges in the switching waveforms where the
scope won’t quite sync up. Technically speaking, this
(usually) harmless jitter is unstable operation, since the
switching frequency is now non-constant. As the
capacitor quality is reduced, the jitter becomes more
pronounced and the load-transient output voltage
waveform starts looking ragged at the edges.
Eventually, the load-transient waveform has enough
ringing on it that the peak noise levels exceed the
allowable output voltage tolerance. Note that even with
zero phase margin and gross instability present, the
output voltage noise never gets much worse than IPEAK
x RESR (under constant loads, at least).
Designers of RF communicators or other noise-sensitive analog equipment should be conservative and
stick to the guidelines. Designers of notebook computers and similar commercial-temperature-range digital
systems can multiply the RESR value by a factor of 1.5
without hurting stability or transient response.
The output voltage ripple is usually dominated by the
ESR of the filter capacitor and can be approximated as
IRIPPLE x RESR. There is also a capacitive term, so the
full equation for ripple in the continuous mode is
VNOISE(p-p) = IRIPPLE x (RESR + 1 / (2 x pi x f x CF)). In
idle mode, the inductor current becomes discontinuous
with high peaks and widely spaced pulses, so the
noise can actually be higher at light load compared to
full load. In idle mode, the output ripple can be calculated as:
0.02 x RESR
VNOISE(p-p) = —————— +
RSENSE
0.0003 x L x [1 / VOUT + 1 / (VIN - VOUT)]
———————————————————
(RSENSE)2 x CF
Transformer Design
(MAX796/MAX799 Only)
Buck-plus-flyback applications, sometimes called “coupled-inductor” topologies, need a transformer in order to
generate multiple output voltages. The basic electrical
design is a simple task of calculating turns ratios and
adding the power delivered to the secondary in order to
calculate the current-sense resistor and primary inductance. However, extremes of low input-output differentials, widely different output loading levels, and high turns
ratios can complicate the design due to parasitic transformer parameters such as inter-winding capacitance,
secondary resistance, and leakage inductance. For
examples of what is possible with real-world transformers,
see the graphs of Maximum Secondary Current vs. Input
Voltage in the Typical Operating Characteristics.
Power from the main and secondary outputs is lumped
together to obtain an equivalent current referred to the
main output voltage (see Inductor L1 for definitions of
parameters). Set the value of the current-sense resistor
at 80mV / ITOTAL.
PTOTAL = the sum of the output power from all outputs
ITOTAL = PTOTAL / VOUT = the equivalent output current referred to VOUT
VOUT (VIN(MAX) - VOUT)
L(primary) = —————————————
VIN(MAX) x f x ITOTAL x LIR
VSEC + VFWD
Turns Ratio N = ——————————————
VOUT(MIN) + VRECT + VSENSE
where: VSEC is the minimum required rectified secondary-output voltage
VFWD is the forward drop across the secondary
rectifier
VOUT(MIN) is the minimum value of the main
output voltage (from the Electrical
Characteristics)
VRECT is the on-state voltage drop across the
synchronous-rectifier MOSFET
VSENSE is the voltage drop across the sense
resistor
In positive-output (MAX796) applications, the transformer secondary return is often referred to the main
output voltage rather than to ground in order to reduce
the needed turns ratio. In this case, the main output
voltage must first be subtracted from the secondary
voltage to obtain VSEC.
______________________________________________________________________________________
21
MAX796/MAX797/MAX799
These equations are “worst-case” with 45 degrees of
phase margin to ensure jitter-free fixed-frequency operation and provide a nicely damped output response for
zero to full-load step changes. Some cost-conscious
designers may wish to bend these rules by using less
expensive (lower quality) capacitors, particularly if the
load lacks large step changes. This practice is tolerable,
provided that some bench testing over temperature is
done to verify acceptable noise and transient response.
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
______Selecting Other Components
MOSFET Switches
The two high-current N-channel MOSFETs must be
logic-level types with guaranteed on-resistance specifications at VGS = 4.5V. Lower gate threshold specs are
better (i.e., 2V max rather than 3V max). Drain-source
breakdown voltage ratings must at least equal the maximum input voltage, preferably with a 20% derating factor. The best MOSFETs will have the lowest
on-resistance per nanocoulomb of gate charge.
Multiplying RDS(ON) x QG provides a meaningful figure
by which to compare various MOSFETs. Newer MOSFET process technologies with dense cell structures
generally give the best performance. The internal gate
drivers can tolerate >100nC total gate charge, but
70nC is a more practical upper limit to maintain best
switching times.
In high-current applications, MOSFET package power
dissipation often becomes a dominant design factor.
I2R power losses are the greatest heat contributor for
both high- and low-side MOSFETs. I2R losses are distributed between Q1 and Q2 according to duty factor
(see the equations below). Switching losses affect the
upper MOSFET only, since the Schottky rectifier clamps
the switching node before the synchronous rectifier
turns on. Gate-charge losses are dissipated by the driver- er and don’t heat the MOSFET. Ensure that both
MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the
temperature rise according to package thermal-resistance specifications. The worst-case dissipation for the
high-side MOSFET occurs at the minimum battery voltage, and the worst-case for the low-side MOSFET
occurs at the maximum battery voltage.
PD (upper FET) = ILOAD2 x RDS(ON) x DUTY
(
VIN x CRSS
+ VIN x ILOAD x f x ––––––––––– +20ns
IGATE
)
PD (lower FET) = ILOAD2 x RDS(ON) x (1 - DUTY)
DUTY = (VOUT + VQ2) / (VIN - VQ1)
where: On-state voltage drop VQ_ = ILOAD x RDS(ON)
CRSS = MOSFET reverse transfer capacitance
IGATE = DH driver peak output current capability
(1A typically)
20ns = DH driver inherent rise/fall time
Under output short circuit, the synchronous-rectifier
MOSFET suffers extra stress and may need to be oversized if a continuous DC short circuit must be tolerated.
22
During short circuit, Q2’s duty factor can increase to
greater than 0.9 according to:
Q2 DUTY (short circuit) = 1 - [VQ2 / (VIN(MAX) - VQ1)]
where the on-state voltage drop VQ = (120mV / RSENSE)
x RDS(ON).
Rectifier Diode D1
Rectifier D1 is a clamp that catches the negative inductor swing during the 110ns dead time between turning
off the high-side MOSFET and turning on the low-side.
D1 must be a Schottky type in order to prevent the
lossy parasitic MOSFET body diode from conducting. It
is acceptable to omit D1 and let the body diode clamp
the negative inductor swing, but efficiency will drop one
or two percent as a result. Use an MBR0530 (500mA
rated) type for loads up to 1.5A, a 1N5819 type for
loads up to 3A, or a 1N5822 type for loads up to 10A.
D1’s rated reverse breakdown voltage must be at least
equal to the maximum input voltage, preferably with a
20% derating factor.
Boost-Supply Diode D2
A signal diode such as a 1N4148 works well for D2 in
most applications. If the input voltage can go below 6V,
use a small (20mA) Schottky diode for slightly improved
efficiency and dropout characteristics. Don’t use large
power diodes such as 1N5817 or 1N4001, since high
junction capacitance can cause VL to be pumped up to
excessive voltages.
Rectifier Diode D3
(Transformer Secondary Diode)
The secondary diode in coupled-inductor applications
must withstand high flyback voltages greater than 60V,
which usually rules out most Schottky rectifiers.
Common silicon rectifiers such as the 1N4001 are also
prohibited, as they are far too slow. This often makes
fast silicon rectifiers such as the MURS120 the only
choice. The flyback voltage across the rectifier is related to the VIN-VOUT difference according to the transformer turns ratio:
VFLYBACK = VSEC + (VIN - VOUT) x N
where: N is the transformer turns ratio SEC/PRI
VSEC is the maximum secondary DC output voltage
VOUT is the primary (main) output voltage
Subtract the main output voltage (VOUT) from VFLYBACK
in this equation if the secondary winding is returned to
VOUT and not to ground. The diode reverse breakdown
rating must also accommodate any ringing due to leakage inductance. D3’s current rating should be at least
twice the DC load current on the secondary output.
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
Low input voltages and low input-output differential voltages each require some extra care in the design. Low
absolute input voltages can cause the VL linear regulator
to enter dropout, and eventually shut itself off. Low input
voltages relative to the output (low VIN-VOUT differential)
can cause bad load regulation in multi-output flyback
applications. See the design equations in the Transformer
Design section. Finally, low VIN-VOUT differentials can also
cause the output voltage to sag when the load current
changes abruptly. The amplitude of the sag is a function
of inductor value and maximum duty factor (an Electrical
Characteristics parameter, 93% guaranteed over temperature at f = 150kHz) as follows:
(ISTEP)2 x L
VSAG = ———————————————
2 x CF x (VIN(MIN) x DMAX - VOUT)
The cure for low-voltage sag is to increase the value of
the output capacitor. For example, at VIN = 5.5V, VOUT
= 5V, L = 10µH, f = 150kHz, a total capacitance of
660µF will prevent excessive sag. Note that only the
capacitance requirement is increased and the ESR
requirements don’t change. Therefore, the added
capacitance can be supplied by a low-cost bulk
capacitor in parallel with the normal low-ESR capacitor.
__________Applications Information
Heavy-Load Efficiency Considerations
The major efficiency loss mechanisms under loads are,
in the usual order of importance:
• P(I2R), I2R losses
• P(gate), gate-charge losses
• P(diode), diode-conduction losses
• P(tran), transition losses
• P(cap), capacitor ESR losses
• P(IC), losses due to the operating supply current
of the IC
Inductor-core losses are fairly low at heavy loads
because the inductor’s AC current component is small.
Therefore, they aren’t accounted for in this analysis.
Ferrite cores are preferred, especially at 300kHz, but
powdered cores such as Kool-mu can work well.
Efficiency = POUT / PIN x 100%
= POUT / (POUT + PTOTAL) x 100%
PTOTAL = P(I2R) + P(gate) + P(diode) + P(tran) +
P(cap) + P(IC)
P(I2R) = (ILOAD)2 x (RDC + RDS(ON) + RSENSE)
where RDC is the DC resistance of the coil, RDS(ON) is
the MOSFET on-resistance, and RSENSE is the current-
Table 4. Low-Voltage Troubleshooting
SYMPTOM
CONDITION
ROOT CAUSE
SOLUTION
Sag or droop in VOUT
under step load change
Low VIN-VOUT differential, Limited inductor-current slew
<1.5V
rate per cycle.
Increase bulk output capacitance per
formula above. Reduce inductor value.
Dropout voltage is too
high (VOUT follows VIN as
VIN decreases)
Low VIN-VOUT differential, Maximum duty-cycle limits
<1V
exceeded.
Reduce f to 150kHz. Reduce MOSFET
on-resistance and coil DCR.
Unstable—jitters between
two distinct duty factors
Inherent limitation of fixed-freLow VIN-VOUT differential,
quency current-mode SMPS
<1V
slope compensation.
Reduce L value. Tolerate the remaining
jitter (extra output capacitance helps
somewhat).
Secondary output won’t
support a load
Not enough duty cycle left to
Reduce f to 150kHz. Reduce secondary
Low VIN-VOUT differential,
initiate forward-mode operation.
impedances—use Schottky if possible.
VIN < 1.3 x VOUT(main)
Small AC current in primary can’t
Stack secondary winding on main output.
(MAX796/MAX799 only)
store energy for flyback operation.
High supply current,
poor efficiency
Low input voltage, <5V
VL linear regulator is going into
dropout and isn’t providing
good gate-drive levels.
Won’t start under load or
quits before battery is
completely dead
Low input voltage, <4.5V
VL output is so low that it hits the Supply VL from an external source other
VL UVLO threshold at 4.2V max. than VBATT, such as the system 5V supply.
Use a small 20mA Schottky diode for
boost diode D2. Supply VL from an
external source.
______________________________________________________________________________________
23
MAX796/MAX797/MAX799
____________Low-Voltage Operation
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
sense resistor value. The RDS(ON) term assumes identical MOSFETs for the high- and low-side switches
because they time-share the inductor current. If the
MOSFETs aren’t identical, their losses can be estimated by averaging the losses according to duty factor.
P(gate) = gate-driver loss = qG x f x VL
where VL is the MAX796 internal logic supply voltage
(5V), and qG is the sum of the gate-charge values for
low- and high-side switches. For matched MOSFETs,
qG is twice the data sheet value of an individual MOSFET. If VOUT is set to less than 4.5V, replace VL in this
equation with VBATT. In this case, efficiency can be
improved by connecting VL to an efficient 5V source,
such as the system +5V supply.
P(diode) = diode conduction losses
= ILOAD x VFWD x tD x f
where tD is the diode conduction time (110ns typ) and
VFWD is the forward voltage of the Schottky.
PD(tran) = transition loss =
VBATT x CRSS
VBATT x ILOAD x f x (——————— + 20ns)
IGATE
where CRSS is the reverse transfer capacitance of the
high-side MOSFET (a data sheet parameter), IGATE is
the DH gate-driver peak output current (1A typ), and
20ns is the rise/fall time of the DH driver (20ns typ).
P(cap) = input capacitor ESR loss = (IRMS)2 x RESR
where IRMS is the input ripple current as calculated in the
Input Capacitor Value section of the Design Procedure.
Light-Load Efficiency Considerations
Under light loads, the PWM operates in discontinuous
mode, where the inductor current discharges to zero at
some point during the switching cycle. This causes the
AC component of the inductor current to be high compared to the load current, which increases core losses
and I2R losses in the output filter capacitors. Obtain best
light-load efficiency by using MOSFETs with moderate
gate-charge levels and by using ferrite, MPP, or other
low-loss core material. Avoid powdered iron cores; even
Kool-mu (aluminum alloy) is not as good as ferrite.
__PC Board Layout Considerations
Good PC board layout is required to achieve specified
noise, efficiency, and stability performance. The PC
board layout artist must be provided with explicit
instructions, preferably a pencil sketch of the placement of power switching components and high-current
routing. See the evaluation kit PC board layouts in the
MAX796 and MAX797 EV kit manuals for examples. A
24
ground plane is essential for optimum performance. In
most applications, the circuit will be located on a multilayer board and full use of the four or more copper layers is recommended. Use the top layer for high-current
connections, the bottom layer for quiet connections
(REF, SS, GND), and the inner layers for an uninterrupted ground plane. Use the following step-by-step guide.
1) Place the high-power components (C1, C2, Q1, Q2,
D1, L1, and R1) first, with their grounds adjacent.
Priority 1: Minimize current-sense resistor trace
lengths (see Figure 10).
Priority 2: Minimize ground trace lengths in the
high-current paths (discussed below).
Priority 3: Minimize other trace lengths in the highcurrent paths. Use >5mm wide traces.
C1 to Q1: 10mm max length.
D1 cathode to Q2: 5mm max length
LX node (Q1 source, Q2 drain, D1 cathode, inductor): 15mm max length
Ideally, surface-mount power components are
butted up to one another with their ground terminals
almost touching. These high-current grounds (C1-,
C2-, source of Q2, anode of D1, and PGND) are
then connected to each other with a wide filled zone
of top-layer copper, so that they don’t go through
vias. The resulting top-layer “sub-ground-plane” is
connected to the normal inner-layer ground plane at
the output ground terminals. This ensures that the
analog GND of the IC is sensing at the output terminals of the supply, without interference from IR
drops and ground noise. Other high-current paths
should also be minimized, but focusing ruthlessly
on short ground and current-sense connections
eliminates about 90% of all PC layout
headaches. See the evaluation kit PC board layouts
for examples.
2) Place the IC and signal components. Keep the main
switching node (LX node) away from sensitive analog components (current-sense traces and REF and
SS capacitors). Placing the IC and analog components on the opposite side of the board from the
power-switching node is desirable. Important: the
IC must be no farther than 10mm from the currentsense resistor. Keep the gate-drive traces (DH, DL,
and BST) shorter than 20mm and route them away
from CSH, CSL, REF, and SS.
3) Employ a single-point star ground where the input
ground trace, power ground (sub-ground-plane),
and normal ground plane all meet at the output
ground terminal of the supply.
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
MAIN CURRENT PATH
SENSE RESISTOR
MAX796
MAX797
MAX799
Figure 10. Kelvin Connections for the Current-Sense Resistor
_________________________________________________________Application Circuits
VIN (6.5V TO 18V)
+15V
AT
250mA
22μF, 35V
2
7
SECFB
ON/OFF
6
C2
4.7μF
11
VL
FB
V+
DH
SHDN
BST
10
16
D1
CMPSH
-3A
210k, 1%
C2
4.7μF
Si9410
0.01μF
D2
EC11FS1
14
49.9k, 1%
C3
15μF
2.5V
18V
1/4 W
+5V
AT 3A
0.1μF
MAX796
1
LX
DL
SS
15
13
T1
15μH
2.2:1
Si9410
20mΩ
1N5819
220μF
6.3V
PGND 12
0.01μF
(OPTIONAL)
4
GND
CSH
CSL
SYNC
5
8
22Ω*
9
4700pF*
REF
T1 = TRANSPOWER TTI5870
* = OPTIONAL, MAY NOT BE NEEDED
3
0.33μF
Figure 11. +5V/+15V Dual-Output Application (MAX796)
______________________________________________________________________________________
25
MAX796/MAX797/MAX799
FAT, HIGH-CURRENT TRACES
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
____________________________________________Application Circuits (continued)
33μF, 35V
VIN (8V TO 18V AS SHOWN)
102k, 1%
100k, 1%
10
2
V+
11
1N4148
VL
SECFB
BST
ON/OFF
6
SHDN
DH
LX
4.7μF
14
16
T1
1:1.5
Q1
0.1μF
15
+5V
AT
500mA
MBR0502L
47μF
10μH
+3.3V
AT 2A
25mΩ
1N5819
MAX796
DL
PGND
13
Q2
Q3
330μF
12
1N5817
1
SS
CSH
CSL
0.01μF
(OPTIONAL)
GND
SYNC
REF
4
3
FB
102k
1%
8
33.2k
1%
9
7
T1 = TDK 1:1.5 TRANSFORMER
PC40EEM 12.7/13.7 - A160 CORE
BEM 12.7/13.7 BOBBIN
PRIMARY = 8 TURNS 24 AWG
SECONDARY = 12 TURNS 24 AWG
DESIGN FOR TIGHT MAGNETIC COUPLING
Q1-Q2 = Si9410 or EQUIVALENT
Q3 = Si9955 or EQUIVALENT (50V)
5
0.33μF
49.9k
1%
Figure 12. +3.3V/+5V Dual-Output Application (MAX796)
VIN (9V TO 18V)
22μF, 35V
107k, 1%
1000pF
221k, 1%
1μF
3
11
REF
5
2
SECFB
VL
SYNC
-5.5V OUT
(-5.5V AT 200mA)
1N4148
V+
DH
BST
EQ11FS1
10
4.7μF
22μF
10V
16
1/2
Si9936
14
0.1μF
LX
MAX799
ON/OFF
6
DL
PGND
SHDN
CSH
CSL
GND
4
SS
FB
15
13
1/2 Si9936
1N5819
T1
15μH
1:1.3
50mΩ
12
8
9
7
T1 = TRANSPOWER TTI5926
1
0.01μF
(OPTIONAL)
Figure 13. ±5V Dual-Output Application (MAX799)
26
______________________________________________________________________________________
+5V OUT
(+5V AT 1A)
220μF
10V
Step-Down Controllers with
Synchronous Rectifier for CPU Power
MAX797
INPUT
4.5V
TO 30V
V+
STANDARD 3.3V
CIRCUIT
+3.3V
MAIN OUTPUT
MAIN
3.3V
OUTPUT
(CSL)
REF
(2.505V)
VL (5V)
82pF
1k
Q1
Si9433DY
OR MMSF4P01
MAX473
100k, 1%
1.5k
+2.9V OUTPUT
AT 2A
20pF
16k, 1%
10μF
10μF
SANYO OS-CON
Figure 14. 2.9V Low-Dropout Linear Regulator with Fast Transient Response
0.033Ω
VIN
2.5V TO 5.25V
CSH
C1
100μF
L1
5μH
CSL
+5V AT 1A
DL
REF
Q1
DH
0.33μF
GND
MAX797
LX
PGND
SKIP
D1
C2
C3
100μF 100μF
V+
BST
4.7μF
0.1μF
SHDN
VL
100k
FB
SYNC
100k
33k
1N4148
1N4148
2N7002
0.01μF
+3.3V
(EXTERNAL)
190kHz - 340kHz
L1 = SUMIDA CDRH125, 5μH
D1 = MOTOROLA MBR130
C1 - C3 = AVX TPS 100μF, 10V
Q1 = SILICONIX Si9936 (BOTH SECTIONS)
OR MOTOROLA MMDF3N03L
OPTIONAL SYNC AND LOW-VOLTAGE
START-UP CIRCUIT
Figure 15. Low-Noise Boost Converter for Cellular Phones
______________________________________________________________________________________
27
MAX796/MAX797/MAX799
____________________________________________Application Circuits (continued)
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
____________________________________________Application Circuits (continued)
0.01Ω
VIN
4.75V TO 6V
CSH
SYNC
C1
220μF
+12V AT 2A
D1
Q1
DH
REF
LX
MAX797
0.33μF
L1
5μH
CSL
GND
PGND
C2
C3
150μF 150μF
SKIP
V+
191k
SHDN
BST
4.7μF
FB
VL
SS
49.9k
L1 = 2x SUMIDA CDRH125-100 IN PARALLEL
D1 = MOTOROLA MBR640
Q1 = MOTOROLA MTD20N03HDL
C1 = SANYO OS-CON 220μF, 10V
C2, C3 = SANYO OS-CON 150μF, 16V
0.01μF
Figure 16. 5V-to-12V PWM Boost Converter
INPUT
3V TO 6.5V
OUTPUT
+5V AT 500mA
33mΩ
CMPSH-3A
T1
CSH
CSL
100μF
BST
220μF
LX
4.7μF
MAX797
DL
Q2
PGND
HI EFF
LOW IQ
220μF
Q1
DH
VL
SKIP
V+
SHDN
SYNC
REF
GND
200k
FB
200k
0.33μF
Q1, Q2 = Si9410DY
T1 = COILTRONIX CTX 10-4
10μH PRIMARY, 1:1
START-UP SUPPLY VOLTAGE = 3.5V TYP
Figure 17. 90% Efficient, Low-Voltage PWM Flyback Converter (4 Cells to 5V)
28
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
INPUT
V+
VL
SHDN
4.7μF
BST
Q1
DH
OUTPUT
3.3V ±1.8%
L1
LX
SKIP
MAX797
RSENSE
Q2
DL
REMOTE
SENSE
POINT
PGND
CSH
SS
0.01μF
CSL
FB
GND
SYNC
REF
51k
5%
R1
VOUT = VREF 1 + –––
R2
ADJUST RANGE = 2.5V TO 4V AS SHOWN.
(
)
R1
63.4k
0.1%
200k
5%
R2
200k
0.1%
TO
VL
0.33μF
OMIT R2 FOR VOUT = 2.5V.
51k
5%
1000pF
USE EXTERNAL REFERENCE
(MAX872) FOR BETTER ACCURACY.
10k
MAX495
Figure 18. High-Accuracy Adjustable-Output Application
INPUT
4.5V TO 25V
V+
FB
1N4148
VL
BST
0.1μF
4.7μF
Si9410
22μF
22μF
DH
SHDN
1N5819
-5V AT 1.5A
LX
L1
MAX797
Si9410
CSH
150μF
150μF
0.025Ω
CSL
DL
GND
PGND
SYNC
REF
SKIP
0.33μF
L1 = DALE LPE6562-A093
Figure 19. Negative-Output (Inverting Topology) Power Supply
______________________________________________________________________________________
29
MAX796/MAX797/MAX799
____________________________________________Application Circuits (continued)
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
____________________________________________Application Circuits (continued)
INPUT
0.1μF
1N4148
V+
VL
4.7μF
C1
2x 22μF
BST
Q1
DH
SHDN
T1
0.1μF
10μH
LX
MAX797
+5V OUTPUT
AT 3A
Q2
DL
PGND
D1
1N5819
100k
1%
1N4148
100k
1%
C2
220μF
SS
FB
CSH
0.01μF
SKIP
GND
1.91Ω, 1%
CSL
SYNC
REF
T1 = 1:70 5mm SURFACE-MOUNT TRANSFORMER
DALE LPE-3325-A087
Q1, Q2 = MMSF5N03 OR Si9410DY
0.33μF
Figure 20. Buck Converter with Low-Loss SMT Current-Sense Transformer
INPUT
4.75V
TO 5.5V
C1
220μF
OS-CON
0.1μF
D1
V+
VL
N1
DH
C3
0.1μF
LX
ON/OFF
N2
DL
SHDN
MAX797
4.7μF
BST
L1
3.3μH
N1 = N2 = MTD20N03HDL
L1 = COILCRAFT DO3316-332
R1
12mΩ
1.5V OUTPUT
AT 5A
C2
2 x 220μF
OS-CON
D2
1N5820
PGND
CSH
SS
C6
0.01μF
CSL
R6
49.9k
FB
C7
330pF
SYNC
R5
150k
R7
124k
REF
R3
66.5k
1%
SKIP GND
C5
0.33μF
R4
100k
1%
TO
VL
MAX495
REMOTE SENSE LINE
Figure 21. 1.5V GTL Bus Termination Supply
30
______________________________________________________________________________________
Step-Down Controllers with
Synchronous Rectifier for CPU Power
10
6
VIN
10.5V to
28V
V+
VL
SHDN
2X
22μF
35V
BST
SKIP
DH
LX
MAX797
CSH
REF
1
SS
CSL
0.01μF
DL
5
SYNC
FB
7
PGND
GND
4
11
14
2
D1
0.01μF
4.7μF
16
Q1
15
D3
8
T1
3
L1
10μH
1.7Ω
9
13
Q2
3X
100μF
16V
D2
IOUT
2.5A
12
0.025Ω
0.33μF
6
1.0k
MAX495
4
D1, D3 CENTRAL SEMI. CMPSH-3
D2 NIEC EC10QS02L, SCHOTTKY RECT.
L1 DALE IHSM-4825 10μH 15%
T1 DALE LPE-3325-A087, CURRENT TRANSFORMER, 1:70
Q1, Q2 MOTOROLA MMSF5N03HD
3
7
0.1μF
0.33μF
2
39k
Figure 22. Battery-Charger Current Source
______________________________________________________________________________________
31
MAX796/MAX797/MAX799
____________________________________________Application Circuits (continued)
MAX796/MAX797/MAX799
Step-Down Controllers with
Synchronous Rectifier for CPU Power
_Ordering Information (continued)
PART
TEMP RANGE
PIN-PACKAGE
0°C to +70°C
16 Plastic DIP
MAX797CPE+
0°C to +70°C
16 Plastic DIP
MAX797CSE
0°C to +70°C
16 Narrow SO
MAX797CPE
MAX797CSE+
0°C to +70°C
16 Narrow SO
MAX797C/D
0°C to +70°C
Dice*
MAX797C/D+
0°C to +70°C
Dice*
MAX797EPE
-40°C to +85°C
16 Plastic DIP
MAX797EPE+
-40°C to +85°C
16 Plastic DIP
MAX797ESE
-40°C to +85°C
16 Narrow SO
MAX797ESE+
-40°C to +85°C
16 Narrow SO
MAX797MJE
-55°C to +125°C
16 CERDIP
MAX797MJE+
-55°C to +125°C
16 CERDIP
MAX799CPE
0°C to +70°C
16 Plastic DIP
MAX799CSE
0°C to +70°C
16 Narrow SO
MAX799C/D
0°C to +70°C
Dice*
MAX799EPE
-40°C to +85°C
16 Plastic DIP
MAX799ESE
-40°C to +85°C
16 Narrow SO
MAX799MJE
-55°C to +125°C
*Contact factory for dice specifications.
16 CERDIP
___________________Chip Topography
SS
DH
LX
SKIP
(SECFB)
BST
REF
DL
GND
PGND
0.16O"
(4.064mm)
SYNC
VL
SHDN
V+
FB
CSH
CSL
0.085"
(2.159mm)
( ) ARE FOR MAX796/MAX799 ONLY.
TRANSISTOR COUNT: 913
SUBSTRATE CONNECTED TO GND
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
32 __________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 (408) 737-7600
© 2005 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products, Inc.
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