TPS65140

TPS65140
TPS65140, TPS65141
TPS65145
www.ti.com
SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
TRIPLE OUTPUT LCD SUPPLY WITH LINEAR REGULATOR AND POWER GOOD
FEATURES
•
•
•
•
•
•
•
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•
•
•
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•
•
DESCRIPTION
2.7-V to 5.8-V Input Voltage Range
1.6-MHz Fixed Switching Frequency
3 Independent Adjustable Outputs
Main Output up to 15 V With <1% Typical
Output Voltage Accuracy
Virtual Synchronous Converter Technology
Negative Regulated Charge Pump Driver VO2
Positive Charge Pump Converter VO3
Auxiliary 3.3-V Linear Regulator Controller
Internal Soft Start
Internal Power-On Sequencing
Fault Detection of all Outputs (TPS65140/45)
No Fault Detection (TPS65141)
Thermal Shutdown
System Power Good
Available in TSSOP-24 and QFN-24
PowerPAD™ Packages
The main output VO1 is a 1.6-MHz fixed frequency
PWM boost converter providing the source drive
voltage for the LCD display. The device is available
in two versions with different internal switch current
limits to allow the use of a smaller external inductor
when lower output power is required. The
TPS65140/41 has a typical switch current limit of 2.3
A and the TPS65145 has a typical switch current
limit of 1.37 A. A fully integrated adjustable charge
pump doubler/tripler provides the positive LCD gate
drive voltage. An externally adjustable negative
charge pump provides the negative gate drive
voltage. Due to the high 1.6-MHz switching
frequency of the charge pumps, inexpensive and
small 220-nF capacitors can be used.
Additionally, the TPS6514x series has a system
power good output to indicate when all supply rails
are acceptable. For LCD panels powered by 5 V the
device has a linear regulator controller using an
external transistor to provide a regulated 3.3 V
output for the digital circuits. For maximum safety,
the TPS65140/45 goes into shutdown as soon as
one of the outputs is out of regulation. The device
can be enabled again by toggling the input or the
enable (EN) pin to GND. The TPS65141 does not
enter shutdown when one of its outputs is below its
power good threshold.
APPLICATIONS
•
•
•
•
•
•
The TPS6514x series offers a compact and small
power supply solution to provide all three voltages
required by thin film transistor (TFT) LCD displays.
The auxiliary linear regulator controller can be used
to generate a 3.3-V logic power rail for systems
powered by a 5-V supply rail only.
TFT LCD Displays for Notebooks
TFT LCD Displays for Monitors
Portable DVD Players
Tablet PCs
Car Navigation Systems
Industrial Displays
TPS6514x
Vin
2.7 V to 5.8 V
Boost
Converter
Vo1
Up to 15 V / 400 mA
Positive Charge
Pump
Vo3
Up to 30 V / 20 mA
Negative
Charge Pump
Vo2
Up to 12 V / 20 mA
Vcom Buffer
Vcom
Linear Regulator
Controller
Vo4
3.3 V
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2003–2006, Texas Instruments Incorporated
TPS65140, TPS65141
TPS65145
www.ti.com
SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
TYPICAL APPLICATION CIRCUIT
VI
2.7 V to 5.8 V
L1
4.2 µH
C3
22 µF
TPS65140
VIN
C13
10 nF
C1
VO2
Up to 12 V/20 mA
C6
0.22 µF
D2
C12
0.22 µF
0.22 µF
D3
R3
C5
C4
22 µF
R1
SW
COMP
SW
GND
FB1
EN
ENR
SUP
C1+
C1−
C2−/MODE
DRV
FB3
C1
C2+
R2
0.22 µF
VO3
Up to 30 V/20 mA
OUT3
FB2
PG
REF
FB4
PGND
BASE
VO1
Up to 15 V/350 mA
D1
C7
0.22 µF
R5
PGND
GND
R4
R6
Q1
BCP68
C11
100 nF
VI
VO4
3.3 V
VI
R7
33 kΩ
C9
4.7 µF
C9
1 µF
System Power
Good
ORDERING INFORMATION
TA
-40°C to
85°C
(1)
(2)
2
PACKAGE (1) (2)
LINEAR REGULATOR
OUTPUT VOLTAGE
MINIMUM SWITCH
CURRENT LIMIT
TSSOP
QFN
PACKAGE
MARKING
3.3 V
1.6 A
TPS65140PWP
TPS65140RGE
TPS65140
3.3 V
1.6 A
TPS65141PWP
TPS65141RGE
TPS65141
3.3 V
0.96 A
TPS65145PWP
TPS65145RGE
TPS65145
The PWP and RGE packages are available taped and reeled. Add an R suffix to the device type (TPS65100PWPR) to order the device
taped and reeled. The PWPR package has quantities of 2000 devices per reel, and the the RGER package has 3000 devices per reel.
Without the suffix, the PWP package only, is shipped in tubes with 60 devices per tube.
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
Web site at www.ti.com.
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TPS65140, TPS65141
TPS65145
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
UNIT
Voltages on pin VIN (2)
-0.3 V to 6 V
(2)
-0.3 V to 15.5 V
ENR (2)
-0.3 V to VI + 0.3 V
Voltages on pin VO1, SUP, PG
Voltages on pin EN, MODE,
Voltage on pin SW (2)
20 V
Power good maximum sink current (PG)
1 mA
Continuous power dissipation
See Dissipation Rating Table
Operating junction temperature range
-40°C to 150°C
Storage temperature range
-65°C to 150°C
Lead temperature (soldering, 10 sec)
(1)
(2)
260°C
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating
conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltage values are with respect to network ground terminal.
DISSIPATION RATINGS
PACKAGE
RΘJA
TA ≤ 25°C
POWER RATING
TA = 70°C
POWER RATING
TA = 85°C
POWER RATING
24-Pin TSSOP
30.13 C°/W (PWP soldered)
3.3 W
1.83 W
1.32 W
24-Pin QFN
30 C°/W
3.3 W
1.8 W
1.3 W
RECOMMENDED OPERATING CONDITIONS
MIN
TYP
MAX
VIN
Input voltage range
L
Inductor (1)
TA
Operating ambient temperature
-40
85
°C
TJ
Operating junction temperature
-40
125
°C
(1)
2.7
UNIT
5.8
V
µH
4.7
See the application information section for further information.
ELECTRICAL CHARACTERISTICS
Vin = 3.3 V, EN = VIN, VO1 = 10 V, TA= -40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY CURRENT
Vi
Input voltage range
IQ
Quiescent current into VIN
IQCharge
Charge pump quiescent
current into SUP
IQEN
2.7
5.5
V
mA
ENR = GND, VO3 = 2 x VO1,
Boost converter not switching
0.7
0.9
VO1 = SUP = 10 V, VO3 = 2 x VO1
1.7
2.7
VO1 = SUP = 10 V, VO3 = 3 x VO1
3.9
6
LDO controller quiescent
current into Vin
ENR = VIN, EN = GND
300
800
µA
ISD
Shutdown current into VIN
EN = ENR = GND
1
10
µA
VUVLO
Undervoltage lockout
threshold
VI falling
2.2
2.4
V
Thermal shutdown
Temperature rising
160
mA
°C
LOGIC SIGNALS EN, ENR
VIH
High level input voltage
VIL
Low level input voltage
II
Input leakage current
1.5
EN = GND or VIN
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V
0.01
0.4
V
0.1
µA
3
TPS65140, TPS65141
TPS65145
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
ELECTRICAL CHARACTERISTICS (continued)
Vin = 3.3 V, EN = VIN, VO1 = 10 V, TA= -40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
MAIN BOOST CONVERTER
VO1
Output voltage range
5
VO1-Vin
Minimum input to output
voltage difference
1
VREF
Reference voltage
1.205
1.13
1.219
V
VFB
Feedback regulation
voltage
1.136
1.146
1.154
V
IFB
Feedback input bias
current
10
100
nA
rDS(on)
N-MOSFET on-resistance
(Q1)
VO1 = 10 V, Isw = 500 mA
195
290
VO1 = 5 V, Isw = 500 mA
285
420
ILIM
N-MOSFET switch current
limit (Q1)
TPS65140, TPS65141
rDS(on)
P-MOSFET on-resistance
(Q2)
IMAX
Maximum P-MOSFET peak
switch current
Ileak
Switch leakage current
fSW
Oscillator frequency
15
V
V
mΩ
1.6
2.3
2.6
A
0.96
1.37
1.56
A
VO1 = 10 V, Isw = 100 mA
9
15
VO1 = 5 V, Isw = 100 mA
14
22
TPS65145
Vsw = 15 V
1
A
µA
1
10
0°C ≤ TA ≤ 85°C
1.295
1.6
2.1
-40°C ≤ TA ≤ 85°C
1.191
1.6
2.1
Line regulation
2.7 V ≤ VI ≤ 5.7 V; Iload = 100 mA
Load regulation
0 mA ≤ IO ≤ 300 mA
Ω
MHz
0.012
%/V
0.2
%/A
NEGATIVE CHARGE PUMP VO2
VO2
Output voltage range
Vref
Reference voltage
VFB
Feedback regulation
voltage
IFB
Feedback input bias
current
rDS(on)
IO
Q8 P-Channel switch
rDS(on)
Q9 N-Channel switch
rDS(on)
-2
V
1.205
1.213
1.219
-36
0
36
mV
10
100
nA
4.3
8
2.9
4.4
Ω
IO = 20 mA
Maximum output current
V
20
mA
Line regulation
7 V ≤ VO1 ≤ 15 V, Iload =10 mA,
VO2 = -5 V
0.09
%/V
Load regulation
1 mA ≤ IO ≤ 20 mA, VO2 = -5 V
0.126
%/mA
POSITIVE CHARGE PUMP VO3
4
VO3
Output voltage range
30
V
Vref
Reference voltage
1.205
1.213
1.219
V
VFB
Feedback regulation
voltage
1.187
1.214
1.238
V
IFB
Feedback input bias
current
10
100
nA
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TPS65145
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
ELECTRICAL CHARACTERISTICS (continued)
Vin = 3.3 V, EN = VIN, VO1 = 10 V, TA= -40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
Q3 P-Channel switch
rDS(on)
rDS(on)
Q4 N-Channel switch
rDS(on)
IO
Maximum output current
9.9
15.5
1.1
1.8
4.6
8.5
1.2
2.2
610
720
UNIT
Ω
Q6 N-Channel switch
rDS(on)
D1 – D4 Shottky diode
forward voltage
MAX
IO = 20 mA
Q5 P-Channel switch
rDS(on)
Vd
TYP
ID1-D4 = 40 mA
20
mV
mA
Line regulation
10 V ≤ VO1 ≤ 15 V, Iload = 10 mA,
VO3 = 27 V
0.56
%/V
Load regulation
1 mA ≤ IO ≤ 20 mA, VO3 = 27 V
0.05
%/mA
LINEAR REGULATOR CONTROLLER VO4
VO4
4.5 V≤VI≤ 5.5 V; 10 mA ≤ IO ≤ 500 mA
Output voltage
IBASE
Vin-VO4-VBE ≥ 0.5 V (1)
Maximum base drive
current
Vin-VO4-VBE ≥ 0.75 V
3.2
3.3
13.5
19
20
27
(1)
Line regulation
4.75 V ≤ VI≤ 5.5 V, Iload = 500 mA
0.186
Load regulation
1 mA ≤ IO ≤ 500 mA, VI = 5 V
0.064
Start up current
VO4 ≤ 0.8 V
3.4
V
mA
%/V
%/A
11
20
25
mA
-12
-8.75% VO1
-6
V
-13
-9.5% VO2
-5
V
-11
-8% VO3
-5
V
SYSTEM POWER GOOD (PG)
V(PG, Vo1)
V(PG, Vo2)
Power good threshold (2)
V(PG, Vo3)
VOL
PG output low voltage
I(sink) = 500 µA
IL
PG output leakage current
VPG = 5 V
V
1
uA
0.001
With VI = supply voltage of the TPS6514x, VO4 = output voltage of the regulator, VBE = basis emitter voltage of external transistor.
The power good goes high when all 3 outputs (VO1, VO2, VO3) are above their threshold. The power good goes low as soon as one of
the outputs is below their threshold.
DEVICE INFORMATION
PWP PACKAGE
TOP VIEW
7
8
18
17
9
16
10
15
11
14
12
13
C1+
DRV
FB1
FB4
5
14
BASE
6
13
9 10 11 12
ENR
EN
7
8
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C2−/MODE
C2+
OUT3
FB3
GND
PG
SUP
6
C1−
20
19
24 23 22 21 20 19
1
18
2
17
Exposed
3
16
Thermal Die*
4
15
COMP
PGND
5
GND
21
PGND
22
4
REF
3
EN
ENR
COMP
FB2
REF
GND
DRV
C1−
C1+
C2−/MODE
C2+
OUT3
SW
23
FB2
24
2
SW
1
RGE PACKAGE
TOP VIEW
VIN
FB1
FB4
BASE
VIN
SW
SW
PGND
PGND
SUP
PG
GND
FB3
Thermal PAD*
(1)
(2)
0.3
5
TPS65140, TPS65141
TPS65145
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
DEVICE INFORMATION (continued)
Terminal Functions
TERMINAL
NO.
(PWP)
NO.
(RGE)
I/O
VIN
4
7
I
Input voltage pin of the device.
EN
24
3
I
Enable pin of the device. This pin should be terminated and not be left floating. A logic
high enables the device and a logic low shuts down the device.
COMP
22
1
PG
10
13
O
Open drain output indicating when all outputs VO1, VO2, VO3 are within 10% of their
nominal output voltage. The output goes low when one of the outputs falls below 10%
of their nominal output voltage.
ENR
23
2
I
Enable pin of the linear regulator controller. This pin should be terminated and not be
left floating. Logic high enables the regulator and a logic low puts the regulator in
shutdown.
C1+
16
19
C1-
17
20
DRV
18
21
O
External charge pump driver
FB2
21
24
I
Feedback pin of negative charge pump
REF
20
23
O
Internal reference output typically 1.23 V
FB4
2
5
I
Feedback pin of the linear regulator controller. The linear regulator controller is set to a
fixed output voltage of 3.3 V or 3 V depending on the version.
BASE
3
6
O
Base drive output for the external transistor
GND
NAME
Compensation pin for the main boost converter. A small capacitor is connected to this
pin.
Positive terminal of the charge pump flying capacitor
Negative terminal of the charge pump flying capacitor
11, 19
14, 22
PGND
7, 8
10, 11
FB3
12
15
I
Feedback pin of positive charge pump
OUT3
13
16
O
Positive charge pump output
C2-/MODE
15
18
Negative terminal of the charge pump flying capacitor and charge pump MODE pin. If
the flying capacitor is connected to this pin, the converter operates in a voltage tripler
mode. If the charge pump needs to operate in a voltage doubler mode, the flying
capacitor is removed and the C2-/MODE pin needs to be connected to GND.
C2+
14
17
Positive terminal for the charge pump flying capacitor. If the device runs in voltage
doubler mode, this pin needs to be left open.
12
I
Supply pin of the positive, negative charge pump, boost converter, and gate drive
circuit. This pin needs to be connected to the output of the main boost converter and
cannot be connected to any other voltage source. For performance reasons, it is not
recommended for a bypass capacitor to be connected directly to this pin.
SUP
9
Ground
Power ground
FB1
1
4
I
Feedback pin of the boost converter
SW
5, 6
8, 9
I
Switch pin of the boost converter
PowerPAD
™ /Thermal
Die
6
DESCRIPTION
The PowerPAD or exposed thermal die needs to be connected to power ground pins
(PGND)
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TPS65145
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
FUNCTIONAL BLOCK DIAGRAM
VIN
SW
SW
Q2
Main boost
converter
EN
FB1
FB2
FB3
Bias Vref = 1.213 V
Thermal Shutdown
Start−Up Sequencing
Undervoltage Detection
Overvoltage Detection
Short Circuit Protection
D
S
Current Limit
and
Soft Start
1.6-MHz
Oscillator
SUP
Control Logic
Gate Drive Circuit
D
Q1
COMP
S
Comparator
Sawtooth
Generator
FB1
VFB
1.146 V
SUP
FB3
SUP
(VO)
Positive
Charge Pump
GM Amplifier
Low Gain
SUP
D
VFB
1.146 V
Current
Control
S
Gain Select
(Doubler or
Tripler Mode)
D
Vref
1.214 V
Q3
C1−
S
Q4
SUP
Negative
Charge Pump
SUP
Soft Start
D
S
DRV
D
Q7
S
D
Q9
C1+
Current
Control
Soft Start
Q8
SUP
Vo3
S
D2
D
S
D1
D4
Q5
C2+
FB2
D
Vref
0V
D3
Q6
S
C2−
Reference
Output
Vref
REF
1.213 V
Soft Start
Iref = 20 mA
Short Circuit
Detect
Vref
1.213 V
Vin
System Power
Good
PG
FB1
~1 V
D
FB2
FB4
D
Linear
Regulator
Controller
S
Q10
Vref
1.213 V
Logic and
1-µs Glitch
Filter
S
FB3
ENR
BASE
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GND
GND
PGND
PGND
7
TPS65140, TPS65141
TPS65145
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
Main Boost Converter
Efficiency, main boost converter VO1
vs Load current
1
Efficiency, main boost converter VO1
vs Load current
2
Efficiency
vs Input voltage
3
fsw
Switching frequency
vs Free-air temperature
4
rDS(on)
rDS(on) N-Channel main switch Q1
vs Free-air temperature
5
η
PWM operation continuous mode
6
PWM operation, discontinuous (light load)
7
Load transient response, CO = 22 µF
8
Load transient response, CO = 2 x 22 µF
9
Power-up sequencing
10
Soft start VO1
11
Negative Charge Pump
Imax
VO2 maximum load current
vs Output voltage VO1
12
Positive Charge Pump
Imax
VO3 maximum load current
vs Output voltage VO1 (doubler mode)
13
Imax
VO3 Maximum load current
vs Output voltage VO1 (tripler mode)
14
EFFICIENCY
vs
LOAD CURRENT
100
100
90
90
Vo1 = 15 V
40
30
70
Vo1 = 10 V
60
50
Vo1 = 15 V
40
Vo1 = 6 V
90
Vo1 = 10 V
85
Vo1 = 15 V
80
30
VI = 3.3 V
Vo2, Vo3 = No Load, Switching
20
10
10
100
IL − Load Current − mA
Figure 1.
1k
75
VI = 5 V
Vo2, Vo3 = No Load, Switching
20
10
1
8
ILoad at Vo1 = 100 mA
Vo2, Vo3 = No Load, Switching
Efficiency - %
Efficiency − %
60
50
100
80
Vo1 = 10 V
70
EFFICIENCY
vs
INPUT VOLTAGE
95
Vo1 = 6 V
80
Efficiency − %
EFFICIENCY
vs
LOAD CURRENT
1
10
100
IL − Load Current − mA
Figure 2.
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1k
70
2.5
3
3.5
4
4.5
5
VI - Input V oltage - V
Figure 3.
5.5
6
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TPS65145
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
SWITCHING FREQUENCY
vs
FREE-AIR TEMPERATURE
rDS(on) N-CHANNEL MAIN SWITCH
vs
FREE-AIR TEMPERATURE
350
r
− N−Channel Main Switch − mΩ
DS(on)
Switching Frequency − MHz
1.9
1.8
VI = 2.7 V
1.7
VI = 3.3 V
1.6
VI = 5.8 V
1.5
1.4
1.3
−40
PWM OPERATION CONTINUOUS
MODE
−20
0
20
40
60
80
100
VSW
10 V/div
300
Vo1 = 5 V
250
VO
50 mV/div
200
Vo1 = 10 V
150
IL
1 A/div
Vo1 = 15 V
100
−40
−20
0
20
40
60
80
VI = 3.3 V
VO = 10 V/300 mA
100
250 ns/div
TA − Free-Air Temperature − °C
TA − Free-Air Temperature − °C
Figure 4.
Figure 5.
Figure 6.
PWM OPERATION AT LIGHT LOAD
LOAD TRANSIENT RESPONSE
LOAD TRANSIENT RESPONSE
VI = 3.3 V
Vo1 = 10 V, CO= 2*22 µF
Vo1
200 mV/div
VSW
10 V/div
Vo1
100 mV/div
VO
50 mV/div
VI = 3.3 V
VO = 10 V/10 mA
IO
50 mA to 250 mA
IL
500 mA/div
VI = 3.3 V
Vo1 = 10 V, CO= 22 µF
IO
50 mA to 250 mA
100 µs/div
100 µs/div
250 ns/div
Figure 7.
Figure 8.
POWER-UP SEQUENCING
SOFT START VO1
Figure 9.
VO2 MAXIMUM LOAD CURRENT
0.20
VI = 3.3 V
VO = 10 V,
IO = 300 mA
Vo2 = −8 V
0.18
I O − Output Current − A
Vo1
5 V/div
Vo1
5 V/div
Vo2
5 V/div
Vo3
10 V/div
VI = 3.3 V
VO = 10 V,
500 µs/div
TA = −40°C
0.16
II
500
mA/div
0.14
TA = 85°C
0.12
0.10
TA = 25°C
0.08
0.06
0.04
0.02
500 µs/div
0
8.8
9.8
10.8
11.8
12.8
13.8
14.8
Vo1 − Output Voltage − V
Figure 10.
Figure 11.
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Figure 12.
9
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SLVS497C – SEPTEMBER 2003 – REVISED APRIL 2006
VO3 MAXIMUM LOAD CURRENT
VO3 MAXIMUM LOAD CURRENT
0.14
0.12
TA = −40°C
Vo3 = 18 V (Doubler Mode)
0.10
TA = −40°C
I O − Output Current − A
I O − Output Current − A
0.12
TA = 85°C
0.10
TA = 25°C
0.08
0.06
0.04
TA = 25°C
0.08
TA = 85°C
0.06
0.04
0.02
0.02
Vo3 = 28 V (Tripler Mode)
0
9
10
11
12
13
14
15
0
9
10
Vo1 − Output Voltage − V
Figure 13.
11
12
13
14
15
Vo1 − Output Voltage − V
Figure 14.
DETAILED DESCRIPTION
The TPS6514x series consists of a main boost converter operating with a fixed switching frequency of 1.6 MHz
to allow for small external components. The boost converter output voltage VO1 is also the input voltage,
connected via the pin SUP, for the positive and negative charge pump. The linear regulator controller is
independent from this system with its own enable pin. This allows the linear regulator controller to continue to
operate while the other supply rails are disabled or in shutdown due to a fault condition on one of their outputs.
Refer to the functional block diagram for more information.
Main Boost Converter
The main boost converter operates with PWM and a fixed switching frequency of 1.6 MHz. The converter uses a
unique fast response, voltage mode controller scheme with input voltage feedforward. This achieves excellent
line and load regulation (0.2% A load regulation typical) and allows the use of small external components. To
add higher flexibility to the selection of external component values, the device uses external loop compensation.
Although the boost converter looks like a nonsynchronous boost converter topology operating in discontinuous
mode at light load, the TPS6514x series maintains continuous conduction even at light load currents. This is
accoplished using the Virtual Synchronous Converter Technology for improved load transient response. This
architecture uses an external Schottky diode and an integrated MOSFET in parallel connected between SW and
SUP (see the functional block diagram). The integrated MOSFET Q2 allows the inductor current to become
negative at light load conditions. For this purpose, a small integrated P-channel MOSFET with typically 10 Ω
rDS(on) is sufficient. When the inductor current is positive, the external Schottky diode with the lower forward
voltage conducts the current. This causes the converter to operate with a fixed frequency in continuous
conduction mode over the entire load current range. This avoids the ringing on the switch pin as seen with a
standard nonsynchronous boost converter and allows a simpler compensation for the boost converter.
Power-Good Output
The TPS6514x sereis has an open-drain power-good output with a maximum sink capability of 1 mA. The
power-good output goes high as soon as the main boost converter VO1 and the negative and the positive charge
pumps are within regulation. The power-good output goes low as soon as one of the outputs is out of regulation.
In this case, the device goes into shutdown at the same time. See the electrical characteristics table for the
power-good thresholds.
Enable and Power-On Sequencing (EN, ENR)
The device has two enable pins. These pins should be terminated and not left floating to prevent faulty
operation. Pulling the enable pin (EN) high enables the device and starts the power-on sequencing with the main
boost converter VO1 coming up first, then the negative and positive charge pump. The linear regulator has an
independent enable pin (ENR). Pulling this pin low disables the regulator, and pulling this pin high enables this
regulator.
If the enable pin (EN) is pulled high, the device starts its power-on sequencing. The main boost converter starts
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up first with its soft start. If the output voltage has reached 91.25% of its output voltage, the negative charge
pump comes up next. The negative charge pump starts with a soft start and when the output voltage has
reached 91% of the nominal value, the positive charge pump comes up with the soft start. Pulling the enable pin
low shuts down the device. Dependent on load current and output capacitance, each of the outputs comes
down.
Positive Charge Pump
The TPS6514x series has a fully regulated integrated positive charge pump generating VO3. The input voltage
for the charge pump is applied to the SUP pin that is equal to the output of the main boost converter VO1. The
charge pump is capable of supplying a minimum load current of 20 mA. Higher load currents are possible
depending on the voltage difference between VO1 and VO3. See Figure 13 and Figure 14.
Negative Charge Pump
The TPS6514x sereis has a regulated negative charge pump using two external Schottky diodes. The input
voltage for the charge pump is applied to the SUP pin that is connected to the output of the main boost
converter VO1. The charge pump inverts the main boost converter output voltage and is capable of supplying a
minimum load current of 20 mA. Higher load currents are possible depending on the voltage difference between
VO1 and VO2. See Figure 12.
Linear Regulator Controller
The TPS6514x series includes a linear regulator controller to generate a 3.3-V rail which is useful when the
system is powered from a 5-V supply. The regulator is independent from the other voltage rails of the device and
has its own enable (ENR). Since most of the systems require this voltage rail to come up first it is recommended
to use a R-C delay on EN. This delays the start-up of the main boost converter which reduces the inrush current
as well.
Soft Start
The main boost converter as well as the charge pumps and linear regulator have an internal soft start. This
avoids heavy voltage drops at the input voltage rail or at the output of the main boost converter VO1 during
start-up. See Figure 10 and Figure 11. During softstart of the main boost converter VO1 the internal current limit
threshold is increased in three steps. The device starts with the first step where the current limit is set to 2/5 of
the typical current limit (2/5 of 2.3A) for 1024 clock cycles then increased to 3/5 of the current limit for 1024 clock
cycles and the 3rd step is the full current limit. The TPS65141 has an extended softstart time where each step is
2048 clock cycles.
Fault Protection
All of the outputs of the TPS65140/45 have short-circuit detection and cause the device to go into shutdown.
The TPS65141, as an exception, does not enter shutdown in case one of the outputs falls below its power good
threshold. The main boost converter has overvoltage and undervoltage protection. If the output voltage VO1 rises
above the overvoltage protection threshold of typically 5% of VO1, then the device stops switching, but remains
operational. When the output voltage falls below this threshold, the converter continues operation. When the
output voltage falls below the undervoltage protection threshold of typically 8.75% of VO1, because of a
short-circuit condition, the TPS65140/45 goes into shutdown. Because there is a direct pass from the input to
the output through the diode, the short-circuit condition remains. If this condition needs to be avoided, a fuse at
the input or an output disconnect using a single transistor and resistor is required. The negative and positive
charge pumps have an undervoltage lockout (UVLO) to protect the LCD panel of possible latch-up conditions
due to a short-circuit condition or faulty operation. When the negative output voltage is typically above 9.5% of
its output voltage (closer to ground), then the device enters shutdown. When the positive charge pump output
voltage, VO3, is below 8% typical of its output voltage, the device goes into shutdown. See the fault protection
thresholds in the electrical characteristics table. The device is enabled by toggling the enable pin (EN) below 0.4
V or by cycling the input voltage below the UVLO of 1.7 V. The linear regulator reduces the output current to 20
mA typical under a short-circuit condition when the output voltage is typically < 1 V. See the functional block
diagram. The linear regulator does not go into shutdown under a short-circuit condition.
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Thermal Shutdown
A thermal shutdown is implemented to prevent damage due to excessive heat and power dissipation. Typically,
the thermal shutdown threshold is 160°C. If this temperature is reached, the device goes into shutdown. The
device can be enabled by toggling the enable pin to low and back to high or by cycling the input voltage to GND
and back to VI again.
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APPLICATION INFORMATION
BOOST CONVERTER DESIGN PROCEDURE
The first step in the design procedure is to calculate the maximum possible output current of the main boost
converter under certain input and output voltage conditions. Below is an example for a 3.3-V to 10-V conversion:
Vin = 3.3 V, Vout = 10 V, Switch voltage drop Vsw = 0.5 V, Schottky diode forward voltage VD = 0.8 V
1. Duty cycle:
Vout V V
D
in 10 V 0.8 V 3.3 V 0.73
D
V out V V sw
10 V 0.8 V 0.5 V
D
2. Average inductor current:
I
I out 300 mA 1.11 A
L
1 0.73
1D
3. Inductor peak-to-peak ripple current:
i L
Vin Vsw D
fs L
(3.3 V 0.5 V) 0.73
304 mA
1.6 MHz 4.2 H
4. Peak switch current:
i
I
I L 1.11 A 304 mA 1.26 A
swpeak
L
2
2
The integrated switch, the inductor, and the external Schottky diode must be able to handle the peak switch
current. The calculated peak switch current has to be equal or lower than the minimum N-MOSFET switch
current limit as specified in the electrical characteristics table (1.6 A for the TPS65140/41 and 0.96 A for the
TPS65145). If the peak switch current is higher, then the converter cannot support the required load current.
This calculation must be done for the minimum input voltage where the peak switch current is highest. The
calculation includes conduction losses like switch rDS(on) (0.5 V) and diode forward drop voltage losses (0.8 V).
Additional switching losses, inductor core and winding losses, etc., require a slightly higher peak switch current
in the actual application. The above calculation still allows for a good design and component selection.
Inductor Selection
Several inductors work with the TPS6514x. Especially with the external compensation, the performance can be
adjusted to the specific application requirements. The main parameter for the inductor selection is the saturation
current of the inductor which should be higher than the peak switch current as calculated above with additional
margin to cover for heavy load transients and extreme start-up conditions. Another method is to choose the
inductor with a saturation current at least as high as the minimum switch current limit of 1.6 A for the
TPS65140/41 and 0.96 A for the TPS65145. The different switch current limits allow selection of a physically
smaller inductor when less output current is required. The second important parameter is the inductor DC
resistance. Usually, the lower the DC resistance, the higher the efficiency. However, the inductor DC resistance
is not the only parameter determining the efficiency. Especially for a boost converter where the inductor is the
energy storage element, the type and material of the inductor influences the efficiency as well. Especially at high
switching frequencies of 1.6 MHz, inductor core losses, proximity effects, and skin effects become more
important. Usually, an inductor with a larger form factor yields higher efficiency. The efficiency difference
between different inductors can vary between 2% to 10%. For the TPS6514x, inductor values between 3.3 µH
and 6.8 µH are a good choice but other values can be used as well. Possible inductors are shown in Table 1.
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APPLICATION INFORMATION (continued)
Table 1. Inductor Selection
DEVICE
TPS65140
TPS65145
INDUCTOR VALUE
COMPONENT SUPPLIER
DIMENSIONS / mm
ISAT/DCR
4.7 µH
Coilcraft DO1813P-472HC
8,89 x 6,1 x 5
2.6 A/54 mΩ
4.2 µH
Sumida CDRH5D28 4R2
5,7 x 5,7 x 3
2.2 A/23 mΩ
4.7 µH
Sumida CDC5D23 4R7
6 x 6 x 2,5
1.6 A/48 mΩ
3.3 µH
Wuerth Elektronik 744042003
4,8 x 4,8 x 2
1.8 A/65 mΩ
4.2 µH
Sumida CDRH6D12 4R2
6,5 x 6, 5 x 1,5
1.8 A/60 mΩ
3.3 µH
Sumida CDRH6D12 3R3
6,5 x 6,5 x 1,5
1.9 A/50 mΩ
3.3 µH
Sumida CDPH4D19 3R3
5,1 x 5,1 x 2
1.5 A/26 mΩ
3.3 µH
Coilcraft DO1606T-332
6,5 x 5,2 x 2
1.4 A/120 mΩ
3.3 µH
Sumida CDRH2D18/HP 3R3
3,2 x 3,2 x 2
1.45 A/69 mΩ
4.7 µH
Wuerth Elektronik 744010004
5,5 x 3,5 x 1
1 A/260 mΩ
3.3 µH
Coilcraft LPO6610-332M
6,6 x 5,5 x 1
1.3 A/160 mΩ
Output Capacitor Selection
For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low
ESR value but depending on the application, tantalum capacitors can be used as well. A 22-µF ceramic output
capacitor works for most of the applications. Higher capacitor values can be used to improve load transient
regulation. See Table 2 for the selection of the output capacitor. The output voltage ripple can be calculated as:
Ip L
I
V out out 1 I p ESR
Cout
f s Vout V V
d
in
with:
IP = Peak switch current as calculated in the previous section with ISW(peak).
L = Selected inductor value
IOUT = Normal load current
fs = Switching frequency
Vd = Rectifier diode forward voltage (typical 0.3 V)
COUT = Selected output capacitor
ESR = Output capacitor ESR value
Input Capacitor Selection
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 22-µF ceramic input capacitor
is sufficient for most of applications. For better input voltage filtering, this value can be increased. See Table 2
and the typical applications for input capacitor recommendations.
Table 2. Input and Output Capacitors Selection
CAPACITOR
VOLTAGE RATING
COMPONENT SUPPLIER
COMMENTS
22 µF/1210
16 V
Taiyo Yuden EMK325BY226MM
CO
22 µF/1206
6.3 V
Taiyo Yuden JMK316BJ226
CI
Rectifier Diode Selection
To achieve high efficiency, a Schottky diode should be used. The voltage rating should be higher than the
maximum output voltage of the converter. The average forward current should be equal to the average inductor
current of the converter. The main parameter influencing the efficiency of the converter is the forward voltage
and the reverse leakage current of the diode; both should be as low as possible. Possible diodes are: On
Semiconductor MBRM120L, Microsemi UPS120E, and Fairchild Semiconductor MBRS130L.
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Converter Loop Design and Stability
The TPS6514x converter loop can be externally compensated and allows access to the internal
transconductance error amplifier output at the COMP pin. A small feedforward capacitor across the upper
feedback resistor divider speeds up the circuit as well. To test the converter stability and load transient
performance of the converter, a load step from 50 mA to 250 mA is applied and the output voltage of the
converter is monitored. Applying load steps to the converter output is a good tool to judge the stability of such a
boost converter.
Design Procedure Quick Steps
1. Select the feedback resistor divider to set the output voltage.
2. Select the feedforward capacitor to place a zero at 50 kHz.
3. Select the compensation capacitor on pin COMP. The smaller the value, the higher the low frequency gain.
4. Use a 50-kΩ potentiometer in series to Cc and monitor Vout during load transients. Fine tune the load
transient by adjusting the potentiometer. Select a resistor value that comes closest to the potentiometer
resistor value. This needs to be done at the highest Vin and highest load current because stability is most
critical at these conditions.
Setting the Output Voltage and Selecting the Feedforward Capacitor
The output voltage is set by the external resistor divider and is calculated as:
V out 1.146 V 1 R1
R2
Across the upper resistor, a bypass capacitor is required to speed up the circuit during load transients as shown
in Figure 15.
VO1
Up to 10 V/150 mA
D1
C8
6.8 pF
SW
SW
FB1
SUP
C2+
C2−/MODE
C2
0.22 µF
R1
430 kΩ
C4
22 µF
R2
56 kΩ
Figure 15. Feedforward Capacitor
Together with R1 the bypass capacitor C8 sets a zero in the control loop at approximately 50 kHz:
1
1
C8 2 f z R1
2 50 kHz R1
A value closest to the calculated value should be used. Larger feedforward capacitor values reduce the load
regulation of the converter and cause load steps as shown in Figure 16.
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Load Step
Figure 16. Load Step Caused By A Too Large Feedforward Capacitor Value
Compensation
The regulator loop can be compensated by adjusting the external components connected to the COMP pin. The
COMP pin is connected to the output of the internal transconductance error amplifier. A typical compensation
scheme is shown in Figure 17.
CC
RC
15 kΩ
VIN
COMP
1 nF
Figure 17. Compensation Network
The compensation capacitor Cc adjusts the low frequency gain, and the resistor value adjusts the high frequency
gain. The following formula calculates at what frequency the resistor increases the high frequency gain.
1
fz 2 Cc Rc
Lower input voltages require a higher gain and a lower compensation capacitor value. A good start is Cc = 1 nF
for a 3.3-V input and Cc = 2.2 nF for a 5-V input. If the device operates over the entire input voltage range from
2.7 V to 5.8 V, a larger compensation capacitor up to 10 nF is recommended. Figure 18 shows the load transient
with a larger compensation capacitor, and Figure 19 shows a smaller compensation capacitor.
CC = 4.7 nF
Figure 18. CC = 4. 7 nF
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CC = 1 nF
Figure 19. CC = 1 nF
Lastly, Rc needs to be selected. A good practice is to use a 50-kΩ potentiometer and adjust the potentiometer
for the best load transient where no oscillations should occur. These tests have to be done at the highest Vin and
highest load current because the converter stability is most critical under these conditions. Figure 20, Figure 21,
and Figure 22 show the fine tuning of the loop with Rc.
Figure 20. Overcompensated (Damped Oscillation), RC Is Too Large
Figure 21. Undercompensated (Loop Is Too Slow), RC Is Too Small
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Figure 22. Optimum, RC Is Ideal
Negative Charge Pump
The negative charge pump provides a regulated output voltage by inverting the main output voltage, VO1. The
negative charge pump output voltage is set with external feedback resistors.
The maximum load current of the negative charge pump depends on the voltage drop across the external
Schottky diodes, the internal on resistance of the charge pump MOSFETS Q8 and Q9, and the impedance of
the flying capacitor, C12. When the voltage drop across these components is larger than the voltage difference
from VO1 to VO2, the charge pump is in drop out, providing the maximum possible output current. Therefore, the
higher the voltage difference between VO1 and VO2, the higher the possible load current. See Figure 12 for the
possible output current versus boost converter voltage VO1 and the calculations below.
Voutmin = -(VO1 - 2 VD - IO (2 x rDS(on)Q8 + 2 x rDS(on)Q9 + Xcfly))
Setting the output voltage:
VOUT = -VREF x R3 = -1.213 V x R3
R4
R4
|VOUT|
|VOUT|
R3 = R4 x
= R4 x
VREF
1.213
The lower feedback resistor value, R4, should be in a range between 40 kΩ to 120 kΩ or the overall feedback
resistance should be within 500 kΩ to 1 MΩ. Smaller values load the reference too heavy and larger values may
cause stability problems. The negative charge pump requires two external Schottky diodes. The peak current
rating of the Schottky diode has to be twice the load current of the output. For a 20 mA output current, the dual
Schottky diode BAT54 or similar is a good choice.
Positive Charge Pump
The positive charge pump can be operated in a voltage doubler mode or a voltage tripler mode depending on
the configuration of the C2+ and C2-/MODE pins. Leaving the C2+ pin open and connecting C2-/MODE to GND
forces the positive charge pump to operate in a voltage doubler mode. If higher output voltages are required the
positive charge pump can be operated as a voltage tripler. To operate the charge pump in the voltage tripler
mode, a flying capacitor needs to be connected to C2+ and C2-/MODE.
The maximum load current of the positive charge pump depends on the voltage drop across the internal
Schottky diodes, the internal on-resistance of the charge pump MOSFETS, and the impedance of the flying
capacitor. When the voltage drop across these components is larger than the voltage difference VO1 x 2 to VO3
(doubler mode) or VO1 x 3 to VO3 (tripler mode), then the charge pump is in dropout, providing the maximum
possible output current. Therefore, the higher the voltage difference between VO1 x 2 (doubler) or VO1 x 3
(tripler) to VO3, the higher the possible load current. See Figure 13 and Figure 14 for output current versus boost
converter voltage, VO1, and the following calculations.
Voltage doubler:
VO3max = 2 x VO1 - (2 VD + 2 x IO x (2 x rDS(on)Q5 + rDS(on)Q3 + rDS(on)Q4 + XC1))
Voltage tripler:
VO3max = 3 x VO - (3 x VD + 2 x IO x (3 x rDS(on)Q5 + rDS(on)Q3 + rDS(on)Q4 + XC1 + XC2))
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The output voltage is set by the external resistor divider and is calculated as:
V out 1.214 1 R5
R6
R5 R6 Vout
V
FB
1
R6 V out
1
1.214
Linear Regulator Controller
The TPS6514x includes a linear regulator controller to generate a 3.3-V rail when the system is powered from a
5-V supply. Because an external npn transistor is required, the input voltage of the TPS6514x applied to VIN
needs to be higher than the output voltage of the regulator. To provide a minimum base drive current of 13.5
mA, a minimum internal voltage drop of 500 mV from Vin to Vbase is required. This can be translated into a
minimum input voltage on VIN for a certain output voltage as the following calculation shows:
VI(min) = VO4 + VBE + 0.5 V
The base drive current together with the hFE of the external transistor determines the possible output current.
Using a standard npn transistor like the BCP68 allows an output current of 1 A and using the BCP54 allows a
load current of 337 mA for an input voltage of 5 V. Other transistors can be used as well, depending on the
required output current, power dissipation, and PCB space. The device is stable with a 4.7-µF ceramic output
capacitor. Larger output capacitor values can be used to improve the load transient response when higher load
currents are required.
Thermal Information
An influential component of the thermal performance of a package is board design. To take full advantage of the
heat dissipation abilities of the PowerPAD or QFN package with exposed thermal die, a board that acts similar to
a heatsink and allows for the use of an exposed (and solderable) deep downset pad should be used. For further
information. see Texas Instrumens application notes (SLMA002) PowerPAD Thermally Enhanced Package, and
(SLMA004) Power Pad Made Easy. For the QFN package, see the application report (SLUA271) QFN/SON
PCB Attachement. Especially for the QFN package it is required to solder down the Thermal Pad to achieve the
required thermal resistance.
Layout Considerations
For all switching power supplies, the layout is an important step in the design, especially at high-peak currents
and switching frequencies. If the layout is not carefully designed, the regulator might show stability and EMI
problems. Therefore, the traces carrying high-switching currents should be routed first using wide and short
traces. The input filter capacitor should be placed as close as possible to the input pin VIN of the IC. See the
evaluation module (EVM) for a layout example.
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L1
3.3uH
Vin
3.3V
C3
22uF
C13
1n
TPS65140
R7
15k
VIN
C1
0.22u
D2
C12 0.22u
Vo2
−5 V / 20 mA
C6
0.22u
Vo1
10V / 150 mA
D1
COMP
GND
EN
ENR
C1+
C1−
DRV
R3
620k
C2
0.22u
C2+
C2−/MODE
OUT3
R1
430
C4
22uF
R2
56k
Vo3
up to 23V/20mA
FB3
PG
PGND
PGND
GND
FB2
REF
FB4
BASE
D3
C5
6.8pF
SW
SW
FB1
SUP
R5
1M
R4
150k
C7
0.22u
R6
56k
C11
220nF
Vin
R7
33k
System Power
Good
Figure 23. Typical Application, Notebook Supply
L1
4.7uH
Vin
5.0 V
C3
22uF
C13
2.2n
TPS65140
R7
4.3k
VIN
C1
0.22u
D2
C12 0.22u
Vo2
−7 V / 20 mA
C6
0.22u
Vo1
13.5V / 400 mA
D1
D3
R3
750k
COMP
GND
EN
ENR
C1+
C1−
DRV
C5
3.3pF
SW
SW
FB1
SUP
Vo3
up to 23V/20mA
FB3
PG
PGND
PGND
GND
R5
1M
R4
130k
C11
220nF
C7
0.22u
R6
56k
Q1
BCP68
Vin
C9
1uF
Vo4
3.3V/500mA
Vin
C10
4.7uF
Figure 24. Typical Application, Monitor Supply
20
C4
22uF
R2
75k
C2+
C2−/MODE
OUT3
FB2
REF
FB4
BASE
R1
820
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R7
33k
System Power
Good
PACKAGE OPTION ADDENDUM
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18-Sep-2008
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
TPS65140PWP
ACTIVE
HTSSOP
PWP
24
60
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65140PWPG4
ACTIVE
HTSSOP
PWP
24
60
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65140PWPR
ACTIVE
HTSSOP
PWP
24
2000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65140PWPRG4
ACTIVE
HTSSOP
PWP
24
2000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65140RGER
ACTIVE
VQFN
RGE
24
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65140RGERG4
ACTIVE
VQFN
RGE
24
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65141PWP
ACTIVE
HTSSOP
PWP
24
60
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65141PWPG4
ACTIVE
HTSSOP
PWP
24
60
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65141PWPR
ACTIVE
HTSSOP
PWP
24
2000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65141PWPRG4
ACTIVE
HTSSOP
PWP
24
2000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65141RGER
ACTIVE
VQFN
RGE
24
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65141RGERG4
ACTIVE
VQFN
RGE
24
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65145PWP
ACTIVE
HTSSOP
PWP
24
60
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65145PWPG4
ACTIVE
HTSSOP
PWP
24
60
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65145PWPR
ACTIVE
HTSSOP
PWP
24
2000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65145PWPRG4
ACTIVE
HTSSOP
PWP
24
2000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65145RGER
ACTIVE
VQFN
RGE
24
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS65145RGERG4
ACTIVE
VQFN
RGE
24
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
18-Sep-2008
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF TPS65140, TPS65145 :
• Automotive: TPS65140-Q1, TPS65145-Q1
NOTE: Qualified Version Definitions:
• Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Mar-2008
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
TPS65140PWPR
Package Package Pins
Type Drawing
SPQ
HTSSOP
Reel
Reel
Diameter Width
(mm) W1 (mm)
PWP
24
2000
330.0
16.4
A0 (mm)
B0 (mm)
K0 (mm)
P1
(mm)
W
Pin1
(mm) Quadrant
6.95
8.3
1.6
8.0
16.0
Q1
TPS65140RGER
VQFN
RGE
24
3000
330.0
12.4
4.3
4.3
1.5
8.0
12.0
Q2
TPS65141PWPR
HTSSOP
PWP
24
2000
330.0
16.4
6.95
8.3
1.6
8.0
16.0
Q1
TPS65141RGER
VQFN
RGE
24
3000
330.0
12.4
4.3
4.3
1.5
8.0
12.0
Q2
TPS65145PWPR
HTSSOP
PWP
24
2000
330.0
16.4
6.95
8.3
1.6
8.0
16.0
Q1
TPS65145RGER
VQFN
RGE
24
3000
330.0
12.4
4.3
4.3
1.5
8.0
12.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Mar-2008
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPS65140PWPR
HTSSOP
PWP
24
2000
346.0
346.0
33.0
TPS65140RGER
VQFN
RGE
24
3000
346.0
346.0
29.0
TPS65141PWPR
HTSSOP
PWP
24
2000
346.0
346.0
33.0
TPS65141RGER
VQFN
RGE
24
3000
346.0
346.0
29.0
TPS65145PWPR
HTSSOP
PWP
24
2000
346.0
346.0
33.0
TPS65145RGER
VQFN
RGE
24
3000
346.0
346.0
29.0
Pack Materials-Page 2
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