Automotive Power Electronics.pdf

Automotive Power Electronics.pdf
Automotive
Power Semiconductor Applications
Philips Semiconductors
CHAPTER 5
Automotive Power Electronics
5.1 Automotive Motor Control
(including selection guides)
5.2 Automotive Lamp Control
(including selection guides)
5.3 The TOPFET
5.4 Automotive Ignition
421
Automotive
Power Semiconductor Applications
Philips Semiconductors
Automotive Motor Control
(including selection guides)
423
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.1.1 Automotive Motor Control with Philips MOSFETS
The trend for comfort and convenience features in today’s
cars means that more electric motors are required than ever
- a glance at Table 1 will show that up to 30 motors may be
used in top of the range models, and the next generation
of cars will require most of these features as standard in
middle of the range models.
Wound field DC Commutator Motors
Traditionally motors with wound stator fields, a rotor supply
fed via brushes and a multi-segment commutator - see
Fig. 1 - have been widely used. Recently, however, they
have been largely replaced by permanent magnet motors.
Characteristically they are found with square frames. They
may be Series wound (with high torque at start up but tend
to ’run away’ on no-load), Shunt wound (with relatively flat
speed/torque characteristics) or (rarely) Compound wound.
All these motors need to be activated and deactivated,
usually from the dashboard; that requires a lot of copper
cable in the wiring harnesses - up to 4km in overall length,
weighing about 20 kg. Such a harness might contain over
1000 wires, each requiring connectors at either end and
taking up to six hours to build. Not only does this represent
a cost and weight penalty, it can also create major
’bottlenecks’ at locations such as door hinges, where it
becomes almost impossible to physically accommodate the
70-80 wires required. Now, if the motor switching, reversing
or speed control were to be done at the load by
semiconductor switches, these in turn can be driven via
much thinner, lighter wiring thus alleviating the bottlenecks.
Even greater savings - approaching the weight of a
passenger - can be achieved by incorporating multiplex
wiring controlled by a serial bus.
DC
field
winding
air gap
commutator
Types of motors used in automobiles
Motor design for automotive applications represents an
attempt at achieving the optimum compromise between
conflicting requirements. The torque/speed characteristic
demanded by the application must be satisfied while taking
account of the constraints of the materials, of space and of
cost.
stator
slotted rotor
with windings
There are four main families of DC motors which are, or
which have the potential to be used in automobiles.
brushes
Fig. 1 Wound Field DC Commutator Motor
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Automotive
motor
application
Power Semiconductor Applications
Philips Semiconductors
typical
power (W)
nominal
current (A)
typical
number of
such motors
type of drive
typical
number of
switches per
motor
proposed
MOSFET *
standard
BUK-
L2FET
BUK-
comments
airconditioning
300
25
1
unidirectional,
variable speed
1
456
556
Active suspension may
also require such high
power motors
radiator fan
120-240
10-20
1
unidirectional,
variable speed
1
455
555
These motors may go
brushless, requiring 3
to 6 lower rated
switches
fuel pump
100
8
1
unidirectional
1
453
553
1-2
unidirectional,
variable speed
wipers:
front
rear
60-100
5-8
headlamp
1
1
452/453
552/553
undirectional
1
452
552
2
washers:
front
1-2
30-60
2.5-5
rear
1-2
window lifter
25-120
2-10
2-4
reversible
4
452/455
552/555
sun-roof
40-100
3.5-8
1
reversible
4
452/453
552/553
50
4
4-16
reversible
4
453
553
seat belt
50
4
2-4
reversible
4
453
553
pop-up
headlamp
50
4
2
reversible
4
453
553
radio aerial
25
2
1
reversible
4
452
552
12-36
1-3
6-9
reversible
4
451/452
551/552
12
1
2
reversible
4
451
551
seat
adjustment
(slide,
recline, lift,
lumbar)
door lock
mirror
adjustment
*
Reversing action is at
present mechanical.
This could be done
electronically using 2 or
4 switches
These are meant for guidance only. Specific applications should be checked against individual users requirements. In addition to standard and
L2FETs, FredFETs and low and high side TOPFETs might be considered. Also a variety of isolated, non-isolated and surface mount package
options are available
Table 1 Typical motor and switch requirements in top of range car.
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Automotive
Power Semiconductor Applications
Philips Semiconductors
Permanent Magnet (PM) DC Commutator
Motors
PM Brushless DC Motors
Although common in EDP systems, brushless DC motors
are not yet used extensively in cars. They are under
consideration for certain specialised functions, e.g. fuel
pump where their ’arc free’ operation makes them
attractive. They have a wound stator field and a permanent
magnet rotor - Fig. 4. As their name suggests they have
neither mechanical commutator nor brushes, thus
eliminating brush noise/wear and associated maintenance.
Instead they depend on electronic commutation and they
require a rotor position monitor, which may incorporate Hall
effect sensors, magneto resistors or induced signals in the
non energised winding. Thanks to their lightweight, low
inertia rotor they offer high efficiency, high power density,
high speed operation and high acceleration. They can be
used as servos.
These are now the most commonly used motors in modern
cars. The permanent magnet forms the stator, the rotor
consists of slotted iron containing the copper windings - see
Fig. 2. They have a lighter rotor and a smaller frame size
than wound field machines. Typical weight ratios between
a PM and a wound field motor are:
Copper
Magnets
Rotor
Case
1:10
1:7
1:2.5
1:1
PM motors have a linear torque/speed characteristic - see
Fig. 3 for typical curves relating torque, speed, current and
efficiency. (Philips 4322 010 76130). They are generally
used below 5000 rpm. Their inductance (typically 100 500 µH) is much lower than wound field machines. New
materials (e.g. neodymium iron boron compounds) offer
even more powerful fields in smaller volumes.
3-phase sinewave
or squarewave
slotted stator
with windings
DC
commutator
permanent
magnets
on rotor
permanent
magnets
air gap
Fig. 4 Permanent Magnet Brushless Motor
air gap
slotted rotor
with windings
Switched Reluctance Motors
stator
brushes
These motors - see Fig. 5 - are the wound field equivalent
to the PM brushless DC machine, with similar advantages
and limitations. Again, not yet widely used, they have been
proposed for some of the larger motor applications such as
radiator and air conditioning fans, where their high
power/weight ratio makes them attractive. They can also
be used as stepper motors in such applications as ABS and
throttle control.
Fig. 2 Permanent Magnet Commutator Motor
Motor drive configurations
The type of motor has a considerable influence on the
configuration of the drive circuit. The two families of DC
motors, commutator and brushless need different drive
circuits. However suitably chosen MOSFETs can be used
to advantage with both.
Fig. 3 Performance Curves for PM Commutator Motor
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Automotive
Power Semiconductor Applications
Philips Semiconductors
When a motor is switched off, it may or may not be running.
If it is, then the motor acts as a voltage source and the
rotating mechanical energy must be dissipated either by
friction or by being transformed into electrical energy and
returned to the supply via the inherent anti-parallel diode
of the MOSFET. If it is not turning, then the motor appears
as purely an inductance and for a low side switch the voltage
transient developed will take the MOSFET into avalanche.
Now, depending on the magnitude of the energy stored in
the field and the avalanche capability of the MOSFETs, a
diode in parallel with the motor may or may not be required.
switched DC
salient poles
with
field windings
rotor with
salient poles
As a first approximation, if
1
.L I 2 < WDSS
2 m m
air gap
then a diode may not be needed.
Fig. 5 Switch Reluctance Motor
+ V BAT
Commutator Motors
TR1
TR4
Both permanent magnet and wound field commutator
motors can be controlled by a switch in series with the DC
supply - Fig. 6. Traditionally relays have been used, but
they are not considered to be very reliable, particularly in
high vibration environments. Semiconductors offer an
attractive alternative, providing:
TR2
TR3
• low on-state voltage drop.
• low drive power requirements.
• immunity from vibration.
The Power MOSFET scores on all counts, offering ON
resistances measured in mΩ and requiring only a few volts
(at almost zero current) at the gate, to achieve this.
Fig. 7 H Bridge using MOSFETs
Reversing the polarity of the supply, to a commutator motor,
reverses the direction of rotation. This usually requires an
H bridge of semiconductors, see Fig. 7. In this case the built
in diodes, inherent in MOSFETs, mean that no extra diodes
are necessary. It should be noted that there are now two
devices in series with the motor. So, to maintain the same
low level of on-state voltage drop, each MOSFET must be
doubled in area. With four devices in all, this means a
reversing H bridge requires 8 x the crystal area needed by
a unidirectional drive.
+ V BAT
freewheel
diode
N-channel
Chopping the supply, controls the mean voltage applied to
the motor, and hence its speed. In the case of the H bridge
TR1 and TR4 might be used to control direction, while a
chopping signal (typically 20kHz) is applied to TR3 or TR2.
When reversing the direction of rotation, it is preferable to
arrange the gating logic so that the system goes through a
condition where TR1, TR2, TR3 and TR4 are all off.
MOSFET
Fig. 6 Commutating Motor Switch
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Automotive
Power Semiconductor Applications
Philips Semiconductors
Switched Field Motors
+ V BAT
PM brushless motors typically require 6 switches to
generate the rotating field, see Fig. 8. Although there are
motors, which operate at lower power density, which can
be driven from 3 switches. The circuit in Fig. 9 shows a low
side switch version of such a drive. A similar arrangement
with high side switches would be possible.
+
+ V BAT
TR1
TR3
TR5
TR2
TR4
TR6
+
Fig. 10 4 MOSFET Switched Reluctance Motor Drive
PWM speed control pushes up the required switching
speed even further. Philips MOSFETs are designed so that
both switch and inbuilt diode are capable of efficient
switching at the highest frequencies and voltages
encountered in automotive applications.
Fig. 8 MOSFET Brushless Motor Drive
High side drivers
Often, in automobiles, there is a requirement for the switch
to be connected to the positive battery terminal with the
load connected via the common chassis to negative.
Negative earth reduces corrosion and low side load is safer
when loads are being worked on or replaced. Also, when
H bridges are considered the upper arms are of course high
side switches.
+ V BAT
TR1
TR2
TR3
+V
BAT
p-channel
MOSFET
Fig. 9 3 MOSFET Brushless Motor Drive
Switched reluctance motors may use as few as 4 or as
many as 12 switches to generate the rotating field, a 4
switch version is shown in Fig. 10.
The speed and direction of all switched field motors is
controlled by the timing of the field pulses. In the case of
brushless DC machines these timing pulses can be derived
from a dedicated IC such as the Philips NE5570. Rotor
position sensing is required - using, for example,
magnetoresistive sensors - to determine which windings
should be energised. Compared with a DC commutator
motor, the power switches for a brushless motor have to
be fast, because they must switch at every commutation.
Fig. 11 P-channel high side switch
There are two MOSFET possibilities for high side switches:
429
Automotive
Power Semiconductor Applications
Philips Semiconductors
v
An alternative approach for H bridge choppers is to use
the MOSFETs themselves to generate the drive voltage
with a bootstrap circuit as shown in Fig. 13. This circuit
works well over a range of mark-space ratios from 5% to
95%. Zener diodes should be used in this circuit to limit
the transients that may be introduced onto the auxiliary
line.
charge pump
+ V BAT
8
VCC
RES
4
6
5
DIS
THR
CV
AU7555D
N-channel
7
O/P
TRIG
GND
3
MOSFET
+ V BAT
2
status
1
input
BUK202-50Y
TOPFET
Fig. 12 N-channel high side switch with charge pump
• P-channel switches. These simplify the drive circuit
which only needs referencing to the positive supply, see
Fig. 11. Unfortunately p-channel devices require almost
three times the silicon area to achieve the same on
resistance as n-channel MOSFETs, which increases cost.
Also P-channel devices that can be operated from logic
level signals are not readily available.
Fig. 14 High Side TOPFET
+ V BAT
TR1
High Side TOPFET
TR4
+
+
TR2
The ideal high side switch to drive motor loads would be
one which could be switched on and off by a ground
referenced logic signal, is fully self-protected against short
circuit motors and over temperatures and is capable of
reporting on the load status to a central controller.
TR3
The Philips response to these requirements is a range of
high side TOPFETs. The range contains devices with
RDS(ON) from 38 to 220 mΩ, with and without internal ground
resistors. All the devices feature on board charge pump and
level shifting, short circuit and thermal protection and status
reporting of such conditions as open or short circuit load.
As can be seen in Fig. 14, the use of a TOPFET makes the
circuit for a protected high side drive for a motor very simple.
Fig. 13 Bootstrap bridge drive
• N-channel switches. To ensure that these are fully turned
on, the gate must be driven 10 V higher than the positive
supply for conventional MOSFETs or 5 V higher for Logic
Level types. This higher voltage might be derived from an
auxiliary supply, but the cost of ’bussing’ this around the
vehicle is considerable.
Currents in motor circuits
There are 5 classes of current that can flow in a motor
circuit:-
The additional drive can be obtained locally from a charge
pump, an example in shown in Fig. 12. An oscillator (e.g
Philips AU7555D) free runs to generate a rectangular 12 V
waveform, typically at around 100kHz. A voltage doubler
then raises this to around twice the battery voltage. This
arrangement is equally suitable for ’DC’ or chopper drives.
• nominal - this is the maximum steady state current that
will flow when the motor is performing its function under
normal conditions. It is characterised by its relatively low
level and its long duration.
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Automotive
Power Semiconductor Applications
Philips Semiconductors
• overload - this is the current which flows when the motor
is driving a load greater than it is capable of driving
continuously, but is still performing its function i.e not
stalled. This is not necessarily a fault condition - some
applications where the motor is used infrequently and for
only a short time, use a smaller motor, than would be
needed for continuous operation, and over-run it. In these
cases the nominal current is often the overload current.
Overload currents tend to be about twice the nominal
current and have a duration between 5 and 60 seconds.
• inrush - or starting currents are typical 5 to 8 times the
nominal current and have a duration of around 100 ms,
see Fig. 15. The starting torque of a motor is governed
by this current so if high torque is required then the control
circuit must not restrict the current. Conversely if starting
torque is not critical, then current limiting techniques can
be employed which will allow smaller devices to be used
and permit sensitive fault thresholds to be used.
Fig. 16 Current in stalled 2 A Motor
It is important that the devices, selected for the control
circuit, can operate reliably with all of these currents. With
some types of switching device, it is necessary to select on
the basis of the absolute maximum current alone. Often this
results in a large and expensive device being used. The
characteristics of MOSFETs, in particular their thermally
limited SOAR (no second breakdown), allows the designer
to specify a much smaller device whose performance more
closely matches the needs of the circuit.
Device requirements
Voltage
The highest voltage encountered under normal operation
is 16 V, under jump start this can rise to 22 V. In the case
where the battery becomes disconnected with the
alternator running the voltage can rise to 50 V (assuming
external protection is present) or 60 V in the case of 24 V
vehicles see Table 2. Thus the normal voltage requirement
is 50/60v, however the power supply rail in a vehicle is
particularly noisy. The switching of the numerous inductive
loads generates local voltage spikes and surges of both
polarities. These can occur singly or in bursts, have
magnitudes of 100 V or more and durations of the order of
1ms.
Fig. 15 Start-up Current in 2 A Motor
• stall - if the motor cannot turn then the current is limited
only by the series resistance of the motor windings and
the switch. In this case, a current of 5-8 times the running
current can flow through the combination. Fig. 16 shows
the current that flows through a stalled 2 A motor - the
current gradually falls as the temperature, and
consequently the resistance, of the motor and the
MOSFET rises.
It is important to chose MOSFETs capable of withstanding
these stresses, either by ensuring VDS exceeds the value
of the transients or by selecting 50/60 V devices with
sufficient avalanche energy capability to absorb the pulse.
For transients in excess of these values it is necessary to
provide external protection.
• short circuit - if the motor is shorted out then the current
is limited only by the resistance of the switch and the
wiring. The normal protection method, in this case, is a
fuse. Unless other current control methods are used then
it is the I2t rating of the fuse which determines how long
the current will flow.
However, the TOPFET range of devices, both low and high
side, have overvoltage protection on chip. As a
consequence they are rated to withstand very much higher
transient energies.
431
Automotive
Voltage Range
>50 (60)*
30 to 50
22 to 30
16 to 22
(32 to 40)*
10.5 to 16
(20 to 32)*
8 to 10.5
6 to 8
(9 to 12)*
0 to 6
(0 to 6)*
negative
*
Power Semiconductor Applications
Philips Semiconductors
L2FETs
Cause
coupling of spurious spikes
clamped load dump
voltage surge on cut-off of inductive
loads
jump start or regulator degraded
The supply voltage in an automobile derived from the
battery is only 12 V (nominal). This can vary from 10.5 V to
16 V under normal operation. It is important that the
MOSFET switches be fully turned on under these
conditions, not forgetting that for high side switches it may
be necessary to derive the gate drive from a charge pump
or bootstrap.
normal operating condition
alternator degraded
starting a petrol engine
Whilst a gate source voltage of 6 V is usually sufficient to
turn a conventional MOSFET on, to achieve the lowest on
resistance, 10 V is required. Thus the margin between
available and required gate drive voltage may be quite tight
in automotive drive applications.
starting a diesel engine
negative
peaks
connected battery
or
reverse
One way to ease the problem is to use Logic Level
MOSFETs (L2FET), such as the BUK553-60A or
BUK555-60A, which achieve a very low on resistance state
with only 5 V gate-source.
24 V supply
Table 2 Conditions Affecting Abnormal Supply Voltages
Temperature
Conclusions
The ambient temperature requirement in the passenger
compartment is -40 to +85˚C , and -40 to +125˚C under the
bonnet. All Philips MOSFETs shown in Table 1 have
Tjmax = 175˚C.
There is an increasing demand for low cost, reliable
electronic switching of motors in automobiles. Despite the
wide variety of motor types and drive configurations there
is a Philips Power MOSFET solution to all of these
demands. The broad range of types includes standard and
logic level FETs, FredFETs, high and low side TOPFETs.
The combination of low on-state resistance, ease of drive
and ruggedness makes them an attractive choice in the
arduous automotive environment.
The TOPFETs have a maximum operating Tj of 150˚C
because above this temperature the on chip protection
circuits may react and turn the device off. This prevents the
device from damage that could result from over dissipation.
This protection eases the problems of the thermal design
by reducing the need for large safety margins.
432
Automotive
Power Semiconductor Applications
Philips Semiconductors
Automotive Lamp Control
(including selection guides)
433
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.2.1 Automotive Lamp Control with Philips MOSFETS
• Overloads - relays can also prove to be unreliable under
high transient load conditions. The arcing which occurs
when switching high currents and voltages causes contact
wear leading eventually to high resistance or even the
contacts welding together.
The modern motor vehicle, with its many features, is a
complex electrical system. The safe and efficient operation
of this system calls for sophisticated electronic control. A
significant part of any control system is the device which
switches the power to the load. It is important that the right
type of device is chosen for this job because it can have a
major influence on the overall system cost and
effectiveness. This choice should be influenced by the
nature of the load. This article will discuss the features of
the various types of switching device - both mechanical and
solid state. These factors will be put into the context of the
needs of a device for the control of resistive loads like lamps
and heaters. It will be shown that solid state devices allow
the designer a greater degree of control than mechanical
switches and that the features of Power MOSFETs make
them well suited to use in automotive applications.
• Hazardous Materials - to achieve the prefered switching
performance, relays need to use materials like cadmium.
The use of such materials is becoming restricted by
legislation on health and safety grounds.
• Noise - the operation of a relay is not silent. This is proving
to be unacceptably intrusive when relays are sited in the
passenger compartment.
Solid-state switches can overcome these limitations but can
also give the designer the option of introducing the following
useful features:• Current limiting - a relay has two states - on or off so the
current which flows depends only on the load. There is no
mechanism which allows a relay to regulate the current
which flows through it. The best that a relay can do is to
try and turn off, when a high current is detected, but
because they are so slow, very large currents may be
flowing before the relay can react and damage may have
already been caused. However the characteristics of solid
state devices like MOSFETs and bipolar transistors allow
them to control the current. This allows designers the
chance to introduce systems which can handle faults in a
safe and controlled manner.
Choice of switch type
Mechanical or solid-state
Designers of automotive systems now have the choice of
either mechanical or solid-state switches. Although
mechanical switches can prove be a cheap solution they
do have their limitations. Solid-state switches overcome
these limitations and provide the designer with several
useful additional features.
Areas where the limitations of relays become apparent
include:-
• Control of switching rate - the lack control that a relay
has over the current proves to be a limitation not only
during fault conditions but also during normal switching.
Without control, the rate at which current changes, dI/dt,
depends only on the external circuit and extremely high
rates can result. The combination of high dI/dt and the
contact bounce that relays are prone to, creates an
’electrically’ noisy environment for surrounding systems.
The control available with solid-state switches permits the
designer to restrain the current and produce ’soft’
switching eliminating any possible EMC problems.
• Reliability - to achieve the required levels of sensitivity
and efficiency means that relay coils have to be wound
with many turns of very fine wire. This wire is susceptible
to damage under conditions of high mechanical stress vibration and shock.
• Mounting - special assembly techniques are needed
when dealing with automotive relays. Their outlines are
not compatible with the common methods of automated
assembly like auto insertion and surface mounting.
• Dissipation - the power loss in the coil of a relay is not
negligible - the resulting temperature rise makes it unwise
to mount other components in close proximity. In some
multiple relay applications it is necessary to provide
cooling by ventilation.
Power MOSFET or Bipolar Transistor
All solid-state switches have significant advantages over
relays but there are different types of solid-state switch and
their particular characteristics need to be taken into account
if an optimum choice is to be made. There are two major
types of solid-state switches which are suitable for use in
automotive applications - power MOSFETs and bipolar
transistors - and several factors need to be considered if
the optimum choice is to be made.
• Temperature - the maximum operating temperature of
relays is typically in the range 70˚C - 85˚C.
• Corrosion - the unsealed mechanism of relays are
vulnerable in contaminating and corrosive environments.
435
Automotive
Power Semiconductor Applications
Philips Semiconductors
• Overload - The choice of device type can be influenced
by the magnitude and duration of overload currents
associated with the application - for example the inrush
current of lamps. This factor is particularly important
because the maximum current that can be safely
conducted by a bipolar transistor is independent of its
duration. Whereas the safe operating area of a MOSFET
allows it to handle short duration currents very much
greater than its DC rating.
Low Side Switch
In this arrangement the load is permanently connected
(perhaps via a fuse and the ignition switch) to the positive
supply. The switching device is connected between the
negative terminal of the load and the vehicle ground. This,
together with the almost universal practice of referencing
control signals to the vehicle ground, makes the
implementation of a low side switch with MOSFETs
extremely simple. The circuit shown in Fig. 1 shows a
MOSFET connected as a low side switch to a lamp load.
The Source terminal of the MOSFET is connected to ground
so the control signal, which is also referenced to ground,
can be connected to the Gate.
• Drive power - There can be a significant difference
between the total power needed to drive bipolar and MOS
transistors. A MOSFET’s oxide insulation makes it a
voltage controlled device whereas a bipolar needs current
drive. However, most control circuits are voltage rather
than current orientated and the conversion to current
operation often involves the used of loss inducing
resistors.
+ V BAT
• Reverse protection - If the switching device is required
to survive reverse conduction conditions then it is
necessary to have a diode, connected in anti parallel,
around it. If the device is a bipolar transistor then an extra
component will be needed. However the device is a
MOSFET then it has an inherent body / drain diode which
will perform this function without the additional
expenditure in components or board space.
N-channel
MOSFET
Logic level and standard mosfets
The battery voltage in a car is a nominal 12 V. This can vary
from 10.5 V to 16 V under normal operation and can fall as
low as 6 V during starting. It is important that MOSFET
switches be fully turned on at these voltages, bearing in
mind that for a high-side switches it may be necessary to
derive the gate voltage from a charge pump circuit. While
a VGS of 6 V is usually sufficient to turn a standard MOSFET
on, 10 V is required to achieve the lowest on-state
resistance, RDS(ON). Thus the margin between available and
required gate drive voltage may be quite tight in automotive
drive applications. One way to overcome this problem is to
use L2FETs such as the BUK553-60A or BUK555-60A,
which achieve a very low RDS(ON) with a VGS of only 5 V.
Fig. 1 Low side switch with N-channel MOSFET
High Side Drivers
Often, however, there is a requirement for the switch to be
connected to the positive battery terminal with the load
connected via the common chassis to the negative. This
arrangement reduces electrochemical corrosion and the
risk of accidentally activating the device during
maintenance.
One method of creating such a high side switch is to use
P-channel rather than N-channel MOSFETs. A typical
arrangement is shown in Fig. 2. In this the source is
connected to the +ve feed and the drain to the load. The
MOSFET can be turned ON by taking the control line to
zero and it will be OFF when the gate is at +ve supply
voltage. Unfortunately P-channel MOSFETs require almost
three times the silicon area to achieve the same low on-state
resistance as N-channel types and so are much more
expensive. An additional problem is the difficulty of
obtaining P-channel devices with low enough gate
threshold voltage to operate reliably at low battery voltages.
Switch configuration
A load’s control circuit can be sited in either its positive or
negative feeds. These are referred to as high side and low
side switching respectively. Which configuration is chosen
often depends on the location of the load/switch and the
wiring scheme of the vehicle but other factors like safety
can be overriding. The use of semiconductor switches
introduces another element into the decision process
because of the need to ensure that they are being driven
correctly.
436
Automotive
Power Semiconductor Applications
Philips Semiconductors
of the supply being dropped across the it, the gate has to
taken to a voltage higher than the supply voltage. This
higher voltage might be derived from an auxiliary supply,
but the cost of ’bussing’ this around the vehicle would be
high. Figure 3 shows how this auxiliary supply could be
produced locally. It consists of an oscillator - based around
the Philips AU7555D - running at approximately 100 kHz
which is driving a charge pump which nearly doubles the
supply voltage.
+ V BAT
P-channel
MOSFET
An alternative approach, which can be used when the
device doesn’t have to be continuously ON, for example
PWM lamp dimming or lamp flashing, is shown in Fig. 4. In
this bootstrap arrangement capacitor C is charged to the
supply voltage when the MOSFET is OFF. When the
MOSFET is turned ON, its source terminal, and the negative
end of C, rises to the supply voltage. The potential of the
positive end of C is now higher than the +ve supply and
diode D is reverse biased preventing C from being
discharged. C can now act as the high voltage supply for
the gate. The inevitable leakages will tend to discharge C
and hence reduce the gate/source voltage, but with good
components it is easy to ensure that a voltage high enough
to keep the MOSFET fully ON is available for several
seconds.
Fig. 2 P-channel high side switch.
Using N-channel devices overcomes these problems but
involves a more complicated drive circuit.
To ensure that a n-channel MOSFET is fully turned on, the
gate must be driven 10 V higher than its source, for
conventional MOSFETs, or 5 V higher for Logic Level (L2)
FETs. With the source connected to the load and with most
v
charge pump
+ V BAT
N-channel
MOSFET
4
R
2
TR
3
Q
AU7555D
5 CV
DIS
THR
7
6
GND
1
Fig. 3 N-channel High-side switch with charge pump
437
Automotive
Power Semiconductor Applications
Philips Semiconductors
The normal operating temperature of a heater is not as high
as that of a lamp, so the inrush current is rarely greater than
twice the nominal current and often less. The duration of
the ’inrush’ can, however, last for many minutes and it may
be this current which is used to define the ’normal’ operating
condition.
+ V BAT
Being essentially resistive, lamps and heaters have very
low inductance. This means that the current in the load will
rise as quickly as the rest of the wiring will let it. This can
lead to serious interference problems.
R1
BC337
N-channel
R2
MOSFET
C
10uF
BC327
Lamp Current (10A/div.)
BC337
Fig. 4 Bootstrap High side driver
Inrush current
Any circuit or device which is intended to drive either a lamp
or a heater must be able to handle not only the normal
running current but also the inrush current at start up. All
lamps and many heaters are essentially resistors made
from metal conductors whose resistivity will increase with
temperature.
0
10ms/div
In the case of lamps, the extremely high operating
temperature (3000 K) means that the hot to cold resistance
ratio is large. Typical values for a 60 W headlamp bulb are:filament
resistance
current
cold (-40˚C)
0.17 Ω
70 A
hot
2.4 Ω
5A
Fig. 5 Current in 60 W lamp during start up
Switch rate
The inductance associated with the supply wires in a car,
is not negligible - a figure of 5µH is often quoted. This
inductance, combined with the high rates of change of
current associated with the switching of resistive loads and
lamps, results in transient voltage appearing on the supply
leads. The magnitude of the transient is given by:-
The figures given for the currents assume that there is 12 V
across the lamp, in practice wiring and switch resistance
will reduce the cold current somewhat, but the ratio will still
be large. The actual ratio depends upon the size and
construction of the lamp but figures between 10 and 14 are
common. For safety, the higher figure should be used.
Vtransient = −L.
dI
dt
For example a current which rises as slowly as 2 A/µs will
cause a 10 V dip in the supply to the switching circuit. This
effect can be clearly seen in the waveforms of Fig. 6a. Such
a perturbation can have an effect in two ways. In the first
case the control circuit may be upset by having its supply
reduced to only 2 V and may, if not specifically designed to
cope with it, fail to function correctly. In the second case, it
is easy for a transient as large as this, with its significant
high frequency content, to be transmitted into adjacent
conductors in the wiring loom. If some of the conductors
are signal wires then false triggering of other functions could
result.
The low thermal mass and the high power dissipation
(850 W peak in 60W lamp) means that the lamp heats up
very quickly. This means that the current falls from its peak
value equally quickly. The time it takes for the current to fall
back to its normal value depends on the size and
construction of the lamp - the larger the lamp the longer it
will take to heat up. Typically the current will have an
exponentially decay with a time constant of 1 - 10 ms. The
waveforms in Fig. 5 show the typical inrush current for a
60 W lamp being switched on by a MOSFET. The initial
temperature of the lamp filament was 25˚C.
438
Automotive
Power Semiconductor Applications
Philips Semiconductors
a) Low impedance gate drive
Supply Voltage (5V/div)
Gate
Drive
0
Lamp Current (10A/div)
0
peak dI/dt=2A/us
Fig. 7 Gate supply networks for switching rate control
20us/div
unlikely that the voltage would exceed 30 V. Transient
voltages of this magnitude are relatively common in the
automotive environment and all circuits should be able to
withstand them. It is still worthwhile keeping the turn off
transient under control by ensuring that the dI/dt is low
enough - a figure of <1 A/µs is standard.
b) 47kΩ gate drive resistor
Supply Voltage (5V/div)
0
Soft turn off, like soft turn on, is easy to implement if the
controlling device is a Power MOSFET. In fact the same
series resistor can be used to limit both the turn on and turn
off rates. With a lamp load, however, this method will give
a much slower turn off than is really necessary because of
the large difference between the current at turn on and turn
off. If this is a problem then an additional resistor and diode
put in parallel with the first resistor - see Fig. 7 - will speed
up the turn off.
Lamp Current (10A/div)
0
peak dI/dt=0.5A/us
20us/div
Fig. 6 Effect of high dI/dt on supply voltage
MOSFET selection
The dip will be reduced to manageable proportions if the
dI/dt can be held to 0.5 A/µs. Since the loads are resistive,
achieving this means reducing the rate that the voltage is
applied to the load. This type of ’soft’ starting is relatively
easy to implement when the controlling device is a Power
MOSFET. All that is needed is to put resistance in series
with the gate drive.
The type of device chosen for a particular application
depends upon the features that the control circuit needs to
have. Table 3 lists the available MOSFET types and some
of their features that would be useful in automotive
applications.
Having chosen the type of MOSFET it becomes necessary
to decided on the size of device. With MOSFETs this
decision is made easier because, in its on-state, a MOSFET
can be treated as a resistance and because its safe
operating area (SOAR) is set by thermal considerations
only (no second breakdown effects). The first stage of the
selection process is to chose a device on the basis of the
nominal current requirement. The next stage is to check
that the inrush current, of the particular application and the
drive method used, does not result in the MOSFET
exceeding the transient thermal ratings. Having selected a
device that is capable of switching the load the designer
can then use the quoted values for the on-state resistance
(RDS(ON)) to check that any on-state voltage drop
The plots shown in Fig. 6b illustrate the effect inserting
47 kΩ in series with the gate supply of a BUK455-60A. The
load for these tests was a 60W lamp being supplied from
a battery via a 5 µH inductor. The dip in voltage due to dI/dt
is now lost in the voltage drop from the wiring resistance.
The rate at which current falls at turn off is also important.
High negative dI/dt will result in a large positive spike on
the supply rails. As with the negative dip, this spike could
cause interference in adjacent wires but it could also cause
overvoltage damage. Unlike the turn on dip which can never
be greater than 12 V, the magnitude of the turn off spike is
potentially unlimited. In practice, however, it is extremely
439
Automotive
Power Semiconductor Applications
Philips Semiconductors
expected. For this reason it is necessary to assume that
the circuits and devices will have to work in an ambient
temperature of 85˚C.
requirements are being met. Tables 3 and 4 lists many of
the different of lamps and resistive loads found in cars and
suggests MOSFET types that can be used to control them.
MOSFET
Type
Standard
L2FET
Voltage
Features
It is possible to split the voltage conditions that can occur
into two groups - Normal and Abnormal. ’Normal’ conditions
are essentially those which can be present for very long
periods of time. Under such conditions it is reasonable to
expect devices and circuits to be completely operational
and to suffer no ill effects. ’Abnormal’ conditions are
characterised by their temporary nature. They are not
expected to persist for long periods and during them, some
loss in device / circuit performance can be expected and,
in some cases, is allowable.
Wide range of current ratings from 5 to
>100 A.
Wide range of package styles
Fast recovery anti-parallel diode (60 /
100 V types)
Extremely fast switching.
as standard
+
Fully operational with low voltage supply
Low side
TOPFET
as L2FET
+
overvoltage protection
overload protection
over temperature protection
3 and 5 pin versions
linear and switching control
High side
TOPFET
Single component providing:high side switch (on chip charge pump
and level shifting)
device protection
load protection
status reporting
CMOS compatible input
Normal voltages
When considering the ’Normal’ environment it is important
to included both the typical and extreme cases. The crucial
condition for most devices and circuits is when the engine
is running. At this time the supply voltage can be anywhere
between 10.5 and 16 V in ’12 V’ systems or between 20
and 32 V in ’24 V’ systems.
The other significant ’normal’ operating mode is when
engine not running. In this state the supply voltage could
be very low but voltages below some level must be
considered as a fault condition. However some circuits will
have to operate with voltages as low as 6 V.
Voltage Level
Cause
12 V systems 24 V systems
TABLE 1 MOSFET Types and Features
40 V - 50 V
30 V - 40 V
22 V - 30 V
16 V - 22 V
16 V - 22 V
8 V - 10.5 V
6V-8V
0V-6V
The automotive environment
The environment that circuits and devices can be subjected
to in automotive applications can prove to be extremely
severe. Knowledge of the conditions that can exist is
necessary to ensure that suitable devices and circuits are
chosen. The two most stressful aspects of the environment
are the temperature and voltage.
60 V - 75 V
50 V - 60 V
22 V - 30 V
32 V - 40 V
32 V - 40 V
12 V - 20 V
9 V - 12 V
0V-6V
external spikes
clamped load dump
inductive load switch off
jump start
faulty regulator
faulty alternator
starting a petrol engine
starting a diesel engine
Table 2 Abnormal Supply Voltages
Temperature
Abnormal voltages
The lowest temperature that is likely to be reached is -40˚C.
This is related to the minimum outside temperature and
may be lower under some special circumstances. The
maximum temperature depends to a great extent upon the
siting of circuits. The general ambient temperature in the
engine compartment can be quite high and it is reasonable
to assume that devices will see temperatures of 125˚C.
Within the passenger area, conditions are somewhat more
benign, but in areas where heat is generated and air flow
is restricted, the temperature will be higher than might be
It is possible to envisage a situation in which nearly any
voltage could appear on the supply wires of a vehicle. How
extreme the voltages get depends to a great extent upon
the protection, both deliberate and incidental, built into the
system. The actual voltage that appears at the terminals of
a circuit is also influenced strongly by its location and the
location of the protection. Analysis of the automotive
environment has produced a list of expected abnormal
conditions. The values of voltage that these conditions can
be expected to produce are shown in Table 2.
440
Automotive
Load
Power Semiconductor Applications
Philips Semiconductors
Typical
Power
Nominal
Current
Peak
Inrush
Current
60 W
55 W
45 W
40 W
5A
4.6 A
3.8 A
3.3 A
70 A
64 A
53 A
47 A
spotlight
55 W
4.6 A
front fog light
55 W
rear fog light
21 W
front sidelight
headlamp
Number
of lamps
/car
Recommended MOSFET1
Standard FET
Logic Level FET
SOT186
TO220
SOT186
TO220
2
BUK445-60A
BUK455-60A
BUK545-60A
BUK555-60A
64 A
2
BUK445-60A
BUK455-60A
BUK545-60A
BUK555-60A
4.6 A
64 A
2
BUK445-60A
BUK455-60A
BUK545-60A
BUK555-60A
1.8 A
25 A
2
BUK442-60A
BUK443-60A2
BUK452-60A
BUK453-60A2
BUK542-60A
BUK543-60A2
BUK552-60A
BUK553-60A2
5W
0.4 A
6A
2
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
rear sidelight
5W
10 W
0.42 A
0.83 A
5.8 A
12 A
2
2
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
brake light
21 W
1.8 A
25 A
2
BUK442-60A
BUK443-60A2
BUK452-60A
BUK453-60A2
BUK542-60A
BUK543-60A2
BUK552-60A
BUK553-60A2
direction indicator
light
21 W
1.8 A
25 A
4
BUK442-60A
BUK443-60A2
BUK452-60A
BUK453-60A2
BUK542-60A
BUK543-60A2
BUK552-60A
BUK553-60A2
side marker light
3W
4W
5W
0.25 A
0.33 A
0.42 A
3.5 A
4.7 A
5.8 A
4
4
4
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
license plate light
3W
5W
0.25 A
0.42 A
3.5 A
5.8 A
2
1
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
reversing /
backup light
21 W
1.8 A
25 A
2
BUK442-60A
BUK443-60A2
BUK452-60A
BUK453-60A2
BUK542-60A
BUK543-60A2
BUK552-60A
BUK553-60A2
instrument panel
light
2.2 W
0.18 A
2.5 A
5
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
courtesy light
2.2 W
0.18 A
2.5 A
4
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
door light
2.2 W
0.18 A
2.5 A
4
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
boot / bonnet light
2.2 W
0.18 A
2.5 A
4
BUK441-60A
BUK451-60A
BUK541-60A
BUK551-60A
Notes
1
These are meant for general guidance only. Specific applications should be checked against individual users’
requirements. In addition to standard and logic level MOSFETs, high and low side TOPFETs might also be considered.
2
This device can be used to control two bulbs simultaneously.
TABLE 3 Automotive lamps - characteristics and recommended MOSFET drivers
441
Automotive
Load
screen heater
seat heater
Power Semiconductor Applications
Philips Semiconductors
Recommended MOSFET1
SOT186(A)
F-pack
Typical
Power
Nominal
Current
Number
/car
TO220
300-600 W
25-50 A
1
2 x BUK556-60H
100-120 W
8-10 A
2
2
BUK452-60A
Comments
Devices connected in parallel
2
BUK442-60A
Notes
1
These are meant for general guidance only. Specific applications should be checked against individual users’
requirements. In addition to standard MOSFETs, L2FETs, low and high side TOPFETs might also be considered.
2
To achieve an on-state voltage drop of <1 V the BUKxx3-60A device should be used.
TABLE 4 Automotive Resistive Loads - characteristics and recommended MOSFET drivers
442
Automotive
Power Semiconductor Applications
Philips Semiconductors
The TOPFET
443
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.1 An Introduction to the 3 pin TOPFET
Variants of this configuration with differing input resistor
values (higher or lower) will be produced to suit different
application requirements.
The TOPFET (Temperature and Overload Protected
MOSFET) concept has been developed by Philips
Semiconductors and is achieved by the addition of a series
of dedicated on-chip protection circuits to a low voltage
power MOSFET. The resulting device has all the
advantages of a conventional power MOSFET (low RDS(on),
logic level or standard gate voltage drive) with the additional
benefit of integrated protection from hazardous overstress
conditions.
DRAIN
O/V
CLAMP
TOPFETs are designed for operation in low voltage power
applications, particularly automotive electronic systems.
The operation and protection features of the TOPFET range
of devices also make them suitable for other low voltage
power systems. TOPFETs can be used for all common load
types currently controlled by conventional power
MOSFETs.
INPUT
POWER
MOSFET
LOGIC AND
PROTECTION
The first generation of TOPFET devices are summarised
in Table 1.
SOURCE
Protection strategy
D
A functional block diagram and the circuit symbol of the first
generation 3-pin TOPFETs are shown in Fig. 1. The
functional block diagram indicates that the logic and
protection circuits are supplied directly from the input pin.
This places a requirement on the user that the input voltage
must be sufficiently high to ensure that the protection
circuits are being correctly driven.
TOPFET
I
The TOPFET includes an internal resistance between the
input pin and the power MOSFET gate. This is required to
ensure that the protection circuits are supplied even under
conditions when the circuits have been activated to turn off
the power MOSFET stage. The value of this resistance has
been chosen to be a suitable compromise between the
requirements of switching speed and drive capability.
P
S
Fig. 1 Schematic diagram and circuit of 3-pin TOPFET
TOPFET
Package
VDS (V)
RDS(ON) (mΩ)
at VIS = (V)
BUK100-50GL
BUK100-50GS
BUK101-50GL
BUK101-50GS
BUK102-50GL
BUK102-50GS
TO220
TO220
TO220
TO220
TO220
TO220
50
50
50
50
50
50
125
100
60
50
35
28
5
10
5
10
5
10
Table 1. 3-pin TOPFET type range
445
Automotive
Power Semiconductor Applications
Philips Semiconductors
Overtemperature protection
Overvoltage protection
TOPFETs include an on-chip protection circuit which
measures the absolute temperature of the device. If the
chip temperature rises to a dangerous level then the
overtemperature protection circuit operates to turn off the
power MOSFET stage. Once tripped, the device remains
protected until it is reset via the input pin. In the tripped
condition the gate of the power MOSFET stage is pulled
down by the control logic and so some current is drawn by
the input pin of the TOPFET. If the overtemperature
condition persists after the gate has been reset then the
protection circuit is reactivated.
Transient overvoltage protection is an additional feature of
the TOPFET range. This is achieved by a combination of
a rugged avalanche breakdown characteristic in the
PowerMOS stage and an internal dynamic clamp circuit.
Operation is guaranteed by an overvoltage clamping
energy rating for the TOPFET. Overvoltage protection gives
guarantees against fault conditions in the system as well
as the ability for unclamped inductive load turn-off.
ESD protection
Short circuit protection
In the case of short circuit faults the rate of rise of
temperature in a MOSFET switch can be very rapid.
Guaranteed protection under this type of condition is best
achieved using the on-chip protection strategy which is
implemented in the TOPFET range of devices. The short
circuit protection circuit acts rapidly to protect the device if
the temperature of the TOPFET rises excessively.
The input pin of the TOPFET is protected with an ESD
protection zener. This device protects the PowerMOS gate
and the TOPFET circuit from ESD transients. The energy
in the ESD pulse is dissipated in the ESD source rather
than in the TOPFET itself. This input zener diode cannot
be used in the continuous breakdown mode and so is the
determining factor in setting the maximum allowable
TOPFET input voltage.
The TOPFET does not limit the current in the power circuit
under normal operation. This ensures that the TOPFET
does not affect the operation of circuits where large inrush
currents are required. As with the overtemperature
protection circuit, the short circuit protection circuit turns off
the power MOSFET gate via the control logic and is reset
by taking the input pin low.
One feature of the implementation of the protection circuits
used in the first generation TOPFET devices is that the input
cannot be reverse biased with respect to the source. This
must be adhered to at all times. When the TOPFET is in
reverse conduction the protection circuits are not active.
446
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.2 An Introduction to the 5 pin TOPFET
chip temperature rises to a dangerous level then the
overtemperature protection circuit operates to turn off the
power MOSFET stage. Once tripped the device remains
protected until it is reset via the protection supply pin.
The TOPFET (Temperature and Overload Protected
MOSFET) concept has been developed by Philips
Semiconductors and is achieved by the addition of a series
of dedicated on-chip protection circuits to a low voltage
power MOSFET. The resulting device has the advantages
of a conventional power MOSFET (low RDS(on), logic level
gate voltage drive) with the additional benefit of integrated
protection from hazardous overstress conditions.
DRAIN
PROTECTION SUPPLY
FLAG
TOPFETs are designed for operation in low voltage power
applications, particularly automotive electronic systems.
The operation and protection features of the TOPFET range
of devices also make them suitable for other low voltage
power systems. TOPFETs can be used for all common load
types currently controlled by conventional power
MOSFETs.
O/V
CLAMP
INPUT
POWER
MOSFET
LOGIC AND
PROTECTION
The second generation of TOPFET devices offers
enhanced protection and drive capabilities making them
suitable for a wide variety of applications, including those
requiring fast switching (eg PWM control) or linear control.
The circuit diagram for the 5-pin TOPFET types is shown
in Fig. 1. The key features of these devices are:
SOURCE
D
TOPFET
• Overtemperature protection
• Short circuit load protection
• Overvoltage protection
• Full ESD protection
• Direct access to the gate of the Power MOSFET.
• Flag signal reporting of certain fault conditions
• Separate protection circuit supply
P
F
I
P
The 5-pin TOPFET range is summarised in Table 1.
Overtemperature protection
S
TOPFETs include an on-chip protection circuit which
measures the absolute temperature of the device. If the
Fig. 1 Schematic diagram and circuit of 5-pin TOPFET
TOPFET
Package
VDS (V)
RDS(ON) (mΩ)
at VIS (V)
for VPSP > (V)
BUK105-50L
SOT263
50
60
50
5
7
4
4.4
BUK105-50S
SOT263
50
60
50
5
7
5
5.4
Table 1. 5-pin TOPFET type range
447
Automotive
Power Semiconductor Applications
Philips Semiconductors
In the tripped condition the gate of the power MOSFET
stage is pulled down by the control logic and so current is
drawn by the input pin of the TOPFET. A minimum value
of external gate drive resistor is specified in order that the
protection circuit can turn off the PowerMOS stage and thus
protect the device. The flag pin gives a logic high output to
indicate that a fault has occurred. If the overtemperature
condition persists after the protection supply has been reset
then the protection circuit is reactivated.
Overvoltage protection
Transient overvoltage protection is an additional feature of
the TOPFET range. This is achieved by a combination of
a rugged avalanche breakdown characteristic in the
PowerMOS stage and an internal dynamic clamp circuit.
ESD protection
The input pin, flag pin and protection supply pins of the
TOPFET are all protected with ESD protection zeners.
These devices protect the PowerMOS gate and the
TOPFET circuits from ESD transients. The protection
devices cannot be used in continuous breakdown.
Short circuit protection
In the case of short circuit faults the rate of rise of
temperature in a MOSFET switch can be very rapid.
Guaranteed protection under this type of condition is best
achieved using the on-chip protection strategy which is
implemented in the TOPFET range of devices. The short
circuit protection circuit acts rapidly to protect the device if
the temperature of the TOPFET rises excessively.
Protection supply
An error condition is recorded and the flag signal is activated
if the protection supply is absent. Valid protection is only
guaranteed once the protection supply is in excess of VPSP
(See Table 1).
The TOPFET does not limit the current in the power circuit
under normal operation. This ensures that the TOPFET
does not affect the operation of circuits where large inrush
currents are required. As with the overtemperature
protection circuit the short circuit protection circuit turns off
the power MOSFET gate via the control logic and provides
a flag signal. The TOPFET is reset by taking the protection
supply pin low.
One feature of the implementation of the protection circuits
used in this generation of TOPFET devices is that the input,
flag or protection supply pins cannot be reverse biased with
respect to the source. This must be adhered to at all times.
When the TOPFET is in reverse conduction the protection
circuits are not active.
448
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.3 BUK101-50DL - a Microcontroller compatible TOPFET
The TOPFET version BUK101-50DL can be directly
controlled by the port outputs of standard microcontrollers
and other high impedance driver stages. This member of
the TOPFET family has the same functional features as its
predecessors BUK101-50GS and BUK101-50GL. All these
versions are 3-pin devices for the replacement of Power
MOSFETs or partially protected Power MOSFETs. They
are internally protected against over temperature, short
circuit load, overvoltage and electrostatic discharge. For
more information concerning the basic technical concept of
TOPFET see Philips Technical Publication ’TOPFET - A
NEW CONCEPT IN PROTECTED MOSFET’. This section
covers the special features of the BUK101-50DL version,
criteria for driver stage design and application.
V
CC
DRAIN
R out
TOPFET
R IG
INPUT
LOGIC AND
PROTECTION
V IS
Overview on BUK101-50 versions
SOURCE
The GS, GL and DL versions of the BUK101-50 TOPFET
each have the same functionality but differ in their input
characteristics. Table 1 gives an overview on these
characteristics.
Type
Nominal
Input
Voltage
(V)
Normal
Input
Current
(mA)
Latched
Input
Current
(mA)
Max. Input
Voltage
(V)
GS
GL
DL
10
5
5
1.0
0.35
0.35
4.0
2.0
0.65
11
6
6
T off
Fig. 1 Diagram of 3-pin TOPFET input
For all versions the internal circuits for over temperature
and short circuit load protection are supplied from the input
pin. This determines the input current IIS under normal
conditions, i.e. the Power MOS transistor is on and Toff in
Fig. 1 is off. To ensure proper function of the protection
circuits, a minimum input voltage VIS = 4V has to be applied.
If, however, the device has turned off due to over
temperature or short circuit load (i.e. transistor Toff in Fig. 1
is on), a minimum of VIS = 3.5V is required to keep the device
in its ’latched’ state. Latched means that the device will stay
off even if the error condition has disappeared. Figure 1
indicates that under this condition the input current IISL will
be increased due to the additional current that has to be
sourced into resistor RIG. RIG allows the Power MOS gate
to be pulled down internally while the input pin is at high
level. The typical value of RIG in the GL version is 4kΩ, while
for the DL version this value has been increased to 30kΩ.
Thus the maximum input current has been reduced to allow
for high impedance driver stages such as microcontroller
port outputs.
Table 1. Comparison of GS, GL and DL versions
Table 1 shows that the GS version (S for Standard type) is
specified for 10V driver outputs while the GL and DL
versions (L for Logic Level type) are specified for 5V logic
level driver outputs. The two logic level types differ in the
input current, IISL, which flows when the device is in its
’latched’ state i.e. shutdown has occurred due to over
temperature or short circuit load. The GL version is suitable
for pulsed applications up to 1kHz and needs a push-pull
driver stage while the DL version is optimised for high
impedance drive circuits and can handle pulsed
applications up to 100Hz.
The criteria stated above result in the following
requirements on the driver stage output resistance Rout:
Criteria for choice/design of driver stage
Normal: Rout
≤
(1)
Figure 1 shows a simplified circuit diagram for the input of
a 3-pin TOPFET. Also indicated is the high level output
impedance of the driver stage Rout.
Vcc − 4V
IIS (VIS = 4V)
Latched: Rout
≤
Vcc − 3.5V
IIS (VIS = 3.5V)
(2)
449
Automotive
Power Semiconductor Applications
Philips Semiconductors
the port output demands a minimum value for Rpull-up. For
the 80C51 microcontroller family a maximum output voltage
of Vout,low = 0.45V is specified at a sink current of 1.6mA for
ports 1 to 3 and 3.2mA for port 0. This voltage level is safely
below the minimum turn-on threshold VIS(TO) = 1V of the
TOPFET. Considering VCC = 5V and the above specification
of the port output, the minimum value for Rpull-up is:
The maximum input currents of the BUK101-50DL are
specified as follows:
IIS,max = 270µA at VIS = 4V
IISL,max = 430µA at VIS = 3.5V
Considering a 5V supply, equation (2) leads to a maximum
output resistance Rout,max = 3.5kΩ.
Rpullup
Vbatt
VCC = 5V
80C51 or
Derivative
Port
Output
≥
5V − 0.45V
1.6mA
= 2.8kΩ
Thus a value of 3kΩ meets the requirements.
Other applications for the BUK101-50DL
Load
Logic IC as driver
3k
P
Besides microcontroller port outputs the BUK101-50DL can
also be driven by standard 5V logic IC families. Table 2
gives an overview on these families and states - if necessary
- the minimum value for a pull-up resistor.
BUK101-50DL
Fig. 2. Direct control by 80C51 microcontroller
Application example - 80C51
microcontroller as TOPFET driver
Family
Rpull-up min
TTL
LSTTL
STTL
HE4000B
HCMOS
ACL
300Ω
620Ω
240Ω
no Rpull-up required
no Rpull-up required
no Rpull-up required
Table 2. 5 V logic IC families driving the BUK101-50DL
Figure 2 shows an application that takes advantage of the
low input current of the BUK101-50DL. As has been shown
above, the external pull-up resistor Rpull-up in this circuit
should have a maximum value of 3.5kΩ at VCC = 5V for safe
operation of the TOPFET protection circuits. An additional
requirement is that the TOPFET must be off when the port
output is at low level. Thus the limited sinking capability of
High Side driver
The low input current of the BUK101-50DL is also
advantageous, when using the device as a high side switch.
In this configuration the low drive requirements mean that
smaller capacitors are needed in charge pump or bootstrap
circuits. This subject is described more fully in section 5.3.6.
450
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.4 Protection with 5 pin TOPFETs
TOPFETs in the 5-pin SOT263 outline extend the range of
application of TOPFET to circuits requiring faster switching
or protected linear operation. 3-pin TOPFETs are ideal for
use in DC and low frequency switching applications but the
need to generate the protection supply from the input is a
limitation. Providing a separate pin for the protection supply
gives the designer freedom to control the input / MOSFET
gate in the way he chooses.
the MOSFET partially ON. In an overtemperature or
overload situation the TOPFET will turn on the gate
pull-down transistor and attempt to turn itself OFF.
The flag indicates when the TOPFET has been tripped by
an overtemperature, overload or short circuit condition. It
will also indicate if the protection supply is absent, for
example during a reset. It should be pointed out that the
flag low state does not mean that the protection supply is
high enough, just that it is present.
This note will look at the organisation of the 5-pin devices
and then discuss some of the more important operational
considerations. Application examples will be presented in
the later sections in this chapter.
The flag is the open drain of a MOS transistor which is OFF
to indicate a fault. It is intended that the flag pin is connected
to a 10 kΩ pull-up resistor. This arrangement gives the flag
a failsafe characteristic.
Functional description
Operational considerations
The logic and protection circuits within this device are
similar to those in the 3-pin TOPFETs but the configuration
has been modified (see Fig. 1) to give greater operational
versatility.
DRAIN
PROTECTION SUPPLY
FLAG
Supplying the protection circuits from their own pin, rather
than sharing a pin with the MOSFET gate drive, has several
beneficial effects. One is that it allows the MOSFET gate
to be independently controlled without adversely affecting
the protection features. This is particularly useful when
TOPFET is being used as a linear controller.
O/V
CLAMP
INPUT
The removal of the input gate resistor gives the designer
the opportunity of selecting the most appropriate value. It
is important to understand that if TOPFET is to protect itself,
it needs to control its gate by overriding the external drive
circuit. It can only do this if the impedance of the driver is
high enough. The conditions for satisfactory operation are
given in Table 1.
POWER
MOSFET
LOGIC AND
PROTECTION
Minimum driver impedance
SOURCE
Protection level
Fig. 1 Elements of a 5-pin TOPFET
These devices use pin 2 of the SOT263 as a flag and pin
4 as the supply / reset to the logic and protection circuits.
Separating the protection supply from the input has allowed
the internal input gate resistor to be removed. (In a 3-pin
TOPFET, this resistor is needed to maintain the protection
supply during latched fault conditions).
The operation of the protection circuits has not been
changed. If there is an overvoltage between drain and
source, the overvoltage protection circuit will still try to turn
ON drive
OFF drive
5V
10 V
input /
source
Full self protection
1 kΩ
2 kΩ
100 Ω
Overvoltage
protection only
0Ω
0Ω
100 Ω
Overtemperature,
overload and short
circuit protection only
1 kΩ
2 kΩ
0Ω
Table 1. Driver impedance and protection level
451
Automotive
Power Semiconductor Applications
Philips Semiconductors
The simplest way of satisfying the self protection
requirements is to fit a 2 kΩ resistor between the driver and
the input pin. This is simple in a linear controller but may
not be feasible in a switching controller where this
resistance will result in a significant turn OFF delay. An
alternative may be to have an ON drive via 2 kΩ and an
OFF drive via 100 Ω.
VPSP / V
BUK105-50L/S
10
8
min
BUK105-50S
6
If lower turn ON drive impedance is needed then the
approach would be to use the flag output to control the
signal being fed to the driver circuit. It should be noted that
to have overvoltage protection the turn OFF impedance
must still be > 100 Ω.
4
BUK105-50L
2
The S and L versions differ only in the protection supply
voltage range. The L types are designed to be supplied
from the output of 5 V logic ICs, like the 74HC/HCT families.
The S types are intended to be supplied with a nominal 10 V
from either HEF4000 type logic, linear ICs (e.g operational
amplifiers) or discrete circuits.
0
0
2
4
6
VIS / V
8
10
Fig. 2 Minimum protection supply for shorted load
protection
One additional benefit of the independent protection supply
is that, unlike 3-pin L types, the input of a 5-pin L type can
be as high as 11 V, allowing a significantly lower RDS(ON) to
be achieved.
The input, flag and protection supply pins are all protected
against the effects of ESD by special diodes between the
pin and source. It is important to realise that these devices
are not designed to run in continuous forward or reverse
conduction. This means that the continuous voltage
between these pins and source should be > 0 and < 11 V.
It is important to realise that, at high levels of input voltage,
the MOSFET transfer characteristic of both L and S types
will allow a very high current to flow during shorted load
situations. This current, flowing through the resistance in
the connections between the chip’s source metalisation and
the source pad on the pcb, will give a significant volt drop.
Since the return for the protection supply will be to the pcb
source bond pad, the volt drop will subtract from the
effective protection supply voltage. To compensate for this
effect, the minimum protection supply voltage, VPSP, is
increased at high levels of input voltage, VIS. For example
the minimum VPSP of the BUK105-50L is 4 V if VIS ≤ 5 V. If,
however, the input is taken to 7 V, to achieve an RDS(ON) of
50 mΩ, VPSP must be ≥ 4.4 V. A curve in data (reproduced
as Fig. 2) gives minimum VPSP values for VIS from 0 to 11 V.
Reverse Battery
There is always a risk that the car’s battery could be
reversed. If this happened to a system where a TOPFET
is fitted then the TOPFET will survive provided:
the current flowing through the body drain diode is
restricted by the load to a level which does not cause
the TOPFET to over dissipate,
the current flowing out of the input, flag and protection
supply pins is < 10 mA.
452
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.5 Driving TOPFETs
The output of a TOPFET is similar to that of a Power
MOSFET. However, the TOPFET’s protection features
make the input characteristics significantly different. As a
consequence, TOPFETs have different drive requirements.
This fact sheet describes these requirements and suggests
suitable drivers for the different TOPFET versions.
is that both the input and the protection pins need to be
supplied. The second difference is that the input resistance
is external and is selected by the designer. One requirement
which remains is the need to keep both the input-source
and protection-source voltages within the range 0 to 11 V.
3-Pin TOPFET
The protection pin driver must be able to keep the voltage
above VPSP when supplying the protection current, IPS. With
the 5-pin device the protection supply is independent, so
the current drawn when TOPFET trips does not change.
Input requirements
3-pin TOPFETs can replace ordinary MOSFETs in many
circuits if the driver can meet certain conditions. The first
of these conditions is the need to keep within the TOPFET’s
VIS ratings and in particular to keep the input positive with
respect to the source. The second is the need to provide
an adequate supply to the protection circuits even when the
TOPFET has tripped and the input current is significantly
higher.
D
Table 1 summarises these requirements. It gives the
limiting values of VIS, the minimum input voltage for valid
protection in normal and latched mode and the normal and
latched input currents for each 3-pin TOPFET.
I
P
S
Drivers
The complementary drive arrangement shown in Fig. 1 is
well suited to the input requirements of 3-pin TOPFETs.
The transistors shown are the output of a cmos IC gate,
which for some TOPFETs may have sufficient drive. If not,
a push pull drive with discrete devices should be used.
Suitable cmos families are given in Table 1.
Fig. 1 Complementary driver for 3-pin TOPFET
5V
The BUK101-50DL has a very low input current
requirement, achieved by increasing the value of the
internal input resistor - at the expense of a significant
increase in switching times. This means that this device can
be driven from the output port of an 80C51 micro controller
as shown in Fig. 2. Designers should be aware that other
high resistance / low current TOPFETs could be produced
if they are requested.
D
80C51
C
3k
I
P
5-Pin TOPFET
Input Requirements
DL
The requirements of a 5-pin TOPFET are somewhat
different to that of a 3-pin device. The first major difference
S
Fig. 2 Micro controller drive for 3-pin TOPFET
453
Automotive
Power Semiconductor Applications
Philips Semiconductors
Input voltage (V)
Type
limiting value
Input Current (mA)
Driver
for valid protection
min.
max.
normal
latched
normal
latched
BUK100-50GS
BUK101-50GS
BUK102-50GS
0
0
0
11
11
11
5
5
5
3.5
3.5
3.5
1.0
1.0
1.0
5.0
4.0
20
HEF / Discrete
HEF / Discrete
Discrete
BUK100-50GL
BUK101-50GL
BUK102-50GL
0
0
0
6
6
6
4
4
4
3.5
3.5
3.5
0.35
0.35
0.35
2.0
2.0
10
HC/HCT
HC/HCT
Discrete
BUK101-50DL
0
6
4
3.5
0.35
0.65
Micro
Table 1 Input parameters of 3-pin TOPFETs
The input pin requirements depend on the mode of
operation chosen by the designer. If the TOPFET is
expected to turn itself off, in overtemperature or shorted
load situations, then the output impedance of the drive
needs to be > 2 kΩ. This will allow the TOPFET’s internal
turn-off transistor to pull the input pin low. If, however, the
circuit uses the TOPFET flag to signal to the driver to turn
off, then driver resistance can be very much lower.
a fault. With one input high, the output of the gate will be
low turning the TOPFET off. The 100 Ω resistor ensures
that the overvoltage clamp is still operational.
D
P
Independent of which method is used for overload turn-off,
there is a separate requirement to ensure adequate
overvoltage clamping. If this feature is needed then the
input to source resistance of the driver - when it is pulling
the input low - needs to be > 100 Ω. If it is lower, then the
TOPFET’s internal clamping drive will be unable to raise
the gate voltage high enough to turn the MOSFET on.
F
I
2k
P
S
Drives
Fig. 3 Self protection driver circuit
The drive for the protection pin can, most conveniently, be
supplied by a cmos IC gate. A 74HC or HCT for L type
devices or a HEF4000 series device for the S type. Care is
needed however to ensure that the minimum protection
voltage, VPSP, requirements are still met when the input
voltage, VIS is high and the load is shorted.
Vcc
D
10k
A typical high impedance drive arrangement, which lets
TOPFET
protect
itself
against
shorted
load,
overtemperature and overvoltage, is shown in Fig. 3.
P
F
P
I
1
One method of creating a fast drive is shown in Fig. 4. In
this arrangement a NOR gate with a low impedance output
stage drives the input via a 100 Ω resistor. One input of the
NOR gate is connected to the flag pin and will be pulled
high by the 10 kΩ pull-up resistor if the TOPFET indicates
100
Fig. 4 Fast driver for 5-pin TOPFET
454
S
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.6 High Side PWM Lamp Dimmer using TOPFET
Although the 3-pin TOPFETs were designed for low side
switch applications, they can, by using standard MOSFET
bootstrap techniques, be used in applications which need
high side control. One such application is the dimming of
automotive headlamps and panel lamps. These
applications need not only a high side switch but also slow,
controlled switching to reduce problems of EMI.
Vbat
100
D1
BAW62
R1
BUK101-50GS
TOPFET
2k2
C1
10uF
P
D2
BAW62
This note will give details of a circuit which fulfils the
operational requirements of this application and, because
it uses a TOPFET, is well protected against shorted load
and overvoltage faults.
D3
c10
C2
3n3
PWM signal
R2
Q1
BC337
(10V peak)
47k
Lamps
R3
10k
Circuit Description
Fig. 1 High side lamp dimmer circuit
The circuit shown in Fig. 1 shows the high side PWM
dimmer circuit. All the main components are shown, the
only exception being the source of the PWM control signal.
This could be either the system controller or a dedicated
oscillator depending on the nature of the overall system.
The circuit of Fig. 1 assumes that the signal is a rectangular
pulse train of the required frequency and duty cycle, with
an amplitude of 10 V.
Component Values
With the components specified the circuit will operate at a
frequency between 50 and 200 Hz and has rise and fall
times of about 300 µs. This slow switching means that the
minimum OFF time, for satisfactory bootstrap operation, is
about 1 ms. At 50 Hz this gives a maximum duty cycle of
95%.
The input signal is attenuated by R2 and R3 and fed to the
base of Q1. The combination of R1 and Q1 will invert and
level shift the signal and feed it to the input of the
BUK101-50GS TOPFET.
The value of C1 has been chosen to ensure that TOPFET
input current does not cause the C1 voltage to fall
significantly during the maximum ON time. This means that
the lowest on state dissipation is being achieved. Lower
values could be used but the voltage droop would be greater
and care would be needed to ensure that the input voltage
does not fall below the VISP of the TOPFET, otherwise the
protection features may not function.
D1, C1 and the TOPFET form the bootstrap circuit. The low
end of C1 is connected to the TOPFET source. When
TOPFET is OFF its source is close to ground, so C1 charges
to Vbat via D1. When TOPFET turns ON, its source rises
to nearly Vbat, lifting the high end of C1 well above Vbat.
C1 can, therefore, provide more than enough voltage to
drive the TOPFET input. In fact, when Vbat is higher than
normal, the voltage would exceed the continuous VIS rating
of the BUK101-50GS, so D3 is included to restrict the input
voltage to below 11 V.
The rate of switching can be changed by adjusting the value
of C2. Larger values would reduce switching speed.
Considerable care is needed when switching times become
very long because while the input voltage is below the VISP
the TOPFET is unprotected. Switching times can be
reduced to about 50 µs by reducing the value of C2 to
470 pF. To reduce the switching times further will mean a
change to the input drive.
Capacitor C2 adds to the Miller capacitance of Q1 and limits
the rate of change of collector voltage. The TOPFET acts
like a source follower circuit, so the load voltage rises and
falls at the same rate as the collector-emitter voltage of Q1.
455
Automotive
Power Semiconductor Applications
Philips Semiconductors
TOPFET input will be negative while its drain-source
voltage is high. This may damage the TOPFET. This
difficulty can be eliminated by the input circuit shown in
Fig. 3. In this circuit diode D4 will turn off if the source
voltage rises. The input is, therefore, no longer clamped by
the drive and can rise with the source, eliminating the risk
of damage.
to Vbat
R4
10k
C2
3n3
D1
PWM signal
R2
BAW62
Q1
BC337
(5V peak)
22k
D5
c5
R1
R3
10k
D4
BUK101-50GS
TOPFET
2k2
C1
P
10uF
Fig. 2 5 V input stage
D2
BAW62
Switching rate, in particular the turn-off rate, is also
influenced by the amplitude of the input signal. R2 and R3
have been chosen to give similar rise and fall times with an
input of 10 V. If the input amplitude is lower the fall time
would increase. This can be compensated for by lower
attenuation. An input modified for 5 V input is shown in
Fig. 2. This arrangement also includes D5 to clamp the input
voltage to 5 V and R4 to allow the use of an open collector
or drain drivers.
D3
c10
Fig. 3 Protection against negative input
If the load becomes short circuit, TOPFET will trip as soon
as the temperature of the power part of the chip becomes
too high. Since this circuit is a PWM controller the TOPFET
will be reset at the end of the ON period. During the period
between tripping and the start of the next cycle the TOPFET
will cool. It will, therefore, turn on when the input goes high
again and the short circuit current will flow until TOPFET is
tripped once more.
Operation in fault conditions
TOPFET will protect itself against high voltage supply line
transients by partially turning on and restricting the applied
voltage to about 60 V. In high side applications the
remainder of high voltage may appear across the load. In
many systems the grounding and smoothing arrangements
will ensure that this will not be problem. In some
configurations the TOPFET source will rise above ground
while the input is held at ground. This means that the
TOPFET is able to withstand this type of operation for a
considerable period of time but not necessarily indefinitely.
The dissipation is considerable, the temperatures could be
high and operating life may be affected. It is advisable,
therefore, that short circuit operation is evaluated.
456
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.7 Linear Control with TOPFET
Although the pulse width modulation, PWM, method of
motor speed control is often preferred over the linear
method it is not without problems. Some of these are totally
eliminated in linear controllers. However, linear control
techniques have their own limitations. By using a Philips
TOPFET as the power device some of the disadvantages
are removed giving a fully protected, linear control system.
TOPFET in Linear Control
The circuit shown in Fig. 10. is a linear controller for a car
heater fan based around a BUK105-50S. TOPFET is well
suited to this application because it is a real power device
in a real power package giving it good thermal
characteristics and low RDS(ON). The 5-pin TOPFET is used
because the protection circuits need to be supplied
independently from the input. The on-chip overtemperature
protection feature of TOPFET is precisely the protection
strategy needed in this type of high dissipation application.
This note will compare linear and PWM controllers. It will
then give details of a circuit based around a BUK105-50S
which shows that, with a TOPFET, it is simple to produce
a fully protected, linear controller for adjusting the speed of
a car heater fan.
Input Pin
Linear and PWM Control
PWM is often selected as the method of controlling the
speed of a brush motor because it is more efficient than
linear control. The reduction in energy loss results from a
reduction in the loss in the controlling power device. The
loss is lower because the device is only transiently in the
high dissipation state of being partially ON. To keep the
loss as low as possible, the transition time needs to be kept
short, implying fast switching and high values of dV/dt and
dI/dt. It is these fast switching rates which create the
electrical noise that can be such a problem in automotive
applications.
In this circuit the input of the TOPFET is connected, via R1
and D1, to the output of an operational amplifier. The
TOPFET drain voltage is attenuated by R2/R3 and fed to
the positive input of the amplifier. The negative input is
connected to the wiper of the speed setting potentiometer.
This
TOPFET/op-amp
arrangement
creates
a
non-inverting amplifier with a gain of
Linear control does not create this noise because it holds
the output at a steady value. The power device is
continuously in the partially ON state and its dissipation is
high. If, however, this heat can be handled and the
inefficiency is acceptable then linear control may be the
better choice.
In such a low frequency system the presence of R1 at
2.2 kΩ will not have a significant effect on normal operation.
However, if TOPFET is tripped, its internal gate source
transistor will be turned on and, because R1 is greater than
the 2 kΩ needed for self protection (see RI in the data
sheet), the MOSFET gate will be pulled down and the
TOPFET will be OFF.
gain =
Device Selection Factors
(R2 + R3)
R3
Diode D1 prevents the input of the TOPFET being pulled
negative with respect to the source.
In PWM control, on-state dissipation is the major energy
loss, so RDS(ON) is the main selection criterion. In linear
control, maximum dissipation occurs when half the supply
voltage is being dropped across the device. In this state
RDS(ON) is not relevant as dissipation is being controlled by
the load and the supply. The limiting factor in this case is
the need to dissipate the energy and keep the junction
temperature to a safe value. The selection, therefore, is
based on junction to mounting base and mounting base to
heatsink thermal resistance. RDS(ON) cannot be ignored,
however, because it sets the residual voltage loss at
maximum speed which can be important.
Protection Supply
To ensure that the overtemperature and shorted load
protection circuits work, the protection supply pin needs to
be connected to an adequate supply. To allow TOPFET to
be reset, provision has to be made to switch this so it can
fall below the minimum reset voltage, VPSR. Possibly the
easiest way to achieve this is by feeding the protection
supply from a CMOS gate.
457
Automotive
Power Semiconductor Applications
Philips Semiconductors
Vbat
100
10k
+
c10
Fan motor
D2
reset
10k
100k
D4
R4
LED
+
D3
100k
R2
22k
100k
10k
R3
10k
D1
P
F
P
I
R1
2k2
+
-
BUK105-50S
TOPFET
47k
47n
Fig. 1 Linear speed controller circuit
Two version of BUK105-50 are available, ’S’ and ’L’. They
differ in their protection supply requirements. L devices are
designed to operate from a nominal 5 V. This makes them
compatible with 5 V logic families like the 74HC and HCT
series. L types can be driven at 10 V but as curves in the
data show the protection characteristics are affected. On
the other hand, S devices are designed to work with a
nominal 10 V such as is available from HEF4000 logic
gates.
controller could use this information to initiate a reset
sequence or perhaps shut down the circuit and record the
fact in a maintenance record store.
In this simpler system the flag output feeds the input of an
op-amp wired as a comparator which in turn indicates a
fault by lighting a LED. The output is also fed via D3 to the
input of the speed controller op-amp. This overrides the
signal from the speed adjusting potentiometer and takes
the TOPFET input low. This arrangement has been used even though the circuit has been designed to allow the
TOPFET to self protect - to prevent the TOPFET from
turning back on when there is no protection supply, for
example during reset.
If this circuit were part of a larger system then it is likely that
such a gate would be available. In the circuit given here the
protection pin is connected to the output of an op-amp wired
as a non-inverting buffer. The buffer input is pulled up to
the +ve rail with 10 kΩ. The protection supply can be taken
low - to reset the TOPFET - by a pushbutton which grounds
the input of the buffer.
Drain pin
Freewheel diode D4 is needed if the energy stored in the
motor inductance exceeds the TOPFET’s non-repetitive
inductive turn-off energy rating at the designed operating
junction temperature. The overvoltage clamping of the
TOPFET is still needed, however, to protect against supply
line transients.
Flag pin
In this circuit the flag pin is connected to a 10 kΩ pull-up
resistor, R4. In a more sophisticated system this signal
could then be fed to the input of a logic gate and used to
inform the system controller of a fault condition. The
458
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.8 PWM Control with TOPFET
Resistor R15 is included between the driver T4/T5 and the
TOPFET input to ensure proper function of the TOPFET’s
internal overvoltage protection. This overvoltage protection
is an active clamp circuit that will try to pull up the gate of
the TOPFET’s power MOSFET (i.e. the input pin) if the
drain-source voltage exceeds 50V. A minimum resistance
of 100Ω between input and ground is needed for the active
clamp to succeed.
Speed control of permanent magnet dc motors is required
in many automotive and industrial applications, such as
blower fan drives. The need for protected load outputs in
such systems can be met by using a TOPFET with its
inherent protection against short circuit, overtemperature,
overvoltage and ESD. In section 5.3.7 the two basic
methods for speed control, linear and PWM, are compared
and discussed and a circuit example for linear control is
given. This section gives an example of a PWM drive circuit
using a 5-pin TOPFET.
If the load is shorted or the TOPFET’s junction temperature
is too high, the internal sensors of the TOPFET will detect
it and inform the protection logic which will turn off the
internal flag transistor. The flag pin, which is connected to
the drain of this transistor, will be pulled high by resistor
R16. This will turn on transistor T3 pulling the input to the
driver stage, T4/T5, low and hence turning the TOPFET off.
Circuit Description
The circuit shown in Fig. 1 contains all the elements needed
to produce a PWM circuit which can control the speed of a
heater fan motor. The power device, because it is a
TOPFET, can survive if the load is partially or completed
shorted, if overvoltage transients appear on the supply lines
or if the cooling is, or becomes, insufficient.
The TOPFET will remain in this state - even if the error
condition disappears - until a reset is applied. The 5-pin
TOPFETs are reset by taking the protection SUPPLY pin
below VPSR. In this circuit this is done by closing the reset
switch, pulling the protection pin to ground. In this state
there is no protection supply so the TOPFET is unprotected.
However, the TOPFET indicates the absence of a
protection supply by the flag transistor remaining off. In this
circuit this causes the drive to the TOPFET to be low hence
the TOPFET will stay off. The TOPFET will resume normal
operation when the reset switch is opened and the
protection supply is re-established.
In a PWM control system the supply to the motor has to be
switched periodically at a frequency significantly above its
mechanical time constant. The net armature voltage and
thus the motor speed is controlled by the duty cycle, i.e.
on-time/period, of the control signal. With the component
values shown, the circuit operates at a frequency of 20kHz.
This means that any mechanical noise created by the
switching is ultrasonic. The main building blocks of the
circuit are the PWM generator, the power driver and the
interface between the two.
PWM Generator
Power Stage
In Fig. 1, OP1 together with T1 and T2 form a saw-tooth
generator, whose frequency is determined by R1 and C1.
OP2 compares the saw-tooth voltage waveform at its
inverting input with the voltage determined by the
potentiometer P1. The output of OP2 is high as long as the
saw-tooth voltage is less than the P1 voltage. As a result,
the higher the voltage at P1, the longer the positive pulse
width and thus the higher the duty cycle of the signal at the
output of OP2.
In this circuit, the main power switch is a BUK105-50L which
has an RDS(ON) of 60 mΩ @ VIS = 5 V. The L version of the
BUK105 has been chosen so that the protection supply can
be fed from the available 5 V supply. The maximum
protection supply current, IPS, is 350 µA, the voltage drop
across R17 could be 0.42 V. Even if the voltage is regulated
as low as 4.5 V, the protection supply will still be > 4 V, the
minimum VPSP for valid protection with a VIS of 5 V.
If a lower RDS(ON) were needed this could be achieved by
modifying the circuit to give a higher VIS on the TOPFET.
A VIS = 7 V would give an RDS(ON) = 50 mΩ. An input voltage
as high as 10 V could be used but any increase must be
accompanied by an increase in the protection supply
voltage. A curve showing the required VPSP for the full range
of input voltage is given in the data sheet.
Interface PWM Generator - TOPFET
The output signal of OP2 is fed to emitter-followers T4 and
T5. These act as a low impedance driver for the input of
the TOPFET. The drive is needed to achieve the short
switching times which keep the dynamic switching losses
of the TOPFET below the on-state losses.
459
Automotive
Power Semiconductor Applications
Philips Semiconductors
The given circuit can be used in both 12V and 24V systems
because, with an input voltage of 5V, the TOPFET is short
circuit protected up to a supply voltage of 35V. However, if
a supply this high is expected then the dissipation and
voltage rating of the regulator would need to be studied.
repetitively clamping at 20 kHz is very high, much higher
than any power switch of this size would be able to handle.
R18/C4 are optional devices that slow down switching,
reducing dV/dt and hence RF noise emission.
D1 is a freewheel diode across the motor load which must
be present even though the TOPFET has an internal clamp
circuit. This is because the dissipation resulting from
Capacitor C5 helps to decouple the circuit from the supply
and prevents excessive dI/dt on the power lines and the
excessive voltage spikes it would produce.
R17
10
Vbat
LM78L05
R2
33k
R6
8k2
R10
12k
C2
22u
R11
1k
C3
1u5
+
T4
BC
548
OP1
-
R5
1k2
R7
6k8
+
C1
10n
T1
BC
548
P1
10k
D
P
F
OP2
I
T5
BC
558
R8
6k8
R3
22k
R17
1k2
R15
100
T2
BC
548
R4
22
R16
10k
T3
BC
548
RESET
OP1,OP2: LM 393
Fig. 1. PWM Control Circuit using TOPFET
460
R18
47
P
TOPFET
BUK10550L
R12
8k2
D1
BYV
28
M
R1
6k8
C5
1000u
63V
S
C4
470p
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.9 Isolated Drive for TOPFET
An isolated drive for a power transistor is required if an
electronic replacement of an electromechanical relay is to
be realised. By using a TOPFET, with its integrated
protection functions, in combination with an isolated input
drive, the following advantages over an electromechanical
relay can be achieved:
o
o
o
o
o
Transformer Isolated Drives
As with photo-cells, pulse transformers provide a means of
transferring energy from the control pin to the input of the
power device. However, the transfer efficiency of a pulse
transformer is much higher, so the protection circuits of a
TOPFET can be supplied satisfactorily.
Permanent short circuit protection
Over temperature protection
Active clamping at inductive turn-off
Logic level control
Higher switching frequency
Extremely small pulse transformers are now available, and
some outlines are suitable for surface mount. It is, therefore,
realistic and practical to use this method to create a relay
replacement for high and low side configurations.
This section presents a complete circuit example of a
transformer isolated drive. It also discusses other isolation
techniques particularly in relation to meeting TOPFET’s
specific requirements.
Basic Methods for Isolated INPUT Drives
Circuit Description
Opto-Isolated Drives
Figure 1 shows a transformer-isolated drive circuit for
TOPFET. As discussed above, a TOPFET in combination
with this drive circuit can be employed either in high side
or low side configuration without modifications on the driver
side. The drive signal on the transformer’s primary side is
a pulse train that is rectified on the secondary side to provide
a continuous input voltage VIS for the TOPFET. For the given
dimensioning, a pulse rate in the range of 100kHz is well
suited. A high pulse rate is advantageous as it allows the
dimensions of the transformer and smoothing capacitor,
C2, to be minimised.
For this method a light emitter (e.g. LED or lamp) and a
photo-device is needed. The latter can be subdivided into
two groups:
Photo Resistors/Transistors
With these devices a ’switch’ can be built to control the input
voltage of a TOPFET. They cannot provide the power
needed to drive the input so a separate supply is needed.
In low side configurations this can be the main supply
directly. In high side configurations an input voltage above
main supply level is needed which could be generated by
a charge pump. However, the supply connection needed
for this type of opto-isolated drive is not needed with an
electromechanical relay. So an opto-isolated drive with
photo resistors/transistors cannot serve as a universal relay
replacement.
On the primary side, a voltage is applied to the transformer
when T1 is on. The positive pulse amplitude is limited by
D7 on the secondary side. The drain current of T1 and the
transformer current are limited by R1.
During the off period of T1, the transformer’s primary current
freewheels through D1 and D2. Thus the absolute
maximum value for the negative pulse amplitude on the
primary winding is equal to the sum of breakdown voltage
of zener diode D2 and forward diode drop across D1. At a
duty cycle of 50%, this value should be at least as high as
the positive pulse amplitude. This allows the primary current
to reach zero and thus the magnetic flux in the core to be
reset while T1 is off. The maximum off-state drain-source
voltage of T1 occurs if the secondary winding of the
transformer is left open. It is the sum of supply voltage VP,
zener voltage of D2 and forward voltage drop across D1.
Photo Cells
The drive energy from a control pin can be transferred to
the input pin of a power device by means of photo cells.
This would eliminate the need for the additional supply
connection. Integrated devices exist that combine an LED
and a chain of photo-cells. They are designed to drive
ordinary power MOSFETs so their output current, due to
the low efficiency of the photo-cells, is only a few µA. This
is not enough to supply the protection circuits of a TOPFET
so this method cannot be used to provide isolated drive for
a TOPFET.
461
Automotive
Power Semiconductor Applications
Philips Semiconductors
the minimum supply voltage VP on the primary side, the
transformer ratio, and the diode voltage drops at the bridge
rectifier. Zener diode D7 ensures that VIS cannot exceed
the upper voltage limit.
TOPFET
BUK101-50GL
VP
D3...D6
4x1N4148
P
R1
The time constant of R2 and smoothing capacitor C2
determine the fall time of VIS after the control input at the
primary side goes low. A fall time significantly longer than
that chosen here should be avoided for the following
reason.
D7
5V1
240
C2
10n
D1
1N4148
Control
Input
After a TOPFET has turned off to protect itself, it is latched
off so it stays in the off-state as long as VIS is high. To reset
the TOPFET, VIS must go low. In this circuit this happens
when the control input at the primary side goes low,
disconnecting the drive pulses from the gate of T1. On the
secondary side, this allows C2 to be discharged by R2 and
hence VIS to decrease. When VIS has fallen below the
protection reset voltage level VISR, the fault latch will reset
and an internal transistor, which holds the gate low, will turn
off. The gate voltage will now rise to the C2 voltage and the
TOPFET’s output MOSFET will conduct again. The
MOSFET will be fully off when VIS falls below the TOPFET
threshold voltage VIS(TO). In the range between VISR and
VIS(TO) (max. 3.5V-1V for the BUK101-50GL) the output
MOSFET may conduct while the protection circuits are
non-active. For safe reset of a latched TOPFET with a
shorted load, this VIS-range must be passed through within
a limited time interval. With the dimensioning of R2 and C2
shown in Fig. 1, this time interval is approximately 130 µs.
The BUK101-50GL is guaranteed to withstand a hard short
circuit for > 300 µs at a battery voltage of 35V and VIS=5V.
So the chosen values of R2 and C2 ensure safe turn-off of
the TOPFET.
Pulse Transformer
e.g. PE5163X
D2
15v
100kHz
R2
5k6
T1
&
BST72A
R3
1k
Fig. 1 Transformer-Isolated Drive Circuit for TOPFET
Using a bridge rectifier on the secondary side makes use
of both positive and negative pulses to generate the
input-to-source voltage VIS for driving the TOPFET. This
increases the efficiency. It also reduces the ripple on VIS,
therefore the ripple on the load current and hence the
electromagnetic noise emission.
The minimum value for VIS is set by the need to have enough
voltage for correct operation of the TOPFET’s overload
protection circuits. The maximum is determined by the
breakdown voltage of the ESD protection diode at the input
pin. Taking this into account, VIS should be within the range
of 4V-6V in the case of the TOPFET type BUK101-50GL.
In the given circuit the lower limit of VIS is determined by
462
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.10 3 pin and 5 pin TOPFET Leadforms
The TOPFET (Temperature and Overload Protected
MOSFET) range of devices from Philips Semiconductors
is based on conventional vertical power MOSFET
technology with the advantages of on-chip protection
circuitry. Using this approach the devices are able to
achieve the very low values of RDS(on) which are required
in applications for automotive and other power circuits.
TOPFET devices are currently available in two topologies
for maximum compatibility with the requirements of circuit
designers.
3-pin TOPFETs
TO220
Rthj-mb
(K/W)
5-pin TOPFETs
SOT263
Rthj-mb
(K/W)
BUK100-50GL
BUK100-50GS
BUK101-50GL
BUK101-50GS
BUK102-50GL
BUK102-50GS
3.1
3.1
1.67
1.67
1.0
1.0
BUK104-50L
BUK104-50S
BUK105-50L
BUK105-50S
BUK106-50L
BUK106-50S
3.1
3.1
1.67
1.67
1.0
1.0
Although these outlines are industry standards, on
occasions users have the need to form the leads of the
devices to accommodate a variety of assembly
requirements. Philips Semiconductors can offer a number
of standard pre-formed leadbend options to make the
purchase and specification of leadformed devices easier.
These pre-forms satisfy the basic rules concerning the
bending and forming of copper leads and ensure that, for
example, the bend radius is not less than the thickness of
the lead and that there is sufficient material at the base of
the plastic moulding to enable the act of pre-forming to take
place without damage to the crystal or its die attach and
wire-bonding.
10.3
max
4.5 min
1.3
3.6
2.8
5.9
min
Table 1. 3-pin and 5-pin TOPFET type ranges
15.8
max
2.4
max
10,3
max
4,5 min
1,3
3,7
2,8
13.5
min
5,9
min
0.6
min (4 x)
1.7
15,8
max
(4 x)
0.9 max
0.6
2.4
(5 x)
Fig. 2 SOT263
5,1
13,5
min
Figure 3 shows leadform option L02 for a TO220 type. A
device with this standard leadbend can be ordered by
specifying /L02 as the suffix for the device type. For
example, a BUK101-50GL with this leadbend is specified
by ordering type BUK101-50GL/L02.
1,3 max (2x)
0,9 max (3x)
2,54 2,54
0,6
2,4
Fig. 1 TO220AB
In addition to this, there is often the necessity to crop the
tab off the device to make a low profile version, when height
above the pcb is restricted. Again, without control, there is
a risk of fracturing the crystal during this process but Philips
Semiconductors can offer this option (SOT226), shown in
Fig. 4, which can be ordered by specifying the suffix /CR
to the device type number, eg. BUK102-50GS/CR.
The 3-pin TOPFETs are assembled in the standard
TO220-AB package (Fig. 1), which is also sometimes
known as SOT78. The 5-pin versions are assembled in the
SOT263 PENTAWATT package (Fig. 2). Depending upon
the load and the application the devices can be operated
in free air or attached to a heatsink. When using a heatsink
the advantage of these outlines lies in the very low thermal
impedance which can be achieved. Table 1 shows the
thermal resistances for the range of TOPFET devices.
For surface mountable TOPFETs, the leadbend option L06
means that the device can be used in applications where
a low profile is required. With this option an electrical contact
463
Automotive
Power Semiconductor Applications
Philips Semiconductors
from the pcb to the tab of the device is possible. The device
is shown in Fig. 5 and, for the BUK100-50GS would be
specified as the BUK-100-50GS/CRL06.
1.3
0.125 +0.125
-
For the 5-pin TOPFET the device is available in the leadbent
SOT263 outline as standard (Fig. 6). For the leadform
option the device type number is modified by the addition
of the suffix P to the SOT263 type name, eg BUK104-50L
(SOT263) becomes BUK104-50LP (leadbent SOT263).
2.5 +- 0.3
3.4 +-0.2
1.3 +0.2
0.72 +0.15
FLAT
1.3 MIN. TINNED
Bend radius 1.0
30o
Bend radius 0.5
Fig. 5 SOT226, L06 leadbend
12 +- 0.5
2.54 +- 0.3
Fig. 3 TO220, L02 leadbend
10.3
max
1.3
4.5 min
1.3
3.6
2.8
5.9
min
15.8
max
2.4
max
5.6
9.75
5
0.6
min (4 x)
12.7 min
1.7
(4 x)
4.5
0.9 max
(5 x)
Fig. 4 SOT226
0.6
2.4
8.2
Fig. 6 SOT263, leadbend
464
All radii >0.5
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.11 TOPFET Input Voltage
With an input potential lower than the source potential, the
input acts as an emitter, the drain as a collector and the
source as a base, so the potential difference will act as bias
for the parasitic transistor. The diffusion concentrations
used to create a good ESD protection diode create a
transistor with a limited forward SOA. The characteristics
of the transistor mean that it can be damaged if its VCE is
greater than 30 V when its base is forward biased. For the
TOPFET this means that damage could be caused only if
the input goes negative while the drain voltage is > 30 V.
Low side TOPFET data sheets specify that the voltage
between the input and source pins should not be less than
0 V, in other words should not go negative. In many
circumstances, sound layout using normal logic gates will
ensure that this condition is always satisfied. However, in
some situations it is difficult to design a circuit in which this
condition is met under all conditions. This section explains
the reason for the quoted rating and shows that it is a limit
in only a few circumstances. The paper will also illustrate
how negative inputs can be generated. Section 5.1.12
shows how negative inputs can be prevented and
recommends a simple method of stopping a TOPFET being
damaged if negative inputs do occur.
It should be noted that the conditions which may damage
the transistor assume the impedance of the bias supply is
low. If the bias is restricted the limits of SOA are different
so the drain voltage needed for damage will be different. In
any event at drain voltages < 30 V, a negative input will
cause the parasitic transistor to conduct but will not cause
damage.
Reason for specification limit
All the pins of a low side TOPFET are protected against
ESD. The input pin - the most sensitive pin of a normal
MOSFET - is protected by a special diode connected
between the input and the source. In the presence of an
ESD pulse, this diode conducts and clamps the voltage on
the input pin to a safe level.
Conditions creating negative input
The most obvious effect of the minimum VIS is to preclude
the use of negative drive to speed up turn off. However, this
technique is only justifiable in very high frequency circuits
and TOPFET is intended for use in DC or low frequency
applications, so it is unlikely that this type of drive will be
under consideration. The typical TOPFET driver stage will
be unipolar using gates or discrete transistors from positive
supply rails only. These drivers will turn the TOPFET off
either by removing the drive and allowing TOPFET to turn
itself off, via its internal pull down resistor, or by pulling the
input to zero volts. It would appear, therefore, that negative
inputs should not occur, but in some situations and with
some circuit configurations they can.
The diode is formed by an area of n++ in a p+ region which
is diffused into the n- epi layer, see Fig. 1. The input pin is
connected to the n++ region and then to the rest of the
circuits. The p+ region is connected by the metalisation to
the source area of the power MOSFET part of the TOPFET.
However, the p+ region also connects to the n- epi layer
and hence to the drain via the n+ substrate. The ESD diode
is formed by the n++ / p+ junction. However, the n++ and
p+ diffusions in the n- epi also create a parasitic npn
transistor. It is the presence of this transistor which makes
the negative input rating necessary.
anode of
ESD diode
to source
p++
from input pin
to other circuits
n++
p+
High side circuits
cathode of
ESD diode
A negative input can be created if an overvoltage transient
is applied to an off-state TOPFET being used as a high side
switch. A TOPFET will start to conduct if a supply line
voltage transient exceeds its clamping voltage. The current
now flowing through the TOPFET will also flow through the
low side load, raising the source potential above ground.
The driver stage may be designed to turn the TOPFET off
by pulling the input to ground as in Fig. 2. If it is, then the
conditions for harmful negative input have been created the drain voltage is > 30 V, the input is at ground and the
source potential is higher, so the input is negative.
p++
n- epi
n+ substrate
Fig. 1 Cross section of TOPFET ESD diode
465
Automotive
Power Semiconductor Applications
Philips Semiconductors
may a considerable distance from the TOPFET. The
resistance of the wiring will be low but even 20 mΩ may be
significant if the current is high.
Vbatt
Vbatt
TOPFET
VCC
LOAD
TOPFET
LOAD
signal
ground
power
ground
Fig. 3 Low side switch - separate grounds
Fig. 2 Driver taking input to ground - high side
There are two occasions when a large enough current could
be flowing. The first is during the turn-on of a load with a
high inrush current, for example a cold incandescent lamp.
The second is when the load is shorted out. If the TOPFET
turns off while this current is flowing, the energy in the
inductance of the wiring from the load to the TOPFET drain
would raise the drain voltage, possibly to greater than 30 V.
The high current, as high as 60 A, in the source to ground
wiring, say 20 mΩ, would raise the source 1.2 V above
ground. So, the combination of conditions which may
damage a TOPFET have been created.
Low side circuits
In some circumstances it is possible to create negative input
in a low side configuration. In the previous example it was
a small current in relatively large resistance that raised the
source above ground. The same effect can be created by
a large current in the low, but not negligible, resistance of
the wiring between the source pin and ground.
Systems are often configured with separate power and
signal grounds and it is possible that the driver will be
referenced to signal ground, see Fig. 3. In this case the
TOPFET input will be pulled to signal ground potential when
it is being turned off. The source will be connected to power
ground and the common connection between the grounds
The circuits and circumstances mentioned in this paper are
only examples and other hazardous negative input
situations will exist. Methods of preventing negative input
and of stopping a TOPFET being damaged, if negative
inputs do occur, is presented in section 5.1.12.
466
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.12 Negative Input and TOPFET
The second method is to arrange the drive so that it turns
the TOPFET off by pulling the input to the source rather
than to ground. The circuit shown in Fig. 1 shows a high
side drive in which this has been achieved. The TOPFET
is turned off by a pnp transistor being turned on and pulling
the input to the source.
Low side TOPFET data sheets specify that the voltage
between the input and source pins should not be less than
0 V, ie. should not go negative. This limit is needed to
prevent the parasitic transistor, formed by the input ESD
protection diode in the n- epi, being damaged in some
circumstances. The reason for the limit and the causes of
potentially damaging conditions are discussed more fully
in section 5.3.11. This section will show how damaging
negative inputs can be prevented and recommend a simple
method of stopping low side TOPFETs being damaged if
negative inputs do occur.
Vbatt
VCC
Vbatt
LOAD
TOPFET
TOPFET
Fig. 2 Low side driver taking input to source
Figure 2 shows a low side drive where the GND pin of the
cmos gate is connected as close as possible to the TOPFET
source pin. Once more the effect is to turn off the TOPFET
by pulling the input to source.
LOAD
If negative inputs cannot be avoided
Fig. 1 High side driver taking input to source
The technique of referencing drivers to the source pin helps
prevent negative inputs being generated. It is used in most
power MOSFET switching situations and should be used
with TOPFET wherever possible. If negative inputs cannot
be eliminated there are ways of preventing them from
causing damage to a TOPFET.
Avoiding negative input
Section 5.3.11 gave examples of high and low side drive
configurations which could, in some circumstances,
generate a potentially damaging negative input. There are
two ways to prevent the input from being taken too low. The
first is to fit a diode in series with the input pin. The cathode
of the diode would be connected to the TOPFET. The diode
would conduct while the driver output was high but would
turn off and isolate the input pin when the driver tried to pull
the input low. The driver would now not be driving the
TOPFET off but would be allowing it to turn itself off via its
internal input - source resistor.
Although published data gives 0 V as the lower limit of VIS,
lower values can be acceptable. The VIS limit of 0 V ensures
that the SOA of the parasitic transistor associated with the
ESD diode is never exceeded. The arrangement shown in
Fig. 3. can be used to ensure this. This shows the parasitic
npn transistor of the TOPFET and two additional
anti-parallel diodes in series with the input.
467
Automotive
R
Power Semiconductor Applications
Philips Semiconductors
A
D1
INPUT
ESD
diode
During a normal turn-off the gate discharge current will flow
through the forward biased D2. No special measures are
needed to cope with D2’s voltage drop because 0.5 V is
well below the TOPFET’s threshold voltage so it will be
properly turned off if point A is taken to 0 V.
DRAIN
D2
Vbatt
SOURCE
D3
If the drive voltage goes negative, the diode D1 (see fig. 3)
is reverse biased and diodes D2 and D3 are forward biased.
The voltage between Source and point A is limited by D3
and the current is limited by R. This voltage is divided
between D2 and the base-emitter junction of the ESD diode.
The current flowing through the ESD diode’s base-emitter
junction is therefore negligible and so the SOA of this
transistor is not exceeded. This means that all the
conditions needed to damage the device can be avoided
and the TOPFET is protected against negative input.
R
TOPFET
Fig. 5 Low side drive with series anti-parallel diodes
Figure 4 shows the high side drive of Fig. 1 modified to
include the series anti-parallel diodes, D1 and D2. D3 is
already present in the form of the input voltage limiting zener
so the only extra components are the series anti-parallel
diodes. A modified low side drive is shown in Fig. 5. Here
D1 and D2 are fitted between the output of a cmos gate
and the TOPFET input pin. In this circuit, diode limiting is
provided by the bipolar parasitic diode inherent in cmos
output stages.
TOPFET
D1
D3
D1
D2
Vbatt
R
LOAD
VCC
Fig. 3 Equivalent circuit of protected TOPFET input
D2
Series resistor values
The recommended minimum resistor values are,
LOAD
Types
Over-voltage
transient
Minimum series
resistor
3-Pin
< 200 V for 2 ms
50 Ω
3-Pin
< 300 V for 2 ms
300 Ω
5-Pin
< 100 V for 2 ms
200 Ω
Fig. 4 High side drive with series anti-parallel diodes
5-Pin
< 200 V for 2 ms
1000 Ω
In the normal on state, D1 will be forward biased but it will
create a voltage drop of about 0.5 V between point A and
the TOPFET input. To enable a 3 pin TOPFET to protect
itself, its input must be >4.0 V so the designer needs to
ensure that the voltage at A is >4.5 V.
5-Pin
< 300 V for 2 ms
2000 Ω
If the negative voltage between point A and the source is
present for a longer period of time than 2 ms then a larger
value of series resistor may be required.
468
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.13 Switching Inductive Loads with TOPFET
If there is current flowing in the coil of a solenoid or a relay
then there is energy stored in the inductance. At turn-off
this energy has to be removed from the coil and dissipated
somewhere. During this process, an extremely high voltage
will be generated unless measures are taken to limit it. This
voltage can lead to breakdown and, beyond a certain
energy level, damage to the switching transistor. Common
methods of controlling this voltage are a freewheel diode
in parallel with the inductor or a suppressor diode in parallel
with the switching transistor.
Saving of external overvoltage protection
The TOPFET clamp feature is the only voltage limiting
required if the energy associated with turn-off, Eclamp, does
not increase the TOPFET’s junction temperature too far.
The following section shows how to estimate Eclamp. Limiting
values for the energies EDSM for non-repetitive clamping and
EDRM for repetitive clamping are stated in the data sheet.
EDSM relates to a peak junction temperature of 225˚C
reached during clamping which is acceptable if it occurs
only a few times in the lifetime of a device. Thus EDSM should
only be used when deciding on the necessity for external
protection against overvoltage transients that occur
extremely rarely.
A TOPFET with its overvoltage clamping feature can save
these extra elements, provided that its limiting values are
not exceeded during the turn-off procedure. This section
shows a simplified method of estimating the dissipated
energy and the junction temperature rise in a TOPFET at
inductive turn-off. The equations given here are first order
approximations. They act as an aid in determining the need
for an external freewheel or suppressor element.
However, when switching inductive loads, absorbing Eclamp
is a normal condition. So to achieve the best longterm
reliability, the peak junction temperature should not exceed
150˚C. A method for estimating the peak junction
temperature is given later.
In this type of repetitive clamping application, the EDRM rating
in the data sheet can be compared with Eclamp to give an
initial indication of need for external voltage limiting. This
initial assessment should be followed by a temperature
calculation to find the maximum allowable mounting base
temperature and thus the heatsink requirements.
V bat
TOPFET Drain-Source Voltage
Equivalent
to TOPFET
at turn-off
V
(CL)DSS
Drain
Estimation of clamping energy
0
I D0
Input
Source
TOPFET Drain Current
The energy stored in the coil of a solenoid valve or a relay
with the inductance L at a current I is:
0
Figure 1a
EL
Figure 1b
Fig. 1 TOPFET. Switching an inductive load
=
1 2
LI
2
(1)
The clamping energy Eclamp in the TOPFET during an
inductive turn-off follows from equation (1) and the fact that,
during clamping, the battery also delivers energy to the
TOPFET:
TOPFET behaviour
Eclamp
Figure 1 shows an equivalent circuit diagram and the
shapes of drain current ID and drain-source voltage VDS
versus time for a TOPFET switching an inductor. The
overvoltage clamp feature of TOPFET is represented by a
zener diode that drives the output power MOSFET into
conduction if VDS rises too high. In this state the TOPFET
acts as an active clamp element, limiting its own VDS to
typically 60V.
=
1 2 V(CL)DSS
LI
2 D0 V(CL)DSS -Vbat
(2)
In (2) ID0 is the drain current at start of turn-off, V(CL)DSS the
TOPFET’s typical drain-source clamping voltage, Vbat the
battery voltage and L the load inductance. Equation (2)
assumes an inductor with no resistance. In practice, there
will be some resistance, which will dissipate a fraction of
Eclamp. Therefore, (2) represents a worst case situation.
469
Automotive
Power Semiconductor Applications
Philips Semiconductors
If these calculations indicate that the peak junction
temperature is less than Tj max, then external voltage limiting
is not needed.
Estimation of junction temperature
The peak junction temperature during clamping can be
estimated by adding the maximum temperature rise ∆Tj to
the average junction temperature, Tj0.
= Tj0 + ∆Tj
Tj, pk
Calculation examples
(3)
Both examples are carried out for Vbat=13 V and a
BUK101-50GS with a clamping voltage of 60 V. For
calculation of on-state losses, the maximum RDS(ON) at
Tj=150˚C of 87.5 mΩ is taken.
Measurements have shown that ∆Tj can be approximated
by
∆Tj
=
5
V
⋅I ⋅Z
6 (CL)DSS D0 th
(4)
Example 1:
An inductor with L=10 mH is switched off
non-repetitively at a dc current ID0=7 A.
Where Zth is the transient thermal impedance for a pulse
1
width of 3 of the time in clamping, which, for a coil resistance
(5) gives tclamp = 1.5 ms. The BUK101 data curve indicates
of zero Ohms, is:
L ⋅ ID0
=
V(CL)DSS − Vbat
tclamp
a Zth of about 0.28 K/W at tclamp/3 = 500 µs. (4) then gives
a ∆Tj of about 100 K. It is a non repetitive application so
(5)
use (7) to find Tj0, which indicates that Tj is about 7˚C above
Tmb due to on-state losses. From the ∆Tj and Tj0 figures it
Average dissipation will make Tj0 higher than the mounting
base temperature Tmb, which can be assumed as constant,
if the TOPFET is mounted on a heatsink. In repetitive
switching applications, both on-state losses and turn-off
losses contribute to the average dissipation. So Tj0 will be:
can be inferred:
Tj,pk < 150˚C for Tmb < (150-100-7)˚C = 43˚C.
Example 2:
(6)
An inductor with L=3 mH is switched at
ID0=4 A and a frequency of 100Hz and a duty cycle of 0.5.
In (6) IRMS is the root mean square value of the load current
and RDS(ON) is the on-state resistance of the TOPFET. In
non repetitive applications, the average dissipation is the
on state dissipation so Tj0 is:
(2) yields a clamp energy of 31 mJ, which is less than the
EDRM rating of the BUK101 of 40 mJ so repetitive clamping
is allowed. (6) yields that Tj0 will be about 8 K above Tmb.
From (4) and (5), ∆Tj can be estimated to be < 30 K. These
Tj0
=
Tj0
Tmb +  Eclamp
⋅ f +I
2
RMS
⋅ RDS(ON)
⋅ Rth, j − mb
2
= Tmb + ID0
⋅ RDS(ON) ⋅ Rth, j − mb
figures imply that this load can be safely driven if the Tmb of
the BUK101-50GS is < 112˚C.
(7)
470
Automotive
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Philips Semiconductors
5.3.14 Driving DC Motors with TOPFET
the nominal current. This current will cause overheating in
the motor which may damage the winding insulation or
demagnetize the stator magnets.
Examples for motor drive circuits using low side TOPFET
have already been given in section 5.3.7: "Linear Control
with TOPFET", and section 5.3.8: "PWM Control with
TOPFET". This section discusses the characteristics of DC
motors that have to be considered when designing a drive
circuit with low side TOPFET and gives examples of some
basic drive circuits.
The current would also cause extra dissipation in the driver
but a TOPFET, with its over temperature protection, would
survive a permanent stall condition. In addition, with careful
thermal design, the TOPFET can also be used to prevent
damage to the motor.
Important motor characteristics
The permanent magnet motor is the most common type of
motor for driving a wide range of applications including
small industrial drives, cooling fans and model cars.
Therefore, the following discussions are based on this type.
The equivalent circuit of these motors is shown in Fig. 1,
where RA and LA represent the resistance and inductance
of the armature.
L A
Inductive kick back at turn-off
The energy stored in the armature inductance, LA, has to
be removed when the motor is turning off. As in the case
of inductive loads such as solenoid valves and relay coils,
this is usually done by a freewheel diode. Provided that the
energy is within its EDRM rating, a TOPFET’s overvoltage
protection feature can be used instead of a freewheel diode.
Section 5.3.13: "Switching Inductive Loads with TOPFET",
covers this topic in more detail and gives a simple
calculation method to assess the need for a freewheel
diode. If overtemperature shutdown due to a stalled motor
can occur, a freewheel diode is generally recommended.
Without freewheel diode the TOPFET would have to absorb
a very high energy at a junction temperature of at least
150 ˚C.
RA
IA
EA
VM
In the case of pulsed operation of the motor (e.g. pulse
width modulation for speed control), the use of a freewheel
diode is advisable. Without it, motor current ripple would be
higher and the loss in the switching device could be as high
as it would be in a linear control circuit.
Fig. 1 Equivalent circuit for PM DC motor
Inrush current
Special effects of back EMF
Correct operation of some mechanical loads creates a
special starting torque requirement for the motor. Since
motor torque is proportional to motor current, high starting
torque can only be achieved if the inrush current is allowed
to be high. The TOPFETs BUK100...BUK106 do not use
current limiting techniques to provide overload protection,
so the inrush current they can deliver to a motor is limited
only by the forward transconductance gfs. To meet extreme
starting torque requirements, an ’S’type with 10 V control
is to be preferred over an ’L’ type with 5 V control because
’S’ types can deliver approximately twice the current of ’L’
types. Typical currents can be judged from the data sheet
ID(SC) in the section TRANSFER CHARACTERISTICS.
Effects at running out
The back EMF, EA, of a motor is proportional to the rotational
speed. When the TOPFET is turned off, the motor acts as
a generator and EA can serve as the feedback signal in a
PWM control system.
Although the back EMF voltage of many motors is, during
normal running, below its terminal voltage, in some
situations and with some motors the peak back EMF can
exceed the terminal voltage. Shortly after turn-off these
EMF peaks may even exceed the battery voltage plus one
diode drop. In this case the EMF can supply current into
the battery circuit by forward biasing the TOPFET’s
Source-Drain diode (see Fig. 2a). As a result of the internal
structure of a low side TOPFET, the Source-Drain diode
current will create a conduction path from the Input to the
Drain. The current through this path can be limited to a safe
Stall current
The stall current of a dc motor is limited by the armature
resistance, RA in Fig. 1, and can reach values of 5-8 times
471
Automotive
Power Semiconductor Applications
Philips Semiconductors
value by including a series resistance Ri as shown in
Fig. 2a. Recommended values for Ri are 100Ω for 5V
drivers and 220Ω for drivers above 6V.
circuit opens before the motor has stopped. This behaviour
will not damage the TOPFET. However, Figs 2b and 2c
show ways of avoiding it if it is not acceptable.
For 5 pin TOPFETs a path is also created from the
Protection Supply and Flag pins. In this case, sufficient
current limiting is often provided by the resistors that are
fitted to connect the Flag and Protection Supply pins to Vcc
(see Fig. 2c). The actual resistor values must be
determined from consideration of the TOPFET and control
circuit data sheets.
The first method (Fig. 2b) is to avoid forward biasing of
TOPFET’s Source-Drain diode by means of a series diode
D1. An alternative path for the generator current is provided
by zener diode D2. (It is worth noting that interruption of the
current path with D1 will be required in applications where
reverse battery must not activate the motor.) If the inclusion
of a power diode into the motor circuit is not acceptable the
alternative shown in Fig. 2c can be used. In this approach
the flag signal sets an external latch when the TOPFET is
tripped by the short circuit. In this way the TOPFET status
is stored even when its Source-Drain diode is forward
biased. If the TOPFET is being driven from a
microcontroller, the ’latch’ function could be implemented
in software.
Effects of intermittent short circuit
When a TOPFET’s short circuit protection has tripped due
to a short circuited motor, the motor will continue to turn. In
this situation the motor acts as a generator and its current
is reversed. The motor will lose rotational energy and, if the
short circuit remains long enough, will stop. In practice
however, contact sparking can cause intermittent short
circuits. In this case the short circuit may be interrupted
before the motor has stopped. After the interruption the
generator current will continue to flow, forced by the
armature inductance LA. A path for this negative current into
the battery is provided via TOPFET’s Source-Drain diode.
As described in the above section, currents into Input,
Protection Supply and Flag terminals should then be limited
by means of series resistances.
Vbat
Vbat
Vbat
D1
D2
Vcc
M
D
I
Ctrl
P
S
a) Simplest
circuit
D
I
Ctrl
Ri
Besides this, TOPFET’s internal circuits are non-active
while its Source-Drain diode is forward biased and a
previous overload shutdown will not stay latched. As a
consequence, a TOPFET that has turned off due to a short
circuit across its motor load may turn on again if the short
M
M
Ctrl
P
Ri
b) Reverse
blocking
D
Latch
P
F
I
P
S
S
c) External
latching
Fig. 2 Basic motor drive circuits with TOPFET
472
Automotive
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Philips Semiconductors
5.3.15 An Introduction to the High Side TOPFET
The introduction of high side TOPFETs enhances the range
of protected power MOSFETs available from Philips. These
devices combine the real power handling ability of low
RDS(ON) MOSFETs with protection circuits and the interfacing
to allow ground referenced logic signals to control a high
side switch.
developed across the TOPFET will cause the short circuit
detector to react and latch the TOPFET off until it is reset
by toggling the input. Both modes of overload turn-off are
reported by pulling the status pin low.
Type range
If the battery to ground voltage is too low for its circuits to
work correctly a high side TOPFET will turn off.
Supply undervoltage lockout -
Table 1 shows the range of high side TOPFETs. Included
in the range are devices with on-state resistance in the
range 38 to 220 mΩ. For each of the types an ’X’ or ’Y’
variant can be supplied (’Y’ types have an additional internal
resistor in the ground line). All the devices are 50 V types
designed for use in 12 V automotive systems.
Type
Open load detection TOPFET monitors its own on-state voltage drop. If the drop
is too low, indicating that the current is very small probably
because the load is open circuit, TOPFET will report this
by pulling the status pin low.
RDS(ON)
( mΩ)
BUK200-50X / BUK200-50Y
100
BUK201-50X / BUK201-50Y
60
BUK202-50X / BUK202-50Y
38
BUK203-50X / BUK203-50Y
220
Quiescent current One factor of great importance, particularly as the number
of devices in a car increases, is quiescent current. In
TOPFETs, the supply which feeds the circuits is turned off
when the input is low. This reduces off state current
consumption from typically 25 µA to less than 1 µA.
Ground resistor -
Table 1. High side TOPFET type range
For the fullest protection against the harsh automotive
electrical environment, it is often necessary to fit a resistor
between the ground pin of a high side device and module
ground. To help with this the Y types of the TOPFET range
have this resistor integrated on the chip. Apart from the
obvious saving in component count, this approach has the
advantage that the resistor is now in a package where its
dissipation can be easily handled. (This feature is
particularly useful when long duration reverse battery
situations are being considered).
Features
Particular care has been taken during the development of
the high side TOPFET to make a device which closely
matches the requirements of the automotive designer.
Overload Protection High side TOPFETs are protected from the full range of
overload conditions. Low level overloads which result in
higher than expected dissipation can cause the TOPFET
to overheat. In this case the overtemperature sensor will
trip and the TOPFET will turn itself off until the chip
temperature falls below the reset point. In the event of a
medium level overload, which could allow a high current to
flow, TOPFET will limit the current, and hence dissipation,
to a level which allows the overtemperature sensor time to
react and turn the TOPFET off until it cools sufficiently. In
high overload situations, like hard short circuits, the voltage
Inductive load turn-off clamping TOPFETs have a network between the MOSFET gate and
the ground pin. This network sets the maximum negative
potential between the load and ground pins. If the potential
tries to exceed this figure, for example during inductive load
turn-off, TOPFET will partially turn on, clamping the voltage
at the load pin.
473
Automotive
Power Semiconductor Applications
Philips Semiconductors
improvement can be obtained by simply looking at the
current in the ground pin with an AC coupled current probe.
Waveforms for the ground pin current of a Philips TOPFET
and another manufacturer’s high side switch are shown in
Fig. 1.
EMC
Electromagnetic compatibility is an increasingly important
factor in all electronic designs. EMC covers the immunity
and the emissions, both conducted and radiated, of
electronic units and systems. The directives and tests are
rarely applicable to individual electronic components
although the behaviour of devices can have a significant
influence on EMC performance. In recognition of this,
TOPFET has been designed to create as few EMC
problems as possible.
Pulse 1a
Test Voltage
Pulse width
-100 V
0.05 ms
Pulse 1b
Pulse 2a
a) Philips TOPFET
Average ground pin current (0.5 mA/div)
2 ms
+100 V
0.05 ms
Pulse 2b
0.5 ms
Pulse 3a
-200 V
0.1 µs
Pulse 3b
+200 V
0.1 µs
Pulse 5
+46.5 V
400 ms
time (1 us/div)
b) Manufacturer ’B’
Table 2. TOPFET transient tests
Average ground pin current (0.5 mA/div)
Conducted immunity One area where TOPFET helps with EMC is with its inherent
immunity to conducted transients. The voltage supply of a
vehicle is notorious for its transients and circuits and
systems have to be designed to handle them. On the
TOPFET chip are separate circuits which allow the output
MOSFET and the control circuits to withstand transients
between the battery and both the load and ground pins. The
range of transients which high side TOPFETs can survive
is shown in Table 2.
time (1 us/div)
Low emission -
Fig. 11. High side switch Ground pin current
High side switch devices generate their gate drive voltage
with oscillators and charge pumps running at high
frequency - often in excess of 1 MHz. Unless care is taken
in the basic design of the device, emissions at the oscillator
frequency or its harmonics can appear at the ground and
load pins.
Conclusions
High side TOPFETs are real power devices designed for
controlling a wide range of automotive loads. The care
taken during their design means that TOPFETs are
compatible with circuit designers’ protection and EMC
requirements.
The TOPFET designers have taken the necessary care.
The appropriate choice of oscillator and charge pump
circuits and the inclusion of on-chip filtering have reduced
emissions considerably. Some indication of the
474
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.3.16 High Side Linear Drive with TOPFET
increases the gate-source voltage turning the TOPFET on
harder. The voltage drop will now reduce, returning the
source - hence motor voltage - to its original value but at a
higher current level. All of this means that even without an
external feedback network, motor speed is inherently
stable, although not absolutely constant, under the full
range of motor loads.
This section describes a complete high side linear drive
circuit using a TOPFET. A low side linear TOPFET drive
circuit is described in section 5.3.7 and the principal pros
and cons of linear versus PWM drivers are discussed there.
The most important differences between high and low side
linear drives are:
- The high side drive needs a charge pump circuit to provide
an input voltage higher than the battery voltage.
Transistor T2 works as a current generator and supplies
the protection circuit of the TOPFET. T2 is switchable via
transistor T1 and Schottky diode D3. If the potentiometer
is in position A, transistors T1 and T2 are switched off
allowing R11 to pull the protection supply voltage to 0 V.
This feature means the TOPFET, if it has tripped due to
over temperature or overload, can be reset by turning the
potentiometer to position A.
- In the high side drive the load provides negative feedback
for the output transistor. Therefore, the control loop circuit
needed to maintain stability in a low side drive can be
saved.
The circuit described in this paper was designed for and
tested with a 200W fan motor for cars.
Circuit description
Position A is also the standby mode. With both transistors
switched off, the drive circuit has a very small current
consumption. This means that in standby the current
consumption of the whole circuit (drive and charge pump)
is about 0.3mA.
The complete high side linear drive circuit can be split up
into two blocks:
- The drive circuit
- The charge pump
Drive circuit
R10
240
Vcp
Figure 1 shows the drive circuit. Motor speed is controlled
by changing the TOPFET’s input voltage and therefore its
voltage drop. A 5-pin TOPFET is used because this type
allows the protection circuit to be supplied independently
of the input. This is necessary because in this application
the input-source voltage may become too low to supply the
protection circuit of a 3-pin TOPFET.
R8
7.5k
Vbat
T2
BC558
D6
D4
R7
15k
B
The TOPFET’s input voltage and therefore the speed of the
fan motor is determined by potentiometer R5. The TOPFET
is operating as a source follower. The inherent negative
feedback of this configuration will automatically ensure that
the source potential will equal the input potential (minus the
gate-source voltage) no matter what current is flowing in
the motor.
BAW62
R12
100k
Z2
BZX79/
C10
D
P
BUK10650L
P
F
R5
1M
A
D5
I
2x
BAW62
R13
100k
R6
R9
R11
1M
110k
100k
T1
BC548
An increase in motor load will tend to slow the motor
reducing its back EMF and creating a demand for extra
current. The extra current would increase the voltage drop
across the TOPFET, lowering the source potential. Since
the input potential has been set, the lower source potential
D7
D3
BAS83
0V
Fig. 1 Drive circuit
475
S
Fan
Motor
Automotive
Power Semiconductor Applications
Philips Semiconductors
For correct operation of TOPFET’s active protection
circuits, sufficient voltage has to be applied to its protection
pin. The minimum protection supply voltage for the
BUK106-50L is 4V for input voltages Vis up to 6.5V (see
data sheet Fig. 17). For the circuit presented and the
component values given, this requirement is met with a
battery voltage as low as 8 V. If operation at a lower battery
voltage is needed then a voltage tripler charge pump could
be used in place of the voltage doubler proposed in this
paper.
TOPFET interface
Negative potentials are not permitted between a TOPFET’s
protection supply (P), input (I) or flag (F) and its source.
This must be considered, especially when designing high
side drivers, where the source potential is determined by
the load voltage.
If an overvoltage pulse occurs at the supply terminal while
the TOPFET is off, the source potential will rise with the
overvoltage as soon as the TOPFET’s clamp voltage is
exceeded. At this time the P,F and I pins should not be
clamped with reference to ground, but should be allowed
to rise with the source potential. In this circuit this is
achieved by diodes D5 and D6 in the feeds to the I,P and
F pins.
Vbat
Zener diode Z2 limits the maximum protection supply and
flag voltages to about 10 V and, via D4, the input-source
voltage to about 10.6 V. Resistor R7 has a value high
enough to allow the TOPFET’s internal protection circuits
to turn off the device in the event of an over temperature or
short circuit load.
D2
BAW62
Vcp
D1
+
+
C4
10u/
22V
C5
10u/
22V
BAW62
R1
200
Charge pump
C3
22n
R2
330k
Figure 2 shows the charge pump circuit. IC1 works as an
astable pulse generator at a frequency of 20 kHz which,
together with D1, D2, C4, C5 produces a voltage doubler.
The ICM7555 is a type with low current consumption. This
is an important feature because the circuit consumes
current, even when the driver circuit is in standby mode.
R4
4
7
Z1
BZD23/
C15
R3
160k
IC1
6
ICM
7555
3
C1
10u/
22V
In its normal operating mode, the drive circuit has a typical
current consumption of 1.5mA which determines the values
of C4 and C5. R4 is included to limit the output current of
IC1. The charge pump generates an output voltage of about
22V at a battery voltage of 12.6V. Z1, R1 and C1 will smooth
and limit the supply to the circuit and provide protection
from voltage spikes.
160
8
2
5
1
C2
100p
0V
Fig. 2 Charge pump circuit
476
Automotive
Power Semiconductor Applications
Philips Semiconductors
Automotive Ignition
477
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.4.1 An Introduction to Electronic Automotive Ignition
The function of an automotive ignition circuit is to provide
a spark of sufficient energy to ignite the compressed air-fuel
mixture at the appropriate time. Increasingly, electronics is
being used to optimise the ignition event. This is now
necessary to ensure conformance with emission
regulations and to achieve maximum engine performance,
fuel economy and engine efficiency. This section will look
at some important aspects of the power stage of an
electronic ignition system. Other sections in this chapter will
look more closely at the power devices for this application.
-
+
Battery
Ignition
Switch
Coil
Dynamic
Clamp
Ignition
Electronic
Switch
Controller
Electronic ignition circuit
Fig. 1 Typical automotive ignition circuit
There are several different configurations for electronic
ignition. Some are still being studied and there are several
already in use. But by far the most common configuration
for the power stage is that shown in Fig. 1. With this
arrangement there is no distributor. The circuit shown is for
a four cylinder engine and has two separate power circuits
each feeding two cylinders. Extra power stages can be
added for 6 and 8 cylinder engines. When one power stage
fires, both plugs will spark but, by choosing pairs of cylinders
which are 360˚ out of phase in the 4-stroke cycle, only one
will have a mixture that can be ignited - the other will be
approaching tdc at the end of the exhaust stroke.
Spark energy
Under ideal conditions the mixture can be ignited with a
spark energy of 0.3 mJ but, for reliable ignition under all
possible engine conditions, spark energies in the range
60mJ to 150mJ are needed. The energy comes from the
coil and is the energy stored as flux generated by the current
that was allowed to build up in the primary. The energy
stored in the magnetic field of the coil is:
1
Eprim = Lprimi 2
2
(2)
Operation
Timing
During normal operation the transistor will be turned on
some time before the spark is needed (t1 in Fig. 2). Current
will now rise at a rate given by the equation
rateofrise =
di V
=
dt L
The timing of the spark is one of the most critical factors in
achieving optimum engine performance. The controller
uses information about engine speed, temperature, fuel etc.
to decide how far before tdc the spark is needed. It then
uses data from crankshaft position sensors to decide when
to signal for a spark.
(1)
where V is the voltage across the primary of the coil. When
the spark is needed (t3), the transistor is turned off. The
current in the inductance will try to stop flowing but it can
only change at the rate given by (1). This means that voltage
on the primary is forced to become large and negative.
Transformer action increases the secondary voltage until
it reaches the voltage needed to create a spark at the plugs
- minimum 5 kV but may be 10 to 30 kV. Current now flows
through the spark and the secondary winding, the voltage
now falls back to that necessary to maintain the current in
the spark, t5. When all the coil energy has been delivered,
t6, the voltage at the collector falls to the battery voltage.
One factor which the controller cannot control is the delay
between it issuing the command to spark and the spark
being generated. Part of this delay is the time it takes the
transistor to start turning off together with the rate at which
the transistor voltage rises. The controller can make
allowance for this delay but in many systems this is no more
than a fixed offset. In practice the delay will vary with
variations in the drive circuit, temperature and between
devices - with some transistor types being more susceptible
to variation than others.
479
Automotive
Power Semiconductor Applications
Philips Semiconductors
low particularly during engine cranking. Ensuring that the
circuit operates reasonably well at these low voltages
means keeping the transistor voltage drop as low as
possible.
I(A)
6.0
4.0
t3
2.0
Fault conditions
t2
t4
Automotive systems must be reliable. Achieving high
reliability means designing systems that can survive all the
operating environments that the automobile can produce.
Some of the harshest conditions are the fault conditions.
0
2.0
4.0
6.0
8.0
t(ms)
a) Coil current
Vce(V)
300
Open circuit secondary
t3 t4
Disconnection of a spark plug lead means that the stored
coil energy cannot be dissipated in the spark. Unless steps
are taken to prevent it, the voltage will be forced higher until
it reaches the breakdown voltage of the transistor. The
combination of high current and voltage would probably
destroy the device. The solution to this problem is to operate
the transistor in dynamic clamping. This can be achieved
either by connecting a network between collector and the
gate/base or by using a device with the network already
integrated into it. With this arrangement the voltage rises
to the clamping voltage, the transistor then turns on
partially, with enough drive to allow the coil current to flow
at a collector emitter voltage equal to the clamping voltage.
The clamping voltage is set higher than the voltage normally
needed to generate the spark.
200
100
t5
t1
t6
0
2.0
4.0
6.0
8.0
t(ms)
b) Transistor collector voltage
Fig. 2 Ignition circuit waveforms
Dwell
As mentioned earlier, proper ignition means there must be
enough energy stored in the coil when the spark is needed,
so the transistor must be turned on soon enough to allow
time for the current to reach the required level. However,
turning on too soon will mean that the current is higher than
it needs to be. Although proper spark timing and energy is
more important, optimum coil current is also significant.
Higher currents create higher loss which reduces efficiency
and increases the problems of thermal management. They
can also reduce the life and reliability of the coil and create
major difficulties when designing for survival under fault
conditions like open circuit secondary.
Reverse Battery
Another condition which must be survived is when the
battery connections are reversed. Ideally no current should
flow and this can be achieved with some transistors which
have a reverse blocking voltage greater than the battery
voltage. With many transistors, however, reverse blocking
is not guaranteed and to block the current means adding a
diode in series. This is rarely acceptable because the diode
forward voltage drop adds too much to the effective voltage
drop. The alternatives are to allow the current to flow either
by using a transistor which is rated to operate with reverse
current or by fitting a diode in anti-parallel with the transistor.
The time to turn on the transistor is governed by (1). Coil
inductance is an attribute of the coil but the primary voltage
depends on battery voltage and the voltage drop across the
transistor. Battery voltage can vary widely and can be very
480
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.4.2 IGBTs for Automotive Ignition
This publication describes a range of power transistors for
automotive
ignition
applications
from
Philips
Semiconductors. This range of IGBTs has been specifically
optimised for the demanding conditions of ignition circuits.
The IGBT is a voltage controlled, low loss, high power
transistor which gives the ease of drive and low conduction
losses that are required in automotive ignition circuits. The
Philips range of ignition IGBTs includes conventional IGBT
devices with standard gate drive input. It also includes a
range of standard and logic level input protected IGBTs with
integral gate drain and gate source clamping diodes.
Input Voltage
IGBTs, like MOSFETs, can have standard or logic level
gate sensitivity. A standard device has a threshold voltage
of typically 3.5 V - the threshold voltage is the gate voltage
needed to allow the IGBT to conduct 1 mA, i.e just started
to turn on. To be fully on, with an acceptable low VCE, the
gate voltage needs to be 8.5 V. In some situations, such
as engine cranking, the battery voltage falls to less than 6 V
and achieving adequate drive may be a problem.
An alternative would be to use a logic level IGBT which has
a threshold of typically 1.5 V and is fully on with 5 V.
Introduction to the IGBT
The structure of an IGBT is similar to that of a Power
MOSFET, both being created by the parallel connection of
many thousands of identical cells. Figure 1 shows the cross
section of one IGBT cell. The only difference between this
drawing and one for a MOSFET would be the polarity of
the substrate - the MOSFET would be n+ rather than the
p+ of the IGBT. Since the gate structures are identical the
IGBT like the MOSFET is a voltage driven device with an
extremely high input impedance.
Another factor in the choice between standard or logic level,
is that of noise immunity. In this application it can be very
important that the IGBT is fully off, in a very low leakage
state, when the driver stage output is LOW. Unfortunately,
the LOW that a driver produces may not create a gate to
emitter voltage of 0 V.
The threshold voltage of an IGBT falls as temperature rises.
So the gate emitter voltage of a logic level IGBT (at
Tj = 120˚C) needs to be < 0.7V to ensure that it is off. A
standard level part, with its higher threshold, has more
immunity and it would still be off if the voltage was < 1.4 V.
Gate
Emitter
p
Emitter
n
n
p
Turn off control
n-
The time between the gate signal arriving at the IGBT and
the collector voltage rising is known as the delay time, td.
An ignition system produces a spark when the collector
voltage rises. Since the timing of the spark is critical, it is
advantageous to have good control of td. With the IGBT,
unlike some other ignition switches, td is dominated by gate
charge and so can be very low and is easily controlled by
the resistance of the driver circuit.
p+
Collector
Fig. 1 Cross section of IGBT cell
Bipolar operation
Where the IGBT differs is in the characteristics of the output
device. Conduction in a MOSFET is by majority carriers
only but the p substrate silicon used for IGBTs promotes
injection and gives bipolar conduction with both majority
and minority carriers. The effect of this is to make the on
state voltage drop of a high voltage IGBT much lower than
that of the same size and voltage MOSFET. This feature is
particularly useful in automotive ignition where high voltage
devices are needed which can operate from low voltage
supplies. In recognition of its combination of MOSFET input
and bipolar output, the terminals of an IGBT are called
Collector, Emitter and Gate.
Safe Operating Area
One of the worst situations for creating IGBT latch up is
inductive turn off. Such a turn off takes place in electronic
ignition. IGBTs, for ignition applications, are specified with
a safe operating area (SOA) and limiting value of collector
current that can be safely switched under clamped inductive
load conditions (ICLM). Providing that the device is operated
within its safe operating area (SOA) dynamic latch-up (or
SOA failure) cannot occur. Philips ignition IGBTs have a
large turn-off SOA and a large energy handling capability
making them easy to use in ignition circuits.
481
Automotive
Power Semiconductor Applications
Philips Semiconductors
Feature
processing stages which allows polysilicon diodes, of
known breakdown voltage, to be integrated with the IGBT
structure. A short chain of diodes is connected between the
gate and the emitter. This gives ESD protection by clamping
the voltage, which can be applied across the gate emitter
oxide, to a safe value.
Advantage
IGBTs
•Voltage driven
-Low gate drive power
-Simple gate circuit
•Logic level capability
-Low battery operation
•Bipolar operation
-Low conduction loss
-Small device size
•PowerMOS/bipolar
structure
-Negligible Storage time
-Reverse blocking
-Energy handling
•Large SOA
-No snubber required
-Design flexibility
A much longer chain, with a combined breakdown voltage
of several hundreds of volts, is connected between the
collector and the gate. This chain makes the IGBT into a
dynamic clamp - possibly the best way of ensuring survival
during ignition faults like open circuit secondary. The
position of the diode chains is shown in the circuit symbol,
see Fig. 2.
collector
Clamped IGBTs
•Integral clamp diodes
-Design simplicity
-Overvoltage protection
Clamp voltage control
-Improved reliability
-ESD protection
gate
collector
gate
emitter
emitter
Fig. 2 IGBT circuit symbols
Table 1. Advantages of IGBTs
Reverse Battery
Conclusions
The n- p+ junction, see Fig. 1, which is inherent in the
structure of the IGBT, creates a reverse blocking junction.
This junction, although unable to support very high reverse
voltages, is able to block voltages in excess of a battery
voltage. This gives the IGBT a reverse battery blocking
capability which ensures that reverse battery fault
conditions will not give rise to high currents which could
damage the IGBT or any other components in the ignition
circuit.
The Philips Semiconductors BUK854-500IS ignition IGBTs
and clamped IGBTs BUK856-400IZ and BUK856-450IX
are specifically designed to give a low loss, easy to drive
and rugged solution to the demanding applications of
automotive ignition circuits. IGBTs require the minimum of
external components in the gate drive circuit and give
negligible drive losses. The energy handling and reverse
blocking capabilities of the device make it suitable for use
in automotive environments - even under fault conditions.
Voltage clamping and ESD protection give ease of design
and use, improved reliability and performance in the ignition
controller circuit.
Clamped IGBT
A refinement of the conventional IGBT is the clamped or
protected IGBT. This is produced by adding extra
482
Automotive
Power Semiconductor Applications
Philips Semiconductors
5.4.3 Electronic Switches for Automotive Ignition
Earlier sections in this chapter have discussed the nature
of automotive electronic ignition and looked at a range of
IGBTs which have been optimised for use in this type of
application. This section will compare ignition IGBTs with
ignition darlington transistors and come to the conclusions
that IGBTs have several advantages which would be useful
to the automotive designer.
The low on-state voltage drop of a bipolar device is the
result of minority carrier injection. However, the minority
carrier injection also introduces ’stored charge’ into the
device which must be removed at turn-off. The charge is
extracted, at least partially, as negative base current during
the period known as the storage time, ts. How long this
takes, depends on the amount of stored charge and the
rate it is extracted. The amount of charge varies from device
to device, with the level of the current and with temperature.
The rate of extraction depends on the drive circuit and
whether a ’simple’ circuit like that of Fig. 1 is used or one
which uses negative drive to remove the charge more
quickly.
Darlington transistors
In the past, the darlington transistor has been the favoured
power transistor for ignition applications. The darlington
connection is, in fact, a cascade of two separate bipolar
transistors. The combination increases the gain allowing
the high voltage device to be controlled by a relatively low
power driver stage.
Vbat
Coil
Primary
Vbat
Coil
Secondary
Coil
i
V
in
Vin
RG
C
v
CE
Spark
gap
BUK854-500IS
IGBT
BUV90
Fig. 2 Ignition IGBT circuit
Storage time adds to the delay between the input changing
state and the spark being produced and the uncertainty in
storage time, which results from the large number of
variables, adds to the inaccuracy of the ignition timing.
Fig. 1 Typical ignition circuit with darlington switch
As the darlington is a bipolar device it has a relatively low
on-state voltage drop even though it can block a high
voltage. The low voltage drop keeps conduction losses low
and allows the ignition circuit to function at low battery
voltages.
Typical ignition darlingtons often include an internal
antiparallel diode connected across the main
emitter-collector terminals as shown in Fig. 1. This diode is
not necessary for the normal operation of the ignition circuit
and its function is simply to protect the darlington from
reverse battery faults. However, during this condition, the
diode does allow large reverse currents to flow through the
ignition circuit.
The disadvantage of a darlington is the complexity and cost
of the base drive. Even though the gain is improved, by the
darlington connection, a large gate current is still needed
(approx. 100 mA) a circuit similar to that shown in Fig. 1 will
be needed. It would be inefficient and costly to supply a
current this large from the stabilised 5 V rail, so the supply
could be the battery. This means that the drive dissipation
when the transistor is on, is about 1.2 W and the average
dissipation about 0.5 W. This level of dissipation requires
a special driver IC or a circuit using discretes. All of this
adds to the cost complexity and thermal problems of the
ignition system.
IGBTs
The IGBT is a combination of bipolar transistor and Power
MOSFET technologies. It has the advantage of the low
on-state voltage drop of a bipolar darlington and can also
be voltage driven in the same way as a Power MOSFET.
This gives a highly efficient, easy to drive, minimum loss
solution for the switching transistor in an ignition circuit.
483
Automotive
Power Semiconductor Applications
Philips Semiconductors
A typical ignition circuit using the Philips BUK854-500IS
IGBT is shown in Fig. 2; the saving in gate drive
components is self evident. The need for a special driver
stage is eliminated because drive dissipation for an IGBT
will be approximately 10 µW which can be easily supplied
by standard ICs. The BUK854-500IS has a voltage rating
of 500V and standard gate threshold voltage, and is
assembled in the TO220 package.
Table 1 shows a breakdown of the ignition system losses
and demonstrates that whilst the device losses are slightly
higher in the IGBT, the overall losses are higher in the
darlington circuit due to extra loss in the base drive.
Power loss (W)
Clamped IGBTs
One of the most exciting features of IGBT technology is the
ability to integrate protection functions into the IGBT to give
significant advantages to the designer of power circuits.
The BUK856-400IZ and BUK856-450IX are two such
devices which have been specifically designed for
automotive ignition circuits. The BUK856-400IZ is a logic
level device, the BUK856-450IX has a standard gate
threshold. The nominal clamp voltages are 400V and 450V
respectively.
Vclamp=400V, ICmax=6.0A
100Hz, (3000rpm)
IGBT
darlington
Conduction
1.37
1.16
Switching
0.71
0.79
Drive
0.00001
0.5
Total
2.08
2.45
Table 1. IGBT and darlington ignition circuit losses
Conclusion
In these devices the dynamic clamp network shown in Fig. 2
is fabricated directly onto the IGBT. This gives guaranteed
clamping of the IGBT at a fixed clamp voltage without the
need for an external circuit. The clamp voltage is held to
very tight tolerances over the full temperature range (-40˚C
to +150˚C) required in automotive applications.
Table 2 summarises the comparison between the IGBT and
the darlington as the power switch in automotive ignition.
The comparison shows that the darlington is good in the
application but that the IGBT has some clear advantages
making it significantly better.
In both these devices gate-source protection diodes have
also been incorporated into the structure of the devices to
give full protection against ESD damage during handling
and assembly of the device into engine management units.
Driver component count
Speed of response, ’time to
spark’
Total loss
Drive power
Logic level operation
Open circuit load
Reverse blocking
Package size
Inbuilt voltage clamp
Inbuilt protection
IGBTs and darlingtons - A performance
comparison
The performance of the BUK856-400IZ ignition IGBT has
been compared with that of a typical ignition darlington in
the ignition circuit of Fig. 1.
At turn-off the darlington switched considerably slower than
the IGBT. The time between the input going low and the
spark was 32 µs for the darlington and only 19 µs for the
IGBT.
IGBT
darlington
Low
Fast
High
Slow
Better
Low
Yes
Yes
Yes
Small
Possible
Yes
Good
High
Yes
Yes
No
Large
Possible
No
Table 2. Performance comparison
484
Preface
Power Semiconductor Applications
Philips Semiconductors
Acknowledgments
We are grateful for all the contributions from our colleagues within Philips and to the Application Laboratories in Eindhoven
and Hamburg.
We would also like to thank Dr.P.H.Mellor of the University of Sheffield for contributing the application note of section 3.1.5.
The authors thank Mrs.R.Hayes for her considerable help in the preparation of this book.
The authors also thank Mr.D.F.Haslam for his assistance in the formatting and printing of the manuscripts.
Contributing Authors
N.Bennett
D.J.Harper
J.Oosterling
M.Bennion
W.Hettersheid
N.Pichowicz
D.Brown
J.v.d.Hooff
W.B.Rosink
C.Buethker
J.Houldsworth
D.C. de Ruiter
L.Burley
M.J.Humphreys
D.Sharples
G.M.Fry
P.H.Mellor
H.Simons
R.P.Gant
R.Miller
T.Stork
J.Gilliam
H.Misdom
D.Tebb
D.Grant
P.Moody
H.Verhees
N.J.Ham
S.A.Mulder
F.A.Woodworth
C.J.Hammerton
E.B.G. Nijhof
T.van de Wouw
This book was originally prepared by the Power Semiconductor Applications Laboratory, of the Philips Semiconductors
product division, Hazel Grove:
M.J.Humphreys
D.Brown
C.J.Hammerton
R.Miller
L.Burley
It was revised and updated, in 1994, by:
N.J.Ham
C.J.Hammerton
D.Sharples
Preface
Power Semiconductor Applications
Philips Semiconductors
Preface
This book was prepared by the Power Semiconductor Applications Laboratory of the Philips Semiconductors product
division, Hazel Grove. The book is intended as a guide to using power semiconductors both efficiently and reliably in power
conversion applications. It is made up of eight main chapters each of which contains a number of application notes aimed
at making it easier to select and use power semiconductors.
CHAPTER 1 forms an introduction to power semiconductors concentrating particularly on the two major power transistor
technologies, Power MOSFETs and High Voltage Bipolar Transistors.
CHAPTER 2 is devoted to Switched Mode Power Supplies. It begins with a basic description of the most commonly used
topologies and discusses the major issues surrounding the use of power semiconductors including rectifiers. Specific
design examples are given as well as a look at designing the magnetic components. The end of this chapter describes
resonant power supply technology.
CHAPTER 3 describes motion control in terms of ac, dc and stepper motor operation and control. This chapter looks only
at transistor controls, phase control using thyristors and triacs is discussed separately in chapter 6.
CHAPTER 4 looks at television and monitor applications. A description of the operation of horizontal deflection circuits is
given followed by transistor selection guides for both deflection and power supply applications. Deflection and power supply
circuit examples are also given based on circuits designed by the Product Concept and Application Laboratories (Eindhoven).
CHAPTER 5 concentrates on automotive electronics looking in detail at the requirements for the electronic switches taking
into consideration the harsh environment in which they must operate.
CHAPTER 6 reviews thyristor and triac applications from the basics of device technology and operation to the simple design
rules which should be followed to achieve maximum reliability. Specific examples are given in this chapter for a number
of the common applications.
CHAPTER 7 looks at the thermal considerations for power semiconductors in terms of power dissipation and junction
temperature limits. Part of this chapter is devoted to worked examples showing how junction temperatures can be calculated
to ensure the limits are not exceeded. Heatsink requirements and designs are also discussed in the second half of this
chapter.
CHAPTER 8 is an introduction to the use of high voltage bipolar transistors in electronic lighting ballasts. Many of the
possible topologies are described.
Contents
Power Semiconductor Applications
Philips Semiconductors
Table of Contents
CHAPTER 1 Introduction to Power Semiconductors
General
3
1.1.1 An Introduction To Power Devices ............................................................
Power MOSFET
1.2.1
1.2.2
1.2.3
1.2.4
1.2.5
1.2.6
1.2.7
1.2.8
1.2.9
PowerMOS Introduction .............................................................................
Understanding Power MOSFET Switching Behaviour ...............................
Power MOSFET Drive Circuits ..................................................................
Parallel Operation of Power MOSFETs .....................................................
Series Operation of Power MOSFETs .......................................................
Logic Level FETS ......................................................................................
Avalanche Ruggedness .............................................................................
Electrostatic Discharge (ESD) Considerations ..........................................
Understanding the Data Sheet: PowerMOS ..............................................
High Voltage Bipolar Transistor
1.3.1
1.3.2
1.3.3
1.3.4
1
Introduction To High Voltage Bipolar Transistors ......................................
Effects of Base Drive on Switching Times .................................................
Using High Voltage Bipolar Transistors .....................................................
Understanding The Data Sheet: High Voltage Transistors .......................
CHAPTER 2 Switched Mode Power Supplies
Using Power Semiconductors in Switched Mode Topologies
2.1.1 An Introduction to Switched Mode Power Supply Topologies ...................
2.1.2 The Power Supply Designer’s Guide to High Voltage Transistors ............
2.1.3 Base Circuit Design for High Voltage Bipolar Transistors in Power
Converters ...........................................................................................................
2.1.4 Isolated Power Semiconductors for High Frequency Power Supply
Applications .........................................................................................................
Output Rectification
2.2.1 Fast Recovery Epitaxial Diodes for use in High Frequency Rectification
2.2.2 Schottky Diodes from Philips Semiconductors ..........................................
2.2.3 An Introduction to Synchronous Rectifier Circuits using PowerMOS
Transistors ...........................................................................................................
i
5
17
19
29
39
49
53
57
61
67
69
77
79
83
91
97
103
105
107
129
141
153
159
161
173
179
Contents
Power Semiconductor Applications
Philips Semiconductors
Design Examples
2.3.1 Mains Input 100 W Forward Converter SMPS: MOSFET and Bipolar
Transistor Solutions featuring ETD Cores ...........................................................
2.3.2 Flexible, Low Cost, Self-Oscillating Power Supply using an ETD34
Two-Part Coil Former and 3C85 Ferrite ..............................................................
Magnetics Design
2.4.1 Improved Ferrite Materials and Core Outlines for High Frequency Power
Supplies ...............................................................................................................
Resonant Power Supplies
2.5.1. An Introduction To Resonant Power Supplies ..........................................
2.5.2. Resonant Power Supply Converters - The Solution For Mains Pollution
Problems ..............................................................................................................
CHAPTER 3 Motor Control
AC Motor Control
3.1.1 Noiseless A.C. Motor Control: Introduction to a 20 kHz System ...............
3.1.2 The Effect of a MOSFET’s Peak to Average Current Rating on Invertor
Efficiency .............................................................................................................
3.1.3 MOSFETs and FREDFETs for Motor Drive Equipment .............................
3.1.4 A Designers Guide to PowerMOS Devices for Motor Control ...................
3.1.5 A 300V, 40A High Frequency Inverter Pole Using Paralleled FREDFET
Modules ...............................................................................................................
DC Motor Control
3.2.1 Chopper circuits for DC motor control .......................................................
3.2.2 A switched-mode controller for DC motors ................................................
3.2.3 Brushless DC Motor Systems ....................................................................
Stepper Motor Control
3.3.1 Stepper Motor Control ...............................................................................
CHAPTER 4 Televisions and Monitors
Power Devices in TV & Monitor Applications (including selection
guides)
4.1.1 An Introduction to Horizontal Deflection ....................................................
4.1.2 The BU25XXA/D Range of Deflection Transistors ....................................
ii
185
187
199
205
207
217
219
225
241
243
245
251
253
259
273
283
285
293
301
307
309
317
319
321
331
Contents
Power Semiconductor Applications
Philips Semiconductors
4.1.3 Philips HVT’s for TV & Monitor Applications ..............................................
4.1.4 TV and Monitor Damper Diodes ................................................................
TV Deflection Circuit Examples
4.2.1 Application Information for the 16 kHz Black Line Picture Tubes ..............
4.2.2 32 kHz / 100 Hz Deflection Circuits for the 66FS Black Line Picture Tube
SMPS Circuit Examples
4.3.1 A 70W Full Performance TV SMPS Using The TDA8380 .........................
4.3.2 A Synchronous 200W SMPS for 16 and 32 kHz TV ..................................
Monitor Deflection Circuit Example
4.4.1 A Versatile 30 - 64 kHz Autosync Monitor .................................................
CHAPTER 5 Automotive Power Electronics
Automotive Motor Control (including selection guides)
5.1.1 Automotive Motor Control With Philips MOSFETS ....................................
Automotive Lamp Control (including selection guides)
5.2.1 Automotive Lamp Control With Philips MOSFETS ....................................
The TOPFET
5.3.1 An Introduction to the 3 pin TOPFET .........................................................
5.3.2 An Introduction to the 5 pin TOPFET .........................................................
5.3.3 BUK101-50DL - a Microcontroller compatible TOPFET ............................
5.3.4 Protection with 5 pin TOPFETs .................................................................
5.3.5 Driving TOPFETs .......................................................................................
5.3.6 High Side PWM Lamp Dimmer using TOPFET .........................................
5.3.7 Linear Control with TOPFET ......................................................................
5.3.8 PWM Control with TOPFET .......................................................................
5.3.9 Isolated Drive for TOPFET ........................................................................
5.3.10 3 pin and 5 pin TOPFET Leadforms ........................................................
5.3.11 TOPFET Input Voltage ............................................................................
5.3.12 Negative Input and TOPFET ...................................................................
5.3.13 Switching Inductive Loads with TOPFET .................................................
5.3.14 Driving DC Motors with TOPFET .............................................................
5.3.15 An Introduction to the High Side TOPFET ...............................................
5.3.16 High Side Linear Drive with TOPFET ......................................................
iii
339
345
349
351
361
377
379
389
397
399
421
423
425
433
435
443
445
447
449
451
453
455
457
459
461
463
465
467
469
471
473
475
Contents
Power Semiconductor Applications
Philips Semiconductors
Automotive Ignition
477
5.4.1 An Introduction to Electronic Automotive Ignition ......................................
5.4.2 IGBTs for Automotive Ignition ....................................................................
5.4.3 Electronic Switches for Automotive Ignition ...............................................
CHAPTER 6 Power Control with Thyristors and Triacs
Using Thyristors and Triacs
6.1.1
6.1.2
6.1.3
6.1.4
479
481
483
485
487
Introduction to Thyristors and Triacs .........................................................
Using Thyristors and Triacs .......................................................................
The Peak Current Handling Capability of Thyristors ..................................
Understanding Thyristor and Triac Data ....................................................
Thyristor and Triac Applications
489
497
505
509
521
6.2.1 Triac Control of DC Inductive Loads ..........................................................
6.2.2 Domestic Power Control with Triacs and Thyristors ..................................
6.2.3 Design of a Time Proportional Temperature Controller .............................
Hi-Com Triacs
523
527
537
547
6.3.1 Understanding Hi-Com Triacs ...................................................................
6.3.2 Using Hi-Com Triacs ..................................................................................
CHAPTER 7 Thermal Management
Thermal Considerations
549
551
553
555
7.1.1 Thermal Considerations for Power Semiconductors .................................
7.1.2 Heat Dissipation .........................................................................................
CHAPTER 8 Lighting
557
567
575
Fluorescent Lamp Control
577
8.1.1 Efficient Fluorescent Lighting using Electronic Ballasts .............................
8.1.2 Electronic Ballasts - Philips Transistor Selection Guide ............................
8.1.3 An Electronic Ballast - Base Drive Optimisation ........................................
iv
579
587
589
Index
Power Semiconductor Applications
Philips Semiconductors
Index
Airgap, transformer core, 111, 113
Anti saturation diode, 590
Asynchronous, 497
Automotive
fans
see motor control
IGBT, 481, 483
ignition, 479, 481, 483
lamps, 435, 455
motor control, 425, 457, 459, 471, 475
resistive loads, 442
reverse battery, 452, 473, 479
screen heater, 442
seat heater, 442
solenoids, 469
TOPFET, 473
Avalanche, 61
Avalanche breakdown
thyristor, 490
Avalanche multiplication, 134
Bridge circuits
see Motor Control - AC
Brushless motor, 301, 303
Buck-boost converter, 110
Buck converter, 108 - 109
Burst firing, 537
Burst pulses, 564
Capacitance
junction, 29
Capacitor
mains dropper, 544
CENELEC, 537
Charge carriers, 133
triac commutation, 549
Choke
fluorescent lamp, 580
Choppers, 285
Clamp diode, 117
Clamp winding, 113
Commutation
diode, 164
Hi-Com triac, 551
thyristor, 492
triac, 494, 523, 529
Compact fluorescent lamp, 585
Continuous mode
see Switched Mode Power Supplies
Continuous operation, 557
Converter (dc-dc)
switched mode power supply, 107
Cookers, 537
Cooling
forced, 572
natural, 570
Crest factor, 529
Critical electric field, 134
Cross regulation, 114, 117
Current fed resonant inverter, 589
Current Mode Control, 120
Current tail, 138, 143
Baker clamp, 138, 187, 190
Ballast
electronic, 580
fluorescent lamp, 579
switchstart, 579
Base drive, 136
base inductor, 147
base inductor, diode assisted, 148
base resistor, 146
drive transformer, 145
drive transformer leakage inductance, 149
electronic ballast, 589
forward converter, 187
power converters, 141
speed-up capacitor, 143
Base inductor, 144, 147
Base inductor, diode assisted, 148
Boost converter, 109
continuous mode, 109
discontinuous mode, 109
output ripple, 109
Bootstrap, 303
Breakback voltage
diac, 492
Breakdown voltage, 70
Breakover current
diac, 492
Breakover voltage
diac, 492, 592
thyristor, 490
Damper Diodes, 345, 367
forward recovery, 328, 348
losses, 347
outlines, 345
picture distortion, 328, 348
selection guide, 345
Darlington, 13
Data Sheets
High Voltage Bipolar Transistor, 92,97,331
MOSFET, 69
i
Index
Power Semiconductor Applications
Philips Semiconductors
dc-dc converter, 119
Depletion region, 133
Desaturation networks, 86
Baker clamp, 91, 138
dI/dt
triac, 531
Diac, 492, 500, 527, 530, 591
Diode, 6
double diffused, 162
epitaxial, 161
schottky, 173
structure, 161
Diode Modulator, 327, 367
Disc drives, 302
Discontinuous mode
see Switched Mode Power Supplies
Domestic Appliances, 527
Dropper
capacitive, 544
resistive, 544, 545
Duty cycle, 561
ESD, 67
see Protection, ESD
precautions, 67
ETD core
see magnetics
F-pack
see isolated package
Fall time, 143, 144
Fast Recovery Epitaxial Diode (FRED)
see epitaxial diode
FBSOA, 134
Ferrites
see magnetics
Flicker
fluorescent lamp, 580
Fluorescent lamp, 579
colour rendering, 579
colour temperature, 579
efficacy, 579, 580
triphosphor, 579
Flyback converter, 110, 111, 113
advantages, 114
clamp winding, 113
continuous mode, 114
coupled inductor, 113
cross regulation, 114
diodes, 115
disadvantages, 114
discontinuous mode, 114
electronic ballast, 582
leakage inductance, 113
magnetics, 213
operation, 113
rectifier circuit, 180
self oscillating power supply, 199
synchronous rectifier, 156, 181
transformer core airgap, 111, 113
transistors, 115
Flyback converter (two transistor), 111, 114
Food mixer, 531
Forward converter, 111, 116
advantages, 116
clamp diode, 117
conduction loss, 197
continuous mode, 116
core loss, 116
core saturation, 117
cross regulation, 117
diodes, 118
disadvantages, 117
duty ratio, 117
ferrite cores, 116
magnetics, 213
magnetisation energy, 116, 117
EFD core
see magnetics
Efficiency Diodes
see Damper Diodes
Electric drill, 531
Electronic ballast, 580
base drive optimisation, 589
current fed half bridge, 584, 587, 589
current fed push pull, 583, 587
flyback, 582
transistor selection guide, 587
voltage fed half bridge, 584, 588
voltage fed push pull, 583, 587
EMC, 260, 455
see RFI, ESD
TOPFET, 473
Emitter shorting
triac, 549
Epitaxial diode, 161
characteristics, 163
dI/dt, 164
forward recovery, 168
lifetime control, 162
operating frequency, 165
passivation, 162
reverse leakage, 169
reverse recovery, 162, 164
reverse recovery softness, 167
selection guide, 171
snap-off, 167
softness factor, 167
stored charge, 162
technology, 162
ii
Index
Power Semiconductor Applications
Philips Semiconductors
operation, 116
output diodes, 117
output ripple, 116
rectifier circuit, 180
reset winding, 117
switched mode power supply, 187
switching frequency, 195
switching losses, 196
synchronous rectifier, 157, 181
transistors, 118
Forward converter (two transistor), 111, 117
Forward recovery, 168
FREDFET, 250, 253, 305
bridge circuit, 255
charge, 254
diode, 254
drive, 262
loss, 256
reverse recovery, 254
FREDFETs
motor control, 259
Full bridge converter, 111, 125
advantages, 125
diodes, 126
disadvantages, 125
operation, 125
transistors, 126
Heat sink compound, 567
Heater controller, 544
Heaters, 537
Heatsink, 569
Heatsink compound, 514
Hi-Com triac, 519, 549, 551
commutation, 551
dIcom/dt, 552
gate trigger current, 552
inductive load control, 551
High side switch
MOSFET, 44, 436
TOPFET, 430, 473
High Voltage Bipolar Transistor, 8, 79, 91,
141, 341
‘bathtub’ curves, 333
avalanche breakdown, 131
avalanche multiplication, 134
Baker clamp, 91, 138
base-emitter breakdown, 144
base drive, 83, 92, 96, 136, 336, 385
base drive circuit, 145
base inductor, 138, 144, 147
base inductor, diode assisted, 148
base resistor, 146
breakdown voltage, 79, 86, 92
carrier concentration, 151
carrier injection, 150
conductivity modulation, 135, 150
critical electric field, 134
current crowding, 135, 136
current limiting values, 132
current tail, 138, 143
current tails, 86, 91
d-type, 346
data sheet, 92, 97, 331
depletion region, 133
desaturation, 86, 88, 91
device construction, 79
dI/dt, 139
drive transformer, 145
drive transformer leakage inductance, 149
dV/dt, 139
electric field, 133
electronic ballast, 581, 585, 587, 589
Fact Sheets, 334
fall time, 86, 99, 143, 144
FBSOA, 92, 99, 134
hard turn-off, 86
horizontal deflection, 321, 331, 341
leakage current, 98
limiting values, 97
losses, 92, 333, 342
Miller capacitance, 139
operation, 150
Gate
triac, 538
Gate drive
forward converter, 195
Gold doping, 162, 169
GTO, 11
Guard ring
schottky diode, 174
Half bridge, 253
Half bridge circuits
see also Motor Control - AC
Half bridge converter, 111, 122
advantages, 122
clamp diodes, 122
cross conduction, 122
diodes, 124
disadvantages, 122
electronic ballast, 584, 587, 589
flux symmetry, 122
magnetics, 214
operation, 122
synchronous rectifier, 157
transistor voltage, 122
transistors, 124
voltage doubling, 122
Heat dissipation, 567
iii
Index
Power Semiconductor Applications
Philips Semiconductors
optimum drive, 88
outlines, 332, 346
over current, 92, 98
over voltage, 92, 97
overdrive, 85, 88, 137, 138
passivation, 131
power limiting value, 132
process technology, 80
ratings, 97
RBSOA, 93, 99, 135, 138, 139
RC network, 148
reverse recovery, 143, 151
safe operating area, 99, 134
saturation, 150
saturation current, 79, 98, 341
secondary breakdown, 92, 133
smooth turn-off, 86
SMPS, 94, 339, 383
snubber, 139
space charge, 133
speed-up capacitor, 143
storage time, 86, 91, 92, 99, 138, 144, 342
sub emitter resistance, 135
switching, 80, 83, 86, 91, 98, 342
technology, 129, 149
thermal breakdown, 134
thermal runaway, 152
turn-off, 91, 92, 138, 142, 146, 151
turn-on, 91, 136, 141, 149, 150
underdrive, 85, 88
voltage limiting values, 130
Horizontal Deflection, 321, 367
base drive, 336
control ic, 401
d-type transistors, 346
damper diodes, 345, 367
diode modulator, 327, 347, 352, 367
drive circuit, 352, 365, 406
east-west correction, 325, 352, 367
line output transformer, 354
linearity correction, 323
operating cycle, 321, 332, 347
s-correction, 323, 352, 404
TDA2595, 364, 368
TDA4851, 400
TDA8433, 363, 369
test circuit, 321
transistors, 331, 341, 408
waveforms, 322
Ignition
automotive, 479, 481, 483
darlington, 483
Induction heating, 53
Induction motor
see Motor Control - AC
Inductive load
see Solenoid
Inrush current, 528, 530
Intrinsic silicon, 133
Inverter, 260, 273
see motor control ac
current fed, 52, 53
switched mode power supply, 107
Irons, electric, 537
Isolated package, 154
stray capacitance, 154, 155
thermal resistance, 154
Isolation, 153
J-FET, 9
Junction temperature, 470, 557, 561
burst pulses, 564
non-rectangular pulse, 565
rectangular pulse, composite, 562
rectangular pulse, periodic, 561
rectangular pulse, single shot, 561
Lamp dimmer, 530
Lamps, 435
dI/dt, 438
inrush current, 438
MOSFET, 435
PWM control, 455
switch rate, 438
TOPFET, 455
Latching current
thyristor, 490
Leakage inductance, 113, 200, 523
Lifetime control, 162
Lighting
fluorescent, 579
phase control, 530
Logic Level FET
motor control, 432
Logic level MOSFET, 436
Magnetics, 207
100W 100kHz forward converter, 197
100W 50kHz forward converter, 191
50W flyback converter, 199
core losses, 208
core materials, 207
EFD core, 210
ETD core, 199, 207
IGBT, 11, 305
automotive, 481, 483
clamped, 482, 484
ignition, 481, 483
iv
Index
Power Semiconductor Applications
Philips Semiconductors
flyback converter, 213
forward converter, 213
half bridge converter, 214
power density, 211
push-pull converter, 213
switched mode power supply, 187
switching frequency, 215
transformer construction, 215
Mains Flicker, 537
Mains pollution, 225
pre-converter, 225
Mains transient, 544
Mesa glass, 162
Metal Oxide Varistor (MOV), 503
Miller capacitance, 139
Modelling, 236, 265
MOS Controlled Thyristor, 13
MOSFET, 9, 19, 153, 253
bootstrap, 303
breakdown voltage, 22, 70
capacitance, 30, 57, 72, 155, 156
capacitances, 24
characteristics, 23, 70 - 72
charge, 32, 57
data sheet, 69
dI/dt, 36
diode, 253
drive, 262, 264
drive circuit loss, 156
driving, 39, 250
dV/dt, 36, 39, 264
ESD, 67
gate-source protection, 264
gate charge, 195
gate drive, 195
gate resistor, 156
high side, 436
high side drive, 44
inductive load, 62
lamps, 435
leakage current, 71
linear mode, parallelling, 52
logic level, 37, 57, 305
loss, 26, 34
maximum current, 69
motor control, 259, 429
modelling, 265
on-resistance, 21, 71
package inductance, 49, 73
parallel operation, 26, 47, 49, 265
parasitic oscillations, 51
peak current rating, 251
Resonant supply, 53
reverse diode, 73
ruggedness, 61, 73
safe operating area, 25, 74
series operation, 53
SMPS, 339, 384
solenoid, 62
structure, 19
switching, 24, 29, 58, 73, 194, 262
switching loss, 196
synchronous rectifier, 179
thermal impedance, 74
thermal resistance, 70
threshold voltage, 21, 70
transconductance, 57, 72
turn-off, 34, 36
turn-on, 32, 34, 35, 155, 256
Motor, universal
back EMF, 531
starting, 528
Motor Control - AC, 245, 273
anti-parallel diode, 253
antiparallel diode, 250
carrier frequency, 245
control, 248
current rating, 262
dc link, 249
diode, 261
diode recovery, 250
duty ratio, 246
efficiency, 262
EMC, 260
filter, 250
FREDFET, 250, 259, 276
gate drives, 249
half bridge, 245
inverter, 250, 260, 273
line voltage, 262
loss, 267
MOSFET, 259
Parallel MOSFETs, 276
peak current, 251
phase voltage, 262
power factor, 262
pulse width modulation, 245, 260
ripple, 246
short circuit, 251
signal isolation, 250
snubber, 276
speed control, 248
switching frequency, 246
three phase bridge, 246
underlap, 248
Motor Control - DC, 285, 293, 425
braking, 285, 299
brushless, 301
control, 290, 295, 303
current rating, 288
v
Index
Power Semiconductor Applications
Philips Semiconductors
drive, 303
duty cycle, 286
efficiency, 293
FREDFET, 287
freewheel diode, 286
full bridge, 287
half bridge, 287
high side switch, 429
IGBT, 305
inrush, 430
inverter, 302
linear, 457, 475
logic level FET, 432
loss, 288
MOSFET, 287, 429
motor current, 295
overload, 430
permanent magnet, 293, 301
permanent magnet motor, 285
PWM, 286, 293, 459, 471
servo, 298
short circuit, 431
stall, 431
TOPFET, 430, 457, 459, 475
topologies, 286
torque, 285, 294
triac, 525
voltage rating, 288
Motor Control - Stepper, 309
bipolar, 310
chopper, 314
drive, 313
hybrid, 312
permanent magnet, 309
reluctance, 311
step angle, 309
unipolar, 310
Mounting, transistor, 154
Mounting base temperature, 557
Mounting torque, 514
Power MOSFET
see MOSFET
Proportional control, 537
Protection
ESD, 446, 448, 482
overvoltage, 446, 448, 469
reverse battery, 452, 473, 479
short circuit, 251, 446, 448
temperature, 446, 447, 471
TOPFET, 445, 447, 451
Pulse operation, 558
Pulse Width Modulation (PWM), 108
Push-pull converter, 111, 119
advantages, 119
clamp diodes, 119
cross conduction, 119
current mode control, 120
diodes, 121
disadvantages, 119
duty ratio, 119
electronic ballast, 582, 587
flux symmetry, 119, 120
magnetics, 213
multiple outputs, 119
operation, 119
output filter, 119
output ripple, 119
rectifier circuit, 180
switching frequency, 119
transformer, 119
transistor voltage, 119
transistors, 121
Qs (stored charge), 162
RBSOA, 93, 99, 135, 138, 139
Rectification, synchronous, 179
Reset winding, 117
Resistor
mains dropper, 544, 545
Resonant power supply, 219, 225
modelling, 236
MOSFET, 52, 53
pre-converter, 225
Reverse leakage, 169
Reverse recovery, 143, 162
RFI, 154, 158, 167, 393, 396, 497, 529, 530,
537
Ruggedness
MOSFET, 62, 73
schottky diode, 173
Parasitic oscillation, 149
Passivation, 131, 162
PCB Design, 368, 419
Phase angle, 500
Phase control, 546
thyristors and triacs, 498
triac, 523
Phase voltage
see motor control - ac
Power dissipation, 557
see High Voltage Bipolar Transistor loss,
MOSFET loss
Power factor correction, 580
active, boost converted, 581
Safe Operating Area (SOA), 25, 74, 134, 557
forward biased, 92, 99, 134
reverse biased, 93, 99, 135, 138, 139
vi
Index
Power Semiconductor Applications
Philips Semiconductors
Saturable choke
triac, 523
Schottky diode, 173
bulk leakage, 174
edge leakage, 174
guard ring, 174
reverse leakage, 174
ruggedness, 173
selection guide, 176
technology, 173
SCR
see Thyristor
Secondary breakdown, 133
Selection Guides
BU25XXA, 331
BU25XXD, 331
damper diodes, 345
EPI diodes, 171
horizontal deflection, 343
MOSFETs driving heaters, 442
MOSFETs driving lamps, 441
MOSFETs driving motors, 426
Schottky diodes, 176
SMPS, 339
Self Oscillating Power Supply (SOPS)
50W microcomputer flyback converter, 199
ETD transformer, 199
Servo, 298
Single ended push-pull
see half bridge converter
Snap-off, 167
Snubber, 93, 139, 495, 502, 523, 529, 549
active, 279
Softness factor, 167
Solenoid
TOPFET, 469, 473
turn off, 469, 473
Solid state relay, 501
SOT186, 154
SOT186A, 154
SOT199, 154
Space charge, 133
Speed-up capacitor, 143
Speed control
thyristor, 531
triac, 527
Starter
fluorescent lamp, 580
Startup circuit
electronic ballast, 591
self oscillating power supply, 201
Static Induction Thyristor, 11
Stepdown converter, 109
Stepper motor, 309
Stepup converter, 109
Storage time, 144
Stored charge, 162
Suppression
mains transient, 544
Switched Mode Power Supply (SMPS)
see also self oscillating power supply
100W 100kHz MOSFET forward converter,
192
100W 500kHz half bridge converter, 153
100W 50kHz bipolar forward converter, 187
16 & 32 kHz TV, 389
asymmetrical, 111, 113
base circuit design, 149
boost converter, 109
buck-boost converter, 110
buck converter, 108
ceramic output filter, 153
continuous mode, 109, 379
control ic, 391
control loop, 108
core excitation, 113
core loss, 167
current mode control, 120
dc-dc converter, 119
diode loss, 166
diode reverse recovery effects, 166
diode reverse recovery softness, 167
diodes, 115, 118, 121, 124, 126
discontinuous mode, 109, 379
epitaxial diodes, 112, 161
flux swing, 111
flyback converter, 92, 111, 113, 123
forward converter, 111, 116, 379
full bridge converter, 111, 125
half bridge converter, 111, 122
high voltage bipolar transistor, 94, 112, 115,
118, 121, 124, 126, 129, 339, 383, 392
isolated, 113
isolated packages, 153
isolation, 108, 111
magnetics design, 191, 197
magnetisation energy, 113
mains filter, 380
mains input, 390
MOSFET, 112, 153, 33, 384
multiple output, 111, 156
non-isolated, 108
opto-coupler, 392
output rectifiers, 163
parasitic oscillation, 149
power-down, 136
power-up, 136, 137, 139
power MOSFET, 153, 339, 384
pulse width modulation, 108
push-pull converter, 111, 119
vii
Index
Power Semiconductor Applications
Philips Semiconductors
RBSOA failure, 139
rectification, 381, 392
rectification efficiency, 163
rectifier selection, 112
regulation, 108
reliability, 139
resonant
see resonant power supply
RFI, 154, 158, 167
schottky diode, 112, 154, 173
snubber, 93, 139, 383
soft start, 138
standby, 382
standby supply, 392
start-up, 391
stepdown, 109
stepup, 109
symmetrical, 111, 119, 122
synchronisation, 382
synchronous rectification, 156, 179
TDA8380, 381, 391
topologies, 107
topology output powers, 111
transformer, 111
transformer saturation, 138
transformers, 391
transistor current limiting value, 112
transistor mounting, 154
transistor selection, 112
transistor turn-off, 138
transistor turn-on, 136
transistor voltage limiting value, 112
transistors, 115, 118, 121, 124, 126
turns ratio, 111
TV & Monitors, 339, 379, 399
two transistor flyback, 111, 114
two transistor forward, 111, 117
Switching loss, 230
Synchronous, 497
Synchronous rectification, 156, 179
self driven, 181
transformer driven, 180
Thermal characteristics
power semiconductors, 557
Thermal impedance, 74, 568
Thermal resistance, 70, 154, 557
Thermal time constant, 568
Thyristor, 10, 497, 509
’two transistor’ model, 490
applications, 527
asynchronous control, 497
avalanche breakdown, 490
breakover voltage, 490, 509
cascading, 501
commutation, 492
control, 497
current rating, 511
dI/dt, 490
dIf/dt, 491
dV/dt, 490
energy handling, 505
external commutation, 493
full wave control, 499
fusing I2t, 503, 512
gate cathode resistor, 500
gate circuits, 500
gate current, 490
gate power, 492
gate requirements, 492
gate specifications, 512
gate triggering, 490
half wave control, 499
holding current, 490, 509
inductive loads, 500
inrush current, 503
latching current, 490, 509
leakage current, 490
load line, 492
mounting, 514
operation, 490
overcurrent, 503
peak current, 505
phase angle, 500
phase control, 498, 527
pulsed gate, 500
resistive loads, 498
resonant circuit, 493
reverse characteristic, 489
reverse recovery, 493
RFI, 497
self commutation, 493
series choke, 502
snubber, 502
speed controller, 531
static switching, 497
structure, 489
switching, 489
Temperature control, 537
Thermal
continuous operation, 557, 568
intermittent operation, 568
non-rectangular pulse, 565
pulse operation, 558
rectangular pulse, composite, 562
rectangular pulse, periodic, 561
rectangular pulse, single shot, 561
single shot operation, 561
Thermal capacity, 558, 568
viii
Index
Power Semiconductor Applications
Philips Semiconductors
switching characteristics, 517
synchronous control, 497
temperature rating, 512
thermal specifications, 512
time proportional control, 497
transient protection, 502
trigger angle, 500
turn-off time, 494
turn-on, 490, 509
turn-on dI/dt, 502
varistor, 503
voltage rating, 510
Thyristor data, 509
Time proportional control, 537
TOPFET
3 pin, 445, 449, 461
5 pin, 447, 451, 457, 459, 463
driving, 449, 453, 461, 465, 467, 475
high side, 473, 475
lamps, 455
leadforms, 463
linear control, 451, 457
motor control, 430, 457, 459
negative input, 456, 465, 467
protection, 445, 447, 451, 469, 473
PWM control, 451, 455, 459
solenoids, 469
Transformer
triac controlled, 523
Transformer core airgap, 111, 113
Transformers
see magnetics
Transient thermal impedance, 559
Transient thermal response, 154
Triac, 497, 510, 518
400Hz operation, 489, 518
applications, 527, 537
asynchronous control, 497
breakover voltage, 510
charge carriers, 549
commutating dI/dt, 494
commutating dV/dt, 494
commutation, 494, 518, 523, 529, 549
control, 497
dc inductive load, 523
dc motor control, 525
dI/dt, 531, 549
dIcom/dt, 523
dV/dt, 523, 549
emitter shorting, 549
full wave control, 499
fusing I2t, 503, 512
gate cathode resistor, 500
gate circuits, 500
gate current, 491
gate requirements, 492
gate resistor, 540, 545
gate sensitivity, 491
gate triggering, 538
holding current, 491, 510
Hi-Com, 549, 551
inductive loads, 500
inrush current, 503
isolated trigger, 501
latching current, 491, 510
operation, 491
overcurrent, 503
phase angle, 500
phase control, 498, 527, 546
protection, 544
pulse triggering, 492
pulsed gate, 500
quadrants, 491, 510
resistive loads, 498
RFI, 497
saturable choke, 523
series choke, 502
snubber, 495, 502, 523, 529, 549
speed controller, 527
static switching, 497
structure, 489
switching, 489
synchronous control, 497
transformer load, 523
transient protection, 502
trigger angle, 492, 500
triggering, 550
turn-on dI/dt, 502
varistor, 503
zero crossing, 537
Trigger angle, 500
TV & Monitors
16 kHz black line, 351
30-64 kHz autosync, 399
32 kHz black line, 361
damper diodes, 345, 367
diode modulator, 327, 367
EHT, 352 - 354, 368, 409, 410
high voltage bipolar transistor, 339, 341
horizontal deflection, 341
picture distortion, 348
power MOSFET, 339
SMPS, 339, 354, 379, 389, 399
vertical deflection, 358, 364, 402
Two transistor flyback converter, 111, 114
Two transistor forward converter, 111, 117
Universal motor
back EMF, 531
ix
Index
Power Semiconductor Applications
Philips Semiconductors
starting, 528
Vacuum cleaner, 527
Varistor, 503
Vertical Deflection, 358, 364, 402
Voltage doubling, 122
Water heaters, 537
Zero crossing, 537
Zero voltage switching, 537
x
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