A Design Study of a Future 10 kW Converter Sebastian Fant

A Design Study of a Future 10 kW Converter  Sebastian Fant

A Design Study of a Future 10 kW Converter

Examensarbete utfört i elektroniksystem av

Sebastian Fant

LiTH-ISY-EX--08/4093--SE

Linköping 2008

Examinator: Kent Palmkvist

Handledare: Jonny Lindgren

A Design Study of a Future 10 kW Converter

Examensarbete utfört i elektroniksystem vid Linköpings tekniska högskola av

Sebastian Fant

LiTH-ISY-EX--08/4093--SE

Presentationsdatum

2008-03-18

Publiceringsdatum (elektronisk version)

Institution och avdelning

Institutionen för systemteknik

Department of Electrical Engineering

Språk

Svenska x Annat (ange nedan)

Engelska

Antal sidor

93

Typ av publikation

Licentiatavhandling x Examensarbete

C-uppsats

D-uppsats

Rapport

Annat (ange nedan)

ISBN (licentiatavhandling)

ISRN LiTH-ISY-EX--08/4093--SE

Serietitel (licentiatavhandling)

Serienummer/ISSN (licentiatavhandling)

URL för elektronisk version http://www.ep.liu.se

Publikationens title

A Design Study of a Future 10 kW Converter

Författare

Sebastian Fant

Sammanfattning

This master thesis aim to design and evaluate a high power 3-phase DC/AC and AC/AC converter. The purpose is to use it for an electric motor in an aircraft possibly driving electric actuators or a propeller in an UAV or a small vehicle. Factors such as power loss and weight are of importance and will be estimated using known models supplied by various manufacturers of components. Different topologies of semiconductors suitable for this purpose are examined and presented. Extensive resources have been put to properly select the most suitable switching device according to their power loss and weight.

The need for filters and protective circuits will be estimated according to regulations of common military avionic standards and will be included in the resulting estimation along with simulations to evaluate their need and importance.

Snubber circuits will be presented and their specific ability to reduce voltage transients and switching losses will be examined along with some simulations to illustrate their performance.

In the final part an estimation of efficiency and weight of higher and lower power models of the same inverter has been made using the same procedure as presented in this paper. Engineering rules have been formed from these estimations to simply be able to calculate the proportions of a future inverter of arbitrary rated power.

Antal sidor: 93

Nyckelord

Inverter design, Power electronics, Snubber circuits, filter, IGBT

Abstract

This master thesis aim to design and evaluate a high power 3-phase DC/AC and AC/AC converter. The purpose is to use it for an electric motor in an aircraft possibly driving electric actuators, a propeller in an UAV or a small vehicle. Factors such as power loss and weight are of importance and will be estimated using known models supplied by various manufacturers of components. Different topologies of semiconductors suitable for this purpose are examined and presented. Extensive resources have been put to properly select the most suitable switching device according to their power loss and weight.

The need for filters and protective circuits will be estimated according to regulations of common military avionic standards and will be included in the resulting estimation along with simulations to evaluate their need and importance. Snubber circuits will be presented and their specific ability to reduce voltage transients and switching losses will be examined along with some simulations to illustrate their performance. In the final part an estimation of efficiency and weight of higher and lower power models of the same inverter has been made using the same procedure as presented in this paper. Engineering rules have been formed from these estimations to simply be able to calculate the proportions of a future converter of arbitrary rated power.

Keywords: Inverter design, Power electronics, Snubber circuits, filter, IGBT

I

Sammanfattning

Det här examensarbetet syftar till att designa och utvärdera en högeffektskonverter kapabel att konvertera lik eller trefas växelspänning till en trefas växelspänning med högre frekvens. Syftet är att driva en elektrisk motor för möjligtvis en domkraft eller en propeller för en obemannad eller småskalig flygfarkost. Faktorer såsom förlusteffekt och vikt är viktigt och kommer att uppskattas med hjälp av välkända modeller framtagna av flertalet tillverkare av komponenter. Olika topologier av halvledarkomponenter passande

ändamålet är undersökta och presenterade. Mycket tid läggs på att hitta de perfekta halvledarkomponenterna för ändamålet med tyngdpunkt på dess förlusteffekt och vikt.

Behovet av filter och skyddskretsars omfattning är uppskattade enligt standardiserade bestämmelser för flygfarkoster och kommer i slutskedet att bidra med både vikt och effektförluster. Simuleringar utförs i Matlab och Simulink för att visa dess behov och prestanda. Olika typer av snubberkretsar presenteras med dess unika egenskaper som syftar till att undertrycka spänningstransienter over de känsliga halvledarkomponenterna.

Simuleringar utförs i Pspice för att illustrera varje snubberkrets respektive för och nackdelar samt prestanda. I slutskedet av rapport har det genomförts en uppskattning av verkningsgraden och vikten gällande för konverters av högre och lägre märkeffekt. Det har gjorts analogt med det tillvägagångssätt som använts i denna rapport. Ingenjörsregler har blivit approximerade för att enkelt kunna uppskatta vikt och förlusteffekt för godtycklig märkeffekt.

Nyckelord: Inverter design, kraft elektronik, snubberkretsar, högeffektsfilter, IGBT

II

Acknowledgements

This report is a result of my master-thesis project carried out at SAAB Aerosystems in

Linköping under supervision of the department of Electrical Engineering at the

University of Linköping. As a student at the programme for applied physics and electro engineering I chose the alignment which felt the most natural and rewarding to me, electronics. This master-thesis was chosen among several available projects due to my interest in analog and high power electronics although the amount of the courses taken in the field was few. As I expect the need for this competence will grow significantly in the near I made the tactical choice by accepting the proposal offered to me.

Looking back over the time spent at SAAB Aerosystems it has been very interesting, rewarded and has completely fulfilled my expectations. I have been given opportunities to reflect the daily life of an avionic electrical engineer and the problems they face.

Great insight into the avionic industry has been given by my brilliant supervisor Lars

Austrin and I would like to thank him dearly for that. I would also like to thank Eduardo

Figueroa for giving me many discussions about the future life as an engineer and where it might lead. I would like to thank Andreas Johansson for his revising and the many practical hints he gave me. I would like to thank my co-worker Daniel Eidborn for a job well done and I wish you all the luck in the future.

Thank you SAAB and especially the TDG department for providing the finances and office necessary for completing this master-thesis.

Last but not the least I would like to thank my family and my friends for supporting me and my studies throughout the years.

Linköping March 2008

Sebastian Fant

III

Table of contents

LIST OF FIGURES.................................................................................................................................... VI

LIST OF TABLES...................................................................................................................................... VI

1.

INTRODUCTION............................................................................................................................... 1

1.1.

O

BJECTIVES

................................................................................................................................. 1

1.2.

B

ACKGROUND

.............................................................................................................................. 1

1.3.

C

HALLENGES

............................................................................................................................... 1

1.4.

R

ESEARCH METHOD

..................................................................................................................... 2

1.5.

D

ELIMITATIONS

........................................................................................................................... 2

1.6.

S

TRUCTURE OF REPORT

................................................................................................................ 2

2.

THEORY ............................................................................................................................................. 5

2.1.

C

ONVERTER PRINCIPLES

............................................................................................................... 5

2.2.

P

RINCIPLES FOR

C

ONTROL

........................................................................................................... 7

2.2.1.

Switching frequency................................................................................................................ 7

2.3.

P

OWER COMPONENTS

................................................................................................................... 8

2.3.1.

Switching transistors .............................................................................................................. 8

2.3.1.1.

IGBT ............................................................................................................................................ 8

2.3.1.2.

MOSFET.................................................................................................................................... 12

2.3.1.3.

BJT............................................................................................................................................. 13

2.3.2.

Free-wheeling-diodes ............................................................................................................13

2.3.2.1.

Silicon Schottky......................................................................................................................... 14

2.3.2.2.

Silicon Carbide Schottky............................................................................................................ 14

2.3.3.

Rectifier .................................................................................................................................14

2.4.

F

ILTERS

.......................................................................................................................................16

2.4.1.

Input filters ............................................................................................................................16

2.4.2.

DC-bus filter ..........................................................................................................................19

2.4.2.1.

Capacitor bank ........................................................................................................................... 19

2.4.2.2.

Choke inductor........................................................................................................................... 25

2.4.2.3.

Output filters .....................................................................................................................26

2.5.

S

NUBBER CIRCUITS

.....................................................................................................................30

2.5.1.

Increased SOA .......................................................................................................................30

2.5.2.

Reducing losses......................................................................................................................35

2.6.

P

RINCIPLES OF COOLING

..............................................................................................................36

2.6.1.

Thermal resistance ................................................................................................................36

2.6.2.

Heat transfer..........................................................................................................................36

2.7.

C

HASSIS DESIGN

..........................................................................................................................37

2.7.1.

Cooling scenarios ..................................................................................................................39

2.7.1.1.

Exterior heatsink ........................................................................................................................ 39

2.7.1.2.

Interior heat sink ........................................................................................................................ 42

2.7.1.3.

Internal fluid cooling.................................................................................................................. 43

2.8.

T

RANSISTOR DRIVES

...................................................................................................................43

2.9.

P

OWER SUPPLY

............................................................................................................................45

2.10.

C

ONTROLLER

..............................................................................................................................46

3.

LOSSES ..............................................................................................................................................47

3.1.

S

WITCHING LOSSES

.....................................................................................................................47

3.1.1.

Transistor ..............................................................................................................................47

3.1.2.

Diode .....................................................................................................................................48

3.2.

C

ONDUCTION LOSSES

..................................................................................................................49

3.2.1.

Transistor ..............................................................................................................................49

3.2.2.

Diode .....................................................................................................................................49

3.3.

L

OWEST LOSS ESTIMATION OF

IGBT

AND

FWD .........................................................................50

3.4.

R

ECTIFIER

...................................................................................................................................51

IV

3.5.

F

ILTERS

.......................................................................................................................................52

3.5.1.

Input filter ..............................................................................................................................52

3.5.2.

DC-bus filter ..........................................................................................................................52

3.5.3.

Output filter ...........................................................................................................................53

3.6.

C

ONTROLLER

..............................................................................................................................53

3.7.

T

RANSISTOR DRIVES

...................................................................................................................53

3.8.

I

NTERNAL POWER SUPPLY

...........................................................................................................54

4.

RESULTS ...........................................................................................................................................55

4.1.

P

OWER DENSITY

..........................................................................................................................55

4.2.

S

CALABILITY

..............................................................................................................................56

4.3.

E

FFICIENCY

.................................................................................................................................56

4.4.

R

ELIABILITY

...............................................................................................................................57

4.5.

S

IMULATIONS

..............................................................................................................................58

5.

CONCLUSIONS ................................................................................................................................59

6.

FUTURE WORK ...............................................................................................................................60

BIBLIOGRAPHY .......................................................................................................................................61

A

PPENDIX

A1 – O

UTPUT

C

HARACTERISTICS USING

S

URGE

C

URRENT

L

IMITER

.........................................63

A

PPENDIX

A2 – O

UTPUT

C

HARACTERISTICS USING

NO S

URGE

C

URRENT

L

IMITER

..................................64

A

PPENDIX

A3 – R

ECTIFIER

/IGBT C

URRENTS USING

S

URGE

C

URRENT

L

IMITER

.......................................65

A

PPENDIX

A4 – R

ECTIFIER

/IGBT C

URRENTS USING

N

O

S

URGE

C

URRENT

L

IMITER

.................................66

A

PPENDIX

B – S

YSTEM

S

IMULATION

M

ODEL

............................................................................................67

A

PPENDIX

C1 – M

AIN

G

RID

V

OLTAGE

H

ARMONICS WITHOUT

F

ILTER

......................................................68

A

PPENDIX

C2 – M

AIN

G

RID

C

URRENT

H

ARMONICS WITHOUT

F

ILTER

......................................................69

A

PPENDIX

C3 – M

AIN

G

RID

V

OLTAGE

H

ARMONICS WITH

F

ILTER

.............................................................70

A

PPENDIX

C4 – M

AIN

G

RID

V

OLTAGE

H

ARMONICS WITH

12-P

ULSE

R

ECTIFIER

.......................................71

A

PPENDIX

C5 – M

AIN

G

RID

C

URRENT

H

ARMONICS USING

12-P

ULSE

R

ECTIFIER

......................................72

A

PPENDIX

D1 – IGBT

CHART

1.................................................................................................................73

A

PPENDIX

D2 – IGBT

CHART

2.................................................................................................................74

A

PPENDIX

D3 – M

OSFET

/D

IODE CHART

....................................................................................................75

A

PPENDIX

E1 – D

ECOUPLING

C

APACITOR

S

NUBBER

C

IRCUIT

S

CHEMATICS

..............................................76

A

PPENDIX

E2 – R

ESTRICTED

D

ECOUPLING

C

APACITOR

S

NUBBER

C

IRCUIT

S

CHEMATICS

.........................76

A

PPENDIX

E3 – RCD C

HARGE

/D

ISCHARGE

S

NUBBER

C

IRCUIT

S

CHEMATICS

...........................................77

A

PPENDIX

E4 – RCD C

LAMP

-S

NUBBER

C

IRCUIT

S

CHEMATICS

.................................................................77

A

PPENDIX

F – L

AYOUT OF COMPONENTS WITHIN ENCLOSURE

...................................................................78

V

List of figures

F

IGURE

1 : I

NVERTER OVERVIEW

..................................................................................................................... 5

F

IGURE

2 : R

ECTIFIER STAGE

........................................................................................................................... 5

F

IGURE

3 : S

MOOTHING STAGE

........................................................................................................................ 5

F

IGURE

4 : S

WITCHING STAGE

......................................................................................................................... 6

F

IGURE

5 : F

ILTERING OF PULSE TRAIN IN OUTPUT STAGE

............................................................................... 7

F

IGURE

6 : S

IMPLIFIED MODEL

........................................................................................................................ 8

F

IGURE

7 : S

YMBOL LAYOUT

........................................................................................................................... 9

F

IGURE

8: T

URN

-

OFF BEHAVIOR

...................................................................................................................... 9

F

IGURE

9: I

NTERNAL STRAY CAPACITANCES IN AN

IGBT...............................................................................10

F

IGURE

10 : T

YPCIAL

IGBT T

URN

-

ON

............................................................................................................11

F

IGURE

11 : T

YPICAL

IGBT T

URN

-O

FF

..........................................................................................................12

F

IGURE

12: IGBT

WITH

FWD ........................................................................................................................13

F

IGURE

13: O

VERVEIW OF THE RECTIFIER

......................................................................................................15

F

IGURE

14 : I

NPUT FILTER

..............................................................................................................................17

F

IGURE

15 : T

YPICAL IMPEDANCE FOR INPUT FILTER VS

.

HARMONICS OF GRID FREQUENCY

...........................17

F

IGURE

16 : P

ROTECTIVE CIRCUITS

:

SIMPLE

,

SERIES AND PARALLEL CONNECTION

.........................................21

F

IGURE

17 : C

HARGE AND

D

ISCHARGE TIME OF CAPACITOR BANK

.................................................................22

F

IGURE

18 : E

LECTRICAL LAYOUT OF CAPACITOR BANK

................................................................................24

F

IGURE

20 : O

UTPUT FILTER

...........................................................................................................................26

F

IGURE

21 : D

ECOUPLING AND RESTRICTED DECOUPLING CAPACITOR

...........................................................30

F

IGURE

22 : S

WITCING DEVICE VOLTAGE WITHOUT SNUBBER CIRCUIT

...........................................................31

F

IGURE

23 : D

EVICE VOLTAGE WITH DECOUPLING CAPACITOR

.......................................................................32

F

IGURE

24 : D

EVICE VOLTAGE WITH DISCHARGE RESTRICTED DECOUPLING CAPACITOR

................................33

F

IGURE

25 : RCD C

HARGE

-D

ISCHARGE

S

NUBBER AND

RCD C

LAMP

-S

NUBBER

............................................34

F

IGURE

26 : RCD C

HARGE

-D

ISCHARGE

S

NUBBER

.........................................................................................34

F

IGURE

27 : RCD C

LAMP

-S

NUBBER

...............................................................................................................35

F

IGURE

28 : S

CETCH OF CHASSIS

....................................................................................................................40

F

IGURE

29 : T

HERMAL RESISTIVITY ESTIMATION

...........................................................................................40

F

IGURE

30 : H

EATSINK LAYOUT

.....................................................................................................................41

F

IGURE

31: T

HERMAL RESISTANCE FACTOR VS

.

AIRFLOW

..............................................................................42

F

IGURE

32 : D

ESCRIPTIVE DRIVER INTERNAL CIRCUIT

(IGBT

WITHIN DASHED LINE

) .....................................43

F

IGURE

33 : A

PPROXIMATE OUTPUT CHARACTERISTICS

.................................................................................49

List of tables

T

ABLE

1 : I

NPUT CURRENT HARMONICS

..........................................................................................................16

T

ABLE

2 : V

OLTAGE HARMONICS ON GRID

.....................................................................................................18

T

ABLE

3 : I

NPUT FILTER COMPONENTS

...........................................................................................................19

T

ABLE

4 : DC-

BUS FILTER COMPONENTS

........................................................................................................26

T

ABLE

5 : O

UTPUT FILTER COMPONENTS

........................................................................................................29

T

ABLE

6 : S

NUBBER CHARACTERISTICS

..........................................................................................................36

T

ABLE

7 :

SWITCHING CHARACTERISTICS VS

.

DRIVER DIMENSIONS

................................................................44

T

ABLE

8 : D

RIVER COMPONENTS

....................................................................................................................45

T

ABLE

9 : P

OWER SUPPLY SPECIFICATIONS AND DIMENSIONS

........................................................................46

T

ABLE

10 : C

ONTROLLER COMPONENTS

.........................................................................................................46

T

ABLE

11 : S

UITABLE DISCRETE

IGBT

S

.........................................................................................................50

T

ABLE

12: S

UITABLE

IGBT

MODULES

...........................................................................................................51

T

ABLE

13 : S

UITABLE RECTIFIERS

..................................................................................................................52

T

ABLE

14 : DC-

BUS FILTER LOSSES WHEN USING

230/400 VAC

INPUT

..........................................................52

T

ABLE

15 : DC-

BUS FILTER LOSSES WHEN USING

115/200 VAC

INPUT

..........................................................53

T

ABLE

16 : A

PPROXIMATE SEPARATED WEIGHT FOR

230/400VAC

APPLICATIONS

.........................................55

T

ABLE

17 : A

PPROXIMATE SEPARATED WEIGHT FOR

115/200VAC

APPLICATIONS

.........................................55

T

ABLE

18 : A

PPROXIMATE SEPARATED LOSSES

@ 6

K

H

Z SWITCHING FREQUENCY

.........................................57

VI

T

ABLE

19 : A

PPROXIMATE SEPARATED LOSSES

@ 10

K

H

Z SWITCHING FREQUENCY

.......................................57

T

ABLE

20 : A

PPROXIMATE SEPARATED LOSSES

@14

K

H

Z SWITCHING FREQUENCY

.......................................57

VII

List of symbols

V

DC

The voltage achieved on the DC-bus after the following smoothing

V

AC

Input/Output voltage in alternating mode

Collector current I

C

V

CD

V

GE

R

DS

V

DS

I

D

Abbreviations

PWM Pulse Width Modulation

FET Field Effect Transistor

IGBT Insulated Gate Bipolar Transistor

PMSM Permanent Magnet Synchronous Motor

SOA Safe Operating Area

FWD Free Wheeling Diode

LC Inductive-Capacitive

RLC Resistive-Inductive-Capacitive

MTBF Minimum Time Before Failure

HFE small signal Forward Current Gain

MASL Meters Above Sea Level

VIII

1. Introduction

As we all get more conscious about the damage the combustion of fossil fuels inflicts on the environment more and more alternative solutions come to light. Hybrid drive, biological petrol and power cells are terms most people are familiar with. Electric propulsion in automotives is being promoted throughout the world as the most environmental friendly solution on the market. Once electricity is produced the electric propulsion is completely clean and it is easier to produce clean electricity in a controlled environment such as a power plant. However, the environment issue is not the only benefit. The increased control, performance and efficiency achieved are in some applications the most wanted benefit, especially in avionics.

1.1. Objectives

This master thesis aims to design and evaluate a 10kW three-phase converter handling both AC/DC and DC/DC. The function of the converter is to drive a permanent magnetized synchronized motor possibly working as a starter motor or driving an electric actuator possibly in an aircraft. A main focus will be to pinpoint the parts responsible for the largest power loss, how to reduce this magnitude and estimate the weight of the product. As an extension the characteristics for a few imaginary converters with different power ratings will be estimated. This to in the end form up approximate equations giving the power loss and weight based of a few input parameters.

1.2. Background

Today it is not uncommon to use highly efficient petrol driven generators which supplies electronic equipment and motors with power and in the end reach a higher efficiency compared driving the equipment directly with combustion engines. The middle step between the engine and the power source is the converter, a device not seldom responsible for a large loss due to inefficiency, demanding cooling and thereby introducing further weight.

A new concept in aviation is the “More Electric Aircraft” which aims to replace all hydraulics with an electric correspondence. This is to reach higher efficiency which will reduce weight but also increase control and improved behavior.

With numerous motors demanding an individual converter its losses and weight becomes prominent and has to be optimized. The long term goal is to remove the compressor and improving the generator driven by the combustion engine to supply the electric power needed.

1.3. Challenges

The environments and conditions in an aircraft such as JAS 39 GRIPEN are very different from sea-level. This complicates the objective as electronic devices behave differently or possibly not at all when exposed to high temperature, cosmic rays and mechanical stress.

1

Cooling gets more difficult when air density decreases as it does on higher altitudes. A higher altitude also increases the failure rate due to cosmic radiation further extending the need for effective cooling. Since weight is an emphasized problem and cooling is responsible for most of it a well considered tradeoff is needed.

How can we maintain a low weight while still having a high reliability? How much will new technology reduce power dissipation? Is it possible to use the aircrafts built-in cooling system and will it improve overall efficiency?

• Literature study

• Examine which switching topology and which components to select for the optimum solution.

• Estimate the amounts of power dissipated, simulate if possible to confirm

• Estimate cooling needs and its size/weight

• Design and simulate filters to follow standard regulations

• Evaluate the operating area of the components and which counter measures to take to guarantee reliability and functionality.

• Form engineering rules to model scalability, power density and efficiency

1.5. Delimitations

This master thesis will not cover the physical construction or commissioning of the converter. The programming required in the controller will not be carried out. Some very modern applications will be neglected because it is too time consuming to estimate the gain over conventional solutions. Modeling of reliability will not be carried out very detailed

1.6. Structure of report

This report will be divided into seven chapters and below is a short description of each chapter.

1. Introduction –

2. Theory – A general introduction to power inverters is given, how they operate, difficulties etc. Also a light discussion and explanation of the concepts and behavior of the different components and parts building up the converter is carried out as well as some general calculations in aspect of cooling and weight of an enclosure.

3. Losses – A more specified estimation of the losses of the different components is carried out accompanied by simulations to hopefully confirm these estimations and

2

illustrate their effectiveness. Loss in filter will be estimated and suggestions for improvement will be carried out with respect to setbacks. Different snubber circuits will be evaluated with their pros and cons.

4. Results – Results with aspect to losses and its respective cooling are presented for a few suitable solutions along with their weight, reliability and cost.

A way of scaling is presented with their respective parameters.

5. Conclusions – Presents the conclusions of the work as well as gives answers to the challenges mentioned concerning the converter. General thoughts on outlooks of these applications are presented with regards to ongoing research in this field.

Also an evaluation of how the work has proceeded is included, which problems have risen and how they were handled.

6. Future work – Discusses improvements and work to be carried out to further optimize the converter. Which parts have been neglected etc?

7. Appendix – Various plots, spreadsheets and graphs from simulations as well as schematics from simulations.

3

4

2. Theory

• Overview

Figure 1 : Inverter overview

1. Rectifier

In the first step of the inverter the rectifier rectifies the tri-phase figure 2:

Figure 2 : Rectifier stage

2. Smoothing stage

The oscillating rectified DC-bus voltage will cause problems further on and has do be dealt with, this is done with a LC-filter (inductor-capacitor) which is low pass filtering the voltage but also providing an extra current buffer. As explained in 2.3.3. the oscillations can be as much as 9% of the DC-bus voltage and has to be decreased otherwise the noise would spread further and cause loss and misbehavior. A change in frequency or controlling the average DC-bus voltage with thyristors would alter the need for a filter. A higher frequency would make the ripple more frequent requiring a smaller capacitor but if the voltage is controlled a larger ripple would occur requiring larger capacitors, more about this in 2.4.1. After the DC-bus filters the voltage will look like it does in figure 3, with less ripple:

Figure 3 : Smoothing stage

5

If the DC-input would be used it would connected after the bypassed rectifier but still use the DC-filter to relieve the source in the same manner as the V chapter 2.4.1.

AC input filter covered in

3. Switching stage

Now as the transistor bridge has a much more stable voltage to process it can start chopping the voltage and forming the pulse width modulated sinus. The way of chopping and switching is decided by a controller which gives signals to specified drives for the switching transistors forming the pulse width modulated sinus wave. The transistor bridge output wave can look as described in figure 4 where the voltage is measured between one phase and another:

Figure 4 : Switching stage

4. Filtering and output stage

This rough sinus wave formed by the transistor bridge would inflict noise and EMI both in wires and in the load due to its step nature and has to be low pass filtered at least just before the load but rather directly after the transistor bridge. Having long wires with large voltage spikes would create high magnitude radio frequency disturbance which is very unwanted, especially in an aircraft. This filtration can be done using a three phase LCfilter but also in some cases the inductive coils of the motor can be used as a low pass filter if the converter is close enough to the load.

In the picture 5 the pulse train has been filtered with a LC-filter. As can be seen it resembles a sinus wave but with higher frequent noise. To remove this relative small noise additional high power coils and capacitors has to be included further increasing weight and size of the unit, again a tradeoff has to be made.

6

Figure 5 : Filtering of pulse train in output stage

2.2. Principles for Control

2.2.1. Switching frequency

Calculating and choosing a proper switching frequency of a PWM inverter is not always trivial, there as various advantages and disadvantages of choosing a too high or a too low switching frequency.

Low m f

(

m f

=

f sw

21 ) [12]. (2.2.1.1)

f out

Where f sw

is the switching frequency at which the transistors operate and f out intended maximum frequency at which the load operates, f out filtered version of f sw

. The major setback of a too low m f

is the

is thereby a low-pass and too low switching frequency is that the voltage-ripple on the sine-wave otherwise becomes prominent which causes inefficient ripple within the load, as seen in figure 5. A synchronous PWM is strongly advised since synchronizing the PWM means that the present output voltage is fed back to the controller via a voltage-divider and then errors are compensated for, greatly reducing the voltage-ripple and thus improving the efficiency. Synchronizing the circuit requires m f

to be an odd integer as well as more competent circuitry, including an

A/D-converter and perhaps hardware-support for arithmetic operations, however most modern circuitry includes this. These voltage ripples or harmonics mainly cause loss in effect but also can become a burden to people and mechanical devices since equipment and especially motors can start to vibrate significantly but also generate other non desirable side-effects. These vibrations can cause reduced lifetime in equipment due to stress but also emit harmful and annoying sound in often heard frequencies. However if a highly competent filter on the output is implemented many of the setbacks of a low m f can be avoided, the current ripple in the load can be reduced heavily as well as reducing the amount of emitted electromagnetic waves. The setback here is the additional weight and possibly significant additional loss

High m f

(

m f

=

f sw

21 ) [12]. (2.2.1.2)

f out

When the switching frequency increases the need for a synchronized PWM decreases since the amplitudes of the harmonics are very small and high frequent and thus somewhat insignificant. Although when controlling an AC-motor with variable frequency

7

the currents at very low output frequencies can become prominent and undesired, despite the very low amplitudes. As a result, synchronous PWM is almost always advised.

However, even if the overall negative effect of harmonics decreases with high switching frequency the switching loss in the switching device increases in proportion to the increased frequency which always is unwanted.

In this part mostly the semi conductive components responsible for the main part of the loss are considered, those are the main supply rectifier and the transistor-bridge.

Mainly two types of transistors have been considered, the IGBT and the MOSFET and their advantages and disadvantages are explained further on. A third option exists, the bipolar junction transistor (BJT) but as this one is current controlled it would require a driver capable of providing a high current while switching very fast. A combination which is very difficult to realize.

2.3.1. Switching transistors

2.3.1.1. IGBT

The Insulated Gated Bipolar Transistor is a device that combines the low forward conduction loss, especially at high voltages, of a Bipolar Junction Transistor (BJT) and the short switching times of the Metal Oxide Semiconductor Field Effect Transistor

(MOSFET). MOSFET on its own has very high conduction loss at high voltages while

BJT turns off and on much slower [12]. The two figures 6 and 7 illustrate a simplified model of an IGBT and the symbol layout:

Figure 6 : Simplified model

8

Figure 7 : Symbol layout

The upper figure consist of a Darlington coupled pnp-doped BJT and an n-channel

MOSFET with a resistor that corresponds to the drain drift region in the IGBT.

It is also common to see an npn-doped transistor between the base if the BJT and the emitter, this is to reduce the turn of tail illustrated in figure 8:

Figure 8: Turn-off behavior

As visible in figure 8, apart from this turn off tail the switching characteristics of an

IGBT resembles the ones of a MOSFET expect for the MOSFET usually being a lot faster.

To design and dimension the drive stage knowledge of the internal stray capacitances are required to reach a sufficient turn on and turn off time without a too large over-shoot and to maintain within the safe operating area (SOA). Although the stray capacitances vary with the voltage supplied over them making it a bit harder but there are usually provided some graphs displaying this in the datasheet. To understand the switching characteristics a more detailed explanation has to be done, however the IGBT-symbol can be simplified for switching evaluation as described in figure 9:

9

Figure 9: Internal stray capacitances in an IGBT

• IGBT Turn-On

When a positive voltage is applied on the gate (V

GE the gate through the gate resistor R

G

charging the C ge exponentially over the capacitor until reaching V

GE(th)

)at this point does not contribute much.

) a current (I

G

) will start to flow into

capacitor and the voltage rises

. The Miller effect capacitance

(C gc

Beyond this point the collector current (I

C

) starts to increase quickly and linearly to an over-shoot level depending on the semiconductor structure and the external circuit where it stabilizes as the V

GE instead of C ge

.

The gate current decreases to a level

reaches the Miller plateau since the C gc

now gets charged due to the low voltage at the collector. Since the voltage on the collector is decreasing the voltage on the gate remains rather constant when charging the C gc increases again after the V

CE reaches the V

CE(sat).

but

To finally stop at the maximum V

GE when both the gate capacitors are fully charged. The speed of the whole Turn-On process is directly linked to the gate resistor R g

, a smaller resistor speeds up the process while causing excessive oscillations or voltage spikes in the circuit. If a snubber circuit is used it can help to resize the components through filtering out unwanted parts of the signal and then be able to reducing the gate resistor and make the switching process faster. Although minimizing stray inductances in wiring and coils is the most effective way of reducing noise without particular setbacks except practical. On the other hand a larger resistor slows down the circuit but causes much less noise and voltage transients [9].

10

Figure 10 : Typcial IGBT Turn-on

The dissipated energy during each Turn-On can be calculated from the triangle infolded by the collector current I

C and the collector-emitter voltage V

CE

times the time period.

• IGBT Turn-Off

During Turn-Off the gate voltage turn to zero and current start to flow from the gate through the gate resistor, discharging both the gate capacitances, C ge

and C ce

until the

Miller plateau is reached. Changing the gate resistor does not change the time of the process like it did for the Turn-On except in a pure MOSFET where it is possible to decrease Turn-Off time by reducing this resistor [9]. Then the collector-emitter voltage

(V

CE

) starts to increase until reaching the DC-bus voltage. The gate-emitter voltage (V continues to decrease until passing the threshold voltage (V

GE(th)

GE

)

) and turning the IGBT off. Due to the bipolar part of the IGBT a current tail will arise as shown in the figure below, inflicting additional power loss. The current tail is highly unwanted but is very hard (impossible today), to eliminate completely.

11

Figure 11 : Typical IGBT Turn-Off

The losses are calculated in the same manner as during Turn-On, the triangular area infolded by the collector current and the collector-emitter voltage. In addition there is the current-tail area multiplied by the collector-emitter voltage. In datasheets the dissipated energy due to the current-tail is often included in the total Turn-Off energy.

2.3.1.2. MOSFET

A power Metal Oxide Semiconductor Field Effect Transistor (MOSFET) is just like a regular small signal MOSFET but larger in every sense. Larger current and higher voltages causes the internal capacitances and other critical parameters to suffer increases both switching and conduction losses. Even though slower than a signal MOSFET it is definitively faster than any IGBT. The appearance of the Turn-On and Turn-Off graphs for the MOSFET is very similar to the ones of the IGBT except for a much faster process and there are no current tails during Turn-Off. In addition, altering the gate resistor can reduce both Turn-On and Turn-Off time unlike the case with the IGBT where the latter was somewhat unchangeable. Even if the MOSFET is much faster it suffers large losses during forward conduction, at least when operating in high voltage applications, this partly due to the internal resistance growing exponentially with the rated V

DS described by 2.3.1.1.

as

R

DS

(

on

)

=

R

0

α

V

DS

(2.3.1.1)

Where

α

1 .

6 , V

DS

is the maximum rated voltage and R

0

the initial resistance [9].

This resistance along with the current forms up the voltage drop over the junction as in

2.3.1.2.

V

DS

(

sat

)

=

I

D

R

DS

(

on

)

=

I

D

R

0

α

V

DS

(2.3.1.2)

12

As seen the forward voltage drop increases very quickly with increasing current and especially the V

DS

voltage. The consequence of this is the inability to reach high efficiency while operating under high voltages and large currents since the resistance does not become small enough to compensate for the vast

2

I term. This loss is the dissipated power as shown as in equation 2.3.1.3.

P cond

=

I

2

D

R

DS

(

on

)

(2.3.1.3)

2.3.1.3. BJT

The power Bipolar Junction Transistor (BJT) is one of the components forming the

IGBT. The benefits of using a BJT are its capability of handling high currents and high forward voltages even if the reverse voltage capabilities are limited. The forward saturation voltage is almost independent on the current which keeps the conduction loss at a low level. In opposite to the MOSFET and the IGBT the BJT is a current controlled device and high power devices usually have a very low HFE

1

, usually a value around 10 for a 10kW application.

This demands very high currents from the driver to saturate the BJT as an unsaturated device will result in an unwanted high power dissipation most likely to cause failure in the device. The high base current along with a switching speed near the one of the IGBT makes the BJT an unsuitable device for this application.

2.3.2. Free-wheeling-diodes

A Free Wheeling Diode is an electronic component used to avoid damage to switching transistors by reversing load current induction. When switching off an inductive load, the current cannot go to zero in zero time since there is some energy stored in the magnetic field. The coil produces a high voltage large enough to let the current continue to flow over the contact gap, possibly causing permanent damage to the transistor as well as radiating radio waves. The free wheeling diode is connected anti-parallel with the transistor and by doing so it doesn't conduct normally as illustrated in figure 12:

Figure 12: IGBT with FWD

If the coil is switched off, the voltage across the coil reverses to maintain the direction of the current. Now the diode carries the current until the energy is consumed by the inner resistance of the coil and the forward voltage drop of the diode. This dissipated energy in the diode is depending on the forward voltage as well as the switching characteristics of the diode. Because of this, low forward voltage and small stray capacitances are wanted

1

Forward current gain

13

as well as low reverse recovery time which are characteristics that usually contradict each other. The reverse recovery time is the time taken from forward conduction to blocking in the reverse direction, this time directly causes loss on the circuit.

2.3.2.1. Silicon Schottky

Usually Schottky diodes are used which have very low reverse recovery time, slightly lower forward voltage drop and being much faster (much lower stray capacitance) compared to conventional diodes, although they have low maximum reverse voltage and a relatively high reverse leakage current that also increases with increasing temperature which makes them a bad choice in high voltage and high temperature applications.

2.3.2.2. Silicon Carbide Schottky

Since some ten years back other interesting materials are being researched. Diodes made of Silicon Carbide have proven to have excellent characteristics for high voltage, high frequency and high temperature. The reverse leaking current is up to 40 times less than for a regular Shottky, directly reducing losses, reverse voltage up to 1200 V and extremely low reverse charge as a result of junction capacitance, not stored charge. The setback is high price and a relatively high saturation voltage, introducing increased loss when conducting. By having high thermal conductivity and nearly no thermal runaway also makes the Silicon Carbide the best choice in applications with high temperature.

With special packing junction operating temperatures as high as 500 °K (227 °C) is made possible which opens up for a wide range of applications. The reverse recovery loss is usually a significant part of the total switching loss in a hard switched

2

IGBT and by almost reducing it to zero great reductions in dissipated effect and heat can be made.

2.3.3. Rectifier

The rectifier forms a direct current from an alternating current, in this application from three phase shifted sources of alternating current. In this case a 6-pulse rectifier model has been chosen due to its simplicity. However, when handling disturbances on the main grid a 12-pulse rectifier bridge is to prefer since it heavily reduces harmonics which otherwise will require large filters. A simulation has been made using Matlab with

Simulink where a 6-pulse rectifier is used and the resulting voltage frequency spectrum on the grid is measured, see Appendix C1 and C2. Analogous the same measures are done using a 12-pulse rectifier where it is observed that the noise due to harmonics is significantly lower, see Appendix C4 and C5. The setback of the 12-pulse bridge is the needed high power transformer and an additional 6- pulse bridge, together largely contributing to additional weight. The mean voltage archived on the DC side is calculated as a combination of all the input voltages as seen in equation 2.3.3.1 [12]:

V

DC

_

AVE

=

3

×

2

π

×

V

L

L =

3

×

2

π

×

400

≈540 V (2.3.3.1)

2

Switching with no snubber circuits or filters and an inductive load

14

This is an average voltage and it will actually oscillate between 490 and 566 volts at the input frequency multiplied by six demanding a filter to provide a fixed voltage. For a

10kW application the maximum DC-bus current will be as large as 18.5A with reservation for the result of the DC-bus filter temporarily capable of supplying more than

18.5A The choice of a rectifier bridge is mainly focused on low energy loss and weight where low energy loss needs less cooling and therefore less weight although in some applications disturbances and harmonics will cause problems and has to be given higher priority. However the weight and size of the rectifier itself has in some cases proven to differ a lot. The ability to withstand heat and to remove this heat should also be taken into consideration, represented by the thermal resistance of the package. The layout of the rectifier diodes can be found in figure 13 (encircled):

Figure 13: Overveiw of the rectifier

15

2.4. Filters

2.4.1. Input filters

The non-linear nature of the converter with especially its rectifier and the inductive load will form a load on the main grid that is far from ideal

3

. Noise generated back on the grid is a problem for all the other connected equipment but will also emit RMI if not protected either by shielded wires or an input filter that compensating for this behavior. Usually it exist a lot of regulations concerning how inductive a load can be and how much noise a load can inflict, especially in scenarios where the main supply is weak

4

and where it drives sensitive electronic equipment, such as in an aircraft or other vehicles.

In the scenario covered by this master-thesis a 400 Hz supply is given which implies that it should not contain high magnitude components at other frequencies. Because of this a band-pass filter has to be implemented at the input to reduce the magnitude of unwanted components reflected back to the supply. The components are the fundamental frequency component and its harmonics of order 6 k

±1

5

. This periodic order due to the switched operation of the line commutated rectifier, in this case a three-phase diode bridge. The currents harmonics can further be resolved into sequences according to the following table:

Sequence Harmonics

Positive 1,7,13,19,…

Negative 5,11,17,23,…

Table 1 : Input current harmonics

Usually it is trivial to filter out these harmonics but the high power application along with the low weight goal makes it difficult to remain within the SOA of all the components.

In figure 14 is the schematic of a simple input filter that attenuates the first two harmonic components of the current as well as a wide range of high frequency noise. More filters can be added to filter out additional harmonics but will add weight and volume to the device. How much attenuation of respective harmonics is demanded is usually set by the environment. In an aircraft there is usually very sensitive equipment on a weak power source increasing the needs for good filtering. Most of the regulations used and followed by SAAB AB are found in MIL-STD-704 produced by the US department of defense

[19]. Too some extent high frequency components emitting radio waves can be contained using shielded cables.

3

Ideal is a purely sinusoidal current

4

The source is considered weak if the voltage is reduced significantly when loaded

5

K is any positive integer, which means the orders can be as in table 1

16

Figure 14 : Input filter

As it is hard to analytically calculate the behavior of currents and voltages in the filter components an iterative approach using extensive simulations is usually taken. This is due to the analytical result being valid only during the steady state and not during start up or shutdown where surge currents will occur. These surge currents can cause failure or reduce lifetime in the filter components and especially in the electrolyte capacitor if not dimensioned properly.

Figure 15 : Typical impedance for input filter vs. harmonics of grid frequency

The amounts of harmonics reflected back from the inverter to the grid is also dependant on the LC-filter between the rectifier and the switching bridge. This because currents through the diodes is dependant on the charging currents of the capacitor which itself is dependant on the choke inductor and the surge limiter. These harmonics are also hard to analytically estimate and a system level simulation is favored. To estimate the harmonics in this converter Matlab with Simulink will be used where the entire converter is modeled and the frequency spectrum on the input is measured. When having the high amplitude harmonics determined a band-stop filter for each undesired harmonic has to be implemented with appropriate attenuations. This frequency spectrum measurement can be seen in Appendix C1 where the harmonics can easily be identified. The maximum allowed disturbance can be found in the MIL-STD-704D military standard document

17

[19]. In table 2 the larger harmonics in comparison with the maximum allowed levels and the required attenuation can be viewed:

Harmonic # Frequency(hz) Amplitude peak/RMS V Allowed level RMS Attenuation dB fundamental 400 550/388 -

11 4400 20/14

13 5200 14/10

17 7800 8/5.7

20

20

20

16

12

2.3

-

-

-

-

Table 2 : Voltage harmonics on grid

As in table 2 the given voltages arise due to harmonic currents through the source usually represented by a generator the respective current components can be filtered instead. This is carried out as explained in figure 14 with serial RLC circuit connected to next phase or virtual ground. The impedance of the RLC circuit is usually calculated as in 2.4.1.1:

RLC

Z

=

R

2

+

ω

L

1

ω

C

2

(2.4.1.1)

The RLC impedance is at minimum at the resonance frequency

ω

=

1

LC

where it assumes the value R. This value R limits the current at the filtered frequency preventing it to become unnecessary large. The L and C values can be chosen somewhat arbitrary as long as the resonance frequency is the correct one. The current harmonics can be found in appendix C along with the voltage harmonics.

As seen in table 2 the voltage at the 5 th

harmonic caused by its respective harmonic current has to be reduced to 59% relying on the filter to consume this excessive current.

It is also the only component needed to be individually reduced, the higher frequency components are attenuated with a high-pass filter as illustrated in figure 14. Given the

RMS voltage 34 V and the RMS current 2.52 A at the frequency 2000 Hz of which the filter will consume 41 % gives the value of R according to 2.4.1.2:

R

=

I

V harmonics

_

harmonic

_

rms rms

×

0 .

41

=

2 .

52

34

×

0 .

41

=

32 .

9

Ω

(2.4.1.2)

Assuming the value 100µH for the inductor L gives along the resonance frequency 2000

Hz the capacitor C a value of 63 µF. The estimated voltage and current give a power loss of 34W. Although this is per phase and should be multiplied by three when added to the complete system. This has to be done with the components as well, adding weight and space. The combined weight of the 5 th

harmonic filter per phase is estimated to 120g which gives a total of 360g. The dimension can be seen in the layout sketch, in Appendix

F

18

The high-pass filters are calculated in the same manner but when calculating loss a few more harmonic components are added, although smaller in magnitude. A 10Ω with a

100nF capacitor in series nearly does not interfere with the 5 th

harmonic filter but filter higher order harmonics rather effectively. The accumulated power loss is estimated to be about 3 watts per phase which will all be dissipated in the resistor. The estimated weight per phase is 11 g, giving a total of 33g.

Component Filter

R 5 th harmonic

L 5 th harmonic

C 5 th harmonic

Ratings Manufacturer Mass and Model (g)

100μH Pulse

L100

PEH200

Volume

(mm

3

)

3*19 3*(15x25x24)

10Ω VITROHM 3*5 3*(8x8x22)

KH

Power loss (W)

3*34

3*3

100nF 3*7 3*(11x22x26)

PHE845

Table 3 : Input filter components

2.4.2. DC-bus filter

2.4.2.1. Capacitor bank

Because of the nature of the rectifier the voltage on the DC-bus will vary with time, but not only because of this but from the pulse shape of the output current as well. Due to this a LC-filter is needed to stabilize the voltage as well as providing a current buffer. The performance of the filter is limited by the weight and cost always wanted to be kept low.

As electrolyte capacitors are used the lifetime is limited and estimated using a simple model to assure lifetime long enough according to regulations.

The dimensioning of the components requires some extensive calculations which in detail will be explained in this chapter starting with reviewing the provided parameters.

• Main power grid: 400VAC, 400Hz three-phase, 560VDC maximum

• Capacity: 10kW with 10kHz switching frequency

• Full wave bridge rectifier: 400*6 Hz ripple frequency

• Maximum allowed ripple voltage: 3% of average DC-bus voltage i.e. 16 V.

• Common MTBF: 5000 hours of flight/operation

Assuming all the energy is stored in the capacitor bank a calculation gives according to the well known formula of the potential energy in a capacitor

E cap

=

1

2

C tot

×

V

2

(2.4.2.1.1)

19

and its extension to fit the actual case.

P out

×

t ripple

=

1

2

C tot

×

(

V

DC

_

MAX

2

(

V

DC

_

MAX

V ripple

)

2

)

(2.4.2.1.2)

Where P out

is the rated power of the inverter, V

DC_MAX

is the maximum voltage of the rectified output, V ripple

is the maximum ripple allowed and t ripple

is the period time of the ripple. The ripple frequency for a full wave bridge is the main grid frequency multiplied by six. Solving for C tot

in (2.4.2.1.3) with insertion of the proper values gives:

10000

×

1

2400

=

1

2

C tot

×

(

566

2

( 566

16 )

2

)

C tot

=

4 , 67

×

10

4

= 467 μF (2.4.2.1.3)

As very few capacitors exist capable of handling this high voltage and still maintain a large capacitance multiple capacitors has to be connected in series. Although when serial connected a resistor has to be in parallel with each capacitor as explained more in detail later in this chapter. A common way is to use two legs in parallel with two identical serial connected capacitors maintaining the same capacitance as a single one while doubling the maximum voltage. This precaution due to the voltage peaks during start-up when the voltage over the capacitor can rise to a level between 1.4 and 1.8 times normal depending on the size of the choke inductor and the protective circuits. The surge current appear because during start-up the capacitor bank is virtually short circuited and the choke inductor will try to maintain this current, thus inflicting a high voltage transient. The surge current and voltage over-shoot can be reduced or almost eliminated with these protective circuits as illustrated in figure 16. The switch in this case is a high power transistor, most likely an IGBT with a low saturation voltage. As the resistor will initially limit the charge time of the capacitor it will also decrease the over-shoot. The surge limiter can be expanded into several levels reducing the resistance sequentially although many copies of this circuit in series will accumulate a large on-resistance in steady state and therefore add a non negligible loss. However, if connected in parallel they can replace each other instead and will end up with only one transistor in series causing a smaller on-resistance and a lower power loss. When this rather fast switching behavior has occurred it can be relieved by a relay, these are usually much slower but have a much smaller on-resistance than any semiconductor almost eliminating the added loss. A bonus of this circuit is that it limits the converter inrush current significantly from a level that would probably destroy the rectifier to a reasonable current. Simulations of this behavior with and without the surge limiter can be seen in appendix A which is based on a system simulation done in Matlab and Simulink, the circuitry can be found in appendix B.

20

Figure 16 : Protective circuits: simple, series and parallel connection

Not only does the capacitor require a wide margin for voltage but also for the capacitance where tolerance (

±20%) and wear-out (-10% can reduce the capacitance with up to 28%

(1-0,8*0,9).

Taking these precautions suggests the Evox Rifa PEH200 electrolyte capacitor with the ratings 400V and 680 µF with an individual weight of 180g. The four combined capacitors get a rating of 800V maximum and still has the same capacitance as a single one but of course weights four times as much i.e. 720g.

A doubling in the rated power of the inverter would result in at least doubling the capacitance required suggesting the 1500 µF version which would weight 430g each increasing the weight of the capacitor bank by at least 140%.

However, the next step would be to calculate the ripple currents from the AC-line and from to the load by first calculating the capacitor charging time with 2.4.2.1.4 [15]:

T

C

=

V

arccos

2

×

DC

π

_

MAX

V

DC

×

f

V ripple

_

MAX grid

= arccos

566

566

16

2

×

π

×

400

=94.8 μs (2.4.2.1.4)

The capacitor charge time can be illustrated and defined as in figure 17:

21

Figure 17 : Charge and Discharge time of capacitor bank

With the charge time and the period time of the ripple voltage it is easy to derive the capacitor discharge time as in equation 2.4.2.1.5.

T

DC

=

T ripple

T

C

=

1 −

2400

0 , 0000948 =321.8 μs (2.4.2.1.5)

Based on the change in voltage vs. time (dV/dt), the peak and RMS charge current (the rectifier output current) through each capacitor leg can now be calculated as in equation

2.4.2.1.6 and 2.4.2.1.7.

I

Cpeak

=

C

×

dV dT

C

=

39.4

A (2.4.2.1.6)

Where C is the capacitance in each leg, which is

C tot

2

due to the serial connection.

I

Crms

=

I

Cpeak

2

×

T

C

×

f ripple

=

18.8

A (2.4.2.1.7)

In the same manner the peak and RMS discharge currents can be calculated using the discharge time instead as in equation 2.4.2.1.8 and 2.4.2.1.9:

I

DCpeak

=

C

×

dV dT

DC

=

11.6

A (2.4.2.1.8)

I

DCrms

=

I

DCpeak

2

×

T

DC

×

f ripple

=

10.2

A (2.4.2.1.9)

With calculations done in equation 14 and 15 the ripple current resulting from the rectification of the grid can now be calculated for each branch of the capacitor bank as in equation 2.4.2.1.10.

I rms

=

2

I

Crms

+

2

I

DCrms

=

21.4

A (2.4.2.1.10)

Thus, through the whole capacitor bank it flows twice as much since it has two legs adding up to 42.8 A.

22

The load ripple current for the whole capacitor bank is calculated as in equation

2.4.2.1.11.

I load

_

rms

=

⎜⎜

V

DX

_

MAX

+

(

V

P rated

DC

2

_

MAX

V ripple

)

⎟⎟

=

10000

566

+

2

550

=

17.92

A

(2.4.2.1.11)

Which gives a load current of 8.96 A per leg. The total makeup of the current through the capacitor bank is 17.92 A @ 10 kHz and 42.8 A @ 2400 Hz.

The next thing is to determine the power loss achieved in the capacitor bank and whether it is to large enough to require extensive cooling or a large choke inductor to maintain sufficient lifetime. The power loss can be calculated by somewhat altering the well known formula P = U x I as done in equation 2.4.2.1.12.

P loss

=

U

×

I

=

R

×

I

2

=

I rms

2

×

ERS

(2.4.2.1.12)

Where the ERS is the equivalent series resistance which can be found in diagrams within the datasheet describing the capacitor. This ESR is strongly dependant on the frequency and the hotspot

6

temperature (T ht

) which is typically 20º C warmer than ambient temperature(T a

) of 70º C [15].

These expected typical values are found to be: ESR(2400Hz) = 13.5 mΩ

ESR(10kHz) = 12.5 mΩ

Using the estimated ESR values and the calculated current at each frequency two components of power loss can be estimated for each capacitor, trivially four times this loss would give the total loss of the capacitor bank excluding the choke inductor.

P loss

( 2400

Hz

)

=

21 .

4

2 ×

0 .

0135

=

6.18

W

P loss

( 10000

Hz

)

=

8 .

96

2

×

0 .

0125

=

1 W (2.4.2.1.13)

This gives each capacitor a loss of about 7 W which is dissipated through heat. Whether the capacitor can withstand this produced heat is dependant on the thermal resistance between hotspot and the ambient air as well as the temperature of this ambient air.

A too high temperature at the hotspot will result in a reduced lifetime and this temperature can be calculated with equation 2.4.2.1.14.

P loss

=

T ht

R ht

_

T a a

T ht

=

P loss

×

R ht

_

a

+

T a

=

7 .

18

×

3 .

4

+

70

=

94 .

4

°

C

(2.4.2.1.14)

The thermal resistance 3.4ºC/W between the hotspot and the ambience (R ht_a

) can be found in the datasheets under a variety of conditions, this one is without any heatsink or

6

The center of the capacitor where the highest temperature is likely to arise.

23

airflow. As it can be seen the temperature at the hotspot is very close to where the ESR was specified negating the need for further more detailed calculations concerning the loss. However, if this increase in temperature would affect the lifetime too much the choke inductor can be designed to extensively filter the high frequency components.

Usually it is just meant for suppressing the high frequent ripple.

Suppose the temperature would be too high and the power loss needed to be reduced.

To keep the hotspot temperature at 90 º C 1.3 W is needed to be transferred to the choke inductor as explained in chapter 2.4.2.2.

However, for the intended voltage sharing of the two serial capacitors to work correctly a resistor has to be connected in parallel with each capacitor. The resistance of this resistor is dependent on the capacitor value according to equation 2.4.2.1.15 [15]:

R vsr

=

0 .

1000

015

×

C

[ ]

=143 kΩ (

2.4.2.1.15)

As a precaution the power rating of the resistor should be at least 50% higher than estimated and the tolerance should be kept below 5% to prevent failure. The electrical layout is illustrated as in figure 18, the physical layout on the other hand should not be taken to easily as the inductance has to be kept low.

Figure 18 : Electrical layout of capacitor bank

According to the equation for lifetime the capacitor will live for at least 30 kH which is far beyond the limit of 5 kH. The uncertainty is due to few values of expected rate of failure where an average had to be taken. Although in the worst case the lifetime is still enough by far.

(

85

T ht

) (

85

94

)

LOP

=

A

×

2

C

=

6

×

10

4 ×

2

12

≈36000 h

(

2.4.2.1.16)

Where A is the lifetime at reference temperature and C is the rise in degrees required for cutting the lifetime in half, both of these available from the data sheet.

24

However, this lifetime can still prove uncertain in avionic applications as the thermal resistance due to convection will increase with altitude resulting in a higher hotspot temperature. The air pressure will also affect the lifetime of the capacitor as the electrolyte degrade faster.

2.4.2.2. Choke inductor

The choke inductor is mainly for reducing the transients capable of causing fatal failure in the capacitor bank. However it can be designed to filter some of the high frequency components of the charge current resulting in transferring the loss from the capacitor to the inductor

Suppose the 94.4 ºC is too high and need to be reduced to 90 ºC, based on previous calculations a reduction of 1.29 W is needed in each capacitor. This combined loss reduction achieved in the four capacitors will be withdrawn from the inbound charge current. A rough estimation for the needed reduction in voltage can be done by using the maximum ripple voltage over the capacitors and the reduction in power as explained by equation 2.4.2.2.1.

V choke

=

V ripple

P reduction

P initial

=

16

×

1 .

29

6 .

18

=

3 .

34

V pk

pk

(

2.4.2.2.1)

V choke

_

rms

=

V pk

pk

2 .

11

=

1 .

58

V rms

(

2.4.2.2.2)

The 2.11 factor in (is different than the normal 2

×

2 since the voltage is not pure sinusoidal to its form. The remaining fundamental ripple current can be estimated using the new power loss and the ESR of the capacitors as in equation 2.4.2.2.3.

I ripple

=

P loss

_

branch

ESR branch

×

n branches

=

2

×

2

(

6 .

18

×

1 .

29

0 .

0135

)

×

2 =38.1 A

(

2.4.2.2.3)

This ripple current is a small reduction from the former value of 39.4 A and now the proper inductance can be calculated as in equation 2.4.2.2.4:

X

L

=

2

×

π

×

f ripple

×

L

=

U

I

L

=

2

×

π

×

U f ripple

×

I

=

2

×

π

×

1 .

58

2400

×

38 .

1

=2.75 μH (2.4.2.2.4)

The power capabilities of the choke inductor are usually done with some empirically established rules actually meant for transformers. However the figures given above are enough to choose an inductor from an arbitrary manufacturer. Consider the DC3-56G from WILCO, it weights approximately 30g each and measures 28 x 21.3 mm, to cover the current two are used in parallel, hence the inductance of 5.6 µH.

Component Ratings Manufacturer and Model

Mass

(g)

Volume

(mm

3

)

Power loss (W)

25

L 5.6μH WILCO

DC3-56G

2*30 3*(28x22) 4*1.3

PEH200

Table 4 : DC-bus filter components

If there are long wires between the output of the inverter and the inductive load the stray impedance (usually mostly capacitive) can induce annoying behavior both in the motor and the driving transistors. The long wires would also serve as a good antennae emitting

EMI caused by the transistor transients when switching fast. Although, the problem with emitted EMI can be treated with proper shielding of the conducting wire. In the following chapter advantages/disadvantages and guidelines for an approximate design of an output filter is carried out under the assumption that the cable is no longer than 30 meters. Near and above this limit several phenomenons occur such as standing waves due to impedance mismatch between the motor and the cable. Such standing waves can cause serious over voltages and current oscillations too strong to be managed by a LC-filter.

Many motor manufacturers have recently published maximum dV/dt ratings for their product, usually around 5 V/ns while the commutation of state of the art transistor can be far above this rating, up to 10 V/ns or even more. Here a mean to reduce dV/dt at the motor terminals are required and such a mean is often a simple LC-filter. It has also been proven that placing the filter near the motor terminals can further improve this misbehavior compared to placing it on the inverters board. However, as the task at hand is to evaluate a complete inverter solution the choice is to have the filter within the inverter unit, this filter will also be designed without any snubber circuits implemented.

In figure 20 is an illustration presented where a LC-filter has been added between the output of the transistor-bridge and the motor, although only one phase is represented. The motor has been modeled as a large inductor L

2 winding.

corresponding to the phase/neutral

Figure 19 : Output filter

26

Where V

1

represents the DC-bus voltage and the two capacitors represents the Collector-

Emitter-capacitor (C ce

) which also plays an important role during commutation.

Obviously the design of the filter is very simple and by assuming I peak current, C

1

can be chosen as in equation 2.4.2.3.1.

is the peak motor

C

1

=

I peak

(2.4.2.3.1)

dV

/ dt max

C

1

softens the switching from the IGBTs so that the maximum dV/dt at the motor terminals is kept within limits supplied by the manufacturer or the EMI specifications are met. However, C

1 cannot be left alone as there would run very large current peaks through the IGBT leg during turn most likely to trigger the over-current protection handled by the controlling circuit. This is where the inductor L

1

comes in to block this high charge current and the dimensioning of the inductor will be such that current peaks in the IGBT (or C

1

) does not trigger the over-current protection, at least. The over-current protection is dimensioned in advance with respect to the switching devices and their current handling capabilities. As so far this is a simple LC circuit and it is driven by very high dV/dt it will soon start go generate oscillations higher than the DC-bus voltage which is far from what the filter components and the motor are designed for. A damping resistor in series with the capacitor would suppress this behavior and in addition removing un-useful power dissipation otherwise dissipated in the capacitor. When this L

1

C

1 follows by equation 2.4.2.3.2 for the initial dV/dt:

and is involved the expression in equation 23 is obsolete due to the resonance between L

1

. Thus the actual dV/dt will also be determined by L

1

giving a new expression as

dV dt

=

L

V

DC

1

×

C

1

(2.4.2.3.2)

The value of L

1

will be designed taking into account that each IGBT sees the phase current of the motor, the recovery current of the opposite recovery diode and the peak current of the filter. Although a fourth component could exist if the commutation causes cross talk but it is not very common due to very potent controlling circuits and drivers.

After considering over-current limits and the peak recovery current the allowed peak current contribution of the output filter can be determined. Excluding the damping resistor R

2 the Z

C

the current through the filter can be calculated as in equation 2.4.2.3.3 where

is the characteristic impedance of the filter as determined in equation 2.4.2.3.4.

I filter

_ max

=

V

DC

Z

C

(2.4.2.3.3)

Z

C

=

L

1

C

1

(2.4.2.3.4)

27

But as explained earlier the damping cannot be zero and thus the effect of R

2 considered. Critically damping is achieved when R could be chosen due to C

8

and C

9

. Suppose R

2

2

is equal to Z

is chosen to Z

C

C the IGBTs due to the output filter would be as determined in equation 2.4.2.3.5:

has to be

but a lower value

, then the current peak in

I filter

_ max

=

V

2

×

DC

(2.4.2.3.5)

Z

C

As current limiting is not the only constraint for L1 it also has to be designed to ensure enough build-up voltage in the motor’s coils during a very low duty cycle [19]. The minimum duty cycle time should at least be twice as long as the resonance cycle caused by the LC circuit. Neglecting the damping caused by R

2 gives the proper value of L

1 is known.

and adding some margins instead

as in equation 2.4.2.3.6 assuming the minimum turn-on time

π

×

L

1

×

C

1

T on

_ min

(2.4.2.3.6)

Combining the 2.4.2.3.2 and 2.4.2.3.6 an expression for the maximum dV/dt related to

V

DC

and the minimum turn-on time is achieved by equation 2.4.2.3.7.

dV dt

=

V

DC

T on

_

× min

π

(2.4.2.3.7)

The result achieved in 2.4.2.3.7 seems highly logical as it would require a shorter turn-on time to get a higher dV/dt. This overall gain in less EMI and enough slow dV/dt for the motor does not come without loss, both in more weight and in more direct power loss.

The loss is dissipated in the damping resistor and cannot be neglected, in the case of R

2 being equal to Z

C

the power loss can be estimated as in equation 2.4.2.3.8 [19]:

P output

_

filter

_

loss

=

2

4

V

DC

×

R

2

×

T on

_ min

×

f sw

(2.4.2.3.8)

Applying the above given theory to the specific case examined in this paper gives some guidelines for values and physical size of the filter components. As they should be applied to each phase conductor pf course three of each has to be included.

Assuming as before the maximum dV/dt for the motor is the common 5V/ns and in the data sheet for the IRG4PSH71UD it can be found that the typical dV/dt is 30 V/ns which is much too fast. As the peak voltage through the IGBT is 20.5A ( 2

×

14 .

5 ) the value of

C

1

can be calculated using 2.4.2.3.1, which gives the value 4.1nF. According to 2.4.2.3.7 the T on_min

is 350 ns which along with equation 2.4.2.3.6 gives the value for L

1 the characteristic impedance Z

C

27Ω which is also the minimum value of R

2

, actually R

2

, 3μH. As

is calculated as in equation 2.4.2.3.4 it receives the value

can be chosen a bit larger to suppress some overshoot but it should remain between Z

C

and 2

×

Z

C

. Suppose the

28

maximum current due to opposite diode reverse recovery is 20.5 A and the maximum current through the filter being according to 2.4.2.3.4 (if R

2

= Z

C

) 10 A. This along with the well known maximum current through and IGBT gives a maximum current of 51 A, this is what the over-current protection level should be set to. The power loss mainly dissipated in the resistor is calculated using equation 33 and is estimated to be 10 W. Yet again, this is for only one phase and the loss and weight of components has to be tripled to make up the actual figures. When of-the-shelf products are considered the weight estimated to be 60g for L

1

, 6g for C

1

and 7g for R

2

adding up to73g per phase.

Filter component and model

μH Pulse - nF RIFA

PHE845

Table 5 : Output filter components

29

The main cause of implementing snubber circuits is to make sure the switching power components remain within their safe-operating-area (SOA) both at turn-on and turn-off.

This is very important to ensure the longevity of the devices as well as reducing the amounts of EMI emitted from the high power components. When introducing a snubber it is not uncommon to be able to reduce switching losses in the switching devices, in particular the turn-on switching loss. However, in general the losses removed from the transistor are transferred to the snubber circuits instead while these components are usually less sensitive to voltage spikes and are able to perform satisfactory in a wider temperature span. As there are numerous variants of snubber circuits for different purposes only the ones estimated to prove suitable will be presented below along with their corresponding advantages and disadvantages. The simulations carried out are done using Cadence Pspice Student Edition, the circuits can be viewed in Appendix E.

Although the simulations will not be used very much for estimating good snubber circuits to obtain reduced power loss as much as ensuring longevity to the switching devices.

2.5.1. Increased SOA

Due to the non avoidable stray inductance in the DC-bus loop in addition to the internal stray inductances in most of the components causes along with the very fast current switching of modern semiconductors a large voltage over-shoot when switched off. To some extent there is an over-shoot during turn-on as well but not nearly as large. The faster switching and larger stray inductance the larger voltage over-shoot. The easiest and least complex method to manage this over-shoot is to compensate for the stray inductance in the DC-bus loop by connecting a small, fast and low inductance capacitor as close to the IGBT terminals as possible as shown in figure 20 (left). Actually increasing the gate resistor to slow down the switching process would work as well but switching losses would rise according to it.

Figure 20 : Decoupling and restricted decoupling capacitor

There are capacitors manufactured for this purpose with very low self inductance and designated connections to properly fit the concerned IGBT module. However when using discrete components this gets harder as the components are separated which also causes higher stray inductances. As the separation is one of the causes of choosing discrete

30

components the inductance is a large trade-off. The difference when using only the decoupling capacitor can be seen in the figure 23.

Figure 21 : Switching device voltage without snubber circuit

As seen in figure 22 the transient at turn-off is easily large enough to cause failure in the switching device, it is nearly twice as large as the DC-bus voltage. By attaching the decoupling capacitor the over-shoot decreases to about 123% of DC-bus voltage but instead an oscillating voltage will occur over the transistor as seen in figure 23. This oscillating voltage will inflict a large current through the capacitor which often is designed for small currents, with low self inductance as a benefit. The capacitor value can be estimated with equation 2.5.1.1 [20].

C sn

=

(

V pk

L s

×

I

0

2

V

DC

)

2

(2.5.1.1)

Where I

0

is the maximum switched current, L s

being the DC-loop inductance and V pk

the maximum allowed peak voltage over the switching device. This oscillating behavior is seen in 23.

31

Figure 22 : Device voltage with decoupling capacitor

This oscillating behavior causing high RMS current through the capacitor is the major disadvantage if the decoupling capacitor. The more inductive DC-loop and higher output current the higher RMS current will run through the decoupling capacitor. This makes this type of snubber circuit suitable for low current applications. The estimated power loss due to in the capacitor can be estimated with 2.5.1.2 [20].

P loss

_

decoupling

=

ESR

×

I

RMS

2

=

f sw

×

L

S

×

I

0

2

(2.5.1.2)

This would not result in a very high power loss but still it can be too high for these kinds of capacitors as they often are very delicate. The decoupling capacitor is a good solution when having a switching frequency of only a few kHz. The loss and oscillating voltage can be reduced by adding a diode and a resistor to restrict the de-charging and reduce oscillations. The circuit can be seen in figure 21 (right). Although to get a small enough over-shoot the capacitance needs to higher as the diode and the resistor introduce further inductance to the loop. The diode should be of fast and soft recovery type to avoid severe oscillations following V pk

2.5.1.3.

at turn off. The resistor can be calculated using equation

32

R sn

=

(

6

×

C sn

1

×

f sw

)

(2.5.1.3)

Figure 23 : Device voltage with discharge restricted decoupling capacitor

However, as this circuit adds more components and therefore more stray inductance a larger capacitor will be needed, although the power lost will mainly be dissipated in the resistor and the diode as explained in equation 2.5.1.4.

P loss

_

restriced

_

decoupling

=

1

2

×

C sn

×

(

V pk

2

V

DC

2

)

×

f sw

(2.5.1.4)

The RCD Clamp-Snubber seen in figure 24 (right) does have a very favorable effect on both the Turn-On and Turn-Off transients but have limited possibilities to reduce the switching loss in the transistor. However, the loss dissipated in the snubber circuits is much less than in the RCD Charge-Discharge Snubber (seen in figure 24, left) which is calculated as in equation 2.5.1.5 [20].

P loss

_

RCD

_

Ch

arg

e

_

Disch

arg

e

=

1

2

×

C sn

×

V pk

2

×

f sw

(2.5.1.5)

Although the RCD Charge-Discharge Snubber inflicts some power loss it can help to speed up both the Turn-On and in particular the Turn-Off processes directly reducing power lost in the IGBT and diode. But it can be difficult finding component capable of sustaining the internal loss while maintaining a low self inductance [20].

33

Figure 24 : RCD Charge-Discharge Snubber and RCD Clamp-Snubber

In figure 26 and 27 a switch has been simulated and plotted illustrating the difference between the RCD Charge-Discharge Snubber and the RCD Clamp Snubber. As it can bee seen they Turn-ON rather similar and cause about the same amount of over-shoot but the

RCD Charge-Discharge Snubber has a much quicker Turn-Off. As the Turn-Off with its tail is the main part of the switching loss the total loss can be vastly reduced using a RCD

Charge-Discharge Snubber even without reducing the gate resistor. Reducing the resistor would also reduce the losses but would do so by causing the over-shoot to increase once again.

Figure 25 : RCD Charge-Discharge Snubber

34

Figure 26 : RCD Clamp-Snubber

2.5.2. Reducing losses

As the snubbers circuits help to attenuate the voltage transients the switching process can be faster causing reduced switching losses by reducing the gate resistor. However, precaution has to be taken to ensure the driver can withstand this higher peak current.

Although not actually reducing the losses as much as moving them is done with some snubber as they consume then unwanted energy and dissipates it within its own components. If this is a diode or a resistor they can withstand a lot of power loss at a high temperature implying little need for excessive cooling.

Type Advantages

Decoupling

Capacitor

Low snubber losses

Directly and favorably effects Turn-

Off and Turn-On on voltage stress

Special solutions of casings for modules

Disadvantages

Produces voltage and current oscillations in the DC-bus, forcing usage of capacitor with high RMS current limit

Only a good choice in low current regions

Discharge restricted decoupling

Capacitor

Low snubber loss

Directly reduces the Turn-Off voltage over-shoots but also has as a favorable effect on Turn-On voltage transients.

Much quieter switching as diode blocks of oscillations.

Additional circuitry increase snubber inductance, making protection less effective.

Snappy diode could produce high recovery voltage spikes and dv/dts across IGBT pair.

Good in medium current ranges

RCD Very high snubber losses

35

Charge-

Discharge snubber circuit

Reduces Turn-Off voltage over-shoots Requires more components

Could substantially reduce Turn-Off loss in switching device

Clamp

Snubber

NO oscillations on the DC-bus

Directly reduces the Turn-Off voltage over-shoots but also has as a favorable

Difficult components selection

Good for high current and low DC-bus voltage applications

RCD Low snubber losses effect on Turn-On voltage transients.

NO oscillations on the DC-bus Requires more components

Good for medium to high current applications

Table 6 : Snubber characteristics

A quick glance gives the impression that the RCD Charge-Discharge Snubber would be a good choice for this inverter as the chosen transistor already has a very small gate resistor as default. In this way loss could be reduced without adventuring a fatal over-shoot.

2.6. Principles of cooling

2.6.1. Thermal resistance

A common design problem is to remove heat enough to guarantee satisfactory operation.

Then the thermal conductance between to adjacent mediums should be as high as possible to in the end dump as much heat as possible in a large buffer, usually the surrounding air.

The thermal resistance is as usual the inverse of the conductance and is as a normal resistance additive when connected in series respectively. An equation describing the temperature where the power is lost in comparison to the ambient temperature in respect to thermal resistance can be derived from Newton’s law of cooling and known relations.

Δ

Δ

Q t

=

k

×

A

Δ

×

x

Δ

T

k

Δ

Δ

Q

Δ

×

Δ

T t

A x

=

=

T

=

j

P heat

R

1

θ

ja

T a

P heat

=

T j

R

θ

ja

T a

(2.6.1.1)

The different components can be fragmented into many if there are several layers of different mediums between the source and the buffer. This makes it rather simple to estimate heat sink needs when most manufacturers state the estimated thermal resistance of the device. However, when having different cooling topologies a slight different approach has to be taken.

2.6.2. Heat transfer

When using a cooling bed with circulating air or fuel as cooling medium the case is somewhat different, a lower limit of coolant weight at a certain temperature has to be derived. The allowed rise in temperature is dependant on the following devices in the

36

chain and how high temperature they can withstand. A higher flow of coolant will cause it to warm up less in each device while a higher pressure would suppress this behavior as more coolant can absorb more heat. The energy delivered to the cooling bed can be calculated using equation 23 with the respective thermal resistance. This energy is to be removed by the coolant and depending on the specific heat capacity and the throughput of the coolant the cooling bed can be held at a reasonable temperature. Assuming it is known how many degrees the whole chain devices can raise the temperature and how many devices there are; an average per device of allowed temperature raise can be derived. This along with the total power loss needed to be removed and the nature of the coolant the needed throughput can be calculated as in equation 2.6.1.2:

Δ

Q

=

m

×

C

P

× Δ

T

m

=

C p

Δ

Q

× Δ

T

Δ

Q

Δ

t

=

P

m

× Δ

t

=

P

× Δ

T

(2.6.2.1)

C p

Where C p is the specific heat capacity, ΔQ is the energy differential and ΔT is the temperature differential. Suppose a 2°C raise in temperature is allowed and 200 watts needs to be removed from the cooling bed by air. Using the formula above gives a minimum flow of 100g/s. From consulting with the engineer Lars Austrin at SAAB AB it was given that the onboard cooler is slightly less efficient than a heat sink with forced convection at sea level.

The main purpose of the chassis usually is to provide a safe and protected environment for the usually sensitive equipment. The challenges are many and not always obvious, for instance the most common challenges are:

• Mechanical shock

• Electrical hazard both to components and to environment

• Dust and other small particles such as sand

• Humidity and liquids

• Electromagnetic environment

• Cosmic radiation

• Temperature

The mechanical shock is everything from a bump to the slightest vibration, both possible to cause stress resulting in either short-circuit or interrupted conduction. As it can be assumed that everything in an aircraft is very fixed the main issue is still vibration, the remedy can be to mount everything elastically or just the chassis it self. Easily understood it is more practical to only mount the chassis on suspensions but as it usually is much heavier than smaller components these can suffer from a vibration of higher frequency than the one filtered by the chassis. It is also of highly importance that the chassis does not submit to fatigue especially when the temperature can vary in a wide span and chemicals introduced in the environment possibly causing some material to decrease in tensile strength.

Different materials can be used with their own advantages and disadvantages such as polymers and metals, especially alloys of either aluminum or steel.

37

Polymers are usually very light and can be constructed in any imaginable form and shape, however, it is very likely to submit to fatigue either from unfavorable temperature, strong light or chemicals such as acids or different hydrocarbons. Metals and usually aluminum alloys are heavier and less moldable but are much more resistant to external influences and are much stronger. Within a great temperature span it is not affected, strong light can also be neglected but some chemicals can cause corrosion, at least for untreated nonprecious metals. A big advantage with a metallic chassis is it being thermal conductive all over. More characteristics for both metals and polymers will be explained further on.

Electrical hazard to components can be avoided by both protecting it from physical touch with an enclosure and providing a fixed support for the internal conducting parts making them firmly separated from each other. A chassis made of polymer is usually a very good insulator which makes it very unlikely to be exposed to electric hazard by touching the chassis in the case of an internal malfunction. In a fully conducting chassis a fatal accident to devices or humans can occur in the case of an error making the chassis electrifying. It also has some practical advantages due to its insulating nature when it comes to mounting conduction parts on the chassis making individual insulators unnecessary.

In the case of forced convection using a fan, dust and other particles can collect in the fan or in a filter reducing its capacity and even eventually causing a failure, not to mention the documented limited lifetime of a fan further increasing risk of failure.

Even in an external heat sink massive amount of dust can prevent natural airflow and radiation also reducing the capacity, even if the airflow is high enough this problem is unlikely to arise in either solution; however it is worth a thought. If a fully sealed enclosure with no external heat sink can be used this problem is eliminated. This sealed enclosure would further help reducing the amount of water and moist inside the chassis which in some cases can prove fatal to circuitry and even make metals to corrode possibly inflicting problems such as connection faults and reduced tensile strength.

A very common problem is when high power circuitry is mixed with logics. Due to inductive and capacitive load high energy electromagnetic waves at radiofrequency is radiated causing disturbances and noise in logic. In some extreme cases it can even destroy sensitive devices. The noise can be generated by the device itself or generated by other devices in the environment, in either way the susceptibility and emitted noise should be kept to a minimal. Except from considering the cabling a shielding cage can be made out of the chassis, it will work as the renowned Faraday cage and will keep noise generated inside in and noise generated on the outside out. A metallic chassis automatically inherits this benefit but a chassis made out of polymer has to have a film or fine-meshed grating on either the inside or outside.

At high altitudes the failure rate in power-devices increases rapidly due to cosmic radiation. Except from reducing the voltage over the device or the junction temperature not much can be made to reducing the influence of cosmic radiation except forming an

38

enclosure out of a thick heavy material such as lead or concrete. However, as this task is to optimize the power/weight-ratio this is a highly unsuitable solution.

2.7.1. Cooling scenarios

Besides the issues stated above the volume is of course critical but the main factor for choice is how to implement cooling depending on which environment and which cooling types that can be inherited from the surroundings.

In the case of an avionic application such as a fighter aircraft there are mainly three possible solutions for cooling and each have to be evaluated individually:

• Exterior heat sink integrated in chassis relying on a sufficient natural airflow around the device

• Interior heat sink relying on either a chassis mounted fan or airflow from external system

• Internal fluid cooling using coolant from external system

2.7.1.1. Exterior heatsink

To use a passively cooled heat sink to remove the heat is by far the heaviest and most space consuming solution but it is also completely independent of an external system, has no mechanical parts that wear out and cannot suffer from leakage or condense which is possible in fluid cooling. In an aircraft such as JAS 39 GRIPEN the temperature within the hull can easily reach 70°C and considering operating at 10000-15000 MASL

7

the convection contribution becomes less as the air becomes less dense. For instance from a properly designed heat sink relying on natural convection at sea-level about 70% of the heat is transferred by natural convection and 30% by radiation but at about 15000 MASL the ratio is about the opposite. This further increase the size required for the heat sink since the radiation part is not increasing, it is only the convection part that is decreasing.

This inverted ratio increases the overall thermal resistance to about 233% of the original since the convection part of the heat transfer reduces to about 18% of the original while the radiation part is constant [15].

In power equipment it is very common to make heat sinks form up the long side walls with high loss components internally mounted on them. Then the short side is made of sheet-metal making a suitable place for mounting indicators, connectors, switches or even a fan. In the case of a fan an inlet or an outlet has to be made on the opposite side to permit airflow, usually done by perforating the sheet.

The power-components should be spread evenly over the heat sink to reach the best performance in respect to cooling. As the heat sinks usually are very sturdy, the other sides can be made out of thinner material, reducing the weight and this principle is illustrated in the sketch given in figure 28. A more detailed sketch with components fitted can be viewed in appendix F.

7

Meters Above Sea Level

39

Figure 27 : Scetch of chassis

The weight and volume for an enclosure of this solution is highly dependant on the actual heat sink whose figures are estimated from the dissipated energy and the different thermal resistance of the different components as illustrated in figure 26. However this is only for the top-side transistors since the two sides will be mounted on separate heat sinks along with one out of two rectifiers. The transistors used in the calculations for thermal resistivity are the IRG4PSH71UD with a built-in diode, the rectifier is from Semikron and the thermal resistance values are given in their datasheets.

Figure 28 : Thermal resistivity estimation

R

θ

TOT

=

R

θ

sa

1

//

R

θ

sa

3

//

R

θ

sa

5

//

R

θ

sarect

0 .

39

8

8

“//” is equal to parallel coupling

40

(2.7.1.1.1)

The internal thermal resistances can be found in the datasheets while the thermal resistance of the heat sink has to be calculated for each one of the components and then joined using a parallel coupling as seen in 2.7.1.1.1. The temperatures given where the ambient temperature T a

= 70°C and the maximum junction temperature T j

= 125°C. This way of calculating is done because some of the components have different thermal resistance and dissipated energy and then requires different cooling than the next one.

The amount of cooling and space on the heat sink for each component will be determined the easiest way with their specific thermal conductivity

9

required from the heat sink, for instance, the rectifier requires a thermal conductivity of about 0.59 which is about 23% of the total 2.63 R th

-1

. The layout of the heat sink used in the sketch of the chassis, placing of the components and the actual distribution of the cooling area on the heat sink can be viewed in figure 27.

Figure 29 : Heatsink layout

The heat sinks used in the model illustrated in figure 27 is from H S Marston, it measures

150x200x40mm and is optimized for natural convection having a thermal resistance of

0.4°C/W which is sufficient. However, this is at sea level only; an increase in altitude will increase the thermal resistance significantly as explained before. The weight of 1.44 kg for each side gives a rough estimation of the weight of the chassis since the steel sheets forming the rest of the enclosure will weight much less.

Using the heat sinks as two sturdy walls and steel sheets of 250x150x1mm

10

as the other walls an internal volume of 7.5dm

3

is reached, estimated to fit all the devices, circuitry, filters and the capacitor-bank. The bottom and top plate can be made up from similar steel sheets but the bottom should be slightly thicker to provide better support, thus doubling the thickness. A 250x200x2mm and a 250x200x1mm along with the other walls

9

The conductivity is the inverse of the resistivity

10

Width x height x thickness,

41

made of aluminum weights about 610g which when added with the heat sink the whole enclosure weights 3.5kg.

2.7.1.2. Interior heat sink

When mounting the heat sink inside the chassis the air circulation would be very limited if not increased via an external source, the ambient temperature in the chassis along with the junction temperature would probably rise to a critical level and the system would eventually shut-down or burn. However even with a small fan airflow can be produced through the interior raising the performance of the heat sink, often many times over as illustrated in the figure below.

Figure 30: Thermal resistance factor vs. airflow

Where F is a factor of which the thermal resistance of a heat sink is reduced depending on the airflow across the heat sink. The curve resulting in a horizontal line is mostly due to the fact that airflow does not increase the radiated heat.

This would allow a reduction in weight due to removal of excess heat sink at the cost of a few watts of lost effect for the fan and the possibility of fan-breakdown, causing a fatal temperature build-up. The fan would at best push in air at the temperature up to 70°C and the slightly warmer air coming out would blend with the ambient air and if the compartment is not too small it would not raise the environment temperature significantly. The increased internal temperature has to be included in the calculations where size of the heat sink is determined. This solution would also suffer in cooling capacity due to a decreasing air density but it can at least be reduced by serial-coupling many fans which can create a strong pressure in the chassis by having a smaller outlet.

However it would not be nearly enough to compensate for the loss in pressure at higher altitudes since even if the convection could be doubled or tripled it initially represents such a small share of the total the reduction in thermal resistivity would be insignificant.

Most aircrafts and particularly JAS 39 GRIPEN have equipment for providing pressurized air at a few degrees above 0°C and this can strongly reduce the need for a large heat sink at the cost of a less stand-alone device. Since the air can contain moist and particles the air cannot be sprayed directly on a heat sink or the component but is meant to be ran through a cooling block similar to the one of conventional fluid cooling; although not nearly as effective due to the difference in specific heat capacity. Here a failure in the cooling system would inevitable lead to a failure in the inverter as well as all other equipment depending on the same cooling system. It has to be considered here

42

whether a loss or a gain in weight is achieved depending on the efficiency of the cooler, if it requires more weight and power to produce this very cooled pressurized air than required by a passive heat sink a total gain in weight is achieved which was not the final task.

2.7.1.3. Internal fluid cooling

A third way of cooling in an aircraft such as the JAS 39 GRIPEN is to use its fuel as a coolant before it is combusted. The fuel is pumped around and reaches a temperature of maximum 100°C keeping a cooling bed at the same temperature which is equivalent to a very large heat sink with the same temperature.

This system is also very dependant on the external cooling system and will fail or become able to only supply as much smaller load in case of a failure in the external cooling system. The main advantage here is the reduction in weight but also no mechanical parts.

Designing the chassis also require less consideration since no wind-tunnels has to be made. In the case of using this system each heat sink in the sketch is replaced with a cooling bed of similar size although very likely to significantly remove weight of the unit.

The design of the transistor drive circuit is another subsystem which deserves attention as a thoroughly designed driver along with snubber circuits can reduce switching losses extensively. However, not only the losses but all switching behavior can be altered and this is wanted as there is no universal perfect switching. All applications require individual design, for example a highly inductive load such as a motor will cause a large voltage over-shoot if the switching is too fast. This over-shoot is very unwanted as it may bring the IGBT outside its safe-operating-area (SOA).

Figure 31 : Descriptive driver internal circuit (IGBT within dashed line)

The main factions to alter for a satisfying behavior are the positive bias voltage, the negative bias voltage, or the gate resistor. However, the gate resistor can be in parallel with a diode and a resistor in series as this will change the turn-on and turn-off behavior independently.

The +V

GE

is often set on a level as close to the maximum gate voltage as possible with regard to tolerance which can be as much as 10% in some cases. This because a high gate voltage during the ON-state minimizes the ON-resistance (or ON-state V

CE

saturation

43

voltage) while a too high gate voltage may permanently damage the device. Respectively a low -V

GE

will increase the OFF-resistance causing less loss during the OFF-state but will increase chance of shoot through currents if too small [7].

Setting +V

GE

voltages high (and -V

GE

low) while keeping the gate resistor value constant will result in a low switching time and thus low switching loss. However, a faster behavior will as stated before cause unwanted surge currents and voltage spikes over the devices possibly destroying the device.

A rule of thumb is to set the +V

GE

and –V

GE

values to 15 and -5 respectively while mainly altering the gate resistor to achieve a satisfactory switching behavior. Below is a table describing common transistor behavior when changing driver parameters.

Main characteristics +V

GE

rise -V

GE

rise RG rise

V

CE(sat) t on

E on

Fall - Rise t off

E on

Turn-ON surge voltage Rise - Fall

Turn-OFF surge voltage - dV/dT malfunction Rise

Rise

Fall

Fall

Fall

Current limit value

Short circuit withstand capability

Rise

Radiational EMI noise Rise

-

Fall - Rise

-

Fall

Fall

Table 7 : switching characteristics vs. driver dimensions

Using the values recommended by the rule of thumb the maximum current through the gate resistor can be estimated and then the proper driver circuit can be chosen as well as a sufficient power supply. The gate resistor value has been chosen as the default value given in the datasheet for the transistor irg4psh71u from International Rectifier, with this value the switching loss has also been estimated.

I

G

_

peak

=

+

V

GE

R

G

+

+

V

GE

R g

=

15

+

5

5

= 4 A (2.8.1)

The R g

is an internal gate resistor common in larger transistor modules but for the models represented in this thesis this resistance is negligible.

As well as the peak current a sufficient continuous current will be required from the driver and the power supply which can be calculated as in equation 2.8.2.

I

G

=

f sw

×

(

Q g

+

C ies

× −

V

GE

)

=

10000

×

(

570

nC

+

6200

pF

×

5

)

=

0.006 A (2.8.2)

44

Where f sw

is the carrier frequency, Q g

is the gate charge from 0V to +V

GE

and C ies

is the input capacitance. As these currents are per driver and each transistor has its own driver the average current supplied by the power supply has to be at least six times the above stated value. The high peak current can be managed by a buffer capacitor but then the power supply has to be slightly bigger. Without the buffer capacitor the power supply has to be able to provide the peak current which will give a vast loss and a heavy device.

If all the power losses are completely consumed by the gate resistor then the power required from the driver is shown in equation 2.8.3.

P g

_

loss

(

on

)

=

f sw

×

1

2

Q g

+

V

GE

+

1

2

C ies

× −

V

GE

2

=

⎛ 570

nC

×

15

2

+

6200

pF

×

2

25 ⎞

=

=0.0435 W (2.8.3)

Since the gate turn-ON turn is as large as the turn-OFF loss the total loss is simply the double turn-ON loss as in equation 2.8.4.

P g

=

f sw

×

(

Q g

+

V

GE

+

C ies

× −

V

GE

2

)

=

10000

×

(

570

nC

_

loss

×

15

+

6200

pF

×

25

)

=

0.087 W (2.8.4)

Selecting the Half Bridge Gate Driver IR2114SSPbF from International Rectifier easily match the driving needs for each half bridge with the least amount of external components.

The internal maximum loss of this driver is about 1.5 W each which adds up to the dynamic loss as explained earlier. To provide the best performance both positive and negative power supply is recommended, although it’s not crucial.

Driver component model (mm)

Buffer capacitor

43 nF Jamicon 3*2 3*(5x11) -

Resistor 5 Phoenix 6*0.1 6*0.087

Table 8 : Driver components

To have a versatile solution being able to provide the controlling system with power regardless of whether the AC-source or DC-source is used a high voltage step-down converter is necessary. It has to be connected to the DC-bus and output a double voltage capable of supplying both the controller and the driver. A fly-back solution from

Fairchild has proven to be an excellent choice since with a high voltage IGBT transistor instead of the common MOSFET it is capable of handling the voltage spikes common in fly-back circuits [17]. This along with a transformer with double secondary windings two different voltages with different polarity is achieved.

45

Input voltage: < 707V

Efficiency: ~80%

Output voltage:

-5V,+15V

±5%

Maximum output power: <25W

Weight: ~70g

Size: ~90x50x30mm

Table 9 : Power supply specifications and dimensions

The physical figures are estimated from the application note [17] using the same transformer core and the same transistor as used in the example in the same application note. However a few components such as rectifier and smoothing capacitor can be removed since it is fed with a direct current.

2.10. Controller

The controller is logic device responsible for handling all signals and take appropriate action to ensure operation. It handles all fault signals and causes a halt if a serious problem arises. It also observes the actual voltage levels and steers it towards the proper level. The controller is of the less crucial components when it comes to power losses or weight and thus it is not examined in detail. Still, a suitable off-the-shelf component has to chosen to have some guidelines. The IRMCK203 from International Rectifier is a fully capable controller requiring minimal external circuits. However since it requires a 3.3V power supply a fixed regulator has to be used to decrease the voltage from the power supply.

Since the lost power due to the fixed regulator is dependant on the voltage drop and the current through it the lowest voltage from the power supply should be used.

The above chosen controller dissipates 1.2 W @3.3V typically according to the data sheet which gives a current of 400 mA. The combined loss of the controller and the regular are thus 2 W

(

400

mA

×

(

1 .

7

V

+

3 .

3

V

) )

. The space and weight consumed by the controller is not of vast proportions and could easily be shared with the power supply and the driver on the same circuit board.

Controller compoenents

Controller - and model (g)

Dimensions

(mm)

Loss

(w)

Rectifier

IRMCK203

Voltage regulator

3.3 V National

Semiconductor

LP3852

9 10x21x5 0.8

Buffer capacitor

25 V

10 μF

-

Table 10 : Controller components

46

3. Losses

In this part mostly the components responsible for the main part of the loss are considered, those are the main supply rectifier and the transistor-bridge. To some extent the input filter and the capacitor bank will also add up to the losses. Mainly two types of transistors have been considered, the IGBT and the MOSFET. The switching loss in the rectifier is in this case ignored due to the low frequency and the low switching energies in the diodes, which leave only the forward conduction loss remaining.

However in the IGBT-bridge the largest part of the loss is due to the high switching frequency in the slow IGBT-transistors, a minor loss also arise due to forward voltage drop. In the transistor-bridge the free wheeling diode is usually included further contributing to power loss but in the cases of missing a built-in diode the IGBT-bridge and a suitable diode will be evaluated individually. There will also be some loss in the filter capacitances and the snubber circuit depending on which type chosen and the severity of the compensation. Some minor power losses will occur in a switched voltage supply and the control circuit but it is almost insignificant in amount and hard to optimize.

In appendix D a spreadsheet has been formed over suitable transistors to ease the choosing of the most suitable transistor, different technologies such as IGBT and

MOSFET has been evaluated as have different types of semi conducting materials. In the spreadsheet there is a majority of IGBT before MOSFET, this is due to the difficulty finding a MOSFET with a V

DS

rating of about 1200V, this is the recommended voltage when having a line voltage of 400V due to large voltage peaks. We were able to find one

MOSFET with a V

DS

of 800V, however voltage peaks are likely to destroy this device and the fact that is has a very large R

DSon energy makes this device not suitable.

that generates a vast amount of dissipated

3.1.1. Transistor

The switching losses can be derived from the switching energies found in datasheets but usually requires some conversion since the rated currents and bus voltages might differ from the application specific currents and voltages. The conversion can be done for the current using equation 3.1.1.1 and 3.1.1.2 [7]:

E on

=

E on

'

(

I ave

/

ratedI ave

)

α

(3.1.1.1)

E off

=

E off

'

(

I ave

/

ratedI ave

)

β

(3.1.1.2)

47

Where α and β are multipliers depending on the device and E on’

and E off’

are the energies at the rated average current which is calculated according to equation 3.1.1.3.

I ave

=

2

π

2

×

I

0

(3.1.1.3)

The same should be done analogous with the voltage since it usually differs as well.

• Turn-On loss (P on

)

P on

=

f c

E on

(

I ave

) (3.1.1.4)

Where f c

is the switching frequency and E current I ave

[7]. on

(I ave

) is the Turn-On energy at the average

• Turn-Off loss (P off

)

P off

=

f c

E off

(

I ave

) (3.1.1.5)

Where fc is the switching frequency and E off average current I ave

[7].

(I ave

) is the Turn-Off energy at the

3.1.2. Diode

FWDrr

=

E rr

(

I ave

)

×

f c

2

(3.1.2.1)

E rr

=

E rr

'

(

I ave

/

ratedI ave

)

χ

(3.1.2.2)

I ave

=

2

π

2

×

I

0

(3.1.2.3)

E rr

is the reverse recovery energy per switch which with the switching frequency f c

gives the switching loss of the diode. Usually all characteristic values for the diode can be found in the datasheet; however it may require some interpolation between values in graphs to determine the appropriate value for the specific application, such as estimated junction temperature.

It is very common for an IGBT or a power MOSFET to be co-packed with a silicon FWD which gives some benefits such as shared cooling and less wiring reducing capacitive and inductive noise but it also makes it slightly more difficult to experiment with other types of diodes. A few IGBTs suitable for this application are available with and without a

FWD making interesting experiments possible, especially to use a SiC FWD to optimize switching loss.

48

3.2.1. Transistor

The losses in the IGBT can be approximated using the information supplied in the datasheet and some knowledge of the application specifics such as switching frequency and the power factor of the load [13].

• The ON-state power loss in an IGBT transistor can be calculated as in equation

54:

P sat

=

2

I

0

2

π

V

0

+

π

4

×

R

×

2

×

I

0

+

π

4

× cos

φ

×

⎜⎜

V

0

+

8

R

3

π

×

2

×

I

0

⎟⎟

(3.2.1.1)

Where cos

φ is the power factor of the load. I phase output current and both V

0

0

is the root mean square value (RMS) of the

and R can be derived from the following picture illustrating the output characteristics of an IGBT [13].

Figure 32 : Approximate output characteristics

• Total power loss (P losstot

)

The total power dissipation can be calculated as in equation 3.2.1.2 [13]:

P losstot

=

P on

+

P off

+

P sat

(3.2.1.2)

3.2.2. Diode

The total loss and dissipated heat in the diode is dependent on many factors such as switching frequency, characteristics of the diode and how inductive the load is. An easy approximate model for power loss calculations is described in equation 3.2.2.1 [13]:

FWDsat

=

2

I

0

2

π

V

0

+

π

4

×

R

×

2

×

I

0

π

4

× cos

φ

×

⎜⎜

V

8

R

3

π

×

2

×

I

0

⎟⎟

(3.2.2.1)

Where cos

φ is the power factor, I

0 is the forward RMS current, R and V

0

form up the voltage drop of the diode in the same manner as for the IGBT but probably with slightly different values. Usually all characteristic values for the diode can be found in the

49

datasheet; however it may require some interpolation between values in graphs to determine the appropriate value for the specific application, such as estimated junction temperature.

• Total power loss in the FWD (P losstot

)

The total power dissipation can be calculated as in equation 3.2.2.2 [13]:

P losstot

=

P on

+

P off

+

P sat

(3.2.2.2)

In appendix D it can be found a MOSFET in comparison with various IGBT with almost the same voltages and currents and as predicted the power loss in the MOSFET is greatly exceeding any IGBT regardless of model. This is easily explained by the high voltage ratings.

On the other hand, in other applications in example as an inverter with a DC-bus supplied from a regular car battery (12V) which is consuming 50A an off-the-shelf IGBT with

V

CEsat

= 2.5V has the efficiency of 79%. An of-the-shelf MOSFET with R

DS(on)

= 0.007 Ω has an efficiency of 97% and if we add switching losses the benefit of the MOSFET is increased further more. The conclusion looks like choosing the IGBT with the lowest power losses are the best choice without a doubt but there are other concerns as well.

By viewing the chart of suitable transistors/diodes again and removing all those which may have a problem with either too high peak currents or too high peak voltages

11 a few good transistor solutions remain, in this case with respect to power loss per IGBT with

FWD. Calculations are done using the equation 55 and 57 with the parameters cos

φ=1,

I

0

=15 A and the model specific parameters can be found in appendix D where all the parameters have been normalized to fit the actual values. The switching frequency is assumed to be 10kHz giving it a m f

= 15.

Manufacturer Model P

LOSSTOT

12

Technology

R

θ

JC

+

R

θ

CS

SI IGBT, 2 x SiC FWD 0.36+0.24,1.1+0.24

CREE

C2D05120

2 x CID100512 40W 2 x SI IGBT, 2 x SiC FWD 1.15+0.24

Table 11 : Suitable discrete IGBTs

Although even if the discrete transistors CO-packed with free-wheeling-diodes above seems to outperform the modules below in respect to loss the cabling has to be considered since it will contribute with some capacitance and possibly some inductance. In some cases the effects of this is large enough to be ineligible in favor of the big modules.

The discrete transistors also have the possibility to be mounted on a separate heatsink or to displace the dissipated heat over a larger area making cooling easier in some

11

12

The rated voltage and current should be at least twice the nominal to handle overshoots [7].

Conduction loss and switching loss for one IGBT with FWD

50

applications. However, the modules are much easier to handle, one big bulky brick with optimized internal wiring prepared for mounting a PCB on top along with built-in brake

IGBT and a high performing rectifier-bridge results in a lot less wiring and complexity.

On the other hand a module requires a larger heatsink or equal device capable of removing a lot of heat from a relatively small area.

Manufacturer Model P

LOSSTOT

4

Technology

R

θ

JC

+

R

θ

CS

Infineon

FUJI

Infineon

FS25R12W1T4

29,6W SI IGBT, SI FWD

7MBR35UB120

13

FP25R12KT3

5

33W SI

30,7W SI IGBT, SI FWD

0.66 + 0.88

(0.8,1.35) + 0.3

Table 12: Suitable IGBT modules

In the end a more detailed approach than only the power loss has to be considered for choosing the most suitable solution, mostly depending on the choice of cooling but also on how much the transistor-performance can be increased with the help of snubber circuits.

3.4. Rectifier

The total conduction loss can be derived from the average current running through a leg times the forward voltage drop of two diodes since the current always runs through two.

This voltage drop is somewhat depending in some grade on the current and the temperature of the junction and the proper value can be found in the datasheet. The average current is simply the output current divided by three, notice that this value will not set the voltage drop since there is momentarily a higher current running through the diode when actually conducting, they are just not conducting more than a half period at a time.

P

LOSSCOND

=

2

×

V

SAT

×

I avg

×

3 (3.4.1)

The “2” since the current is running through two diodes and the “3” since there are three legs in the rectifier bridge as illustrated in figure 13. Mainly three rectifiers from three different suppliers have been considered, FUJI, International Rectifier and Semikron.

They have all been chosen from a wide range of models and only the best suiting rectifier for this application has been considered. Their main properties are described in the table below.

13

The module is equipped with a built-in rectifier and brake function

51

Manufacturer Model

Fuji

IRF

Semikron

V

@14,5A

SAT

125° C

6RI30E 0,87

36MT80 0,9

SKD 33 1,21

P

LOSSTOT

35,7 W

36,9 W

49,6 W

Weight Size(packet)

100g

20g

30g

52 x 22 x 26

64 x 30 x 21

R

θ

JC

+

0.417

R

Table 13 : Suitable rectifiers

As seen in the table above a tradeoff has to be made, the dissipated energy is lower in some of the cases but it is also harder to remove the heat due to the larger thermal resistance. For instance the dissipated energy in the Semikron diode bridge is approximately 39% higher compared to the Fuji, however the Semikron has only about

46% of the thermal resistance making it a lot easier to cool. If an external efficient cooling system is available then the Fuji would be the best choice to maximize efficiency but with worse conditions for cooling the Semikron would be the best choice neglecting the slightly lower efficiency.

θ

CS

0.80+ 0.10

29 x 26 x 21 1.16 + 0.2

3.5. Filters

3.5.1. Input filter

As the current and voltage harmonics will not change in frequency with increasing rated power of the inverter the components values can be scaled proportional as long as the main voltage is considered fixed. Along with this proportional increase comes the power loss. The factor that would have largest effect on the input filter components and loss would be to change rectifier complexity. Stepping up from the present 6-pulse rectifier bridge to at least a 12-pulse bridge would reduce initial harmonics significantly and therefore almost completely removing the need for additional filter except for EMI.

However the EMI filter does barely represent any significant loss except in some additional weight.

However if the 10kW input to a 6-pulse rectifier needs to filtered to an acceptable level including a simple EMI filter it requires filters worth of 131g and 37 W of loss per phase, which is not negligible. It adds up to a total of 393g and 111W.

3.5.2. DC-bus filter

As the losses already have been derived and explained some estimated values for different output capacities and switching frequencies has been collected. A 3% ripple voltage of the maximum DC-bus voltage is allowed in all scenarios.

Output power/switching frequency @ 230/400VAC

6kHz 10kHz 14kHz

Table 14 : DC-bus filter losses when using 230/400 VAC input

52

Output power/switching frequency @ 115/200VAC

6kHz 10kHz 14kHz

Table 15 : DC-bus filter losses when using 115/200 VAC input

The loss due to the surge limiting circuit is not included since it is only active during start-up. A 12-pulse rectifying bridge would also here prove to be favorable as it decreases the voltage ripple significantly doubling its frequency, relieving the capacitor from much ripple currents which will require a smaller capacitor bank.

3.5.3. Output filter

The power loss in the output filter is depending on several factors where the largest is the

DC-bus voltage V

DC

, f sw

, I peak

and the maximum allowable dV/dt in the motor which can be calculated using the simplified equation 3.5.3.1 from chapter 2.4.3.

P output

_ _

=

V

DC

2

×

I peak

×

f sw

×

π

(3.5.3.1)

filter loss

4

×

dV dt

Considering the maximum dV/dt for the motor rather fixed would make this model only consist of fundamental parameters and straight to calculate.

Using the parameters for the model examined in this paper a power loss per phase is estimated to be 10 W, giving a total of 30 W.

3.6. Controller

The controller is responsible for an insignificant power loss in comparison to the high power components. But as it is trivial to estimate the maximum loss there are no real reason to neglect it. This loss can also be considered fixed regardless of the output power or the switching frequency as it mainly depends on the internal clock frequency.

As it can be found in the datasheet the maximum loss is [email protected] but as explained before a voltage regulator is needed, further contributing to an additional 0.8 W loss.

The power loss in the transistor drives consists of one rather fixed component and one minor depending on the switching frequency. However if the transistor configuration needs to be changed due to higher power it would alter the dynamic power loss as well.

Suppose the same transistor would be used in parallel to obtain higher power rating then the dynamic power loss would increase with a factor corresponding to the number of parallel transistors. Otherwise it would increase proportional to the switching frequency as explained in chapter 2.7.

53

The 10kW inverter examined in this paper would accumulate about 5W of power loss due to the three drives ( 3

×

1 .

5 W) and the dynamic 0.5 W assuming the switching frequency is 10 kHz. Even if the switching frequency would be changed to 6 kHz or 14 kHz it would not change the total significantly, (4.8 W, 5 W, 5.2 W).

A parallel coupled transistor would double this loss as well as requiring a stronger driver.

The power loss of the internal power supply is dependant on the devices it provides with power as it usually has a rather fix percentage of loss. As the transistor drives have a high peak current while low RMS current a buffering capacitor is favorable to use instead of over sizing the power supply. Since the efficiency in the power supply is estimated to

80% the other 20% of the input power is pure loss dissipated in the supply.

As stated in chapter 3.7 and 3.8 the power loss from the controller and drives sums up to

7 W giving the power supply a loss of 1.75 W adding up to the total of 8.75 W put in to the power supply.

54

4. Results

The power density is defined as the ratio between the rated output power and the weight of the device, the higher the better. The power density is very much related to the efficiency of the device as thoroughly explained in this report. The weight of three different power levels, 2000W, 10kW and 20kW will be estimated using the same method as used in the paper for 10kW. Two different input voltages are used, the 400V three phase input used in this paper and a 200V three phase input which is common in aircrafts. The weight is divided in the respective segment of the inverter to pinpoint the specific weight. As can be seen some parts grow with rated output while some parts remain constant. The switching frequency is considered constant at 10kHz for this evaluation.

Weight (g) @ 230/400VAC / 540VDC

Rated effect/component 2000W 10kW 20kW

Input filter 72 393 720

Dc-bus filter w. surge limiter 240

Power supply 70

Controller and driver

IGBT and FWD

60

54

Output filter

Internal wiring

39

130

780

70

60

54

219

240

1872

70

60

54

403

370

Table 16 : Approximate separated weight for 230/400VAC applications

Weight (g) @ 115/200VAC / 270VDC

Rated effect/component 2000W 10kW 20kW

Input filter 98 480 1230

Dc-bus filter w. surge limiter 90

Power supply 70

Controller and driver

IGBT and FWD

60

54

Output filter

Internal wiring

53

173

130

70

60

54

365

360

585

70

60

54

623

470

Table 17 : Approximate separated weight for 115/200VAC applications

55

4.2. Scalability

The ability and simplicity to up-scale or down-scale a solution to fit the actual needs without any unnecessary weight is the definition of scalability. Of course some weight is added to but it is far from a linear relation. Some parts will remain fixed when adding or removing output effect while some will be abundant.

From the tables in chapter 4.3. a mathematical approximation can be derived to estimate the loss of an arbitrary converter with some basic input characteristics. The inputs would be power in watts, switching frequency in hertz and the phase to phase voltage on the input. The estimation has a maximum error according to the table of 7% and should be precise enough to cover at least 20% outside the regions studied.

P loss

(

P in

,

V

ACin

,

f sw

)

=

⎜⎜

400

V

ACin

⎟⎟

0 .

59

(

121

+

0 .

0034

(

f sw

6000

)

1 .

15

+

0 .

0084

(

P in

2000

)

1 .

1515

)

(4.2.1)

The same approximation can be done for the estimated mass using the tables in chapter

4.1. Input parameters are the same and the result is given in grams.

Mass

(

P in

,

V

ACin

,

f sw

)

=

⎜⎜

400

V

ACin

⎟⎟

0 .

0 .

24

(

2290

+

0 .

27

(

f sw

6000

)

0 .

772 +

0 .

305

(

P in

2000

)

1 .

059

)

(4.2.2)

The method used to fit the equations to the values is to assume the relation is exponential and then start from the lowest value. In this scenario the 200 VAC, 2 kW and 6 KHz version is used as base. The increase in weight and loss is averaged on the values considered as fixed when the value at hand is growing, in this case in three steps. Doing this for all three input values and adding their components makes up (4.2.1) and (4.2.2).

4.3. Efficiency

The estimated amount of loss compared to the rated output power. As can be seen in the following tables a lot of combinations are given to estimate the power loss for corresponding combination. For the lower voltage case the lower voltage version of the transistor and FWD are used as they both have one. For the lower and higher effects similar components but with different power ratings are taken of-the-shelf usually from the same series and manufacturer as the components in the example.

Rated effect/component

@ 6kHz

230/400VAC / 540VDC

2kW 10kW 20kW

115/200VAC / 270VDC

2kW 10kW 20kW

23 111 220 37 170 460

Rectifier 7 45 85

Dc-bus filter w. surge limiter 2.6 27 54

14 89.6

5 42 80

1.5 1.5 1.5 1.5 1.5 1.5

2+4.8 2+4.8 2+4.8 2+4.8 2+4.8 2+4.8

56

Snubber circuits

Sum of loss:

36 109 278 43 172 363

4 21 42 7 39 67

0.05 0.5

81 322

1.476

689

0.074 0.74 2.21

114.4 521 1165

Table 18 : Approximate separated losses @ 6 kHz switching frequency

Rated effect/component

@ 10kHz

230/400VAC / 540VDC

2kW 10kW 20kW

115/200VAC / 270VDC

2kW 10kW 20kW

23 111 220 37 170 460

Rectifier 7 45 85

Dc-bus filter w. surge limiter 3.2 33 67

14 89.6

6.3 52 100

Controller driver

IGBT and FWD

1.75 1.75 1.75 1.75 1.75 1.75

2+5 2+5 2+5 2+5 2+5 2+5

44.6 145 355 59 188 397

5 30 56 9 49 76

0.083 0.123 1.23 3.68

92 373 794 134 558 1230.8

Table 19 : Approximate separated losses @ 10 kHz switching frequency

Rated effect/component

@ 14kHz

230/400VAC / 540VDC

2kW 10kW 20kW

115/200VAC / 270VDC

2kW 10kW 20kW

Rectifier

23 111 220 37 170 460

7 45 85 14 89.6

Dc-bus filter w. surge limiter 3.8 41 81 8.2 68 120

1.85 1.85 1.85 1.85 1.85 1.85

Snubber circuits

Sum of loss :

2+5.3 2+5.3 2+5.3 2+5.3 2+5.3 2+5.3

53 175 433 74 204 431

8 47 93 12 67 107

0.117 1.167 3.44 0.173 1.73 5.16

104 429 924.6 154.52 609.5 1317.3

Table 20 : Approximate separated losses @14 kHz switching frequency

4.4. Reliability

The reliability is dependant on margins and precautions taken to ensure the longevity of the device. Functions implemented to protect the device during errors or unwanted conditions are a necessary factor to prevent permanent damage to the device. Also how effective the components are cooled will result in a reliability factor as higher temperature reduces lifetime in most semiconductors and especially electrolyte capacitors. It may be needed to pinpoint every component that is subject to high power loss and evaluate which temperature it will reach and how it will affect its lifetime.

Protective measures such as snubber circuits which will reduce voltage transients over already very warm components can improve lifetime significantly as heat, junction voltage and cosmic radiation are the main factors of reducing semiconductor lifetime

[21].

57

4.5. Simulations

Some of the simulations in this master-thesis are simply to illustrate the qualitative behavior of some circuits while some are just to measure noise and disturbances. A few

Simulations has also been done to verify the calculated values of some components. For the snubber considerations only Pspice have been used to illustrate the effects a snubber circuits can inflict on the voltage transients. Any tries to reduce power loss has been disregarded as it is very time consuming to find components with values that actually can be realized.

For component value verification Matlab and Simulink have been used, the layout of the system can be viewed in appendix B. A fully working system using the surge current limiter can be found in appendix A1 where the graphs shows the growing voltage of the

DC-bus, the switched voltage from an IGBT couple and the same low-pass filtered voltage. Under the same circumstances the currents running through the diode bridge and the transistor bridge have been measured and the results are presented in appendix A2.

First is an averaged form of the following diode currents. The average current seen in the first graph eventually corresponds to the steady-state value as expected. However the instantaneous current seen in the second graph is of larger interest as in the case of no surge current limiter used these currents as seen in the top graph in appendix A4 rise to a dangerous level. The current running through the IGBT bridge can be seen being regardless of using the surge current limiter or not as seen in the bottom graph in appendix A3 and A4.

In appendix C1 the frequency spectrum at the input terminal has been measured using the model set up as in appendix B. The voltage spectrum consists of the fundamental frequency and its harmonics without any filtering. In appendix C2 the current harmonics have been measured. The very same voltage measurement has been done in appendix C3 but this time it is filtered. The most disturbing harmonics as well as the total harmonic distortion has been reduced significantly. In appendix C4 the same voltage measurement has been carried out but this time a 12-pulse rectifier is used. In addition the current harmonics have been measured in appendix C5. As can be sent the harmonics are much higher in frequency but much lower in magnitude making it much easier to filter if needed, in this scenario it is not.

58

5. Conclusions

The most obvious conclusion from performing this master-thesis is that the selection of switching transistors should be given some time as the amount of power loss can be reduced significantly with a delicate choice of components. However, as seen in this thesis there are other parts of the converter contributing to a vast amounts of loss and in particular weight not foreseen in the beginning of the thesis. In the converter studied in this master-thesis the filters add up to almost 25% of the total weight, although in some other configurations it becomes more or less prominent. The power lost in the filters is in the same size as in the switching devices making them worth further more attention.

Altogether it would make good choice to address the filters the same detailed approach as the switching devices.

The choice of using a SiC FWD ended in slightly less calculated loss in the switching devise due to the negligible reverse recovery energy; however this should be simulated extensively under appropriate conditions to be verified. Although uncertain loss it can easily be said that the SiC technology requires less cooling as it can work under very high temperatures. However using a FWD not integrated in the package introduces additional capacitance and inductance possibly slows down the switching process canceling the reduction gained. In addition the need for snubbers circuits increase when having higher stray inductance.

As it turns out, using cooling relying on passive convection only in an aircraft a high altitude would be too inefficient and the built-in cooling system would be the only choice; regardless of which coolant used. In an aircraft flying on low altitude on the other hand the high velocity winds could be used to cool the equipment.

59

6. Future work

Several of the solutions explained in this master-thesis is today obsolete and can be replaced with better performing and more sophisticated solutions. However most of these solutions is by far too time consuming to evaluate and are excluded in favor of the less sophisticated ones. More attention should be put to minimize the filter need as they consume both power and space/weight. Additional time can be put to optimize and evaluate the snubber circuits as they now only are evaluated in the sense of suppressing voltage transients. All the wires with their respective stray inductance and capacitance have to be estimated to avoid noise and transients. A light weight transformer could be used to form a 12-pulse rectifier to remove the need for extensive filter on the input.

When having further knowledge of both the generator supplying power and the motor consuming power more detailed filters could be designed. All parts also has to be verified and tested extensively using both simulation and under real circumstances.

60

Bibliography

[1] Martin Gårdman Andreas Johansson. Commissioning and Evaluation of an Inverter

Prototype. Dept. of industrial Electrical Engineering and Automation. Lund

University. 2007

[2] Julius Luukko. Direct torque control of permanent magnet synchronous machines- analysis and implementation. Lappeenranta 2000.

[3] István Schmidt, Katalin Vincze, Károly Veszprémi and Balázs Seller. Adaptive hysteresis current vector control of synchronous servo driver with different tolerance areas. Department of electrical machines and drives. Budapest University of technology and economics. June 29, 2000

[4] Michael O’Neill. The benefits of using a Cree inc. IGBT/SiC Schottky co-pack in

AC inverter applications. Cree inc. September 2006.

[5] Jim Richmond. Hard-switched silicon GBTs? Cut switching losses in half with silicon carbide Schottky diodes. Cree Inc.

[6] S. Vieillard, R. Meuret, High efficiency, high reliability 2 kW inverter for aeronautical application. Hispano-Suiza, rond point René Ravaud, Moissy-

Camayel, France. 2007. ISBN: 9789075815108

[7] Fuji IGBT modules application manual. Fuji electric device technology co., LTD.

February 2004.

[8] Laszlo Balogh. Design and application guide for high speed MOSFET gate drive circuits. http://www.powersystems.eetchina.com/PDF/2007JUL/PSCOL_2007JUL26_DRO

P_TA_101.PDF

[9] International Rectifier. Applications note AN-990. Application characterization of

IGBTs

[10] International Rectifier. Applications note AN-983. IGBT characteristics.

[11] Alf Alfredsson, Karl Axel Jaconbsson, Anders Rejminger. Elkrafthandboken,

Elmaskiner, Studentlitteratur, ISBN: 91-47-00066

[12] Mohan, Undeland, Robbins, Power Electronics, Wiley, ISBN:0-471-22693-9

[13] IGBT MODULE APPLICATION Manual Hitachi, Ltd. Ref.No.IGBT-01 (Rev.2)

[14] Mats Alaküla, Power Electronic Control, KFS Lund AB

61

[15] Designing LC-filters for AC-drives, Bravo Electric Components, Inc.

[16] Electrolytic capacitors application guide, Evox Rifa.

[17] AN9011High Input Voltage, Off-line Flyback Switching Power Supply using FSC

IGBT (SGL5N150UF), Fairchild semiconductor

[18] International Rectifier. Application note AN-1095. Design of the Inverter Output

Filter for Motor Drives with IRAMS Power Modules

[19] MIL-STD-704, Military Standard, Aircraft Electric Power Characteristics,

Department of Defense USA

[20] Snubber Considerations for IGBT Applications, International Rectifier

[21] Failure Rates of HiPak Modules Due to Cosmic Rays, AN-5SYA 2042-02ABB,

ABB, Nando Kaminski

62

Appendix A1 - Output Characteristics using Surge Current Limiter

Vdc

600

0

600

400

200

0

500

400

300

200

100

-200

-400

-600

500

400

300

200

100

0

-100

-200

-300

-400

-500

0

Time offset: 0

0.005

0.01

Vab inverter

Vab Load

0.015

NO over-shoot

0.02

0.025

Appendix A2 - Output Characteristics using NO Surge Current Limiter

Vdc

600

0

600

400

200

0

500

400

300

200

100

-200

-400

-600

500

400

300

200

100

0

-100

-200

-300

-400

-500

0

Time offset: 0

0.002

Slight over-shoot

0.004

0.006

Vab inverter

Vab Load

0.008

0.01

0.012

0.014

10

5

0

-5

-10

-15

-20

0

Time offset: 0

18

Appendix A3 - Rectifier/IGBT Currents using Surge Current Limiter

I Diodes 1 3 5 ave

10

8

6

0

40

4

2

16

14

12

35

30

25

20

15

10

5

0

-5

20

I Diodes 1 3 5

I IGBT 1 3 5

Steady state current level correspond to the calculated value

15

0.005

0.01

0.015

0.02

0.025

140

Appendix A4 - Rectifier/IGBT Currents using No Surge Current Limiter

I Diodes 1 3 5 ave

150

100

50

0

-50

20

15

10

5

0

-5

-10

-15

0

Time offset: 0

120

100

80

60

40

Plot available after one cycle when calculating RMS

20

0

300

Critical over-shoot

250

200

0.002

0.004

I Diodes 1 3 5

I IGBT 1 3 5

0.006

0.008

0.01

0.012

Appendix B - System

Simulation Model

B

A

C

Appendix C1 - Main Grid Voltage Harmonics without Filter

AppendixC

Appendix C2 - Main Grid Current Harmonics without Filter

Mag (% of Fundamental)

Appendix C3 - Main Grid Voltage Harmonics with Filter

Mag

Appendix C4 - Main Grid Voltage Harmonics with 12-Pulse Rectifier

12-PulseRectifier Harmonicswith

Mag

GridVoltage -Main AppendixC4

Appendix C5 - Main Grid Current Harmonics using 12-Pulse Rectifier

Mag

Appendix D1 - IGBT chart 1

Vd @15A

Rd @ 15A

Vt @15A

Rt @15A

FWD

Plosstot @14kHz

Plosstot @10kHz

Plosstot @6kHz

[email protected]

[email protected]@kHz

[email protected]

Pcond @ 15A 125°C 50% Dutycycle (W)

Err @125°C (mJ/cycle) 15A 600V

Eoff @125°C (mJ/cycle) 15A 600V

Eon @125°C (mJ/cycle) (ns) 15A

600V

Fsw

Vf FWD @ 15A 125°C

Vcesat @125°C & 15A

Ic @100°C

Ic @25°C

Vcemax

Ptot @25°C

Package

Manufacturer

Modelname IGBT

Appendix D2 - IGBT chart 2

Vd @15A

Rd @ 15A

Vt @15A

Rt @15A

FWD

Plosstot @14kHz

Plosstot @10kHz

Plosstot @6kHz

[email protected]

[email protected]@kHz

[email protected]

Pcond @ 15A 125°C 50% Duty-cycle (W)

Err @125°C (mJ/cycle) 15A 600V

Eoff @125°C (mJ/cycle) 15A 600V

Eon @125°C (mJ/cycle) (ns) 15A 600V

Fsw

Vf FWD @ 15A 125°C

Vcesat @125°C & 15A

Ic @100°C

Ic @25°C

Vcemax

Ptot @25°C

Package

Manufacturer

Modelname IGBT (continue)

Appendix D3 - Mosfet/Diode chart

Semiconductor

Plosstot @14kHz

Plosstot @10kHz

Plosstot @6kHz

[email protected]

[email protected]@kHz

[email protected]

Conduction-loss @ 15A 125°C 50%

Duty-cycle (W)

Err @125°C (mJ/cycle)

Eoff @125°C (mJ/cycle)

Eon @125°C (mJ/cycle) (ns) Semiconductor

Switching Speed

Vf FWD @ 15A 125°C

Rdson @ 125 °C 15A

Id @100°C

Id @25°C

Q

Vf 160°C @ 10A

Vf 25°C @ 10A

Id 160°C

Id 125°C

Vds

Ptot

Vds

Ptot

Package

Manufacturer

Package

Manufacturer

Modelname MOSFET Diode

Appendix E1 - Decoupling Capacitor Snubber Circuit Schematics

Appendix E2 - Restricted Decoupling Capacitor Snubber Circuit Schematics

Appendix F - Layout of components within enclosure

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