Protecting Instrumentation Amplifiers

Protecting Instrumentation Amplifiers
All data acquisition board designs have to contend with ESD, EMI, and
overvoltages. Can one solution protect the circuitry
against all three hazards?
y their nature, instrumentation amplifiers are used in applications where their inputs are
often exposed to electromagnetic interference (EMI), electrostatic discharge (ESD), and
overvoltage events. Although these conditions are fundamentally different, the circuits used
to protect against each of them are similar. This implies the possibility of a single application
circuit that can protect against all three conditions.
In this article, we look at each interference source and propose suitable protection circuitry as we work
toward a universal input-protection circuit. As is often the case, there’s a tradeoff—in this case, between
the level of protection and the corresponding effect on circuit performance.
Overvoltage Protection
Whenever an amplifier’s input voltage goes outside its supply range, the unit can be damaged, usually
by high current flow, even when it is turned off. Typical maximum ratings limit the allowable input voltage to the positive and negative supplies, or
possibly 0.3 V outside the supplies. But
some devices incorporate internal, currentlimiting resistors, which let the input voltage exceed the supplies by varying degrees.
Furthermore, some instrumentation amplifiers can tolerate—and indeed are designed
to operate under—high-input, commonmode voltages.
To avoid damage, engineers generally use
clamping to limit the input current to
either 10 mA or 20 mA, depending on the
device. This value is a conservative rule of
thumb based on metal trace widths in a typical amplifier input stage. Higher currents
can cause metal migration, which will
eventually lead to an open trace. Migration
is a cumulative effect that might not lead to
a failure for a long time. Failure can be
caused by multiple overvoltage events,
making the failure modes difficult to identify. Even though an amplifier may appear
to withstand overvoltage currents well
above this level for a short time, limiting
James Bryant,
Walt Kester,
Chuck Kitchin,
Eamon Nash,
Analog Devices, Inc.
Figure 1. The equivalent circuit for the input stage of Analog Devices’ AD620 during an overvoltage event
includes two 400 Ω series resistors. Up to 20 mA can safely flow through the resistors. Adding external
series resistors (RLIMIT) provides additional protection.
Device Input Noise Max input Rext for 10%
Rext for 40%
additional noise additional noise
AD620 9 nV/√Hz
AD623 35 nV/√Hz
AD627 38 nV/√Hz
20 mA
10 mA
20 mA
348 Ω
8.08 Ω
10.0 Ω
2.49 kΩ
40.2 kΩ
45.3 kΩ
Figure 2. External series resistors combine with internal electrostatic discharge diodes to create an overvoltage, current-limiting clamping circuit. Higher currents can be carried by external Schottky diodes,
allowing smaller series resistors. Zener diodes or TransZorb diodes provide differential overvoltage protection if RLIMIT is small or is omitted.
the current is important to guarantee longterm reliability.
Figure 1 shows an equivalent input circuit
for the input stage of Analog Devices’
AD620 during an overvoltage condition.
The unit has internal 400 Ω resistors in
series, with the input transistor junctions
and their protection diodes. The AD620
series was designed to handle maximum
input currents of 20 mA, so the internal
diodes protect the unit from input voltages 8
V greater than either supply voltage
(20 mA × 0.4 kΩ ). So, for ±15 V supplies,
the maximum safe input level is ±23 V.
In addition, the differential input voltage
should also be given a value that limits the
maximum input current to 10 mA. The
equivalent circuit shows that the input current flows through two external RLIMIT resistors, the two internal RS resistors, the gainsetting resistor RG, and two diode drops (D1
and the Vbe junction of Q2). For a given
differential input voltage, the input current
is a function of RG and the amplifier gain.
Therefore, a gain resistor of 49.9 Ω (gain =
1000) affects the input current more than
does an RG of 5.49 kΩ (gain = 10).
A generalized external protection circuit
using Schottky diodes and external currentlimiting resistors can ensure input protection (see Figure 2, page 64). The circuit
incorporates protection against both differential- and common-mode overvoltages. If
the amplifier has internal protection diodes
to the supplies (as shown), the diodes conduct at about 0.6 V forward drop above or
below the supply rails. So the internal
diodes—whose primary function is to protect against ESD—actually serve a dual role
because they also clamp the input voltage to
0.6 V beyond the supply rails. In such cases,
external diodes may be unnecessary.
You must choose the external current-limiting resistor so that the maximum input
current is limited to 20 mA. This can cause
large values of R LIMIT, and the resulting
increase in resistor (Johnson) noise might
not be acceptable. Resistors contribute
noise according to the following equation:
Noise (nV/√Hz) = √4KRT × 109
K = Boltzmann’s constant (1.38 × 10–23)
R = resistance in ohms
T = temperature in kelvin (~300 K at
room temperature)
For example, a 1 kΩ resistor has a Johnson
noise of 4 nV/√Hz at room temperature.
Because the protection circuit includes two
equal resistors, whose noise is uncorrelated—
that is, the two noise sources are independent
of each other—the above result must be multiplied by the square root of 2 (the root sum
square of the two noise voltages).
So you have to find a reasonable balance
between the protection provided and the
increased resistor noise introduced. Circuits
that use amplifiers with relatively high noise
are able to tolerate more series protection
without experiencing a serious deterioration
of performance. A good rule of thumb for
choosing protection resistors is to select values that contribute no more than an additional 10% to 30% to the total circuit noise.
For example, a circuit using an amplifier
with a rated noise level of 20 nV/√Hz can
tolerate an additional 2–6 nV/√Hz of
Johnson noise. Because the lower value is
root sum squared with the higher value, the
effect of the lower value on the total noise is
almost negligible.
In situations where the required protection resistor generates too much noise, you
can use the external Schottky protection
diodes shown in Figure 2. These begin to
conduct at about 0.3 V, so the overvoltage
current is shunted through them to the supply rails rather than through the internal
diodes. Therefore, you can set RLIMIT by the
maximum allowable diode current, which
can be much larger than the internal limit
of 10 mA or 20 mA. For instance, a 500 Ω
RLIMIT resistor would limit the diode current to 200 mA for a VIN of 100 V.
Unfortunately, most ordinary diodes (e.g.,
Schottky, silicon, and germanium) have
high leakage currents that cause large offset
errors at the amplifier’s input, and the leakage increases directly in proportion to temperature. This tends to rule out the use of
external diodes in applications in which the
amplifier is used with high-impedance
Figure 3. The IEC 1000-4-2 ESD test pulse can be up to 16 kV in amplitude (air-gap discharge) and has a
much faster rise time than the Human Body Model pulse.
sources. For most applications, limiting
resistors alone provide adequate protection
against overvoltage conditions.
A series protection resistor also produces a
voltage drop because of the amplifier bias
current flowing through it. This drop
appears as an increase in the circuit offset
voltage (and as offset drift if the bias current
changes with temperature). But because the
same series protection resistance is generally
used in both the amplifier’s inputs, the
effect is serious only with amplifiers that
have large input offset currents (offset current is the difference between the amplifier’s two bias currents).
Input series resistors also limit current
when a differential overvoltage condition
occurs. In low-noise applications, you
achieve differential overvoltage protection
using Zener or TransZorb diodes, as shown
in Figure 2. The diodes in Figure 2 can
limit the maximum differential input voltage to less than |VPOS – VNEG|, if required.
Two-op-amp instrumentation amplifiers
don’t generally need such differential protection because their input current is not a
function of the gain-setting resistor, as is the
case with a three-op-amp instrumentation
amplifier configuration.
Electrostatic Discharge
The high voltages and high peak currents
generated by ESD can partially or permanently damage an IC. Several specifications
related to the safe handling of semiconductors—such as MIL-STD-883B, METHOD
3015.7 (also called Human Body Model, or
HBM), the Machine Model (MM), and the
Charged Device Model (CDM)—are well
known in the semiconductor business. But
only recently have system-level standards for
ESD susceptibility emerged.
Some devices require higher ESD immunity than others. For example, ICs on a PC
board surrounded by other circuits are generally far less likely to be subject to ESD
than are signal-conditioning devices (e.g.,
amplifiers that interface with sensors in the
outside world).
Since 1996, all electronic equipment sold
in the European community must be
immune to ESD, as defined in specification
IEC1000-4-2. This is a system-level specification applying to the end equipment, not
to the individual ICs. The practical reality is
that the signal-conditioning components
(e.g., amplifiers, which are the “first line of
fire”) must be protected from specified ESD
or must exhibit immunity to it. However,
precision analog circuits, which often feature low bias currents, actually tend to be
more susceptible to damage than do common digital circuits. This is mainly because
they cannot use the traditional ESD protec-
tion structures that tend to increase input
leakage currents.
Traditional ESD test methods don’t fully
test a product’s susceptibility to the type of
discharge specified in IEC1000-4-2. Figure 3
(page 65) shows the profile of the test pulses
used for the two types of testing. There are
some important differences between the
HBM method and the IEC specification.
The IEC specification is far more stringent
in terms of discharge energy. The peak current injected is more than four times greater.
The current rise time is significantly faster in
the IEC test, and the IEC test is conducted
with the device under power.
IEC1000-4-2 specifies ESD compliance
testing using two coupling methods: contact
discharge and air-gap discharge. Contact
discharge, which has a maximum discharge
voltage of 8 kV, calls for a direct connection
to the unit being tested. Air-gap discharge
uses a higher test voltage (16 kV) but does
not make direct contact with the test unit.
With air-gap discharge, the ESD gun moves
toward the test unit, developing an arc
across the air gap. This method is influenced by humidity, temperature, barometric pressure, distance, and the discharge
gun’s closure rate. The contact-discharge
method, although less realistic, is more
repeatable and is gaining acceptance in
preference to the air-gap method.
Figure 4 (page 66) shows a simple technique for protecting amplifiers against highvoltage ESD. The technique is quite simple
and relies again on input-series-resistor protection to limit the current flowing into the
device during the ESD. Carbon resistors,
which are noninductive, should be used as
protection resistors (R PROT in Figure 4)
instead of devices made of metal film or carbon film. There is, however, a tradeoff here,
because carbon film resistors are imprecise
and can add noise.
Reducing EMI
Radio frequency interference (also called
EMI) can seriously affect the DC performance of high-accuracy circuits. Because of
their relatively low bandwidth, amplifiers
don’t accurately amplify RF signals in the
Figure 4. External series resistors augment internal ESD protection resistors by limiting fault current flow
into the instrumentation amplifier’s inputs.
MHz range. But these out-of-band signals
(either differential mode or common mode)
can couple into the precision amplifier
through its input, output, or power-supply
pins. Then, through various junctions in
the amplifier, an unexplained and unwanted DC offset at the output can result.
Fortunately, proper filtering can minimize
or prevent these errors.
Precision amplifiers are particularly sensitive to common-mode RFI. Figure 5 (page
68) shows recommended filtering component values, along with equations for calculating appropriate components. The figure
shows both common-mode filtering
(R1/C1, R2/C2) and differential-mode filtering (R1 + R2, and C3).
If R1/R2 and C1/C2 are not well
matched, some of the input common-mode
signal at VIN will be converted to a differential signal at the amplifier’s inputs. For this
reason, C1 and C2 should be matched to
within at least 5% of each other. R1 and R2
should be 1% metal film resistors to ensure
matching. Capacitor C3 helps attenuate the
differential signal that can result from im-
perfect matching of the common-mode filters. In this type of filter, C3 should be
much larger than C1 or C2 so that any differential signal caused by mismatching of
the common-mode signals will be sufficiently attenuated. You can use either traditional 5% silver-mica capacitors, miniature
micas, or Panasonic ± 2% PPS film capacitors. The overall filter bandwidth should be
at least 100 × the input signal bandwidth.
For optimum filter performance, the EMI
filter components should be symmetrically
mounted on a PC board with a large ground
plane and should be close to the amplifier
input. Figure 6 (page 68) shows a typical layout for a standard 8-pin amplifier. This type
of layout is easiest to achieve if you use surface-mount components. The board should
have at least two layers, ideally with one continuous ground plane running under the
component plane (as shown). Multiple vias
should run between the sections of ground
plane on the component layer and the continuous ground plane (either on the other
side of the circuit board or in a buried layer).
We tested the common-mode rejection of
C1/ C2
402 kΩ
1000 pF
0047 µF
10 kΩ
1000 pF
0022 µF
20 kΩ
1000 pF
0022 µF
Figure 5. Protection against electromagnetic interference requires attention to both differential-mode and
common-mode interference. R1/C1 and R2/C2 time constants should be closely matched to preserve highfrequency common-mode rejection.
Figure 6. Symmetrical layout of EMI filter components increases the effectiveness of the filter. A continuous ground plane should run under the component layer (either on the other side of the circuit board or in
a buried layer) with plenty of vias connecting to the sections of ground plane on the component layer.
the AD620 in the circuit shown in Figure 5
by applying a 1 V p-p common-mode signal
to the input. The AD620 gain was 1000. We
measured the RTI offset voltage as the frequency of the sine-wave source varied from
DC to 20 MHz. The maximum input offset
voltage shift was 1.5 µV. The filter bandwidth was ~400 Hz.
Common-Mode Chokes
Common-mode chokes offer a simple, onecomponent alternative to RC passive filters.
Because their DC resistance is low (a few
ohms), they attenuate RFI and add little noise
as compared with RC networks. Selecting the
proper common-mode choke is critical, however. The choke used in the circuit shown in
Figure 7 (page 69) is a Pulse Engineering
B4001 designed for Digital Subscriber Line
data receivers (through-hole mount). The
B4003 is an equivalent surface-mount choke.
The maximum RTI offset shift measured
from DC to 20 MHz was 4.5 µV. Unlike the
RC filter of Figure 5, the choke-based filter
offers no differential-mode filtration.
But you can add an RC differential-mode
filter after the choke. The advantage of this
configuration is that there is no penalty of
potential common-mode–to–differentialmode conversion, as there is with the combination RC networks discussed earlier.
In addition to filtering the inputs and the
power pins, you have to protect amplifier
outputs from RFI, especially if they must
drive long lengths of cable. RFI on the output can couple into the amplifier, where it is
rectified and appears again on the output as a
DC offset shift. A resistor or ferrite bead in
series with the output is the simplest output
filter. Adding a capacitor and a resistor (as
shown in Figure 7, page 69) improves the filter. In general, however, you shouldn’t connect such a capacitor to the resistor’s amplifier side because it could cause the amplifier
to become unstable.
A Multipurpose Protection Circuit
It is clear from this discussion and from
the examples that a simple circuit can protect against more than one type of interference. For example, the circuit in Figure 5
provides some protection against all three
types of interference—the input series resistors limiting the current during ESD and
overvoltage, as well as constituting part of a
low-pass differential and common-mode filter. In this example, you don’t need to protect against large differential voltages
because the large series resistors limit the
worst-case current (for a 160 V differential
overvoltage) to around 20 mA.
For Further Reading
Amplifier Applications Guide. 1992. Analog
Devices, Inc., Norwood, MA, Section
EIAJ ED-4701 Test Method C-111,
Electrostatic Discharges (available from the
Japan Electronics Bureau, New York, NY).
ESD Association Draft Standard DS5.3 for
Electrostatic Discharge (ESD) Sensitivity
Testing—Charged Device Model (CDM)
Component Testing (available from the
ESD Association Inc., Rome, NY).
ESD Association Standard S5.2 for Electrostatic Discharge (ESD) Sensitivity
Testing—Machine Model (MM)—Component Level (available from the ESD
Association Inc., Rome, NY).
ESD Prevention Manual. 1986. Analog
Devices, Inc., Norwood, MA.
How to Reliably Protect CMOS Circuits
Against Power Supply Overvoltaging,
Figure 7. Common-mode chokes offer a simple, one-component alternative to RC filters. A simple RC filter
can protect the amplifier’s output from EMI.
Application Note AN-311, Analog
Devices, Inc., Norwood, MA.
Linear Design Seminar. 1994. Analog
Devices, Inc., Norwood, MA, Section
Lyne, N. 1995. Electrical Overstress
Damage to CMOS Converters, Application Note AN-397, Analog Devices, Inc.,
Norwood, MA.
MIL-STD-883 Method 3015, Electrostatic
Discharge Sensitivity Classification (available from Standardization Document
Order Desk, Philadelphia, PA).
Pulse Engineering, Inc., San Diego, CA.
©Reprinted from SENSORS, April 2000
Systems Applications Guide. 1992. Analog
Devices, Inc., Norwood, MA, Section
1:1.37-1.55 and 56-72.
TransZorbs (available from General
Semiconductor, Inc., Melville, NY). ■
James Bryant is European Applications
Manager, Walt Kester is Corporate Staff
Applications Engineer, Chuck Kitchin is
Technical Support Engineer, and Eamon Nash
is an Applications Engineer, Standard Linear
Products Div., Analog Devices, Inc., 804
Woburn St., MS-125, Wilmington, MA 018873462; 781-937-1292, fax 781-937-1024.
Printed in U.S.A.
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