Volume 44, Number 4

Volume 44, Number 4
Volume 44, Number 4, 2010
A forum for the exchange of circuits, systems, and software for real-world signal processing
In This Issue
2 Editors’ Notes; New Product Introductions
3 Two Ways to Measure Temperature Feature Simplicity, Accuracy, and Flexibility
9 Inertial Sensors Facilitate Autonomous Operation in Mobile Robots
13 Multichannel DDS Enables Phase-Coherent FSK Modulation
15 High-Side Current Sensing with Wide Dynamic Range: Three Solutions
20 Boost Supply and High-Voltage DAC Provide Tuning Signal for Antennas and Filters
23 High-Performance Difference Amplifier with Precision Supply-Referenced Level Shift
www.analog.com/analogdialogue
Editors’ Notes
PRODUCT INTRODUCTIONS: Volume 44, number 4
IN THIS ISSUE
Two Ways to Measure Temperature Feature Simplicity, Accuracy,
and Flexibility
Thermocouples are widely used for measuring temperature.
This article suggests two signal conditioning solutions. The
first combines reference-junction compensation and signal
conditioning in a single analog IC for convenience and ease of use;
the second separates the reference-junction compensation from
the signal conditioning to provide greater flexibility and accuracy
in a digital solution. Page 3.
Inertial Sensors Facilitate Autonomous Operation in Mobile Robots
Ground-based robot systems must often handle dull, dirty, and
dangerous tasks in missions where direct human involvement
is expensive, dangerous, or ineffective. Some robotic platforms
can operate autonomously, using navigation systems to monitor
and control their motion. Accuracy is key for useful autonomous
operation. MEMS gyroscopes provide a mechanism for optimizing
navigation performance. Page 9.
Multichannel DDS Enables Phase-Coherent FSK Modulation
Common single-channel direct-digital synthesizers produce
phase-continuous frequency transitions. In Doppler radar,
NMR/MRI spectrometry, and other applications, however,
phase-coherent transitions are preferred. This article demonstrates
how to configure the AD9958/AD9959 multichannel DDS as a
robust phase-coherent frequency-shift keyed (FSK) modulator by
summing the DDS outputs. Page 13.
High-Side Current Sensing with Wide Dynamic Range:
Three Solutions
Precision current sensing allows designers to measure motor
torque, dc-to-dc converter efficiency, bias current in a power
transistor, and other critical parameters in the presence of
high common-mode voltages. This article describes three
solutions—discrete, integrated, and application-optimized—
that provide high-accuracy, high-resolution current sensing
for a variety of applications. Page 15.
Boost Supply and High-Voltage DAC Provide Tuning Signal for
Antennas and Filters
Antenna arrays and filters are often tuned by varying the voltage
on a barium strontium titanate (BST) capacitor. Voltages
applied to this ferroelectric material cause small variations in the
crystal structure, changing the dielectric constant and, thus, the
capacitance. The capacitor is tuned by applying a voltage of up to
30 V. This article shows an easy way to generate a high-voltage
tuning signal in a low-voltage system. Page 20.
High-Performance Difference Amplifier with Precision SupplyReferenced Level Shift
Designed on small-geometry processes, high-performance ADCs
typically run on single 1.8-V to 5-V supplies. When processing
±10-V or larger signals, an amplifier circuit ahead of the ADC can
attenuate the signal to keep it from saturating the ADC inputs. A
difference amplifier (diff amp) is commonly used when the signal
includes a large common-mode voltage. Page 23.
Dan Sheingold [[email protected]]
Scott Wayne [[email protected]]
2
Data sheets for all ADI products can be found by entering the part
number in the search box at www.analog.com.
October
Amplifier, difference, high-voltage, zero-drift ....................... AD8207
DAC, voltage-output, 20-bit, ±1-LSB INL ............................ AD5791
Driver, backlight, 7-channel, charge pump ......................... ADP8870
Driver, current, programmable, industrial . ........................... AD5749
Driver, MOSFET, dual, high-speed, 4-A ........................... ADP3654
Multiplexer, CMOS, 4-channel, differential....................... ADG5409
Multiplexer, CMOS, 4-channel, latch-up proof ................. ADG5404
Multiplexer, CMOS, 8-channel, latch-up proof . ................ ADG5408
Sensor, angular rate, precision . ....................................... ADIS16133
Transceiver, FSK/GFSK/OOK/MSK/GMSK ................... ADF7023
Translator, clock, integer-N .................................................. AD9550
Transmitter, HDMI, low-power, CEC ............................ ADV7524A
November
ADC, pipelined, 14-bit, 80-MSPS ......................................... AD9641
ADC, SAR, 8-channel, 12-bit, 1-MSPS . ............................... AD7298
Amplifier, audio, Class-D, 3-W mono ................................. SSM2375
Amplifiers, thermocouple, with CJC..................................... AD849x
Controller, hot-swap/digital power monitor ....................... ADM1275
Regulators, low-dropout, 150-mA loads ............... ADP160/ADP162
December
ADC, pipelined, 16-bit, 200-MSPS/250-MSPS ..................... AD9467
ADC, SAR, 8-channel, 10-bit, 1-MSPS . ........................... AD7298-1
Amplifier, audio, Class-D, 2 × 2-W stereo ......................... SSM2380
Amplifier, audio, Class-D, 2.4-W mono, digital-input ......... SSM2517
Amplifier, difference, pin-selectable attenuation ................... AD8475
Amplifier, variable-gain, digitally programmable, dual........... AD8366
Codec, audio, 28-/56-bit, 2 ADCs/2 DACs ....................... ADAU1961
Comparator, fast, low-power, automotive-grade . ................. AD8468
Converter, dc-to-dc, 6-MHz, step-down, 500-mA . ........... ADP2125
Converter, dc-to-dc, isolated, 5 kV rms ........................... ADuM6000
DACs, octal, 12-/16-bit, 5-ppm/°C reference . .... AD5629R/AD5669R
DACs, unbuffered-voltage output,
12-/16-bit, 1-μs ................................................ AD5512A/AD5542A
Drivers, ADC, ultralow-noise ........................................ ADA4930-x
Front Ends, analog, 8-channel, medical/auto .......... AD9278/AD9279
Generator, clock, low-jitter, 14-output/29-output . ............ AD9523-1
Isolators, digital, 2-channel, 5-kV rms ........ ADuM2210/ADuM2211
Isolators, digital, 2-channel, 5-kV, 400-mW dc-to-dc . .... ADuM620x
Isolators, digital, 4-channel, transformer driver .............. ADuM347x
Microcontroller, precision analog, ARM7, 12-bit ............ ADuC7126
Mixer, balanced, 2200-MHz to 2700-MHz ........................ ADL5353
Receiver, broadband, 100-MHz to 1000-MHz ................ ADRF6850
Regulators, low-dropout, triple-output, 200-mA .... ADP322/ADP323
Sensor, inertial, six-degrees-of-freedom . .......................... ADIS16375
Sensor, vibration, digital, 3-axis ...................................... ADIS16227
Supervisors, microprocessor,
programmable 2-supply ................................. ADM6305/ADM6306
Switches, CMOS, quad SPST,
power-off protection ........................................ ADG4612/ADG4613
Switches, CMOS, triple/quad SPDT,
latch-up proof .................................................. ADG5433/ADG5434
Transistors, matched, quad NPN .......................................... MAT14
Analog Dialogue, www.analog.com/analogdialogue, the technical magazine of
Analog Devices, discusses products, applications, technology, and techniques
for analog, digital, and mixed-signal processing. Published continuously
for 44 years—starting in 1967—it is currently available in two versions.
Monthly editions offer technical articles; timely information including recent
application notes, new-product briefs, pre-release products, webinars and
tutorials, and published articles; and potpourri, a universe of links to important
and relevant information on the Analog Devices website, www.analog.com.
Printable quarterly issues feature collections of monthly articles. For history
buffs, the Analog Dialogue archive includes all regular editions, starting with
Volume 1, Number 1 (1967), and three special anniversary issues. If you wish
to subscribe, please go to www.analog.com/analogdialogue/subscribe.html.
Your comments are always welcome; please send messages to dialogue.
[email protected] or to Dan Sheingold, Editor [[email protected]]
or Scott Wayne, Publisher and Managing Editor [[email protected]].
ISSN 0161-3626 ©Analog Devices, Inc. 2011
Two Ways to Measure Temperature
Using Thermocouples Feature
Simplicity, Accuracy, and Flexibility
By Matthew Duff and Joseph Towey
Introduction
The thermocouple is a simple, widely used component for
measuring temperature. This article provides a basic overview
of thermocouples, describes common challenges encountered
when designing with them, and suggests two signal conditioning
solutions. The first solution combines both reference-junction
compensation and signal conditioning in a single analog IC for
convenience and ease of use; the second solution separates the
reference-junction compensation from the signal conditioning to
provide digital-output temperature sensing with greater flexibility
and accuracy.
Thermocouple Theory
A thermocouple, shown in Figure 1, consists of two wires of
dissimilar metals joined together at one end, called the measurement
(“hot”) junction. The other end, where the wires are not joined,
is connected to the signal conditioning circuitry traces, typically
made of copper. This junction between the thermocouple metals
and the copper traces is called the reference (“cold”) junction.*
THERMOCOUPLE
METAL A
WIRING TO SIGNAL
CONDITIONING
CIRCUITRY
METAL B
MEASUREMENT
JUNCTION
REFERENCE
JUNCTION
Figure 1. Thermocouple.
The voltage produced at the reference junction depends on the
temperatures at both the measurement junction and the reference
junction. Since the thermocouple is a differential device rather
than an absolute temperature measurement device, the reference
junction temperature must be known to get an accurate absolute
temperature reading. This process is known as reference junction
compensation (cold junction compensation.)
Thermocouples have become the industry-standard method for
cost-effective measurement of a wide range of temperatures with
reasonable accuracy. They are used in a variety of applications
up to approximately +2500°C in boilers, water heaters, ovens,
and aircraft engines—to name just a few. The most popular
thermocouple is the type K, consisting of Chromel® and Alumel®
(trademarked nickel alloys containing chromium, and aluminum,
manganese, and silicon, respectively), with a measurement range
of –200°C to +1250°C.
*We use the terms “measurement junction” and “reference junction”
rather than the more traditional “hot junction” and “cold junction.” The
traditional naming system can be confusing because in many applications the
measurement junction can be colder than the reference junction.
Analog Dialogue Volume 44 Number 4
Why Use a Thermocouple?
Advantages
•Temperature range: Most practical temperature ranges,
from cryogenics to jet-engine exhaust, can be served using
thermocouples. Depending on the metal wires used, a
thermocouple is capable of measuring temperature in the range
–200°C to +2500°C.
•Robust: Thermocouples are rugged devices that are immune
to shock and vibration and are suitable for use in hazardous
environments.
•Rapid response: Because they are small and have low thermal
capacity, thermocouples respond rapidly to temperature
changes, especially if the sensing junction is exposed. They
can respond to rapidly changing temperatures within a few
hundred milliseconds.
•No self heating: Because t her mocouples require no
excitation power, they are not prone to self heating and are
intrinsically safe.
Disadvantages
•Complex signal conditioning: Substantial signal conditioning
is necessary to convert the thermocouple voltage into a usable
temperature reading. Traditionally, signal conditioning has
required a large investment in design time to avoid introducing
errors that degrade accuracy.
•Accuracy: In addition to the inherent inaccuracies in
thermocouples due to their metallurgical properties, a
thermocouple measurement is only as accurate as the reference
junction temperature can be measured, traditionally within
1°C to 2°C.
•Susceptibility to corrosion: Because thermocouples consist of
two dissimilar metals, in some environments corrosion over
time may result in deteriorating accuracy. Hence, they may
need protection; and care and maintenance are essential.
•Susceptibility to noise: When measuring microvolt-level
signal changes, noise from stray electrical and magnetic
fields can be a problem. Twisting the thermocouple wire pair
can greatly reduce magnetic field pickup. Using a shielded
cable or running wires in metal conduit and guarding can
reduce electric field pickup. The measuring device should
provide signal filtering, either in hardware or by software,
with strong rejection of the line frequency (50 Hz/60 Hz)
and its harmonics.
Difficulties Measuring with Thermocouples
It is not easy to transform the voltage generated by a thermocouple
into an accurate temperature reading for many reasons: the
voltage signal is small, the temperature-voltage relationship is
nonlinear, reference junction compensation is required, and
thermocouples may pose grounding problems. Let’s consider
these issues one by one.
Voltage signal is small: The most common thermocouple
types are J, K, and T. At room temperature, their voltage
varies at 52 μV/°C, 41 μV/°C, and 41 μV/°C, respectively. Other
less-common types have an even smaller voltage change with
temperature. This small signal requires a high gain stage before
analog-to-digital conversion. Table 1 compares sensitivities of
various thermocouple types.
3
Table 1. Voltage Change vs. Temperature Rise
(Seebeck Coefficient) for Various Thermocouple Types at 25°C.
Thermocouple Seebeck Coefficient
Type
(𝛍V/°C)
E
61
J
52
K
41
N
27
R
9
S
6
T
41
Because the voltage signal is small, the signal-conditioning
circuitry typically requires gains of about 100 or so—fairly
straightforward signal conditioning. What can be more difficult
is distinguishing the actual signal from the noise picked up on the
thermocouple leads. Thermocouple leads are long and often run
through electrically noisy environments. The noise picked up on
the leads can easily overwhelm the tiny thermocouple signal.
Two approaches are commonly combined to extract the signal from
the noise. The first is to use a differential-input amplifier, such as
an instrumentation amplifier, to amplify the signal. Because much
of the noise appears on both wires (common-mode), measuring
differentially eliminates it. The second is low-pass filtering, which
removes out-of-band noise. The low-pass filter should remove
both radio-frequency interference (above 1 MHz) that may cause
rectification in the amplifier and 50 Hz/60 Hz (power-supply) hum.
It is important to place the filter for radio frequency interference
ahead of the amplifier (or use an amplifier with filtered inputs).
The location of the 50-Hz/60-Hz filter is often not critical—it can
be combined with the RFI filter, placed between the amplifier and
ADC, incorporated as part of a sigma-delta ADC, or it can be
programmed in software as an averaging filter.
Reference junction compensation: The temperature of the
thermocouple’s reference junction must be known to get an
accurate absolute-temperature reading. When thermocouples were
first used, this was done by keeping the reference junction in an
ice bath. Figure 2 depicts a thermocouple circuit with one end at
an unknown temperature and the other end in an ice bath (0°C).
This method was used to exhaustively characterize the various
thermocouple types, thus almost all thermocouple tables use 0°C
as the reference temperature.
+
There are three common ways to compensate for the nonlinearity
of the thermocouple.
Choose a portion of the curve that is relatively f lat and
approximate the slope as linear in this region—an approach
that works especially well for measurements over a limited
temperature range. No complicated computations are needed.
One of the reasons the K- and J-type thermocouples are popular
is that they both have large stretches of temperature for which the
incremental slope of the sensitivity (Seebeck coefficient) remains
fairly constant (see Figure 3).
70
V
–
IRON
REFERENCE
JUNCTIONS
ICE BATH
REFERENCE
Figure 2. Basic iron-constantan thermocouple circuit.
But keeping the reference junction of the thermocouple in an ice
bath is not practical for most measurement systems. Instead most
systems use a technique called reference-junction compensation,
(also known as cold-junction compensation). The reference
junction temperature is measured with another temperaturesensitive device—typically an IC, thermistor, diode, or RTD
4
Voltage signal is nonlinear: The slope of a thermocouple
response curve changes over temperature. For example, at 0°C, a
T-type thermocouple output changes at 39 μV/°C, but at 100°C,
the slope increases to 47 μV/°C.
+
CONSTANTAN
THERMOCOUPLE
–
A variety of sensors are available for measuring the reference
temperature:
1.Thermistors: They have fast response and a small package;
but they require linearization and have limited accuracy,
especially over a wide temperature range. They also require
current for excitation, which can produce self-heating, leading
to drift. Overall system accuracy, when combined with signal
conditioning, can be poor.
2.Resistance temperature-detectors (RTDs): RTDs are
accurate, stable, and reasonably linear, however, package size
and cost restrict their use to process-control applications.
3.Remote thermal diodes: A diode is used to sense the
temperature near the thermocouple connector. A conditioning
chip converts the diode voltage, which is proportional to
temperature, to an analog or digital output. Its accuracy is
limited to about ±1°C.
4.Integrated temperature sensor: An integrated temperature
sensor, a standalone IC that senses the temperature locally,
should be carefully mounted close to the reference junction,
and can combine reference junction compensation and signal
conditioning. Accuracies to within small fractions of 1°C can
be achieved.
J-TYPE
60
SEEBECK COEFFICIENT (𝛍𝛍V/∙C)
UNKNOWN
TEMPERATURE
COPPER WIRE
(resistance temperature-detector). The thermocouple voltage
reading is then compensated to reflect the reference junction
temperature. It is important that the reference junction be read
as accurately as possible—with an accurate temperature sensor
kept at the same temperature as the reference junction. Any
error in reading the reference junction temperature will show
up directly in the final thermocouple reading.
50
K-TYPE
T-TYPE
40
30
20
10
0
–200
0
200
400
600
800
1000
TEMPERATURE (∙C)
Figure 3. Variation of thermocouple sensitivity with
temperature. Note that K-type’s Seebeck coefficient is
roughly constant at about 41 μV/°C from 0°C to 1000°C.
Analog Dialogue Volume 44 Number 4
Another approach is to store in memory a lookup table that matches
each of a set of thermocouple voltages to its respective temperature.
Then use linear interpolation between the two closest points in
the table to get other temperature values.
A third approach is to use higher order equations that model
the behavior of the thermocouple. While this method is the
most accurate, it is also the most computationally intensive.
There are two sets of equations for each thermocouple. One
set converts temperature to thermocouple voltage (useful for
reference junction compensation). The other set converts
t her mocouple volt age to temper at u re. T her mocouple
tables and the higher order thermocouple equations can be
found at http://srdata.nist.gov/its90/main/. The tables and
equations are all based on a reference-junction temperature
of 0°C. Reference-junction compensation must be used if the
reference-junction is at any other temperature.
Grounding requirements: Thermocouple manufacturers make
thermocouples with both insulated and grounded tips for the
measurement junction (Figure 4).
INSULATED
GROUNDED
EXPOSED
all three tip cases if the amplifier’s common-mode range has some
ability to measure below ground in the single-supply configuration.
To deal with the common-mode limitation in some single-supply
systems, biasing the thermocouple to a midscale voltage is useful.
This works well for insulated thermocouple tips, or if the overall
measurement system is isolated. However, it is not recommended
for nonisolated systems that are designed to measure grounded
or exposed thermocouples.
Practical thermocouple solutions: Thermocouple signal
conditioning is more complex than that of other temperature
measurement systems. The time required for the design and
debugging of the signal conditioning can increase a product’s
time to market. Errors in the signal conditioning, especially in
the reference junction compensation section, can lead to lower
accuracy. The following two solutions address these concerns.
The first details a simple analog integrated hardware solution
combining direct thermocouple measurement with referencejunction compensation using a single IC. The second solution
details a software-based reference-junction compensation scheme
providing improved accuracy for the thermocouple measurement
and the flexibility to use many types of thermocouples.
Measurement Solution 1: Optimized for Simplicity
Fi g u r e 6 s how s a sc hem at ic for me a su r i n g a K- t y pe
thermocouple. It is based on using the AD8495 thermocouple
amplifier, which is designed specifically to measure K-type
thermocouples. This analog solution is optimized for minimum
design time: It has a straightforward signal chain and requires
no software coding.
Figure 4. Thermocouple measurement junction types.
The thermocouple signal conditioning should be designed so as to
avoid ground loops when measuring a grounded thermocouple, yet
also have a path for the amplifier input bias currents when measuring
an insulated thermocouple. In addition, if the thermocouple tip is
grounded, the amplifier input range should be designed to handle
any differences in ground potential between the thermocouple tip
and the measurement system ground (Figure 5).
WHEN USING ISOLATED
THERMOCOUPLE TIPS
MEASUREMENT
JUNCTION
PCB
TRACES
THERMOCOUPLE
RFI
FILTER
10k𝛀𝛀
10k𝛀𝛀
1M𝛀𝛀
REFERENCE
JUNCTION
THERMOCOUPLE
AMPLIFIER
5V
1nF
10nF
1nF
AD8495
REF
FILTER FOR
50Hz/60Hz
5mV/∙C
100k𝛀𝛀
1𝛍𝛍F
fC = 1.6Hz
INCLUDES
GROUND COMMON MODE REFERENCECONNECTION
fC = 16kHz
JUNCTION
COMPENSATION
DIFFERENTIAL
fC = 1.3kHz
Figure 6. Measurement Solution 1: optimized for simplicity.
WHEN USING EXPOSED OR
GROUNDED THERMOCOUPLE TIPS
ELECTRICAL CONNECTION
OCCURS AT TIP. VOLTAGE MUST
STAY IN COMMON-MODE INPUT
RANGE OF AMPLIFIER
WHEN THERMOCOUPLE TIP TYPE IS
UNKNOWN
1M𝛀𝛀
Figure 5. Grounding options when using different tip types.
For nonisolated systems, a dual-supply signal-conditioning system
will typically be more robust for grounded tip and exposed tip
types. Because of its wide common-mode input range, a dualsupply amplifier can handle a large voltage differential between
the PCB (printed-circuit board) ground and the ground at the
thermocouple tip. Single-supply systems can work satisfactorily in
Analog Dialogue Volume 44 Number 4
How does this simple signal chain address the signal conditioning
requirements for K-type thermocouples?
Gain and output scale factor: The small thermocouple signal
is amplified by the AD8495’s gain of 122, resulting in a 5-mV/°C
output signal sensitivity (200°C/V).
Noise reduction: High-frequency common-mode and differential
noise are removed by the external RFI filter. Low frequency
common-mode noise is rejected by the AD8495’s instrumentation
amplifier. Any remaining noise is addressed by the external post filter.
Reference-junction compensation: The AD8495, which
includes a temperature sensor to compensate for changes in
ambient temperature, must be placed near the reference junction
to maintain both at the same temperature for accurate referencejunction compensation.
Nonlinearity correction: The AD8495 is calibrated to
give a 5 mV/°C output on the linear portion of the K-type
thermocouple curve, with less than 2°C of linearity error
in the –25°C to +400°C temperature range. If temperatures
beyond this range are needed, Analog Devices Application
Note AN-1087 describes how a lookup table or equation could
be used in a microprocessor to extend the temperature range.
5
Table 2. Solution 1 (Figure 6) Performance Summary
Thermocouple Measurement-Junction
Type
Temperature Range
–25°C to +400°C
K
Reference-Junction
Temperature Range
±3°C (A grade)
±1°C (C grade)
0°C to 50°C
Handling insulated, grounded, and exposed thermocouples:
Figure 5 shows a 1-MΩ resistor connected to ground, which allows
for all thermocouple tip types. The AD8495 was specifically
designed to be able to measure a few hundred millivolts below
ground when used with a single supply as shown. If a larger ground
differential is expected, the AD8495 can also be operated with
dual supplies.
Power
Consumption
Accuracy at 25°C
1.25 mW
Table 2 summarizes the performance of the integrated hardware
solution using the AD8495:
Measurement Solution 2: Optimized for Accuracy and Flexibility
Figure 8 shows a schematic for measuring a J-, K-, or T-type
thermocouple with a high degree of accuracy. This circuit includes
a high-precision ADC to measure the small-signal thermocouple
voltage and a high-accuracy temperature sensor to measure the
reference-junction temperature. Both devices are controlled using
an SPI interface from an external microcontroller.
More about the AD8495: Figure 7 shows a block diagram of
the AD8495 thermocouple amplifier. Amplifiers A1, A2, and
A3—and the resistors shown—form an instrumentation amplifier
that amplifies the K-type thermocouple’s output with a gain
appropriate to produce an output voltage of 5 mV/°C. Inside
the box labeled “Ref junction compensation” is an ambient
temperature sensor. With the measurement junction temperature
held constant, the differential voltage from the thermocouple
will decrease if the reference junction temperature rises for any
reason. If the tiny (3.2 mm × 3.2 mm × 1.2 mm) AD8495 is in
close thermal proximity to the reference junction, the referencejunction compensation circuitry injects additional voltage into
the amplifier, so that the output voltage stays constant, thus
compensating for the reference temperature change.
How does this configuration address the signal conditioning
requirements mentioned earlier?
Remove noise and amplify voltage: The AD7793, shown in
detail in Figure 9—a high-precision, low-power analog front end—
is used to measure the thermocouple voltage. The thermocouple
output is filtered externally and connects to a set of differential
inputs, AIN1(+) and AIN1(–). The signal is then routed through
a multiplexer, a buffer, and an instrumentation amplifier—which
amplifies the small thermocouple signal—and to an ADC, which
converts the signal to digital.
GND
AVDD
REFIN(+)/AIN3(+) REFIN(–)/AIN3(–)
–IN
1M𝛀𝛀
ESD AND
OVP
AD8494/AD8495/
AD8496/AD8497
REF JUNCTION
COMPENSATION
ESD AND
OVP
BAND GAP
REFERENCE
A3
AIN1(+)
AIN1(–)
AIN2(+)
AIN2(–)
OUT
MUX
BUF
AVDD
A1
IOUT1
IN-AMP
INTERNAL
CLOCK
REF
FILTERING
THERMOCOUPLE
MEASUREMENT
JUNCTION
AD7793
DOUT/RDY
DIN
SCLK
CS
DVDD
CLK
Figure 7. AD8495 functional block diagram.
REFERENCE
JUNCTION
𝚺𝚺-𝚫𝚫
ADC
GND
IOUT2
Figure 9. AD7793 functional block diagram.
𝚺𝚺-𝚫𝚫 ADC
AIN1+
VDD
DECOUPLING
VDD
AD7793
AIN1–
TEMPERATURE
SENSOR
ADT7320
SPI_CLK
SPI_MISO
SPI_MOSI
SPI_SEL_A
GND
AVDD
THERMOCOUPLE
+IN
VBIAS
A2
SERIAL INTERFACE
AND CONTROL LOGIC
SENSE
SCLK
DOUT
DIN
CS
VDD
GND
EP
INT
CT
VDD
GND
MICROCONTROLLER
DOUT
SPI_MISO
DIN
SPI_MOSI
SCLK
CS
SPI_SEL_A
VDD
DECOUPLING
VDD
SPI_CLK
SPI_SEL_B
SPI_SEL_A GND
Figure 8. Measurement Solution 2: Optimized for accuracy and flexibility.
6
Analog Dialogue Volume 44 Number 4
Table 3. Solution 2 (Figure 8) Performance Summary
Thermocouple Measurement-Junction Reference-Junction
Type
Temperature Range
Temperature Range Accuracy
J, K, T
Full Range
–10°C to +85°C
–20°C to +105°C
C ompensate for reference -ju nct ion temper at u re:
The ADT7320 (detailed in Figure 10), if placed close enough
to the reference junction, can measure the reference-junction
temperature accurately, to ±0.2°C, from –10°C to +85°C. An
on-chip temperature sensor generates a voltage proportional to
absolute temperature, which is compared to an internal voltage
reference and applied to a precision digital modulator. The
digitized result from the modulator updates a 16-bit temperature
value register. The temperature value register can then be
read back from a microcontroller, using an SPI interface, and
combined with the temperature reading from the ADC to effect
the compensation.
Correct nonlinearity: The ADT7320 provides excellent linearity
over its entire rated temperature range (–40°C to +125°C),
requiring no correction or calibration by the user. Its digital
output can thus be considered an accurate representation of the
reference-junction state.
To determine the actual thermocouple temperature, this
reference-temperature measurement must be converted into
an equivalent thermoelectric voltage using equations provided
by the National Institute of Standards and Technolog y
(NIST). This voltage then gets added to the thermocouple
voltage measured by the AD7793; and the summation is then
translated back into a thermocouple temperature, again using
NIST equations.
Handle insulated and grounded thermocouples: Figure 8
shows a thermocouple with an exposed tip. This provides the best
response time, but the same configuration could also be used with
an insulated-tip thermocouple.
±0.2°C
±0.25°C
Power
Consumption
3 mW
3 mW
Table 3 summarizes the performance of the software-based
reference-junction measurement solution, using NIST data:
Conclusion
Thermocouples offer robust temperature measurement over a
quite wide temperature range, but they are often not a first choice
for temperature measurement because of the required trade-offs
between design time and accuracy. This article proposes costeffective ways of resolving these concerns.
The first solution concentrates on reducing the complexity of the
measurement by means of a hardware-based analog referencejunction compensation technique. It results in a straightforward
signal chain with no software programming required, relying on
the integration provided by the AD8495 thermocouple amplifier,
which produces a 5-mV/°C output signal that can be fed into the
analog input of a wide variety of microcontrollers.
The second solution provides the highest accuracy measurement
and also enables the use of various thermocouple types. A
software-based reference-junction compensation technique, it
relies on the high-accuracy ADT7320 digital temperature sensor
to provide a much more accurate reference-junction compensation
measurement than had been achievable until now. The ADT7320
comes fully calibrated and specified over the –40°C to +125°C
temperature range. Completely transparent, unlike a traditional
thermistor or RTD sensor measurement, it neither requires a
costly calibration step after board assembly, nor does it consume
processor or memory resources with calibration coefficients or
linearization routines. Consuming only microwatts of power,
it avoids self-heating issues that undermine the accuracy of
traditional resistive sensor solutions.
SCLK 1
DOUT 2
SPI
INTERFACE
DIN 3
INTERNAL
REFERENCE
CS 4
TEMPERATURE
VALUE
REGISTER
CONFIGURATION
AND STATUS
REGISTER
THYST
REGISTER
TCRT
REGISTER
THIGH
REGISTER
TLOW
REGISTER
INTERNAL
OSCILLATOR
ADT7320
TCRIT
TEMPERATURE
SENSOR
𝚺𝚺-𝚫𝚫
MODULATOR
THIGH
FILTER
LOGIC
TLOW
6 CT
5 INT
7
GND
8
VDD
Figure 10. ADT7320 functional block diagram.
Analog Dialogue Volume 44 Number 4
7
Appendix
Use of NIST Equation to Convert ADT7320 Temperature to Voltage
The thermocouple reference-junction compensation is based on
the relationship:
(1)
where:
ΔV = thermocouple output voltage
V @ J1 = voltage generated at the thermocouple junction
V @ J 2 = voltage generated at the reference junction
Authors
For this compensation relationship to be valid, both terminals
of the reference junction must be maintained at the same
temperature. Temperature equalization is accomplished with an
isothermal terminal block that permits the temperature of both
terminals to equalize while maintaining electrical isolation.
After the reference-junction temperature is measured, it must be
converted into the equivalent thermoelectric voltage that would
be generated with the junction at the measured temperature. One
technique uses a power series polynomial. The thermoelectric
voltage is calculated:
(2)
where:
E = thermoelectric voltage (microvolts)
an = thermocouple-type-dependent polynomial coefficients
T = temperature (°C)
n = order of polynomial
8
NIST publishes tables of polynomial coefficients for each type
of thermocouple. In these tables are lists of coefficients, order
(the number of terms in the polynomial), valid temperature
ranges for each list of coefficients, and error range. Some types
of thermocouples require more than one table of coefficients to
cover the entire temperature operating range. Tables for the power
series polynomial are listed in the main text.
Matthew Duff [[email protected]]
joined Analog Devices in 2005 as an applications
engineer in the Integrated Amplifier Products
Group. Prior to joining Analog Devices, Matt
worked for National Instr uments in both
design and project management positions on
instrumentation and automotive products. He
received his BS from Texas A&M and MS from Georgia Tech,
both in electrical engineering.
Jose ph Towey [ joe.towe y @ a n a log.com]
joined Analog Devices in 2002 as a senior
test development engineer with the Thermal
Sensing Group. Joe is currently applications
manager for the Thermal Sensing and Switch/
Multiplexer Group. Prior to joining Analog
Devices, Joe worked for Tellabs and Motorola
in both test development and project management positions.
He is qualified with a BSc (Hons) degree in computer science
and a diploma in electronic engineering.
Analog Dialogue Volume 44 Number 4
Inertial Sensors Facilitate
Autonomous Operation in
Mobile Robots
By Mark Looney
The key steps in developing a navigation system start with a good
understanding of each function, with particular emphasis on its
operational goals and limitations. Each function typically has
some clearly defined and easily executed aspects, but also offers
challenging limitations that need to be managed. In some cases,
this process can be iterative, where identifying and dealing with
limitations enables new opportunities for optimization. The best
way to describe this process is through an example.
Introduction
Adept MobileRobots Seekur
The Adept MobileRobots Seekur2 is an autonomous robot that
uses an inertial navigation system (INS) similar to the one shown in
Figure 3. This vehicle has a 4-wheel drive system, with independent
steering and speed control for each wheel, providing the flexibility
to move the platform in any horizontal direction. This ability is
valuable for robotic vehicles in such emerging applications as
warehouse delivery systems, hospital specimen/supply delivery
systems, and military force augmentation systems.
ROBOT
TRIP
BODY
PLAN COMMANDS
TRAJECTORY
PLANNER
–
TRIP PROGRESS
INERTIAL CONDITIONS
Ground-based robot systems must often handle “the dull, the
dirty, and the dangerous” tasks, according to Seth Allen, project
manager at Adept MobileRobots.1 In other words, robot systems
are typically used for missions where direct human involvement is
too expensive, too dangerous, or just ineffective. In many cases, the
ability of robotic platforms to operate autonomously is a valuable
feature, using navigation systems to monitor and control their
motion when moving from one location to the next. Accuracy in
managing position and motion is a key factor in enabling really
useful autonomous operation, and MEMS (microelectromechanical
system) gyroscopes provide a feedback sensing mechanism that
can be very useful in optimizing navigation system performance.
The Seekur® robot system, shown in Figure 1, is an example of
an autonomous system that employs advanced MEMS devices to
improve navigation performance.
FORWARD KINEMATICS
WHEEL #1
DRIVE AND
STEERING
MOTORS
WHEEL #2
DRIVE AND
STEERING
MOTORS
WHEEL #3
DRIVE AND
STEERING
MOTORS
WHEEL #4
DRIVE AND
STEERING
MOTORS
STEERING
AND DRIVE
ENCODERS
STEERING
AND DRIVE
ENCODERS
STEERING
AND DRIVE
ENCODERS
STEERING
AND DRIVE
ENCODERS
KALMAN FILTER
GPS
LASER
INVERSE KINEMATICS
MEMS
GYRO
Figure 3. Adept MobileRobots Seekur navigation system.
Forward Control
Figure 1. Adept MobileRobots Seekur system.
Robot Navigation Overview
A robot movement typically starts with a position change request
from the central processor that is managing the progress of the
robot’s overall mission. The navigation system begins executing a
position change request by developing a trip plan or trajectory.
The trip plan considers available paths, known obstacle locations,
robot capability, and any relevant mission objectives. (For
example, delivery time can be critical for a specimen delivery
robot in a hospital.) The trip plan is fed into a controller, which
produces drive and direction profiles for navigational control.
These profiles result in motion and progress with respect to the
plan. The motion is typically monitored by a number of sensing
systems, each of which produces feedback signals; the feedback
controller combines and translates them into updated trip plans
and conditions. Figure 2 is a basic block diagram of a generic
navigation system.
DESTINATION
INPUT
ROUTE
PLANNER
–
FORWARD
CONTROL
FEEDBACK
CONTROL
DRIVE AND
DIRECTION
INERTIAL
SENSING
Figure 2. Generic navigation system.
Analog Dialogue Volume 44 Number 4
Robot body commands, the main error signals, represent the
difference between the trip plan provided by the trajectory
planner and trip progress updates produced by the feedback
sensing system. They are fed into the inverse kinematics system,
which translates the robot body commands into steering and
velocity profiles for each individual wheel. These profiles
are calculated using Ackermann steering relationships,* which
incorporate tire diameter, surface contact area, spacing, and
other important geometrical features. Ackermann steering
principles and relationships enable these robot platforms to
create electronically linked steering angle profiles similar to
those of the mechanical rack-and-pinion systems used in many
automotive steering systems. Incorporating these relationships
remotely, without requiring the axles to be mechanically linked,
helps minimize friction and tire slip, provides the benefits
of reduced tire wear and energy loss, and allows motion not
possible with simple mechanical linkages.
Wheel Drive and Steering System
Each wheel has a drive shaft that is mechanically coupled to its drive
motor through a gear box and—through another gear box—to an
optical encoder, which is an input to the odometry feedback system.
The steering shaft couples the axle to another servo motor, which
establishes the wheel’s steering angle. The steering shaft also
couples to a second optical encoder through a gear box—which
provides another input to the odometry feedback system.
*Patented by Rudolph Ackermann in 1817(!)
9
Feedback Sensing and Control
The navigation system uses an extended Kalman filter3 to estimate
the pose of the robot on the map by combining data from multiple
sensors. The odometry data on the Seekur is derived from
the wheel traction and steering encoders—which provide the
translation—and a MEMS gyro, which provides the rotation.
Odometry
The odometry feedback system estimates robot position, heading,
and speed using optical-encoder measurements of drive- and
steering shaft rotation. In optical encoders, a disk blocks an internal
light source or allows it to shine on a light sensor via thousands of
tiny openings. As the disk rotates, it produces a series of electrical
pulses that are typically fed into a counter circuit. The number
of counts per rotation is equal to the number of slots in the disc,
which allows the number of rotations (including fractional) to be
calculated from the encoder circuit’s pulse count. Figure 4 provides
a graphical reference and relationship for translating the drive
shaft’s rotation count into linear displacement (position) changes.
L
L = 𝛑𝛑 ∙ D ∙
D
NE
NC
(TIRE DIAMETER)
Figure 5. Laser mapping.
For outdoor applications, the Seekur uses global positioning systems
(GPS) for position measurements (Figure 6). These systems
use flight times of radio signals from at least four satellites to
triangulate a position on the earth’s surface. When available,
they can provide levels of accuracy to within 1 m. However, these
systems are limited by the line of sight requirements, which can
be impeded by buildings, trees, bridges, tunnels, and many other
types of objects. In some cases, where outdoor object locations and
features are known (urban canyons), radar and sonar can also be
used to supplement the position estimates during GPS outages.
Even so, effectiveness is often limited when dynamic conditions
exist, such as cars passing by or construction.
NC = NUMBER OF ENCODER COUNTS PER ROTATION.
NE = NUMBER OF ENCODER COUNTS READ INTO INVERSE KINEMATICS.
Figure 4. Odometry linear-displacement relationship.
The drive-axle and steering-shaft encoder measurements for each
wheel are combined in the forward-kinematics processor, using the
Ackermann steering formulas, which produce heading, turn rate,
position, and linear velocity measurements.
The advantage of this measurement system is that its sensing
function is directly coupled to the drive and steering control
systems, so their state is accurately known. However, its accuracy
in terms of the actual speed and direction of the vehicle is limited
unless reference to a set of real-world coordinates is available. The
primary limitations, or error sources, are in the tire-geometry
consistency (the accuracy and variation of D in Figure 4) and
breaks in the contact between the tire and the ground surface.
Tire geometry is dependent on tread consistency, air pressure,
temperature, and weight—all conditions that can change during
normal robot use. Tire slip depends on turn radii, velocity, and
surface consistency.
Position Sensing
The Seekur system uses various range sensors. For indoor
applications, it employs a 270° laser scanner to build a map
of its environment. The laser system measures object shapes,
sizes, and distance from the laser source using returned-energy
patterns and signal-return times. When in its mapping mode, it
characterizes its workspace by combining scan results from many
different positions in the workspace (Figure 5). This produces
a map of object locations, sizes, and shapes, which is used as a
reference for run-time scans. When used in conjunction with the
mapping information, the laser scanner function provides accurate
position information. If used by itself, it would bear limitations
that include the stop time for scans and an inability to manage a
changing environment. In a warehouse environment, people, lift
trucks, pallet jacks, and many other objects change position often,
which could potentially impact speed to a destination and, indeed,
accuracy of achieving the correct destination.
10
DISTANCE
DISTANCE
DISTANCE
DISTANCE
Figure 6. GPS position sensing.
MEMS Angular Rate Sensing
The MEMS gyroscope used in the Seekur system provides a
direct measurement of the Seekur’s rate of rotation about the yaw
(vertical) axis, which is normal to the earth’s surface in the Seekur
navigational reference frame. The mathematical relationship for
calculating a relative heading is a simple integration of the angular
rate measurement over a fixed period (t1 to t2).
One of the key advantages of this approach is that the gyroscope,
being attached to the robot frame, measures the vehicle’s actual
motion without relying on gear ratios, backlash, tire geometry,
or surface contact integrity. However, the heading estimate does
rely on sensor accuracy, which is a function of the following key
parameters: bias error, noise, stability, and sensitivity. Fixed bias
error translates into a heading drift rate, as shown in the following
relationship that includes the bias error, ωBE:
Analog Dialogue Volume 44 Number 4
Bias error can be broken down into two categories: current and
condition-dependent. The Seekur system estimates current bias
errors when it is not in motion. This requires the navigation
computer to recognize when no position change commands are
being executed and facilitate data-collection bias estimate and
correction-factor updates. The accuracy of this process depends
on sensor noise and the amount of time available to collect data
and formulate an error estimate. The Allan variance curve provides
a convenient relationship between bias accuracy and averaging
time, as shown in Figure 7, which captures the relationship
for the ADIS16265—an iSensor® MEMS device similar to the
gyroscope currently used in the Seekur system. In this case, the
Seekur can reduce the bias error to less than 0.01°/sec, averaged
over 20 seconds, and can optimize the estimate by averaging over
about 100 seconds.
ROOT ALLAN VARIANCE (∙/sec)
1
0.1
Application Examples:
Warehouse Inventory Delivery
Warehouse automation currently uses lift trucks and belt
systems to move materials for organizing inventory and fulfilling
demands. The lift trucks require direct human control, and the
belt system requires regular maintenance attention. In order to
achieve maximum warehouse value, many warehouses are being
reconfigured, a process that opens the door for autonomous robot
platforms. Instead of a substantial construction effort to revise lift
trucks and belt systems, a fleet of robots requires only software
changes and retraining the robot’s navigation system for its new
mission. The key performance requirement in a warehouse delivery
system is the robot’s ability to maintain a consistent pattern of
travel and maneuver safely in a dynamic environment, where
obstacles move and human safety cannot be compromised. In order
to demonstrate the value of MEMS gyroscope feedback on the
Seekur in this type of application, Adept MobileRobots conducted
an experiment to find out how well the Seekur would maintain a
repetitive path, without (Figure 8) and with (Figure 9) MEMS
gyroscopic feedback. It is important to note that this experiment
was run without GPS or laser-scanning correction—for the
purpose of studying the impact of MEMS gyroscopic feedback.
MEAN
+1𝛔𝛔
0.01
–1𝛔𝛔
0.001
0.1
1
10
100
1k
𝛕𝛕 (Seconds)
Figure 7. ADIS16265 Allan variance curve.
The Allan variance 4 relationship also offers insights into the
optimal integration time (τ = t2 – t1). The minimum point on this
curve is typically identified as the in-run bias stability. The heading
estimates are optimized by setting the integration time, τ, equal
to the integration time associated with the minimum point on the
Allan variance curve for the gyroscope in use.
Figure 8. Seekur path accuracy, no MEMS gyroscope
feedback.
Because they influence performance, condition-dependent errors,
such as bias temperature coefficient, can determine how often the
robot must stop to update its bias correction. Using precalibrated
sensors can help address the most common error sources, such
as temperature- and power-supply changes. For example, a
change from the ADIS16060 to the precalibrated ADIS16265
may incrementally increase size, price, and power, but offers 18×
better stability with respect to temperature. For a 2°C change in
temperature, the maximum bias of 0.22°/sec with the ADIS16060
is reduced to 0.012°/sec with the ADIS16265.
The sensitivity error source is proportional to the actual change in
heading, as shown in the following relationship:
Commercial MEMS sensors often provide sensitivity error
specifications that range from ±5% to above ±20%, so they will
need calibration to minimize these errors. Precalibrated MEMS5
gyroscopes, such as the ADIS16265 and ADIS16135, provide
specifications of less than ±1%—with even better performance
in controlled environments.
Analog Dialogue Volume 44 Number 4
Figure 9. Seekur path accuracy, MEMS gyroscope
feedback enabled.
The difference in maintaining the path accuracy is easy to see when
comparing the path traces in Figure 8 and Figure 9. It is important
to note that these experiments were run on early generation
MEMS technology that supported ~0.02°/sec stability. Current
gyroscopes enable 2× to 4× performance improvement at the same
cost, size, and power levels. As this trend continues, the ability to
maintain accurate navigation on repetitive paths will continue to
improve—opening up additional markets and applications, such
as specimen and supply delivery in hospitals.
11
Supply Convoys
Current DARPA initiatives continue to call for more robot
technology to help in force multiplication. Supply convoys are
an example of this type of application, where military convoys
are exposed to opposing threats while forced to move in slow,
predictable patterns. Accurate navigation enables robots, like
the Seekur, to take on more responsibility in supply convoys,
reducing human exposure to threats along their paths. One
key performance metric, where the MEMS gyroscope heading
feedback is particularly helpful, is in managing GPS outage
conditions. The latest Seekur navigation effort, geared towards
this environment, employs MEMS inertial measurement units
(IMUs)6 for better accuracy and their ability to incorporate
future integration advances—for terrain management and other
functional areas.
In order to test how well this system localizes—with and without
the IMU—the error of an outdoor path was recorded and analyzed.
Figure 10 shows the comparison of the errors—with respect to the
true path (from the GPS)—of the odometry only with the errors
when the odometry and the IMU are combined in the Kalman filter.
The positional accuracy was nearly 15× better in the latter case.
450
WITHOUT IMU
WITH IMU
SYSTEM ACCURACY WITH
ADIS16362 FEEDBACK DISABLED
400
350
ERROR (mm)
300
250
200
SYSTEM ACCURACY
WITH ADIS16362
ENABLED
150
50
0
500
1000
1500
2000
2500
3000
TIME (Seconds)
Figure 10. Seekur position error using odometry/IMU (green)
vs. odometry only (blue).
Conclusion
Robot platform developers are finding that MEMS gyroscope
technology provides cost-effective methods for improving
directional estimation and overall accuracy in their navigation
12
Acknowledgment
Analog Devices would like to thank Seth Allen, George Paul, and
the entire team at Adept MobileRobots for the contributions that
they made to this article.
References
(Information on all ADI components can be found at www.analog.com.)
1
http://go.adept.com/content/adeptacqmobrob06142010.
2
w w w.mobilerobots.com /researchrobots/researchrobots/
seekurugv.aspx.
3
http://en.wikipedia.org/wiki/Kalman_filter.
4
http://en.wikipedia.org/wiki/Allan_Variance.
5
www.analog.com/librar y/analogdialogue/archives/43 - 09/
EDCh%203%20sensors.pdf.
6
www.analog.com/en/mems/imu/products/index.html.
Author
100
0
systems. The availability of precalibrated, system-ready devices
enables simple functional integration, which leads to early success
in the development process and allows engineers to concentrate on
system optimization. As MEMS technology continues to improve
gyroscope noise, stability, and accuracy specifications, it will
continue to enable higher levels of accuracy and control, which will
likely continue to open new markets for autonomous robot platforms.
Next-generation development for systems such as the Seekur could
move from gyroscopes to fully integrated MEMS IMU/6-degreesof-freedom (6DoF) sensors. While the yaw-oriented approach is
useful, the world isn’t flat; many other applications, existing and
future, can incorporate MEMS IMUs for terrain management and
for additional accuracy refinement, with three gyroscopes enabling
full alignment feedback and correction.
Mark Looney [[email protected]] is
an iSensor applications engineer at Analog
Devices in Greensboro, NC. Since joining
ADI in 1998, he has accumulated experience
in sensor signal processing, high-speed analogto-digital converters, and dc-to-dc power
conversion. He earned BS (1994) and MS
(1995) degrees in electrical engineering from
the University of Nevada, Reno, and has published several
articles. Prior to joining ADI, he helped start IMATS, a vehicle
electronics and traffic-solutions company, and worked as a
design engineer for Interpoint Corporation.
Analog Dialogue Volume 44 Number 4
Multichannel DDS
Enables Phase-Coherent
FSK Modulation
full scale. Logic low on the profile pins shuts off the sine wave
while logic high passes it to the output. The operation requires
two complementary input data streams to alternate between
frequencies. Note that the two DDS channels continuously
generate F1 and F2. The off feature mutes the appropriate DDS
output, thereby producing a phase-coherent FSK signal.
PHASE-COHERENT FSK MODULATOR
By David Brandon
Common single-channel direct-digital synthesizers (DDS)
produce phase-continuous frequency transitions, as shown in
Figure 1. In applications such as coherent pulse Doppler radar
and NMR/MRI spectrometry for medical and material analysis,
however, phase-coherent transitions are preferred. This article
demonstrates how to configure the AD9958/AD9959 multichannel
DDS as a robust phase-coherent frequency-shift keyed (FSK)
modulator by summing the DDS outputs.
A multichannel DDS virtually eliminates channel-to-channel
temperature and timing issues encountered when synchronizing
multiple single-channel devices. Multichannel DDS outputs,
though independent, share the same system clock, allowing them
to track better over temperature and power-supply deviations
than the outputs of multiple single-channel devices. As a result, a
multichannel DDS is better suited for producing phase-coherent
frequency transitions at the summed output.
SIGNAL A
SIGNAL B
PHASECONTINUOUS
SWITCH
PHASECOHERENT
SWITCH
Figure 1. Phase-continuous and phase-coherent
frequency transitions.
AD9520
AD9958
LPF
CH0
LVPECL
XTAL
LVPECL
500MSPS
REF CLK
REF CLK
CH1
SYNC_CLK
PS3
(ON/OFF)
PS2
(ON/OFF)
CLK
DATA
SOURCE
PRS
EDGES MUST MEET SETUP/
HOLD TIMES TO SYNC_CLK
Figure 2. Setup for phase-coherent FSK modulator.
The AD9959 4-channel DDS produced the results shown in
Figure 3. Its two additional channels serve as a phase reference
for the two switched frequencies at the summed output, making
it easier to demonstrate the phase-coherent switching. The
summed output, shown in the upper trace, exhibits phase-coherent
switching. The middle two traces show reference signals F1
and F2. The bottom trace shows the pseudo-random sequence
(PRS) data stream that selects between the two frequencies. Note
the edges of the PRS data stream do not align exactly with the
frequency transitions of the summed outputs due to the pipeline
delay within the device.
TEK STOP: 5.0GSPS
𝚫𝚫: 478ns
@: 996ns
4
C1 + DUTY
33.34%
LOW SIGNAL
AMPLITUDE
Circuit Description
The AD9520 clock distribution device drives the AD9958 DDS
with a high-performance reference clock, while providing the
same clock to the source for the FSK data stream. The AD9520
provides multiple output logic choices and adjustable delays to
meet setup- and hold times between the FSK data stream and
SYNC_CLK of the multichannel DDS.
The AD9958’s two independent channels operate at preprogrammed
frequencies F1 and F2. Wiring the outputs together sums them.
The profile pins, which drive the multiplier at each DAC input to
control the output amplitude, switch the channel outputs on or off
to select the desired frequency. To accomplish this, each multiplier
has two preprogrammed profile-selectable settings, zero- and
Analog Dialogue Volume 44 Number 4
2
3
1
CH1 5.0V
CH2 200mV 𝛀𝛀
CH3 200mV 𝛀𝛀 CH4 200mV 𝛀𝛀
M100ns
CH1
2.30V
Figure 3. Measured phase-coherent FSK transition.
13
Fig u re 4 shows a n exa mple of phase - cont i nuous FSK
switching, also produced by the A D9959. T his ty pe of
operation requires less bandwidth, but does not maintain
phase memory between transitions.
TEK STOP: 5.0GSPS
4
2
3
and configurable output buffers, it performs with subpicosecond
jitter. Four options provide an on-chip VCO with center frequencies
ranging from 1.45 GHz to 2.95 GHz; a fifth option operates with
an external VCO at frequencies up to 2.4 GHz. Accepting one
differential- or two single-ended references, at frequencies up
to 250 MHz, the devices provide four groups of three LVPECL
clocks at frequencies up to 1.6 GHz. Programmable dividers, with
a divide ratio of 1 to 32, set the output frequency and the coarse
delay for each group. Each LVPECL output can be reconfigured to
provide two 250-MHz CMOS outputs. Operating on a single 3.3-V
supply, the AD9520-x consume 1.5 W max; separate output-driver
and charge-pump supplies can be used for logic compatibility and
to support VCOs having an extended tuning range. Available in
64-lead LFCSP packages, they are specified from –40°C to +85°C
and priced at $12.65 in 1000s.
OPTIONAL
CH1 5.0V
CH2 200mV 𝛀𝛀
CH3 200mV 𝛀𝛀 CH4 200mV 𝛀𝛀
M100ns
CH1
REF1
REFIN
1.90V
Figure 4. Measured phase-continuous FSK transition.
REFIN
Analog Devices offers a variety of direct digital synthesizers, clock
distribution chips, and clock buffers to build a DDS-based clock
generator. Refer to www.analog.com/dds and www.analog.com/
clock for more information.
CLK
(2)
500MSPS
AD9958
DDS CORES
10-BIT
DAC
RECONSTRUCTED
SINE WAVE
10-BIT
DAC
RECONSTRUCTED
SINE WAVE
MODULATION CONTROL
SYSTEM
CLOCK
SOURCE
REF CLOCK
INPUT CIRCUITRY
TIMING AND
CONTROL
USER INTERFACE
Figure 5. AD9958 functional block diagram.
Clock Generator Has 12 LVPECL/24 CMOS Outputs
The AD9520-x clock generator (Figure 6) derives up to 12
LVPECL or 24 CMOS clocks from a single reference frequency.
Integrating a complete PLL with VCO, programmable dividers,
14
DIVIDER
AND MUXES
VCO
ZERO
DELAY
LVPECL/
CMOS
Multichannel, 10-Bit, 500-MSPS Direct Digital Synthesizers
The 2-channel AD9958 (Figure 5) and 4-channel AD9959 direct
digital synthesizers (DDS) include two/four 10-bit, 500-MSPS
current-output DACs. All channels share a common system clock,
providing inherent synchronization; interconnecting multiple
devices enables higher channel counts. Independent control of
each channel’s frequency, phase, and amplitude allows the devices
to correct for system-related mismatches. All parameters can
be swept linearly; or 16 levels can be chosen for FSK, PSK, or
ASK modulation. Output sine wave tuning has 32-bit frequency
resolution, 14-bit phase resolution, and 10-bit amplitude
resolution. Operating with a 1.8-V core supply, plus a 3.3-V I/O
supply for logic compatibility, the AD9958/AD9959 consume
315 mW/540 mW with all channels on, and 13 mW in power-down
mode. Specified from –40°C to +85°C, they are available in 56-lead
LFCSP packages and priced at $20.48/$37.59 in 1000s.
REF2
LF
STATUS
MONITOR
PLL
1
SWITCHOVER
AND MONITOR
CP
DIV/Φ
OUT0
OUT1
OUT2
DIV/Φ
OUT3
OUT4
OUT5
DIV/Φ
OUT6
OUT7
OUT8
DIV/Φ
OUT9
OUT10
OUT11
SPI/I2C CONTROL
PORT AND
DIGITAL LOGIC
EEPROM
AD9520
Figure 6. AD9520 functional block diagram.
Further Reading
1.AN-837 Application Note, DDS-Based Clock Jitter Performance
vs. DAC Reconstruction Filter Performance.
2.Kester, Walt. The Data Conversion Handbook. Analog Devices.
Chapters 6 and 7. 2005.
3.Kester, Walt. High Speed System Applications. Analog Devices.
Chapters 2 and 3. 2006.
4.MT-101 Tutorial, Decoupling Techniques.
5.MT-031 Tutorial, Grounding Data Converters and Solving the
Mystery of AGND and DGND.
Author
David Brandon [[email protected]]
has supported DDS products since the first DDS
released back in 1995. His career spans 28 years
at ADI, with the last 11 years as an applications
engineer in the Clock and Signal Synthesis Group.
He has authored a number of application notes
and a couple of magazine articles.
Analog Dialogue Volume 44 Number 4
High-Side Current Sensing
with Wide Dynamic Range:
Three Solutions
2.Attain an increased level of integration by using a high
voltage bidirectional current shunt monitor, such as the
AD8210, with additional external components to extend
dynamic range and accuracy.
3. Employ an application-optimized device such as the newly
available AD8217, an easy-to-use, highly integrated zerodrift current sensor with an input common-mode voltage
range of 4.5 V to 80 V.
By Neil Zhao, Wenshuai Liao, and Henri Sino
Introduction
Current sensing is a critical function, necessary for precision
closed-loop control in applications such as motor control, solenoid
control, communications infrastructure, and power management.
End uses range from safety-critical automotive and industrial
applications to handheld devices, where power and efficiency are
essential. Precision current monitoring allows designers to obtain
critical instantaneous information, such as motor torque—based on
motor current—efficiency of a dc-to-dc converter, or bias current in
a base station’s LDMOS (laterally diffused MOS) power transistor,
or diagnostic information, such as shorts to ground.
Configuring a Standard Op Amp for High-Side Current Sensing
Figure 1 shows an op-amp-based discrete solution employing
the AD8628. The same setup is valid with other op amps as
well, but specific features that are required include low input
offset voltage, low offset voltage drift, low input bias current,
and rail-to-rail input- and output swing capability, if possible.
Other recommended amplifiers include the AD8538, AD8571,
and AD8551.
BUS VOLTAGE
To understand the key trade-offs, options, and challenges faced by
system designers when choosing the most accurate, cost-effective
current sensor for a circuit board, we take a close look at current
sensing in LDMOS bias current monitoring in cellular base station
power amplifiers and other relevant applications.
Cur rent monitor ing is necessar y in base station power
amplifiers, especially with the more complex modulation
methods used in 3G and LTE, where the peak-to-average power
ratio varies from 3.5 dB (about 2.2 to 1) for 3G W-CDMA to
8.5 dB (about 7.1 to 1) for LTE OFDM—compared with 3 dB
(about 2 to 1) for the most popular 2G single-carrier GSM. One of
the control-loop functions is to monitor the LDMOS bias current,
which allows the bias of the LDMOS to be properly modulated
for a given power output. Typically, this dc bias current has a
wide dynamic range based on operating, maximum, or off-peak
operation. To the designer, this means that an accurate current
sensor is needed to monitor a current that can range from 50 mA
(or as little as 15 mA)1 to 20 A, while the drain of the LDMOS is
biased at a high voltage ranging from 28 V to 60 V. Using a shunt
resistor to monitor this current means that the designer is limited
to a very small shunt that will not dissipate too much power when
the LDMOS current is 20 A. As an example, even a 10 mΩ shunt
will dissipate 4 W at maximum current.
While shunt resistors are available to handle this power, lower
power dissipation might be a requirement of the board. But
the choice of such low resistance values means that at low
currents, say 50 mA, the voltage across the 10 mΩ shunt is
extremely small (500 μV), making it a challenge to monitor
accurately—with a circuit that must also withstand high
common-mode voltage.
This article will focus on providing current-sensing solutions that
can help designers accurately monitor wide-ranging dc currents in
the presence of high common-mode voltages. Special attention will
also be devoted to temperature performance, a critical parameter
that is often not easy to calibrate but must be faced in the case of
outdoor power amplifiers. Three optional solution approaches are
described here—in order of decreasing design complexity—that
provide viable high-accuracy, high-resolution current sensing for
a variety of applications.
1. Use discrete components, such as op amps, resistors, and
Zener diodes, to build a current sensor. This solution employs
the AD8628 zero-drift amplifier as its key component.
1
According to Antenna Interface Standards Group (AISG) 1.1.
Analog Dialogue Volume 44 Number 4
RG
I
RSHUNT
R1
V+
AD8628
V–
LOAD
RBIAS
OUTPUT
RL
Figure 1. Discrete high-current sensing solution using an
operational amplifier.
This circuit monitors the high-side current, I. The amplifier
is biased on via the Zener diode, which, in this case, is rated at
5.1 V. Its use ensures that the amplifier operates safely at the
high common-mode level and that its supply voltage remains
steady and within the allowed supply limits, while the output
is converted to a current by the MOSFET—and to a groundreferenced voltage by resistor R L . The output voltage can thus
feed converters, analog processors, and other ground-referenced
components—such as op amps or comparators—for further
signal conditioning.
In this configuration, the voltage across RG is equal to the
voltage across R SHUNT because feedback via the MOSFET
maintains both high-impedance op-amp inputs at the same
voltage. The current through RG flows through the FET and
R L to develop VOUTPUT. The relationship between the current,
I, flowing through the shunt resistor and VOUTPUT is expressed
by Equation 1:
(1)
R SHUNT Selection: The maximum value of RSHUNT is limited
by the allowable power consumption at maximum current. The
minimum value of RSHUNT is limited by the input range and error
budget of the op amp. Normally, the value of RSHUNT is from 1 mΩ
to 10 mΩ for monitoring currents greater than 10 A. If a single
resistor cannot meet the power consumption requirement or is too
large for the PCB, RSHUNT may have to be made up of multiple
resistors in parallel.
15
RG Selection: RG is used to translate a current proportional to
the high-side current to the low-side. The maximum RG is limited
by the drain-source leakage current of the P-channel MOSFET.
For example, consider the common P-channel enhancement-mode
vertical DMOS transistor BSS84. The maximum IDSS at various
conditions is shown in Table 1.
accuracy over temperature, especially when the monitored current
is low. The drift of the Zener diode will impact the stability of the
amplifier’s power supply, so the amplifier used should have high
power-supply rejection (PSR).
Conditions
Maximum IDSS
VGS = 0 V; V DS = –40 V; T J = 25°C
–100 nA
Further, designers must take into account the low power efficiency
of this solution: A lot of power is consumed by R BIAS. For example,
if the bus common-mode voltage is 28 V, the Zener-diode voltage
output is 5.1 V, and R BIAS is a 1000-Ω resistor, the circuit will
dissipate more than 0.52 W of undesirable power. This adds to the
power-consumption budget and must be taken into account.
VGS = 0 V; V DS = –50 V; T J = 25°C
–10 µA
High-Side Current Sensing with the AD8210 and External Components
Table 1. Drain-Source Leakage Current
VGS = 0 V; V DS = –50 V; T J = 125°C –60 µA
Consider the LDMOS drain current monitoring example, with
28-V common mode and IDSS of 100 nA. The mirror of the
minimum current through R L should be at least 20 times IDSS .
This results in
Figure 2a shows a simplified block diagram of the AD8210
integrated high-voltage bidirectional current shunt monitor;
Figure 2b shows a unidirectional application with an external
voltage reference.
ISHUNT
RSHUNT
The minimum RG is limited by allowable mirror current power
consumption at maximum load current
VS
R2
R1
AD8210
A1
Q1
R BIAS Selection: The current through R BIAS divides to produce
the op amp’s quiescent current and the essentially constant Zenerdiode voltage, VZ , (which determines the op amp’s supply voltage).
Make sure the current flowing through the Zener diode does not
exceed its maximum regulated current, IZ_MAX, when the amplifier
current, ISUPPLY, is essentially zero and V IN is the maximum:
VREF1
Q2
VOUT = (ISHUNT ∙ RSHUNT) ∙ 20
1
+
(V
+ VREF2)
2 REF1
A2
R4
R3
VREF2
GND
(a)
I
To ensure a stable diode voltage, the current flowing through it
should be higher than its minimum operating current, IZ_ MIN,
when ISUPPLY is the maximum and V IN is the minimum:
LOAD
AD8210
–IN
GND
VREF2
NC
The Zener diode and R BIAS are the key components in this solution
because they take away the high common-mode voltage from the
following circuits and allow the use of a low-voltage precision op
amp. For best voltage stability, the Zener diode should have low
dynamic resistance and low temperature drift.
R1 Selection: R1 is used to limit the amplifier input current if
input transients exceed the op amp’s power supply voltage. A
10-kΩ resistor is recommended.
The offset voltage, VOS , and offset current, IOS , of the chosen op
amp are critical, especially with low values of shunt resistance
and at low load currents. VOS + IOS × R1 must be smaller than
I MIN × R SHUNT, or the amplifier may be saturated. Therefore,
a rail-to-rail input amplifier with zero crossover distortion is
preferred for optimal performance.
Another issue to consider for this discrete solution is temperature
drift. Even if an amplifier with zero drift is used, it is very difficult
or costly to optimize drifts caused by discrete components:
the Zener diode, MOSFET, and resistors. From Table 1, the
maximum IDSS of the MOSFET changes from –10 μA to –60 μA
as the operating temperature changes from 25°C to 125°C with
VGS = 0 V and V DS = –50 V. This drift will degrade the system
16
BUS
VOLTAGE
RSHUNT
+IN
5V
5V
5V
VREF1
AD8603
+
V+
OUT
+
–
VOUT
R1
20k𝛀𝛀
R2
0.5k𝛀𝛀
(b)
Figure 2. (a) AD8210 high-voltage bidirectional current
shunt monitor. (b) Wide-range unidirectional application
with an external reference.
The AD8210 amplifies a small differential input voltage generated
by a positive or negative current flowing through the shunt resistor.
The AD8210 rejects high common-mode voltages (up to 65 V)
and provides a ground-referenced buffered output.
As shown in Figure 2a, it comprises two main blocks, a differential
amplifier, and an instrumentation amplifier. The input terminals
are connected to differential amplifier A1 via R1 and R 2. A1 nulls
the voltage appearing across its own input terminals by adjusting
the small currents through R1 and R 2 with Q1 and Q2. When
the input signal to the AD8210 is 0 V, the currents in R1 and R 2
are equal. When the differential signal is nonzero, the current
increases through one of the resistors and decreases in the other.
The current difference is proportional to the size and polarity of
the input signal.
Analog Dialogue Volume 44 Number 4
The differential currents through Q1 and Q2 are converted
into a differential voltage by R 3 and R4. A2 is configured as an
instrumentation amplifier. The differential voltage is converted
into a single-ended output voltage by A2. The gain is internally
set to 20 V/V with precision-trimmed, thin film resistors.
The output reference voltage is easily adjusted using the V REF1 and
V REF2 pins. In a typical configuration for handling bidirectional
current flow, V REF1 is connected to VCC while V REF2 is connected
to GND. In this case, the output is centered at VCC/2 when the
input signal is 0 V, so with a 5-V supply the output is centered at
2.5 V. The output will be greater or less than 2.5 V, depending on
the direction of the current through the shunt resistor.
This configuration works well for charge/discharge applications,
but if the user needs to utilize the entire output range to measure
a unidirectional current flow, then the circuit of Figure 2b shows
a typical way that an external source may be used to set the range.
Here a resistive divider is buffered by an op amp to drive the V REF1
and V REF2 pins, connected together, to offset the output.
a maximum of 8 µV/°C and 20 ppm/°C, respectively. And if, for
example, the AD8603 were used as the buffer, it would contribute
an offset of only 1 µV/°C, which can be neglected compared
with the AD8210’s already low offset-voltage drift. The power
consumption of the divider, R1 and R 2, is
or only 1.2 mW, using the parameters in Figure 2b.
High-Side Current Monitoring Using the Zero-Drift AD8217
Recently, Analog Devices introduced the AD8217, a high-voltage
current sensor featuring zero drift and 500-kHz bandwidth,
designed specifically to enhance resolution and accuracy over
wide temperature-, input common-mode-, and differential voltage
ranges. Figure 3a shows a simplified block diagram of the device;
Figure 3b shows it in a typical application.
It is difficult for the amplifier alone to monitor the load current as
it comes close to zero. With a 5-V supply, the AD8210 specifies
a linear output range with a minimum output of 50 mV and a
maximum output of 4.9 V. Consider an application where the
shunt resistance is 10 mΩ. The minimum current through it must
be greater than 250 mA to ensure that the output of the AD8210
is above its lowest point of 50 mV.
R4
AD8217
–IN
+IN
R1
OUT
R2
R3
LDO
GND
(a)
BUS
VOLTAGE
The configuration shown in Figure 2b adds an offset to allow
smaller currents to be measured. The relationship between output
voltage and monitor current, based on an amplifier gain of 20 V/V,
can be calculated as Equation 2:
In fact, using this configuration, the designer can offset the
output of the AD8210 to any point within its supply range in
order to handle arbitrary current ranges having any degree of
asymmetry. An op amp to buffer the voltage divider is desirable
because precision-trimmed resistances are connected internally
to the reference inputs—so, for best results, those inputs should
be driven at low impedance. Precision and low cost op amps
that can be used to buffer the external reference include—for
example—the AD8541, AD8601, AD8603, AD8605, AD8613,
AD8691, and AD8655.
Compared with the discrete solution, this integrated solution
requires that the current shunt monitor have high common-mode
voltage range and an output offset—if the output voltage range
cannot meet the current-detection range requirements. But it
can handle bidirectional current monitoring, and it avoids the
temperature drift and power consumption issues described above.
The offset drift and gain drift of the AD8210 are guaranteed to be
Analog Dialogue Volume 44 Number 4
+IN
–IN
NC
NC
NC
NC
GND
OUT
LOAD
+
–
VOUT
(b)
For example, using resistances, R1 and R2, of 9800 Ω and 200 Ω,
respectively, the offset voltage will be 100 mV. When the differential
input is 0 V, the AD8210 output will now be 100 mV, safely in the
linear range. If the range of shunt current is from 50 mA to 20 A,
with R SHUNT = 10 mΩ, the input range would be 0.5 mV to
200 mV; and the output range of the AD8210 is 10 mV to 4 V
plus the offset voltage, or 0.11 V to 4.1 V, well within the AD8210’s
specified linear range.
RSHUNT
AD8217
(2)
I
4.5V TO 80V
Figure 3. (a) The AD8217 high-resolution, zero-drift
current-shunt monitor. (b) High-side current sensing
using the AD8217.
For measuring very small currents through a small shunt resistor,
the AD8217 features a minimum 20-mV output range over
temperature, an improvement over the AD8210’s 50-mV range.
Thus, if the minimum load current being monitored across the
shunt produces 20-mV minimum output, which is 1-mV minimum
input, from the current sensor, then the user can choose the
AD8217 configured as shown in Figure 3b. The relationship
between the output voltage of the AD8217 and the input current
can be calculated as Equation 3:
(3)
The AD8217 features an internal low-dropout regulator (LDO),
which provides a constant-voltage supply to the amplifier. The
LDO withstands the high common-mode voltage that can vary
from 4.5 V to 80 V, essentially performing a similar function to
that of the Zener diode in Figure 1.
The AD8217 features a factory-set gain of 20 V/V, with a maximum
±0.35% gain error over the entire temperature range. The initial
offset, specified ±300 µV over temperature, and the miniscule
temperature drift, ±100 nV/°C, will enhance any error budget.
The buffered output voltage directly interfaces with any typical
analog-to-digital converter. Regardless of the common mode,
17
the AD8217 provides the correct output voltage when the input
differential is at least 1 mV. Using a 10-mΩ shunt resistor, as above,
the minimum current can be as low as 100 mA.
The single-chip solution avoids the temperature drift and power
consumption issues of the discrete solution.
Figure 6 is a linearity plot of the output voltage of the AD8217 vs.
low-end values of input current flowing through R SHUNT in the
circuit of Figure 3b. RSHUNT is 10 mΩ, and the load resistance is
50 Ω. The maximum relative error is 0.088%, and the average is
0.025% after linear correction.
140
Performance Results Compared
Figure 4 is a linearity plot of the output voltage across R L as
a function of low-end values of input current flowing through
R SHUNT, measured using the circuit of Figure 1. RSHUNT is 10 mΩ;
RG is 13 Ω; R BIAS is 100 Ω; R1 is 10 kΩ; load resistance is 200 Ω; R L
is 200 Ω; the Zener diode output is 5.1 V; the op amp is AD8628;
the MOSFET is BSS84. The maximum relative error is 0.69%,
and the average is 0.21% after calibration.
19
VOUT (mV)
13
11
7
AD8628
BEST-FIT STRAIGHT LINE
OF THE RESULTS (AD8628)
0
50
100
150
200
IIN (mA)
Figure 4. Low-current test result with AD8628 in Figure 1.
Figure 5 is a linearity plot of the output voltage of the AD8210
as a function of low-end values of input current flowing through
R SHUNT, measured using the circuit of Figure 2b. R SHUNT is
10 mΩ; R1 is 20 kΩ; R2 is 0.5 kΩ; the load resistance is 200 Ω. The
external reference buffer is an AD8603. The maximum relative
error is 0.03%, and the average is 0.01% after calibration.
160
y = 0.2109x + 123.79
R2 = 1
155
40
20
100
AD8217
BEST-FIT STRAIGHT LINE
OF THE RESULTS (AD8217)
200
400
500
600
700
IIN (mA)
Table 2. Maximum and Average Error at Different
Temperatures Using the Same Correction Factor
Solution Circuit
–40°C
+25°C
+85°C
AD8210
AD8217
Max error (%) 11.982
2.117
0.271
Avg error (%)
4.929
2.059
0.171
Max error (%) 1.806
0.075
0.103
Avg error (%)
0.228
0.039
0.022
Max error (%) 6.632
3.800
0.918
Avg error (%)
3.498
0.421
AD8628
5.769
If a temperature sensor is available for use in the system,
different correction factors can be used to calibrate the data
at different temperatures, but with increased component and
manufacturing cost. Table 3 shows the maximum relative error
and average error when different correction factors are used at
–40°C, +25°C, and +85°C.
145
Solution Circuit
140
–40°C
135
+25°C
130
125
300
Table 3. Maximum and Average Error at Different
Temperatures Using Different Correction Factors
150
VOUT (mV)
60
Temperature experiments were also done for each solution at
–40°C, +25°C, and +85°C. Table 2 shows the maximum relative
error and average error when using the same correction factor at
+25°C to calibrate the data at –40°C and +85°C.
9
AD8210
BEST-FIT STRAIGHT LINE
OF THE RESULTS (AD8210)
0
50
100
150
200
IIN (mA)
Figure 5. Low-current test result with AD8210 in Figure 2b.
18
80
Note that the tests were necessarily concentrated at the low end
of the range and did not cover the full 50 mA to 20 A range. The
reason is that the linearity challenge is mainly at the low output
voltage (low unipolar current) portion of the range.
15
5
100
Figure 6. Low-current test result with AD8217 in Figure 3b.
y = 0.1019x + 2.2726
R2 = 1
17
y = 0.2083x – 0.2418
R2 = 1
112
VOUT (mV)
The following section will display test results obtained comparing
the three different methods. The input current through the
shunt was adjusted by changing both the input voltage and load
resistance. In the data, initial calibration has been performed in
order to remove the initial gain and offset errors associated with
all the parts used in our board.
+85°C
AD8628
AD8210
AD8217
Max error (%) 1.981
0.022
0.114
Avg error (%)
0.303
0.009
0.023
Max error (%) 1.806
0.075
0.103
Avg error (%)
0.228
0.039
0.022
Max error (%) 1.844
0.038
0.075
Avg error (%)
0.013
0.020
0.241
The temperature experiments show the high accuracy available
over wide temperature ranges with devices using auto-zero
technology, especially in the case of the AD8217.
Analog Dialogue Volume 44 Number 4
Conclusion
20
18
VOUT (mV)
16
14
12
10
8
–40°C
+25°C
+85°C
6
4
20
40
60
80
100
120
140
160
IIN (mA)
Figure 7. Temperature experiment using discrete solution
with AD8628.
170
VOUT (mV)
Acknowledgment
Mr. Ryan Du contributed to the discrete solution setup
and measurement portion of this article while interning at
Analog Devices.
150
140
Authors
130
–40°C
+25°C
+85°C
120
20
70
120
170
IIN (mA)
Figure 8. Temperature experiment using integrated solution
with AD8210.
140
Neil Zhao ([email protected]) is an
applications engineer in ADI’s Micromachined
Products Division in Beijing, China. Previously,
he w a s a f ield appl ic at ion s e n g i ne e r i n
A DI’s China Applications Suppor t Team,
where he had been working for nearly three
years. Neil graduated in January 2008 from
Beihang University with a master’s degree in
communication and information systems.
Wenshuai Liao ([email protected])
is a marketing engineer in ADI’s Integrated
Amplifier Products (IAP) Group in Beijing,
China. After earning a master’s degree in optical
engineering from Tsinghua University, Wenshuai
spent three years as a TD-SCDMA Node B RF
engineer at Datang Telecommunications Group.
He joined ADI in 2002.
120
100
VOUT (mV)
Based on these test results, the AD8217 solution is the most
suitable of the three choices for wide dynamic range unidirectional
high-side current sensing and monitoring. We also note that an
AD8210 solution offers operation right down to 0-V inputs—which
could be beneficial for short-to-ground conditions. Note also that
the AD8210 is capable of single-chip monitoring of bipolar current
flow, as in charge/discharge applications.
Calibration and temperature sensing are recommended in practical
system designs requiring best system performance.
160
80
60
40
20
100
The test results show that all three solutions can be used for wide
dynamic range high-side current sensing: the outputs are linear in
all three solutions, while the solution using the AD8217 leads to the
best error performance without the necessity of a standalone power
supply. The ±100-nV/°C offset drift characteristic also makes it
ideal for the most accurate performance throughout the –40°C
to +125°C temperature range. From a system design perspective,
the single-chip solution can save PCB area, ease PCB layout,
decrease system cost, and improve reliability. These findings apply
particularly to unidirectional current sensing applications, where
the load-current range is wide and dynamic range is critical.
–40°C
+25°C
+85°C
200
300
400
500
600
700
IIN (mA)
Figure 9. Temperature experiment using single-chip solution
with AD8217.
Analog Dialogue Volume 44 Number 4
Hen r i Si no [ hen r i.si [email protected] a na log.com] i s
an applications engineer in the Integrated
Amplifier Products (IAP) Group at Analog
Devices in Wilmington, MA. Henri has been
with ADI for six years, since obtaining a BSEE
from Worcester Polytechnic Institute. During
his time at ADI, Henri has mainly focused on
supporting products and customers relating to
the automotive and communications markets.
19
Antenna arrays and filters are often tuned by varying the voltage
on a barium strontium titanate (BST) capacitor. When this
ferroelectric material is used in capacitors, an applied voltage
causes a small variation in the crystal structure, which changes the
dielectric constant and, therefore, the capacitance. Electronically
tunable BST capacitors can handle higher power and larger signal
amplitude than the conventional varactor diodes.
In typical applications, the tuning capacitor compensates for
component tolerance, adjusts the cutoff frequency of a filter, or
matches the network impedance of a tunable antenna. The BST
capacitor is tuned by applying a voltage between 0 V and 30 V.
As power supplies in modern electronic devices trend toward
lower voltages, 3.3-V, 2.5-V, or even 1.8-V supplies are common,
especially in battery-powered applications. Despite the benefits
of tuning, it does not always make sense to add a separate highvoltage supply for this function alone. Thus, a convenient way to
generate the power supply is required.
In this application, for example, a 3-V power supply is available,
but the BST capacitors require voltages in excess of 20 V for full
control. The two main circuit blocks are the ADP1613 step-up
switching converter and the AD5504 high-voltage DAC. The
circuit shown in Figure 1 generates DAC output voltages up
to 30 V. The DAC outputs set the bias voltages for the BST
capacitors, thus adjusting the antenna response.
The AD1613 step-up dc-to-dc switching converter (Figure 4)
integrates a power switch capable of providing an output as
high as 20 V. Higher voltages can be achieved by using external
components. As shown, the ADP1613 generates a 32-V output
from a 3-V input. The ADIsimPower ™ tool1 provides an easy way
for designers to determine the appropriate components based on
the input requirements.
VDD
1.5𝛍𝛍F
22𝛍𝛍F
6.3V
10nF
10pF
VDD
97.6k𝛀𝛀
ADP1613
COMP
SS 8
2
FB
RT 7
3
SD
IN 6
4
GND
SW 5
AD5504 OUTPUT
(CONTROL VOLTAGE)
BST
CAP
110nH
2.2𝛍𝛍F
50V
VDD
0.1𝛍𝛍F
50V
16
14
VLOGIC
VDD
13 R_SEL
140k𝛀𝛀
TO MICROCONTROLLER
10𝛍𝛍F
50V
0.1𝛍𝛍F
50V
AD5504
VDD
1𝛍𝛍F
5.62k𝛀𝛀
BST
CAP
Circuits such as that of Figure 1 can benefit next-generation mobile
phones, which are being pressured by two opposing forces. On
one side is the ever-present requirement to reduce size and power
consumption. On the other is the need to increase performance,
utilizing more frequency bands by inserting more antennas
and radio systems into a smaller volume. Antenna designers
are reaching physical design limits with regard to volume and
efficiency, as decreasing antenna volume decreases efficiency.
Tunable antennas solve this problem in multiband, multimode
phones and can extend the operating frequency range of a cell
phone, switching from US GSM850 to European GSM900,
for example, while maintaining size and efficiency. In multiuse
devices, different head and hand positions used while texting,
talking, or surfing the Web present the antenna with different load
impedances, detuning the antenna and decreasing signal quality. A
tunable impedance matching network can adapt for these varying
conditions and recover the detuned signals.
SI4346DY-T1-E3
G
S
DC
BLOCK
4.62nH
Figure 2. BST capacitor equivalent circuit.
1𝛍𝛍F
50V
D
ANTENNA
DC
BLOCK
RF SIGNAL
1N5819HW-7-F
10𝛀𝛀
1
Figure 2 shows the equivalent circuit of a BST capacitor used as
a tunable matching network. Figure 3 shows the transfer function
of the BST capacitance vs. voltage and the antenna response. BST
capacitors can be obtained from suppliers such as Agile RF.2
+
By Ken Kavanagh
The 32-V output from the ADP1613 powers the AD5504 quad,
12-bit, high-voltage DAC (Figure 5), which can provide up to
60 V on each of its four outputs. The voltage on the R_SEL pin
determines its full-scale output. In this application, R_SEL is
connected to V DD, setting the full-scale output to 30 V. The DAC
registers are updated via the 3-V compatible serial interface. All
four DACs are updated simultaneously by pulsing the load pin
(LDAC) low, thus allowing four BST capacitances to be changed
at the same time.
+
Boost Supply and High-Voltage
DAC Provide Tuning Signal for
Antennas and Filters
1
CLR
2
SYNC
3
SCLK
4
SDI
5
SDO
6
LDAC
VOUTA 12
0V TO 30V
VOUTB 11
0V TO 30V
VOUTC 10
0V TO 30V
VOUTD 9
0V TO 30V
DGND
AGND
16
14
Figure 1. Boost supply and high-voltage DAC provide tuning signal for BST capacitors.
20
Analog Dialogue Volume 44 Number 4
Step-Up DC-to-DC Switching Converter Operates at 650 kHz/1300 kHz
The ADP1613 step-up converter is capable of supplying over
150 mA at voltages as high as 20 V, while operating with a
single 2.5-V to 5.5-V supply. Integrating a 2-A, 0.13-Ω power
switch with a current-mode, pulse-width modulated regulator,
its output varies less than 1% with changes in input voltage,
load current, and temperature. The operating frequency is pinselectable and can be optimized for high efficiency or minimum
external component size: at 650 kHz it provides 90% efficiency;
at 1.3 MHz its circuit implementation occupies the smallest
space, making it ideal for space-constrained environments in
portable devices and liquid-crystal displays. The adjustable
soft-start circuit minimizes inrush currents, ensuring safe,
predictable start-up conditions. The ADP1613 consumes
2.2 mA in the switching state, 700 µA in the nonswitching
state, and 10 nA in shutdown mode. Available in an 8-lead
MSOP package, it is specified from – 40°C to +85°C and
priced at $0.70 in 1000s.
Quad, 12-Bit DACs Provide High-Voltage Outputs
T he A D5504 quad, 12-bit, high-voltage DAC has pinselectable output ranges of 0 V to 30 V and 0 V to 60 V.
Fu nct ionally complete, it i ncludes a precision voltage
reference, temperature sensor, four double-buffered DACs,
and four high-voltage amplifiers. Upon power-up, the digital
section is enabled and set to a known state; the analog
section remains disabled until a power-up command is issued
via the SPI port. The temperature sensor disconnects the
analog outputs and sets an alarm f lag if the die temperature
exceeds 110°C. The AD5504 specifies 1-LSB max differential
nonlinearity (DNL) and 3-LSB max integral nonlinearity
(INL) in 30 V mode. Operating on 10-V to 62-V and 2.3-V
to 5.5-V supplies, it consumes 2 mA in normal mode and
30 µA in power-down mode. Available in a 16-lead TSSOP
package, it is specified from – 40°C to +105°C and priced at
$9.92 in 1000s.
References
(Information on all ADI components can be found at www.analog.com.)
1
www.analog.com/adisimpower.
2
www.agilerf.com.
Author
Ken Kavanagh [[email protected]]
is an applications engineer in ADI’s Precision
DAC Group. Currently responsible for providing
applications support for the nanoDAC ® and
denseDAC ® portfolios, Ken has worked in
applications since 1994. He earned a BEng from
the University of Limerick in 1999.
VOLTAGE VARIABLE CAPACITANCE
1.2
–4
1.0
–8
0.8
–12
S11 (dB)
NORMALIZED CAPACITANCE
ANTENNA RESPONSE
0
0.6
–16
–20
0.4
–24
0.2
0
SEE ANTENNA RESPONSE #1
SEE ANTENNA RESPONSE #2
SEE ANTENNA RESPONSE #3
0
5
10
BIAS VOLTAGE (V)
15
ANTENNA RESPONSE #1
ANTENNA RESPONSE #2
ANTENNA RESPONSE #3
–28
20
–32
780
800
820
840
860
880
900
920
940
960
980
FREQUENCY (MHz)
Figure 3. Bias voltage vs. BST capacitance; resulting antenna response.
Analog Dialogue Volume 44 Number 4
21
L1
VIN
>1.6V
CIN
VIN
<0.3V
6
D
COMPARATOR
VOUT
FB
ERROR
AMPLIFIER
PWM
COMPARATOR
SW
COUT
5𝛍𝛍A
S
R
TSD
COMPARATOR
5𝛍𝛍A
SS
VOUT
DREF
UVLOREF
VSS
CCOMP
CURRENT
SENSING
UVLO
COMPARATOR
VIN
1
RCOMP
5
OSCILLATOR
VBG
COMP
A
+
2
R2
D1
+
VIN
R1
FREQ
7
8
CSS
N1
DRIVER
TSENSE
SOFT
START
Q
BG
BAND GAP
RESET
TREF
AGND
ADP1612/AD1613
1.1M𝛀𝛀
EN
3
AGND
GND
>1.6V
4
<0.3V
Figure 4. ADP1613 functional block diagram.
CLR R_SEL VLOGIC
VDD
LDAC
REFERENCE
122.36k𝛀𝛀
SDI
12
SDO
SCLK
SYNC
INPUT
CONTROL
LOGIC
ALARM
POWER-ON
RESET
INPUT
REGISTER
A
DAC
REGISTER
A
INPUT
REGISTER
B
DAC
REGISTER
B
INPUT
REGISTER
C
DAC
REGISTER
C
INPUT
REGISTER
D
DAC
REGISTER
D
12
DAC A
DAC B
DAC C
–
VOUTB
+
–
VOUTC
+
1713k𝛀𝛀
122.36k𝛀𝛀
12
VOUTA
+
1713k𝛀𝛀
122.36k𝛀𝛀
12
–
1713k𝛀𝛀
122.36k𝛀𝛀
12
1713k𝛀𝛀
DAC D
–
VOUTD
+
POWER-DOWN
CONTROL LOGIC
AD5504
TEMPERATURE
SENSOR
DGND
AGND
Figure 5. AD5504 functional block diagram.
22
Analog Dialogue Volume 44 Number 4
High-Performance Difference
Amplifier with Precision
Supply-Referenced Level Shift
By Moshe Gerstenhaber and Michael O’Sullivan
max gain drift, 1000-μV max offset, and 74-dB min commonmode rejection. Both grades specify –110-dB harmonic distortion,
15-MHz bandwidth, and 30-V/μs slew rate. This combination of
speed and precision makes the device ideal for instrumentation
amplifiers, driving ADCs, level shifting, and automatic test
equipment. Operating on a single 5-V to 36-V supply, or dual
±2.5-V to ±18-V supplies, the AD8271 draws 2.3 mA. Available
in a 10-lead MSOP package, it is specified from –40°C to +85°C
and priced from $1.25 in 1000s.
R1
Designed on small-geometry processes, high-performance ADCs
typically run on single 1.8-V to 5-V supplies. When processing
±10-V or larger signals, an amplifier circuit ahead of the ADC can
attenuate the signal to keep it from saturating the ADC inputs. A
difference amplifier (diff amp) is commonly used when the signal
includes a large common-mode voltage.
IN–
IN+
Precision Programmable-Gain Difference Amplifier
The AD8271 low-distortion, programmable-gain difference
amplifier comprises a precision op amp and seven laser-trimmed
gain-setting resistors, enabling user-selectable differential gains
of 0.5, 1, or 2. It can also be configured in over 40 single-ended
configurations, with gains ranging from −2 to +3. Two grades are
available: the B-grade specifies 0.02% max gain error, 2-ppm/°C
max gain drift, 600-μV max offset, and 80-dB min common-mode
rejection; the A-grade specifies 0.05% max gain error, 10-ppm/°C
Analog Dialogue Volume 44 Number 4
VOUT
R4
VCC
R
= VCC/2
R
Figure 1. Single-supply difference amplifier with
midsupply output.
Early diff amps, like many other analog ICs, typically operated on
dual ±5-V to ±15-V supplies. As ADCs and other components
moved to lower supply voltages, for a time the only circuit requiring
dual supplies was the diff amp at the front end. But adding a
negative supply for this single circuit was quite inconvenient.
Figure 2 shows an alternative solution that offers lower cost and
higher performance by using the AD8271 difference amplifier,
with its multiple integrated precision-trimmed resistors. The
on-chip resistors set the device output to midsupply. The resistors
are all manufactured from the same low-drift thin-film material, so
their ratio match over temperature is excellent; they are trimmed
to match the other resistors in the circuit, so they do not degrade
the excellent CMR.
R2
R3
The ability of a diff amp to reject a common-mode voltage is
determined by the ratio match of the gain-setting resistors; the
closer the match, the higher the common-mode rejection (CMR).
For discrete amplifiers with 0.1% external resistors, CMR is limited
to 54 dB. ICs that integrate precision laser-trimmed resistors with
an op amp can achieve CMR better than 80 dB.
New diff amps can operate on single 2.7-V to 15-V supplies, but
the op amp inputs and output would all be pinned to the negative
rail (ground) under some operating conditions. To measure signals
containing negative common-mode voltages, the common-mode
input must be raised off the negative rail. To measure negative
signals, the amplifier output must be raised off the negative rail.
Both of these level shifts can be accomplished by applying a positive
voltage to the reference pin. With a single 5-V supply, for example,
a 2.5-V source on the reference pin sets the output to midsupply
and raises the common-mode voltage seen at the op amp inputs.
The source must be low impedance to avoid degrading the CMR
and low drift to maintain accuracy over temperature. Figure 1
shows a typical solution that uses two external precision resistors
and a low-drift precision op amp.
VCC
AD8271
10k𝛀𝛀
VIN
IN–
10k𝛀𝛀
IN+
10k𝛀𝛀
10k𝛀𝛀
10k𝛀𝛀
VEE
VCC
20k𝛀𝛀
VOUT = (VIN + VCC/2)
20k𝛀𝛀
VCC
Figure 2. AD8271 shifts output to midsupply with
no external components.
Authors
Moshe Gerstenhaber [[email protected]
analog.com] is a Division Fellow at Analog
Devices. He began his career at ADI in 1978
and has held various senior positions over the
years in manufacturing, product engineering,
and product design. Moshe is currently the
design manager of the Integrated Amplifier
Products Group. He has made significant contributions in the
field of amplifier design, especially very-high-precision specialty
amplifiers such as instrumentation and difference amplifiers.
Michael O’Sullivan [[email protected]
analog.com] has worked at Analog Devices since
2004. Currently the product and test engineering
manager of the Integrated Amplifier Products
Group, he supports product characterization
and release of very-high-precision specialty
a mpl i f ie r s suc h a s i n st r u ment at ion a nd
difference amplifiers. Mike worked as a product
engineer in the semiconductor field for over 14 years.
23
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©2011 Analog Devices, Inc. All rights reserved.
Trademarks and registered trademarks are the property
of their respective owners.
M02000444-0-2/11
www.analog.com/analogdialogue
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