SECTION 10 HARDWARE DESIGN TECHNIQUES Walt Kester

SECTION 10 HARDWARE DESIGN TECHNIQUES Walt Kester
HARDWARE DESIGN TECHNIQUES
SECTION 10
HARDWARE DESIGN TECHNIQUES
Walt Kester
RESISTOR AND THERMOCOUPLE ERRORS IN HIGH
ACCURACY SYSTEMS
Walt Kester, Walt Jung, and James Bryant
Resistor accuracy is crucial in precision systems. The circuit element called a
resistor should not be taken for granted! Figure 10.1 shows a simple non-inverting
op amp where the gain of 100 is set by the external resistors R1 and R2. The
temperature coefficients of the two resistors are a somewhat obvious source of error.
Assume that the op amp gain errors are negligible, and that the resistors are
perfectly matched at +25ºC. If the temperature coefficients of the resistors differ by
only 25ppm/ºC, the gain of the amplifier will change by 250ppm for a 10ºC
temperature change. This is about 1 LSB in a 12-bit system, and a major disaster in
a 16-bit system.
RESISTOR TEMPERATURE COEFFICIENT MISMATCHES
CAUSE GAIN VARIATION WITH TEMPERATURE
G=1+
+
R1
= 100
R2
R1 = 9.9kΩ, 1/4 W
_
TC = +25ppm/°c
R2 = 100Ω, 1/4 W
TC = +50ppm/°c
Temperature change of 10°C causes gain change of 250ppm
This is 1LSB in a 12-bit system and a disaster in a 16-bit system
Figure 10.1
Even if the temperature coefficients are identical, there still may be significant
errors. Suppose R1 and R2 have identical temperature coefficients of +25ppm/ºC
and are both ¼W resistors. If the signal input in Figure 10.2 is zero, the resistors
will dissipate no heat, but if it is 100mV there will be 9.9V across R1 which will
dissipate 9.9mW and experience a temperature rise of 1.24ºC (the thermal
resistance of a ¼W resistor is 125ºC/W). The 1.24ºC rise causes a resistance change
10.1
HARDWARE DESIGN TECHNIQUES
of 31ppm, and a corresponding change in gain. R2, with only 100mV across it, is
only heated 0.0125ºC, which is negligible. The 31ppm gain error represents a
fullscale error of ½ LSB at 14-bits, and is a disaster for a 16-bit system.
RESISTOR SELF-HEATING EVEN IN MATCHED RESISTORS
CAN CAUSE GAIN VARIATION WITH INPUT LEVEL
+100mV
G=1+
+
R1
= 100
R2
+10V
R1 = 9.9kΩ, 1/4 W
TC = +25ppm/°c
_
Assume TC of R1 = TC of R2
R2 = 100Ω, 1/4 W
TC = +25ppm/°c
R1, R2 Thermal Resistance = 125°c / W
Temperature of R1 will rise by 1.24°C, PD = 9.9mW
Temperature rise of R2 is negligible, PD = 0.1mW
Gain is altered by 31ppm, or 1/2 LSB @ 14-bits
Figure 10.2
These, and similar errors, are avoided by selecting critical resistors that are
accurately matched for both value and temperature coefficient, and ensuring tight
thermal coupling between resistors whose matching is important. This is best
achieved by using a resistor network on a single substrate - such a network may be
within an IC or may be a separately packaged thin-film resistor network.
Another more subtle problem with resistors is the thermocouple effect, sometimes
referred to as thermal EMF. Wherever there is a junction between two different
conductors there is a thermoelectric voltage. If two junctions are present in a circuit,
we have a thermocouple, and if these two junctions are at different temperatures,
there will be a net voltage in the circuit. This effect is used to measure temperature,
but is a potential source of inaccuracy in low level circuits, since wherever two
different conductors meet, we have a thermocouple, whether we like it or not. This
will cause errors if the various junctions are at different temperatures. The effect is
hard to avoid, even if we are only making connections with copper wire, since a
copper-to-copper junction formed by copper wire from two different manufacturers
may have a thermoelectric voltage of up to 0.2µV/ºC.
Consider the resistor model shown in Figure 10.3. The connections between the
resistor material and the leads form two thermocouple junctions. The thermocouple
EMF can be as high as 400µV/ºC for carbon composition resistors and as low as
10.2
HARDWARE DESIGN TECHNIQUES
0.05µV/ºC for specially constructed resistors (Reference 1). Metal film resistors (RNtypes) are typically about 20µV/ºC.
RESISTORS CONTAIN THERMOCOUPLES
+
T1
RESISTOR
MATERIAL
+
T2
+
RESISTOR LEADS
TYPICAL RESISTOR THERMOCOUPLE EMFs
CARBON COMPOSITION
≈ 400 µV/ °C
METAL FILM
≈ 20 µV/ °C
EVENOHM OR
MANGANIN WIREWOUND
≈ 2 µV/ °C
RCD Components HP-Series
≈ 0.05 µV/ °C
Figure 10.3
These thermocouple effects are unimportant at AC or where the resistor is at a
uniform temperature, but if the dissipation in a resistor, or its orientation with
respect to heat sources, can cause one of its ends to be warmer than the other, then
there will be a net thermocouple voltage differential, which will introduce a DC
error into the circuit. For instance, using ordinary metal film resistors, a
temperature differential of 1ºC will cause a thermocouple voltage of 20µV which is
quite significant when compared to the input offset voltage of truly precision op
amps such as the OP177 or the AD707, and extremely significant when compared to
chopper-stabilized op amps.
Figure 10.4 shows how resistor orientation can make a difference in the net
thermocouple voltage. Standing the resistor on end in order to conserve board space
will invariable cause a temperature gradient across the resistor, especially if it is
dissipating any significant power. Placing the resistor flat on the PC board will
eliminate this problem unless there is airflow across the resistor parallel to its axis.
Orienting the resistor axis perpendicular to the airflow will minimize the error,
since this tends to force the resistor ends towards the same temperature.
Figure 10.5 shows how to orient the resistor on a vertically mounted PC board,
where the convection cooling air currents flow up the board. Again, the thermal axis
of the resistor should be perpendicular to the convection flow to minimize the effect.
10.3
HARDWARE DESIGN TECHNIQUES
Because of their small size, the thermocouple effect in surface mount resistors is
generally less than leaded types because of the tighter thermal coupling between
the ends of the resistor.
AVOIDING THERMAL GRADIENTS MINIMIZES
THERMOCOUPLE ERROR VOLTAGES
∆T
WRONG
RIGHT
Figure 10.4
PROPER ORIENTATION OF SURFACE MOUNT RESISTORS
MINIMIZES THERMOCOUPLE ERROR VOLTAGE
∆T
WRONG
RIGHT
Figure 10.5
10.4
HARDWARE DESIGN TECHNIQUES
A simple circuit shown in Figure 10.6 will further illustrate the parasitic
thermocouple problem. Here, we have a remote bridge driving an instrumentation
amplifier which has current limiting resistors in each lead. Each resistor has four
thermocouples: two are internal to the resistor, and two are formed where the
resistor leads connect to the copper wires. Another pair of thermocouples is formed
where the copper wire connects to the Kovar pins of the in-amp. The Copper/Kovar
junction has a thermocouple voltage of about 35µV/ºC. Most molded plastic ICs use
copper leadframes which would be an order of magnitude or so less (e.g., the AD620
in-amp).
In addition, the copper wire has a resistance temperature coefficient (TC of 30 gage
copper wire is approximately 0.385%/ºC) which can introduce error if the
temperature of the wires is significantly different, or if they are different lengths. In
this example, however, this error is negligible because there is minimal current flow
in the wires.
Obviously, this simple circuit must have a good thermal as well as electrical design
in order to maintain microvolt precision. Some good design precautions include
minimizing number of thermocouple junctions, minimizing thermal gradients by
proper layout or blocking airflow to critical devices using metallic or plastic shields,
minimizing power dissipation in sensitive devices, proper selection of precision
resistors, and matching the number of junctions in each half of a differential signal
path by adding "dummy" components if required. Sockets, connectors, switches, or
relays in the critical signal path can introduce unstable contact resistances as well
as "unknown" thermocouple junctions which may not track to the required accuracy.
They should be avoided if possible.
PARASITIC THERMOCOUPLES IN SIMPLE CIRCUIT
THERMOCOUPLE DIFFERENTIAL EMF AT LEAD JUNCTIONS
≈ 35µV/ °C • ∆T
Cu
Cu
+
REMOTE
BRIDGE
IN-AMP
Cu
Cu
–
RESISTOR-TO-LEAD
CONNECTIONS
KOVAR
PINS
Figure 10.6
10.5
HARDWARE DESIGN TECHNIQUES
REFERENCES: RESISTOR AND THERMOELECTRIC
ERRORS
1.
RCD Components, Inc., 520 E. Industrial Park Drive, Manchester NH,
03109, 603-669-0054, http://www.rcd-comp.com.
2.
Steve Sockolov and James Wong, High-Accuracy Analog Needs More
Than Op Amps, Electronic Design, Oct.1, 1992, p.53.
3.
Doug Grant and Scott Wurcer, Avoiding Passive Component Pitfalls,
The Best of analog Dialogue, Analog Devices, 1991, p. 143.
4.
Brian Kerridge, Elegant Architectures Yield Precision Resistors,
EDN, July 20, 1992.
10.6
HARDWARE DESIGN TECHNIQUES
GROUNDING IN MIXED SIGNAL SYSTEMS
Walt Kester, James Bryant
Today's signal processing systems generally require mixed-signal devices such as
analog-to-digital converters (ADCs) and digital-to-analog converters (DACs) as well
as fast digital signal processors (DSPs). Requirements for processing analog signals
having wide dynamic ranges increases the importance of high performance ADCs
and DACs. Maintaining wide dynamic range with low noise in hostile digital
environments is dependent upon using good high-speed circuit design techniques
including proper signal routing, decoupling, and grounding.
In the past, "high precision, low-speed" circuits have generally been viewed
differently than so-called "high-speed" circuits. With respect to ADCs and DACs, the
sampling (or update) frequency has generally been used as the distinguishing speed
criteria. However, the following two examples show that in practice, most of today's
signal processing ICs are really "high-speed," and must therefore be treated as such
in order to maintain high performance. This is certainly true of DSPs, and also true
of ADCs and DACs.
All sampling ADCs (ADCs with an internal sample-and-hold circuit) suitable for
signal processing applications operate with relatively high speed clocks with fast
rise and fall times (generally a few nanoseconds) and must be treated as high speed
devices, even though throughput rates may appear low. For example, the 12-bit
AD7892 successive approximation (SAR) ADC operates on an 8MHz internal clock,
while the sampling rate is only 600kSPS.
Sigma-delta (Σ-∆) ADCs also require high speed clocks because of their high
oversampling ratios. The AD7722 16-bit ADC has an output data rate (effective
sampling rate) of 195kSPS, but actually samples the input signal at 12.5MSPS (64times oversampling). Even high resolution, so-called "low frequency" Σ-∆ industrial
measurement ADCs (having throughputs of 10Hz to 7.5kHz) operate on 5MHz or
higher clocks and offer resolution to 24-bits (for example, the Analog Devices
AD7730 and AD7731).
To further complicate the issue, mixed-signal ICs have both analog and digital
ports, and because of this, much confusion has resulted with respect to proper
grounding techniques. Digital and analog design engineers tend to view these
devices from different perspectives, and the purpose of this section is to develop a
general grounding philosophy that will work for most mixed signal devices, without
having to know the specific details of their internal circuits.
Ground and Power Planes
The importance of maintaining a low impedance large area ground plane is critical
to all analog circuits today. The ground plane not only acts as a low impedance
return path for decoupling high frequency currents (caused by fast digital logic) but
10.7
HARDWARE DESIGN TECHNIQUES
also minimizes EMI/RFI emissions. Because of the shielding action of the ground
plane, the circuits susceptibility to external EMI/RFI is also reduced.
Ground planes also allow the transmission of high speed digital or analog signals
using transmission line techniques (microstrip or stripline) where controlled
impedances are required.
The use of "buss wire" is totally unacceptable as a "ground" because of its
impedance at the equivalent frequency of most logic transitions. For instance, #22
gauge wire has about 20nH/inch inductance. A transient current having a slew rate
of 10mA/ns created by a logic signal would develop an unwanted voltage drop of
200mV at this frequency flowing through 1 inch of this wire:
∆v = L
10mA
∆i
= 20 nH ×
= 200mV.
ns
∆t
For a signal having a 2V peak-to-peak range, this translates into an error of about
200mV, or 10% (approximate 3.5-bit accuracy). Even in all-digital circuits, this error
would result in considerable degradation of logic noise margins.
Figure 10.7 shows an illustration of a situation where the digital return current
modulates the analog return current (top figure). The ground return wire
inductance and resistance is shared between the analog and digital circuits, and
this is what causes the interaction and resulting error. A possible solution is to
make the digital return current path flow directly to the GND REF as shown in the
bottom figure. This is the fundamental concept of a "star," or single-point ground
system. Implementing the true single-point ground in a system which contains
multiple high frequency return paths is difficult because the physical length of the
individual return current wires will introduce parasitic resistance and inductance
which can make obtaining a low impedance high frequency ground difficult. In
practice, the current returns must consist of large area ground planes for low
impedance to high frequency currents. Without a low-impedance ground plane, it is
therefore almost impossible to avoid these shared impedances, especially at high
frequencies.
All integrated circuit ground pins should be soldered directly to the low-impedance
ground plane to minimize series inductance and resistance. The use of traditional
IC sockets is not recommended with high-speed devices. The extra inductance and
capacitance of even "low profile" sockets may corrupt the device performance by
introducing unwanted shared paths. If sockets must be used with DIP packages, as
in prototyping, individual "pin sockets" or "cage jacks" may be acceptable. Both
capped and uncapped versions of these pin sockets are available (AMP part
numbers 5-330808-3, and 5-330808-6). They have spring-loaded gold contacts which
make good electrical and mechanical connection to the IC pins. Multiple insertions,
however, may degrade their performance.
10.8
HARDWARE DESIGN TECHNIQUES
DIGITAL CURRENTS FLOWING IN ANALOG
RETURN PATH CREATE ERROR VOLTAGES
ID
IA
+
VD
INCORRECT
+
VA
ANALOG
CIRCUITS
VIN
GND
REF
IA + ID
DIGITAL
CIRCUITS
ID
ID
IA
+
VD
+
VA
GND
REF
VIN
CORRECT
ANALOG
CIRCUITS
DIGITAL
CIRCUITS
IA
ID
Figure 10.7
Power supply pins should be decoupled directly to the ground plane using low
inductance ceramic surface mount capacitors. If through-hole mounted ceramic
capacitors must be used, their leads should be less than 1mm. The ceramic
capacitors should be located as close as possible to the IC power pins. Ferrite beads
may be also required for additional decoupling.
Double-Sided vs. Multilayer Printed Circuit Boards
Each PCB in the system should have at least one complete layer dedicated to the
ground plane. Ideally, a double-sided board should have one side completely
dedicated to ground and the other side for interconnections. In practice, this is not
possible, since some of the ground plane will certainly have to be removed to allow
for signal and power crossovers, vias, and through-holes. Nevertheless, as much
area as possible should be preserved, and at least 75% should remain. After
completing an initial layout, the ground layer should be checked carefully to make
sure there are no isolated ground "islands," because IC ground pins located in a
ground "island" have no current return path to the ground plane. Also, the ground
plane should be checked for "skinny" connections between adjacent large areas
which may significantly reduce the effectiveness of the ground plane. Needless to
say, auto-routing board layout techniques will generally lead to a layout disaster on
a mixed-signal board, so manual intervention is highly recommended.
10.9
HARDWARE DESIGN TECHNIQUES
Systems that are densely packed with surface mount ICs will have a large number
of interconnections; therefore multilayer boards are preferred. This allows a
complete layer to be dedicated to ground. A simple 4-layer board would have
internal ground and power plane layers with the outer two layers used for
interconnections between the surface mount components. Placing the power and
ground planes adjacent to each other provides additional inter-plane capacitance
which helps high frequency decoupling of the power supply.
GROUND PLANES ARE MANDATORY!
■ Use Large Area Ground (and Power) Planes for Low Impedance
Current Return Paths (Must Use at Least a Double-Sided Board!)
■ Double-Sided Boards:
◆ Avoid High-Density Interconnection Crossovers and
Feedthroughs Which Reduce Ground Plane Area
◆ Keep > 75% Board Area on One Side for Ground Plane
■ Multilayer Boards
◆ Dedicate at Least One Layer for the Ground Plane
◆ Dedicate at Least One Layer for the Power Plane
■ Use at Least 30% to 40% of PCB Connector Pins for Ground
■ Continue the Ground Plane on the Backplane Motherboard to
Power Supply Return
Figure 10.8
Multicard Mixed-Signal Systems
The best way of minimizing ground impedance in a multicard system is to use a
"motherboard" PCB as a backplane for interconnections between cards, thus
providing a continuous ground plane to the backplane. The PCB connector should
have at least 30-40% of its pins devoted to ground, and these pins should be
connected to the ground plane on the backplane mother card. To complete the
overall system grounding scheme there are two possibilities:
1. The backplane ground plane can be connected to chassis ground at numerous
points, thereby diffusing the various ground current return paths. This is commonly
referred to as a "multipoint" grounding system and is shown in Figure 10.9.
2. The ground plane can be connected to a single system "star ground" point
(generally at the power supply).
10.10
HARDWARE DESIGN TECHNIQUES
MULTIPOINT GROUND CONCEPT
VA
PCB
VD
VA
GROUND PLANE
PCB
VD
GROUND PLANE
BACKPLANE
GROUND PLANE
CHASSIS
GROUND
POWER
SUPPLIES
VA
VD
Figure 10.9
The first approach is most often used in all-digital systems, but can be used in
mixed-signal systems provided the ground currents due to digital circuits are
sufficiently diffused over a large area. The low ground impedance is maintained all
the way through the PC boards, the backplane, and ultimately the chassis.
However, it is critical that good electrical contact be made where the grounds are
connected to the sheet metal chassis. This requires self-tapping sheet metal screws
or "biting" washers. Special care must be taken where anodized aluminum is used
for the chassis material, since its surface acts as an insulator.
The second approach ("star ground") is often used in high speed mixed-signal
systems having separate analog and digital ground systems and warrants
considerable further discussion.
Separating Analog and Digital Grounds
In mixed-signal systems with large amounts of digital circuitry, it is highly
desirable to physically separate sensitive analog components from noisy digital
components. It may also be beneficial to use separate ground planes for the analog
and the digital circuitry. These planes should not overlap in order to minimize
capacitive coupling between the two. The separate analog and digital ground planes
are continued on the backplane using either motherboard ground planes or "ground
10.11
HARDWARE DESIGN TECHNIQUES
screens" which are made up of a series of wired interconnections between the
connector ground pins. The arrangement shown in Figure 10.10 illustrates that the
two planes are kept separate all the way back to a common system "star" ground,
generally located at the power supplies. The connections between the ground planes,
the power supplies, and the "star" should be made up of multiple bus bars or wide
copper brads for minimum resistance and inductance. The back-to-back Schottky
diodes on each PCB are inserted to prevent accidental DC voltage from developing
between the two ground systems when cards are plugged and unplugged. Schottky
diodes are used because of their low capacitance to prevent coupling between the
analog and digital ground planes. However, Schottky diodes begin to conduct at
about 300mV, so if the total differential peak-to-peak voltage (the sum of the AC
and DC components) between the two ground planes exceeds this value, additional
diodes in series should be used.
SEPARATING ANALOG AND DIGITAL GROUND PLANES
VA
ANALOG
GROUND
PLANE
PCB
VD
VA
DIGITAL
GROUND
PLANE
ANALOG
GROUND
PLANE
D
A
PCB
VD
DIGITAL
GROUND
PLANE
D
A
DIGITAL GROUND PLANE
BACKPLANE
ANALOG GROUND PLANE
POWER
SUPPLIES
SYSTEM
STAR
GROUND
VA
VD
Figure 10.10
Grounding and Decoupling Mixed-Signal ICs
Sensitive analog components such as amplifiers and voltage references are always
referenced and decoupled to the analog ground plane. The ADCs and DACs (and
other mixed-signal ICs) should generally be treated as analog components and also
grounded and decoupled to the analog ground plane. At first glance, this may seem
somewhat contradictory, since a converter has an analog and digital interface and
usually pins designated as analog ground (AGND) and digital ground (DGND). The
diagram shown in Figure 10.11 will help to explain this seeming dilemma.
10.12
HARDWARE DESIGN TECHNIQUES
PROPER GROUNDING OF MIXED-SIGNAL ICs
VA
VD
FERRITE BEAD
A
LP
CSTRAY
ANALOG
CIRCUITS
LP
R
B
CSTRAY
IA
ID
AGND
A
RP
DIGITAL
CIRCUITS DATA
A
RP
D
LP
RP
AIN/
OUT
A
VD
VA
SHORT
CONNECTIONS
BUFFER
GATE OR
REGISTER
DATA
BUS
CIN ≈ 10pF
RP
LP
DGND
A
A = ANALOG GROUND PLANE
VNOISE
D
D = DIGITAL GROUND PLANE
Figure 10.11
Inside an IC that has both analog and digital circuits, such as an ADC or a DAC,
the grounds are usually kept separate to avoid coupling digital signals into the
analog circuits. Figure 10.11 shows a simple model of a converter. There is nothing
the IC designer can do about the wirebond inductance and resistance associated
with connecting the bond pads on the chip to the package pins except to realize it's
there. The rapidly changing digital currents produce a voltage at point B which will
inevitably couple into point A of the analog circuits through the stray capacitance,
CSTRAY. In addition, there is approximately 0.2pF unavoidable stray capacitance
between every pin of the IC package! It's the IC designer's job to make the chip
work in spite of this. However, in order to prevent further coupling, the AGND and
DGND pins should be joined together externally to the analog ground plane with
minimum lead lengths. Any extra impedance in the DGND connection will cause
more digital noise to be developed at point B; it will, in turn, couple more digital
noise into the analog circuit through the stray capacitance. Note that connecting
DGND to the digital ground plane applies VNOISE across the AGND and DGND
pins and invites disaster!
The name "DGND" on an IC tells us that this pin connects to the digital ground of
the IC. This does not imply that this pin must be connected to the digital ground of
the system.
It is true that this arrangement will inject a small amount of digital noise onto the
analog ground plane. These currents should be quite small, and can be minimized
by ensuring that the converter output does not drive a large fanout (they normally
10.13
HARDWARE DESIGN TECHNIQUES
can't, by design). Minimizing the fanout on the converter's digital port will also keep
the converter logic transitions relatively free from ringing and minimize digital
switching currents, and thereby reducing any potential coupling into the analog port
of the converter. The logic supply pin (VD) can be further isolated from the analog
supply by the insertion of a small lossy ferrite bead as shown in Figure 10.11. The
internal digital currents of the converter will return to ground through the VD pin
decoupling capacitor (mounted as close to the converter as possible) and will not
appear in the external ground circuit. These decoupling capacitors should be low
inductance ceramic types, typically between 0.01µF and 0.1µF.
Treat the ADC Digital Outputs with Care
It is always a good idea (as shown in Figure 10.11) to place a buffer register
adjacent to the converter to isolate the converter's digital lines from noise on the
data bus. The register also serves to minimize loading on the digital outputs of the
converter and acts as a Faraday shield between the digital outputs and the data
bus. Even though many converters have three-state outputs/inputs, this isolation
register still represents good design practice.
The series resistors (labeled "R" in Figure 10.11) between the ADC output and the
buffer register input help to minimize the digital transient currents which may
affect converter performance. The resistors isolate the digital output drivers from
the capacitance of the buffer register inputs. In addition, the RC network formed by
the series resistor and the buffer register input capacitance acts as a lowpass filter
to slow down the fast edges.
A typical CMOS gate combined with PCB trace and through-hole will create a load
of approximately 10pF. A logic output slew rate of 1V/ns will produce 10mA of
dynamic current if there is no isolation resistor:
∆I = C
1V
∆v
= 10pF ×
= 10mA .
ns
∆t
A 500Ω series resistors will minimize this output current and result in a rise and
fall time of approximately 11ns when driving the 10pF input capacitance of the
register:
t r = 2.2 × τ = 2.2 × R ⋅ C = 2.2 × 500Ω × 10 pF = 11ns.
TTL registers should be avoided, since they can appreciably add to the dynamic
switching currents because of their higher input capacitance.
The buffer register and other digital circuits should be grounded and decoupled to
the digital ground plane of the PC board. Notice that any noise between the analog
and digital ground plane reduces the noise margin at the converter digital interface.
Since digital noise immunity is of the orders of hundreds or thousands of millivolts,
this is unlikely to matter. The analog ground plane will generally not be very noisy,
but if the noise on the digital ground plane (relative to the analog ground plane)
exceeds a few hundred millivolts, then steps should be taken to reduce the digital
10.14
HARDWARE DESIGN TECHNIQUES
ground plane impedance, thereby maintaining the digital noise margins at an
acceptable level.
Separate power supplies for analog and digital circuits are also highly desirable.
The analog supply should be used to power the converter. If the converter has a pin
designated as a digital supply pin (VD), it should either be powered from a separate
analog supply, or filtered as shown in the diagram. All converter power pins should
be decoupled to the analog ground plane, and all logic circuit power pins should be
decoupled to the digital ground plane as shown in Figure 10.12. If the digital power
supply is relatively quiet, it may be possible to use it to supply analog circuits as
well, but be very cautious.
In some cases it may not be possible to connect VD to the analog supply. Some of the
newer, high speed ICs may have their analog circuits powered by +5V, but the
digital interface powered by +3V to interface to 3V logic. In this case, the +3V pin of
the IC should be decoupled directly to the analog ground plane. It is also advisable
to connect a ferrite bead in series with power trace that connects the pin to the +3V
digital logic supply.
GROUNDING AND DECOUPLING POINTS
FERRITE
BEAD
VA
A
A
VD
VA
VD
A
VA
D
R
ADC
OR
DAC
AMP
A
VA
AGND
A
R
A
A
SAMPLING
CLOCK
GENERATOR
A
TO OTHER
DIGITAL
CIRCUITS
DGND
A
VOLTAGE
REFERENCE
BUFFER
GATE
OR
REGISTER
D
VA
A
A
ANALOG
GROUND PLANE
D
DIGITAL
GROUND PLANE
Figure 10.12
The sampling clock generation circuitry should be treated like analog circuitry and
also be grounded and heavily-decoupled to the analog ground plane. Phase noise on
the sampling clock produces degradation in system SNR as will be discussed
shortly.
10.15
HARDWARE DESIGN TECHNIQUES
The Origins of the Confusion about Mixed-Signal Grounding:
Applying Single-Card Grounding Concepts to Multicard Systems
Most ADC, DAC, and other mixed-signal device data sheets discuss grounding
relative to a single PCB, usually the manufacturer's own evaluation board. This has
been a source of confusion when trying to apply these principles to multicard or
multi-ADC/DAC systems. The recommendation is usually to split the PCB ground
plane into an analog one and a digital one. It is then further recommended that the
AGND and DGND pins of a converter be tied together and that the analog ground
plane and digital ground planes be connected at that same point. This essentially
creates the system "star" ground at the mixed-signal device. While this approach
will generally work in a simple system with a single PCB and single ADC/DAC, it is
not optimum for multicard mixed-signal systems. In systems having several ADCs
or DACs on different PCBs (or on the same PCB, for that matter), the analog and
digital ground planes become connected at several points, creating the possibility of
ground loops and making a single-point "star" ground system impossible. These
ground loops can also occur if there is more than one mixed-signal device on a single
PCB. For these reasons, this grounding approach is not recommended for multicard
systems, and the approach previously discussed should be used.
Sampling Clock Considerations
In a high performance sampled data system a low phase-noise crystal oscillator
should be used to generate the ADC (or DAC) sampling clock because sampling
clock jitter modulates the analog input/output signal and raises the noise and
distortion floor. The sampling clock generator should be isolated from noisy digital
circuits and grounded and decoupled to the analog ground plane, as is true for the
op amp and the ADC.
The effect of sampling clock jitter on ADC Signal-to-Noise Ratio (SNR) is given
approximately by the equation:
 1 
SNR = 20 log10 
,
 2π ft j 
where SNR is the SNR of a perfect ADC of infinite resolution where the only source
of noise is that caused by the RMS sampling clock jitter, tj. Note that f in the above
equation is the analog input frequency. Just working through a simple example, if tj
= 50ps RMS, f = 100kHz, then SNR = 90dB, equivalent to about 15-bits dynamic
range.
It should be noted that tj in the above example is the root-sum-square (RSS) value
of the external clock jitter and the internal ADC clock jitter (called aperture jitter).
However, in most high performance ADCs, the internal aperture jitter is negligible
compared to the jitter on the sampling clock.
Since degradation in SNR is primarily due to external clock jitter, steps must be
taken to ensure the sampling clock is as noise-free as possible and has the lowest
possible phase jitter. This requires that a crystal oscillator be used. There are
several manufacturers of small crystal oscillators with low jitter (less than 5ps
10.16
HARDWARE DESIGN TECHNIQUES
RMS) CMOS compatible outputs. (For example, MF Electronics, 10 Commerce Dr.,
New Rochelle, NY 10801, Tel. 914-576-6570.)
Ideally, the sampling clock crystal oscillator should be referenced to the analog
ground plane in a split-ground system. However, this is not always possible because
of system constraints. In many cases, the sampling clock must be derived from a
higher frequency multi-purpose system clock which is generated on the digital
ground plane. It must then pass from its origin on the digital ground plane to the
ADC on the analog ground plane. Ground noise between the two planes adds
directly to the clock signal and will produce excess jitter. The jitter can cause
degradation in the signal-to-noise ratio and also produce unwanted harmonics. This
can be remedied somewhat by transmitting the sampling clock signal as a
differential signal using either a small RF transformer as shown in Figure 10.13 or
a high speed differential driver and receiver IC. If an active differential driver and
receiver are used, they should be ECL to minimize phase jitter. In a single +5V
supply system, ECL logic can be connected between ground and +5V (PECL), and
the outputs AC coupled into the ADC sampling clock input. In either case, the
original master system clock must be generated from a low phase noise crystal
oscillator.
SAMPLING CLOCK DISTRIBUTION FROM
DIGITAL TO ANALOG GROUND PLANES
DIGITAL GROUND PLANE
VD
VD
LOW PHASE
NOISE
MASTER CLOCK
D
ANALOG GROUND PLANE
SAMPLING
CLOCK
SYSTEM CLOCK
GENERATORS
VD
METHOD 1
D
D
A
VA
VD
DSP OR MICROPROCESSOR
+
SAMPLING
CLOCK
_
METHOD 2
D
SNR = 20 log10
D
1
2π f tj
A
tj = Sampling Clock Jitter
f = Analog Input Frequency
Figure 10.13
10.17
HARDWARE DESIGN TECHNIQUES
Some PC Board Layout Guidelines for Mixed-Signal Systems
It is evident that noise can be minimized by paying attention to the system layout
and preventing different signals from interfering with each other. High level analog
signals should be separated from low level analog signals, and both should be kept
away from digital signals. We have seen elsewhere that in waveform sampling and
reconstruction systems the sampling clock (which is a digital signal) is as
vulnerable to noise as any analog signal, but is as liable to cause noise as any digital
signal, and so must be kept isolated from both analog and digital systems.
The ground plane can act as a shield where sensitive signals cross. Figure 10.14
shows a good layout for a data acquisition board where all sensitive areas are
isolated from each other and signal paths are kept as short as possible. While real
life is rarely as tidy as this, the principle remains a valid one.
ANALOG AND DIGITAL CIRCUITS
SHOULD BE PARTITIONED ON PCB LAYOUT
SAMPLING
CLOCK GENERATOR
REFERENCE
ANALOG
ADC
CONTROL
LOGIC
BUFFER
REGISTER
DEMULTIPLEXER
DIGITAL
FILTER
DSP
OR
µP
AMPLIFIER
POWER
TIMING
CIRCUITS
MULTIPLE
ANALOG GROUNDS
INPUT
DATA
BUS
ADDRESS
BUS
BUFFER
MEMORY
MULTIPLE
GROUNDS
Figure 10.14
There are a number of important points to be considered when making signal and
power connections. First of all a connector is one of the few places in the system
where all signal conductors must run in parallel - it is therefore imperative to
separate them with ground pins (creating a faraday shield) to reduce coupling
between them.
Multiple ground pins are important for another reason: they keep down the ground
impedance at the junction between the board and the backplane. The contact
resistance of a single pin of a PCB connector is quite low (of the order of 10mΩ)
when the board is new - as the board gets older the contact resistance is likely to
10.18
HARDWARE DESIGN TECHNIQUES
rise, and the board's performance may be compromised. It is therefore well
worthwhile to allocate extra PCB connector pins so that there are many ground
connections (perhaps 30-40% of all the pins on the PCB connector should be ground
pins). For similar reasons there should be several pins for each power connection,
although there is no need to have as many as there are ground pins.
Manufacturers of high performance mixed-signal ICs like Analog Devices offer
evaluation boards to assist customers in their initial evaluations and layout. ADC
evaluation boards generally contain an on-board low-jitter sampling clock oscillator,
output registers, and appropriate power and signal connectors. They also may have
additional support circuitry such as the ADC input buffer amplifier and external
reference.
The layout of the evaluation board is optimized in terms of grounding, decoupling,
and signal routing and can be used as a model when laying out the ADC PC board
in the system. The actual layout is usually available from the ADC manufacturer in
the form of computer CAD files (Gerber files).
10.19
HARDWARE DESIGN TECHNIQUES
REFERENCES ON GROUNDING:
1.
William C. Rempfer, Get All the Fast ADC Bits You Pay For,
Electronic Design, Special Analog Issue, June 24, 1996, p.44.
2.
Mark Sauerwald, Keeping Analog Signals Pure in a Hostile Digital
World, Electronic Design, Special Analog Issue, June 24, 1996, p.57.
3.
Jerald Grame and Bonnie Baker, Design Equations Help Optimize
Supply Bypassing for Op Amps, Electronic Design, Special Analog
Issue, June 24, 1996, p.9.
4.
Jerald Grame and Bonnie Baker, Fast Op Amps Demand More Than
a Single-Capacitor Bypass, Electronic Design, Special Analog Issue,
November 18, 1996, p.9.
5.
Walt Kester and James Bryant, Grounding in High Speed Systems,
High Speed Design Techniques, Analog Devices, 1996, Chapter 7, p. 7-27.
6.
Jeffrey S. Pattavina, Bypassing PC Boards: Thumb Your Nose at Rules
of Thumb, EDN, Oct. 22, 1998, p.149.
7.
Henry Ott, Noise Reduction Techniques in Electronic Systems,
Second Edition, New York, John Wiley and Sons, 1988.
8.
Howard W. Johnson and Martin Graham, High-Speed Digital Design,
PTR Prentice Hall, 1993.
9.
Paul Brokaw, An I.C. Amplifier User's Guide to Decoupling, Grounding
and Making Things Go Right for a Change, Application Note,
Analog Devices, Inc., http://www.analog.com.
10.
Walt Kester, A Grounding Philosophy for Mixed-Signal Systems,
Electronic Design Analog Applications Issue, June 23, 1997, p. 29.
11.
Ralph Morrison, Grounding and Shielding Techniques, Fourth Edition,
John Wiley, 1998.
12.
Ralph Morrison, Solving Interference Problems in Electronics,
John Wiley, 1995.
13.
C. D. Motchenbacher and J. A. Connelly, Low Noise Electronic System
Design, John Wiley, 1993.
14.
Crystal Oscillators: MF Electronics, 10 Commerce Drive, New Rochelle,
NY, 10801, 914-576-6570.
10.20
HARDWARE DESIGN TECHNIQUES
POWER SUPPLY NOISE REDUCTION AND
FILTERING
Walt Jung, Walt Kester, Bill Chestnut
Precision analog circuitry has traditionally been powered from well regulated, low
noise linear power supplies. During the last decade however, switching power
supplies have become much more common in electronic systems. As a consequence,
they also are being used for analog supplies. Good reasons for the general popularity
include their high efficiency, low temperature rise, small size, and light weight.
In spite of these benefits, switchers do have drawbacks, most notably high output
noise. This noise generally extends over a broad band of frequencies, resulting in
both conducted and radiated noise, as well as unwanted electric and magnetic
fields. Voltage output noise of switching supplies are short-duration voltage
transients, or spikes. Although the fundamental switching frequency can range
from 20kHz to 1MHz, the spikes can contain frequency components extending to
100MHz or more. While specifying switching supplies in terms of RMS noise is
common vendor practice, as a user you should also specify the peak (or p-p)
amplitudes of the switching spikes, at the output loading of your system.
The following section discusses filter techniques for rendering a switching regulator
output analog ready, that is sufficiently quiet to power precision analog circuitry
with relatively small loss of DC terminal voltage. The filter solutions presented are
generally applicable to all power supply types incorporating switching element(s) in
their energy path. This includes various DC-DC converters as well as popular 5V
(PC type) supplies.
An understanding of the EMI process is necessary to understand the effects of
supply noise on analog circuits and systems. Every interference problem has a
source, a path, and a receptor [Reference 1]. In general, there are three methods for
dealing with interference. First, source emissions can be minimized by proper
layout, pulse-edge rise time control/reduction, filtering, and proper grounding.
Second, radiation and conduction paths should be reduced through shielding and
physical separation. Third, receptor immunity to interference can be improved, via
supply and signal line filtering, impedance level control, impedance balancing, and
utilizing differential techniques to reject undesired common-mode signals. This
section focuses on reducing switching power supply noise with external post filters.
Tools useful for combating high frequency switcher noise are shown by Figure 10.15.
They differ in electrical characteristics as well as practicality towards noise
reduction, and are listed roughly in an order of priorities. Of these tools, L and C
are the most powerful filter elements, and are the most cost-effective, as well as
small in size.
10.21
HARDWARE DESIGN TECHNIQUES
SWITCHING REGULATOR NOISE REDUCTION TOOLS
■ Capacitors
■ Inductors
■ Ferrites
■ Resistors
■ Linear Post Regulation
■ Proper Layout and Grounding Techniques
■ PHYSICAL SEPARATION FROM SENSITIVE
ANALOG CIRCUITS!!
Figure 10.15
Capacitors are probably the single most important filter component for switchers.
There are many different types of capacitors, and an understanding of their
individual characteristics is absolutely mandatory to the design of effective practical
supply filters. There are generally three classes of capacitors useful in 10kHz100MHz filters, broadly distinguished as the generic dielectric types; electrolytic,
organic, film, and ceramic. These can in turn can be further sub-divided. A
thumbnail sketch of capacitor characteristics is shown in the chart of Figure 10.16.
TYPES OF CAPACITORS
Aluminum
Electrolytic
(General
Purpose)
Aluminum
Electrolytic
(Switching
Type)
Tantalum
Electrolytic
OS-CON
Electrolytic
Polyester
(Stacked
Film)
Ceramic
(Multilayer)
Size
100 µF
120 µF
120 µF
100 µF
1 µF
0.1 µF
Rated
Voltage
25 V
25 V
20 V
20 V
400 V
50 V
0.6 Ω @
100 kHz
0.18 Ω @
100 kHz
0.12 Ω @
100 kHz
0.02 Ω @
100 kHz
0.11 Ω @
1 MHz
0.12 Ω @
1 MHz
≅ 100 kHz
≅ 500 kHz
≅ 1 MHz
≅ 1 MHz
≅ 10 MHz
≅ 1 GHz
ESR
Operating
Frequency
(*)
(*) Upper frequency strongly size and package dependent
Figure 10.16
10.22
HARDWARE DESIGN TECHNIQUES
With any dielectric, a major potential filter loss element is ESR (equivalent series
resistance), the net parasitic resistance of the capacitor. ESR provides an ultimate
limit to filter performance, and requires more than casual consideration, because it
can vary both with frequency and temperature in some types. Another capacitor loss
element is ESL (equivalent series inductance). ESL determines the frequency where
the net impedance characteristic switches from capacitive to inductive. This varies
from as low as 10kHz in some electrolytics to as high as 100MHz or more in chip
ceramic types. Both ESR and ESL are minimized when a leadless package is used.
All capacitor types mentioned are available in surface mount packages, preferable
for high speed uses.
The electrolytic family provides an excellent, cost-effective low-frequency filter
component, because of the wide range of values, a high capacitance-to-volume ratio,
and a broad range of working voltages. It includes general purpose aluminum
electrolytic types, available in working voltages from below 10V up to about 500V,
and in size from 1 to several thousand µF (with proportional case sizes). All
electrolytic capacitors are polarized, and thus cannot withstand more than a volt or
so of reverse bias without damage. They also have relatively high leakage currents
(up to tens of µA, and strongly dependent upon design specifics).
A subset of the general electrolytic family includes tantalum types, generally
limited to voltages of 100V or less, with capacitance of up to 500µF [Reference 3]. In
a given size, tantalums exhibit a higher capacitance-to-volume ratios than do
general purpose electrolytics, and have both a higher frequency range and lower
ESR. They are generally more expensive than standard electrolytics, and must be
carefully applied with respect to surge and ripple currents.
A subset of aluminum electrolytic capacitors is the switching type, designed for
handling high pulse currents at frequencies up to several hundred kHz with low
losses [Reference 4]. This capacitor type competes directly with tantalums in high
frequency filtering applications, with the advantage of a broader range of values.
A more specialized high performance aluminum electrolytic capacitor type uses an
organic semiconductor electrolyte [Reference 5]. The OS-CON capacitors feature
appreciably lower ESR and higher frequency range than do other electrolytic types,
with an additional feature of low low-temperature ESR degradation.
Film capacitors are available in a very broad range of values and an array of
dielectrics, including polyester, polycarbonate, polypropylene, and polystyrene.
Because of the low dielectric constant of these films, their volumetric efficiency is
quite low, and a 10µF/50V polyester capacitor (for example) is actually the size of
your hand. Metalized (as opposed to foil) electrodes do help to reduce size, but even
the highest dielectric constant units among film types (polyester, polycarbonate) are
still larger than any electrolytic, even using the thinnest films with the lowest
voltage ratings (50V). Where film types excel is in their low dielectric losses, a factor
which may not necessarily be a practical advantage for filtering switchers. For
example, ESR in film capacitors can be as low as 10mΩ or less, and the behavior of
films generally is very high in terms of Q. In fact, this can cause problems of
spurious resonance in filters, requiring damping components.
10.23
HARDWARE DESIGN TECHNIQUES
Film capacitors using a wound layer-type construction can be inductive. This can
limit their effectiveness for high frequency filtering. Obviously, only non-inductively
made film caps are useful for switching regulator filters. One specific style which is
non-inductive is the stacked-film type, where the capacitor plates are cut as small
overlapping linear sheet sections from a much larger wound drum of dielectric/plate
material. This technique offers the low inductance attractiveness of a plate sheet
style capacitor with conventional leads [see References 4, 5, 6]. Obviously, minimal
lead length should be used for best high frequency effectiveness. Very high current
polycarbonate film types are also available, specifically designed for switching
power supplies, with a variety of low inductance terminations to minimize ESL
[Reference 7].
Dependent upon their electrical and physical size, film capacitors can be useful at
frequencies to well above 10MHz. At the highest frequencies, only stacked film
types should be considered. Some manufacturers are now supplying film types in
leadless surface mount packages, which eliminates the lead length inductance.
Ceramic is often the capacitor material of choice above a few MHz, due to its
compact size, low loss, and availability up to several µF in the high-K dielectric
formulations (X7R and Z5U), at voltage ratings up to 200V [see ceramic families of
Reference 3]. NP0 (also called COG) types use a lower dielectric constant
formulation, and have nominally zero TC, plus a low voltage coefficient (unlike the
less stable high-K types). NP0 types are limited to values of 0.1µF or less, with
0.01µF representing a more practical upper limit.
Multilayer ceramic “chip caps” are very popular for bypassing/ filtering at 10MHz or
higher, simply because their very low inductance design allows near optimum RF
bypassing. For smaller values, ceramic chip caps have an operating frequency range
to 1GHz. For high frequency applications, a useful selection can be ensured by
selecting a value which has a self-resonant frequency above the highest frequency of
interest.
All capacitors have some finite ESR. In some cases, the ESR may actually be helpful
in reducing resonance peaks in filters, by supplying “free” damping. For example, in
most electrolytic types, a nominally flat broad series resonance region can be noted
by the impedance vs. frequency plot. This occurs where |Z| falls to a minimum
level, nominally equal to the capacitor’s ESR at that frequency. This low Q
resonance can generally cover a relatively wide frequency range of several octaves.
Contrasted to the very high Q sharp resonances of film and ceramic caps, the low Q
behavior of electrolytics can be useful in controlling resonant peaks.
In most electrolytic capacitors, ESR degrades noticeably at low temperature, by as
much as a factor of 4-6 times at –55°C vs. the room temperature value. For circuits
where ESR is critical to performance, this can lead to problems. Some specific
electrolytic types do address this problem, for example within the HFQ switching
types, the –10°C ESR at 100kHz is no more than 2× that at room temperature. The
OSCON electrolytics have a ESR vs. temperature characteristic which is relatively
flat.
10.24
HARDWARE DESIGN TECHNIQUES
As noted, all real capacitors have parasitic elements which limit their performance.
The equivalent electrical network representing a real capacitor models both ESR
and ESL as well as the basic capacitance, plus some shunt resistance (see Figure
10.17). In such a practical capacitor, at low frequencies the net impedance is almost
purely capacitive. At intermediate frequencies, the net impedance is determined by
ESR, for example about 0.12Ω to 0.4Ω at 125kHz, for several types. Above about
1MHz these capacitor types become inductive, with impedance dominated by the
effect of ESL. All electrolytics will display impedance curves similar in general
shape to that of Figure 10.18. The minimum impedance will vary with the ESR, and
the inductive region will vary with ESL (which in turn is strongly effected by
package style).
CAPACITOR EQUIVALENT CIRCUIT
AND PULSE RESPONSE
i
IPEAK = 1A
INPUT
CURRENT
v
ESR = 0.2Ω
Ω
di
1A
=
dt 100ns
Equivalent f = 3.5MHz
0
ESL = 20nH
di
dt
+ ESR • IPEAK = 400 m V
VPEAK = ESL •
C = 100µF
OUTPUT
VOLTAGE
XC = 0.0005Ω
Ω
@ 3.5MHz
ESR • IPEAK = 200mV
0
Figure 10.17
Regarding inductors, Ferrites (non-conductive ceramics manufactured from the
oxides of nickel, zinc, manganese, or other compounds) are extremely useful in
power supply filters [Reference 9]. At low frequencies (<100kHz), ferrites are
inductive; thus they are useful in low-pass LC filters. Above 100kHz, ferrites
become resistive, an important characteristic in high-frequency filter designs.
Ferrite impedance is a function of material, operating frequency range, DC bias
current, number of turns, size, shape, and temperature. Figure 10.19 summarizes a
number of ferrite characteristics, and Figure 10.20 shows the impedance
characteristic of several ferrite beads from Fair-Rite (http://www.fair-rite.com).
10.25
HARDWARE DESIGN TECHNIQUES
ELECTROLYTIC CAPACITOR
IMPEDANCE VERSUS FREQUENCY
C (100µF)
REGION
LOG
|Z|
ESL (20nH)
REGION
Ω)
ESR (0.2Ω)
REGION
Ω
ESR = 0.2Ω
10kHz
1MHz
LOG FREQUENCY
Figure 10.18
FERRITES SUITABLE FOR HIGH FREQUENCY FILTERS
■ Ferrites Good for Frequencies Above 25kHz
■ Many Sizes and Shapes Available Including Leaded "Resistor
Style"
■ Ferrite Impedance at High Frequencies Primarily Resistive -Ideal for HF Filtering
■ Low DC Loss: Resistance of Wire Passing Through Ferrite is
Very Low
■ High Saturation Current Versions Available
■ Choice Depends Upon:
◆ Source and Frequency of Interference
◆ Impedance Required at Interference Frequency
◆ Environmental: Temperature, AC and DC Field Strength,
Size / Space Available
■
Always Test the Design!
Figure 10.19
10.26
HARDWARE DESIGN TECHNIQUES
IMPEDANCE OF FERRITE BEADS
80
#43
MATERIAL
#64
MATERIAL
60
Z
Ω
#73
MATERIAL
40
20
0
1
10
100
1000
FREQUENCY (MHz)
Courtesy: Fair-Rite Products Corp, Wallkill, NY
(http://www.fair-rite.com)
Figure 10.20
Several ferrite manufacturers offer a wide selection of ferrite materials from which
to choose, as well as a variety of packaging styles for the finished network (see
References 10 and 11). A simple form is the bead of ferrite material, a cylinder of
the ferrite which is simply slipped over the power supply lead to the decoupled
stage. Alternately, the leaded ferrite bead is the same bead, pre-mounted on a
length of wire and used as a component (see Reference 11). More complex beads
offer multiple holes through the cylinder for increased decoupling, plus other
variations. Surface mount beads are also available.
PSpice ferrite models for Fair-Rite materials are available, and allow ferrite
impedance to be estimated [see Reference 12]. These models have been designed to
match measured impedances rather than theoretical impedances.
A ferrite’s impedance is dependent upon a number of inter-dependent variables, and
is difficult to quantify analytically, thus selecting the proper ferrite is not
straightforward. However, knowing the following system characteristics will make
selection easier. First, determine the frequency range of the noise to be filtered.
Second, the expected temperature range of the filter should be known, as ferrite
impedance varies with temperature. Third, the peak DC current flowing through
the ferrite must be known, to ensure that the ferrite does not saturate. Although
models and other analytical tools may prove useful, the general guidelines given
above, coupled with some experimentation with the actual filter connected to the
supply output under system load conditions, should lead to a proper ferrite
selection.
10.27
HARDWARE DESIGN TECHNIQUES
Using proper component selection, low and high frequency band filters can be
designed to smooth a noisy switcher’s DC output to produce an analog ready 5V
supply. It is most practical to do this over two (and sometimes more) stages, each
stage optimized for a range of frequencies. A basic stage can be used to carry all of
the DC load current, and filter noise by 60dB or more up to a 1-10MHz range. This
larger filter is used as a card entry filter providing broadband filtering for all power
entering a PC card. Smaller, more simple local filter stages are also used to provide
higher frequency decoupling right at the power pins of individual stages.
Switching Regulator Experiments
In order to better understand the challenge of filtering switching regulators, a series
of experiments were conducted with a representative device, the ADP1148
synchronous buck regulator with a 9V input and a 3.3V/1A output.
In addition to observing typical input and output waveforms, the objective of these
experiments was to reduce the output ripple to less than 10mV peak-to-peak, a
value suitable for driving most analog circuits.
Measurements were made using a Tektronix wideband digitizing oscilloscope with
the input bandwidth limited to 20MHz so that the ripple generated by the switching
regulators could be more readily observed. In a system, power supply ripple
frequencies above 20MHz are best filtered locally at each IC power pin with a low
inductance ceramic capacitor and perhaps a series-connected ferrite bead.
Probing techniques are critical for accurate ripple measurements. A standard
passive 10X probe was used with a "bayonet" probe tip adapter for making the
ground connection as short as possible (see Figure 10.21). Use of the "ground clip
lead" is not recommended in making this type of measurement because the lead
length in the ground connection forms an unwanted inductive loop which picks up
high frequency switching noise, thereby corrupting the signal being measured.
Note: Schematic representation of proper physical grounding is almost impossible. In
all the following circuit schematics, the connections to ground are made to the
ground plane using the shortest possible connecting path, regardless of how they are
indicated in the actual circuit schematic diagram.
The circuit for the ADP1148 9V to 3.3V/1A buck regulator is shown in Figure 10.22.
The output waveform of the ADP1148 buck regulator is shown in Figure 10.23. The
fundamental switching frequency is approximately 150kHz, and the output ripple is
approximately 40mV.
Adding an output filter consisting of a 50µH inductor and a 100µF leaded tantalum
capacitor reduced the ripple to approximately 3mV.
10.28
HARDWARE DESIGN TECHNIQUES
PROPER PROBING TECHNIQUES
PROBE
"GROUND CLIP"
CONNECTOR
SLIP-ON
"BAYONET"
GROUND
ADAPTER
"GROUND CLIP"
LEAD
(DO NOT USE!!)
SIGNAL
CONTACT
GROUND PLANE
CONTACT
IC
Figure 10.21
VIN, 9V
ADP1148 BUCK REGULATOR CIRCUIT
1µF
VIN
10nF
+
220µF/25V
IRF7204
C1
INT VCC
P-DRIVE
P-CH
ADP1148-3.3
L, 50µH
RSENSE
0.1Ω
Ω
SHUTDOWN
ITH
SENSE (+)
CT
SENSE (–)
RC, 1kΩ
Ω
CC
3300pF
CT
470pF
C2
+
1000pF
N-CH
N-DRIVE
SGND
VOUT
3.3V/1A
10BQ040
100µF
20V
PGND
IRF7403
C1 = 220µF/25V GEN PURPOSE AL ELECTROLYTIC
+ 1µF CERAMIC
L=COILTRONICS CTX-50-4
C2 = 100µF/20V LEADED TANTALUM, KEMET T356-SERIES, ESR = 0.6Ω
Ω
Figure 10.22
10.29
HARDWARE DESIGN TECHNIQUES
ADP1148 BUCK OUTPUT WAVEFORM
VOUT
3.3V
1A
VIN
9V
40mV p-p
ADP1148
BUCK
REG
CIRCUIT
+
1µF 220µF
25V
C1 = 220 µF
+1 µF
+
100µF
20V
C2 = 100µF/20V
VERTICAL SCALE:
HORIZ. SCALE:
10mV / DIV
5µs / DIV
C1 = 1µF CERAMIC + 220µF/25V GENERAL PURPOSE AL ELECTROLYTIC
C2 = 100µF/20V LEADED TANTALUM, KEMET T356-SERIES (ESR = 0.6Ω
Ω)
Figure 10.23
ADP1148 BUCK FILTERED OUTPUT
LF
50µH
VIN
9V
ADP1148
BUCK
REG
CIRCUIT
+
1µF 220µF
25V
C1 = 220 µF
+1 µF
VOUT
3.3V
1A
+
+
100µF
20V
CF
100µF
20V
3mV p-p
C2 = 100µF/20V
VERTICAL SCALE:
HORIZ. SCALE:
10mV / DIV
5µs / DIV
C1 = 1µF CERAMIC + 220µF/25V GENERAL PURPOSE AL ELECTROLYTIC
C2 = 100µF/20V LEADED TANTALUM, KEMET T356-SERIES (ESR = 0.6Ω
Ω)
OUTPUT FILTER
LF=COILTRONICS CTX-50-4
CF = 100µF/20V LEADED TANTALUM, KEMET T356-SERIES
Figure 10.24
Linear regulators are often used following switching regulators for better regulation
and lower noise. Low dropout (LDO) regulators such as the ADP3310 are desirable
in these applications because they require only a small input-to-output series
10.30
HARDWARE DESIGN TECHNIQUES
voltage to maintain regulation. This minimizes power dissipation in the pass device
and may eliminate the need for a heat sink. Figure 10.25 shows the ADP1148 buck
regulator configured for a 9V input and a 3.75V/1A output. The output drives an
ADP3310 linear LDO regulator configured for 3.75V input and 3.3V/1A output. The
input and output of the ADP3310 is shown in Figure 10.26. Notice that the
regulator reduces the ripple from 40mV to approximately 5mV.
ADP1148 BUCK REGULATOR DRIVING
ADP3310 LOW DROPOUT REGULATOR
VIN, 9V
WAVEFORMS
1µF
VIN
10nF
IRF7204
P-DRIVE
IFR7404
C1
INT VCC
P-CH
ADP1148
+
220µF
35V
L, 68µH
RSENSE
0.1Ω
3.75V
SD
ITH
IN
SENSE (+)
2200pF
1000pF
CT
CT
470pF
C2
SENSE (–)
N-CH
N-DRIVE
SGND
OUT
ADP3310-3.3
RC, 1kΩ
CC
GATE
+
100µF
20V
10BQ040
FB PGND
GND
3.3V
1A
C3
10µF
35V
R1
20kΩ
R2
10kΩ
IRF7403
Figure 10.25
WAVEFORMS FOR ADP1148 BUCK REGULATOR
DRIVING ADP3310 LOW DROPOUT REGULATOR
ADP1148 OUTPUT
(ADP3310 INPUT)
ADP3310 OUTPUT
40mV p-p
VERTICAL SCALE: 10mV/DIV
HORIZ. SCALE:
5µs/DIV
5mV p-p
VERTICAL SCALE: 10mV/DIV
HORIZ. SCALE:
5µs/DIV
Figure 10.26
10.31
HARDWARE DESIGN TECHNIQUES
There are many tradeoffs in designing power supply filters. The success of any
filter circuit is highly dependent upon a compact layout and the use of a large area
ground plane. As has been stated earlier, all connections to the ground plane should
be made as short as possible to minimize parasitic resistance and inductance.
Output ripple can be reduced by the addition of low ESL/ESR capacitors to the
output. However, it may be more efficient to use an LC filter to accomplish the
ripple reduction. In any case, proper component selection is critical. The inductor
should not saturate under the maximum load current, and its DC resistance should
be low enough as not to induce significant voltage drop. The capacitors should have
low ESL and ESR and be rated to handle the required ripple current.
Low dropout linear post regulators provide both ripple reduction as well as better
regulation and can be effective, provided the sacrifice in efficiency is not excessive.
Finally, it is difficult to predict the output ripple current analytically, and there is
no substitute for a prototype using the real-world components. Once the filter is
proven to provide the desired ripple attenuation (with some added safety margin),
care must be taken that parts substitutions or vendor changes are not made in the
final production units without first testing them in the circuit for equivalent
performance.
SWITCHING SUPPLY FILTER SUMMARY
■ Proper Layout and Grounding (using Ground Plane) Mandatory
■ Low ESL/ESR Capacitors Give Best Results
■ Parallel Capacitors Lower ESR/ESL and Increase Capacitance
■ External LC Filters Very Effective in Reducing Ripple
■ Linear Post Regulation Effective for Noise Reduction and Best
Regulation
■ Completely Analytical Approach Difficult, Prototyping is
Required for Optimum Results
■ Once Design is Finalized, Do Not Switch Vendors or Use Parts
Substitutions Without First Verifying Their Performance in
Circuit
■ High Frequency Localized Decoupling at IC Power Pins is Still
Required
Figure 10.27
Localized High Frequency Power Supply Filtering
The LC filters described in the previous section are useful in filtering switching
regulator outputs. However, it may be desirable to place similar filters on the
individual PC boards where the power first enters the board. Of course, if the
switching regulator is placed on the PC board, then the LC filter should be an
integral part of the regulator design.
10.32
HARDWARE DESIGN TECHNIQUES
Localized high frequency filters may also be required at each IC power pin (see
Figure 10.28). Surface mount ceramic capacitors are ideal choices because of their
low ESL. It is important to make the connections to the power pin and the ground
plane as short as possible. In the case of the ground connection, a via directly to the
ground plane is the shortest path. Routing the capacitor ground connection to
another ground pin on the IC is not recommended due to the added inductance of
the trace. In some cases, a ferrite bead in series with the power connection may also
be desirable.
LOCALIZED DECOUPLING TO GROUND PLANE
USING SHORTEST PATH
CORRECT
POWER
SUPPLY
TRACE
OPTIONAL
FERRITE BEADS
DECOUPLING
CAPACITOR
INCORRECT
POWER
SUPPLY
TRACE
DECOUPLING
CAPACITOR
V+
IC
V+
VIAS TO
GROUND
PLANE
GND
IC
PCB
TRACE
GND
VIA TO
GROUND
PLANE
Figure 10.28
The following list summarizes the switching power supply filter layout/construction
guidelines which will help ensure that the filter does the best possible job:
(1) Pick the highest electrical value and voltage rating for filter capacitors which is
consistent with budget and space limits. This minimizes ESR, and maximizes filter
performance. Pick chokes for low ∆L at the rated DC current, as well as low DCR.
(2) Use short and wide PCB tracks to decrease voltage drops and minimize
inductance. Make track widths at least 200 mils for every inch of track length for
lowest DCR, and use 1 oz or 2 oz copper PCB traces to further reduce IR drops and
inductance.
(3) Use short leads or better yet, leadless components, to minimize lead inductance.
This minimizes the tendency to add excessive ESL and/or ESR. Surface mount
packages are preferred. Make all connections to the ground plane as short as
possible.
10.33
HARDWARE DESIGN TECHNIQUES
(4) Use a large-area ground plane for minimum impedance.
(5) Know what your components do over frequency, current and temperature
variations! Make use of vendor component models for the simulation of prototype
designs, and make sure that lab measurements correspond reasonably with the
simulation. While simulation is not absolutely necessary, it does instill confidence in
a design when correlation is achieved (see Reference 15).
Filtering the AC Power Lines
The AC power line can also be an EMI entry/exit path! To remove this noise path
and reduce emissions caused by the switching power supply or other circuits, a
power line filter is required.
Figure 10.29 is an example of a hybrid power transient protection network
commonly used in many applications where lightning transients or other power-line
disturbances are prevalent. These networks can be designed to provide protection
against transients as high as 10kV and as fast as 10ns. Gas discharge tubes
(crowbars) and large geometry zener diodes or Transient Voltage Suppressers
(TVSs) are used to provide both differential and common-mode protection. Metaloxide varistors (MOVs) can be substituted for the zener diodes or TVSs in less
critical, or in more compact designs. Chokes are used to limit the surge current until
the gas discharge tubes fire.
POWER LINE DISTURBANCES CAN GENERATE EMI
Reprinted from EDN Magazine (January 20, 1994), © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
GAS DISCHARGE
TUBES
"CROWBARS"
CHOKES
TRANSIENT
SUPPRESSORS
BIG ZENERS
OR MOVs
V
LINE
LOAD
N
G
COMMON-MODE AND DIFFERENTIAL MODE PROTECTION
Figure 10.29
Commercial EMI filters, as illustrated in Figure 10.30, can be used to filter less
catastrophic transients or high-frequency interference. These EMI filters provide
both common-mode and differential mode filtering. An optional choke in the safety
ground can provide additional protection against common-mode noise. The value of
10.34
HARDWARE DESIGN TECHNIQUES
this choke cannot be too large, however, because its resistance may affect power-line
fault clearing. These filters work in both directions: they not only protect the
equipment from surges on the power line but also prevent transients from the
internal switching power supplies from corrupting the power line.
SCHEMATIC FOR A COMMERCIAL POWER LINE FILTER
Reprinted from EDN Magazine (January 20, 1994), © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
HOT
HOT
LINE
LOAD
NEU
NEU
GND
OPTIONAL
NOTE: OPTIONAL CHOKE ADDED FOR COMMON-MODE PROTECTION
Figure 10.30
Transformers provide the best common-mode power line isolation. They provide
good protection at low frequencies (<1MHz), and for transients with rise and fall
times greater than 300ns. Most motor noise and lightning transients are in this
range, so isolation transformers work well for these types of disturbances. Although
the isolation between input and output is galvanic, isolation transformers do not
provide sufficient protection against extremely fast transients (<10ns) or those
caused by high-amplitude electrostatic discharge (1 to 3ns). Isolation transformers
can be designed for various levels of differential- or common-mode protection. For
differential-mode noise rejection, the Faraday shield is connected to the neutral,
and for common-mode noise rejection, the shield is connected to the safety ground.
It is important to remember that AC line power can potentially be lethal! Do not
experiment without proper equipment and training! All components used in power
line filters should be UL approved, and the best way to provide this is to specify a
packaged UL approved filter. It should be installed in such a manner that it is the
first circuit the AC line sees upon entering the equipment. Standard three wire IEC
style line cords are designed to mate with three terminal male connectors integral
to many line filters. This is the best way to achieve this function, as it automatically
grounds the third wire to the shell of the filter and equipment chassis via a low
inductance path.
10.35
HARDWARE DESIGN TECHNIQUES
Commercial power line filters are quite effective in reducing AC power-line noise.
This noise generally has both common-mode and differential-mode components.
Common-mode noise is noise that is found on any two of the three power
connections (black, white, or green) with the same amplitude and polarity. In
contrast, differential-mode noise is noise found only between two lines. By design,
most commercially available filters address both noise modes (see Reference 16).
10.36
HARDWARE DESIGN TECHNIQUES
REFERENCES: NOISE REDUCTION AND FILTERING
1.
EMC Design Workshop Notes, Kimmel-Gerke Associates, Ltd.,
St. Paul, MN. 55108, (612) 330-3728.
2.
Walt Jung, Dick Marsh, Picking Capacitors, Parts 1 & 2, Audio,
February, March, 1980.
3.
Tantalum Electrolytic and Ceramic Capacitor Families, Kemet
Electronics, Box 5928, Greenville, SC, 29606, (803) 963-6300.
4.
Type HFQ Aluminum Electrolytic Capacitor and type V Stacked
Polyester Film Capacitor, Panasonic, 2 Panasonic Way, Secaucus,
NJ, 07094, (201) 348-7000.
5.
OS-CON Aluminum Electrolytic Capacitor 93/94 Technical Book,
Sanyo, 3333 Sanyo Road, Forrest City, AK, 72335, (501) 633-6634.
6.
Ian Clelland, Metalized Polyester Film Capacitor Fills High Frequency
Switcher Needs, PCIM, June 1992.
7.
Type 5MC Metallized Polycarbonate Capacitor, Electronic Concepts, Inc.,
Box 1278, Eatontown, NJ, 07724, (908) 542-7880.
8.
Walt Jung, Regulators for High-Performance Audio, Parts 1 and 2,
The Audio Amateur, issues 1 and 2, 1995.
9.
Henry Ott, Noise Reduction Techniques in Electronic Systems,
2d Ed., 1988, Wiley.
10.
Fair-Rite Linear Ferrites Catalog, Fair-Rite Products, Box J, Wallkill,
NY, 12886, (914) 895-2055, http://www.fair-rite.com.
11.
Type EXCEL leaded ferrite bead EMI filter, and type EXC L leadless
ferrite bead, Panasonic, 2 Panasonic Way, Secaucus, NJ, 07094,
(201) 348-7000.
12.
Steve Hageman, Use Ferrite Bead Models to Analyze EMI Suppression,
The Design Center Source, MicroSim Newsletter, January, 1995.
10.37
HARDWARE DESIGN TECHNIQUES
13.
Type 5250 and 6000-101K chokes, J. W. Miller, 306 E. Alondra Blvd.,
Gardena, CA, 90247, (310) 515-1720.
14.
DIGI-KEY, PO Box 677, Thief River Falls, MN, 56701-0677,
(800) 344-4539.
15.
Tantalum Electrolytic Capacitor SPICE Models, Kemet Electronics,
Box 5928, Greenville, SC, 29606, (803) 963-6300.
16.
Eichhoff Electronics, Inc., 205 Hallene Road, Warwick, RI., 02886,
(401) 738-1440, http://www.eichhoff.com.
17.
Practical Design Techniques for Power and Thermal Management,
Analog Devices, 1998, Chapter 8.
10.38
HARDWARE DESIGN TECHNIQUES
PREVENTING RFI RECTIFICATION
Walt Kester, Walt Jung, Chuck Kitchin
High frequency radio frequency interference (RFI) can seriously affect the DC
performance of high accuracy circuits. Because of their relatively low bandwidth,
precision operational amplifiers and instrumentation amplifiers will not accurately
amplify RF signals in the MHz range. However, if these out-of-band signals are
allowed to couple into the precision amplifier through either its input, output, or
power supply pins, they can be rectified by various junctions in the amplifier and
ultimately cause an unexplained and unwanted DC offset at the output. An
excellent analysis of the phenomenon is found in Reference 1, but the purpose here
is to show how proper filtering can be used to minimize or prevent these errors.
We have previously discussed how proper power supply decoupling techniques will
minimize RFI on the IC power pins. Further discussion is required with respect to
the amplifier inputs and outputs.
The best way to prevent rectification due to input RFI is to use a filter located close
to the op amp input as shown in Figure 10.31. In the case of the inverting op amp,
the filter capacitor C1 is placed between R1 and R2. The DC closed loop gain of the
circuit is –R3/(R1+R2). C1 is not connected directly to the inverting input of the op
amp because that would result in instability. The filter bandwidth is chosen to be at
least 100 times larger than the actual signal bandwidth to prevent signal
attenuation. For the non-inverting configuration, the filter capacitor can be
connected directly to the op amp input as shown.
It should be noted that a ferrite bead can be used instead of R1, however ferrite
bead impedance is not well controlled and is generally no greater than 100Ω at
10MHz to 100MHz. This requires a large value capacitor to attenuate the lower
frequencies.
Precision instrumentation amplifiers are particularly sensitive to common-mode
RFI. Proper filtering is shown in Figure 10.32. Note that there is both commonmode filtering (R1/C1, R2/C2) and differential mode filtering (R1+R2, and C3). If
R1/R2 and C1/C2 are not well matched, some of the input common-mode signal at
VIN will be converted to a differential one at the in-amp inputs. For this reason, C1
and C2 should be matched to within at least 5% of each other. R1 and R2 should be
1% metal film resistors to insure matching. Capacitor C3 attenuates the differential
signal which can result from imperfect matching of the common-mode filters. In this
type of filter, C3 should be much larger than C1 or C2 in order to ensure that any
differential signal due to mismatching of the common-mode signals is sufficiently
attenuated.
The overall filter bandwidth should be chosen to be at least 100 times the input
signal bandwidth. The components should be symmetrically mounted on a PC board
with a large area ground plane and placed very close to the in-amp input for
optimum performance of the filter.
10.39
HARDWARE DESIGN TECHNIQUES
FILTERING AMPLIFIER INPUTS
TO PREVENT RFI RECTIFICATION
R1 = R
R3
R1 = 2R
R2 = 2R
+
C1
_
_
C1
+
R3
R2
1
2π R C1
FILTER BANDWIDTH =
> 100 × SIGNAL BANDWIDTH
Figure 10.31
FILTERING IN-AMP INPUTS
R1
+
VIN R1 = R2
C1 = C2
R2
C1
C3
IN-AMP
C2
_
τDIFF = (R1 + R2) C3
τCM = R1·C1 = R2·C2
τDIFF >> τCM
R1·C1 SHOULD MATCH R2·C2
R1 = R2 SHOULD BE 1% RESISTORS
C1 = C2 SHOULD BE 5% CAPACITORS
DIFFERENTIAL
FILTER BANDWIDTH
1
=
2π (R1 + R2)
Figure 10.32
10.40
C1·C2
C1 + C2 + C3
HARDWARE DESIGN TECHNIQUES
Figure 10.33 shows an actual filter for use with the AD620 in-amp. The commonmode rejection was tested by applying a 1V p-p common-mode signal to the input
resistors. The AD620 gain was 1000. The RTI offset voltage of the in-amp was
measured as the frequency of the sinewave source was varied from DC to 20MHz.
The maximum RTI input offset voltage shift was 1.5µV. The filter bandwidth was
approximately 400Hz.
Common-mode chokes offer a simple, one component alternative to RC passive
filters. Selecting the proper common-mode choke is critical, however. The choke
used in the circuit of Figure 10.34 was a Pulse Engineering B4001 designed for
XDSL data receivers (through-hole mount). The B4003 is an equivalent surface
mount choke. The maximum RTI offset shift measured from DC to 20MHz was
4.5µV. Unlike the RC filter of Figure 10.32, the choke-based filter offers no
differential mode filtration, as shown.
COMMON AND DIFFERENTIAL MODE FILTER WITH AD620
RG = 49.9Ω
Ω G = 1000
R1
DC TO 20MHz
1V p-p
C1
4.02kΩ
Ω
1% 1000pF, 5%
SINEWAVE
SOURCE
C2
R2
50Ω
Ω
+
C3
0.047µF
1000pF, 5%
AD620
_
4.02kΩ
Ω
1%
FILTER BANDWIDTH ≈ 400Hz
OFFSET SHIFT RTI < 1.5 µV
Figure 10.33
In addition to filtering the inputs and the power pins, amplifier outputs need to be
protected from RFI, especially if they must drive long lengths of cable. RFI on the
output can couple into the amplifier where it is rectified and appears again on the
output as a DC offset shift. A resistor or ferrite bead in series with the output is the
simplest output filter. Adding a capacitor as shown in Figure 10.35 improves this
filter, but the capacitor should not be connected to the op-amp side of the resistor
because it may cause the amplifier to become unstable. Many amplifiers are
sensitive to direct output capacitive loads, so this condition should be avoided unless
the amplifier data sheet clearly specifies that the output is insensitive to capacitive
loading.
10.41
HARDWARE DESIGN TECHNIQUES
COMMON MODE CHOKE WITH AD620
Ω G = 1000
RG = 49.9Ω
DC TO 20MHz
1V p-p
+
SINEWAVE
SOURCE
AD620
_
Ω
50Ω
COMMON MODE CHOKE:
PULSE ENGINEERING B4001
http://www.pulseeng.com
OFFSET SHIFT RTI < 4.5 µV
Figure 10.34
FILTERING AMPLIFIER OUTPUTS PROTECTS
AGAINST EMI/RFI EMISSION AND SUSCEPTIBILITY
RESISTOR OR
FERRITE BEAD
AMP
RESISTOR OR
FERRITE BEAD
AMP
C
RESISTOR OR
FERRITE BEAD
AMP
C
Figure 10.35
10.42
MAY CAUSE
INSTABILITY
HARDWARE DESIGN TECHNIQUES
REFERENCES ON RFI RECTIFICATION
1.
System Applications Guide, Analog Devices, Inc., 1993, Section 1,
pp. 1.37-1.55.
2.
Pulse Engineering, Inc., 12220 World Trade Drive, San Diego, CA
92128, 619-674-8100, http://www.pulseeng.com.
10.43
HARDWARE DESIGN TECHNIQUES
DEALING WITH HIGH SPEED LOGIC
Much has been written about terminating printed circuit board traces in their
characteristic impedance to avoid reflections. A good rule-of-thumb to determine
when this is necessary is as follows: Terminate the line in its characteristic
impedance when the one-way propagation delay of the PCB track is equal to or
greater than one-half the applied signal rise/fall time (whichever edge is faster). A
conservative approach is to use a 2 inch (PCB track length)/nanosecond (rise-, falltime) criterion. For example, PCB tracks for high-speed logic with rise/fall time of
5ns should be terminated in their characteristic impedance if the track length is
equal to or greater than 10 inches (including any meanders). The 2 inch/nanosecond
track length criterion is summarized in Figure 10.36 for a number of logic families.
LINE TERMINATION SHOULD BE USED WHEN
LENGTH OF PCB TRACK EXCEEDS 2 inches/ns
Reprinted from EDN Magazine (January 20, 1994), © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
DIGITAL IC
FAMILY
GaAs
t r , ts
(ns)
0.1
PCB TRACK LENGTH
(inches)
0.2
PCB TRACK LENGTH
(cm)
0.5
ECL
0.75
1.5
3.8
Schottky
3
6
15
FAST
3
6
15
AS
3
6
15
AC
4
8
20
ALS
6
12
30
LS
8
16
40
TTL
10
20
50
HC
18
36
90
tr = rise time of signal in ns
tf = fall time of signal in ns
For analog signals @ fmax, calculate tr = tf = 0.35 / fmax
Figure 10.36
This same 2 inch/nanosecond rule of thumb should be used with analog circuits in
determining the need for transmission line techniques. For instance, if an amplifier
must output a maximum frequency of fmax, then the equivalent risetime, tr, can be
calculated using the equation tr = 0.35/fmax.
10.44
HARDWARE DESIGN TECHNIQUES
The maximum PCB track length is then calculated by multiplying the risetime by 2
inch/nanosecond. For example, a maximum output frequency of 100MHz
corresponds to a risetime of 3.5ns, and a track carrying this signal greater than 7
inches should be treated as a transmission line.
Equation 10.1 can be used to determine the characteristic impedance of a PCB track
separated from a power/ground plane by the board’s dielectric (microstrip
transmission line):
Z o (Ω) =
87
 5.98d 
ln 

ε r + 1.41
 0.89w + t 
Eq. 10.1
where εr = dielectric constant of printed circuit board material;
d = thickness of the board between metal layers, in mils;
w = width of metal trace, in mils; and
t = thickness of metal trace, in mils.
The one-way transit time for a single metal trace over a power/ground plane can be
determined from Eq.10.2:
t pd (ns / ft) = 1.017 0.475ε r + 0.67
Eq. 10.2
For example, a standard 4-layer PCB board might use 8-mil wide, 1 ounce (1.4 mils)
copper traces separated by 0.021" FR-4 (εr=4.7) dielectric material. The
characteristic impedance and one-way transit time of such a signal trace would be
88Ω and 1.7ns/ft (7"/ns), respectively.
The best ways to keep sensitive analog circuits from being affected by fast logic are
to physically separate the two and to use no faster logic family than is dictated by
system requirements. In some cases, this may require the use of several logic
families in a system. An alternative is to use series resistance or ferrite beads to
slow down the logic transitions where the speed is not required. Figure 10.37 shows
two methods. In the first, the series resistance and the input capacitance of the gate
form a lowpass filter. Typical CMOS input capacitance is 10pF. Locate the series
resistor close to the driving gate. The resistor minimizes transient currents and may
eliminate the necessity of using transmission line techniques. The value of the
resistor should be chosen such that the rise and fall times at the receiving gate are
fast enough to meet system requirement, but no faster. Also, make sure that the
resistor is not so large that the logic levels at the receiver are out of specification
because of the source and sink current which must flow through the resistor.
10.45
HARDWARE DESIGN TECHNIQUES
SLOW DOWN FAST LOGIC EDGES
TO MINIMIZE EMI/RFI PROBLEMS
LOGIC
GATE
R
< 2 inches
LOGIC
GATE
CIN
Risetime = 2.2 R·CIN
> 2 inches
LOGIC
GATE
R
LOGIC
GATE
C
CIN
Risetime = 2.2 R·(C + CIN)
Figure 10.37
10.46
HARDWARE DESIGN TECHNIQUES
A REVIEW OF SHIELDING CONCEPTS
The concepts of shielding effectiveness presented next are background material.
Interested readers should consult References 1,3, and 4 cited at the end of the
section for more detailed information.
Applying the concepts of shielding requires an understanding of the source of the
interference, the environment surrounding the source, and the distance between the
source and point of observation (the receptor or victim). If the circuit is operating
close to the source (in the near-, or induction-field), then the field characteristics are
determined by the source. If the circuit is remotely located (in the far-, or radiationfield), then the field characteristics are determined by the transmission medium.
A circuit operates in a near-field if its distance from the source of the interference is
less than the wavelength (λ) of the interference divided by 2π, or λ/2π. If the
distance between the circuit and the source of the interference is larger than this
quantity, then the circuit operates in the far field. For instance, the interference
caused by a 1ns pulse edge has an upper bandwidth of approximately 350MHz. The
wavelength of a 350MHz signal is approximately 32 inches (the speed of light is
approximately 12"/ns). Dividing the wavelength by 2π yields a distance of
approximately 5 inches, the boundary between near- and far-field. If a circuit is
within 5 inches of a 350MHz interference source, then the circuit operates in the
near-field of the interference. If the distance is greater than 5 inches, the circuit
operates in the far-field of the interference.
Regardless of the type of interference, there is a characteristic impedance associated
with it. The characteristic, or wave impedance of a field is determined by the ratio
of its electric (or E-) field to its magnetic (or H-) field. In the far field, the ratio of the
electric field to the magnetic field is the characteristic (wave impedance) of free
space, given by Zo = 377Ω. In the near field, the wave-impedance is determined by
the nature of the interference and its distance from the source. If the interference
source is high-current and low-voltage (for example, a loop antenna or a power-line
transformer), the field is predominately magnetic and exhibits a wave impedance
which is less than 377Ω. If the source is low-current and high-voltage (for example,
a rod antenna or a high-speed digital switching circuit), then the field is
predominately electric and exhibits a wave impedance which is greater than 377Ω.
Conductive enclosures can be used to shield sensitive circuits from the effects of
these external fields. These materials present an impedance mismatch to the
incident interference because the impedance of the shield is lower than the wave
impedance of the incident field. The effectiveness of the conductive shield depends
on two things: First is the loss due to the reflection of the incident wave off the
shielding material. Second is the loss due to the absorption of the transmitted wave
within the shielding material. Both concepts are illustrated in Figure 10.38. The
amount of reflection loss depends upon the type of interference and its wave
impedance. The amount of absorption loss, however, is independent of the type of
interference. It is the same for near- and far-field radiation, as well as for electric or
magnetic fields.
10.47
HARDWARE DESIGN TECHNIQUES
REFLECTION AND ABSORPTION ARE THE TWO
PRINCIPAL SHIELDING MECHANISMS
Reprinted from EDN Magazine (January 20, 1994), © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
INCIDENT RAY
REFLECTED RAY
SHIELD
MATERIAL
TRANSMITTED
RAY
ABSORPTIVE
REGION
Figure 10.38
Reflection loss at the interface between two media depends on the difference in the
characteristic impedances of the two media. For electric fields, reflection loss
depends on the frequency of the interference and the shielding material. This loss
can be expressed in dB, and is given by:
 σ

r 
R e (dB) = 322 + 10log10 
 µ f 3 r2 
 r

Eq. 10.3
where σr = relative conductivity of the shielding material, in Siemens per meter;
µr = relative permeability of the shielding material, in Henries per meter;
f = frequency of the interference, and
r = distance from source of the interference, in meters
For magnetic fields, the loss depends also on the shielding material and the
frequency of the interference. Reflection loss for magnetic fields is given by:
 f r2 σ 
r
R m (dB) = 14.6 + 10log10 
 µr 


10.48
Eq. 10.4
HARDWARE DESIGN TECHNIQUES
and, for plane waves ( r > λ/2π), the reflection loss is given by:
σ 
R pw (dB) = 168 + 10log10  r 
µr f 
Eq. 10.5
Absorption is the second loss mechanism in shielding materials. Wave attenuation
due to absorption is given by:
A (dB) = 3.34 t σ r µ r f
Eq. 10.6
where t = thickness of the shield material, in inches. This expression is valid for
plane waves, electric and magnetic fields. Since the intensity of a transmitted field
decreases exponentially relative to the thickness of the shielding material, the
absorption loss in a shield one skin-depth (δ) thick is 9dB. Since absorption loss is
proportional to thickness and inversely proportional to skin depth, increasing the
thickness of the shielding material improves shielding effectiveness at high
frequencies.
Reflection loss for plane waves in the far field decreases with increasing frequency
because the shield impedance, Zs, increases with frequency. Absorption loss, on the
other hand, increases with frequency because skin depth decreases. For electric
fields and plane waves, the primary shielding mechanism is reflection loss, and at
high frequencies, the mechanism is absorption loss. For these types of interference,
high conductivity materials, such as copper or aluminum, provide adequate
shielding. At low frequencies, both reflection and absorption loss to magnetic fields
is low; thus, it is very difficult to shield circuits from low-frequency magnetic fields.
In these applications, high-permeability materials that exhibit low-reluctance
provide the best protection. These low-reluctance materials provide a magnetic
shunt path that diverts the magnetic field away from the protected circuit. Some
characteristics of metallic materials commonly used for shielded enclosures are
shown in Figure 10.39.
A properly shielded enclosure is very effective at preventing external interference
from disrupting its contents as well as confining any internally-generated
interference. However, in the real world, openings in the shield are often required to
accommodate adjustment knobs, switches, connectors, or to provide ventilation.
Unfortunately, these openings may compromise shielding effectiveness by providing
paths for high-frequency interference to enter the instrument.
10.49
HARDWARE DESIGN TECHNIQUES
CONDUCTIVITY AND PERMEABILITY FOR
VARIOUS SHIELDING MATERIALS
MATERIAL
RELATIVE
CONDUCTIVITY
RELATIVE
PERMEABILITY
Copper
1
1
Aluminum
1
0.61
Steel
0.1
1,000
Mu-Metal
0.03
20,000
Conductivity: Ability to Conduct Electricity
Permeability: Ability to Absorb Magnetic Energy
Figure 10.39
The longest dimension (not the total area) of an opening is used to evaluate the
ability of external fields to enter the enclosure, because the openings behave as slot
antennas. Equation 10.7 can be used to calculate the shielding effectiveness, or the
susceptibility to EMI leakage or penetration, of an opening in an enclosure:
 λ 
Shielding Effectiveness (dB) = 20 log10 

 2 ⋅ L
Eq. 10.7
where λ = wavelength of the interference and
L = maximum dimension of the opening
Maximum radiation of EMI through an opening occurs when the longest dimension
of the opening is equal to one half-wavelength of the interference frequency (0dB
shielding effectiveness). A rule-of-thumb is to keep the longest dimension less than
1/20 wavelength of the interference signal, as this provides 20dB shielding
effectiveness. Furthermore, a few small openings on each side of an enclosure is
preferred over many openings on one side. This is because the openings on different
sides radiate energy in different directions, and as a result, shielding effectiveness
is not compromised. If openings and seams cannot be avoided, then conductive
gaskets, screens, and paints alone or in combination should be used judiciously to
limit the longest dimension of any opening to less than 1/20 wavelength. Any cables,
wires, connectors, indicators, or control shafts penetrating the enclosure should
have circumferential metallic shields physically bonded to the enclosure at the point
of entry. In those applications where unshielded cables/wires are used, then filters
are recommended at the point of shield entry.
10.50
HARDWARE DESIGN TECHNIQUES
General Points on Cables and Shields
Although covered in more detail later, the improper use of cables and their shields
is a significant contributor to both radiated and conducted interference. Rather than
developing an entire treatise on these issues, the interested reader should consult
References 1,2, 4, and 5. As illustrated in Figure 10.40, effective cable and enclosure
shielding confines sensitive circuitry and signals within the entire shield without
compromising shielding effectiveness. As shown in the diagram, the enclosures and
the shield must be grounded properly, otherwise they will act as an antenna and
make the radiated and conducted interference problem worse.
"ELECTRICALLY LONG" OR "ELECTRICALLY SHORT"
APPLICATION
Reprinted from EDN Magazine (January 20, 1994), © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
SHIELDED ENCLOSURE B
SHIELDED ENCLOSURE A
LENGTH
SHIELDED
CABLE
FULLY SHIELDED ENCLOSURES CONNECTED BY FULLY
SHIELDED CABLE KEEP ALL INTERNAL CIRCUITS AND
SIGNAL LINES INSIDE THE SHIELD.
TRANSITION REGION: 1/20 WAVELENGTH
Figure 10.40
Depending on the type of interference (pickup/radiated, low/high frequency), proper
cable shielding is implemented differently and is very dependent on the length of
the cable. The first step is to determine whether the length of the cable is
electrically short or electrically long at the frequency of concern. A cable is
considered electrically short if the length of the cable is less than 1/20 wavelength of
the highest frequency of the interference, otherwise it is electrically long. For
example, at 50/60Hz, an electrically short cable is any cable length less than 150
miles, where the primary coupling mechanism for these low frequency electric fields
is capacitive. As such, for any cable length less than 150 miles, the amplitude of the
interference will be the same over the entire length of the cable.
In those applications where the length of the cable is electrically long, or protection
against high-frequency interference is required, then the preferred method is to
connect the cable shield to low-impedance points at both ends (direct connection at
the driving end, and capacitive connection at the receiver). Otherwise, unterminated
10.51
HARDWARE DESIGN TECHNIQUES
transmission lines effects can cause reflections and standing waves along the cable.
At frequencies of 10MHz and above, circumferential (360°) shield bonds and metal
connectors are required to main low-impedance connections to ground.
In summary, for protection against low-frequency (<1MHz), electric-field
interference, grounding the shield at one end is acceptable. For high-frequency
interference (>1MHz), the preferred method is grounding the shield at both ends,
using 360° circumferential bonds between the shield and the connector, and
maintaining metal-to-metal continuity between the connectors and the enclosure.
Grounding the shield at both ends, however, can create a low frequency ground loop
in a practical situation as shown in Figure 10.41.
GROUND LOOPS IN SHIELDED TWISTED PAIR CABLE
A1
GND 1
A2
IN
VN
GND 2
VN Causes Current in Shield (Usually 50/60Hz)
Differential Error Voltage is Produced at Input of A2 Unless:
A1 Output is Perfectly Balanced and
A2 Input is Perfectly Balanced and
Cable is Perfectly Balanced
Figure 10.41
As discussed above, cable shields can be subjected to both low and high frequency
interference. Good design practice requires that the shield be grounded at both ends
if the cable is electrically long to the interference frequency, as is usually the case
with RF interference.
When two systems A1 and A2 are remote from each other, however, there is usually
a difference in the ground potentials at each system. The frequency of this potential
difference is generally the line frequency (50Hz or 60Hz) and multiples thereof. If
the shield is grounded at both ends as shown, however, a noise current flows
through the shield. In a perfectly balanced system, the common-mode rejection of
the system is infinite, and this current produces no differential error at the receiver
A2. However, perfect balance is never achieved in the driver, its impedance, the
cable, or the receiver, so a certain portion of the shield current will appear as a
10.52
HARDWARE DESIGN TECHNIQUES
differential signal at the input of A2. The following examples illustrate the correct
way to ground the shield under various conditions.
Figure 10.42 shows a remote passive RTD sensor connected to a bridge and
conditioning circuit by a shielded cable. The proper grounding method is shown in
the upper part of the figure, where the shield is grounded at the receiving end.
However, safety considerations may require that the remote end of the shield be
grounded. If this is the case, the receiving end can be grounded with a low
inductance ceramic capacitor (0.01µF to 0.1µF) . The capacitor acts as a ground to
RF signals on the shield but blocks line frequency current to flow in the shield. This
technique is often referred to as a hybrid ground.
In the case of an active remote sensor (Figure 10.43), the hybrid ground is also
appropriate, either for a balanced or single-ended driver. The capacitor breaks the
DC ground loop in both cases. In both cases, the line is driven from an impedance of
RS, split between legs. In the case of the bottom diagram, the RS/2 resistor in the
return leg can only be used for applications with a balanced receiver, as shown.
GROUNDING SHIELDED CABLE WITH
REMOTE PASSIVE SENSOR
BRIDGE
AND
CONDITIONING
CIRCUITS
RTD
NC
BRIDGE
AND
CONDITIONING
CIRCUITS
RTD
"HYBRID
GROUND
C
Figure 10.42
Coaxial cables are different from shielded twisted pair cables in that the signal
return current path is through the shield. For this reason, the ideal situation is to
ground the shield at the driving end and allow the shield to float at the differential
receiver (A2) as shown in Figure 10.44. For this technique to work, however, the
receiver must be differential and have good high frequency common mode rejection.
If the receiver is a single-ended type, there is no choice but to ground the coaxial
cable shield at both ends.
10.53
HARDWARE DESIGN TECHNIQUES
GROUNDING SHIELDED CABLE
WITH REMOTE ACTIVE SENSOR
A1
RS/2
A2
RS/2
C
A1
RS/2
A2
RS/2
C
Figure 10.43
COAXIAL CABLE GROUNDING
COAX CABLE
A1
A2
DIFF
AMP
Shield Carries Signal Return Current
A1
A2
SINGLEENDED
AMP
Figure 10.44
10.54
HARDWARE DESIGN TECHNIQUES
Digital Isolation Techniques
Another way to break ground loops is to use isolation techniques. Analog isolation
amplifiers find many applications where a high degree of isolation is required, such
as in medical instrumentation. Digital isolation techniques offer a reliable method
of transmitting digital signals over interfaces without introducing ground noise.
Optoisolators are useful and available in a wide variety of styles and packages.
Current is applied to an LED transmitter as shown in Figure 10.45. The light
output is received by a phototransistor. Isolation voltages range from 5000V to
7000V. In the circuit, the LED is driven with a current of approximately 10mA. This
produces a light output sufficient to saturate the phototransistor. Although
excellent for digital signals, optoisolators are too nonlinear for most analog
applications. One should realize that since the phototransistor is operated in a
saturated mode, rise and fall-times can range from 10µs to 20µs in some slower
devices, so the proper optoisolator for the application must be selected.
ISOLATION USING OPTOISOLATORS
+V, (5V) 425Ω
CMOS
GATE
G1
10kΩ
IOUT
IIN
HIGH VOLTAGE
ISOLATION BARRIER
+V, (5V)
VOUT
G2
Uses Light for Transmission Over a High Voltage Barrier
The LED is the Transmitter, and the Phototransistor is the Receiver
High Voltage Isolation: 5000V to 7000V
Non-Linear -- Best for Digital or Frequency Information
Rise and Fall-times can be 10 to 20µs in Slower Devices
Example: Siemens IL-1 (http://www.siemens.com)
Figure 10.45
The AD260/AD261 family of digital isolators isolates five digital control signals
to/from high speed DSPs, microcontrollers, or microprocessors. The AD260 also has
a 1.5W transformer for a 3.5kV isolated external DC/DC power supply circuit.
10.55
HARDWARE DESIGN TECHNIQUES
Each line of the AD260 can handle digital signals up to 20MHz with a propagation
delay of only 14ns which allows for extremely fast data transmission. Output
waveform symmetry is maintained to within ±1ns of the input so the AD260 can be
used to accurately isolate time-based pulse width modulator (PWM) signals.
A simplified schematic of one channel of the AD260/AD261 is shown in Figure
10.46. The data input is passed through a schmitt trigger circuit, through a latch,
and a special transmitter circuit which differentiates the edges of the digital input
signal and drives the primary winding of a proprietary transformer with a "sethigh/set-low" signal. The secondary of the isolation transformer drives a receiver
with the same "set-hi/set-low" data which regenerates the original logic waveform.
An internal circuit operates in the background which interrogates all inputs about
every 5µs and in the absence of logic transitions, sends appropriate "set-hi/set-low"
data across the interface. Recovery time from a fault condition or at power-up is
thus between 5µs and 10µs.
The power transformer (available on the AD260) is designed to operate between
150kHz and 250kHz and will easily deliver more than 1W of isolated power when
driven push-pull (5V) on the transmitter side. Different transformer taps, rectifier
and regulator schemes will provide combinations of ±5V, 15V, 24V, or even 30V or
higher. The output voltage when driven with a low voltage-drop drive will be
37V p-p across the entire secondary with a 5V push-pull drive.
AD260/AD261 DIGITAL ISOLATORS
DATA
IN
SCHMITT
TRIGGER
LATCH
D
XMTR
RCVR
TRI STATE
DATA
OUT
E
ENABLE
ENABLE
CONTINUOUS
UPDATE
CIRCUIT
ISOLATED POWER
XFMR (AD260)
37V p-p, 1.5W
NOTE: SINGLE DATA CHANNEL SHOWN
3.5kV RMS ISOLATION BARRIER
(AD260B/AD261B)
Figure 10.46
10.56
HARDWARE DESIGN TECHNIQUES
AD260/AD261 DIGITAL ISOLATOR KEY SPECIFICATIONS
■ Isolation Test Voltage to 3.5kV RMS (AD260B/AD261B)
■ Five Isolated Digital Lines Available in 6 Input/Output Configurations
■ Logic Signal Frequency: 20MHz Max.
■ Isolated Power Transformer: 37V p-p, 1.5W (AD260)
■ Waveform Edge Transmission Symmetry: ±1ns
■ Propagation Delay: 14ns
■ Rise and Fall-Times < 5ns
Figure 10.47
REFERENCES ON EMI/RFI AND SHIELDING
1.
EDN’s Designer’s Guide to Electromagnetic Compatibility, EDN,
January, 20, 1994, material reprinted by permission of Cahners Publishing
Company, 1995.
2.
Designing for EMC (Workshop Notes), Kimmel Gerke Associates, Ltd., 1994.
3.
Systems Application Guide, Chapter 1, pg. 21-55, Analog Devices,
Incorporated, Norwood, MA, 1994.
4.
Henry Ott, Noise Reduction Techniques In Electronic Systems,
Second Edition, New York, John Wiley & Sons, 1988.
5.
Ralph Morrison, Grounding And Shielding Techniques In
Instrumentation, Fourth Edition, New York, John Wiley & Sons, 1998.
6.
Amplifier Applications Guide, Chapter XI, pg. 61, Analog Devices,
Incorporated, Norwood, MA, 1992.
7.
B.Slattery and J.Wynne, Design and Layout of a Video Graphics
System for Reduced EMI, Analog Devices Application Note AN-333.
10.57
HARDWARE DESIGN TECHNIQUES
8.
Paul Brokaw, An IC Amplifier User Guide To Decoupling, Grounding,
And Making Things Go Right For A Change, Analog Devices
Application Note, Order Number E1393-5-590.
9.
A. Rich, Understanding Interference-Type Noise, Analog Dialogue, 16-3,
1982, pp. 16-19.
10.
A. Rich, Shielding and Guarding, Analog Dialogue, 17-1, 1983, pp. 8-13.
11.
EMC Test & Design, Cardiff Publishing Company, Englewood, CO.
An excellent, general purpose trade journal on issues of EMI and EMC.
12.
A. Rich, Understanding Interference-Type Noise, Analog Dialogue,
16-3, 1982, pp. 16-19.
13.
James Bryant and Herman Gelbach, High Frequency Signal
Contamination, Analog Dialogue, Vol. 27-2, 1993.
14.
Walt Jung, System RF Interference Prevention, Analog Dialogue,
Vol. 28-2, 1994.
15.
Neil Muncy, Noise Susceptibility in Analog and Digital Signal
Processing Systems, presented at 97th Audio Engineering Society
Convention, Nov. 1994.
16.
Ralph Morrison, Solving Interference Problems in Electronics,
John Wiley, 1995.
17.
Siemens Optoisolator Products, http://www.siemens.com.
10.58
HARDWARE DESIGN TECHNIQUES
OVERVOLTAGE PROTECTION
Walt Kester, Wes Freeman, Joe Buxton
Op amps and instrumentation amplifiers must often interface to the outside world,
which may entail handling voltages that exceed their absolute maximum ratings.
Sensors are often placed in an environment where a fault may connect the sensor to
high voltages: if the sensor is connected to an amplifier, the amplifier inputs may
see voltages exceeding its power supplies. Whenever its input voltage goes outside
its supply range, an op amp may be damaged, even when they are turned off.
Almost all op amps' input absolute maximum ratings limit the maximum allowable
input voltage to the positive and negative supplies or possibly 0.3V outside these
supplies. A few exceptions to this rule do exist, which can be identified from
individual data sheets, but the vast majority of amplifiers require input protection if
over-voltage can possibly occur.
Any op amp input will break down to the positive rail or the negative rail if it
encounters sufficient over-voltages. The breakdown voltage is entirely dependent on
the structure of the input stage. It may be equivalent to a diode drop of 0.7V or to a
process breakdown voltage of 50V or more. The danger of an over-voltage is that
when conduction occurs large currents may flow, which can destroy the device. In
many cases, over-voltage results in current well in over 100mA, which can destroy a
part almost instantly.
To avoid damage, input current should be limited to less than 5mA unless otherwise
stated on the relevant data sheet. This value is a conservative rule of thumb based
on metal trace widths in a typical op amp input stage. Higher levels of current can
cause metal migration, which will eventually lead to an open trace. Migration is a
cumulative effect that may not result in a failure for a long period of time. Failure
may occur due to multiple over-voltages, which is a difficult failure mode to identify.
Thus, even though an amplifier may appear to withstand over-voltage currents well
above 5mA for a short period of time, it is important to limit the current to
guarantee long term reliability.
Two types of conduction occur in over-voltage conditions, forward biasing of PN
junctions inherent in the structure of the input stage or, given enough voltage,
reverse junction breakdown. The danger of forward biasing a PN-junction is that
excessive current will flow and damage the part. As long as the current is limited no
damage should occur. However, when the conduction is due to the reverse
breakdown of a PN junction, the problem can be more serious. In the case of a baseemitter junction break down, even small amounts of current can cause degradation
in the beta of the transistor. After a breakdown occurs, input parameters such as
offset and bias current may be well out of specification. Diode protection is needed
to prevent base-emitter junction breakdown. Other junctions, such as base-collector
junctions and JFET gate-source junctions do not exhibit the same degradation in
performance on break down, and for these the input current should be limited to
5mA, unless the data sheet specifies a larger value.
10.59
HARDWARE DESIGN TECHNIQUES
INPUT OVERVOLTAGE
■ INPUT SHOULD NOT EXCEED ABSOLUTE MAXIMUM RATINGS
(Usually Specified With Respect to Supply Voltages)
■ A Common Specification Requires the Input Signal Remain
Within 0.3V of the Supply Rails
■ Input Stage Conduction Current Should Be Limited
(Rule of Thumb: < 5mA Unless Otherwise Specified)
■ Avoid Reverse Bias Junction Breakdown in Input Stage Junctions
■ Differential and Common Mode Ratings May Differ
■ No Two Amplifiers are Exactly the Same
■ Some ICs Contain Input Protection (Voltage Clamps, Current Limits,
or Both), but Absolute Maximum Ratings Must Still Be Observed
Figure 10.48
A generalized external protection circuit using two Schottky diodes and an external
current limiting resistor can be used to ensure input protection as shown in Figure
10.49. If the op amp has internal protection diodes to the supplies, they will conduct
at about 0.6V forward drop above or below the supply rails. The external current
limit resistor must be chosen so that the maximum amount of input current is
limited to 5mA. This can result in large values of RLIMIT, and the resulting
increase in noise and offset voltage may not be acceptable. For instance, to protect
against a 100V input at VIN, RLIMIT must be greater than 20kΩ (assuming a worst
case condition where the supply voltages are at zero volts). The external Schottky
protection diodes will begin to conduct at about 0.3V, and overvoltage current is
shunted through them to the supply rails rather than through the internal ones.
This allows RLIMIT to be set by the maximum allowable diode current, which can
be much larger than the internal limit of 5mA. For instance, a 500Ω RLIMIT
resistor would limit the diode current to 200mA for a VIN of 100V.
A protection resistor in series with an amplifier input will also produce a voltage
drop due to the amplifier bias current flowing through it. This drop appears as an
increase in the circuit offset voltage (and, if the bias current changes with
temperature, offset drift). In amplifiers where bias currents are approximately
equal, a resistor in series with each input will tend to balance the effect and reduce
the error.
When using external Schottky clamp diodes to protect operational amplifier inputs,
the effects of diode junction capacitance and leakage current should be evaluated in
the application. Diode junction capacitance and RLIMIT will add an additional pole
in the signal path, and diode leakage currents will double for every 10°C rise in
ambient temperature. Therefore, low leakage diodes should be used such that, at
the highest ambient temperature for the application, the total diode leakage current
is less than one-tenth of the input bias current for the device at that temperature.
Another issue with regard to the use of Schottky diodes is the change in their
10.60
HARDWARE DESIGN TECHNIQUES
forward voltage drop as a function of temperature. These diodes do not, in fact, limit
the signal to ±0.3V at all ambient temperatures, but if the Schottky diodes are at
the same temperature as the op amp, they will limit the voltage to a safe level, even
if they do not limit it at all times to within the data sheet rating. This is true if
over-voltage is only possible at turn-on, when the diodes and the op amp will always
be at the same temperature. If the op amp is warm when it is repowered, however,
steps must be taken to ensure that diodes and op amp are at the same temperature.
GENERALIZED EXTERNAL OVERVOLTAGE
PROTECTION CIRCUIT FOR OP AMPS
VPOS
D1
VIN
RLIMIT
+
I LIMIT
IIN(MAX)
< 5mA
VOUT
D2
_
RFB
VNEG
Figure 10.49
A simplified schematic of the AD620 instrumentation amplifier is shown in Figure
10.50. The 400Ω input resistors are thin-film, and therefore do not behave as
junctions, as would be the case with diffused resistors. The input transistors Q1 and
Q2 have diodes D1 and D2 across their base-emitter junctions to prevent reverse
breakdown. Figure 10.51 shows an equivalent input circuit for an overvoltage
condition. The common-mode voltage at +VIN or –VIN should be limited to 0.3V
above VPOS and 0.3V below VNEG. In addition, the differential input voltage
should be limited to a value which limits the input current to 10mA maximum. The
equivalent circuit shows that the input current flows through the two external
RLIMIT resistors, the two internal RS resistors, the gain-setting resistor RG, and
two diode drops (Q2 and D1). For a given differential input voltage, the input
current is a function of RG and hence the gain. For a gain of 1000, RG = 49.9Ω, and
therefore has more of an impact on the input current than for a gain of 10, where
RG = 5.49kΩ.
10.61
HARDWARE DESIGN TECHNIQUES
AD620 SIMPLIFIED SCHEMATIC
VPOS
49.4kΩ
Ω
RG = G – 1
VB
_
+
_
+
A1
A2
10kΩ
Ω
10kΩ
Ω
Q1
24.7kΩ
Ω
400Ω
Ω
24.7kΩ
Ω
10kΩ
Ω
_
A3
+
10kΩ
Ω
Q2
VREF
400Ω
Ω
RG
D2
–VIN D1
VO
+VIN
VNEG
Figure 10.50
AD620 EQUIVALENT INPUT CIRCUIT WITH OVERVOLTAGE
RLIMIT
+VIN
RS
Q2
400Ω
Ω
D2
AT +VIN AND –VIN:
VNEG – 0.3V < VCM < VPOS + 0.3V
I IN(MAX)
< 10mA
VDIFF
RLIMIT
RS
IIN
RG
D1
Q1
–VIN
400Ω
Ω
VDIFF = I IN( 2RS +2RLIMIT + RG ) + 1.2V
V DIFF(MAX) < I IN(MAX)( 2RS +2RLIMIT + RG ) + 1.2V
Figure 10.51
10.62
HARDWARE DESIGN TECHNIQUES
A generalized external voltage protection circuit for an in-amp is shown in Figure
10.52. The RLIMIT resistors are chosen to limit the maximum current through the
diodes connected to VPOS and VNEG. The Zener diodes or Transient Voltage
Suppressers (TVSs, or TransZorbs™) are selected to limit the maximum differential
input voltage to less than |VPOS – VNEG| if required.
GENERALIZED EXTERNAL PROTECTION
FOR INSTRUMENTATION AMPLIFIER INPUTS
VPOS
RLIMIT
+
*
VOUT
RG
VIN
IN-AMP
VREF
RLIMIT
_
*ZENER DIODES
OR TVSs (TransZorbs™)
LIMIT VDIFF
IF REQUIRED
VNEG
Figure 10.52
The two op amp instrumentation (see Figure 10.53) can generally be protected with
external Schottky diodes to the supplies and current limit resistors. The input
current is not a function of the gain-setting resistor as in the case of the three op
amp in-amp configuration.
ADCs whose input range falls between the supply rails can generally be protected
with external Schottky diodes and a current limit resistor as shown in Figure 10.54.
Even if internal ESD protection diodes are provided, the use of the external ones
allows smaller values of RLIMIT and lower noise and offset errors. ADCs with thinfilm input attenuators, such as the AD7890-10 (see Figure 10.55), can be protected
with Zener diodes on TVSs with an RLIMIT resistor to limit the current through
them.
10.63
HARDWARE DESIGN TECHNIQUES
INPUT PROTECTION FOR TWO OP AMP IN-AMP (AD627)
VPOS
RLIMIT
VIN
+
RLIMIT
VOUT
A2
+
_
A1
R1'
_
V2
R1
R2'
R2
R2
2R2
G = 1 + R1 + R
G
VREF
RG
VNEG
Figure 10.53
INPUT PROTECTION FOR ADCs WITH INPUT RANGES
WITHIN SUPPLY VOLTAGES
VPOS
*
VIN
RLIMIT
AIN
*
INTERNAL
ESD PROTECTION
DIODES (0.6V)
IIN
ADC
VNEG
Choose RLIMIT to Limit IIN Current to 5mA
*Additional External Schottky Diodes Allow Lower
Values of RLIMIT
Figure 10.54
10.64
HARDWARE DESIGN TECHNIQUES
INPUT PROTECTION FOR SINGLE-SUPPLY ADCs WITH
THIN FILM RESISTOR INPUT ATTENUATORS
VPOS = +5V
AD7890-10
RLIMIT << 30kΩ
RLIMIT
VIN
±10V
RANGE
±17V
ABS. MAX.
1N4745
16V, 1W
OR
TVSs
+2.5V
REF
7.5kΩ
Ω
Ω
30kΩ
10kΩ
Ω
VNEG
Figure 10.55
Overvoltage Protection Using CMOS Channel Protectors
The ADG465/ADG466/ADG467 are CMOS channel protectors which are placed in
series with the signal path. The channel protector will protect sensitive components
from voltage transients whether the power supplies are present or not. Because the
channel protection works whether the supplies are present or not, the channel
protectors are ideal for use in applications where correct power sequencing cannot
always be guaranteed (e.g., hot-insertion rack systems) to protect analog inputs.
Each channel protector (see Figure 10.56) has an independent operation and
consists of four MOS transistors - two NMOS and two PMOS. One of the PMOS
devices does not lie directly in the signal path but is used to connect the source of
the second PMOS device to its backgate. This has the effect of lowering the
threshold voltage and so increasing the input signal range of the channel for normal
operation. The source and backgate of the NMOS devices are connected for the same
reason.
The channel protector behaves just like a series resistor (60Ω to 80Ω) during normal
operation, i.e., (VSS + 2V) < VD < (VDD – 1.5V). When a channel's analog input
voltage exceeds this range, one of the MOSFETs will switch off, clamping the output
at either VSS + 2V or VDD – 1.5V. Circuitry and signal source protection is
provided in the event of an overvoltage or power loss. The channel protectors can
withstand overvoltage inputs from VSS – 20V to VDD + 20V with power on (VDD –
VSS = 44V maximum). With power off (VDD = VSS = 0V), maximum input voltage
is ±35V. The channel protectors are very low power devices, and even under fault
conditions, the supply current is limited to sub microampere levels. All transistors
10.65
HARDWARE DESIGN TECHNIQUES
are dielectrically isolated from each other using a trench isolation method thereby
ensuring that the channel protectors cannot latch up.
Figure 10.58 shows a typical application that requires overvoltage and power supply
sequencing protection. The application shows a hot-insertion rack system. This
involves plugging a circuit board or module into a live rack via an edge connector. In
this type of application it is not possible to guarantee correct power supply
sequencing. Correct power supply sequencing means that the power supplies should
be connected before any external signals. Incorrect power sequencing can cause a
CMOS device to latch up. This is true of most CMOS devices regardless of the
functionality. RC networks are used on the supplies of the channel protector to
ensure that the rest of the circuit is powered up before the channel protectors. In
this way, the outputs of the channel protectors are clamped well below VDD and
VSS until the capacitors are charged. The diodes ensure that the supplies on the
channel protector never exceed the supply rails when it is being disconnected. Again
this ensures that signals on the inputs of the CMOS devices never exceed the
supplies.
ADG465, ADG466, and ADG467
SINGLE, TRIPLE, AND OCTAL CHANNEL PROTECTORS
VDD
VD
VS
VDD – 1.5V
VSS
VSS + 2V
VSS
PMOS
NMOS
NMOS
VD
VS
PMOS
VDD
VSS
Figure 10.56
10.66
VDD
HARDWARE DESIGN TECHNIQUES
ADG465, ADG466, and ADG467
CHANNEL PROTECTORS KEY SPECIFICATIONS
■ Low On-Resistance (50Ω for ADG465, 80Ω for ADG466/467)
■ On-Resistance Match: 3%
■ 44V Maximum Supply Voltage, VDD – VSS
■ Fault and Overvoltage Protection up to ±40V
■ Positive Overvoltages Clamped at VDD – 1.5V
■ Negative Overvoltages Clamped at VSS + 2V
■ Signal Paths Open-Circuit with Power Off
■ Latch-Up Proof Construction
Figure 10.57
OVERVOLTAGE AND POWER SUPPLY SEQUENCING
PROTECTION USING THE ADG466
EDGE CONNECTOR
VDD
+5V
VSS
–5V
ANALOG IN
–2.5V TO +2.5V
ADC
LOGIC
CONTROL
LOGIC
LOGIC
GND
ADG466
Figure 10.58
10.67
HARDWARE DESIGN TECHNIQUES
ELECTROSTATIC DISCHARGE
Walt Kester, Wes Freeman, James Bryant
Electrostatic discharge is a single, fast, high current transfer of electrostatic charge
that results from:
(1) Direct contact transfer between two objects at different potentials (sometimes
called contact discharge), or
(2) A high electrostatic field between two objects when they are in close proximity
(sometimes called air discharge)
The prime sources of static electricity are mostly insulators and are typically
synthetic materials, e.g., vinyl or plastic work surfaces, insulated shoes, finished
wood chairs, Scotch tape, bubble pack, soldering irons with ungrounded tips, etc.
Voltage levels generated by these sources can be extremely high since their charge
is not readily distributed over their surfaces or conducted to other objects.
The generation of static electricity caused by rubbing two substances together is
called the triboelectric effect. Examples are shown in Figure 10.59.
EXAMPLES OF ELECTROSTATIC CHARGE GENERATION
■ Walking Across a Carpet
◆ 1000V - 1500V Generated
■ Walking Across a Vinyl Floor
◆ 150V - 250V Generated
■ Handling Material Protected by Clear Plastic Covers
◆ 400V - 600V Generated
■ Handling Polyethylene Bags
◆ 1000V - 2000V Generated
■ Pouring Polyurethane Foam Into a Box
◆ 1200V - 1500V Generated
■ Note: Assume 60% RH. For Low RH (30%), Generated
Voltages Can Be >10 Times Those Listed Above
Figure 10.59
Integrated circuits can be damaged by the high voltages and high peak currents
that can be generated by electrostatic discharge. Precision analog circuits, which
often feature very low bias currents, are more susceptible to damage than common
digital circuits, because the traditional input-protection structures which protect
against ESD damage also increase input leakage.
10.68
HARDWARE DESIGN TECHNIQUES
For the design engineer or technician, the most common manifestation of ESD
damage is a catastrophic failure of the IC. However, exposure to ESD can also cause
increased leakage or degrade other parameters. If a device appears to not meet a
data sheet specification during evaluation, the possibility of ESD damage should be
considered.
UNDERSTANDING ESD DAMAGE
■ ESD Failure Mechanisms:
◆ Dielectric or junction damage
◆ Surface charge accumulation
◆ Conductor fusing
■ ESD Damage Can Cause:
◆ Increased leakage
◆ Degradation in performance
◆ Functional failures of ICs.
■ ESD Damage is often Cumulative:
◆ For example, each ESD "zap" may increase junction
damage until, finally, the device fails.
Figure 10.60
All ESD sensitive devices are shipped in protective packaging. ICs are usually
contained in either conductive foam or antistatic tubes. Either way, the container is
then sealed in a static-dissipative plastic bag. The sealed bag is marked with a
distinctive sticker, such as that shown in Figure 10.61, which outlines the
appropriate handling procedures. In addition, the data sheets for ESD sensitive ICs
generally have a statement to that effect (see Figure 10.62).
Once ESD-sensitive devices are identified, protection is relatively easy. Obviously,
keeping ICs in their original protective packaging as long as possible is the first
step. The second step is to discharge potential ESD sources before damage to the IC
can occur. Discharging a potentially dangerous voltage can be done quickly and
safely through a high impedance.
The key component required for safe ESD handling is a workbench with a staticdissipative surface, as shown in Figure 10.63. This surface is connected to ground
through a 1MΩ resistor, which dissipates static charge while protecting the user
from electrical shock hazards caused by ground faults. If existing bench tops are
nonconductive, a static-dissipative mat should be added, along with a discharge
resistor.
10.69
HARDWARE DESIGN TECHNIQUES
RECOGNIZING ESD SENSITIVE DEVICES
All static sensitive devices are sealed in
protective packaging and marked with
special handling instructions
CAUTION
CAUTION
SENSITIVE ELECTRONIC DEVICES
SENSITIVE ELECTRONIC DEVICES
DO NOT SHIP OR STORE NEAR STRONG
ELECTROSTATIC, ELECTROMAGNETIC,
MAGNETIC, OR RADIOACTIVE FIELDS
DO NOT OPEN EXCEPT AT
APPROVED FIELD FORCE
PROTECTIVE WORK STATION
Figure 10.61
ESD STATEMENT ON DATA SHEETS
OF MOST LINEAR AND MIXED-SIGNAL ICs
WARNING!
ESD SENSITIVE DEVICE
CAUTION
ESD (Electrostatic Discharge) sensitive device. Electrostatic charges as
high as 4000 V readily accumulate on the human body and test equipment
and can discharge without detection. Although the ADXXX features
proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high energy electrostatic discharges. Therefore,
proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Figure 10.62
10.70
HARDWARE DESIGN TECHNIQUES
WORKSTATION FOR HANDLING
ESD-SENSITIVE DEVICES
PERSONNEL
GROUND STRAP
ESD PROTECTIVE
TRAYS, SHUNTS, ETC.
ESD PROTECTIVE
TABLE TOP
ESD
PROTECTIVE
FLOOR OR
MAT
BUILDING FLOOR
Note: Conductive Table Top Sheet Resistance
Ω/
≈ 1MΩ
Figure 10.63
Notice that the surface of the workbench has a moderately high sheet resistance. It
is neither necessary nor desirable to use a low-resistance surface (such as a sheet of
copper-clad PC board) for the work surface. Remember, a high peak current may
flow if a charged IC is discharged through a low impedance. This is precisely what
happens when a charged IC contacts a grounded copper clad board. When the same
charged IC is placed on the surface shown in Figure 10.63, however, the peak
current is not high enough to damage the device.
A conductive wrist strap is also recommended while handling ESD-sensitive
devices. The wrist strap ensures that normal tasks, such as peeling tape off of
packages, will not cause damage to ICs. Again, a 1MΩ resistor, from the wrist strap
to ground, is required for safety.
When building prototype breadboards or assembling PC boards which contain ESDsensitive devices, all passive components should be inserted and soldered before the
ICs. This procedure minimizes the ESD exposure of the sensitive devices. The
soldering iron must, of course, have a grounded tip.
Protecting ICs from ESD requires the participation of both the IC manufacturer and
the customer. IC manufacturers have a vested interest in providing the highest
possible level of ESD protection for their products. IC circuit designers, process
engineers, packaging specialists and others are constantly looking for new and
improved circuit designs, processes, and packaging methods to withstand or shunt
ESD energy.
10.71
HARDWARE DESIGN TECHNIQUES
A complete ESD protection plan, however, requires more than building-ESD
protection into ICs. Users of ICs must also provide their employees with the
necessary knowledge of and training in ESD handling procedures (Figure 10.64).
ESD PROTECTION REQUIRES A PARTNERSHIP
BETWEEN THE IC SUPPLIER AND THE CUSTOMER
ANALOG DEVICES:
■
↓
↓
■
↓
Circuit Design and Fabrication Design and manufacture products with the highest level of ESD
protection consistent with required analog and digital performance.
Pack and Ship Pack in static dissipative material. Mark packages with ESD warning.
CUSTOMERS:
■
↓
■
↓
■
↓
↓
■
Incoming Inspection Inspect at grounded workstation. Minimize handling.
Inventory Control Store in original ESD-safe packaging. Minimize handling.
Manufacturing Deliver to work area in original ESD-safe packaging. Open packages only at
grounded workstation. Package subassemblies in static dissipative packaging.
Pack and Ship Pack in static dissipative material if required. Replacement or optional
boards may require special attention.
Figure 10.64
Special care should be taken when breadboarding and evaluating ICs. The effects of
ESD damage can be cumulative, so repeated mishandling of a device can eventually
cause a failure. Inserting and removing ICs from a test socket, storing devices
during evaluation, and adding or removing external components on the breadboard
should all be done while observing proper ESD precautions. Again, if a device fails
during a prototype system development, repeated ESD stress may be the cause.
The key word to remember with respect to ESD is prevention. There is no way to
undo ESD damage, or to compensate for its effects.
ESD Models and Testing
Some applications have higher ESD sensitivity than others. ICs which are located
on a PC board surrounded by other circuits are generally much less susceptible to
ESD damage than circuits which must interface with other PC boards or the outside
world. These ICs are generally not specified or guaranteed to meet any particular
ESD specification (with the exception of MIL-STD-883 Method 3015 classified
devices). A good example of an ESD-sensitive interface is the RS-232 port on a
computer (see Figure 10.65). The RS-232 driver and receiver ICs are directly in the
firing line for voltage transients as well as ESD. In order to guarantee ESD
performance for such devices, the test methods and limits must be specified.
10.72
HARDWARE DESIGN TECHNIQUES
RS-232 PORT IS VERY SUSCEPTIBLE TO ESD
■ I-O Transceiver Is Directly in the Firing Line for Transients - RS-232
Port Is Particularly Vulnerable
■ I-O Port Is an Open Gateway in the Enclosure
■ Harmonised Standards Are Now Mandatory Requirements in
European Community
Figure 10.65
A host of test waveforms and specifications have been developed to evaluate the
susceptibility of devices to ESD. The three most prominent of these waveforms
currently in use for semiconductor or discrete devices are: The Human Body Model
(HBM), the Machine Model (MM), and the Charged Device Model (CDM). Each of
these models represents a fundamentally different ESD event, consequently,
correlation between the test results for these models is minimal.
Since 1996, all electronic equipment sold to or within the European Community
must meet Electromechanical Compatibility (EMC) levels as defined in specification
IEC1000-4-x. This does not apply to individual ICs, but to the end equipment. These
standards are defined along with test methods in the various IEC1000 specifications
shown in Figure 10.66.
IEC1000-4-2 specifies compliance testing using two coupling methods, contact
discharge and air-gap discharge. Contact discharge calls for a direct connection to
the unit being tested. Air-gap discharge uses a higher test voltage, but does not
make direct contact with the unit under test. With air discharge, the discharge gun
is moved toward the unit under test, developing an arc across the air gap, hence the
term air discharge. This method is influenced by humidity, temperature, barometric
pressure, distance and rate of closure of the discharge gun. The contact-discharge
method, while less realistic, is more repeatable and is gaining acceptance in
preference to the air-gap method.
10.73
HARDWARE DESIGN TECHNIQUES
IEC 1000-4-x BASIC IMMUNITY STANDARDS
FOR ELECTRONIC EQUIPMENT (NOT ICs!)
■ IEC1000-4
Electromagnetic Compatibility EMC
■ IEC1000-4-1 Overview of Immunity Tests
■ IEC1000-4-2 Electrostatic Discharge Immunity (ESD)
■ IEC1000-4-3 Radiated Radio-Frequency Electromagnetic
Field Immunity
■ IEC1000-4-4 Electrical Fast Transients (EFT)
■ IEC1000-4-5 Lightening Surges
■ IEC1000-4-6 Conducted Radio Frequency Disturbances
above 9kHz
■ Compliance Marking:
Figure 10.66
Although very little energy is contained within an ESD pulse, the extremely fast
risetime coupled with high voltages can cause failures in unprotected ICs.
Catastrophic destruction can occur immediately as a result of arcing or heating.
Even if catastrophic failure does not occur immediately, the device may suffer from
parametric degradation, which may result in degraded performance. The
cumulative effects of continuous exposure can eventually lead to complete failure.
I-O lines are particularly vulnerable to ESD damage. Simply touching or plugging in
an I-O cable can result in a static discharge that can damage or completely destroy
the interface product connected to the I-O port (such as RS-232 line drivers and
receivers). Traditional ESD test methods such as MIL-STD-883B Method 3015.7 do
not fully test a product's susceptibility to this type of discharge. This test was
intended to test a product's susceptibility to ESD damage during handling. Each pin
is tested with respect to all other pins. There are some important differences
between the MIL-STD-883B Method 3015.7 test and the IEC test:
(1) The IEC test is much more stringent in terms of discharge energy. The peak
current injected is over four times greater.
(2) The current risetime is significantly faster in the IEC test.
(3) The IEC test is carried out while power is applied to the device.
10.74
HARDWARE DESIGN TECHNIQUES
It is possible that ESD discharge could induce latch-up in the device under test.
This test is therefore more representative of a real-world I-O discharge where the
equipment is operating normally with power applied. For maximum confidence,
however, both tests should be performed on interface devices, thus ensuring
maximum protection both during handling, and later, during field service.
A comparison of the test circuit values for the IEC1000-4-2 model versus the MILSTD-883B Method 3015.7 Human Body Model is shown in Figure 10.67, and the
ESD waveforms are compared in Figure 10.68.
MIL STD 883B METHOD 3015.7 HUMAN BODY MODEL
VERSUS IEC 1000-4-2 ESD TESTING
HIGH
VOLTAGE
GENERATOR
R1
R2
DEVICE
UNDER TEST
C1
ESD TEST METHOD
R2
C1
Human Body Model MIL STD 883B
Method 3015.7
1.5kΩ
100pF
IEC 1000-4-2
330Ω
150pF
NOTE: CONTACT DISCHARGE VOLTAGE SPEC FOR IEC 1000-4-2 IS ±8kV
Figure 10.67
Suitable ESD-protection design measures are relatively easy to incorporate, and
most of the over-voltage protection methods previously discussed in this section will
help. Additional protection can be obtained by the addition of TransZorbs at
appropriate places in the system. For RS-232 and RS-485 line drivers and receivers,
the ADMXXX-E series is supplied with guaranteed 15kV (HBM) ESD specifications.
10.75
HARDWARE DESIGN TECHNIQUES
MIL-STD-883B, METHOD 3015.7 HUMAN BODY MODEL
AND IEC 1000-4-2 ESD WAVEFORMS
HUMAN BODY MODEL
MIL-STD-883B, METHOD 3015.7
IEC 1000-4-2
100%
90%
100%
90%
%
IPEAK
%
IPEAK
36.8%
10%
10%
Time
tRL
Time
0.1to 1 ns
tDL
30ns
60ns
■ Voltage : 8 kV
■ Peak Current :
◆ MIL-883B, Method 3015.7 HBM : 5 A
◆ IEC 1000-4-2 : 25 A
Figure 10.68
CUSTOMER DESIGN PRECAUTIONS FOR ICs WHICH
MUST OPERATE AT ESD-SUSCEPTIBLE INTERFACES
■ Observe all Absolute Maximum Ratings on Data Sheet!
■ Follow General Overvoltage Protection Recommendations
◆ Add Series Resistance to Limit Currents
◆ Add Zeners or Transient Voltage Supressors (TVS) TransZorbs™
for Extra Protection (http://www.gensemi.com)
■ Purchase ESD-Specified Digital Interface Devices Such as
◆ ADMXXX-E Series of RS-232 / RS-485 Drivers / Receivers
(MIL-883B, Method 3015.7: 15kV, IEC 1000-4-2: 8kV)
■ Read AN-397, "Electrically Induced Damage to Standard Linear
Integrated Circuits: The Most Common Causes and the Associated
Fixes to Prevent Reocurrence," by Niall Lyne - Available from Analog
Devices, http://www.analog.com
Figure 10.69
10.76
HARDWARE DESIGN TECHNIQUES
REFERENCES ON ESD AND OVERVOLTAGE:
1.
Amplifier Applications Guide, Section XI, pp. 1-10, Analog Devices,
Incorporated, Norwood, MA, 1992.
2.
Systems Applications Guide, Section 1, pp. 56-72, Analog Devices,
Incorporated, Norwood, MA, 1993.
3.
Linear Design Seminar, Section 1, pp. 19-22, Analog Devices,
Incorporated, Norwood, MA, 1994.
4.
ESD Prevention Manual, Analog Devices, Inc.
5.
MIL-STD-883 Method 3015, Electrostatic Discharge Sensitivity
Classification. Available from Standardization Document Order Desk,
700 Robbins Ave., Building #4, Section D, Philadelphia, PA 19111-5094.
6.
EIAJ ED-4701 Test Method C-111, Electrostatic Discharges. Available from
the Japan Electronics Bureau, 250 W 34th St., New York NY 10119, Attn.:
Tomoko.
7.
ESD Association Standard S5.2 for Electrostatic Discharge (ESD) Sensitivity
Testing -Machine Model (MM)- Component Level. Available from the ESD
Association, Inc., 200 Liberty Plaza, Rome, NY 13440.
8.
ESD Association Draft Standard DS5.3 for Electrostatic Discharge (ESD)
Sensitivity Testing - Charged Device Model (CDM) Component Testing.
Available from the ESD Association, Inc., 200 Liberty Plaza, Rome, NY
13440.
9.
Niall Lyne, Electrical Overstress Damage to CMOS Converters, Application
Note AN-397, Analog Devices, 1995, http://www.analog.com.
10.
How to Reliably Protect CMOS Circuits Against Power Supply
Overvoltaging, Application Note AN-311, Analog Devices,
http://www.analog.com
11.
ADM3311E RS-232 Port Transceiver Data Sheet, Analog Devices, Inc.,
http://www.analog.com.
12.
TransZorbs Available from General Semiconductor, Inc., 10 Melville
Park Road, Melville, NY, 11747-3113, 516-847-3000,
http://www.gensemi.com.
10.77
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