Fundamentals of Analog to Digital Conversion

Fundamentals of Analog to Digital Conversion
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
ANALOG-DIGITAL CONVERSION
1. Data Converter History
2. Fundamentals of Sampled Data Systems
2.1
2.2
2.3
2.4
2.5
Coding and Quantizing
Sampling Theory
Data Converter AC Errors
General Data Converter Specifications
Defining the Specifications
3. Data Converter Architectures
4. Data Converter Process Technology
5. Testing Data Converters
6. Interfacing to Data Converters
7. Data Converter Support Circuits
8. Data Converter Applications
9. Hardware Design Techniques
I. Index
ANALOG-DIGITAL CONVERSION
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
CHAPTER 2
FUNDAMENTALS OF SAMPLED DATA
SYSTEMS
SECTION 2.1: CODING AND QUANTIZING
Walt Kester, Dan Sheingold, James Bryant
Analog-to-digital converters (ADCs) translate analog quantities, which are
characteristic of most phenomena in the "real world," to digital language, used in
information processing, computing, data transmission, and control systems. Digital-toanalog converters (DACs) are used in transforming transmitted or stored data, or the
results of digital processing, back to "real-world" variables for control, information
display, or further analog processing. The relationships between inputs and outputs of
DACs and ADCs are shown in Figure 2.1.
VREF
MSB
DIGITAL
INPUT
N-BITS
+FS
N-BIT
DAC
ANALOG
OUTPUT
LSB
RANGE
(SPAN)
0 OR –FS
VREF
MSB
+FS
RANGE
(SPAN)
0 OR –FS
ANALOG
INPUT
N-BIT
ADC
DIGITAL
OUTPUT
N-BITS
LSB
Figure 2.1: Digital-to-Analog Converter (DAC) and Analog-to-Digital Converter
(ADC) Input and Output Definitions
Analog input variables, whatever their origin, are most frequently converted by
transducers into voltages or currents. These electrical quantities may appear (1) as fast or
slow "dc" continuous direct measurements of a phenomenon in the time domain, (2) as
modulated ac waveforms (using a wide variety of modulation techniques), (3) or in some
combination, with a spatial configuration of related variables to represent shaft angles.
2.1
ANALOG-DIGITAL CONVERSION
Examples of the first are outputs of thermocouples, potentiometers on dc references, and
analog computing circuitry; of the second, "chopped" optical measurements, ac strain
gage or bridge outputs, and digital signals buried in noise; and of the third, synchros and
resolvers.
The analog variables to be dealt with in this chapter are those involving voltages or
currents representing the actual analog phenomena. They may be either wideband or
narrowband. They may be either scaled from the direct measurement, or subjected to
some form of analog pre-processing, such as linearization, combination, demodulation,
filtering, sample-hold, etc.
As part of the process, the voltages and currents are "normalized" to ranges compatible
with assigned ADC input ranges. Analog output voltages or currents from DACs are
direct and in normalized form, but they may be subsequently post-processed (e.g., scaled,
filtered, amplified, etc.).
Information in digital form is normally represented by arbitrarily fixed voltage levels
referred to "ground," either occurring at the outputs of logic gates, or applied to their
inputs. The digital numbers used are all basically binary; that is, each "bit," or unit of
information has one of two possible states. These states are "off," "false," or "0," and
"on," "true," or "1." It is also possible to represent the two logic states by two different
levels of current, however this is much less popular than using voltages. There is also no
particular reason why the voltages need be referenced to ground—as in the case of
emitter-coupled-logic (ECL), positive-emitter-coupled-logic (PECL) or low-voltagedifferential-signaling logic (LVDS) for example.
Words are groups of levels representing digital numbers; the levels may appear
simultaneously in parallel, on a bus or groups of gate inputs or outputs, serially (or in a
time sequence) on a single line, or as a sequence of parallel bytes (i.e., "byte-serial") or
nibbles (small bytes). For example, a 16-bit word may occupy the 16 bits of a 16-bit bus,
or it may be divided into two sequential bytes for an 8-bit bus, or four 4-bit nibbles for a
4-bit bus.
Although there are several systems of logic, the most widely used choice of levels are
those used in TTL (transistor-transistor logic) and, in which positive true, or 1,
corresponds to a minimum output level of +2.4 V (inputs respond unequivocally to "1"
for levels greater than 2.0 V); and false, or 0, corresponds to a maximum output level of
+0.4 V (inputs respond unequivocally to "0" for anything less than +0.8 V). It should be
noted that even though CMOS is more popular today than TTL, CMOS logic levels are
generally made to be compatible with the older TTL logic standard.
A unique parallel or serial grouping of digital levels, or a number, or code, is assigned to
each analog level which is quantized (i.e., represents a unique portion of the analog
range). A typical digital code would be this array:
a7 a6 a5 a4 a3 a2 a1 a0 = 1 0 1 1 1 0 0 1
It is composed of eight bits. The "1" at the extreme left is called the "most significant bit"
(MSB, or Bit 1), and the one at the right is called the "least significant bit" (LSB, or
2.2
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
bit N: 8 in this case). The meaning of the code, as either a number, a character, or a
representation of an analog variable, is unknown until the code and the conversion
relationship have been defined. It is important not to confuse the designation of a
particular bit (i.e., Bit 1, Bit 2, etc.) with the subscripts associated with the "a" array. The
subscripts correspond to the power of 2 associated with the weight of a particular bit in
the sequence.
The best-known code (other than base 10) is natural or straight binary (base 2). Binary
codes are most familiar in representing integers; i.e., in a natural binary integer code
having N bits, the LSB has a weight of 20 (i.e., 1), the next bit has a weight of 21 (i.e., 2),
and so on up to the MSB, which has a weight of 2N–1 (i.e., 2N/2). The value of a binary
number is obtained by adding up the weights of all non-zero bits. When the weighted bits
are added up, they form a unique number having any value from 0 to 2N – 1. Each
additional trailing zero bit, if present, essentially doubles the size of the number.
In converter technology, full-scale (abbreviated FS) is independent of the number of bits
of resolution, N. A more useful coding is fractional binary which is always normalized to
full-scale. Integer binary can be interpreted as fractional binary if all integer values are
divided by 2N. For example, the MSB has a weight of ½ (i.e., 2(N–1)/2N = 2–1), the next bit
has a weight of ¼ (i.e., 2–2), and so forth down to the LSB, which has a weight of 1/2N
(i.e., 2–N). When the weighted bits are added up, they form a number with any of 2N
values, from 0 to (1 – 2–N) of full-scale. Additional bits simply provide more fine
structure without affecting full-scale range. The relationship between base-10 numbers
and binary numbers (base 2) are shown in Figure 2.2 along with examples of each.
WHOLE NUMBERS:
Number10 = aN–12N–1 + aN –22N–2 + … +a121 + a020
MSB
LSB
Example: 10112 = (1×23) + (0×22)+ (1×21)+ (1×20)
= 8
+ 0 + 2 + 1
= 1110
FRACTIONAL NUMBERS:
Number10 = aN–12–1 + aN–2 2–2 + … + a12–(N–1) + a02–N
MSB
LSB
Example: 0.10112 = (1×0.5) + (0×0.25) + (1×0.125) + (1×0.0625)
= 0.5 +
0
+ 0.125 + 0.0625 = 0.687510
Figure 2.2: Representing a Base-10 Number with a Binary Number (Base 2)
Unipolar Codes
In data conversion systems, the coding method must be related to the analog input range
(or span) of an ADC or the analog output range (or span) of a DAC. The simplest case is
2.3
ANALOG-DIGITAL CONVERSION
when the input to the ADC or the output of the DAC is always a unipolar positive voltage
(current outputs are very popular for DAC outputs, much less for ADC inputs). The most
popular code for this type of signal is straight binary and is shown in Figure 2.3 for a
4-bit converter. Notice that there are 16 distinct possible levels, ranging from the allzeros code 0000, to the all-ones code 1111. It is important to note that the analog value
represented by the all-ones code is not full-scale (abbreviated FS), but FS – 1 LSB. This
is a common convention in data conversion notation and applies to both ADCs and
DACs. Figure 2.3 gives the base-10 equivalent number, the value of the base-2 binary
code relative to full-scale (FS), and also the corresponding voltage level for each code
(assuming a +10 V full-scale converter. The Gray code equivalent is also shown, and will
be discussed shortly.
BASE 10
NUMBER
+15
+14
+13
+12
+11
+10
+9
+8
+7
+6
+5
+4
+3
+2
+1
0
SCALE
+FS – 1LSB = +15/16 FS
+7/8 FS
+13/16 FS
+3/4 FS
+11/16 FS
+5/8 FS
+9/16 FS
+1/2 FS
+7/16 FS
+3/8 FS
+5/16 FS
+1/4 FS
+3/16 FS
+1/8 FS
1LSB = +1/16 FS
0
+10V FS
BINARY
GRAY
9.375
8.750
8.125
7.500
6.875
6.250
5.625
5.000
4.375
3.750
3.125
2.500
1.875
1.250
0.625
0.000
1111
1110
1101
1100
1011
1010
1001
1000
0111
0110
0101
0100
0011
0010
0001
0000
1000
1001
1011
1010
1110
1111
1101
1100
0100
0101
0111
0110
0010
0011
0001
0000
Figure 2.3: Unipolar Binary Codes, 4-bit Converter
Figure 2.4 shows the transfer function for an ideal 3-bit DAC with straight binary input
coding. Notice that the analog output is zero for the all-zeros input code. As the digital
input code increases, the analog output increases 1 LSB (1/8 scale in this example) per
code. The most positive output voltage is 7/8 FS, corresponding to a value equal to FS –
1 LSB. The mid-scale output of 1/2 FS is generated when the digital input code is 100.
The transfer function of an ideal 3-bit ADC is shown in Figure 2.5. There is a range of
analog input voltage over which the ADC will produce a given output code; this range is
the quantization uncertainty and is equal to 1 LSB. Note that the width of the transition
regions between adjacent codes is zero for an ideal ADC. In practice, however, there is
always transition noise associated with these levels, and therefore the width is non-zero.
It is customary to define the analog input corresponding to a given code by the code
center which lies halfway between two adjacent transition regions (illustrated by the
black dots in the diagram). This requires that the first transition region occur at ½ LSB.
The full-scale analog input voltage is defined by 7/8 FS, (FS – 1 LSB).
2.4
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
FS
7/8
3/4
ANALOG
OUTPUT
5/8
1/2
3/8
1/4
1/8
0
000
001
010
011
100
101
110
111
DIGITAL INPUT (STRAIGHT BINARY)
Figure 2.4: Transfer Function for Ideal Unipolar 3-bit DAC
111
110
DIGITAL
OUTPUT
(STRAIGHT
BINARY)
101
100
1 LSB
011
1/2 LSB
010
001
000
0
1/8
1/4
3/8
1/2
5/8
3/4
7/8
FS
ANALOG INPUT
Figure 2.5: Transfer Function for Ideal Unipolar 3-bit ADC
2.5
ANALOG-DIGITAL CONVERSION
Gray Code
Another code worth mentioning at this point is the Gray code (or reflective-binary) which
was invented by Elisha Gray in 1878 (Reference 1) and later re-invented by Frank Gray
in 1949 (see Reference 2). The Gray code equivalent of the 4-bit straight binary code is
also shown in Figure 2.3. Although it is rarely used in computer arithmetic, it has some
useful properties which make it attractive to A/D conversion. Notice that in Gray code, as
the number value changes, the transitions from one code to the next involve only one bit
at a time. Contrast this to the binary code where all the bits change when making the
transition between 0111 and 1000. Some ADCs make use of it internally and then convert
the Gray code to a binary code for external use.
One of the earliest practical ADCs to use the Gray code was a 7-bit, 100 kSPS electron
beam encoder developed by Bell Labs and described in a 1948 reference (Reference 3).
The basic electron beam coder concepts are shown in Figure 2.6 for a 4-bit device. The
early tubes operated in the serial mode (A). The analog signal is first passed through a
sample-and-hold, and during the "hold" interval, the beam is swept horizontally across
the tube. The Y-deflection for a single sweep therefore corresponds to the value of the
analog signal from the sample-and-hold. The shadow mask is coded to produce the
proper binary code, depending on the vertical deflection. The code is registered by the
collector, and the bits are generated in serial format. Later tubes used a fan-shaped beam
(shown in Figure 2.6B), creating a "flash" converter delivering a parallel output word.
X Deflectors
(A) SERIAL MODE
Y Deflectors
Shadow Mask
Collector
Electron gun
(B) PARALLEL MODE
Y Deflectors
Shadow Mask
Collector
Electron gun
Figure 2.6: The Electron Beam Coder: (A) Serial Mode and (B) Parallel
or "Flash" Mode
2.6
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
Early electron tube coders used a binary-coded shadow mask, and large errors can occur
if the beam straddles two adjacent codes and illuminates both of them. The way these
errors occur is illustrated in Figure 2.7A, where the horizontal line represents the beam
sweep at the mid-scale transition point (transition between code 0111 and code 1000).
For example, an error in the most significant bit (MSB) produces an error of ½ scale.
These errors were minimized by placing fine horizontal sensing wires across the
boundaries of each of the quantization levels. If the beam initially fell on one of the
wires, a small voltage was added to the vertical deflection voltage which moved the beam
away from the transition region.
(A) 4-BIT BINARY CODE
SHADOW MASK
(B) 4-BIT REFLECTED-BINARY CODE
(GRAY CODE)
1111
1110
1101
1100
1011
1010
1001
1000
0111
0110
0101
0100
0011
0010
0001
0000
MSB
LSB
SHADOW MASK
1000
1001
1011
1010
1110
1111
1101
1100
0100
0101
0111
0110
0010
0011
0001
0000
MSB
LSB
Figure 2.7: Electron Beam Coder Shadow Masks for
Binary Code (A) and Gray Code (B)
The errors associated with binary shadow masks were eliminated by using a Gray code
shadow mask as shown in Figure 2.7B. As mentioned above, the Gray code has the
property that adjacent levels differ by only one digit in the corresponding Gray-coded
word. Therefore, if there is an error in a bit decision for a particular level, the
corresponding error after conversion to binary code is only one least significant bit
(LSB). In the case of mid-scale, note that only the MSB changes. It is interesting to note
that this same phenomenon can occur in modern comparator-based flash converters due
to comparator metastability. With small overdrive, there is a finite probability that the
output of a comparator will generate the wrong decision in its latched output, producing
the same effect if straight binary decoding techniques are used. In many cases, Gray
code, or "pseudo-Gray" codes are used to decode the comparator bank. The Gray code
output is then latched, converted to binary, and latched again at the final output.
As a historical note, in spite of the many mechanical and electrical problems relating to
beam alignment, electron tube coding technology reached its peak in the mid-l960s with
an experimental 9-bit coder capable of 12-MSPS sampling rates (Reference 4). Shortly
thereafter, however, advances in all solid-state ADC techniques made the electron tube
technology obsolete.
2.7
ANALOG-DIGITAL CONVERSION
Other examples where Gray code is often used in the conversion process to minimize
errors are shaft encoders (angle-to-digital) and optical encoders.
ADCs which use the Gray code internally almost always convert the Gray code output to
binary for external use. The conversion from Gray-to-binary and binary-to-Gray is easily
accomplished with the exclusive-or logic function as shown in Figure 2.8.
BINARY
GRAY
MSB
1
1
1
0
1
1
0
1
GRAY
1
BINARY
MSB
1
1
0
1
0
1
1
Figure 2.8: Binary-to-Gray and Gray-to-Binary Conversion
Using the Exclusive-Or Logic Function
Bipolar Codes
In many systems, it is desirable to represent both positive and negative analog quantities
with binary codes. Either offset binary, twos complement, ones complement, or sign
magnitude codes will accomplish this, but offset binary and twos complement are by far
the most popular. The relationships between these codes for a 4-bit systems is shown in
Figure 2.9. Note that the values are scaled for a ±5-V full-scale input/output voltage
range.
For offset binary, the zero signal value is assigned the code 1000. The sequence of codes
is identical to that of straight binary. The only difference between a straight and offset
binary system is the half-scale offset associated with analog signal. The most negative
value (–FS + 1 LSB) is assigned the code 0001, and the most positive value
(+FS – 1 LSB) is assigned the code 1111. Note that in order to maintain perfect
symmetry about mid-scale, the all-zeros code (0000) representing negative full-scale
(–FS) is not normally used in computation. It can be used to represent a negative offrange condition or simply assigned the value of the 0001 (–FS + 1 LSB).
2.8
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
BASE 10
NUMBER
SCALE
±5V FS
OFFSET
BINARY
TWOS
COMP.
+7
+6
+5
+4
+3
+2
+1
0
–1
–2
–3
–4
–5
–6
–7
–8
+FS – 1LSB = +7/8 FS
+3/4 FS
+5/8 FS
+1/2 FS
+3/8 FS
+1/4 FS
+1/8 FS
0
– 1/8 FS
– 1/4 FS
– 3/8 FS
–1/2 FS
–5/8 FS
–3/4 FS
– FS + 1LSB = –7/8 FS
– FS
+4.375
+3.750
+3.125
+2.500
+1.875
+1.250
+0.625
0.000
–0.625
–1.250
–1.875
–2.500
–3.125
–3.750
–4.375
–5.000
1111
1110
1101
1100
1011
1010
1001
1000
0111
0110
0101
0100
0011
0010
0001
0000
0111
0110
0101
0100
0011
0010
0001
0000
1111
1110
1101
1100
1011
1010
1001
1000
NOT NORMALLY USED
IN COMPUTATIONS (SEE TEXT)
*
0+
0–
ONES
COMP.
0111
0110
0101
0100
0011
0010
0001
*0 0 0 0
1110
1101
1100
1011
1010
1001
1000
ONES
COMP.
0000
1111
SIGN
MAG.
0111
0110
0101
0100
0011
0010
0001
*1 0 0 0
1001
1010
1011
1100
1101
1110
1111
SIGN
MAG.
0000
1000
Figure 2.9: Bipolar Codes, 4-bit Converter
The relationship between the offset binary code and the analog output range of a bipolar
3-bit DAC is shown in Figure 2.10. The analog output of the DAC is zero for the zerovalue input code 100. The most negative output voltage is generally defined by the 001
code (–FS + 1 LSB), and the most positive by 111 (+FS – 1 LSB). The output voltage for
the 000 input code is available for use if desired, but makes the output non-symmetrical
about zero and complicates the mathematics.
The offset binary output code for a bipolar 3-bit ADC as a function of its analog input is
shown in Figure 2.11. Note that zero analog input defines the center of the mid-scale
code 100. As in the case of bipolar DACs, the most negative input voltage is generally
defined by the 001 code (–FS + 1 LSB), and the most positive by 111 (+FS – 1 LSB). As
discussed above, the 000 output code is available for use if desired, but makes the output
non-symmetrical about zero and complicates the mathematics.
Twos complement is identical to offset binary with the most-significant-bit (MSB)
complemented (inverted). This is obviously very easy to accomplish in a data converter,
using a simple inverter or taking the complementary output of a "D" flip-flop. The
popularity of twos complement coding lies in the ease with which mathematical
operations can be performed in computers and DSPs. Twos complement, for conversion
purposes, consists of a binary code for positive magnitudes (0 sign bit), and the twos
complement of each positive number to represent its negative. The twos complement is
formed arithmetically by complementing the number and adding 1 LSB. For example,
–3/8 FS is obtained by taking the twos complement of +3/8 FS. This is done by first
complementing +3/8 FS, 0011 obtaining 1100. Adding 1 LSB, we obtain 1101.
2.9
ANALOG-DIGITAL CONVERSION
+FS
+3/8
+1/4
ANALOG
OUTPUT
+1/8
0
–1/8
–1/4
–3/8
–FS
000
001
010
011
100
101
110
111
DIGITAL INPUT (OFFSET BINARY)
Figure 2.10: Transfer Function for Ideal Bipolar 3-bit DAC
111
110
DIGITAL
OUTPUT
(OFFSET
BINARY)
101
100
011
010
001
000
–FS
–3/8
–1/4
–1/8
0
+1/8 +1/4
+3/8
+FS
ANALOG INPUT
Figure 2.11: Transfer Function for Ideal Bipolar 3-bit ADC
2.10
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
Twos complement makes subtraction easy. For example, to subtract 3/8 FS from 4/8 FS,
add 4/8 to –3/8, or 0100 to 1101. The result is 0001, or 1/8, disregarding the extra carry.
Ones complement can also be used to represent negative numbers, although it is much
less popular than twos complement and rarely used today. The ones complement is
obtained by simply complementing all of a positive number's digits. For instance, the
ones complement of 3/8 FS (0011) is 1100. A ones complemented code can be formed by
complementing each positive value to obtain its corresponding negative value. This
includes zero, which is then represented by either of two codes, 0000 (referred to as 0+)
or 1111 (referred to as 0–). This ambiguity must be dealt with mathematically, and
presents obvious problems relating to ADCs and DACs for which there is a single code
which represents zero.
Sign-magnitude would appear to be the most straightforward way of expressing signed
analog quantities digitally. Simply determine the code appropriate for the magnitude and
add a polarity bit. Sign-magnitude BCD is popular in bipolar digital voltmeters, but has
the problem of two allowable codes for zero. It is therefore unpopular for most
applications involving ADCs or DACs.
Figure 2.12 summarizes the relationships between the various bipolar codes: offset
binary, twos complement, ones complement, and sign-magnitude and shows how to
convert between them.
To Convert From
To
Sign Magnitude
2's Complement
Sign Magnitude
No
Change
If MSB = 1,
complement
other bits,
add 00…01
2's Complement
If MSB = 1,
complement
other bits,
add 00…01
No
Change
Offset binary
Complement MSB
If new MSB = 0
complement
other bits,
add 00…01
Complement
MSB
1's Complement
If MSB = 1,
complement
other bits
If MSB = 1,
add 11…11
Offset Binary
Complement MSB
If new MSB = 1,
complement
other bits,
add 00…01
Complement
MSB
No
Change
1's Complement
If MSB = 1,
complement
other bits
If MSB = 1,
add 00…01
Complement MSB
If new MSB = 0,
add 00…01
Complement MSB
If new MSB = 1,
add 11…11
No
Change
Figure 2.12: Relationships Among Bipolar Codes
The last code to be considered in this section is binary-coded-decimal (BCD), where each
base-10 digit (0 to 9) in a decimal number is represented as the corresponding 4-bit
straight binary word as shown in Figure 2.13. The minimum digit 0 is represented as
2.11
ANALOG-DIGITAL CONVERSION
0000, and the digit 9 by 1001. This code is relatively inefficient, since only 10 of the 16
code states for each decade are used. It is, however, a very useful code for interfacing to
decimal displays such as in digital voltmeters.
BASE 10
NUMBER
SCALE
+15 +FS – 1LSB = +15/16 FS
+7/8 FS
+14
+13/16 FS
+13
+3/4 FS
+12
+11/16 FS
+11
+5/8 FS
+10
+9/16 FS
+9
+1/2 FS
+8
+7/16 FS
+7
+3/8 FS
+6
+5/16 FS
+5
+1/4 FS
+4
+3/16 FS
+3
+1/8 FS
+2
1LSB = +1/16 FS
+1
0
0
+10V FS
9.375
8.750
8.125
7.500
6.875
6.250
5.625
5.000
4.375
3.750
3.125
2.500
1.875
1.250
0.625
0.000
DECADE 1 DECADE 2 DECADE 3 DECADE 4
1001
1000
1000
0111
0110
0110
0101
0101
0100
0011
0011
0010
0001
0001
0000
0000
0011
0111
0001
0101
1000
0010
0110
0000
0011
0111
0001
0101
1000
0010
0110
0000
0111
0101
0010
0000
0111
0101
0010
0000
0111
0101
0010
0000
0111
0101
0010
0000
0101
0000
0101
0000
0101
0000
0101
0000
0101
0000
0101
0000
0101
0000
0101
0000
Figure 2.13: Binary Coded Decimal (BCD) Code
Complementary Codes
Some forms of data converters (for example, early DACs using monolithic NPN quad
current switches), require standard codes such as natural binary or BCD, but with all bits
represented by their complements. Such codes are called complementary codes. All the
codes discussed thus far have complementary codes which can be obtained by this
method. A complementary code should not be confused with a ones complement or a
twos complement code.
In a 4-bit complementary-binary converter, 0 is represented by 1111, half-scale by 0111,
and FS – 1 LSB by 0000. In practice, the complementary code can usually be obtained by
using the complementary output of a register rather than the true output, since both are
available.
Sometimes the complementary code is useful in inverting the analog output of a DAC.
Today many DACs provide differential outputs which allow the polarity inversion to be
accomplished without modifying the input code. Similarly, many ADCs provide
differential logic inputs which can be used to accomplish the polarity inversion.
DAC and ADC Static Transfer Functions and DC Errors
The most important thing to remember about both DACs and ADCs is that either the
input or output is digital, and therefore the signal is quantized. That is, an N-bit word
represents one of 2N possible states, and therefore an N-bit DAC (with a fixed reference)
2.12
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
can have only 2N possible analog outputs, and an N-bit ADC can have only 2N possible
digital outputs. As previously discussed, the analog signals will generally be voltages or
currents.
The resolution of data converters may be expressed in several different ways: the weight
of the Least Significant Bit (LSB), parts per million of full-scale (ppm FS), millivolts
(mV), etc. Different devices (even from the same manufacturer) will be specified
differently, so converter users must learn to translate between the different types of
specifications if they are to compare devices successfully. The size of the least significant
bit for various resolutions is shown in Figure 2.14.
RESOLUTION
N
2N
VOLTAGE
(10V FS)
ppm FS
% FS
dB FS
2-bit
4
2.5 V
250,000
25
– 12
4-bit
16
625 mV
62,500
6.25
– 24
6-bit
64
156 mV
15,625
1.56
– 36
8-bit
256
39.1 mV
3,906
0.39
– 48
10-bit
1,024
9.77 mV (10 mV)
977
0.098
– 60
12-bit
4,096
2.44 mV
244
0.024
– 72
14-bit
16,384
610 µV
61
0.0061
– 84
16-bit
65,536
153 µV
15
0.0015
– 96
18-bit
262,144
38 µV
4
0.0004
– 108
20-bit
1,048,576
9.54 µV (10 µV)
1
0.0001
– 120
22-bit
4,194,304
2.38 µV
0.24
0.000024
– 132
16,777,216
596 nV*
0.06
0.000006
– 144
24-bit
*600nV is the Johnson Noise in a 10kHz BW of a 2.2kΩ Resistor @ 25°C
Remember: 10-bits and 10V FS yields an LSB of 10mV, 1000ppm, or 0.1%.
All other values may be calculated by powers of 2.
Figure 2.14: Quantization: The Size of a Least Significant Bit (LSB)
Before we can consider the various architectures used in data converters, it is necessary
to consider the performance to be expected, and the specifications which are important.
The following sections will consider the definition of errors and specifications used for
data converters. This is important in understanding the strengths and weaknesses of
different ADC/DAC architectures.
The first applications of data converters were in measurement and control where the
exact timing of the conversion was usually unimportant, and the data rate was slow. In
such applications, the dc specifications of converters are important, but timing and ac
specifications are not. Today many, if not most, converters are used in sampling and
reconstruction systems where ac specifications are critical (and dc ones may not be)—
these will be considered in Section 2.3 of this chapter.
Figure 2.15 shows the ideal transfer characteristics for a 3-bit unipolar DAC and a 3-bit
unipolar ADC. In a DAC, both the input and the output are quantized, and the graph
consists of eight points—while it is reasonable to discuss the line through these points, it
2.13
ANALOG-DIGITAL CONVERSION
is very important to remember that the actual transfer characteristic is not a line, but a
number of discrete points.
DAC
FS
ADC
111
110
ANALOG
OUTPUT
DIGITAL
OUTPUT
101
100
011
QUANTIZATION
UNCERTAINTY
010
QUANTIZATION
UNCERTAINTY
001
000
000
001
010
011
100
101
DIGITAL INPUT
110
111
ANALOG INPUT
FS
Figure 2.15: Transfer Functions for Ideal 3-Bit DAC and ADC
The input to an ADC is analog and is not quantized, but its output is quantized. The
transfer characteristic therefore consists of eight horizontal steps. When considering the
offset, gain and linearity of an ADC we consider the line joining the midpoints of these
steps—often referred to as the code centers.
For both DACs and ADCs, digital full-scale (all "1"s) corresponds to 1 LSB below the
analog full-scale (FS). The (ideal) ADC transitions take place at ½ LSB above zero, and
thereafter every LSB, until 1½ LSB below analog full-scale. Since the analog input to an
ADC can take any value, but the digital output is quantized, there may be a difference of
up to ½ LSB between the actual analog input and the exact value of the digital output.
This is known as the quantization error or quantization uncertainty as shown in Figure
2.15. In ac (sampling) applications this quantization error gives rise to quantization noise
which will be discussed in Section 2.3 of this chapter.
As previously discussed, there are many possible digital coding schemes for data
converters: straight binary, offset binary, 1's complement, 2's complement, sign
magnitude, gray code, BCD and others. This section, being devoted mainly to the analog
issues surrounding data converters, will use simple binary and offset binary in its
examples and will not consider the merits and disadvantages of these, or any other forms
of digital code.
The examples in Figure 2.15 use unipolar converters, whose analog port has only a single
polarity. These are the simplest type, but bipolar converters are generally more useful in
real-world applications. There are two types of bipolar converters: the simpler is merely a
unipolar converter with an accurate 1 MSB of negative offset (and many converters are
arranged so that this offset may be switched in and out so that they can be used as either
unipolar or bipolar converters at will), but the other, known as a sign-magnitude
converter is more complex, and has N bits of magnitude information and an additional bit
which corresponds to the sign of the analog signal. Sign-magnitude DACs are quite rare,
2.14
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
and sign-magnitude ADCs are found mostly in digital voltmeters (DVMs). The unipolar,
offset binary, and sign-magnitude representations are shown in Figure 2.16.
UNIPOLAR
OFFSET BIPOLAR
FS – 1 LSB
FS – 1 LSB
0
SIGN MAGNITUDE
BIPOLAR
CODE
FS – 1 LSB
CODE
0
ALL
"1"s
CODE
0
ALL
"1"s
1 AND ALL "0"s
–FS
–(FS – 1 LSB)
Figure 2.16: Unipolar and Bipolar Converters
The four dc errors in a data converter are offset error, gain error, and two types of
linearity error (differential and integral). Offset and gain errors are analogous to offset
and gain errors in amplifiers as shown in Figure 2.17 for a bipolar input range. (Though
offset error and zero error, which are identical in amplifiers and unipolar data converters,
are not identical in bipolar converters and should be carefully distinguished.)
The transfer characteristics of both DACs and ADCs may be expressed as a straight line
given by D = K + GA, where D is the digital code, A is the analog signal, and K and G
are constants. In a unipolar converter, the ideal value of K is zero; in an offset bipolar
converter it is –1 MSB. The offset error is the amount by which the actual value of K
differs from its ideal value.
The gain error is the amount by which G differs from its ideal value, and is generally
expressed as the percentage difference between the two, although it may be defined as the
gain error contribution (in mV or LSB) to the total error at full-scale. These errors can
usually be trimmed by the data converter user. Note, however, that amplifier offset is
trimmed at zero input, and then the gain is trimmed near to full-scale. The trim algorithm
for a bipolar data converter is not so straightforward.
The integral linearity error of a converter is also analogous to the linearity error of an
amplifier, and is defined as the maximum deviation of the actual transfer characteristic of
the converter from a straight line, and is generally expressed as a percentage of full-scale
(but may be given in LSBs). For an ADC, the most popular convention is to draw the
straight line through the mid-points of the codes, or the code centers. There are two
2.15
ANALOG-DIGITAL CONVERSION
common ways of choosing the straight line: end point and best straight line as shown in
Figure 2.18.
+FS
+FS
ACTUAL
ACTUAL
IDEAL
IDEAL
0
0
ZERO ERROR
OFFSET
ERROR
ZERO ERROR
NO GAIN ERROR:
ZERO ERROR = OFFSET ERROR
–FS
–FS
WITH GAIN ERROR:
OFFSET ERROR = 0
ZERO ERROR RESULTS
FROM GAIN ERROR
Figure 2.17: Bipolar Data Converter Offset and Gain Error
END POINT METHOD
BEST STRAIGHT LINE METHOD
OUTPUT
LINEARITY
ERROR = X
INPUT
LINEARITY
ERROR ≈ X/2
INPUT
Figure 2.18: Method of Measuring Integral Linearity Errors
(Same Converter on Both Graphs)
In the end point system, the deviation is measured from the straight line through the
origin and the full-scale point (after gain adjustment). This is the most useful integral
linearity measurement for measurement and control applications of data converters (since
error budgets depend on deviation from the ideal transfer characteristic, not from some
arbitrary "best fit"), and is the one normally adopted by Analog Devices, Inc.
The best straight line, however, does give a better prediction of distortion in ac
applications, and also gives a lower value of "linearity error" on a data sheet. The best fit
straight line is drawn through the transfer characteristic of the device using standard
2.16
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
curve fitting techniques, and the maximum deviation is measured from this line. In
general, the integral linearity error measured in this way is only 50% of the value
measured by end point methods. This makes the method good for producing impressive
data sheets, but it is less useful for error budget analysis. For ac applications it is better to
specify distortion than dc linearity, so it is rarely necessary to use the best straight line
method to define converter linearity.
The other type of converter nonlinearity is differential nonlinearity (DNL). This relates to
the linearity of the code transitions of the converter. In the ideal case, a change of 1 LSB
in digital code corresponds to a change of exactly 1 LSB of analog signal. In a DAC, a
change of 1 LSB in digital code produces exactly 1 LSB change of analog output, while
in an ADC there should be exactly 1 LSB change of analog input to move from one
digital transition to the next. Differential linearity error is defined as the maximum
amount of deviation of any quantum (or LSB change) in the entire transfer function from
its ideal size of 1 LSB.
Where the change in analog signal corresponding to 1 LSB digital change is more or less
than 1 LSB, there is said to be a DNL error. The DNL error of a converter is normally
defined as the maximum value of DNL to be found at any transition across the range of
the converter. Figure 2.19 shows the non-ideal transfer functions for a DAC and an ADC
and shows the effects of the DNL error.
DAC
FS
ADC
111
110
ANALOG
OUTPUT
DIGITAL
OUTPUT
NON-MONOTONIC
101
100
MISSING CODE
011
010
001
000
000
001
010
011
100
101
110
111
ANALOG INPUT
FS
DIGITAL INPUT
Figure 2.19: Transfer Functions for Non-Ideal 3-Bit DAC and ADC
The DNL of a DAC is examined more closely in Figure 2.20. If the DNL of a DAC is
less than –1 LSB at any transition, the DAC is non-monotonic i.e., its transfer
characteristic contains one or more localized maxima or minima. A DNL greater than +1
LSB does not cause non-monotonicity, but is still undesirable. In many DAC applications
(especially closed-loop systems where non-monotonicity can change negative feedback
to positive feedback), it is critically important that DACs are monotonic. DAC
monotonicity is often explicitly specified on data sheets, although if the DNL is
guaranteed to be less than 1 LSB (i.e., |DNL| ≤ 1 LSB) then the device must be
monotonic, even without an explicit guarantee.
2.17
ANALOG-DIGITAL CONVERSION
FS
d
1 LSB,
DNL = 0
BIT 2 IS 1 LSB HIGH
BIT 1 IS 1 LSB LOW
2 LSB,
DNL = +1 LSB
ANALOG
OUTPUT
1 LSB,
DNL = 0
–1 LSB,
DNL = –2 LSB
1 LSB,
DNL = 0
2 LSB,
DNL = +1 LSB
NON-MONOTONIC IF
DNL < –1 LSB
1 LSB,
DNL = 0
d
000
001
010 011
100 101 110
DIGITAL CODE INPUT
111
Figure 2.20: Details of DAC Differential Nonlinearity
In Figure 2.21, the DNL of an ADC is examined more closely on an expanded scale.
ADCs can be non-monotonic, but a more common result of excess DNL in ADCs is
missing codes. Missing codes in an ADC are as objectionable as non-monotonicity in a
DAC. Again, they result from DNL < –1 LSB.
MISSING CODE (DNL < –1 LSB)
DIGITAL
OUTPUT
CODE
1 LSB,
DNL = 0
0.25 LSB,
DNL = –0.75 LSB
1.5 LSB,
DNL = +0.5 LSB
0.5 LSB,
DNL = –0.5 LSB
1 LSB,
DNL = 0
ANALOG INPUT
Figure 2.21: Details of ADC Differential Nonlinearity
2.18
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
Not only can ADCs have missing codes, they can also be non-monotonic as shown in
Figure 2.22. As in the case of DACs, this can present major problems—especially in
servo applications.
MISSING CODE
DIGITAL
OUTPUT
CODE
NON-MONOTONIC
ANALOG INPUT
Figure 2.22: Non-Monotonic ADC with Missing Code
In a DAC, there can be no missing codes—each digital input word will produce a
corresponding analog output. However, DACs can be non-monotonic as previously
discussed. In a straight binary DAC, the most likely place a non-monotonic condition can
develop is at mid-scale between the two codes: 011…11 and 100…00. If a nonmonotonic conditions occurs here, it is generally because the DAC is not properly
calibrated or trimmed. A successive approximation ADC with an internal non-monotonic
DAC will generally produce missing codes but remain monotonic. However it is possible
for an ADC to be non-monotonic—again depending on the particular conversion
architecture. Figure 2.22 shows the transfer function of an ADC which is non-monotonic
and has a missing code.
ADCs which use the subranging architecture divide the input range into a number of
coarse segments, and each coarse segment is further divided into smaller segments—and
ultimately the final code is derived. This process is described in more detail in Chapter 4
of this book. An improperly trimmed subranging ADC may exhibit non-monotonicity,
wide codes, or missing codes at the subranging points as shown in Figure 2.23 A, B, and
C, respectively. This type of ADC should be trimmed so that drift due to aging or
temperature produces wide codes at the sensitive points rather than non-monotonic or
missing codes.
2.19
ANALOG-DIGITAL CONVERSION
(A) NON-MONOTONIC
(B) WIDE CODES
(C) MISSING CODES
NON-MONOTONIC
MISSING
CODE
WIDE
CODE
NON-MONOTONIC
MISSING
CODE
WIDE
CODE
ANALOG INPUT
Figure 2.23: Errors Associated with Improperly Trimmed Subranging ADC
Defining missing codes is more difficult than defining non-monotonicity. All ADCs
suffer from some inherent transition noise as shown in Figure 2.24 (think of it as the
flicker between adjacent values of the last digit of a DVM). As resolutions and
bandwidths become higher, the range of input over which transition noise occurs may
approach, or even exceed, 1 LSB. High resolution wideband ADCs generally have
internal noise sources which can be reflected to the input as effective input noise summed
with the signal. The effect of this noise, especially if combined with a negative DNL
error, may be that there are some (or even all) codes where transition noise is present for
the whole range of inputs. There are therefore some codes for which there is no input
which will guarantee that code as an output, although there may be a range of inputs
which will sometimes produce that code.
CODE TRANSITION NOISE
DNL
TRANSITION NOISE
AND DNL
ADC
OUTPUT
CODE
ADC INPUT
ADC INPUT
ADC INPUT
Figure 2.24: Combined Effects of Code Transition Noise and DNL
2.20
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.1 CODING AND QUANTIZING
For low resolution ADCs, it may be reasonable to define no missing codes as a
combination of transition noise and DNL which guarantees some level (perhaps 0.2 LSB)
of noise-free code for all codes. However, this is impossible to achieve at the very high
resolutions achieved by modern sigma-delta ADCs, or even at lower resolutions in wide
bandwidth sampling ADCs. In these cases, the manufacturer must define noise levels and
resolution in some other way. Which method is used is less important, but the data sheet
should contain a clear definition of the method used and the performance to be expected.
A complete discussion of effective input noise follows in Section 2.3 of this chapter.
The discussion thus far has only dealt with the most important dc specifications
associated with data converters. Other less important specifications require only a
definition. For specifications not covered in this section, the reader is referred to Section
2.5 of this chapter for a complete alphabetical listing of data converter specifications
along with their definitions.
2.21
ANALOG-DIGITAL CONVERSION
REFERENCES:
2.1 CODING AND QUANTIZATION
1.
K. W. Cattermole, Principles of Pulse Code Modulation, American Elsevier Publishing Company,
Inc., 1969, New York NY, ISBN 444-19747-8. (An excellent tutorial and historical discussion of data
conversion theory and practice, oriented towards PCM, but covers practically all aspects. This one is
a must for anyone serious about data conversion!
2.
Frank Gray, "Pulse Code Communication," U.S. Patent 2,632,058, filed November 13, 1947, issued
March 17, 1953. (detailed patent on the Gray code and its application to electron beam coders).
3.
R. W. Sears, "Electron Beam Deflection Tube for Pulse Code Modulation," Bell System Technical
Journal, Vol. 27, pp. 44-57, Jan. 1948. (describes an electon-beam deflection tube 7-bit,100kSPS flash
converter for early experimental PCM work).
4.
J. O. Edson and H. H. Henning, "Broadband Codecs for an Experimental 224Mb/s PCM Terminal,"
Bell System Technical Journal, Vol. 44, pp. 1887-1940, Nov. 1965. (summarizes experiments on
ADCs based on the electron tube coder as well as a bit-per-stage Gray code 9-bit solid state ADC. The
electron beam coder was 9-bits at 12MSPS, and represented the fastest of its type).
5.
Dan Sheingold, Analog-Digital Conversion Handbook, 3rd Edition, Analog Devices and PrenticeHall, 1986, ISBN-0-13-032848-0. (the defining and classic book on data conversion).
2.22
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.2 SAMPLING THEORY
SECTION 2.2: SAMPLING THEORY
Walt Kester
This section discusses the basics of sampling theory. A block diagram of a typical realtime sampled data system is shown in Figure 2.25. Prior to the actual analog-to-digital
conversion, the analog signal usually passes through some sort of signal conditioning
circuitry which performs such functions as amplification, attenuation, and filtering. The
lowpass/bandpass filter is required to remove unwanted signals outside the bandwidth of
interest and prevent aliasing.
fs
fa
LPF
OR
BPF
N-BIT
ADC
AMPLITUDE
QUANTIZATION
fs
DSP
LPF
OR
BPF
N-BIT
DAC
DISCRETE
TIME SAMPLING
fa
ts=
1
fs
t
Figure 2.25: Sampled Data System
The system shown in Figure 2.25 is a real-time system, i.e., the signal to the ADC is
continuously sampled at a rate equal to fs, and the ADC presents a new sample to the
DSP at this rate. In order to maintain real-time operation, the DSP must perform all its
required computation within the sampling interval, 1/fs, and present an output sample to
the DAC before arrival of the next sample from the ADC. An example of a typical DSP
function would be a digital filter.
In the case of FFT analysis, a block of data is first transferred to the DSP memory. The
FFT is calculated at the same time a new block of data is transferred into the memory, in
order to maintain real-time operation. The DSP must calculate the FFT during the data
transfer interval so it will be ready to process the next block of data.
Note that the DAC is required only if the DSP data must be converted back into an
analog signal (as would be the case in a voiceband or audio application, for example).
There are many applications where the signal remains entirely in digital format after the
initial A/D conversion. Similarly, there are applications where the DSP is solely
responsible for generating the signal to the DAC. If a DAC is used, it must be followed
by an analog anti-imaging filter to remove the image frequencies. Finally, there are
2.23
ANALOG-DIGITAL CONVERSION
slower speed industrial process control systems where sampling rates are much lower—
regardless of the system, the fundamentals of sampling theory still apply.
There are two key concepts involved in the actual analog-to-digital and digital-to-analog
conversion process: discrete time sampling and finite amplitude resolution due to
quantization. An understanding of these concepts is vital to data converter applications.
The Need for a Sample-and-Hold Amplifier (SHA) Function
The generalized block diagram of a sampled data system shown in Figure 2.25 assumes
some type of ac signal at the input. It should be noted that this does not necessarily have
to be so, as in the case of modern digital voltmeters (DVMs) or ADCs optimized for dc
measurements, but for this discussion assume that the input signal has some upper
frequency limit fa.
Most ADCs today have a built-in sample-and-hold function, thereby allowing them to
process ac signals. This type of ADC is referred to as a sampling ADC. However many
early ADCs, such as Analog Devices' industry-standard AD574, were not of the sampling
type, but simply encoders as shown in Figure 2.26. If the input signal to a SAR ADC
(assuming no SHA function) changes by more than 1 LSB during the conversion time
(8µs in the example), the output data can have large errors, depending on the location of
the code. Most ADC architectures are subject to this type of error—some more, some
less—with the possible exception of flash converters having well-matched comparators.
ANALOG INPUT
2N
v(t) = q
2
sin (2π f t )
N-BIT
SAR ADC ENCODER
CONVERSION TIME = 8µs
2N
dv
q
2π f cos (2π f t )
=
2
dt
dv
q 2(N–1) 2π f
dt max =
fmax =
fmax =
dv
dt max
2(N–1) 2π q
dv
dt max
N
fs = 100 kSPS
EXAMPLE:
dv = 1 LSB = q
dt = 8µs
N = 12, 2N = 4096
fmax = 9.7 Hz
qπ 2N
Figure 2.26: Input Frequency Limitations of Non-Sampling ADC (Encoder)
Assume that the input signal to the encoder is a sinewave with a full-scale amplitude
(q2N/2), where q is the weight of 1 LSB.
v(t) = q (2N/2) sin (2π f t).
2.24
Eq. 2.1
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.2 SAMPLING THEORY
Taking the derivative:
dv/dt = q 2πf (2N/2) cos (2π f t).
Eq. 2.2
The maximum rate of change is therefore:
dv/dt |max = q 2πf (2N/2).
Eq. 2.3
f = (dv/dt |max )/(q π 2N).
Eq. 2.4
Solving for f:
If N = 12, and 1 LSB change (dv = q) is allowed during the conversion time (dt = 8µs),
then the equation can be solved for fmax, the maximum full-scale signal frequency that can
be processed without error:
fmax = 9.7 Hz.
This implies any input frequency greater than 9.7 Hz is subject to conversion errors, even
though a sampling frequency of 100 kSPS is possible with the 8-µs ADC (this allows an
extra 2µs interval for an external SHA to re-acquire the signal after coming out of the
hold mode).
To process ac signals, a sample-and-hold function is added as shown in Figure 2.27. The
ideal SHA is simply a switch driving a hold capacitor followed by a high input
impedance buffer. The input impedance of the buffer must be high enough so that the
capacitor is discharged by less than 1 LSB during the hold time. The SHA samples the
signal in the sample mode, and holds the signal constant during the hold mode. The
timing is adjusted so that the encoder performs the conversion during the hold time. A
sampling ADC can therefore process fast signals—the upper frequency limitation is
determined by the SHA aperture jitter, bandwidth, distortion, etc., not the encoder. In the
example shown, a good sample-and-hold could acquire the signal in 2 µs, allowing a
sampling frequency of 100 kSPS, and the capability of processing input frequencies up to
50 kHz. A complete discussion of the SHA function including these specifications
follows later in this chapter.
It is important to understand a subtle difference between a true sample-and-hold amplifier
(SHA) and a track-and-hold amplifier (T/H, or THA). Strictly speaking, the output of a
sample-and-hold is not defined during the sample mode, however the output of a trackand-hold tracks the signal during the sample or track mode. In practice, the function is
generally implemented as a track-and-hold, and the terms track-and-hold and sampleand-hold are often used interchangeably. The waveforms shown in Figure 2.27 are those
associated with a track-and-hold.
In order to better understand the types of ac errors an ADC can make without a sampleand-hold function, consider Figure 2.28. The photos show the reconstructed output of an
8-bit ADC (flash converter) with and without the sample-and-hold function. In an ideal
flash converter the comparators are perfectly matched, and no sample-and-hold is
required. In practice, however, there are timing mismatches between the comparators
2.25
ANALOG-DIGITAL CONVERSION
which cause high-frequency inputs to exhibit nonlinearities and missing codes as shown
in the right-hand photos. The data was taken by driving a DAC with the ADC output. The
DAC output is a low frequency aliased sinewave corresponding to the difference between
the sampling frequency (20 MSPS) and the ADC input frequency (19.98 MHz). In this
case, the alias frequency is 20 kHz. (Aliasing is explained in detail in the next section).
SAMPLING
CLOCK
TIMING
ANALOG
INPUT
ADC
ENCODER
SW
CONTROL
N
C
ENCODER CONVERTS
DURING HOLD TIME
HOLD
SW
CONTROL
SAMPLE
SAMPLE
Figure 2.27: Sample-and-Hold Function Required for Digitizing AC Signals
WITH SHA
WITHOUT SHA
fs = 20 MSPS, fa = 19.98 MHz, fs – fa = 20kHz
Figure 2.28: 8-bit, 20-MSPS Flash ADC With and Without Sample-and-Hold
2.26
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.2 SAMPLING THEORY
The Nyquist Criteria
A continuous analog signal is sampled at discrete intervals, ts = 1/fs,which must be
carefully chosen to ensure an accurate representation of the original analog signal. It is
clear that the more samples taken (faster sampling rates), the more accurate the digital
representation, but if fewer samples are taken (lower sampling rates), a point is reached
where critical information about the signal is actually lost. The mathematical basis of
sampling was set forth by Harry Nyquist of Bell Telephone Laboratories in two classic
papers published in 1924 and 1928, respectively. (See References 1 and 2 as well as
Chapter 1 of this book). Nyquist's original work was shortly supplemented by R. V. L.
Hartley (Reference 3). These papers formed the basis for the PCM work to follow in the
1940s, and in 1948 Claude Shannon wrote his classic paper on communication theory
(Reference 4).
Simply stated, the Nyquist criteria requires that the sampling frequency be at least twice
the highest frequency contained in the signal, or information about the signal will be lost.
If the sampling frequency is less than twice the maximum analog signal frequency, a
phenomena known as aliasing will occur.
A signal with a maximum frequency fa must be sampled at a rate fs > 2fa
or information about the signal will be lost because of aliasing.
Aliasing occurs whenever fs < 2fa
The concept of aliasing is widely used in communications applications
such as direct IF-to-digital conversion.
A signal which has frequency components between fa and fb
must be sampled at a rate fs > 2 (fb – fa) in order to prevent alias
components from overlapping the signal frequencies.
Figure 2.29: Nyquist's Criteria
In order to understand the implications of aliasing in both the time and frequency
domain, first consider the case of a time domain representation of a single tone sinewave
sampled as shown in Figure 2.30. In this example, the sampling frequency fs is not at
least 2fa, but only slightly more than the analog input frequency fa—the Nyquist criteria is
violated. Notice that the pattern of the actual samples produces an aliased sinewave at a
lower frequency equal to fs – fa.
The corresponding frequency domain representation of this scenario is shown in Figure
2.31B. Now consider the case of a single frequency sinewave of frequency fa sampled at
a frequency fs by an ideal impulse sampler (see Figure 2.31A). Also assume that fs > 2fa
as shown. The frequency-domain output of the sampler shows aliases or images of the
original signal around every multiple of fs, i.e. at frequencies equal to |± Kfs ± fa|, K = 1,
2, 3, 4, .....
2.27
ANALOG-DIGITAL CONVERSION
ALIASED SIGNAL = fs – fa
INPUT = fa
1
fs
t
NOTE: fa IS SLIGHTLY LESS THAN fs
Figure 2.30: Aliasing in the Time Domain
A
fa
fs
0.5fs
1st NYQUIST
ZONE
B
I
I
2nd NYQUIST
ZONE
0.5fs
1.5fs
3rd NYQUIST
ZONE
fa
I
2fs
4th NYQUIST
ZONE
I
fs
I
I
I
I
1.5fs
2fs
Figure 2.31: Analog Signal fa Sampled @ fs Using Ideal Sampler
Has Images (Aliases) at |± Kfs ± fa|, K = 1, 2, 3, . . .
The Nyquist bandwidth is defined to be the frequency spectrum from dc to fs/2. The
frequency spectrum is divided into an infinite number of Nyquist zones, each having a
width equal to 0.5fs as shown. In practice, the ideal sampler is replaced by an ADC
followed by an FFT processor. The FFT processor only provides an output from dc to
fs/2, i.e., the signals or aliases which appear in the first Nyquist zone.
Now consider the case of a signal which is outside the first Nyquist zone (Figure 2.31B).
The signal frequency is only slightly less than the sampling frequency, corresponding to
the condition shown in the time domain representation in Figure 2.30. Notice that even
though the signal is outside the first Nyquist zone, its image (or alias), fs – fa, falls inside.
Returning to Figure 2.31A, it is clear that if an unwanted signal appears at any of the
2.28
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.2 SAMPLING THEORY
image frequencies of fa, it will also occur at fa, thereby producing a spurious frequency
component in the first Nyquist zone.
This is similar to the analog mixing process and implies that some filtering ahead of the
sampler (or ADC) is required to remove frequency components which are outside the
Nyquist bandwidth, but whose aliased components fall inside it. The filter performance
will depend on how close the out-of-band signal is to fs/2 and the amount of attenuation
required.
Baseband Antialiasing Filters
Baseband sampling implies that the signal to be sampled lies in the first Nyquist zone. It
is important to note that with no input filtering at the input of the ideal sampler, any
frequency component (either signal or noise) that falls outside the Nyquist bandwidth in
any Nyquist zone will be aliased back into the first Nyquist zone. For this reason, an
antialiasing filter is used in almost all sampling ADC applications to remove these
unwanted signals.
Properly specifying the antialiasing filter is important. The first step is to know the
characteristics of the signal being sampled. Assume that the highest frequency of interest
is fa. The antialiasing filter passes signals from dc to fa while attenuating signals above fa.
Assume that the corner frequency of the filter is chosen to be equal to fa. The effect of the
finite transition from minimum to maximum attenuation on system dynamic range is
illustrated in Figure 2.32A.
fa
B
A
fs - fa
fa
Kfs - fa
DR
fs
fs
2
STOPBAND ATTENUATION = DR
TRANSITION BAND: fa to fs - fa
Kfs
2
STOPBAND ATTENUATION = DR
TRANSITION BAND: fa to Kfs - fa
CORNER FREQUENCY: fa
CORNER FREQUENCY: fa
Kfs
Figure 2.32: Oversampling Relaxes Requirements
on Baseband Antialiasing Filter
2.29
ANALOG-DIGITAL CONVERSION
Assume that the input signal has full-scale components well above the maximum
frequency of interest, fa. The diagram shows how full-scale frequency components above
fs – fa are aliased back into the bandwidth dc to fa. These aliased components are
indistinguishable from actual signals and therefore limit the dynamic range to the value
on the diagram which is shown as DR.
Some texts recommend specifying the antialiasing filter with respect to the Nyquist
frequency, fs/2, but this assumes that the signal bandwidth of interest extends from dc to
fs/2 which is rarely the case. In the example shown in Figure 2.32A, the aliased
components between fa and fs/2 are not of interest and do not limit the dynamic range.
The antialiasing filter transition band is therefore determined by the corner frequency fa,
the stopband frequency fs – fa, and the desired stopband attenuation, DR. The required
system dynamic range is chosen based on the requirement for signal fidelity.
Filters become more complex as the transition band becomes sharper, all other things
being equal. For instance, a Butterworth filter gives 6-dB attenuation per octave for each
filter pole (as do all filters). Achieving 60 dB attenuation in a transition region between 1
MHz and 2 MHz (1 octave) requires a minimum of 10 poles—not a trivial filter, and
definitely a design challenge.
Therefore, other filter types are generally more suited to applications where the
requirement is for a sharp transition band and in-band flatness coupled with linear phase
response. Elliptic filters meet these criteria and are a popular choice. There are a number
of companies which specialize in supplying custom analog filters. TTE is an example of
such a company (Reference 5).
From this discussion, we can see how the sharpness of the antialiasing transition band can
be traded off against the ADC sampling frequency. Choosing a higher sampling rate
(oversampling) reduces the requirement on transition band sharpness (hence, the filter
complexity) at the expense of using a faster ADC and processing data at a faster rate.
This is illustrated in Figure 2.32B which shows the effects of increasing the sampling
frequency by a factor of K, while maintaining the same analog corner frequency, fa, and
the same dynamic range, DR, requirement. The wider transition band (fa to Kfs – fa)
makes this filter easier to design than for the case of Figure 2.32A.
The antialiasing filter design process is started by choosing an initial sampling rate of 2.5
to 4 times fa. Determine the filter specifications based on the required dynamic range and
see if such a filter is realizable within the constraints of the system cost and performance.
If not, consider a higher sampling rate which may require using a faster ADC. It should
be mentioned that sigma-delta ADCs are inherently highly oversampled converters, and
the resulting relaxation in the analog anti-aliasing filter requirements is therefore an
added benefit of this architecture.
The antialiasing filter requirements can also be relaxed somewhat if it is certain that there
will never be a full-scale signal at the stopband frequency fs – fa. In many applications, it
is improbable that full-scale signals will occur at this frequency. If the maximum signal at
the frequency fs – fa will never exceed X dB below full-scale, then the filter stopband
attenuation requirement can be reduced by that same amount. The new requirement for
2.30
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.2 SAMPLING THEORY
stopband attenuation at fs – fa based on this knowledge of the signal is now only DR – X
dB. When making this type of assumption, be careful to treat any noise signals which
may occur above the maximum signal frequency fa as unwanted signals which will also
alias back into the signal bandwidth.
There are a number of companies which specialize in supplying custom analog filters.
TTE is an example of such a company (Reference 5). As an example, the normalized
response of the TTE, Inc., LE1182 11-pole elliptic antialiasing filter is shown in Figure
2.33. Notice that this filter is specified to achieve at least 80 dB attenuation between fc
and 1.2fc. The corresponding passband ripple, return loss, delay, and phase response are
also shown in Figure 2.33. This custom filter is available in corner frequencies up to
100 MHz and in a choice of PC board, BNC, or SMA with compatible packages.
Reprinted with Permission of TTE, Inc.,
11652 Olympic Blvd., Los Angeles CA 90064
http://www.tte.com
Figure 2.33: Characteristics of 11-Pole Elliptical Filter (TTE, Inc., LE1182-Series)
Undersampling (Harmonic Sampling, Bandpass Sampling, IF Sampling,
Direct IF-to-Digital Conversion)
Thus far we have considered the case of baseband sampling, i.e., all the signals of interest
lie within the first Nyquist zone. Figure 2.34A shows such a case, where the band of
sampled signals is limited to the first Nyquist zone, and images of the original band of
frequencies appear in each of the other Nyquist zones.
Consider the case shown in Figure 2.34B, where the sampled signal band lies entirely
within the second Nyquist zone. The process of sampling a signal outside the first
Nyquist zone is often referred to as undersampling, or harmonic sampling. Note that the
image which falls in the first Nyquist zone contains all the information in the original
signal, with the exception of its original location (the order of the frequency components
within the spectrum is reversed, but this is easily corrected by re-ordering the output of
the FFT).
2.31
ANALOG-DIGITAL CONVERSION
A
ZONE 1
I
I
0.5fs
fs
I
I
1.5fs
I
I
2.5fs
2fs
3.5fs
3fs
ZONE 2
B
I
I
fs
0.5fs
I
1.5fs
I
I
3fs
2.5fs
2fs
I
3.5fs
ZONE 3
C
I
I
0.5fs
I
fs
1.5fs
I
2fs
I
2.5fs
I
3fs
3.5fs
Figure 2.34: Undersampling and Frequency Translation Between Nyquist Zones
Figure 2.34C shows the sampled signal restricted to the third Nyquist zone. Note that the
image that falls into the first Nyquist zone has no frequency reversal. In fact, the sampled
signal frequencies may lie in any unique Nyquist zone, and the image falling into the first
Nyquist zone is still an accurate representation (with the exception of the frequency
reversal which occurs when the signals are located in even Nyquist zones). At this point
we can clearly restate the Nyquist criteria:
A signal must be sampled at a rate equal to or greater than twice its bandwidth in order
to preserve all the signal information.
Notice that there is no mention of the absolute location of the band of sampled signals
within the frequency spectrum relative to the sampling frequency. The only constraint is
that the band of sampled signals be restricted to a single Nyquist zone, i.e., the signals
must not overlap any multiple of fs/2 (this, in fact, is the primary function of the
antialiasing filter).
Sampling signals above the first Nyquist zone has become popular in communications
because the process is equivalent to analog demodulation. It is becoming common
practice to sample IF signals directly and then use digital techniques to process the signal,
thereby eliminating the need for an IF demodulator and filters. Clearly, however, as the
IF frequencies become higher, the dynamic performance requirements on the ADC
become more critical. The ADC input bandwidth and distortion performance must be
adequate at the IF frequency, rather than only baseband. This presents a problem for most
ADCs designed to process signals in the first Nyquist zone, therefore an ADC suitable for
undersampling applications must maintain dynamic performance into the higher order
Nyquist zones.
2.32
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.2 SAMPLING THEORY
Antialiasing Filters in Undersampling Applications
Figure 2.35 shows a signal in the second Nyquist zone centered around a carrier
frequency, fc, whose lower and upper frequencies are f1 and f2. The antialiasing filter is a
bandpass filter. The desired dynamic range is DR, which defines the filter stopband
attenuation. The upper transition band is f2 to 2fs – f2, and the lower is f1 to fs – f1. As in
the case of baseband sampling, the antialiasing filter requirements can be relaxed by
proportionally increasing the sampling frequency, but fc must also be increased so that it
is always centered in the second Nyquist zone.
fs - f1
f1
f2
2fs - f 2
fc
DR
SIGNALS
OF
INTEREST
IMAGE
0
IMAGE
IMAGE
0.5fS
fS
BANDPASS FILTER SPECIFICATIONS:
1.5fS
2fS
STOPBAND ATTENUATION = DR
TRANSITION BAND: f2 TO 2fs - f2
f1 TO f s - f 1
CORNER FREQUENCIES: f1, f2
Figure 2.35: Antialiasing Filter for Undersampling
Two key equations can be used to select the sampling frequency, fs, given the carrier
frequency, fc, and the bandwidth of its signal, ∆f. The first is the Nyquist criteria:
fs > 2∆f .
Eq. 2.5
The second equation ensures that fc is placed in the center of a Nyquist zone:
fs =
4f c
,
2 NZ − 1
Eq. 2.6
where NZ = 1, 2, 3, 4, .... and NZ corresponds to the Nyquist zone in which the carrier
and its signal fall (see Figure 2.36).
NZ is normally chosen to be as large as possible while still maintaining fs > 2∆f. This
results in the minimum required sampling rate. If NZ is chosen to be odd, then fc and its
signal will fall in an odd Nyquist zone, and the image frequencies in the first Nyquist
zone will not be reversed. Tradeoffs can be made between the sampling frequency and
2.33
ANALOG-DIGITAL CONVERSION
the complexity of the antialiasing filter by choosing smaller values of NZ (hence a higher
sampling frequency).
ZONE NZ - 1
ZONE NZ
ZONE NZ + 1
I
∆f
I
fc
0.5fs
f s > 2∆ f
0.5fs
0.5fs
fs =
4fc
, NZ = 1, 2, 3, . . .
2NZ - 1
Figure 2.36: Centering an Undersampled Signal within a Nyquist Zone
As an example, consider a 4-MHz wide signal centered around a carrier frequency of
71 MHz. The minimum required sampling frequency is therefore 8 MSPS. Solving Eq.
2.6 for NZ using fc = 71 MHz and fs = 8 MSPS yields NZ = 18.25. However, NZ must be
an integer, so we round 18.25 to the next lowest integer, 18. Solving Eq. 2.6 again for fs
yields fs = 8.1143 MSPS. The final values are therefore fs = 8.1143 MSPS, fc = 71 MHz,
and NZ = 18.
Now assume that we desire more margin for the antialiasing filter, and we select fs to be
10 MSPS. Solving Eq. 2.6 for NZ, using fc = 71 MHz and fs = 10 MSPS yields NZ =
14.7. We round 14.7 to the next lowest integer, giving NZ = 14. Solving Eq. 2.6 again for
fs yields fs = 10.519 MSPS. The final values are therefore fs = 10.519 MSPS,
fc = 71 MHz, and NZ = 14.
The above iterative process can also be carried out starting with fs and adjusting the
carrier frequency to yield an integer number for NZ.
2.34
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.2 SAMPLING THEORY
REFERENCES:
2.2 SAMPLING THEORY
1.
H. Nyquist, "Certain Factors Affecting Telegraph Speed," Bell System Technical Journal, Vol. 3,
April 1924, pp. 324-346.
2.
H.. Nyquist, Certain Topics in Telegraph Transmission Theory, A.I.E.E. Transactions, Vol. 47, April
1928, pp. 617-644.
3.
R.V.L. Hartley, "Transmission of Information," Bell System Technical Journal, Vol. 7, July 1928,
pp. 535-563.
4.
C. E. Shannon, "A Mathematical Theory of Communication," Bell System Technical Journal, Vol.
27, July 1948, pp. 379-423 and October 1948, pp. 623-656.
5.
TTE, Inc., 11652 Olympic Blvd., Los Angeles, CA 90064, http://www.tte.com.
2.35
ANALOG-DIGITAL CONVERSION
NOTES:
2.36
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
SECTION 2.3: DATA CONVERTER AC ERRORS
Walt Kester, James Bryant
This section examines the ac errors associated with data converters. Many of the errors
and specifications apply equally to ADCs and DACs, while some are more specific to
one or the other. All possible specifications are not discussed here, only the most
common ones. Section 2.4 of this Chapter contains a comprehensive listing of converter
specifications as well as their definitions, including some not discussed in this section.
Theoretical Quantization Noise of an Ideal N-Bit Converter
The only errors (dc or ac) associated with an ideal N-bit data converter are those related
to the sampling and quantization processes. The maximum error an ideal converter makes
when digitizing a signal is ±½ LSB. The transfer function of an ideal N-bit ADC is
shown in Figure 2.37. The quantization error for any ac signal which spans more than a
few LSBs can be approximated by an uncorrelated sawtooth waveform having a peak-topeak amplitude of q, the weight of an LSB. Although this analysis is not precise, it is
accurate enough for most applications. W. R. Bennett of Bell Laboratories analyzed the
actual spectrum of quantization noise in his classic 1948 paper (Reference 1). With
certain simplifying assumptions, his detailed mathematical analysis simplifies to that of
Figure 2.37. Other significant papers on converter noise (References 2-5) followed
Bennett's classic publication.
DIGITAL
OUTPUT
ANALOG
INPUT
ERROR
(INPUT – OUTPUT)
q = 1 LSB
Figure 2.37: Ideal N-bit ADC Quantization Noise
2.37
ANALOG-DIGITAL CONVERSION
The quantization error as a function of time is shown in Figure 2.38. Again, a simple
sawtooth waveform provides a sufficiently accurate model for analysis. The equation of
the sawtooth error is given by
e(t) = st, –q/2s < t < +q/2s.
Eq. 2.7
The mean-square value of e(t) can be written:
e2 (t) =
s + q / 2s 2
(st ) dt .
q ∫− q / 2s
Eq. 2.8
Performing the simple integration and simplifying,
e2 (t) =
q2
.
12
Eq. 2.9
The root-mean-square quantization error is therefore
rms quantization noise = e 2 ( t ) =
q
.
12
Eq. 2.10
e(t)
+q
2
SLOPE = s
t
–q
2
+q
2s
–q
2s
ERROR = e(t) = st,
–q
+q
<t<
2s
2s
s
MEAN-SQUARE ERROR = e2(t) =
q
ROOT-MEAN-SQUARE ERROR =
∫
+q/2s
(st) 2 dt
=
–q/2s
e2(t) =
q2
12
q
√ 12
Figure 2.38: Quantization Noise as a Function of Time
As Bennett points out (Reference 1), this noise is approximately Gaussian and spread
more or less uniformly over the Nyquist bandwidth dc to fs/2. The underlying assumption
here is that the quantization noise is uncorrelated to the input signal. Under certain
conditions where the sampling clock and the signal are harmonically related, the
quantization noise becomes correlated and the energy is concentrated at the harmonics of
the signal—the rms value remains approximately q/√12.
2.38
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
The theoretical signal-to-noise ratio can now be calculated assuming a full-scale input
sinewave:
Input FS Sinewave = v ( t ) =
q2 N
sin(2πft ).
2
Eq. 2.11
The rms value of the input signal is therefore
q2 N
.
rms value of FS input =
2 2
Eq. 2.12
The rms signal-to-noise ratio for an ideal N-bit converter is therefore
SNR = 20 log10
rms value of FS input
rms value of quantization noise
 q2 N / 2 2 
SNR = 20 log10 
 = 6.02 N + 1.76 dB , over dc to fs/2 bandwidth.
 q / 12 
Eq. 2.13
Eq. 2.14
These relationships are summarized in Figure 2.39.
FS INPUT =
v(t) =
q 2N
sin (2π f t )
2
RMS Value of FS Sinewave =
q 2N
2√2
RMS Value of Quantization Noise =
SNR = 20 log10
q
√ 12
RMS Value of FS Sinewave
RMS Value of Quantization Noise
= 20 log102N + 20 log10
3
2
SNR = 6.02N + 1.76dB
(Measured over the Nyquist Bandwidth : DC to fs/2)
Figure 2.39: Theoretical Signal-to-Quantization Noise Ratio
of an Ideal N-Bit Converter
Bennett's paper shows that although the actual spectrum of the quantization noise is quite
complex to analyze—the simplified analysis which leads to Eq. 2.14 is accurate enough
for most purposes. However, it is important to emphasize again that the rms quantization
noise is measured over the full Nyquist bandwidth, dc to fs/2. In many applications, the
actual signal of interest occupies a smaller bandwidth, BW. If digital filtering is used to
2.39
ANALOG-DIGITAL CONVERSION
filter out noise components outside the bandwidth BW, then a correction factor (called
process gain) must be included in the equation to account for the resulting increase in
SNR. The process of sampling a signal at a rate which is greater than twice its bandwidth
is often referred to as oversampling. In fact, oversampling in conjunction with
quantization noise shaping and digital filtering is a key concept in sigma-delta converters.
SNR = 6.02 N + 1.76 dB + 10 log10
fs
, over bandwidth BW.
2 ⋅ BW
Eq. 2.15
q
NOISE
SPECTRAL
DENSITY
RMS VALUE =
q = 1 LSB
12
MEASURED OVER DC TO
fs
2
q / 12
fs / 2
fs
BW
2
SNR = 6.02N + 1.76dB + 10log10
fs
2•BW
FOR FS SINEWAVE
Process Gain
Figure 2.40: Quantization Noise Spectrum
The significance of process gain can be seen from the following example. In many digital
basestations or other wideband receivers the signal bandwidth is composed of many
individual channels, and a single ADC is used to digitize the entire bandwidth. For
instance, the analog cellular radio system (AMPS) in the U.S. consists of 416 30-kHz
wide channels, occupying a bandwidth of approximately 12.5 MHz. Assume a 65-MSPS
sampling frequency, and that digital filtering is used to separate the individual 30-kHz
channels. The process gain due to oversampling is therefore given by:
65 × 106
fs
= 10 log10
= 30.3 dB .
Process Gain = 10 log10
2 ⋅ BW
2 × 30 × 103
Eq. 2.16
The process gain is added to the ADC SNR specification to yield the actual SNR in the
30-kHz bandwidth. In the above example, if the ADC SNR specification is 65 dB (dc to
fs/2), then it is increased to 95.3 dB in the 30-kHz channel bandwidth (after appropriate
digital filtering).
Figure 2.41 shows an application which combines oversampling and undersampling. The
signal of interest has a bandwidth BW and is centered around a carrier frequency fc. The
sampling frequency can be much less than fc and is chosen such that the signal of interest
2.40
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
is centered in its Nyquist zone. Analog and digital filtering removes the noise outside the
signal bandwidth of interest, and therefore results in process gain per Eq. 2.16.
ZONE 3
ZONE 1
BW
0.5fs
fs
fc
1.5fs
SNR = 6.02N + 1.76dB + 10log10
2fs
fs
2•BW
2.5fs
3fs
3.5fs
FOR FS SINEWAVE
Process Gain
Figure 2.41: Undersampling and Oversampling
Combined Results in Process Gain
Although the rms value of the noise is accurately approximated by q/√12, its frequency
domain content may be highly correlated to the ac-input signal. For instance, there is
greater correlation for low amplitude periodic signals than for large amplitude random
signals. Quite often, the assumption is made that the theoretical quantization noise
appears as white noise, spread uniformly over the Nyquist bandwidth dc to fs/2.
Unfortunately, this is not true in all cases. In the case of strong correlation, the
quantization noise appears concentrated at the various harmonics of the input signal, just
where you don't want them. Bennett (Reference 1) has an extensive analysis of the
frequency content contained in the quantization noise spectrum in his classic 1948 paper.
In most practical applications, the input to the ADC is a band of frequencies (always
summed with some unavoidable system noise), so the quantization noise tends to be
random. In spectral analysis applications (or in performing FFTs on ADCs using
spectrally pure sinewaves—see Figure 2.42), however, the correlation between the
quantization noise and the signal depends upon the ratio of the sampling frequency to the
input signal. This is demonstrated in Figure 2.43, where the output of an ideal 12-bit
ADC is analyzed using a 4096-point FFT. In the left-hand FFT plot, the ratio of the
sampling frequency to the input frequency was chosen to be exactly 32, and the worst
harmonic is about 76 dB below the fundamental. The right hand diagram shows the
effects of slightly offsetting the ratio to 4096/127 = 32.25196850394, showing a
relatively random noise spectrum, where the SFDR is now about 92 dBc. In both cases,
the rms value of all the noise components is approximately q/√12, but in the first case, the
noise is concentrated at harmonics of the fundamental.
2.41
ANALOG-DIGITAL CONVERSION
fs
ANALOG
INPUT
fa
IDEAL
N-BIT
ADC
N
BUFFER
MEMORY
M-WORDS
M POINT
M-POINT
2
FFT
PROCESSOR SPECTRAL
OUTPUT
Figure 2.42: Dynamic Performance Analysis of an Ideal N-bit ADC
fs / fa = 32
M = 4096
fs / fa = 4096 / 127
Figure 2.43: Effect of Ratio of Sampling Clock to Input Frequency
on SFDR for Ideal 12-bit ADC
Note that this variation in the apparent harmonic distortion of the ADC is an artifact of
the sampling process and the correlation of the quantization error with the input
frequency. In a practical ADC application, the quantization error generally appears as
random noise because of the random nature of the wideband input signal and the
additional fact that there is a usually a small amount of system noise which acts as a
dither signal to further randomize the quantization error spectrum.
It is important to understand the above point, because single-tone sinewave FFT testing
of ADCs is one of the universally accepted methods of performance evaluation. In order
to accurately measure the harmonic distortion of an ADC, steps must be taken to ensure
that the test setup truly measures the ADC distortion, not the artifacts due to quantization
noise correlation. This is done by properly choosing the frequency ratio and sometimes
by injecting a small amount of noise (dither) with the input signal. The exact same
precautions apply to measuring DAC distortion with an analog spectrum analyzer.
Figure 2.44 shows the FFT output for an ideal 12-bit ADC. Note that the average value of
the noise floor of the FFT is approximately 100 dB below full-scale, but the theoretical
SNR of a 12-bit ADC is 74 dB. The FFT noise floor is not the SNR of the ADC, because
the FFT acts like an analog spectrum analyzer with a bandwidth of fs/M, where M is the
number of points in the FFT. The theoretical FFT noise floor is therefore 10log10(M/2)
dB below the quantization noise floor due to the processing gain of the FFT. In the case
2.42
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
of an ideal 12-bit ADC with an SNR of 74 dB, a 4096-point FFT would result in a
processing gain of 10log10(4096/2) = 33 dB, thereby resulting in an overall FFT noise
floor of 74 + 33 = 107 dBc. In fact, the FFT noise floor can be reduced even further by
going to larger and larger FFTs; just as an analog spectrum analyzer's noise floor can be
reduced by narrowing the bandwidth. When testing ADCs using FFTs, it is important to
ensure that the FFT size is large enough so that the distortion products can be
distinguished from the FFT noise floor itself.
(dB)
0
ADC FULLSCALE
N = 12-BITS
M = 4096
20
40
74dB = 6.02N + 1.76dB
60
RMS QUANTIZATION NOISE LEVEL
80
74dB
M
33dB = 10log 10 2
( )
100
FFT NOISE FLOOR
120
BIN SPACING =
fs
4096
107dB
fs
2
Figure 2.44: Noise Floor for an Ideal 12-bit ADC Using 4096-point FFT
Noise in Practical ADCs
A practical sampling ADC (one that has an integral sample-and-hold), regardless of
architecture, has a number of noise and distortion sources as shown in Figure 2.45. The
wideband analog front-end buffer has wideband noise, nonlinearity, and also finite
bandwidth. The SHA introduces further nonlinearity, bandlimiting, and aperture jitter.
The actual quantizer portion of the ADC introduces quantization noise, and both integral
and differential nonlinearity. In this discussion, assume that sequential outputs of the
ADC are loaded into a buffer memory of length M and that the FFT processor provides
the spectral output. Also assume that the FFT arithmetic operations themselves introduce
no significant errors relative to the ADC. However, when examining the output noise
floor, the FFT processing gain (dependent on M) must be considered.
Equivalent Input Referred Noise
Wideband ADC internal circuits produce a certain amount of rms noise due to resistor
noise and "kT/C" noise. This noise is present even for dc-input signals, and accounts for
the fact that the output of most wideband (or high resolution) ADCs is a distribution of
codes, centered around the nominal value of a dc input (see Figure 2.46). To measure its
value, the input of the ADC is either grounded or connected to a heavily decoupled
voltage source, and a large number of output samples are collected and plotted as a
2.43
ANALOG-DIGITAL CONVERSION
histogram (sometimes referred to as a grounded-input histogram). Since the noise is
approximately Gaussian, the standard deviation of the histogram is easily calculated (see
Reference 6), corresponding to the effective input rms noise. It is common practice to
express this rms noise in terms of LSBs rms, although it can be expressed as an rms
voltage referenced to the ADC full-scale input range.
fs
ANALOG
INPUT
ADC
TO MEMORY
SAMPLE
AND
HOLD
BUFFER
NOISE
DISTORTION
BAND LIMITING
ENCODER
N-BITS
QUANTIZATION NOISE
DIFFERENTIAL NON-LINEARITY
INTEGRAL NON-LINEARITY
NOISE
DISTORTION
BAND LIMITING
APERTURE JITTER
M POINT
2
N
M-POINT
FFT
PROCESSOR
BUFFER
MEMORY
TEST
SYSTEM
M- WORDS
SPECTRAL
OUTPUT
PROCESSING GAIN = 10log10
( M2 )
ROUND OFF ERROR (NEGLIGIBLE)
Figure 2.45: ADC Model Showing Noise and Distortion Sources
NUMBER OF
OCCURRENCES
P-P INPUT NOISE
≈
6.6 × RMS NOISE
STANDARD DEVIATION
= RMS NOISE (LSBs)
n–4
n–3
n–2
n–1
n
n+1
n+2
n+3
n+4
OUTPUT CODE
Figure 2.46: Effect of Input-Referred Noise on ADC "Grounded Input" Histogram
2.44
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
Noise-Free (Flicker-Free) Code Resolution
The noise-free code resolution of an ADC is the number of bits beyond which it is
impossible to distinctly resolve individual codes. The cause is the effective input noise
(or input-referred noise) associated with all ADCs and described above. This noise can be
expressed as an rms quantity, usually having the units of LSBs rms. Multiplying by a
factor of 6.6 converts the rms noise into peak-to-peak noise (expressed in LSBs peak-topeak). The total range of an N-bit ADC is 2N LSBs. The noise-free (or flicker-free)
resolution can be calculated using the equation:
Noise-Free Code Resolution = log2 (2N/Peak-to-Peak Noise).
Eq. 2.17
The specification is generally associated with high-resolution sigma-delta measurement
ADCs, but is applicable to all ADCs.
The ratio of the FS range to the rms input noise is sometimes used to calculate resolution.
In this case, the term effective resolution is used. Note that under identical conditions,
effective resolution is larger than noise-free code resolution by log2(6.6), or
approximately 2.7 bits.
Effective Resolution = log2 (2N/RMS Input Noise)
Effective Resolution = Noise-Free Code Resolution + 2.7 bits.
Eq. 2.18
Eq. 2.19
The calculations are summarized in Figure 2.47.
Effective Input Noise
= en rms
Peak-to-Peak Input Noise = 6.6 en rms
Noise-Free Code Resolution = log2
= log2
"Effective Resolution" = log2
= log2
Peak-to-Peak Input Range
Peak-to-Peak Input Noise
2N
Peak-to-Peak Input Noise (LSBs)
Peak-to-Peak Input Range
RMS Input Noise
2N
RMS Input Noise (LSBs)
= Noise-Free Code Resolution + 2.7 bits
Figure 2.47: Calculating Noise-Free (Flicker-Free) Code Resolution
from Input-Referred Noise
2.45
ANALOG-DIGITAL CONVERSION
Dynamic Performance of Data Converters
There are various ways to characterize the ac performance of ADCs. Before the 1970s,
there was little standardization with respect to ac specifications, and measurement
equipment and techniques were not well understood or available. Over nearly a 30 year
period, manufacturers and customers have learned more about measuring the dynamic
performance of converters, and the specifications shown in Figure 2.48 represent the
most popular ones used today. Practically all the specifications represent the converter’s
performance in the frequency domain. The FFT is the heart of practically all these
measurements and is discussed in more detail in Chapter 6 of this book.
Integral and Differential Nonlinearity Distortion Effects
One of the first things to realize when examining the nonlinearities of data converters is
that the transfer function of a data converter has artifacts which do not occur in
conventional linear devices such as op amps or gain blocks. The overall integral
nonlinearity of an ADC is due to the integral nonlinearity of the front-end and SHA as
well as the overall integral nonlinearity in the ADC transfer function. However,
differential nonlinearity is due exclusively to the encoding process and may vary
considerably dependent on the ADC encoding architecture. Overall integral nonlinearity
produces distortion products whose amplitude varies as a function of the input signal
amplitude. For instance, second-order intermodulation products increase 2 dB for every
1-dB increase in signal level, and third-order products increase 3 dB for every 1-dB
increase in signal level.
Harmonic Distortion
Worst Harmonic
Total Harmonic Distortion (THD)
Total Harmonic Distortion Plus Noise (THD + N)
Signal-to-Noise-and-Distortion Ratio (SINAD, or S/N +D)
Effective Number of Bits (ENOB)
Signal-to-Noise Ratio (SNR)
Analog Bandwidth (Full-Power, Small-Signal)
Spurious Free Dynamic Range (SFDR)
Two-Tone Intermodulation Distortion
Multi-tone Intermodulation Distortion
Noise Power Ratio (NPR)
Adjacent Channel Leakage Ratio (ACLR)
Noise Figure
Settling Time, Overvoltage Recovery Time
Figure 2.48: Quantifying Data Converter Dynamic Performance
2.46
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
The differential nonlinearity in the ADC transfer function produces distortion products
which not only depend on the amplitude of the signal but the positioning of the
differential nonlinearity errors along the ADC transfer function. Figure 2.49 shows two
ADC transfer functions having differential nonlinearity. The left-hand diagram shows an
error which occurs at mid-scale. Therefore, for both large and small signals, the signal
crosses through this point producing a distortion product which is relatively independent
of the signal amplitude. The right-hand diagram shows another ADC transfer function
which has differential nonlinearity errors at 1/4 and 3/4 full-scale. Signals which are
above 1/2 scale peak-to-peak will exercise these codes and produce distortion, while
those less than 1/2 scale peak-to-peak will not.
OUT
OUT
IN
IN
MIDSCALE DNL
(A)
1/4FS, 3/4FS DNL
(B)
Figure 2.49: Typical ADC/ DAC DNL Errors (Exaggerated)
Most high-speed ADCs are designed so that differential nonlinearity is spread across the
entire ADC range. Therefore, for signals which are within a few dB of full-scale, the
overall integral nonlinearity of the transfer function determines the distortion products.
For lower level signals, however, the harmonic content becomes dominated by the
differential nonlinearities and does not generally decrease proportionally with decreases
in signal amplitude.
Harmonic Distortion, Worst Harmonic, Total Harmonic Distortion (THD), Total
Harmonic Distortion Plus Noise (THD + N)
There are a number of ways to quantify the distortion of an ADC. An FFT analysis can be
used to measure the amplitude of the various harmonics of a signal. The harmonics of the
input signal can be distinguished from other distortion products by their location in the
frequency spectrum. Figure 2.50 shows a 7-MHz input signal sampled at 20 MSPS and
the location of the first 9 harmonics. Aliased harmonics of fa fall at frequencies equal to
|±Kfs ± nfa|, where n is the order of the harmonic, and K = 0, 1, 2, 3,.... The second and
third harmonics are generally the only ones specified on a data sheet because they tend to
be the largest, although some data sheets may specify the value of the worst harmonic.
2.47
ANALOG-DIGITAL CONVERSION
Harmonic distortion is normally specified in dBc (decibels below carrier), although at
audio frequencies it may be specified as a percentage. Harmonic distortion is generally
specified with an input signal near full-scale (generally 0.5 to 1 dB below full-scale to
prevent clipping), but it can be specified at any level. For signals much lower than fullscale, other distortion products due to the DNL of the converter (not direct harmonics)
may limit performance.
RELATIVE
AMPLITUDE
fa = 7MHz
HARMONICS AT: |±Kfs±nfa|
fs = 20MSPS
n = ORDER OF HARMONIC, K = 0, 1, 2, 3, . . .
HARMONICS
3
6
1
HARMONICS
2
2
9
8
5
3
4
5
4
6
7
8
7
9
10
FREQUENCY (MHz)
Figure 2.50: Location of Distortion Products: Input
Signal = 7 MHz, Sampling Rate = 20 MSPS
Total harmonic distortion (THD) is the ratio of the rms value of the fundamental signal to
the mean value of the root-sum-square of its harmonics (generally, only the first 5 are
significant). THD of an ADC is also generally specified with the input signal close to
full-scale, although it can be specified at any level.
Total harmonic distortion plus noise (THD + N) is the ratio of the rms value of the
fundamental signal to the mean value of the root-sum-square of its harmonics plus all
noise components (excluding dc). The bandwidth over which the noise is measured must
be specified. In the case of an FFT, the bandwidth is dc to fs/2. (If the bandwidth of the
measurement is dc to fs/2, THD + N is equal to SINAD—see below).
Signal-to-Noise-and-Distortion Ratio (SINAD), Signal-to-Noise Ratio (SNR), and
Effective Number of Bits (ENOB)
SINAD and SNR deserve careful attention, because there is still some variation between
ADC manufacturers as to their precise meaning. Signal-to-Noise-and Distortion (SINAD,
or S/(N + D) is the ratio of the rms signal amplitude to the mean value of the root-sumsquare (rss) of all other spectral components, including harmonics, but excluding dc (see
Figure 2.50). SINAD is a good indication of the overall dynamic performance of an ADC
as a function of input frequency because it includes all components which make up noise
(including thermal noise) and distortion. It is often plotted for various input amplitudes.
SINAD is equal to THD + N if the bandwidth for the noise measurement is the same. A
typical plot for the AD9226 12-bit, 65-MSPS ADC is shown in Figure 2.52.
2.48
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
SINAD (Signal-to-Noise-and-Distortion Ratio):
The ratio of the rms signal amplitude to the mean value of the
root-sum-squares (RSS) of all other spectral components,
including harmonics, but excluding DC.
ENOB (Effective Number of Bits):
ENOB =
SINAD – 1.76dB
6.02
SNR (Signal-to-Noise Ratio, or Signal-to-Noise Ratio Without
Harmonics:
The ratio of the rms signal amplitude to the mean value of the
root-sum-squares (RSS) of all other spectral components,
excluding the first 5 harmonics and DC
Figure 2.51: SINAD, ENOB, and SNR
ANALOG INPUT FREQUENCY (MHz)
Figure 2.52: AD9226 12-bit, 65-MSPS ADC SINAD and ENOB
for Various Input Full-Scale Spans (Range)
The SINAD plot shows where the ac performance of the ADC degrades due to highfrequency distortion and is usually plotted for frequencies well above the Nyquist
frequency so that performance in undersampling applications can be evaluated. SINAD is
often converted to effective-number-of-bits (ENOB) using the relationship for the
theoretical SNR of an ideal N-bit ADC: SNR = 6.02N + 1.76 dB. The equation is solved
for N, and the value of SINAD is substituted for SNR:
ENOB =
SINAD − 1.76 dB
.
6.02
Eq. 2.20
2.49
ANALOG-DIGITAL CONVERSION
Signal-to-noise ratio (SNR, or SNR-without-harmonics) is calculated the same as SINAD
except that the signal harmonics are excluded from the calculation, leaving only the noise
terms. In practice, it is only necessary to exclude the first 5 harmonics since they
dominate. The SNR plot will degrade at high frequencies, but not as rapidly as SINAD
because of the exclusion of the harmonic terms.
Many current ADC data sheets somewhat loosely refer to SINAD as SNR, so the
engineer must be careful when interpreting these specifications.
Analog Bandwidth
The analog bandwidth of an ADC is that frequency at which the spectral output of the
fundamental swept frequency (as determined by the FFT analysis) is reduced by 3 dB. It
may be specified for either a small signal (SSBW—small signal bandwidth), or a fullscale signal (FPBW—full power bandwidth), so there can be a wide variation in
specifications between manufacturers.
Like an amplifier, the analog bandwidth specification of a converter does not imply that
the ADC maintains good distortion performance up to its bandwidth frequency. In fact,
the SINAD (or ENOB) of most ADCs will begin to degrade considerably before the input
frequency approaches the actual 3-dB bandwidth frequency. Figure 2.53 shows ENOB
and full-scale frequency response of an ADC with a FPBW of 1 MHz, however, the
ENOB begins to drop rapidly above 100 kHz.
FPBW = 1MHz
GAIN (FS INPUT)
GAIN
ENOB (FS INPUT)
ENOB
ENOB (-20dB INPUT)
10
100
1k
10k
100k
1M
10M
ADC INPUT FREQUENCY (Hz)
Figure 2.53: ADC Gain (Bandwidth) and ENOB Versus Frequency Shows
Importance of ENOB Specification
2.50
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
Spurious Free Dynamic Range (SFDR)
Probably the most significant specification for an ADC used in a communications
application is its spurious free dynamic range (SFDR). SFDR of an ADC is defined as
the ratio of the rms signal amplitude to the rms value of the peak spurious spectral
content measured over the bandwidth of interest. Unless otherwise stated, the bandwidth
is assumed to be the Nyquist bandwidth dc to fs/2.
Occasionally the frequency spectrum is divided into an in-band region (containing the
signals of interest) and an out-of-band region (signals here are filtered out digitally). In
this case there may be an in-band SFDR specification and an out-of-band SFDR
specification, respectively.
SFDR is generally plotted as a function of signal amplitude and may be expressed
relative to the signal amplitude (dBc) or the ADC full-scale (dBFS) as shown in Figure
2.54.
FULL SCALE (FS)
INPUT SIGNAL LEVEL (CARRIER)
SFDR (dBFS)
dB
SFDR (dBc)
WORST SPUR LEVEL
FREQUENCY
fs
2
Figure 2.54: Spurious Free Dynamic Range (SFDR)
For a signal near full-scale, the peak spectral spur is generally determined by one of the
first few harmonics of the fundamental. However, as the signal falls several dB below
full-scale, other spurs generally occur which are not direct harmonics of the input signal.
This is because of the differential nonlinearity of the ADC transfer function as discussed
earlier. Therefore, SFDR considers all sources of distortion, regardless of their origin.
The AD6645 is a 14-bit, 80-MSPS wideband ADC designed for communications
applications where high SFDR is important. The single-tone SFDR for a 69.1-MHz input
and a sampling frequency of 80 MSPS is shown in Figure 2.55. Note that a minimum of
89-dBc SFDR is obtained over the entire first Nyquist zone (dc to 40 MHz).
2.51
ANALOG-DIGITAL CONVERSION
ALIASED:
fs – fin = 80 – 61.1 = 10.9MHz
FREQUENCY (MHz)
Figure 2.55: AD6645 14-bit, 80-/105-MSPS ADC SFDR for 69.1-MHz Input
SFDR as a function of signal amplitude is shown in Figure 2.56 for the AD6645. Notice
that over the entire range of signal amplitudes, the SFDR is greater than 90 dBFS. The
abrupt changes in the SFDR plot are due to the differential nonlinearities in the ADC
transfer function. The nonlinearities correspond to those shown in Figure 2.49B, and are
offset from mid-scale such that input signals less than about 65 dBFS do not exercise any
of the points of increased DNL. It should be noted that the SFDR can be improved by
injecting a small out-of-band dither signal—at the expense of a slight degradation in
SNR.
Figure 2.56: AD6645 14-bit, 80-/105-MSPS ADC SFDR vs.
Input Power Level for 69.1-MHz Input
2.52
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
SFDR is generally much greater than the ADCs theoretical N-bit SNR (6.02N + 1.76 dB).
For example, the AD6645 is a 14-bit ADC with an SFDR of 90 dBc and a typical SNR of
73.5 dB (the theoretical SNR for 14-bits is 86 dB). This is because there is a fundamental
distinction between noise and distortion measurements. The process gain of the FFT
(33 dB for a 4096-point FFT) allows frequency spurs well below the noise floor to be
observed. Adding extra resolution to an ADC may serve to increase its SNR but may or
may not increase its SFDR.
Two Tone Intermodulation Distortion (IMD)
Two tone IMD is measured by applying two spectrally pure sinewaves to the ADC at
frequencies f1 and f2, usually relatively close together. The amplitude of each tone is set
slightly more than 6 dB below full-scale so that the ADC does not clip when the two
tones add in-phase. The location of the second and third-order products are shown in
Figure 2.57. Notice that the second-order products fall at frequencies which can be
removed by digital filters. However, the third-order products 2f2 – f1 and 2f1 – f2 are close
to the original signals and are more difficult to filter. Unless otherwise specified, twotone IMD refers to these third-order products. The value of the IMD product is expressed
in dBc relative to the value of either of the two original tones, and not to their sum.
2 = SECOND ORDER IMD PRODUCTS
f1
3
f2
= THIRD ORDER IMD PRODUCTS
NOTE: f1 = 5MHz, f2 = 6MHz
2
2
f2 - f1
f2 + f1
2f 2
2f 1
3
3
2f1 - f2
1
4
5
2f2 - f1
6 7
10 11 12
3
3f 1
3
2f2 + f1
2f1 + f2
3f 2
15 16 17 18
FREQUENCY: MHz
Figure 2.57: Second and Third-Order Intermodulation Products
for f1 = 5 MHz, f2 = 6 MHz
Note, however, that if the two tones are close to fs/4, then the aliased third harmonics of
the fundamentals can make the identification of the actual 2f2 – f1 and 2f1 – f2 products
difficult. This is because the third harmonic of fs/4 is 3fs/4, and the alias occurs at
fs – 3fs/4 = fs/4. Similarly, if the two tones are close to fs/3, the aliased second harmonics
may interfere with the measurement. The same reasoning applies here; the second
harmonic of fs/3 is 2fs/3, and its alias occurs at fs – 2fs/3 = fs/3.
2.53
ANALOG-DIGITAL CONVERSION
Second and Third-Order Intercept Points, 1 dB Compression Point
Third-order IMD products are especially troublesome in multi-channel communications
systems where the channel separation is constant across the frequency band. Third-order
IMD products can mask out small signals in the presence of larger ones.
In amplifiers, it is common practice to specify the third-order IMD products in terms of
the third order intercept point, as is shown by Figure 2.58. Two spectrally pure tones are
applied to the system. The output signal power in a single tone (in dBm) as well as the
relative amplitude of the third-order products (referenced to a single tone) are plotted as a
function of input signal power. The fundamental is shown by the slope = 1 curve in the
diagram. If the system nonlinearity is approximated by a power series expansion, it can
be shown that second-order IMD amplitudes increase 2 dB for every 1 dB of signal
increase, as represented by slope = 2 curve in the diagram.
Similarly, the third-order IMD amplitudes increase 3 dB for every 1-dB of signal
increase, as indicated by the slope = 3 plotted line. With a low level two-tone input
signal, and two data points, one can draw the second and third order IMD lines as they
are shown in Figure 2.58 (using the principle that a point and a slope define a straight
line).
SECOND ORDER
INTERCEPT
IP2
OUTPUT
POWER
(PER TONE)
dBm
THIRD ORDER
INTERCEPT
IP3
1dB
1 dB COMPRESSION
POINT
FUNDAMENTAL
(SLOPE = 1)
SECOND
ORDER IMD
(SLOPE = 2)
THIRD ORDER IMD
(SLOPE = 3)
INPUT POWER (PER TONE), dBm
Figure 2.58: Definition of Intercept Points and 1-dB Compression
Points for Amplifiers
Once the input reaches a certain level however, the output signal begins to soft-limit, or
compress. A parameter of interest here is the 1-dB compression point. This is the point
where the output signal is compressed 1 dB from an ideal input/output transfer function.
This is shown in Figure 2.58 within the region where the ideal slope = 1 line becomes
dotted, and the actual response exhibits compression (solid).
2.54
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
Nevertheless, both the second- and third-order intercept lines may be extended, to
intersect the (dotted) extension of the ideal output signal line. These intersections are
called the second and third-order intercept points, respectively, or IP2 and IP3. These
power level values are usually referenced to the output power of the device delivered to a
matched load (usually, but not necessarily 50 Ω) expressed in dBm.
It should be noted that IP2, IP3, and the 1-dB compression point are all a function of
frequency, and as one would expect, the distortion is worse at higher frequencies.
For a given frequency, knowing the third order intercept point allows calculation of the
approximate level of the third-order IMD products as a function of output signal level.
The concept of second- and third-order intercept points is not valid for an ADC, because
the distortion products do not vary in a predictable manner (as a function of signal
amplitude). The ADC does not gradually begin to compress signals approaching fullscale (there is no 1-dB compression point); it acts as a hard limiter as soon as the signal
exceeds the ADC input range, thereby suddenly producing extreme amounts of distortion
because of clipping. On the other hand, for signals much below full-scale, the distortion
floor remains relatively constant and is independent of signal level. This is shown
graphically in Figure 2.59.
OUTPUT
POWER*
(PER TONE)
dBm
ADC HARD LIMITS
IN THIS REGION,
LARGE IMD
PRODUCTS RESULT
*ANALOG EQUIVALENT
OF DIGITAL SIGNAL LEVEL
MEASURED BY ADC
INTERSECTION
HAS NO PRACTICAL
SIGNIFICANCE
FUNDAMENTAL
(SLOPE = 1)
IMD PRODUCTS
CONSTANT
IN THIS REGION
(SLOPE = 1)
IMD PRODUCTS MAY
START TO INCREASE
IN THIS REGION
INPUT POWER (PER TONE), dBm
Figure 2.59: Intercept Points for Data Converters Have No Practical Significance
The IMD curve in Figure 2.59 is divided into three regions. For low level input signals,
the IMD products remain relatively constant regardless of signal level. This implies that
as the input signal increases 1 dB, the ratio of the signal to the IMD level will increase
1 dB also. When the input signal is within a few dB of the ADC full-scale range, the IMD
may start to increase (but it might not in a very well-designed ADC). The exact level at
which this occurs is dependent on the particular ADC under consideration—some ADCs
may not exhibit significant increases in the IMD products over their full input range,
2.55
ANALOG-DIGITAL CONVERSION
however most will. As the input signal continues to increase beyond full-scale, the ADC
should function act as an ideal limiter, and the IMD products become very large.
For these reasons, the 2nd and 3rd order IMD intercept points are not specified for ADCs.
It should be noted that essentially the same arguments apply to DACs. In either case, the
single- or multi-tone SFDR specification is the most accepted way to measure data
converter distortion.
Multi-Tone Spurious Free Dynamic Range
Two-tone and multi-tone SFDR is often measured in communications applications. The
larger number of tones more closely simulates the wideband frequency spectrum of
cellular telephone systems such as AMPS or GSM. Figure 2.60 shows the 2-tone
intermodulation performance of the AD6645 14-bit, 80-/105-MSPS ADC. The input
tones are at 55.25 MHz and 56.25 MHz and are located in the second Nyquist Zone.
The aliased tones therefore occur at 23.75 MHz and 24.75 MHz in the first Nyquist Zone.
High SFDR increases the receiver's ability to capture small signals in the presence of
large ones, and prevents the small signals from being masked by the intermodulation
products of the larger ones. Figure 2.61 shows the AD6645 two-tone SFDR as a function
of input signal amplitude for the same input frequencies.
ALIASED
80 – 56.25
= 23.75MHz
80 – 55.25
= 24.75MHz
ALIASED
ALIASED
Figure 2.60: Two-Tone SFDR for AD6645 14-bit, 80-/105-MSPS ADC,
Input Tones: 55.25 MHz and 56.25 MHz
2.56
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
Figure 2.61: Two-Tone SFDR vs. Input Amplitude
for AD6645 14-bit, 80-/105-MSPS ADC
Wideband CDMA (WCDMA) Adjacent Channel Power Ratio (ACPR) and
Adjacent Channel Leakage Ratio (ADLR)
A wideband CDMA channel has a bandwidth of approximately 3.84 MHz, and channel
spacing is 5 MHz. The ratio in dBc between the measured power within a channel
relative to its adjacent channel is defined as the adjacent channel power ratio (ACPR).
The ratio in dBc between the measured power within the channel bandwidth relative to
the noise level in an adjacent empty carrier channel is defined as adjacent channel
leakage ratio (ACLR).
ALIASED: 2fs – fin = 153.6 – 140 = 13.6MHz
ACLR
= 70dB
Figure 2.62: Wideband CDMA (WCDMA)
Adjacent Channel Leakage Ratio (ACLR)
2.57
ANALOG-DIGITAL CONVERSION
Figure 2.62 shows a single wideband CDMA channel centered at 140 MHz sampled at a
frequency of 76.8 MSPS using the AD6645. This is a good example of undersampling
(direct IF-to-digital conversion).The signal lies within the fourth Nyquist zone: 3fs/2 to
2fs (115.2 MHz to 153.6 MHz). The aliased signal within the first Nyquist zone is
therefore centered at 2fs – fa = 153.6 – 140 = 13.6 MHz. The diagram also shows the
location of the aliased harmonics. For example, the second harmonic of the input signal
occurs at 2 × 140 = 280 MHz, and the aliased component occurs at 4fs – 2fa =
4 × 76.8 – 280 = 307.2 – 280 = 27.2 MHz.
Noise Power Ratio (NPR)
Noise power ratio has been used extensively to measure the transmission characteristics
of Frequency Division Multiple Access (FDMA) communications links (see Reference
7). In a typical FDMA system, 4-kHz wide voice channels are "stacked" in frequency
bins for transmission over coaxial, microwave, or satellite equipment. At the receiving
end, the FDMA data is demultiplexed and returned to 4-kHz individual baseband
channels. In an FDMA system having more than approximately 100 channels, the FDMA
signal can be approximated by Gaussian noise with the appropriate bandwidth. An
individual 4-kHz channel can be measured for "quietness" using a narrow-band notch
(bandstop) filter and a specially tuned receiver which measures the noise power inside the
4-kHz notch (see Figure 2.63).
GAUSSIAN
NOISE
SOURCE
GAUSSIAN
NOISE
SOURCE
LPF
NOTCH
FILTER
LPF
NOTCH
FILTER
TRANSMISSION
SYSTEM
NARROWBAND
RECEIVER
N
ADC
BUFFER
MEMORY
AND FFT
PROCESSOR
fs
RMS
NOISE
LEVEL
(dB)
NPR
FREQUENCY
0.5f s
Figure 2.63: Noise Power Ratio (NPR) Measurements
Noise Power Ratio (NPR) measurements are straightforward. With the notch filter out,
the rms noise power of the signal inside the notch is measured by the narrowband
receiver. The notch filter is then switched in, and the residual noise inside the slot is
2.58
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
measured. The ratio of these two readings expressed in dB is the NPR. Several slot
frequencies across the noise bandwidth (low, midband, and high) are tested to
characterize the system adequately. NPR measurements on ADCs are made in a similar
manner except the analog receiver is replaced by a buffer memory and an FFT processor.
The NPR is plotted as a function of rms noise level referred to the peak range of the
system. For very low noise loading level, the undesired noise (in non-digital systems) is
primarily thermal noise and is independent of the input noise level. Over this region of
the curve, a 1-dB increase in noise loading level causes a 1-dB increase in NPR. As the
noise loading level is increased, the amplifiers in the system begin to overload, creating
intermodulation products which cause the noise floor of the system to increase. As the
input noise increases further, the effects of "overload" noise predominate, and the NPR is
reduced dramatically. FDMA systems are usually operated at a noise loading level a few
dB below the point of maximum NPR.
In a digital system containing an ADC, the noise within the slot is primarily quantization
noise when low levels of noise input are applied. The NPR curve is linear in this region.
As the noise level increases, there is a one-for-one correspondence between the noise
level and the NPR. At some level, however, "clipping" noise caused by the hard-limiting
action of the ADC begins to dominate. A theoretical curve for 10, 11, and 12-bit ADCs is
shown in Figure 2.64 (see References 8 and 21).
ADC RANGE = ±VO
NPR
(dB)
62.7dB
VO
k=
σ
60
σ = RMS NOISE LEVEL
IT
S
57.1dB
11
-B
IT
S
12
-B
55
51.6dB
10
-B
IT
S
50
45
–30
–25
–20
–15
–10
RMS NOISE LOADING LEVEL = –20log10(k) dB
Figure 2.64: Theoretical NPR for 10, 11, 12-bit ADCs
Figure 2.65 shows the maximum theoretical NPR and the noise loading level at which the
maximum value occurs for 8 to 16-bit ADCs. The ADC input range is 2VO peak-to-peak.
The rms noise level is σ, and the noise-loading factor k (crest factor) is defined as VO/σ ,
the peak-to-rms ratio (k is expressed either as numerical ratio or in dB).
2.59
ANALOG-DIGITAL CONVERSION
BITS
k OPTIMUM
k(dB)
MAX NPR (dB)
8
3.92
11.87
40.60
9
4.22
12.50
46.05
10
4.50
13.06
51.56
11
4.76
13.55
57.12
12
5.01
14.00
62.71
13
5.26
14.41
68.35
14
5.49
14.79
74.01
15
5.72
15.15
79.70
16
5.94
15.47
85.40
ADC Range = ±Vo
k = Vo / σ
σ = RMS Noise Level
Figure 2.65: Theoretical Maximum NPR for 8 to 16-bit ADCs
In multi-channel high frequency communication systems, where there is little or no phase
correlation between channels, NPR can also be used to simulate the distortion caused by
a large number of individual channels, similar to an FDMA system. A notch filter is
placed between the noise source and the ADC, and an FFT output is used in place of the
analog receiver. The width of the notch filter is set for several MHz as shown in Figure
2.66 for the AD9430 12-bit 170-/210-MSPS ADC. The notch is centered at 19 MHz, and
the NPR is the "depth" of the notch. An ideal ADC will only generate quantization noise
inside the notch, however a practical one has additional noise components due to
additional noise and intermodulation distortion caused by ADC imperfections. Notice
that the NPR is about 57 dB compared to 62.7-dB theoretical.
NPR =
57dB
INPUT FREQUENCY
Figure 2.66: AD9430 12-bit,170-/210-MSPS ADC NPR
Measures 57 dB (62.7 dB Theoretical)
2.60
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
Noise Factor (F) and Noise Figure (NF)
Noise figure (NF) is a popular specification among RF system designers. It is used to
characterize RF amplifiers, mixers, etc., and widely used as a tool in radio receiver
design. Many excellent textbooks on communications and receiver design treat noise
figure extensively (see Reference 9, for example)—it is not the purpose of this discussion
to discuss the topic in much detail, but only how it applies to data converters.
Since many wideband operational amplifiers and ADCs are now being used in RF
applications, the inevitable day has come where the noise figure of these devices becomes
important. As discussed in Reference 10, in order to determine the noise figure of an op
amp correctly, one must not only know op amp voltage and current noise, but the exact
circuit conditions—closed-loop gain, gain-setting resistor values, source resistance,
bandwidth, etc. Calculating the noise figure for an ADC is even more of a challenge as
will be seen.
Figure 2.67 shows the basic model for defining the noise figure of an ADC. The noise
factor, F, is simply defined as the ratio of the total effective input noise power of the
ADC to the amount of that noise power caused by the source resistance alone. Because
the impedance is matched, the square of the voltage noise can be used instead of noise
power. The noise figure, NF, is simply the noise factor expressed in dB, NF = 10log10F.
SOURCE
fs
B = Filter
Noise Bandwidth
ADC
FILTER
R
B
PFS(dBm)
R*
N
*May be external
(TOTAL EFFECTIVE INPUT NOISE) 2
NOISE FACTOR (F) =
NOISE FIGURE (NF) = 10log10
(TOTAL INPUT NOISE DUE TO SOURCE R) 2
(TOTAL EFFECTIVE INPUT NOISE) 2
(TOTAL INPUT NOISE DUE TO SOURCE R) 2
Note: Noise Must be Measured Over the Filter Noise Bandwidth, B
Figure 2.67: Noise Figure for ADCs: Use with Caution!
This model assumes the input to the ADC comes from a source having a resistance, R,
and that the input is band-limited to fs/2 with a filter having a noise bandwidth equal to
fs/2. It is also possible to further band-limit the input signal resulting in oversampling and
process gain, and this condition will be discussed shortly.
2.61
ANALOG-DIGITAL CONVERSION
It is also assumed that the input impedance to the ADC is equal to the source resistance.
Many ADCs have a high input impedance, so this termination resistance may be external
to the ADC or used in parallel with the internal resistance to produce an equivalent
termination resistance equal to R. The full-scale input power is the power of a sinewave
whose peak-to-peak amplitude fills the entire ADC input range. The full-scale input
sinewave given by the following equation has a peak-to-peak amplitude of 2VO
corresponding to the peak-to-peak input range of the ADC:
v(t) = VO sin 2πft
Eq. 2.21
The full-scale power in this sinewave is given by:
PFS =
( VO / 2 ) 2 VO 2
=
R
2R
Eq. 2.22
It is customary to express this power in dBm (referenced to 1 mW) as follows:
 P 
PFS(dBm) = 10 log10  FS  .
1 mW 
Eq. 2.23
The noise bandwidth of a non-ideal brick wall filter is defined as the bandwidth of an
ideal brick wall filter which will pass the same noise power as the non-ideal filter.
Therefore, the noise bandwidth of a filter is always greater than the 3-dB bandwidth of
the filter by a factor which depends upon the sharpness of the cutoff region of the filter.
Figure 2.68 shows the relationship between the noise bandwidth and the 3-dB bandwidth
for Butterworth filters up to 5 poles. Note that for two poles, the noise bandwidth and
3-dB bandwidth are within 11% of each other, and beyond that the two quantities are
essentially equal.
NUMBER OF POLES
NOISE BW / 3dB BW
1
1.57
2
1.11
3
1.05
4
1.03
5
1.02
Figure 2.68: Relationship Between Noise Bandwidth
and 3-dB Bandwidth for Butterworth Filter
2.62
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
The first step in the NF calculation is to calculate the effective input noise of the ADC
from its SNR. The SNR of the ADC is given for a variety of input frequencies, so be sure
and use the value corresponding to the input frequency of interest. Also, make sure that
the harmonics are not included in the SNR number—some ADC data sheets may confuse
SINAD with SNR. Once the SNR is known, the equivalent input rms voltage noise can
be calculated starting from the equation:
 VFS RMS 
SNR = 20 log10 

 VNOISE RMS 
Eq. 2.24
Solving for VNOISE RMS:
VNOISE RMS = VFS RMS ⋅ 10 −SNR / 20
Eq. 2.25
This is the total effective input rms noise voltage at the carrier frequency measured over
the Nyquist bandwidth, dc to fs/2. Note that this noise includes the source resistance
noise. These results are summarized in Figure 2.69.
Start with the SNR of the ADC measured at the carrier frequency
(Note: this SNR value does not include the harmonics of the
fundamental and is measured over the Nyquist bandwidth, dc to fs/2)
SNR = 20 log10
VFS-RMS
VNOISE-RMS
VNOISE-RMS = VFS-RMS 10 –SNR / 20
This is the total ADC effective input noise at the carrier frequency
measured over the Nyquist bandwidth, dc to fs/2
Figure 2.69: Calculating ADC Total Effective Input Noise from SNR
The next step is to actually calculate the noise figure. In Figure 2.70 notice that the
amount of the input voltage noise due to the source resistance is the voltage noise of the
source resistance √(4kTBR) divided by two, or √(kTBR) because of the 2:1 attenuator
formed by the ADC input termination resistor.
The expression for the noise factor F can be written:
F=
VNOISE RMS 2
kTRB
 VFS RMS2  1
1
   10 −SNR / 10  
=
 B 
R

  kT 


[
]
Eq. 2.26
2.63
ANALOG-DIGITAL CONVERSION
The noise figure is obtained by converting F into dB and simplifying:
NF = 1010logF = PFS(dBm) + 174 dBm – SNR – 1010logB,
Eq. 2.27
Where SNR is in dB, B in Hz, T = 300 K, k = 1.38 × 10–23 J/K.
÷2
fs
4kTBR
kTBR
~
FILTER
R
f
B= s
2
NF =
VNOISE-RMS
kTRB
2
=
B = Filter
Noise Bandwidth
R
PFS(dBm)
= VFS-RMS 10
VNOISE-RMS
F=
ADC
VFS-RMS 2
1
R
kT
–SNR / 20
10
–SNR / 10
1
B
10 log10F = PFS(dBm) + 174dBm – SNR – 10 log10B ,
where SNR is in dB, B in Hz,
T = 300K, k = 1.38 × 10–23 J/K
Figure 2.70: ADC Noise Figure in Terms of SNR,
Sampling Rate, and Input Power
Oversampling and filtering can be used to decrease the noise figure as a result of the
process gain as has been previously discussed. In this case, the signal bandwidth B is less
than fs/2. Figure 2.71 shows the correction factor which results in the following equation:
NF = 1010logF = PFS(dBm) + 174 dBm – SNR – 10 log10[fs/2B] – 10 log10B. Eq. 2.28
Figure 2.72 shows an example NF calculation for the AD6645 14-bit, 80-MSPS ADC. A
52.3-Ω resistor is added in parallel with the AD6645 input impedance of 1 kΩ to make
the net input impedance 50 Ω. The ADC is operating under Nyquist conditions, and the
SNR of 74 dB is the starting point for the calculations using Eq. 2.28 above. A noise
figure of 34.8 dB is obtained.
2.64
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
fs
ADC
FILTER
R
NF =
f
B< s
2
R
PFS(dBm)
PFS(dBm) + 174dBm – SNR – 10 log10
Measured
DC to fs / 2
where SNR is in dB, B in Hz,
B = Filter
Noise Bandwidth
fs / 2
– 10 log10B,
B
Process
Gain
T = 300K, k = 1.38 × 10–23 J/K
Figure 2.71: Effect of Oversampling and Process Gain on ADC Noise Figure
1:1
TURNS RATIO
50Ω
AD6645
50Ω
FILTER
B = 40MHz
52.3Ω
VFS P-P = 2.2V
fs = 80MSPS
SNR = 74dB
Input 3dB BW = 250MHz
1kΩ
VFS P-P = 2.2V
fs = 80MSPS
VFS-RMS = 0.778V
PFS =
(0.778) 2
50
= 12.1mW
PFS(dBm) = +10.8dBm
NF = PFS(dBm) + 174dBm – SNR – 10 log10B
= +10.8dBm + 174dBm – 74dB – 10 log10(40 × 106)
= 34.8dB
Figure 2.72: Example Calculation of Noise Figure
Under Nyquist Conditions for AD6645
Figure 2.73 shows how using an RF transformer with voltage gain can improve the noise
figure. Figure 2.73A shows a 1:1 turns ratio, and the noise figure (from Figure 2.72) is
34.8. Figure 2.73B shows a transformer with a 1:2 turns ratio. The 249-Ω resistor in
parallel with the AD6645 internal resistance results in a net input impedance of 200 Ω.
2.65
ANALOG-DIGITAL CONVERSION
The noise figure is improved by 6 dB because of the "noise-free" voltage gain of the
transformer. Figure 2.73C shows a transformer with a 1:4 turns ratio. The AD6645 input
is paralleled with a 4.02-kΩ resistor to make the net input impedance 800 Ω. The noise
figure is improved by another 6 dB. Transformers with higher turns ratios are not
generally practical because of bandwidth and distortion limitations.
1:1 TURNS RATIO
(A)
50Ω
B=
40MHz
50Ω
AD6645
52.3Ω
1kΩ
VFS P-P = 2.2V
fs = 80MSPS
SNR = 74dB
Input 3dB BW = 250MHz
NF = 34.8dB
PFS(dBm) = +10.8dBm
1:2 TURNS RATIO
(B)
50Ω
B=
40MHz
200Ω
AD6645
249Ω
1kΩ
NF = 28.8dB
PFS(dBm) = +4.8dBm
1:4 TURNS RATIO
(C)
50Ω
B=
40MHz
800Ω
AD6645
4.02kΩ
1kΩ
NF = 22.8dB
PFS(dBm) = –1.2dBm
Figure 2.73: Using RF Transformers to Improve Overall ADC Noise Figure
Even with the 1:4 turns ratio transformer, the overall noise figure for the AD6645 was
still 22.8 dB, still relatively high by RF standards. The solution is to provide low-noise
high-gain stages ahead of the ADC. Figure 2.74 shows how the Friis equation is used to
calculate the noise factor for cascaded gain stages. Notice that high gain in the first stage
reduces the contribution of the noise factor of the second stage—the noise factor of the
first stage dominates the overall noise factor.
Figure 2.75 shows the effects of a high-gain (25 dB) low-noise (NF = 4 dB) stage placed
in front of a relatively high NF stage (30 dB)—the noise figure of the second stage is
typical of high performance ADCs. The overall noise figure is 7.53 dB, only 3.53 dB
higher than the first stage noise figure of 4 dB.
2.66
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
G1
F1
RS
G2
F2
FT = F1 +
F2 – 1
+
G1
G3
F3
F3 – 1
G1•G2
+
G4
F4
RL
F4 – 1
+...
G1•G2•G3
High gain in the first stage reduces the
contribution of the NF of the second stage
NF of the first stage dominates the total NF
NFT = 10 log10FT
Figure 2.74: Cascaded Noise Figure Using the Friis Equation
RS
G1dB = 25dB
G2dB = 0dB
NF1 = 4dB
NF2 = 30dB
RL
G1 = 10 25/10 = 10 2.5 = 316, F1 = 10 4/10 = 10 0.4 = 2.51
F2 = 10 30/10 = 10 3 = 1000
G2 = 1,
FT = F1 +
F2 – 1
G1
= 2.51 + 1000 – 1 = 2.51 + 3.16 = 5.67
316
NFT = 10 log105.67 = 7.53dB
The first stage dominates the overall NF
It should have the highest gain possible with the lowest NF possible
Figure 2.75: Example of Two-Stage Cascaded Network
In summary, applying the noise figure concept to characterize wideband ADCs must be
done with extreme caution to prevent misleading results. Simply trying to minimize the
noise figure using the equations can actually increase circuit noise.
For instance, NF decreases with increasing source resistance according to the
calculations, but increased source resistance increases circuit noise. Also, NF decreases
with increasing ADC input bandwidth if there is no input filtering. This is also
contradictory, because widening the bandwidth increases noise. In both these cases, the
circuit noise increases, and the NF decreases. The reason NF decreases is that the source
2.67
ANALOG-DIGITAL CONVERSION
noise makes up a larger component of the total noise (which remains relatively constant
because the ADC noise is much greater than the source noise); therefore according to the
calculation, NF decreases, but actual circuit noise increases.
It is true that on a stand-alone basis ADCs have relatively high noise figures compared to
other RF parts such as LNAs or mixers. In the system the ADC should be preceded with
low-noise gain blocks as shown in the example of Figure 2.75. Noise figure
considerations for ADCs are summarized in Figure 2.76.
NF decreases with increasing source resistance.
NF decreases with increasing ADC input bandwidth if there is no
input filtering.
In both cases, the circuit noise increases, and the NF decreases.
The reason NF decreases is that the source noise makes up a
larger component of the total noise (which remains relatively
constant because the ADC noise is much greater than the source
noise).
In practice, input filtering is used to limit the input noise bandwidth
and reduce overall system noise.
ADCs have relatively high NF compared to other RF parts. In the
system the ADC should be preceded with low-noise gain blocks.
Exercise caution when using NF!
Figure 2.76: Noise Figure Considerations for ADCs: Summary and Caution
Aperture Time, Aperture Delay Time, and Aperture Jitter
Perhaps the most misunderstood and misused ADC and sample-and-hold (or track-andhold) specifications are those that include the word aperture. The most essential dynamic
property of a SHA is its ability to disconnect quickly the hold capacitor from the input
buffer amplifier as shown in Figure 2.77. The short (but non-zero) interval required for
this action is called aperture time (or sampling aperture), ta. The actual value of the
voltage that is held at the end of this interval is a function of both the input signal slew
rate and the errors introduced by the switching operation itself. Figure 2.77 shows what
happens when the hold command is applied with an input signal of two arbitrary slopes
labeled as 1 and 2. For clarity, the sample-to-hold pedestal and switching transients are
ignored. The value that is finally held is a delayed version of the input signal, averaged
over the aperture time of the switch as shown in Figure 2.77. The first-order model
assumes that the final value of the voltage on the hold capacitor is approximately equal to
the average value of the signal applied to the switch over the interval during which the
switch changes from a low to high impedance (ta).
The model shows that the finite time required for the switch to open (ta) is equivalent to
introducing a small delay te in the sampling clock driving the SHA. This delay is constant
and may either be positive or negative. The diagram shows that the same value of te
works for the two signals, even though the slopes are different. This delay is called
2.68
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
effective aperture delay time, aperture delay time, or simply aperture delay, te. In an
ADC, the aperture delay time is referenced to the input of the converter, and the effects of
the analog propagation delay through the input buffer, tda and the digital delay through
the switch driver, tdd, must be considered. Referenced to the ADC inputs, aperture time,
te', is defined as the time difference between the analog propagation delay of the front-end
buffer, tda, and the switch driver digital delay, tdd, plus one-half the aperture time, ta/2.
ANALOG
DELAY, tda
INPUT
SIGNAL
INPUT
SAMPLING
CLOCK
APERTURE
TIME, ta
ta = APERTURE TIME
tda = ANALOG DELAY
tdd = DIGITAL DELAY
te = ta / 2 = APERTURE DELAY
TIME FOR tda = tdd
te
2
CHOLD
SWITCH
DRIVER
VOLTAGE ON
HOLD CAPACITOR
SWITCH
INPUT SIGNALS
1
DIGITAL
DELAY, tdd
te' = APERTURE DELAY
TIME REFERENCED TO INPUTS
1
2
SWITCH
DRIVER OUTPUT
ta
t
te' = tdd – tda + a
2
HOLD
SAMPLE
Figure 2.77: Sample-and-Hold Waveforms and Definitions
The effective aperture delay time is usually positive, but may be negative if the sum of
one-half the aperture time, ta/2, and the switch driver digital delay, tdd, is less than the
propagation delay through the input buffer, tda. The aperture delay specification thus
establishes when the input signal is actually sampled with respect to the sampling clock
edge.
Aperture delay time can be measured by applying a bipolar sinewave signal to the ADC
and adjusting the synchronous sampling clock delay such that the output of the ADC is
mid-scale (corresponding to the zero-crossing of the sinewave). The relative delay
between the input sampling clock edge and the actual zero-crossing of the input sinewave
is the aperture delay time (see Figure 2.78).
Aperture delay produces no errors (assuming it is relatively short with respect to the hold
time), but acts as a fixed delay in either the sampling clock input or the analog input
(depending on its sign). However, in simultaneous sampling applications or in direct I/Q
demodulation where two or more ADCs must be well matched, variations in the aperture
delay between converters can produce errors on fast slewing signals. In these
applications, the aperture delay mismatches must be removed by properly adjusting the
phases of the individual sampling clocks to the various ADCs.
2.69
ANALOG-DIGITAL CONVERSION
If, however, there is sample-to-sample variation in aperture delay (aperture jitter), then a
corresponding voltage error is produced as shown in Figure 2.79. This sample-to-sample
variation in the instant the switch opens is called aperture uncertainty, or aperture jitter
and is usually measured in rms picoseconds. The amplitude of the associated output error
is related to the rate-of-change of the analog input. For any given value of aperture jitter,
the aperture jitter error increases as the input dv/dt increases. The effects of phase jitter
on the external sampling clock (or the analog input for that matter) produce exactly the
same type of error.
+FS
ZERO CROSSING
ANALOG INPUT
SINEWAVE
0V
-FS
–t e '
+te '
SAMPLING
CLOCK
t e'
Figure 2.78: Effective Aperture Delay Time
Measured with Respect to ADC Input
dv
∆v =
dt
ANALOG
INPUT
dv
= SLOPE
dt
∆t
∆ v RMS = APERTURE JITTER ERROR
NOMINAL
HELD
OUTPUT
∆ t RMS = APERTURE JITTER
HOLD
TRACK
Figure 2.79: Effects of Aperture Jitter and Sampling Clock Jitter
2.70
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
The effects of aperture and sampling clock jitter on an ideal ADCs SNR can be predicted
by the following simple analysis. Assume an input signal given by
v(t) = VO sin 2πft
Eq. 2.29
The rate of change of this signal is given by:
dv/dt = 2πfVO cos 2πft.
Eq. 2.30
The rms value of dv/dt can be obtained by dividing the amplitude, 2πfVO, by √2:
dv/dt| rms = 2πfVO / √2.
Eq. 2.31
Now let ∆vrms = the rms voltage error and ∆t = the rms aperture jitter tj, and substitute:
∆vrms / tj = 2πfVO / √2.
Eq. 2.32
∆vrms = 2πfVO tj/ √2.
Eq. 2.33
Solving for ∆vrms :
The rms value of the full-scale input sinewave is VO/√2, therefore the rms signal to rms
noise ratio is given by
 V / 2 
 1 
V / 2 
O
=
SNR = 20 log10  O
20
log
=
20
log


.

10
10 
 2πfVO t j / 2 
 2πf t j 
 ∆v rms 
Eq. 2.34
This equation assumes an infinite resolution ADC where aperture jitter is the only factor
in determining the SNR. This equation is plotted in Figure 2.80 and shows the serious
effects of aperture and sampling clock jitter on SNR, especially at higher input/output
frequencies. Therefore, extreme care must be taken to minimize phase noise in the
sampling/reconstruction clock of any sampled data system.
This care must extend to all aspects of the clock signal: the oscillator itself (for example,
a 555 timer is absolutely inadequate, but even a quartz crystal oscillator can give
problems if it uses an active device which shares a chip with noisy logic); the
transmission path (these clocks are very vulnerable to interference of all sorts), and phase
noise introduced in the ADC or DAC. As discussed, a very common source of phase
noise in converter circuitry is aperture jitter in the integral sample-and-hold (SHA)
circuitry, however the total rms jitter will be composed of a number of components—the
actual SHA aperture jitter often being the least of them.
2.71
ANALOG-DIGITAL CONVERSION
tj = 50fs
120
SNR = 20log 10
tj = 0.1ps
1
2π ft j
16
tj = 1ps
100
18
14
tj = 10ps
80
SNR
(dB)
12
10
tj = 100ps
60
ENOB
8
tj = 1ns
40
6
4
20
1
3
100
10
30
FULL-SCALE SINEWAVE ANALOG INPUT FREQUENCY (MHz)
Figure 2.80: Theoretical SNR and ENOB Due to Jitter vs.
Fullscale Sinewave Input Frequency
A Simple Equation for the Total SNR of an ADC
A relatively simple equation for the ADC SNR in terms of sampling clock and aperture
jitter, DNL, effective input noise, and the number of bits of resolution is shown in Figure
2.81. The equation combines the various error terms on an rss basis. The average DNL
error, ε, is computed from histogram data. This equation is used in Figure 2.82 to predict
the SNR performance of the AD6645 14-bit, 80-MSPS ADC as a function of sampling
clock and aperture jitter.
SAMPLING
CLOCK JITTER
SNR = – 20log10 (2π × fa × tj rms) 2 +
fa
tj rms
QUANTIZATION
NOISE, DNL
2 1+ε
3
2
2N
+
EFFECTIVE
INPUT NOISE
2 × √2 × VNOISErms
2
1
2
2N
= Analog input frequency of fullscale input sinewave
= Combined rms jitter of internal ADC and external clock
ε
= Average DNL of the ADC (typically 0.41 LSB for AD6645)
N
= Number of bits in the ADC
VNOISErms = Effective input noise of ADC (typically 0.9LSB rms for AD6645)
If tj = 0, ε = 0, and VNOISErms = 0, the above equation reduces to the familiar:
SNR = 6.02 N + 1.76dB
Figure 2.81: Relationship Between SNR, Sampling Clock Jitter,
Quantization Noise, DNL, and Input Noise
2.72
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
Figure 2.82: AD6645 SNR Versus Jitter
Before the 1980s, most sampling ADCs were generally built up from a separate SHA and
ADC. Interface design was difficult, and a key parameter was aperture jitter in the SHA.
Today, almost all sampled data systems use sampling ADCs which contain an integral
SHA. The aperture jitter of the SHA may not be specified as such, but this is not a cause
of concern if the SNR or ENOB is clearly specified, since a guarantee of a specific SNR
is an implicit guarantee of an adequate aperture jitter specification. However, the use of
an additional high-performance SHA will sometimes improve the high-frequency ENOB
of even the best sampling ADC by presenting "dc" to the ADC, and may be more costeffective than replacing the ADC with a more expensive one.
ADC Transient Response and Overvoltage Recovery
Most high speed ADCs designed for communications applications are specified primarily
in the frequency domain. However, in general purpose data acquisition applications the
transient response (or settling time) of the ADC is important. The transient response of
an ADC is the time required for the ADC to settle to rated accuracy (usually 1 LSB) after
the application of a full-scale step input. The typical response of a general-purpose 12 bit,
10-MSPS ADC is shown in Figure 2.83, showing a 1 LSB settling time of less than
40 ns. The settling time specification is critical in the typical data acquisition system
application where the ADC is being driven by an analog multiplexer as shown in Figure
2.84. The multiplexer output can deliver a full-scale sample-to-sample change to the
ADC input. If both the multiplexer and the ADC have not both settled to the required
accuracy, channel-to-channel crosstalk will result, even though only dc or low frequency
signals are present on the multiplexer inputs.
2.73
ANALOG-DIGITAL CONVERSION
Figure 2.83: ADC Transient Response (Settling Time)
Figure 2.84: Settling Time is Critical in Multiplexed Applications
Most ADCs have settling times which are less than 1/fs max , even if not specified.
However sigma-delta ADCs have a built in digital filter which can take several output
sample intervals to settle. This should be kept in mind when using sigma-delta ADCs in
multiplexed applications.
The importance of settling time in multiplexed systems can be seen in Figure 2.85, where
the ADC input is modeled as a single-pole filter having a corresponding time constant,
τ = RC. The required number of time constants to settle to a given accuracy (1 LSB) is
shown. A simple example will illustrate the point.
Assume a multiplexed 16-bit data acquisition system uses an ADC with a sampling
frequency fs = 100 kSPS. The ADC must settle to 16-bit accuracy for a full-scale step
function input in less than 1/fs = 10 µs. The chart shows that 11.09 time constants are
required to settle to 16-bit accuracy. The input filter time constant must therefore be less
2.74
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
than τ = 10 µs/11.09 = 900 ns. The corresponding risetime tr = 2.2τ = 1.98 µs. The
required ADC full power input bandwidth can now be calculated from BW = 0.35/tr =
177 kHz. This neglects the settling time of the multiplexer and second-order settling time
effects in the ADC.
RESOLUTION,
# OF BITS
LSB (%FS)
# OF TIME
CONSTANTS
6
1.563
4.16
8
0.391
5.55
10
0.0977
6.93
12
0.0244
8.32
14
0.0061
9.70
16
0.00153
11.09
18
0.00038
12.48
20
0.000095
13.86
22
0.000024
15.25
Figure 2.85: Settling Time as a Function of Time Constant
for Various Resolutions
Overvoltage recovery time is defined as that amount of time required for an ADC to
achieve a specified accuracy, measured from the time the overvoltage signal re-enters the
converter's range, as shown in Figure 2.86. This specification is usually given for a signal
which is some stated percentage outside the ADC's input range. Needless to say, the
ADC should act as an ideal limiter for out-of-range signals and should produce either the
positive full-scale code or the negative full-scale code during the overvoltage condition.
Some converters provide over- and under-range flags to allow gain-adjustment circuits to
be activated. Care should always be taken to avoid overvoltage signals which will
damage an ADC input.
ADC SHOULD
READ FS CODE
OVERVOLTAGE
RECOVERY
TIME
Figure 2.86: Overvoltage Recovery Time
2.75
ANALOG-DIGITAL CONVERSION
ADC Sparkle Codes, Metastable States, and Bit Error Rate (BER)
A primary concern in the design of many digital communications systems using ADCs is
the bit error rate (BER). Unfortunately, ADCs contribute to the BER in ways that are not
predictable by simple analysis. This section describes the mechanisms within the ADCs
that can contribute to the error rate, ways to minimize the problem, and methods for
measuring the BER.
Random noise, regardless of the source, creates a finite probability of errors (deviations
from the expected output). Before describing the error code sources, however, it is
important to define what constitutes an ADC error code. Noise generated prior to, or
inside the ADC can be analyzed in the traditional manner. Therefore, an ADC error code
is any deviation from the expected output that is not attributable to the equivalent input
noise of the ADC. Figure 2.87 illustrates an exaggerated output of a low-amplitude
sinewave applied to an ADC that has error codes. Note that the noise of the ADC creates
some uncertainty in the output. These anomalies are not considered error codes, but are
simply the result of ordinary noise and quantization. The large errors are more significant
and are not expected. These errors are random and so infrequent that an SNR test of the
ADC will rarely detect them. These types of errors plagued a few of the early ADCs for
video applications, and were given the name sparkle codes because of their appearance
on a TV screen as small white dots or "sparkles" under certain test conditions. These
errors have also been called rabbits or flyers. In digital communications applications, this
type of error increases the overall system bit error rate (BER).
(SPARKLE CODES,
FLYERS, RABBITS)
LOW-AMPLITUDE
DIGITIZED SINEWAVE
Figure 2.87: Exaggerated Output of ADC Showing Error Codes
In order to understand the causes of the error codes, we will first consider the case of a
simple flash converter. The comparators in a flash converter are latched comparators
usually arranged in a master-slave configuration. If the input signal is in the center of the
threshold of a particular comparator, that comparator will balance, and its output will take
a longer period of time to reach a valid logic level after the application of the latch strobe
than the outputs of its neighboring comparators which are being overdriven. This
phenomenon is known as metastability and occurs when a balanced comparator cannot
reach a valid logic level in the time allowed for decoding. If simple binary decoding logic
is used to decode the thermometer code, a metastable comparator output may result in a
2.76
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
large output code error. Consider the case of a simple 3 bit flash converter shown in
Figure 2.88. Assume that the input signal is exactly at the threshold of Comparator 4 and
random noise is causing the comparator to toggle between a "1" and a "0" output each
time a latch strobe is applied. The corresponding binary output should be interpreted as
either 011 or 100. If, however, the comparator output is in a metastable state, the simple
binary decoding logic shown may produce binary codes 000, 011, 100, or 111. The codes
000 and 111 represent a one-half scale departure from the expected codes.
Figure 2.88: Metastable Comparator Output States
May Cause Error Codes in Data Converters
The probability of errors due to metastability increases as the sampling rate increases
because less time is available for a metastable comparator to settle.
Various measures have been taken in flash converter designs to minimize the metastable
state problem. Decoding schemes described in References 12 to 15 minimize the
magnitude of these errors. Optimizing comparator designs for regenerative gain and small
time constants is another way to reduce these problems.
Metastable state errors may also appear in successive approximation and subranging
ADCs which make use of comparators as building blocks. The same concepts apply,
although the magnitudes and locations of the errors may be different.
The test system shown in Figure 2.89 may be used to test for BER in an ADC. The
analog input to the ADC is provided by a high stability low noise sinewave generator.
The analog input level is set slightly greater than full-scale, and the frequency such that
there is always slightly less than 1 LSB change between samples as shown in Figure 2.90.
2.77
ANALOG-DIGITAL CONVERSION
E = Number of Errors in Interval T
BER =
E
2 T fs
Figure 2.89: ADC Bit Error Rate Test Setup
The test set uses series latches to acquire successive codes A and B. A logic circuit
determines the absolute difference between A and B. This difference is then compared to
the error limit, chosen to allow for expected random noise spikes and ADC quantization
errors. Errors which cause the difference to be larger than the limit will increment the
counters. The number of errors, E, are counted over a period of time, T. The error rate is
then calculated as BER = E/2Tfs. The factor of 2 in the denominator is required because
the hardware records a second error when the output returns to the correct code after
making the initial error. The error counter therefore is incremented twice for each error. It
should be noted that the same function can be accomplished in software if the ADC
outputs are stored in a memory and analyzed by a computer program.
The input frequency must be carefully chosen such that there is at least one sample taken
per code. Assume a full-scale input sinewave having an amplitude of 2N/2:
v( t ) =
2N
sin 2πft
2
Eq. 2.35
The maximum rate of change of this signal is
dv 
≤ 2 N πf .

dt  max
Eq. 2.36
Letting dv = 1 LSB, dt = 1/fs, and solving for the input frequency:
f in ≤
2.78
fs
2N π
.
Eq. 2.37
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
Choosing an input frequency less than this value will ensure that there is at least one
sample per code.
fin ≤
fs
2N π
1
fs
Figure 2.90: ADC Analog Signal for Low Frequency BER Test
The same test can be conducted at high frequencies by applying an input frequency
slightly offset from fs/2 as shown in Figure 2.91. This causes the ADC to slew full-scale
between conversions. Every other conversion is compared, and the "beat" frequency is
chosen such that there is slightly less than 1 LSB change between alternate samples. The
equation for calculating the proper frequency for the high frequency BER test is derived
as follows.
Assume an input full-scale sinewave of amplitude 2N/2 whose frequency is slightly less
than fs/2 by a frequency equal to ∆f.
v( t ) =
2N
 f

sin 2π s − ∆f  t  .
2

 2
Eq. 2.38
The maximum rate of change of this signal is
dv 
f

≤ 2 N π s − ∆f  .

dt  max
2

Eq. 2.39
Letting dv = 1 LSB and dt = 2/fs, and solving for the input frequency ∆f:
f 
1 
∆f ≤ s  1 −
.
2  2 ⋅ 2N π 
Eq. 2.40
2.79
ANALOG-DIGITAL CONVERSION
fin =
∆f ≤
fs
2
fs
2
– ∆f
1–
1
2·2Nπ
2
fs
Figure 2.91: ADC Analog Input for High Frequency BER Test
Establishing the BER of a well-behaved ADC is a difficult, time-consuming task; a single
unit can sometimes be tested for days without an error. For example, tests on a typical 8bit flash converter operating at a sampling rate of 75 MSPS yield a BER of
approximately 3.7×10–12 (1 error per hour) with an error limit of 4 LSBs. Meaningful
tests for longer periods of time require special attention to EMI/RFI effects (possibly
requiring a shielded screen room), isolated power supplies, isolation from soldering irons
with mechanical thermostats, isolation from other bench equipment, etc. Figure 2.92
shows the average time between errors as a function of BER for a sampling frequency of
75 MSPS. This illustrates the difficulty in measuring low BER because the long
measurement times increase the probability of power supply transients, noise, etc.
causing an error.
Figure 2.92: Average Time Between Errors Versus BER
when Sampling at 75 MSPS
2.80
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
DAC Dynamic Performance
The ac specifications which are most likely to be important with DACs are settling time,
glitch impulse area, distortion, and Spurious Free Dynamic Range (SFDR).
DAC Settling Time
The input to output settling time of a DAC is the time from a change of digital code (t =
0) to when the output comes within and remains within some error band as shown in
Figure 2.93. With amplifiers, it is hard to make comparisons of settling time, since their
specified error bands may differ from amplifier to amplifier, but with DACs the error
band will almost invariably be specified as ±1 or ±½ LSB. Note that in some cases, the
output settling time may be of more interest, in which case it is referenced to the time the
output first leaves the error band.
The input to output settling time of a DAC is made up of four different periods: the
switching time or dead time (during which the digital switching, but not the output, is
changing), the slewing time (during which the rate of change of output is limited by the
slew rate of the DAC output), the recovery time (when the DAC is recovering from its
fast slew and may overshoot), and the linear settling time (when the DAC output
approaches its final value in an exponential or near-exponential manner). If the slew time
is short compared to the other three (as is usually the case with current output DACs),
then the settling time will be largely independent of the output step size. On the other
hand, if the slew time is a significant part of the total, then the larger the step, the longer
the settling time.
ERROR BAND
SETTLING
TIME (OUTPUT)
t=0
ERROR BAND
DEAD
TIME
SLEW
TIME
RECOVERY
TIME
LINEAR
SETTLING
SETTLING TIME (INPUT TO OUTPUT)
Figure 2.93: DAC Settling Time
Settling time is especially important in video display applications. For example a
standard 1024×768 display updated at a 60-Hz refresh rate must have a pixel rate of
2.81
ANALOG-DIGITAL CONVERSION
1024×768×60 Hz = 47.2 MHz with no overhead. Allowing 35% overhead time increases
the pixel frequency to 64 MHz corresponding to a pixel duration of 1/(64×106) = 15.6 ns.
In order to accurately reproduce a single fully-white pixel located between two black
pixels, the DAC settling time should be less than the pixel duration time of 15.6 ns.
Higher resolution displays require even faster pixel rates. For example, a 2048×2048
display requires a pixel rate of approximately 330 MHz at a 60-Hz refresh rate.
Glitch Impulse Area
Ideally, when a DAC output changes it should move from one value to its new one
monotonically. In practice, the output is likely to overshoot, undershoot, or both (see
Figure 2.94). This uncontrolled movement of the DAC output during a transition is
known as a glitch. It can arise from two mechanisms: capacitive coupling of digital
transitions to the analog output, and the effects of some switches in the DAC operating
more quickly than others and producing temporary spurious outputs.
IDEAL TRANSITION
TRANSITION WITH
DOUBLET GLITCH
t
t
TRANSITION WITH
UNIPOLAR (SKEW)
GLITCH
t
Figure 2.94: DAC Transitions (Showing Glitch)
Capacitive coupling frequently produces roughly equal positive and negative spikes
(sometimes called a doublet glitch) which more or less cancel in the longer term. The
glitch produced by switch timing differences is generally unipolar, much larger and of
greater concern.
Glitches can be characterized by measuring the glitch impulse area, sometimes
inaccurately called glitch energy. The term glitch energy is a misnomer, since the unit for
glitch impulse area is volt-seconds (or more probably µV-sec or pV-sec. The peak glitch
area is the area of the largest of the positive or negative glitch areas. The glitch impulse
area is the net area under the voltage-versus-time curve and can be estimated by
approximating the waveforms by triangles, computing the areas, and subtracting the
negative area from the positive area as shown in Figure 2.95.
The mid-scale glitch produced by the transition between the codes 0111...111 and
1000...000 is usually the worst glitch because all switches are changing states. Glitches at
other code transition points (such as 1/4 and 3/4 full-scale) are generally less. Figure 2.96
shows the mid-scale glitch for a fast low-glitch DAC. The peak and net glitch areas are
estimated using triangles as described above. Settling time is measured from the time the
2.82
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
waveform leaves the initial 1 LSB error band until it enters and remains within the final
1-LSB error band. The step size between the transition regions is also 1 LSB.
+V1
A1
t
A2
–V2
t1
t2
PEAK GLITCH IMPULSE AREA = A1 ≈
NET GLITCH IMPULSE AREA
V1· t1
2
= A1 – A2 ≈
V1· t1
2
–
V2· t2
2
Figure 2.95: Calculating Net Glitch Impulse Area
SETTLING TIME
= 4.5ns
NET GLITCH AREA = 1.34 pV-s
PEAK GLITCH AREA = 1.36 pV-s
2 mV/DIVISION
1 LSB
1 LSB
1 LSB
5 ns/DIVISION
≈ 4.5ns
Figure 2.96: DAC Mid-scale Glitch Shows 1.34pV-s
Net Impulse Area and Settling Time of 4.5ns
DAC SFDR and SNR
DAC settling time is important in applications such as RGB raster scan video display
drivers, but frequency-domain specifications such as SFDR are generally more important
in communications.
2.83
ANALOG-DIGITAL CONVERSION
If we consider the spectrum of a waveform reconstructed by a DAC from digital data, we
find that in addition to the expected spectrum (which will contain one or more
frequencies, depending on the nature of the reconstructed waveform), there will also be
noise and distortion products. Distortion may be specified in terms of harmonic
distortion, Spurious Free Dynamic Range (SFDR), intermodulation distortion, or all of
the above. Harmonic distortion is defined as the ratio of harmonics to fundamental when
a (theoretically) pure sine wave is reconstructed, and is the most common specification.
Spurious free dynamic range is the ratio of the worst spur (usually, but not necessarily
always a harmonic of the fundamental) to the fundamental.
Code-dependent glitches will produce both out-of-band and in-band harmonics when the
DAC is reconstructing a digitally generated sinewave as in a Direct Digital Synthesis
(DDS) system. The mid-scale glitch occurs twice during a single cycle of a reconstructed
sinewave (at each mid-scale crossing), and will therefore produce a second harmonic of
the sinewave, as shown in Figure 2.97. Note that the higher order harmonics of the
sinewave, which alias back into the Nyquist bandwidth (dc to fs/2), cannot be filtered.
+ FULL SCALE
fO = 3MHz
MIDSCALE
fS = 10MSPS
– FULL SCALE
AMPLITUDE
CANNOT
BE FILTERED
fO
fS – 2fO
0
1
2
3
4
fS
fS – fO
2fO
5
fS
2
6
7
8
FREQUENCY (MHz)
9
10
Figure 2.97: Effect of Code-Dependent Glitches on Spectral Output
It is difficult to predict the harmonic distortion or SFDR from the glitch area specification
alone. Other factors, such as the overall linearity of the DAC, also contribute to distortion
as shown in Figure 2.98. In addition, certain ratios between the DAC output frequency
and the sampling clock cause the quantization noise to concentrate at harmonics of the
fundamental thereby increasing the distortion at these points.
It is therefore customary to test reconstruction DACs in the frequency domain (using a
spectrum analyzer) at various clock rates and output frequencies as shown in Figure 2.99.
Typical SFDR for the 16-bit AD9777 Transmit TxDAC is shown in Figure 2.100. The
clock rate is 160 MSPS, and the output frequency is swept to 50 MHz. As in the case of
2.84
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
ADCs, quantization noise will appear as increased harmonic distortion if the ratio
between the clock frequency and the DAC output frequency is an integer number. These
ratios should be avoided when making the SFDR measurements.
Resolution
Integral Non-Linearity
Differential Non-Linearity
Code-Dependent Glitches
Ratio of Clock Frequency to Output Frequency (Even in an Ideal
DAC)
Mathematical Analysis is Difficult !
Figure 2.98: Contributors to DDS DAC Distortion
PARALLEL OR
SERIAL PORT
DDS
SYSTEM
PC
N
N
DAC
LATCH
fo
STABLE
FREQUENCY
REFERENCE
fc
SPECTRUM
ANALYZER
Figure 2.99: Test Setup for Measuring DAC SFDR
There are nearly an infinite combination of possible clock and output frequencies for a
low distortion DAC, and SFDR is generally specified for a limited number of selected
combinations. For this reason, Analog Devices offers fast turnaround on customerspecified test vectors for the Transmit TxDAC family. A test vector is a combination of
amplitudes, output frequencies, and update rates specified directly by the customer for
SFDR data on a particular DAC.
2.85
ANALOG-DIGITAL CONVERSION
DAC OUTPUT FREQUENCY (MHz)
Figure 2.100: AD9777 16-bit TxDAC® SFDR, Data Update Rate = 160 MSPS
Measuring DAC SNR with an Analog Spectrum Analyzer
Analog spectrum analyzers are used to measure the distortion and SFDR of high
performance DACs. Care must be taken such that the front end of the analyzer is not
overdriven by the fundamental signal. If overdrive is a problem, a bandstop filter can be
used to filter out the fundamental signal such that the spurious components can be
observed.
Spectrum analyzers can also be used to measure the SNR of a DAC provided attention is
given to bandwidth considerations. SNR of an ADC is normally defined as the signal-tonoise ratio measured over the Nyquist bandwidth dc to fs/2. However, spectrum analyzers
have a resolution bandwidth which is less than fs/2—this therefore lowers the analyzer
noise floor by the process gain equal to 10 log10[fs/(2·BW)], where BW is the resolution
noise bandwidth of the analyzer (see Figure 2.101).
dB
NOISE FLOOR
BW
(NOISE BANDWIDTH)
SWEEP
fs
2
f
BW = ANALYZER RESOLUTION NOISE BANDWIDTH
SNR = NOISE FLOOR – 10 log10
fs/2
BW
Figure 2.101: Measuring DAC SNR with an Analog Spectrum Analyzer
2.86
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
It is important that the noise bandwidth (not the 3-dB bandwidth) be used in the
calculation, however from Figure 2.68 the error is small assuming that the analyzer
narrowband filter is at least two poles. The ratio of the noise bandwidth to the 3-dB
bandwidth of a one-pole Butterworth filter is 1.57 (causing an error of 1.96 dB in the
process gain calculation). For a two-pole Butterworth filter, the ratio is 1.11 (causing an
error of 0.45 dB in the process gain calculation).
DAC Output Spectrum and sin (x)/x Frequency Rolloff
The output of a reconstruction DAC can be represented as a series of rectangular pulses
whose width is equal to the reciprocal of the clock rate as shown in Figure 2.102. Note
that the reconstructed signal amplitude is down 3.92 dB at the Nyquist frequency, fc/2.
An inverse sin(x)/x filter can be used to compensate for this effect in most cases. The
images of the fundamental signal occur as a result of the sampling function and are also
attenuated by the sin(x)/x function.
SAMPLED
SIGNAL
t
RECONSTRUCTED
SIGNAL
t
1
–3.92dB
A
1
fc
sin
A=
IMAGES
IMAGES
πf
fc
πf
fc
IMAGES
f
0
0.5fc
fc
1.5fc
2fc
2.5fc
3fc
Figure 2.102: DAC sin x/x Roll Off (Amplitude Normalized)
Oversampling Interpolating DACs
In ADC-based systems, oversampling can ease the requirements on the antialiasing filter.
In a DAC-based system (such as DDS), the concept of interpolation can be used in a
similar manner. This concept is common in digital audio CD players, where the basic
update rate of the data from the CD is 44.1 kSPS. Early CD players used traditional
binary DACs and inserted "Zeros" into the parallel data, thereby increasing the effective
update rate to 4-times, 8-times, or 16-times the fundamental throughput rate . The 4×, 8×,
or 16× data stream is passed through a digital interpolation filter which generates the
extra data points. The high oversampling rate moves the image frequencies higher,
thereby allowing a less complex filter with a wider transition band. The sigma-delta 1-bit
2.87
ANALOG-DIGITAL CONVERSION
DAC architecture uses a much higher oversampling rate and represents the ultimate
extension of this concept and has become popular in modern CD players.
The same concept of oversampling can be applied high speed DACs used in
communications applications, relaxing the requirements on the output filter as well as
increasing the SNR due to process gain.
Assume a traditional DAC is driven at an input word rate of 30 MSPS (see Figure
2.103A). Assume the DAC output frequency is 10 MHz. The image frequency
component at 30 – 10 = 20 MHz must be attenuated by the analog antialiasing filter, and
the transition band of the filter is 10 to 20 MHz. Assume that the image frequency must
be attenuated by 60 dB. The filter must therefore go from a passband of 10 MHz to 60 dB
stopband attenuation over the transition band lying between 10 and 20 MHz (one octave).
A filter gives 6-dB attenuation per octave for each pole. Therefore, a minimum of 10
poles is required to provide the desired attenuation. Filters become even more complex as
the transition band becomes narrower.
A
ANALOG LPF
fCLOCK = 30MSPS
dB
fo
IMAGE
IMAGE
10
20
30
40
IMAGE
50
IMAGE
60
70
80
FREQUENCY (MHz)
B
fCLOCK = 60MSPS
dB
fo
ANALOG
LPF
IMAGE
IMAGE
10
20
30
40
50
60
70
80
Figure 2.103: Analog Filter Requirements for fo = 10 MHz:
(A) fc = 30 MSPS, and (B) fc = 60 MSPS
Assume that we increase the DAC update rate to 60 MSPS and insert a "zero" between
each original data sample. The parallel data stream is now 60 MSPS, but we must now
determine the value of the zero-value data points. This is done by passing the 60-MSPS
data stream with the added zeros through a digital interpolation filter which computes the
additional data points. The response of the digital filter relative to the 2-times
oversampling frequency is shown in Figure 2.103B. The analog antialiasing filter
transition zone is now 10 to 50 MHz (the first image occurs at 2fc – fo = 60 – 10 =
50 MHz). This transition zone is a little greater than 2 octaves, implying that a 5- or 6pole filter is sufficient.
2.88
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
The AD9773/AD9775/AD9777 (12/14/16-bit) series of Transmit DACs (TxDAC) are
selectable 2×, 4×, or 8× oversampling interpolating dual DACs, and a simplified block
diagram is shown in Figure 2.103. These devices are designed to handle 12/14/16-bit
input word rates up to 160 MSPS. The output word rate is 400 MSPS maximum. For an
output frequency of 50 MHz, an input update rate of 160 MHz, and an oversampling ratio
of 2×, the image frequency occurs at 320 MHz – 50 MHz = 270 MHz. The transition
band for the analog filter is therefore 50 MHz to 270 MHz. Without 2× oversampling, the
image frequency occurs at 160 MHz – 50 MHz = 110 MHz, and the filter transition band
is 50 MHz to 110 MHz.
N
fc
LATCH
N
PLL
DIGITAL
N
INTERPOLATION
FILTER
K•fc
N
LATCH
DAC
LPF
TYPICAL APPLICATION: fc = 160MSPS
fo = 50MHz
K=2
Image Frequency = 320 – 50 = 270MHz
fo
Figure 2.104: Oversampling Interpolating TxDAC® Simplified Block Diagram
Notice also that an oversampling interpolating DAC allows both a lower frequency input
clock and input data rate, which are much less likely to generate noise within the system.
2.89
ANALOG-DIGITAL CONVERSION
REFERENCES:
2.3 DATA CONVERTER AC ERRORS
1.
W. R. Bennett, "Spectra of Quantized Signals," Bell System Technical Journal, Vol. 27, July 1948,
pp. 446-471.
2.
B. M. Oliver, J. R. Pierce, and C. E. Shannon, "The Philosophy of PCM," Proceedings IRE, Vol. 36,
November 1948, pp. 1324-1331.
3.
W. R. Bennett, "Noise in PCM Systems," Bell Labs Record, Vol. 26, December 1948, pp. 495-499.
4.
H. S. Black and J. O. Edson, "Pulse Code Modulation," AIEE Transactions, Vol. 66, 1947, pp. 895899.
5.
H. S. Black, "Pulse Code Modulation," Bell Labs Record, Vol. 25, July 1947, pp. 265-269.
6.
Steve Ruscak and Larry Singer, Using Histogram Techniques to Measure A/D Converter Noise,
Analog Dialogue, Vol. 29-2, 1995.
7.
M.J. Tant, The White Noise Book, Marconi Instruments, July 1974.
8.
G.A. Gray and G.W. Zeoli, Quantization and Saturation Noise due to A/D Conversion, IEEE Trans.
Aerospace and Electronic Systems, Jan. 1971, pp. 222-223.
9.
Kevin McClaning and Tom Vito, Radio Receiver Design, Noble Publishing, 2000, ISBN 1-88-493207-X.
10. Walter G. Jung, editor, Op Amp Applications, Analog Devices, Inc., 2002, ISBN 0-916550-26-5, pp.
6.144-6.152.
11. Brad Brannon, Aperture Uncertainty and ADC System Performance, Application Note AN-501,
Analog Devices, Inc., January 1998. (available for download at http://www.analog.com)
12. Christopher W. Mangelsdorf, A 400-MHz Input Flash Converter with Error Correction, IEEE
Journal of Solid-State Circuits, Vol. 25, No. 1, February 1990, pp. 184-191.
13. Charles E. Woodward, A Monolithic Voltage-Comparator Array for A/D Converters, IEEE Journal of
Solid State Circuits, Vol. SC-10, No. 6, December 1975, pp. 392-399.
14. Yukio Akazawa et. al., A 400MSPS 8 Bit Flash A/D Converter, 1987 ISSCC Digest of Technical
Papers, pp. 98-99.
15. A.. Matsuzawa et al., An 8b 600MHz Flash A/D Converter with Multi-stage Duplex-gray Coding,
Symposium VLSI Circuits, Digest of Technical Papers, May 1991, pp. 113-114.
16. Ron Waltman and David Duff, Reducing Error Rates in Systems Using ADCs, Electronics Engineer,
April 1993, pp. 98-104.
17. K. W. Cattermole, Principles of Pulse Code Modulation, American Elsevier Publishing Company,
Inc., 1969, New York NY, ISBN 444-19747-8. (An excellent tutorial and historical discussion of data
conversion theory and practice, oriented towards PCM, but covers practically all aspects. This one is
a must for anyone serious about data conversion!
18. Robert A. Witte, Distortion Measurements Using a Spectrum Analyzer, RF Design, September, 1992,
pp. 75-84.
19. Walt Kester, Confused About Amplifier Distortion Specs?, Analog Dialogue, 27-1, 1993, pp. 27-29.
2.90
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.3 DATA CONVERTER AC ERRORS
20. Dan Sheingold, Editor, Analog-to-Digital Conversion Handbook, Third Edition, Prentice-Hall,
1986.
21. Fred H. Irons, "The Noise Power Ratio—Theory and ADC Testing," IEEE Transactions on
Instrumentation and Measurement, Vol. 49, No. 3, June 2000, pp. 659-665.
2.91
ANALOG-DIGITAL CONVERSION
NOTES:
2.92
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.4 GENERAL DATA CONVERTER SPECIFICATIONS
SECTION 2.4: GENERAL DATA CONVERTER
SPECIFICATIONS
James Bryant
Overall Considerations
Data converters, as we have observed, have a digital port and an analog port, and like all
integrated circuits they require power supplies and will draw current from those supplies.
Data converter specifications will therefore include the usual specifications common to
any integrated circuit, including supply voltage and supply current, logic interfaces,
power on and standby timing, package and thermal issues and ESD. We shall not
consider these at any length, but there are some issues which may require a little
consideration.
An over-riding piece of advice here is read the data sheet. There is no excuse for being
unaware of the specifications of a device for which one owns a data sheet—and it is often
possible to deduce extra information which is not printed on it by understanding the
issues and conventions involved in preparing it.
Traditional precision analog integrated circuits (which include amplifiers, converters and
other devices) were designed for operation from supplies of ±15 V, and many (but not
all—it is important to check with the data sheet) would operate within specification over
quite a wide range of supply voltages. Today the processes used for many, but by no
means all, modern converters have low breakdown voltages and absolute maximum
ratings of only a few volts. Converters built with these processes may only work to
specification over a narrow range of supply voltages.
It is therefore important when selecting a data converter to check both the absolute
maximum supply voltage(s) and the range of voltages where correct operation can be
expected. Some low-voltage devices work equally well with both 5-V and 3.3-V supplies,
others are sold in 5-V and 3.3-V versions with different suffixes on their part numbers—
with these it is important to use the correct one.
Absolute maximum ratings are ratings which can never be exceeded without grave risk of
damage to the device concerned—they are not safe operating limits. But they are
conservative—integrated circuit manufacturers try to set absolute maximum ratings so
that every device that they manufacture will survive brief exposures to absolute
maximum conditions. As a result many devices will, in fact, appear to operate safely and
continuously outside the permitted limits. Good engineers do not take advantage of this
for three reasons: (1) components are not tested outside their absolute maximum limits
so, although they may be operating, they may not be operating at their specified accuracy.
Also the damage done by incorrect operation may not be immediately fatal, but may
cause low levels of disruption which, in turn, may (2) shorten the device's life, or (3) may
affect its subsequent accuracy even when it is operated within specification again. None
of these effects is at all desirable and absolute maximum ratings should always be
respected.
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ANALOG-DIGITAL CONVERSION
The supply current in a data converter specification is usually the no-load current – i.e.
the current consumption when the data converter output is driving a high impedance or
open-circuit load. CMOS logic, and to a lesser extent some other types, have current
consumption which is proportional to clock speed so a CMOS data converter current may
be defined at a specific clock frequency and will be higher if the clock runs faster.
Current consumption will also be higher when the output (or the reference output if there
is one, or both) is loaded. There may be another figure for "standby" current—the current
which flows when the data converter is connected to a power supply but is internally shut
down into a non-operational low power state to conserve current.
When power is first applied to some data converters they may take several tens, hundreds
or even thousands of microseconds for their reference and amplifier circuitry to stabilize
and, although this is less common, some may even take a long time to "wake up" from a
power saving standby mode. It is therefore important to ensure that data converters which
have such delays are not used in applications where full functionality is required within a
short time of power-up or wake-up.
All integrated circuits are vulnerable to electrostatic discharge (ESD), but precision
analog circuits are, on the whole, more vulnerable than some other types. This is because
the technologies available for minimizing such damage also tend to degrade the
performance of precision circuitry, and there is a necessary compromise between
robustness and performance. It is always a good idea to ensure that when you handle
amplifiers, converters and other vulnerable circuits you take the necessary steps to avoid
ESD.
Specifications of packages, operating temperature ranges and similar issues, although
important, do not need further discussion here.
Logic Interface Issues
As it is important to read and understand power supply specifications so it is equally
important to read and understand logic specifications. In the past most integrated circuit
logic circuitry (with the exception of emitter-coupled logic or ECL) operated from 5-V
supplies and had compatible logic levels—with a few exceptions 5-V logic would
interface with other 5-V logic. Today, with the advent of low voltage logic operating with
supplies of 3.3 V, 2.7 V, or even less, it is important to ensure that logic interfaces are
compatible. There are several issues which must be considered—absolute maximum
ratings, worst case logic levels, and timing. The logic inputs of integrated circuits
generally have absolute maximum ratings, as do most other inputs, of 300 mV outside the
power supply. Note that these are instantaneous ratings. If an IC has such a rating and is
currently operating from a +5-V supply then the logic inputs may be between –0.3 V and
+5.3 V—but if the supply is not present then that input must be between +0.3 V and
–0.3 V, not the –0.3 V to +5.3 V which are the limits once the power is applied—ICs
cannot predict the future.
The reason for the rating of 0.3 V is to ensure that no parasitic diode on the IC is ever
turned on by a voltage outside the IC’s absolute maximum rating. It is quite common to
protect an input from such over-voltage with a Schottky diode clamp. At low
temperatures the clamp voltage of a Schottky diode may be a little more than 0.3 V, and
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FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.4 GENERAL DATA CONVERTER SPECIFICATIONS
so the IC may see voltages just outside its absolute maximum rating. Although, strictly
speaking, this subjects the IC to stresses outside its absolute maximum ratings and so is
forbidden, this is an acceptable exception to the general rule provided the Schottky diode
is at a similar temperature to the IC that it is protecting (say within ±10°C).
Some low voltage devices, however, have inputs with absolute maximum ratings which
are substantially greater than their supply voltage. This allows such circuits to be driven
by higher voltage logic without additional interface or clamp circuitry. But it is important
to read the data sheets and ensure that both logic levels and absolute maximum voltages
are compatible for all combinations of high and low supplies.
This is the general rule when interfacing different low-voltage logic circuitry—it is
always necessary to check that at the lowest value of its power supply (a) the logic 1
output from the driving circuit applied to its worst-case load is greater than the specified
minimum logic 1 input for the receiving circuit, and (b) with its output sinking maximum
allowed current, the logic 0 output is less than the specified logic 0 input of the receiver.
If the logic specifications of your chosen devices do not meet these criteria it will be
necessary to select different devices, use different power supplies, or use additional
interface circuitry to ensure that the required levels are available. Note that additional
interface circuitry introduces extra delays in timing.
It is not sufficient to build an experimental set-up and test it. In general logic thresholds
are generously specified and usually logic circuits will work correctly well outside their
specified limits—but it is not possible to rely on this in a production design. At some
point a batch of devices near the limit on low output swing will be required to drive some
devices needing slightly more drive than usual—and will be unable to do so.
Data Converter Logic: Timing and other Issues
It is not the purpose of this brief section to discuss logic architectures, so we shall not
define the many different data converter logic interface operations and their timing
specifications except to note that data converter logic interfaces may be more complex
than you expect—read the data sheet—and do not expect that because there is a pin with
the same name on memory and interface chips it will behave in exactly the same way in a
data converter. Also some data converters reset to a known state on power-up but many
more do not.
But it is very necessary to consider general timing issues. The new low voltage processes
which are used for many modern data converters have a number of desirable features.
One which is often overlooked by users (but not by converter designers!) is their higher
logic speed. DACs built on older processes frequently had logic that was orders of
magnitude slower than the microprocessors that they interfaced with, and it was
sometimes necessary to use separate buffers, or multiple WAIT instructions, to make the
two compatible. Today it is much more common for the write times of DACs to be
compatible with those of the fast logic with which they interface.
Nevertheless not all DACs are speed compatible with all logic interfaces, and it is still
important to ensure that minimum data set-up times and write pulse widths are observed.
Again, experiments will often show that devices work with faster signals than their
2.95
ANALOG-DIGITAL CONVERSION
specification requires—but at the limits of temperature or supply voltage some may not,
and interfaces should be designed on the basis of specified rather than measured timing.
2.96
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
SECTION 2.5: DEFINING THE SPECIFICATIONS
Dan Sheingold, Walt Kester
The following list, in alphabetical order, should prove helpful regarding specifications
and their definitions. Some of the most popular ones are discussed in other places in the
text as well as here. Many of the application-specific specifications are defined where
they are mentioned in the text and are not repeated here. The original source for these
definitions was provided by Dan Sheingold from Chapter 11 in his classic book—
Analog-to-Digital Conversion Handbook, Third Edition, Prentice-Hall, 1986.
Accuracy, Absolute. Absolute accuracy error of a DAC is the difference between actual
analog output and the output that is expected when a given digital code is applied to the
converter. Error is usually commensurate with resolution, i.e., less 1/2 LSB of full-scale,
for example. However, accuracy may be much better than resolution in some
applications; for example, a 4-bit DAC having only 16 discrete digitally chosen levels
would have a resolution of 1/16, but it might have an accuracy to within 0.01 % of each
ideal value.
Absolute accuracy error of an ADC at a given output code is the difference between the
actual and the theoretical analog input voltages required to produce that code. Since the
code can be produced by any analog voltage in a finite band (see Quantizing
Uncertainty), the "input required to produce that code" is usually defined as the midpoint
of the band of inputs that will produce that code. For example, if 5 volts, ±1.2 mV, will
theoretically produce a 12-bit half-scale code of 1000 0000 0000, then a converter for
which any voltage from 4.997 V to 4.999 V will produce that code will have absolute
error of (1/2)(4.997 + 4.999) – 5 volts = +2 mV.
Sources of error include gain (calibration) error, zero error, linearity errors, and noise.
Absolute accuracy measurements should be made under a set of standard conditions with
sources and meters traceable to an internationally accepted standard.
Accuracy, Logarithmic DACs. The difference (measured in dB) between the actual
transfer function and the ideal transfer function, as measured after calibration of gain
error at 0 dB.
Accuracy, Relative. Relative accuracy error, expressed in %, ppm, or fractions of 1 LSB,
is the deviation of the analog value at any code (relative to the full analog range of the
device transfer characteristic) from its theoretical value (relative to the same range), after
the full-scale range (FSR) has been calibrated (see Full-Scale Range).
Since the discrete analog values that correspond to the digital values ideally lie on a
straight line, the specified worst case relative accuracy error of a linear
ADC or DAC can be interpreted as a measure of end-point nonlinearity (see Linearity).
The "discrete points" of a DAC transfer characteristic are measured by the actual analog
outputs. The "discrete points" of an ADC transfer characteristic are the midpoints of the
quantization bands at each code (see Accuracy, Absolute).
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ANALOG-DIGITAL CONVERSION
Acquisition Time. The acquisition time of a track-and-hold circuit for a step change is the
time required by the output to reach its final value, within a specified error band, after the
track command has been given. Included are switch delay time, the slewing interval, and
settling time for a specified output voltage change.
Adjacent Channel Power Ratio (ACPR). The ratio in dBc between the measured power
within a channel relative to its adjacent channel. See Adjacent Channel Leakage Ratio
(ACLR).
Adjacent Channel Leakage Ratio (ACLR). The ratio in dBc between the measured power
within the carrier bandwidth relative to the noise level in an adjacent empty carrier
channel. Both ACPR and ACLR are Wideband-CDMA (WCDMA) specifications. The
channel bandwidth for WCDMA is approximately 3.84 MHz with 5-MHz spacing
between channels.
Aliasing. A signal within a bandwidth fa must be sampled at a rate fs > 2fa in order to
avoid the loss of information. If fs < 2fa, a phenomenon called aliasing, inherent in the
spectrum of the sampled signal, will cause a frequency equal to fs – fa, called an alias, to
appear in the Nyquist bandwidth, dc to fs/2. For example, if fs = 4 kSPS and fa = 3 kHz, a
1-kHz alias will appear. Note also that for fa = 1 kHz (within the dc to fs/2 bandwidth), an
alias will occur at 3 kHz (outside the dc to fs/2 bandwidth). Since noise is also aliased, it
is essential to provide low pass (or band pass) filtering prior to the sampling stage to
prevent out-of-band noise on the input signal from being aliased into the signal range and
thereby degrading the SNR.
Analog Bandwidth. For an ADC, the analog input frequency at which the spectral power
of the fundamental frequency (as determined by the FFT analysis) is reduced by 3 dB.
This can be specified as full-power bandwidth, or small signal bandwidth. See also
(Bandwidth, Full Linear and Bandwidth, Full Power).
Analog Bandwidth, 0.1dB. For an ADC, the analog input frequency at which the spectral
power of the fundamental frequency (as determined by the FFT analysis) is reduced by
0.1 dB. This is a popular video specification. See also (Bandwidth, Full Linear and
Bandwidth, Full Power).
Aperture Time. (classic definition) Aperture time in a sample-and-hold is defined as the
time required for the internal switch to switch from the closed position (zero resistance)
to the fully open position (infinite resistance). A first order analysis which neglects nonlinear effects assumes that the input signal is averaged over this time interval to produce
the final output signal. The analysis shows that this does not introduce an error as long as
the switch opens in a repeatable fashion, and as long as the aperture time is reasonably
short with respect to the hold time. There exists an effective sampling point in time which
will cause an ideal sample-and-hold to produce the same held voltage. The difference
between this effective sampling point and the actual sampling point is defined as
effective aperture delay time.
Aperture Delay Time, or Effective Aperture Delay Time. In a sample-and-hold or trackand-hold, there exists an effective sampling point in time which will cause an ideal
2.98
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
sample-and-hold to produce the same held voltage. The difference between this effective
sampling point and the actual sampling point is defined as the aperture delay time or
effective aperture delay time. In a sampling ADC, aperture delay time can be measured
by sampling the zero-crossing of a sinewave with a sampling clock locked to the
sinewave. The phase of the sampling clock sampling clock is adjusted until the output of
the ADC is 100…00. The time difference between the leading edge of the sampling clock
and the zero-crossing of the sinewave—referenced to the analog input—is the effective
aperture delay time. A dual trace oscilloscope can be used to make the measurement.
Aperture Uncertainty (or Aperture Jitter) is the sample-to-sample variation in the
sampling point because of jitter. Aperture jitter is expressed as an rms quantity and
produces a corresponding rms voltage error in the sample-and-hold output. In an ADC it
is caused by internal noise and jitter in the sampling clock path from the sampling clock
input pin to the internal switch. Jitter in the external sampling clock produces the same
type of error.
Automatic Zero. To achieve zero stability in many integrating-type converters, a time
interval is provided during each conversion cycle to allow the circuitry to compensate for
drift errors. The drift error in such converters is substantially zero. A similar function
exists in many high-resolution sigma-delta ADCs.
Bandwidth, Full-Linear. The full-linear bandwidth of an ADC is the input frequency at
which the slew-rate limit of the sample-and-hold amplifier is reached. Up to this point,
the amplitude of the reconstructed fundamental signal will have been attenuated by less
than 0.1 dB. Beyond this frequency, distortion of the sampled input signal increases
significantly.
Bandwidth, Full-Power (FPBW). The full-power bandwidth is that input frequency at
which the amplitude of the reconstructed fundamental signal (measured using FFTs) is
reduced by 3 dB for a full-scale input. In order to be meaningful, the FPBW must be
examined in conjunction with the signal-to-noise ratio (SNR), signal-to-noise-plusdistortion ratio (SINAD), effective number of bits (ENOB), and harmonic distortion in
order to ascertain the true dynamic performance of the ADC at the FPBW frequency.
Bandwidth, Analog Input Small-Signal. Analog input bandwidth is measured similarly to
FPBW at a reduced analog input amplitude. This specification is similar to the smallsignal bandwidth of an op amp. The amplitude of the input signal at which the smallsignal bandwidth is measured should be specified on the data sheet.
Bandwidth, Effective Resolution (ERB). Some ADC manufacturers define the frequency
at which SINAD drops 3 dB as the effective resolution bandwidth (ERB). This is the same
frequency at which the ENOB drops ½ bit. This specification is a misnomer, however,
since bandwidth normally is associated with signal amplitude.
Bias Current is the zero-signal dc current required from the signal source by the inputs of
many semiconductor circuits. The voltage developed across the source resistance by bias
current constitutes an (often negligible) offset error. When an instrumentation amplifier
performs measurements of a source that is remote from the amplifier's power-supply,
there must be a return path for bias currents. If it does not already exist and is not
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ANALOG-DIGITAL CONVERSION
provided, those currents will charge stray capacitances, causing the output to drift
uncontrollably or to saturate. Therefore, when amplifying outputs of "floating" sources,
such as transformers, insulated thermocouples, and ac-coupled circuits, there must be a
high-impedance dc leakage path from each input to common, or to the driven-guard
terminal (if present). If a dc return path is impracticable, an isolator must be used.
Bipolar Mode. (See Offset).
Bipolar Offset. ( See Offset).
Bus. A bus is a parallel path of binary information signals—usually 4, 8, 16, 32, or 64bits wide. Three common types of information usually found on buses are data,
addresses, and control signals. three-state output switches (inactive, high, and low) permit
many sources—such as ADCs—to be connected to a bus, while only one is active at any
time.
Byte. A byte is a binary digital word, usually 8-bits wide. A byte is often part of a longer
word that must be placed on an 8-bit bus in two stages. The byte containing the MSB is
called the high byte; that containing the LSB is called the low byte. A 4-bit byte is called
a nibble on an 8-bit or greater bus.
Channel-to-Channel Isolation. In multiple DACs, the proportion of analog input signal
from one DAC's reference input that appears at the output of the other DAC, expressed
logarithmically in dB. See also crosstalk.
Charge Transfer, Charge Injection (or Offset Step), the principal component of sampleto-hold offset (or pedestal), is the small charge transferred to the storage capacitor via
interelectrode capacitance of the switch and stray capacitance when switching to the hold
mode. The offset step is directly proportional to this charge:
Offset error = Incremental Charge/Capacitance = ∆Q/C.
It can be reduced somewhat by lightly coupling an appropriate polarity version of the
hold signal to the capacitor for first-order cancellation. The error can also be reduced by
increasing the capacitance, but this increases acquisition time.
Code Width. This is a fundamental quantity for ADC specifications. In an ADC where the
code transition noise is a fraction of an LSB, it is defined as the range of analog input
values for which a given digital output code will occur. The nominal value of a code
width (for all but the first and last codes) is the voltage equivalent of 1 least significant
bit (LSB) of the full-scale range, or 2.44 mV out of 10 volts for a 12-bit ADC. Because
the full-scale range is fixed, the presence of excessively wide codes implies the existence
of narrow and perhaps even missing codes. Code transition noise can make the
measurement of code width difficult or impossible. In wide bandwidth and high
resolution ADCs additional noise modulates the effective code width and appears as
input-referred noise. Many ADCs have input-referred noise which spans several code
widths, and histogram techniques must be used to accurately measure differential
linearity.
2.100
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
Common-Mode Range. Common-mode rejection usually varies with the magnitude of the
range through which the input signal can swing, determined by the sum of the commonmode and the differential voltage. Common-mode range is that range of total input
voltage over which specified common-mode rejection is maintained. For example, if the
common-mode signal is ±5 V and the differential signal is ±5 V, the common-mode
range is ±10 V.
Common-Mode Rejection (CMR) is a measure of the change in output voltage when both
inputs are changed by equal amounts of ac and/or dc voltage. Common-mode rejection is
usually expressed either as a ratio (e.g., CMRR = 1,000,000:1) or in decibels: CMR =
20log10CMRR; if CMRR = 106, CMR = 120 dB. A CMRR of 106 means that 1 volt of
common mode is processed by the device as though it were a differential signal of
1-µV at the input.
CMR is usually specified for a full-range common-mode voltage change (CMV), at a
given frequency, and a specified imbalance of source impedance (e.g., 1 kΩ source
unbalance, at 60 Hz). In amplifiers, the common-mode rejection ratio is defined as the
ratio of the signal gain, G, to the common-mode gain (the ratio of common-mode signal
appearing at the output to the CMV at the input.
Common-Mode Voltage (CMV). A voltage that appears in common at both input
terminals of a differential-input device, with respect to its output reference (usually
"ground"). For inputs, V1 and V2, with respect to ground, CMV = ½(V1 + V2). An ideal
differential-input device would ignore CMV. Common-mode error (CME) is any error at
the output due to the common-mode input voltage. The errors due to supply-voltage
variation, an internal common-mode effect, are specified separately.
Compliance-Voltage Range. For a current-output DAC, the maximum range of (output)
terminal voltage for which the device will maintain the specified current-output
characteristics.
Conversion Complete. An ADC digital output signal which indicates the end of
conversion. When this signal is in the opposite state, the ADC is considered to be "busy."
Also called end-of-conversion (EOC), data ready, or status in some converters.
Conversion Time and Conversion Rate. For an ADC without a sample-and-hold, the time
required for a complete measurement is called conversion time. For most converters
(assuming no significant additional systemic delays), conversion time is essentially
identical with the inverse of conversion rate. For simple sampling ADCs, however, the
conversion rate is the inverse of the conversion time plus the sample-and-hold's
acquisition time. However, in many high-speed converters, because of pipelining, new
conversions are initiated before the results of prior conversions have been determined;
thus, there can one, two, three, or more clock cycles of conversion delay (plus a fixed
delay in some cases). Once a train of conversions has been initiated, as in signalprocessing applications, the conversion rate can therefore be much faster than the
conversion time would imply.
Crosstalk. Leakage of signals, usually via capacitance between circuits or channels of a
multi-channel system or device, such as a multiplexer, multiple input ADC, or multiple
2.101
ANALOG-DIGITAL CONVERSION
DAC. Crosstalk is usually determined by the impedance parameters of the physical
circuit, and actual values are frequency-dependent. See also channel-to-channel isolation.
Multiple DACs have a digital crosstalk specification: the spike (sometimes called a
glitch) impulse appearing at the output of one converter due to a change in the digital
input code of another of the converters. It is specified in nanovolt- or picovolt-seconds
and measured at VREF = 0 V.
Data Ready. (See Conversion Complete).
Deglitcher (See Glitch). A device that removes or reduces the effects of time-skew in
D/A conversion. A deglitcher normally employs a track-and-hold circuit, often
specifically designed as part of the DAC. When the DAC is updated, the deglitcher holds
the output of the DAC's output amplifier constant at the previous value until the switches
reach equilibrium, then acquires and tracks the new value.
DAC Glitch. A glitch is a switching transient appearing in the output during a code
transition. The worst-case DAC glitch generally occurs when the DAC is switched
between the 011…111 and 100…000 codes. The net area under the glitch is referred to as
glitch impulse area and is measured in millivolt-nanoseconds, nanovolt-seconds, or
picovolt-seconds. Sometimes the term glitch energy is used to describe the net area under
the glitch—this terminology is incorrect because the unit of measurement is not energy.
Differential Analog Input Resistance, Differential Analog Input Capacitance, and
Differential Analog Input Impedance. The real and complex impedances measured at
each analog input port of an ADC. The resistance is measured statically and the
capacitance and differential input impedances are measured with a network analyzer.
Differential Analog Input Voltage Range. The peak-to-peak differential voltage that must
be applied to the converter to generate a full-scale response. Peak differential voltage is
computed by observing the voltage on a single pin and subtracting the voltage from the
other pin, which is 180 degrees out of phase. Peak-to-peak differential is computed by
rotating the inputs phase 180 degrees and taking the peak measurement again. Then the
difference is computed between both peak measurements.
Differential Gain (∆G). A video specification which measures the variation in the
amplitude (in percent) of a small amplitude color subcarrier signal as it is swept across
the video range from black to white.
Differential Phase (∆ϕ). A video specification which measures the phase variation (in
degrees) of a small amplitude color subcarrier signal as it is swept across the video range
from black to white.
Droop Rate. When a sample-and-hold circuit using a capacitor for storage is in hold, it
will not hold the information forever. Droop rate is the rate at which the output voltage
changes (by increasing or decreasing), and hence gives up information. The change of
output occurs as a result of leakage or bias currents flowing through the storage capacitor.
The polarity of change depends on the sources of leakage within a given device. In
integrated circuits with external capacitors, it is usually specified as a (droop or drift)
2.102
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
current, in ICs having internal capacitors, a rate of change. Note: dv/dt (volts/second) =
I/C (picoamperes/picofarads).
Dual-Slope Converter. An integrating ADC in which the unknown signal is converted to
a proportional time interval, which is then measured digitally. This is done by integrating
the unknown for a predetermined length of time. Then a reference input is switched to the
integrator, which integrates "down" from the level determined by the unknown until the
starting level is reached. The time for the second integration process, as determined by
the counter, is proportional to the average of the unknown signal level over the
predetermined integrating period. The counter provides the digital readout.
Effective Input Noise. (See Input-Referred Noise).
Effective Number of Bits (ENOB). With a sinewave input, Signal-to-Noise-and-Distortion
(SINAD) can be expressed in terms of the number of bits. Rewriting the theoretical SNR
formula for an ideal N-bit ADC and solving for N:
N = (SNR – 1.76 dB)/6.02
The actual ADC SINAD is measured using FFT techniques, and ENOB is calculated
from:
ENOB = (SINAD – 1.76 dB)/6.02
Effective Resolution. (See Noise-Free Code Resolution)
Encode Command. (See Encode, Sampling Clock).
Encode (Sampling Clock) Pulsewidth/Duty Cycle. Pulsewidth high is the minimum
amount of time that the ENCODE pulse should be left in Logic 1 state to achieve rated
performance; pulsewidth low is the minimum time ENCODE or pulse should be left in
low state. See timing implications of changing the width in the text of high speed ADC
data sheets. At a given clock rate, these specs define an acceptable ENCODE duty cycle.
Feedthrough. Undesirable signal-coupling around switches or other devices that are
supposed to be turned off or provide isolation, e.g., feedthrough error in a sample-andhold, multiplexer, or multiplying DAC. Feedthrough is variously specified in percent, dB,
parts per million, fractions of 1 LSB, or fractions of 1 volt, with a given set of inputs, at a
specified frequency.
In a multiplying DAC, feedthrough error is caused by capacitive coupling from an ac
VREF to the output, with all switches off. In a sample-and-hold, feedthrough is the
fraction of the input signal variation or ac input waveform that appears at the output in
hold. It is caused by stray capacitive coupling from the input to the storage capacitor,
principally across the open switch.
Flash Converter. A converter in which all the bit choices are made at the same time. It
requires 2N – 1 voltage-divider taps and comparators and a comparable amount of priority
encoding logic. A scheme that gives extremely fast conversion, it requires large numbers
2.103
ANALOG-DIGITAL CONVERSION
of nearly identical components, hence it is well suited to integrated-circuit form for
resolutions up to 8 bits. Several flash converters are often used in multistage sub-ranging
converters, to provide high resolution at somewhat slower speed than pure flash
conversion.
Four-Quadrant. In a multiplying DAC, "four quadrant" refers to the fact that both the
reference signal and the number represented by the digital input may be of either positive
or negative polarity. Such a DAC can be thought of as a gain control for ac signals
("reference" input) with a range of positive and negative digitally controlled gains. A
four-quadrant multiplier is expected to obey the rules of multiplication for algebraic sign.
Frequency-to-Voltage Conversion (FVC). The input of a FVC device is an ac
waveform—usually a train of pulses (in the context of conversion); the output is an
analog voltage, proportional to the number of pulses occurring in a given time. FVC is
usually performed by a voltage-to-frequency converter in a feedback loop. Important
specifications, in addition to the accuracy specs typical of VFCs (see Voltage-toFrequency conversion), include output ripple (for specified input frequencies), threshold
(for recognition that another cycle has been initiated, and for versatility in interfacing
several types of sensors directly), hysteresis, to provide a degree of insensitivity to noise
superimposed on a slowly varying input waveform, and dynamic response (important in
motor control).
Full-Scale Input Power (ADC). Expressed in dBm (power level referenced to 1 mW).
Computed using the following equation, where V Full Scale rms is in volts, and Zinput is in Ω.
 V 2 Full Scale rms 


Z Input


Power Full Scale = 10 log10 
.
0.001






Full-Scale Range (FSR). For binary ADCs and DACs, that magnitude of voltage, current,
or—in a multiplying DAC—gain, of which the MSB is specified to be exactly one-half or
for which any bit or combination of bits is tested against its (their) prescribed ideal
ratio(s). FSR is independent of resolution; the value of the LSB (voltage, current, or gain)
is 2–N FSR. There are several other terms, with differing meanings, that are often used in
the context of discussions or operations involving full-scale range. They are:
Full-scale—similar to full-scale range, but pertaining to a single polarity. Thus,
full-scale for a unipolar device is twice the prescribed value of the MSB and has the same
polarity. For a bipolar device, positive or negative full-scale is that positive or negative
value, of which the next bit after the polarity bit is tested to be one-half.
Span—the scalar voltage or current range corresponding to FSR.
All-1's—All bits on, the condition used, in conjunction with all-zeros, for gain
adjustment of an ADC or DAC, in accordance with the manufacturer's instructions. Its
magnitude, for a binary device, is (l – 2–N) FSR. All-1's is a positive-true definition of a
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FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
specific magnitude relationship; for complementary coding the "all-l's" code will actually
be all zeros. To avoid confusion, all-l's should never be called "full-scale;" FSR and FS
are independent of the number of bits, all-l's isn't.
All-0's—All bits off, the condition used in offset (and gain) adjustment of a DAC
or ADC, according to the manufacturer's instructions. All-0's corresponds to zero output
in a unipolar DAC and negative full-scale in an offset bipolar DAC with positive output
reference. In a sign-magnitude device, all-0's refers to all bits after the sign bit.
Analogous to "all-l's," "all-0's" is a positive-true definition of the all-bits-off condition; in
a complementary-coded device, it is expressed by all ones. To avoid confusion, all-0's
should not be called "zero" unless it accurately corresponds to true analog zero output
from a DAC.
Gain. The "gain" of a converter is that analog scale factor setting that establishes the
nominal conversion relationship, e.g., 10 volts full-scale. In a multiplying DAC or
ratiometric ADC, it is indeed a gain. In a device with fixed internal reference, it is
expressed as the full-scale magnitude of the output parameter (e.g., 10 V or 2 mA). In a
fixed-reference converter, where the use of the internal reference is optional, the
converter gain and the reference may be specified separately. Gain and zero adjustment
are discussed under zero.
Glitch. Transients associated with code changes generally stem from several sources.
Some are spikes, known as digital-to-analog feedthrough, or charge transfer, coupled
from the digital signal (clock or data) to the analog output, defined with zero reference.
These spikes are generally fast, fairly uniform, code-independent, and hence filterable.
However, there is a more insidious form of transient, code-dependent, and difficult to
filter, known as
the "glitch."
If the output of a counter is applied to the input of a DAC to develop a "staircase"
voltage, the number of bits involved in a code change between two adjacent codes
establish "major" and "minor" transitions. The most major transition is at ½-scale, when
the DAC switches all bits, i.e., from 011. . .111 to 100. . .000. If, for digital inputs having
no skew, the switches are faster to switch off than on, this means that, for a short time, the
DAC will seek zero output, and then return to the required 1 LSB above the previous
reading. This large transient spike is commonly known as a "glitch." The better matched
the input transitions and the switching times, the faster the switches, the smaller will be
the area of the glitch. Because the size of the glitch is not proportional to the signal
change, linear filtering may be unsuccessful and may, in fact, make matters worse. (See
also Deglitcher.)
The severity of a glitch is specified by glitch impulse area, the product of its duration and
its average magnitude, i.e., the net area under the curve. This product will be recognized
as the physical quantity, impulse (electromotive force × ∆time); however, it has also been
incorrectly termed "glitch energy" and "glitch charge." Glitch impulse area is usually
expressed, for fast converters, in units of pV-s or mV-ns.
The glitch can be minimized through the use of fast, non-saturating logic, such as ECL,
LVDS, matched latches, and non-saturating CMOS switches.
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ANALOG-DIGITAL CONVERSION
Glitch Charge, Glitch Energy, Glitch Impulse, Glitch Impulse Area. (See Glitch).
Harmonic Distortion, 2nd.The ratio of the rms signal amplitude to the rms value of the
second harmonic component, reported in dBc.
Harmonic Distortion, 3rd. The ratio of the rms signal amplitude to the rms value of the
third harmonic component, reported in dBc.
Harmonic Distortion, Total (THD). The ratio of the rms signal amplitude to the rms sum
of all harmonics (neglecting noise components). In most cases, only the first five
harmonics are included in the measurement because the rest have negligible contribution
to the result. The THD can be derived from the FFT of the ADC's output spectrum. For
harmonics that are above the Nyquist frequency, the aliased component is used.
Harmonic Distortion, Total, Plus Noise (THD + N). Total harmonic distortion plus noise
(THD + N) is the ratio of the rms signal amplitude to the rms sum of all harmonics and
noise components. THD + N can be derived from the FFT of the ADC's output spectrum
and is a popular specification for audio applications.
Impedance, Input. The dynamic load of an ADC presented to its input source. In
unbuffered CMOS switched-capacitor ADCs, the presence of current transients at the
converter's clock frequency mandates that the converter be driven from a low impedance
(at the frequencies contained in the transients) in order to accurately convert. For
buffered-input ADCs, the input impedance is generally represented by a resistive and
capacitive component.
Input-Referred Noise (Effective Input Noise). Input-referred noise can be viewed as the
net effect of all internal ADC noise sources referred to the input. It is generally expressed
in LSBs rms, but can also be expressed as a voltage. It can be converted to a peak-to-peak
value by multiplying by the factor 6.6. The peak-to-peak input-referred noise can then be
used to calculate the noise-free code resolution. (See noise-free code resolution).
Intermodulation Distortion (IMD). With inputs consisting of sinewaves at two
frequencies, f1 and f2, any device with nonlinearities will create distortion products of
order (m + n), at sum and difference frequencies of mf1 ± nf2, where m, n = 0, 1, 2, 3, ….
Intermodulation terms are those for which m or n is not equal to zero. For example, the
second-order terms are (f1 + f2) and (f2 – f1), and the third-order terms are (2f1 + f2),
(2f1 – f2), (f1 + 2f2), and (f1 – 2f2). The IMD products are expressed as the dB ratio of the
rms sum of the distortion terms to the rms sum of the measured input signals.
Latency. (See pipelining).
Leakage Current, Output. Current which appears at the output terminal of a DAC with all
bits "off." For a converter with two complementary outputs (for example, many fast
CMOS DACs), output leakage current is the current measured at OUT 1, with all digital
inputs low—and the current measured at OUT 2, with all digital inputs high.
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FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
Least-Significant Bit (LSB). In a system in which a numerical magnitude is represented
by a series of binary (i.e. two-valued) digits, the least-significant bit is that digit (or "bit")
that carries the smallest value, or weight. For example, in the natural binary number 1101
(decimal 13, or (1 × 23) + (1 × 22) + (0 × 21) + (1 × 20)), the rightmost digit is the LSB. Its
analog weight, in relation to full-scale (see Full-Scale Range), is 2–N, where N is the
number of binary digits. It represents the smallest analog change that can be resolved by
an n-bit converter.
In data converter nomenclature, the LSB is bit N; in bus nomenclature (integer binary), it
is Data Bit 0.
Left-Justified Data. When a 12-bit word is placed on an 8-bit bus in two bytes, the high
byte contains the 4- or 8- most-significant bits. If 8, the word is said to be left justified; if
4 (plus filled-in leading sign bits), the word is said to be right justified.
Linearity. (See also Nonlinearity.) Linearity error of a converter (also, integral
nonlinearity—see Linearity, Differential), expressed in % or parts per million of fullscale range, or (sub)multiples of 1 LSB, is a deviation of the analog values, in a plot of
the measured conversion relationship, from a straight line. The straight line can be either
a "best straight line," determined empirically by manipulation of the gain and/or offset to
equalize maximum positive and negative deviations of the actual transfer characteristic
from this straight line; or, it can be a straight line passing through the end points of the
transfer characteristic after they have been calibrated, sometimes referred to as "endpoint" linearity. "End-point" nonlinearity is similar to relative accuracy error (see
Accuracy, Relative). It provides an easier method for users to calibrate a device, and it is
a more conservative way to specify linearity.
For multiplying DACs, the analog linearity error, at a specified analog gain (digital
code), is defined in the same way as for analog multipliers, i.e., by deviation from a "best
straight line" through the plot of the analog output-input response.
Linearity, Differential. In a DAC, any two adjacent digital codes should result in
measured output values that are exactly 1 LSB apart (2–N of full-scale for an N-bit
converter). Any positive or negative deviation of the measured "step" from the ideal
difference is called differential nonlinearity, expressed in (sub)multiples of 1 LSB. It is
an important specification, because a differential linearity error more negative than
–1 LSB can lead to non-monotonic response in a DAC and missed codes in an ADC
using that DAC.
Similarly, in an ADC, midpoints between code transitions should be 1 LSB apart.
Differential nonlinearity is the deviation between the actual difference between midpoints
and 1 LSB, for adjacent codes. If this deviation is equal to or more negative than –1 LSB,
a code will be missed (see missing codes).
Often, instead of a maximum differential nonlinearity specification, there will
be a simple specification of "monotonicity" or "no missing codes", which implies that the
differential nonlinearity cannot be more negative than –1 for any adjacent pair of codes.
However, the differential linearity error may still be more positive than
+1 LSB.
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ANALOG-DIGITAL CONVERSION
Linearity, Integral. (See Linearity). While differential linearity deals with errors in step
size, integral linearity has to do with deviations of the overall shape of the conversion
response. Even converters that are not subject to differential linearity errors.(e.g.,
integrating types) have integral linearity (sometimes just "linearity") errors.
Maximum Conversion Rate. The maximum sampling (encode) rate at which parametric
testing is performed.
Minimum Conversion (Sampling) Rate. The encode rate at which the SNR of the lowest
analog signal frequency drops by no more than 3 dB below the guaranteed limit.
Missing Codes. An ADC is said to have missing codes when a transition from one
quantum of the analog range to the adjacent one does not result in the adjacent digital
code, but in a code removed by one or more counts. Missing codes can be caused by
large negative differential linearity errors, noise, or changing inputs during conversion. A
converter's proclivity towards missing codes is also a function of the architecture and
temperature.
Monotonicity. An DAC is said to be monotonic if its output either increases or remains
constant as the digital input increases, with the result that the output will always be a
single-valued function of the input. The condition "monotonic" requires that the
derivative of the transfer function never change sign. Monotonic behavior requires that
the differential nonlinearity be more positive than –1 LSB. The same basic definition
applies to an ADC—the digital output code either increases or remains constant as the
digital input increases. In practice, however, noise will cause the ADC output code to
oscillate between two code transitions over a small range of analog input. Input-referred
noise can make this effect worse, so histogram techniques are often used to measure
ADC monotonicity in these situations.
Most Significant Bit (MSB). In a system in which a numerical magnitude is represented
by a series of binary (i.e., two-valued) digits, the most-significant bit is that digit (or
"bit") that carries the greatest value or weight. For example, in the natural binary number
1101 (decimal 13, or (1 × 23) + (1 × 22) + (0 × 21) + (1 × 20 )), the leftmost "1" is the
MSB, with a weight of ½ nominal peak-to-peak full-scale (full-scale range). In bipolar
devices, the sign bit is the MSB.
In converter nomenclature, the MSB is bit 1; in bus nomenclature, it is Data Bit (N – 1).
Multiplying DAC. A multiplying DAC differs from the conventional fixed-reference
DAC in being designed to operate with varying (or ac) reference signals. The output
signal of such a DAC is proportional to the product of the "reference" (i.e., analog input)
voltage and the fractional equivalent of the digital input number. See also FourQuadrant.
Multitone Spurious Free Dynamic Range (SFDR). The ratio of the rms value of an input
tone to the rms value of the peak spurious component. The peak spurious component may
or may not be an intermodulation distortion (IMD) product. May be reported in dBc (dB
relative to the carrieror in dBFS (dB relative to full-scale). The amplitudes of the
2.108
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
individual tones are equal and chosen such that the ADC is not overdriven when they add
in-phase.
Noise-Free (Flicker-Free) Code Resolution. The noise-free code resolution of an ADC is
the number of bits beyond which it is impossible to distinctly resolve individual codes.
The cause is the effective input noise (or input-referred noise) associated with all ADCs.
This noise can be expressed as an rms quantity, usually having the units of LSBs rms.
Multiplying by a factor of 6.6 converts the rms noise into peak-to-peak noise (expressed
in LSBs peak-to-peak). The total range of an N-bit ADC is 2N. The noise-free (or flickerfree) resolution can be calculated using the equation:
Noise-Free Code Resolution = log2(2N/Peak-to-Peak Noise).
The specification is generally associated with high-resolution sigma-delta measurement
ADCs, but is applicable to all ADCs.
The ratio of the FS range to the rms input noise is sometimes used to calculate resolution.
In this case, the term effective resolution is used. Note that effective resolution is larger
than noise-free code resolution by log2(6.6), or approximately 2.7 bits.
Effective Resolution = log2(2N/RMS Input Noise).
Noise, Peak and RMS. Internally generated random noise is not a major factor in DACs,
except at extreme resolutions and dynamic ranges. Random noise is characterized by rms
specifications for a given bandwidth, or as a spectral density (current or voltage per root
hertz); if the distribution is Gaussian, the probability of peak-to-peak values exceeding
6.6 × the rms value is less than 0.l%.
Of much greater importance in DACs is interference, in the form of high-amplitude, lowenergy (hence low-rms) spikes appearing at a DAC's output, caused by coupling of
digital signals in a surprising variety of ways; they include coupling via stray capacitance,
via power supplies, via inadequate ground systems, via feedthrough, and by glitchgeneration (see Glitch). Their presence underscores the necessity for maximum
application of the designer's art, including layout, shielding, guarding, grounding,
bypassing, and deglitching.
Noise in ADCs in effect narrows the region between transitions. Sources of noise include
the input sample-and-hold, resistor noise, "KT/C" noise, the reference, the analog signal
itself, and pickup in infinite variety.
Noise Power Ratio (NPR). In this measurement, wideband Gaussian noise (bandwidth
< fs/2) is applied to an ADC through a narrowband notch filter. The notch filter removes
all noise within its bandwidth. The output of the ADC is examined with a large FFT. The
ratio of the rms noise level to the rms noise level inside the notch (due to quantization
noise, thermal noise, and intermodulation distortion) is defined as the noise power ratio
(NPR). The rms noise level at the input to the ADC is generally adjusted to give the best
NPR value.
No Missing Codes Resolution. (See Resolution, No Missing Codes).
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ANALOG-DIGITAL CONVERSION
Nonlinearity--or "gain nonlinearity"—is defined as the deviation from a straight line on
the plot of output vs. input. The magnitude of linearity error is the maximum deviation
from a "best straight line," with the output swinging through its full-scale range.
Nonlinearity is usually specified in percent of full-scale output range.
Normal Mode. For an amplifier used in instrumentation, the normal-mode signal is the
actual difference signal being measured. This signal often has noise associated with it.
Signal conditioning systems and digital panel instruments usually contain input filtering
to remove high frequency and line frequency noise components. Normal-mode rejection
(NMR), is a logarithmic measure of the attenuation of normal-mode noise components at
specified frequencies in dB.
Offset, Bipolar. For the great majority of bipolar converters (e.g., ±10-V output), negative
currents are not actually generated to correspond to negative numbers; instead, a unipolar
DAC is used, and the output is offset by half full-scale (1 MSB). For best results, this
offset voltage or current is derived from the same reference supply that determines the
gain of the converter.
Because of nonlinearity, a device with perfectly calibrated end points may have offset
error at analog zero.
Offset Step. (See Pedestal).
Output Propagation Delay. For an ADC having a single-ended sampling (or ENCODE)
clock input, the delay between the 50% point of the sampling clock and the time when all
output data bits are within valid logic levels. For an ADC having differential sampling
clock inputs, the delay is measured with respect to the zero-crossing of the differential
sampling clock signal.
Output Voltage Tolerance. For a reference, the maximum deviation from the normal
output voltage at 25°C and specified input voltage, as measured by a device traceable to a
recognized fundamental voltage standard.
Overload. An input voltage exceeding the ADC's full-scale input range producing an
overload condition.
Overvoltage Recovery Time. Overvoltage recovery time is defined as the amount of time
required for an ADC to achieve a specified accuracy after an overvoltage (usually 50%
greater than full-scale range), measured from the time the overvoltage signal reenters the
converter's range. The ADC should act as an ideal limiter for out-of-range signals,
producing a positive or negative full-scale code during the overvoltage condition. Some
ADCs provide over- and under-range flags to allow gain-adjustment circuits to be
activated.
Overrange, Overvoltage. An input signal that exceeds the full-scale input range of an
ADC, but is less than an overload.
2.110
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
Pedestal, or Sample-to-Hold Offset Step. In sample/track-and-hold amplifiers, a shift in
level between the last value in sample and the value settled-to in hold; in devices having
fixed internal capacitors, it includes charge transfer, or offset step. However, for devices
that may use external capacitors, it is often defined as the residual step error after the
charge transfer is accounted for and/or cancelled. Since it is unpredictable in magnitude
and may be a function of the signal, it is also known as offset nonlinearity.
Pipelining. A pipelined converter is a multistage converter which is capable of accepting
a new signal before it has completed the conversion of one or more previous ones. A new
signal arrives while others are still "in the pipeline." This is a technique used where a fast
conversion rate is desired and the latency of individual conversions is relatively
unimportant.
Power-Supply Rejection Ratio (PSRR). The ratio of a change in dc power supply voltage
to the resulting change in the specified device error, expressed in percentage, parts per
million, or fractions of 1 LSB. It may also be expressed logarithmically, in dB,
PSR = 20 log10 (PSRR).
Quad-Slope Converter. This is an integrating analog-to-digital converter that goes
through two cycles of dual-slope conversion, once with zero input and once with the
analog input being measured. The errors determined during the first cycle are subtracted
digitally from the result in the second cycle. The scheme can result in high-accuracy
conversion.
Quantizing Uncertainty (or" Quantization Error"). The analog continuum is partitioned
into 2N discrete ranges for N-bit conversion and processing. All analog values within a
given quantum are represented by the same digital code, usually assigned to the nominal
mid-range value. There is, therefore, an inherent quantization uncertainty of ±½ LSB, in
addition to the actual conversion errors. In integrating ADCs, this "error" is often
expressed as " ±1 count." Depending on the system context, it may be interpreted as a
truncation (round-off) error or as noise.
Ratiometric. The output of an ADC is a digital number proportional to the ratio of (some
measure of) the input to a reference voltage. Most requirements for conversions call for
an absolute measurement, i.e., against a fixed reference; but this presumes that the signal
applied to the converter is either reference-independent or in some way derived from
another fixed reference. However, real references are not truly fixed; the references for
both the converter and the signal source vary with time, temperature, loading, etc.
Therefore, if the converter is used with signal sources that also rely on references (for
example, strain-gage bridges, RTDs, thermistors), it makes sense to replace this
multiplicity of references by a single system reference. In this case, reference-caused
errors will tend to cancel out. This can be done by using the converter's internal reference
(if it has one) as the system reference. Another way is to use a separate external system
reference, which also becomes the reference for a ratiometric converter. For instance, if a
bridge is excited with the same voltage used for the ADC reference, ratiometric operation
is achieved, and the ADC output code is not a function of the reference. This is because
the bridge output signal is proportional to the same voltage which defines the ADC input
range.
2.111
ANALOG-DIGITAL CONVERSION
Resolution. An N-bit binary converter has N digital data inputs (DAC) or N digital data
outputs (ADC). A converter that satisfies this criterion is said to have a resolution of N
bits.
Resolution, No Missing Codes. The no missing code resolution of an ADC is the
maximum number of bits of resolution beyond which the ADC will have missing codes.
For instance, if an 18-bit ADC has a no missing code resolution of 16 bits, then there will
be no missing codes if only the 16 MSBs are utilized. Codes may be missed at the 17and 18-bit level.
The smallest output change that can be resolved by a linear DAC is 2–N of the full-scale
span. Thus, for example, the resolution of an 8-bit DAC would be 2–8, or 1/256. On the
other hand, a nonlinear device, such as the AD7111 LOGDAC, can ideally achieve a
dynamic range of 89.625 dB, or 30,000:1, in 0.375-dB steps, using only 8 bits of digital
resolution.
Right-Justified Data. When a 12-bit word is placed on an 8-bit bus in two stages, the high
byte contains the 4- or 8- most-significant bits. If 8, the word is said to be left justified; if
4 (plus filled-in leading sign bits), the word is said to be right justified.
Sample-to-Hold Offset.(See Pedestal).
Sampling ADC. A sampling ADC includes a sample-and-hold function which acquires
the input value at a given instant and holds it throughout the conversion time (or until the
converter is ready for the next sample point). Flash ADCs and sigma-delta ADCs are
inherently sampling devices.
Sampling Clock. (See Encode Command).
Sampling Frequency. The rate at which an ADC converts an analog input signal into
digital outputs, not to be confused with conversion time.
Serial Output. A bit-serial output consists of a series of bits clocked out on a single line.
There must be some means of identifying the beginning and ends of words; this can be
accomplished via an additional clock line, by using synchronized clocks, and/or by
providing a consistent identifying signature for the beginning of a word. Byte-serial
consists of a series of bytes transmitted in sequence on a bus (see Byte).
Settling Time—ADC. The time required, following an analog input step change (usually
full-scale), for the digital output of the ADC to reach and remain within a given fraction
(usually ± ½LSB).
Settling Time—DAC. The time required, following a prescribed data change, for the
output of a DAC to reach and remain within an error band (usually ±½ LSB) of the final
value. Typical prescribed changes are full-scale, 1 MSB, and 1 LSB at a major carry.
Settling time of current-output DACs is quite fast. The major share of settling time of a
voltage-output DAC is usually contributed by the settling time of the output op-amp.
DAC settling time can also be defined with respect to the output. Output settling time is
the time measured from the point the output signal leaves an error band referenced to the
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FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
initial output value until the time the signal enters and remains within the error band
referenced to the final output value.
Signal-to-Noise-and-Distortion Ratio (SINAD). The ratio of the rms signal amplitude (set
1 dB below full-scale to prevent overdrive) to the rms value of the sum of all other
spectral components, including harmonics but excluding dc.
Signal-to-Noise Ratio (without Harmonics).The ratio of the rms signal amplitude (set at
1 dB below full-scale to prevent overdrive) to the rms value of the sum of all other
spectral components, excluding the first five harmonics and dc. Technically, all
harmonics should be excluded, but in practice, only the first five are generally significant.
Single-Slope Conversion. In the single-slope converter, a reference voltage is integrated
until the output of the integrator is equal to the input voltage. The time period required
for the integrator to go from zero to the level of the input is proportional to the magnitude
of the input voltage and is measured by an internal clock. Measurement accuracy is
sensitive to clock speed and integrating capacitance, as well as the reference accuracy.
Slew(ing) Rate. A limitation in the rate of change of output voltage, usually imposed by
some basic circuit consideration, such as limited current to charge a capacitor. The output
slewing speed of a voltage-output DAC is usually limited by the slew rate of the
amplifier used at its output.
Spurious-Free Dynamic Range (SFDR). The ratio of the rms signal amplitude to the rms
value of the peak spurious spectral component. The peak spurious component may or
may not be a harmonic. May be reported in dBc (i.e., degrades as signal level is lowered)
or dBFS (related back to converter full-scale).
Stability. In a well-designed, intelligently applied converter, dynamic stability is not an
important question. The term stability usually applies to the insensitivity of the
converter's characteristics to time, temperature, etc. All measurements of stability are
difficult and time consuming, but stability vs. temperature is sufficiently critical in most
applications to warrant universal inclusion in tables of specifications (see Temperature
Coefficient).
Staircase. A voltage or current, increasing in equal increments as a function of time and
having the appearance of a staircase (in a time plot); it is generated by applying a pulse
train to a counter, and the output of the counter to the input of a DAC.
Subranging ADCs. In this type of converter, a fast converter produces the mostsignificant portion of the output word. This portion is stored in a holding register and also
converted back to analog with a fast, high-accuracy DAC. The analog result is subtracted
from the input, and the resulting residue is amplified, converted to digital at high speed,
and combined with the results of the earlier conversion to form the output word. In
digitally corrected subranging (DCS) ADCs, the two conversions are combined in a
manner that corrects for the error of the LSB of the most significant bits. For example,
using 8-bit and 5-bit conversion, plus this technique and a great deal of video-speed
converter expertise, a full-accuracy high-speed 12-bit ADC can be built. Many pipelined
subranging ADCs use more than two stages with error correction between each stage.
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Successive Approximation. Successive approximation is a method of conversion by
comparing an unknown against a group of weighted references. The operation of a
successive-approximation ADC is generally similar to the orderly weighing of an
unknown quantity on a precision balance, using a set of weights, such as 1 gram ½ gram,
¼ gram, etc. The weights are tried in order, starting with the largest. Any weight that tips
the scale is removed. At the end of the process, the sum of the weights remaining on the
scale will be within 1 LSB of the actual weight (±½ LSB, if the scale is properly biased—
see Zero). The successive approximation ADC is often called a SAR ADC, because the
logic block which controls the conversion process is known as a successive
approximation register (SAR).
Switching Time. In a DAC, the switching time is the time taken for an analog switch to
change to a new state from the previous one. It includes propagation delay time, and rise
time from 10% to 90%, but does not include settling time.
Temperature Coefficient. In general, temperature instabilities are expressed as %/°C,
ppm/°C, fractions of 1 LSB per degree C, or as a change in a parameter over a specified
temperature range. Measurements are usually made at room temperature (25°C) and at
the extremes of the specified range, and the temperature coefficient (tempco, TC) is
defined as the change in the parameter, divided by the corresponding temperature change.
Parameters of interest include gain, linearity, offset (bipolar), and zero.
a. Gain Tempco: Two factors principally affect converter gain stability with
temperature. In fixed-reference converters, the reference voltage will vary with
temperature. The reference circuitry and switches (and comparator in ADCs) will also
contribute to the overall gain TC.
b. Linearity Tempco: Sensitivity of linearity (integral and/or differential linearity)
to temperature, in % FSR/°C or ppm FSR/°C, over the specified range. Monotonic
behavior in DACs is achieved if the differential nonlinearity is less than 1 LSB at any
temperature in the range of interest. The differential nonlinearity temperature coefficient
may be expressed as a ratio, as a maximum change over a temperature range, and/or
implied by a statement that the device is monotonic over the specified temperature range.
To avoid missing codes in noiseless ADCs, it is sufficient that the differential
nonlinearity error be less than –1 LSB at any temperature in the range of interest. The
differential nonlinearity temperature coefficient is often implied by the statement that
there are no missed codes when operating within a specified temperature range. In DACs,
the differential nonlinearity TC is often implied by the statement that the DAC is
monotonic over a specified temperature range.
c. Zero TC (unipolar converters): The temperature stability of a unipolar
fixed-reference DAC, measured in % FSR/°C or ppm FSR/°C, is principally affected by
current leakage (current-output DAC), and offset voltage and bias current of the output
op amp (voltage-output DAC). The zero stability of an ADC is dependent on the zero
stability of the DAC or integrator and/or the input buffer and the comparator. It is
typically expressed in µV/°C or in percent or ppm of full-scale range (FSR) per degree C.
2.114
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
d. Offset Tempco: The temperature coefficient of the all-DAC-switches-off
(minus full-scale) point of a bipolar converter (in % FSR/°C or ppm FSR/°C) depends on
three major factors—the tempco of the reference source, the voltage zero-stability of the
output amplifier, and the tracking capability of the bipolar-offset resistors and the gain
resistors. In an ADC, the corresponding tempco of the negative full-scale point depends
on similar quantities—the tempco of the reference source, the voltage stability of the
input buffer and the sample-and-hold, and the tracking capabilities of the bipolar offset
resistors and the gain-setting resistors.
Thermal Tail. The slow drift of an amplifier having a thermally induced offset due to
self-heating as it settles to a final electrical equilibrium value corresponding to internal
thermal equilibrium.
Total Unadjusted Error. A comprehensive specification on some devices which includes
full-scale error, relative-accuracy and zero-code errors, under a specified set of
conditions.
Transient Response. (See settling time).
Two Tone SFDR. The ratio of the rms value of either input tone to the rms value of the
peak spurious component. The peak spurious component may or may not be an
intermodulation distortion (IMD) product. May be reported in dBc (i.e., degrades as
signal level is lowered) or in dBFS (always related back to converter full-scale).
Worst Other Spur. The ratio of the rms signal amplitude to the rms value of the worst
spurious component (excluding the second and third harmonic) reported in dBc.
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ANALOG-DIGITAL CONVERSION
General References on Data Conversion and Related Topics
1.
Alfred K. Susskind, Notes on Analog-Digital Conversion Techniques, John Wiley, 1957.
2.
David F. Hoeschele, Jr., Analog-to-Digital/Digital-to-Analog Conversion Techniques, John Wiley
and Sons, 1968.
3.
K. W. Cattermole, Principles of Pulse Code Modulation, American Elsevier Publishing Company,
Inc., 1969, New York NY, ISBN 444-19747-8.
4.
Hermann Schmid, Electronic Analog/Digital Conversions, Van Nostrand Reinhold Co., 1970.
5.
Dan Sheingold, Analog-Digital Conversion Handbook, First Edition, Analog Devices, 1972.
6.
Donald B. Bruck, Data Conversion Handbook, Hybrid Systems Corporation, 1974.
7.
Eugene R. Hnatek, A User's Handbook of D/A and A/D Converters, John Wiley, New York, 1976,
ISBN 0-471-40109-9.
8.
Nuggehally S. Jayant, Waveform Quantizing and Coding, John Wiley-IEEE Press, 1976, ISBN 087942-074-X.
9.
Dan Sheingold, Analog-Digital Conversion Notes, Analog Devices, 1977.
10. C. F. Kurth, editor, IEEE Transactions on Circuits and Systems Special Issue on Analog/Digital
Conversion, CAS-25, No. 7, July 1978.
11. Daniel J. Dooley, Data Conversion Integrated Circuits, John Wiley-IEEE Press, 1980, ISBN 0-47108155-8.
12. Bernard M. Gordon, The Analogic Data-Conversion Systems Digest, Fourth Edition, Analogic
Corporation, 1981.
13. Eugene L. Zuch, Data Acquisition and Conversion Handbook, Datel-Intersil, 1982.
14. Frank F. E. Owen, PCM and Digital Transmission Systems, McGraw-Hill, 1982, ISBN 0-07047954-2.
15. Dan Sheingold, Analog-Digital Conversion Handbook, Analog Devices/Prentice-Hall, 1986, ISBN
0-13-032848-0.
16. Matthew Mahoney, DSP-Based Testing of Analog and Mixed-Signal Circuits, IEEE Computer
Society Press, 1987, ISBN 0-8186-0785-8.
17. Michael J. Demler, High-Speed Analog-to-Digital Conversion, Academic Press, Inc., 1991, ISBN 012-209048-9.
18. J. C. Candy and Gabor C. Temes, Oversampling Delta-Sigma Data Converters, IEEE Press, ISBN
0-87942-258-8, 1992.
19. David F. Hoeschele, Jr., Analog-to-Digital and Digital-to-Analog Conversion Techniques, Second
Edition, John Wiley and Sons, 1994, ISBN-0-471-57147-4.
20. Rudy van de Plassche, Integrated Analog-to-Digital and Digital-to-Analog Converters, Kluwer
Academic Publishers, 1994, ISBN 0-7923-9436-4.
21. David A. Johns and Ken Martin, Analog Integrated Circuit Design, John Wiley, 1997, ISBN 0-47114448-7.
2.116
FUNDAMENTALS OF SAMPLED DATA SYSTEMS
2.5 DEFINING THE SPECIFICATIONS
22. Mikael Gustavsson, J. Jacob Wikner, and Nianxiong Nick Tan, CMOS Data Converters for
Communications, Kluwer Academic Publishers, 2000, ISBN 0-7923-7780-X.
23. R. Jacob Baker, CMOS Circuit Design Volumes I and II, John Wiley-IEEE Computer Society, 2002,
ISBN 0-4712-7256-6.
24. Rudy van de Plassche, CMOS Integrated Analog-to-Digital and Digital-to-Analog Converters,
Second Edition, Kluwer Academic Publishers, 2003, ISBN 1-4020-7500-6.
Analog Devices' Seminar Series:
25. Walt Kester, Practical Analog Design Techniques, Analog Devices, 1995, ISBN 0-916550-16-8,
available for download at http://www.analog.com.
26. Walt Kester, High Speed Design Techniques, Analog Devices, 1996, ISBN 0-916550-17-6, available
for download at http://www.analog.com.
27. Walt Kester, Practical Design Techniques for Power and Thermal Management, Analog Devices,
1998, ISBN 0-916550-19-2, available for download at http://www.analog.com.
28. Walt Kester, Practical Design Techniques for Sensor Signal Conditioning, Analog Devices, 1999,
ISBN 0-916550-20-6, available for download at http://www.analog.com.
29. Walt Kester, Mixed-Signal and DSP Design Techniques, Analog Devices, 2000, ISBN 0-916550-222, available for download at http://www.analog.com.
30. Walt Kester, Mixed-Signal and DSP Design Techniques, Analog Devices and Newnes (An Imprint
of Elsevier Science), ISBN 0-75067-611-6, 2003.
31. Walter G. Jung, Op Amp Applications, Analog Devices, 2002, ISBN 0-916550-26-5.
2.117
ANALOG-DIGITAL CONVERSION
NOTES:
2.118
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