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VOLTAGE
A Test Engineering Perspective
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Analog-to-Digital Converter
Clock Optimization:
Δv
By Rob Reeder, Wayne Green, and Robert Shillito
System clock optimization can be both challenging and rewarding.
It may be relatively easy to design an analog-to-digital converter
encode circuit with a respectable 350 femtoseconds (fs) of jitter,
but is this adequate for today’s high speed requirements? For
example, when testing an AD9446-1001—a 16-bit, 100-MHz
ADC—at Nyquist with a 100-MHz sample clock, 350 fs of jitter
the same device is tested at the 3rd Nyquist zone with a 105-MHz
the clock jitter to a more tolerable 100 fs or less, the designer
needs to understand where the clock jitter is coming from,
as well as how much jitter the ADC can tolerate. It can be
quite discouraging to realize—too late—that the clock-circuit
performance is jitter-limited, and that this problem could have
been more easily prevented during the design phase.
We will consider here the relevant clock specifications and means
of achieving the expected performance of a high speed converter—
employing a little know-how and experience. Starting with a
typical ADC clocking scheme, such as that shown in Figure 1, we
will highlight techniques that can be used to optimize the clock
at each point in the signal chain—and identify some commonly
used techniques that should be avoided.
SOURCE
DRIVER
CONDITIONER
ANALOG
INPUT
DIGITAL
OUTPUT
Figure 1. Typical clock signal chain.
Δt
TIME
Analog Dialogue 42-02, February (2008)
b) HIGH FREQUENCY
Figure 2. Conversion error as a function of clock jitter and
Since this relationship is intuitively obvious, the engineer will
ultimately determine how much jitter is acceptable by relating the
ADC’s performance to the jitter of the encode clock. Equation 1
defines the SNR (dB)—with frequency—of a perfect ADC having
infinite resolution, while Equation 2 is the SNR (dB) of a perfect
ADC with N- (10, 12, 14, or 16) bit resolution.
(1)
(see diagonal lines of Figure 3)
(2)
(see horizontal lines of Figure 3)
Figure 3 combines these two equations. The intersections allow
the user to determine the amount of total clock jitter that can be
the accuracy is limited by the resolution of the converter. As the
input frequency increases, however, a point is reached beyond
which the performance of the ADC is dominated by the total
clock jitter of the system. For input frequencies to the left of the
intersections, lower jitter is unlikely to be of concern.
120
350fs JITTER
95
SNR (dBFS)
Jitter is variation in the placement of a clock edge; it will produce a
timing error, leading directly to errors in conversion amplitude
increases the slope of the input signal, which magnifies the
conversion error (Figure 2b). It is important to note that the
magnitude of the conversion error is relative—a 0.5-LSB (leastsignificant-bit) conversion error for a 10-bit device is the equivalent
of 32 LSBs of error for a 16-bit device. This means that jitter
becomes more of a concern as both ADC resolution and analog
input frequency increase.
TIME
a) LOW FREQUENCY
What Is Jitter?
Jitter is probably the most important parameter in developing a
good system clock circuit, so it is important to review some basics
and understand what is meant by the term. Many technical papers
describe the mathematics of jitter to the nth degree; however,
design for good converter performance is not all about the exact
description of jitter. One must also understand how it can get into
the system and how to minimize its impact.
Δt
100fs JITTER
16 BITS
14 BITS
70
12 BITS
10 BITS
45
20
8 BITS
1
10
100
1000
10000
INPUT FREQUENCY (MHz)
and jitter.
http://www.analog.com/analogdialogue
1
For example, if a 14-bit ADC is tested using a clock that has
350 fs of jitter, the analog input frequency must be limited to
frequencies below 35 MHz (the intersection of the 14-bit level
and the 350 fs slope) to avoid significantly reduced performance.
If the jitter can be reduced to 100 fs, input frequencies as high
as 125 MHz can be handled.
In practice, this simplified model, using these first-order
approximations, loses validity as the analog test frequency
approaches the intersections. In order to fully understand the effect
that clock jitter has on the ADC’s performance, the quantization
to the resolution (Equation 3, based on Further Reading 9).
SAMPLE-CLOCK
JITTER
QUANTIZATION
NOISE, DNL
EFFECTIVE
INPUT NOISE
(3)
where
SNR
=Signal-to-noise ratio in dB.
fa
tj rms
=Combined rms internal ADC jitter and external clock jitter.
ε
=Average differential nonlinearity (DNL) of the ADC in LSBs.
N
V NOISE rms =Effective input noise of ADC.
If tj rms = 0, ε = 0, and V NOISE rms = 0, the above equation reduces
to the familiar
SNR = 6.02N + 1.76dB
For example, assume that an ADC has 0.5-LSB quantization
0.5 dB below full scale. Figure 4, combining Equation 2 and
Equation 3, shows that the encode clock jitter will affect the SNR
performance at lower frequencies than in the simplified model.
NOISE = 0.5 LSB
LEVEL = –0.5dB
120
350fs JITTER
SNR (dBFS)
14 BITS
12 BITS
70
10 BITS
8 BITS
45
20
1
10
100
1000
10000
INPUT FREQUENCY (MHz)
clock jitter, and quantization noise.
The earlier example showed that a clock with 350-fs jitter would
approached 35 MHz. However, when the effects of quantization
noise, input frequency, and input amplitude are considered, input
frequencies as low as 10 MHz should be of concern. Likewise,
100 fs of jitter on the clock will also cause SNR degradation at
frequencies lower than 100 MHz.
2
Now that the basics of jitter have been reviewed, we can consider
sources of jitter. Anything that can modulate the edge transition
of the ADC’s clock will introduce or affect jitter. These include
crosstalk, EMI (electromagnetic interference), ground effects,
and supply noise.
Crosstalk-induced jitter can occur in any two adjacent traces. If
one trace carries a signal, and a nearby parallel trace carries a
varying current, a voltage will be induced in the signal trace; if
it is a clock signal, the time at which the clock edge occurs will
be modulated.
Jitter can also be induced by EMI radiation on sensitive signal
traces. EMI is produced by switching power supplies, high-voltage
power lines, RF signals, and other similar sources. EMI produces
similar effects to crosstalk by means of electrical or magnetic
coupling that modulates the signal or clock timing.
Figure 5 illustrates the effects of electromagnetic interference on
SNR. The blue curve represents the baseline SNR-vs.-frequency
of the AD9446, with an external clock and a linear power supply.
The clock is not otherwise attached to the evaluation board in any
way. The red curve shows the degradation that occurs when the
same clock circuit is fixed or soldered onto the board, which is
substantial improvement in the converter’s performance can be
obtained if the oscillator is choked and filtered off this supply.
85
BASELINE—OSCILLATOR OFF BOARD
80
OSCILLATOR ON BOARD—CHOKED
75
70
OSCILLATOR ON BOARD
65
60
0
20
40
60
80
100
120
140
160
180
FREQUENCY (MHz)
Figure 5. Converter performance vs. oscillator supply
configuration and frequency.
Bouncing grounds due to switching currents or improper ground
connections can also bring about jitter. Switching currents can
become large when many gates are switching at the same time.
This can induce current spikes on power and ground planes,
level-shifting the threshold voltages on clock-circuit or analoginput signals. Consider the following example:
CLOSER TO 10MHz
95 16 BITS
Keeping the Jitter Out
SNRFS (dBFS)
If the analog input frequency is near or to the right of an intersection,
however, the frequency or resolution must be reduced—or the
jitter specification must be improved. Thus, as the jitter intervals
increase, the point where the SNR performance is dominated by
the clock system jitter occurs at ever-lower frequencies.
Assume a gate output has a 10-pF combined load from the
PCB trace and input of the receiver gate. When the gate
switches, 10 mA of dynamic current can flow into or out of
each output. [10 mA is derived from 10 pF × 1 V/ns, the
typical slew rate of a CMOS gate (I = C dV/dt).] A midscale
transition could thus account for 120 mA of dynamic current
if 12 gates are switching simultaneously. This would cause
a large current spike to be drawn through the supply leads,
one of which may be ground. The transient voltage drop
(bounce) due to the lead resistance will affect all circuits
that rely on it to be at ground potential.
To diminish the jitter caused by these sources, good layout
practices and proper circuit partitioning should be employed. It
is essential to restrict analog circuits and digital circuits to their
respective domains! This principle should be observed on every
Analog Dialogue 42-02, February (2008)
layer to ensure good isolation. It is important to understand how
return currents flow relative to their source, and to avoid any
encroachments or crossovers between analog and digital circuitry.
away from other circuitry and traces that can influence them in
undesired ways.
85
SNRFS (dBFS)
Improving Jitter Means Improving Slew
Now that the basics of jitter and its possible deleterious influences
have been covered, one might ask, “How do I make improvements
to my system clock or clock circuit so as to reduce jitter?”
Recalling the initial discussion, jitter or noise can only corrupt
the ADC’s timing when present during the transition or threshold
period of the clock, shown in Figure 6. Making this edge (and
hence the threshold period) faster by increasing the slew rate will
inevitably lessen the amount of time that noise can be present
during the threshold period and effectively lessen the amount of
rms (root-mean-square) jitter introduced to the system.
THRESHOLD
RANGE
Figure 6. Expanded view of the threshold/transition region
of a differential clock.
Keep in mind that increased slew rate doesn’t affect the original
signal quality, only the transition time through the threshold
region. To confirm this statement, refer to Figure 2b. Notice that
with this faster signal swing less time is spent in the transition
region. Figure 7 illustrates the inverse relationship between jitter
and slew rate. Relating this to the earlier example, a 12-bit ADC
requires a slew rate of 1 V/ns.
200
RMS JITTER (fs)
VENDOR 1
70
VENDOR 2
VENDOR 3
65
55
VENDOR 4
0
20
40
60
80
100
120
140
160
180
FREQUENCY (MHz)
Figure 8. AD9446-80 performance is affected by the
choice of oscillator source.
Typically, a custom high-performance clock oscillator is used to
characterize the baseline performance achieved by Analog Devices
ADCs (blue trace). Not all users of these high-speed converters
can afford the cost or space required by a high-performance,
oven-controlled, low-jitter oscillator, however, but available costeffective oscillators can achieve reasonable performance, even
performance with some affordable devices.
An important point is that care should be taken when selecting
an “off-the-shelf” oscillator, since oscillator vendors do not all
tend to specify or measure jitter in the same way. A pragmatic
way to determine which oscillator is best for the particular
application is to collect a handful and test them in the system
directly. By making this choice the only variable, a prediction of
performance can be made (assuming that the oscillator vendor
maintains reasonable standards of quality control). Better yet is
to contact the oscillator manufacturer to obtain jitter- or phasenoise data, and get suggestions as to how to best terminate the
device. Improper oscillator termination can seriously degrade the
converter’s spurious-free dynamic range (SFDR).
Further Improvements
If the best oscillator available, based on price and performance,
is still not adequate, one might consider using frequency
division and/or filtering. Equation 4 describes the output of
a sine-wave oscillator:
(4)
Two parameters affect the slew rate—signal frequency ( f ) and
amplitude (A). Increasing either of these will increase the slew
rate and reduce the system clock jitter to a more desirable number.
It is generally easier to increase the clock frequency. Frequency
division will then be used to produce the desired converter clock
rate, as well as to feed the other stages in the system clock tree.
100
50
0
75
60
150
BASELINE
80
0
2
4
6
Frequency dividers do add cost in terms of circuit components
and power requirements. They also add jitter. Each active
component added to the clock signal chain will increase the
total jitter.
INPUT SLEW RATE (V/ns)
Figure 7. RMS jitter vs. slew rate.
Thus, minimizing jitter means improving the slew rate of the clock
edge. One way this can be done is to improve the clock source
itself. Figure 8 compares a number of different “off-the-shelf”
oscillators when used as a clock source for one of ADI’s highest
Analog Dialogue 42-02, February (2008)
(5)
When using a divider, all the relevant specifications must be
considered. Typical among ADI’s clock-divider products is the
to having the dividing function built right in, features such as clock
distribution and duty-cycle control are also available.
3
It is worth noting that clock dividers must contribute, however
minimally, to the overall jitter on an absolute basis; but because of
the frequency reduction they provide, their output jitter becomes
a smaller fraction of the output period, and thus introduces less
error. For example, if a 100-MHz clock source and other members
of the chain contribute 800 fs of jitter (about 12.5% of the 10-ns
period), and a clock divider reduces the frequency to 10 MHz,
while introducing 250 fs of jitter, the resulting 840 fs of jitter is
less than 1% of the 100-ns output period.
Reducing Phase Noise
As Equation 5 indicates, total jitter is the root-sum-square (RSS)
of the jitter from the clock cleanup circuitry, as well as the jitter
in the source and any other intervening components. Thus, if
the divider circuit is driven by an extremely noisy source, the full
potential of the divider circuit may not be fully realized, simply
because the largest jitter term dominates the equation. In this
situation, consider using a passive, narrow-band filter between the
clock source and the divider circuit.
To illustrate the advantages of filtering, consider a source having
a jitter specification of 800 fs. If a clock divider circuit is placed
between the source and the converter, the jitter can be reduced to
roughly 500 fs even though the divider circuit is capable of much
better performance. However, by placing a 5% LC band-pass filter
between the source and divider circuit the jitter can be reduced
to 250 fs. (See Figure 9).
A = 10log10 (A1+ A2 + A3 + A4)
RMS JITTER (SECONDS) Ƽ
A2
10k
A3
A4
1M
10M
FREQUENCY OFFSET (Hz)
100M
1G
Figure 10. Calculating jitter from phase noise.
Consider now a source with 800-fs jitter. Plotting the phase
noise of the source (Figure 11) makes it easy to determine where
in the frequency domain most of the jitter is coming from.
In the case of the clock with 800-fs jitter, it can be seen that
the dominant part of the jitter in the spectrum is wideband.
Therefore, emphasis in reducing wideband noise is paramount
in sampled-type systems.
–100
–110
–120
–130
–140
–150
–160
38fs
46fs
85fs
787fs
–170
0.01
0.1
1
10
100
Figure 11a. Phase-noise plot of an 800-fs source.
–100
600
NOISY SOURCE
LIMITS PERFORMANCE
500
–110
–120
400
300
MATCHES THE
DATA SHEET
200
SOURCE
ONLY
WITH
DIVIDER
WITH DIVIDER
AND FILTER
–130
(dBc)
JITTER (fs)
2Ãfo
PERCENT OF FUNDAMENTAL FREQUENCY
700
–140
–150
–160
SMGU
Figure 9. Jitter reduction with clock division and filtering.
In order to understand how a filter can improve the jitter of a
sinusoidal source, it can be useful to think of jitter in the frequency
domain and estimate its value from a phase-noise plot. Although
the calculation is straightforward and provides a good method of
comparison, it doesn’t take into account nonlinear factors such
as slew rate. As a result, this model will often predict more jitter
than is actually present.
To perform the calculation, the phase-noise plot is divided into
frequency regions, and the integrated noise power of each region is
calculated, as shown in Figure 10. This permits identification of the
jitter contribution from each region, as well as the total jitter of the
source (by RSS summation). For these equations, f0 is the carrier
frequency. The integrated phase noise is multiplied by the square
root of 2 because the plot represents one of the two sidebands.
4
100k
–180
0.001
800
0
2 ò 10A/10
INTEGRATE TO Ƽ2fo = 200MHz
900
100
2 ò 10A/10
fo = OSCILLATOR FREQUENCY (100MHz)
A1
(dBc)
As a consequence of Equation 5, since the largest contributor
dominates the overall jitter, the maximum jitter of the clock source
should be no more than one-third of the largest contributor, but
not necessarily a great deal less. The actual choices depend on the
application’s performance requirements—such as for SNR over
a given frequency range—the characteristics of available system
components, and the usual limitations of size and cost.
A = AREA = INTEGRATED PHASE NOISE POWER (dBc)
–170
–180
0.001
38fs
46fs
0.01
85fs
0.1
241fs
1
10
100
PERCENT OF FUNDAMENTAL FREQUENCY
Figure 11b. Phase noise of the 800-fs source with a band-pass
LC multipole filter with a 5% pass band applied.
The use of a simple band-pass LC multipole filter with a 5% pass
band (5% LCBP) on the output of the clock source can greatly
improve the performance, as shown in Figure 11b. Note the
improvement from 800 fs to less than 300 fs. That corresponds
to an SNR improvement of over 12 dB.
Five-percent LCBP filters can be easily obtained, but they can
be big and expensive. An alternative is to use a crystal-type filter.
Figure 12 shows the improvement in phase noise from 800 fs to
less than 100 fs. This represents an additional 3-dB improvement
over the 5% LCBP filter’s 12 dB, for a total of 15 dB!
Analog Dialogue 42-02, February (2008)
Finally, some crystal filters may require external components
for impedance matching. Filters can do the job, but they require
additional parts, tricky matching, and extra cost.
–100
–110
–120
A quick summary of divider and filter solutions to improve the
slew rate is shown in Table 1.
(dBc)
–130
It is desirable to clip the signal before it approaches the ADC
clock inputs using back-to-back Schottky diodes. This allows the
source amplitude to be increased, thus increasing the slew rate,
yet keeping the clock’s amplitude at a level compatible with the
converter’s clock inputs.
–140
–150
–160
–170
38fs
16fs
4fs
–180
0.001
0.01
44fs
0.1
1
10
100
PERCENT OF FUNDAMENTAL FREQUENCY
Figure 12. Phase-noise of an 800-fs source with a crystal filter.
To demonstrate the effectiveness of cascading crystal filters
with a noisy source, an experiment was performed using an
old benchtop pulse generator to clock the 16-bit, 100-MHz
than 4 ps of jitter, resulting in an SNR degradation of over
30 dB. With the crystal filter applied the calculated jitter was
close to 50 fs, providing an improvement in SNR approaching
data-sheet typical performance (Figure 13).
0
LECROY UNFILTERED
LECROY CRYSTAL FILTERED
–20
POWER (dBc)
–40
If the clocking system is small or the last stage has short trace
lengths, consider using a transformer in concert with the clipping
diode. The transformer is passive and won’t add jitter to the overall
clock signal. Transformers can also be used to provide gain for
the oscillator’s signal voltage, increasing the A term (amplitude)
in Equation 4. Lastly, transformers inherently provide pass-band
filtering. Those with gain (1:2 or 1:4 impedance ratios) have
narrower bandwidth, providing even better filtering of the clock
signal. Transformers can also convert that single-ended signal
into a differential signal—common and highly recommended in
Keep in mind that not all diodes will perform equally well
(Figure 14). The “baseline” condition is the performance of
the best-performing diode, relative to all the other diodes in
this test batch, measured under the same conditions. Read the
specifications carefully and pay particularly close attention to the
dynamic resistance and total capacitance specifications. Diodes
with low R and C values can improve clipping speed.
82.5
–60
82.0
47dBFS SNR
81.5
–80
VENDOR 1
VENDOR 4
VENDOR 3
NO DIODE
VENDOR 2
81.0
80.5
SNR (dBFS)
–100
–120
–140
BASELINE
78dBFS SNR
0
5
10
15
20
25
30
35
40
45
FREQUENCY (MHz)
Figure 13. Crystal filters are helpful—even with noisy sources.
Crystal filters, with their very narrow pass-band region—usually
<1%—can reduce jitter from many sources to less than 100 fs,
but they add expense and are bulkier than active filters. It is also
worth noting that crystal filters have a limited input/output range
of 5 dBm to 10 dBm. Pushing them beyond their specified range
80.0
79.5
79.0
78.5
78.0
77.5
77.0
76.5
0
20
40
60
80
100
120
140
160
180
Figure 14. AD9446-80 performance is affected by choice
of clipping Schottky diodes.
Table 1. Summary of Divider and Filter Trade-Offs
Divider
5% LCBP Filter
Pros
• Low cost (\$5 to \$20).
• High slew rate at low frequencies.
• Can vary the duty cycle.
• Clock distribution chips = more
outputs available.
• With proper source, can achieve <100 fs jitter. • Ultralow jitter for all sources.
• Very small (also available 50 Ω matched).
• High max input power.
Cons
• Best case jitter ~ 200 fs to 250 fs.
• Encode limited to the pass-band frequency.
• Duty cycle limited to 50%.
• More expensive than dividers (~\$300).
• Encode limited to the pass-band frequency.
• Duty cycle limited to 50%.
• Custom cost results in a 50% cost increase over LCBP.
• Dividers can make matters worse.
• Max output power is limited by filter
insertion loss and max specified input power.
• Max output power is limited by filter insertion loss and
max specified input power.
• Request high max power when ordering custom filters.
Don’t • For best performance, place
Forget
a band-pass filter before the
divider.
Analog Dialogue 42-02, February (2008)
Crystal Filter
5
platform; the only change was in the source of back-to-back diodes.
The circuit used for this evaluation is shown in Figure 15.
800
600
0.1¿F
0.1¿F
ANALOG
INPUT
DIGITAL
OUTPUT
CLK+ CLK–
SCHOTTKY
DIODES:
HSM2812
JITTER (fs)
100±
50±
FILTER ONLY
DIVIDER/FILTER
Mini-Circuits®
0.1¿F
0.1¿F
XFMR
CLK+
RAW SOURCE
700
500
400
300
200
Figure 15. AD9446 clock circuit for data of Figure 14.
100
Jitter Reduction in Clock Hardware Interfaces
0
There are many circuits and solutions that can be used when
interfacing to the ADC’s clock input pins. However, a review of
Equation 5
reminds us that a valid expectation is that each active component
(oscillator source, driver or fanout gate, divider, etc.) in the signal
chain will increase the total amount of jitter presented to the
ADC’s clock input pins. Figure 16 shows that the addition of two
gates, each contributing 700 fs of jitter, to a source with 300 fs
of jitter can degrade resolution from about 12 bits to less than
10 bits at 140 MHz.
SOURCE DRIVER
0.3ps rms 0.7ps rms
SNR = 20 × log
1
2Ãfsignaltjitter
for fsignal = 140MHz
OUTPUT =
(0.32 + 0.72) = 0.76ps rms
SNR = 63.5dB
OUTPUT =
(0.32 + 0.72) = 0.76ps rms
SNR = 63.5dB
SOURCE DRIVER
0.7ps rms 0.3ps rms
SINE SOURCE A
SINE SOURCE B
Figure 17. Gates will add jitter.
Another common approach leads to manifest inability to achieve
data sheet performance. A flexible gate driver can be achieved
fairly simply using an FPGA (often with a digital clock manager—
DCM, which provides clock division). However, as Figure 18
shows, this approach has huge costs in degradation of SNR using
ENOB, for example. The high-performance oscillator establishes
the baseline SNR performance over a range of frequencies, as
shown by the red curve. The green curve shows the difference
in performance using the same clock, but with an FPGA as the
gate driver between the high-performance oscillator and the
converter. At 40 MHz, the FPGA reduces the SNR to 52 dB
(8.7-bit performance) while the DCM contributes an additional
8-dB (1.3-bit) reduction of SNR. That performance difference is
pretty alarming with 29-dB degradation in SNR, which means
that the FPGA driver gate alone has an additive jitter of roughly
10 ps using Equation 1!
90
DRIVER
SOURCE DRIVER
0.3ps rms 0.7ps rms 0.7ps rms
80
(0.32 + 0.72 + 0.72) = 1.03ps rms
SNR = 60.9dB
9.8 BITS!
Figure 16. Multiple driver gates increase jitter and reduce SNR.
Thus, minimizing the number of components in the clock signal
chain can help keep the total RSS jitter low.
The type of clock gates chosen is also worth noting. Simple logic
gates are probably not the best choice when seeking to obtain good
read the data sheets of candidate devices and understand the
pertinent specifications, such as jitter and skew. This is especially
important when they are to work with sources that have extremely
low jitter. For example, in Figure 17, Source A has 800 fs of jitter
and Source B has 125 fs of jitter. With a crystal filter, the respective
jitter levels can be reduced to 175 fs and 60 fs. However, a divider
(or a gate with comparable jitter specifications) can increase the
jitter to above 200 fs in both cases. This underscores the fact that
proper selection and placement of clock drivers in the clock signal
chain is important.
6
70
SNR (dBFS)
OUTPUT =
60
50
40
30
BASELINE OSCILLATOR
BASELINE OSCILLATOR THROUGH FPGA
BASELINE OSCILLATOR THROUGH FPGA WITH DCM
VENDOR 1 BASELINE
VENDOR 1 THROUGH FPGA
VENDOR 1 THROUGH FPGA WITH DCM
20
10
0
20
40
60
80
100
120
140
160
180
FREQUENCY (MHz)
Figure 18. AD9446-80 performance is affected by FPGA
gate drive circuits.
Choosing the best clock driver gate can be difficult. Table 2 gives a
rough comparison of the additive jitter of a number of driver gates
on the market. The suggestions on the lower half of the table may
Analog Dialogue 42-02, February (2008)
Table 2. Summary of Clock-Driver Gates and
1
2
Logic Family
FPGA
33 ps to 50 ps (driver gates
only, not including internal
gates of DLL/PLL)1
74LS00
4.94 ps2
74HCT00
2.2 ps2
74ACT00
0.99 ps2
MC100EL16 PECL
0.7 ps1
0.22 ps1
8. HSMS-2812 Data Sheet.
9. Kester, Walt. Analog-Digital Conversion. Analog Devices
(2004). Section 2.3, p. 2.72, Fig. 2.81.
10.K&L Filter Data Sheets.
11.Mercer, Doug, Steve Reine, and David Carr. Application
Note AN-642, “Coupling a Single-Ended Clock Source to
the Differential Clock Input of Third-Generation TxDAC
and TxDAC+ Products.”
12.Monolithic Crystal Filters (used for majority of evaluations)
• QuartzCom (www.quartzcom.com).
13.Smith, Paul. Application Note AN-741. “Little-Known
Characteristics of Phase Noise.”
NBSG16, Reduced Swing
ECL (0.4 V)
0.2 ps1
ACKNOWLEDGEMENTS
Driver Family
0.1 ps1
Manufacturer’s specification.
CONCLUSION
It is critical to understand the entire clock system in order to
achieve the best possible performance of the converter. Figure 3
and Equations 1 and 2 are helpful guides to clock requirements for
applying either a jitter-limited ADC having very high resolution or
below the intersection of these lines, one must consider a clock
source and associated circuitry with reduced jitter.
Decreasing the jitter of the system clock circuit can be achieved in
many ways, including improving the clock source, filtering, and/
or frequency-dividing, as well as proper choice of clock circuit
hardware. Remember to pay attention to the slew rate of the clock.
This will determine the amount of noise that can corrupt the
converter during the transition time. Minimizing this transition
time can improve the converter’s performance.
Use only necessary circuitry to drive and distribute the clock
because each component in the signal chain will increase the
overall jitter. Finally, don’t use “cheap” hardware gates; their
performance is likely to be disappointing. One can’t expect
championship performance from a \$70,000 car outfitted with
\$20 tires.
3. Barrow, Jeff. “Reducing Ground Bounce in DC-to-DC
Converters—Some Grounding Essentials.” Analog Dialogue,
vol. 41, no. 2 (2007). pp. 3–7.
4. Brannon, Brad. Application Note AN-756, “Sampled
Systems and the Effects of Clock Phase Noise and Jitter.”
5. Brannon, Brad and Allen Barlow. Application Note AN-501,
“Aperture Uncertainty and ADC System Performance.”
6. Curtin, Mike, and Paul O’Brien. “Phase-Locked Loops
for High-Frequency Receivers and Transmitters—Part 2.”
Analog Dialogue, vol. 33, no. 1 (1999). pp. 13–17.
7. Custom Discrete Crystal Filters
• Filtronetics (www.filtro.net)
• Anatech Electronics, Inc. (www.anatechelectronics.com).
Analog Dialogue 42-02, February (2008)
The authors would like to acknowledge and thank Yi Wang,
Brad Brannon, and Walt Kester for their help and the benefits of
their experience in this and related fields of study—and Ben Beasley
for data collection in the laboratory.
REFERENCES—VALID AS OF FEBRUARY 2008
1
2
THE AUTHORS
Rob Reeder [[email protected]] is
a senior converter-applications engineer
working in the High-Speed Converter
group in Greensboro. He has published
numerous papers on conver ters and
converter testing. Formerly, Rob was a
group, designing converter modules for
space- and military applications. Rob
received his MSEE and BSEE in 1998 and
1996, respectively, from Northern Illinois University in DeKalb,
IL. In his spare time he enjoys mixing music, art-airbrushing, and
building Hot Wheels® race tracks with his two boys.
Wayne Green [[email protected]]
works as a test development engineer
for the High-Speed Converter group
in Greensboro. Wayne has developed
test solut ions for d ig it a l-to - a na log
converters, high-speed comparators, pin
drivers, and high-speed analog-to-digital
converters. Currently his development
work involves test solutions for ADI’s
family of multichannel, multiresolution,
Robert Shillito [[email protected]]
is a product-test development engineer
with the High-Speed Converters Group
in Greensboro. In June 2006, he joined