Chapter 11: Overvoltage Effects on Analog Integrated Circuits

Chapter 11: Overvoltage Effects on Analog Integrated Circuits
OVERVOLTAGE, EMI AND RFI
CHAPTER 11: OVERVOLTAGE EFFECTS
ON ANALOG INTEGRATED CIRCUITS
SECTION 11.1: OVERVOLTAGE EFFECTS
AMPLIFIER INPUT STAGE OVERVOLTAGE
AMPLIFIER OUTPUT VOLTAGE PHASE REVERSAL
SECTION 11.2: ELECTROSTATIC DISCHARGE (ESD)
UNDERSTANDING AND PROTECTING INTEGRATED CIRCUITS
FROM ELECTROSTATIC DISCHARGE (ESD)
SECTION 11.3: EMI/RFI CONSIDERATIONS
A PRIMER ON EMI REGULATIONS
COMMERCIAL EQUIPMENT
MILITARY EQUIPMENT
MEDICAL EQUIPMENT
INDUSTRIAL AND PROCESS-CONTROL EQUIPMENT
AUTOMOTIVE EQUIPMENT
EMC REGULATIONS' IMPACT ON DESIGN
PASSIVE COMPONENTS: YOUR ARSENAL AGAINST EMI
RADIO FREQUENCY INTERFERENCE (RFI)
GROUND REDUCES EFFECTIVENESS
SOLUTIONS FOR POWER-LINE DISTURBANCES
PRINTED CIRCUIT BOARD DESIGN FOR EMI PROTECTION
A REVIEW OF SHIELDING CONCEPTS
GENERAL POINTS ON CABLES AND SHIELDS
EMI TROUBLE SHOOTING PHILOSOPHY
REFERENCES
11.1
11.1
11.4
11.11
11.11
11.23
11.23
11.23
11.24
11.24
11.24
11.25
11.25
11.25
11.30
11.33
11.35
11.37
11.42
11.47
11.49
11.50
BASIC LINEAR DESIGN
OVERVOLTAGE EFFECTS ON ANALOG INTEGRATED CIRCUITS
CHAPTER 11: OVERVOLTAGE EFFECTS ON
ANALOG INTEGRATED CIRCUITS
SECTION 11.1: OVERVOLTAGE EFFECTS
One of the most commonly asked applications questions is: “What happens if external
voltages are applied to an analog integrated circuit with the supplies turned off?” This
question describes situations that can take on many different forms: from lightning strikes
on cables which propagate very high transient voltages into signal conditioning circuits,
to walking across a carpet and then touching a printed circuit board full of sensitive
precision circuits. Regardless of the situation, the general issue is the effect of
overvoltage stress (and, in some cases, abuse) on analog integrated circuits. The
discussion which follows will be limited in general to operational amplifiers, because it is
these devices that most often interface to the outside world. The principles developed
here can and should be applied to all analog integrated circuits which are required to
condition or digitize analog waveforms. These devices include (but are not limited to)
instrumentation amplifiers, analog comparators, sample-and-hold amplifiers, analog
switches and multiplexers, and analog-to-digital converters.
Amplifier Input Stage Overvoltage
In real world signal conditioning, sensors are often used in hostile environments where
faults can and do occur. When these faults take place, signal conditioning circuitry can be
exposed to large voltages which exceed the power supplies. The likelihood for damage is
quite high, even though the components’ power supplies may be turned on. Published
specifications for operational amplifier absolute maximum ratings state that applied input
signal levels should never exceed the power supplies by more than 0.3 V or, in some
devices, 0.7 V. Exceeding these levels exposes amplifier input stages to potentially
destructive fault currents which flow through internal metal traces and parasitic P-N
junctions to the supplies. Without some type of current limiting, unprotected input
differential pairs (BJTs or FETs) can be destroyed in a matter of microseconds. There
are, however, some devices with built-in circuitry that can provide protection beyond the
supply voltages, but in general, absolute maximum ratings must still be observed.
Although more recent vintage operational amplifiers designed for single-supply or railto-rail operation are now including information with regard to input stage overvoltage
effects, there are very many amplifiers available today without such information provided
by the manufacturer. In those cases, the circuit designer using these components must
ascertain the input stage current-voltage characteristic of the device in question before
11.1
BASIC LINEAR DESIGN
steps can be taken to protect it. All amplifiers will conduct current to the
positive/negative supply, provided the applied input voltage exceeds some internal
threshold. This threshold is device dependent, and can range from 0.7 V to 30 V,
depending on the internal construction of the input stage. Regardless of the threshold
level, externally generated fault currents should be limited to no more than ±5 mA.
„ INPUT SHOULD NOT EXCEED ABSOLUTE MAXIMUM RATINGS
(Usually Specified With Respect to Supply Voltages)
„ A Common Specification Requires the Input Signal < |Vs| ± 0.3 V
„ Input Voltage Should be Held Near Zero in the Absence of Supplies
„ Input Stage Conduction Current Needs to be Limited (Rule of Thumb: ≤ 5 mA)
„ Avoid Reverse Bias Junction Breakdown in Input Stage Base - Emitter
Junctions
„ Differential and Common-Mode Ratings may Differ
„ No Two Amplifiers are exactly the Same
„ Some Op Amps Contain Input Protection (Voltage Clamps, Current Limits, or
Both), but Absolute Maximum Ratings Must Still be Observed
Figure 11.1: Input Stage Overvoltage
Many factors contribute to the current-voltage characteristic of an amplifier’s input stage:
internal differential clamping diodes, current-limiting series resistances, substrate
potential connections, and differential input stage topologies (BJTs or FETs). Input
protection diodes used as differential input clamps are typically constructed from the
base-emitter junctions of NPN transistors. These diodes usually form a parasitic P-N
junction to the negative supply when the applied input voltage exceeds the negative
supply. Current-limiting series resistances used in the input stages of operational
amplifiers can be fabricated from three types of material: N- or P-type diffusions,
polysilicon, or thin-films (SiCr, for example). Polysilicon and thin-film resistors are
fabricated over thin layers of oxide which provide an insulating barrier to the substrate;
as such, they do not exhibit any parasitic P-N junctions to either supply. Diffused
resistors, on the other hand, exhibit P-N junctions to the supplies because they are
constructed from either P- or N-type diffusion regions. The substrate potential of the
amplifier is the most critical component, for it will determine the sensitivity of an
amplifier’s input current-voltage characteristic to supply voltage.
The configuration of the amplifier’s input stage also plays a large role in the currentvoltage characteristic of the amplifier. Input differential pairs of operational amplifiers
are constructed from either bipolar transistors (NPN or PNP) or field-effect transistors
(junction or MOS, N- or P-channel). While the bipolar input differential pairs do not have
any direct path to either supply, FET differential pairs do. For example, an N-channel
JFET forms a parasitic P-N junction between its backgate and the P-substrate that
11.2
OVERVOLTAGE EFFECTS ON ANALOG INTEGRATED CIRCUITS
energizes when VIN + 0.7 V < VNEG. As mentioned previously, many manufacturers of
analog integrated circuits do not provide any details with regard to the behavior of the
device’s input structure. Either simplified schematics are not provided or, if they are
shown, the behavior of the input stage under an overvoltage condition is omitted.
Therefore, other measures must be taken in order to identify the conduction paths.
A standard transistor curve tracer can be configured to determine the current-voltage
characteristic of any amplifier regardless of input circuit topology. Both amplifier supply
pins are connected to ground, and the collector output drive is connected to one of the
amplifier’s inputs. The curve tracer applies a DC ramp voltage and measures the current
flowing through the input stage. In the event that a transistor curve tracer is not available,
a DC voltage source and a multimeter can be substituted for the curve tracer. A 10 kΩ
resistor should be used between the DC voltage source and the amplifier input for
additional protection. Ammeter readings from the multimeter at each applied DC voltage
will yield the same result as that produced by the curve tracer. Although either input can
be tested (both inputs should), it is recommended that the unused input is left open;
otherwise, additional junctions could come into play and would complicate matters
further. Evaluations of current feedback amplifier input stages are more difficult because
of the lack of symmetry between the inputs. As a result, both inputs should be
characterized for their individual current-voltage characteristics.
„ Junctions may be Forward Biased if the Current is Limited
„ In General a Safe Current Limit is 5mA
„ Reverse Bias Junction Breakdown is Damaging Regardless of the Current
Level
„ When in Doubt, Protect with External Diodes and Series Resistances
„ Curve Tracers Can be Used to Check the Overvoltage Characteristics of a
Device
„ Simplified Equivalent Circuits in Data Sheets do not tell the Entire Story!!!
Figure 11.2: Overvoltage Effects
Once the input current-voltage characteristic has been determined for the device in
question, the next step is to determine the minimum level of resistance required to limit
fault currents to ± 5 mA. Equation 11.1 illustrates the computation for Rs when the input
overvoltage level is known:
Rs =
VIN(MAX) − VSUP P LY
5 mA
Eq. 11-1
The worst case condition for overvoltage would be when the power supplies are initially
turned off or disconnected. In this case, VSUPPLY is equal to zero. For example, if the
11.3
BASIC LINEAR DESIGN
input overvoltage could reach 100 V under some type of fault condition, then the external
resistor should be no smaller than 20kΩ. Most operational amplifier applications only
require protection at one input; however, there are a few configurations (difference
amplifiers, for example) where both inputs can be subjected to overvoltage and both must
be protected. The need for protection on both inputs is much more common with
instrumentation amplifiers.
„ Sometimes Occurs in FET and Bipolar Input (Especially SingleSupply) Op Amps when Input Exceeds Common Mode Range
„ Does Not Harm Amplifier, but may be Disastrous in Servo
Systems!
„ Not Usually Specified on Data Sheet, so Amplifier Must be Checked
„ Easily Prevented:
BiFETs:
Add Appropriate Input Series Resistance
(Determined Empirically, Unless Provided in
Data Sheet)
Bipolars:
Use Schottky Diode Clamps to the Supply
Rails.
Figure 11.3: BEWARE OF AMPLIFIER OUTPUT PHASE REVERSAL
Amplifier Output Voltage Phase Reversal
Some operational amplifiers exhibit output voltage phase reversal when one or both of
their inputs exceeds their input common-mode voltage range. Phase reversal is usually
associated with JFET (n- or p-channel) input amplifiers, but some bipolar devices
(especially single-supply amplifiers operating as unity-gain followers) may also be
susceptible. In the vast majority of applications, output voltage phase reversal does not
harm the amplifier nor the circuit in which the amplifier is used. Although a number of
operational amplifiers suffer from phase reversal, it is rarely a problem in system design.
However, in servo loop applications, this effect can be quite hazardous. Fortunately, this
is only a temporary condition. Once the amplifier’s inputs return to within its normal
operating common-mode range, output voltage phase reversal ceases. It may still be
necessary to consult the amplifier manufacturer, since phase reversal information rarely
appears on device data sheets.
In BiFET operational amplifiers, phase reversal may be prevented by adding an
appropriate resistance in series with the amplifier’s input to limit the current. Bipolar
11.4
OVERVOLTAGE EFFECTS ON ANALOG INTEGRATED CIRCUITS
input devices can be protected by using a Schottky diode to clamp the input to within a
few hundred millivolts of the negative rail. For a complete description of the output
voltage phase reversal effect, please consult Reference 1.
VPOS
*
D3
*
B
E
R3
R4
B
D5
D7
E
5k Ω
IN+
V+
D8
Q1
D1
Q3
Q2
Q4
D2
IN5k Ω
*
D6
D4
R1
*
R2
VNEG
* D3 - D6: SUBSTRATE PNPs (COLLECTORS TO VNEG)
Figure 11.4: A Closer Look at the OP-X91 Input Stage
Reveals Additional Devices
Rail-to-rail operational amplifiers present a special class of problems to the integrated
circuit designer, because these types of devices should not exhibit any abnormal behavior
throughout the entire input common-mode range. In fact, it is desirable that devices used
in these applications also not exhibit any abnormal behavior if the applied input voltages
exceed the power supply range. One of the more recent vintage rail-to-rail input/output
operational amplifiers, the OPX91 family (the OP191, the OP291, and the OP491),
includes additional components that prevent overvoltage and damage to the device. As
shown in Figure 11.4, the input stage of the OPX91 devices use six diodes and two
resistors to clamp the input terminals to each other and to the supplies. D1 and D2 are
base-emitter NPN diodes which are used to protect the bases of Q 1- Q2 and Q3 - Q4
against avalanche breakdown when the applied differential input voltage to the device
exceeds 0.7 V. Diodes D3 - D6 are diodes formed from substrate PNP transistors that
clamp the applied input voltages on the OPX91 to the supply rails.
An interesting benefit from using substrate PNPs as clamp diodes is that their collectors
are connected to the negative supply; thus, when the applied input voltage exceeds either
supply rail, the diodes energize, and the fault currents are diverted directly to the supply
and not through or into the device’s input stage. There are also 5kΩ resistors in series
with each of the inputs to the OPX91 to limit the fault current through D1 and D2 when
11.5
BASIC LINEAR DESIGN
the differential input voltage exceeds 0.7 V. Note that these 5 kΩ resistors are p-type
diffusions placed inside an n-well, which is then connected to the positive supply. When
the applied input voltage exceeds the positive supply, some of the fault current generated
is also diverted to VPOS and away from the input stage. As a result of these measures,
the input overvoltage characteristic of the OPX91 is well behaved as shown in Figure
11.6. Note that the combination of the 5 kΩ resistors and clamp diodes safely limits the
input current to less than 2 mA, even when the inputs of the device exceed the supply
rails by 10 V.
IIN
2mA
1mA
-10V
-5V
5V
10V
VIN
-1mA
-2mA
Figure 11.5:
Internal 5 kΩ Resistors Plus Input Clamp Diodes Combine to
Protect OP-X91 Devices Against Overvoltage
As an added safety feature, an additional pair of diodes is used in the input stage across
Q3 and Q4 to prevent subsequent stages internal to the OPX91 from collapsing (that is,
forced into cutoff). If these stages were forced into cutoff, then the amplifier would
undergo output voltage phase reversal when the inputs exceeded the positive input
common mode voltage. An illustration of the diodes’ effectiveness is shown in
Figure 11.6. Here, the OPX91 family can safely handle a 20 Vp-p input signal on ±5 V
supplies without exhibiting any sign of output voltage phase reversal or other anomalous
behavior. With these amplifiers, no external clamping diodes are required.
11.6
OVERVOLTAGE EFFECTS ON ANALOG INTEGRATED CIRCUITS
+5V
VIN
20Vp-p
3
8
+
1/2
2 OP291
4
VOUT
VIN - 2.5V/DSIV
VOUT - 2V/DSIV
-5V
TIME - 200μs/DIV
TIME - 200μs/DIV
Figure 11.6: Addition of Two Clamp Diodes Protects OP-X91
Devices Against Output Phase Reversal
VPOS
RFB
D1
VIN
RS
VOUT
+
D2
Value for RS provided by manufacturer or determined empirically
RFB may be required for high bias current devices
D1 and D2 can be Schottky diodes (Check their capacitance and
leakage current first)
Figure 11.7: Generalized External Protection Schemes Against Input
Overvoltage Abuse and Output Voltage Phase Reversal
in Single Supply Op Amps
For those amplifiers where external protection is clearly required against both
overvoltage abuse and output phase reversal, a common technique is to use a series
11.7
BASIC LINEAR DESIGN
resistance, Rs, to limit fault current, and Schottky diodes to clamp the input signal to the
supplies, as shown in Figure 11.7.
The external input series resistance, Rs, will be provided by the manufacturer of the
amplifier, or determined empirically by the user with the method previously shown in
Figure 11.2 and Eq. 11.1. More often than not, the value of this resistor will provide
enough protection against output voltage phase reversal, as well as limiting the fault
current through the Schottky diodes.
It is evident that whenever resistance is added in series with an amplifier’s input, its
offset and noise performance will be affected. The effects of this series resistance on
circuit noise can be calculated using the following equation.
E n ,t ot a l =
(e n ,op a m p )2 + (e n ,R s ) 2 + (R s ⋅ i n ,op a m p )2
Eq. 11-2
The thermal noise of the resistor, the voltage noise due to amplifier noise current flowing
through the resistor, and the input noise voltage of the amplifier are added together (in
root-sum-square manner, since the noise voltages are uncorrelated) to determine the total
input noise and may be compared with the input voltage noise in the absence of the
protection resistor.
A protection resistor in series with an amplifier input will also produce a voltage drop
due to the amplifier bias current flowing through it. This drop appears as an increase in
the circuit offset voltage (and, if the bias current changes with temperature, offset drift).
In amplifiers where bias currents are approximately equal, a resistor in series with each
input will tend to balance the effect and reduce the error. The effects of this additional
series resistance on the circuit’s overall offset voltage can be calculated:
Vos ( t ot a l ) = Vos + I b R s
Eq. 11-3
For the case where RFB = Rs or where the source impedance levels are balanced, then
the total circuit offset voltage can be expressed as:
Vos(t ot a l) = Vos + I os R s
Eq. 11-4
To limit the additional noise of RFB, it can be shunted with a capacitor.
When using external clamp diodes to protect operational amplifier inputs, the effects of
diode junction capacitance and leakage current should be evaluated in the application.
Diode junction capacitance and Rs will add an additional pole in the signal path, and
diode leakage currents will double for every 10°C rise in ambient temperature. Therefore,
low leakage diodes should be used such that, at the highest ambient temperature for the
application, the total diode leakage current is less than one-tenth of the input bias current
for the device at that temperature. Another issue with regard to the use of Schottky diodes
is the change in their forward voltage drop as a function of temperature. These diodes do
11.8
OVERVOLTAGE EFFECTS ON ANALOG INTEGRATED CIRCUITS
not, in fact, limit the signal to ±0.3 V at all ambient temperatures, but if the Schottky
diodes are at the same temperature as the op amp, they will limit the voltage to a safe
level, even if they do not limit it at all times to within the data sheet rating. This is true if
over-voltage is only possible at turn-on, when the diodes and the op amp will always be
at the same temperature. If the op amp is warm when it is repowered, however, steps
must be taken to ensure that diodes and op amp are at the same temperature.
11.9
BASIC LINEAR DESIGN
Notes:
11.10
ELECTROSTATIC DISCHARGE (ESD)
SECTION 11.2: ELECTROSATIC DISCHARGE (ESD)
Understanding and Protecting Integrated Circuits from Electrostatic
Discharge (ESD)
Integrated circuits can be damaged by the high voltages and high peak currents that can
be generated by electrostatic discharge. Precision analog circuits, which often feature
very low bias currents, are more susceptible to damage than common digital circuits,
because the traditional input-protection structures which protect against ESD damage
also increase input leakage.
The keys to eliminating ESD damage are: (1) awareness of the sources of ESD voltages,
and (2) understanding the simple handling steps that will discharge potential voltages
safely.
„
ESD (Electrostatic Discharge):
„ A single fast, high current transfer of electrostatic charge.
„ Direct contact between two objects at different potentials.
„ A high electrostatic field between two objects when they are in close
proximity.
„
ESD Failure Threshold:
„ The highest voltage level at which all pins on a device can be subjected
to ESD zaps without failing any 25°C data sheet limits.
Figure 11.8: ESD Definitions
The basic definitions relating to ESD are given in Figure 11.8. Notice that the ESD
Failure Threshold level relates to any of the IC data sheet limits, and not simply to a
catastrophic failure of the device. Also, the limits apply to each pin of the IC, not just to
the input and output pins.
The generation of static electricity caused by rubbing two substances together is called
the triboelectric effect. Static charge can be generated either by dissimilar materials (for
example, rubber-soled shoes moving across a rug) or by separating similar materials (for
example, pulling transparent tape off of a roll).
11.11
BASIC LINEAR DESIGN
A wide variety of common human activities can create high electrostatic charge. Some
examples are given in Figure 11.9. The values shown will occur with a fairly high
relative humidity. Low humidity, such as can occur indoors during cold weather, can
generate voltages 10 times (or more) greater than the values shown.
„
„ Person walks across a typical carpet.
1000 - 1500V generated
„
„ Person walks across a typical vinyl floor.
150 - 250V generated
„
„ Person handles instructions protected by clear plastic covers.
400 - 600V generated
„
„ Person handles polyethylene bags.
1000 - 1200V generated
„
„ Person pours polyurethane foam into a box.
1200 - 1500V generated
„
„ An IC slides down a grounded handler chute.
50 - 500V generated
„
„ An IC slides down an open conductive shipping tube.
25 - 250V generated
Note: Above values can occur in a high (≈60%) RH environment. For low
RH (≈30%), generated voltages can be >10 times those listed above!
Figure 11.9: Examples of ESD Generation
In an effort to standardize the testing and classification of integrated circuits for ESD
robustness, ESD models have been developed (Figure 11.10). These models attempt to
simulate the source of ESD voltage. The assumptions underlying the three commonlyused models are different, so results are not directly comparable.
„ Three Models:
1.
Human Body Model (HBM)
2.
Machine Model (MM)
3.
Charged Device Model (CDM)
„ Model Correlation:
„ Low - Assumptions are Different
Figure 11.10:
11.12
Modeling Electrostatic Potential
ELECTROSTATIC DISCHARGE (ESD)
The most-often encountered ESD model is the Human Body Model (HBM). This model
simulates the approximate resistance and capacitance of a human body with a simple RC
network. The capacitor is charged through a high voltage power supply (HVPS) and then
discharged (using a high voltage switch) through a series resistor. The RC values for
different individuals will, of course, vary. However, the HBM has been standardized by
MIL-STD-883 Method 3015 Electrostatic Discharge Sensitivity Classification, which
specifies R-C combinations of 1.5 kΩ and 100 pF. (R, C, and L values for all three ESD
models are shown in Figure 11.12.)
„ Human Body Model (HBM)
Simulates the discharge event that occurs when a person
charged to either a positive or negative potential touches
an IC at a different potential.
RLC:
R = 1.5kΩ,
L ≈ 0nH,
C =100pF
„ Machine Model (MM)
Non-real-world Japanese model based on worst-case HBM.
RLC:
R ≈ 0 Ω,
L ≈ 500nH,
C = 200pF
„ Charged Device Model (CDM)
Simulates the discharge that occurs when a pin on an IC,
charged to either a positive or negative potential,
contacts a conductive surface at a different (usually ground)
potential.
RLC:
R=1Ω
L ≈ 0nH,
C = 1 - 20pF
Figure 11.11:
ESD Models Applicable to ICs
The Machine Model (MM) is a worst-case Human Body Model. Rather than using an
average value for resistance and capacitance of the human body, the MM assumes a
worst-case value of 200 pF and 0Ω. The 0Ω output resistance of the MM is also intended
to simulate the discharge from a charged conductive object (for example, a charged DUT
socket on an automatic test system) to an IC pin, which is how the Machine Model
earned its name. However, the MM does not simulate many known real-world ESD
events. Rather, it models the ESD event resulting from a ideal voltage source (in other
words, with no resistance in the discharge path). EIAJ Specification ED-4701 Test
Method C-111 Condition A and ESD Association Specification S5.2 provide guidelines
for MM testing.
The Charged Device Model (CDM) originated at AT&T. This model differs from the
HBM and the MM, in that the source of the ESD energy is the IC itself. The CDM
assumes that the integrated circuit die, bond wires, and lead frame are charged to some
potential (usually positive with respect to ground). One or more of the IC pins then
contacts ground, and the stored charge rapidly discharges through the leadframe and
11.13
BASIC LINEAR DESIGN
bond wires. Typical examples of triboelectric charging followed by a CDM discharge
include:
1. An IC slides down a handler chute and then a corner pin contacts a grounded stop bar.
2. An IC slides down an open conductive shipping tube and then a corner pin contacts a
conductive surface.
R1
HBM
+
S1
1 0 MΩ
HVPS
C1
R1
MM
+
R2
0
1.5kΩ
L1
500nH
1 0 MΩ
C1
I
HBM
DUT
100pF
S1
HVPS
L1
t
R2
I
0
DUT
200pF
MM
CDM +
R1
1 0 MΩ
HVPS
-
DUT
S1
L1
R2
0
1Ω
I
t
CDM
CPKG
1-20pF
t
Figure 11.12: Schematic Representations of ESD Models
and Typical Discharge Waveforms
The basic concept of the CDM is different than the HBM and MM in two ways. First, the
CDM simulates a charged IC discharging to ground, while the HBM and MM both
simulate a charged source discharging into the IC. Thus, current flows out of the IC
during CDM testing, and into the IC during HBM and MM testing. The second difference
is that the capacitor in the CDM is the capacitance of the package, while the HBM and
MM use a fixed external capacitor.
Unlike the HBM and MM, CDM ESD thresholds may vary for the same die in different
packages. This occurs because the device under test (DUT) capacitance is a function of
the package. For example, the capacitance of an 8-pin package is different than the
capacitance of a 14-pin package. CDM capacitance values can vary from about 1 to
20 pF. The device capacitance is discharged through a 1 Ω resistor.
11.14
ELECTROSTATIC DISCHARGE (ESD)
Schematic representations of the three models are shown in Figure 11.13. Notice that C1
in the HBM and MM are external capacitors, while CPKG in the CDM is the internal
capacitance of the DUT.
The HBM discharge waveform is a predicable unipolar RC pulse, while the MM
discharge shows ringing because of the parasitic inductance in the discharge path
(typically 500 nH.). Ideally, the CDM waveform is also a single unipolar pulse, but the
parasitic inductance in series with the 1 Ω resistor slows the rise time and introduces
some ringing.
MODEL:
Simulate:
Origin:
Real World?
RC:
Rise Time
Ipeak at 400V
Energy:
Package
Dependent:
Standard:
*
**
HBM
Human Body
US Military, Late
1960s
Yes
MM
SOCKETED CDM
Machine
Japan, 1976
Charged Device
AT&T,1974
Generally
Yes
0 Ω, 200 pF
14 ns*
5.8 A*
High
1 Ω , 1 – 20 pF
400 ps**
2.1 A**
Low
No
No
Yes
MIL-STD-883
Method 3015
ESD Association
Std. S5.2;
EIAJ Std. ED-4701,
Method C-111
ESD Association
Draft Std. DS5.3
1.5 k Ω, 100 pF
<10 ns (6-9 ns typ )
0.27 A
Moderate
These values per ESD Association Std. S5.2.
EIAJ Std. ED-4701, Method C-111 includes no waveform specifications.
These values are for the direct charging (socketed) method.
Figure 11.13: Comparison of HBM, MM, AND CDM ESD Models
The significant features of each ESD model are summarized in Figure 11.13. The peak
currents shown for each model are based on a test voltage of 400 V. Peak current is
lowest for the HBM because of the relatively high discharge resistance. The CDM
discharge has low energy because device capacitance is only in the range of 1 pF to 20
pF, but peak current is high. The MM has the highest energy discharge, because it has the
highest capacitance value (Power = 0.5 CV2).
Figure 11.14 compares 400 V discharge waveforms of the CDM, MM, and HBM, with
the same current and time scales.
11.15
BASIC LINEAR DESIGN
AMPS
6
4
Socketed CDM
2
0
t
AMPS
6
4
2
MM
0
t
-2
AMPS
-4
HBM
2
0
HORIZONTAL SCALES
20 ns/div
t
Figure 11.14: Relative Comparison of 400 V HBM, MM, AND CDM Discharges
The CDM waveform corresponds to the shortest known real-world ESD event. The
waveform has a rise time of <1 ns, with the total duration of the CDM event only about
2 ns. The CDM waveform is essentially unipolar, although some ringing occurs at the
end of the pulse that results in small negative-going peaks. The very short duration of the
overall CDM event results in an overall discharge of relatively low energy, but peak
current is high.
The MM waveform consists of both positive- and negative-going sinusoidal peaks, with a
resonance frequency of 10 MHz to 15 MHz. The initial MM peak has a typical rise time
of 14 ns, and the total pulse duration is about 150 ns. The multiple high current, moderate
duration peaks of the MM result in an overall discharge energy that is by far the highest
of the three models for a given test voltage.
The risetime for the unipolar HBM waveform is typically 6 - 9 ns, and the waveform
decays exponentially towards 0 V with a fall time of approximately 150 ns. (Method
3015 requires a rise time of <10 ns and a delay time of 150 ns ± 20 ns, with decay time
defined as the time for the waveform to drop from 100% to 36.8% of peak current). The
peak current for the HBM is 400 V/1500 Ω or 0.267 A, which is much lower than is
11.16
ELECTROSTATIC DISCHARGE (ESD)
produced by 400 V CDM and MM events. However, the relatively long duration of the
total HBM event still results in an overall discharge of moderately high energy.
As previously noted, the MM waveform is bipolar while HBM and CDM waveforms are
primarily unipolar. However, HBM and CDM testing is done with both positive and
negative polarity pulses. Thus all three models stress the IC in both directions.
MIL-STD-883 Method 3015 classifies ICs for ESD failure threshold. The classification
limits, shown in Figure 11.15, are derived using the HBM shown in Figure 11.13.
Method 3015 also mandates a marking method to denote the ESD classification. All
military grade Class 1 and 2 devices have their packages marked with one or two “Δ”
symbols, respectively, while class 3 devices (with a failure threshold >4 kV) do not have
any ESD marking. Commercial and industrial grade IC packages may not be marked with
any ESD classification symbol.
HBM ESD CLASS
FAILURE THRESHOLD
1
<2 kV
2
2 kV – <4 kV
3
>4 kV
MARKING
None
Note: Commercial and Industrial ICs are not marked for ESD
Figure 11.15: Classifying and Marking ICs for ESD
Per MIL-883C, METHOD 3015
Notice that the Class 1 limit includes all devices which do not pass a 2 kV threshold.
However, a Class 1 rating does not imply that all devices within that class will pass
1,999 V. In any event, the emphasis must be placed on eliminating ESD exposure, not on
attempting to decide how much ESD exposure is ‘safe.’
„ ESD Failure Mechanisms:
‹ Dielectric or junction damage
‹ Surface charge accumulation
‹ Conductor fusing.
„ ESD Damage Can Cause:
‹ Increased leakage
‹ Reduced performance
‹ Functional failures of ICs.
„ ESD Damage is often Cumulative:
‹ For example, each ESD "zap" may increase junction damage until, finally,
the device fails.
Figure 11.16:
Understanding ESD Damage
11.17
BASIC LINEAR DESIGN
A detailed discussion of IC failure mechanisms is beyond the scope of this seminar, but
some typical ESD effects are shown in Figure 11.17.
„ ESD DAMAGE CANNOT BE “CURED”!
„ Circuits cannot be tweaked, nulled, adjusted, etc.,
to compensate for ESD damage.
ESD DAMAGE MUST BE PREVENTED!
Figure 11.17:
The Most Important Thing to Remember about ESD Damage
For the design engineer or technician, the most common manifestation of ESD damage is
a catastrophic failure of the IC. However, exposure to ESD can also cause increased
leakage or degrade other parameters. If a device appears to not meet a data sheet
specification during evaluation, the possibility of ESD damage should be considered.
Special care should be taken when breadboarding and evaluating ICs. The effects of ESD
damage can be cumulative, so repeated mishandling of a device can eventually cause a
failure. Inserting and removing ICs from a test socket, storing devices during evaluation,
and adding or removing external components on the breadboard should all be done while
observing proper ESD precautions. Again, if a device fails during a prototype system
development, repeated ESD stress may be the cause.
The key word to remember with respect to ESD is prevention. There is no way to un-do
ESD damage, or to compensate for its effects.
Two key elements in protecting circuits from
ESD damage are:
„
Recognizing ESD-sensitive products
„
Always handling ESD-sensitive products at a
grounded workstation
Figure 11.18: Preventing ESD Damage to ICs
Since ESD damage can not be undone, the only cure is prevention. Luckily, prevention is
a simple two-step process. The first step is recognizing ESD-sensitive products, and the
second step is understanding how to handle these products.
11.18
ELECTROSTATIC DISCHARGE (ESD)
All static sensitive devices are sealed in
protective packaging and marked with
special handling instructions
CAUTION
CAUTION
SENSITIVE ELECTRONIC DEVICES
SENSITIVE ELECTRONIC DEVICES
DO NOT SHIP OR STORE NEAR STRONG
ELECTROSTATIC, ELECTROMAGNETIC,
MAGNETIC, OR RADIOACTIVE FIELDS
DO NOT OPEN EXCEPT AT
APPROVED FIELD FORCE
PROTECTIVE WORK STATION
Figure 11.19 Recognizing ESD Sensitive Devices
All static sensitive devices are shipped in protective packaging. ICs are usually contained
in either conductive foam or in antistatic tubes. Either way, the container is then sealed in
a static-dissipative plastic bag. The sealed bag is marked with a distinctive sticker, such
as is shown in Figure 11.20, which outlines the appropriate handling procedures.
PERSONNEL
GROUND STRAP
ESD PROTECTIVE
TRAYS, SHUNTS,
ETC.
ESD PROTECTIVE
TABLE TOP
COMMON
GROUND
POINT
ESD PROTECTIVE
FLOOR OR MAT
BUILDING FLOOR
GROUND
Note: Conductive table top sheet resistance » 106 Ω /
Figure 11.20: Workstation for Handling ESD-Sensitive Devices
Once ESD-sensitive devices are identified, protection is easy. Obviously, keeping ICs in
their original protective packaging as long as possible is the first step. The second step is
to discharge potential ESD sources before damage to the IC can occur. The HBM
capacitance is only 100pF, so discharging a potentially dangerous voltage can be done
quickly and safely through a high impedance. Even with a source resistance of 10MΩ,
the 100pF will be discharged in less than 100milliseconds.
11.19
BASIC LINEAR DESIGN
The key component required for safe ESD handling is a workbench with a staticdissipative surface, as shown in Figure 11.20. This surface is connected to ground
through a 1 MΩ resistor, which dissipates static charge while protecting the user from
electrical shock hazards caused by ground faults. If existing bench tops are
nonconductive, a static-dissipative mat should be added, along with a discharge resistor.
„
Analog Devices is committed to helping our customers prevent ESD damage
by:
‹
‹
‹
Building products with the highest level of ESD protection commensurate
nsurate
with performance requirements
Protecting products from ESD during shipment
Helping customers to avoid ESD exposure during manufacture
Figure 11.21:
Analog Devices Commitment
Notice that the surface of the workbench has a moderately high sheet resistance. It is
neither necessary nor desirable to use a low-resistance surface (such as a sheet of copperclad PC board) for the work surface. Remember, the CDM assumes that a high peak
current will flow if a charged IC is discharged through a low impedance. This is precisely
what happens when a charged IC contacts a grounded copper clad board. When the same
charged IC is placed on the surface shown in Figure 11.20, however, the peak current is
not high enough to damage the device.
A conductive wrist strap is also recommended while handling ESD-sensitive devices.
The wrist strap ensures that normal tasks, such as peeling tape off of packages, will not
cause damage to ICs. Again, a 1 MΩ resistor, from the wrist strap to ground, is required
for safety.
When building prototype breadboards or assembling PC boards which contain ESDsensitive devices, all passive components should be inserted and soldered before the ICs.
This procedure minimizes the ESD exposure of the sensitive devices. The soldering iron
must, of course, have a grounded tip.
Protecting ICs from ESD requires the participation of both the IC manufacturer and the
customer. IC manufacturers have a vested interest in providing the highest possible level
of ESD protection for their products. IC circuit designers, process engineers, packaging
specialists and others are constantly looking for new and improved circuit designs,
processes, and packaging methods to withstand or shunt ESD energy (Figure 11.22)
11.20
ELECTROSTATIC DISCHARGE (ESD)
ANALOG DEVICES:
„
↓
↓
„
↓
Circuit Design and Fabrication Design and manufacture products with the h ighest level of ESD protection
consi stent with requir ed analog and digital performance.
Pack and Ship Pack in static diss ipative material. Mar k pack ages with ESD warning.
CUSTOMERS:
„
↓
„
↓
„
↓
↓
„
Incoming Inspection Inspe ct at grounded workstation. Minimize handling.
Inventory Control Store in origin al ESD -safe packag ing. Minimize handling.
Manufacturing Deliver to work area in original ESD -safe pack aging. Open packa ges only at
grounded workst ation. Pack age subassemblies in static dissipative packaging.
Pack and Ship Pack in static diss ipative material if r equired. R eplacement or optional
boards may require s pecial attention.
Figure 11.22 ESD Protection Requires a Partnership Between
the IC Supplier and the Customer
A complete ESD protection plan, however, requires more than building-ESD protection
into ICs. Users of ICs must also provide their employees with the necessary knowledge
of and training in ESD handling procedures.
11.21
BASIC LINEAR DESIGN
Notes:
11.22
EMI/RFI CONSIDERATIONS
SECTION 11.3: EMI/RFI CONSIDERATIONS
Electromagnetic interference (EMI) has become a hot topic in the last few years among
circuit designers and systems engineers. Although the subject matter and prior art have
been in existence for over the last 50 years or so, the advent of portable and highfrequency industrial and consumer electronics has provided a comfortable standard of
living for many EMI testing engineers, consultants, and publishers. With the help of EDN
Magazine and Kimmel Gerke Associates, this section will highlight general issues of
EMC (electromagnetic compatibility) to familiarize the system/circuit designer with this
subject and to illustrate proven techniques for protection against EMI.
A Primer on EMI Regulations
The intent of this section is to summarize the different types of electromagnetic
compatibility (EMC) regulations imposed on equipment manufacturers, both voluntary
and mandatory. Published EMC regulations apply at this time only to equipment and
systems, and not to components. Thus, EMI hardened equipment does not necessarily
imply that each of the components used (integrated circuits, especially) in the equipment
must also be EMI hardened.
Commercial Equipment
The two driving forces behind commercial EMI regulations are the FCC (Federal
Communications Commission) in the U. S. and the VDE (Verband Deutscher
Electrotechniker) in Germany. VDE regulations are more restrictive than the FCC’s with
regard to emissions and radiation. The European Community added immunity to RF,
electrostatic discharge, and power-line disturbances to the VDE regulations in 1996. In
Japan, commercial EMC regulations are covered under the VCCI (Voluntary Control
Council for Interference) standards and, implied by the name, are much looser than their
FCC and VDE counterparts.
All commercial EMI regulations primarily focus on radiated emissions, specifically to
protect nearby radio and television receivers, although both FCC and VDE standards are
less stringent with respect to conducted interference (by a factor of 10 over radiated
levels). The FCC Part 15 and VDE 0871 regulations group commercial equipment into
two classes: Class A, for all products intended for business environments; and Class B,
for all products used in residential applications. For example, Table 11.1 illustrates the
electric-field emission limits of commercial computer equipment for both FCC Part 15
and VDE 0871 compliance.
In addition to the already stringent VDE emission limits, the European Community EMC
standards (IEC and IEEE) require mandatory compliance to these additional EMI threats:
Immunity to RF fields, electrostatic discharge, and power-line disturbances. All
equipment/systems marketed in Europe must exhibit an immunity to RF field strengths of
11.23
BASIC LINEAR DESIGN
1-10V/m (IEC standard 801-3), electrostatic discharge (generated by human contact or
through material movement) in the range of 10 kV to 15 kV (IEC standard 801-2), and
power-line disturbances of 4kV EFTs (extremely fast transients, IEC standard 801-4) and
6 kV lightning surges (IEEE standard C62.41).
Radiated Emission Limits for Commercial Computer Equipment
Frequency (MHz)
30 to 88
88 to 216
216 to 1000
Class A
( at 3 m)
300 µV/m
500 µV/m
700 µV/m
Class B
(at 3 m)
100 µV/m
150 µV/m
200 µV/m
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Table 11.1
Military Equipment
The defining EMC specification for military equipment is MIL-STD-461 which applies
to radiated equipment emissions and equipment susceptibility to interference. Radiated
emission limits are very typically 10 to 100 times more stringent than the levels shown in
Table 11.1. Required limits on immunity to RF fields are typically 200 times more
stringent (RF field strengths of 5 to 50 mV/m) than the limits for commercial equipment.
Medical Equipment
Although not yet mandatory, EMC regulations for medical equipment are presently being
defined by the FDA (Food and Drug Administration) in the USA and the European
Community. The primary focus of these EMC regulations will be on immunity to RF
fields, electrostatic discharge, and power-line disturbances, and may very well be more
stringent than the limits spelled out in MIL-STD-461. The primary objective of the
medical EMC regulations is to guarantee safety to humans.
11.24
EMI/RFI CONSIDERATIONS
Automotive Equipment
Perhaps the most difficult and hostile environment in which electrical circuits and
systems must operate is that found in the automobile. All of the key EMI threats to
electrical systems exist here. In addition, operating temperature extremes, moisture, dirt,
and toxic chemicals further exacerbate the problem. To complicate matters further,
standard techniques (ferrite beads, feedthrough capacitors, inductors, resistors, shielded
cables, wires, and connectors) used in other systems are not generally used in automotive
applications because of the cost of the additional components.
Presently, automotive EMC regulations, defined by the very comprehensive SAE
Standards J551 and J1113, are not yet mandatory. They are, however, very rigorous. SAE
standard J551 applies to vehicle-level EMC specifications, and standard J1113
(functionally similar to MIL-STD-461) applies to all automotive electronic modules. For
example, the J1113 specification requires that electronic modules cannot radiate electric
fields greater than 300nV/m at a distance of 3 meters. This is roughly 1000 times more
stringent than the FCC Part 15 Class-A specification. In many applications, automotive
manufacturers are imposing J1113 RF field immunity limits on each of the active
components used in these modules. Thus, in the very near future, automotive
manufacturers will require that IC products comply with existing EMC standards and
regulations.
EMC Regulations’ Impact on Design
In all these applications and many more, complying with mandatory EMC regulations
will require careful design of individual circuits, modules, and systems using established
techniques for cable shielding, signal and power-line filtering against both small- and
large-scale disturbances, and sound multilayer PCB layouts. The key to success is to
incorporate sound EMC principles early in the design phase to avoid time-consuming and
expensive redesign efforts.
Passive Components: Your Arsenal Against EMI
Minimizing the effects of EMI requires that the circuit/system designer be completely
aware of the primary arsenal in the battle against interference: passive components. To
successfully use these components, the designer must understand their non-ideal
behavior. For example, Figure 11.47 illustrates the real behavior of the passive
components used in circuit design. At very high frequencies, wires become transmission
lines, capacitors become inductors, inductors become capacitors, and resistors behave as
resonant circuits.
11.25
BASIC LINEAR DESIGN
B EH AVIO R:
RESISTIVE
LOW FREQUENCY –
RESISTIVE
MEDIUM FREQUENCY – INDUCTIVE
HIGH FREQUENCY TRANSMISSION LINE
AND ANTENNA EFFECTS
TYPICAL ROUND
WIRE
IMPEDANCE
GROUND
PLANE
FREQUENCY
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.23: Impedance Comparison: Wire vs. Ground Plane
A specific case in point is the frequency response of a simple wire compared to that of a
ground plane. In many circuits, wires are used as either power or signal returns, and there
is no ground plane. A wire will behave as a very low resistance (less than 0.02 Ω/ft for
22-gauge wire) at low frequencies, but because of its parasitic inductance of
approximately 20 nH/ft, it becomes inductive at frequencies above 160 kHz.
Furthermore, depending on size and routing of the wire and the frequencies involved, it
ultimately becomes a transmission line with an uncontrolled impedance. From our
knowledge of RF, unterminated transmission lines become antennas with gain, as
illustrated in Figure 11.26. On the other hand, large area ground planes are much more
well-behaved, and maintain a low impedance over a wide range of frequencies. With a
good understanding of the behavior of real components, a strategy can now be developed
to find solutions to most EMI problems.
With any problem, a strategy should be developed before any effort is expended trying to
solve it. This approach is similar to the scientific method: initial circuit misbehavior is
noted, theories are postulated, experiments designed to test the theories are conducted,
and results are again noted. This process continues until all theories have been tested and
expected results achieved and recorded. With respect to EMI, a problem solving
framework has been developed. As shown in Figure 11.24, the model suggested by
Kimmel-Gerke in [Reference 1] illustrates that all three elements (a source, a receptor or
victim, and a path between the two) must exist in order to be considered an EMI problem.
The sources of electromagnetic interference can take on many forms, and the everincreasing
number
of
portable
instrumentation
and
personal
communications/computation equipment only adds the number of possible sources and
receptors.
11.26
EMI/RFI CONSIDERATIONS
Interfering signals reach the receptor by conduction (the circuit or system
interconnections) or radiation (parasitic mutual inductance and/or parasitic capacitance).
In general, if the frequencies of the interference are less than 30 MHz, the primary means
by which interference is coupled is through the interconnects. Between 30 MHz and
300 MHz, the primary coupling mechanism is cable radiation and connector leakage. At
frequencies greater than 300 MHz, the primary mechanism is slot and board radiation.
There are many cases where the interference is broadband, and the coupling mechanisms
are combinations of the above.
ANY INTERFERENCE PROBLEM CAN BE BROKEN DOWN INTO:
„ The SOURCE of interference
„ The RECEPTOR of interference
„ The PATH coupling the source to the receptor
Microcontroller
Analog
Digital
„
„
Communications
Receivers
„
„
„
Conducted
Signal
Power
Ground
„
ESD
Communications
Transmitters
Power
Disturbances
Lightning
Radiated
EM Fields
Crosstalk
Capacitive
Inductive
Other Electronic
Systems
„
„
„
Microcontroller
Analog
Digital
RECEPTORS
„
PATHS
SOURCES
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.24:
A Diagnostic Framework for EMI
When all three elements exist together, a framework for solving any EMI problem can be
drawn from Figure 11.25. There are three types of interference with which the circuit or
system designer must contend. The first type of interference is that generated by and
emitted from an instrument; this is known as circuit/system emission and can be either
conducted or radiated. An example of this would be the personal computer. Portable and
desktop computers must pass the stringent FCC Part 15 specifications prior to general
use.
The second type of interference is circuit or system immunity. This describes the behavior
of an instrument when it is exposed to large electromagnetic fields, primarily electric
fields with an intensity in the range of 1 to 10 V/m at a distance of 3 meters. Another
term for immunity is susceptibility, and it describes circuit/system behavior against
radiated or conducted interference.
11.27
BASIC LINEAR DESIGN
HANDHELD
TRANSMITTER
RADIO
TRANSMITTER
RADIATED
EMISSIONS
INTERNAL
ELECTRONICS
LIGHTNING
CONDUCTED
EMISSIONS
HUMAN ESD
POWER
DISTURBANCE
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.25: Three Types of Interference:
Emissions – Immunity – Internal
The third type of interference is internal. Although not directly shown on the figure,
internal interference can be high speed digital circuitry within the equipment which
affects sensitive analog (or other digital circuitry), or noisy power supplies which can
contaminate both analog and digital circuits. Internal interference often occurs between
digital and analog circuits, or between motors or relays and digital circuits. In mixed
signal environments, the digital portion of the system often interferes with analog
circuitry. In some systems, the internal interference reaches such high levels that even
very high speed digital circuitry can affect other low-speed digital circuitry as well as
analog circuits.
In addition to the source-path-receptor model for analyzing EMI-related problems,
Kimmel Gerke Associates have also introduced the FAT-ID concept [Reference 1].
FAT-ID is an acronym that describes the five key elements inherent in any EMI problem.
These five key parameters are: frequency, amplitude, time, impedance, and distance.
The frequency of the offending signal suggests its path. For example, the path of lowfrequency interference is often the circuit conductors. As the interference frequency
increases, it will take the path of least impedance, usually stray capacitance. In this case,
the coupling mechanism is radiation.
Time and frequency in EMI problems are interchangeable. In fact, the physics of EMI
have shows that the time response of signals contains all the necessary information to
construct the spectral response of the interference. In digital systems, both the signal rise
time and pulse repetition rate produce spectral components according to the following
relationship:
f E MI =
11.28
1
π ⋅ t r ise
Eq. 11-5
EMI/RFI CONSIDERATIONS
INPUTS P ICK UP HIGH FRE QUENCY ENERGY ON
SIGNAL LINE, WHICH IS DETECTED BY THE AMPLIFIER
VCC
OUTPUT DRIV ERS CAN BE JAMMED, T OO: EN ERGY
COUPLES BACK TO INPUT VIA VCC OR SIGNAL LINE
L
AND THEN IS DETECTED OR AMPLIFIED
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.26: RFI Can Cause Rectification in Sensitive Analog Circuits
For example, a pulse having a 1 ns rise time is equivalent to an EMI frequency of over
300 MHz. This time-frequency relationship can also be applied to high speed analog
circuits, where slew rates in excess of 1000 V/µs and gain-bandwidth products greater
than 500 MHz are not uncommon.
When this concept is applied to instruments and systems, EMI emissions are again
functions of signal rise time and pulse repetition rates. Spectrum analyzers and high
speed oscilloscopes used with voltage and current probes are very useful tools in
quantifying the effects of EMI on circuits and systems.
Another important parameter in the analysis of EMI problems is the physical dimensions
of cables, wires, and enclosures. Cables can behave as either passive antennas (receptors)
or very efficient transmitters (sources) of interference. Their physical length and their
shield must be carefully examined where EMI is a concern. As previously mentioned, the
behavior of simple conductors is a function of length, cross-sectional area, and
frequency. Openings in equipment enclosures can behave as slot antennas, thereby
allowing EMI energy to affect the internal electronics.
11.29
BASIC LINEAR DESIGN
Radio Frequency Interference (RFI)
The world is rich in radio transmitters: radio and TV stations, mobile radios, computers,
electric motors, garage door openers, electric jackhammers, and countless others. All this
electrical activity can affect circuit/system performance and, in extreme cases, may
render it inoperable. Regardless of the location and magnitude of the interference,
circuits/systems must have a minimum level of immunity to radio frequency interference
(RFI). The next section will cover two general means by which RFI can disrupt normal
instrument operation: the direct effects of RFI sensitive analog circuits, and the effects of
RFI on shielded cables.
Two terms are typically used in describing the sensitivity of an electronic system to RF
fields. In communications, radio engineers define immunity to be an instrument’s
susceptibility to the applied RFI power density at the unit. In more general EMI analysis,
the electric-field intensity is used to describe RFI stimulus. For comparative purposes,
Equation 11-6 can be used to convert electric-field intensity to power density and viceversa:
r ⎛ V⎞
E ⎜ ⎟ = 61.4
⎝m⎠
⎛ mW ⎞
⎟⎟
P T ⎜⎜
⎝ cm 2 ⎠
Eq. 11-6
where:
E = Electric Field Strength, in volts per meter, and
PT = Transmitted power, in milliwatts per cm2.
From the standpoint of the source-path-receptor model, the strength of the electric field,
E, surrounding the receptor is a function of transmitted power, antenna gain, and
distance from the source of the disturbance. An approximation for the electric-field
intensity (for both near- and far-field sources) in these terms is given by Equation 11-7:
⎛ PT ⋅ G A
r ⎛ V⎞
E ⎜ ⎟ = 5.5 ⎜⎜
⎝m⎠
d
⎝
⎞
⎟
⎟
⎠
Eq. 11-7
where:
E = Electric field intensity, in V/m;
PT = Transmitted power, in mW/cm2;
GA = Antenna gain (numerical); and
d = distance from source, in meters
For example, a 1 W hand-held radio at a distance of 1 meter can generate an electric-field
of 5.5 V/m, whereas a 10 kW radio transmission station located 1 km away generates a
field smaller than 0.6 V/m.
11.30
EMI/RFI CONSIDERATIONS
Analog circuits are generally more sensitive to RF fields than digital circuits because
analog circuits, operating at high gains, must be able to resolve signals in the
microvolt/millivolt region. Digital circuits, on the other hand, are more immune to RF
fields because of their larger signal swings and noise margins. As shown in Figure 11.27,
RF fields can use inductive and/or capacitive coupling paths to generate noise currents
and voltages which are amplified by high impedance analog instrumentation. In many
cases, out-of-band noise signals are detected and rectified by these circuits. The result of
the RFI rectification is usually unexplained offset voltage shifts in the circuit or in the
system.
LOCAL
VPOS
REMOTE
VNEG
Decouple all voltage supplies to analog chip with high-frequency capacitors
Use high-frequency filters on all lines that leave the board
Use high-frequency filters on the voltage reference if it is not grounded
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.27: Keeping RFI Away from Analog Circuits
There are techniques that can be used to protect analog circuits against interference from
RF fields (see Figure 11.28). The three general points of RFI coupling are signal inputs,
signal outputs, and power supplies. At a minimum, all power supply pin connections on
analog and digital ICs should be decoupled with 0.1 µF ceramic capacitors. As was
shown in Reference 3, low-pass filters, whose cutoff frequencies are set no higher than
10 to 100 times the signal bandwidth, can be used at the inputs and the outputs of signal
conditioning circuitry to filter noise.
Care must be taken to ensure that the low-pass filters (LPFs) are effective at the highest
RF interference frequency expected. As illustrated in Figure 11.28, real low-pass filters
may exhibit leakage at high frequencies. Their inductors can lose their effectiveness due
to parasitic capacitance, and capacitors can lose their effectiveness due to parasitic
inductance. A rule of thumb is that a conventional low-pass filter (made up of a single
capacitor and inductor) can begin to leak when the applied signal frequency is 100 to
1000 higher than the filter’s cutoff frequency. For example, a 10 kHz LPF would not be
considered very efficient at filtering frequencies above 1 MHz.
11.31
BASIC LINEAR DESIGN
TYPICALLY 100 - 1000 f3dB
FILTER
ATTENUATION
f3dB
FREQUENCY
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.28: Single Low Power Low Pass Filter Loses Effectiveness
at 100 – 1000 f3dB
Rather than use one LPF stage, it is recommended that the interference frequency bands
be separated into low-band, mid-band, and high-band, and then use individual filters for
each band. Kimmel Gerke Associates use the stereo speaker analogy of woofermidrange-tweeter for RFI low-pass filter design illustrated in Figure 11.29. In this
approach, low frequencies are grouped from 10 kHz to 1 MHz, mid-band frequencies are
grouped from 1 MHz to 100 MHz, and high frequencies grouped from 100 MHz to 1
GHz. In the case of a shielded cable input/output, the high frequency section should be
located close to the shield to prevent high-frequency leakage at the shield boundary. This
is commonly referred to as feed-through protection. For applications where shields are
not required at the inputs/outputs, then the preferred method is to locate the high
frequency filter section as close the analog circuit as possible. This is to prevent the
possibility of pickup from other parts of the circuit.
FEEDTHROUGH
CAPACITOR
FERRITE
BEAD
.01 µf
TWEETER
IRON
CORE
1 µf
0.1 µf
MIDRANGE
WOOFER
STEREO SPEAKER ANALOGY
Figure 11.29: Multistage Filters Are More Effective
11.32
EMI/RFI CONSIDERATIONS
FILTER
HF
ENERGY
HF
ENERGY
BOND IMPEDANCE
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.30: Non-Zero (Inductive and/or Resistive) Filter
Ground Reduces Effectiveness
Another cause of filter failure is illustrated in Figure 11.31. If there is any impedance in
the ground connection (for example, a long wire or narrow trace connected to the ground
plane), then the high frequency noise uses this impedance path to bypass the filter
completely. Filter grounds must be broadband and tied to low impedance points or planes
for optimum performance. High frequency capacitor leads should be kept as short as
possible, and low inductance surface-mounted ceramic chip capacitors are preferable.
SHIELD
EQUIVALENT
CIRCUIT
ICM
ICM
ICM = COMMON-MODE CURRENT
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.31: “Shielded” Cable Can Carry High Frequency Current and
Behaves as an Antenna
11.33
BASIC LINEAR DESIGN
In the first part of this discussion on RF immunity, circuit level techniques were
discussed. In this next section, the second strategic concept for RF immunity will be
discussed: all cables behave as antennas. As shown in Figure 11.31, pigtail terminations
on cables very often cause systems to fail radiated emissions tests because high
frequency noise has coupled into the cable shield, generally through stray capacitance. If
the length of the cable is considered electrically long (a concept to be explained later) at
the interference frequency, then it can behave as a very efficient quarter-wave antenna.
The cable pigtail forms a matching network, as shown in the figure, to radiate the noise
which coupled into the shield. In general, pigtails are only recommended for applications
below 10 kHz, such as 50 Hz/60 Hz interference protection. For applications where the
interference is greater than 10 kHz, shielded connectors, electrically and physically
connected to the chassis, should be used. In applications where shielding is not used,
filters on input/output signal and power lines work well. Small ferrites and capacitors
should be used to filter high frequencies, provided that: (1) the capacitors have short
leads and are tied directly to the chassis ground, and (2) the filters are physically located
close to the connectors to prevent noise pickup.
„
Radio-Frequency Interference is a Serious Threat
Radio‹ Equipment causes interference to nearby radio and television
‹ Equipment upset by nearby transmitters
„
RF-Failure Modes
RF‹ Digital circuits prime source of emissions
‹ Analog circuits more vulnerable to RF than digital circuits
„
Two Strategic Concepts
‹ Treat all cables as antennas
‹ Determine the most critical circuits
„
RF Circuit Protection
‹ Filters and multilayer boards
‹ Multistage filters often needed
„
RF Shielding
‹ Slots and seams cause the most problems
„
RF Cable Protection
‹ HighHigh-quality shields and connectors needed for RF protection
Figure 11.32: Summary of Radio Frequency Interference
and Protection Techniques
11.34
EMI/RFI CONSIDERATIONS
GAS DISCHARGE
TUBES
"CROWBARS"
CHOKES
TRANSIENT
SUPPRESSORS
BIG ZENERS
OR MOVs
V
LINE
N
LOAD
G
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.33: Power Line Disturbances Can Generate EMI
The key issues and techniques described in this section on solving RFI related problems
are summarized in Figure 11.32. Some of the issues were not discussed in detail, but are
equally important. For a complete treatment of this issue, the interested reader should
consult References 1 and 2. The main thrust of this section was to provide the reader with
a problem-solving strategy against RFI and to illustrate solutions to commonly
encountered RFI problems.
Solutions for Power-Line Disturbances
The goal of this next section is not to describe in detail all the circuit/system failure
mechanisms which can result from power-line disturbances or faults. Nor is it the intent
of this section to describe methods by which power-line disturbances can be prevented.
Instead, this section will describe techniques that allow circuits and systems to
accommodate transient power-line disturbances.
Figure 11.34 is an example of a hybrid power transient protection network commonly
used in many applications where lightning transients or other power-line disturbances are
prevalent. These networks can be designed to provide protection against transients as
high as 10 kV and as fast as 10 ns. Gas discharge tubes (crowbars) and large geometry
zener diodes (clamps) are used to provide both differential and common-mode protection.
Metal-oxide varistors (MOVs) can be substituted for the zener diodes in less critical, or
in more compact designs. Chokes are used to limit the surge current until the gas
discharge tubes fire.
11.35
BASIC LINEAR DESIGN
HOT
HOT
LINE
LOAD
NEU
NEU
GND
OPTIONAL
NOTE: OPTIONAL CHOKE ADDED FOR COMMON-MODE PROTECTION
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.34: Schematic for a Commercial Power Line Filter
Commercial EMI filters, as illustrated in Figure 11.34, can be used to filter less
catastrophic transients or high frequency interference. These EMI filters provide both
common-mode and differential mode filtering. An optional choke in the safety ground
can provide additional protection against common-mode noise. The value of this choke
cannot be too large, however, because its resistance may affect power-line fault clearing.
STANDARD TRANSFORMER - NO SHIELD
NOTE CONNECTION FROM SECONDARY
TO SAFETY GROUND TO ELIMINATE
GROUND-TO-NEUTRAL VOLTAGE
SINGLE FARADAY SHIELD
CONNECT TO SAFETY GROUND FOR
COMMON-MODE PROTECTION
SINGLE FARADAY SHIELD
CONNECT TO NOISY-SIDE NEUTRAL
WIRE FOR DIFFERENTIAL-MODE
PROTECTION
TRIPLE FARADAY SHIELD
CONNECT TO SAFETY GROUND FOR
COMMON MODE
CONNECT TO NEUTRALS FOR
DIFFERENTIAL MODE
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.35: Faraday Shields in Isolation Transformers
Provide Increasing Levels of Protection
11.36
EMI/RFI CONSIDERATIONS
Transformers provide the best common-mode power line isolation. They provide good
protection at low frequencies (<1 MHz), or for transients with rise and fall times greater
than 300 ns. Most motor noise and lightning transients are in this range, so isolation
transformers work well for these types of disturbances. Although the isolation between
input and output is galvanic, isolation transformers do not provide sufficient protection
against extremely fast transients (<10 ns) or those caused by high-amplitude electrostatic
discharge (1 ns to 3 ns). As illustrated in Figure 11.36, isolation transformers can be
designed for various levels of differential- or common-mode protection. For differentialmode noise rejection, the Faraday shield is connected to the neutral, and for commonmode noise rejection, the shield is connected to the safety ground.
COUPLING TO I/O VIA
CROSSTALK OR RADIATION
RADIATION FROM
POWER WIRING
COUPLING VIA COMMON
POWER IMPEDANCE
COUPLING VIA COMMON
GROUND IMPEDANCE
RADIATION
FROM I/O
WIRING
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.36: Methods by Which High Frequency Energy Couples
and Radiates Into Circuitry Via Placements
Printed Circuit Board Design for EMI Protection
This section will summarize general points regarding the most critical portion of the
design phase: the printed circuit board layout. It is at this stage where the performance of
the system is most often compromised. This is not only true for signal-path performance,
but also for the system’s susceptibility to electromagnetic interference and the amount of
electromagnetic energy radiated by the system. Failure to implement sound PCB layout
techniques will very likely lead to system/instrument EMC failures.
Figure 11.37 is a real-world printed circuit board layout which shows all the paths
through which high-frequency noise can couple/radiate into/out of the circuit. Although
the diagram shows digital circuitry, the same points are applicable to precision analog,
high speed analog, or mixed analog/digital circuits. Identifying critical circuits and paths
helps in designing the PCB layout for both low emissions and susceptibility to radiated
and conducted external and internal noise sources.
11.37
BASIC LINEAR DESIGN
A key point in minimizing noise problems in a design is to choose devices no faster than
actually required by the application. Many designers assume that faster is better: fast
logic is better than slow, high bandwidth amplifiers are clearly better than low bandwidth
ones, and fast DACs and ADCs are better, even if the speed is not required by the system.
Unfortunately, faster is not better, but worse where EMI is concerned.
FERRITE
BEAD
VCC
FERRITE BEAD OR
10 - 33Ω RESISTOR
GND
MICROPROCESSOR
OR OTHER HIGH-SPEED
CLOCKED CIRCUIT
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.37: Power Supply Filtering and Signal Line Snubbing
Greatly Reduces EMI Emissions
Many fast DACs and ADCs have digital inputs and outputs with rise and fall times in the
nanosecond region. Because of their wide bandwidth, the sampling clock and the digital
inputs and can respond to any form of high frequency noise, even glitches as narrow as
1 ns to 3 ns. These high speed data converters and amplifiers are easy prey for the high
frequency noise of microprocessors, digital signal processors, motors, switching
regulators, hand-held radios, electric jackhammers, etc. With some of these high-speed
devices, a small amount of input/output filtering may be required to desensitize the
circuit from its EMI/RFI environment. Adding a small ferrite bead just before the
decoupling capacitor as shown in Figure 11.38 is very effective in filtering high
frequency noise on the supply lines. For those circuits that require bipolar supplies, this
technique should be applied to both positive and negative supply lines.
To help reduce the emissions generated by extremely fast moving digital signals at DAC
inputs or ADC outputs, a small resistor or ferrite bead may be required at each digital
input/output.
Once the system’s critical paths and circuits have been identified, the next step in
implementing sound PCB layout is to partition the printed circuit board according to
circuit function. This involves the appropriate use of power, ground, and signal planes.
Good PCB layouts also isolate critical analog paths from sources of high interference
11.38
EMI/RFI CONSIDERATIONS
(I/O lines and connectors, for example). High frequency circuits (analog and digital)
should be separated from low frequency ones. Furthermore, automatic signal routing
CAD layout software should be used with extreme caution, and critical paths routed by
hand.
BEFORE
AFTER
Route
Power
Power
Route
Ground
Route
Route
Ground
„
„
„
„ Advantages of Embedding
Lower impedances, therefore lower emissions and crosstalk
Reduction in emissions and crosstalk is significant above 50MHz
Traces are protected
„
„
„
„ Disadvantages of Embedding
Lower interboard capacitance, harder to decouple
Impedances may be too low for matching
Hard to prototype and troubleshoot buried traces
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.38: “To Embed or Not To Embed” That Is the Question
Properly designed multilayer printed circuit boards can reduce EMI emissions and
increase immunity to RF fields by a factor of 10 or more compared to double-sided
boards. A multilayer board allows a complete layer to be used for the ground plane,
whereas the ground plane side of a double-sided board is often disrupted with signal
crossovers, etc.
The preferred multilayer board arrangement is to embed the signal traces between the
power and ground planes, as shown in Figure 11.39. These low impedance planes form
very high frequency stripline transmission lines with the signal traces. The return current
path for a high frequency signal on a trace is located directly above and below the trace
on the ground/power planes. The high frequency signal is thus contained inside the PCB,
thereby minimizing emissions. The embedded signal trace approach has an obvious
disadvantage: debugging circuit traces that are hidden from plain view is difficult.
11.39
BASIC LINEAR DESIGN
GaAs
0.1
PCB TRACK
LENGTH
(inches)
0.2
ECL
0.75
1.5
3.8
Schottky
3
6
15
FAST
3
6
15
AS
3
6
15
AC
4
8
20
ALS
6
12
30
LS
8
16
40
TTL
10
20
50
HC
18
36
90
DIGITAL IC
FAMILY
t r, t f
(ns)
PCB TRACK
LENGTH
(cm)
0.5
tr = rise time of signal in ns
tf = fall time of signal in ns
J For analog signals @ fmax, calculate tr = tf = 0.35 / fmax
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.39: Line Termination Should Be Used When the
Length of the PCB Trace Exceeds 2 inches/ns
Much has been written about terminating printed circuit board traces in their
characteristic impedance to avoid reflections. A good rule-of-thumb to determine when
this is necessary is as follows: Terminate the line in its characteristic impedance when
the one-way propagation delay of the PCB track is equal to or greater than one-half the
applied signal rise/fall time (whichever edge is faster). A conservative approach is to use
a 2 inch (PCB track length)/nanosecond (rise-, fall-time) criterion. For example, PCB
tracks for high speed logic with rise/fall time of 5 ns should be terminated in their
characteristic impedance and if the track length is equal to or greater than 10 inches
(including any meanders). The 2 inch/nanosecond track length criterion is summarized in
Figure 11.38 for a number of logic families
This same 2 inch/nanosecond rule should be used with analog circuits in determining the
need for transmission line techniques. For instance, if an amplifier must output a
maximum frequency of fmax, then the equivalent risetime, tr, can be calculated using the
equation tr = 0.35/fmax. The maximum PCB track length is then calculated by
multiplying the rise time by 2 inch/nanosecond. For example, a maximum output
11.40
EMI/RFI CONSIDERATIONS
frequency of 100 MHz corresponds to a rise time of 3.5 ns, and a track carrying this
signal greater than 7 inches should be treated as a transmission line.
“ALL EMI PROBLEMS BEGIN AND END AT A CIRCUIT”
„ Identify critical, sensitive circuits
„ Where appropriate, choose ICs no faster than needed
„ Consider and implement sound PCB design
„ Spend time on the initial layout (by hand, if necessary)
„ Power supply decoupling (digital and analog circuits)
„ High-speed digital and high-accuracy analog don't mix
„ Beware of connectors for input / output circuits
„ Test, evaluate, and correct early and often
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.40: Circuit Board Design and EMI
Equation 9.4 can be used to determine the characteristic impedance of a PCB track
separated from a power/ground plane by the board’s dielectric (microstrip transmission
line):
Z o (Ω) =
⎡ 5.98d
ln ⎢
ε r + 1.41
⎣ 0.89w +
87
⎤
t ⎥⎦
Eq. 11-8
where:
εr = dielectric constant of printed circuit board material;
d = thickness of the board between metal layers, in mils;
w = width of metal trace, in mils; and
t = thickness of metal trace, in mils.
The one-way transit time for a single metal trace over a power/ground plane can be
determined from Eq. 9.5:
t pd ( n s / ft ) = 1.017 0.475 ε r + 0.67
Eq. 11-9
11.41
BASIC LINEAR DESIGN
For example, a standard 4-layer PCB board might use 8-mil wide, 1 ounce (1.4 mils)
copper traces separated by 0.021" FR-4 (εr = 4.7) dielectric material. The characteristic
impedance and one-way transit time of such a signal trace would be 88 Ω and 1.7 ns/ft
(7 "/ns), respectively. Transmission lines can be effectively terminated in several ways
depending on the application..
Figure 11.41 is a summary of techniques that should be applied to printed circuit board
layouts to minimize the effects of electromagnetic interference, both emissions and
immunity.
INCIDENT RAY
REFLECTED RAY
SHIELD
MATERIAL
TRANSMITTED
RAY
ABSORPTIVE
REGION
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.41 Reflection and Absorption Are the Two Principal
Shielding Mechanisms
A Review of Shielding Concepts
The concepts of shielding effectiveness presented next are background material.
Interested readers should consult References 1, 3, and 4 cited at the end of the section for
more detailed information.
Applying the concepts of shielding requires an understanding of the source of the
interference, the environment surrounding the source, and the distance between the
source and point of observation (the receptor or victim). If the circuit is operating close to
the source (in the near-, or induction-field), then the field characteristics are determined
by the source. If the circuit is remotely located (in the far-, or radiation-field), then the
field characteristics are determined by the transmission medium.
A circuit operates in a near-field if its distance from the source of the interference is less
than the wavelength (λ) of the interference divided by 2π, or π/2λ. If the distance
between the circuit and the source of the interference is larger than this quantity, then the
11.42
EMI/RFI CONSIDERATIONS
circuit operates in the far field. For instance, the interference caused by a 1ns pulse edge
has an upper bandwidth of approximately 350 MHz. The wavelength of a 350 MHz
signal is approximately 32 inches (the speed of light is approximately 12"/ns). Dividing
the wavelength by 2π yields a distance of approximately 5 inches, the boundary between
near- and far-field. If a circuit is within 5 inches of a 350 MHz interference source, then
the circuit operates in the near-field of the interference. If the distance is greater than 5
inches, the circuit operates in the far-field of the interference.
Regardless of the type of interference, there is a characteristic impedance associated with
it. The characteristic, or wave impedance of a field is determined by the ratio of its
electric (or E-) field to its magnetic (or H-) field. In the far field, the ratio of the electric
field to the magnetic field is the characteristic (wave impedance) of free space, given by
Zo = 377 Ω. In the near field, the wave-impedance is determined by the nature of the
interference and its distance from the source. If the interference source is high-current
and low-voltage (for example, a loop antenna or a power-line transformer), the field is
predominately magnetic and exhibits a wave impedance which is less than 377 Ω. If the
source is low current and high voltage (for example, a rod antenna or a high speed digital
switching circuit), then the field is predominately electric and exhibits a wave impedance
which is greater than 377 Ω.
VENTILATORS
SEAMS
SWITCHES
DISPLAY
PANEL
DATA
CABLES
POWER
CABLES
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.42: Any Opening in an Enclosure Can Act as an EMI Waveguide
by Compromising Shielding Effectiveness
Conductive enclosures can be used to shield sensitive circuits from the effects of these
external fields. These materials present an impedance mismatch to the incident
interference because the impedance of the shield is lower than the wave impedance of the
incident field. The effectiveness of the conductive shield depends on two things: First is
the loss due to the reflection of the incident wave off the shielding material. Second is the
loss due to the absorption of the transmitted wave within the shielding material. Both
concepts are illustrated in Figure 11.42. The amount of reflection loss depends upon the
type of interference and its wave impedance. The amount of absorption loss, however, is
11.43
BASIC LINEAR DESIGN
independent of the type of interference. It is the same for near- and far-field radiation, as
well as for electric or magnetic fields.
Reflection loss at the interface between two media depends on the difference in the
characteristic impedances of the two media. For electric fields, reflection loss depends on
the frequency of the interference and the shielding material. This loss can be expressed in
dB, and is given by:
⎡ σ
⎤
r
⎥
R e (dB ) = 322 + 10log10 ⎢
⎢μ f 3 r 2 ⎥
⎣ r
⎦
Eq. 11.10
where
σr = relative conductivity of the shielding material, in Siemens per meter;
µr = relative permeability of the shielding material, in Henries per meter;
f = frequency of the interference, and
r = distance from source of the interference, in meters
For magnetic fields, the loss depends also on the shielding material and the frequency of
the interference. Reflection loss for magnetic fields is given by:
⎡f r 2 σ
r
R m (dB ) = 14.6 + 10log10 ⎢
⎢ μr
⎣
⎤
⎥
⎥
⎦
Eq. 11.11
and, for plane waves ( r > λ/2π), the reflection loss is given by:
⎡σ ⎤
R pw (dB ) = 168 + 10log10 ⎢ r ⎥
⎣μr f ⎦
Eq. 11.12
Absorption is the second loss mechanism in shielding materials. Wave attenuation due to
absorption is given by:
A (dB ) = 3.34 t σ r μ r f
Eq. 11.13
where t = thickness of the shield material, in inches. This expression is valid for plane
waves, electric and magnetic fields. Since the intensity of a transmitted field decreases
exponentially relative to the thickness of the shielding material, the absorption loss in a
shield one skin-depth (δ) thick is 9 dB. Since absorption loss is proportional to thickness
and inversely proportional to skin depth, increasing the thickness of the shielding
material improves shielding effectiveness at high frequencies.
Reflection loss for plane waves in the far field decreases with increasing frequency
because the shield impedance, Zs, increases with frequency. Absorption loss, on the other
hand, increases with frequency because skin depth decreases. For electric fields and plane
11.44
EMI/RFI CONSIDERATIONS
waves, the primary shielding mechanism is reflection loss, and at high frequencies, the
mechanism is absorption loss. For these types of interference, high conductivity
materials, such as copper or aluminum, provide adequate shielding. At low frequencies,
both reflection and absorption loss to magnetic fields is low; thus, it is very difficult to
shield circuits from low frequency magnetic fields. In these applications, high
permeability materials that exhibit low reluctance provide the best protection. These low
reluctance materials provide a magnetic shunt path that diverts the magnetic field away
from the protected circuit. Some characteristics of metallic materials commonly used for
shielded enclosures are shown in Table 11.2.
Impedance and Skin Depths for Various Shielding Materials
Material
Conductivity
σr
Permeability
µr
Shield
Impedance
|Zs|
Cu
1
1
3.68E - 7 ⋅ f
2.6
4.71E - 7 ⋅ f
f
3.3
3.68E - 5 ⋅ f
f
0.26
3E - 4 ⋅ f
f
0.11
Al
Steel
µ Metal
1
0.1
0.03
0.61
1000
20,000
Skin
depth
δ (inch)
f
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Table 9.2
Where:
σo = 5.82 × 107 S/m
µo = 4π × 10-7 H/m
εo = 8.85 × 10-12 F/m
A properly shielded enclosure is very effective at preventing external interference from
disrupting its contents as well as confining any internally generated interference.
However, in the real world, openings in the shield are often required to accommodate
adjustment knobs, switches, connectors, or to provide ventilation (see Figure 11.43).
Unfortunately, these openings may compromise shielding effectiveness by providing
paths for high-frequency interference to enter the instrument.
The longest dimension (not the total area) of an opening is used to evaluate the ability of
external fields to enter the enclosure, because the openings behave as slot antennas.
Equation 9.10 can be used to calculate the shielding effectiveness, or the susceptibility to
EMI leakage or penetration, of an opening in an enclosure:
11.45
BASIC LINEAR DESIGN
⎛ λ ⎞
Sh ieldin g E ffect iven ess (dB ) = 20 log10 ⎜
⎟
⎝ 2 ⋅L⎠
Eq. 11.14
where:
λ = wavelength of the interference and
L = maximum dimension of the opening
Maximum radiation of EMI through an opening occurs when the longest dimension of
the opening is equal to one half-wavelength of the interference frequency (0 dB shielding
effectiveness). A rule of thumb is to keep the longest dimension less than 1/20
wavelength of the interference signal, as this provides 20 dB shielding effectiveness.
Furthermore, a few small openings on each side of an enclosure are preferred over many
openings on one side. This is because the openings on different sides radiate energy in
different directions, and as a result, shielding effectiveness is not compromised. If
openings and seams cannot be avoided, then conductive gaskets, screens, and paints
alone or in combination should be used judiciously to limit the longest dimension of any
opening to less than 1/20 wavelength. Any cables, wires, connectors, indicators, or
control shafts penetrating the enclosure should have circumferential metallic shields
physically bonded to the enclosure at the point of entry. In those applications where
unshielded cables/wires are used, then filters are recommended at the point of shield
entry.
SHIELDED ENCLOSURE B
SHIELDED ENCLOSURE A
LENGTH
SHIELDED
CABLE
FULLY SHIELDED ENCLOSURES CONNECTED BY FULLY
SHIELDED CABLE KEEP ALL INTERNAL CIRCUITS AND
SIGNAL LINES INSIDE THE SHIELD.
TRANSITION REGION: 1/20 WAVELENGTH
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.43 Length of Shielded Cables Determines as “Electrically Long”
or “Electronically Short” Applications
11.46
EMI/RFI CONSIDERATIONS
General Points on Cables and Shields
Although covered in more detail later, the improper use of cables and their shields is a
significant contributor to both radiated and conducted interference. Rather than
developing an entire treatise on these issues, the interested reader should consult
References 1, 2, 4, and 5. As illustrated in Figure 11.44 effective cable and enclosure
shielding confines sensitive circuitry and signals within the entire shield without
compromising shielding effectiveness.
CAPACITIVE COUPLING
TO CABLE
RECEIVER
CABLE SHIELD
GROUNDED AT LOAD
RECEIVER
en
en
en
EQUIVALENT
CIRCUITS
en
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.44: Connect the Shield at One Point at the Load to Protect Against
Low Frequency (50 Hz/60 Hz) Threats
Depending on the type of interference (pickup/radiated, low/high frequency), proper
cable shielding is implemented differently and is very dependent on the length of the
cable. The first step is to determine whether the length of the cable is electrically short or
electrically long at the frequency of concern. A cable is considered electrically short if
the length of the cable is less than 1/20 wavelength of the highest frequency of the
interference, otherwise it is electrically long. For example, at 50 Hz/60 Hz, an
electrically short cable is any cable length less than 150 miles, where the primary
coupling mechanism for these low frequency electric fields is capacitive. As such, for any
cable length less than 150 miles, the amplitude of the interference will be the same over
the entire length of the cable. To protect circuits against low frequency electric-field
pickup, only one end of the shield should be returned to a low impedance point. A
generalized example of this mechanism is illustrated in Figure 11.45.
11.47
BASIC LINEAR DESIGN
„ Diagnose before you fix
„
„
„
„
„ Ask yourself:
What are the symptoms?
What are the causes?
What are the constraints?
How will you know you have fixed it?
„ Use available models for EMI to identify source - path - victim
„ Start at low frequency and work up to high frequency
„
„
„
„
„
„
„
„ EMI doctor's bag of tricks:
Aluminum foil
Conductive tape
Bulk ferrites
Power line filters
Signal filters
Resistors, capacitors, inductors, ferrites
Physical separation
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.45: EMI Troubleshooting Philosophy
In this example, the shield is grounded at the receiver. An exception to this approach
(which will be highlighted again later) is the case where line-level (>1 V rms) audio
signals are transmitted over long distances using twisted pair, shielded cables. In these
applications, the shield again offers protection against low-frequency interference, and an
accepted approach is to ground the shield at the driver end (LF and HF ground) and
ground it at the receiver with a capacitor (HF ground only).
In those applications where the length of the cable is electrically long, or protection
against high frequency interference is required, then the preferred method is to connect
the cable shield to low impedance points at both ends (direct connection at the driving
end, and capacitive connection at the receiver). Otherwise, unterminated transmission
lines effects can cause reflections and standing waves along the cable. At frequencies of
10 MHz and above, circumferential (360°) shield bonds and metal connectors are
required to main low-impedance connections to ground.
In summary, for protection against low frequency (<1 MHz), electric-field interference,
grounding the shield at one end is acceptable. For high frequency interference (>1 MHz),
the preferred method is grounding the shield at both ends, using 360° circumferential
bonds between the shield and the connector, and maintaining metal-to-metal continuity
between the connectors and the enclosure. Low-frequency ground loops can be
eliminated by replacing one of the DC shield connections to ground with a low
inductance 0.01 µF capacitor. This capacitor prevents low frequency ground loops and
shunts high frequency interference to ground.
11.48
EMI/RFI CONSIDERATIONS
EMI Trouble Shooting Philosophy
System EMI problems often occur after the equipment has been designed and is operating
in the field. More often than not, the original designer of the instrument has retired and is
living in Tahiti, so the responsibility of repairing it belongs to someone else who may not
be familiar with the product. Figure 11.46 summarizes the EMI problem solving
techniques discussed in this section and should be useful in these situations.
SINGLE-ENDED OR
DIFFERENTIAL?
PASSIVE
OR
ACTIVE
SENSORS?
SHIELDED
CABLE
MEASUREMENT
SYSTEM
?
?
DC OR AC GROUND?
ONE OR BOTH ENDS?
G1
G2
Reprinted from EDN Magazine (January 20, 1994) © CAHNERS PUBLISHING COMPANY 1995, A Division of Reed Publishing USA
Figure 11.46: Precision Sensors and Cable Shielding
11.49
BASIC LINEAR DESIGN
REFERENCES:
1.
“EDN’s Designer’s Guide to Electromagnetic Compatibility,” EDN, January, 20, 1994,
material reprinted by permission of Cahners Publishing Company, 1995.
2.
Designing for EMC (Workshop Notes), Kimmel Gerke Associates, Ltd., 1994.
3.
Systems Application Guide, Chapter 1, pg. 21-55, Analog Devices, Incorporated,
Norwood, MA, 1994.
4.
Henry Ott, Noise Reduction Techniques in Electronic Systems, Second Edition, New York,
ohn Wiley & Sons, 1988.
5.
Ralph Morrison, Grounding and Shielding Techniques in Instrumentation, Third Edition,
New York, John Wiley & Sons, 1986.
6.
Amplifier Applications Guide, Chapter XI, pg. 61, Analog Devices, Incorporated,
Norwood, MA, 1992.
7.
B. Slattery and J. Wynne, “Design and Layout of a Video Graphics System for Reduced EMI,”
Analog Devices Application Note AN-333.
8.
Paul Brokaw, “An IC Amplifier User Guide to Decoupling, Grounding, and Making Things Go
Right for a Change”, Analog Devices Application Note AN-202.
9.
A. Rich, “Understanding Interference-Type Noise,” Analog Dialogue, 16-3, 1982, pp. 16-19.
10.
A. Rich, “Shielding and Guarding,” Analog Dialogue, 17-1, 1983, pp. 8-13.
11.
EMC Test & Design, Cardiff Publishing Company, Englewood, CO.
An excellent, general-purpose trade journal on issues of EMI and EMC.
12.
Amplifier Applications Guide, Section XI, pp. 1-10, Analog Devices, Incorporated, Norwood,
MA, 1992.
13.
Systems Applications Guide, Section 1, pp. 56-72, Analog Devices, Incorporated, Norwood,
MA, 1993.
14.
Linear Design Seminar, Section 1, pp. 19-22, Analog Devices, Incorporated, Norwood, MA,
1994.
15.
ESD Prevention Manual, Analog Devices, Inc.
16.
MIL-STD-883 Method 3015, Electrostatic Discharge Sensitivity Classification. Available
from Standardization Document Order Desk, 700 Robbins Ave., Building #4, Section D,
Philadelphia, PA 19111-5094.
17.
EIAJ ED-4701 Test Method C-111, Electrostatic Discharges. Available from the Japan
Electronics Bureau, 250 W 34th St., New York NY 10119, Attn.: Tomoko.
18.
ESD Association Standard S5.2 for Electrostatic Discharge (ESD) Sensitivity Testing Machine Model (MM)- Component Level. Available from the ESD Association, Inc., 200
Liberty Plaza, Rome, NY 13440.
11.50
EMI/RFI CONSIDERATIONS
19.
ESD Association Draft Standard DS5.3 for Electrostatic Discharge (ESD) Sensitivity Testing
- Charged Device Model (CDM) Component Testing. Available from the ESD Association,
Inc., 200 Liberty Plaza, Rome, NY 13440.
20.
Niall Lyne, “Electrical Overstress Damage to CMOS Converters,” Application Note AN-397,
Analog Devices, 1995.
11.51
BASIC LINEAR DESIGN
11.52
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