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Staying Well Grounded
Sigma-delta (Σ-∆) ADCs also require high-speed clocks because
of their high oversampling ratios. Even high resolution, so-called
“low frequency” industrial measurement ADCs (such as the
AD77xx-series), with throughputs of 10 Hz to 7.5 kHz, operate on
5-MHz or higher-frequency clocks and offer resolutions to 24 bits.
By Hank Zumbahlen
Grounding is undoubtedly one of the most difficult subjects in
system design. While the basic concepts are relatively simple, implementation is very involved. Unfortunately, there is no “cookbook”
approach that will guarantee good results, and there are a few things
that, if not done well, will probably cause headaches.
For linear systems, the ground is the reference against which we
base our signal. Unfortunately, it has also become the return path
for the power-supply current in unipolar supply systems. Improper
application of grounding strategies can cripple performance in
high-accuracy linear systems.
Grounding is an issue for all analog designs, and it is a fact that
proper implementation is no less essential in PCB-based circuits.
Fortunately, certain principles of quality grounding, especially
the use of ground planes, are intrinsic to the PCB environment.
Since this factor is one of the more significant advantages to PCBbased analog designs, appreciable discussion here is focused on it.
Some other aspects of grounding that must be managed include
the control of spurious ground and signal return voltages that
can degrade performance. These voltages can be due to external
signal coupling, common currents, or, simply, excessive IR drops
in ground conductors. Proper conductor routing and sizing, as
well as differential signal handling and ground isolation techniques,
enable control of such parasitic voltages.
An important topic to be discussed is grounding techniques
appropriate for a mixed-signal, analog/digital environment.
Indeed, the single issue of quality grounding can—and must—
influence the entire layout philosophy of a high-performance
mixed-signal PCB design.
Today’s signal processing systems generally require mixedsignal devices, such as analog-to-digital converters (ADCs) and
digital-to-analog converters (DACs), as well as fast digital signal
processors (DSPs). Requirements for processing analog signals
that have a wide dynamic range impose the need to use highperformance ADCs and DACs. Maintaining a wide dynamic
range with low noise in a hostile digital environment is dependent
upon using good high-speed circuit design techniques, including
proper signal routing, decoupling, and grounding.
In the past, “high-precision, low-speed” circuits have generally
been viewed differently than so-called “high-speed” circuits. With
respect to ADCs and DACs, the sampling (or update) frequency
has generally been used as the distinguishing speed criterion.
However, the following two examples show that, in practice, most
of today’s signal processing ICs are really “high-speed,” and must,
therefore, be treated as such in order to maintain high performance.
While certainly true of DSPs, it is also true for ADCs and DACs.
All sampling ADCs (those employing an internal sample-and-hold
circuit) suitable for signal processing applications operate with
relatively high-speed clocks with fast rise and fall times (generally
a few nanoseconds), so they must be treated as high-speed devices,
even though throughput rates may appear low. For example, a
medium-speed 12-bit successive-approximation (SAR) ADC
may operate from a 10-MHz internal clock, while the sampling
rate is only 500 kSPS.
Analog Dialogue 46-06, June (2012)
To further complicate the issue, mixed-signal ICs have both
analog and digital ports, adding to the confusion with respect
to proper grounding techniques. In addition, some mixed-signal
ICs have relatively low digital currents, while others have high
digital currents. In many cases, these two types require different
treatment for optimum grounding.
Digital and analog design engineers tend to view mixed-signal
devices from different perspectives, so the purpose of this article
is to describe a general grounding philosophy that will work for
most mixed-signal devices, without the need to know the specific
details of their internal circuits.
From the above, it should be clear that the issue of grounding
cannot be handled in a “cookbook” approach. Unfortunately, we
cannot provide a list of things to do that will guarantee success.
We can say that not doing certain things will probably lead to
difficulties. And what works in one frequency range may not
necessarily work in another frequency range. And, often, there
are competing requirements. The key to handling grounding is
to understand how the currents flow.
Star Ground
The “star” ground philosophy builds on the theory that all
voltages in a circuit are referred to a single ground point, known
as the star ground point. This can be better understood by a visual
analogy—the multiple conductors extending radially from the
common schematic ground to resemble a star. The star point need
not look like a star—it may be a point on a ground plane—but
the key feature of the star ground system is that all voltages are
measured with respect to a particular point in the ground network,
not just to an undefi ned “ground” (wherever one can clip a probe).
The star grounding philosophy, while reasonable theoretically, is
difficult to implement in practice. For example, if we design a star
ground system, drawing out all signal paths to minimize signal
interaction and the effects of high impedance signal or ground
paths, implementation problems arise. When the power supplies
are added to the circuit diagram, they either add unwanted
ground paths, or their supply currents flowing in the existing
ground paths are large enough, or noisy enough (or both), to
corrupt the signal transmission. This particular problem can
often be avoided by having separate power supplies (and, thus,
separate ground returns) for the various portions of the circuit.
For example, separate analog and digital supplies with separate
analog and digital grounds, joined at the star point, are common
in mixed-signal applications.
Separate Analog and Digital Grounds
It is a fact of life that digital circuitry is noisy. Saturating logic,
such as TTL and CMOS, draws large, fast current spikes from its
supply during switching. Logic stages, with hundreds of millivolts
(or more) of noise immunity, usually have little need for high levels
of supply decoupling. On the other hand, analog circuitry is quite
vulnerable to noise—on both power supply rails and grounds—so
it is sensible to separate analog and digital circuitry to prevent
digital noise from corrupting analog performance. Such separation
involves separation of both ground returns and power rails—which
can be inconvenient in a mixed-signal system.
www.analog.com/analogdialogue
1
Nevertheless, if a high-accuracy mixed-signal system is to deliver
full performance, it is essential to have separate analog and digital
grounds and separate power supplies. The fact that some analog
circuitry will “operate” (function) from a single +5-V supply does
not mean that it may optimally be operated from the same noisy
+5-V supply as the microprocessor, dynamic RAM, electric fan,
and other high-current devices! The analog portion must operate
at full performance from such a supply, not just be functional. By
necessity, this distinction will require very careful attention to
both the supply rails and the ground interfacing.
If the logic supply to the converter is isolated with a small resistance,
and decoupled to analog ground with a local 0.1-μF (100-nF)
capacitor, all the fast-edge digital currents of the converter will
return to ground through the capacitor and will not appear in
the external ground circuit. If a low-impedance analog ground is
maintained—as it should be for adequate analog performance—
additional noise due to the external digital ground current should
rarely present a problem.
Note that the analog and digital grounds in a system must be
joined at some point to allow signals to be referred to a common
potential. This star point, or analog/digital common point, is
carefully chosen so as not to introduce digital currents into the
ground of the analog part of the system—it is often convenient to
make the connection at the power supplies.
Related to the star ground system discussed earlier is the use of a
ground plane. To implement a ground plane, one side of a doublesided PCB (or one layer of a multilayer one) is made of continuous
copper and used as ground. The theory behind this is that the
large amount of metal will have as low a resistance as is possible.
Because of the large flattened conductor pattern, it will also have
as low an inductance as possible. It then offers the best possible
conduction, in terms of minimizing spurious ground difference
voltages, across the conducting plane.
Many ADCs and DACs have separate analog ground (AGND) and
digital ground (DGND) pins. On the device data sheets, users are
often advised to connect these pins together at the package. This
seems to confl ict with the advice to connect analog and digital
ground at the power supplies, and, in systems with more than one
converter, with the advice to join the analog and digital ground
at a single point.
There is, in fact, no confl ict. The labels, “analog ground” and
“digital ground,” on these pins refer to the internal parts of the
converter to which the pins are connected and not to the system
grounds to which they must go. For an ADC, these two pins
should generally be joined together and to the analog ground of
the system. It is not possible to join the two pins within the IC
package because the analog part of the converter cannot tolerate
the voltage drop resulting from the digital current flowing in the
bond wire to the chip. But they can be tied together externally.
Figure 1 illustrates this concept of ground connections for an
ADC. If these pins are connected in this way, the digital noise
immunity of the converter is diminished, somewhat, by the
amount of common-mode noise between the digital and analog
system grounds. However, since digital noise immunity is often
of the order of hundreds or thousands of millivolts, this factor is
unlikely to be important.
The analog noise immunity is diminished only by the external
digital currents of the converter itself flowing in the analog
ground. These currents should be kept quite small, and they can
be minimized by ensuring that the converter outputs don’t see
heavy loads. A good way to do this is to use a low input current
buffer at the ADC output, such as a CMOS buffer-register IC.
R
DIGITAL
OUTPUTS
CONVERTER
AGND
EXTERNAL DIGITAL
CURRENT RETURNS
THROUGH LOW
IMPEDANCE AGND
INTERNAL
DIGITAL
CURRENT
DGND
SYSTEM
ANALOG
GROUND
Figure 1. Analog (AGND) and digital ground
(DGND) pins of a data converter should be
returned to system analog ground.
2
+VS
Ground Planes
Note that the ground plane concept can also be extended to include
voltage planes. A voltage plane offers advantages similar to a ground
plane— a very low impedance conductor—but is dedicated to one
(or more) of the system supply voltages. A system can thus have
more than one voltage plane, as well as a ground plane.
While ground planes solve many ground impedance problems,
they aren’t a panacea. Even a continuous sheet of copper foil
has residual resistance and inductance; in some circumstances,
these can be enough to prevent proper circuit function. Designers
should be wary of injecting very high currents in a ground plane
because they can produce voltage drops that interfere with sensitive
circuitry.
Maintaining a low impedance, large area ground plane is of critical
importance to all analog circuits today. The ground plane not only
acts as a low impedance return path for decoupling high-frequency
currents (caused by fast digital logic) but also minimizes EMI/RFI
emissions. Because of the shielding action of the ground plane,
the circuit’s susceptibility to external EMI/RFI is also reduced.
Ground planes also allow the transmission of high-speed digital
or analog signals using transmission line techniques (microstrip
or stripline), where controlled impedances are required.
The use of “bus wire” is totally unacceptable as a “ground” because
of its impedance at the equivalent frequency of most logic transitions.
For instance, #22 gauge wire has about 20 nH/in inductance.
A transient current having a slew-rate of 10 mA/ns created by a
logic signal would develop an unwanted voltage drop of 200 mV
when flowing through one inch of this wire:
∆v = L
10 mA
∆i
= 20 nH ×
* ns = 200 mV
∆t
(1)
For a signal having a 2-V peak-to-peak range, this translates
into an error of about 200 mV, or 10% (approximately “3.5-bit
accuracy”). Even in all-digital circuits, this error would result in
considerable degradation of the logic noise-margins.
Figure 2 shows a situation where the digital return current
modulates the analog return current (top figure). The ground
return wire inductance and resistance is shared between the analog
and digital circuits; this causes the interaction and resulting error.
Analog Dialogue 46-06, June (2012)
A possible solution is to make the digital return current path
directly to the GND REF, as shown in the bottom figure. This
illustrates the fundamental concept of a “star,” or single-point
ground system. Implementing the true single-point ground in
a system that contains multiple high-frequency return paths is
difficult. The physical length of the individual return current
wires will introduce parasitic resistance and inductance, making
it difficult to obtain a low-impedance ground at high frequencies.
In practice, the current returns must consist of large area ground
planes to obtain low impedance to high-frequency currents.
Without a low-impedance ground plane, it is almost impossible
to avoid these shared impedances, especially at high frequencies.
All integrated circuit ground pins should be soldered directly to the
low-impedance ground plane to minimize series inductance and
resistance. The use of traditional IC sockets is not recommended
with high-speed devices. The extra inductance and capacitance of
even “low profi le” sockets may corrupt the device performance by
introducing unwanted shared paths. If sockets must be used with
DIP packages, as in prototyping, individual “pin sockets” or “cage
jacks” may be acceptable. Both capped and uncapped versions
of these pin sockets are available. They have spring-loaded gold
contacts, which make good electrical and mechanical connection
to the IC pins. However, multiple insertions may degrade their
performance.
ID
IA
INCORRECT
VD +
VA +
VIN
GND
REF
DIGITAL
CIRCUITS
ANALOG
CIRCUITS
IA + ID
ID
ID
IA
CORRECT
VD +
VA +
VIN
GND
REF
ANALOG
CIRCUITS
DIGITAL
CIRCUITS
IA
ID
Figure 2. Digital currents flowing in analog
return path create error voltages.
Power supply pins should be decoupled directly to the ground
plane using low-inductance, ceramic surface-mount capacitors.
If through-hole mounted ceramic capacitors must be used, their
lead length should be less than 1 mm. The ceramic capacitors
should be as close as possible to the IC power pins. Ferrite beads
may also be required for noise fi ltering.
So, the more ground the better—right? Ground planes solve many
ground impedance problems, but not all. Even a continuous sheet
of copper foil has residual resistance and inductance, and in some
circumstances, these can be enough to prevent proper circuit
function. Figure 3 shows such a problem—and a possible solution.
Analog Dialogue 46-06, June (2012)
10cm
10cm
POWER CONNECTIONS THIS END
POWER CONNECTIONS THIS END
PRECISION
ANALOG
CIRCUITRY
(MAY BE
AFFECTED BY
0.7mV/cm IN
GROUND
PLANE)
0.038mm
THICK
GROUND
PLANE
VOLTAGE DROP
0.7mV/cm RIGHT
ACROSS BOARD
15A POWER
OUTPUT STAGE
GROUND
PLANE
PRECISION
ANALOG
CIRCUITRY
(ISOLATED
FROM
VOLTAGE
DROP IN
GROUND
PLANE)
SLIT IN
GROUND
PLANE
TO ISOLATE
PRECISION
CIRCUITRY
HEAT SINK
VOLTAGE DROP
1.0mV/cm
BUT NOT IN
REGION OF
PRECISION
CIRCUITRY
15A POWER
OUTPUT STAGE
HEAT SINK
Figure 3. A slit in the ground plane can
reconfigure current flow for better accuracy.
Due to the realities of the mechanical design, the power input
connector is on one side of the board, and the power output
section—which needs to be near the heat sink—is on the other
side. The board has a 100-mm wide ground plane and a power
amplifier that draws 15 A. If the ground plane is 0.038-mm thick
and 15 A flows in it, there will be a voltage drop of 68 μV/mm.
This voltage drop would cause serious problems for the groundreferenced precision analog circuitry sharing the PCB. The ground
plane can be slit so that high current does not flow in the precision
circuitry region; instead, it is forced to flow around the slit. This
can prevent a grounding problem (which in this case it does), even
though the voltage gradient increases in those parts of the ground
plane where the current flows.
One thing to defi nitely avoid in multiple ground plane systems
is overlapping the ground planes, especially analog and digital
grounds. This will cause capacitive coupling of noise from
one (probably digital ground) into the other. Remember that a
capacitor is made up of two conductors (the two ground planes)
separated by an insulator (the PC board material).
Grounding and Decoupling Mixed-Signal ICs
with Low Digital Currents
Sensitive analog components, such as amplifiers and voltage
references, are always referenced and decoupled to the analog
ground plane. The ADCs and DACs (and other mixed-signal ICs)
with low digital currents should generally be treated as analog
components and also grounded and decoupled to the analog ground
plane. At first glance, this may seem somewhat contradictory since
a converter has analog and digital interfaces and usually has pins
designated analog ground (AGND) and digital ground (DGND).
Figure 4 will help to explain this apparent dilemma.
3
VA
VD
FERRITE BEAD
A
VA
AIN/
OUT
D
VD
LP
LP
RP
RP
CSTRAY
ANALOG
CIRCUITS
DIGITAL
CIRCUITS DATA
A
CSTRAY
IA
ID
SHORT
CONNECTIONS
BUFFER
GATE OR
REGISTER
DATA
BUS
CIN ≈ 10pF
RP
AGND
A
R
B
RP
LP
SEE
TEXT
LP
DGND
A
A = ANALOG GROUND PLANE
VNOISE
D
D = DIGITAL GROUND PLANE
Figure 4. Proper grounding of mixed-signal
ICs with low internal digital currents.
Inside an IC that has both analog and digital circuits (an ADC or a
DAC, for example), the grounds are usually kept separate to avoid
coupling digital signals into the analog circuits. Figure 4 shows a
simple model of a converter. There is nothing the IC designer can
do about the wire bond inductance and resistance associated with
connecting the bond pads on the chip to the package pins, except
to realize it’s there. The rapidly changing digital currents produce
a voltage at Point B that will inevitably couple into Point A of the
analog circuits through the stray capacitance, CSTRAY. In addition,
there is approximately 0.2 pF of unavoidable stray capacitance
between every adjacent pin-pair of the IC package! It’s the IC
designer’s job to make the chip work in spite of this. However, in
order to prevent additional coupling, the AGND and DGND pins
should be joined together externally to the analog ground plane
with minimum lead lengths. Any extra impedance in the DGND
connection will cause more digital noise to be developed at Point B
which will, in turn, couple more digital noise into the analog circuit
through the stray capacitance. Note that connecting DGND to
the digital ground plane applies V NOISE across the AGND and
DGND pins, inviting disaster!
The name “DGND” tells us that this pin connects to the digital
ground of the IC. This does not imply that this pin must be
connected to the digital ground of the system. It could be better
described as the IC’s internal “Digital Return.”
It is true that the grounding arrangement described may inject a
small amount of digital noise onto the analog ground plane, but
these currents should be quite small and can be minimized by
ensuring that the converter’s output does not drive a large fanout
(they normally can’t, by design). Minimizing the fanout on the
converter’s digital port (which, in turn, means lower currents) also
keeps the converter’s logic transition waveforms relatively free of
ringing, minimizes digital switching currents, and thereby reduces
any coupling into the analog port of the converter. The logic supply
pin (V D) can be further isolated from the analog supply by the
insertion of a small lossy ferrite bead, as shown in Figure 4. The
internal transient digital currents of the converter will flow in
the small loop from V D through the decoupling capacitor and to
DGND (this path is shown in red on the diagram). The transient
digital currents will, therefore, not appear on the external analog
ground plane but are confi ned to the loop. The V D pin decoupling
4
capacitor should be mounted as close to the converter as possible
to minimize parasitic inductance. The decoupling capacitors
should be low inductance ceramic types, typically between 0.01 μF
(10 nF) and 0.1 μF (100 nF).
Again, no single grounding scheme is appropriate for all
applications. However, by understanding the options and planning
ahead, problems can be minimized.
Treat the ADC Digital Outputs with Care
It is always a good idea to place a data buffer adjacent to the
converter to isolate the digital output from data bus noise (Figure 4).
The data buffer also serves to minimize loading on the converter’s
digital outputs and acts as a Faraday shield between the digital
outputs and the data bus (Figure 5). Even though many converters
have three-state outputs/inputs, these registers are on the die;
they allow data pin signals to couple into sensitive areas, so the
isolation buffer still represents good design practice. In some cases,
it may even be desirable to provide an additional data buffer on
the analog ground plane next to the converter output to provide
greater isolation.
ADC
IC
ANALOG
INPUT
PORT(S)
NOISY
DATA
BUS
Figure 5. A high-speed ADC using a buffer/latch
at the output shows enhanced immunity to digital
data bus noise.
The series resistors (labeled “R” in Figure 4) between the ADC
output and the buffer register input help to minimize the digital
transient currents, which may affect converter performance. The
resistors isolate the digital output drivers from the capacitance of
the buffer register inputs. In addition, the RC network formed by
the series resistor and the buffer register’s input capacitance acts
as a low-pass fi lter to slow down the fast edges.
A typical CMOS gate, combined with PCB trace and a throughhole, will create a load of approximately 10 pF. A logic output slew
rate of 1 V/ns will produce 10 mA of dynamic current if there is
no isolation resistor:
∆i = C
1V
∆v
= 10 pF ×
*
= 10 mA
ns
∆t
(2)
A 500 Ω series resistor will minimize the transient output current
and result in rise- and fall-times of approximately 11 ns when
driving the 10 pF input capacitance of the register:
t r = 2.2 × t = 2.2 × R × C = 2.2 × 500 Ω × 10 pF = 11 ns
(3)
Analog Dialogue 46-06, June (2012)
VA FERRITE
BEAD
VA
A
A
SEE
TEXT
VA
The effect of sampling clock jitter on ADC SNR is given
approximately by Equation 4:
VD
D
VD
ADC
OR
DAC
AMP
A
VA
A
VOLTAGE
REFERENCE
A
R
AGND
DGND
A
A
SAMPLING
CLOCK
GENERATOR
A
BUFFER
GATE
OR
REGISTER
TO OTHER
DIGITAL
CIRCUITS
D
VA
A
A
D
=
ANALOG
GROUND PLANE
=
DIGITAL
GROUND PLANE
Figure 6. Grounding and decoupling points.
TTL registers should be avoided; they can appreciably add to the
dynamic switching currents because of their higher input capacitance.
The buffer register and other digital circuits should be grounded and
decoupled to the digital ground plane of the PC board. Notice that
any noise between the analog and digital ground planes reduces the
noise margin at the converter digital interface. Since digital noise
immunity is of the order of hundreds or thousands of millivolts, this
is unlikely to matter. The analog ground plane will generally not be
very noisy, but if the noise on the digital ground plane (relative to
the analog ground plane) exceeds a few hundred millivolts, then
steps should be taken to reduce the digital ground plane impedance
to maintain the digital noise margins at an acceptable level. Under
no circumstances should the voltage between the two ground planes
exceed 300 mV, or the ICs may be damaged.
Separate power supplies for analog and digital circuits are also
highly desirable. The analog supply should be used to power the
converter. If the converter has a pin designated as a digital supply
pin (V D), it should either be powered from a separate analog supply
or fi ltered, as shown in Figure 6. All converter power pins should
be decoupled to the analog ground plane, and all logic circuit
power pins should be decoupled to the digital ground plane, as
shown in Figure 6. If the digital power supply is relatively quiet,
it may be possible to use it to supply analog circuits as well, but
be very cautious.
In some cases, it may not be possible to connect V D to the analog
supply. Some high-speed ICs may have their analog circuits
powered by +5 V, but the digital interface is powered by +3.3 V or
less, to interface to external logic. In this case, the +3.3-V pin of
the IC should be decoupled directly to the analog ground plane. It
is also advisable to connect a ferrite bead in series with the power
trace that connects the pin to the +3.3-V digital logic supply.
The sampling clock-generation circuitry should be treated like
analog circuitry and also be grounded and heavily decoupled to the
analog ground plane. Phase noise on the sampling clock degrades
system signal-to-noise ratio (SNR); this will be discussed shortly.
Sampling Clock Considerations
In a high-performance sampled-data system, a low-phase-noise
crystal oscillator should be used to generate the ADC (or DAC)
sampling clock, because sampling clock jitter modulates the analog
input/output signal and raises the noise-and-distortion floor. The
sampling clock generator should be isolated from noisy digital
circuits and grounded and decoupled to the analog ground plane,
along with the op amp and the ADC.
Analog Dialogue 46-06, June (2012)
ª 1 º
20 log10 «
»
«¬ 2πft j »¼
SNR
R
(4)
where f is the analog input frequency, SNR is that of a perfect ADC
of infinite resolution, and the only source of noise is rms sampling
clock jitter, tj. Working through a simple example, if tj = 50 ps (rms),
and f = 100 kHz, then SNR = 90 dB, equivalent to approximately
15-bit dynamic range.
It should be noted that tj in the above example is actually the
root-sum-square (rss) value of the external clock jitter and the
internal ADC clock jitter (called aperture jitter). However, in most
high-performance ADCs, the internal aperture jitter is negligible
compared to the jitter on the sampling clock.
Since degradation in SNR is primarily due to external clock jitter,
steps must be taken to render the sampling clock as noise-free as
possible with the lowest possible phase jitter. This requires that
a crystal oscillator be used. There are several manufacturers of
small crystal oscillators with low-jitter (less than 5 ps rms) CMOScompatible outputs.
Ideally, the sampling clock crystal oscillator should be referenced to
the analog ground plane in a split-ground system. However, system
constraints may not permit this. In many cases, the sampling clock
must be derived from a higher frequency multipurpose system
clock that is generated on the digital ground plane. It must then
pass from its origin on the digital ground plane to the ADC on
the analog ground plane. Ground noise between the two planes
adds directly to the clock signal and will produce excess jitter.
The jitter can cause degradation in the signal-to-noise ratio and
produce unwanted harmonics.
DIGITAL GROUND PLANE
VD
LOW-PHASENOISE MASTER
CLOCK
D
ANALOG GROUND PLANE
VD
SAMPLING
CLOCK
SYSTEM CLOCK
GENERATORS
VD
METHOD 1
D
D
A
VD
VA
D
A
DSP OR MICROPROCESSOR
SAMPLING
CLOCK
METHOD 2
D
SNR = 20 log10
ª« 1 º»
¬«2π f tj ¼»
tj = SAMPLING CLOCK JITTER
f = ANALOG INPUT FREQUENCY
Figure 7. Sampling clock distribution from digital
to analog ground planes.
This can be relieved somewhat by transmitting the sampling
clock signal as a differential signal, using either a small RF
transformer—as shown in Figure 7—or a high-speed differential
driver and receiver. If the latter are used, they should be ECL to
minimize phase jitter. In a single +5-V supply system, ECL logic
can be connected between ground and +5 V (PECL), with the
outputs ac-coupled into the ADC sampling clock input. In either
case, the original master system clock must be generated from a
low-phase-noise crystal oscillator.
5
The Origins of the Confusion About Mixed-Signal Grounding
Most data sheets for ADCs, DACs, and other mixed-signal
devices discuss grounding relative to a single PCB, usually the
manufacturer’s own evaluation board. This has been a source of
confusion when trying to apply these principles to multicard or
multi-ADC/DAC systems. The recommendation is usually to split
the PCB ground plane into an analog plane and a digital plane,
with the further recommendation that the AGND and DGND
pins of a converter be tied together and that the analog ground
plane and digital ground planes be connected at that same point,
as shown in Figure 8. This essentially creates the system “star”
ground at the mixed-signal device. All noisy digital currents flow
through the digital power supply to the digital ground plane and
back to the digital supply; they are isolated from the sensitive
analog portion of the board. The system star ground occurs where
the analog and digital ground planes are joined together at the
mixed-signal device.
While this approach will generally work in a simple system, with
a single PCB and a single ADC/DAC, it is not optimum for
multicard mixed-signal systems. In systems having several ADCs
or DACs on different PCBs (or even on the same PCB), the analog
and digital ground planes become connected at several points,
creating the possibility of ground loops and making a singlepoint “star” ground system impossible. For these reasons, this
grounding approach is not recommended for multicard systems;
the approach discussed earlier should be used for mixed-signal
ICs with low digital currents.
the ground return between Via 1 and Via 2 would be zero, and
the voltage appearing across the current source would also be zero.
AC + DC
U-SHAPED TOP CONDUCTOR
SCHEMATIC
Figure 9. Schematic and layout of current source
with U-shaped trace on PCB and return through
ground plane.
This simple schematic hardly begins to show the underlying
subtleties, but an understanding of how the current flows in the
ground plane from Via 1 to Via 2 discloses the realities and shows
how ground noise in high-frequency layouts can be avoided.
VIA 2
VIA 1
DC CURRENT PATH
VA
VD
MIXEDSIGNAL
DEVICE
ANALOG
CIRCUITS
AGND
SYSTEM
STAR
GROUND
A
GROUND
PLANE
ON BACK
DC
VD
VA
SOLID SHEET OF
GROUND PLANE
ON BOTTOM
VIA 1
VIA 2
IN
GND
PLANE
ON TOP TRACE
DIGITAL
CIRCUITS
DGND
A
D
D
AREA OF LOOP FOR
CALCULATING INDUCTANCE
ANALOG
GROUND PLANE
DIGITAL
GROUND PLANE
A
D
ANALOG
SUPPLY
DIGITAL
SUPPLY
Figure 8. Grounding mixed-signal ICs: single PCB
(typical evaluation/test board).
Figure 10. DC current flow for PCB shown in Figure 9.
The dc current flows in the manner shown in Figure 10, taking
the path of least resistance from Via 1 to Via 2 in the ground
plane. Some current spreading occurs, but little current flows at
a substantial distance from this path. In contrast, the ac current
takes the path of least impedance, which, in turn, depends on
inductance.
I
OPPOSING
OUTSIDE
LOOP
Grounding for High-Frequency Operation
The “ground plane” layer is often advocated as the best return
for power and signal currents, while providing a reference node
for converters, references, and other subcircuits. However, even
extensive use of a ground plane does not ensure a high-quality
ground reference for an ac circuit.
The simple circuit of Figure 9, built on a two-layer printed circuit
board, has an ac + dc current source on the top layer connected to
Via 1 at one end and to Via 2 by way of a single U-shaped copper
trace. Both vias go through the circuit board and connect to the
ground plane. Ideally, the impedance in the top connector and in
6
FLUX
LINES
AROUND
BOTTOM
WIRE
FLUX LINES
AROUND
TOP WIRE
REINFORCING
INSIDE
LOOP
OPPOSING
OUTSIDE
LOOP
I
Figure 11. Magnetic field lines and inductive loop
(right hand rule).
Analog Dialogue 46-06, June (2012)
GROUND PLANE
ON BACK
VIA 2
VIA 1
AC
VIA 2
VIA 1
GROUND-PLANE
CURRENT PATH
AC
GROUND-PLANE CURRENT PATH
PATH IS UNDER TOP TRACE
AREA OF INDUCTOR LOOP
TOP TRACE CURRENT PATH
TOP TRACE CURRENT PATH
Figure 12. AC current path without resistance (left) and with resistance (right) in the ground plane.
Inductance is proportional to the area of the loop made by the
current flow; the relationship can be illustrated by the right hand
rule and the magnetic field shown in Figure 11. Inside the loop,
current along all parts of the loop produces magnetic field lines
that add constructively. Away from the loop, however, field lines
from different parts add destructively, thus the field is confi ned
principally within the loop. The larger the loop, the greater
the inductance, which means that, for a given current level, a
larger loop has more stored magnetic energy (Li2) and greater
impedance (XL = jωL), and, hence, will develop more voltage at
a given frequency.
Which path will the current choose in the ground plane? Naturally,
the lowest-impedance path. Considering the loop formed by the
U-shaped surface lead and the ground plane, and neglecting
resistance, the high-frequency ac current will follow the path with
the least inductance, hence the least area.
In the example shown, the loop with the least area is quite
evidently formed by the U-shaped top trace and the portion of
the ground plane directly underneath it. So while Figure 10 shows
the dc current path, Figure 12 shows the path that most of the ac
current takes in the ground plane, where it fi nds minimum area,
directly under the U-shaped top trace. In practice, the resistance
in the ground plane causes the low- and mid-frequency current
to flow somewhere between straight back and directly under the
top conductor. However, the return path is nearly under the top
trace at frequencies as low as 1 MHz or 2 MHz.
two signal conductors, and the two ground return currents, flowing
in opposite sides of the ground plane as a result of skin effects, do
not interact.
With a multilayer board, both the crossover and the continuous
ground plane can be accommodated without the need for a wire
link. Multilayer boards are expensive and harder to troubleshoot
than simpler double-sided boards, but they do offer even better
shielding and signal routing. The principles involved remain
unchanged, but the range of layout options is increased.
The use of double-sided or multilayer PCBs with at least
one continuous ground plane is undoubtedly one of the most
successful design approaches for high-performance mixedsignal circuitry. Often the impedance of such a ground plane
is sufficiently low to permit the use of a single ground plane for
both analog and digital parts of the system. However, whether
or not this is possible does depend upon the resolution and
bandwidth required, and the amount of digital noise present
in the system.
‹ THIS VIEW FROM PCB
CONDUCTOR (TRACK) SIDE
‹ NOTE: RETURN CURRENTS
A AND B MAY INTERACT
BREAK IN GROUND PLANE
CROSSOVER “B” ON
GROUND PLANE SIDE
SIGNAL
CURRENT A
Be Careful with Ground Plane Breaks
Wherever there is a break in the ground plane beneath a conductor,
the ground plane return current must, by necessity, flow around
the break. As a result, both the inductance and the vulnerability
of the circuit to external fields are increased. This situation is
diagrammed in Figure 13, where Conductors A and B must cross
one another.
Where such a break is made to allow a crossover of two perpendicular
conductors, it would be far better if the second signal were carried
across both the first signal and the ground plane by means of a
piece of wire. The ground plane then acts as a shield between the
Analog Dialogue 46-06, June (2012)
SIGNAL CURRENT B
RETURN CURRENT B
DIVERTS AROUND
GROUND PLANE
BREAK, RAISING
INDUCTANCE
RETURN CURRENT A DIVERTS
AROUND GROUND PLANE BREAK,
RAISING INDUCTANCE
Figure 13. A ground-plane break raises circuit inductance and increases vulnerability to external fields.
7
(a)
(b)
Figure 14. AD8001AR evaluation board—top view (a) and bottom view (b).
NO CAPACITOR
WITH CAPACITOR
T
T
1
1
CH1 100mV𝛀
M 10.0ns
CH1
–66mV
CH1 100mV𝛀
M 10.0ns
CH1
–66mV
VERTICAL SCALE: 100mV/DIV
HORIZONTAL SCALE: 10ns/DIV
Figure 15. Effects of 10-pF stray capacitance on the inverting input on amplifier (AD8001) pulse response.
In one other instance, less is more. High-frequency, currentfeedback amplifiers are very sensitive to capacitance around their
inverting inputs. An input trace running next to a ground plane
can have just the sort of capacitance that may cause problems.
Remember that a capacitor consists of two conductors (the trace
and the ground plane) separated by an insulator (the board and
possible solder mask). To that end, ground planes should be cut
back from the input pins, as shown in Figure 14, which is an
evaluation board for the AD8001 high-speed current feedback
amplifier. The effect of even small capacitance on the input of
a current feedback amplifier is shown in Figure 15. Note the
ringing on the output.
It is mandatory that at least one layer of the PC board be dedicated
to ground plane! The initial board layout should provide for
nonoverlapping analog and digital ground planes, but pads and
vias should be provided at several locations for the installation of
back-to-back Schottky diodes or ferrite beads, if required—and
also so that the analog and digital ground planes can be connected
together with jumpers if required.
Grounding Summary
The AGND pins of mixed-signal devices should, in general, always
be connected to the analog ground plane. An exception to this
is DSPs with internal phase-locked loops (PLLs), such as the
ADSP-21160 SHARC® processor. The ground pin for the PLL
is labeled AGND but should be connected directly to the digital
ground plane for the DSP.
There is no single grounding method that will guarantee
optimum performance 100% of the time. This article presents a
number of possible options, depending upon the characteristics
of the particular mixed-signal devices in question. When laying
out the initial PC board, it is helpful to provide for as many
options as possible.
The material presented in this article was compiled from
many contributors, including James Bryant, Mike Byrne,
Walt Jung, Walt Kester, Ray Stata, and the engineering staff
at Analog Devices.
8
Acknowledgment
Analog Dialogue 46-06, June (2012)
References
Barrow, Jeff. “Avoiding Ground Problems in High Speed Circuits.” RF Design, July 1989.
Morrison, Ralph. Grounding and Shielding Techniques. 4th Edition.
John Wiley & Sons, Inc., 1998. ISBN: 0471245186.
Morrison, Ralph. Solving Interference Problems in Electronics.
John Wiley & Sons, Inc., 1995.
Barrow, Jeff. “Reducing Ground Bounce in DC-to-DC Converters—Some Grounding Essentials.” Analog Dialogue. Vol. 41,
No. 2, pp. 3-7, 2007.
Motchenbacher, C. D. and J. A. Connelly. Low Noise Electronic
System Design. John Wiley & Sons, Inc., 1993.
Bleaney, B & B.I. Electricity and Magnetism. Oxford at the
Clarendon Press, 1957: pp. 23, 24, and 52.
Op Amp Applications Handbook. Edited by Walt Jung. Newnes,
2005. ISBN 0-7506-7844-5.
Brokaw, Paul. AN-202 Application Note. An IC Amplifier User’s
Guide to Decoupling, Grounding and Making Things Go Right for a
Change. Analog Devices, 2000.
Ott, Henry W. Noise Reduction Techniques in Electronic Systems.
2nd Edition. John Wiley & Sons, Inc., 1988. ISBN: 0-471-85068-3.
Brokaw, Paul and Jeff Barrow. AN-345 Application Note.
Grounding for Low- and High-Frequency Circuits. Analog Devices.
The Data Conversion Handbook. Edited by Walt Kester. Newnes,
2005. ISBN 0-7506-7841-0.
Johnson, Howard W. and Martin Graham. High-Speed Digital
Design. PTR Prentice Hall, 1993. ISBN: 0133957241.
Kester, Walt. “A Grounding Philosophy for Mixed-Signal Systems.”
Electronic Design Analog Applications Issue, June 23, 1997: pp. 29.
Kester, Walt and James Bryant. “Grounding in High Speed
Systems.” High Speed Design Techniques. Analog Devices, 1996:
Chapter 7, pp. 7-27.
Linear Circuit Design Handbook. Edited by Hank Zumbahlen.
Newnes, February 2008. ISBN 978-0-7506-8703-4.
Montrose, Mark. EMC and the Printed Circuit Board. IEEE Press,
1999 (IEEE Order Number PC5756).
Analog Dialogue 46-06, June (2012)
Rempfer, William C. “Get All the Fast ADC Bits You Pay For.”
Electronic Design. Special Analog Issue, June 24, 1996: pp. 44.
Rich, Alan. “Shielding and Guarding.” Analog Dialogue. Vol. 17,
No. 1, pp. 8, 1983.
Sauerwald, Mark. “Keeping Analog Signals Pure in a Hostile
Digital World.” Electronic Design. Special Analog Issue, June 24,
1996: pp. 57.
Author
Louis “Hank” Zumbahlen [[email protected]] has
worked at ADI since 1989, originally as a field applications
engineer based in California. For the last several years, he has
been involved with training and seminar development as a senior
staff applications engineer. Previously, he held a
similar position at Signetics (Philips)—and
positions as a design engineer at several companies,
primarily in the test and measurement areas.
Hank has a BSEE from the University of Illinois.
He is the author of Linear Circuit Design Handbook
(Newnes-Elsevier 2008).
9
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