a AD7714/AD7715 Instrumentation Converter FAQs: Analog Performance

a  AD7714/AD7715 Instrumentation Converter FAQs: Analog Performance
AD7714/AD7715 Instrumentation Converter FAQs:
Analog Performance
How is the programmable gain function implemented?
Programmable gain is implemented using a combination of multiple input sampling per modulator cycle and scaling of the
ratio of reference capacitor to input capacitor. Programmable gain is implemented using switched capacitor techniques. This
is achieved by altering the sample rate of the input capacitor in the analog modulator. This technique is shown below.
V ref
Φ 1 C ref Φ 2
C int
V in
C in
Programmable Gain Implementation.
In this circuit, the reference capacitor Cref is used to sample the reference voltage Vref using non-overlapping clocks φ1 and
φ2 at the modulator sampling frequency fmod. The input capacitor Cin samples the analog input voltage again using nonoverlapping clocks φ3 and φ4 with a sampling frequency of fin. If fmod is equal to fin, the implemented gain is equal to the
ratio of Cin to Cref. With the timing shown in figure 3, fin = 2fmod, thereby implementing a gain of 2 when Cin = Cref.
What is the function of the CLK bit in the Clock register and how does it influence the performance of the
Refers to the AD7715 only, the AD7714 does not have a CLK bit. The CLK bit should be set to ‘1’ if the Master clock
frequency is greater than 2MHz and should be cleared to ‘0’ if the master clock frequency is equal to or less than 2MHz. If
this bit is set as recommended in the datasheet, the part will meet all the datasheet specifications in terms of power
consumption, noise, accuracy and output update rate. If the CLK bit is set incorrectly, the part will not meet the datasheet
The CLK bit is used to boost the current within the ADC when it is operating with the higher master clock so that the device
will meet the datasheet specifications. If the part is used with a master clock less than 2 MHz and the CLK bit is set to 1, the
part will still operate but it will draw more power than necessary. Also, the filter notches may deviate from the locations
described in the datasheet. If the AD7714/15 is operated with a master clock in excess of 2MHz and the CLK bit equals 0, the
modulator will not be supplied with enough current and the settling time for the modulator will increase. At frequencies
slightly over 2MHz (with CLK = 0), gain errors will occur and the noise will increase. As the frequency is increased further,
the part will eventually stop working as the modulator will refuse to operate with the reduced current.
The datasheet mentions that the analog input of the ADC can accept bipolar inputs of +/-30mV centered
around 0V when the buffer is not used. Can a 3/5V single supply part really do this?
Yes!! A basic outline of an analog input channel is shown in Figure 3 of the AD7714/AD7715 datasheet. Not shown in the
figure are ESD protection diodes that are connected from each analog input to the power supply rails (VDD and GND). The
input stage is truely differential so, in theory, it can tolerate negative input voltages. However, leakage through the ESD
protection diodes limits the ability to handle negative voltages. The diodes are Schottky type and will turn on when
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Analog Performance
approximately 300 mVis across them. This turn on voltage is temperature dependant. The effect of leakage through these
diodes is to increase the noise at the input and hence reduce the overall resolution.
Analog Devices guarantee full datasheet performance for analog inputs of +-30 mV over the full industrial temperature range
and, after performing some additional testing, guarantee performance for analog inputs of +-200 mV at 25 deg C.
When the ADC is operated with the inputs shorted, different output codes are obtained depending on
whether the part in operated in buffered or unbuffered mode. Is this normal?
If an internal offset and gain calibration is performed when the buffer is selected or de-selected, the same code should be
obtained from the ADC in both buffered and unbuffered mode within the noise limit of the part. If a calibration is not
performed, it is quite possible to see a different output code in buffered and unbuffered mode for the same input voltage. The
buffer is similar to an operational amplifier, therefore it will have an offset voltage. When the user switches between
buffered and unbuffered mode, a calibration is essential to remove this offset.
It’s important to remember that a calibration must be performed whenever the filter word, input range or the mode of the
buffer is changed. The datasheet includes noise tables that list the expected noise and code spread for the various operating
I’m measuring the input leakage current in unbuffered mode. It is significantly higher than the 1nA typ.
quoted in the datasheet. Why is this?
Static leakage current is specified in the datasheet. The leakage current is typically 1 nA in buffered mode or in unbuffered
mode when the ADC is in idle mode. The dynamic leakage current (the leakage current when the ADC is converting) is not
If you put the ADC into unbuffered mode and measure the input current, you will observe a combination of leakage current
and the dynamic current charging and discharging the sampling capacitor. If the dynamic charging current is likely to be a
problem in your application, for example, if you have a high source impedance, you can use the on-chip buffer to isolate the
input from these dynamic charging currents. However, this will result in reduced input common mode voltage. The buffer
input common mode range is 1.5V below VDD and 50mV above AGND compared to a common mode range of 30mV above
VDD and 30mV below AGND for unbuffered mode.
The datasheet mentions that large external capacitors between the input and ground can affect
measurement accuracy in unbuffered mode. Can you explain why this is the case? Does it only affect AC
input signals or are DC input signals affected too?
If you use the ADC in unbuffered mode, large RC constants on the input can interact with the internal sampling capacitor,
and effectively starve the sampling cap of charging current. This will cause gain errors in the ADC. The solution is either to
use buffered mode or ensure that you respect the maximum RC values given in Table XIV on page 16 of the datasheet.
The reference inputs are also unbuffered so, as with using the analog inputs in unbuffered mode, the RC loading on the
reference inputs must be sufficiently low to avoid introducing errors into the conversion process.
How does switching between channels on the AD7714 effect throughput?
The output data rate, which is listed in the datasheet, is the rate at which valid conversions are available when continuous
conversions are being performed on a single channel. When the user switches to another channel, additional time is required
for the sigma delta modulator and digital filter to settle. The settling time associated with these converters is the time it takes
the output data to reflect the input voltage following a channel change. To accurately reflect the analog input following a
channel change, the digital filter must be flushed of all data pertaining to the previous analog input. The digital filter on these
converters is a sinc^3 filter. Therefore, it takes three times the programmed update rate (conversion time) to clear the filter.
Therefore, if the output data rate is 5 Hz, for example, the time required to generate a valid conversion after switching
channels is (1/(3*5 Hz)).
When a channel change occurs, the digital filter and modulator are automatically reset, DRDY goes high and will remain high
until a valid conversion is available from the ‘new’ analog input channel. Therefore, following a channel change, DRDY will
remain high until the digital filter has calculated a valid conversion i.e. it will remain high for 3 conversion cycles.
When a step change occurs (on the analog input channel being converted), the ADC is not reset. The ADC continues to
output conversions and DRDY continues to pulse when a conversion is valid. However, the conversions will not be valid as
the digital filter will require 3 conversion periods to generate a digital word relevant to the altered analog input. If the step
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Analog Performance
change occurs at the beginning of a conversion cycle, the ADC will output a valid word 3 conversion cycles later. However,
if the step change occurs asynchronously so that it occurs in the middle of a conversion, the ADC needs to complete the
present conversion and then perform 3 more conversions to generate an output valid to the ‘new’ analog input. Therefore, it
may take 4 conversion cycles from the instant at which the step change occurs to the instant at which a valid conversion is
In summary, the channel switching speed is one-third the data output rate. Therefore, in switching applications such as data
acquisition systems, it is important to realise that the rate at which conversions are available is three times less than the
conversion rate achieved when continuously sampling a single channel.
How does the AD7714 compare with the AD7731 in multiplexed applications?
The AD7731 offers much faster channel-to-channel throughput rates and better noise performance than the AD7714. The
max throughput rate achievable with the AD7714 in channel switching applications is 3 ms giving a performance level of
between 8 and 8.5 bits peak-peak across its input ranges. The AD7731 offers a max throughput rate in excess of 2 kHz with
11 bits peak to peak performance on all input ranges. The following tables show the relative performance of the two parts for
selected input conditions.
Maximum Throughput Time for 16-Bit Peak-Peak Performance
+/-1.28V I/P Range
+/-20mV I/P Range
Peak-to-Peak Resolution @ 5ms Throughput Rate
+/-1.28V I/P Range
AD7731 (CHP=0)
+/-20mV I/P Range
In CHP=1 mode, the AD7731 offers significant noise improvement over the AD7714 for a given data update rate. The
following table shows the comparison between the AD7714 and AD7731 for a 50Hz update rate.
Rms Noise Performance at 50Hz Update Rate
+/-1.28V I/P Range
+/-20mV I/P Range
The AD7731 reference input includes the supply voltages (5V) while on the AD7714, the reference input voltage is limited to
2.5V with a 5V power supply. The AD7731 also offers a reference detect circuit to indicate when a reference voltage is not
present at the input. The AD7714 does not offer this feature. Another feature that the AD7731 offers is a fast step mode. The
fast step mode is a filter option used to give a fast indication of the input at low resolution.
The advantages that the AD7714 offers over the AD7731 are in power consumption and lower operating power supply
voltages. In order to achieve the lower noise and higher performance, the AD7731 takes much higher current in its analog
section. The normal mode power dissipation on the AD7731 is 76.5mW. The normal mode power dissipation of the AD7714
can be as low as 1.65mW. The AD7731 only operates with a 5V power supply while the AD7714 is specified with supplies
from 3V to 5V. The AD7714 therefore offers significant power benefits for smart transmitter or low power applications.
Is there any suggested protection schemes against ESD that should be considered with these products?
These converters are manufactured on a standard CMOS process and, therefore, all standard practices and protection schemes
apply to these devices as with all other CMOS devices. There are ESD protection diodes on all the inputs that protect the
device from possible ESD hits due to handling and production. These ESD protection diodes will act to clamp the voltage at
any pin to within 0.5V of the supplies. They can carry quite high currents but only for a short period of time so, they can
protect the IC from large pulses of short duration (the total energy is still quite low). The Latchup current is typically 100mA
on all pins.
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The maximum DC current that these protection diodes can withstand is 10mA. Therefore, the maximum current that can be
applied to any input is 10 mA. If it is possible for a current in excess of 10 mA to be applied to a pin due to an overvoltage,
external protection is required. Protection schemes that can be applied include transzorbs on the power supply lines, series
resistors on digital input lines, and resistors and diodes on analog inputs. For example, the external protection could be a
resistor in series with the input pin to limit the current into the pin to less than 10 mA. For example, if the maximum
overvoltage applied to a pin will be 5V, a 1kOhm series resistor in each line will limit the current to 5 mA.
There are a number of application notes and seminar material etc available on this topic. These are available on the Analog
Devices web site:
1) AN-202 : IC Amplifier User’s Guide to Decoupling, Grounding and Making Things go Right for a Change.
2) AN-311 : How to reliably protect CMOS circuits against power supply overvoltage.
3) AN-397 : Electrically Induced Damage to Standard Linear Integrated Circuits.
4) Overvoltage Affects on Analog ICs
What about susceptibility to conducted and radiated electromagnetic emissions?
Any sigma delta ADC will be susceptible to conducted RF into either the inputs, the power supply pins or into the reference.
The reason is that spurious RF signals and their harmonics can be averaged by the sigma-delta modulator and show up as a
DC offset or an increase in the noise floor. Radiated RF is a little more difficult to discuss but, similar problems can occur
and, there are situations where it is necessary to shield the sigma-delta ADC in a system from large RF fields generated
locally within the system.
The amount of protection required will depend on the strength of the local field. There are no hard and fast rules when
designing for EMC compatibility as every system will be different but, there are general guidelines that can be followed.
Consider the inputs, reference and power supply pins and ensure that each of these lines are properly filtered up to the
required maximum frequency. Decoupling capacitors on the power supply, mounted locally to the IC, possibly a small
inductor between the analog and digital supplies, filtering on the Reference and the inputs is also critical. A solid low
impedance ground plane, and separation of the analog and digital grounds - all the usual good practices with the ground plane
running under the whole of the IC. The evaluation board provides a good starting point.
Occasionally it is necessary to provide a Faraday shield for an ADC if the part is operating in the presence of high EM fields
such as next to a power supply or relay or RF transmitter but, this is an exceptional case.
As a component manufacturer, Analog Devices do not perform EMC testing as a general rule since EMC is a system level
specification rather than a component specification. It is the responsibility of the PCB designer to ensure that sensitive parts
of the circuitry are protected from spurious signals. We don’t have guaranteed bullet proof EMC design that we can give to
customers but if you use the evaluation board and follow standard practices for layout, grounding and decoupling, it is
possible to design a system which meets the CE mark and beyond without expending too much design effort. The final
chapter in all our seminar books is dedicated to hardware design techniques and deals with such issues as grounding,
decoupling, parasitic thermocouples and good PCB design.
Explain the converter noise with respect to the noise tables in the data sheets and the sources of this
The noise tables in the data sheet show the output rms noise for the selectable notch and output data rate for the part. The
numbers given are for the bipolar input ranges with a specified reference and VDD supply used. These noise numbers are
typical and are generated at an analog input voltage of 0V based on 1000 conversion results at the specified update rate. The
rms noise numbers are also converted to effective resolution in bits. These numbers can be represented as effective resolution
in bits rms or bits peak-to-peak. Effective resolution in bits rms is defined as the magnitude of the output rms noise with
respect to the input fullscale (2*Vref/Gain). It is important to note that the peak-to-peak numbers represent the resolution for
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Analog Performance
which there will be no code flicker. They are not calculated based on rms noise but on peak-to-peak noise. Peak to peak noise
is 6.6* Rms noise. Bits (peak-to-peak) = Effective Bits (rms)-2.5.
The numbers given are for bipolar input ranges. For the unipolar ranges, the rms noise numbers will be the same as the bipolar
range but the peak-to-peak resolution is now based on half the signal range that effectively means loosing 1 bit of resolution.
The output noise comes from two sources. The first is the electrical noise in the semiconductor devices (device noise) used in
the implementation of the modulator. Secondly, when the analog input is converted into the digital domain, quantization
noise is added. The device noise is at a low level and is independent of frequency. The quantization noise starts at an even
lower level but rises rapidly with increasing frequency to become the dominant noise source.
For example, consider the AD7714 operating with a 10 Hz update rate, a 5 V power supply and a 2.5V reference. With a
gain of 128, the rms noise equals 0.36 uV as given in the datasheet.
When operated in bipolar mode, the fullscale analog input is 5V/128=39mV. The peak-to-peak noise is 6.6 * 0.36uV = 2.376
uV peak to peak. Therefore, the ratio of full-scale input to peak-to-peak noise is 39 mV / 2.376 uV = 16414 which is the
resolution in counts. Converting this to bits peak to peak, log (16414) / log 2 = 14 bits peak-to-peak rounded to the nearest 0.5
What is the peak-to-peak resolution if the ADC is interfaced to a Transducer which generates an analog
output of 0 to 15 mV?
(fullscale range=0 to 20mV in unipolar mode)15 mV / 6.6 * 0.36 uV = 6313 counts which equates to 12.6bits peak to peak.
Therefore there will be no code flicker at the ADC’s digital output to the 12-bit level.
Are there benefits to post filtering the data from a converter?
Improvements in noise performance can be obtained using post filtering. The on-chip modulator of these converters
provides samples at 19.2 kHz to the digital filter when fCLK IN is 2.4576 MHz. The on-chip digital filter decimates these
samples to provide data at an output rate that corresponds to the programmed output rate of the filter. Since the output data
rate is higher than the Nyquist criterion (which states that the output data rate must be at least twice the bandwidth), the
output rate for a given bandwidth will satisfy most application requirements. However, there may be some applications that
require a higher data rate for a given bandwidth and noise performance. Applications that need this higher data rate will
require some post-filtering following the digital filter.
For example, if the required bandwidth is 7.86 Hz but the required update rate is 100 Hz, the data can be taken from
the converter at the 100 Hz rate giving a -3dB bandwidth of 26.2 Hz. Post-filtering can then be applied to reduce the
bandwidth and output noise to the 7.86 Hz bandwidth level, while maintaining an output rate of 100 Hz.
Post-filtering can also be used to reduce the output noise from the device for bandwidths below 1.3 Hz. At a gain of 128 and
a bandwidth of 1.3 Hz, the output rms noise is 260 nV when using the AD7714 converter in buffered mode. This is
essentially device noise or white noise that has primarily a flat frequency response. By reducing the bandwidth below 1.3
Hz, the noise in the resultant passband can be reduced. A reduction in bandwidth by a factor of 2 results in a reduction of
approximately √2 in the output rms noise. This additional filtering will reduce the system throughput.
What output coding is used in unipolar and bipolar mode?
The output coding is straight binary in unipolar mode and offset binary in bipolar mode.
Unipolar mode (Binary Coding): With an analog input voltage of 0V, the output
code is 000000Hex for the AD7714 and 0000Hex for the AD7715.
With an.analog input voltage of Vref/Gain, the output code is FFFFFFHex for the AD7714 and is FFFFHex
for the AD7715.
The output code for any analog input voltage can be represented as follows:
Code = (AIN * GAIN *2n)/Vref
Where AIN is the analog input voltage and n = 16 for 16-bit operation
(AD7715) and 24 for 24-bit operation (AD7714).
Bipolar mode (offset Binary Coding): With an analog input voltage of (-Vref/gain), the output code is
000000 Hex for the AD7714 and 0000 Hex for the AD7715
With an analog input voltage of 0V, the output code is 800000Hex for the AD7714 and 8000Hex for the
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With an analog input voltage of (+Vref/gain), the output code is FFFFFF Hex for the AD7714 and
FFFFHex for the AD7715.
Note that the analog inputs are pseudo bipolar inputs and the absolute analog input voltage must remain
within the common mode input range at all times.
The output code for any analog input voltage can be represented as follows:
Code = 2n-1*[(AIN * GAIN/Vref)+1]
Where AIN is the analog input voltage and n = 16 in 16-bit mode (AD7715) and n = 24 in 24-bit mode
What is the difference between fully- differential and pseudo-differential operation on the AD7714?
The analog inputs to the AD7714 can be configured to operate as either 3-fully differential input channels or 5 pseudo
differential input channels. In fully differential input mode, the independent input channels are AIN1/AIN2, AIN3/AIN4 and
AIN5/AIN6. In this operating mode, the input signal is represented by the difference between the two input terminals
representing the input channel. The common mode voltage can sit anywhere within the range of the ADC and the converter
will convert the signal seen between the two inputs. For example, if the AD7714 is configured for a fully differential input
and converting the signal from a bridge transducer excited with 5V, the common mode voltage will typically be 2.5V with
the displacement voltage usually a few mV sitting on this. The AD7714 in fully differential mode will only convert the
differential voltage and the common mode voltage is rejected. When the AD7714 is operated with three fully differential
channels, each of the inputs can have a different common mode voltage and the AD7714 converts only the difference
between the two input nodes.
The AD7714 can also be configured for operation as 5 pseudo differential channels. With this configuration, all 5 input
channels are referred to a common input - AIN6. All channels operate in a differential fashion but, since they are all
referenced to a single common point, they all must operate with the same common mode voltage. Pseudo differential
operation offers the advantage of having 5 input channels in an application provided that they are all referenced to a common
input point.
The user must ensure that the analog input voltage in either pseudo or fully differential mode is within the common mode
input range for the ADC for the given configuration selected. Pseudo-differential or fully differential mode of operation is
configured using the CH bits in the Communications register.
CH 5
CH 4
CH 3 CH 2 CH 1
Pseudo Differential
Fully Differential
Pseudo-Differential Vs Fully Differential Input Configuration
What is the recommended common mode input range for the analog and reference inputs?
The common mode range for the reference input is from GND to AVDD. The reference input is
unbuffered and, therefore, the common mode input range includes the supplies. The recommended
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reference is 2.5V when the device is operated with a 5V supply and 1.225 V for 3V operation. These
ADCs will not operate with REFIN(+) at 5V and REFIN(-) at GND.
For the analog inputs, the common mode range is GND – 30 mV to AVDD + 30 mV in unbuffered mode. In buffered mode,
there is a restriction on the maximum and minimum input voltage due to limitations on the input buffer. This restricts the
absolute and common mode input voltage to AGND + 50 mV to AVDD - 1.5 V.
How is the input range for the AD7714/AD7715 set?
The input range is dependent on the reference voltage, unipolar /bipolar mode selection and the gain setting chosen.
In unipolar mode, the input range is given as [ Vref(+) - Vref (-) ] / GAIN.
In bipolar mode, the input range is given as + [ Vref(+) - Vref (-) ] / GAIN.
In pseudo-differential mode (AD7714 only), input AIN6 is used as a common analog input and the voltages on inputs AIN1
to AIN4 are referenced to the voltage on AIN6. Again, the common mode voltage can be varied using AIN(6) provided the
full-scale and zero-scale voltages for each channel remain within the absolute input voltages (between
Can an analog input range other than those specified in the datasheet be used?
Yes – the analog input range can be altered. There are two methods to do this:
A system calibration can be performed during which the user applies the user-specifed zero-scale and
full-scale voltages to the ADC during the calibration process. The user must ensure that the ful-lscale
value used is within 0.8VREF/gain and 1.05VREF/gain for specified operation. The ADC will then
continue to meet the noise values listed in the datasheet for the original voltage range. For example, the
AD7714 has an rms noise spec of 1.5 uV in buffered mode when operated with a 5V power supply, a 10
Hz update rate and a gain of 1 which results in a signal range of 0 to 2.5 mV in unipolar mode. If the
new full-scale value is within 0.8 * 2.5V and 1.05 * 2.5V, the rms noise will continue to be 1.5 uV.
2) The reference voltage can be altered. For example, if an analog input range of 0 to 1.5V is required,
using a reference voltage of 1.5V will result in an analog input of 0 giving an output code of 000000Hex
and an analog input of 1.5 V giving a code of FFFFFFHex. Following a self-calibration with the 1.5 V
reference supplying the ADC, the rms noise will be the same as that for a 2.5 V reference that is given in
the datasheet. Using the example above, the AD7714 has an rms noise of 1.5 uV when operated with a
10 Hz update rate, 5 V power supply, a 2.5 V reference and buffered mode. If the reference voltage is
changed to 1.5 V and a self-calibration is performed, the rms noise will continue to be 1.5 uV assuming
the update rate, power supply, etc remain unchanged. The peak-to-peak resolution equals 18 bits when
operated with a 2.5V reference. With a 1.5 V reference, the peak to peak resolution equals log (2 V / 6.6
* 1.5uV) / log 2 = 17.5. The reduction in peak-to-peak resolution is due to the reduced input signal span.
What crystals do ADI recommend using with the sigma-delta ADCs and which parameters are important?
A low drift, high accuracy crystal should be used with the AD7714 and AD7715. With any sigma-delta ADC, the output data
rate and –3 dB point are directly related to the master clock frequency. The initial accuracy determines the output data rate
and, hence, the notch locations as the filter notches occur at integer multiples of the output data rate. Low drift ensures that
the output data rate and, hence, the filter notch locations do not move considerably from the desired locations. For example,
with a 10 Hz output data rate, the AD7714 has notches at 50 Hz and 60 Hz that allows the device to reject mains-injected
noise. If the crystal frequency changes, these notches will move and the rejection at 50 Hz and 60 Hz may degrade. A
company such as CMAC manufacture low drift crystals with high initial accuracy. .
Can I use a ceramic resonator to drive a sigma delta ADC?
Resonators can be used with the AD7714 and AD7715. However, resonators are a poor relation to crystals. The frequency
accuracy and temperature drift is much worse on resonators. Also, all the filter notches scale in proportion to the master clock
frequency. Therefore, if a notch is placed at 50Hz and the master clock frequency moves with temperature, the notch
frequency will also move with temperature. However, some customers are using Analog Devices’ sigma delta converters
with ceramic resonators and can live with the poor drift performance.
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I am using the AD7714/AD7715 in my system. When I look at the output of the ADC, I don’t get the full
resolution; a few of the LSBs are flickering. Why is this happening?
While the AD7714 has a resolution of 24 bits and the AD7715 has a resolution of 16 bits, the accuracy of the part varies with
update rate and gain. The datasheet lists the rms noise and achievable accuracy for different update rates and gains. For
example, the AD7715 has a peak-to-peak resolution of 16 bits when the gain is 1 (analog input range of +VREF), an update
rate of 50Hz and operated in bipolar mode. If the update rate is changed to 250 Hz, the peak-to-peak resolution is reduced to
13 bits. If the gain is now changed to 128 (+20 mV input range), the peak-to-peak resolution is reduced to 12 bits.
The accuracy of an ADC is specified as effective resolution or peak-to-peak resolution. The effective resolution is calculated
using the rms noise that is given in the datasheet. The effective resolution equals log (input span / rms noise) / Log 2. The
peak-to-peak resolution is the number of bits which do not flicker and is calculated using the peak-to-peak noise which
equals 6.6 * rms noise. Therefore, the peak-to-peak noise equals log (input span / (6.6 * rms noise)) / Log 2.
The datasheet rms noise values are measured with the chosen analog input channel shorted to some voltage such as
VREF(both terminals of the analog input channel are connected to VREF). Therefore, the user should short the analog input
on their system board using a similar method to ensure that the best performance is being obtained from the part. After
performing a calibration, commence conversions. Using several thousand samples, the rms noise can be calculated and, from
these,.the peak to peak resolution can be determined. The obtained value should be compared with the values given in the
datasheet. If the accuracy specified in the datasheet is not obtained, this is due to noise on the circuit board, for example,
ensure that ground loops do not exist and ensure that the power supply is adequately decoupled using a 10uF tantalum
capacitor in parallel with a 0.1uF ceramic capacitor from each power supply to its respective ground. The capacitors should
be placed as close as possible to the ADC’s pins.
With a constant DC input, the output of the ADC drifts with temperature. Why?
The performance of any ADC varies with temperature. When a calibration is performed at a temperature, the offset error and
gain error are minimized at the temperature at which the calibration is performed. However, the offset error and gain error
vary with temperature. For example, if you short the analog inputs to some voltage such as the references voltage and
perform a calibration, the peak to peak resolution given in the datasheet for the specific update rate and gain should be met.
If the temperature varies, the ADC output will vary as the offset error and gain error have drifted with temperature. These
drifts are specified in the datasheet.
There will also be drift in the remainder of the signal chain. For example, the reference voltage will vary with temperature
also. This variation is specified in the voltage reference datasheet. Resistance values also will vary with temperature.
Therefore, it is important to use an accurate reference that has low drift along with resistors with tight tolerances.
Solder joints on a circuit board will also have thermal qualities. Each solder joint connects two dissimilar metals which
generates a small thermocouple. Therefore, signal paths for analog signals such as the analog inputs should be kept identical.
This will ensure that affects outside the ADC on the analog input lines due to temperature variation are matched and will be
removed as a common mode affect.
I plan to use the AD7714 in a 32-channel data acquisition system with 2 of the AD7714 inputs being used
for system calibration. There are four 8 channel multiplexers connected to the Ain1-Ain4 inputs (pseudo
differential ) with channel AIN5 being used as the pseudo-ground connection and AIN6 being used for the
system calibration. This means that system calibration is performed on one of the Ain channels. What
errors can I expect when using the calibration coefficients from the channel for all the other channels
An 8-channel multipliexer is connected to channels AIN1 to AIN4. Channel AIN5 is used as the pseudo-ground connection
while channel AIN6 is used for calibration. If all the channels use the same gain and update rate, a self-calibration on
channel AIN6 will be valid for channels AIN1 to AIN4 also. For a system calibration, the offset and gain coefficients will be
the same for all channels also if each analog input sees the same external circuitry. Therefore, identical multiplexers should
be connected to channels AIN1 to AIN4. An identical multiplexer should also be used on channel AIN6.
If all the channels use the same external circuitry, the main source of error will be mismatch between the multiplexers and the
temperature drift mismatch of any external signal conditioning components such as resistors. The AD7714 datasheet gives
the expected performance for the different gains and update rates. If all the channels were matched identically, these noise
specifications would be obtained. By using resistors with low temperature coefficients and identical multiplexers, the errors
introduced by the external components can be minimized.
The integral non linearity is expressed as % of Full-scale. For what gain setting is this valid?
The INL spec refers to the ADC only (not the PGA) so this assumes a gain of 1.
Instrumentation Converter Tech Note
Ver 1.06 10/03
AD7714/AD7715 Instrumentation Converter FAQs:
Analog Performance
What is the link between random conversion noise, peak-peak and RMS noise given in the datasheet?
Assume that the noise is truly random and is described by a normal distribution (white noise). Then:
Vnoise (peak to peak) = Vnoise (rms) x 6.6; for 99.9% of the time.
Refer to the Technical Note ‘Peak to Peak Resolution vs. Effective Resolution at
for more detailed information.
Do I need an anti-alias filter for my sigma-delta converter?
Yes, an anti-alias filter is required. However, because a sigma-delta converter oversamples the analog input, the design of the
anti-alias filter is greatly simplified compared to an ADC that samples at the Nyquist rate (Maximum signal Bandwidth * 2).
For example, the AD7714’s modulator samples the analog input at a frequency of Master Clock) / 64 for a gain of 1 which
equals 38.4kHz for a Master Clock Frequency of 2.4576MHz (see table XIV in the AD7714 datasheet).
The AD7714 and the AD7715 have a programmable low pass digital filter. Figure 4 in the AD7714 datasheet shows the
frequency response for an output data rate of 10 Hz. As this is a digital filter, the frequency response is reflected around the
sampling frequency. This means that the filter will provide 0dB of attenuation at frequencies which are integer multiples of
the sampling frequency. Therefore, an anti-alias filter in the analog domain is required to adequately attenuate these
frequencies; usually a single pole (possibly a 2 pole) RC filter is all that is required. For example, if the 3dB bandwidth of
the AD7714 is set to 10Hz and the sampling frequency is at 10kHz, a single pole RC filter would give 60dB of attenuation at
the sampling frequency.
When the sigma-delta converter is operated in unbuffered mode, the inputs look directly into the sampling capacitor of the
modulator. The modulator is continually charging and discharging the sampling capacitor. If the time constant of the antialiasing filter is too large, the modulator may be unable to fully charge the sampling capacitor and gain errors will result. To
prevent the R-C combination from introducing errors, the datasheet for each ADC specifies the maximum allowable R and C
values that can be used for the different gain settings in unbuffered mode.
Which reference should be used with the AD7714/AD7715?
A low noise reference source is required to achieve the best performance from the ADC. When using the ADC with a 5V
power supply, suitable references include the AD780, REF43 and REF192. When operating the ADC in 3V mode, the
AD589 or AD1580 are suitable. It is recommended to decouple the output of these references to further reduce the noise
In applications such as pressure measurement systems or weighscales, the excitation voltage for the bridge can be used to
derive the reference voltage for the ADC also. The excitation voltage can be divided using a resistor network to generate a
2.5V or 1.225V reference. In these applications, the affect of the noise in the excitation voltage will be removed as the
application is ratiometric.
Ver 1.06 10/03
Instrumentation Converter Tech Note
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