ANALYSIS AND DESIGN OF A ZERO VOLTAGE ... DC-DC BOOST CONVERTER FOR PHOTOVOLTAIC (PV) ENERGY SYSTEM

ANALYSIS  AND DESIGN  OF A ZERO  VOLTAGE ... DC-DC BOOST CONVERTER  FOR PHOTOVOLTAIC  (PV) ENERGY  SYSTEM
ANALYSIS AND DESIGN OF A ZERO VOLTAGE TRANSITION
DC-DC BOOST CONVERTER FOR PHOTOVOLTAIC (PV)
ENERGY SYSTEM
A THESIS SUBMITTED IN PARTIAL FULFILMENTS OF THE
REQUIREMENTS FOR THE AWARD OF THE DEGREE OF
Master of Technology
in
Industrial Electronics
Department of Electrical Engineering
By
VEMULA ANUSHA
Roll No: 212EE5259
Under the Guidance of
Dr. MONALISA PATTNAIK
Dr. B. CHITTI BABU
Department of Electrical Engineering
National Institute Technology, Rourkela-769008
1
NATIONAL INSTITUTE OF TECHNOLOGY
ROURKELA
CERTIFICATE
This is to certify that the project entitled “ANALYSIS AND DESIGN OF A ZERO
VOLTAGE TRANSITION DC-DC BOOST CONVERTER FOR PHOTOVOLTAIC (PV)
ENERGY SYSTEM” submitted by Vemula Anusha (212ee5259) in partial fulfillment of the
requirements for the award of Master of Technology degree in Industrial Electronics,
Department of Electrical Engineering at National Institute of Technology, Rourkela is an
authentic work carried out by her under my supervision and guidance.
To the best of my knowledge the matter embodied in this thesis has not been submitted to
any other university/Institute for the award of any Degree.
Date:
(Dr. Monalisa Pattnaik)
Place: Rourkela
Department of Electrical Engineering
NIT Rourkela
2
ACKNOWLEDGEMENT
On the submission of my thesis entitled “Analysis and Design of a Zero Voltage
Transition DC-DC Boost Converter for PV Energy System” I would like to extend my
gratitude and sincere thanks to my supervisor Dr. Monalisa Pattnaik, Asst. professor, Dept. of
Electrical Engineering for her constant motivation and support during the course work. I am very
thankful to her for giving me good basics in PV during the course work, which makes a good
part of the project.
I am sincerely thankful to Dr. B. Chitti Babu, for helping me give a good start for the
work. I truly appreciate and value his esteemed guidance and encouragement in the beginning.
I would like to thank all others who have consistently encouraged and gave me moral
support, without whose help it would be difficult to finish this project.
I would like to thank my parents and friends for their consistent support throughout.
3
ABSTRACT
India being the world’s third largest power producer and consumer is still considered to
have unreliable electrical infrastructure. It is estimated that about 27% of the energy generated is
stolen or lost in transmission. During the 2012 grid failure, some villages that were not
connected to grid were not affected, such as Meerwada located in Madhya Pradesh because it has
a 14KW solar power station. The photovoltaic (PV) energy systems are gaining popularity
because the systems are being developed and designed to extract maximum energy from the sun
in most efficient way and feed it to the loads without affecting their performance.
In this thesis, a boost converter operating all the switching devices under Zero Voltage
Transition is studied and a model converter which can supply a load of 250W is designed and is
used in a PV energy system. In this converter topology, a part of the circuit resonates for a small
portion of the switching cycle of the converter, known as the auxiliary circuit that enhances the
soft transition from ON state to OFF state and vice versa, thus improving the converter efficiency
by reducing the dominating portion of in losses i.e. the losses that occur due to hard transition of
the switches. Due to reduced losses during switching transitions heating effect of MOSFETs is
reduced and they have a longer life. The comparative study between the new topology and
conventional hard switching converter is analyzed in terms of improvement of efficiency and
reduction of switching losses.
4
INDEX
Abstract………………………………………………………………………………………….. 4
Index……………………………………………………………………………………………... 5
List of the tables made…………………………………………………………………………… 8
List of the figures shown...……………………………………………………………………...... 9
List of abbreviations used……………………………………………………………………..... 11
List of the symbols used...……………………………………………………………..……...... 12
CHAPTER-1
1.1
Introduction……………………………………………………………………………... 15
1.2
Literature Review…………………………………………………………………….......16
1.3
Motivation………………………………………………………………………………. 17
1.4
Objectives…………………………………………………………………………...….. 18
CHAPTER-2
2.1
Boost converter for PV energy system…………………………………….…………… 19
2.2
Losses in hard-switching converters……………………………………………………. 20
2.3
Soft switching techniques………………………………………………………………. 21
2.4
2.3.1
Zero Current Switching…………………………………………………. 21
2.3.2
Zero Voltage Switching………………………………………………… 22
ZVT converters…………………………………………………………………………. 23
5
CHAPTER-3
ZVT DC-DC BOOST CONVERTER
3.1
Circuit description and its novelty……………………………………………………… 25
3.2
Circuit operation………………………………………………………………………... 26
3.3
Theoretical waveforms………………………………………………………………….. 31
3.4
Converter design………………………………………………………………………... 33
3.5
3.4.1
Design of the power circuit……………………………………………... 34
3.4.2
Design of the auxiliary circuit…………………………………………... 36
Simulation of converter…………………………………………………………………. 39
CHAPTER-4
PHOTO VOLTAIC ARRAY
4.1
4.2
4.3
Introduction…………………………………………………………………………….. 40
4.1.1
PV cell………………………………………………………………………….. 40
4.1.2
PV module……………………………………………………………………… 41
4.1.3
PV array………………………………………………………………………… 41
Modeling of PV system………………………………………………………………… 42
4.2.1
PV cell modeling……………………………………………………………….. 42
4.2.2
PV array modeling……………………………………………………………… 44
PV array simulation…………………………………………………………………….. 46
CHAPTER-5
MAXIMUM POWER POINT TRACKING
5.1
Introduction……………………………………………………………………………... 47
5.2
Different types of MPPT algorithms……………………………………………………. 48
6
5.3
P&O algorithm………………………………………………………………………….. 48
5.4
Flow chart……………………………………………………………………………..... 49
CHAPTER-6
RESULTS AND DISCUSSION
6.1
Simulation results of converter ………………………………………………………… 50
6.2
Loss calculation and comparative study……………………………………………….. 55
6.2.1
Losses in soft switching converter……………………………………………… 55
6.2.2
Losses in conventional hard switching converter………………………………. 57
6.3
Simulation results of PV array and MPPT...…………………………………………..... 58
6.4
Conclusions……………………………………………………………………………... 60
REFERENCES
7
LIST OF TABLES
Table.No.
Name of the table
Page.No.
3.1
converter specifications
33
3.2
component specifications for simulation
39
4.1
Parameters of the simulated PV module
46
6.1
Duty cycle variation w.r.t. input voltage
54
6.2
Auxiliary RMS current at various voltages
54
6.3
component values for conventional boost converter
55
6.4
comparison of soft switching with hard switching topology
58
8
LIST OF FIGURES
Name of the Figure
Fig.No.
Page.No.
2.1
Block diagram of DC-DC converter with PV energy system
19
2.2
Switching losses in hard switching converters
20
2.3
(a) ZCS turn OFF using negative voltage
22
(b) Switching waveforms of hard switching and ZCS during turn OFF
22
(a) ZVS turn ON using negative current
23
(b) Switching waveforms of hard switching and ZVS during turn ON
23
3.1
Schematic diagram of the converter
25
3.2
converter circuit in simple boost converter mode [t<t0 ]
26
3.3
Equivalent circuit for interval [t0 -t1 ]
27
3.4
Equivalent circuit for interval [t1 -t2 ]
28
3.5
Equivalent circuit for interval [t2 -t3 ]
28
3.6
Equivalent circuit for interval [t3 -t4 ]
29
3.7
Equivalent circuit for interval [t4 -t5 ]
29
3.8
Equivalent circuit for interval [t5 -t6 ]
30
3.9
Equivalent circuit for interval [t6 -t7 ]
30
3.10
Theoretical waveforms of the converter
31
3.11
ZVS interval for S1
32
4.1
photocurrent generation
40
4.2
Equivalent circuit of a practical PV cell
42
4.3
PV Module- representation of series parallel combination
44
2.4
9
4.4
Typical I-V curve
45
4.5
Typical P-V curve
45
5.1
Concept of maximum power point tracking
47
5.2
Flow chart for MPPT P&O algorithm
49
6.1
Auxiliary inductor current
50
6.2
Auxiliary capacitor voltage
50
6.3
Feed-forward capacitor voltage
51
6.4
Main switch voltage
51
6.5
Main switch current
51
6.6
ZVS turn ON of the S1
52
6.7
Reduced voltage turn OFF of S 1
52
6.8
ZCS turn ON of S2
53
6.9
ZVS turn OFF of S2
53
6.10
IV characteristics of the PV array
58
6.11
PV characteristics of the PV array
59
6.12
MPPT result
59
10
LIST OF ABBREVIATIONS
PV
Photo Voltaic
ZVT
Zero Voltage Transition
DC
Direct Current
MOSFET
Metal Oxide Semiconductor Field Effect transistor
PWM
Pulse Width Modulation
EMI
Electro-Magnetic Interference
ZVS
Zero Voltage Switching
ZCS
Zero Current Switching
MATLAB
MATrix LABoratory
MPPT
Maximum Power Point Tracking
P&O
Perturb and Observe
IC
Incremental Conductance
11
LIST OF SYMBOLS
CS
Snubber capacitance
VGS
Gate to source voltage of the MOSFET
VDS
Drain to Source voltage
ID
Drain current
Ton
Turn-ON time of the MOSFET
Toff
Turn-OFF time of the MOSFET
PSW
Switching Power losses
VS
Voltage blocked by the switch
IS
Flow of current allowed by the switch
fS
Switching frequency of the converter
Vin
Input voltage
S1
Main switch/boost switch
S2
Auxiliary switch
Lin
Input boost inductor
Lr
Auxiliary resonant inductor
Cr
Auxiliary resonant capacitor
Cb
Energy feed forward capacitor
Cs
Parasitic capacitance of the switch
D1 -D5
Diodes
C0
Output capacitor
t 0 -t 7
Time instants in one switching cycle
ILr
Current flowing through the auxiliary circuit
12
Vcr
Voltage blocked by the resonant capacitor
Vcb
Voltage blocked by the feed forward capacitor
Pout
Output power rating
Vout
Output voltage rating
Vrp
Output voltage ripple
ΔIrp
Input peak current ripple
η
Efficiency of the converter
Iin_pk
Input peak current
D
Duty cycle
fr
resonant frequency
Vb
Base voltage
Ib
Base current
Zr
Resonant impedance
Zr_pu
Per unit resonant impedance
Zrb
Base resonant impedance
Tr
Resonating time period of the auxiliary circuit
t rr
Reverse recovery time of the diode in the boost circuit
VS2_pk
Peak voltage across S2
K
constant
IS2_pk
Peak current of S2
IS2_rms
RMS current of S2
IS1_rms
RMS current of S1
IPV
PV current
I0
Diode’s reverse saturation current
VT
Diode’s thermal voltage
13
a
Diode’s ideality factor
KI
Temperature coefficient of Isc
KV
Temperature coefficient of Voc
G
Irradiance on the surface of the cell
G STC
Irradiance under STC
IPV_STC
Photocurrent generated under Standard Test Conditions (STC)
Eg
Energy gap of the semiconductor
I0_STC
Nominal saturation current
ISC_STC
Nominal value of Isc
VOC_STC
Nominal value of Voc
Ns
Number of modules in a serial connected string
Np
Number of strings that are connected in parallel
Vmp
Operating Voltage at MPP
Imp
Operating Current at MPP
Pmax
Maximum power
14
CHAPTER-1
1.1
INTRODUCTION
Usually, the converters operating under Zero Voltage Transition (ZVT) help solving the problem
of prohibitive Electromagnetic Interference (EMI) either by using a diode whose recovery
characteristics are not fast, to increase the turn OFF time switch present in the boost circuit,
which increases the switching losses [3], or by using passive snubber circuits which increase the
conduction losses [4] [5], thus reducing the converter efficiency and limiting the switching
frequency. So the problem of EMI is solved only at the cost of reduced efficiency. So there is a
need for highly efficient converters with reduced EMI.
The most important thing in the converter design is the positioning of the auxiliary
switch. If the source terminal of the switching MOSFET is not connected to the common point of
grounding in the circuit, we will need a floating gate drive, which demands an effective gate
voltage greater than the input voltage. A reduced stress of voltage and current peaks on the
switching devices is always recommended for safety of devices.
The principle of ZVT is that the auxiliary circuit carries a current higher than the input
current flowing through the boost inductor just for a fractional part of switching time, in order to
attain soft turn ON and OFF transitions of the main and auxiliary switches. So, these converters
have higher ohmic losses than the simple or conventional converters that do not operate under
soft transitions of switching. But the efficiency of the soft switching converters is high as the
losses due to hard switching in the soft switching converters are very low as compared to the
conventional hard switching converters. Also, as the auxiliary circuit it-self is soft switching and
due to the creative placing of the snubber capacitor which controls the ON to OFF transition of
the switch in the boost circuit, this converter reduces the EMI and increases the efficiency.
15
1.2
LITERATURE REVIEW
In conventional hard switching converters, the conduction losses are very low. But due to high
switching losses, the efficiency of these converters is low. So the technique of soft switching is
introduced to make the switching transitions at either zero voltage condition or zero current
condition, so that the dominating portion of losses (the ones caused due to switching under high
voltages or currents) can also be reduced and the efficiency of the converters can be highly
improved.
In [3] it has been made clear that the sources of major losses in a boost converter
operating in CCM are the diode and the inductor present in the part of the circuit that boosts the
voltage and the MOSFET used for switching. The efficiency is also affected by the input boost
inductor’s refined design and the OFF transition losses of the main switch. As the turn ON losses
of the switch depend on the reverse recovery characteristics of the diode in the boost circuit,
adopting a fast recovery diode reduces the turn-ON losses. But, use of a fast recovery diode
increases ringing effect and EMI. Further we need to use EMI filters which would increase the
complexity and may affect the improvement in efficiency achieved by using fast recovery diode.
In [4] the process of active turn-ON snubbing for the ZVT is studied. The use of an extra
switch to make easy discharge of resonant inductor to reduce voltage stress on main switch is an
alternative for passive snubbing. This configuration of active snubbing is usually referred to as
the ZVT boost converter. This paper has provided the basic idea of the implementing boost
converter with zero voltage transitions. But in this circuit only the boost circuit switch operates
in zero voltage transitions. But the auxiliary circuit switch operates on zero current transitions.
The resonant energy that has been stored in the inductance of the resonating circuit is circulated
to the load.
Many papers have been referred only to study the different methods adopted in the turnoff process of the boost switch present in the main circuit. In [2] the snubber capacitance C s
controls turns off process of the boost switch by controlling the rate of rise of voltage across the
switch. But the energy that has been stored in this capacitor is always dissipated in the auxiliary
circuit. This increased the conduction losses of the converter. In [5] the conduction losses are
16
slightly reduced by using an energy feed-forward circuit. In this circuit, some part of the resonant
energy is allowed to be directly fed to the connected load by making use of a transformer. The
circuit topology presented in this thesis has all its resonant energy dissipated into the load.
1.3
MOTIVATION
In recent days of continuously and rapidly growing power demand and but very slowly
improving supplying capacity, there are more chances of power outage or grid failures like in the
case of 2012 grid failure in India. Also most of the remote areas are not connected to the grid and
they do not have power supply. These areas can generate power on their own using renewable
resources such as solar energy.
The efficiency of the PV energy system solely depends on the PV panels, power
converter and the Maximum Power Point Tracking system. The efficiency of a single PV cell is
very low. The efficiency of hard switching converters is low. So they can be replaced by the soft
switching converters that have very less losses and high efficiency. Use of good and efficient
MPPT algorithm also improves the system efficiency. In this thesis, improvement of converter
efficiency is most focused on.
Usually ZVT converters solve the problem of EMI either by using slow recovery diodes
or by using passive snubbers which increase the conduction losses. So the reduction of EMI is
achieved only at the cost of reduced efficiency. So there is a need for more efficient converters
with less EMI. Also most of the soft switching converters involving auxiliary switch does not
attain soft switching of the auxiliary switch. So an auxiliary circuit which helps in soft switching
and also is soft switching is required.
17
1.4
OBJECTIVES
The main objective of this thesis is to study and analyze the operation of converter and design
the converter that can satisfy all the requirements that have motivated to take up this work. In
short the objectives of the thesis are listed as follows
1. To design a converter that has
a. Reduced switching losses
b. Reduced conduction losses
c. Less EMI
d. Reduced stress of voltage and current on the devices
2. Study the operation of the converter and verify it by waveforms and study the soft
switching of both the switches.
3. Implementation in PV environment along with MPPT
4. Calculation of losses and efficiency of the converter and comparative study with a simple
boost converter
18
CHAPTER-2
2.1
BOOST CONVERTER FOR PV ENERGY SYSTEM
The efficiency of a photovoltaic system is very low since the output of the PV array depends on
various environmental conditions most likely to be temperature and solar irradiation. Therefore,
there is a need for a system to condition the power output of the PV array before supplying it to
the domestic loads.
Figure 2.1: block diagram of DC-DC converter with PV energy system
Figure 2.1 represents a block diagram showing the use of a converter for PV energy
system. The PV array’s output is supplied to the load after being conditioned by the ZVT DCDC boost converter. The switching of the MOSFETs constituting the circuit is controlled by a
maximum power point tracking (MPPT) algorithm which tracks that operating point of the PV
array that meets the DC load line (including the effect of converter).
19
2.2
POWER LOSSES IN HARD-SWITCHING CONVERTERS
In the switching converters, when the switching device is in ON state, as the voltage blocked by
the switch is zero, the power losses are zero. When the switch remains in the off state, as the
current allowed by the switch is zero, the power losses are zero. But during the transition of the
switch from both ON state to OFF state and OFF state to ON state, if there is no mechanism to
make either voltage or current zero, power losses occur. This is in the case of hard-switching
converters.
In the hard switching converters, power losses will occur when there will be a
simultaneous non-zero voltage applied across and non-zero current flowing through the switch.
When the switching device turns ON or OFF, the device voltage and current are high in
simultaneous cases resulting in high losses. This is shown as waveforms in figure 2.2, (i)
showing control pulse given to the switching device, (ii) the device voltage and current and (iii)
power losses per switching cycle.
Figure 2.2: switching losses in hard switching converters
20
The power losses corresponding to a single switching transition are the product of the
voltage that appears across the terminals of the switch and the current flowing through the
switch. The entire switching losses are the product of energy or power lost per switching
transition and the switching frequency. The power losses that occur due to these switching
transitions are referred to as switching losses.
The switching losses in one switching cycle can be denoted in equation 2.1
 Ton  Toff
Psw  Vs I s f s 
2




(2.1)
From the above equation, the switching losses in any semiconductor device vary linearly
with switching frequency and delay times. Therefore such hard switching converters cannot be
used for high frequency switching applications. Though use of passive snubbers across the
switch reduces voltage stresses, the efficiency cannot be improved due to high switching losses.
From the equation of switching losses, it can be observed that the switching losses can be
reduced in 2 ways
i.
By reducing the delay times during turn ON and turn OFF, by using faster and more
efficient switches in converter.
ii.
By making the voltage across or current through the switch zero before turning it
ON/OFF, the concept of soft switching converters.
2.3
SOFT SWITCHING TECHNIQUES
There are two basic methods to attain soft switching, zero current switching (ZCS) and zero
voltage switching (ZVS), based on the parameter that is made zero, either the voltage or current
through the device.
21
2.3.1
ZERO CURRENT SWITCHING
A switch operating with ZCS has an inductor and a blocking diode in series with it. The
switch turns ON under ZCS as the rate of rise of current after the voltage becomes zero is
controlled by the inductor. As the inductor does not allow sudden change in current, it rises
linearly from zero.
When a negative voltage is made to appear across the combination of inductor and switch
using a resonant circuit, the current flowing through the switch is naturally reduced to zero which
results in the turn OFF of the switch under zero current switching.
Figure 2.3: (a) ZCS turn OFF using negative voltage
(b) Switching waveforms of hard switching and ZCS during turn OFF
2.3.2
ZERO VOLTAGE SWITCHIING
A switch operating with ZVS has an anti-parallel diode and a capacitor across it. During turn
OFF as the current reduces to zero, the rate of voltage rise that takes place across the switch is
controlled by the capacitor. As the capacitor does not allow sudden change in voltage, it rises
linearly from zero.
22
The turn OFF characteristics of the switch are controlled by a capacitor connected across
it. This capacitor reduces the voltage rise rate as current flow reduces to zero.
(a)
(b)
Figure 2.4: (a) ZVS turn ON using negative current
(b) Switching waveforms of hard switching and soft switching
2.4
ZERO VOLTAGE TRANSITION CONVERTERS
The ZVT converters accomplish zero voltage switching during both turn-ON and turn-OFF
transitions of the primary or boost switch.
The zero voltage transition in zero voltage switching converters is accomplished by
turning OFF the switch which has capacitor and a diode connected in parallel with it. As the
flow of current through the switch falls to zero, the capacitor maintains zero voltage across
the switch. Where as in zero voltage transition, as the switch turns OFF, the current in the
switch is transferred to the capacitor connected in parallel to it.
The turn ON transition in zero voltage switching is accomplished by discharging the
capacitor connected in parallel by making use of the energy stored in a magnetic circuit
23
element like a transformer winding or an inductor coil. The switch is turned ON after the
parallel diode enters into the state of conduction. This ensures a zero voltage across the
switch during transition.
There are various zero voltage switching techniques. Each one differs from other in the
techniques used to control and modulate to attain regulation and also in the mechanism of
storing energy to attain zero voltage turn ON.
24
CHAPTER-3
ZERO VOLTAGE TRANSITION DC-DC BOOST CONVERTER
3.1
CIRCUIT DESCRIPTION AND ITS NOVELTY
The circuit schematic of the zero voltage transition DC-DC boost converter is shown in Figure
3.1. It is just a simple boost converter with a diode D1 , input boost inductor Lin , main switch S1
and an output capacitor C 0 across a load Rload. In addition to the boost circuit, it also constitutes
of an additional circuit that resonates, consisting of an inductor Lr, a capacitor C r, diodes D2 -D5
and a capacitor C b to feed the resonant energy to the load. The capacitance C s shown across the
main switch S1 is its parasitic capacitance and not an external capacitance.
Figure 3.1: Schematic diagram of the ZVT dc-dc boost converter
The basic principle of Zero Voltage Transitions is that the auxiliary circuit carries a
current higher than that of the input supply current, for a small portion of the entire switching
cycle in order to attain soft switching of the switching elements present in the converter.
Therefore Zero Voltage Transition converters have higher ohmic losses than that of those
25
converters that operate under hard-switching. But the efficiency of the converters that operate
under soft switching is inflated unlike the hard switching converters on account of diminished
switching losses.
The innovative part of this circuit lies in the reduction of conduction losses along with
soft switching of all the switches main and the auxiliary ones, and also all the diodes. This is
achieved by using a capacitor that controls the turn-off characteristics of the main switch, whose
resonant energy all dissipated into the load. From figure 3.1 it is clear that the capacitor C b is the
feed-forward capacitor that controls the turn-off of the main/boost switch. The energy retained
by this capacitor is completely discharged into the load through the boost diode, either D1 or D3 .
So the rms current carried by the auxiliary circuit is reduced and so happens with the conduction
power losses, unlike the other topologies mentioned in the literature review.
3.2
CIRCUIT OPERATION
The operation of the circuit is explained for one complete switching cycle, which is split into
seven parts for easy understanding. Each of these cases is explained along with equivalent circuit
for that interval. Initially the boost diode D1 supplies the output current and the circuit acts as a
simple pulse width modulated boost converter. The circuit under this condition is shown in
figure 3.2.
Figure 3.2: [t<t0 ] converter circuit in simple boost converter mode
26
Interval 1 [t0 -t1 ]:
At instant t0 the switch S2 is switched ON with Zero Current Transition owing to the existence of
auxiliary resonant inductor serially connected to it. The current slowly starts to divert from the
diode D1 to that part of the circuit which supplements the main circuit, which eventually starts to
resonate. The resonant inductor decelerates the turn-off current rate through D1 that turns off
under ZCS by the end of this interlude. By this time the auxiliary current flowing through boost
inductor equals the input boost current. The equivalent circuit for this interval is manifested in
figure 3.3.
Figure 3.3: Equivalent circuit for interval [t0 -t1 ]
Interval 2 [t1 -t2 ]:
The auxiliary circuit current keeps on increasing all interval long. But the input current supplied
is assumed to be constant due to large inductance which does not allow sudden change in
current. So the parasitic capacitance C s of the main switch S1 starts discharging into the auxiliary
circuit in order to supply the increased portion of auxiliary current. By the end of this interval the
capacitance discharges completely. The equivalent circuit for this interval is manifested in figure
3.4.
27
Figure 3.4: Equivalent circuit for interval [t1 -t2 ]
Interval 3 [t2 -t3 ]:
After the instant t2 the diode internally present in the main switch connected anti parallel to it
starts conducting, which causes the voltage blocked by the main switch S1 to be zero. This is the
Zero Voltage duration during which the switch S1 must be supplied with the trigger. By the end
of this interval the current carried by the auxiliary circuit equals the input supply current and the
main switch is in a condition of about to start conduction. The equivalent circuit of the converter
for this mode is manifested in figure 3.5.
Figure 3.5: Equivalent circuit for interval [t2 -t3 ]
Interval 4 [t3 -t4 ]:
The auxiliary current declines the input supply current and the residue of the input supply current
after supplying with the auxiliary current starts flowing through the switch S 1 . By the end of this
28
interval the flow of current in the auxiliary circuit becomes zero. The equivalent circuit of the
converter for this mode is manifested in figure 3.6.
Figure 3.6: Equivalent circuit for interval [t3 -t4 ]
Interval 5 [t4 -t5 ]:
In this interval, the direction of current flow in the auxiliary part of the circuit changes and the
negative portion of the resonant cycle starts at this instant. Diode D4 which is in series the switch
makes the branch unidirectional and does not allow the switch S 2 to conduct and so this current
passes through the diode D5 creating a Zero Voltage turn-off condition for S2 . Meanwhile the
current from the diode D2 is rerouted to the capacitor C b which starts getting charged. The
equivalent circuit for this interval is manifested in figure 3.7.
Figure 3.7: Equivalent circuit for interval [t4 -t5 ]
29
Interval 6 [t5 -t6 ]:
As this interval starts at instant t5 , the auxiliary current becomes zero and the resonant cycle ends
here. The circuit begins to runs identical to a PWM boost converter operating in its charging
state. The equivalent circuit of the converter in this interval is manifested in figure 3.8.
Figure 3.8: Equivalent circuit for interval [t5 -t6 ]
Interval 7 [t6 -t7 ]:
At the beginning of this interval at instant t6 S1 is turned-off. The feed forward capacitor C b is
responsible for the slow voltage rise across S1 . Voltage across the capacitor C b reverse biases the
boost diode D1 and it cannot conduct. So the energy stored in the capacitor C b during the
resonant cycle i.e. the auxiliary circuit energy discharges through diode D3 and when this voltage
reaches zero the diode D1 starts conducting and the succeeding switching cycle gets initiated.
The equivalent circuit of the converter for this mode is shown in figure 3.9.
VS 1  0
Figure 3.9: Equivalent circuit for interval [t6 -t7 ]
30
3.3
THEORETICAL WAVEFORMS
Figure 3.10 shows the theoretical waveforms for the operation of the converter showing each
interval in the entire switching cycle. The variations of the resonant circuit current and voltage of
the inductor and capacitor respectively, switch voltages and currents (both main and auxiliary),
the feed forward capacitor voltage in each interval are shown clearly.
VG2
VG1
VS1
Is1
ILr
Vcr
Vcb
t0 t1
t2
t3
t4
t5
t6 t7
Figure 3.10: Hypothetical waveforms of the converter
31
It is to be noted that the resonant cycle (The time during which the auxiliary circuit
supplements the main circuit) is just a small part of the entire switching cycle. For easy
understanding the operation in resonant cycle is more focused on. The simple boost converter
operation interval i.e. [t5 -t6 ] is time compressed. Only that part of the waveform is compressed in
which there will be no change in the waveforms, or the variables remains constant.
For further detailing in the ZVS turn ON of the S1 , all the waveforms are put together on
common time axis and shown. Figure 3.11 shows the zero voltage interval during which the
main switch has to be turned ON. The waveforms of current carried by the resonant inductor,
voltage blocked by the resonant capacitor, voltage blocked by the feed forward capacitor and the
main switch voltage are overlapped for easy understanding.
Figure 3.11: ZVS interval for S1
32
3.4
CONVERTER DESIGN
Design objectives:
The converter is designed to meet the following objectives:
1. Minimize the switching losses the main switch.
2. Minimize the turn-off losses the main switch.
3. Reduce the EMI of the boost diode.
4. The auxiliary circuit resonant cycle must be kept as short as possible because with the
increase in the cycle length the losses in the auxiliary circuit increase.
Design specifications:
The specifications for the design of the converter are given in table 3.1. The specifications
include converter output power rating, input voltage range, output voltage, allowable ripple
percentage in current and voltage etc.
Sl.No.
Parameter
Specification
Value
1
Output power
P0ut
250W
2
Output voltage
V0ut
400V
3
Input voltage
Vin
90-265V
4
Switching frequency
Fsw
100kHz
5
Output voltage ripple
Vrp
1%
6
Input current peak ripple
ΔIrpp
20%
Table 3.1 Converter specifications for design
DESIGN PROCEDURE
The design procedure of this converter is divided into two parts:
I.
II.
Design of the boost or power circuit which is active for entire switching cycle.
Design of the auxiliary resonant circuit which is active for a resonant cycle which is just a
minor part of the switching cycle.
33
3.4.1 DESIGN OF POWER CIRCUIT
The power circuit consists of the main switch, boost diode, input inductor and the output
capacitor. Calculation of each circuit element values is shown very clearly.
Input inductor Lin:
The numerical value of the input inductor Lin , must be decided first because its value sets the
peak input current which the converter switches have to withstand and therefore this current is
necessary to decide the rating of other power circuit components. The maximum current without
ripple is
2
I in _ pk 
P0ut

Vin
250
0.95  4.135 A
90
2

(3.1)
The maximum peak-peak ripple current is
I rpp  I pk _ max  I  4.135  20%  0.827 A
(3.2)
Therefore the maximum peak input current with ripple is
I rpk _ max  I pk _ max 
I rpp
2
 4.135 
0.827
 4.55 A
2
(3.3)
The duty ratio of the converter when the maximum current occurs is
D pk  1 
2  Vin _ min
V0
 1
2  90
 0.682
400
(3.4)
The input inductor value is calculated as follows
Lin 
2  Vin _ min  D pk
I rpp  Fsw

2  90  0.682
 1050H
0.827  100kHz
34
(3.5)
Where Fsw is the switching frequency
Output capacitor:
The output capacitor acts as an energy storage element. It stores energy when the input voltage
and current are near their peak and provides this energy to the output load when the line is low.
The point of reference for selection of this capacitor is the endurable ripple in the output voltage.
The peak charging current of the capacitor is
I chg _ pk 
P0ut 250

 0.625 A
V0ut 400
(3.6)
The voltage ripple across C 0 is
Vchg _ pk 
C0 
I chg _ pk
2    f r  C0
I chg _ pk
2    f r  Vchg _ pk

0.625
 207 F
2    120 Hz  (0.01  400)
(3.7)
Boost diode:
The maximum voltage across the boost diode will be the output voltage V0 =400V which appears
across the diode when the main switch remains in the conducting state. The peak current that
flows through the diode is the peak with ripple of the current flowing through boost inductor i.e.
I rpk _ max  4.55 A . The average current flowing through the diode is
I D1 _ avg 
P0 250

 0.625 A
V0 400
(3.8)
The peak current rating of the main boost switch depends upon the auxiliary circuit. So it is
designed after the auxiliary circuit.
35
3.4.2 DESIGN OF THE AUXILIARY CIRCUIT AND MAIN SWITCH
Base values:
The base voltage is defined as:
Vb  V0  400V
(3.9)
The base current is defined as:
I b  I pk _ max 
I rpp
2
 4.135 
0.827
 3.722
2
(3.10)
Therefore the base impedance is defined as:
Z rb 
Vb
400

 107.48
I b 3.722
(3.11)
The base time is defined as the natural resonant cycle of the auxiliary circuit and is given as:
Tr  2    Lr  Cr
(3.12)
The worst case condition is where the ZVS interval will be least which occurs when the input
boost current is at its maximum peak. At this value of peak current the impedance Zrb will be 1
p.u. and the auxiliary circuit has to be designed for this value only.
Resonant Inductor Lr:
The selection of the inductor Lr is made keeping in mind that the D1 ’s reverse recovery current is
to be made zero. Therefore selection of resonant inductor depends on the boost diode’s turn-off
di/dt and this can be controlled by slowly rerouting the current flowing through it to resonant
inductor.
With increase in the value of the inductor the rise time of the current flowing through it
increases which in turn decreases the reverse recovery current of D1 . But this increases the
resonant cycle Tr , which results in an increase of ohmic losses due to increase in the rms currents
of the auxiliary circuit. So a compromise must be made in the selection of resonant inductor.
Lr is chosen such that it lets the auxiliary current to increase up to the input peak current
Iin_pk within 3 times the reverse recovery time t rr of D1 that is specified.
36
The boost diode must be an ultra-fast recovery diode with as low value of t rr as possible
as a slower diode requires a larger value of Lr.. So an ultra-fast diode which will satisfy all
voltage and current requirements and have minimum t rr is selected. Assuming the value of
trr=30ns the value of Lr can be calculated as follows
Lr 
3  t rr  Vs 2 _ pk
Ib

3  30ns  (0.7  400)
 5.8H  6H
3.722
(3.13)
Where Vs2_pk is the peak voltage that appears across S2 and is assumed 0.7 pu [1].
Resonant Capacitor C r :
The value of resonant capacitor Cr is selected from the graph of ZVS interval vs. resonant
impedance Zr. We have to choose Cr that will give adequate ZVS turn-on interval as well as
good turn-off. For proper design we select the curve K=3 and Zr=0.21 pu [1] and the value of Cr
can be determined as follows
Zr 
Lr
L
 C r  r2
Cr
Zr
Z r  Z r _ pu  Z rb  0.21  107.469  22.568  C r  11nF
(3.14)
Auxiliary capacitor C b:
ZVS at turn-off is provided by capacitor Cb. the selection of this capacitor is easy and is as
follows
K
Cr
3
Cb
Cb 
Cr
 3.66nF
K
(3.15)
37
Rating of the auxiliary switch:
From the values of K and Zr chosen it is found that the peak voltage blocked by auxiliary switch
Vs2_pk is found to be
Vs 2 _ pk  0.64 pu  400V  256V
(3.16)
The peak current flowing through the switch is found to be
I s 2 _ pk  1.61 pu.  3.722  5.99 A
(3.17)
The rms current of the switch is found to be
I s 2 _ rms  ( I ss2 _ rms , pu)  I b  Tr  Fsw  (0.53)  3.722  1.58s  100kHz  0.786 A
(3.18)
Rating of the auxiliary circuit diodes:
The auxiliary circuit diodes have the same voltage rating as that of boost diode. The two series
diodes D2 and D4 will conduct the same peak current as auxiliary switch S 2 . The peak current
carried by diode D5 is also somewhat same as the above. The peak current carried by diode D3 is
the peak current with ripple Irpk_max that flows in the converter was found to be 4.55A. The
average current through diode D2 is found to be
I D 2 _ avg  ( I D 2 _ avg , pu)  I b  Tr  Fsw  (0.21 pu)  (3.722 A)  1.587s  100kHz  0.12 A
(3.19)
Rating of the main switch:
The maximum voltage that this switch must be able to handle is the output voltage V0 with
ripple. The ripple present in the output voltage can be found as
Vchg _ pk  0.01  400  4V
(3.20)
Thus the switch S1 must handle 404 volts
38
The peak current carried by the main switch is 2.27 pu  3.722  8.448 A
(3.21)
The maximum rms current for the switch is found to be
I s1 _ rms  I pk _ max 
1 4  Vin _ min  2
1 4  90  2

 3.722 

 2.25 A
2
3    V0
2 3    400
(3.22)
The voltage across the capacitor C b for K=3 and Zr=0.21 is -0.87 pu. This means that the voltage
blocked by switch S1 during turn-off is 0.13 pu. Therefore voltage blocked by switch S1 is found
to be
0.13 pu  400  52V
Therefore the turn-off losses are also greatly reduced.
3.5
SIMULATION OF THE CONVERTER
The converter circuit shown in figure-1 is simulated in MATLAB simulink environment by
taking the component values shown in table 3.2.
Sl.no.
Circuit component
Symbol
specification
1
Resonant inductor
Lr
6µH
2
Resonant capacitor
Cr
15Nf
3
Feed-forward capacitor
Cb
3.5Nf
4
Boost inductor
Lin
1050µH
5
Output capacitor
C0
470µF
6
Input voltage
Vin
265V
7
Switching frequency
Fsw
100kHz
Table 3.2 Component specifications for simulation
39
CHAPTER-4
PHOTOVOLTAIC ARRAY MODELLING
4.1
INTRODUCTION
A Photovoltaic system plies solar modules or panels for converting solar energy to electrical
energy. The basic unit of a PV array is a PV cell.
4.1.1 PHOTOVOLTAIC CELL
This is similar to simple P-N junction devices. When sunlight hits the surface of the PV cell, the
photons are absorbed by the atoms in the semiconductor material and electrons are freed from
the negative layer. When this cell is connected to an external circuit, the free electrons find a
path to reach the positive layer. The current generation process is shown in figure 4.1
Figure 4.1 photocurrent generation
Detailed construction and working of a PV Cell:
PV cells are usually manufactured from various types of semiconductor materials using disparate
processes. In present days, the monocrystalline and polycrystalline are mostly found. Si cells
have a Si film that is connected to terminals of other devices. One side of the layer undergoes a
process of addition of impurities, usually called doping to materialize a
thin grid (metallic) is planted on the top of the PV cell which faces the sun.
40
P-N junction. A very
As the light is incident on the surface of the cell, charge carriers are generated, which
originates an electric current when the cell becomes a part of a loop or is connected to a load. As
the energy of the incident photon becomes sufficient to break the covalent bond and detach the
electrons of the semiconductor, charge carriers are generated. Photons that have lower energies
than the energy gap of PV cell are not of any use and they help generating no voltage. Whereas
Photons that have energy surpassing the band gap can produce electricity, but the energy
associated with the band gap is only made use of. The remaining energy will be dissipated in the
form of heat [10].
4.1.2
PHOTOVOLTAIC MODULE
The voltage generated by a single cell is very low around 0.5 volts. So a number of cells should
be connected in serial and parallel to achieve the desired output. Diodes may be needed in order
to avoid reverse current in the array, in case of partial shading.
4.1.3
PHOTOVOLTAIC ARRAY
The power generated by a single module may not be sufficient to supply the most of the
appliances. So a group of modules are connected in series which is generally used for high
voltage applications and in the same way they are connected in parallel, the connection which is
useful for high current applications.
41
4.2
4.2.1
MODELING OF PV SYSTEM
PV CELL MODELING
The single diode model of a single PV cell is manifested in the figure 4.2. It includes a current
source, a diode, parallel connected to the current source which represents the photocurrent, a
series resistance Rs and a parallel resistance Rsh .
An accurate single diode model is depicted in the above figure. Equation 3.1 represents the
current generated from the cell.
  V  IRS
I  I PV  I 0 exp 
  aVT
   V  IRS
  1  
   RP



(4.1)
Figure 4.2 Analytical circuit of a practical PV cell
Where
I0 is the diode’s reverse saturation current
VT is the diode’s thermal voltage
a is the ideality factor of the diode
The equation of a PV current as a concomitant of changing environmental conditions, the
temperature and irradiance can be written as
I PV  I PV _ STC  K I T 
G
GSTC
(4.2)
Where
42
IPV_STC is the photocurrent under Standard Test Conditions (STC)
ΔT=T-TSTC (in Kelvin) and TSTC=25°C
G is the irradiance on the surface of the cell
G STC is the irradiance under STC (1000W/m²)
KI is the short circuit current coefficient (generally provided by the manufacturer)
The equation for the saturation current of the diode is given as
 qE g  1
1 
T 

I 0  I 0 _ STC  STC  exp 
 
 T 
 ak  TSTC T 
3
(4.3)
Where
Eg is the energy gap of the semiconductor
I0_STC is the nominal saturation current
The reverse saturation current equation can be further improved as a function of temperature as
follows
I0 
I
SC _ STC
 K I T 
exp VOC _ STC  KV T  / aVT   1
(4.4)
Where
KV is the temperature coefficient of open circuit voltage
ISC_STC is the nominal short circuit current
VOC_STC is the nominal open circuit voltage
43
4.2.2 PV ARRAY MODELING
All the above equations are applicable for a single PV cell. But in a typical installation of a PV
power station, PV modules are used in which series and parallel connected PV cells are used in
order to bridge the supply demand gap. Series combination of the cells increases the voltage and
the parallel combination of the cells increases the current of the entire module
In such case, the output equation can be written as follows
 
 NS    
 N 
    V  IRS  S  
  V  IRS 

 N P    1  
 NP  
I  I PV N P  I 0 N P exp 




aVT N S
 N S  



 
   RP  N  
 P 
  
 
(4.5)
The configuration of modules in a series parallel structure is shown in figure 4.3.
NP
RS
NS
NP
Id NP
I PV N P
NS
RP
NS
NP
Figure 4.3: PV Module- representation of series parallel combination
44
The typical characteristics of a PV energy system are shown in the figures below. The
Photovoltaic characteristics include the current voltage (I-V) characteristics and the power
voltage (P-V) characteristics. The typical curves of these characteristics are shown. The point
Pmax in the I-V curve shows the point at which maximum power can be coerced. It represents the
voltage at the maximum power point from the curve. This maximum power point is tracked from
the P-V curve using different tracking techniques.
Figure 4.4: Typical I-V curve
Figure 4.5: Typical P-V curve
The peak point of the P-V curve gives the maximum power point of a PV cell. This point
will be different for different cells.
But the maximum power point tracking system tracks the
MPP of the system on a whole, i.e. all the cells connected. Recently research is going on
implementing MPP tracking devices for each cell.
45
4.3
PV ARRAY SIMULATION
The PV module model is simulated in MATLAB simulink using the above equations. The
parameters used for simulating the PV module are as shown in the table 4.1
Sl.No.
PARAMETER
SYMBOL
VALUE
1
Current at maximum power
Imp
7.61 A
2
Voltage at maximum power
Vmp
26.3 V
3
Short circuit current
Isc
8.21 A
4
Maximum power
Pmax
200.143 W
5
Open circuit voltage
Voc
32.9 V
6
Temperature coefficient of V
Kv
-0.1230 V/K
7
Temperature coefficient of I
Ki
0.0032 A/K
Table 4.1 Parameters of the simulated PV module
When a single module is simulated, the open circuit voltage of the module is found to be
around 30 volts. But this voltage is found not to be present within the input voltage range of the
designed converter. This voltage cannot be fed to the converter to check its operation in the PV
environment. So 6 modules of similar kind are connected in serial so that the output voltage
increases. When 6 modules are serial connected, then the output voltage of the array is raised to
almost 180 volts. This voltage of 180 volts is within the input range of the designed Zero Voltage
Transition DC-DC boost converter which is 90-265 volts.
46
CHAPTER-5
MAXIMUM POWER POINT TRACKING
5.1
INTRODUCTION
As the PV panel has non linear characteristics of voltage and current, MPPT algorithms are
required to improve the efficiency of the PV system by setting the operating point to MPP of the
characteristic curve. The only 3 major components of the PV energy system are the PV panels,
the converter and the MPP tracker. The efficiency improvement of the first two components is
that easy as it depends on the technology used and it may involve lot of cost. So the efficiency of
the entire PV system can be improved cost-effectively by using MPPT algorithms [10].
Fig 5.1: Concept of maximum power point tracking
Maximum power point (MPP) is an operating point on either IV or PV characteristic
curve of a PV array. It is that point of operation, where the power supplied to the load is
maximum. From maximum power transfer theorem, when the load of the converter is fixed and
the duty cycle of the converter is varied which in turns varies the effective load on the PV
47
system, maximum power can be coerced from the PV energy system. In this way changing the
slope of the load line and shifting the operating point and fixing it at the MPP, maximum power
can be coerced from the PV array. The concept of maximum power point tracking is shown in
figure 5.1
5.2 TYPES OF MPPT TECHNIQUES/ALGORITHMS
In recent years, many algorithms have been introduced to track maximum power point. They
differ from one another in aspects like complexity, efficiency and cost. Some of them are
1. Perturb and observe
2. Incremental conductance
3. Fuzzy logic control
4. Neural networks
5. Fractional open circuit voltage
6. Fractional short circuit current
7. Current sweep
8. Maximum power point current and voltage computation
9. State based MPP tracking technique
From all the above techniques, we use Perturb and observe algorithm for simplicity.
5.3 PERTURB AND OBSERVE ALGORITHM
The Perturb and Observe algorithm is a kind of hill climbing technique. In hill-climbing, the duty
cycle of the converter is perturbed and in P&O the DC link operating voltage or the voltage at
the PV panel output or the converter input terminals is perturbed.
In this technique, the present perturbation is decided by the signs of the previous
perturbations and increments. If the power is incremented by the last perturbation, then the
perturbation should be in the same direction, where as if it results in the decrement of power, the
48
direction of the perturbation should be changed. The perturbations are repeatedly carried out
until the MPP is attained.
5.4
FLOW CHART
The flow chart for the MPPT algorithm using P&O method is shown below in figure 5.2
D  D  D
D  D  D
D  D  D
Fig 5.2: Flow chart for MPPT P&O algorithm
49
D  D  D
CHAPTER-6
RESULTS AND DISCUSSION
6.1
SIMULATION RESULTS OF CONVERTER
The operation of the converter is verified and the waveforms of the auxiliary circuit elements for
a resonant cycle and the main switch current and voltage waveforms for one switching cycle are
shown below.
Fig 6.1: Auxiliary inductor current
Fig 6.2: Auxiliary capacitor voltage
50
Figures 6.1 and 6.2 show the resonating circuit waveforms for one resonant cycle which is a part
of the switching cycle. These waveforms are compared with the analytical waveforms shown in
the figure 3.10. Figure 6.3 shows the feed forward capacitor voltage waveform.
Fig 6.3: Feed-forward capacitor voltage
Fig 6.4: Main switch voltage
51
Fig 6.5: Main switch current
Figure 6.4 manifests the main switch voltage waveform and the figure 6.5 shows the
main switch current waveform. Superimposing one on the other, the zero voltage turn ON of the
main switch is manifested in figure 6.6 and the reduced voltage turn-off of the main switch is
manifested in figure 6.7. The voltage during turn-OFF is measured to be 80 volts.
Fig 6.6: ZVS turn ON of S1
52
Fig 6.7: Reduced voltage turn OFF of S1
The switching transitions of the auxiliary switch S 2 ZCS turn-ON and the ZVS turn-OFF
are shown in the figures 5.8 and 5.9 respectively.
Fig 6.8: ZCS turn ON of S2
Fig 6.9: ZVS turn OFF of S2
53
The circuit is run under different input conditions with input voltage ranging from 100265V and the circuit is found to give an output voltage of 400V for different values of duty
cycles ranging from 35-81.2% and is shown in table 6.1.
Input voltage Vin (V)
Duty cycle δ (%)
Input voltage Vin (V)
Duty cycle δ (%)
265
35.00
185
56.45
260
36.25
180
57.80
255
37.60
175
59.25
250
38.90
170
60.70
245
40.20
165
62.10
240
41.60
160
63.60
235
42.85
155
65.10
230
44.20
150
66.54
225
45.56
145
68.10
220
46.90
140
69.70
215
48.30
135
71.3
210
49.60
130
73
205
51.00
125
74.7
200
52.30
120
76.6
195
53.70
115
78.65
190
55.00
110
81.2
Table 6.1 Duty cycle corresponding to input voltage for 400V output
Also the RMS current flowing through the auxiliary switch is measured for different
values of input voltage and is tabulated below in table 04
Input voltage (volts)
Duty cycle (%)
Rms current of switch-2(Amps)
265
35.00
0.135
200
52.30
0.243
150
66.54
0.415
110
81.20
0.634
Table 6.2 RMS current of auxiliary switch
54
6.2
LOSS CALCULATION AND COMPARATIVE STUDY
In order to carry out the comparative study, a conventional hard switching converter is designed
for the same specifications. A boost converter with the component values given in table 6.1 is
simulated and the losses of both the converters are compared.
Sl.No.
Component
Symbol
Value
1
Boost inductor
Lin
5.58mH
2
Output capacitor
Co
1.21µF
Table 6.3 components for conventional hard switching converter
The switching losses of any switch is calculated using the following formula
 Ton  Toff
Psw  V0  I 0  Fsw  
2

6.2.1



(6.1)
LOSSES IN THE STUDIED SOFT SWITCHING CONVERTER
The main switch’s turn ON transition takes place under zero voltage. Therefore from the above
formula the switching losses of the main switch during turn ON are zero.
Psw1on  0W
The main switch’s turn OFF transition takes place at reduced voltage and the voltage during turn
OFF is measured and is found to be 80V and the peak current that is carried by this switch is
measured to be 4A.
The switching losses during turn OFF time is calculated as follows
 0  100  10 9 
  1.6W
Psw1off  80  4  105  
2


The auxiliary switch’s turn ON transition takes place under ZCS and its turn OFF transition takes
place under ZVS. Therefore the switching losses of S2 are zero.
55
Psw2on  Psw2off  0W
The total switching losses are
Psw  Psw1  Psw2  1.6  0  1.6W
The conduction losses of the switches are calculated using the formula
Psw _ cond  1.8  I s _ rms  Ron
2
(6.2)
The rms current of the main switch is measured to be 2.3481Amperes and the conduction losses
are calculated as follows
Pcond _ s1  1.8  2.34812  0.85  8.4357W
The rms current of the auxiliary switch is measured to be 0.786 Amperes and the conduction
losses are calculated as follows
Pcond _ s 2  1.8  0.7862  0.85  0.947W
The total conduction losses of the switches are
Pcond _ s  8.4357  0.947  9.3827W
The conduction losses of the diode are product of the forward voltage drop across the diode and
the average current flowing through it. The forward voltage drop is measured to be 0.8027 Volts
and the current flowing through it is the load current which is 0.625 Amperes. So the conduction
losses of the diode are calculated as follows.
PD _ cond  VF  I D _ avg  0.8027  0.625  0.5016  0.502W
The total losses in the converter are
Plosses  Psw  Pcond  PD  1.6  9.3827  0.502  11.4847W
56
The efficiency of the converter is calculated as follows


P0
250


  100  
  100  95.6%
 250  11.484 
 P0  Plosses 
  
6.2.2
LOSSES IN CONVENTIONAL HARD SWITCHING CONVERTER
The only switch in this converter is hard switched and the switching losses are calculated using
the equation 6.1. The voltage across the switch during ON and OFF conditions is 400V and the
peak current is measured to be 4.6A. The switching losses of the switch in conventional boost
converter are
 100  10 9  100  10 9 
  18.4W
Psw  400  4.6  105  
2


The conduction losses in the hard switching converter are calculated using the formula
given in equation 6.2. The rms current flowing through the switch is measured to be 2.2632A.
The conduction losses are calculated as
Psw _ cond  1.8  2.26322  0.85  7.836W
The conduction losses of the diode remain same as that of the soft switching converter.
PD _ cond  VF  I D _ avg  0.8027  0.625  0.5016  0.502W
The total losses in the converter are
Plosses  Psw  Pcond  PD _ cond  18.4  7.386  0.502  26.288W
The efficiency of the converter is calculated as follows


P0
250


  100  
  100  90.4%
 250  26.288 
 P0  Plosses 
  
57
All the results obtained from the comparative study are tabulated and shown in table 6.2
Sl.No.
Specification
Hard switching
Soft switching
converter
converter
1
Psw
18.4
1.6
2
Pcond
7.386
9.3827
3
PD
0.502
0.502
4
%η
90.4
95.6
Table 6.4
6.3
SIMULATION RESULTS OF PV ARRAY
The simulation results of the PV array are shown below. The IV characteristics and PV
characteristics are in figures 6.10 and 6.11 respectively. The open circuit voltage of the simulated
PV array is 180 volts and the short circuit current is 2.25 amperes.
Fig 6.10: IV characteristics of the PV array
58
Fig 6.11: PV characteristics of the PV array
The tracked operating point at which maximum power can be extracted using an MPPT
algorithm is shown below in figure 6.12
Fig 6.12: Output power of the PV array after MPPT
59
6.4
CONCLUSIONS
 The main switch losses of conventional converter are much greater than those of the softswitched converter.
 The auxiliary switch losses are zero in both converters since no auxiliary switch in
conventional converter and in the new converter it is soft switched.
 The diode conduction losses remain same in both the cases
 The conduction losses vary by the RMS current carried by the switches. It is found to be
more in soft switched converter since the auxiliary circuit losses are added up to
conduction losses.
But the switching loss contribution of the hard switching converter dominates in the
calculation of total losses and hence the soft-switched converter is found to be more efficient
than the conventional hard-switched converters.
6.5
FUTURE SCOPE
Work may be done on further reducing the voltage during turn-off transition of the main
switch or making it zero without increasing the circuit complexity.
60
REFERENCES
[1]
Nikhil Jain, Praveen K.Jain and Geza Joos, “A Zero Voltage Transition Boost Converter
Employing a Soft Switching Auxiliay Circuit With Reduced Conduction Losses,” in
IEEE Transactions on power electronics, vol.19, no.1, January 2004.
[2]
In-beom Song, Doo-yong Jung, Young-hyok Ji, Seong-chon Choi, Yong-chae Jung and
Chung-yuen Won, “A Soft Switching Boost Converter using an Auxiliary Resonant
Circuit for a PV System,” International Conference on Power Electronics - ECCE Asia
May 30-June 3, 2011.
[3]
Basu S., Undeland T.M., “Diode recovery characteristics considerations for optimizing
EMI performance of continuous mode PFC converters,” Power Electronics and
Applications, 2005 European Conference, pp.9 pp.,P.9, doi: 10.1109/EPE.2005.219496.
[4]
John Bazinet and John A.O’Connor, “Analysis and Design of a Zero Voltage Transition
Power Factor Correction Circuit,” Unitode Integrated Circuits Merrimack, NH 03054.
[5]
G.Moschopoulos, P.Jain, Y.Liu and Geza Joos, “A Zero Voltage Switched PWM Boost
Converter with an Energy Feedforward Auxiliary Circuit,” IEEE Transactions on Power
Electronics, vol.14, paper 653-662, July 1999.
[6]
Sang-Hoon Park, Gil-Ro Cha, Yong-Chae Jung and Chung-Yuen Won, “Design and
Application fro PV Generation System Using a Soft-Switching Boost Converter with
SARC,” IEEE Transactions on Industrial Electronics, vol.57, no.2, February 2010.
[7]
Saravana Selvan. D, “Modeling and Simulation of Incremental Conductance MPPT
Algorithm
for
Photovoltaic
Applications,”
International
Journal
of
Scientific
Engineering and Technology, Vol.2, no.7, paper no: 681-685, July 2013.
[8]
M G Villalva, J R Gazoli and E R Filho, “ Comprehensive Approach to Modeling and
Simulation of PV Arrays,” IEEE Transactions on Power Electronics, vol.24, no.5, May
2009.
[9]
T Salmi, M Bouzguenda, A Gastli and A Masmoudi, “Matlab/Simulink based Modeling
of solar Photovoltaic cell,” International Journal of Renewable Energy Research, vol.2,
no.2, 2012.
61
[10]
David Sanz Morales, “Maximum power point tracking algorithms for PV applications,”
student paper, Alto university.
[11]
J. Surya Kumari, Dr. Ch. Sai Babu, A. Kamalakar babu, “Design and analysis of P&O
and IP&O MPPT techniques for PV system,” International Journal of Modern
Engineering research, vol.2, issue.4, paper 2174-2180, July-Aug. 2012.
[12]
Salam Z., Ishaque K. and Taheri H., “An improved two-diode photovoltaic (PV) model
for PV system,” Power Electronics, Drives and Energy Systems (PEDES) & 2010 Power
India, 2010 Joint International Conference, vol., no., pp.1,5, 20-23 December 2010.
[13]
S.
Sheik
Mohammed,
MATLAB/Simulink,”
“Modeling and
International
Simulation of Photovoltaic module using
Journal
of
Chemical
and
Environmental
Engineering, Vol.2, No.5, October 2011.
[14]
Jenifer A., N. R. Newlin, G. Rohini, and V. Jamuna, “Development of Matlab Simulink
model for photovoltaic arrays,” International Conference on Computing, Electronics and
Electrical Technologies (ICCEET), 2012, pp. 436-442. IEEE, 2012.
[15]
Ismail Aksoy, Haci Bodur, and A. Faruk Bakan, “A New ZVT-ZCT-PWM DC–DC
Converter,” IEEE transactions on power electronics, Vol. 25, N0. 8, August 2010.
[16]
Aryuanto Soetedjo, Abraham Lomi, Yusuf Ismail Nakhoda, Awan Uji Krismanto,
“Modeling of Maximum Power Point Tracking Controller for Solar Power System,”
TELKOMNIKA (Indonesian Journal of Electrical Engineering), Vol.10, No.3, pp. 419430, July 2012.
62
Was this manual useful for you? yes no
Thank you for your participation!

* Your assessment is very important for improving the work of artificial intelligence, which forms the content of this project

Download PDF

advertisement