# Development of Radar Pulse Compression Techniques Using Computational Intelligence Tools Ajit Kumar Sahoo

Development of Radar Pulse Compression Techniques Using Computational Intelligence Tools Ajit Kumar Sahoo Roll - 507EC005 Department of Electronics and Communication Engineering National Institute of Technology Rourkela Rourkela – 769 008, India Development of Radar Pulse Compression Techniques Using Computational Intelligence Tools Thesis submitted in partial fulfillment of the requirements for the degree of Doctor of Philosophy in Electronics and Communication Engineering by Ajit Kumar Sahoo (Roll - 507EC005) under the guidance of Prof. Ganapati Panda Department of Electronics and Communication Engineering National Institute of Technology Rourkela Rourkela, Orissa, 769008, India Electronics and Communication Engineering National Institute of Technology, Rourkela Rourkela-769 008, Orissa, India. Dr. Ganapati Panda FNAE, FNASc. Professor March 17, 2012 Certificate This is to certify that the thesis entitled “Development of Radar Pulse Compression Techniques Using Computational Intelligence Tools” by Ajit Kumar Sahoo, submitted to the National Institute of Technology, Rourkela for the degree of Doctor of Philosophy, is a record of an original research work carried out by him in the department of Electronics and Communication Engineering under my supervision. I believe that the thesis fulfills part of the requirements for the award of degree of Doctor of Philosophy. Neither this thesis nor any part of it has been submitted for any degree or academic award elsewhere. Ganapati Panda Acknowledgement I take the opportunity to express my reverence to my supervisor Prof. G. Panda for his guidance, inspiration and innovative technical discussions during the course of this work. He encouraged, supported and motivated me throughout the work. I always had the freedom to follow my own ideas for which I am very grateful. I am thankful to Prof. S. Meher, Prof. K. K. Mahapatra, Prof. S. K. Patra of Electronics and Communication Engg. department and Prof. K. B. Mohanty of Electrical Engg. department for extending their valuable suggestions and help whenever I approached. My special thanks to Dr. D. P. Acharya and Dr. Sitanshu Sahu for their constant inspiration and encouragement during my research. My hearty thanks to Jagganath, Trilochan, Upendra, Sudhansu, Prakash, Pyari, Nithin, Vikas and Yogesh for their help, cooperation and encouragement. I acknowledge all staff, research scholars and juniors of ECE department, NIT Rourkela for helping me. I am also grateful to Prof. S. K. Sarangi, Director NIT Rourkela for providing me adequate infrastructure and other facilities to carry out the investigations for my research work. I take this opportunity to express my regards and obligation to my family members whose support and encouragement I can never forget in my life. I am indebted to many people who contributed through their support, knowledge and friendship to this work and made my stay in Rourkela an unforgettable and rewarding experience. Ajit Kumar Sahoo Abstract Pulse compression techniques are used in radar systems to avail the benefits of large range detection capability of long duration pulse and high range resolution capability of short duration pulse. In these techniques a long duration pulse is used which is either phase or frequency modulated before transmission and the received signal is passed through a filter to accumulate the energy into a short pulse. Usually, a matched filter is used for pulse compression to achieve high signal-to-noise ratio (SNR). However, the matched filter output i.e. autocorrelation function (ACF) of a modulated signal is associated with range sidelobes along with the mainlobe. These sidelobes are unwanted outputs from the pulse compression filter and may mask a weaker target which is present nearer to a stronger target. Hence, these sidelobes affect the performance of the radar detection system. In this thesis, few investigations have been made to reduce the range sidelobes using computational intelligence techniques so as to improve the performance of radar detection system. In phase coded signals a long pulse is divided into a number of sub pulses each of which is assigned with a phase value. The phase assignment should be such that the ACF of the phase coded signal attain lower sidelobes. A multiobjective evolutionary approach is proposed to assign the phase values in the biphase code so as to achieve low sidelobes. Basically, for a particular length of code mismatch filter is preferred over matched filter to get better peak to sidelobe ratio (PSR). Recurrent neural network (RNN) and recurrent radial basis function (RRBF) structures are proposed as mismatch filters to achieve better PSR values under various noise conditions, Doppler shift and multiple target environment. Polyphase and linear frequency modulated (LFM) codes yield lower sidelobes compared to biphase codes. Various weighing functions are used to further suppress the sidelobes of polyphase and LFM codes. In this thesis, convolutional windows are used as weighing function to achieve high PSR magnitude at different Doppler shift conditions. In high range resolution radar wide bandwidth signals are used for transmission. The conventional narrowband hardware may not support the instantaneous wide bandwidth. Therefore, the wide bandwidth signal is split into several narrowband signals which are transmitted and recombined coherently at the receiver to get the effect of the wideband signal. However, the ACF of such narrow band pulse train suffers from grating lobes and hence reduce the range resolution capability of the pulse train. In this work, evolutionary computation algorithms are proposed to optimally choose the parameters of stepped frequency LFM pulse train to achieve reduced grating lobes, low peak sidelobe and narrow mainlobe width. Keywords: Pulse Compression, Matched filter, Sidelobes, ACF, Multiobjective, RNN, RRBF, LFM, Polyphase Codes, Convolutional Windows, Grating Lobes. vi Contents Certificate iii Acknowledgement iv Abstract v List of Figures x List of Tables xiv List of Acronyms xv 1 Introduction 1 1.1 Pulse compression . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 1.2 Matched filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 1.2.1 Matched filter for a narrow bandpass signal . . . . . . . . . . 7 1.2.2 Matched filter response to Doppler shifted signal . . . . . . . . 8 1.2.3 Properties of ambiguity function . . . . . . . . . . . . . . . . . 9 1.2.4 Cuts through ambiguity function . . . . . . . . . . . . . . . . 10 1.3 Radar signals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 1.3.1 Frequency modulated signal . . . . . . . . . . . . . . . . . . . 11 1.3.2 Phase coded signal . . . . . . . . . . . . . . . . . . . . . . . . 12 1.4 Background and scope of the thesis . . . . . . . . . . . . . . . . . . . 15 1.5 Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 1.6 Objective of the thesis . . . . . . . . . . . . . . . . . . . . . . . . . . 17 1.6.1 1.7 Structure and chapter wise contribution of the thesis . . . . . 18 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 vii 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm 22 2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 2.2 Merit measures and problem formulation . . . . . . . . . . . . . . . . 24 2.3 Techniques used . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 2.4 2.3.1 Genetic algorithm . . . . . . . . . . . . . . . . . . . . . . . . . 25 2.3.2 Multi objective GA . . . . . . . . . . . . . . . . . . . . . . . . 29 Generation of pulse compression codes . . . . . . . . . . . . . . . . . 35 2.4.1 Using genetic algorithm . . . . . . . . . . . . . . . . . . . . . 35 2.4.2 Using NSGA-II . . . . . . . . . . . . . . . . . . . . . . . . . . 38 2.5 Simulation results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39 2.6 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction 43 3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43 3.2 Problem formulation . . . . . . . . . . . . . . . . . . . . . . . . . . . 45 3.3 Techniques used . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 3.4 3.5 3.3.1 Adaptive linear combiner . . . . . . . . . . . . . . . . . . . . . 47 3.3.2 Artificial neural network . . . . . . . . . . . . . . . . . . . . . 51 Simulation results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62 3.4.1 Sidelobe suppression using adaptive linear combiner . . . . . . 64 3.4.2 Sidelobe suppression using MLP, RNN, RBF, RRBF . . . . . 64 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows 76 4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76 4.2 LFM and polyphase codes . . . . . . . . . . . . . . . . . . . . . . . . 79 4.3 4.2.1 LFM signal . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79 4.2.2 Polyphase codes . . . . . . . . . . . . . . . . . . . . . . . . . . 81 Problem formulation . . . . . . . . . . . . . . . . . . . . . . . . . . . 88 viii 4.3.1 For LFM signal . . . . . . . . . . . . . . . . . . . . . . . . . . 88 4.3.2 For polyphase codes . . . . . . . . . . . . . . . . . . . . . . . 89 4.4 Windows used for sidelobe suppression . . . . . . . . . . . . . . . . . 90 4.5 Simulation results 4.6 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92 4.5.1 Analysis for LFM signals . . . . . . . . . . . . . . . . . . . . . 93 4.5.2 Analysis for polyphase codes . . . . . . . . . . . . . . . . . . . 98 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques 104 5.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 104 5.2 LFM pulse train . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106 5.3 Problem formulation . . . . . . . . . . . . . . . . . . . . . . . . . . . 110 5.4 5.5 5.3.1 Problem formulation -1 . . . . . . . . . . . . . . . . . . . . . . 110 5.3.2 Problem formulation -2 . . . . . . . . . . . . . . . . . . . . . . 110 5.3.3 Problem formulation -3 . . . . . . . . . . . . . . . . . . . . . . 111 Techniques used . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111 5.4.1 Particle swarm optimization . . . . . . . . . . . . . . . . . . . 112 5.4.2 NSGA-II . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 114 Determination of parameters of LFM pulse train . . . . . . . . . . . . 115 5.5.1 Using PSO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115 5.5.2 Using NSGA-II . . . . . . . . . . . . . . . . . . . . . . . . . . 116 5.6 Simulation results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 117 5.7 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 126 6 Conclusion and Future Work 127 6.1 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127 6.2 Future work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 129 Bibliography 130 Dissemination of Work 141 ix List of Figures 1.1 Pulsed radar waveform . . . . . . . . . . . . . . . . . . . . . . . . . . 2 1.2 Transmitter and receiver ultimate signals . . . . . . . . . . . . . . . . 3 1.3 Block diagram of a pulse compression radar system . . . . . . . . . . 4 1.4 Block diagram of matched filter . . . . . . . . . . . . . . . . . . . . . 5 1.5 The instantaneous frequency of the LFM waveform over time . . . . . 11 1.6 Phase modulated waveform . . . . . . . . . . . . . . . . . . . . . . . 12 1.7 Matched filter output of different signals . . . . . . . . . . . . . . . . 13 2.1 Crossover . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 2.2 Mutation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 2.3 Flow chart for GA . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 2.4 NSGA-II procedure . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 2.5 Flow chart for NSGA-II . . . . . . . . . . . . . . . . . . . . . . . . . 37 2.6 Crossover using binary bits 1 and -1 . . . . . . . . . . . . . . . . . . . 38 2.7 Mutation using binary bits 1 and -1 . . . . . . . . . . . . . . . . . . . 38 3.1 Adaptive linear combiner . . . . . . . . . . . . . . . . . . . . . . . . . 47 3.2 Single neuron structure . . . . . . . . . . . . . . . . . . . . . . . . . . 52 3.3 Mlutilayer perceptron network . . . . . . . . . . . . . . . . . . . . . . 55 3.4 Block diagram of recurrent neural nerwork . . . . . . . . . . . . . . . 58 3.5 Architecture of radial basis function network . . . . . . . . . . . . . . 59 3.6 Architecture of recurrent radial basis function network . . . . . . . . 61 3.7 26 different possible input sequences for 13-bit Barker codes . . . . . 63 x 3.8 Filter response in dB for 13-bit Barker code obtained using (a)ACF (b)LMS (c)RLS (d)Modified RLS algorithms 3.9 . . . . . . . . . . . . . 65 Filter response in dB for 35-bit Barker code obtained using (a)ACF (b)LMS (c)RLS (d)Modified RLS . . . . . . . . . . . . . . . . . . . . 66 3.10 Convergence graphs of different structures for (a)13-bit (b)35-bits Barker codes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68 3.11 Compressed waveforms for 13 bit Barker code using (a)MLP (b)RNN (c)RBF (d)RRBF structures . . . . . . . . . . . . . . . . . . . . . . . 69 3.12 Input waveform on addition of two 5-DA 13-bit Barker sequence having same magnitude (a)Left shift (b)Right shift (c)Added waveform (d)Waveform after flip about the vertical axis . . . . . . . . 72 3.13 Compressed waveforms for 13-bit Barker code having same IMR and 5 DA for (a)MLP (b)RNN (c) RBF (d)RRBF structures . . . . . . . 73 4.1 Real and imaginary part of the chirp signal for T B = 50 . . . . . . . 80 4.2 Amplitude spectrum of chirp signal for T B = 50 4.3 Compressed envelope . . . . . . . . . . . . . . . . . . . . . . . . . . . 81 4.4 Matched filter output and phase values of 100 element Frank code . . 83 4.5 Matched filter output and phase values of 100 element P1 code . . . . 84 4.6 Matched filter output and phase values of 100 element P2 code . . . . 85 4.7 Matched filter output and phase values of 100 element P3 code . . . . 87 4.8 Matched filter output and phase values of 100 element P4 code . . . . 87 4.9 Frequency response curve . . . . . . . . . . . . . . . . . . . . . . . . . 92 . . . . . . . . . . . 81 4.10 Matched filter output with Hamming weighing at the receiver . . . . 93 4.11 Effect on sidelobes due to Doppler shift . . . . . . . . . . . . . . . . 94 4.12 Compressed waveforms for T B = 50 for amplitude tapering (α = 0.1) 97 4.13 Compressed waveforms for T B = 50 for cubic phase distortion (∆B = 0.75B and ∆T = 1 ) B . . . . . . . . . . . . . . . . . . . . . . . . . . . 97 4.14 Matched filter output for 100 element Frank code using Hamming window . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99 xi 4.15 Matched filter output of P3 code under different Doppler shift . . . . 100 4.16 Matched filter output of P4 code under different Doppler shift . . . . 100 4.17 Effect on sidelobes due to Doppler shift . . . . . . . . . . . . . . . . 101 5.1 Stepped frequency LFM pulse train . . . . . . . . . . . . . . . . . . . 105 5.2 Stepped frequency LFM pulse for Tp ∆f = 3, Tp B = 4.5 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . 109 5.3 Stepped frequency LFM pulse for Tp ∆f = 3, Tp B = 0 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . 109 5.4 Stepped frequency LFM pulse for Tp ∆f = 2.5, Tp B = 12.5 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . 118 5.5 Stepped frequency LFM pulse for Tp ∆f = 4, Tp B = 16 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . 118 5.6 Pareto front obtained using NSGA-II for Tp ∆f ∈ [2, 10], c ∈ [2, 10] and N = 8 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 120 5.7 Stepped frequency LFM pulse for Tp ∆f = 2, c = 5, Tp B = 12 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 121 5.8 Stepped frequency LFM pulse for Tp ∆f = 2, c = 5.1412, Tp B = 12.2824 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 121 5.9 Stepped frequency LFM pulse for Tp ∆f = 2.8721, c = 5.0978, Tp B = 17.5135 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122 5.10 Pareto front obtained using NSGA-II for Tp ∆f ∈ [2, 10] c ∈ [2, 10], ǫ = 0.01 and N = 8 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122 5.11 Pareto front obtained using NSGA-II for Tp ∆f ∈ [5, 30], c ∈ [2, 10], ǫ = 0.01 and N = 8 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123 5.12 Pareto front obtained using NSGA-II for Tp ∆f ∈ [2, 10], c ∈ [2, 5], ǫ = 0.01 and N = 8 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123 xii 5.13 Stepped frequency LFM pulse for Tp ∆f = 9.0188, c = 3.5502, Tp B = 41.0373 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 5.14 Stepped frequency LFM pulse for Tp ∆f = 4.9667, c = 4.0720, Tp B = 25.1911 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124 5.15 Stepped frequency LFM pulse for Tp ∆f = 3.6048, c = 4.6129, Tp B = 20.2334 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 125 5.16 Stepped frequency LFM pulse for Tp ∆f = 3, c = 5, Tp B = 18 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 125 xiii List of Tables 1.1 Barker codes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 2.1 Sequences obtained using GA . . . . . . . . . . . . . . . . . . . . . . 40 2.2 Sequences obtained using NSGA-II . . . . . . . . . . . . . . . . . . . 41 3.1 PSRs obtained using various learning algorithms. . . . . . . . . . . . 64 3.2 PSRs obtained by various structures . . . . . . . . . . . . . . . . . . 68 3.3 Comparison of PSRs in dB at different SNRs for 13-bit Barker code . 70 3.4 Comparison of PSRs in dB at different SNRs for 35-bit Barker code . 70 3.5 Comparison of range resolution ability for 13-bit Barker code of two targets having same IMR and DA. . . . . . . . . . . . . . . . . . . . . 71 3.6 Comparison of range resolution ability for 35-bit Barker code of two targets having same IMR and DA. . . . . . . . . . . . . . . . . . . . . 71 3.7 Comparison of range resolution ability for 13-bit Barker code of two targets having different IMR and DA. . . . . . . . . . . . . . . . . . . 74 3.8 Comparison of 35-bit Barker code for range resolution ability of two targets having same IMR and DA . . . . . . . . . . . . . . . . . . . . 74 3.9 Doppler shift performance . . . . . . . . . . . . . . . . . . . . . . . . 74 4.1 Comparison of PSR for different Doppler shift for T B = 50 4.2 PSR using amplitude tapering . . . . . . . . . . . . . . . . . . . . . . 96 4.3 PSR using cubic phase distortion . . . . . . . . . . . . . . . . . . . . 98 4.4 Comparison of PSR for different Doppler shift . . . . . . . . . . . . . 102 5.1 Values of Tp ∆f , Tp B obtained for N = 8 and f1 = 0 . . . . . . . . . . 119 xiv . . . . . 95 List of Acronyms Radar RAdio Detection And Ranging CW Continuous Waveform TB Time-Bandwidth PCR Pulse Compression Ratio TR Transreceiver AWGN Additive White Gaussian Noise SNR Signal-to-Noise Ratio PSD Power Spectral Density ACF Autocorrelation Function AF Ambiguity Function LFM Linear Frequency Modulated PSL Peak Sidelobe Level PSR Peak to Sidelobe Ratio ISR Integrated Sidelobe Ratio MPS Minimum Peak Sidelobe CI Computational Intelligence xv EA Evolutionary Algorithm MF Merit Factor GA Genetic Algorithm NSGA Nondominated Sorting Genetic Algorithm NSGA-II Nondominated Sorting Genetic Algorithm -II VEGA Vector Evaluated Genetic Algorithm MOGA Multiobjective Genetic Algorithm VLSI Very Large Scale Integrated ISL Integrated Sidelobe Level LS Least Square LP Linear Programming ALC Adaptive Linear Combiner FIR Finite Impulse Response MSE Mean Square Error LMS Least Mean Square RLS Recursive Least Square MLP Multilayer Perceptron BP Back Propagation RNN Recurrent Neural Network RBF Radial Basis Function xvi RRBF Recurrent Radial Basis Function ANN Artificial Neural Network NN Neural Network DA Delay Apart IMR Input Magnitude Ratio NLFM Nonlinear Frequency Modulated DFT Discrete Fourier Transform PSO Particle Swarm Optimization pdf probability distribution function xvii Chapter 1 Introduction Radar an acronym for RAdio Detection And Ranging. It is an electromagnetic system used to detect and locate the object by transmitting the electromagnetic signals and receiving the echoes from the objects within its coverage [1]. The echoes are used to extract the information about the target such as range, angular position, velocity and other identifying characteristics. A continuous waveform (CW) is the simplest radar waveform which is transmitted continuously while receiving target echoes on a separate antenna. The advantage of CW is the unambiguous Doppler measurement. However, due to continuous nature of the waveform the target range measurement is entirely ambiguous. Most of the modern radar systems employ a pulsed waveform which provides range information accurately. The primary advantage of pulsed radar is that the transmitter and receiver can share the same antenna due to pulsating nature of the waveform. A pulsed waveform is shown in Figure 1.1, where Tp is the pulse duration and Tr is the pulse repetition time. The unambiguous range Ru that can be measured by this waveform as described in [2] is Ru = cTr 2 (1.1) where c is the speed of light. Two important factors to be considered for radar waveform design are range resolution and maximum range detection. Range resolution is the ability of the 1 Introduction Chapter 1 Figure 1.1: Pulsed radar waveform radar to separate closely spaced targets and it is related to the pulse width of the waveform. The narrower the pulse width the better is the range resolution. But, if the pulse width is decreased, the amount of energy in the pulse is decreased and hence maximum range detection gets reduced. To overcome this problem pulse compression techniques are used in the radar systems. 1.1 Pulse compression The maximum detection range depends upon the strength of the received echo. To get high strength reflected echo the transmitted pulse should have more energy for long distance transmission since it gets attenuated during the course of transmission. The energy content in the pulse is proportional to the duration as well as the peak power of the pulse. The product of peak power and duration of the pulse gives an estimate of the energy of the signal. A low peak power pulse with long duration provides the same energy as achieved in case of high peak power and short duration pulse. Shorter duration pulses achieve better range resolution. The range resolution rres is expressed [2] as rres = c 2B (1.2) where B is the bandwidth of the pulse. For unmodulated pulse the time duration is inversely proportional to the bandwidth. If the bandwidth is high, then the duration of the pulse is short and hence this offers a superior range resolution. Practically, the pulse duration cannot be reduced indefinitely. According to Fourier theory a signal with bandwidth B cannot have duration shorter than 1/B i.e. its time-bandwidth (T B) product cannot be less than 2 Introduction Chapter 1 unity. A very short pulse requires high peak power to get adequate energy for large distance transmission. However, to handle high peak power the radar equipment become heavier, bigger and hence cost of this system increases. Therefore peak power of the pulse is always limited by the transmitter. A pulse having low peak power and longer duration is required at the transmitter for long range detection. At the output of the receiver, the pulse should have short width and high peak power to get better range resolution. Figure 1.2 illustrates two pulses having same energy with different pulse width and peak power. To get the advantages of larger range detection ability of long pulse and better range resolution ability of short pulse, pulse compression [3] techniques are used in radar systems. The range resolution depends on the bandwidth of a pulse but not necessarily on the Figure 1.2: Transmitter and receiver ultimate signals duration of the pulse [4]. Some modulation techniques such as frequency and phase modulation are used to increase the bandwidth of a long duration pulse to get high range resolution having limited peak power. In pulse compression technique a pulse having long duration and low peak power is modulated either in frequency or phase before transmission and the received signal is passed through a filter to accumulate the energy in a short pulse. The pulse compression ratio (P CR) is defined as P CR = width of the pulse before compression width of the pulse after compression 3 (1.3) Introduction Chapter 1 The block diagram of a pulse compression radar system is shown in Figure 1.3. The transmitted pulse is either frequency or phase modulated to increase the bandwidth. Transreceiver (TR) is a switching unit helps to use the same antenna as transmitter and receiver. The pulse compression filter is usually a matched filter whose frequency response matches with the spectrum of the transmitted waveform. The filter performs a correlation between the transmitted and the received pulses. The received pulses with similar characteristics to the transmitted pulses are picked up by the matched filter whereas other received signals are comparatively ignored by the receiver. Figure 1.3: Block diagram of a pulse compression radar system 1.2 Matched filter In radar applications the reflected signal is used to determine the existence of the target. The reflected signal is corrupted by additive white Gaussian noise (AWGN). The probability of detection is related to signal-to-noise ratio (SNR) rather than exact shape of the signal received. Hence it is required to maximize the SNR rather 4 Introduction Chapter 1 than preserving the shape of the signal. A filter which maximizes the output SNR is called matched filter [5]. A matched filter is a linear filter whose impulse response is determined for a signal in such way that the output of the filter yields maximum SNR when the signal along with AWGN is passed through the filter. An input signal s(t) along with AWGN is given as input to the matched filter as shown in Figure 1.4. Let N0 /2 be the two sided power spectral density (PSD) of AWGN. It is required to find out the impulse response h(t) or the frequency response H(f ) (Fourier transform of h(t)) that yields maximum SNR at a predetermined delay t0 . In other words, h(t) or H(f ) is determined to maximize the output SNR which is given by Figure 1.4: Block diagram of matched filter SP NP = out |s0 (t0 )|2 n20 (t) (1.4) where SP is the signal power, N P is the output noise power, s0 (t0 ) is the value of the output signal s0 (t) at t = t0 and n20 (t) is the mean square value of the noise. If S(f ) is the Fourier transform of s(t), then s0 (t) is obtained as Z ∞ s0 (t) = H(f )S(f )ej2πf t df (1.5) −∞ The value of s0 (t) at t = t0 is s0 (t0 ) = Z ∞ H(f )S(f )ej2πf t0 df (1.6) −∞ The mean square value n20 (t) of the noise is evaluated as Z N0 ∞ 2 |H(f )|2 df n0 (t) = 2 −∞ 5 (1.7) Introduction Chapter 1 Substituting (1.6) and (1.7) in (1.4) yields 2 R ∞ j2πf t0 df −∞ H(f )S(f )e SP R = N0 ∞ N P out |H(f )|2 df 2 −∞ Using Schwarz inequality the numerator of (1.8) can be written as Z ∞ 2 Z ∞ Z ∞ 2 j2πf t 0 |H(f )| df |S(f )ej2πf t0 |2 df df ≤ H(f )S(f )e −∞ −∞ (1.8) (1.9) −∞ In (1.9) the equality holds good if H(f ) = K1 [S(f )ej2πf t0 ]∗ = K1 S ∗ (f )e−j2πf t0 (1.10) where K1 is an arbitrary constant and ∗ stands for complex conjugate. Using the equality sign of (1.9), which corresponds to maximum output SNR, in (1.8) R∞ |S(f )|2 df 2E SP (1.11) = −∞ N0 = N P out N 0 2 where E is the energy of the finite time signal and defined as Z ∞ Z ∞ 2 E= |s(t)| dt = |S(f )|2 df −∞ (1.12) −∞ From (1.11) it is obvious that the maximum SNR is a function of the energy of the signal but not the shape. Taking inverse Fourier transform of (1.10) the impulse response of matched filter is obtained as h(t) = K1 s∗ (t0 − t) (1.13) From (1.13) it is clear that the impulse response of matched filter is a delayed mirror image of the conjugate of the input signal. From (1.6) and (1.10) the output at t = t0 is given as s0 (t0 ) = K1 = R∞ −∞ R∞ K1 −∞ S(f )S ∗ (f )e−j2πf t0 ej2πf t0 df |S(f )|2 df = K1 E 6 (1.14) Introduction Chapter 1 Equation (1.14) states that regardless of the type of waveform, at the predefined delay t = t0 the output is the energy of the waveform for K1 = 1. The output of the matched filter is evaluated as s0 (t) = s(t) ⊗ h(t) R∞ = −∞ s(τ )h(t − τ )dτ R∞ = −∞ s(τ )K1 s∗ (τ − t + t0 )dτ R∞ = K1 =1,t0 =0 −∞ s(τ )s∗ (τ − t)dτ (1.15) where ⊗ denotes the linear convolution operation. The right hand side of (1.15) is known as autocorrelation function (ACF) of the input signal s(t). 1.2.1 Matched filter for a narrow bandpass signal Most of the radar signals are narrow bandpass signals. A narrowband signal s(t) [5] can be represented as 1 1 s(t) = u(t)ej2πf0 t + u∗ (t)e−j2πf0 t 2 2 (1.16) where u(t) is the complex envelope of s(t) and f0 is the carrier frequency. From (1.15) and (1.16) R∞ s0 (t) = K41 −∞ [u(τ )ej2πf0 τ + u∗ (τ )e−j2πf0 τ ] ∗ u (τ − t + t0 )e−j2πf0 (τ −t+t0 ) + u(τ − t + t0 )ej2πf0 (τ −t+t0 ) dτ (1.17) Evaluating the products, (1.17) is represented as R∞ s0 (t) = K41 ej2πf0 (t−t0 ) −∞ u(τ )u∗ (τ − t + t0 )dτ R∞ + K41 e−j2πf0 (t−t0 ) −∞ u∗ (τ )u(τ − t + t0 )dτ R∞ + K41 ej2πf0 (t−t0 ) −∞ u∗ (τ )u∗ (τ − t + t0 )e−j4πf0 τ dτ R∞ + K41 e−j2πf0 (t−t0 ) −∞ u(τ )u(τ − t + t0 )ej4πf0 τ dτ (1.18) of first and third terms respectively. So it can be written as o n R K1 ∗ j2πf0 (t−t0 ) ∞ s0 (t) = 2 Re e u(τ )u (τ − t + t0 )dτ −∞ o n R ∞ + K21 Re ej2πf0 (t−t0 ) −∞ u∗ (τ )u∗ (τ − t + t0 )e−j4πf0 τ dτ (1.19) In (1.18) the second and fourth terms of right hand side are the complex conjugate 7 Introduction Chapter 1 The second term on the right hand side of (1.19) is the Fourier transform of u∗ (τ )u∗ (τ − t + t0 ) evaluated at f = 2f0 , which is at much higher frequency than the spectrum of the complex envelope u(t). So neglecting the second term the expression in (1.19) becomes o R∞ ej2πf0 (t−t0 ) −∞ u(τ )u∗ (τ − t + t0 )dτ o i nh R∞ = Re K21 e−j2πf0 t0 −∞ u(τ )u∗ (τ − t + t0 )dτ ej2πf0 t s0 (t) = K1 Re 2 n (1.20) The expression inside the square bracket of (1.20) is defined as new complex envelope u0 (t) which is expressed as u0 (t) = K2 Z ∞ −∞ where K2 = u(τ )u∗ (τ − t + t0 )dτ (1.21) K1 −j2πf0 t0 e . 2 The output of the matched filter is s0 (t) = Re u0 (t)ej2πf0 t (1.22) From (1.21) and (1.22) it is observed that the matched filter output of narrow bandpass signal has a complex envelope u0 (t) which is obtained by passing the complex envelope u(t) through its own matched filter. 1.2.2 Matched filter response to Doppler shifted signal Most of the targets in the environment are non stationary. So the frequency of the reflected signal from a target experiences Doppler shift. The Doppler shifted complex envelope is represented as uD (t) = u(t)ej2πfd t (1.23) where fd is the Doppler shift. Substituting uD (t) for first u(t) in (1.21) and choosing t0 = 0 and K2 = 1 Z ∞ u(τ )ej2πfd τ u∗ (τ − t)dτ u0 (t, fd ) = −∞ 8 (1.24) Introduction Chapter 1 Reversing the operations of τ and t a modified expression obtained as Z ∞ u(t)u∗ (t − τ )ej2πfd t dt χ(τ, fd ) = (1.25) −∞ Equation (1.25) is one of the versions of the ambiguity function (AF). The AF describes the output of the matched filter if the input signal is delayed by τ and Doppler shifted by fd relative to the values for which the matched filter is designed. The AF was introduced by Woodward [6] which is an important tool for radar signal analysis. But the AF expressions given in [2,4–6] differ in the sign of τ and fd . τ gives the information whether the target is farther from or nearer to the reference and fd gives the information whether the target is moving towards or moving away from the radar. A standard form of AF which is used in most of the radar systems is Z |χ(τ, fd )| = ∞ ∗ j2πfd t u(t)u (t + τ )e −∞ dt (1.26) where a positive τ corresponds to the target being farther from the radar and a positive fd corresponds to the target moving towards the radar. 1.2.3 Properties of ambiguity function Some of the important properties of AF [5] are explained below where energy of u(t) normalized to unity . 1. It has maximum value at origin (0,0) i.e. |χ(τ, fd )| ≤ |χ(0, 0)| = 1 (1.27) 2. The total volume under AF is unity and independent of signal waveform. Z ∞Z ∞ |χ(τ, fd )|2 dτ dfd = 1 (1.28) −∞ −∞ 3. AF is symmetrical with respect to origin |χ(τ, fd )| = |χ(−τ, −fd )| 9 (1.29) Introduction Chapter 1 4. If a complex envelope u(t) has AF |χ(τ, fd )| then addition of linear frequency modulation, which is equivalent to a quadratic phase modulation, makes the AF as 2 u(t)ejπkt ⇔ |χ(τ, fd − kτ )| 1.2.4 (1.30) Cuts through ambiguity function 1. Cuts along the delay axis The cut along the delay axis is obtained by setting fd = 0 in (1.26) i.e. Z ∞ ∗ |χ(τ, 0)| = u(t)u (t + τ )dt = |R(τ )| (1.31) −∞ where R(τ ) is the autocorrelation function of u(t). 2. Cuts along the Doppler axis Setting τ = 0 in (1.26) yields Z |χ(0, fd )| = ∞ −∞ |u(t)|2 ej2πfd t dt (1.32) Equation (1.32) states that the cut along the Doppler axis yields the Fourier transform of the magnitude of the square of the complex envelope u(t). 1.3 Radar signals In radar system a particular waveform is first determined for a given application and it is used to design the optimum detection system. The waveform should provide least amount of uncertainty or ambiguity when the reflected signal is used to extract the information about the range, the velocity and the number of true targets present in the environment. The different types of signals those are mostly used in radar systems are discussed in sequel. 10 Introduction Chapter 1 1.3.1 Frequency modulated signal Linear frequency modulated (LFM) signals are used in most of the radar systems to achieve wide operating bandwidth. In this case the frequency increases (up chirp) or decreases (down chirp) linearly across the pulse. The instantaneous phase of the chirp signal is expressed as 1 φ(t) = 2π(f0 t + kt2 ) 2 (1.33) where f0 is the carrier frequency and k is the frequency sweep rate related to pulse duration Tp and bandwidth B as k= B Tp (1.34) The instantaneous frequency is given by f (t) = d 1 (f0 t + kt2 ) = f0 + kt dt 2 (1.35) Equation (1.35) states that the instantaneous frequency is a linear function of Figure 1.5: The instantaneous frequency of the LFM waveform over time time, and hence is called as linear frequency modulation. Figure 1.5 illustrates the instantaneous frequency of LFM waveform that sweeps from f0 to f1 . The matched filter responses of an unmodulated pulse (duration 10µs) and an LFM pulse (duration 10µs and bandwidth 3MHz) are depicted in Figures 1.7(a) and 1.7(b) respectively. From these figures it is evident that the matched filter output 11 Introduction Chapter 1 of LFM signal has narrow mainlobe width and hence has better range resolution capability. However it is associated with sidelobes which are unwanted in output from the filter. The compressed pulse width of LFM signal is 1/B and the PCR is obtained as P CR = BTp 1.3.2 (1.36) Phase coded signal The increase in bandwidth can also be achieved by phase modulation. In this case a long pulse width Tp is divided into a number of sub pulses each of width tb as shown in Figure 1.6. Each sub pulse is assigned with a phase value φi , where i = 1, 2, ...N . The received echo is passed through a filter to get a single output peak. The most popular phase coding is biphase or binary coding. A biphase code consists of a sequence of +1 and -1. The phases of the transmitted waveform is 00 for +1 and 1800 for -1. The coded signal is discontinuous at the point of phase reversal. The matched filter response of a randomly assigned 10-bit biphase code ([1 -1 1 -1 1 -1 -1 1 1 -1]) is shown in Figure 1.7(c). It is evident from the figure that phase coded signals are also associated with the sidelobes. The PCR of phase coded pulse is obtained as Figure 1.6: Phase modulated waveform P CR = Tp tb (1.37) Figure 1.7 shows that the modulated signals provide better range resolution as compared to unmodulated signals but the matched filter output of the modulated signals suffer from the sidelobes. These sidelobes may hide the small targets or may cause false target detection. The sidelobe having largest amplitude is called peak 12 Introduction Chapter 1 1 0.8 |ACF| 0.6 0.4 0.2 0 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 0.4 0.6 0.8 1 Delay time −5 x 10 (a) Matched filter response for unmodulated pulse 1 Mainlobe |ACF| 0.8 0.6 Sidelobes 0.4 0.2 0 −1 −0.8 −0.6 −0.4 −0.2 0 Delay time 0.2 0.4 0.6 0.8 1 −5 x 10 (b) Matched filter response for frequency modulated pulse (T B = 30) 1 Mainlobe 0.8 |ACF| 0.6 0.4 Sidelobes 0.2 0 0 2 4 6 8 10 12 14 16 18 Delay time (c) Matched filter response for phase modulated pulse Figure 1.7: Matched filter output of different signals 13 20 Introduction Chapter 1 Table 1.1: Barker codes Code length Coded signal PSR in dB 2 1-1,-11 -6 3 11-1 -9.5 4 11-11,111-1 -12 5 111-11 -14 7 111-1-11-1 -16.9 11 111-1-1-11-1-11-1 -20.8 13 11111-1-111-11-11 -22.3 sidelobe. The lower the peak sidelobe level (PSL) the better is the code. To quantify the the waveform characteristics peak to sidelobe ratio (PSR) and integrated sidelobe ratio (ISR) are used as measures of performance in radar systems. These are defined as peak sidelobe power mainlobe power total power in sidelobes ISR = 10 log10 mainlobe power P SR = 10 log10 (1.38) (1.39) In biphase codes the selection of random phase 0 or π is a difficult task. The phases are selected so that the matched filter output of the code has lower sidelobes. Barker codes are the special type of binary codes having sidelobes of unity magnitude. Exhaustive computer based search reveals that the Barker codes are available for the length of 2, 3, 4, 5, 7, 11 and 13 only. The Barker codes along with their PSR values are listed in Table 1.1. The Barker code have maximum compression ratio is 13 and highest PSR magnitude is 22.3 dB. A longer code is required for many radar application to achieve high pulse compression ratio. One way to obtain a longer code having lower sidelobe level is by nesting two Barker codes using Kronecker product. This type of code is called compound Barker code. If one Barker code has length l1 and that of other is l2 , then the compound Barker code is of length l1 l2 and the compression ratio is l1 l2 . For example a 35-bit compound Barker code is generated by taking the Kronecker tensor product of 5-bit and 7-bit Barker codes and the resultant code is [1 1 1 -1 -1 14 Introduction Chapter 1 1 -1 1 1 1 -1 -1 1 -1 1 1 1 -1 -1 1 -1 -1 -1 -1 1 1 -1 1 1 1 1 -1 -1 1 -1]. Although a larger compression ratio is achieved by compound Barker code, the peak sidelobes are not proportionally decreased. The codes those yield minimum peak sidelobe level but do not meet the Barker condition (i.e. maximum PSL is unity) are called minimum peak sidelobe (MPS) level codes. If the pulse is allowed to take more than two values, it is known as a polyphase code. The phases of the polyphase code are chosen in such way that its ACF should have lower sidelobes. However the polyphase codes are sensitive to Doppler shift. To overcome this problem the polyphase codes are derived from the phase history of the frequency modulated pulses. The details of polyphase codes and their pulse compression methods are discussed in Chapter 4. 1.4 Background and scope of the thesis A lot of research work has been carried out over past few decades to achieve low sidelobes and high range resolution in the radar pulse detection system [5]. Biphase coding techniques are preferred in pulse compression techniques owing to their easy implementation. The phases of biphase codes are assigned randomly to different bits of a certain length of code according to different measure of performance. So efficient techniques are required to assign the phases of biphase codes such that it would provide better performance indices. Practically the mismatch filters are used to provide better PSR, with some SNR loss, than matched filter. Various mismatch filters such as adaptive linear combiner (ALC) and neural networks are used to suppress the sidelobes [38, 39, 41]. However, the convergence of the neural network is slow during the training period. Hence new efficient structures and the corresponding learning algorithms having faster convergence are required for pulse compression. Apart from biphase codes, polyphase codes [80, 81] and frequency modulated codes are also used in radar systems. In the literature different type of windows are used as weighing function to suppress the sidelobes [89, 90] of polyphase codes 15 Introduction Chapter 1 and LFM waveforms [71]. Under the Doppler shift conditions the PSR magnitude provided by the windows are low. Hence efficient amplitude weighing techniques are needed to achieve lower sidelobes in Doppler shift conditions. In phased array radar, wide bandwidth waveforms are used to acquire high range resolution. Generation of such types of waveforms increases the overall cost and complexity of the system. The conventional hardware designed for narrowband signals in radar systems may not sustain instantaneous wide bandwidth. To overcome such limitation the wide bandwidth signal is split into a set of narrowband signals which are transmitted and received separately. The effect of wideband signal is obtained by coherently combining the narrowband signals. Such type of narrowband signals taken together is called ‘synthetic wideband waveform’ or ‘stepped frequency waveform’ or ‘frequency jumped train’. However, the matched filter output i.e. ACF of such signals suffers from grating lobes due to constant frequency step. Therefore there is a need to design a signal having wide bandwidth but can be processed by the hardware for narrow band signals and its ACF has suppressed grating lobe, low peak sidelobes and narrow mainlobe width. 1.5 Motivation Substantial effort has been made to suppress the sidelobes of the different waveforms using computational intelligence (CI) tools such as evolutionary computing techniques and neural networks. • Several existing evolutionary computing techniques have been employed to assign the phase to different bits of biphase codes using weighted sum of PSL and merit factor (MF) as cost function [14, 15]. The problem associated with these methods is to choose the appropriate values of the weights. • The matched filter does not provide adequate PSR for many radar applications. Hence to obtain improved PSR, the mismatch filters have been introduced whose weights are adapted using known input output data. Various mismatch 16 Introduction Chapter 1 filters using multilayer perceptron (MLP) and radial basis function (RBF) networks have been reported in the literature. However these filters provide poor convergence performance and hence the magnitude of PSR is less during detection. Thus there is need to design improved mismatch filters. • For polyphase and LFM waveforms the amplitude weighing techniques are used at the receiver to suppress sidelobes. The targets in the environment are not always stationary. If the target is in motion, the reflected waveform is Doppler shifted version of the transmitted waveform. When this Doppler shifted waveforms are passed through the weighted receiver matched filter the PSR degrades. Under such situations it is required to improve the PSR. • The matched filter output i.e. ACF of wide bandwidth stepped frequency LFM pulse train suffers from grating lobes due to constant frequency step. Several methods have been implemented to suppress the grating lobes in [113, 114]. These methods generally ignore the PSL and mainlobe width which are also important measures of the performance for target detection. Therefore, algorithms need to be developed to choose the parameters of stepped frequency waveform such that the output of the matched filter provides high range resolution, lower grating lobes and reduced sidelobes. Based on the aforementioned motivations, the objectives of the research work of this thesis is developed. The thesis employs evolutionary, soft computing and signal processing techniques to solve these problems of pulse compression. 1.6 Objective of the thesis The main objective of present research work is to propose efficient pulse compression techniques for different radar signals. The various objectives may be listed as: • To generate pulse compression biphase codes having lower peak sidelobes and better MF using multiobjective algorithm. 17 Introduction Chapter 1 • To develop efficient sidelobe reduction structures using neural networks which converge faster during the training time as well as provide higher magnitude of PSR. • To introduce and assess amplitude weighing technique for LFM waveform and polyphase codes which is expected to provide better PSR at higher Doppler shifts. • To select appropriate parameters of LFM pulse train to achieve reduced grating lobes, low peak sidelobe level and narrow mainlobe width. 1.6.1 Structure and chapter wise contribution of the thesis Chapter-1 The concept of pulse compression, matched filter, ambiguity function and different radar signals are introduced in this chapter. The motivation behind the application of evolutionary, neural network and signal processing techniques for pulse compression is outlined. The summary of framework of the research and contributions are also included. Chapter-2 In this chapter the biphase codes having lower PSL and better MF in their ACFs are generated. Genetic algorithm (GA) is used to optimize a cost function consisting of weighted combination of PSL and MF. However there is difficulty in selection of proper weight value to optimize the combination. In order to overcome this difficulty a multiobjective algorithm (based on nondominated sorting genetic algorithm-II (NSGA-II) ) is proposed which simultaneously optimize the PSL and MF. The proposed algorithm provides a set of nondominated solutions. Simulations have been carried out using proposed algorithm to generate pulse compression biphase codes for length 49 to 59. NSGA-II provides more than one nondominated codes for each length. A particular code of specified length is chosen in accordance to the requirement of the environmental condition. If the environmental condition is 18 Introduction Chapter 1 dominated by distributed clutter then the code having high MF is preferred. On the other hand if the application requires the detection of target in presence of large discrete clutter the code having low PSL is chosen. Chapter-3 Several mismatch filters are investigated in this chapter which provide better PSR values as compared to the matched filter. The best binary codes available in the literature are known as Barker codes having maximum sidelobe level of unity amplitude. The largest Barker code available is of length 13 having a PSR of magnitude 22.3 dB which is not adequate for many radar applications. The Barker codes of larger length are generated by taking Kronecker product of existing Barker codes. To obtain higher PSR value the mismatch filters such as adaptive linear combiner, multilayer perceptron (MLP) and radial basis function (RBF) along with their learning algorithms are investigated. The convergence performance of MLP and RBF structures is very slow. Therefore recurrent neural network (RNN) and recurrent RBF (RRBF) structures capable of yielding faster convergence are proposed for the pulse compression filter. The shifted version of 13-bit and 35-bit Barker codes are used as input to the different networks. The desired output of the network is always zero except at one point corresponding to the presence of target. The convergence rate during training for RNN and RRBF are compared to that of MLP and RBF networks. After the training is complete the networks are used for pulse radar detection. The PSR values of RRN and RRBF for different noise conditions, presence of multiple target and under Doppler shift condition are evaluated and compared with those of MLP and RBF. Chapter-4 This chapter presents pulse compression for LFM waveforms and polyphase codes. The LFM and polyphase codes have lower sidelobes compared to the biphase codes. LFM waveforms are more Doppler tolerant than phase coded waveforms. Polyphase codes are derived from the LFM waveform to get the advantage of the Doppler tolerant property of the LFM waveform. The matched filter output of the LFM 19 Introduction Chapter 1 waveform yields PSR of -13.2 dB. Different weighing functions are used in the receiver to achieve high PSR magnitude and the LFM waveform is amplitude tapered or phase distorted before transmission to get even higher PSR magnitude. The weighing functions are also used for sidelobe suppression of polyphase codes. If the target is in motion then the reflected signal is Doppler shifted version of the transmitted signal. In this chapter convolutional windows are proposed to use as weighing function for LFM and polyphase codes to achieve better PSR values in Doppler shift conditions. Simulation study is carried out to assess the performance of the convolutional windows and is compared to those of conventional windows. Chapter-5 In this chapter evolutionary computing techniques are proposed to determine the parameters of stepped frequency LFM pulse train. In case of high range resolution radar the required bandwidth is very high. The conventional narrowband hardware may not support the instantaneous wide bandwidth. Therefore, the wide bandwidth signal is split into narrowband signals which are transmitted and combined coherently at receiver to get the effect of the wideband signal. But the ACF of such narrow band pulse train suffers from grating lobes and hence destroys the range resolution capability of the pulse train. In the proposed work the particle swarm optimization (PSO) technique is used to determine the parameters of the LFM pulse train such that all the grating lobes are cancelled. Apart from cancellation or suppression of grating lobe, minimization of mainlobe width and peak sidelobe level of ACF are also important for the radar systems. In this chapter NSGA-II algorithm is proposed to choose the parameters of stepped frequency LFM pulse train to accomplish reduced grating lobes, low peak sidelobe and narrow mainlobe width. Chapter-6 In this chapter the overall contributions of the thesis are reported. This chapter also contains the details of further research work which can be attempted subsequently. 20 Introduction Chapter 1 1.7 Conclusion This chapter provides a brief introduction to radar, pulse compression technique and different signals used in radar. The merits and demerits of the pulse compression technique are studied. It also systematically outlines the scope, the motivation behind this work and the objectives of the thesis. In essence, this chapter provides an overview of the thesis in a comprehensive manner. 21 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm 2.1 Introduction In a pulse radar system the transmitted pulse width should be as long as possible to increase the sensitivity of the system and as small as possible at the receiver for better range resolution. Range resolution is the ability of the radar receiver to discriminate nearby targets. The performance of range resolution radar would be optimal, if the coded waveform has impulsive ACF. Biphase coded waveforms support better range resolution compared to LFM pulses because the windowing functions used with LFM pulses to lower time sidelobes cause a broadening in the mainlobe. But the ACF of biphase coded waveforms contain higher range sidelobes, which have a negative influence on the detection performance of radar systems. A desirable property of the compressed pulse is that it should have low sidelobes in order to prevent a weaker target from being masked in the sidelobes of a nearby stronger target. The lower the sidelobes relative to the mainlobe peak, the better the main peak can be distinguished and hence the better is the corresponding code. The selection of a pulse compression code depends on the application and the environmental conditions. If the application is radar designed for a scenario dominated by distributed clutter, then integrated sidelobe level (ISL) is very important. On the other hand if the 22 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm application requires detection of targets in the presence of large discrete clutter, then the PSL is more important. If the desired ISL or PSL performance is not achieved with a matched filter, a mismatch filter is used to achieve the desired sidelobe level with some SNR loss. Binary pulse compression codes [7] such as the Barker codes [8] or maximal-length sequences [9] are extensively used in radar systems. The Barker codes which are known as the best ideal waveform can provide a maximum PCR of 13. Many practical applications require longer codes to achieve higher PCRs much greater than 13. Therefore sequences with the lowest possible sidelobes at the longer length are needed. There is no analytical technique available to construct a sequence for a given PSL. Time consuming and money consuming exhaustive computer search program are generally used to generate best possible sequences. By exhaustive computer search program, Lindner [10] found all binary sequences up to length 40 with minimum PSL. With an improved algorithm Cohen et al. [11] further extended those results to sequence length 48. For larger sequences some heuristic methods, such as neural network (NN) and evolutionary algorithms (EAs) are proposed to search the binary sequences with good aperiodic autocorrelation [12–15]. Using an NN approach, Hu et al. [12] obtained useful binary sequences for lengths 49 up to 100. An objective function which consists of weighted sum of PSL and merit factor (MF) is optimized using genetic algorithm (GA) to generate codes from 49 to 100 [15]. The demerit of this type of objective function is to choose the accurate weight values. It is also required to run the program repeatedly for different combinations of weight values. To overcome this problem, in the proposed work a multiobjective algorithm is introduced in which PSL and MF are used as two different objective functions to generate the biphase codes. 23 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 2.2 Merit measures and problem formulation Let an L length binary sequence is given by S = {s1 , s2 , s3 , · · · , sL } (2.1) where each element of S has a value either +1 or -1. The ACF of S for positive delays is given as Ck (S) = L−k X si si+k (2.2) i=1 where k = 0, 1, 2, · · · , L − 1. Ideally, the range resolution radar signal should have high ACF value for zero shift and zero value for nonzero shift. A significant problem inherent in biphase pulse compression is that the ACF does not yield a perfect impulse, that means it does not produce Ck (S) = 0 for k 6= 0. Any non zero value of Ck (S) for k 6= 0 is referred to as sidelobe where as the zero-offset correlation value C0 (S) is called the mainlobe. The difference between a pulse compression waveform and a simple pulse waveform lies in the existence and magnitude of these sidelobes. These sidelobes limit the usefulness of a code regardless of the strength of the mainlobe. Codes are chosen for a given application based on their length and sidelobe levels. There are two main criteria [16, 17] used to decide the goodness of a pulse compression code. The first one is the PSL which is the largest sidelobe in the ACF of the code and defined as P SL = M ax |(Ck (S))| , k 6= 0 (2.3) The second one is the merit factor MF which is defined as the ratio of energy in the main peak of the ACF to the total energy in the sidelobes. As the signal is real valued the ACF is real and symmetric about the zero delay. The MF is represented as MF = C02 (S) PL−1 2 2 k=1 Ck (S) 24 (2.4) Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 The denominator of (2.4) is known as ISL. For a good sequence or code the PSL should be low and MF should be high. To optimize simultaneously PSL and MF using GA, the fitness function is defined as f= α + βM F P SL (2.5) The fitness function f is maximized using suitable values of α and β such that α + β = 1. However it is a difficult task to choose a proper combination of α and β to get a optimized code. Hence in the proposed work nondominated sorting genetic algorithm -II (NSGA-II) is used to optimize multiple objective functions PSL and MF simultaneously to generate biphase pulse compression codes. 2.3 Techniques used The techniques which are used to generate pulse compression codes are described in this section. 2.3.1 Genetic algorithm The GA is a programming technique that mimics biological evolution process and uses the genetic operators such as selection, crossover and mutation for problem solving strategy. GAs are based on Darwin’s theory of evolution i.e. the strong survivors have better opportunity to transfer their genes to future generations through reproduction. Species those carry correct combination in their genes are dominant in the population. Sometimes during the process of evolution, random changes may occur in genes. If these changes render advantages for the survival, new species evolve from the old ones. In other words unsuccessful changes are eliminated by the natural selection. The GA was originally proposed by J. Holland [18] in 1975 which imitates nature’s robust way of evolving successful organisms. Afterwards it became popular due to the publication of D. Goldberg’s book in 1989 [19]. Since then the GAs have been used in a wide range of applications where optimization is needed. In the GA, 25 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm a solution is called as an individual or chromosome and each element of chromosome called as genes. The GA works on a set of chromosomes called as a population. As the search evolves, the population have fitter and fitter solutions. The various steps involved in the GA are follows 1. Population initialization Population size is the number of chromosomes required in one generation. If there are too few chromosomes, GA have a few possibilities to perform crossover and only a small part of the search space is explored. On the other hand, if there are too many chromosomes, GA process will slow down. A chromosome is represented in such a way that it should contain information about the solution. The chromosomes are presented in real numbers such as 0.5, -0.3, 1.5 etc or encoded to binary form, using an encoding process, such as ‘1001010101’, ‘1001001010’ etc. This chapter is dealt with only binary representation of the chromosomes. M number of chromosomes are randomly initialized with binary forms. 2. Fitness function evaluation The initialized population is used to evaluate the objective function which is to be optimized. This is known as fitness function evaluation since the objective function value corresponds to the fitness of that chromosome. 3. Selection Chromosomes from the population are selected by using a mechanism to enter into a mating pool. Chromosomes from the mating pool are used to produce offspring which form the basis for the next generation. As the genes of the chromosome are to be inherited to the next generation, it is desirable the mating pool should contains good chromosomes. So a selection procedure in GA is used to select better individuals in the population for the mating pool. The selection pressure is the degree to which the better chromosomes are favored. The higher the selection 26 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm pressure, the more the better individuals are favored. Selection pressure helps GA to improve the population fitness over succeeding generations. There are many selection methods available in the literature such as roulette wheel, tournament, rank and steady state selection [20] etc. In this thesis a binary tournament selection is used to choose a chromosome from the population. In this selection a tournament size consisting of two chromosomes are randomly chosen and the winner of the two is the chromosome with the highest fitness value. The winner is entered into the mating pool. As the mating pool comprised of tournament winners, it has a higher average fitness than average population fitness. This fitness difference provides the selection pressure, which helps GA to improve the fitness of each succeeding generation. 4. Genetic operators Genetic operators such as crossover and mutation are used to explore and exploit new and better solutions from the existing solutions in the search space. The operators are explained below. a. Crossover In this operation two chromosomes, called parents, are selected using binary tournament selection from the existing population and combined together to form new chromosomes. These newly formed chromosomes are called offspring. It is always expected that offspring inherits good genes from the parent. A single point crossover and a two point crossover are shown in Figure 2.1. In case of single point crossover a point is randomly selected and all the genes after this point are swapped between the two parent chromosomes to form two offspring. Similarly, in case of two point crossover two points are randomly selected and the genes in between the two points are swapped between the two parents to form two offspring. This operation is carried out with certain probability called as crossover probability which indicates how often crossover will be performed. If there is a crossover, offspring is made from parts of parents chromosome. If crossover probability is 100%, then all offspring is 27 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm (a) Single Point Crossover (b) Two Point Crossover Figure 2.1: Crossover made by crossover. If it is 0%, whole new generation is made from exact copies of chromosomes from old population. Crossover is made in hope that new chromosomes will contain good parts of the old chromosomes and therefore the new chromosomes are better. However it is good to leave some part of the old population survive to next generation. b. Mutation It takes place at the gene level. It introduces random changes into the features of chromosomes. In GA the probability of mutation is smaller in comparison to the probability of crossover. If there is no mutation, offspring is taken after crossover (or copy) without any change. If mutation is performed, part of chromosome is changed. If mutation probability is 100%, whole chromosome is changed and if it is 0%, nothing is changed. Mutation reintroduces the genetic diversity back 28 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 into the population and helps to escape from the local minima. In case of binary representation of codes a randomly chosen bit is switched from 1 to 0 or 0 to 1 as shown in Figure 2.2. Figure 2.2: Mutation 5. Recombination and selection This process is used to weed out the weaker chromosomes from the population so that the more productive chromosomes will be used in the next generation. In most of the cases the fitness function value of a chromosome decides its survival for the next generation. The current generation population is combined with the offspring population and the fitness values of each chromosome of the combined population is evaluated. The best M chromosomes are selected according to the fitness value to carryout the next generation. A flow chart for GA operation is depicted in Figure 2.3. 2.3.2 Multi objective GA In single objective problems one has to find out the best solution which is usually the global maximum or minimum relying on the problem. In practice, the optimization problem is associated with multiple, possibly conflicting, objectives and this type of problem may not have one best solution with respect to all the objectives. A set of solutions exists in the search space which are superior to rest of the solutions with respect to all the objectives but are inferior among themselves with respect to one or more objectives. These solutions are called as nondominated solutions or Pareto optimal solutions. None of the nondominated solutions is better than the 29 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Figure 2.3: Flow chart for GA 30 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 other or in other words every solution is an acceptable solution. The superiority of one solution over the other depends upon the knowledge of the problem and its application. Thus, a solution chosen by a designer may not be accepted by another. A multiple objectives problem can be solved as a single objective problem by assigning a weight wi to each objective as follows: minimize z = w1 z1 (x) + w2 z2 (x) + ..... + wk zk (x) where z1 (x), z2 (x), ....zk (x) are the objective functions and (2.6) Pk i=1 wi = 1. In this approach a single set of weight vector produces a single solution. If multiple solutions are required the problem has to run repeatedly for different set of weights. The drawback of this type of approach is judicious selection of a weight vector for each solution, which is a difficult task. To overcome this difficulty many multiobjective evolutionary algorithms are found in literature to get a set of nondominated solution in a single run. In [21–26], the evolutionary algorithms are amply demonstrated and it is found that these are efficient to find multiple and diversified nondominated solutions. The difference between single objective GA and multiobjective GA (MOGA) is the concept of dominance used directly or indirectly in the selection phase of MOGA. The effective MOGA approximates the true Pareto front and maintains diversity in the population [21]. Schaffer [22] has proposed the first practical multiobjective algorithm, called as vector evaluated GA (VEGA). This algorithm solves each objective separately and then combines sub solution of each objective. One of the demerits of this algorithm is that it is biased towards some of the Pareto optimal solutions. In [23], an MOGA is proposed which explores the solution in all possible directions in the search space. Subsequently many GAs have been proposed by many researchers to find out improved nondominated solutions in the objective space. These algorithms are efficient in terms of complexity, rate of convergence, diversity among the nondominated solution and the interval distance from the Pareto optimal front. Deb and Srinivas [25] have proposed a robust popular nondominated sorting genetic algorithm (NSGA) to solve multiobjective optimization problems. But this algorithm involves high computational complexity, 31 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 lacks elitism and difficulty in choosing the optimal value of sharing parameter. An improved version of NSGA, called NSGA-II, is dealt in [26] which uses the concept of elitism and does not use the sharing parameter. The various steps of NSGA-II algorithm are given below. 1. Population initialization: The population contains a set of M chromosomes. Each chromosome is initialized randomly with binary bits. 2. Fitness function evaluation: The fitness functions which are to be optimized are evaluated for each chromosome. 3. Nondominated sort: The initialized population is sorted according to nondomination. The sorting algorithm [26] is given below. • for each solution x in the main population X do the following – the domination counter nx , the number of solution that dominate the solution x, is initialized as zero i.e nx = 0. – Sx , a set which contains all the solutions to those the solution x dominates, is initialized as an empty set φ i.e. Sx = φ – for each solution y in X ∗ if x dominates y · y is added to the set Sx i.e Sx = Sx ∪ {y}. ∗ else if y dominates x then · the domination counter of x is incremented i.e. nx = nx + 1. – if nx = 0 i.e. no solution dominates x then it belongs to the first front. Assign rank one to the solution i.e. xrank = 1. The first front is updated by adding x to it i.e. F1 = F1 ∪ {x}. 32 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 • This process is executed for all solutions in X. • The front counter i is initialized to one i.e. i = 1. • The following steps will be executed until ith front is non empty i.e.Fi 6= φ. – A set Y = φ is defined to store the solutions for next front. – for each solution x in front Fi ∗ for each solution y in Sx · ny = ny − 1, the domination count for solution y is decreased. · if ny = 0, then y belongs to the next front. Hence yrank = i + 1. The set Y is updated as Y = Y ∪ {y}. – The front counter incremented by one i.e. i = i + 1. – Y is set as next front i.e. Fi = Y . 4. Crowding distance: An efficient multiobjective algorithm not only converges to the true Pareto optimal set but also requires good spread or diversity among the obtained solutions. The original NSGA [25] uses a sharing parameter to achieve the diversity among the solutions. The difficulties of this algorithm are choosing the sharing parameter value and associated heavy computational complexity. These difficulties are overcome in NSGA-II by providing better diversity among the solutions using the concept of crowding distance. It does not require any user defined parameter to maintain the diversity among the solutions. The crowding distance is calculated front wise as follows. • For any front Fi , l is the number of solutions i.e. |Fi | = l. – The distance of all the solutions are initialized to zero i.e. Fi (Dj ) = 0, where the index j corresponds to j th solution in front Fi . – for each objective function m 33 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 ∗ The solutions in front Fi are sorted in ascending order according to the objective function value i.e. I = sort(Fi , m) ∗ Infinite distance value is assigned to the boundary solutions of front Fi i.e. Fi (D1 ) = Fi (Dl ) = ∞ ∗ for j = 2 to l − 1 Fi (Dj ) = Fi (Dj ) + I(j+1)·m−I(j−1)·m max −f min fm m where I(j) · m is the mth objective function value of j th solution in min max I. fm and fm are minimum and maximum value of mth objective function. • The above procedure is carried out for all the fronts. The solution that has large crowding distance value means it is far away from others, hence it is selected first. 5. Selection: a. Based on nondomination rank: A solution is selected if its nondomination rank is lower than other. b. Based on crowding distance: If two solutions belong to the same front, the solution having higher crowding distance is selected. The selection procedure is used during binary tournament selection and population reduction phase. 6. Genetic operators: These operators are used to produce offspring from the parent. The process of crossover and mutation are carried out as explained in Section 2.3.1 7. Recombination and selection: The current generation population is combined with the offspring population and selection is carried out to choose the best M solutions for next generation. In this 34 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm algorithm elitism is ensured as all the previous population is added with the current population for selection process. The algorithm for selection process is given below. • Let Ag and Bg are current and offspring population respectively of g th generation. The combined population Cg = Ag + Bg . • Sort the population Cg according to nondominated sort and determine all the fronts (F1 , F2 , ....). • Initialize i = 1 • The following is carried out until (|Cg+1 | + |Fi |) ≤ M – Assign the crowding distance to the solutions of the front Fi . – Update Cg+1 by adding all solutions of Fi to it i.e. Cg+1 = Cg+1 ∪ Fi . – i = i + 1. • Sort the solutions of front Fi in descending order of the crowding distance. • Update Cg+1 by adding first (M − |Cg+1 |) solutions of Fi i.e. Cg+1 = Cg+1 ∪ Fi [1 : (M − |Cg+1 |)]. The new population Cg+1 having M chromosomes is used for next generation. The NSGA-II procedure and a flow chart is presented in Figures 2.4 and 2.5 respectively. 2.4 Generation of pulse compression codes 2.4.1 Using genetic algorithm The fitness function defined in (2.5) is maximized to generate the biphase codes. The various steps are 1. The codes are to be generated are biphase codes i.e. they are consists of 1 or -1. So the chromosomes which are initialized must in the form of 1 or -1. 35 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Figure 2.4: NSGA-II procedure M number of chromosomes are randomly initialized with binary bits (1 or -1) with each chromosome of length same as the code length to be generated. 2. The ACF, PSL and MF for each chromosome are calculated according to (2.2), (2.3) and (2.4) respectively. The objective function value which is to be maximized is evaluated as given in (2.5). 3. The chromosomes are selected according to the binary tournament selection as described in Section 2.3.1. The selected chromosomes are used for off-spring generation. 4. The offspring are generated using the genetic operators such as crossover and mutation as explained in Section 2.3.1. In this case the binary bits with 1 and -1 are used as shown in Figure 2.6 and 2.7. 5. The current generation population is combined with the parent population and best M chromosomes are selected according to the fitness value to carryout the next generation. Steps from 3 to 5 are carried out until the maximum number of generation is met. 36 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Figure 2.5: Flow chart for NSGA-II 37 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 (a) Single Point Crossover (b) Two Point Crossover Figure 2.6: Crossover using binary bits 1 and -1 Figure 2.7: Mutation using binary bits 1 and -1 2.4.2 Using NSGA-II In the proposed work NSGA-II algorithm is employed to optimize the two objective functions PSL and MF as explained in Section 2.2. The different steps are 38 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm 1. The population is initialized with random binary bits (1 or -1) as in case of GA. 2. The two objective functions are evaluated as given in (2.3) and (2.4). PSL is minimized and MF is maximized simultaneously using the NSGA-II algorithm. 3. The chromosomes are sorted according to nondominated sort and found out all possible fronts as described in Section 2.3.2. 4. The crowding distance for chromosomes in each front are evaluated according to the procedure explained in Section 2.3.2. 5. The chromosomes are selected using binary tournament selection according to Section 2.3.2. 6. The selected chromosomes undergo genetic operations such as crossover and mutation to produce off-springs as explained in case of GA. 7. The off-spring population is combined with parent population and the best M chromosome selected for next generation as described in Section 2.3.2 The steps from 5 to 7 are carried out until the maximum number of generations is met. 2.5 Simulation results The GA dealt in Section 2.3.1 is used to maximize the fitness function f given in (2.5) to obtain the desired binary string. The population size M is chosen as 250 for each code generation. Each chromosome is randomly initialized and the fitness function f is calculated for all the chromosomes for a given combination of α and β. The process of selection, crossover and mutation are carried out to produce offspring. The two point crossover is used to generate the offspring. The probability of crossover and mutation are chosen to be 0.8 and 0.2 respectively. The current generation population is mixed with the offspring population and the best M chromosomes are 39 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 Table 2.1: Sequences obtained using GA Seq Len 49 50 51 52 53 α 0.9 0.8 0.7 0.6 0.5 0.9 0.8 0.7 0.6 0.5 0.9 0.8 0.7 0.6 0.5 0.9 0.8 0.7 0.6 0.5 0.9 0.8 0.7 0.6 0.5 β 0.1 0.2 0.3 0.4 0.5 0.1 0.2 0.3 0.4 0.5 0.1 0.2 0.3 0.4 0.5 0.1 0.2 0.3 0.4 0.5 0.1 0.2 0.3 0.4 0.5 PSL 5 5 6 6 6 5 5 6 6 7 5 6 6 6 7 5 6 6 6 8 6 6 6 7 7 MF 3.5309 4.0557 4.2271 4.2875 4.6173 3.1807 3.6232 3.7994 3.8941 3.9936 3.6429 3.6841 3.7264 3.7696 3.9529 3.9075 4.0479 4.2516 4.3057 4.7943 3.9075 3.6767 3.7553 4.2561 4.5306 Sequences 1101011100101010110110100001101100111111101100111 1001001110010111100000001010100001000011001011001 1100010111011100101110101001111001101000000100101 0101011000011110011100000010111101110111110110011 1111111100011010001110100010101101001100111100100 11010000100100000010100101110101110000010011011100 11101101100010010000101010001001111001000011001111 10011000011001000110010101001010011111010000010000 01010100010100101101110110110011000110001111100000 10100011001100101001111010001111101111010100000110 100011100101011010001100000001011111000100100100100 101101000101100000100101000011010111110011001100111 001011011111110110100110010101010000110111000011110 010101100011000101010111000000111110111011011011010 011100101110000010011100001001110111101101100101001 1110011110011001000011111000000010010010101010101101 1011111011001011111110000110000101001010110011001100 0011111101100011100010010001111110101101001101110101 1010101001110001110100100110011111100000000100100001 1101011010000000010011110000111011100010111010011001 10011111101100011101101011000111010111111010011010001 00101101110011101110110101111000011000001011111010100 01010010110110110000011110001100010001111110111011101 11000110100101100101100110011000000001111101010111101 01010100011000101111001101101100111101011011110100000 selected to carry out the next generation. The algorithm is run for 100 generations for each code. For different combinations of α and β the obtained codes from 49 to 53 are tabulated in Table 2.1. 0’s are used in place of -1’s to conserve space. From Table 2.1 it is observed that by giving different weightage to PSL and MF different codes are obtained. A particular code is selected according the requirement of application such as low PSL or high MF. If it requires low PSL, then a high value of α is required. If it requires high MF, then a high value of β is required. It is too difficult to choose the appropriate values for α and β to get an optimized code. The combination α = 0.8 and β = 0.2 produces better code as compared to the combination α = 0.9 and β = 0.1. Because in both the cases PSL is same but for the weight combination α = 0.8 and β = 0.2 provides better MF. 40 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm Chapter 2 Table 2.2: Sequences obtained using NSGA-II Seq Len 49 50 51 52 53 54 55 56 57 58 59 PSL 4 5 6 5 6 7 5 6 5 6 5 6 5 6 7 5 6 5 6 7 5 6 7 5 6 5 6 7 MF 4.2875 5.0869 5.1746 3.7994 3.8941 4.4484 4.3788 5.1403 4.4768 4.7943 3.9675 4.0129 3.8880 3.9728 4.1538 3.6446 3.6800 3.4087 4.0412 4.2609 3.1240 4.1441 4.6151 3.7798 4.8195 3.5020 4.2555 4.4288 Sequences 0011111000111111011000000100101100111001101010101 0011111100111111011000000100101100110100101010101 0011111100111111011010000100111100110100101010101 00010010011100111100100101000110111010101111111010 10010000011100111100100101000110111010101111111010 10010000111100111100100101000110111010101111111010 110000010000000110100110011100010001011110010101101 100010010000000110100110111100010001011110010101001 0000110011001100001111000000001010001011010101101101 0101001110001110100101110110110100000001101100010011 11001001101001000000010001001110101111000010101110001 11111011011101101000011100111001000010111010101100100 011110101101001000100000110000111010100010011101110100 110110101101001000100000110000111010100010011101110000 010110101101001000100000110000111010100010011101110000 1000111100010001011111001000110101011111110101101001001 1001101100010001111110001000011101011111110101111001011 00111000100011011100011010001000000010011011010000111101 00111010100011011100011010001000000010111011010000100101 00111010100011011100011010001000000010011011010000100101 110011010111110101110110101110000100010010000111111011000 110110010111110101110110101110001100010010000111111011000 110110010111010101110100101110001100010010000101111011000 1001010001110111000101011111000011101001110110010000000010 1001010001110011000101011011000011101001110110010000000000 00000100101100001111111100010101100011110100011001001110101 00000101101101001111101100010101110011110100011011001110100 00000100101101001111111100010101110011110100011001001110101 To overcome the difficulty of choosing the values of α and β and to get the optimized code in a single run, the code is generated by using multiobjective GA. PSL and MF are the two objective functions used for optimization using NSGA-II algorithm as explained in Section 2.4.2. The population size M is taken as 250 and the chromosomes are initialized randomly. The PSL and MF for each chromosome are found out and the population is sorted based on nondomination. Each chromosome in the first front have a rank value of 1 and the chromosome in the second front is assigned a rank value of 2 and so on. Crowding distance is assigned front wise to each chromosome. Parents are selected from the population 41 Chapter 2 Generation of Pulse Compression Codes Using Multiobjective Genetic Algorithm using binary tournament selection based on rank and crowding distance. The selected population generate offspring using crossover and mutation operations. The probability of crossover and mutation are chosen same as earlier. In this case also two point cross over is used to generate offspring. The offspring population is recombined with the current population and the best M chromosomes are selected for next generation. The number of generations is taken as 100. The sequences found by the proposed method for length 49 to 59 are listed in Table 2.2. In this case all the possible nondominated solutions can be achieved in a single run. For code length 49 the lowest PSL obtained is 4 and corresponding MF is 4.2875 which is better the lowest PSL obtained using GA i.e. 5 and corresponding MF is 3.5309. For a particular length, NSGA-II provides more than one solution. A solution is chosen according to the requirement of the application such as better PSL or better MF. 2.6 Conclusion By using NSGA-II algorithm a list of biphase sequences of length 49-59 has been generated and is listed in Table 2.2 along with their PSL and MF values. The results reveal that the proposed method performs better than the weighted sum approach in GA. The search for optimum sequence depends on the selection of the initial population of parent sequences. As the sequence length increases the search procedure requires more time for obtaining a good solution. The quality of solution improves with increase in the number of generations. 42 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction 3.1 Introduction The objective of pulse compression technique is to achieve appreciable PSR and acceptable range sidelobes in an economical manner. The types of waveforms used in this technique decide the cost and complexity of the radar system. The binary phase codes have better range resolution as compared to frequency coded waveforms. This advantage is obtained at the cost of high range sidelobes [27,28]. The reduction or elimination of range sidelobes can be achieved in a pulse compression radar by the use of a pulse which is coded in both amplitude and phase [29]. However, these techniques are seldom used because of the expensive amplitude modulation circuitry. With the increase in compactness and decrease in cost of the digital circuits due to the revolution of very large scale integrated (VLSI) circuits, it is appropriate to implement complex techniques offering improved performance. Therefore, efforts were made to devise alternative methods which could provide acceptable range sidelobes. The range sidelobes of binary phase codes are reduced by using different types 43 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction of digital filters whose parameters are determined by use of various algorithms. The efficiency of different approaches are evaluated by the ISL, PSL, hardware complexity involved and loss in SNR as compared to matched filter. Two methods are generally used to suppress the sidelobes. First one employs an additional weighing network after the matched filter. Rihaczek and Golden [30] have synthesized a filter in frequency domain for 13-bit Barker code which is a simple network and is able to suppress the sidelobes to an acceptable level. In this technique the complexity of the digital processor is reduced due to presence of few tap weights in the tap-delay filter. The second method is to design a mismatched filter directly [31, 32] instead of placing a weighing filter after the matched filter. In [31], an optimum mismatch filter is developed for 13-bit Barker code in least square (LS) sense, which gives optimal performance in terms of ISL. The weights of this filter is designed in such a way that the response which approximates (in least square sense) to a sequence, has all its elements as zeros except for one non zero element present at the central position. Zoraster [32] has used the linear programming (LP) to determine the filter weights for reduction of the PSL of 13-bit Barker code. To achieve satisfactory peak sidelobe and integrated sidelobes, the length of the LP filter should be very large which effectively increases the hardware cost. Hua and Oksman [33] have combined the advantages of [30] and [32] to obtain a new algorithm which provides lower peak sidelobes for 13-bit Barker code. In this method the transfer function of the sidelobe suppression filter is fitted with a polynomial expansion series in frequency domain, which consists of some unknown expansion coefficients. By applying inverse Fourier transform and LP, the coefficients of the transfer function can then be determined. A mismatch filter in cascade with a finite impulse response (FIR) filter is used in [34] to suppress the side lobes of Barker codes which requires less multipliers and adders due to its symmetry. With the advancement of adaptive signal processing and neural networks, researchers have put their efforts to design the sidelobe reduction filters using these techniques. Sidelobe reduction using adaptive filters are discussed in [35, 36] where 44 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction various algorithms like the least mean square (LMS), recursive least square (RLS) and modified RLS algorithms have been used to reduce the peak sidelobe level. Among these algorithms the modified RLS yields the highest PSR magnitude of 25.7 dB for 13-bit Barker code which is not suitable for many radar applications. This necessatiates the use of improved techniques like the neural networks. The neural network such as MLP has been applied [37–39] for pulse compression which provides PSR magnitude more than 40 dB for 13-bit Barker code. The weights of the MLP have been determined by training the network with all possible input patterns. Kwan and Lee [38] have employed back propagation (BP) algorithm and achieved acceptably good results. But the convergence speed of the BP algorithm is inherently slow and the network is sensitive to Doppler shift [39]. To overcome this drawbacks, the recurrent neural network (RNN) and recurrent radial basis function (RRBF) networks are proposed for pulse compression. 3.2 Problem formulation In biphase codes the transmitted pulse of duration Tp is divided into N sub pulses each of duration tb = Tp . N The ACF of transmitted code mathematically is represented [40] as N −|k| 1 X xi xi+|k| yk = N i=1 k = −N + 1, ....., N − 1 (3.1) xi = 1 for phase=0 and xi = −1 for phase=π. In matrix form (3.1) is written as y−N +1 y−N +2 ... y−1 y0 y1 ... yN −2 yN −1 x1 x2 ... xN −1 = 1 xN N 0 ... 0 0 0 x1 ... 0 0 ... xN −2 xN −1 xN ... 0 0 xN −3 xN −2 xN −1 ... 0 0 45 ... ... ... ... ... ... ... ... ... 0 0 ... x1 x2 x3 ... xN 0 0 0 ... 0 x1 x2 ... xN −1 xN xN xN −1 xN −2 ... ... ... ... x2 x1 (3.2) Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction The vector in the right hand side of (3.2) is the replica of the transmitted signal. In other words they are the weighing sequence for the received signal. So (3.2) can be expressed as y= 1 Xw1 N (3.3) where X is the matrix formed by shifting the input sequence {xi } and w1 is the weight vector. It is observed from (3.2) that X has 2N − 1 number of patterns. However, an additional null sequence {0} is considered for no input signal. So there is 2N number of patterns that are used as the input to the pulse compression filter. The desired output of the pulse compression filter for an input sequence is modeled as a all zero vector except at one point at which the desired response is nonzero. Thus the desired response is represented as d = [0 0 0 . . . 1 0 0 0 . . .]T (3.4) where [.]T denotes the transpose operation. The nonzero component represents the mainlobe. The problem is to design a suitable network for the pulse compression using input output pairs so as to get better performance in terms of PSR for range resolution, detection in presence of noise and Doppler shift. Different networks such as the adaptive linear combiner (ALC), MLP, RNN, RBF and RRBF are used as pulse compression network described in next section. The adaptive network contains connecting weights which are trained by various learning algorithms such as the LMS, RLS and BP etc. The weights of the filter, which provide input output relationship, are determined in an iterative manner. 3.3 Techniques used In this section various models such as ALC, MLP, RNN, RBF, RRBF and their learning algorithms are discussed. These models are used as mismatch filter for radar pulse compression. 46 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Figure 3.1: Adaptive linear combiner 3.3.1 Adaptive linear combiner An ALC or nonrecursive adaptive filter is a computational device that attempts to model the relationship between input and output in an iterative manner. The general form of an ALC [42, 43] is depicted in Figure 3.1. The input signals to the ALC are patterns and the nth pattern is represented as x(n) = [x1 (n), x2 (n), .....xN (n)]T (3.5) where N represents the number of elements in each pattern. The ALC contains a set of adjustable weights given by w = [w1 , w2 , .....wN ]T (3.6) The estimated output of nth pattern is y(n) = wT x(n) (3.7) The training algorithms for ALC are explained below. (a)Least mean square algorithm There are many algorithms found in literature to train various adaptive models. The performance of these algorithms depend on the rate of convergence, training 47 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction time, computational complexity and residual mean square error (MSE). The LMS algorithm is mostly used to train the weights of adaptive filters because of its low computational complexity and ease of implementation. From Figure 3.1 the error signal for nth pattern is obtained as e(n) = d(n) − y(n) (3.8) where d(n) is the desired output for nth pattern. The cost function to be optimized is n 1 1X e2 (n) ξ= 2 n=1 (3.9) where n1 is the number of patterns. The weights associated with the filter are adjusted in such a way that the cost function is minimized. The proposed study uses an epoch based adaptation for weight updation. The ALC is trained with all n1 patterns and the change in weight for each pattern is stored. These change in weights are used for a single update of the filter weights which in turn constitutes an epoch. The new weights are used to carry out the next epoch. The LMS algorithm uses the gradient descent technique to minimize the cost function and the weights are updated [43] as n1 X ∂ξ e(n)x(n) = wk (m) + µ wk (m + 1) = wk (m) − µ ∂wk (m) n=1 (3.10) where k = 1, 2....N and m is the epoch index. (b)Recursive least square algorithm The algorithm such as LMS is derived by using some approximation made in the estimate of the performance function gradient. This type of algorithm have the disadvantages that they are slow to obtain the optimum weight vector and once close to it, usually “rattle around” the optimal vector rather than actually converging to it. To overcome this difficulty, another efficient approach known as RLS algorithm has been discussed in this section. The advantage gained by the use of the RLS 48 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction algorithm is at the expense of an increase in computational complexity. Steps involved in RLS algorithm (a) Accept new pattern x(n) and corresponding desired output d(n). (b) Compute the estimated output y(n) as expressed in (3.7). Initially assume the weight vector as zero. (c) Compute the error e(n) as given in (3.8). (d) Compute filtered information vector z(n): z(n) = R−1 (m)x(n) (3.11) where R is the autocorrelation matrix of input pattern. The R−1 (m) is assumed to be exist, where m is epoch index. Initially R−1 (m) is taken as ηI, where I is an identity matrix of size N × N . The value of η taken as very large i.e. about 104 . (e) Compute normalized error power q: q = xT (n)z(n) (3.12) (f) Compute gain constant v: v= 1 1+q (3.13) (g) Compute the normalized information vector ẑ(n): ẑ = vz(n) (3.14) (h) Compute the change in weight vector as ∆w(n) = e(n)ẑ (3.15) (i) Compute the change in inverse correlation matrix ∆R−1 (n) = −ẑ(n)zT (n) 49 (3.16) Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction The steps from (a) to (i) are carried out for all n1 patterns. The weights are updated as w(m + 1) = w(m) + n1 X ∆w(n) (3.17) n=1 and the inverse correlation matrix is updated as −1 −1 R (m + 1) = R (m) + n1 X ∆R−1 (n) (3.18) n=1 The updated values of the weights and the inverse correlation matrix are used to carry out next epoch. (c)Modified RLS algorithm The modified RLS algorithm is derived by using the condition |e(n)| ≥ Th (3.19) where Th represents a threshold value. The instantaneous error is compared with the threshold value. If the instantaneous error is greater than the threshold value then steps from (d) to (i) of the RLS algorithm are evaluated otherwise not. If |e(n)| < Th , then values of ∆R−1 (n) and ∆w(n) are zero. The weights and the inverse correlation matrix are updated as given in (3.17) and (3.18) respectively. Initially the threshold value is chosen as very small and later it is updated for each epoch based on the maximum error value at that epoch. The updation of threshold at mth epoch is given by maxerrm = maximum|em (n)| (3.20) Thm = δ ∗ maxerrm (3.21) where em (n) is the error vector at mth epoch and maxerrm is the maximum value of all the errors in error vector. δ, a constant whose value is close or equal to 1, affects the rate of convergence. 50 Chapter 3 3.3.2 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Artificial neural network An artificial neural network (ANN) is an information processing paradigm that is inspired by the way biological nervous system, such as the brain, process the information [44, 45]. The first artificial neuron was developed in 1943 by the neurophysiologist Warren McCulloch and the logician Walter Pits. But the technology available at that time did not allow them to proceed further. In past few decades the ANN has emerged as a powerful learning tool to perform complex tasks in highly nonlinear dynamic environment. The ANN is capable of performing nonlinear mapping between the input and output space due to its large parallel interconnection between different layers and the nonlinear processing characteristic. Therefore, the ANN is used extensively in the field of communication, control, instrumentation and forecasting [46–48]. ANN technique is also used for classification, modeling and optimization problems [49, 50]. An artificial neuron basically consists of a computing element that performs the weighted sum of the input signal and the connecting weight. The sum is added with the bias or threshold and the resultant signal is then passed through a nonlinear function of sigmoid or hyperbolic tangent type. Each neuron is associated with three parameters whose learning can be adjusted. These are the connecting weights, the bias and the slope of the nonlinear function. For the structural point of view a neural network (NN) may be single layer or it may be multilayer. In MLP there is a number of layers and each layer contains one or many artificial neurons. Each neuron of the one layer is connected to each and every neuron of the next layer. A trained neural network can be thought of as an “expert” in the category of information it has been given to analyze. The advantages of ANN are (a) Adaptive learning: It is the ability of the network to learn how to do tasks based on the data given for training or initial experience. (b) Self-organization: An ANN can create its own organization or representation of the information as it receives during learning time. 51 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction (c) Real time operation: The ANN computations may be carried out in parallel, and special hardware devices are being designed and manufactured which take advantage of this capability. (d) Fault tolerance via redundant information coding: Partial destruction of a network leads to the corresponding degradation in performance. However, some network capabilities may be retained even with major network damage. Single neuron structure A neuron is an information processing unit for the operation of a neural network. The operation in a single neuron involves the computation of the weighted sum of inputs and threshold. The resultant signal is then passed through a nonlinear activation function. The basic structure of a single neuron is shown in Figure 3.2. The output associated with the neuron is computed as Figure 3.2: Single neuron structure y=f " N X wi x i + b i=1 # (3.22) where xi , i = 1, 2...N , are inputs to the neuron, wi is the synaptic weights of the ith input, b is the bias and f is the nonlinear activation function. The activation functions generally used in neural computation are discussed below. Activation functions The activation or transfer function may be a linear or a nonlinear in nature. A particular transfer function is chosen to satisfy some specification of the problem that the neuron is attempting to solve. Some of the activation functions are explained below. 52 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction 1. Log-sigmoid function This transfer function takes the input and squashes the output into the range of 0 to 1, according to expression given below f (x) = 1 1 + e−x (3.23) 2. Hyperbolic tangent Sigmoid: This function is expressed as f (x) = tanh(x) = ex − e−x ex + e−x (3.24) 3. Signum Function: The expression for this activation function 1 if f (x) = 0 if −1 if is given by x>0 x=0 (3.25) x<0 4. Threshold function This function is given by the expression 1 if x ≥ 0 f (x) = 0 if x < 0 (3.26) This function is represented as 1 f (x) = x 0 (3.27) 5. Piecewise linear function if x > 0.5 if − 0.5 ≤ x ≤ 0.5 if 53 x < 0.5 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction The different structures of neural network and their learning algorithms are described below. (i) Multilayer perceptron The MLP is a feed forward network having an input layer, one or more hidden layer and an output layer. The layer to which the input data is given called as input layer and the layer from which output is taken called as output layer. All the intermediate layers are called as hidden layers. The layers are fully interconnected i.e. each neuron is connected to every neuron in previous and succeeding layers. The input signal propagates through the network on a layer by layer basis. This network have been applied successfully to solve many nonlinear and complex problems in several fields. The structure of a three layer MLP is shown in Figure 3.3 which consists of input layer, one hidden layer and output layer. i, j and k are the indices used for input, hidden and output layer respectively. x(n) = [x1 (n), x2 (n), ..., xN (n)]T is the input to the network for nth pattern and wji is the synaptic weight connecting input xi (n) to the hidden neuron j. Similarly wkj is the synaptic weight connecting output of j th hidden neuron output to the k th neuron of output layer. bj and bk are the biases to the hidden layer and output layer respectively. f represents the nonlinear activation function for both hidden and output layer. The activation functions can be different for different layers. The output of j th hidden neuron for nth pattern is yj (n) = f (aj (n)) where aj (n) = N X wji xi (n) + bj i=1 The response of the k th (3.28) ! (3.29) output node is yk (n) = f (ak (n)) where ak (n) = n2 X wkj yj (n) + bk j=1 54 (3.30) ! (3.31) Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Figure 3.3: Mlutilayer perceptron network and n2 is the number of hidden nodes. Back propagation algorithm This algorithm is used to train the parameters of the MLP to get optimum cost function. Basically BP learning comprises of two passes through the different layers of networks: a forward pass and a backward pass. In forward pass a pattern (input vector) is applied to the input nodes and its effect propagates through the network layer by layer. In forward pass the synaptic weights remain constant. On the other hand during backward pass the synaptic weights and the biases are adjusted according to the error correction rule. The parameters of the neural network are updated by BP on epoch basis. The error of the k th neuron output for nth pattern is ek (n) = dk (n) − yk (n) (3.32) where dk (n) and yk (n) are desired and estimated outputs respectively. The cost 55 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 function which is to be minimized is defined as n n 1 3 X 1X ek (n)2 ξ= 2 k=1 n=1 (3.33) where n3 is the number of neurons in the output layer. According to the gradient descent, each weight change in the network should be proportional to the negative gradient of the cost function with respect to the specific weights. The local gradient for k th output neuron is δk (n) = − ∂ξ ∂ek (n) ∂yk (n) ′ = (dk (n) − yk (n)) f (ak (n)) ∂ek (n) ∂yk (n) ∂ak (n) (3.34) ′ where f (ak (n)) is the first derivative of f (ak (n)) with respect to ak (n). The local gradient for j th the hidden nodes is δj (n) = − n3 X k=1 n 3 ∂ξ ∂ek (n) ∂yk (n) ∂ak (n) ∂yj (n) X ′ δk (n)wkj (m)f (aj (n)) = ∂ek (n) ∂yk (n) ∂ak (n) ∂yj (n) ∂aj (n) k=1 (3.35) where m is the epoch index. The local gradients for all patterns are calculated in each epoch. The change in weights for output layer in mth epoch is ∆wkj (m) = η1 n1 X δk (n)yj (n) (3.36) n=1 where η1 is the learning parameter. The change in biases for output layer is ∆bk (m) = η1 n1 X δk (n) (3.37) n=1 The output layer weights and biases are updated as wkj (m + 1) = wkj (m) + ∆wkj (m) (3.38) bk (m + 1) = bk (m) + ∆bk (m) (3.39) The change in weights for hidden layer is ∆wji (m) = η1 n1 X n=1 56 δj (n)xi (n) (3.40) Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction The change in biases for hidden layer is ∆bj (m) = η1 n1 X δj (n) (3.41) n=1 The hidden layer weights and biases are updated as wji (m + 1) = wji (m) + ∆wji (m) (3.42) bj (m + 1) = bj (m) + ∆bj (m) (3.43) (ii) Recurrent neural network Recently, significant research work has been carried out to demonstrate the effectiveness of recurrent neural network in modeling of nonlinear dynamic systems [51–56]. The RNN has many advantages over static layered networks when used for system identification and feed back controller [51,52]. Moreover, RNN is capable for long range prediction in the presence of measurement noise and also able to filter the noise from the inputs [57]. The RNNs are used to model the plant nonlinearities in more efficient ways as compared to feed forward network. The RNN has at least one feed back loop in its architecture which is not present in feed forward network. Thus in the RNN, there may be one layer with feed back connections as well as there may be neurons with self feed back where output of the neuron is fed back into itself as the input. The presence of feed back loop affects heavily on the learning capability of the network. Contrary to the MLP, the RNN is sensitive and adaptive to past inputs. Among the several ANN architectures available in the literature, ANN having feed back and internal dynamics have been considered more suitable for modeling and control of the nonlinear systems as compared to feed forward network [58]. A block diagram of the RNN is shown in Figure 3.4. The RNN has structure as that of MLP with feedback or recurrent connections. This network has recurrent connections from the hidden neurons to a layer of context units consisting of bank of unit delays [59]. These context units store the outputs of hidden neurons for one time step and feed them back to the input layer. The inputs to the hidden layers are combination of the present inputs and the outputs of the hidden layer which 57 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 Figure 3.4: Block diagram of recurrent neural nerwork are stored from previous time step in context layer. Hence the outputs of the RNN are functions of present state, previous state (that is stored in context units) and present inputs. In this case the the j th hidden layer outputs are calculated as yj (n) = f (aj (n)) where aj (n) = f N X wji xi (n) + i=1 n2 X h=1 (3.44) ujh yj (n − 1) + bj ! (3.45) where n2 is the number of hidden nodes and ujh are the recurrent layer weights. The response of the k th neuron in output layer is yk (n) = f (ak (n)) where ak (n) = n2 X wkj yj (n) + bk j=1 (3.46) ! (3.47) Weight updation for recurrent neural network The hidden node output is used to compute the response of output layer of the RNN as given in (3.46). The local gradient and weight update procedure are same as that of MLP. The change in recurrent layer weights are obtained as n1 X δj (n)yj (n − 1) (3.48) uhj (m + 1) = uhj (m) + ∆uhj (m) (3.49) ∆uhj (m) = η1 n=1 Recurrent layer weights are updated as 58 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Hence all weights are updated based on the corresponding weight correction equations. (iii) Radial basis function network The radial basis function network can be viewed as a feed forward neural network with a single hidden layer which computes the distance between input pattern and the center [60]. It consists of three layers, an input layer, a hidden layer and an output layer. The input layer connects the network to the environment. The second layer is the only hidden layer which transfer the input space nonlinearly using radial basis function. The hidden space is greater than the input space in most of the applications. The response of the network provided by the output layer which is linear in nature. The RBF network is suitable for solving function approximation, system identification and pattern classification because of its simple topological structure and their ability to learn in an explicit manner [61, 62]. The Figure 3.5: Architecture of radial basis function network basic architecture of RBF network is shown in Figure 3.5. Here x(n) is the input to the network and φ represents the radial basis function that perform the nonlinear mapping and M represents the total number of hidden units. Each node has a center vector ck and spread parameter σk , where k = 1, 2, ....M . Radial basis functions The radial basis functions are represented by φ(kx, ck), where k.k represents the Euclidean norm. The radial basis functions which are generally used in various 59 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction applications are 1. Multiquadratics 1 φ(r) = (r2 + c2 ) 2 f or c > 0, r ∈ R (3.50) f or c > 0, r ∈ R (3.51) f or σ > 0, r ∈ R (3.52) wk (m)φ (x(n), ck (m), σk (m))) (3.53) 2. Inverse multiquadratics 1 φ(r) = (r2 + c2 )− 2 3. Gaussian function φ(r) = exp(− r2 ) 2σ 2 Learning algorithm for RBF network The error for the nth pattern is obtained as e(n) = d(n) − M X k=1 where d(n) is the desired output. If the Gaussian function chosen as the radial basis function M X kx(n) − ck (m)k2 e(n) = d(n) − wk (m)exp − σk2 (m) k=1 ! (3.54) The cost function is defined as n 1 1X ξ= e2 (n) 2 n=1 (3.55) where n1 is the number of training patterns. It is required to adjust the free parameters such as weight, center and spread so as to minimize ξ. According to the gradient descent algorithm the free parameters for mth epoch are updated as ∂ξ ∂wk (m) (3.56) ∂ξ ∂ck (m) (3.57) wk (m + 1) = wk (m) − µw ck (m + 1) = ck (m) − µc 60 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction σk (m + 1) = σk (m) − µσ ∂ξ ∂σk (m) (3.58) where µw , µc and µσ are learning parameters and k = 1, 2...M . Finally the updation equations are defined as wk (m + 1) = wk (m) + n1 X µw e(n)φ (x(n), ck (m), σk (m)) (3.59) n=1 ck (m + 1) = ck (m) + n1 X µc n=1 σk (m + 1) = σk (m) + n1 X n=1 where e(n)wk (m) φ (x(n), ck (m), σk (m)) [x(n) − ck (m)] (3.60) σk2 (n) µσ e(n)wk (m) φ (x(n), ck (m), σk (m)) [kx(n) − ck (m)k2 ] σk3 (m) (3.61) kx(n) − ck (m)k2 φ (x(n), ck (m), σk (m)) = exp − σk2 (m) ! (3.62) (iv) Recurrent radial basis function The RRBF [63] combines the advantages of RBF and dynamic representation of time. The RRBF network has been applied for modeling [64], noise cancellation [65, 66] and time series [67] prediction. This network has faster convergence [68] while maintaining the modeling capability of neural networks. The architecture of Figure 3.6: Architecture of recurrent radial basis function network RRBF model is shown in Figure 3.6. The model of RRBF is similar to RBF with an input layer, one hidden layer and an output layer. In this network each output of 61 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 the hidden neurons are fed back to their corresponding input through a delay. The estimated output of the network for nth pattern is y(n) = M X wk (m)φ (x(n), ck (m), σk (m)) (3.63) k=1 where φ (x(n), ck (m), σk (m)) = ! kx(n) − ck (m)k2 + gk (m)φ (x(n − 1), ck (m), σk (m)) (3.64) exp − σk2 (m) Learning algorithm for RRBF In this case the cost function is same as that of RBF as defined in (3.55). wk , ck and σk are updated as that of RBF using the currently defined φ (x(n), ck (m), σk (m)). The recurrent weights are updated as gk (m + 1) = gk (m) − µg ∂ξ ∂gk (m) (3.65) where µg is the learning parameter. n 1 X ∂ξ wk (m)φ (x(n), ck (m), σk (m)) φ (x(n − 1), ck (m), σk (m)) = ∂gk (m) n=1 (3.66) From (3.66) and (3.67) gk (m + 1) = gk (m) − µg n1 X n=1 wk (m)φ (x(n), ck (m), σk (m)) φ (x(n − 1), ck (m), σk (m)) (3.67) where k = 1, 2...M . 3.4 Simulation results This section illustrates the performance of various networks for radar pulse compression. First, the performance of ALC trained with LMS, RLS and Modified 62 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction RLS algorithm is presented. Subsequently the performances of MLP, RBF along with the proposed RNN and RRBF are presented. All the networks are trained with time shifted sequences of the 13-bit and 35-bit Barker codes. The time shifted sequence for 13-bit Barker code is presented in Figure 3.7. A null sequence {0} is added to the shifted sequence that represents radar has not received any information. So there are 26 patterns for 13-bit Barker code. Similarly the number of patterns in case of 35-bit Barker code is 70. In these training sequences the desired output of the network is 1 when the proper Barker code present in the input, otherwise the output is zero. Figure 3.7: 26 different possible input sequences for 13-bit Barker codes 63 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Table 3.1: PSRs obtained using various learning algorithms. Algorithms 3.4.1 13-bit Barker code 35-bit Barker code (PSR in dB) (PSR in dB) ACF -22.27 -13.97 LMS -23.86 -16.6 RLS -24 -16.65 modified RLS -25.52 -18 Sidelobe suppression using adaptive linear combiner The adaptive linear combiner is uniquely defined by its weight coefficients. The length of the weight vector is taken to be same as the input sequence and weights are initialized to zero. 13-bit and 35-bit Barker codes are used as inputs to the filter. Several algorithms like LMS, RLS and modified RLS with suitable parameter values are used for updating the weights of the linear combiner to minimize the error. For LMS algorithm the convergence parameter µ is chosen as 0.01 and for RLS the value of η is chosen as 104 . Similarly for modified RLS, Th = 0 and δ = 0.995 are used. The network is trained for 500 epochs for each algorithm. Once the training is over, the network can be used as pulse compression filter. Output of pulse compression filter for different algorithm for 13-bit and 35-bit Barker codes are depicted in Figures 3.8 and 3.9 respectively. The PSRs for ACF, LMS, RLS and modified RLS using 13-bit and 35-bit Barker codes as input are given in Table 3.1. From the table it is clear that the modified RLS algorithm gives the highest PSR of magnitude 25.52 dB for 13-bit Barker code and 18 dB for 35-bit Barker code. But these low magnitude of PSR values are not suitable for many radar applications. 3.4.2 Sidelobe suppression using MLP, RNN, RBF, RRBF The MLP and RNN consist of input layer one hidden layer and output layer. The log-sigmoid function is used as the activation function in hidden and output layers. 64 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 5 10 15 20 25 15 20 25 15 20 25 15 20 25 Time delay (a) 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 5 10 Time delay (b) 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 5 10 Time delay (c) 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 5 10 Time delay (d) Figure 3.8: Filter response in dB for 13-bit Barker code obtained using (a)ACF (b)LMS (c)RLS (d)Modified RLS algorithms 65 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 10 20 30 40 50 60 70 40 50 60 70 40 50 60 70 50 60 70 Time delay (a) 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 10 20 30 Time delay (b) 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 10 20 30 Filter response in dB (c) 0 Filter response in dB −5 −10 −15 −20 −25 −30 −35 0 10 20 30 40 Time delay (d) Figure 3.9: Filter response in dB for 35-bit Barker code obtained using (a)ACF (b)LMS (c)RLS (d)Modified RLS 66 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction The number of input neurons are same as the length of the input code i.e. 13 for 13-bit Barker code and 35 for 35-bit Barker code. The number of hidden layer and output layer neurons are chosen as three and one respectively. The weights and the biases are randomly initialized. The learning parameter η1 is chosen as 0.8. The RBF and RRBF consist of seven hidden neurons having Gaussian radial basis function and one output neuron is used. Weight(w), centre(c) and spread (σ) parameters are randomly initialized. The values of learning parameters µw , µc and µσ for RBF are chosen as 0.75, 0.8 and 0.75 respectively. Similarly the values of learning parameters µw , µc , µσ and µg for RRBF are chosen as 0.8, 0.8, 0.75 and 0.8 respectively. All the four networks are trained for 500 epochs according to their learning algorithm given in Section 3.3. After completion of the training, the neural network can be used for pulse radar detection by using various set of input sequences. Convergence performance The MSE of all the networks for 13-bit and 35-bit Barker codes are depicted in Figure 3.10. From the figure it is evident that the RRBF based approach offers better convergence speed and very low residual error after training for 13-bit and 35-bit Barker codes as compared to all other networks. PSR performance After the training is over, different inputs are applied to the networks to examine PSR performance. The compressed output of different networks for 13-bit Barker code is shown in Figure 3.11. The PSR values of all the networks for 13-bit and 35-bit Barker codes are listed in Table 3.2. The table shows that the proposed RRBF network have achieved highest PSR magnitude for both 13-bit and 35-bit Barker codes compared to all other approaches. Noise performance Noise is a random signal which interferes with the target echoes. If the noise is very high it may mask the target echo. So it is also required to examine the noise 67 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 −1 10 −2 MLP RNN RBF RRBF Mean square error(log10) 10 −3 10 −4 10 −5 10 −6 10 0 50 100 150 200 250 300 350 400 450 500 Epochs (a) −1 10 −2 MLP RNN RBF RRBF Mean square error(log10) 10 −3 10 −4 10 −5 10 −6 10 0 50 100 150 200 250 300 350 400 450 500 Epochs (b) Figure 3.10: Convergence graphs of different structures for (a)13-bit (b)35-bits Barker codes Table 3.2: PSRs obtained by various structures Structures 13-Bit Barker Code 35-Bit Barker Code (PSR in dB) (PSR in dB) MLP -42.61 -40.87 RNN -45.75 -44.93 RBF -60.43 -56.42 RRBF -64.31 -62.35 68 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 Filter response 1.2 1 0.8 0.6 0.4 0.2 0 0 5 10 15 20 25 30 20 25 30 20 25 30 20 25 30 Time delay (a) Filter response 1.2 1 0.8 0.6 0.4 0.2 0 0 5 10 15 Time delay (b) 1.2 Filter response 1 0.8 0.6 0.4 0.2 0 −0.2 0 5 10 15 Time delay (c) 1.2 Filter response 1 0.8 0.6 0.4 0.2 0 −0.2 0 5 10 15 Time delay (d) Figure 3.11: Compressed waveforms for 13 bit Barker code using (a)MLP (b)RNN (c)RBF (d)RRBF structures 69 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction rejection ability of different networks. The inputs having different SNR ranging from 0 dB to 20 dB are applied to the networks and the output PSR for 13-bit and 35-bit Barker codes are listed in Tables 3.3 and 3.4 respectively. These tables show that as the SNR increases the magnitude of PSR also increases. The RRBF provides highest magnitude of PSR in all SNR values compared to those obtained by all other approaches. Table 3.3: Comparison of PSRs in dB at different SNRs for 13-bit Barker code Structures SNR=0dB SNR=5dB SNR=10dB SNR=15dB SNR=20dB MLP -14.23 -28.61 -36.71 -38.53 -39.82 RNN -17.11 -32.17 -38.35 -40.59 -41.76 RBF -35.28 -45.23 -50.33 -55.77 -57.62 RRBF -40.24 -49.27 -57.30 -60.12 -61.24 Table 3.4: Comparison of PSRs in dB at different SNRs for 35-bit Barker code Structures SNR=0dB SNR=5dB SNR=10dB SNR=15dB SNR=20dB MLP -15.18 -29.17 -32.43 -36.95 -38.12 RNN -19.52 -32.74 -37.83 -40.87 -42.65 RBF -40.25 -48.25 -52.78 -54.44 -55.17 RRBF -42.25 -54.69 -57.47 -58.60 -60.57 Range resolution ability The range resolution is to analyze the ability of a particular network to distinguish between two targets by measurement of their ranges in the radar system. The two targets which are to be resolved must be separated by at least the range equivalent of the width of the processed echo. To compare the range resolution ability two overlapping codes of same length are considered with n-delay apart (DA) having 70 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Table 3.5: Comparison of range resolution ability for 13-bit Barker code of two targets having same IMR and DA. Structures MLP 2-DA 3-DA 4-DA 5-DA (PSR in dB) (PSR in dB) (PSR in dB) (PSR in dB) -36.53 -38.52 -37.32 -36.16 RNN -40.24 -41.23 -39.23 -38.78 RBF -53.32 -55.25 -56.76 -54.23 RRBF -59.72 -58.28 -60.73 -58.71 Table 3.6: Comparison of range resolution ability for 35-bit Barker code of two targets having same IMR and DA. Structures MLP 2-DA 3-DA 4-DA 5-DA (PSR in dB) (PSR in dB) (PSR in dB) (PSR in dB) -34.41 -34.83 -33.75 -32.62 RNN -38.23 -37.79 -36.87 -35.25 RBF -48.34 -47.61 -49.82 -47.13 RRBF -53.72 -55.14 -54.25 -53.23 same or different input magnitude ratio (IMR). The IMR is defined as the magnitude of first pulse train over that of the delayed pulse train. Figure 3.12 shows the added input waveform of equal magnitude (IMR=1) with 5 delay apart for 13-bit Barker code. The compressed output for this input for all the network are shown in Figure 3.13. In this case the PSR is calculated by taking lower value of the two mainlobes. By varying the DA from 2 to 5 the PSR for 13-bit and 35-bit Barker codes are obtained and shown in Tables 3.5 and 3.6 respectively. In Tables 3.7 and 3.8 the PSR for different IMRs and DAs for all the networks are listed. From these tables it is evident that the PSR values for RRBF are the best among those offered by all other networks i.e. RRBF based pulse compression technique have best range resolution ability compared to those of other networks. 71 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction 1 0.8 0.6 0.4 Input 0.2 0 −0.2 −0.4 −0.6 −0.8 −1 0 2 4 6 8 10 12 14 16 18 20 12 14 16 18 20 12 14 16 18 20 Time delay (a) 1 0.8 0.6 0.4 Input 0.2 0 −0.2 −0.4 −0.6 −0.8 −1 0 2 4 6 8 10 Time delay (b) 2 1.5 Added input 1 0.5 0 −0.5 −1 −1.5 −2 0 2 4 6 8 10 Time delay (c) 2 1.5 1 Input 0.5 0 −0.5 −1 −1.5 −2 0 2 4 6 8 10 Time delay 12 14 16 18 20 (d) Figure 3.12: Input waveform on addition of two 5-DA 13-bit Barker sequence having same magnitude (a)Left shift (b)Right shift (c)Added waveform (d)Waveform after flip about the vertical axis 72 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 Filter response 1.2 1 0.8 0.6 0.4 0.2 0 0 5 10 15 20 25 30 20 25 30 20 25 30 20 25 30 Time delay (a) Filter response 1.2 1 0.8 0.6 0.4 0.2 0 0 5 10 15 Time delay (b) 1.2 Filter response 1 0.8 0.6 0.4 0.2 0 −0.2 0 5 10 15 Time delay (c) 1.2 Filter response 1 0.8 0.6 0.4 0.2 0 −0.2 0 5 10 15 Time delay (d) Figure 3.13: Compressed waveforms for 13-bit Barker code having same IMR and 5 DA for (a)MLP (b)RNN (c) RBF (d)RRBF structures 73 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Chapter 3 Table 3.7: Comparison of range resolution ability for 13-bit Barker code of two targets having different IMR and DA. Structures MLP 2-DA 3-DA 4-DA 5-DA 2-IMR 3-IMR 4-IMR 5-IMR (PSR in dB) (PSR in dB) (PSR in dB) (PSR in dB) -38.17 -30.23 -24.16 -12.14 RNN -40.23 -36.56 -31.14 -23.65 RBF -51.38 -49.42 -43.24 -33.18 RRBF -56.36 -55.42 -50.24 -39.37 Table 3.8: Comparison of 35-bit Barker code for range resolution ability of two targets having same IMR and DA Algorithms MLP 2-DA 3-DA 4-DA 5-DA 2-IMR 3-IMR 4-IMR 5-IMR (PSR in dB) (PSR in dB) (PSR in dB) (PSR in dB) -34.46 -27.75 -21.78 -14.54 RNN -39.44 -33.23 -26.74 -20.68 RBF -47.77 -45.24 -38.21 -25.23 RRBF -52.64 -48.71 -43.41 -35.42 Table 3.9: Doppler shift performance Structures 13-bit Barker code 35-bit Barker code (PSR in dB) (PSR in dB) MLP -14.35 -28.34 RNN -30.93 -42.36 RBF -47.45 -46.42 RRBF -55.23 -56.34 74 Chapter 3 Development and Performance Evaluation of New and Efficient ANN Mismatch Filters for Sidelobe Reduction Doppler shift performance The influence of Doppler shift should be accounted for evaluating the detection performance for a moving target. The Doppler tolerance measures the Doppler sensitivity of the pulse compression technique. The Doppler sensitivity is caused by the shifting in phase of the individual elements of the code by the target Doppler. In extreme case the phase shift across the code will be 180o , the last subpulse in the received code is effectively inverted. For 13-bit Barker code at extreme case the input will change from “1 1 1 1 1 -1 -1 1 1 -1 1 -1 1” to “-1 1 1 1 1 -1 -1 1 1 -1 1 -1 1”. For 13-bit and 35-bit Barker codes the extreme case Doppler shift PSR values for different types of network are listed in Table 3.9. From this table it is observed that the MLP has very low Doppler tolerance and RRBF produces the best PSR value of -55.23 dB for 13-bit Barker code. 3.5 Conclusion In this chapter recurrent networks such as RNN and RRBF are proposed for radar pulse compression. The simulation results reveal that the performance of RRBF based pulse compression is much better than MLP, RNN and RBF based pulse compression techniques. The convergence rate of RRBF is higher than that of all other networks and it has low training error. The RRBF approach provides better PSR values in different adverse conditions such as noise and Doppler shift conditions. The range resolution ability of RRBF network is much superior than MLP, RNN and RBF networks. Although the algorithms are applied for 13-bit and 35-bit Barker codes, they can also be used for any other biphase codes. 75 Chapter 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows 4.1 Introduction The pulse compression techniques dealt in Chapter 3 are meant for biphase codes. These codes are easily generated and the correlators for these codes are very simple. But the compressed output of biphase codes are associated with the high time range sidelobes and these codes are more prone to Doppler shift. The application of a pulse compression technique depends on how efficiently it reduces the range sidelobes associated with the compressed waveforms. The number of Barker codes available are very less. Hence, these codes seriously suffer from security problem. Apart from biphase codes, polyphase codes and frequency modulated codes are also used in radar systems. PSL of frequency modulated pulse and polyphase codes are lower than that of the biphase codes. The frequency modulated and polyphase codes are more Doppler tolerant and have less range sidelobes compared to biphase codes. Different windows those are available in the literature are used for reducing the range sidelobes of the compressed output in case of LFM and polyphase codes. In most of the practical radar systems LFM waveform is extensively used because it is more Doppler tolerant than phase coded signals. The matched filter output of a point target for an arbitrary pulse is the ACF which forms a Fourier transform 76 Chapter 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows pair with the energy spectrum of the signal. For rectangular amplitude weighing, the energy spectrum of an LFM signal can be approximated as sin(x)/x or sinc(x) shaped ACF. Hence a compressed LFM signal at the receiver will produce a series of sidelobes surrounding the mainlobe and the first sidelobe occurs at a level of 13 dB below the peak of the mainlobe. Range sidelobes are inherent part of the pulse compression mechanism and they occur due to abrupt rise in the signal spectrum. The conventional method used to suppress these ambiguous sidelobes by modifying the rectangular shape of the chirp spectrum using amplitude weighing. In radar systems, weighing techniques in time or in the frequency domain are mostly employed to reduce these range sidelobes with broadening in the mainlobe. Time domain weighing is preferred to its frequency domain counterpart, as it produces lower peak sidelobe in compressed output [69–71]. Although weighing when used both on transmitter and receiver provides better results, weighing only on receiver is preferred. Weighing on transmitter leads to power loss hence the available transmit power cannot be fully utilized. In low T B product LFM waveforms the Fresnel ripples, which are responsible for producing range sidelobes, are reduced by modifying the chirp waveform before transmission. Amplitude tapering [69] and phase distortion [69, 72] are used to modify the chirp waveform to suppress the peak sidelobe as well as to increase the fall off rate of far sidelobes. Amplitude tapering reduces the far sidelobes effectively. But in most high power radars, the control of the pulse rise time is very difficult. So an appropriate phase distortion function is used in the LFM pulse for short rise time and high power radars. Shennawy et. al. [73] have used an external Hamming window as weighing function in frequency domain to suppress the range sidelobes from a T B product of 50 to 720. Using the weighing technique the dynamic range of the pulse compression system is increased. Hamming weighing has been used to suppress the range sidelobes for rectangular LFM pulses with T B product less than 170 [74] and it is observed from the results that Hamming weighing in time domain produces lower peak sidelobe as compared to Hamming weighing in frequency domain. If 77 Chapter 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows weighing at the receiver is used, the reflected waveform from the object no longer matched to the receiver filter which causes mismatch loss. In place of LFM, nonlinear LFM (NLFM) can be used for transmission which does not need any weighing at the receiver [75–77] to overcome the mismatch loss. If the objects are in motion in the environment then the waveforms reflected are Doppler shifted version of the transmitted signal and the matched filter output of these Doppler shifted signal produces very low PSR magnitude. If the T B product of the transmitted signal is low then the PSR magnitude becomes even worse. The NLFM signals are more affected by Doppler shift and are difficult to design than the LFM signals. Although LFM signal is popularly used in radar, the group of phase coded pulses is also an active research area for particular radar applications. Phase coded waveforms are more compatible for digital generation and compression [78]. However, these waveforms are affected more in the presence of Doppler shift as compared to the LFM signal. To get the Doppler shift advantage of LFM signal various polyphase codes are derived from LFM signal [79–81]. The codes such as Frank [79], P1 and P2 [80] are derived from step approximation to LFM waveform. These codes provide lower peak sidelobes than that offered by the best biphase codes [10] for a particular length. In [81] two more polyphase codes, P3 and P4 are discussed which are derived from the LFM signals. These codes are more Doppler tolerant as compared to P1 and P2 codes. Although polyphase codes have lower sidelobes in their ACF, it is required to further reduce the sidelobes for many radar applications. Many sidelobe reduction techniques for polyphase codes such as amplitude weighing [82,83] and the post compression sliding window techniques [84] are found in the literature . When Doppler shift is zero these techniques substantially reduce the sidelobes of the compressed pulse. Due to Doppler shift, objects with larger velocities experience detection range degradation. Grating lobes are appeared in the ACF of Frank and P1 code with increasing in Doppler shift [85]. Lee and Griffiths [86, 87] have proposed Woo filter for polyphase codes which provides optimum uniform sidelobe level and excellent Doppler shift performance. A modified 78 Chapter 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows P4 code which have better sidelobe and Doppler shift characteristics is presented in [88]. But the mainlobe of the compressed output for this code is broadened by a factor of two. To overcome the problem of mainlobe broadening, Lee [89] has proposed asymmetric weighing at the receiver. Two variants of Woo filter is proposed in [90] for P4 codes which have better resistance for Doppler shift as compared to Woo Filter. In this chapter convolutional windows are proposed to use as the weighing functions at the receiver filter to reduce the sidelobes of LFM and polyphase codes. To assess the performance of convolutional windows under Doppler shift condition, exhaustive simulation studies are carried out under different Doppler shift conditions and the results are compared with the conventional windows. 4.2 LFM and polyphase codes The LFM and polyphase codes that are used in the radar system are explained below. 4.2.1 LFM signal An LFM pulse having rectangular envelope mathematically described as B 2 |t| ≤ Tp /2 t s (t) = exp j2π f0 t + 2Tp (4.1) where f0 = center frequency, B = bandwidth and Tp = pulse duration of s(t). Applying Fourier transform the spectrum of s(t) is calculated as Z ∞ S (f ) = s(t)e−j2πf t dt (4.2) −∞ From (4.1) and (4.2) the spectrum is expressed [73] as r πTp Tp 2 [Z(u2 ) − Z(u1 )] e−j B (f −f0 ) S (f ) = 2B where complex Fresnel integral Z(u) is Z u Z u π π 2 x dx + j x2 dx sin Z(u) = cos 2 2 0 0 79 (4.3) (4.4) Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Amplitude Real part of chirp signal 0.5 0 −0.5 −5 −4 −3 −2 −1 0 Time in µ sec 1 2 3 4 5 2 3 4 5 Imaginary part of chirp signal Amplitude 1 0.5 0 −0.5 −5 −4 −3 −2 −1 0 Time in µ sec 1 Figure 4.1: Real and imaginary part of the chirp signal for T B = 50 The arguments u1 and u2 given by r u1 = −2(f − f0 ) Tp − 2B r Tp B 2 (4.5) r Tp Tp B + (4.6) u2 = −2(f − f0 ) 2B 2 Z(u) is a function of Fresnel integrals cosine C(u) and sine S(u), where Z u π C(u) = cos x2 dx (4.7) 2 0 Z u π (4.8) x2 dx sin S(u) = 2 0 The Fresnel ripple values defined in (4.7) and (4.8) are high at small arguments of u r and vice-versa. The real and imaginary part of the envelope of s(t) and corresponding spectrum for T B = 50 are depicted in Figures 4.1 and 4.2 respectively. From Figure 4.2 it is observed that the spectrum contains the ripples called as Fresnel ripples, which causes the sidelobes after compression. The output of the receiver matched filter or compression filter for T B = 50 is a pulse with Figure 4.3. 80 sin(x) x envelope as shown in Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 80 Amplitude 70 60 50 40 30 20 10 −10 −8 −6 −4 −2 0 2 4 6 8 10 Frequency in MHz Figure 4.2: Amplitude spectrum of chirp signal for T B = 50 Filter response in dB 0 −10 −20 −30 −40 −50 −1 −0.8 −0.6 −0.4 −0.2 0 Time delay × T 0.2 0.4 0.6 0.8 1 p Figure 4.3: Compressed envelope 4.2.2 Polyphase codes The complex envelope of phase coded pulse is expressed as [5] where rect t tb N t − (i − 1)tb 1 X ui rect u(t) = p tb Tp i=1 = 1 for |t| ≤ tb , 2 (4.9) ui = exp(jφi ), tb is the sub pulse width and N is the number of phases given as {φ1 , φ2 ..., φN }. Polyphase codes have harmonically related phases based on a certain fundamental phase increments. These codes have better Doppler tolerance and sidelobe performance than biphase codes. These codes are discrete time sequences having constant magnitude with a variable phase φi . Polyphase codes have more than two elements or phase values. Increasing the number of elements in the sequence enables construction of longer sequences having greater range resolution with a larger compression ratio. But the trade off is that a more complex matched filter as 81 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 compared to the biphase codes. These codes are derived by detecting a frequency modulated pulse compression waveform with either a local oscillator at the band edge of the waveform (single sideband detection) or at band center (double sideband detection) and the resultant in phase I and quadrature phase Q data are sampled at Nyquist rate. Polyphase codes are derived from step approximation to linear frequency modulation, such as Frank, P1 and P2 or from linear frequency modulation like P3 and P4 codes. (a)Frank Code The Frank code is derived from step approximation to a linear frequency modulated waveform having N frequency steps and N samples per frequency [79]. So Frank code having N frequency steps have length Nc = N 2 . The first N samples of the code have zero phase. The second N samples start with zero phase and increase with a phase value of 2π/N from sample to sample. The phase of ith sample of j th frequency step is given by φi,j = 2π (i − 1)(j − 1) N (4.10) where i = 1, 2, 3...N and j = 1, 2, 3...N . The Frank code in N × N 0 0 0 ... 0 matrix form is given as 0 0 ... 1 2 ... 2 4 ... ... ... N −1 2(N − 1) ... ... 0 2(N − 1) ... 2 (N − 1) N −1 (4.11) where the number in matrix represent the multiplying coefficient with the phase angle 2π/N . The Frank polyphase code is formed by concatenating the rows of the Frank matrix and multiplying by fundamental phase increment 2π/N . The 16-element Frank code is given by 82 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 200 70 80 90 100 Time delay (a) Matched filter response 6 Phase in radian 5 4 3 2 1 0 0 10 20 30 40 50 60 Sample number (b) phase Figure 4.4: Matched filter output and phase values of 100 element Frank code 0 0 0 0 0 π 2 π Taking modulo 2π gives 0 0 0 0 0 π2 π 3π 2 0 π 2π 3π 2 0 π 0 π 3π 0 0 3π 2 3π 2 9π 2 3π π π 2 The matched filter response of 100 element Frank code and its phase values are given in Figure 4.4. From the figure it is clear that the peak sidelobe occurs below 30 dB of the main lobe. (b)P1 Code These codes are also derived from step approximation to linear frequency modulation. In case of single sideband detection Frank code is generated and in case of double sideband detection P1 code is generated. P1 code also consists of N × N elements and the phase of the ith element of j th group is given by φi,j = − π [N − (2j − 1)][(j − 1)N + (i − 1)] N (4.12) where i = 1, 2, 3...N and j = 1, 2, 3...N . The matched filter response and phase values of 100 element P1 code is depicted in 83 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 200 70 80 90 100 Time delay (a) Matched filter response 7 Phase in radian 6 5 4 3 2 1 0 0 10 20 30 40 50 60 Sample number (b) phase Figure 4.5: Matched filter output and phase values of 100 element P1 code Figure 4.5. From Figures 4.4(a) and 4.5(a) it is observed that the ACFs of Frank and P1 codes are same. But the difference between P1 code and the Frank code is that P1 code has the highest phase increments from sample to sample at the two ends of the code but the Frank code has the highest phase increments from sample to sample in the center of the code. So when the codes are passed through band pass amplifier of a radar receiver, the P1 code is attenuated mostly at the two ends of the waveform while the Frank code is attenuated most heavily in the center of the waveform. (c)P2 Code In P2 code the starting phases are different from P1 code but the phase increments within each phase group is same as that of P1 codes. The phases of P2 codes of length N 2 is given by φi,j = π π [N − 1] − (i − j)[N + 1 − 2j] 2N N 84 (4.13) Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 where i = 1, 2, 3...N and j = 1, 2, 3...N . The matched filter response and phase value of 100 element P2 code is shown in Figure 4.6. It has same matched filter output as that of Frank and P1 codes. But, the P1 and P2 codes are more precompression bandwidth limit tolerant than the Frank code. Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 200 70 80 90 100 Time delay (a) Matched filter response 7 Phase in radian 6 5 4 3 2 1 0 0 10 20 30 40 50 60 Sample number (b) phase Figure 4.6: Matched filter output and phase values of 100 element P2 code (d)P3 Codes P3 codes [81] are derived from the phase samples LFM signal. These codes are obtained by converting a LFM waveform to base band using a local oscillator on one end of the frequency sweep (single sideband detection) and sampling the I and Q video at Nyquist rate. Let the waveform have pulse duration Tp and the instantaneous frequency is f (t) = f0 + kt 85 (4.14) Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 where k is a constant and defined as k= B Tp (4.15) B is the bandwidth of the signal. So the signal will support a compressed pulse length of tc = 1/B (4.16) The pulse compression ratio is obtained as ρ = Tp /tc = BTp (4.17) If the first samples of I and Q are taken at the leading edge of the waveform, the phases of P3 codes are given by Z (i−1)tc φi = 2π [(f0 + kt) − f0 ]dt = πk(i − 1)2 t2c (4.18) 0 From (4.17) and (4.18) φi = π(i − 1)2 /BTp = π(i − 1)2 /ρ (4.19) The matched filter output and phase values of 100 element P3 code are shown in Figure 4.7. From Figure 4.7(a) it is observed that the peak side lobe occurs below 27 dB from the main peak lobe which is 3 dB inferior from the Frank code. (e)P4 Codes The P4 code is derived from the same waveform as P3 codes but the local oscillator frequency is set at f0 + kTp /2. So the phase of P4 codes are φi = 2π Z 0 (i−1)tc [(f0 + kt) − (f0 + kTp /2)]dt = [π(i − 1)2 /ρ] − π(i − 1) (4.20) The matched filter output and phase values of 100 element P4 code is shown in Figure 4.8. From Figures 4.7(a) and 4.8(a) it is observed that both P3 and P4 codes have the same matched filter output. But P4 code is more tolerant to precompression bandwidth limitation as compared to P3 code. 86 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 70 80 90 Time delay (a) Matched filter response 7 6 Phase angle 5 4 3 2 1 0 0 10 20 30 40 50 60 100 Sample number (b) phase Figure 4.7: Matched filter output and phase values of 100 element P3 code Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 200 70 80 90 100 Time delay (a) Matched filter response 7 Phase angle 6 5 4 3 2 1 0 0 10 20 30 40 50 60 Sample Number (b) phase Figure 4.8: Matched filter output and phase values of 100 element P4 code 87 Chapter 4 4.3 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Problem formulation The LFM signal and polyphase codes are used for transmission and the reflected waveforms are passed through weighted matched filter. If the target is in motion the reflected waveform is Doppler shifted version of the transmitted waveform. The Doppler shifted waveform is passed through the weighted receiver matched filter and PSRs are calculated for different Doppler shift conditions. 4.3.1 For LFM signal The matched filter is given by H(f ) = F [s(t)w(t]∗ (4.21) where w(t) is the window function. The matched filter output is obtained as g(t) = F −1 [S(f )H(f ] (4.22) where S(f ) = F [s(t)]. The Doppler shifted version of the transmitted signal s(t) is represented as B 2 |t| ≤ Tp /2 (4.23) t sd (t) = exp j2π (f0 + fd )t + 2Tp where fd is the Doppler shift. The Doppler shifted signal is passed through the weighted matched filter and the PSR values under different Doppler shift are obtained. To achieve higher magnitude of PSR the transmitted signal is modified using amplitude tapering and phase predistortion function as explained below (I) Amplitude tapering: The Fresnel ripples can be reduced by adding cosine taper of length αTp to the LFM pulse. The amplitude tapered transmitted signal is represented as n s1 (t) = gT (t) exp j2π f0 t + B 2 t 2Tp 88 o |t| ≤ (0.5 + α) Tp (4.24) Chapter 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows where 1 o n p /2 gT (t) = 0.5 1 + cos π |t|−T αTp 0 |t| ≤ Tp/2 Tp/ ≤ |t| ≤ (0.5 + α) Tp 2 elsewhere (4.25) and α is a parameter. (II) Phase distortion: In high power pulsed radars the pulse rise time cannot be controlled easily. So an alternate approach which uses an appropriate phase distortion of the transmitted LFM signal is used for fast rise time transmitters. LFM waveform having cubic phase distortion is given by h n s2 (t) = exp j2π f0 t + B 2 t 2Tp oi + φ(t) |t| ≤ Tp/2 + ∆T (4.26) where ∆B Tp/ )3 2 (−t − 2 3∆T ∆B φ (t) = (t − Tp/2)3 3∆T 2 0 − Tp /2 − ∆T ≤ t < −Tp /2 Tp /2 ≤ t < Tp /2 + ∆T (4.27) elsewhere and ∆B and ∆T are the parameters. 4.3.2 For polyphase codes These codes have discrete values having different phases. The matched filter is designed according to the transmitted polyphase code. The filter is multiplied with the window functions to achieve lower sidelobes. The Doppler shifted signal is modeled by multiplying ej2πifd /B to the transmitted signal, where i = 1, 2, 3...N and N is the code length. The PSR values are calculated using conventional and convolutional windows as weighing function at the receiver end under various Doppler shift conditions. 89 Chapter 4 4.4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Windows used for sidelobe suppression Windows are time domain weighing functions that are used to reduce Gibbs oscillations caused by the truncation of a Fourier series. They are employed in a variety of traditional applications including power spectral estimation, beam forming and digital filter design [91]. Many windows appeared in the literature are not optimal. So the use of a particular window depends upon the application. A number of windows and algorithms are formulated to find an optimal window for a given application [92–96]. The typical windows that are used in the signal processing techniques are based on cosine series such as Hamming, Hanning and Blackman [92]. Although many windows have already been introduced in the literature, research is going on to propose new windows or to parameterize the known windows [97–100]. Window functions are generally categorized as fixed or adjustable. Fixed windows have window length as the parameter which alters the mainlobe width. Adjustable windows have two parameters, namely the window length and a parameter that alters the relative sidelobe amplitude. The best known parametric windows in the literature are Dolph-Chebysev [101] and Kaiser [102] windows. By varying the two adjusting parameters of the Kaiser window it can control the mainlobe width and ripple ratio of the spectrum. Polynomial windows having low computational complexity is presented in [103]. The frequency response of these windows can easily be changed by modifying their coefficients in the time domain. Avci and Nacaroglu [104] have proposed a new class of cosine hyperbolic windows having low computational complexity due to power series expansion in its time domain representation. Convolutional windows are derived by convolving the window with itself. Reljin et. al. [105] have discussed a class of windows that are generated by the time convolution of classical windows to obtain both flat top and high sidelobe attenuation. These windows are suitable for harmonic amplitude evaluation in nonsynchronous sampling case. The convolutional windows from second to eighth order for rectangular window are derived in [106]. These windows are applied for 90 Chapter 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows high accuracy harmonic analysis and parameter estimation of periodic signals. Phase difference algorithm based on Nuttal self-convolutional window is used to eliminate the measurement errors of dielectric loss factor [107]. Dielectric loss factor is caused by non-synchronised sampling and non-integral periodic truncation conditions. A self convolution Hanning window used for complex signal harmonics parameter estimation has been presented in [108]. The convolutional window based phase correction algorithm suppresses the impact of fundamental frequency fluctuation and white noise on harmonic estimation. In this chapter convolutional windows are used for weighing purpose to reduce the range sidelobes that are present in output of pulse compression filter. The windows employed for the analysis are (i) Hamming window: w(n) = 0.54 − 0.46 cos 2πn N −1 (4.28) where n = 0, 1, ........(N − 1) (ii) Hanning window: w(n) = 0.5 − 0.5 cos 2πn N −1 (4.29) (iii) Kaiser window: w(n) = q 2n 2 I0 β 1 − ( N −1 ) I0 (β) (4.30) where I0 is the zeroth order modified Bessel function of the first kind and β is a parameter determines the shape of the window. (iv) Chebysev window: The Dolph-Chebyshev window is in the frequency domain is represented as )]] [β1 cos( πk N −1 cosh[N cosh (β1 )] k cos[N cos W (k) = (−1) −1 (4.31) where β1 = cosh[ N1 cosh−1 (10α1 )] and α1 determines the level of the sidelobe attenuation. The sidelobe level σ(dB) = 20α1 . The Dolph-Chebyshev window 91 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 0 Hamming Conv. Hamming −10 20*log10(amplitude) −20 −30 −40 −50 −60 −70 −80 50 100 150 200 250 Sample number (a) Hamming and Convolutional Hamming window 0 Hamming Conv. Hamming −10 20*log10(amplitude) −20 −30 −40 −50 −60 −70 −80 130 140 150 160 170 180 190 200 210 220 230 Sample number (b) Zoomed version Figure 4.9: Frequency response curve is obtained by taking the inverse DFT of W (k) and scaling the result to have a peak value of 1. The convolutional windows are obtained by convolving a particular window with itself. An N point convolutional window is obtained by convolving two N/2 point windows. After convolution of two N/2 point windows the number of samples is N − 1. So a zero is padded to the convolution result to make the length of the window N and the maximum value is normalized to 1. Frequency responses of Hamming and convolutional Hamming windows are presented in Figure 4.9. From this figure, it is observed that the sidelobes of convolutional Hamming window are lower at the cost of wider mainlobe. 4.5 Simulation results The windows available in the MATLAB library are used for the simulation purpose. 92 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Filter response in dB 0 −10 −20 −30 −40 −50 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 0.4 Time delay × Tp 0.6 0.8 1 Figure 4.10: Matched filter output with Hamming weighing at the receiver 4.5.1 Analysis for LFM signals LFM signal with low T B products is associated with Fresnel ripples. The compressed output of an LFM signal having Hamming weighing at the receiver for T B = 50 is shown in Figure 4.10. The peak sidelobe is approximately 37 dB lower than the main peak. The outputs of matched filter for different Doppler shifts using Hamming and convolutional Hamming window are depicted in Figure 4.11. It is observed that at zero Doppler shift Hamming window yields better PSR value as compared to convolutional window. As the Doppler shift increases sidelobe level affected very less in case of convolutional Hamming window as compared to Hamming window. The PSR values under different Doppler shifts using various windows are presented in Table 4.1. It is observed from the table that at lower Doppler shift the conventional windows yield better PSR values as compared to corresponding convolutional windows. On the other hand for higher Doppler shifts the convolutional windows provide better PSR values than conventional ones. As an illustration for fd B = 0.01 the PSR for Hamming window is -36.2 dB and that of convolutional Hamming window is -34.46. But, for fd B = 0.2 the PSR for Hamming window is -22.2 dB and that of convolutional Hamming window is -29 dB. From Figure 4.11 it is obvious that the mainlobe width in case of convolutional window is wider and the sidelobes near |t| = Tp 2 93 region are not diminished by weighing Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 0 Filter response in dB Hamming Conv. Hamming −10 −20 −30 −40 −50 −1 −0.8 −0.6 −0.4 −0.2 0 Time delay×Tp (a) fd B 0.2 0.4 0.6 0.8 1 =0 0 Filter response in dB Hamming Conv. Hamming −10 −20 −30 −40 −50 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 Time delay×Tp (b) fd B 0.4 0.6 0.8 1 = 0.05 0 Filter response in dB Hamming Conv. Hamming −10 −20 −30 −40 −50 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 0.4 0.6 Time delay×T p (c) fd B = 0.1 Figure 4.11: Effect on sidelobes due to Doppler shift 94 0.8 1 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Table 4.1: Comparison of PSR for different Doppler shift for T B = 50 Doppler Shift( fBd ) PSR using Hamming window in dB PSR using convolutional Hamming window in dB 0.01 0.05 0.1 0.15 0.2 Doppler Shift( fBd ) -36.2 -32.8 -28.6 -25.2 -22.2 PSR using Hanning window in dB -34.46 -33.67 -32.5 -31 -29 PSR using convolutional Hanning window in dB 0.01 0.05 0.1 0.15 0.2 Doppler Shift ( fBd ) -31.68 -30.32 -27.62 -24.47 -22 PSR using Kaiser window in dB (β = 6) -33.67 -33 -31.79 -30.44 -28.7 PSR using convolutional Kaiser window in dB 0.01 0.05 0.1 0.15 0.2 -36.6 -33 -28.7 -25.2 -22.3 -34.2 -33.4 -32.26 -30.86 -29 Doppler Shift( fBd ) 0.01 0.05 0.1 0.15 0.2 PSR using Chebysev PSR using convolutional window in dB (σ = 50) Chebysev window in dB -36 -34.89 -29.89 -26.86 -23.86 -34.3 -33.5 -32.5 -31 -29 technique. Hence to reduce the sidelobes at Tp 2 region as well as peak sidelobe, amplitude tapering and phase predistortion are used . Amplitude tapering with α = 0.1 is used to modify the transmitter signal. In simulation study the amplitude tapers are not used in compression filter. The filter responses for amplitude tapering using Hamming and convolutional Hamming 95 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 windows are depicted in Figure 4.12. From Figure 4.12(a), it is clear that amplitude tapering reduces drastically the sidelobes around |t| = Tp /2 region but the reduction of near in sidelobes is very less. From Figure 4.11, it is evident that the compressed output of convolutional Hamming window produces peak sidelobe around |t| = Tp /2 region. So to get very low overall sidelobe the transmitted signal is amplitude tapered and the receiver is weighed with convolutional Hamming window. The output of pulse compression filter using convolutional Hamming window with amplitude tapered transmitted signal is presented in Figure 4.12(b). The PSR values using Hamming, Kaiser, convolutional Hamming and convolutional Kaiser windows for T B = 50 and T B = 100 for different Doppler shifts are listed in Table 4.2. From the table, it is observed that for higher T B product the PSR value is better and amplitude tapering with convolutional window provides better PSR value compared to that of conventional windows. Table 4.2: PSR using amplitude tapering Doppler Shift fd B 0 0.02 0.04 0.06 0.08 0.1 0.12 0.14 0.16 0.18 0.2 Hamming window PSR in dB TB=50 TB=100 -42.2 -43 -41.2 -42.7 -39 -41.7 -36.6 -39.3 -34.5 -36.8 -32.6 -34.7 -30.8 -32.7 -29.2 -31 -27.8 -29.4 -26.4 -28 -25.2 -26.5 Convolutional Hamming window PSR in dB TB=50 TB=100 -60 -78.7 -58 -75 -55.5 -71.4 -53 -67.3 -50.2 -63.5 -47.7 -60 -45.3 -55.7 -43 -52 -40.5 -48 -38.1 -44.4 -35.7 -41.2 Kaiser window (β = 6) Convolutional Kaiser window PSR in dB TB=50 TB=100 -43.4 -44.5 -41.8 -43.8 -40 -42.5 -38.25 -40.8 -36.6 -39 -35 -37.3 -33.3 -35.7 -31.2 -33.4 -29.3 -31.2 -27.6 -29.8 -26 -28 PSR in dB TB=50 TB=100 -61.2 -79.6 -59.4 -78.2 -56.6 -75.5 -54 -71.38 -51.2 -66.8 -48.6 -62.7 -46.1 -58.5 -43.75 -54.5 -41.3 -50.7 -39 -46.8 -36.5 -43.3 In case of cubic phase distorted transmitted signal the parameter values used are ∆B = 0.75B and ∆T = 1/B. The filter responses for cubic phase distortion using Hamming and convolutional Hamming windows are depicted in Figure 4.13. From 96 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Filter response in dB 0 −20 −40 −60 −80 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 Time delay × Tp 0.4 0.6 0.8 1 (a) Compressed waveform for amplitude tapering with Hamming window Filter response in dB 0 −20 −40 −60 −80 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 Time delay × Tp 0.4 0.6 0.8 1 (b) Compressed waveform for amplitude tapering with convolutional Hamming window Figure 4.12: Compressed waveforms for T B = 50 for amplitude tapering (α = 0.1) Filter response in dB 0 −20 −40 −60 −80 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 0.4 0.6 0.8 1 Time delay× T p (a) Compressed waveform for cubic phase distortion with Hamming window Filter response in dB 0 −20 −40 −60 −80 −1 −0.8 −0.6 −0.4 −0.2 0 0.2 0.4 0.6 0.8 1 Time delay× Tp (b) Compressed waveform for cubic phase distortion with convolutional Hamming window Figure 4.13: Compressed waveforms for T B = 50 for cubic phase distortion (∆B = 0.75B and ∆T = B1 ) 97 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Figure 4.13(a) it is evident that the compressed output of phase distorted signal also reduces the sidelobe level around |t| = Tp /2 region. But in this case the fall off rate of far sidelobes is lesser as compared to amplitude tapering. The output of pulse compression filter using convolutional Hamming window with cubic phase distorted transmitted signal is presented in Figure 4.13(b). The PSR values using Hamming, Kaiser, convolutional Hamming and convolutional Kaiser windows for T B = 50 and T B = 100 for different Doppler shifts are listed in Table 4.3. The table illustrates that convolutional windows provide better PSR values as compared to that of conventional windows. For a particular window, the PSR values for T B = 100 is better than that of T B = 50. Table 4.3: PSR using cubic phase distortion Doppler Shift fd B 0 0.02 0.04 0.06 0.08 0.1 0.12 0.14 0.16 0.18 0.2 4.5.2 Hamming window PSR in dB TB=50 TB=100 -39.8 -41.8 -37.6 -39.2 -35.5 -36.8 -33.6 -34.7 -31.8 -32.8 -30.2 -31 -28.8 -29.4 -27.4 -27.9 -26.2 -26.6 -25 -25.3 -23.8 -24.4 Convolutional Hamming window PSR in dB TB=50 TB=100 -49.7 -55.3 -48.4 -53.6 -46.7 -52 -45 -50.3 -43.4 -48.8 -41.8 -47.2 -40.2 -45.6 -38.6 -43.9 -36.8 -41.2 -34.7 -38.1 -32.4 -35.3 Kaiser window (β = 6) Convolutional Kaiser window PSR in dB TB=50 TB=100 -41.2 -42.2 -39.7 -41.1 -37.8 -39.4 -35.5 -37.3 -33.9 -35.2 -32 -33.1 -30.3 -31.3 -28.7 -29.5 -27.2 -28 -26 -26.4 -24.6 -25 PSR in dB TB=50 TB=100 -61 -79.5 -59.3 -78 -56.5 -75 -53.8 -71 -51.1 -66.4 -48.6 -62.3 -46.1 -58.1 -43.7 -54.1 -41.3 -50.3 -39 -46.5 -36.4 -43 Analysis for polyphase codes Polyphase codes derived from the step approximation to LFM signal (Frank, P1 and P2 ) do not provide satisfactory results using weighing technique. The output of the Hamming weighted matched filter for 100 element Frank code is depicted in Figure 4.14 which shows that the peak sidelobe level 25 dB below the mainlobe peak. From Figures 4.4(a) and 4.14 it is clear that with Hamming weighing the peak sidelobe 98 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 level is increased. So windows are not used for these codes for sidelobe suppression. From Figures 4.7 and 4.8 it is evident that the matched filter output for P3 Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 200 Time delay Figure 4.14: Matched filter output for 100 element Frank code using Hamming window and P4 codes are identical. The matched filter output of P3 and P4 under different Doppler shift is depicted in Figures 4.15 and 4.16 respectively. These figures show that under Doppler shift both P3 and P4 codes provide same performance. So P4 code of length 100 is used for further simulation study and the results are valid also for P3 codes. Figure 4.17 illustrates the output of matched filter under Doppler shift when the receiver filter weighted with Hamming and convolutional Hamming windows. It is observed that at higher Doppler shifts the convolutional windows provide improved results as compared to that of conventional windows. The PSR values under different Doppler shifts using different windows are presented in Table 4.4. It is observed that for lower values of Doppler shift the PSR values for classical windows are better than that of convolutional windows. But as the Doppler shift increases the PSR magnitudes for classical windows drop rapidly as compared to convolutional windows. 99 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 120 140 160 180 Time delay (a) fd B = 0.05 Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 200 Time delay (b) fd B = 0.1 Figure 4.15: Matched filter output of P3 code under different Doppler shift Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 120 140 160 180 Time delay (a) fd B = 0.05 Filter response in dB 0 −10 −20 −30 −40 −50 0 20 40 60 80 100 200 Sample Number (b) fd B = 0.1 Figure 4.16: Matched filter output of P4 code under different Doppler shift 100 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Filter response in dB 0 Hamming Conv. Hamming −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 Time delay (a) fd B =0 Filter response in dB 0 Hamming Conv. Hamming −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 180 Time delay (b) fd B = 0.05 Filter response in dB 0 Hamming Conv. Hamming −10 −20 −30 −40 −50 0 20 40 60 80 100 120 140 160 Time delay (c) fd B = 0.1 Figure 4.17: Effect on sidelobes due to Doppler shift 101 180 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows Chapter 4 Table 4.4: Comparison of PSR for different Doppler shift Doppler Shift( fBd ) PSR using Hamming window in dB PSR using convolutional Hamming window in dB 0.01 0.05 0.1 0.15 0.2 Doppler Shift( fBd ) -40 -37.3 -32 -27 -22.18 PSR using Hanning window in dB -37.17 -37 -36.5 -35.7 -33 PSR using convolutional Hanning window in dB 0.01 0.05 0.1 0.15 0.2 Doppler Shift ( fBd ) -40 -39.6 -37.1 -30 -25 PSR using Kaiser window in dB (β = 6) -36.3 -36.1 -35.7 -35 -33.6 PSR using convolutional Kaiser window in dB 0.01 0.05 0.1 0.15 0.2 Doppler Shift( fBd ) -40.1 -36.8 -37.9 -36.7 -33.8 -36.2 -28.2 -35.4 -23 -34 PSR using Chebysev PSR using convolutional window in dB (σ = 50) Chebysev window in dB 0.01 0.05 0.1 0.15 0.2 4.6 -39 -37.9 -33.3 -27.8 -22.7 -37 -36.8 -36.3 -35.5 -33.8 Conclusion In this chapter the ability of convolutional windows to suppress the sidelobes are analyzed and the results are compared with that of conventional windows. Although the magnitude of PSR at lower Doppler shift in Table 4.1 and 4.4 are better in case of 102 Chapter 4 Effective Sidelobe Suppression of LFM and Polyphase Codes Using Convolutional Windows conventional windows, the convolutional windows provide better PSR value at higher Doppler shift. In case of LFM signal, to decrease the sidelobes around |t| = Tp 2 region the transmitted signal is modified using amplitude tapering or phase distortion. It is further demonstrated that the PSR values of amplitude tapered or phase distorted transmitted signal with convolutional windows are better than that of conventional windows at all Doppler shift conditions. However, the mainlobe width achieved using convolutional windows is wider than that of conventional windows. 103 Chapter 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques 5.1 Introduction In high range resolution radar, signals having wide bandwidth are used to get narrow mainlobe width. Generation of such type of wideband waveforms increases the overall cost and complexity of the system. The conventional narrowband hardware used in the radar system may not sustain instantaneous wide bandwidth. To overcome such limitation the wide bandwidth signal is split into a set of narrowband signals which are transmitted and received separately. The effect of wideband signal is obtained by coherently combining the narrowband signals. Such type of narrowband signals together is called as ‘synthetic wideband waveform’ [109] or ‘stepped frequency waveform’ or ‘frequency jumped train’. Generally a pulse train consists of N pulses each of duration Tp and pulse repetition time Tr . Each pulse has a bandwidth B and center frequency step between the pulses is ∆f . The amplitude and frequency of a stepped frequency LFM pulse train is shown in Figure 5.1. In the proposed work the values of Tp , B and ∆f are assumed to remain constant throughout the pulse train and satisfy the condition 104 Chapter 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques B > ∆f > 0. One of the advantages of this type of signal is that the interval between pulses is utilized to adjust the center frequency of other narrowband components of the radar system. But matched filter output of such signals suffer from grating lobes for cases when Tp ∆f > 1 due to constant frequency step ∆f . These grating lobes reduce the range resolution capability of the signal and hence these are undesirable. Figure 5.1: Stepped frequency LFM pulse train Different techniques for acceptable suppression or complete rejection of grating lobes are dealt in [110–115]. In [110, 111] grating lobes are reduced by varying the pulse width of the pulse train which destroys the periodicity of the waveform. An approach to generate a nonlinear synthetic wideband waveform by distributing the energy nonuniformly over the desired bandwidth is described in [112]. It offers improved performance in terms of lower range sidelobes, higher range resolution and/or reduced grating lobes. Levanon and Mozeson [113] have proposed an analytical technique to establish the relation between parameters of stepped frequency LFM pulse train such that the first two grating lobes are nullified. They have also shown in some cases that nullifying the first two grating lobes leads to removal of all other grating lobes. To establish the required relation between the parameters Tp , B and ∆f using this approach for more than two grating lobes is too difficult. In this chapter PSO based technique is suggested which aims to eliminate 105 Chapter 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques all the grating lobes by judiciously choosing appropriate values of Tp , B and ∆f . The method presented in [113] does not suppress the range sidelobes that occur at the output of the matched filter of the receiver. In [114, 115] the LFM pulses are replaced by nonlinear LFM pulses that suppress the range sidelobes near the mainlobe along with grating lobes. However, the nonlinear LFM signals are not Doppler tolerant. The LFM pulse can be easily generated and more Doppler tolerant than NLFM pulse. Therefore LFM waveforms are widely used in pulse radar systems. But the techniques used in [113–115] to suppress the grating lobes of LFM pulse train ignore the mainlobe width and PSL. The waveform having wide mainlobe width in its ACF has low range resolution capability and the waveform that yields high peak sidelobe in its ACF may hide the small targets or cause false target detection. Hence there is a need to develop an efficient method to determine the parameters of the stepped frequency LFM pulse train by considering grating lobes, mainlobe width and PSL. Keeping this fact in view, in this chapter a new optimization is proposed approach using NSGA-II algorithm to achieve reduced grating lobes, lower sidelobes and narrow mainlobe width. 5.2 LFM pulse train The envelope of a constant frequency or unmodulated pulse of duration Tp is given by 1 u(t) = p rect Tp t Tp (5.1) Frequency modulation is applied to the above constant frequency pulse to get an LFM signal and its complex envelope is represented as t 1 exp(jπkt2 ) u1 (t) = p rect Tp Tp (5.2) where k is the frequency slope of the LFM signal and is defined as k=± 106 B1 Tp (5.3) Chapter 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques B1 is the bandwidth of the single pulse. “+” and “-” signs stands for positive frequency slope and negative frequency slope respectively. Instantaneous frequency of the LFM signal is obtained as f (t) = 1 d(πkt2 ) = kt 2π dt (5.4) A uniform pulse train having N number of LFM pulses separated by Tr ≥ 2Tp is expressed as The multiplication factor N −1 1 X u1 (t − nTr ) uN (t) = √ N n=0 √1 N (5.5) is included in (5.5) to maintain unit energy. Further a slope ks is added to the entire LFM pulse train and the complex envelope is represented as N −1 X 1 2 u1 (t − nTr ) us (t) = uN (t)exp(jπks t ) = √ exp(jπks t ) N n=0 2 (5.6) where ks = ± ∆f Tr ∆f > 0 (5.7) “+” and “-” signs correspond to positive and negative frequency step respectively. In this work “+” sign of k and ks is used, but the results equally hold good for “-” sign also. So the final bandwidth of each pulse in the LFM pulse train is B = (k + ks )Tp (5.8) The total bandwidth of the LFM pulse train is B + (N − 1)∆f . The ACF of the signal us (t) is obtained [5] as |τ | sin(N πτ ∆f ) |τ | sinc Bτ 1 − |R(τ )| = 1 − Tp Tp N sin(πτ ∆f ) (5.9) In (5.9) the expression of R(τ ) consists of product of two terms out of which the first term is the ACF of a single LFM pulse and is given by |τ | |τ | sinc Bτ 1 − |R1 (τ )| = 1 − Tp Tp 107 (5.10) Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 and the second term sin(N πτ ∆f ) |R2 (τ )| = N sin(πτ ∆f ) produces the grating lobes at τg = g ∆f (5.11) where g = 1, 2, 3.... ⌊Tp ∆f ⌋. These grating lobes appear in the form of high spikes and reduce the range resolution potential of the waveform. Nullifying or suppressing these grating lobes essentially depends upon the occurrence of nulls or minima of |R1 (τ )| at τg . In [113] an analysis is provided that sets up simple relations between the pulse time duration Tp , its bandwidth B and frequency step ∆f to nullify first two grating lobes of an LFM pulse train. However, in some cases nullifying two grating lobes also removes all grating lobes. Equation (5.9) can be written as τ sin(N π∆f Tp Tτp ) τ τ τ = 1 − sinc Tp B R 1 − , Tp N sin(πTp ∆f Tτp ) Tp Tp Tp τ ≤1 Tp (5.12) From (5.9), (5.10), (5.11) and (5.12) it is clear that |R(τ )|, |R1 (τ )| and |R2 (τ )| are functions of Tp ∆f and Tp B only for a given value of N . Figure 5.2 shows the plots of |R1 (τ )|, |R2 (τ )| and ACF for Tp ∆f = 3, Tp B = 4.5 and N = 8. It is observed that all the grating lobes are completely removed. For comparison purpose the ACF obtained with fixed frequency pulse train is shown in Figure 5.3 in which the grating lobes are prominent. The nullification of first two grating lobes always does not guarantee that all other grating lobes will be nullified or suppressed because the nulls of |R1 (τ )| do not occur periodically while the peaks of |R2 (τ )| occur with a period 1 . ∆f The mainlobe width depends on the first overall null of the expressions |R1 (τ )| and |R2 (τ )|. The first null of |R2 (τ )| occurs at |R1 (τ )| occurs at 1 Tp B 1 N Tp ∆f and the first null of approximately if Tp B >> 1. So the location of first null of ACF is given by τ1stnull = min Tp 1 1 , Tp B N Tp ∆f (5.13) N ∆f should be always greater than B in order to get a meaningful increase in bandwidth. So delay resolution is principally determined by |R2 (τ )| which is equal to 1 . N Tp ∆f 108 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 | R1(τ) |, | R2(τ) | 1 0.8 0.6 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.6 0.7 0.8 0.9 1 0 ACF in dB −10 −20 −30 −40 −50 −60 0.5 τ/ T p Figure 5.2: Stepped frequency LFM pulse for Tp ∆f = 3, Tp B = 4.5 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) | R1(τ) |,| R2(τ) | 1 0.8 0.6 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ / Tp Figure 5.3: Stepped frequency LFM pulse for Tp ∆f = 3, Tp B = 0 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 109 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 5.3 Problem formulation The grating lobes reduce range resolution ability of the pulse train. So it is required to suppress or nullify these grating lobes. In Section 5.3.1 a problem is formulated in which the PSO is used to determine the parameters of stepped frequency pulse train to nullify the grating lobes. The problem of suppression of grating lobes, minimization of mainlobe width and peak sidelobe level has been formulated in two different ways in a multiobjective framework which are presented in Sections 5.3.2 and 5.3.3. 5.3.1 Problem formulation -1 The function defined in (5.10) must be minimum or zero at τ = τg , so that the grating lobes would be suppressed or nullified. The fitness function which is to be minimized using PSO is defined as X |τ | |τ | g g 1− sinc Bτ 1 − f1 = T T p p g (5.14) subject to N ∆f > B. By choosing suitable values for Tp B and Tp ∆f the grating lobes as well as sidelobes in ACF can be suppressed. The value of Tp B is chosen such that Tp B = (c + 1)Tp ∆f (where c is a positive number) to ensure B > ∆f , so that there will be some frequency overlap between the pulses in spite of the frequency steps. If f1 = 0 then each term in the summation is zero which results in complete elimination of grating lobes otherwise the grating lobes are suppressed to a minimum level. PSO is used to find out the required values of Tp ∆f and c so that f1 is minimized. 5.3.2 Problem formulation -2 The peak sidelobe should be as low as possible compared to the mainlobe so that the target will be easily identified. NSGA-II algorithm is used to choose the values of Tp ∆f and c to achieve reduced grating lobes and minimum peak sidelobe 110 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 level simultaneously. The two objective functions which are to be simultaneously optimized are expressed as follows: Minimize f2 = max|R1 (τg )| where g = 1, 2, 3..... ⌊Tp ∆f ⌋ Minimize f3 = P SR in dB subject to N ∆f > B 5.3.3 Problem formulation -3 The range resolution of stepped frequency LFM pulse train depends upon the mainlobe width of ACF and is given by 1 N Tp ∆f for N Tp ∆f > Tp B. In the literature generally weighing technique is used to suppress the sidelobes of an LFM pulse. The weighing technique adds more emphasis on the center frequencies as compared to the end frequencies. As a result the sidelobes are suppressed and mainlobe is widened, which reduces the range resolution capability of the LFM signal. This effect is also applicable for stepped frequency LFM pulse train as the condition B > ∆f > 0 is assumed. The values of Tp ∆f and c are chosen by using a multiobjective algorithm in such a way that the mainlobe width is lowered (for high range resolution) and the sidelobes are suppressed. The effect of grating lobes is reduced by putting a constraint so that the grating lobes are below a threshold level i.e. |R1 (τg )| < ǫ. The fitness functions which are to be optimized simultaneously are defined as Minimize f3 = P SR in dB Minimize f4 = 1 N Tp ∆f subject to N Tp ∆f > Tp B and |R1 (τg )| < ǫ. 5.4 Techniques used In this chapter single objective evolutionary algorithm, PSO, and multiobjective algorithm, NSGA-II, are used to determine the parameters of the LFM pulse train. An overview of each of the algorithms is presented in sequel. 111 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 5.4.1 Particle swarm optimization The PSO was introduced by Kennedy and Eberhart [116] in 1995 which is a population based and self adaptive search optimization technique. This algorithm was developed based on simulation of animals social behavior such as bird flocking, fish schooling etc. Like other population based evolutionary computation algorithm such as GA, the PSO starts with random initialization of population, called as swarm, in the search space and each individual is called as a particle. Unlike GA, the PSO have not direct combination of genetic materials between the particles during the search. The PSO algorithm employs the social behavior of the particle in the swarm. Hence, it finds the global solution by adjusting the trajectory of each particle towards its own best solution and towards the best particle of the swarm in each generation [116–118]. The PSO is very popular because of the simplicity of implementation of the algorithm and ability to converge quickly to a acceptable good solution. In PSO, the trajectory of each particle in search space is altered according to its own velocity, own flying experience and flying experience of other particles in the swarm. The position of ith particle in D dimensional search space is given by xi = [xi1 , xi2 , ..... xiD ]T (5.15) and the velocity of ith particle is expressed as vi = [vi1 , vi2 , ..... viD ]T (5.16) The fitness function value is found out according to the user defined fitness function which is to be optimized. Let pbesti be the best position i.e. the best fitness value obtained by the ith particle at time t. So pbesti = [pbesti1 , pbesti2 , ..... pbestiD ]T 112 (5.17) Chapter 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques The fittest particle found in the swarm at time t is gbest = [gbest1 , gbest2 , ...... gbestD ]T (5.18) The position and velocity of each particle is updated as vid (t + 1) = wvid (t) + c1 r1 (pbestid (t) − xid (t)) + c2 r2 (gbestd (t) − xid (t)) xid (t + 1) = xid (t) + vid (t + 1) (5.19) (5.20) where d = 1, 2, ....D and w is a positive constant or positive linear or nonlinear function of time [119, 120]. w is called inertia weight which plays the role of balancing the local and global searches. c1 and c2 are two positive constants known as acceleration coefficients and r1 and r2 are two random numbers in between 0 and 1. The first term in the right hand side of (5.19) corresponds to the previous velocity which provides the necessary momentum and the second term stands for the cognitive component which represents the personal thinking of each particle. The cognitive component promotes the particles to move towards their own best position. The third term is called as social component which constitutes the cooperative effect of the particles in finding the global optimal solution. The social component always drags the particle towards the global particle found so far. The population is initialized with random positions and random velocities are assigned to each particle. The fitness function value is evaluated according to defined objective function i.e. to be optimized. At each generation the velocity and position of the each particle are updated according to (5.19) and (5.20) respectively. In a particular generation if a particle finds better position than previously found then its location is stored in the memory. A maximum velocity i.e. Vmaxd is defined for each dimension for the velocity vector vid in order to control the excessive roaming of the particle outside the defined search space. If vid exceeds Vmaxd , then vid is set to Vmaxd . 113 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 5.4.2 NSGA-II The NSGA-II which is described in Chapter 2 dealt with binary coded chromosome. In this chapter the real coded chromosomes are used and the genetic operators such as crossover and mutation are different than binary coded GA. For real coded chromosomes simulated binary crossover and polynomial mutation are used to generate offsprings. The steps such as population initialization, fitness function evaluation, crowding distance assignment, selection, recombination process are same as explained in Section 2.3.2 except the genetic operators. Genetic operators: Genetic operators such as crossover and mutation are used to explore and exploit new and better solution from the existing solutions in the objective space. Real coded NSGA-II uses simulated binary crossover [121, 122] and polynomial mutation [122, 123] to produce offspring. 1. Simulated binary crossover: A random number y is generated between 0 and 1. From a defined probability distribution function (pdf) another variable α is found such that the area under the pdf from 0 to α is equal to y. The pdf is defined as 0.5 (η + 1) αηc c P (α) = 0.5 (ηc + 1) η1+2 α c if α ≤ 1 if α > 1 (5.21) where ηc is the distribution index for crossover. This pdf is obtained by using the transformation α (y) = 1 (2y) (ηc +1) 1 1 [2(1−y)] (ηc +1) if y ≤ 0.5 otherwise (5.22) After obtaining α the off-spring children are computed as 1 [(1 − αk ) x1,k + (1 + αk ) x2,k ] 2 1 = [(1 + αk ) x1,k + (1 − αk ) x2,k ] 2 c1,k = (5.23) c2,k (5.24) 114 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 where αk is the value of α for k th component of the chromosome. ci,k is the k th component of ith child. xi,k is the k th component of ith parent which is selected for crossover. 2. Polynomial mutation: Mutation in GA restores lost or unexpected genetic materials into the solution to avoid convergence of the algorithm into a sub-optimal solution. The polynomial mutation is defined as ck = xk + xuk − xlk δk (5.25) where ck is the mutated child produced from parent xk . xuk and xlk are the upper and lower bound of xk . δk is the small variation which is obtained by (2r ) ηm1+1 − 1 k δk = 1 − [2 (1 − rk )] ηm1+1 if rk ≤ 0.5 otherwise (5.26) where ηm is the distribution index for mutation and rk is a random number in between 0 and 1. 5.5 Determination of parameters of LFM pulse train 5.5.1 Using PSO The fitness function defined in (5.14) is minimized to determine the parameters of LFM pulse train. The various steps are 1. The population of size M is initialized randomly in the given search space and each particle in the population consists of two dimensions corresponds to Tp ∆f and c. Random velocities are assigned to each particle . 2. The fitness function for each chromosome is evaluated according to (5.14). The particle having best fitness value called as gbest. Initially the pbest for a particle assumed as particle position itself. 115 Chapter 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques 3. The velocity and position of each particle are updated as given in (5.19) and (5.20) respectively. 4. The fitness function value is evaluated for the new position for each particle and compared with the corresponding pbest positions fitness value. If for a particle the new position fitness is better than that of pbest then the pbest will be replaced by new particle. The particle having best fitness value among all the pbests is selected as gbest. Steps 3 and 4 repeated until the predefined condition is satisfied. 5.5.2 Using NSGA-II The problems defined in Sections 5.3.2 and 5.3.3 use this algorithm to find the desired parameter values of LFM pulse train. The various steps involved are 1. A population having M chromosomes is randomly initialized and each chromosome contains two random values corresponds to Tp ∆f and c. 2. The fitness function values f2 and f3 (f3 and f4 for problem-3) are evaluated as given in Section 5.3.2. (Section 5.3.3 for problem-3) 3. The chromosomes are sorted using nondominated sort and all possible fronts are obtained as in Section 2.3.2. 4. The crowding distance for chromosomes in each front are evaluated according to the procedure explained in Section 2.3.2. 5. The chromosomes are selected using binary tournament selection according to Section 2.3.2. 6. The selected chromosomes undergo for genetic operations such as crossover and mutation to produce offspring as explained Section 5.4.2. 7. The off-spring population is combined with parent population and the best M chromosome selected for next generation as described in Section 2.3.2 116 Chapter 5 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Steps from 5 to 7 are repeated until the maximum number of generations. 5.6 Simulation results Simulation studies are carried out according to the problem formulation as presented in Section 5.3. The techniques explained in Section 5.4 are used for grating lobe suppression. To carry out PSO based estimation task the swarm size and number of generations are chosen to be 100 and 50 respectively. This choice is based on trial and error so as to achieve the best possible performance. The particles are randomly initialized in the defined search space Tp ∆f ∈ [2, 15] and c ∈ [1, 10] . The values of c1 and c2 are taken as 2. The velocity and position of the particles are updated according to (5.19) and (5.20) respectively. At the end of all the generations if f1 attains a zero value then each term of the right hand side of (5.14) becomes zero which means complete elimination of grating lobes. There are more than one set of Tp ∆f and c present in the defined search space for which f1 = 0. At the end of all the generations the best particle is saved and the program executed repeatedly to get the other distinct best solutions. The values of Tp ∆f , Tp B and B/∆f for N = 8 are listed in Table 5.1 for f1 = 0. Figures 5.4 and 5.5 show the plots of |R1 (τ )|, |R2 (τ )| and |R(τ )| for Tp ∆f = 2.5, Tp B = 12.5 and Tp ∆f = 4, Tp B = 16 respectively. From these figures it is observed that the peaks of the |R2 (τ )| (grating lobes) exactly coincide with the nulls of |R1 (τ )|, as a result there are nulls in |R(τ )| at those points. NSGA-II algorithm is employed for optimization of f2 and f3 associated in problem 2. The population size and the number of generations are taken to be 100 and 50 respectively. The distribution indices for crossover (ηc ) and mutation (ηm ) are chosen as 20 each. The probabilities of crossover and mutation are set to be 0.9 and 0.1 respectively. Tp ∆f and c are the two variables judiciously chosen by the NSGA-II algorithm to get low sidelobe level and reduced grating lobes. The initialized population is sorted based on nondomination and each solution is assigned with a crowding distance. The selection, crossover, mutation and recombination are 117 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 |R (τ)|, |R (τ)| 1 2 0.8 0.6 1 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF [dB] 0 −20 −40 −60 τ/T p Figure 5.4: Stepped frequency LFM pulse for Tp ∆f = 2.5, Tp B = 12.5 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) |R1(τ)|, |R2(τ)| 1 0.8 0.6 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF [dB] 0 −20 −40 −60 τ/Tp Figure 5.5: Stepped frequency LFM pulse for Tp ∆f = 4, Tp B = 16 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 118 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 Table 5.1: Values of Tp ∆f , Tp B obtained for N = 8 and f1 = 0 Tp ∆f 2 2.5 3 3.5 4 5 6 7 9 11 13 15 Tp B = (c + 1)Tp ∆f 4 12.5 9 24.5 16 12.5 36 24.5 40.5 60.5 84.5 112.5 B ∆f =c+1 2 5 3 7 4 2.5 6 3.5 4.5 5.5 6.5 7.5 carried out for each generation according the procedure given in Section 5.4. Figure 5.6 illustrates Pareto front obtained for Tp ∆f ∈ [2, 10], c ∈ [2, 10] and N = 8. This Pareto front provides the trade-off solutions between grating lobe and the PSR. It is evident from Figure 5.6 that for all the solutions the sidelobes are below 30 dB as compared to their respective mainlobes. All the solutions in the Pareto front are nondominant and a particular solution from the front is chosen according to the requirements of the application such as low sidelobe level or low grating lobes. For different values of Tp ∆f and Tp B corresponding values of |R1 (τ )|, |R2 (τ )| and ACF are shown in Figures. 5.7 to 5.9. In Figure 5.7 the nulls of |R1 (τ )| exactly falls on the grating lobes of |R2 (τ )| which means that all the grating lobes are canceled i.e. f2 = 0. In Figure 5.9, the maximum grating lobe amplitude is 0.021 which is prominently observed around τ /Tp = 0.7 in ACF. But the sidelobes occurring in Figure 5.7 are below 30.7753 dB from its mainlobe and that of in Figure 5.9 is 32.5 dB below its mainlobe. Therefore the parameter values of stepped frequency LFM pulse train are chosen according to the requirement of application. If it is required to suppress all the grating lobes below a certain value ǫ, then a constraint i.e. f1 < ǫ is associated with the optimization process. Different Pareto fronts can 119 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 −30.5 PSR in dB (f3) −31 −31.5 −32 −32.5 −33 0 0.005 0.01 0.015 maximum grating lobe amplitude (f ) 0.02 0.025 2 Figure 5.6: Pareto front obtained using NSGA-II for Tp ∆f ∈ [2, 10], c ∈ [2, 10] and N =8 be obtained by varying the upper and lower limits of Tp ∆f and c depending upon their ranges for that particular application. For Tp ∆f ∈ [2, 10], c ∈ [2, 10], ǫ = 0.01, N = 8 and Tp ∆f ∈ [5, 30], c ∈ [2, 10], ǫ = 0.01, N = 8 the Pareto fronts are depicted in Figures 5.10 and 5.11 respectively. The NSGA-II algorithm facilitates for choosing the parameters from a set of available optimal parameters according to the requirements of the system under consideration. Small grating lobes or low sidelobes can be achieved by choosing an appropriate solution from Pareto front. Suitable overlap ratio, i.e. B ∆f = c + 1, is accomplished by properly defining lower and higher limit of c during population initialization. For the third problem the population size and number of generations are chosen to be 200 and 50 for optimizing the values of f3 and f4 . Same set of previously chosen parameters of NSGA-II algorithm is used in this case. The population is randomly initialized for two parameters Tp ∆f and c for the given lower and upper limit. The initialized population is sorted according to the nondomination sorting and the process of selection, crossover, mutation and recombination are carried out for each generation according to the procedure laid down in Section 5.5.2. For Tp ∆f ∈ [2, 10], c ∈ [2, 5], ǫ = 0.01 and N = 8 the Pareto front is shown in the Figure 5.12. This Pareto front provides a trade-off between the grating lobe and mainlobe width. A solution from the Pareto front is chosen according to the requirements of 120 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 0.8 0.6 0.4 1 2 |R (τ)|,|R (τ)| 1 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ/T p Figure 5.7: Stepped frequency LFM pulse for Tp ∆f = 2, c = 5, Tp B = 12 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 0.8 0.6 0.4 1 2 | R (τ), R (τ) | 1 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ / Tp Figure 5.8: Stepped frequency LFM pulse for Tp ∆f = 2, c = 5.1412, Tp B = 12.2824 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 121 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 0.8 2 | R (τ) |, |R (τ) | 1 0.6 1 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ/T p Figure 5.9: Stepped frequency LFM pulse for Tp ∆f = 2.8721, c = 5.0978, Tp B = 17.5135 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) −30.6 −31 3 PSR in dB (f ) −30.8 −31.2 −31.4 −31.6 −31.8 0 0.001 0.002 0.003 0.004 0.005 0.006 0.007 0.008 0.009 0.01 Maximum grating lobe amplitude (f ) 2 Figure 5.10: Pareto front obtained using NSGA-II for Tp ∆f ∈ [2, 10] c ∈ [2, 10], ǫ = 0.01 and N = 8 122 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 −28.4 −28.45 PSR in dB(f3) −28.5 −28.55 −28.6 −28.65 −28.7 −28.75 −28.8 −28.85 0 0.001 0.002 0.003 0.004 0.005 0.006 0.007 0.008 0.009 0.01 Maximum gratiing lobe amplitude (f ) 2 Figure 5.11: Pareto front obtained using NSGA-II for Tp ∆f ∈ [5, 30], c ∈ [2, 10], ǫ = 0.01 and N = 8 −22 −23 −24 −25 f 3 −26 −27 −28 −29 −30 −31 −32 0.01 0.015 0.02 0.025 0.03 0.035 0.04 0.045 f4 Figure 5.12: Pareto front obtained using NSGA-II for Tp ∆f ∈ [2, 10], c ∈ [2, 5], ǫ = 0.01 and N = 8 application under consideration. If the application demands high range resolution, a solution having low value of f4 is chosen and if it requires low sidelobe level, a solution corresponds to high magnitude of f3 is chosen. Figures 5.13 to 5.16 show |R1 (τ )|, |R2 (τ )| and ACF for different values of Tp ∆f , c and ǫ = 0.01. It is evident from the figures that reduction of peak sidelobe is achieved at the cost of increase in mainlobe width or reduction in range resolution. 123 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 0.8 2 | R (τ) |,| R (τ) | 1 0.6 1 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ / Tp Figure 5.13: Stepped frequency LFM pulse for Tp ∆f = 9.0188, c = 3.5502, Tp B = 41.0373 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 0.8 2 |R (τ)|, |R (τ)| 1 0.6 1 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ / Tp Figure 5.14: Stepped frequency LFM pulse for Tp ∆f = 4.9667, c = 4.0720, Tp B = 25.1911 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 124 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Chapter 5 0.8 2 | R (τ) |, | R (τ) | 1 0.6 1 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ / Tp Figure 5.15: Stepped frequency LFM pulse for Tp ∆f = 3.6048, c = 4.6129, Tp B = 20.2334 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 0.8 1 2 | R (τ) |, | R (τ) | 1 0.6 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0.5 0.6 0.7 0.8 0.9 1 ACF in dB 0 −20 −40 −60 τ/T p Figure 5.16: Stepped frequency LFM pulse for Tp ∆f = 3, c = 5, Tp B = 18 and N = 8. Top: |R1 (τ )| (dash) and |R2 (τ )| (solid). Bottom: ACF (in dB) 125 Chapter 5 5.7 Efficient Design of Stepped Frequency Pulse Train Using Evolutionary Computation Techniques Conclusion Frequency stepping technique is mostly used in radar technology to achieve high range resolution by combining the effect of narrowband pulses that span in the desired bandwidth. The drawback of such type of waveform is the presence of the grating lobes. In this chapter the PSO algorithm is used to determine the parameters of LFM pulse train for which all the grating lobes are nullified. Using the PSO various combinations of Tp ∆f and Tp B are found for which all the grating lobes are nullified and listed in Table 5.1. Changing the limits of Tp ∆f and c more combination of Tp ∆f and Tp B can be found for which all the grating lobes are nullified. The multiobjective NSGA-II algorithm has been applied to determine the parameters of stepped frequency pulse train to get reduced grating lobes, low sidelobes and narrow mainlobe width at the matched filter output of the pulse train. The multiobjective optimization algorithm enables to provide trade off solutions between different objectives through Pareto front that contains nondomination solutions. In the proposed work the multiobjective problem has been formulated in two different manners. One formulation provides the trade-off solutions between grating lobes and peak sidelobe and the other provides the trade-off solutions between peak sidelobe and mainlobe width. 126 Chapter 6 Conclusion and Future Work 6.1 Conclusion In this chapter, the conclusion of the whole thesis is presented and future research problems are outlined for further investigation in the same or related topics. In this thesis investigation has been made on developing efficient pulse compression techniques for phase and frequency modulated waveforms. The main contribution of the thesis is the use of neural network structures and evolutionary computation techniques for pulse compression. Biphase codes of longer sequences having low PSL and high MF are important research area in the field of radar signal processing. There is no available technique to generate a certain length code for a given PSL and/or MF. In this thesis a multiobjective algorithm (NSGA-II) is presented to generate the biphase codes of length 49 to 59 using PSL and MF as two different objective functions. The use of NSGA-II algorithm has provided more than one nondominated solutions and a particular code is to be selected depending upon specific situation. This algorithm in general can be applied to generate codes of any length. Mismatch filters are used to provide better PSR than that of matched filter of a given sequence. Several ANN based mismatch filters are used to achieve reduced sidelobes for 13-bit and 35-bit Barker codes at the output of the filter. In this work, the RNN and RRBF structures are proposed to use as a pulse compression filter 127 Conclusion and Future Work Chapter 6 which have been provided better performance in terms of PSR under various adverse conditions such as noise, multiple target and Doppler shift. The performance of the proposed methods is compared to that of MLP and RBF based pulse compression techniques. The comparison study reveals that the RRBF based pulse compression technique performs the best among others. Biphase codes can be easily generated and their matched filters implementation are also simple. These advantages are achieved at the cost of higher sidelobes. Polyphase codes have lower sidelobes compared to biphase codes for a certain length code. The LFM signals are more Doppler tolerant than phase coded signals. Hence, phase codes such as Frank, P1 , P2 , P3 and P4 codes are derived from the LFM signals to get the advantages of the Doppler shift performance of LFM signal. The PSRs offered by LFM and polyphase codes are not adequate for many radar applications. Various windows are used to suppress the sidelobe of LFM and polyphase codes. In this thesis the use of convolutional windows are proposed as the weighing function at the receiver to get better PSR values at higher Doppler shifts compared to that of conventional windows. Amplitude tapering and phase distortion techniques are employed to modify the transmitted LFM signal and the convolutional window is used as the weighing function at the receiver offers better PSR values that of conventional windows. In high range resolution radar the bandwidth of the signal should be large to achieve narrow mainlobe width. To overcome the difficulty of generation and processing, the wideband signal is split into a number of narrowband signals which together called as stepped frequency waveform. The ACF of stepped frequency LFM pulse train suffers from grating lobes for Tp ∆f > 1. Hence the range resolution capability of the waveform is reduced. A PSO algorithm based technique is proposed to choose the parameters of LFM pulse train to nullify or suppress the grating lobes. Further, a widely used multiobjective NSGA-II algorithm based approach is proposed to determine the optimum parameters of LFM pulse train to achieve reduced grating lobes, low sidelobes and narrow mainlobe width. The multiobjective 128 Conclusion and Future Work Chapter 6 problem has been formulated in two different ways. One formulation provides the trade-off solutions between grating lobes and peak sidelobe and the other provides the trade-off between peak sidelobe and mainlobe width. 6.2 Future work The research work presented in the thesis can be further extended in following ways. • Genetic operators such as crossover and mutation are important operations in GA and NSGA-II. 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Panda, “A Multiobjective Optimization Approach to Determine the Parameters of Stepped Frequency Pulse Train,” Aerospace Science and Technology-Elseveir, DOI 10.1016/j.ast.2011.10.008.(Accepted 17-Oct.-2011) 2. A. K. Sahoo and G. Panda “Sidelobe Suppression Using Convolutional Windows in Radar,” Int. J. Signal and Imaging Systems Engineering (IJSISE), Inderscience Publisher.(Accepted 25-Mar-2011) 3. A. K. Sahoo and G. Panda, “Doppler Tolerant Convolutional Windows for Radar Pulse Compression,” Int. J. of Electronics and Communication Engineering, International Research Publication, vol. 4, no.1, pp.145-152, 2011. 4. A. K. Sahoo and G. Panda, “Suppression of grating lobes in stepped frequency LFM pulse train using PSO,” Aerospace Science and Technology-Elseveir.(Communicated) Conferences 1. A. K. Sahoo G. Panda and P.M. Pradhan, “Generation of pulse compression codes using NSGA-II, Annual IEEE India Conference (INDICON),Ahamadabad, pp.1-4, Dec. 18-20, 2009.DOI. 10.1109/INDCON.2009.5409443. 2. A. Sailaja, A. K. Sahoo, G. Panda and V. Baghel, “A recurrent neural network approach to pulse radar detection,” Annual IEEE India Conf. (INDICON), Ahamadabad, pp. 1-4, Dec. 18-20, 2009. DOI. 10.1109/INDCON.2009.5409446. 141 Dissemination of Work 3. A. K. Sahoo and G. Panda, “Sidelobe Reduction of LFM Signal Using Convolutional Windows,” Int. Conf. on Electronics Systems (ICES), NIT Rourkela, pp. 86-89, Jan. 9-11, 2011. 142 Resume Ajit Kumar Sahoo Assistant Professor Department of Electronics & Communication Engineering National Institute of Technology Rourkela Rourkela, Orissa – 769 008, India. Ph: +91-9861370334(M) E-mail: [email protected] Qualification • Ph.D. (Continuing) National Institute of Technology, Rourkela, Orissa, India • M. Tech. (Telematics & Signal Processing) National Institute of Technology Rourkela, Orissa, India. • B.E. (Electronics and Telecommunication Engineering) Biju Patnaik University of Technology, Orissa, India [First division] • +2 (Science) Council of Higher Secondary Education, Orissa, India [First division] • 10th Board of Secondary Education, Orissa, India [First division] Publications • 04 Journal Articles • 05 Conference Papers Date of Birth • 8th April, 1982

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