MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller General Description

MAX1846/MAX1847 High-Efficiency, Current-Mode, Inverting PWM Controller General Description

19-2091; Rev 0; 8/01

High-Efficiency, Current-Mode,

Inverting PWM Controller

General Description

MAX1846/MAX1847 high-efficiency PWM inverting controllers allow designers to implement compact, lownoise, negative-output DC-DC converters for telecom and networking applications. Both devices operate from +3V to +16.5V input and generate -2V to -200V output. To minimize switching noise, both devices feature a current-mode, constant-frequency PWM control scheme. The operating frequency can be set from

100kHz to 500kHz through a resistor.

The MAX1846 is available in an ultra-compact 10-pin

µMAX package. Operation at high frequency, compatibility with ceramic capacitors, and inverting topology without transformers allow for a compact design.

Compatibility with electrolytic capacitors and flexibility to operate down to 100kHz allow users to minimize the cost of external components. The high-current output drivers are designed to drive a P-channel MOSFET and allow the converter to deliver up to 30W.

The MAX1847 features clock synchronization and shutdown functions. The MAX1847 can also be configured to operate as an inverting flyback controller with an Nchannel MOSFET and a transformer to deliver up to

70W. The MAX1847 is available in a 16-pin QSOP package.

Current-mode control simplifies compensation and provides good transient response. Accurate current-mode control and over current protection are achieved through low-side current sensing.

Features

90% Efficiency

+3.0V to +16.5V Input Range

-2V to -200V Output

Drives High-Side P-Channel MOSFET

100kHz to 500kHz Switching Frequency

Current-Mode, PWM Control

Internal Soft-Start

Electrolytic or Ceramic Output Capacitor

The MAX1847 also offers:

Synchronization to External Clock

Shutdown

N-Channel Inverting Flyback Option

PART

MAX1846EUB

MAX1847EEE

Ordering Information

TEMP. RANGE

-40

°C to +85°C

-40

°C to +85°C

PIN-PACKAGE

10 µMAX

16 QSOP

POSITIVE

V

IN

Typical Operating Circuit

Applications

Cellular Base Stations

Networking Equipment

Optical Networking Equipment

SLIC Supplies

CO DSL Line Driver Supplies

Industrial Power Supplies

Automotive Electronic Power Supplies

Servers

VL IN

EXT

MAX1846

MAX1847

COMP

CS

FREQ

PGND

P

NEGATIVE

V

OUT

REF

GND

FB

Pin Configurations appear at end of data sheet.

________________________________________________________________ Maxim Integrated Products 1

For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at

1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

High-Efficiency, Current-Mode,

Inverting PWM Controller

ABSOLUTE MAXIMUM RATINGS

IN, SHDN to GND ...................................................-0.3V to +20V

PGND to GND .......................................................-0.3V to +0.3V

VL to PGND for V

IN

≤ 5.7V...........................-0.3V to (V

IN

+ 0.3V)

VL to PGND for V

IN

> 5.7V .......................................-0.3V to +6V

EXT to PGND ...............................................-0.3V to (V

IN

+ 0.3V)

REF, COMP to GND......................................-0.3V to (VL + 0.3V)

CS, FB, FREQ, POL, SYNC to GND .........................-0.3V to +6V

Continuous Power Dissipation (T

A

= +70°C)

10-Pin µMAX (derate 5.6mW/°C above +70°C) ...........444mW

16-Pin QSOP (derate 8.3mW/°C above +70°C)...........696mW

Operating Temperature Range ...........................-40°C to +85°C

Junction Temperature ......................................................+150°C

Storage Temperature Range .............................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

ELECTRICAL CHARACTERISTICS

(V

SHDN

= V

IN

= +12V, SYNC = GND, PGND = GND, R

FREQ

= 147k

Ω ±1%, C

VL

= 0.47µF, C

REF

= 0.1µF, T

A

= 0°C to +85°C, unless otherwise noted.)

MIN TYP MAX UNITS PARAMETER

PWM CONTROLLER

Operating Input Voltage Range

UVLO Threshold

UVLO Hysteresis

FB Threshold

FB Input Current

Load Regulation

CONDITIONS

V

V

IN

rising

IN

falling

No load

V

FB

= -0.1V

I

C

COMP

= 0.068µF, V

OUT

= -48V,

OUT

= 20mA to 200mA (Note 1)

Line Regulation

C

V

COMP

= 0.068µF, V

OUT

= -48V,

IN

= +8V to +16.5V, I

OUT

= 100mA

Current-Limit Threshold

CS Input Current

Supply Current

Shutdown Supply Current

REFERENCE AND VL REGULATOR

REF Output Voltage

REF Load Regulation

VL Output Voltage

VL Load Regulation

I

I

CS = GND

V

FB

= -0.1V, V

IN

= +3.0V to +16.5V

SHDN = GND, V

IN

= +3.0V to +16.5V

REF

= 50µA

REF

= 0 to 500µA

I

VL

I

VL

= 100µA

= 0.1mA to 2.0mA

3.0

-1

85

1.236

0.04

100

10

16.5

2.8

2.95

2.6

2.74

-12

-50

60

0

-6

12

50

20

0.75

1.2

10

1.25

-2

-20

0

115

25

1.264

-15

3.85

4.25

4.65

-60 mV mV nA mV

µA mA

µA

V

V

%

%

V mV

V mV

2 _______________________________________________________________________________________

High-Efficiency, Current-Mode,

Inverting PWM Controller

ELECTRICAL CHARACTERISTICS (continued)

(V

SHDN

= V

IN

= +12V, SYNC = GND, PGND = GND, R

FREQ

= 147k

Ω ±1%, C

VL

= 0.47µF, C

REF

= 0.1µF, T

A

= 0°C to +85°C, unless otherwise noted.)

OSCILLATOR

Oscillator Frequency

Maximum Duty Cycle

R

R

FREQ

= 500k

Ω ±1%

FREQ

= 147k

Ω ±1%

R

R

FREQ

= 76.8k

Ω ±1%

FREQ

= 500k

Ω ±1%

R

R

FREQ

= 147k

Ω ±1%

FREQ

= 76.8kz

±1%

90 100 110

255 300 345

93

85

500

96

88

80

97

90 kHz

%

SYNC Input Signal Duty-Cycle

Range

7 93 %

Minimum SYNC Input Logic Low

Pulse Width

SYNC Input Rise/Fall Time

SYNC Input Frequency Range

(Note 2)

100

50 200

200

550 ns ns kHz

DIGITAL INPUTS

POL, SYNC, SHDN Input High

Voltage

POL, SYNC, SHDN Input Low

Voltage

POL, SYNC Input Current

SHDN Input Current

SOFT-START

Soft-Start Clock Cycles

Soft-Start Levels

EXT OUTPUT

EXT Sink/Source Current

EXT On-Resistance

POL, SYNC = GND or VL

V

V

V

SHDN

= +5V or GND

SHDN

= +16.5V

IN

= +5V, V

EXT

forced to +2.5V

EXT high or low, tested with 100mA load, V

EXT high or low, tested with 100mA load, V

IN

= +5V

IN

= +3V

2.0

-12

20

-4

1.5

1024

64

1

2

5

0.45

40

0

6

5

10

V

V

µA

µA

Cycles

A

Note 1: Production test correlates to operating conditions.

Note 2: Guaranteed by design and characterization.

_______________________________________________________________________________________ 3

High-Efficiency, Current-Mode,

Inverting PWM Controller

ELECTRICAL CHARACTERISTICS

(V

SHDN

= V

IN

= +12V, SYNC = GND, PGND = GND, R

FREQ

= 147k

Ω ±1%, C

VL unless otherwise noted.) (Note 3)

= 0.47µF, C

REF

= 0.1µF, T

A

= -40°C to +85°C,

MIN MAX UNITS PARAMETER CONDITIONS

PWM CONTROLLER

Operating Input Voltage Range

V

UVLO Threshold

IN

rising

FB Threshold

FB Input Current

V

IN

falling

No load

V

FB

= -0.1V

Load Regulation

I

C

COMP

= 0.068µF, V

OUT

= -48V,

OUT

= 20mA to 200mA (Note 1)

Current Limit Threshold

CS Input Current

Supply Current

Shutdown Supply Current

CS = GND

V

FB

= -0.1V, V

IN

= +3.0V to +16.5V

SHDN = GND, V

IN

= +3.0V to +16.5V

REFERENCE AND VL REGULATOR

REF Output Voltage

REF Load Regulation

VL Output Voltage

VL Load Regulation

OSCILLATOR

I

I

REF

= 50µA

REF

= 0 to 500µA

I

I

VL

= 100µA

VL

= 0.1mA to 2.0mA

Oscillator Frequency

Maximum Duty Cycle

R

R

R

R

FREQ

= 500k

Ω ±1%

FREQ

= 147k

Ω ±1%

FREQ

= 500k

Ω ±1%

FREQ

= 147k

Ω ±1%

SYNC Input Signal Duty-Cycle

Range

Minimum SYNC Input Logic Low

Pulse Width

SYNC Input Rise/Fall Time

SYNC Input Frequency Range

(Note 2)

DIGITAL INPUTS

POL, SYNC, SHDN Input High

Voltage

POL, SYNC, SHDN Input Low

Voltage

3.0

2.6

-20

-50

-2

85

1.225

3.85

84

255

93

84

7

100

2.0

16.5

2.95

20

50

0

115

20

1.2

25

1.275

-15

4.65

-60

116

345

98

93

93

200

200 ns ns

550 kHz

0.45

V

V mV nA mV

µA mA

µA

%

V mV

V mV kHz

%

%

V

V

4 _______________________________________________________________________________________

High-Efficiency, Current-Mode,

Inverting PWM Controller

ELECTRICAL CHARACTERISTICS (continued)

(V

SHDN

= V

IN

= +12V, SYNC = GND, PGND = GND, R

FREQ

= 147k

Ω ±1%, C

VL unless otherwise noted.) (Note 3)

= 0.47µF, C

REF

= 0.1µF, T

A

= -40°C to +85°C,

PARAMETER

POL, SYNC Input Current

SHDN Input Current

CONDITIONS

POL, SYNC = GND or VL

V

V

SHDN

= +5V or GND

SHDN

= +16.5V

MIN

-12

MAX UNITS

40

0

6

µA

µA

EXT OUTPUT

EXT On-Resistance

EXT high or low, I

EXT high or low, I

EXT

= 100mA, V

IN

= +5V

EXT

= 100mA, V

IN

= +3V

7.5

12

Note 3: Parameters to -40°C are guaranteed by design and characterization.

Typical Operating Characteristics

(Circuit references are from Table 1 in the Main Application Circuits section, C

VL

= 0.47µF, C

REF

= 0.1µF, T

A

= +25°C, unless otherwise noted.)

EFFICIENCY vs. LOAD CURRENT

100

90

80

70

60

50

40

30

20

10

0

V

IN

= 5V

V

IN

= 16.5V

1

APPLICATION CIRCUIT A

V

OUT

= -5V

10 100 1000

LOAD CURRENT (mA)

10,000

EFFICIENCY vs. LOAD CURRENT

100

90

80

70

60

50

40

30

20

10

0

V

IN

= 3V

V

V

IN

IN

= 5V

= 3.3V

1

APPLICATION CIRCUIT B

V

OUT

= -12V

10 100 1000

LOAD CURRENT (mA)

10,000

EFFICIENCY vs. LOAD CURRENT

100

90

80

70

60

50

40

30

20

10

0

V

IN

= 12V

V

IN

= 16.5V

1

APPLICATION CIRCUIT C

10 100

LOAD CURRENT (mA)

V

OUT

= -48V

1000

OUTPUT VOLTAGE LOAD REGULATION

-11.90

-11.92

-11.94

-11.96

-11.98

-12.00

-12.02

-12.04

-12.06

-12.08

-12.10

0

APPLICATION CIRCUIT B V

IN

= 5V

100 200 300 400

LOAD CURRENT (mA)

500 600

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0

0 2

SUPPLY CURRENT vs. SUPPLY VOLTAGE

4

V

FB

= -0.1V

6 8 10 12 14 16

V

IN

(V)

1.262

1.258

1.254

1.250

1.246

1.242

1.238

-40 -20

REFERENCE VOLTAGE vs. TEMPERATURE

0 20 40

TEMPERATURE (

°C)

60 80 100

_______________________________________________________________________________________

5

High-Efficiency, Current-Mode,

Inverting PWM Controller

Typical Operating Characteristics (continued)

(Circuit references are from Table 1 in the Main Application Circuits section, C

VL

= 0.47µF, C

REF

= 0.1µF, T

A

= +25°C, unless otherwise noted.)

REFERENCE LOAD REGULATION

VL VOLTAGE vs. TEMPERATURE

VL LOAD REGULATION

1.260

4.27

4.340

4.300

4.26

1.255

4.260

4.25

1.250

4.220

4.24

4.180

1.245

1.240

0 100 200 300

I

REF

(

µA)

400 500

SHUTDOWN SUPPLY CURRENT vs. TEMPERATURE

16

14

12

10

8

V

6

4

2

0

-40 -20

IN

= 16.5V

0 20

V

IN

= 10V

40

V

IN

TEMPERATURE (

°C)

= 3V

60 80 100

302

301

300

299

298

297

296

295

294

-40 -20

SWITCHING FREQUENCY vs. TEMPERATURE

R

FREQ

= 147k

Ω ±1%

0 20 40

TEMPERATURE (

°C)

60 80 100

10

8

6

4

4.140

4.100

-40 -20 0 20 40

TEMPERATURE (

°C)

60

I

VL

= 0

80 100

OPERATING CURRENT vs. TEMPERATURE

14

A: V

IN

= 3V, V

OUT

= -12V

A

12

APPLICATION CIRCUIT A

B: V

IN

= 5V, V

OUT

= -5V

C: V

IN

= 16.5V, V

OUT

= -5V

B

2

C

160

140

120

100

80

60

40

20

0

0

0

-40 -20 0 20 40 60

TEMPERATURE (

°C)

EXT RISE/FALL TIME vs. CAPACITANCE

FALL TIME

RISE TIME

80 100

2000 4000 6000

CAPACITANCE (pF)

V

IN

= 12V

8000 10,000

4.23

4.22

0 0.2

0.4 0.6

0.8 1.0

I

VL

(mA)

1.2 1.4 1.6 1.8 2.0

SWITCHING FREQUENCY vs. R

FREQ

500

400

300

200

100

SHDN

0

V

OUT

0

0

I

L

100 200 300 400 500 600

R

FREQ

(k

Ω)

EXITING SHUTDOWN

APPLICATION CIRCUIT B

1ms/div

6 _______________________________________________________________________________________

MAX1846/7 toc15

5V/div

5V/div

1A/div

High-Efficiency, Current-Mode,

Inverting PWM Controller

Typical Operating Characteristics (continued)

(Circuit references are from Table 1 in the Main Application Circuits section, C

VL

= 0.47µF, C

REF

= 0.1µF, T

A

= +25°C, unless otherwise noted.)

ENTERING SHUTDOWN

MAX1846/7 toc16

HEAVY-LOAD SWITCHING

WAVEFORM

MAX1846/7 toc17

SHDN

V

OUT

5V/div

100mV/div

0

5V/div

1A/div

V

OUT

I

L

I

L

APPLICATION CIRCUIT B

1ms/div

V

OUT

1A/div

LX

LIGHT-LOAD SWITCHING

WAVEFORM

MAX1846/7 toc18

APPLICATION CIRCUIT B

1

µs/div

I

LOAD

= 600mA

100mV/div

1A/div

I

L

10V/div

I

LOAD

V

OUT

LX

LOAD-TRANSIENT RESPONSE

MAX1846/7 toc19

500mV/div

10V/div

APPLICATION CIRCUIT B

1

µs/div

I

LOAD

= 50mA

LOAD-TRANSIENT RESPONSE

MAX1846/7 toc20

I

LOAD

V

OUT

200mV/div

I

L

1A/div

I

L

500mA/div

APPLICATION CIRCUIT B

2ms/div

I

LOAD

= 10mA to 400mA

APPLICATION CIRCUIT C

400

µs/div

I

LOAD

= 4mA to 100mA

_______________________________________________________________________________________ 7

High-Efficiency, Current-Mode,

Inverting PWM Controller

Pin Description

MAX1846

PIN

MAX1847

1

2

3

4

5

8

9

6

7

10

1

2

3

4

5

6

7,9

8

10,11

12

13

14

15

16

NAME FUNCTION

FB

N.C.

SHDN

GND

PGND

CS

EXT

IN

POL

VL

FREQ

COMP

REF

SYNC

Sets polarity of the EXT pin. Connect POL to GND to set EXT for use with an external

PMOS high-side FET. Connect POL to VL to set EXT for use with an external NMOS lowside FET in transformer-based applications.

VL Low-Dropout Regulator. Connect 0.47µF ceramic capacitor from VL to GND.

Oscillator Frequency Set Input. Connect a resistor (R

FREQ

) from FREQ to GND to set the internal oscillator frequency from 100kHz (R

FREQ

= 500k

Ω) to 500kHz (R

FREQ

= 76.8k

Ω).

R

FREQ

is still required if an external clock is used at SYNC. See Setting the Operating

Frequency section.

Compensation Node for Error Amp/Integrator. Connect a series resistor/capacitor network from COMP to GND for loop compensation. See Design Procedure.

1.25V Reference Output. REF can source up to 500µA. Bypass with a 0.1µF ceramic capacitor from REF to GND.

Feedback Input. Connect FB to the center of a resistor-divider connected between the output and REF. The FB threshold is 0.

No Connection

Shutdown Control. Drive SHDN low to turn off the DC-DC controller. Drive high or connect to IN for normal operation.

Analog Ground. Connect to PGND.

Negative Rail for EXT Driver and Negative Current-Sense Input. Connect to GND.

P osi ti ve C ur r ent- S ense Inp ut. C onnect a cur r ent- sense r esi stor ( R

C S

) b etw een C S and

External MOSFET Gate-Driver Output. EXT swings from IN to PGND.

Power-Supply Input

Operating Frequency Synchronization Control. Drive SYNC low or connect to GND to set the internal oscillator frequency with R

FREQ

. Drive SYNC with a logic-level clock input signal to externally set the converter’s operating frequency. DC-DC conversion cycles initiate on the rising edge of the input clock signal. Note that when driving SYNC with an external signal, R

FREQ

must still be connected to FREQ.

8 _______________________________________________________________________________________

3 x 22

µF

10V

High-Efficiency, Current-Mode,

Inverting PWM Controller

Typical Application Circuit

V

IN

+3V to +5.5V

22k

0.22

0.47

µF

µF

150k

10k

VL

2

8

SHDN

IN

15

16

SYNC

EXT

14

CS

13

MAX1847

4

COMP

3

FREQ

5

REF

POL

1

N.C.

PGND

7, 9

12

GND

10, 11

FB

6

0.1

µF

1200pF

FDS6375

CMSH5-40

10

µH

DO5022P-103

0.02

1W

47

µF

16V

R1

95.3k

1%

Sanyo

16TPB47M

R2

10.0k

1%

47

µF

16V

V

OUT

-12V AT 400mA

_______________________________________________________________________________________ 9

High-Efficiency, Current-Mode,

Inverting PWM Controller

Functional Diagram

IN

SHDN

MAX1847 ONLY

POL

SYNC

MAX1847 ONLY

FREQ

COMP

FB

REF

STARTUP

CIRCUITRY

VL

REGULATOR

UNDER-

VOLTAGE

LOCK OUT

OSCILLATOR

CONTROL

CIRCUITRY

ERROR

COMPARATOR

SOFT-START

G

M

ERROR

AMPLIFIER

SLOPE

COMP

CURRENT-

SENSE

AMPLIFIER

REFERENCE

EXT DRIVER

EXT

PGND

VL

MAX1846

MAX1847

CS

PGND

GND

10 ______________________________________________________________________________________

Detailed Description

The MAX1846/MAX1847 current-mode PWM controller use an inverting topology that is ideal for generating output voltages from -2V to -200V. Features include shutdown, adjustable internal operating frequency or synchronization to an external clock, soft-start, adjustable current limit, and a wide (+3V to +16.5V) input range.

PWM Controller

The architecture of the MAX1846/MAX1847 currentmode PWM controller is a Bi-CMOS multi-input system that simultaneously processes the output-error signal, the current-sense signal, and a slope-compensation ramp (Functional Diagram). Slope compensation prevents subharmonic oscillation, a potential result in current-mode regulators operating at greater than 50% duty cycle. The controller uses fixed-frequency, current-mode operation where the duty ratio is set by the input-to-output voltage ratio. The current-mode feedback loop regulates peak inductor current as a function of the output error signal.

Internal Regulator

The MAX1846/MAX1847 incorporate an internal lowdropout regulator (LDO). This LDO has a 4.25V output and powers all MAX1846/MAX1847 functions (excluding EXT) for the primary purpose of stabilizing the performance of the IC over a wide input voltage range

(+3V to +16.5V). The input to this regulator is connected to IN, and the dropout voltage is typically 100mV, so that when V

IN is less than 4.35V, VL is typically V

IN minus 100mV. When the LDO is in dropout, the

MAX1846/MAX1847 still operate with V

IN as low as 3V.

For best performance, it is recommended to connect

VL to IN when the input supply is less than 4.5V.

Undervoltage Lockout

The MAX1846/MAX1847 have an undervoltage lockout circuit that monitors the voltage at VL. If VL falls below the UVLO threshold (2.8V typ), the control logic turns the

P-channel FET off (EXT high impedance). The rest of the

IC circuitry is still powered and operating. When VL increases to 60mV above the UVLO threshold, the IC resumes operation from a start up condition (soft-start).

Soft-Start

The MAX1846/MAX1847 feature a “digital” soft-start that is preset and requires no external capacitor. Upon startup, the FB threshold decrements from the reference voltage to 0 in 64 steps over 1024 cycles of f

OSC or f

SYNC

. See the Typical Operating Characteristics for a scope picture of the soft-start operation. Soft-start is implemented: 1) when power is first applied to the IC,

High-Efficiency, Current-Mode,

Inverting PWM Controller

2) when exiting shutdown with power already applied, and 3) when exiting undervoltage lockout.

Shutdown (MAX1847 only)

The MAX1847 shuts down to reduce the supply current to 10µA when SHDN is low. In this mode, the internal reference, error amplifier, comparators, and biasing circuitry turn off. The EXT output becomes high impedance and the external pullup resistor connected to EXT pulls

V

EXT to V

IN

, turning off the P-channel MOSFET. When in shutdown mode, the converter’s output goes to 0.

Frequency Synchronization

(MAX1847 only)

The MAX1847 is capable of synchronizing its switching frequency with an external clock source. Drive SYNC with a logic-level clock input signal to synchronize the

MAX1847. A switching cycle starts on the rising edge of the signal applied to SYNC. Note that the frequency of the signal applied to SYNC must be higher than the default frequency set by R

FREQ

. This is required so that the internal clock does not start a switching cycle prematurely. If SYNC is inactive for an entire clock cycle of the internal oscillator, the internal oscillator takes over the switching operation. Choose R

FREQ such that f

OSC

= 0.9

✕ f

SYNC

.

EXT Polarity (MAX1847 only)

The MAX1847 features an option to utilize an N-channel

MOSFET configuration, rather than the typical P-channel MOSFET configuration (Figure 1). In order to drive the different polarities of these MOSFETs, the MAX1847 is capable of reversing the phase of EXT by 180 degrees. When driving a P-channel MOSFET, connect

POL to GND. When driving an N-Channel MOSFET, connect POL to VL. These POL connections ensure the proper polarity for EXT. For design guidance in regard to this application, refer to the MAX1856 data sheet.

Design Procedure

Initial Specifications

In order to start the design procedure, a few parameters must be identified: the minimum input voltage expected

(V

IN(MIN)

), the maximum input voltage expected

(V

IN(MAX)

), the desired output voltage (V

OUT

), and the expected maximum load current (I

LOAD

).

Calculate the Equivalent Load Resistance

This is a simple calculation used to shorten the verification equations:

R

LOAD

= V

OUT

/ I

LOAD

______________________________________________________________________________________ 11

High-Efficiency, Current-Mode,

Inverting PWM Controller

V

IN

+12V

0.033

µF

0.47

µF

270k

150k

12

µF

25V

VP1-0190

12.2

µH

1:4

8

2

VL

SHDN

1

POL

16

SYNC

15

IN

MAX1847

EXT

14

13

CS

7, 9

N.C.

IRLL2705

0.05

0.5W

4

COMP

3

FREQ

5

REF

0.1

µF

PGND

FB

GND

10, 11

12

6

CMR1U-02

470

100pF

100V

1800pF

383k

1%

10.0k

1%

V

OUT

-48V AT 100mA

12

µF

100V

Figure 1. Using an N-Channel MOSFET (MAX1847 only)

Calculate the Duty Cycle

The duty cycle is the ratio of the on-time of the MOSFET switch to the oscillator period. This is determined by the ratio of the input voltage to the output voltage. Since the input voltage typically has a range of operation, a minimum (D

MIN

) and maximum (D

MAX

) duty cycle is calculated by:

D

MIN

=

V

IN MAX )

V

V

OUT

SW

+

V

LIM

V

D

V

OUT

+

V

D

D

MAX

=

V

IN MIN )

V

V

SW

OUT

+

V

D

V

LIM

V

OUT

+

V

D where V

D is the forward drop across the output diode,

V

SW is the drop across the external FET when on, and

V

LIM is the current-limit threshold. To begin with, assume V

D

= 0.5V for a Schottky diode, V

SW

= 100mV, and V

LIM

= 100mV. Remember that V

OUT is negative when using this formula.

Setting the Output Voltage

The output voltage is set using two external resistors to form a resistive-divider to FB between the output and

REF (refer to R1 and R2 in Figure 1). V

REF is nominally

1.25V and the regulation voltage for FB is nominally 0.

The load presented to the reference by the feedback resistors must be less than 500µA. This is to guarantee that V

REF is in regulation (see Electrical Characteristics

Table). Conversely, the current through the feedback resistors must be large enough so that the leakage current of the FB input (50nA) is insignificant. Therefore, select R2 so that I

R2 is between 50µA and 250µA.

I

R2

= V

REF

/ R2 where V

REF

= 1.25V. A typical value for R2 is 10k

Ω.

Once R2 is selected, calculate R1 with the following equation:

R1 = R2 x (-V

OUT

/ V

REF

)

Setting the Operating Frequency

The MAX1846/MAX1847 are capable of operating at switching frequencies from 100kHz to 500kHz. Choice of operating frequency depends on a number of factors:

1) Noise considerations may dictate setting (or synchronizing) f

OSC above or below a certain frequency or band of frequencies, particularly in RF applications.

12 ______________________________________________________________________________________

High-Efficiency, Current-Mode,

Inverting PWM Controller

2) Higher frequencies allow the use of smaller value

(hence smaller size) inductors and capacitors.

3) Higher frequencies consume more operating power both to operate the IC and to charge and discharge the gate of the external FET. This tends to reduce the efficiency at light loads.

4) Higher frequencies may exhibit lower overall efficiency due to more transition losses in the FET; however, this shortcoming can often be nullified by trading some of the inductor and capacitor size benefits for lower-resistance components.

5) High-duty-cycle applications may require lower frequencies to accommodate the controller minimum off-time of 0.4µs. Calculate the maximum oscillator frequency with the following formula: f

OSC MAX )

=

V

V

IN MIN )

IN MIN )

V

SW

V

SW

V

LIM

V

LIM

V

OUT

+

V

D

×

1 t

OFF MIN )

Remember that V

OUT is negative when using this formula.

The oscillator frequency is set by a resistor, R

FREQ

, connected from FREQ to GND. The relationship between f

OSC

(in Hz) and R

FREQ

(in

Ω) is slightly nonlinear, as illustrated in the Typical Operating

Characteristics. Choose the resistor value from the graph and check the oscillator frequency using the following formula: f

OSC

=



(

.

×

10

7

)

+

(

1 92

×

10

1

11

)

×

R

FREQ

(

.

×

10

19

)

×

(

R

FREQ

)

2



External Synchronization (MAX1847 only)

The SYNC input provides external-clock synchronization (if desired). When SYNC is driven with an external clock, the frequency of the clock directly sets the

MAX1847’s switching frequency. A rising clock edge on SYNC is interpreted as a synchronization input. If the sync signal is lost, the internal oscillator takes over at the end of the last cycle, and the frequency is returned to the rate set by R

FREQ

. Choose R

FREQ such that f

OSC

= 0.9 x f

SYNC

.

Choosing Inductance Value

The inductance value determines the operation of the current-mode regulator. Except for low-current applications, most circuits are more efficient and economical operating in continuous mode, which refers to continuous current in the inductor. In continuous mode there is a trade-off between efficiency and transient response.

Higher inductance means lower inductor ripple current, lower peak current, lower switching losses, and, therefore, higher efficiency. Lower inductance means higher inductor ripple current and faster transient response. A reasonable compromise is to choose the ratio of inductor ripple current to average continuous current at minimum duty cycle to be 0.4. Calculate the inductor ripple with the following formula:

I

RIPPLE

×

=

I

LOAD MAX )

(

×

(

V

IN MAX )

V

IN MAX )

V

SW

V

LIM

V

SW

V

LIM

)

V

OUT

+

V

D

)

Then calculate an inductance value:

L = (V

IN(MAX)

/ I

RIPPLE

) x (D

MIN

/ f

OSC

)

Choose the closest standard value. Once again, remember that V

OUT is negative when using this formula.

Determining Peak Inductor Current

The peak inductor current required for a particular output is:

I

LPEAK

= I

LDC

+ (I

LPP

/ 2) where I

LDC is the average DC input current and I

LPP is the inductor peak-to-peak ripple current. The I

LDC and

I

LPP terms are determined as follows:

I

LDC

I

LPP

=

=

I

LOAD

(

V

IN MIN )

V

IN MIN )

×

(

V

OUT

V

SW

V

SW

×

OSC

+

V

D

)

V

LIM

×

V

LIM

(

)

×

V

OUT

(

+

V

OUT

V

D

)

+

V

D

) where L is the selected inductance value. The saturation rating of the selected inductor should meet or exceed the calculated value for I

LPEAK

, although most coil types can be operated up to 20% over their saturation rating without difficulty. In addition to the saturation criteria, the inductor should have as low a series resistance as possible. For continuous inductor current, the power loss in the inductor resistance (P

LR

) is approximated by:

P

LR

~ (I

LOAD x V

OUT

/ V

IN

)

2 x R

L where R

L is the inductor series resistance.

______________________________________________________________________________________ 13

High-Efficiency, Current-Mode,

Inverting PWM Controller

Once the peak inductor current is calculated, the current sense resistor, R

CS

, is determined by:

R

CS

= 85mV / I

LPEAK

For high peak inductor currents (>1A), Kelvin-sensing connections should be used to connect CS and PGND to R

CS

. Connect PGND and GND together at the ground side of R

CS

. A lowpass filter between R

CS and

CS may be required to prevent switching noise from tripping the current-sense comparator at heavy loads.

Connect a 100

Ω resistor between CS and the high side of R

CS

, and connect a 1000pF capacitor between CS and GND.

Checking Slope-Compensation Stability

In a current-mode regulator, the cycle-by-cycle stability is dependent on slope compensation to prevent subharmonic oscillation at duty cycles greater than 50%.

For the MAX1846/MAX1847, the internal slope compensation is optimized for a minimum inductor value (L

MIN

) with respect to duty cycle. For duty cycles greater then

50%, check stability by calculating LMIN using the following equation:

L

MIN

=

[

(

V

IN MIN )

×

[

(

2

×

D

MAX

×

R

CS

) (

/

)

M

S

]

D

MAX

)

] where V

IN(MIN) is the minimum expected input voltage,

M s is the Slope Compensation Ramp (41 mV/µs) and

D

MAX is the maximum expected duty cycle. If L

MIN is larger than L, increase the value of L to the next standard value that is larger than L

MIN to ensure slope compensation stability.

Power MOSFET Selection

The MAX1846/MAX1847 drive a wide variety of P-channel power MOSFETs (PFETs). The best performance, especially with input voltages below 5V, is achieved with low-threshold PFETs that specify on-resistance with a gate-to-source voltage (V

GS

) of 2.7V or less.

When selecting a PFET, key parameters include:

1) Total gate charge (Q

G

)

2) Reverse transfer capacitance (C

RSS

)

3) On-resistance

DS(ON)

)

4) Maximum drain-to-source voltage (V

DS(MAX)

)

5) Minimum threshold voltage (V

TH(MIN)

)

At high switching rates, dynamic characteristics (parameters 1 and 2 above) that predict switching losses may have more impact on efficiency than R

DS(ON

), which predicts DC losses. Q

G includes all capacitance associated with charging the gate. In addition, this parameter helps predict the current needed to drive the gate at the selected operating frequency. The power

MOSFET in an inverting converter must have a high enough voltage rating to handle the input voltage plus the magnitude of the output voltage and any spikes induced by leakage inductance.

Choose R

DS(ON)(MAX) specified at V

GS

< V

IN(MIN) to be one to two times R

CS

. Verify that V

IN(MAX)

< V

GS(MAX) and V

DS(MAX)

> V

IN(MAX)

- V

OUT

+ V

D

. Choose the riseand fall-times (t

R

, t

F

) to be less than 50ns.

Output Capacitor Selection

The output capacitor (C

OUT

) does all the filtering in an inverting converter. The output ripple is created by the variations in the charge stored in the output capacitor with each pulse and the voltage drop across the capacitor’s equivalent series resistance (ESR) caused by the current into and out of the capacitor. There are two properties of the output capacitor that affect ripple voltage: the capacitance value, and the capacitor’s

ESR. The output ripple due to the output capacitor’s value is given by:

V

RIPPLE-C

= (I

LOAD

D

MAX

T

OSC

) / C

OUT

The output ripple due to the output capacitor’s ESR is given by:

V

RIPPLE-R

= I

LPP

R

ESR

These two ripple voltages are additive and the total output ripple is:

V

RIPPLE-T

= V

RIPPLE-C

+ V

RIPPLE-R

The ESR-induced ripple usually dominates this last equation, so typically output capacitor selection is based mostly upon the capacitor’s ESR, voltage rating, and ripple current rating. Use the following formula to determine the maximum ESR for a desired output ripple voltage (V

RIPPLE-D

):

R

ESR

= V

RIPPLE-D

/ I

LPP

Select a capacitor with ESR rating less than R

ESR

. The value of this capacitor is highly dependent on dielectric type, package size, and voltage rating. In general, when choosing a capacitor, it is recommended to use low-ESR capacitor types such as ceramic, organic, or tantalum capacitors. Ensure that the selected capacitor has sufficient margin to safely handle the maximum ripple current

(I

LPP

) and the maximum output voltage.

Choosing Compensation Components

The MAX1846/MAX1847 are externally loop-compensated devices. This provides flexibility in designs to accommodate a variety of applications. Proper com-

14 ______________________________________________________________________________________

High-Efficiency, Current-Mode,

Inverting PWM Controller

pensation of the control loop is important to prevent excessive output ripple and poor efficiency caused by instability. The goal of compensation is to cancel unwanted poles and zeros in the DC-DC converter’s transfer function created by the power-switching and filter elements. More precisely, the objective of compensation is to ensure stability by ensuring that the DC-

DC converter’s phase shift is less than 180° by a safe margin, at the frequency where the loop gain falls below unity. One method for ensuring adequate phase margin is to introduce corresponding zeros and poles in the feedback network to approximate a single-pole response with a -20dB/decade slope all the way to unity-gain crossover.

Calculating Poles and Zeros

The MAX1846/MAX1847 current-mode architecture takes the double pole caused by the inductor and output capacitor and shifts one of these poles to a much higher frequency. This makes loop compensation easier. To compensate these devices, we must know the center frequencies of the right-half plane zero (z

RHP

) and the higher frequency pole (p

OUT2

). Calculate the z

RHP frequency with the following formula:

ZRHP

=



(

1

D

MAX

)

2

×

(

2

π

(

V

IN MIN )

×

V

OUT

×

V

OUT

L

)

)

×

R

LOAD



The calculations for p

OUT2 are very complex. For most applications where V

OUT does not exceed -48V (in a negative sense), the p

OUT2 will not be lower than 1/8th of the oscillator frequency and is generally at a higher frequency than z

RHP

. Therefore: p

OUT2

≥ 0.125

✕ f

OSC

A pole is created by the output capacitor and the load resistance. This pole must also be compensated and its center frequency is given by the formula: p = 1 / (2

π

R

LOAD

C

OUT

)

Finally, there is a zero introduced by the ESR of the output capacitor. This zero is determined from the following equation:

OUT1 z

ESR

= 1 / (2

π

C

OUT

R

ESR

)

Calculating the Required Pole Frequency

To ensure stability of the MAX1846/MAX1847, the introduced pole (P

DOM

) by the compensation network must roll-off the error amplifier gain to 1 before z

RHP or

P

OUT2 occurs. First calculate the DC open-loop gain to determine the frequency of the pole to introduce.

A

DC

=

×

G

M

R

×

R

O

LOAD

×

B

×

(

×

R

CS

R

(

1

×

V

LOAD

D



(

MAX

V

2

)

L

)

)

2

+

( )

×

(

V

( )

T

OSC

×

R

CS

(

1

+

V

OUT

D

MAX

M

S 1



)

) where:

B is the feedback divider attenuation factor =

(-V

OUT

/ V

REF

),

G

M is the error amplifier transconductance =

400 µA/V,

R

O is the error amplifier output resistance = 3 M

Ω,

M

S1 is the slope compensation factor =

[(1.636A / µs)

R

CS

],

R

CS is the selected current sense resistor,

L is the selected inductance value

If z

RHP is at a lower frequency than p

OUT2

, the required dominant pole frequency is given by: p

DOM

= z

RHP

/ A

DC

Otherwise the required dominant pole frequency is: p

DOM

= p

OUT2

/ A

DC

Determining the Compensation Component Values

Using p

DOM

, calculate the compensation capacitor required:

C

COMP

= 1 / (2

π

R

O

✕ p

DOM

)

Select the next largest standard value of capacitor and then calculate the compensation resistor required to cancel out the output-capacitor-induced pole (p

OUT1

) determined previously. A zero is needed to cancel the output-induced pole and the frequency of this zero must equal p

OUT1

. Therefore: z

COMP

= p

OUT1

R

COMP

= R

LOAD

C

OUT

/ C

COMP

Choose the nearest lower standard value of the resistor. Now check the final values selected for the compensation components: p

COMP

= 1 / [2

π

C

COMP x (R

O

+ R

COMP

)]

In order for p

COMP to compensate the loop, the openloop gain must reach unity at a lower frequency than the right-half-plane zero or the second output pole, whichever is lower in frequency. If the second output pole and the right-half-plane zero are close together in frequency, the higher resulting phase shift at unity gain

______________________________________________________________________________________ 15

High-Efficiency, Current-Mode,

Inverting PWM Controller

may require a larger compensation capacitor than calculated. It might take more than a couple of iterations to obtain a suitable combination.

Finally, the zero introduced by the output capacitor’s

ESR must be compensated. This is accomplished by placing a capacitor between REF and FB creating a pole directly in the feedback loop. Calculate the value of this capacitor using the frequency of z

ESR and the selected feedback resistor values with the formula:

C

FB

=

R

ESR

×

C

OUT

×

R

1

+

R

2

R

1

×

R

2

Applications Information

Maximum Output Power

The maximum output power that the MAX1846/MAX1847 can provide depends on the maximum input power available and the circuit’s efficiency:

P

OUT(MAX)

= Efficiency

P

IN(MAX)

Furthermore, the efficiency and input power are both functions of component selection. Efficiency losses can be divided into three categories: 1) resistive losses across the inductor, MOSFET on-resistance, currentsense resistor, and the ESR of the input and output capacitors; 2) switching losses due to the MOSFET’s transition region, and charging the MOSFET’s gate capacitance; and 3) inductor core losses. Typically

80% efficiency can be assumed for initial calculations.

The required input power depends on the inductor current limit, input voltage, output voltage, output current, inductor value, and the switching frequency. The maximum output power is approximated by the following formula:

P

MAX

= [V

IN

- (V

LIM

+ I

LIM x R

DS(ON)

)] x I

LIM x

[1 - (LIR / 2)] x [(-V

OUT

+ V

D

) / (V

IN

- V

SW

- V

LIM

- V

OUT

+ V

D

)] where I

LIM is the peak current limit and LIR is the inductor current-ripple ratio and is calculated by:

LIR = I

LPP

/ I

LDC

Again, remember that V

OUT for the MAX1846/

MAX1847 is negative.

Diode Selection

The MAX1846/MAX1847’s high-switching frequency demands a high-speed rectifier. Schottky diodes are recommended for most applications because of their fast recovery time and low forward voltage. Ensure that the diode’s average current rating exceeds the peak inductor current by using the diode manufacturer’s data.

Additionally, the diode’s reverse breakdown voltage must exceed the potential difference between V

OUT and the input voltage plus the leakage inductance spikes. For high output voltages (-50V or more), Schottky diodes may not be practical because of this voltage requirement. In these cases, use an ultrafast recovery diode with adequate reverse-breakdown voltage.

Input Filter Capacitor

The input capacitor (C

IN

) in inverting converter designs reduces the current peaks drawn from the input supply and reduces noise injection. The source impedance of the input supply largely determines the value of C

IN

.

High source impedance requires high input capacitance, particularly as the input voltage falls. Since inverting converters act as “constant-power” loads to their input supply, input current rises as the input voltage falls. Consequently, in low-input-voltage designs, increasing C

IN and/or lowering its ESR can add as much as 5% to the conversion efficiency.

Bypass Capacitor

In addition to C

IN and C

OUT

, other ceramic bypass capacitors are required with the MAX1846/MAX1847.

Bypass REF to GND with a 0.1µF or larger capacitor.

Bypass V

L to GND with a 0.47µF or larger capacitor. All bypass capacitors should be located as close to their respective pins as possible.

PC Board Layout Guidelines

Good PC board layout and routing are required in highfrequency-switching power supplies to achieve good regulation, high efficiency, and stability. It is strongly recommended that the evaluation kit PC board layouts be followed as closely as possible. Place power components as close together as possible, keeping their traces short, direct, and wide. Avoid interconnecting the ground pins of the power components using vias through an internal ground plane. Instead, keep the power components close together and route them in a

“star” ground configuration using component-side copper, then connect the star ground to internal ground using multiple vias.

Main Application Circuits

The MAX1846/MAX1847 are extremely versatile devices.

Figure 2 shows a generic schematic of the MAX1846.

Table 1 lists component values for several typical applications. These component values also apply to the

MAX1847. The first two applications are featured in the

MAX1846/MAX1847 EV Kit.

16 ______________________________________________________________________________________

High-Efficiency, Current-Mode,

Inverting PWM Controller

V

IN

APPLICATION B

ONLY

C

IN

22k

P

D1

C

COMP

R

R

0.47

µF

COMP

FREQ

3

COMP

2

FREQ

4

REF

VL

1 10

IN

EXT

9

8

CS

MAX1846

PGND

7

GND

6

FB

5

0.1

µF

L1

R

CS

C

FB

R1

R2

V

OUT

C

OUT

NOTE: APPLICATIONS A & B USE POS CAPACITORS. APPLICATIONS C & D USE ALUMINUM ELECTROLYTIC CAPACITORS.

Figure 2. MAX1846 Main Application Circuit

Table 1. Component List for Main Application Circuits

CIRCUIT ID

Input (V)

Output (V)

Output (A)

C

COMP

(µF)

C

IN

(µF)

C

OUT

(µF)

C

FB

(pF)

R1 (k

Ω) (1%)

R2 (k

Ω) (1%)

R

COMP

(k

Ω)

R

CS

(

Ω)

R

FREQ

(k

Ω)

D1

L1 (µH)

P1

A

12

-5

2

0.047

3 x 10

2 x 100

390

40.2

10

8.2

0.02

150

CMSH5-40

10

FDS6685

B

3 to 5.5

-12

0.4

0.22

3 x 22

2 x 47

1200

95.3

10

10

0.02

150

CMSH5-40

10

FDS6375

C

12

-48

0.1

0.068

10

47

1800

383

10

150

0.05

150

CMR1U-02

47

IRFR5410

D

12

-72

0.1

0.1

10

33

1800

576

10

1800

0.05

150

CMR1U-02

82

IRFR5410

______________________________________________________________________________________ 17

High-Efficiency, Current-Mode,

Inverting PWM Controller

SUPPLIER

AVX

Central Semiconductor

Coilcraft

Dale

Fairchild

International

Rectifier

IRC

Kemet

On Semiconductor

Panasonic

Sanyo

Siliconix

Sprague

Sumida

Vitramon

COMPONENT

Capacitors

Diodes

Inductors

Resistors

MOSFETs

MOSFETs

Resistors

Capacitors

MOSFETs, Diodes

Capacitors, Resistors

Capacitors

MOSFETs

Capacitors

Inductors

Resistors

PHONE

803-946-0690

516-435-1110

847-639-6400

402-564-3131

408-721-2181

310-322-3331

512-992-7900

864-963-6300

602-303-5454

201-348-7522

619-661-6835

408-988-8000

603-224-1961

847-956-0666

203-268-6261

Component Suppliers

WEBSITE

www.avxcorp.com

www.centralsemi.com

www.coilcraft.com

www.vishay.com/brands/dale/main.html

www.fairchildsemi.com

www.irf.com

www.irctt.com

www.kemet.com

www.onsemi.com

www.panasonic.com

www.secc.co.jp

www.siliconix.com

www.vishay.com/brands/sprague/main.html

www.remtechcorp.com

www.vishay.com/brands/vitramon/main.html

Note: Please indicate that you are using the MAX1846/MAX1847 when contacting these component suppliers.

Pin Configurations

TRANSISTOR COUNT: 2441

Chip Information

PROCESS TECHNOLOGY: BiCMOS

TOP VIEW

VL 1

FREQ 2

COMP 3

REF 4

FB 5

MAX1846

10-PIN

µMAX

10 IN

9

8

7

POL 1

EXT

VL 2

CS

FREQ

PGND

COMP

3

4

6 GND

REF 5

FB

N.C.

6

7

SHDN 8

MAX1847

16-PIN QSOP

16 SYNC

15 IN

14 EXT

13 CS

12 PGND

11 GND

10 GND

9 N.C.

18 ______________________________________________________________________________________

High-Efficiency, Current-Mode,

Inverting PWM Controller

Package Information

______________________________________________________________________________________ 19

High-Efficiency, Current-Mode,

Inverting PWM Controller

Package Information (continued)

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

© 2001 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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