High-Power LED Driver with Integrated MAX16834 High-Side LED Current Sense and

High-Power LED Driver with Integrated MAX16834 High-Side LED Current Sense and
EVALUATION KIT AVAILABLE
MAX16834
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
General Description
Benefits and Features
The MAX16834 is a current-mode high-brightness LED
(HB LED) driver for boost, boost-buck, SEPIC, and highside buck topologies. In addition to driving an n-channel
power MOSFET switch controlled by the switching controller, it also drives an n-channel PWM dimming switch to
achieve LED PWM dimming. The MAX16834 integrates
all the building blocks necessary to implement a fixed-frequency HB LED driver with wide-range dimming control.
The MAX16834 features constant-frequency peak current-mode control with programmable slope compensation to control the duty cycle of the PWM controller.
A dimming driver designed to drive an external n-channel MOSFET in series with the LED string provides
wide-range dimming control up to 20kHz. In addition to
PWM dimming, the MAX16834 provides analog dimming using a DC input at REFI. The programmable
switching frequency (100kHz to 1MHz) allows design
optimization for efficiency and board space reduction.
A single resistor from RT/SYNC to ground sets the
switching frequency from 100kHz to 1MHz while an
external clock signal at RT/SYNC disables the internal
oscillator and allows the MAX16834 to synchronize to
an external clock. The MAX16834’s integrated highside current-sense amplifier eliminates the need for a
separate high-side LED current-sense amplifier in
boost-buck applications.
• Integration Minimizes BOM for High-Brightness LED
Driver with a Wide Dimming Range Saving Space
and Cost
• Constant-Frequency, Peak Current-Mode Control
with Programmable Slope Compensation
• Integrated PWM Dimming MOSFET Driver
• 3000:1 PWM Dimming/Analog Dimming
• Integrated High-Side Current-Sense Amplifier for
LED Current Sense in Boost-Buck Converter
• Internal 7V Low-Dropout Regulator
The MAX16834 operates over a wide supply range of
4.75V to 28V and includes a 3A sink/source gate driver
for driving a power MOSFET in high-power LED driver
applications. It can also operate at input voltages
greater than 28V in boost configuration with an external
voltage clamp. The MAX16834 is also suitable for DCDC converter applications such as boost or boostbuck. Additional features include external enable/
disable input, an on-chip oscillator, fault indicator output (FLT) for LED open/short or overtemperature conditions, and an overvoltage protection sense input
(OVP+) for true overvoltage protection.
The MAX16834 is available in a thermally enhanced
4mm x 4mm, 20-pin TQFN-EP package and in a thermally enhanced 20-pin TSSOP-EP package and is specified
over the automotive -40°C to +125°C temperature range.
Applications
• Single-String LED LCD Backlighting
• Simple Optimization for Efficiency, Board Space and
Input Operating Range
• Supports Boost, Buck-Boost, SEPIC, and HighSide Buck Topologies
• 100kHz to 1MHz Programmable High-Frequency
Operation
• External Clock Synchronization Input
• Wide Input Operating Voltage Range (4.75V to 28V)
• Works for Input Voltage > 28V with External
Voltage Clamp on VIN for Boost Converter
• 20-Pin TQFN-EP and TSSOP-EP Packages
• Protection Features Increase System Reliability
• Programmable UVLO
• Programmable True Differential Overvoltage
Protection
• Fault Output (FLT) for Overvoltage, Overcurrent,
and Thermal-Warning Faults
Simplified Application Circuit
VIN
BOOST LED DRIVER
LED+
NDRV
IN
LEDs
MAX16834
ON
OFF
PWMDIM
ANALOG
DIM
CS
REFI
DIMOUT
PGND
SENSE+
LED-
• Automotive Rear and Front Lighting
• Projection System RGB LED Light Sources
• Architectural and Decorative Lighting (MR16, M111)
• Spot and Ambient Lights
• DC-DC Boost/Boost-Buck Converters
19-4235; Rev 5; 1/15
Ordering Information and Pin Configurations appear at end
of data sheet.
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Absolute Maximum Ratings
IN, HV, LV to SGND................................................-0.3V to +30V
OVP+, SENSE+, DIMOUT, CLV to SGND ..............-0.3V to +30V
SENSE+ to LV........................................................-0.3V to +0.3V
HV, IN to LV ............................................................-0.3V to +30V
OVP+, CLV, DIMOUT to LV ......................................-0.3V to +6V
PGND to SGND .....................................................-0.3V to +0.3V
VCC to SGND..........................................................-0.3V to +12V
NDRV to PGND...........................................-0.3V to (VCC + 0.3V)
All Other Pins to SGND.............................................-0.3V to +6V
NDRV Continuous Current................................................±50mA
DIMOUT Continuous Current..............................................±2mA
VCC Short-Circuit Current to SGND Duration ...........................1s
Continuous Power Dissipation (TA = +70°C)
QFN, TQFN (derate 25.6mW/°C* above +70°C).........2051mW
TSSOP (derate 26.5mW/°C above +70°C) .....................2122mW
Operating Temperature Range .........................-40°C to +125°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Soldering Temperature (reflow) .......................................+260°C
*As per JEDEC51 standard (multilayer board).
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
Package Thermal Characteristics (Note 1)
Junction-to-Ambient Thermal Resistance (θJA)
QFN, TQFN ........................................................................39°C/W
TSSOP .............................................................................37.7°C/W
Junction-to-Case Thermal Resistance (θJC)
QFN, TQFN .....................................................................6°C/W
TSSOP .............................................................................2°C/W
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer
board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Electrical Characteristics
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7μF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
SYMBOL
Input Voltage Range
VIN
Quiescent Supply Current
IQ
Shutdown Supply Current
ISHDN
CONDITIONS
MIN
TYP
MAX
28
V
Excluding ILED
6
10
mA
VUVEN = 0V
30
60
μA
7
7.7
V
4.75
UNITS
INTERNAL LINEAR REGULATOR (VCC)
Output Voltage
VCC
0 ≤ ICC ≤ 50mA, 9.5V ≤ VIN ≤ 28V
Dropout Voltage
VDO
ICC = 35mA (Note 2)
Short-Circuit Current
6.3
0.65
VCC = 0V, VIN = 12V
80
0 ≤ ICLV ≤ 2mA, 6V ≤ VHV ≤ 28V,
6V ≤ V(HV-LV) ≤ 22V
4.7
1
V
300
mA
5.3
V
LINEAR REGULATOR (CLV)
Output Voltage
(VCLV - VLV)
Dropout Voltage
VDO
Short-Circuit Current
5
ICLV = 2mA, 0 ≤ VLV ≤ 23.3V (Note 3)
VCLV = 12V, VIN = 12V, VHV = 24V
2.2
0 ≤ IREF ≤ 1mA, 4.75V ≤ VIN ≤ 28V
3.625
0.5
V
10
mA
REFERENCE VOLTAGE (REF)
Output Voltage
VREF
REF Short-Circuit Current
VREF = 0V
3.70
3.775
30
V
mA
UNDERVOLTAGE LOCKOUT/ENABLE INPUT (UVEN)
UVEN On Threshold Voltage
VUVEN_THUP
1.395
UVEN Threshold Voltage
Hysteresis
Input Leakage Current
ILEAK
VUVEN = 0V
1.435
1.475
V
200
mV
1
μA
PWMDIM
PWMDIM On Threshold Voltage
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VPWMDIM
1.395
1.435
1.475
V
Maxim Integrated | 2
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Electrical Characteristics (continued)
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7μF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
PWMDIM Threshold Voltage
Hysteresis
Input Leakage Current
VPWMDIM = 0V
TYP
MAX
UNITS
200
mV
1
μA
OSCILLATOR
RRT/SYNC = 5kΩ
0.9
1
1.1
MHz
RRT/SYNC = 25kΩ
180
200
220
kHz
Oscillator Frequency Range
(Note 4)
100
1000
kHz
External Sync Input Clock High
Threshold
(Note 4)
2
External Sync Input Clock Low
Threshold
(Note 4)
External Sync Input High Pulse
Width
(Note 4)
Oscillator Frequency
fOSC
V
0.4
200
Maximum External Sync Period
V
ns
50
μs
SLOPE COMPENSATION (SC)
SC Pullup Current
ISCPU
VSC = 100mV
SC Discharge Resistance
RSCD
VSC = 100mV
80
100
120
μA
Ω
8
REFI
REFI Input Bias Current
VREFI = 1V
REFI Input Common-Mode Range
(Note 4)
1
0
μA
2
V
250
μA
mV
SENSE+
SENSE+ Input Bias Current
(VSENSE+ - VLV) = 100mV
HIGH-SIDE LED CURRENT-SENSE AMPLIFIER (VSENSE+ - VLV)
Input Offset Voltage
Voltage Gain
AV
3dB Bandwidth
VLV > 5V, (VSENSE+ - VLV) = 5mV
-2.4
0
+2.4
VLV > 5V, (VSENSE+ - VLV) = 0.2V
9.7
9.9
10.1
V/V
(VSENSE+ - VLV) = 0.1V, no load
1.8
MHz
(VSENSE+ - VLV) = 0.02V, no load
600
kHz
LOW-SIDE LED CURRENT-SENSE AMPLIFIER
Input Offset Voltage
Voltage Gain
AV
VLV < 1V, (VSENSE+ - VLV) = 0V
-2
0
+2
mV
VLV < 1V, (VSENSE+ - VLV) = 0.2V
9.7
9.9
10.1
V/V
3dB Bandwidth
600
kHz
CURRENT ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
Transconductance
gm
Open-Loop DC Gain
AV
VCOMP = 2V, VPWMDIM = 5V
-10
VCOMP
500
600
μS
+10
mV
2.5
V
60
Input Offset Voltage
COMP Voltage Range
400
(Note 4)
0
0.4
dB
PWM COMPARATOR
Input Offset Voltage
Propagation Delay
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0.6
tPD
50mV overdrive
0.65
40
0.70
V
ns
Maxim Integrated | 3
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Electrical Characteristics (continued)
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7μF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
Minimum On-Time
SYMBOL
tON(MIN)
Duty Cycle
CONDITIONS
MIN
On-time includes blanking time
(Note 4)
TYP
MAX
100
90
UNITS
ns
99.5
%
CURRENT PEAK LIMIT COMPARATOR
Trip Threshold Voltage
0.25
Propagation Delay
0.3
50mV overdrive with respect to NDRV
0.35
40
V
ns
OVERVOLTAGE PROTECTION INPUT (OVP+)
OVP+ On Threshold Voltage
VOVP_ON
1.375
OVP+ Hysteresis
OVP+ Input Leakage Current
1.435
1.495
200
(VOVP - VLV) = 1.235V
-1
Off Threshold
VCLV - VLV
4.0
On Threshold
VCLV - VLV
4.1
Error Reject Blankout
fOSC = 500kHz
V
mV
+1
μA
4.3
4.6
V
4.4
4.7
HIGH-SIDE LED SHORT COMPARATOR
256
V
μs
LOW-SIDE LED SHORT COMPARATOR
Off Threshold
0.27
0.30
Error Reject Blankout
0.33
V
5
μs
8.2
ms
3
A
HICCUP TIMER
Hiccup Time
fOSC = 500kHz
GATE-DRIVER OUTPUT (NDRV)
NDRV Peak Pullup Current
VCC = 7V
NDRV Peak Pulldown Current
VCC = 7V
p-Channel MOSFET RDSON
(VCC - VNDRV) = 0.1V
1.2
1.9
Ω
n-Channel MOSFET RDSON
VNDRV = 0.1V
0.9
1.7
Ω
3
A
DIMOUT
DIMOUT Peak Pullup Current
(VCLV - VLV) = 5V
25
50
mA
DIMOUT Peak Pulldown Current
(VCLV - VLV) = 5V
25
50
mA
p-Channel MOSFET RDSON
(VCLV - VDIMOUT) = 0.1V
31
Ω
n-Channel MOSFET RDSON
(VDIMOUT - VLV) = 0.1V
25
Ω
200
ns
PWMDIM to DIMOUT
Propagation Delay
FAULT FLAG (FLT)
FLT Pulldown Current
VFLT = 0.2V
FLT Leakage Current
VFLT = 1.0V
Thermal Warning On Threshold
2
5
10
mA
1
μA
+140
°C
20
°C
Thermal Warning Threshold
Hysteresis
Note 2: Dropout voltage is defined as VIN - VCC, when VCC is 100mV below the value of VCC for VIN = 9.5V.
Note 3: Dropout is defined as VHV - VCLV, when VCLV is 100mV below the value of VCLV for VHV = 8V.
Note 4: Not production tested. Guaranteed by design.
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Maxim Integrated | 4
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Typical Operating Characteristics
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7μF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA = +25°C, unless otherwise noted.)
VREF vs. SUPPLY VOLTAGE
3.74
3.72
3.75
VIN = 12V
3.7015
3.7010
3.70
3.68
VREF (V)
3.70
VREF (V)
VREF (V)
VREF vs. IREF
3.7020
MAX16834 toc02
MAX16834 toc01
3.80
MAX16834 toc03
VREF vs. TEMPERATURE
3.65
3.66
3.7005
3.7000
3.6995
3.60
3.64
3.6990
3.55
3.62
3.6985
VIN = 12V
3.50
12
16
20
24
SUPPLY VOLTAGE (V)
SUPPLY CURRENT
vs. SUPPLY VOLTAGE
SUPPLY CURRENT
vs. TEMPERATURE
12
10
8
6
MAX16834 toc05
8
6
5
1
20
24
SWITCHING FREQUENCY
vs. TEMPERATURE
TEMPERATURE (°C)
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7.16
7.14
7.12
7.10
7.08
7.06
7.04
7.02
7.00
6.98
6.96
6.94
6.92
6.90
10
9
1000
VCC vs. ICC
VIN = 12V
7.2
VIN = 12V
TA = +125°C
TA = +100°C
7.1
VCC (V)
VCC (V)
MAX16834 toc07
-40 -25 -10 5 20 35 50 65 80 95 110 125
8
SWITCHING FREQUENCY (kHz)
VCC vs. ICC
VIN = 12V
7
100
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
605
604
603
602
601
600
599
598
597
596
595
594
593
592
591
590
6
VIN = 12V
1
-40 -25 -10 5 20 35 50 65 80 95 110 125
28
MAX16834 toc08
16
5
10
VIN = 12V
PWMDIM = 0
0
12
4
3
2
8
3
4
2
4
2
7
4
0
1
100
RT (kΩ)
14
0
RT vs. SWITCHING FREQUENCY
9
SUPPLY CURRENT (mA)
16
28
IREF (mA)
10
MAX16834 toc04
PWMDIM = 0
18
SWITCHING FREQUENCY (kHz)
8
TEMPERATURE (°C)
20
SUPPLY CURRENT (mA)
3.6980
4
MAX16834 toc06
-40 -25 -10 5 20 35 50 65 80 95 110 125
MAX16834 toc09
3.60
7.0
TA = +25°C
TA = -40°C
6.9
6.8
0
10 20 30 40 50 60 70 80 90 100
ICC (mA)
0
10 20 30 40 50 60 70 80 90 100
ICC (mA)
Maxim Integrated | 5
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Typical Operating Characteristics (continued)
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7μF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA = +25°C, unless otherwise noted.)
NDRV RISE/FALL TIME
VCC vs. VIN
vs. CAPACITANCE
TA = +25°C
TA = +125°C
TA = -40°C
7.14
VCC (V)
VIN = 12V
40
NDRV RISE TIME (ns)
7.16
50
7.12
7.10
7.08
7.06
MAX16834 toc11
7.18
MAX16834 toc10
7.20
30
RISE TIME
20
FALL TIME
10
7.04
7.02
0
7.00
14
22
18
0
26
1
2
4
5
6
7
8
9
10
VCLV vs. VHV
VCLV vs. ICLV
5.10
MAX16834 toc12
5.50
5.00
4.50
VIN = 12V
5.09
5.08
4.00
5.07
VCLV (V)
3.50
VCLV (V)
3
CAPACITANCE (nF)
VIN (V)
MAX16834 toc13
10
6
3.00
2.50
5.06
5.05
5.04
2.00
5.03
1.50
5.02
1.00
0.50
5.01
VIN = 12V
5.00
0
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
6
10
14
18
22
26
VHV (V)
ICLV (mA)
Pin Description
PIN
NAME
FUNCTION
QFN, TQFN
TSSOP
1
3
OVP+
LED-String Overvoltage Protection Input. Connect a resistive voltage-divider between the
positive output, OVP+, and LV to set the overvoltage threshold. OVP+ has a 1.435V threshold
voltage with a 200mV hysteresis.
2
4
SGND
Signal Ground
3
5
COMP
Error-Amplifier Output. Connect an RC network from COMP to SGND for stable operation. See
the Feedback Compensation section.
4
6
REF
3.7V Reference Output Voltage. Bypass REF to SGND with a 0.1μF to 0.22μF ceramic
5
7
REFI
Current Reference Input. VREFI provides a reference voltage for the current-sense amplifier to
set the LED current.
6
8
SC
Current-Mode Slope Compensation Setting. Connect to an appropriate external capacitor from
SC to SGND to generate a ramp signal for stable operation.
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Maxim Integrated | 6
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Pin Description (continued)
PIN
QFN, TQFN
TSSOP
7
9
NAME
FLT
FUNCTION
Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (FLT) section.
Resistor-Programmable Switching Frequency Setting/Sync Control Input. Connect a resistor
RT/SYNC from RT/SYNC to SGND to set the switching frequency. Drive RT/SYNC to synchronize the
switching frequency with an external clock.
8
10
9
11
10
12
11
13
CS
12
14
PGND
Power Ground
13
15
NDRV
External n-Channel Gate-Driver Output
14
16
VCC
15
17
IN
16
18
HV
17
19
CLV
18
20
DIMOUT
19
1
LV
20
2
SENSE+
—
—
EP
UVEN
Undervoltage-Lockout (UVLO) Threshold/Enable Input. UVEN is a dual-function adjustable
UVLO threshold input with an enable feature. Connect UVEN to VIN through a resistive voltagedivider to program the UVLO threshold. Observe the absolute maximum value for this pin.
PWMDIM PWM Dimming Input. Connect to an external PWM signal for dimming operation.
Current-Sense Amplifier Positive Input. Connect a resistor from CS to PGND to set the inductor
peak current limit.
7V Low-Dropout Voltage Regulator. Bypass to PGND with at least a 1μF low-ESR ceramic
capacitor. VCC provides power to the n-channel gate driver (NDRV).
Positive Power-Supply Input. Bypass to PGND with at least a 0.1μF ceramic capacitor.
High-Side Positive Supply Input Referred to LV. HV provides power to high-side linear regulator
5V High-Side Regulator Output. CLV provides power to the dimming MOSFET driver. Connect a
0.1μF to 1μF ceramic capacitor from CLV to LV for stable operation.
External Dimming MOSFET Gate Driver. DIMOUT is capable of sinking/sourcing 50mA.
High-Side Reference Voltage Input. Connect to SGND for boost configuration. Connect to IN for
boost-buck configuration.
LED Current-Sense Positive Input. Connect a bypass capacitor of at least 0.1μF between
SENSE+ and LV close to the IC.
Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power
dissipation. Do not use as the main IC ground connection. EP must be connected to SGND.
Detailed Description
The MAX16834 is a current-mode, high-brightness LED
(HB LED) driver designed to control a single-string LED
current regulator with two external n-channel MOSFETs.
The MAX16834 integrates all the building blocks necessary to implement a fixed-frequency HB LED driver
with wide-range dimming control. The MAX16834
allows implementation of different converter topologies
such as SEPIC, boost, boost-buck, or high-side buck
current regulator.
The MAX16834 features a constant-frequency, peak-current-mode control with programmable slope compensation to control the duty cycle of the PWM controller. A
dimming driver offers a wide-range dimming control for
the external n-channel MOSFET in series with the LED
string. In addition to PWM dimming, the MAX16834
allows for analog dimming of LED current.
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The MAX16834 switching frequency (100kHz to 1MHz)
is adjustable using a single resistor from RT/SYNC. The
MAX16834 disables the internal oscillator and synchronizes if an external clock is applied to RT/SYNC. The
switching MOSFET driver sinks and sources up to 3A,
making it suitable for high-power MOSFETs driving in
HB LED applications, and the dimming control allows
for wide PWM dimming at frequencies up to 20kHz.
The MAX16834 is suitable for boost and boost-buck
LED drivers (Figures 2 and 3).
The MAX16834 alone operates over a wide 4.75V to
28V supply range. With a voltage clamp that limits the
IN pin voltage to less than 28V, it can operate in boost
configuration for input voltages greater than 28V.
Additional features include external enable/disable
input, an on-chip oscillator, fault indicator output (FLT)
for LED open/short or overtemperature conditions, and
an overvoltage protection circuit for true differential
overvoltage protection (Figure 1).
Maxim Integrated | 7
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
IN
REF
TO
INTERNAL
CIRCUITRY
REFERENCE
TEMPERATURE
SENSE
OT
SGND
VCC
7V
LDO
UVEN
UVLO
VBG
S
Q
R
RT/SYNC
OSC
SC
0.6V
5kΩ
CS
NDRV
RAMP
GENERATOR
PWM
COMP
OR
AND
CURRENT-LIMIT
COMPARATOR
BLANK
NDRVB
PGND
NDRVB
0.3V
VREF
REFI
FLTB
LPF
FLT
ERROR
AMPLIFIER
SENSE+
AV = 9.9
PWMDIM
gm
VLV
FLTA
AND
AND
OT
LED CURRENTSENSE AMPLIFIERS
CLV
COMP
DIMOUT
HV
HIGH-SIDE
5V
REGULATOR
VBG
LV REFERENCE
SWITCH
LV
VLV
REFHI
VIN
VBG
PWMDIM
VBG
FLTB
AND
VLV
0.3V
VHV
OVP+
FLTA
VLV
128 TOSC
ERROR
REJECT
DELAY
4.3V
SENSE+
VREF
REFHI
FLTB
4096 TOSC
HICCUP
TIMER
5μs ERROR
REJECT
DELAY
MAX16834
VBG
VLV
Figure 1. Internal Block Diagram
www.maximintegrated.com
Maxim Integrated | 8
MAX16834
The MAX16834 is also suitable for DC-DC converter
applications such as boost or boost-buck (Figures 6
and 7). Other applications include boost LED drivers
with automotive load dump protection (Figure 4) and
high-side buck LED drivers (Figure 5).
Undervoltage Lockout/Enable
The MAX16834 features an adjustable UVLO using the
enable input (UVEN). Connect UVEN to VIN through a
resistive divider to set the UVLO threshold. The
MAX16834 is enabled when the VUVEN exceeds the
1.435V (typ) threshold. See the Setting the UVLO
Threshold section for more information.
UVEN also functions as an enable/disable input to the
device. Drive UVEN low to disable the output and high
to enable the output.
Reference Voltage (REF)
The MAX16834 features a 3.7V reference output, REF.
REF provides power to most of the internal circuit blocks
except for the output drivers and is capable of sourcing
1mA to external circuits. Connect a 0.1μF to 0.22μF
ceramic capacitor from REF to SGND. Connect REF to
REFI through a resistive divider to set the LED current.
Reference Input (REFI)
The output current is proportional to the voltage at
REFI. Applying an external DC voltage at REFI or using
a potentiometer from REF to SGND allows analog dimming of the output current.
High-Side Reference Voltage Input (LV)
LV is a reference input. Connect LV to SGND for boost
and SEPIC topologies. Connect LV to IN for boost-buck
and high-side buck topologies.
Dimming Driver Regulator
Input Voltage (HV)
The voltage at HV provides the input voltage for the
dimming driver regulator. For boost or SEPIC topology,
connect HV either to IN or to VCC. For boost-buck, connect HV to a voltage higher than IN. The voltage at HV
must not exceed 28V with respect to PGND. For the
high-side buck, connect HV to IN.
Dimming MOSFET Driver (DIMOUT)
The MAX16834 requires an external n-channel MOSFET
for PWM dimming. Connect the gate of the MOSFET to
the output of the dimming driver, DIMOUT, for normal
operation. The dimming driver is capable of sinking or
sourcing up to 50mA of current.
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High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
n-Channel MOSFET Switch Driver (NDRV)
The MAX16834 drives an external n-channel switching
MOSFET. NDRV swings between V CC and PGND.
NDRV is capable of sinking/sourcing 3A of peak current,
allowing the MAX16834 to switch MOSFETs in highpower applications. The average current demanded
from the supply to drive the external MOSFET depends
on the total gate charge (Q G ) and the operating
frequency of the converter, fSW. Use the following equation to calculate the driver supply current I NDRV
required for the switching MOSFET:
INDRV = QG x fSW
Pulse Dimming Inputs (PWMDIM)
The MAX16834 offers a dimming input (PWMDIM) for
pulse-width modulating the output current. PWM dimming can be achieved by driving PWMDIM with a pulsating voltage source. When the voltage at PWMDIM is
greater than 1.435V, the PWM dimming MOSFET turns
on and when the voltage on PWMDIM is below 1.235V,
the PWM dimming MOSFET turns off.
High-Side Linear Regulator (VCLV)
The MAX16834’s 5V high-side regulator (CLV) powers
up the dimming MOSFET driver. VCLV is measured with
respect to LV and sources up to 2mA of current.
Bypass CLV to LV with a 0.1μF to 1μF low-ESR ceramic
capacitor. The maximum voltage on CLV with respect
to PGND must not exceed 28V. This limits the input voltage for boost-buck topology.
Low-Side Linear Regulator (VCC)
The MAX16834’s 7V low-side linear regulator (VCC) powers up the switching MOSFET driver with sourcing capability of up to 50mA. Use at least a 1μF low-ESR ceramic
capacitor from VCC to PGND for stable operation.
LED Current-Sense Input (SENSE+)
The differential voltage from SENSE+ to LV is fed to an
internal current-sense amplifier. This amplified signal is
then connected to the negative input of the transconductance error amplifier. The voltage gain factor of this
amplifier is 9.9 (typ).
Whenever VLV is greater than 5V, the input impedance
of the LED current-sense amplifier seen at the SENSE+
pin is 1kΩ ±30%. In that condition, a bias current of
20μA (±30%) is pulled from SENSE+, in addition to the
current due to the 1kΩ resistor. When VLV is less than
1V, the amplifier input (SENSE+ pin) is in high impedance and the bias current of 20μA (±30%) is pushed
out of that pin.
Always have a bypass capacitor of at least 0.1μF value
between SENSE+ and LV and close to the IC.
Maxim Integrated | 9
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Internal Transconductance Error Amplifier
The MAX16834 has a built-in transconductance amplifier used to amplify the error signal inside the feedback
loop. The amplified current-sense signal is connected
to the negative input of the gm amplifier with the current
reference connected to REFI. The output of the op amp
is controlled by the input at PWMDIM. When the signal
at PWMDIM is high, the output of the op amp connects
to COMP; when the signal at PWMDIM is low, the output of the op amp disconnects from COMP to preserve
the charge on the compensation capacitor. When the
voltage at PWMDIM goes high, the voltage on the compensation capacitor forces the converter into a steady
state. COMP is connected to the negative input of the
PWM comparator with CMOS inputs, which draw very
little current from the compensation capacitor at COMP
and thus prevent discharge of the compensation
capacitor when the PWMDIM input is low.
Internal Oscillator
The internal oscillator of the MAX16834 is programmable from 100kHz to 1MHz using a single resistor at
RT/SYNC. Use the following formula to calculate the
switching frequency:
fOSC (kHz) =
5000kΩ
× (kHz)
RT(kΩ)
where RT is the resistor from RT/SYNC to SGND.
The MAX16834 synchronizes to an external clock signal
at RT/SYNC. The application of an external clock disables the internal oscillator allowing the MAX16834 to
use the external clock for switching operation. The
internal oscillator is enabled if the external clock is
absent for more than 50μs. The synchronizing pulse
minimum width for proper synchronization is 200ns.
Switching MOSFET
Current-Sense Input (CS)
CS is part of the current-mode control loop. The switching control uses the voltage on CS, set by RCS, to terminate the on pulse width of the switching cycle, thus
achieving peak current-mode control. Internal leadingedge blanking is provided to prevent premature turn-off
of the switching MOSFET in each switching cycle.
Slope Compensation (SC)
The MAX16834 uses an internal-ramp generator for
slope compensation. The ramp signal also resets at the
beginning of each cycle and slews at the rate pro-
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grammed by the external capacitor connected at SC.
The current source charging the capacitor is 100μA.
Overvoltage Protection (OVP+)
OVP+ sets the overvoltage threshold limit across the
LEDs. Use a resistive divider between output OVP+
and LV to set the overvoltage threshold limit. An internal
overvoltage protection comparator senses the differential voltage across OVP+ and LV. If the differential voltage is greater than 1.435V, NDRV is disabled and FLT
asserts. When the differential voltage drops by 200mV,
NDRV is enabled and FLT deasserts. The PWM dimming MOSFET is still controlled by the PWMDIM input.
Fault Indicator (FLT)
The MAX16834 features an active-low, open-drain fault
indicator (FLT). FLT asserts when one of the following
occurs:
1) Overvoltage across the LED string
2) Short-circuit condition across the LED string, or
3) Overtemperature condition
When the output voltage drops below the overvoltage
set point minus the hysteresis, FLT deasserts. Similarly
during the short-circuit period, the fault signal
deasserts when the dimming MOSFET is on, which
happens every hiccup cycle during short circuit. During
overtemperature fault, the FLT signal is the inverse of
the PWM input.
Applications Information
Setting the UVLO Threshold
The UVLO threshold is set by resistors R1 and R2 (see
Figure 2). The MAX16834 turns on when the voltage
across R2 exceeds 1.435V, the UVLO threshold. Use
the following equation to set the desired UVLO threshold:
VUVEN = 1.435V(R1 + R2) R2
In a typical application, use a 10kΩ resistor for R2 and
then calculate R1 based on the desired UVLO threshold.
Setting the Overvoltage Threshold
The overvoltage threshold is set by resistors R4 and R9
(see Figure 2). The overvoltage circuit in the MAX16834
is activated when the voltage on OVP+ with respect to
LV exceeds 1.435V. Use the following equation to set
the desired overvoltage threshold:
VOV = 1.435V(R4 + R9) R9
Maxim Integrated | 10
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
VIN
C1
L1
R1
LV
D1
FLT
LED+
C3
IN
Q1
NDRV
LEDs
UVEN
HV
C2
SC
R4
CS
ON
MAX16834
OFF
PWMDIM
R3
LED-
RT/SYNC
C5
Q2
DIMOUT
R2
VCC
C4
REF
SENSE+
OVP+
R6
CLV
R5
REFI
COMP
SGND
PGND
R8
R9
R10
C8
C7
C6
R7
Figure 2. Boost LED Driver
Programming the LED Current
The LED current is programmed using the voltage on
REFI and the LED current-sense resistor R10 (see
Figure 2). The current is given by:
ILED =
VREF × R5
( A)
R10 × (R6 + R5) × 9.9
where VREF is 3.7V and the resistors R5, R6, and R10
are in ohms. The regulation voltage on the LED currentsense resistor must not exceed 0.3V to prevent activation of the LED short-circuit protection circuit.
Boost Configuration
In the boost converter (Figure 2), the average inductor
current varies with the line voltage. The maximum average current occurs at the lowest line voltage. For the
boost converter, the average inductor current is equal
to the input current.
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Calculate maximum duty cycle using the below equation.
D MAX =
VLED + VD − VINMIN
VLED + VD − VFET
where VLED is the forward voltage of the LED string in
volts, VD is the forward drop of the rectifier diode D1 in
volts (approximately 0.6V), VINMIN is the minimum input
supply voltage in volts, and VFET is the average drain to
source voltage of the MOSFET Q1 in volts when it is on.
Use an approximate value of 0.2V initially to calculate
DMAX. A more accurate value of the maximum duty
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.
Use the following equations to calculate the maximum
average inductor current ILAVG, peak-to-peak inductor
current ripple ΔIL, and the peak inductor current ILP in
amperes:
Maxim Integrated | 11
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
ILED
IL AVG =
1 − D MAX
Allowing the peak-to-peak inductor ripple ΔI L to be
±30% of the average inductor current:
ΔIL = IL AVG × 0.3 × 2
Allowing the peak-to-peak inductor ripple (ΔIL) to be
±30% of the average inductor current:
ΔIL = IL AVG × 0.3 × 2
and
IL P = IL AVG +
ΔIL
2
The inductance value (L) of the inductor L1 in henries
(H) is calculated as:
L=
(VINMIN − VFET ) × D MAX
fSW × ΔIL
IL P = IL AVG +
ΔIL
2
The inductance value (L) of the inductor L1 in henries is
calculated as:
L=
(VINMIN − VFET ) × D MAX
fSW × ΔIL
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and ΔIL is in amperes. Choose an
inductor that has a minimum inductance greater than
the calculated value.
Peak Current-Sense Resistor (R8)
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and ΔIL is in amperes.
Choose an inductor that has a minimum inductance
greater than the calculated value. The current rating of
the inductor should be higher than ILP at the operating
temperature.
The value of the switch current-sense resistor R8 for the
boost and boost-buck configurations is calculated as
follows:
Boost-Buck Configuration
where 0.25V is the minimum peak current-sense threshold, ILP is the peak inductor current in amperes, and
the factor 1.25 provides a 25% margin to account for
tolerances. The worst cycle-by-cycle current limiter triggers at 350mV (max). The ISAT of the inductor should
be higher than 0.35V/R8.
In the boost-buck LED driver (Figure 3), the average
inductor current is equal to the input current plus the
LED current.
Calculate maximum duty cycle using the following
equation:
D MAX =
VLED + VD
VLED + VD + VINMIN − VFET
where VLED is the forward voltage of the LED string in
volts, VD is the forward drop of the rectifier diode D1
(approximately 0.6V) in volts, VINMIN is the minimum
input supply voltage in volts, and VFET is the average
drain to source voltage of the MOSFET Q1 in volts when
it is on. Use an approximate value of 0.2V initially to calculate DMAX. A more accurate value of maximum duty
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.
Use the below equations to calculate the maximum
average inductor current ILAVG, peak-to-peak inductor
current ripple ΔIL, and the peak inductor current ILP in
amperes:
IL AVG =
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ILED
1 − D MAX
R8 =
0.25
Ω
(IL P × 1.25)
Output Capacitor
The function of the output capacitor is to reduce the
output ripple to acceptable levels. The ESR, ESL, and
the bulk capacitance of the output capacitor contribute
to the output ripple. In most applications, the output
ESR and ESL effects can be dramatically reduced by
using low-ESR ceramic capacitors. To reduce the ESL
and ESR effects, connect multiple ceramic capacitors
in parallel to achieve the required bulk capacitance. To
minimize audible noise generated by the ceramic
capacitors during PWM dimming, it may be necessary
to minimize the number of ceramic capacitors on the
output. In these cases an additional electrolytic or tantalum capacitor provides most of the bulk capacitance.
Boost and boost-buck configurations: The calculation of the output capacitance is the same for both
boost and boost-buck configurations. The output ripple
is caused by the ESR and the bulk capacitance of the
output capacitor if the ESL effect is considered negligible. For simplicity, assume that the contributions from
Maxim Integrated | 12
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
VIN
C1
L1
D1
R1
LV
HV
IN
NDRV
UVEN
C2
SC
LED+
Q1
LEDs
CS
MAX16834
R3
C3
ON
R4
OFF
PWMDIM
LED-
RT/SYNC
C5
R2
VCC
DIMOUT
REF
SENSE+
Q2
C4
R6
OVP+
R5
REFI
FLT
SGND
CLV
COMP
PGND
R8
R9
R10
C8
C7
C6
R7
VIN
Figure 3. Boost-Buck LED Driver (VLED+ < 28V)
ESR and the bulk capacitance are equal, allowing 50%
of the ripple for the bulk capacitance. The capacitance
is given by:
C OUT ≥
ILED × 2 × D MAX
ΔVOUTRIPPLE × fSW
where ILED is in amperes, COUT is in farads, fSW is in
hertz, and ΔVOUTRIPPLE is in volts. The remaining 50%
of allowable ripple is for the ESR of the output capacitor. Based on this, the ESR of the output capacitor is
given by:
ESR COUT <
ΔVOUTRIPPLE (Ω)
(IL P × 2)
where ILP is the peak inductor current in amperes.
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Use the below equation to calculate the RMS current
rating of the output capacitor:
ICOUT(RMS) =
(IL AVG × (1 - DMAX )) 2 × DMAX
+(IL AVG × DMAX ) 2 × (1 - DMAX )
Input Capacitor
The input filter capacitor bypasses the ripple current
drawn by the converter and reduces the amplitude of
high-frequency current conducted to the input supply.
The ESR, ESL, and the bulk capacitance of the input
capacitor contribute to the input ripple. Use a low-ESR
input capacitor that can handle the maximum input
RMS ripple current from the converter.
For the boost configuration, the input current is the
same as the inductor current. For boost-buck
Maxim Integrated | 13
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
configuration, the input current is the inductor current
minus the LED current. But for both configurations, the
ripple current that the input filter capacitor has to supply is the same as the inductor ripple current with the
condition that the output filter capacitor should be connected to ground for boost-buck configuration. This
reduces the size of the input capacitor, as the inductor
current is continuous with maximum ±30% ripple.
Neglecting the effect of LED current ripple, the calculation of the input capacitor for boost as well as boostbuck configurations is the same.
Neglecting the effect of the ESL, the ESR, and the bulk
capacitance at the input contributes to the input voltage
ripple. For simplicity, assume that the contribution from
the ESR and the bulk capacitance is equal. This allows
50% of the ripple for the bulk capacitance. The capacitance is given by:
CIN ≥
ΔIL
4 × ΔVIN × fSW
where ΔIL is in amperes, CIN is in farads, fSW is in hertz,
and ΔVIN is in volts. The remaining 50% of allowable
ripple is for the ESR of the output capacitor. Based on
this, the ESR of the input capacitor is given by:
source and discharged at the beginning of each switching cycle to generate the slope compensation ramp.
The value of the slope compensation capacitor C2 is
calculated as shown below:
Boost configuration:
C2 =
3 × L × 100 × 10 -6
(VLED - VINMIN ) × R8 × 2
where C2 is in farads, L is the inductance of the inductor L1 in henries, 100μA is the pullup current from SC,
VLED and VINMIN are in volts, and R8 is the switch current-sense resistor in ohms.
Boost-buck configuration:
C2 =
3 × L × 100 × 10 -6
(VLED ) × R8 × 2
where C2 is in farads, L is the inductance of the inductor L1 in henries, 100μA is the pullup current from SC,
VLED is in volts, and R8 is the switch current-sense
resistor in ohms.
Selection of Power Semiconductors
Switching MOSFET
ΔVIN
ESR CIN <
ΔIL × 2
where ΔIL is in amperes, ESRCIN is in ohms, and ΔVIN
is in volts.
Use the below equation to calculate the RMS current
rating of the input capacitor:
ICIN(RMS) =
ΔIL
2 3
Slope Compensation
Slope compensation should be added to converters
with peak current-mode control operating in continuous
conduction mode with more than 50% duty cycle to
avoid current loop instability and subharmonic oscillations. The minimum amount of slope added to the peak
inductor current to stabilize the current control loop is
half of the falling slope of the inductor.
In the MAX16834, the slope compensating ramp is
added to the current-sense signal before it is fed to the
PWM comparator. Connect a capacitor (C2 in the application circuit) from SC to ground for slope compensation. This capacitor is charged with a 100μA current
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The switching MOSFET (Q1) should have a voltage rating sufficient to withstand the maximum output voltage
together with the diode drop of the rectifier diode D1
and any possible overshoot due to ringing caused by
parasitic inductances and capacitances. Use a
MOSFET with a drain-to-source voltage rating higher
than that calculated by the following equations:
Boost configuration:
VDS = ( VLED + VD ) × 1.2
where VDS is the drain-to-source voltage in volts and
VD is the forward drop of the rectifier diode D1. The factor of 1.2 provides a 20% safety margin.
Boost-buck configuration:
VDS = ( VLED + VINMAX + VD ) × 1.2
where VDS is the drain-to-source voltage in volts and
VD is the forward drop of the rectifier diode D1. The factor of 1.2 provides a 20% safety margin.
The continuous drain current rating of the selected
MOSFET, when the case temperature is at +70°C,
should be greater than the value calculated by the fol-
Maxim Integrated | 14
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
lowing equation. The MOSFET must be mounted on a
board as per manufacturer specifications to dissipate
the heat.
The RMS current rating of the switching MOSFET Q1 is
calculated as follows for boost and boost-buck configurations:
⎛
ID RMS = ⎜
⎝
(IL AVG ) 2 × DMAX ⎞⎟⎠ × 1.3
where IDRMS is the MOSFET Q1’s drain RMS current in
amperes.
The MOSFET Q1 will dissipate power due to both
switching losses as well as conduction losses. The conduction losses in the MOSFET is calculated as follows:
PCOND = (IL AVG ) × D MAX × R DSON
2
where RDSON is the on-resistance of Q1 in ohms with
an assumed junction temperature of +100°C, PCOND is
in watts, and ILAVG is in amperes.
Use the following equations to calculate the switching
losses in the MOSFET:
Boost configuration:
⎛ IL
× VLED 2 × C GD × fSW ⎞
PSW = ⎜ AVG
⎟
2
⎝
⎠
⎛ 1
1 ⎞
×⎜
+
⎝ IGON IGOFF ⎟⎠
Boost-buck configuration:
⎛ IL
× (VLED + VINMAX ) 2 × C GD × fSW ⎞
PSW = ⎜ AVG
⎟
2
⎝
⎠
⎛ 1
1 ⎞
+
×⎜
⎝ IGON IGOFF ⎟⎠
Rectifier Diode
Use a Schottky diode as the rectifier (D1) for fast
switching and to reduce power dissipation. The selected Schottky diode must have a voltage rating 20%
above the maximum converter output voltage. The maximum converter output voltage is VLED in boost configuration and VLED + VINMAX in boost-buck configuration.
The current rating of the diode should be greater than
ID in the following equation:
ID = IL AVG × (1 - DMAX ) × 1.5
Dimming MOSFET
Select a dimming MOSFET (Q2) with continuous current
rating at +70°C, higher than the LED current by 30%.
The drain-to-source voltage rating of the dimming
MOSFET must be higher than VLED by 20%.
Feedback Compensation
The LED current control loop comprising of the switching converter, the LED current amplifier, and the error
amplifier should be compensated for stable control of
the LED current. The switching converter small-signal
transfer function has a right half-plane (RHP) zero for
both boost and boost-buck configurations as the inductor current is in continuous conduction mode. The RHP
zero adds a 20dB/decade gain together with a 90°
phase lag, which is difficult to compensate. The easiest
way to avoid this zero is to roll off the loop gain to 0dB
at a frequency less than one-fifth of the RHP zero frequency with a -20dB/decade slope.
The worst-case RHP zero frequency (fZRHP) is calculated as follows:
Boost configuration:
fZRHP =
VLED × (1 - DMAX ) 2
2π × L × ILED
Boost-buck configuration:
where IGON and IGOFF are the gate currents of the
MOSFET Q1 in amperes when it is turned on and
turned off, respectively, VLED and VINMAX are in volts,
ILAVG is in amperes, fSW is in hertz, and CGD is the
gate-to-drain MOSFET capacitance in farads.
Choose a MOSFET that has a higher power rating than
that calculated by the following equation when the
MOSFET case temperature is at +70°C:
PTOT (W) = PCOND (W) + PSW (W)
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fZRHP =
VLED × (1 - DMAX ) 2
2π × L × ILED × DMAX
where fZRHP is in hertz, VLED is in volts, L is the inductance value of L1 in henries (H), and ILED is in amperes.
The switching converter small-signal transfer function
also has an output pole for both boost and boost-buck
configurations. The effective output impedance that
determines the output pole frequency together with the
output filter capacitance is calculated as:
Maxim Integrated | 15
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Boost configuration:
(R LED + R10) × VLED
R OUT =
(R LED + R10) × ILED + VLED
Boost-buck configuration:
R OUT =
(R LED + R10) × VLED
(R LED + R10) × ILED × D MAX + VLED
where RLED is the dynamic impedance (rate of change
of voltage with current) of the LED string at the operating current, R10 is the LED current-sense resistor in
ohms, VLED is in volts, and ILED is in amperes.
The output pole frequency for both boost and boostbuck configurations is calculated as follows:
fP2 =
1
2π × C OUT × R OUT
where fP2 is in hertz, COUT is the output filter capacitance in farads, ROUT is the effective output impedance
in ohms calculated above.
Compensation components R7 and C7 perform two
functions. C7 introduces a low-frequency pole that
introduces a -20dB/decade slope into the loop gain. R7
flattens the gain of the error amplifier for frequencies
above the zero formed by R7 and C7. For compensation, this zero is placed at the output pole frequency fP2
such that it provides a -20dB/decade slope for frequencies above fP2 for the complete loop gain.
The value of R7 needed to fix the total loop gain at fP2
such that the total loop gain crosses 0dB at
-20dB/decade at one-fifth of the RHP zero can be calculated as follows:
R7 =
fZRHP × R8
5 × fP2 × (1 − D MAX ) × R10 × 9.9 × GM COMP
where R7 is the compensation resistor in ohms, fZRHP
and fP2 are in hertz, R8 is the switch current-sense
resistor in ohms, R10 is the LED current-sense resistor
in ohms, factor 9.9 is the gain of the LED current amplifier, and GMCOMP is the transconductance of the error
amplifier in Siemens.
The value of C7 can be calculated as:
www.maximintegrated.com
C7 =
1
2π × R7 × fP2
where C7 is in farads, fP2 is in hertz, and R7 is in ohms.
To minimize switching frequency noise, an additional
capacitor can be added in parallel with the series combination of R7 and C7. The pole from this capacitor and
R7 must be a decade higher than the loop crossover
frequency.
Short-Circuit Protection
Boost Configuration
In the boost configuration (Figure 2), if the LED string is
shorted then the excess current flowing in the LED current-sense resistor will cause NDRV to stop switching.
The input voltage will appear on the output capacitor,
and this causes very high peak currents to flow in the
LED current-sense resistor R10 because the dimming
MOSFET (Q2) is on. Once the voltage across the LED
current-sense resistor exceeds 300mV for more than
5μs, then the dimming MOSFET Q2 turns off and stays
off for 4096 switching clock cycles. At the same time,
NDRV is also off. The MAX16834 goes into the hiccup
mode and recovers from hiccup once the short has
been removed. The power dissipation in the dimming
MOSFET (Q2) is minimized during a short across the
LED string. During the same period, FLT only goes high
when the dimming MOSFET is on.
Boost-Buck Configuration
In the case of the boost-buck configuration (Figure 3),
once an LED string short occurs then the behavior is
different. A short across the LED string causes a high
current spike due to the external capacitors at the output. The regulation loop will cause NDRV to stop
switching. This causes the voltage on HV to drop if its
voltage is derived from LED+. The voltage on CLV will
drop, and this drop is detected after 128 clock cycles.
The dimming MOSFET and the switching MOSFET will
stop switching. It stays off for 4096 clock cycles, and
the cycle repeats itself. The short across the LED string
will cause the MAX16834 to go into a hiccup mode. At
the same time the FLT signal asserts itself for 4096
clock cycles every hiccup cycle. In the case where the
HV voltage is derived from a source different than
LED+, then the LED current will stay in regulation even
during a short across the LED string. In this case, FLT
does not assert itself during the short.
Maxim Integrated | 16
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
VIN
C1
L1
Q3
R1
C8
LV
FLT
IN
NDRV
D1
LED+
D2
24V
C3
Q1
LEDs
UVEN
HV
C2
R4
CS
MAX16834
ON
SC
OFF
PWMDIM
R3
LED-
RT/SYNC
C5
Q2
DIMOUT
R2
VCC
C4
REF
SENSE+
OVP+
R6
CLV
R5
REFI
COMP
SGND
PGND
R8
R9
R10
C9
C7
C6
R7
Figure 4. Boost LED Driver with Automotive Load Dump Protection
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Maxim Integrated | 17
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
VIN
LED+
C1
C3
D1
L1
R1
LV
HV
IN
NDRV
Q1
VLV
LEDs
UVEN
C2
SC
R3
RT/SYNC
CS
MAX16834
ON
OFF
PWMDIM
R4
LED-
C5
R2
VCC
C4
Q2
DIMOUT
REF
SENSE+
R6
OVP+
R5
REFI
CLV
FLT
COMP
SGND
PGND
R8
C7
R9
R10
C8
C6
R7
VLV
VLV
Figure 5. High-Side Buck LED Driver
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Maxim Integrated | 18
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
VIN
C1
L1
VOUT
R1
LV
FLT
IN
NDRV
D1
Q1
UVEN
C3
HV
C2
SC
R3
RT/SYNC
C5
R4
CS
MAX16834
VREF
PWMDIM
DIMOUT
R2
VCC
C4
REF
SENSE+
OVP+
CLV
R6
R5
REFI
COMP
SGND
PGND
OPTIONAL
C6
R9
R10
C7
R8
R7
Figure 6. Boost DC-DC Converter
www.maximintegrated.com
Maxim Integrated | 19
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
VIN
C1
L1
R1
D1
LV
HV
IN
NDRV
VOUT
Q1
UVEN
C2
C3
SC
MAX16834
R4
R11
R9
R10
CS
R3
VREF
RT/SYNC
PWMDIM
C5
R2
VCC
C4
REF
DIMOUT
SENSE+
OVP+
R6
R5
CLV
N.C.
REFI
COMP
FLT
SGND
PGND
C6
R7
C7
R8
VIN
Figure 7. Boost-Buck DC-DC Converter
www.maximintegrated.com
Maxim Integrated | 20
MAX16834
Layout Recommendations
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain current often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a dv/dt source; therefore, minimize the surface area of
the heatsink as much as is compatible with the MOSFET power dissipation or shield it. Keep all PCB traces
carrying switching currents as short as possible to minimize current loops. Use ground planes for best results.
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer
board whenever possible for better noise immunity and
power dissipation. Follow these guidelines for good
PCB layout:
1) Use a large contiguous copper plane under the
MAX16834 package. Ensure that all heat-dissipating components have adequate cooling.
2) Isolate the power components and high-current
path from the sensitive analog circuitry.
3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable, jitter-free operation. Keep switching loops short
such that:
a) The anode of D1 must be connected very close
to the drain of the MOSFET Q1.
www.maximintegrated.com
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
b) The cathode of D1 must be connected very
close to COUT.
c) COUT and the current-sense resistor R8 must be
connected directly to the ground plane.
4) Connect PGND and SGND to a star-point configuration.
5) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs.1oz) to enhance full-load
efficiency.
6) Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer
for the PGND and SGND plane as an EMI shield to
keep radiated noise away from the device, feedback dividers, and analog bypass capacitors.
7) To prevent discharge of the compensation capacitors during the off-time of the dimming cycle,
ensure that the PCB area close to these components has extremely low leakage. Discharge of
these capacitors due to leakage results in reduced
performance of the dimming circuitry.
Maxim Integrated | 21
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Pin Configurations
VCC
NDRV
PGND
CS
TOP VIEW
IN
TOP VIEW
15
14
13
12
11
+
LV 1
20 DIMOUT
SENSE+ 2
PWMDIM
OVP+ 3
9
UVEN
SGND 4
8
RT/SYNC
7
FLT
6
SC
10
HV 16
CLV 17
19 CLV
18 HV
MAX16834
COMP 5
DIMOUT 18
MAX16834
LV 19
2
3
4
5
REF
REFI
OVP+
1
COMP
+
SGND
SENSE+ 20
*EP
17 IN
16 VCC
REF 6
15 NDRV
REFI 7
14 PGND
SC 8
13 CS
FLT 9
12 PWMDIM
RT/SYNC 10
11 UVEN
TSSOP
QFN, TQFN
*EP = EXPOSED PAD.
Ordering Information
PART
TEMP RANGE
Chip Information
PIN-PACKAGE
MAX16834ATP+
-40°C to +125°C
20 TQFN-EP*
MAX16834ATP/V+
-40°C to +125°C
20 TQFN-EP*
MAX16834AUP+
-40°C to +125°C
20 TSSOP-EP*
MAX16834AUP/V+
-40°C to +125°C
20 TSSOP-EP*
MAX16834AGP/VY+
-40°C to +125°C
20 QFN-EP*
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
/V denotes an automotive qualified part.
/VY denotes an automotive qualified part in a “side wettable
flank” package.
www.maximintegrated.com
PROCESS: BiCMOS–DMOS
Package Information
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a
“+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but the
drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE NO.
LAND
PATTERN NO.
20 TQFN
T2044-3
21-0139
90-0037
20 TSSOP
U20E+1
21-0108
90-0114
20 QFN
G2044Y+1
21-0576
90-0360
Maxim Integrated | 22
High-Power LED Driver with Integrated
High-Side LED Current Sense and
PWM Dimming MOSFET Driver
MAX16834
Revision History
PAGES
CHANGED
REVISION
NUMBER
REVISION
DATE
0
8/08
Initial release
1
2/09
Added TSSOP package and automotive version. Also updated Electrical
Characteristics, Pin Description, Detailed Description, and LED CurrentSense Input (SENSE+) section, Pin Configuration and Package Information
2
5/09
Added automotive version of TQFN package
3
1/10
Added requirement for a capacitor on the SENSE+ pin
4
8/11
Added side wettable flank package
5
1/15
Updated Benefits and Features section
DESCRIPTION
—
1, 2, 6, 7, 8, 9, 22
1
2, 3, 4, 7, 9, 11,
13, 17–20
1, 6, 7, 22
1
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent
licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and
max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
© 2015 Maxim Integrated Products, Inc. | 23
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