General Description Benefits and Features
EVALUATION KIT AVAILABLE
MAX15004A/B-MAX15005A/B
General Description
The MAX15004A/B/MAX15005A/B high-performance,
current-mode PWM controllers operate at an automotive
input voltage range from 4.5V to 40V (load dump). The
input voltage can go lower than 4.5V after startup if IN is
bootstrapped to a boosted output voltage. The controllers
integrate all the building blocks necessary for implementing
fixed-frequency isolated/nonisolated power supplies. The
general-purpose boost, flyback, forward, and SEPIC converters can be designed with ease around the MAX15004/
MAX15005.
The current-mode control architecture offers excellent linetransient response and cycle-by-cycle current limit while
simplifying the frequency compensation. Programmable
slope compensation simplifies the design further. A fast
60ns current-limit response time, low 300mV current-limit
threshold makes the controllers suitable for high-efficiency,
high-frequency DC-DC converters. The devices include
an internal error amplifier and 1% accurate reference to
facilitate the primary-side regulated, single-ended flyback
converter or nonisolated converters.
An external resistor and capacitor network programs the
switching frequency from 15kHz to 500kHz (1MHz for
the MAX15005A/B). The MAX15004A/B/MAX15005A/B
provide a SYNC input for synchronization to an
external clock. The maximum FET-driver duty cycle for the
MAX15004A/B is 50%. The maximum duty cycle can be
set on the MAX15005A/B by selecting the right combination of RT and CT.
The input undervoltage lockout (ON/OFF) programs the
input-supply startup voltage and can be used to shutdown
the converter to reduce the total shutdown current down
to 10µA. Protection features include cycle-by-cycle and
hiccup current limit, output over-voltage protection, and
thermal shutdown.
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Benefits and Features
●● Wide Supply Voltage Range Meets Automotive
Power-Supply Operating Requirement Including
“Cold Crank” Conditions
• 4.5V to 40V Operating Input Voltage Range
(Can Operate at Lower Voltage After Startup if Input is Bootstrapped to a Boosted Output)
●● Control Architecture Offers Excellent Performance
While Simplifying the Design
• Current-Mode Control
• 300mV, 5% Accurate Current-Limit Threshold
Voltage
• Programmable Slope Compensation
• 50% (MAX15004) or Adjustable (MAX15005)
Maximum Duty Cycle
●● Accurate, Adjustable Switching Frequency and
Synchronization Avoids Interference with Sensitive
Radio Bands
• Switching Frequency Adjustable from 15kHz to
500kHz (1MHz for the MAX15005A/B)
• RC Programmable 4% Accurate Switching
Frequency
• External Frequency Synchronization
●● Built-In Protection Capability for Improved System
Reliability
• Cycle-by-Cycle and Hiccup Current-Limit
Protection
• Overvoltage and Thermal-Shutdown Protection
• -40°C to +125°C Automotive Temperature Range
• AEC-Q100 Qualified
Ordering Information
PIN-PACKAGE
MAX DUTY CYCLE
16 TSSOP-EP*
50%
The MAX15004A/B/MAX15005A/B are available in spacesaving 16-pin TSSOP and thermally enhanced 16-pin
TSSOP-EP packages. All devices operate over the -40°C
to +125°C automotive temperature range.
MAX15004AAUE/V+ 16 TSSOP-EP*
50%
MAX15004BAUE+
50%
16 TSSOP-EP*
Programmable
Applications
MAX15005AAUE/V+ 16 TSSOP-EP*
Programmable
MAX15005BAUE+
16 TSSOP
Programmable
MAX15005BAUE/V+ 16 TSSOP
Programmable
●● Automotive
●● Vacuum Fluorescent Display (VFD) Power Supply
●● Isolated Flyback, Forward, Nonisolated SEPIC, Boost
Converters
Pin Configuration appears at end of data sheet.
19-0723; Rev 5; 9/15
PART
MAX15004AAUE+
16 TSSOP
MAX15004BAUE/V+ 16 TSSOP
MAX15005AAUE+
50%
Note: All devices are specified over the -40°C to +125°C
temperature range.
+Denotes a lead(Pb)-free/RoHS-compliant package.
/V denotes an automotive qualified part.
*EP = Exposed pad.
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Absolute Maximum Ratings
IN to SGND............................................................-0.3V to +45V
IN to PGND............................................................-0.3V to +45V
ON/OFF to SGND.......................................-0.3V to (VIN + 0.3V)
OVI, SLOPE, RTCT, SYNC, SS, FB, COMP,
CS to SGND...................................... -0.3V to (VREG5 + 0.3V)
VCC to PGND.........................................................-0.3V to +12V
REG5 to SGND........................................................-0.3V to +6V
OUT to PGND........................................... -0.3V to (VCC + 0.3V)
SGND to PGND.....................................................-0.3V to +0.3V
VCC Sink Current (clamped mode).....................................35mA
OUT Current (< 10μs transient)..........................................±1.5A
Continuous Power Dissipation* (TA = +70°C)
16-Pin TSSOP-EP (derate 21.3mW/°C
above +70°C).............................................................1702mW
16-Pin TSSOP (derate 9.4mW/°C above +70°C).............754mW
Operating Junction Temperature Range........... -40°C to +125°C
Junction Temperature.......................................................+150°C
Storage Temperature Range............................. -60°C to +150°C
Lead Temperature (soldering, 10s).................................. +300°C
Soldering Temperature (reflow)........................................+260°C
*As per JEDEC51 Standard, Multilayer Board.
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Electrical Characteristics
(VIN = 14V, CIN = 0.1μF, CVCC = 0.1μF // 1μF, CREG5 = 1μF, VON/OFF = 5V, CSS = 0.01μF, CSLOPE = 100pF, RT = 13.7kΩ, CT = 560pF,
VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical values are at TA = +25°C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
POWER SUPPLY
Input Supply Range
Operating Supply Current
ON/OFF CONTROL
Input-Voltage Threshold
Input-Voltage Hysteresis
Input Bias Current
Shutdown Current
INTERNAL 7.4V LDO (VCC)
Output (VCC) Voltage Set Point
Line Regulation
UVLO Threshold Voltage
UVLO Hysteresis
Dropout Voltage
Output Current Limit
Internal Clamp Voltage
INTERNAL 5V LDO (REG5)
Output (REG5) Voltage Set Point
Line Regulation
VIN
IQ
VON
VON/OFF rising
IB-ON/OFF
VON/OFF = 40V
VHYST-ON
ISHDN
VVCC
V­UVLO-VCC
VHYST-UVLO
IVCC-ILIM
VREG5
www.maximintegrated.com
IREG5-ILIM
1.05
IVCC = 0 to 20mA (sourcing)
7.15
VCC rising
3.15
VIN = 8V to 40V
VCC = 7.5V, IREG5 = 0 to 15mA (sourcing)
VCC = 5.5V to 10V
VCC = 4.5V, IREG5 = 15mA (sourcing)
IREG5 sourcing
V
3.1
mA
1.23
1.40
V
mV
0.5
µA
10
20
µA
7.4
7.60
1
3.5
0.25
3.75
0.5
V
mA
10.0
10.4
10.8
4.75
4.95
5.05
2
32
V
mV
45
0.25
V
mV/V
500
VIN = 4.5V, IVCC = 20mA (sourcing)
IVCC sourcing
40.0
2
75
VON/OFF = 0V
VVCC-CLAMP IVCC = 30mA (sinking)
Dropout Voltage
Output Current Limit
4.5
VIN = 40V, fOSC = 150kHz
V
V
mV/V
0.5
V
mA
Maxim Integrated │ 2
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Electrical Characteristics (continued)
(VIN = 14V, CIN = 0.1μF, CVCC = 0.1μF // 1μF, CREG5 = 1μF, VON/OFF = 5V, CSS = 0.01μF, CSLOPE = 100pF, RT = 13.7kΩ, CT = 560pF,
VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical values are at TA = +25°C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
1000
kHz
OSCILLATOR (RTCT)
Oscillator Frequency Range
RTCT Peak Trip Level
RTCT Valley Trip Level
RTCT Discharge Current
fOSC
fOSC = 2 x fOUT for MAX15004A/B,
fOSC = fOUT for MAX15005A/B
15
VTH,RTCT
VTL,RTCT
IDIS,RTCT
Oscillator Frequency Accuracy
(Note 2)
0.55 x VREG5
VRTCT = 2V
1.30
0.1 x VREG5
1.33
Minimum On-Time
DMAX
tON-MIN
SYNC Lock-In Frequency Range
(Note 4)
SYNC High-Level Voltage
SYNC Low-Level Voltage
SYNC Input Current
+4
RT = 13.7kΩ, CT = 560pF,
fOSC (typ) = 150kHz
-4
+4
RT = 21kΩ, CT = 100pF,
fOSC (typ) = 500kHz
-5
+5
RT = 7kΩ, CT = 100pF,
fOSC (typ) = 1MHz
-7
+7
SYNC Minimum Input Pulse Width
MAX15005A/B,
RT = 13.7kΩ, CT = 560pF,
fOSC (typ) = 150kHz
50
78.5
%
80
81.5
110
170
ns
200
%fOSC
102
2
VSYNC = 0 to 5V
mA
%
VIN = 14V
VIH-SYNC
ISYNC
1.36
-4
RT = 13.7kΩ, CT = 560pF,
fOSC (typ) = 150kHz
VIL-SYNC
V
RT = 13.7kΩ, CT = 4.7nF,
fOSC (typ) = 18kHz
MAX15004A/B
Maximum PWM Duty Cycle
(Note 3)
V
V
-0.5
0.8
V
+0.5
µA
50
ns
ERROR AMPLIFIER/SOFT-START
Soft-Start Charging Current
SS Reference Voltage
SS Threshold for HICCUP Enable
ISS
VSS
VSS = 0V
8
15
21
µA
1.215
1.228
1.240
V
VSS rising
1.1
FB Regulation Voltage
VREF-FB
COMP = FB,
ICOMP = -500µA to +500µA
1.215
FB Input Offset Voltage
VOS-FB
COMP = 0.25V to 4.5V,
ICOMP = -500µA to +500µA,
VSS = 0 to 1.5V
-5
FB Input Current
COMP Sink Current
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ICOMP-SINK
VFB = 0 to 1.5V
VFB = 1.5V, VCOMP = 0.25V
1.228
-300
3
V
1.240
V
+5
mV
+300
5.5
nA
mA
Maxim Integrated │ 3
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Electrical Characteristics (continued)
(VIN = 14V, CIN = 0.1μF, CVCC = 0.1μF // 1μF, CREG5 = 1μF, VON/OFF = 5V, CSS = 0.01μF, CSLOPE = 100pF, RT = 13.7kΩ, CT = 560pF,
VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical values are at TA = +25°C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER
COMP Source Current
SYMBOL
ICOMPSOURCE
CONDITIONS
VFB = 1V, VCOMP = 4.5V
COMP High Voltage
VOH-COMP
VFB = 1V, ICOMP = 1mA (sourcing)
COMP Low Voltage
VOL-COMP
VFB = 1.5V, ICOMP = 1mA (sinking)
Open-Loop Gain
Unity-Gain Bandwidth
Phase Margin
COMP Positive Slew Rate
COMP Negative Slew Rate
AEAMP
MIN
TYP
1.3
2.8
mA
VREG5
- 0.5
VREG5
- 0.2
V
0.1
MAX
0.25
UNITS
V
100
dB
UGFEAMP
1.6
MHz
75
degrees
SR+
0.5
V/µs
SR-
-0.5
V/µs
PMEAMP
PWM COMPARATOR
Current-Sense Gain
ACS-PWM
ΔVCOMP/ΔVCS (Note 5)
PWM Propagation Delay to OUT
tPD-PWM
CS = 0.15V, from VCOMP falling edge:
3V to 0.5V to OUT falling (excluding
leading-edge blanking time)
PWM Comparator Current-Sense
Leading-Edge Blanking Time
tCS-BLANK
2.85
3
3.15
V/V
60
ns
50
ns
CURRENT-LIMIT COMPARATOR
Current-Limit Threshold Voltage
Current-Limit Input Bias Current
VILIM
IB-CS
ILIMIT Propagation Delay to OUT
tPD-ILIM
ILIM Comparator Current-Sense
Leading-Edge Blanking Time
tCS-BLANK
290
OUT= high, 0 ≤ VCS ≤ 0.3V
305
-2
From CS rising above VILIM (50mV
overdrive) to OUT falling (excluding
leading-edge blanking time)
Number of Consecutive ILIMIT
Events to HICCUP
317
mV
+2
µA
60
ns
50
ns
7
HICCUP Timeout
Clock
periods
512
SLOPE COMPENSATION (Note 6)
Slope Capacitor Charging Current
Slope Compensation
Slope Compensation Tolerance
(Note 2)
Slope Compensation Range
www.maximintegrated.com
ISLOPE
VSLOPE = 100mV
9.8
CSLOPE = 100pF
-4
CSLOPE = 100pF
CSLOPE = 22pF
CSLOPE = 1000pF
10.5
11.2
25
+4
110
2.5
µA
mV/µs
%
mV/µs
Maxim Integrated │ 4
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Electrical Characteristics (continued)
(VIN = 14V, CIN = 0.1μF, CVCC = 0.1μF // 1μF, CREG5 = 1μF, VON/OFF = 5V, CSS = 0.01μF, CSLOPE = 100pF, RT = 13.7kΩ, CT = 560pF,
VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical values are at TA = +25°C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
3.5
UNITS
OUTPUT DRIVER
Driver Output Impedance
Driver Peak Output Current
ROUT-N
VCC = 8V (applied externally),
IOUT = 100mA (sinking)
1.7
ROUT-P
VCC = 8V (applied externally),
IOUT = 100mA (sourcing)
3
IOUT-PEAK
OVERVOLTAGE COMPARATOR
Overvoltage Comparator Input
Threshold
Overvoltage Comparator
Hysteresis
Overvoltage Comparator Delay
OVI Input Current
THERMAL SHUTDOWN
Shutdown Temperature
Thermal Hysteresis
VOV-TH
Ω
COUT = 10nF, sinking
1000
COUT = 10nF, sourcing
VOVI rising
IOVI
1.20
TSHDN
THYST
From OVI rising above 1.228V to OUT
falling, with 50mV overdrive
VOVI = 0 to 5V
Temperature rising
mA
750
VOV-HYST
TDOVI
5
1.228
1.26
V
125
mV
1.6
µs
-0.5
+0.5
µA
160
°C
15
°C
Note 1: 100% production tested at +125°C. Limits over the temperature range are guaranteed by design.
Note 2: Guaranteed by design; not production tested.
Note 3: For the MAX15005A/B, DMAX depends upon the value of RT. See Figure 3 and the Oscillator Frequency/External
Synchronization section.
Note 4: The external SYNC pulse triggers the discharge of the oscillator ramp. See Figure 2. During external SYNC, DMAX = 50%
for the MAX15004A/B; for the MAX15005A/B, there is a shift in DMAX with fSYNC/fOSC ratio (see the Oscillator Frequency/
External Synchronization section).
Note 5: The parameter is measured at the trip point of latch, with 0 ≤ VCS ≤ 0.3V, and FB = COMP.
Note 6: Slope compensation = (2.5 x 10-9)/CSLOPE mV/μs. See the Applications Information section.
www.maximintegrated.com
Maxim Integrated │ 5
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Characteristics
VIN = 14V, CIN = 0.1μF, CVCC = 0.1μF // 1μF, CREG5 = 1μF, VON/OFF = 5V, CSS = 0.01μF, CSLOPE = 100pF, RT = 13.7kΩ,
CT = 560pF. TA = +25°C, unless otherwise noted.)
13
10
COUT = 0nF
7
110
135
VCC OUTPUT VOLTAGE
vs. VIN SUPPLY VOLTAGE
IVCC = 0mA
IVCC = 1mA
7.0
IVCC = 20mA
6.5
6.0
5.5
5
10
15 20 25 30 35
VIN SUPPLY VOLTAGE (V)
150
OSCILLATOR FREQUENCY (kHz)
149
148
40
45
10 60 110 160 210 260 310 360 410 460 510
FREQUENCY (kHz)
REG5 OUTPUT VOLTAGE
vs. VCC VOLTAGE
5.000
4.975
IREG5 = 1mA (SOURCING)
4.950
4.925
4.900
4.875
4.850
IREG5 = 15mA (SOURCING)
4.825
4.800
4.775
4.750
4.725
4.700
5.5 6.0 6.5 7.0 7.5 8.0 8.5 9.0 9.5 10.0 10.5
VCC VOLTAGE (V)
OSCILLATOR FREQUENCY (fOSC)
vs. VIN SUPPLY VOLTAGE
RT = 13.7kΩ
CT = 560pF
147
TA = +25°C
146
MAX15005
TA = -40°C
145
144
143
142
141
140
MAX15004 toc06
10
35
60
85
TEMPERATURE (°C)
TA = +125°C
TA = +135°C
5.5 10.5 15.5 20.5 25.5 30.5 35.5 40.5 45.5
VIN SUPPLY VOLTAGE (V)
www.maximintegrated.com
TA = +25°C
TA = -40°C
5
10
40
45
REG5 DROPOUT VOLTAGE
vs. IREG5
0.30
0.28
0.25
VCC = 4.5
VIN = VON/OFF
0.23
0.20
0.18
0.15
0.13
0.10
0.08
0.05
0.03
0
15 20 25 30 35
SUPPLY VOLTAGE (V)
TA = +125°C
TA = +135°C
TA = +25°C
TA = -40°C
0
2
4
6
8
10
IREG5 (mA)
12
14
OSCILLATOR FREQUENCY (fOSC)
vs. RT/CT
1000
OSCILLATOR FREQUENCY (kHz)
-15
1
MAX15004 toc03
MAX15004 toc02
COUT = 10nF
16
TA = +135°C
MAX15004 toc07
19
MAX15004 toc08
-40
7.5
5.0
22
4
REG5 OUTPUT VOLTAGE (V)
20
10
0
25
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
SHUTDOWN SUPPLY CURRENT
vs. SUPPLY VOLTAGE
CT = 100pF
CT = 220pF
CT = 560pF
MAX15004 toc09
50
40
30
28
VIN SHUTDOWN SUPPLY CURRENT (µA)
70
60
MAX15005
VIN = 14V
CT = 220pF
REG5 LDO DROPOUT VOLTAGE (V)
80
31
VIN SUPPLY CURRENT (mA)
MAX15004 toc01
100
90
MAX15004 toc04
VIN UVLO HYSTERESIS (mV)
120
110
VCC OUTPUT VOLTAGE (V)
VIN SUPPLY CURRENT (ISUPPLY)
vs. OSCILLATOR FREQUENCY (fOSC)
VIN UVLO HYSTERESIS
vs. TEMPERATURE
CT = 1000pF
100
CT = 1500pF
CT = 2200pF
CT = 3300pF
10
1
10
100
1000
RT (kΩ)
Maxim Integrated │ 6
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Characteristics (continued)
VIN = 14V, CIN = 0.1μF, CVCC = 0.1μF // 1μF, CREG5 = 1μF, VON/OFF = 5V, CSS = 0.01μF, CSLOPE = 100pF, RT = 13.7kΩ,
CT = 560pF. TA = +25°C, unless otherwise noted.)
80
75
70
65
CT = 3300pF
CT = 2200pF
CT = 1500pF
60
CT = 1000pF CT = 560pF
55
50
10
CT = 220pF
100
OUTPUT FREQUENCY (kHz)
70
55
50
52
51
50
49
48
47
CT = 560pF
RT = 10kΩ
fOSC = fOUT = 180kHz
CRTCT = 220pF
RRTCT = 10kΩ
fOSC = fOUT = 418kHz
-40
-15
10
35
60
85
TEMPERATURE (°C)
110
1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0
fSYNC/fOSC RATIO
GAIN
MAX15004 toc12
75
73
71
69
260
220
180
PHASE
140
1
10
100 1k 10k 100k 1M 10M
FREQUENCY (Hz)
-15
10
35
60
85
TEMPERATURE (°C)
110
135
CS-TO-OUT DELAY vs. TEMPERATURE
MAX15004 toc15
VCS OVERDRIVE = 50mV
80
100
0.1
-40
90
300
60
DRIVER OUTPUT PEAK SOURCE
AND SINK CURRENT
70
60
VCS OVERDRIVE = 190mV
50
40
30
20
10
0
-40
-15
10
35
60
85
TEMPERATURE (°C)
110
135
POWER-UP SEQUENCE THROUGH VIN
MAX15004 toc18
MAX15004 toc17
MAX15004 toc16
77
100
340
40
30
20
OVI TO OUT DELAY THROUGH
OVERVOLTAGE COMPARATOR
79
65
135
MAX15004 toc14
90
80
70
60
50
10
0
-10
81
67
110
100
MAX15004 toc13
MAX15005
CT = 560pF
RT = 13.7kΩ
fOSC = fOUT = 150kHz
83
ERROR AMPLIFIER OPEN-LOOP GAIN
AND PHASE vs. FREQUENCY
65
60
53
45
1000
GAIN (dB)
MAXIMUM DUTY CYCLE (%)
75
85
46
MAXIMUM DUTY CYCLE
vs. fSYNC/fOSC RATIO
80
fOUT = 75kHz
54
MAXIMUM DUTY CYCLE (%)
85
55
PHASE (DEGREES)
CS-TO-OUT DELAY (ns)
MAXIMUM DUTY CYCLE (%)
90
MAX15005 MAXIMUM DUTY CYCLE
vs. TEMPERATURE
MAX15004 MAXIMUM DUTY CYCLE
vs. TEMPERATURE
MAX15004 toc11
CT = 100pF
95
MAXIMUM DUTY CYCLE (%)
100
MAX15004 toc10
MAX15005 MAXIMUM DUTY CYCLE
vs. OUTPUT FREQUENCY (fOUT)
COUT = 10nF
VOUT
VOVI
VOUT
5V/div
VOUT
2V/div
VIN
10V/div
VCC
5V/div
VON/OFF = 5V
REG5
5V/div
IOUT
1A/div
VOVI
500mV/div
VOUT
5V/div
1µs/div
www.maximintegrated.com
400ns/div
2ms/div
Maxim Integrated │ 7
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Characteristics (continued)
VIN = 14V, CIN = 0.1μF, CVCC = 0.1μF // 1μF, CREG5 = 1μF, VON/OFF = 5V, CSS = 0.01μF, CSLOPE = 100pF, RT = 13.7kΩ,
CT = 560pF. TA = +25°C, unless otherwise noted.)
POWER-UP SEQUENCE
THROUGH ON/OFF
POWER-DOWN SEQUENCE THROUGH VIN
POWER-DOWN SEQUENCE
THROUGH ON/OFF
MAX15004 toc21
MAX15004 toc20
MAX15004 toc19
VON/OFF = 5V
ON/OFF
5V/div
VIN
10V/div
ON/OFF
5V/div
VCC
5V/div
VCC
5V/div
VCC
5V/div
REG5
5V/div
REG5
5V/div
REG5
5V/div
VOUT
5V/div
VOUT
5V/div
VOUT
5V/div
400ms/div
1ms/div
4ms/div
LINE TRANSIENT FOR VIN STEP
FROM 14V TO 5.5V
LINE TRANSIENT FOR VIN STEP
FROM 14V TO 40V
MAX15004 toc22
MAX15004 toc23
VIN
10V/div
VCC
5V/div
VIN
20V/div
VCC
5V/div
REG5
5V/div
REG5
5V/div
VOUT
5V/div
VOUT
5V/div
100µs/div
100µs/div
HICCUP MODE FOR FLYBACK CIRCUIT
(FIGURE 7)
DRAIN WAVEFORM IN
FLYBACK CONVERTER (FIGURE 7)
MAX15004 toc25
MAX15004 toc24
ILOAD = 10mA
VCS
200mV/div
10V/div
VANODE
1V/div
ISHORT
500mA/div
10µs/div
www.maximintegrated.com
4µs/div
Maxim Integrated │ 8
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Pin Description
PIN
NAME
1
IN
FUNCTION
2
ON/OFF
3
OVI
4
SLOPE
5
N.C.
6
RTCT
Oscillator-Timing Network Input. Connect a resistor from RTCT to REG5 and a capacitor from RTCT to SGND to
set the oscillator frequency (see the Oscillator Frequency/External Synchronization section).
7
SGND
Signal Ground. Connect SGND to SGND plane.
8
SYNC
External-Clock Synchronization Input. Connect SYNC to SGND when not using an external clock.
9
SS
Soft-Start Capacitor Input. Connect a capacitor from SS to SGND to set the soft-start time interval.
10
FB
Internal Error-Amplifier Inverting Input. The noninverting input is internally connected to SS.
11
COMP
12
CS
13
REG5
5V Low-Dropout Regulator Output. Bypass REG5 with a 1µF ceramic capacitor to SGND.
14
PGND
Power Ground. Connect PGND to the power ground plane.
15
OUT
Gate Driver Output. Connect OUT to the gate of the external n-channel MOSFET.
16
VCC
7.4V Low-Dropout Regulator Output—Driver Power Source. Bypass VCC with 0.1µF and 1µF or higher ceramic
capacitors to PGND. Do not connect external supply or bootstrap to VCC.
—
EP
Exposed Pad (MAX15004A/MAX15005A only). Connect EP to the SGND plane to improve thermal performance.
Do not use the EP as an electrical connection.
Input Power Supply. Bypass IN with a minimum 0.1µF ceramic capacitor to PGND.
ON/OFF Input. Connect ON/OFF to IN for always-on operation. To externally program the UVLO threshold of
the IN supply, connect a resistive divider between IN, ON/OFF, and SGND. Pull ON/OFF to SGND to disable the
controller.
Overvoltage Comparator Input. Connect a resistive divider between the output of the power supply, OVI, and
SGND to set the output overvoltage threshold.
Programmable Slope Compensation Capacitor Input. Connect a capacitor (CSLOPE) to SGND to set the amount
of slope compensation.
Slope compensation = (2.5 x 10-9)/CSLOPE mV/µs with CSLOPE in farads.
No Connection. Not internally connected.
Error-Amplifier Output. Connect the frequency compensation network between FB and COMP.
Current-Sense Input. The current-sense signal is compared to a signal proportional to the error-amplifier output
voltage.
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Maxim Integrated │ 9
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Functional Diagram
IN 1
MAX15004A/B
MAX15005A/B
1.228V
OFF
ON/OFF 2
ON/OFF
COMP
PREREGULATOR
OFF
3.5V
UVLO
REFERENCE
16 VCC
7.4V LDO
REG
UVB
DRIVER
15 OUT
14 PGND
VCC
THERMAL
SHUTDOWN
SET
RESET
OV-COMP
OVI 3
UVB
ILIMIT
COMP
5V LDO
REG
13 REG5
50ns
LEAD
DELAY
12 CS
0.3V
1.228V
PWMCOMP
R
OVRLD
SLOPE 4
SLOPE
COMPENSATION
2R
RTCT 6
OSCILLATOR
10 FB
SGND 7
RESET
SYNC 8
11 COMP
SS_OK
CLK
7
CONSECUTIVE
EVENTS
COUNTER
EAMP
1.228V
9 SS
REF-AMP
OVRLD
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Maxim Integrated │ 10
MAX15004A/B-MAX15005A/B
Detailed Description
The MAX15004A/B/MAX15005A/B are high-performance,
current-mode PWM controllers for wide input-voltage
range isolated/nonisolated power supplies. These controllers are for use as general-purpose boost, flyback,
and SEPIC controllers. The input voltage range of 4.5V
to 40V makes it ideal in automotive applications such as
vacuum fluorescent display (VFD) power supplies. The
internal low-dropout regulator (VCC regulator) enables the
MAX15004A/B/MAX15005A/B to operate directly from an
automotive battery input. The input voltage can go lower
than 4.5V after startup if IN is bootstrapped to a boosted
output voltage.
The undervoltage lockout (ON/OFF) allows the devices
to program the input-supply startup voltage and ensures
predictable operation during brownout conditions.
The devices contain two internal regulators, VCC and
REG5. The VCC regulator output voltage is set at 7.4V
and REG5 regulator output voltage at 5V ±2%. The input
undervoltage lockout (UVLO) circuit monitors the VCC
voltage and turns off the converter when the VCC voltage
drops below 3.5V (typ).
An external resistor and capacitor network programs
the switching frequency from 15kHz to 500kHz. The
MAX15004A/B/MAX15005A/B provide a SYNC input for
synchronization to an external clock. The OUT (FETdriver output) duty cycle for the MAX15004A/B is 50%.
The maximum duty cycle can be set on MAX15005A/B by
selecting the right combination of RT and CT. The RTCT
discharge current is trimmed to 2%, allowing accurate
setting of the duty cycle for the MAX15005. An internal
slope-compensation circuit stabilizes the current loop
when operating at higher duty cycles and can be programmed externally.
The MAX15004/MAX15005 include an internal error
amplifier with 1% accurate reference to regulate the
output in nonisolated topologies using a resistive divider.
The internal reference connected to the noninverting input
of the error amplifier can be increased in a controlled
manner to obtain soft-start. A capacitor connected at SS
to ground programs soft-start to reduce inrush current and
prevent output overshoot.
The MAX15004/MAX15005 include protection features
like hiccup current limit, output overvoltage, and thermal shutdown. The hiccup current-limit circuit reduces
the power delivered to the electronics powered by the
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4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
MAX15004/MAX15005 converter during severe fault conditions. The overvoltage circuit senses the output using
the path different from the feedback path to provide
meaningful overvoltage protection. During continuous
high input operation, the power dissipation into the
MAX15004/MAX15005 could exceed its limit. Internal
thermal shutdown protection safely turns off the converter
when the junction heats up to 160°C.
Current-Mode Control Loop
The advantages of current-mode control overvoltagemode control are twofold. First, there is the feed-forward
characteristic brought on by the controller’s ability to adjust
for variations in the input voltage on a cycle-by-cycle
basis. Secondly, the stability requirements of the currentmode controller are reduced to that of a single-pole
system unlike the double pole in voltage-mode control.
The MAX15004/MAX15005 offer peak current-mode
control operation to make the power supply easy to design
with. The inherent feed-forward characteristic is useful
especially in an automotive application where the input
voltage changes fast during cold-crank and load dump conditions. While the current-mode architecture offers many
advantages, there are some shortcomings. For higher dutycycle and continuous conduction mode operation where
the transformer does not discharge during the off duty
cycle, subharmonic oscillations appear. The MAX15004/
MAX15005 offer programmable slope compensation using
a single capacitor. Another issue is noise due to turn-on
of the primary switch that may cause the premature end
of the on cycle. The current-limit and PWM comparator
inputs have leading-edge blanking. All the shortcomings of
the current-mode control are addressed in the MAX15004/
MAX15005, making it ideal to design for automotive power
conversion applications.
Internal Regulators VCC and REG5
The internal LDO converts the automotive battery voltage
input to a 7.4V output voltage (VCC). The VCC output is
set at 7.4V and operates in a dropout mode at input voltages below 7.5V. The internal LDO is capable of delivering 20mA current, enough to provide power to internal
control circuitry and the gate drive. The regulated VCC
keeps the driver output voltage well below the absolute
maximum gate voltage rating of the MOSFET especially
during the double battery and load dump conditions.
Maxim Integrated │ 11
MAX15004A/B-MAX15005A/B
The second 5V LDO regulator from VCC to REG5 provides power to the internal control circuits. This LDO can
also be used to source 15mA of external load current.
Bypass VCC and REG5 with a parallel combination of
1µF and 0.1µF low-ESR ceramic capacitors. Additional
capacitors (up to 22µF) at VCC can be used although they
are not necessary for proper operation of the MAX15004/
MAX15005.
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
VIN
MAX15004A/B
MAX15005A/B
R1
ON/OFF
1.23V
R2
Startup Operation/UVLO/ON/OFF
The MAX15004A/B/MAX15005A/B feature two undervoltage lockouts (UVLO). The internal UVLO monitors the
VCC-regulator and turns on the converter once VCC rises
above 3.5V. The internal UVLO circuit has about 0.5V
hysteresis to avoid chattering during turn-on. Once the
power is on and the bootstrapped voltage feeds VCC,
IN voltage can drop below 4V. This feature provides
operation at a cold-crank voltage as low as 2.5V.
An external undervoltage lockout can be achieved by
controlling the voltage at the ON/OFF input. The ON/
OFF input threshold is set at 1.23V (rising) with 75mV
hysteresis.
Before any operation can commence, the ON/OFF voltage must exceed the 1.23V threshold.
Calculate R1 in Figure 1 by using the following formula:
 V

=
R1  ON − 1 × R2
 VUVLO

where VUVLO is the ON/OFF’s 1.23V rising threshold,
and VON is the desired input startup voltage. Choose an
R2 value in the 100kΩ range. The UVLO circuits keep
the PWM comparator, ILIM comparator, oscillator, and
output driver shut down to reduce current consumption
(see the Functional Diagram). The ON/OFF input can be
used to disable the MAX15004/MAX15005 and reduce
the standby current to less than 20μA.
Soft-Start
The MAX15004/MAX15005 are provided with an externally adjustable soft-start function, saving a number of
external components. The SS is a 1.228V reference
bypass connection for the MAX15004A/B/MAX15005A/B
and also controls the soft-start period. At startup, after
VIN is applied and the UVLO thresholds are reached, the
device enters soft-start. During soft-start, 15μA is sourced
into the capacitor (CSS) connected from SS to GND
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Figure 1. Setting the MAX15004A/B/MAX15005A/B
Undervoltage Lockout Threshold
causing the reference voltage to ramp up slowly. The
HICCUP mode of operation is disabled during soft-start.
When VSS reaches 1.228V, the output as well as the
HICCUP mode become fully active. Set the soft-start time
(tSS) using following equation:
t SS =
1.23(V) × C SS
15 × 10 −6 (A )
where tSS is in seconds and CSS is in farads.
The soft-start programmability is important to control
the input inrush current issue and also to avoid the
MAX15004/MAX15005 power supply from going into the
unintentional hiccup during the startup. The required softstart time depends on the topology used, current-limit setting, output capacitance, and the load condition.
Oscillator Frequency/External Synchronization
Use an external resistor and capacitor at RTCT to program the MAX15004A/B/MAX15005A/B internal oscillator frequency from 15kHz to 1MHz. The MAX15004A/B
output switching frequency is one-half the programmed
oscillator frequency with a 50% maximum duty-cycle
limit. The MAX15005A/B output switching frequency is
the same as the oscillator frequency. The RC network
connected to RTCT controls both the oscillator frequency and the maximum duty cycle. The CT capacitor
charges and discharges from (0.1 x VREG5) to (0.55 x
VREG5). It charges through RT and discharges through an
internal trimmed controlled current sink. The maximum
duty cycle is inversely proportional to the discharge time
Maxim Integrated │ 12
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
(tDISCHARGE). See Figures 3a and 3b for a coarse
selection of capacitor values for a given switching frequency and maximum duty cycle and then use the following
equations to calculate the resistor value to fine-tune the
switching frequency and verify the worst-case maximum
duty cycle.
The MAX15004A/B is a 50% maximum duty-cycle part,
while the MAX15005A/B is 100% maximum duty-cycle
part.
f OUT =
1
f OSC
2
for the MAX15004A/B and
D
t CHARGE = MAX
f OSC
t DISCHARGE =
f OUT = f OSC
t
RT = CHARGE
0.7 × CT
2.25(V) × RT × CT
for the MAX15005A/B.
(1.33 × 10 −3 (A) × RT) − 3.375(V)
1

...................Use This Equation If f OSC ≤ 500kHz
t
 CHARGE + t DISCHARGE
f OSC = 
1

.......UseThis Equation If f OSC > 500kHz
 t CHARGE + t DISCHARGE + 160ns
where fOSC is the oscillator frequency, RT is a resistor
connected from RTCT to REG5, and CT is a capacitor
connected from RTCT to SGND. Verify that the oscillator frequency value meets the target. Above calculations
could be repeated to fine-tune the switching frequency.
The MAX15004A/B/MAX15005A/B can be synchronized
using an external clock at the SYNC input. For proper
frequency synchronization, SYNC’s input frequency must
be at least 102% of the programmed internal oscillator
frequency. Connect SYNC to SGND when not using an
external clock. A rising clock edge on SYNC is interpreted
as a synchronization input. If the SYNC signal is lost,
the internal oscillator takes control of the switching rate,
returning the switching frequency to that set by RC network connected to RTCT. This maintains output regulation
even with intermittent SYNC signals.
MAX15004A/B (DMAX = 50%)
WITH SYNC
INPUT
WITHOUT
SYNC INPUT
RTCT
CLKINT
SYNC
OUT
D = 50%
D = 50%
MAX15005A/B (DMAX = 81%)
WITH SYNC
INPUT
WITHOUT
SYNC INPUT
RTCT
CLKINT
SYNC
OUT
D = 81.25%
D = 80%
Figure 2. Timing Diagram for Internal Oscillator vs. External SYNC and DMAX Behavior
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Maxim Integrated │ 13
MAX15004A/B-MAX15005A/B
MAX15005 MAXIMUM DUTY CYCLE
vs. OUTPUT FREQUENCY (fOUT)
100
OSCILLATOR FREQUENCY (kHz)
MAXIMUM DUTY CYCLE (%)
90
85
80
75
CT = 3300pF
70
CT = 2200pF
CT = 1500pF
60
CT = 1000pF CT = 560pF
55
50
10
OSCILLATOR FREQUENCY (fOSC)
vs. RT/CT
1000
CT = 100pF
95
65
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
CT = 100pF
CT = 220pF
CT = 560pF
CT = 1000pF
100
CT = 1500pF
CT = 2200pF
CT = 3300pF
CT = 220pF
100
OUTPUT FREQUENCY (kHz)
1000
10
1
10
100
1000
RT (kΩ)
Figure 3a. MAX15005 Maximum Duty Cycle vs. Output
Frequency.
Figure 3b. Oscillator Frequency vs. RT/CT
n-Channel MOSFET Driver
Slope Compensation
OUT drives the gate of an external n-channel MOSFET.
The driver is powered by the internal regulator (VCC),
internally set to approximately 7.4V. The regulated VCC
voltage keeps the OUT voltage below the maximum gate
voltage rating of the external MOSFET. OUT can source
750mA and sink 1000mA peak current. The average
current sourced by OUT depends on the switching
frequency and total gate charge of the external MOSFET.
Error Amplifier
The MAX15004A/B/MAX15005A/B include an internal
error amplifier. The noninverting input of the error amplifier is connected to the internal 1.228V reference and
feedback is provided at the inverting input. High 100dB
open-loop gain and 1.6MHz unity-gain bandwidth allow
good closed-loop bandwidth and transient response.
Moreover, the source and sink current capability of 2mA
provides fast error correction during the output load transient. For Figure 5, calculate the power-supply output
voltage using the following equation:
 R 
VOUT= 1 + A  VREF
 RB 
where VREF = 1.228V. The amplifier’s noninverting input
is internally connected to a soft-start circuit that gradually increases the reference voltage during startup. This
forces the output voltage to come up in an orderly and
well-defined manner under all load conditions.
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The MAX15004A/B/MAX15005A/B use an internal ramp
generator for slope compensation. The internal ramp
signal resets at the beginning of each cycle and slews at
the rate programmed by the external capacitor connected
to SLOPE. The amount of slope compensation needed
depends on the downslope of the current waveform.
Adjust the MAX15004A/B/MAX15005A/B slew rate up to
110mV/μs using the following equation:
Slope compensation (mV µs) =
2.5 × 10 −9 (A)
C SLOPE
where CSLOPE is the external capacitor at SLOPE in
farads.
Current Limit
The current-sense resistor (RCS), connected between the
source of the MOSFET and ground, sets the current limit.
The CS input has a voltage trip level (VCS) of 305mV. The
current-sense threshold has 5% accuracy. Set the currentlimit threshold 20% higher than the peak switch current at
the rated output power and minimum input voltage. Use
the following equation to calculate the value of RS:
=
R S VCS (IPK × 1.2)
where IPRI is the peak current that flows through the
MOSFET at full load and minimum VIN.
Maxim Integrated │ 14
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
When the voltage produced by this current (through the
current-sense resistor) exceeds the current-limit comparator threshold, the MOSFET driver (OUT) quickly
terminates the on-cycle. In most cases, a short-time
constant RC filter is required to filter out the leading-edge
spike on the sense waveform. The amplitude and width
of the leading edge depends on the gate capacitance,
drain capacitance (including interwinding capacitance),
and switching speed (MOSFET turn-on time). Set the RC
time constant just long enough to suppress the leading
edge. For a given design, measure the leading spike at
the highest input and rated output load to determine the
value of the RC filter.
Applications Information
The low 305mV current-limit threshold reduces the power
dissipation in the current-sense resistor. The current-limit
threshold can be further reduced by adding a DC offset
to the CS input from REG5 voltage. Do not reduce the
current-limit threshold below 150mV as it may cause
noise issues. See Figure 4. For a new value of the
current-limit threshold (VILIM_LOW), calculate the value of
R1 using the following equation:
Using the following equation, calculate the minimum
inductor value so that the converter remains in continuous
mode operation at minimum output current (IOMIN):
R1 =
4.75 × R CS
0.290 − VILIM_LOW
CURRENT-LIMIT
COMPARATOR
N
CCS
L MIN =
where:
VIN 2 × D × η
2 × f OUT × VOUT × I OMIN
VOUT + VD − VIN
VOUT + VD − VDS
I OMIN =
(0.1× I O ) to (0.25 × I O )
RCS
0.3V
Inductor Selection in Boost Configuration
and
REG5
R1
The MAX15004A/B/MAX15005A/B can be configured for
step-up conversion. The boost converter output can be
fed back to IN through a Schottky diode (see Figure 5) so
the controller can function during low voltage conditions
such as cold-crank. Use a Schottky diode (DVIN) in the
VIN path to avoid backfeeding the input source. Use the
equations in the following sections to calculate inductor
(LMIN), input capacitor (CIN), and output capacitor (COUT)
when using the converter in boost operation.
D=
VIN
MAX15004A/B
MAX15005A/B
Boost Converter
The higher value of IOMIN reduces the required
inductance; however, it increases the peak and RMS
currents in the switching MOSFET and inductor. Use
IOMIN from 10% to 25% of the full load current. The VD is
the forward voltage drop of the external Schottky diode,
D is the duty cycle, and VDS is the voltage drop across
the external switch. Select the inductor with low DC
resistance and with a saturation current (ISAT) rating
higher than the peak switch current limit of the converter.
RS
Figure 4. Reducing Current-Sense Threshold
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Maxim Integrated │ 15
MAX15004A/B-MAX15005A/B
VIN
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
DVIN
CIN
CVIN
1µF
CREG5
0.1µF
REG5
6
IN
MAX15004A/B
MAX15005A/B
VCC
OUT
CFF
CF
CS
11
10
VOUT
18V
D3
16
COUT
CVCC
4.7µF
RTCT
CT
RF
L
1
13
RT
DVBS
COMP
15
Q
RCS
12
RA
CCS
FB
SLOPE
SS
PGND
9
4
CSLOPE
RS
RB
CSS
Figure 5. Application Schematic
Input Capacitor Selection in Boost
Configuration
The input current for the boost converter is continuous
and the RMS ripple current at the input capacitor is
low. Calculate the minimum input capacitor value and
maximum ESR using the following equations:
C IN =
∆IL × D
4 × f OUT × ∆VQ
ESR =
where :
∆VESR
∆IL
(V − VDS ) × D
∆IL = IN
L × f OUT
VDS is the total voltage drop across the external MOSFET
plus the voltage drop across the inductor ESR. ΔIL is
peak-to-peak inductor ripple current as calculated above.
ΔVQ is the portion of input ripple due to the capacitor
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discharge and ΔVESR is the contribution due to ESR of
the capacitor. Assume the input capacitor ripple contribution due to ESR (ΔVESR) and capacitor discharge (ΔVQ)
is equal when using a combination of ceramic and aluminum capacitors. During the converter turn-on, a large
current is drawn from the input source especially at high
output to input differential. The MAX15004/MAX15005
are provided with a programmable soft-start, however, a
large storage capacitor at the input may be necessary to
avoid chattering due to finite hysteresis.
Output Capacitor Selection in Boost
Configuration
For the boost converter, the output capacitor supplies the
load current when the main switch is on. The required output capacitance is high, especially at higher duty cycles.
Also, the output capacitor ESR needs to be low enough to
minimize the voltage drop due to the ESR while supporting the load current. Use the following equations to calculate the output capacitor, for a specified output ripple. All
ripple values are peak-to-peak.
Maxim Integrated │ 16
MAX15004A/B-MAX15005A/B
ESR =
∆VESR
IO
I × D MAX
C OUT = O
∆VQ × f OUT
IO is the load current, ΔVQ is the portion of the ripple due to
the capacitor discharge, and ΔVESR is the contribution due
to the ESR of the capacitor. DMAX is the maximum duty
cycle at the minimum input voltage. Use a combination
of low-ESR ceramic and high-value, low-cost aluminum
capacitors for lower output ripple and noise.
Calculating Power Loss in Boost Converter
The MAX15004A/MAX15005A devices are available in
a thermally enhanced package and can dissipate up to
1.7W at +70°C ambient temperature. The total power
dissipation in the package must be limited so that the
junction temperature does not exceed its absolute maximum rating of +150°C at maximum ambient temperature;
however, Maxim recommends operating the junction at
about +125°C for better reliability.
The average supply current (IDRIVE-GATE) required by
the switch driver is:
IDRIVE−GATE
= Q g × f OUT
where Qg is total gate charge at 7.4V, a number available
from MOSFET data sheet.
The supply current in the MAX15004A/B/MAX15005A/B
is dependent on the switching frequency. See the Typical
Operating Characteristics to find the supply current
ISUPPLY of the MAX15004A/B/MAX15005A/B at a given
operating frequency. The total power dissipation (PT)
in the device due to supply current (ISUPPLY) and the
current required to drive the switch (IDRIVEGATE) is
calculated using following equation.
PT =VINMAX × (I SUPPLY + IDRIVE −GATE )
MOSFET Selection in Boost Converter
The MAX15004A/B/MAX15005A/B drive a wide variety
of n-channel power MOSFETs. Since VCC limits the OUT
output peak gate-drive voltage to no more than 11V, a
12V (max) gate voltage-rated MOSFET can be used without an additional clamp. Best performance, especially at
low-input voltages (5VIN), is achieved with low-threshold
n-channel MOSFETs that specify on-resistance with a
gate-source voltage (VGS) of 2.5V or less. When selecting the MOSFET, key parameters can include:
1) Total gate charge (Qg).
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4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
2) Reverse-transfer capacitance or charge (CRSS).
3) On-resistance (RDS(ON)).
4) Maximum drain-to-source voltage (VDS(MAX)).
5) Maximum gate frequencies threshold voltage
(VTH(MAX)).
At high switching, dynamic characteristics (parameters 1
and 2 of the above list) that predict switching losses have
more impact on efficiency than RDS(ON), which predicts
DC losses. Qg includes all capacitances associated
with charging the gate. The VDS(MAX) of the selected
MOSFET must be greater than the maximum output voltage setting plus a diode drop. The 10V additional margin
is recommended for spikes at the MOSFET drain due to
the inductance in the rectifier diode and output capacitor
path. In addition, Qg helps predict the current needed to
drive the gate at the selected operating frequency when
the internal LDO is driving the MOSFET.
Slope Compensation in Boost Configuration
The MAX15004A/B/MAX15005A/B use an internal
ramp generator for slope compensation to stabilize the
current loop when operating at duty cycles above 50%. It is
advisable to add some slope compensation even at lower
than 50% duty cycle to improve the noise immunity. The
slope compensations should be optimized because too
much slope compensation can turn the converter into the
voltage-mode control. The amount of slope compensation required depends on the downslope of the inductor
current when the main switch is off. The inductor downslope
depends on the input to output voltage differential of the
boost converter, inductor value, and the switching frequency. Theoretically, the compensation slope should be equal
to 50% of the inductor downslope; however, a little higher
than 50% slope is advised.
Use the following equation to calculate the required
compensating slope (mc) for the boost converter:
mc =
(VOUT
−
VIN ) × R S × 10 −3
(mV µs)
2L
The internal ramp signal resets at the beginning of each
cycle and slews at the rate programmed by the external
capacitor connected to SLOPE. Adjust the MAX15004A/B/
MAX15005A/B slew rate up to 110mV/μs using the following equation:
C SLOPE =
2.5 × 10 −9
mc(mV µs)
where CSLOPE is the external capacitor at SLOPE in
farads.
Maxim Integrated │ 17
MAX15004A/B-MAX15005A/B
Flyback Converter
The choice of the conversion topology is the first stage
in power-supply design. The topology selection criteria
include input voltage range, output voltage, peak currents
in the primary and secondary circuits, efficiency, form factor, and cost.
For an output power of less than 50W and a 1:2 input
voltage range with small form factor requirements, the
flyback topology is the best choice. It uses a minimum
of components, thereby reducing cost and form factor.
The flyback converter can be designed to operate either
in continuous or discontinuous mode of operation. In
discontinuous mode of operation, the transformer core
completes its energy transfer during the off-cycle, while
in continuous mode of operation, the next cycle begins
before the energy transfer is complete. The discontinuous
mode of operation is chosen for the present example for
the following reasons:
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Step 1) Calculate the secondary winding inductance for
guaranteed core discharge within a minimum offtime.
Step 2)Calculate primary winding inductance for sufficient energy to support the maximum load.
Step 3) Calculate the secondary and bias winding turns
ratios.
Step 4) Calculate the RMS current in the primary and
estimate the secondary RMS current.
Step 5)Consider proper sequencing of windings and
transformer construction for low leakage.
Step 1) As discussed earlier, the core must be discharged
during the off-cycle for discontinuous mode operation.
The secondary inductance determines the time required
to discharge the core. Use the following equations to calculate the secondary inductance:
(VOUT + VD ) × (D OFFMIN) 2
● It maximizes the energy storage in the magnetic component, thereby reducing size.
LS ≤
● Simplifies the dynamic stability compensation design
(no right-half plane zero).
D OFF =
● Higher unity-gain bandwidth.
A major disadvantage of discontinuous mode operation is the higher peak-to-average current ratio in the
primary and secondary circuits. Higher peak-to-average
current means higher RMS current, and therefore, higher
loss and lower efficiency. For low-power converters, the
advantages of using discontinuous mode easily surpass
the possible disadvantages. Moreover, the drive capability of the MAX15004/MAX15005 is good enough to drive
a large switching MOSFET. With the presently available
MOSFETs, power output of up to 50W is easily achievable
with a discontinuous mode flyback topology using the
MAX15004/MAX15005 in automotive applications.
Transformer Design
Step-by-step transformer specification design for a discontinuous flyback example is explained below.
Follow the steps below for the discontinuous mode transformer:
www.maximintegrated.com
2 × I OUT × f OUT(MAX)
t OFF
t ON + t OFF
where:
DOFFMIN = Minimum DOFF.
VD = Secondary diode forward voltage drop.
IOUT = Maximum output rated current.
Step 2) The rising current in the primary builds the energy
stored in the core during on-time, which is then released
to deliver the output power during the off-time. Primary
inductance is then calculated to store enough energy
during the on-time to support the maximum output power.
LP =
D=
VINMIN 2 × D MAX 2 × η
2 × POUT × f OUT(MAX)
t ON
t ON + t OFF
DMAX = Maximum D.
Maxim Integrated │ 18
MAX15004A/B-MAX15005A/B
Step 3) Calculate the secondary to primary turns ratio
(NSP) and the bias winding to primary turns ratio (NBP)
using the following equations:
=
N SP
NS
=
NP
LS
LP
and
=
NBP
NBIAS
11.7
=
NP
VOUT + 0.35
The forward bias drops of the secondary diode and the
bias rectifier diode are assumed to be 0.35V and 0.7V,
respectively. Refer to the diode manufacturer’s data sheet
to verify these numbers.
Step 4) The transformer manufacturer needs the RMS
current maximum values in the primary, secondary,
and bias windings to design the wire diameter for the
different windings. Use only wires with a diameter smaller
than 28AWG to keep skin effect losses under control. To
achieve the required copper cross-section, multiple wires
must be used in parallel. Multifilar windings are common
in high-frequency converters. Maximum RMS currents
in the primary and secondary occur at 50% duty cycle
(minimum input voltage) and maximum output power.
Use the following equations to calculate the primary and
secondary RMS currents:
=
IPRMS
I SRMS =
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
former turns ratio and the leakage inductance spike. The
MOSFET’s absolute maximum VDS rating must be higher
than the worst-case (maximum input voltage and output
load) drain voltage.
N

= VINMAX +  P × (VOUT + VD ) + VSPIKE
VDSMAX
N S

Lower maximum VDS requirement means a shorter
channel, lower RDS-ON, lower gate charge, and smaller
package. A lower NP/NS ratio allows a low VDSMAX specification and keeps the leakage inductance spike under
control. A resistor/diode/capacitor snubber network can
be also used to suppress the leakage inductance spike.
The DC losses in the MOSFET can be calculated using the
value for the primary RMS maximum current. Switching
losses in the MOSFET depend on the operating frequency,
total gate charge, and the transition loss during turn-off.
There are no transition losses during turn-on since the
primary current starts from zero in the discontinuous conduction mode. MOSFET derating may be necessary to
avoid damage during system turn-on and any other fault
conditions. Use the following equation to estimate the
power dissipation due to the power MOSFET:
PMOS = (1.4 × R DSON × I 2 PRMS ) + (Q g × VIN × f OUTMAX ) +
×I × t
×f
V
( INMAX PK OFF OUTMAX )
4
POUT
D MAX
×
0.5 × D MAX × η × VINMIN
3
I OUT
D OFFMAX
0.5 × D OFFMAX
3
The bias current for most MAX15004/MAX15005 applications is about 20mA and the selection of wire depends
more on convenience than on current capacity.
Step 5) The winding technique and the windings sequence
is important to reduce the leakage inductance spike at
switch turn-off. For example, interleave the secondary
between two primary halves. Keep the bias winding close
to the secondary, so that the bias voltage tracks the output voltage.
MOSFET Selection
MOSFET selection criteria include the maximum drain
voltage, peak/RMS current in the primary and the maximum-allowable power dissipation of the package without
exceeding the junction temperature limits. The voltage
seen by the MOSFET drain is the sum of the input
voltage, the reflected secondary voltage through trans-
www.maximintegrated.com
+
C DS × VDS 2 × f OUTMAX
2
where:
Qg = Total gate charge of the MOSFET (C) at 7.4V
VIN = Input voltage (V)
tOFF = Turn-off time (s)
CDS = Drain-to-source capacitance (F)
Output Filter Design
The output capacitance requirements for the flyback converter depend on the peak-to-peak ripple acceptable at
the load. The output capacitor supports the load current
during the switch on-time. During the off-cycle, the transformer secondary discharges the core replenishing the lost
charge and simultaneously supplies the load current. The
output ripple is the sum of the voltage drop due to charge
loss during the switch on-time and the ESR of the output
capacitor. The high switching frequency of the MAX15004/
MAX15005 reduces the capacitance requirement.
Maxim Integrated │ 19
MAX15004A/B-MAX15005A/B
An additional small LC filter may be necessary to
suppress the remaining low-energy high-frequency spikes.
The LC filter also helps attenuate the switching frequency
ripple. Care must be taken to avoid any compensation
problems due to the insertion of the additional LC filter.
Design the LC filter with a corner frequency at more than
a decade higher than the estimated closed-loop, unitygain bandwidth to minimize its effect on the phase margin.
Use 1μF to 10μF low-ESR ceramic capacitors and calculate the inductance using following equation:
L≤
1
4 × 10 3 × f C 2 × C
where fC = estimated converter closed-loop unity-gain
frequency.
SEPIC Converter
The MAX15004A/B/MAX15005A/B can be configured for
SEPIC conversion when the output voltage must be lower
and higher than the input voltage when the input voltage
varies through the operating range. The duty-cycle equation:
VO
D
=
VIN 1 − D
indicates that the output voltage is lower than the input for
a duty cycle lower than 0.5 while VOUT is higher than the
input at a duty cycle higher than 0.5. The inherent advantage of the SEPIC topology over the boost converter is a
complete isolation of the output from the source during a
fault at the output. The SEPIC converter output can be fed
back to IN through a Schottky diode (see Figure 6) so the
controller can function during low voltage conditions such
as cold-crank. Use a Schottky diode (DVIN) in the VIN
path to avoid backfeeding the input source.
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
and lower losses. The higher inductance value needed
for a lower ripple current means a larger-sized inductor,
which is a more expensive solution. The inductors L1 and
L2 can be independent, however, winding them on the
same core reduces the ripple currents.
Calculate the maximum duty cycle using the following
equation and choose the RT and CT values accordingly for a given switching frequency (see the Oscillator
Frequency/External Synchronization section).


VOUT + VD
D MAX = 

V
V
V
(V
V
)
+
+
−
+
OUT
D
DS
CS 
 IN−MIN
where VD is the forward voltage of the Schottky diode,
VCS (0.305V) is the current-sense threshold of the
MAX15004/MAX15005, and VDS is the voltage drop
across the switching MOSFET during the on-time.
Inductor Selection in SEPIC Converter
Use the following equations to calculate the inductance
values. Assume both L1 and L2 are equal and that the
inductor ripple current (ΔIL) is equal to 20% of the input
current at nominal input voltage to calculate the inductance value.
V
× D MAX 
=
L L=
=  IN−MIN

1 L2
 2 × f OUT × ∆IL 
0.2 × I OUT −MAX × D MAX 
∆IL =


(1 − D MAX ) × η


where fOUT is the converter switching frequency and
η is the targeted system efficiency. Use the coupled
inductors MSD-series from Coilcraft or PF0553-series from
Pulse Engineering, Inc. Make sure the inductor saturating
current rating (ISAT) is 30% higher than the peak inductor
current calculated using the following equation. Use the
current-sense resistor calculated based on the ILPK value
from the equation below (see the Current Limit section).
The SEPIC converter design includes sizing of inductors,
a MOSFET, series capacitance, and the rectifier diode.
The inductance is determined by the allowable ripple
I

× D MAX
current through all the components mentioned above.
=
ILPK  OUT −MAX
+ I OUT −MAX + ∆IL 
Lower ripple current means lower peak and RMS currents
 (1 − D MAX ) × η

www.maximintegrated.com
Maxim Integrated │ 20
MAX15004A/B-MAX15005A/B
MOSFET, Diode, and Series Capacitor
Selection in a SEPIC Converter
For the SEPIC configuration, choose an n-channel
MOSFET with a VDS rating at least 20% higher than the
sum of the output and input voltages. When operating at
a high switching frequency, the gate charge and switching losses become significant. Use low gate-charge
MOSFETs. The RMS current of the MOSFET is:
=
IMOS −RMS (A)
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
The series capacitor should be chosen for minimum ripple
voltage (ΔVCP) across the capacitor. We recommend
using a maximum ripple ΔVCP to be 5% of the minimum
input voltage (VIN-MIN) when operating at the minimum
input voltage. The multilayer ceramic capacitor X5R and
X7R series are recommended due to their high ripple
current capability and low ESR. Use the following
equation to calculate the series capacitor CP value.
I
× D MAX 
CP =  OUT −MAX

V
f
∆
×
CP OUT


D
(I
) 2 + (ILDC ) 2 + (ILPK × ILDC ) × MAX
 LPK

3
where ILDC = (ILPK - ΔIL).
Use Schottky diodes for higher conversion efficiency. The
reverse voltage rating of the Schottky diode must be higher than the sum of the maximum input voltage (VIN-MAX)
and the output voltage. Since the average current flowing
through the diode is equal to the output current, choose
the diode with forward current rating of IOUT-MAX. The
current sense (RCS) can be calculated using the currentlimit threshold (0.305V) of MAX15004/MAX15005 and
ILPK. Use a diode with a forward current rating more than
the maximum output current limit if the SEPIC converter
needs to be output short-circuit protected.
R CS =
0.305
ILPK
The MAX15004/MAX15005 maximum power dissipation
depends on the thermal resistance from the die to the
ambient environment and the ambient temperature. The
thermal resistance depends on the device package, PCB
copper area, other thermal mass, and airflow.
Calculate the temperature rise of the die using following
equation:
or
− ∆IL

)

where D is the duty cycle at the highest input voltage
(VIN-MAX).
www.maximintegrated.com
Power Dissipation
TJ = TC + (PT x θJC)
Select RCS 20% below the value calculated above.
Calculate the output current limit using the following
equation:
 D
I OUT −LIM = 
× (ILPK
(1 − D)
where ΔVCP is 0.05 x VIN-MIN.
For a further discussion of SEPIC converters, go to
http://pdfserv.maximintegrated.com/en/an/AN1051.
pdf.
TJ = TA + (PT x θJA)
where θJC is the junction-to-case thermal impedance
(3°C/W) of the 16-pin TSSOP-EP package and PT is
power dissipated in the device. Solder the exposed
pad of the package to a large copper area to spread
heat through the board surface, minimizing the case-toambient thermal impedance. Measure the temperature of
the copper area near the device (TC) at worst-case condition of power dissipation and use 3°C/W as θJC thermal
impedance. The case-to-ambient thermal impedance
(θJA) is dependent on how well the heat is transferred
from the PCB to the ambient. Use a large copper area
to keep the PCB temperature low. The θJA is 38°C/W for
TSSOP-16-EP and 90°C/W for TSSOP-16 package with
the condition specified by the JEDEC51 standard for a
multilayer board.
Maxim Integrated │ 21
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
VIN
2.5V TO 16V
L1
L11 = L22 = 7.5mH
C7
6.8µF
VOUT
D1
LL4148
VOUT
(8V/2A)
D2
STP745G
C4
22µF
D3
BAT54C
1
IN
VCC
C1
100nF
C1
6.8µF
16
C2
6.8µF
C5
22µF
C6
22µF
C3
6.8µF
CVCC
1µF
MAX15005A/B
2
ON
ON/OFF
OFF
3
OUT
4
REG5
RT
15kΩ
CT
150pF
6
7
REG5
13
C10
1µF
N.C.
RTCT
CS
RCS
100Ω
12
CCS
100pF
RS
0.025Ω
SGND
COMP
SYNC
8
14
SLOPE
REG5
5
STD20NF06L
OVI
PGND
CSLOPE
47pF
RG
1Ω
15
11
R3
1.8kΩ
SYNC
RSYNC
10kΩ
FB
SS
EP
10
C3
47nF
VOUT
C4
680pF
R2
15kΩ
R1
2.7kΩ
9
CSS
150nF
Figure 6. SEPIC Application Circuit
www.maximintegrated.com
Maxim Integrated │ 22
MAX15004A/B-MAX15005A/B
Layout Recommendations
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain current
often form high di/dt loops. Similarly, the heatsink of the
MOSFET connected to the device drain presents a dv/dt
source; therefore, minimize the surface area of the heatsink as much as possible. Keep all PCB traces carrying
switching currents as short as possible to minimize current loops. Use a ground plane for best results.
Careful PCB layout is critical to achieve low switching losses and clean, stable operation. Refer to the
MAX15005 EV kit data sheet for a specific layout example. Use a multilayer board whenever possible for better
noise immunity. Follow these guidelines for good PCB
layout:
1)Use a large copper plane under the package and
solder it to the exposed pad. To effectively use this
copper area as a heat exchanger between the PCB
and ambient, expose this copper area on the top and
bottom side of the PCB.
2)Do not connect the connection from SGND (pin
7) to the EP copper plane underneath the IC. Use
www.maximintegrated.com
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
midlayer-1 as an SGND plane when using a multilayer
board.
3) Isolate the power components and high-current path
from the sensitive analog circuitry.
4)Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation.
5)Connect SGND and PGND together close to the
device at the return terminal of VCC bypass capacitor.
Do not connect them together anywhere else.
6)Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs. 1oz) to enhance full-load
efficiency.
7) Ensure that the feedback connection to FB is short and
direct.
8)Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer for
SGND as an EMI shield to keep radiated noise away
from the device, feedback dividers, and analog bypass
capacitors.
9) Connect SYNC pin to SGND when not used.
Maxim Integrated │ 23
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Circuits
C12
R7
510Ω 220pF
VIN
(5.5V TO 16V)
C1
330µF
50V
C11
2200pF
100V
R16
10Ω
C18
4700pF
100V
1
R17
100kΩ
1%
R18
47.5kΩ
1%
VCC
2
VIN
R11
182kΩ
1%
C4
100pF
R1
8.45kΩ
1%
5
6
C5
1200pF
7
R19
10kΩ
1
2
VGRID
(60V/12mA)
C15
22µF
60V
16
C3
1µF
16V
R9
NU
C14
NU
R10
36kΩ
FILAMENT+
(3V/650mA)
15
C16
330µF
6.3V
R15
100Ω
R3
50Ω
N
C17
2.2µF
10V
OVI
14
FILAMENTD5
REG5
SLOPE
REG5
N.C.
CS
RTCT
13
C10
1µF
R5
1kΩ
12
C9
560pF
R6
0.06Ω
1%
SGND
COMP
8
D2
D4
PGND
4
R8
100kΩ
ON/OFF
OUT
3
C13
10µF
200V
D2
R2
560Ω
D1
MAX15005A/B
R12
12.1kΩ
1%
REG5
IN
C2
0.1mF
50V
VANODE
(110V/55mA)
11
R2
402kΩ
1%
SYNC
JU1
FB
SS
EP
10
C6
4700pF
9
C8
0.1µF
VANODE
C7
47pF
R13
118kΩ
1%
R14
1.3kΩ
1%
Figure 7. VFD Flyback Application Circuit
www.maximintegrated.com
Maxim Integrated │ 24
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Typical Operating Circuits (continued)
VIN
(4.5V TO 16V)
C1
10µF
25V
L1
10µH/IHLP5050
VISHAY
1
C11
0.1µF
IN
VCC
MAX15005A/B
R11
301kΩ
2
VOUT
R10
100kΩ
R8
153kΩ
D1
B340LB
OUT
3
4
5
R2
13kΩ
6
C3
180pF
7
REG5
N.C.
CS
RTCT
2
14
13
C10
1µF
R3
1kΩ
12
C4
100pF
R4
0.025Ω
SGND
COMP
1
Q
Si736DP
REG5
SLOPE
SYNC
8
C6
56µF/25V
SVP-SANYO
R1
5Ω
15
VOUT
(18V/2A)
OVI
PGND
C2
100pF
C10
1µF/16V
CERAMIC
ON/OFF
R9
10kΩ
REG5
16
11
R5
100kΩ
SYNC
JU1
FB
SS
EP
10
C9
0.1µF
9
C7
0.1µF
VOUT
C8
330pF
R6
136kΩ
R7
10kΩ
Figure 8. Boost Application Circuit
www.maximintegrated.com
Maxim Integrated │ 25
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Pin Configurations
TOP VIEW
IN 1
+
ON/OFF 2
OVI 3
16 VCC
IN 1
15 OUT
ON/OFF 2
+
16 VCC
15 OUT
OVI 3
14 PGND
14 PGND
MAX15004A
13 REG5
MAX15005A
SLOPE 4
RTCT 6
11 COMP
RTCT 6
11 COMP
SGND 7
10 FB
SGND 7
10 FB
SYNC 8
9 SS
SYNC 8
9 SS
SLOPE 4
N.C. 5
12 CS
EP
N.C. 5
PROCESS: BiCMOS
12 CS
Package Information
For the latest package outline information and land patterns
(footprints), go to www.maximintegrated.com/packages. Note
that a “+”, “#”, or “-” in the package code indicates RoHS status
only. Package drawings may show a different suffix character, but
the drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
www.maximintegrated.com
13 REG5
TSSOP
TSSOP-EP
Chip Information
MAX15004B
MAX15005B
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
16 TSSOP
U16+2
21-0066
90-0117
16 TSSOP-EP
U16E+3
21-0108
90-0120
Maxim Integrated │ 26
MAX15004A/B-MAX15005A/B
4.5V to 40V Input Automotive
Flyback/Boost/SEPIC
Power-Supply Controllers
Revision History
REVISION
NUMBER
REVISION
DATE
PAGES
CHANGED
0
1/07
Initial release
1
11/07
Updated Features, revised equations on pages 13, 20, and 21, revised Figure 8 with
correct MOSFET, and updated package outline
1, 13, 20, 21,
25, 28
2
12/10
Added MAX15005BAUE/V+ automotive part, updated Features, updated Package
Information, style edits
1–5, 9, 13, 21,
25–29
3
1/11
Added MAX15004AAUE/V+, MAX15004BAUE/V+, MAX15005AAUE/V+ automotive
parts to the Ordering Information
1
4
1/15
Updated Benefits and Features section
1
5
9/15
Miscellaneous updates
DESCRIPTION
—
1, 6, 9–11,
14–16, 18,
20–22
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
© 2015 Maxim Integrated Products, Inc. │ 27
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