JUNE 1974
© Copr. 1949-1998 Hewlett-Packard Co.
A New Generation in Frequency and Time
This new general-purpose electronic counter refines the
art of frequency and time measurements to an impressive
degree by the application of advanced technology.
by James L. Sorden
THE MODERN HISTORY of measuring time and
frequency with general-purpose electronic in
strumentation dates from the vacuum-tube counters
of the early-1950's. As technology moved from vacu
um tubes to transistors, better input amplifiers and
digital circuits were designed and the frequency lim
its of electronic counters advanced from 10 MHz to
50 MHz. Integrated circuits of the late 1960's brought
the cost of electronic counters down but left perfor
mance relatively unchanged. A significant jump in
the performance of general-purpose machines for the
measurement of time and time-related quantities
awaited the development of wideband digital logic
circuits and high-sensitivity de-coupled input am
In the story of the new Model 5345A Electronic
Counter (Fig. 1), development of these critical cir
cuits fills the largest chapter. However, the story
doesn't end there. At least four major technical con
tributions have gone into this radically new generalpurpose plug-in counter:
High-sensitivity dc amplifiers provide state-ofthe-art front-end signal conditioning for fre
quency and pulse measurements.
500-MHz E2L logic greatly improves resolution
for time interval measurements.
ROM-controlled reciprocal frequency measure
ments provide constant high resolution independ
ent of frequency.
White noise modulation of the time base clock
yields consistent picosecond-resolution time in
terval average measurements.
Model 5345A can measure any frequency from 50
/LiHz to 500 MHz. Above 1 Hz it gives nine-digit res
olution in a one-second measurement. It resolves
one-shot time intervals to two nanoseconds, and
measures the frequencies of RF pulses as brief as 50
nanoseconds. With its improved time interval aver
aging, it can resolve repetitive time intervals to one
picosecond. Frequency averaging, a technique new
with the 5345A, substantially improves precision in
measurements of the frequencies of pulsed micro
wave signals.
Cover: Model 5345A Elec
tronic Counter advances the
art of pulsed RF frequency
measurements with fre
quency averaging, a new
technique that improves
resolution even for very short
gate times. By controlling the
position of the gate (inten
sified section of bursts in cover photo) one can
measure the frequency profile within a burst.
In this Issue:
A New Generation in Frequency and
Time Measurements, by James L.
S o r d e n
p a g e
The 5345A Processor: An Example of
State Machine Design, by Ronald E.
F e l s e n s t e i n
p a g e
Time Interval Averaging: Theory, Prob
lems, and Solutions, by David C. Chu page 12
Third Input Channel Increases
Counter Versatility, by Arthur S. Muto page 16
A Completely Automatic 4-GHz
Heterodyne Frequency Converter, by
A l i B o l o g l u . . . . . p a g e 1 9
Interface Bus Expands Instrument Util
ity, by Bryce E. Jeppsen and Steven E.
S c h u l t z
p a g e
2 2
t Hewlett-Packard Company. 1974
Printed m U S A
© Copr. 1949-1998 Hewlett-Packard Co.
Fig. 1. Model 5345A Electronic
Counter is a new general-purpose
plug-in instrument for frequency
and time measurements. It has an
1 1 -digit display, 10-mV sensitivity,
and 500-MHz direct count capa
bility. The p/ug-in shown, Model
5354A, extends the frequency
measurement range to 4 GHz.
The new counter has two input amplifier chan
nels. Each has 10-mV sensitivity from dc to 500
MHz, is de-coupled, and has switchable input im
pedance: 1 megohm/30 pF or 50 ohms. The 5345A is
the first high-frequency counter that uses standard
oscilloscope probes, either 10:1 compensating highimpedance types or 50-ohm probes.
A new family of high-performance plug-ins is
under design, and two are already available. How
ever, virtually all of the existing plug-ins for the
5345A's predecessor, the widely-used Model 5245L,
function in the 5345A with improved speed and res
One of the two new plug-ins is the first automatic
heterodyne converter that can measure pulsed mi
crowave signals as well as CW signals (see article,
page 19). The second new plug-in adds a third 10-mV
500-MHz input channel, which is especially useful
in analyzing data communications circuits (see ar
ticle, page 16).
Full digital output and remote programming input
are available for the new counter, which is compa
tible with the HP interface bus (see article, page 22).
Frequency Measurements
close the measuring gate at subnanosecond speeds.
A new family of 500-MHz monolithic IC's were de
signed specifically for the 5345A and are manufac
tured by HP.
Time interval averaging is not new with the
5345A. What is new is that for the first time, pico
second-resolution time interval measurements can
be made with no harmonic problems. It can be
shown that averaging reduces the ±1 count (±2 ns)
quantization error to an rms value of 1/VrT count,
where n is the number of samples averaged. How
ever, for true averaging, the counter's gating signal
must maintain a uniform phase distribution around
the counter's clock period. Measurements made
when the distribution is not uniform will lead to an
"average" value that is biased away from the true
value no matter how many intervals are averaged.
This occurs whenever the repetition rate of the in-
500 MHz ± 2 x 10 9
1 10
1 x 10
To make a frequency measurement, the 5345A
first makes a period measurement, then inverts the
period to get frequency, which it displays. Nominal
gate times are set with a front-panel switch, and
these determine resolution. Fig. 2 compares the reso
lution of the new counter with that of conventional
direct counters. The key to the new counter's highfrequency capability and high resolution in brief
measuring periods is its 500-MHz clock coupled
with the reciprocal technique.
Time Interval and T.I. Average Measurements
Time interval measurements are made by the
5345A mainframe. To resolve single time interval
measurements meaningfully to two nanoseconds it
is not enough to have a clock period of two nanosec
onds (500 MHz). It is also necessary to open and
1 x 10 '
1 x 10
1 x 10 -
1 x 10
1 Hz
100 Hz
10kHz 1 MHz
Input Frequency
100 MHz 1 GHz
Fig. 2. Model 5345A is a reciprocal-taking counter. This
diagram compares its resolution with that of conventional
direct frequency counters for a one-second measurement.
© Copr. 1949-1998 Hewlett-Packard Co.
Pulsed RF
Signal f
Event Gate
500 MHz
Fig. 3. Synchronizers track RF bursts in frequency-average
measurements, a technique new with the 5345A. Benefit is
improved resolution over single measurements.
coming signal has any harmonic or submultiple rela
tionship to the counter's clock frequency.
The 5345A is the first counter to recognize this
problem and to guarantee truly useful averaged meas
urements under all circumstances. This is achieved
by deliberately adding jitter to the 500-MHz clock
by modulating it with white noise (see article, page
12). Jitter of slightly greater than one period at 500
MHz is generated by modulating the time base clock
with band-limited Gaussian noise.
Pulsed RF and Frequency Averaging
Because it needs only a brief measurement period,
the 5345A is well suited for frequency measure
ments of pulsed RF signals. Gate openings as brief
as 50 nanoseconds may be selected (one micro
second is usually the minimum found in electronic
counters). To make a single-shot carrier frequency
measurement of an RF burst, all that is necessary is
to select a gate time narrower than the burst. The
counter will trigger on the leading edge of the burst
and measure for the selected time.
With external gating, the counter can measure
the frequency profile within a burst. Or, to improve
resolution for very narrow pulses, it can average
frequency measurements over many pulses. This
capability, new with this instrument, produces res
olution never before attainable.
In frequency-average measurements, synchro
nizers (Fig. 3) track the RF bursts in such a manner
that the averaged sum appears in the event counter
and time counter. The pulsed bursts are clocked
through the event gate flip-flop synchronously with
an externally applied gate pulse. The event gate flipflop output is then used to synchronize the time
counter such that with a zero-biased jittered 500MHz time base clock, the averaging will increase the
resolution of a single shot measurement from ±1
count to an rms value of approximately 1/Vn count,
where n is the number of pulsed RF bursts averaged.
This improvement is qualified by the pulsed RF gate
pulse width and the noise bandwidth of the jittered
clock. Accuracy also improves, the amount of im
provement depending mainly on how precisely the
system has been calibrated.
The jittering is absolutely mandatory. Without
it, it is not possible to assure either unique samples
or a zero-biased answer. Fig. 4 gives theoretical
(colored line) and typical (black line) improve
ments of pulsed RF measurement resolution. The
curve flattening in the log-log resolution improve
ment plot occurs because of the band-limited
nature of the Gaussian noise used to jitter the time
base clock.
Counter Architecture
Fig. 5 is the 5345A's block diagram. System ar
chitecture is designed to support a wide variety of
measurement capabilities, present and future.
The arithmetic processor performs fifteen-digit
addition, subtraction, multiplication, and division.
It also does display formatting on mainframe meas
urements and sum-of-products calculations on
plug-in measurements. The T2L design operates at
a clock rate of 2 MHz. 4096 bits of read-only-memory
Gate Time 100 s 10 s 1s
1 0
1 x 10 » -t
1 x 10-1
© Copr. 1949-1998 Hewlett-Packard Co.
10ns 100ns is 10 ,-s 100,. s 1ms 10ms
Gate Pulse Width
Fig. 4. Resolution in pulsed RF measurements as a function
of external gate width. Colored lines show theoretical reso
lution. Black lines show typical resolution.
Other Controls Same
As in Channel B
Plug-in B
Channel B
Trigger Level
Attenuation Slope Level
or - 1 or x 20
1 Ml!
Function Gate
Strobed 11 Digit
LED Display
Output 10 MHz
1 V rms
into 50!)
Input 1-10 MHz
Gate Time
Minimum 50 ns
100 ns to 1000 s
10 MHz
Decade Steps
Buffer *1
Arbitrary External
Gate Time
Fig. 5. computation 5345/4 Counter's ROM-controlled processor performs various computation and
control elim For averaged measurements, jitter is added to the time base clock to elim
inate coherence errors.
are organized into 128 32-bit instructions. 62 quali
fiers are used in the algorithmic state machine de
sign (see article, page 9). Data storage and manipula
tion in this three-register machine are provided by
three 16 x 4-bit T2L RAMs.
Instruction inputs for the various arithmetic rou
tines and subroutines come from front-panel or re
mote-control function definition, plug-in arithmetic
requirements, output formatting instructions, and
input data (contents of the event counter and time
counter). For a frequency measurement, the event
counter totalizes the number of cycles for a given
unit of time defined by the time counter ±2 ns. Both
event and time counters consist of n-1013 scaling
registers. The highest-speed element is a 500-MHz
Hewlett-Packard E2L binary followed by an E2L dec
ade, the next quinary (25 MHz) is commercial
Schottky T2L and the last -HlO11 dividers consist of
HP PMOS -105 and -nlO6 LSI divider chips.
Input Amplifier and Attenuator Design
The 5345A's 10-mV, 500-MHz input amplifier is
a three-stage integrated-circuit design. Fig. 6 is its
circuit diagram. The 500-MHz bandwidth is a result
of three factors: the use of a fast 1C technology that
had previously been used only for digital circuits,
the somewhat unusual circuit configuration, and fre
quency compensation of the circuit on the 1C wafer.
The first stage is a series-shunt feedback pair, its
low frequency gain proportional to (RE1 + RF1)/REl.
Capacitor CEl shapes the response at high frequen
The second stage is a dc level shifter and low-out
put-impedance emitter-follower impedance convert
er. Its purpose is to minimize loading of the first
stage. The dc level shift is accomplished by using
the emitter-base reverse breakdown voltage of a
transistor, with the collector open to reduce offsets
caused by leakage currents.
The third stage is a compensated cascode ampli
fier that reduces the Miller effect for large load re
sistors. Third stage gain is determined by resistor RE2
and capacitor CE2.
An unusual aspect of this amplifier is the com
pensation of the overall response by adjusting CE1
and CE2 at the second metal layer. The second metal
layer is used to interconnect small MOS capacitors
to obtain a total capacitor value that will compensate
for wafer-to-wafer processing variations. The choice
of capacitor values is governed by measurements of
© Copr. 1949-1998 Hewlett-Packard Co.
38 mA
18 mA
Fig. contributions 5345A. input amplifier is one of the major design contributions in the 5345A.
the characteristics of a test transistor and a test re
sistor on the 1C wafer.
A photomicrograph of the integrated circuit is
shown in Fig. 6. The MOS capacitors are visible as
large rectangular metal areas. Also visible in the
photomicrograph are several metal strips used in
each capacitor, and the test transistor in the lower
right-hand corner of the chip. The input resistance of
the amplifier is greater than 10 kfl and the equivalent
input capacitance is less than 1 pF.1
This monolithic 1C amplifier is incorporated into
the input amplifier and attenuator design shown in
Fig. 7. To meet the requirements of low-frequency,
high-impedance measurements, the attenuator is
constructed with high-impedance components. Fre
quency compensation in the attenuator extends its
useful frequency response to beyond 500 MHz. The
attenuator impedance is changed to 50Ã1 by a
switched, compensated 50Ã1 termination.
To facilitate risetime or pulse width measure
ments, in which start and stop events occur on the
same signal line, the two input amplifiers may be
paralleled. In this mode, good termination is assured
by a 50Ã1 power splitter.
High-Speed Logic
The key to making high-resolution single-shot
time interval and/or fast bit-rate measurements on
computer or communications digital data is the
development of a family of digital logic of com
patible speed. Currently, several manufacturers have
some form of 1C binary capable of toggling at rates
in excess of 500 MHz. The problem that had to be
solved to make a higher-performance counter pos
sible was to develop logic circuits with enough gain
and bandwidth to match the performance of the prescaling binaries.
5345A logic circuits are of E2L (emitter-emitter
logic) type.2 E2L differs from ECL (emitter-coupled
logic) in that the E2L output is an open collector
whose logic swing is approximately 0V and —0.8V
into an external 50ÃÃ load, whereas ECL circuits have
emitter-follower outputs and logic swings of approx
imately -0.8V to -1.6V. By means of a combination
of techniques, including an unusual circuit configu
ration, useful frequency response of a typical E2L gate
has been extended to greater than 650 MHz. E2L cir
cuit outputs are compatible with 50fl microstrip
interconnect techniques, which are mandatory at the
500 megabit/second rates. Fig. 8 shows the 500-MHz
logic circuit on a multilayer printed-circuit board
using microstrip interconnect techniques.
Time Base
The accuracy of any counter depends ultimately
on the stability of the time base. The standard time
base for the 5345A Electronic Counter is a state-ofthe-art 10-MHz oven-stabilized oscillator. The 10
MHz is multiplied to 500 MHz for clock purposes.
Aging rate is under 5 parts in 1010 per day, and short-
© Copr. 1949-1998 Hewlett-Packard Co.
Fig. switchable, or amplifier and attenuator. Input impedance is switchable, 1 Mil/30 pF or 50Ü.
term stability is better than one part in 1011 for a
1-second average. This is the most stable standardequipment oscillator in present-day commercial in
The author is indebted to Merrill Brooksby, Dexter
Hartke, John Dukes, and Darwin Throne for their
leadership and encouragement throughout the years
of development.
In addition to the several engineers who have
authored other articles appearing in this journal,
a special word of thanks must go to Steve Upshinsky
for his design of the input amplifier and to Jose FurIan for his pioneer work on 500 MHz E2L gates. John
Gliever's RF design with associated noise modula
tion scheme made true time interval averaging pos
sible. The product design by Keith Leslie afforded
the electronic designers great latitude.
Successful production has been assured by the
efforts of Tom Coates, who designed most of the
production testing techniques, and Phil Deaver,
whose production engineering effort has been out
standing. Holly Cole and Dick Holmes have assured
us of a fine marketing and service program. Finally,
a personal word of thanks to Dan Lansdon, whose
early encouragement made it possible for me to en
joy the many opportunities at Hewlett-Packard. T
1. R.I. Ollins and S.f. Ratner, "Computer-Aided Design
and Optimization of a Broad-Band High-Frequency Mono
lithic Amplifier", IEEE Journal of Solid-State Circuits.
Vol. SC-7, pp. 487-492, Dec. 1972.
2. M. Brooksby. et. al., Fast Logic Extends Range of HighFrequency Counters". Electronics, Vol. 43, pp. 62-67,
Dec. 7, 1970.
3. "Time Interval Averaging", Hewlett-Packard Applica
tion Note 162-1.
© Copr. 1949-1998 Hewlett-Packard Co.
Fig. 8. 5345A main gate, a 500MHz logic system. Multilayer
printed-circuit boards and microstrip techniques contribute to
the wide bandwidth.
HP Model 5345A Electronic Counter
Frequency. Frequency Average. Period,
Period Average
luency and period are measured Dy measuring the total elapsed limi
i integral number ol cycles N cri me input waveform Computation in
he quantities ol N and T provides direct readout of either frequency o
RANGE. 50 uHz to 500 MHz 2 nsec to 20.000 seconds
MEASUREMENT TIME: Consists ot GATE TIME plus the lime required
to reach the ne« STOP trigger level When m MIN the GATE TIME s
ess than 50 nanoseconds Decade GATE TIME ranges com too nsec to
1000 sec
ACCURACY: Resolution ts nine digits per second o> measurement time With
DISPLAY POSITION switch m AUTO !he least significant digit error is
* 1 count it the most significant digit is i through 4 and 1 2 counls if the most
significan! rj.g.1 is 5 through 9 Accuracy is ± LSD counts - time base accur
acy - Ingger error
Time Interval. Time Interval Average
RANGE: '0 nsec to 20 000 sec
TRIGGER PULSE WIDTH' ' nsec minimum width input at minimum voltage
TIME INTERVAL ± trigger errpr - 2 ns - time Base accuracy
TIME INTERVAL AVERAGING * (Ingger error • 2 nsl «/intervals averaged
•_ 7 nsec * time base accuracy
Hot affected by harmonics of dock frequency
TIME INTERVAL AVERAGE 1 12 nsj v'intervals averaged - 2 DS
Ratio B/A
ANGE Both channels accept dc to 500 MHZ
CCURACY -LSD- trigger error
EASUREMENT TIME Measurement time is equal to the GATE TIME se
•ecled times 500 MHz frequency of channel B mpul
RANGE: dc to 500 MH;
OUTPUT: Output frequency equals input frequency divided by scaling tactoi Rear
aa-ie BNC supplies 80"= duty cycle TTL compatible pulses
Input Channels A and B
RANGE: 0 lo 500 MHz dc coupled SOU and 1 Mil 4 MHz lo 500 MHz ac coupled
501!. 200 Hz to 500 MHz ac coupled 1 Mil
IMPEDANCE- Selectable i Mit shunted by less man 30 pF or sou innmmaij
SENSITIVITY XI 10 mV rms sine wave and 30 mV peak-to-peak pulse X20
200 mV "ns sine wave and 600 mV peak to-peaM pulse
TRIGGER LEVEL: Continuously adjustable 10 more than cover the DYNAMIC
RANGE i • 0 5V dc times the attenuate» setting) Adjustment is nonlinear with
more settattiity in the more sensitive region
PRESET: Centers trigger «vel about dc at 25 C
DfltFT: • lOmVdc ma* ffC to 55 C
OUTPUT. Hear panel BNC connectors Dring Out CHAN A TRIG LEVEL ano CHAN
SLOPE: Independent selection of positive or negative slope
Common Input
In this mod* me s>gnai is applied 10 channel A through a power splitter which
equalizes impedances and delays lo the input amplifiers Channel B input is
disabled Both input impedance switches should be m the same position All
RANGE, ac coupled 50u 4 MHz to 500 MHZ ac coupled 1 Mil 300 Hi to 500
IMPEDANCE: 5011 remains 50!l 1 Mil Becomes 500 Kit Shunted by 60 pF
BOU X1 20 mV rms sine «rave and GO mv peaK-to-peak pulse
X20 400 rnv rms sine wave and ' 2V peak-to-peak pulse
T Mu No change
TRIGGER LEVEL: Continuously adguStaoie over the range of • tv dc m SOU and
•05V dc in 1 Mil multiplied by the attenualor setting
ANGE: Both inputs may have repetition rates from « to 500 MHz
ODES: A A • 6 and A B is determined by a -ea< panel switch
ESOLUTION Mot affected by GATE TtME setting Resolution is one counl up
to e-evei digits
ACCURACY Comentem pulses may b« applied to both inputs One count is re
quired to initiate each input
manually reset or an EXTERNAL ARM signal >s applied Number of readings
per second ml generally be limited by the output device i e 5050B Printer or
9820A Calculator m COMPUTER DUMP mode the counter ca" take up
to several thousand readings pe> second
EXTERNAL ARM INPUT: Counter can be armed Dy a 1 0V signal applied lo the
rear panel 5011 input The signal should be applied tor more than 50 nsec
Minimum lime between EXT ARM and acceptance of start pulse is
EXTERNAL GATE INPUT: Same conations as lor EXT ARM Minimum time
between EXT GATE ant) acceptance of start pulse is - 20 nsec
GATE OUTPUT: -1 volt into SOU
TIME 10 Standard High Statthty Time Base Crystal Frequency. 10 MHi
Aging Rate < 5 • 10 ~10 per day
Short Term I • 10 ~ ' ' tor 1 sec average
For oscillator off time less than 24 hrs
SELF all A 100 MHz signal >s mt«rnaiiy applied for testing all functions Push
mg RESET illuminates all segments of display digits Seven internal diagnostic,
switches are provided 'or verifying the operation of the input amplifiers digital
front end processor and plug-ins
' ill required from source of 1, 2. 25. S Or 10 MHz ±5.0 * 10~^ Input
can be sine or square wave
FREQUENCY STANDARD OUTPUT: -1V rms into SOU at 10 0 MHz sine wave
OPTION 001: Room Temperature Time Base
OPTION 002: Same as 5345A but with no input amplifiers Signal must be applied
through plug -in
OPTION 010: Digital output only HP Interlace Bus format ASCII talk only Useful
with 59301A ASCII -to-Paraliei Converter and 50SOB or 50S5A Digital Printers
OPTION Oil: Digital Inpul Output Full compatibility with HP Interface (ASCIIi
Bus Provides digital output as well as input 'or control over all functions except
input amplifier Allows counter to operate with all ASCII Programmable Modules
i see 59300 seriesi and Conlroners sucr< as card readers calculators and
2100 series computers
Accessories Available
10593A 5345A TO M20A INTERFACE KIT: Inctudes ASCII I O card and PCll
ROM 10631B 10631C documentation and 5345A Self Test Software PacMaoe
To be used with 5345A Option 01 1 and 9820A Calculator
10590A PLUG-IN ADAPTER: increases usefulness Of 5345A Dy providing mte'
they do in the S245L
PRICE IN U.S.A.: 5345A. $3450
5301 Stevens Creek Boulevard
Santa Clara California 9505C USA
James L. Sorden
Jim Sorden is project manager for the 5345A Electronic
Counter. He first joined HP in 1962 after receiving his BSEE
degree from the University of Wisconsin. After a while the
promise of greener pastures lured him elsewhere, but by
October. 1964 he was back to stay. Jim's responsibilities
at HP have been heavily concentrated in digital circuit and
systems design and high-frequency integrated-circuit de
sign. Earlier he served as project leader for a multichannel
analyzer system An amateur winemaker who also enjoys
gardening, sailing, and skiing, Jim is a native of the state
of Wisconsin. He and his wife and two small children live
in Saratoga, California.
© Copr. 1949-1998 Hewlett-Packard Co.
The 5345A Processor: An Example of State
Machine Design
by Ronald E. Felsenstein
THE 5345A PROCESSOR is an example of state
machine design.1 The first step in the design of
such a machine is to draw a flow diagram showing all
processor functions. Fig. 1 is a section of a typical
flow diagram.
The rectangular boxes in the flow diagram repre
sent states. The machine spends sufficient time with
in a state to execute all the commands that are active.
A command might be something very simple, such
as resetting a flip-flop or incrementing a counter. A
more sophisticated example, one used in the 5345A
processor, is a subtraction involving two serial
strings of 16 binary-coded-decimal (BCD) digits. Sub
traction takes a combination of commands. Two com
mands gate 16 clock pulses to shift the data through
a BCD adder/subtractor, one command puts the BCD
adder/subtractor in "subtract" mode, and several
commands route data so the resulting difference is
fed back to one of the subtracting storage elements
while the contents of the other storage element re
main unaltered.
Once sufficient time has elapsed for all of the com
mands to have been executed, one of two new sets of
commands will be executed. Which new set is ex
ecuted depends on the status of a qualifier.
An example of a 5345A qualifier is the status of a
carry flip-flop that determines whether the result of
a subtraction is positive or negative. Some qualifiers
require considerable hardware for generation. For
example, the display is made more readable by an
automatic display formatting algorithm (AUTO),
which gives the displayed digits an annunciator mul
tiplier computed so the decimal point is within the
first three positions to the right of the most significant
digit. The position of the most significant digit varies
according to the number of digits of resolution (in
AUTO, the least significant digit is always right jus
tified within the display). The generation of the quali
fier that determines whether the AUTO criterion has
been met requires a subtracter that outputs the dif
ference between a counter containing the decimal
point code and a four-bit latch storing in binary form
the number of digits of resolution. The output of the
subtracter is then further examined with combina
tional logic to determine whether it ranges between
— \ and -3. If the difference is between -1 and -3,
the AUTO criterion has been met. This qualifier is fur
ther complicated by having to recognize three special
cases where the AUTO criterion of decimal point posi
tioning cannot be met.
Each state box within the flow diagram is coded to
distinguish it from any other state box. These codes
are represented in Fig. 1 by the encircled letters. Thus
if state X occurs at time T, then Y or Z will
occur at time T + l. The process is continuously re
peated as the flow diagram is sequentially executed.
ROM Addressing
When the number of states and commands is large,
as it is in the 5354A, it makes sense to use read-only
memories (ROMs) to store the flow-diagram program.
A block diagram of the first scheme that was consid-
© Copr. 1949-1998 Hewlett-Packard Co.
Y )
States (T-1)
Fig. 1. A segment of the flow diagram of a state
machine. As the commands of the present state are
completed, a qualifier is tested to determine which of
two possible states will be the next state of the machine.
requiring only six ROM outputs (see Fig. 3).
The reduction of ROM outputs was achieved by
coding the STATE (T+l) TRUE and STATE (T + l) FALSE
codes so they differ in only one bit. That bit is estab
lished directly by the selected qualifier. This means
that for STATE (T+i) addressing, only S-l ROM out
puts are required.
The final reduction was achieved by eliminating
the TEST field entirely. This was possible because of
the large number of unique qualifiers that the 5345A
flow diagram definition required. Had a separate TEST
field been used, the number of TEST bits would be
S-l; thus it made sense to address the selector with
the S-l bits directly.
Fig. 2. State-qualifier-pair ROM address structure, a
typical way to implement a state machine.
Command Generation
ered for the 5345A, a state-qualifier-pair ROM ad
dress structure,1 is shown in Fig. 2. The storage cir
cuit contains the code X at state time T. There are
S bits in the state code; these become the inputs or ad
dresses to the ROM. For every address presented, the
ROM has been programmed to produce four output
fields. The COMMANDS field activates all the lines
needed to satisfy the requirements of a flow-diagram
state-box description.
The STATE (T+i) TRUE field contains the S-bit code
representing the state at time T+l if the qualifier fol
lowing state X is true; for Fig. 1 this would be the
code representing state Z . Similarly, the STATE
(T+i) FALSE field contains S bits and represents the
qualifier-false address or the Y code in Fig. 1.
The TEST field addresses a selector (multiplexer)
to select the qualifier required by the flow diagram.
The test field requires T bits such that 2Ts=Q, where
Q is the number of unique qualifiers in the flow dia
gram description.
The qualifier selected then controls the switch so
the STATE (T+i) TRUE code is supplied to the storage
circuit if the qualifier is true, and the STATE (T+i)
FALSE code is supplied if the qualifier is false. Upon
completion of state X , the ROM CLOCK signal ini
tiates the new state, which Will be either Y) or
Z . This cycle repeats for every state box in the
flow diagram until the flow diagram is completed.
The hardware disadvantage of the state-qualifierpair ROM address structure is that many ROM out
puts are used just for stepping through the flow dia
gram. Furthermore, at a given state time, the S-bit
field that does not get selected by the qualifier-con
trolled switch serves no useful purpose. It is unlikely
that any of the STATE (T+i) TRUE or STATE (T+i) FALSE
bits could also serve as command lines.
In the 5345A, the state-qualifier-pair ROM address
structure would have used 20 ROM outputs. The ad
dress structure that was finally used is a modification
When the definition of the 5345A was complete,
the S field was seven bits long and the COMMAND field
was 39 bits long. The only circuits that were com
mercially available in both ROM and PROM versions
had 1024 bits arranged with an eight-bit input ad
dress and four-bit output (256X4 organization). To
convert to a seven-bit input, eight-bit output circuit
(128x8 organization), the ROM is given two ad
dresses at every state time and the ROM outputs dur
ing the first address are stored (see Fig. 4).
A new state code is presented to the ROM every
time a ROM CLOCK pulse occurs and MSB (the most
significant bit of the ROM address) is high. After the
ROM outputs reach steady state, a WORD DOUBLING
CLOCK pulse causes them to be stored and the
MSB = i outputs to be generated. Then MSB goes low,
offering a new address to the ROM and causing a new
set of ROM outputs, called MSB=o outputs, to be
'The in (Programmable Read-Only Memory) saves time and money in the design
stage PROMs are also useful for implementing customers' special requirements.
© Copr. 1949-1998 Hewlett-Packard Co.
Fig. 3. 5345A ROM address structure reduces ROM
output requirements.
Single Pulse
Four 256 >
Three 3 to 8
Line Decoders and
Combinational Logic
As the flow diagram of a state machine is devel
oped, hardware requirements become apparent: stor
age elements, flip-flops, counters, and so on. During
definition of the 5345A, flow diagram and hardware
development went hand in hand, each affecting the
minimization of the other.
The processor was built with TTL circuits. To
handle 15-digit binary-coded-decimal arithmetic,
there are three 16-by-4-bit random-access memories
and a BCD adder/subtractor. Two six-bit counters are
used to store decimal point information. Latches
store the multiplier for the annunciators, the sign,
and the overflow light of the display. Additional
counters, latches, and flip-flops are used for special
purposes such as counting and storing the number of
digits of resolution.
Fig. 4. 5345A command generation scheme.
Because the MSB=I outputs are stored, they have
no transients, but the MSB=o outputs cannot be re
lied upon while MSB is high. To take care of the
MSB=o output transients, a signal called SINGLE
PULSE is generated after the MSB=o outputs achieve
steady state. The MSB = o outputs are gated with
SINGLE PULSE so the hardware looks at these out
puts only when their information is correct.
Even after doubling the number of ROM outputs, if
all the 39 commands had been derived directly from
the ROM in addition to the S-l or six bits for the
STATE (T+i) field, the processor would have re
quired 6K of ROM. This would have increased the
price and power consumption of the processor.
With the aid of a "bit packing" computer program
developed at HP Laboratories, the amount of ROM
was reduced to 4K. The first step in the bit packing
procedure was to feed the program with a complete
listing of the STATE (T) and STATE (T+i) codes, and
the status of each command bit (active, inactive, or
don't care). The program then reduced the number of
ROM outputs required by finding sets of mutually ex
clusive commands that could be derived from a de
coder, and commands that could be generated by
simple logic-gate combinations of other commands,
STATE (T), and/or STATE (T+l) bits.
Originally, there were 45 ROM outputs: 39 com
mands and six STATE (T+i) lines. After bit packing
only 29 outputs were required. On four IK ROMs
with 128x8 organization, three ROM outputs re
main free for possible future use.
Credit for design contributions is due Jim Sorden,
Bob Livengood, and Tommy Thomason. Special
thanks go to Gerry Alonzo for his help in "packing
ROM bits" with the computer program, and Chuck
McWilliams for his idea that made our ROM simula
tor possible. Finally T I would like to thank Tom Coates
for making the processor boards computer testable. T
1. Christopher R. Clare, "Designing Logic Systems Using
State Machines", McGraw-Hill, 1973.
Ronald E. Felsenstein
Ron Felsenstein received his BS degree in electrical engineer
ing from Massachusetts Institute of Technology in 1969 and
joined HP the same year. Designer of the digital processor for
the 5345A counter, he has now been named production
engineer for lasers and logic test instruments. Ron was born
in Montevideo, Uruguay. He and his wife recently returned
there for a visit during a two-month tour of Uruguay, Mexico,
and six other countries in Central and South America. At
home in Santa Clara, California, Ron is currently busy re
building a 1961 Mercedes-Benz, but he also enjoys camp
ing, tennis, hiking, and skiing.
© Copr. 1949-1998 Hewlett-Packard Co.
Time Interval Averaging: Theory,
Problems, and Solutions
by David C. Chu
Non-Averaging Because of Coherence
economical way to increase resolution in meas
uring repetitive time intervals. The idea is quite
simple: the same interval is measured repeatedly
and, given some degree of independence between
measurements, the ±1 count quantization error in
each measurement is statistically reduced if the
average measurement value is used to estimate the
There are many pitfalls in making measurements
this way. My purpose here is to point these out and
describe the approaches used in the 5345A to solve
these problems.
Another fundamental problem of time interval
averaging occurs when the time intervals are re
peated at a rate coherent with the clock frequency.
For example, if the rate is a submultiple of the clock
frequency, the occurrence of the time interval rela
tive to the clock phase is the same for each measure
ment. Hence, all the measurements read exactly the
same and no statistical averaging takes place. In this
case, the quantizing error for a million measure
ments is no different from that for a single
Coherence, unfortunately, is not limited to submultiples. There are other rates at which only partial
averaging takes place. For example, rates given by
ro/(Q + 1/2), where f0 is the clock frequency and Q is a
positive integer, give rise to two alternating discrete
clock phases separated by T0/2. Averaging over a
large number of measurements is no better than aver
aging over two successive measurements.
In general, we can partition the coherent rates into
classes. A "class-M" rate results in the time inter
val's occurring at M discrete phases of the clock. If
one is interested in gaining resolution improvement
by, say, a factor of 100 over the clock period, a
class-M rate where M is under 100 would be unac
ceptable. A class-M rate is given by:
The Folly of Direct Gating
In one all too obvious implementation of time in
terval averaging, the time interval repeatedly en
ables a gate through which the clock pulses are
passed and counted. Unfortunately, when the gate
turns on and off, partial clock pulses are generated
and fed to the counting circuits. The average value
obtained this way is as much a function of the re
sponse of the counter to partial pulses as it is to the
width of the interval. Because this response is diffi
cult to characterize reliably and even harder to con
trol accurately, one can attach no meaningful signi
ficance to the average value so obtained. In general,
for averaging over a large number of measurements,
a biased value, which can differ significantly from
the true time interval value, will be approached. The
difference can be expressed as:
partial pulse bias = Tu
fR(M) =-
M = 1,2,3...
where fR(M) is a class-M rate, Q, L, and M are nonnegative integers, and L«M. Furthermore, L and M are
co-prime, that is, they have no common factors. For
M = 1, the class of submultiples is generated.
These rates are very numerous. In fact, there is an
infinite number of rates for each class.
where T0 is the clock period, d is the duty cycle of
the clock in percent, and r is the fraction of the full
clock pulse below which the counter does not re
spond. The parameter r, which is always between
0 and 1, cannot be controlled precisely.
The use of a synchronizer in HP averaging coun
ters eliminates the partial pulse problem by reducing
the effective duty cycle (d) to zero.1 The decades are
fed full-width pulses under all conditions. The re
mainder of this article assumes the use of synchronizers.
Coherence Bandwidth
If one is interested in gaining resolution by a fac
tor of N, how far must the time interval rate depart
from one of the coherent rates fR to assure proper
averaging? Straightforward analysis of the phase of
the intervals at these frequencies shows that for a
© Copr. 1949-1998 Hewlett-Packard Co.
class-M rate fR, a departure of ±AfR from fR can result
in almost perfect averaging by the factor N, where
AfR is given by:
2 ns
AfR =
The coherence bandwidth is 2AfR.
In terms of fractional frequency stability AfR/fR, the
stability that would cause the non-averaging effect
Conversely, any stability worse than this destroys
the coherence and allows the reduction of measure
ment quantization error by statistics.
The coherence bandwidth is large for high rates
and low M, and non-averaging can often be observed
without instrumentation. Submultiples (M = l), for
example, give rise to measurements always equal to
or close to whole clock periods, an easily discernible
effect. Partial averaging is more subtle, and the ex
perimenter is often led to accept a result which does
not give an adequate interpolation factor.
Rms Phase Variation
Fig. 1. Coherence bias error in time interval averaging is
reduced by increasing random phase modulation of the
counter time base.
error demands just the opposite.
Modulation Bandwidth
Time Base Random Phase Modulation
There are also two opposing requirements on the
phase modulation bandwidth. With a large band
width, time intervals can arrive rapidly and still
maintain relative independence between measure
ments. On the other hand, for very large modula
tion bandwidths and relatively long time intervals,
there may be only limited correlation between the
time base at the start edge and at the stop edge, and
the measurement accuracy may be degraded by time
base short-term instability.
Quantitatively, the relationship is as follows. For
a given time interval T seconds, modulation band
width fc (Hz), and rms phase modulation cr (cycles),
the rms error caused by time base uncertainty in a
single measurement is given by ETB, where
Coherence between the time interval pulse train
and the time base clock pulse train can be destroyed
by introducing random phase modulation to either
or both of the trains, allowing meaningful time inter
val averaging measurements to be made without
regard to the time-interval rate. In the HP 5345A
Counter, phase modulation is deliberately intro
duced into the clock pulse train when making time
interval average and pulsed RF measurements. The
modulation is random Gaussian noise band-limited
to approximately 3 kHz.
The bias errors caused by the non-averaging ef
fect are functions of many parameters, including the
time interval rate, the fractional part of the time in
terval (obtained by subtracting all whole clock
periods from the time interval), and the interval-toclock phase relationship. However, the worst case
bias error, EB, can be expressed simply as a function
of cr, the rms value of the phase modulation:
Jf '
rms value of ETB = Tncr[2(l — e T )]
seconds. For a given time interval, this error is small
er for a smaller modulation bandwidth. Thus a
small modulation bandwidth is desirable.
With averaging, this error is reduced by the factor
1VN, where N is the number of independent mea
surements. Not all measurements averaged are in
dependent if the time interval rate exceeds approxi
mately twice the modulation bandwidth. In this latter
respect, a large bandwidth is desirable because it
makes full use of all measurements made at higher
Notice that ETB increases linearly with cr, the rms
phase modulation, in contrast with the coherence
error, which decreases with cr.
1/2] (5)
where EB is the non-averaging bias error and cr is
the rms value of the phase modulation in units of
2?7 radians or periods of the time base clock.
Fig. 1 shows a plot of bias error EB versus cr in ra
dians, assuming a 2-ns clock period, and a time
interval given by 2(Q + V2) ns. The bias error de
creases rapidly with increasing phase modulation.
This indicates that the phase modulation should be
large. However, as will be shown, another type of
© Copr. 1949-1998 Hewlett-Packard Co.
Normal ±1 Count Quantization Error
Even if the time interval rate is incoherent and
time base uncertainty error is negligible (such as
when measuring short time intervals), an error is
expected in time interval average measurements be
cause of the normal ±1 count quantization error
in each measurement. For a given time interval T =
TO (Q + F), where Q is an integer and F a proper frac
tion, this quantization error can take on only two
fixed values, FT0 and (F-1)T0, with probabilities
(1— F) and F respectively. The mean of this distribu
tion is zero, and the standard deviation (rms value)
is T0[F(1 -F)]1'2. The worst case occurs when F is 1/2,
with the corresponding worst case rms quantization
error of To/2. With proper averaging over N inde
pendent measurements, this rms error is reduced by
the familiar factor 1/VÑ.
10 MHz
10 MHz Tank
with Varactor
Phase Modulator
Fig. 2. Implementation of time-base-clock random phase
modulation in the HP 5345A. The switch is closed only for
time interval average and pulsed RF measurements.
Measurement Error Summary
trigger error, and plain old thermal noise. These
errors are important, but have been excluded from
this discussion, which is limited to those errors that
are unique to the time interval averaging process.
In the 5345A Counter, the phase modulating sig
nal is derived from noise generated by a zener diode.
The noise is amplified and filtered before being used
to modulate the phase of the clock at 10 MHz with an
rms value of approximately 7°. The frequency multi
plier chain effectively increases this value by a factor
of 50 to about 350°. The noise voltage level is accu
rately controlled at all times by a feedback loop. A
block diagram of the implementation scheme is
shown in Fig. 2.
Three types of errors in time interval averaging
have been discussed. They are non-averaging bias
error, time base short-term uncertainty error, and
normal quantization error.
The first error is caused by coherence and can be
as large as one whole clock count. It is independent
of the number of measurements averaged. This error
can be reduced to a negligible level according to
Fig. 1 by randomly phase modulating the time base
clock. In the 5345A Counter, random phase modula
tion of 0.8 cycle rms minimum allows meaningful
time interval average measurements without regard
to the time interval rate.
The second error is a result of time base uncer
tainty caused by the random phase modulation in
troduced. For measuring time intervals less than
7 /us with the 5345A, this time base error is com
pletely dominated by the quantization error, and is
therefore negligible. For measuring time intervals
much larger than 7 /us, the measurement deviation
is increased by a factor of approximately 2.7 above
that due to the normal ±1 count quantization error.
This increase can be nullified by averaging more
The third type of error is a result of quantization
to whole numbers in the counting process and is a
function of many parameters. The worst case occurs
when the time interval has a value half way between
whole clock counts, giving an rms error Tg/2. Like
time base uncertainty error, quantization error is re
duced if averaged over N independent measure
ments by the factor 1/VÑ.
There are other errors in time interval measure
ments, caused primarily by non-ideal circuit com
ponents. Examples are start-stop channel mismatch,
FM versus PM
There are two fundamental reasons why phase
Without Time Base
With Time Base Random
Phase Modulation
12 ns
Time (minutes)
10 ns
Fig. 3. Effectiveness of time base random phase modulation
is demonstrated by this time record of the counter reading
during a time interval average measurement. Time intervals
arrive at a rate of 50 MHz +0.1 Hz, which is nearly coherent
with the 5345A clock frequency of 500 MHz. Without modula
tion, the reading is either 10 ns or 12 ns. Modulation results in
a true reading of approximately 1 1 ns.
r 1 count is actually a misnomer when synchronizers are used, because they make it impossible
to have an error exactly equal to one count.
© Copr. 1949-1998 Hewlett-Packard Co.
base drifts. Zero mean modulation is difficult to ob
tain with nonlinear modulation circuits. With phase
modulation, this zero mean value is not a necessity,
because any constant phase offset is effectively can
celled by the start/stop process.
modulation (PM) of the clock is superior to fre
quency modulation (FM) for time interval averaging.
First of all, it is the phase variations that destroy the
coherence. Of course, phase variations are also
generated by FM, but they decrease at 6 dB/octave as
the modulating frequency is increased. Therefore, to
obtain sizeable and rapid phase variations, relatively
large FM signals must be used.
A more fundamental reason is that with FM, the
mean value of the modulating function must be ab
solutely zero or errors will accumulate as the time
An Experiment
A simple experiment was performed to illustrate
the effectiveness of phase modulation of the clock.
Time intervals arriving at a rate nearly coherent with
the 5345A clock frequency were measured with and
without modulation. Fig. 3 shows the result.
The author received many useful comments from
conversations with Art Muto, Jim Sorden and Ken
Jochim, all of whom had been doing pioneering work
in time interval averaging for some time. The elegant
implementation of the phase modulation is a result of
the ingenuity and expertise of John Gliever and John
Time Interval Estimation in the
Presence of Quantization Error
For measuring short intervals by time interval averaging, only ± 1 count quan
tization error is present An interesting problem is to determine the statistics
of estimating the true time interval from the reading obtained from averaging
N independent measurements. The usual rule-of-thumb estimate gives this
uncertainty as TQ/\ N where T0 is the clock period. A more formal analysis
shows that the actual rms uncertainty of the estimate is not so simply stated. In
fact, computed actual probability density function of the interval can be computed
given The counter reading after averaging N independent measurements The
results can be summarized as follows
An unknown time interval r is measured N independent times with a clock
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' ' ' i . " ' ; ; . - , ; . '
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1. K. Jochim and R. Schmidhauser, "Timer/Counter/DVM:
A Synergistic Prodigy?" and R. Schmidhauser, "Measur
ing Nanosecond Time Intervals by Averaging", HewlettPackard fournarr April 1970.
' ' . . ' , ' â € ¢
and K function integers and 0- K- N. What is the probability density function of e =
(r r) given P, K, and N? Assuming no a priori knowledge of r. maximum likeli
hood estimation may be used, and the result is as follows:
Case I: the = 1,2,3 ..... N-1. The probability density function P(x). that is, the
probability density of (7 - 7) taking on value x, is
rç N'
for -KVN «x<(1-K/N)T0
The mean of this distribution is [ and the standard deviation is <r. where
N(N 4-ZJ
N T s J
Case is K 0. The probability density function P(x) for this case is
for OsxsTo
David C. Chu
The standard deviation of this function is <r. where
Dave Chu holds BS, MS, and PhD degrees in electrical
engineering, the BS from the University of California at
Berkeley in 1 961 and the MS and PhD from Stanford University
in 1962 and 1973. At HP since 1962 (except for a 1967-69
leave of absence to teach physics and mathematics at
Cuttington College in Liberia), Dave has contributed to
the design of many of HP's top-of-the-line counters. A mem
ber of IEEE and the Optical Society of America, he's an ex
pert on computer holography and inventor of the ROACH,
a single transparency with controlled complex transmittance.
He's authored many papers and patent applications in the
fields of optics, information theory, statistics, and electronics.
Dave was born in Hong Kong. He's married, has two children,
and lives in a "shack" — which he's currently remodeling — on
five acres of land in rural Woodside, California. He's active
in his church and political party, and confesses to being
addicted to Balkan and Israeli folkdancing.
These results show that the rms uncertainty in estimating the time interval by
the average measurement value is a function of K as well as of N. The parame
ter K. always an integer, can be obtained directly from the measurement: K. N
is the large part of the normalized measurement value (fT0). For large N.
which is typical for most time interval average measurements, the rms uncer
tainty of equation 2 is reduced to
[11 KuKiT!
(1 N)(N>
For a given N. the largest T. representing the worst case estimation uncer
tainty is when K N is one-half. The corresponding rms uncertainty is
T¡j2\ or The uncertainty becomes less when KN approaches either 0 or 1
The rule-of-thumb estimate of T0 \ Ñ. therefore, represents twice the worst
case of uncertainty for large N One can further use the knowledge of the frac
tional the of the measurement value to increase confidence in estimating the
time interval.
© Copr. 1949-1998 Hewlett-Packard Co.
Third Input Channel Increases Counter
by Arthur S. Muto
tributes several new and useful measurement
capabilities to the 5345A Electronic Counter. It was
conceived primarily as a solution for many measure
ment problems in the new and expanding field of
high-speed digital measurements, but it is useful for
many other applications as well.
The channel C plug-in, Fig. 1, adds a third, 10-millivolt-rms-sensitivity, dc-to-500-megahertz input
channel to the two channels of the mainframe.
Three of the measurement modes of the channel C
plug-in, namely FREQ c, PERIOD C. and RATIO C/A, dupli
cate functions found in the mainframe. As a conse
quence, frequency or period measurements can be
made on two different signals in a systems or pro
duction environment without having to disconnect
coaxial cables or reset trigger levels. One signal can
be measured with the mainframe's channel A and
the other with the plug-in's channel C. Similarly,
the plug-in can be used to make a multiplexed ratio
measurement with channel A by selecting either the
mainframe's RATIO B/A function or the plug-in's RATIO
C/A function.
A fourth function of the channel C plug-in permits
automatic sum and difference frequency measure
ments to be executed by selecting FREQ C+A, or FREQ
C-A. A direct application of this measurement is in
determining a receiver input frequency by separately
measuring the local oscillator frequency (LO) and the
intermediate frequency (IF) and performing the sum
or difference (LO ± IF) calculation. By connecting the
LO to channel C and the IF to channel A, automatic
LO±IF measurements can be performed.
A nulling technique for making frequency adjust
ments is another application of FREQ C-A. With a
standard frequency source connected to channel A,
the frequency of the channel C signal can be adjusted
until the counter displays zero. The frequency of the
channel C input signal is then equal to the frequency
of the standard source.
Events Mode
Still another measurement mode of the 5353A
Channel C Plug-In is called EVENTS C.A to B. In this
mode, channel C events — pulses as narrow as one
nanosecond in width — are totalized during a precise
time interval defined by a start-pulse edge in channel
A and a stop-pulse edge in channel B. The polarity
of the pulse edge is determined by the user's selection
of positive and/or negative trigger slopes (Fig. 2).
In the digital communications field, this EVENTS
measurement is especially useful in measuring
data within a specific burst inside a frame of TDMA
(Time-Division-Multiple-Access) formatted data.
By adding a pulse generator having variable width
and delay capabilities and using the COMMON A in
put feature of the mainframe, any 10 nanosecond or
wider portion of the burst can be isolated for bit pat
tern characterization. This ability to isolate bits is also
useful in troubleshooting digital systems, whenever
searching for specific or even random bits may be
To achieve greater resolution on repetitive sig
nals, automatic averaging of the number of channel C
events over several time intervals can be ac
complished by selecting the AVERAGE EVENTS C, A to B
function. Since the events measurement is basically
Fig. 1. Model 5353A Channel C Plug-in adds a third 10-mVsensitivity, 500-MHz input channel to the two 5345A main
frame channels.
© Copr. 1949-1998 Hewlett-Packard Co.
a given measurement. The plug-in has access to the
full measurement capabilities of the mainframe;
every mainframe measurement can be controlled by
the plug-in through the control interface.
Upon completion of any measurement, the arith
metic processor in the mainframe algebraically com
bines the contents stored in the BCD counters to
create a meaningful result. Through the processor
interface shown in Fig. 3, the plug-in can command
the processor to scale the result before displaying it.
The general form of equation that the processor is
capable of executing is
C l W
Channel C
Channel A
Channel B
Channel B
Channel A/
- Slope
Fig. 2. EVENTS C, A to B mode is especially useful in digital
communications. In this example the counter display would
read "4."
± (K ± XN)
a normalized time-interval measurement, events
averaging is analogous to time-interval averaging,
excluding the time base jittering as described in
David Chu's article in this issue.
All seven 53 53 A functions are programmable via
the optional HP interface bus system. The lamps on
the front-panel indicate when the plug-in is listening
for bus programming instructions (LISTENING lamp)
and when the plug-in is under remote control
(REMOTE lamp). The front-panel LOCAL reset push
button s witch returns the function selection from bus
control to the front-panel FUNCTION switch. Bus pro
gramming can disable the LOCAL reset switch when
where X is the measurement result and K and N are
values supplied by the plug-in when scaling is de
sired. Arithmetic expressions of the general form
± (K ± PI/N)
in which numerical values for K, N and PI are sup
plied by the plug-in, can also be evaluated by the
processor. The algorithm that adds K and the algorithm that multiplies by N will he bypasspH if
values for K or N are not supplied.
Solutions to more complex arithmetic equations
can be calculated by multiple executions of equations
1 and 2 before displaying the final result. Each re
sult calculated by equation 1 or 2 is returned to the
plug-in and can subsequently be returned to the
mainframe as a plug-in variable in later processing
cycles. As a consequence, extremely complex arith
metic processing of measurement results is available
to the plug-in.
Plug-In Interface
The block diagram of the 5353 A Channel C Plug-In
is shown in Fig. 3. Also included in the same figure is
a simplified block diagram of the 5345A Counter
showing the interface between the mainframe instru
ment and all plug-ins, not only the 5353A.
The channel C plug-in uses two of the four RF lines
available to plug-ins. Three of these four lines, the PI
Channel A, PI Channel B, and PI Channel C lines,
represent full-bandwidth, de-coupled, E2L channels
paralleling the two mainframe input channels, A and
B, and the internal 500-MHz clock, respectively. The
fourth RF line is a buffered 10-MHz frequency stand
ard output from the mainframe.
The three PI channels, coupled with the corres
ponding three channels in the mainframe, afford
enormous measurement flexibility to the mainframe/
plug-in combination. Through the control interface
shown in Fig. 3, the mainframe recognizes which
mainframe and/or plug-in RF channels to enable for
Plug-In Operation
The digital control logic of the channel C plug-in
relays to the mainframe the function selected by the
function switch or the optional HP interface bus. In
the case of the FREQ C-A function, for instance, the
mainframe is commanded to perform a frequency
measurement of PI Channel A, which is the plug-in
channel C input signal directed to PI Channel A by
the high-speed digital logic, after digitization by
the input amplifier and trigger circuit. The measure
ment is a standard frequency measurement using
the mainframe 500-MHz clock, and the measurement
result is returned directly to the FREQ C±A digital
logic for storage in the plug-in without additional
processing. As soon as this transmission is completed
a second machine cycle is initiated by the digital
control logic, commanding the mainframe to perform
a frequency measurement of the channel A analog
input signal. During the processing of this second
frequency measurement, the frequency of the channel
In our last issue (May 1974) the result for I0 in Fig. 4 on page 10 should be 6.278 mA.
Also, the second 'user action in the second column on page 13 should be "CHS
following EEX," the third "user action should be "Multiple CHS," and the third
desired result should be Complement mantissa sign, or exponent sign it EEX
has been pressed '
© Copr. 1949-1998 Hewlett-Packard Co.
5345A Electronic Counter
5353A Channel C Plug-in
hannel C
and Trigger
011 HP
Fig. 3. Block diagram of the
5353A plug-in and the 5345A
plug-in interface. The plug-in
controls mainframe measure
ments and processing of meas
ured data.
C input signal is returned to the mainframe as K in
equation 1. The result is then displayed as the posi
tive or negative difference in frequency between the
channel A and channel C input signals.
The other 5353A functions follow similar steps but
require only a single machine cycle to generate a
HP Model 5353A Channel C Plug-In
MODES OF OPERATION: Frequency C, Period C. Ratio C/A, Events C between
A and B, Average Events C between A and B, C-A. C+A.
DC COUPLED: 0 to 500 MHz
AC COUPLED: 10 MHz to 500 MHz
IMPEDANCE: 50!! (nominal), or 1 Mil shunted by less than 30 pF
SENSITIVITY: Variable to 10 mV rms sine wave and 30 mV peak-to-peak pulse.
Attenuator settings are X1 and X20.
TRIGGER LEVEL: Continuously adjustable over dynamic range.
PRESET: Centers trigger level about dc at 25°C.
DRIFT: r 10 mV dc max. 0"C to 55°C.
OUTPUT: Rear panel BNC connectors bring out TRIGGER LEVEL for convenient
DVM monitoring Accuracy is =15 mV.
MAXIMUM INPUT: Same as Channel A and B. separate input.
OPTION 011: Digital Input. Full compatibility with HP Interface (ASCII) Bus.
Provides for digital control over all functions excluding amplifier.
PRICE IN U.S.A.: S850.
5301 Stevens Creek Boulevard
Santa Clara. California 95050 U.S.A.
My associates in the design of the 5353A Channel
C Plug-In were Bryce Jeppsen (T2L digital logic and
HP interface bus option), Keith Leslie (mechanical
design) , and Steve Upshinsky (high-frequency ampli
fier and trigger). The early design efforts of Ron Felsenstein and Hans Trosch are gratefully acknowledged. Z
Arthur S. Muto
Art Muto designed the high-speed gating circuits for the
5345A Counter and was project leader for the 5353A Channel
C Plug-In. A graduate of the University of California at
Berkeley, he received his BSEE degree in 1 967 and his MSEE
in 1969. He joined HP part-time in 1969 and full-time in 1970.
Art is a native Californian. born in Sacramento and raised in
San Francisco. He and his wife and daughter now live in Sara
toga, California. His favorite leisure time pursuits are listening
to classical and rock music, playing table tennis, and camping
and hiking with his family.
© Copr. 1949-1998 Hewlett-Packard Co.
A Completely Automatic 4-GHz
Heterodyne Frequency Converter
by Ali Bologlu
A FREQUENCY CONVERTER translates an un
known high-frequency signal downward in fre
quency by mixing it with a precisely known signal of
slightly lower frequency. This heterodyne process
yields a difference frequency within the basic range
of the counter. Then the counter reading and the
known frequency of the mixing signal are added to
give the unknown.
A new plug-in for the HP 5345A Electronic Counter
completely automates the heterodyne frequency con
verter technique for both CW and pulsed RF input
signals. The 5354A Automatic Converter (see Fig.
1) extends the basic frequency measurement range
of the counter to 4 GHz.
A simplified block diagram is shown in Fig. 2.
The converter sends signals in the 5-to-525-MHz
range directly to the mainframe. Input signals be
tween 510 MHz and 4 GHz are translated by nine
crystal-controlled frequencies to an intermediate
frequency between 10 and 525 MHz, and are then sent
to the mainframe for measurement. The frequency
measured by the mainframe (IF) is then added to the
crystal-controlled frequency used in the translation
process (LO), and the result is displayed. The table
below lists the LO and intermediate frequencies for
various input frequencies.
Input Frequency LO Frequency IF to Counter
( M H z ) ( M H z ) ( M H z )
Operating Modes
The converter has both automatic and manual
modes of operation, selectable by a front-panel
switch. In the automatic mode, the converter auto
matically steps through nine frequency bands repe-
â € ¢
Fig. 1. Model 5354 A Automatic Converter extends the fre
quency measurement range of the 5354A Electronic Counter
to 4 GHz. Operation can be manual or completely automatic,
even for pulsed RF signals.
titively until a signal within the converter range is
detected. The time required for acquisition is 160
microseconds worst case. In the manual mode, the
converter remains in one of the frequency range
bands until the BAND SELECT or RESET switch is pressed.
Each time the BAND SELECT switch is pressed, the
next higher frequency band is selected. Pushing
RESET returns the converter to the zero band.
In the manual mode, if there is no signal or the sig
nal level is inadequate, the mainframe will display
the LO frequency. In both modes, manual and auto
matic, the CONV signal light comes on when the coun
ter is making a correct measurement. One can oper
ate the converter in the automatic mode until a signal
is acquired and then switch to manual. In this case,
the manual mode retains the band information and
will be ready to make a measurement in that particu
lar band within 20 microseconds.
In the automatic mode the converter assumes that
the first signal of sufficient amplitude to be detected
is the correct one. When many frequencies in differ-
© Copr. 1949-1998 Hewlett-Packard Co.
Swit chable
10 MHz from
*• Mainframe
PI Channel A
to Mainframe
15-525 MHz
Fig. 2. The 5354A automatic con
verter heterodynes high-fre
quency input signals with one of
nine crystal-controlled frequen
cies to derive an intermediate
frequency within the range of
the 5345 A.
than 250 nanoseconds can be acquired automatically
in one second.
For narrower pulse widths the manual mode may
be used — the minimum RF pulse width is 50 ns. Con
verter sensitivity is greater in the manual mode; sig
nals of -20 dBm (typical) may be counted. Further
more, the mainframe gate and plug-in signal detec
tors are pre-armed in the manual mode, so actual
measurement times may be as short as 2 ns. The
5354A's pulsed RF capability together with the main
frame's averaging capability allows meaningful highresolution frequency measurements on narrow
pulses. Displaying the auxiliary output of the plugin on an oscilloscope simultaneously with an exter
nal gating signal shows exactly which portion of
the RF pulse is being measured. By sliding the gate
pulse along the down-converted RF signal, frequency
profile measurements may be made.
ent bands are present simultaneously the user may
switch to the appropriate bands manually to count
the higher frequencies.
The rapid acquisition in the automatic mode and
the flexible switching capability in the manual mode
are possible because of the switching filter assembly
shown in Fig. 3.
CW Performance
The 5354A Automatic Converter is very tolerant of
signals that have various kinds of modulation. It can
tolerate large frequency deviations — up to ±250
MHz — at band centers. At band edges it is limited to
deviations of about ±10 MHz. These tolerances are
sufficient for communications applications, in which
carrier frequencies need to be measured under
operating conditions.
Specified sensitivity in the CW mode is - 10 dBm.
Pulsed RF Performance
Special Features
The most unusual feature of the converter is its
ability to measure pulsed frequencies automatically.
Pulsed signals whose modulation width is greater
Besides the RF circuits necessary to process micro
wave signals, the converter also contains digital pro
cessing circuits. These have two main functions. One
is control of the interface to the mainframe and the
other is the algorithms for automatic operation of
the converter. For rapid troubleshooting, a self-test
feature has been incorporated. This is enabled by
putting an internal switch in the test position with
the mainframe in self-check mode. With the plug-in
in manual mode, the mainframe will display the LO
frequency plus 100 MHz. When the 3.5 GHz LO is
selected, the CONV signal light comes on. This rou
tine checks the plug-in/mainframe interface. When
the plug-in is placed in the automatic mode, the
mainframe will immediately display 3.6 GHz with
the CONV light on, thereby checking the automatic
acquisition routine.
Options and Rear-Panel Controls
A buffered version of the down-converted IF signal
is available at the rear-panel AUX OUT connector for
such applications as viewing the down-converted
Fig. 3. Switching filter assembly permits rapid acquisition
of input signals. 760 us is worst case in automatic mode.
© Copr. 1949-1998 Hewlett-Packard Co.
trum-generator modules are hybrids, all contained
in one assembly (Fig. 3). It is this unique structure
that makes the extremely fast acquisition of input
signals possible.
microwave signal on an oscilloscope. Another rearpanel control, the NORM/-L.S. slide switch, controls the
mainframe processing of the converter output sig
nals. In the normal position, the mainframe is instruc
ted to add the frequency of the converter local-oscil
lator signal to the frequency of the converter IF signal
and display the result. In the manual mode the -L.S.
position may be used, causing the IF to be subtracted
from the LO signal frequency. One application for
this control is the measurement of pulsed signals
having narrow pulse widths, when the resulting con
verter IF signal is a low frequency. By switching to the
lower sideband (-L.S.), one can increase the number
of zero crossings per unit time, and therefore the ac
curacy of the measurement.
When the mainframe is under remote control of the
HP interface bus, option Oil for the converter gives
the plug-in its own address and allows all of its func
tions to be remotely programmed.
An optional LO±IF ON-OFF switch controls the
addition or subtraction capability of the converter.
With this option, input signals of microwave re
ceivers can be monitored. The receiver IF signal is
applied to the mainframe channel A input and the receivex local oscillator to the converter input. The
LO±IF ON/OFF switch is set to ON. The +IF/-IF
switch is set to +IF if the receiver local-oscillator freqency is known to be below the received signal fre
quency, or to -IF if the local-oscillator frequency is
above the received frequency.
Special credit is due Hans Trosch and Al Barber for
their design efforts. Larry Jackson initiated the prod
uct design, which was completed by Roland Krevitt.
Thanks are also due Dick Harris for production sup
port and Holly Cole for market introduction. .7
HP Model 5354A Automatic Frequency Converter
RANGE: 15 MHz to 4 GHz
SENSITIVITY: -10 dBm (70 mV rms) to »20 dBm (2.2V rms)
INPUT SIGNAL CAPABILITY: CW signals. Pulsed microwave signals. Signals with
very high FM content.
RF PULSE WIDTH: Determined by counter GATE TIME setting.
FM SENSITIVITY: Maximum deviation at band edges ±10 MHz. Maximum devia
tion at band center: ±250 MHz above 1 GHz and below 500 MHz. ±125 MHz
between 500 MHz and 1 GHz.
OPERATING MODES: Automatic and manual.
AUTOMATIC: Measures lowest frequency signal of sufficient amplitude to
trigger counter.
MANUAL: Measures signal within selected band. Signals of sufficient amplitude
between 15 MHz and 525 MHz will also be counted
AUTOMATIC MODE: cont. wave <160 M sec, pulsed RF <1 sec.
MANUAL MODE: When proper band has been selected: cont. wave <5
/¿sec, pulsed RF <20 nsec.
MIXING FREQUENCIES: 0, 0.5, 0 75, 1.0. 15, 2.0, 25. 3.0, 35 GHz
DAMAGE LEVEL: 25 dSm (4V rms). 10 volts peak-lo-peak.
OPTION 011: Digital Interface Bus for remote control and L.O.±I.F.
PRICE IN U.S.A.: S1950
530I Stevens Creek Boulevard
Santa Clara. California 95050 U.S.A.
LO Details
Crystal-controlled mixing frequencies are generat
ed from a 500-MHz signal derived from the 10-MHz
time base. The 500-MHz signal drives a spectrumgenerator diode whose output goes to a PIN singlepole six-throw switch. Each output of this switch is
connected to a band-pass filter whose center fre
quency is tuned to a harmonic of 500 MHz which, in
turn, is connected to a single-pole six-throw switch.
The output of this switch is connected to the input
mixer module. The PIN switches, filters, and spec
AM Bologlu
Ali Bologlu received the BSEE degree in 1962 and the MSEE
degree in 1963, both from Michigan State University, and
the degree of Electrical Engineer in 1965 from Stanford Uni
versity. At HP since 1963, Ali has contributed to the design of
the 5100A, 5102A, 5103A, 5105A, and 866CVVB Frequency
Synthesizers, and was project leader for the 5354A Con
verter. He now has responsibility for microwave counter
development He's a member of IEEE. Born in Istanbul,
Turkey, Ali now lives in Mountain View, California, where
he devotes much of his free time to the local youth athletics
© Copr. 1949-1998 Hewlett-Packard Co.
Interface Bus Expands Instrument
by Bryce E. Jeppsen and Steven E. Schultz
THE PROLIFERATION OF computer-controlled
instrument systems over the last decade testifies
to the importance of marrying data-input and compu
tational capabilities. However, most systems of this
type have gone into production environments as a
means of reducing test times and simplifying produc
tion decision-making. The cost, complexity of system
integration, and elaborateness of operating pro
cedures has restricted the use of such systems in the
development laboratory. Nevertheless, the engineer
would often prefer to work with computed quanti
ties such as pulse jitter or linearity, rather than raw
time-interval or frequency data.
A pair of recent developments now offer the en
gineer the tools he needs to look at computed quan
tities based on fundamental measurements. One is
the programmable calculator, which brings comput
ing power to the laboratory bench. The other is the
new HP interface bus system, which greatly simpli
fies the assembly of instrument/calculator systems.1
As the applications and ramifications of this "smart"
interface technique become recognized, we can ex
pect to see a profound change in the way bench in
struments are used.
The new 5345A Counter is compatible with the bus
interface. Systems including the 5345A, the 9820A
Calculator, and a new line of ASCII-Programmable
Modules are described, complete with software, in a
new series of Application Notes, AN 174. Because of
space limitations, only two of these systems are dis
cussed here.
ations, and the like may never be investigated.
Using the 5345A Electronic Counter with the new
HP interface bus, this measurement may be made eas
ily with the help of a calculator such as the HP 9820A.
The equipment is connected as shown in Fig. 1.
To program the counter, it is only necessary to enter
the number of measurements desired, the width of
each bin, and the number of bins. Application Note
174-5 tells how this is done. Once these parameters
are defined, measurements may begin.
As data is received, the calculator determines
which bin each measurement falls in. When the
specified number of measurements has been made,
the calculator instructs the plotter to draw the histo
This system has been used to evaluate the overall
quality of digital communications systems by meas
uring the time jitter between frames, and has proved
valuable in PCM communications applications that
require an analysis of the statistical nature of the time
between error pulses from an error-bit-rate detector.
Free Running
1C Timer
5345A - 9820A Histogram
Graph Center - 9.915600E-05
Bar Width - 2.000000E-09
No. of Measurements - 200
Highest Bar - 46
Mean Line - 9.915642000E-05
Pulse-Width Jitter Histograms
A typical counter/calculator system application is
the measurement and analysis of pulse-width jitter,
such as that of any pulsed or clock signal. A histo
gram or probability distribution provides insight into
the statistical nature of the jitter. This insight is use
ful in analyzing the noise characteristics of the signal
source. However, the effort and time required to col
lect and plot this data manually is almost prohibi
tive, so the measurement is rarely performed more
than once, and that one measurement is assumed to
be typical. Effects of temperature, power-supply vari
Fig. 1 . 5345A Counter is compatible with the HP ASCII-coded
interface bus, making it easy to assemble systems like this
one, which plots a histogram showing the distribution of the
pulse-width jitter of an integrated-circuit timer chip.
© Copr. 1949-1998 Hewlett-Packard Co.
ASCII Programmable Modules
The ASCII Programmable Modules make the
5345A Counter more versatile as a systems compo
nent. These modules can be used as "systems glue",
performing necessary systems functions not per
formed by the major instruments. However, they
are also designed to make the basic measurement
instruments like the 5345A Counter more useful in
bench-top environments. The modules may be used
with or without a calculator in the system. Modules
now available are:
59301A ASCII-to-Parallel Converter
59303A Digital-to-Analog Converter
59304A Numeric Display
59306A Relay Actuator
59307A VHP Switch
59308A Timing Generator
59309A ASCII Digital Clock
Model 59301A ASCII-to-Parallel Converter inter
faces the counter to standard digital printers to pro
duce hard-copy records of measured data. Model
59303A Digital-to-Analog Converter can drive a stan
dard strip-chart recorder to provide trend informa
tion. Model 59308A Timing Generator is useful in
the monitoring of crystal warmup characteristics and
similar applications.
Model 59304A Numeric Display can be used to dis
play intermediate results without consuming valu
able calculator processing time. The 59306A Relay
Actuator can control high-frequency switches and at
tenuators, or simply switch bias voltages. Model
59307A VHP Switch is useful for switching highfrequency signals.
VCO Characteristics
An example of the usefulness of the 5345A/9820A
combination in stimulus-response testing is a system
to measure the characteristics of voltage-to-frequency
conversion circuits. Voltage-controlled oscillators
(VCO's) are present throughout electronic designs in
phase-locked loops, modulators, and so on. It is often
desirable to characterize the tracking characteristics
of two similar oscillators. This can be done using the
5345A Counter, the 9820A Calculator, and three
members of the ASCII-Programmable Module family:
the 59303A Digital-to-Analog Converter, the 59304A
Numeric Display, and the 59307A VHP Switch. The
system is described in Application Note 174-4 and
shown in Fig. 2.
The D-to-A converter is told by the calculator to
apply voltages to the VCO's while the VHP switch
alternately applies their frequency outputs to the
5345A Counter. The numeric displays give a real
time presentation of the applied voltage, the fre
quency outputs, and the frequency difference. The
calculator generates a linearity plot for each VCO
and a plot showing how well the two VCO's track.
The key to success in this experiment is the accu
racy of the voltage applied to the VCO's from the digit
al-to-analog converter. This D-to-A converter has
10 mV resolution and accuracy, a dynamic range of
+ 10 volts to -10 volts, and an operating temperature
range of 0 to 55°. At room temperature the 59303A is
typically accurate within 3mV. This D-to-A converter
is also fully programmable. Various digit formats
and output modes can be chosen, including a mode
that adds in an offset for strip-chart recordings.
The authors would like to acknowledge the support
given by John Dukes and Jim Sorden. The pillar of
strength in the 5345A Interface and ASCII Program
mable Module program has been Charlie Trimble,
whose counsel and guidance has greatly contributed
to the program's success. Technical support from
Dave Ricci and product design by Eric Havstad and
Carl Spalding is also appreciated. References
1. G.E. Nelson and D.W. Ricci, "A Practical Inter-
HP Interface Bus
Fig. 2. The ASCII-programmable
Modules complement the 5345A
in systems applications. Three
types are used in this system,
which characterizes the tracking
characteristics of two similar
voltage-tuned oscillators.
© Copr. 1949-1998 Hewlett-Packard Co.
face System for Electronic Instruments", HewlettPackard Journal, October 1972.
2. D.C. Loughry, "A Common Digital Interface for
Programmable Instruments: The Evolution of a Sys
tem", Hewlett-Packard Journal, October 1972.
ASCII Programmable Modules
Converts ASCII coded information to parallel BCD or lirie
per function information. When used with two 562-16C
cables allows direct operation of a 5345A/B Option 010 or
011 and a 5050B Digital Printer. $450.
Converts any three consecutive ASCII digits to a 0.1%
accurate dc voltage within 25 /¿sec. Very useful ior
stability measurements with strip chart recorders and XY
plotters or programming VCO's and power supplies. S850.
Steven E. Schultz (right)
A 1970 BSEE graduate of the University of California at Ber
keley, Steve Schultz is now working towards his master's
degree at Stanford University. At HP since 1971, he helped
design the digital I/O and mainframe boards for the 5345A
Counter, and was project leader for five of the ASCII-Pro
grammable Modules. Steve was born in Oakland, California.
He's married, lives in Menlo Park, California, and says that
most of his spare time is occupied by his master's studies
and other obligations. Whenever he can, though, he indulges
his many other interests which include radio controlled
boats, British cars, sailing, electronics, and riding dirt
Provides auxiliary display when used directly with 5345A/B
Option 010 or 011 or display intermediate results from a
9820A Calculator without consuming calculator display time.
Provides six form "C" relays unrjer front panel or remote
control. Can be used for programming 8761A/B 18 GHz
control switches, 33000 series 18 GHz step attenuators, or
for simply switching bias voltages. $600.
Contains two dc to 500 MHz 50Ã! bi-directional coaxial
switches. Each switch has one input and four outputs and
can be operated either from the front panel or remotely
Useful for scanning time interval signals or switching VHF
signals. $700.
Bryce E. Jeppsen (left)
Bryce Jeppsen graduated from Brigham Young University
in 1 970 with BES and ME degrees, then joined HP and began
working on the 5340A Microwave Counter only to be drafted
into the U.S. Army in October of that year. While in the service
he organized and taught courses in basic semiconductor
electronics. In 1972, his service completed, Bryce returned
to HP and electronic counters. He designed the 10590A
Plug-In Adaptor and the digital circuitry and ASCII control
for the 5353A Plug-In, and has written much of the series
of application notes describing 5345A systems based on the
HP interface bus. Bryce and his wife have three small children
and live in Santa Clara, California, where they're busy work
ing on their new home and participating in church activities.
Provides time intervals and delays to environments where
an accurate time base is necessary Can be used in plots of
crystal warmup characteristics to provide an accurate time I >
b e t w e e n
r e a d i n g s .
U = ^
Provides calendar and time of day output on the HP inter
face bus. Can be used to provide a time axis on a plot, or
used to control the start and stop time of measurements
with a calculator This clock can be set remotely
Three, six and twelve feet respectively Double ended
ASCII connector on each end. $55. $60, and S70
5301 Stevens Creek Boulevard
Santa Clara. California 95050 U.S. A
Bulk Rate
U.S. Postage
Hewlett-Packard Company, 1501 Page Mill
Road, Palo Alto, California 94304
g l g f l
JUNE 1974 Volume 25 • Numt
Technical Information from the Laboratories of
Hewlett-Packard Company
Hewlett-Packard S.A.. CH-1217 Meynn 2
Geneva. Switzerland
Yokogawa-Hewlett-Packard Ltd , Shibuya-Ku
Tokyo 151 Japan
Editorial Director • Howard L. Roberts
Managing Editor • Richard P. Dolan
Contributing Editors • Ross H. Snyder.
Laurence D. Shergalis
Art Director, Photographer • Arvid A. Danielson
An Assistant • Sue M. Reinheimer
Administrative Services • Anne S. LoPresti
European Production Manager • Kurt Hungerbühler
/*^ I
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© Copr. 1949-1998 Hewlett-Packard Co.
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