AN118 - High Voltage, Low Noise, DC/DC Converters

AN118 - High Voltage, Low Noise, DC/DC Converters
Application Note 118
March 2008
High Voltage, Low Noise, DC/DC Converters
A Kilovolt with 100 Microvolts of Noise
Jim Williams
This publication describes a variety of circuits featuring
outputs from 200V to 1000V with output noise below 100µV
measured in a 100MHz bandwidth. Special techniques
enable this performance, most notably power stages
optimized to minimize high frequency harmonic content.
Although sophisticated, all examples presented utilize
standard, commercially available magnetics—no custom
components are required. This provision is intended to
assist the user in quickly arriving at a produceable design.
Circuits and their descriptions are presented beginning
with the next ink.
BEFORE PROCEEDING ANY FURTHER, THE READER
IS WARNED THAT CAUTION MUST BE USED IN THE
CONSTRUCTION, TESTING AND USE OF THE TEXT’S
CIRCUITS. HIGH VOLTAGE, LETHAL POTENTIALS ARE
PRESENT IN THESE CIRCUITS. EXTREME CAUTION
MUST BE USED IN WORKING WITH, AND MAKING
CONNECTIONS TO, THESE CIRCUITS. REPEAT: THESE
CIRCUITS CONTAIN DANGEROUS, HIGH VOLTAGE
POTENTIALS. USE CAUTION.
Resonant Royer Based Converters
The resonant Royer topology is well suited to low noise
operation due to its sinosoidal power delivery1. Additionally, the resonant Royer is particularly attractive because
Figure 1’s resonant Royer topology achieves 100µVP-P
noise at 250V output by minimizing high frequency
harmonic in the power drive stage. The self oscillating
resonant Royer circuitry is composed of Q2, Q3, C1, T1
and L1. Current flow through L1 causes the T1, Q2, Q3,
C1 circuitry to oscillate in resonant fashion, supplying sine
Note 1. This publication sacrifices academic completeness for focus
on the title subject. As such, operating details of the various switching
regulator architectures utilized are not covered. Readers desiring
background tutorial are directed to the References. Resonant Royer theory
appears in Reference 1.
DANGER! Lethal Potentials Present — See Text
250VOUT
1µF, 400V
10k
1µF, 400V
1M*
D1-D4
0.001µF
400V
10
1k
6
5V
T1
5
5V
4
820Ω
2
1
0.22µF
C1
5V
x2
3
D5
Q2
Q3
L1
250µH
= ZDT1048 DUAL
IRLRO24
Q1
L1 = CTX250-4, COILTRONICS
T1 = 210605R, COILTRONICS
1µF = WIMA MKS-4
0.22µF = WIMA MKP-2
D1-D4 = TOSHIBA DUAL DIODE 1SS306.
CONNECT EACH UNIT IN SERIES.
D5-D6 = 2N4393
* = 1% METAL FILM RESISTOR
10k
D6
0.1µF
430k
LT1635
VREF = 0.2V
+
Photomultipliers (PMT), avalanche photodiodes (APD),
ultrasonic transducers, capacitance microphones, radiation detectors and similar devices require high voltage,
low current bias. Additionally, the high voltage must be
pristinely free of noise; well under a millivolt is a common
requirement with a few hundred microvolts sometimes
necessary. Normally, switching regulator configurations
cannot achieve this performance level without employing
special techniques. One aid to achieving low noise is that
load currents rarely exceed 5mA. This freedom permits
output filtering methods that are usually impractical.
transformers originally intended for LCD display backlight
service are readily available. These transformers are multiply sourced, well proven and competitively priced.
–
Introduction
A1
1k
OUTPUT
ADJUST
499Ω*
430k
AN118 F01
Figure 1. Current Fed Resonant Royer Converter Produces
High Voltage Output. A1 Biases Q1 Current Sink, Enforcing
Output Voltage Stabilizing Feedback Loop. A1’s 0.001µF–1kΩ
Network Phase Leads Output Filter, Optimizing Transient
Response. D5-D6, Low Leakage Clamps, Protect A1
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
an118fb
AN118-1
Application Note 118
wave drive to T1’s primary with resultant sine-like high
voltage appearing across the secondary.
The circuit’s low harmonic content combined with the
RC output filter produces a transcendently clean output.
Output noise (Figure 3) is just discernible in the monitoring
instrumentation’s 100µV noise floor3.
T1’s rectified and filtered output is fed back to amplifierreference A1 which biases the Q1 current sink, completing
a control loop around the Royer converter. L1 ensures
that Q1 maintains constant current at high frequency.
Milliampere level output current allows the 10k resistor
in the output filter. This greatly aids filter performance
with minimal power loss.2 The RC path to A1’s negative
input combines with the 0.1µF capacitor to compensate
A1’s loop. D5 and D6, low leakage clamps, protect A1
during start-up and transient events. Although Figure
2’s collector waveforms are distorted, no high frequency
content is present.
Figure 4’s variant of Figure 1 maintains 100µV output noise
while extending input supply range to 32V. Q1 may require
heat sinking at high input supply voltage. Converter and
loop operation is as before although compensation components are re-established to accommodate the LT1431
control element.
Note 2. As previously mentioned, low current requirements permit certain
freedoms in the output filter and feedback network. See Appendix A for
examples and discussion.
Note 3. Measurement technique and instrumentation choice for faithful
low level noise measurement requires diligence. See Appendices B
through E for practical considerations.
DANGER! Lethal Potentials Present — See Text
250VOUT
1µF, 400V
A = 5V/DIV
10k 1µF, 400V
B = 5V/DIV
1M*
D1-D4
0.001µF
400V
20µs/DIV
10
AN118 F02
6
T1
Figure 2. Resonant Royer Collector Waveforms Are Distorted
Sinosoids; No High Frequency Content is Present
1k
5
5V
820Ω
4
1
2
0.22µF
5V
x2
3
D5
D6
5V
5V
L1
250µH
IRLRO24
Q1
= ZDT1048 DUAL
100µV/DIV
AC COUPLED
1k
10k
OUT
0.2µF
L1 = CTX250-4, COILTRONICS
T1 = 210605R, COILTRONICS
1µF = WIMA MKS-4
0.22µF = WIMA MKP-2
D1-D4 = TOSHIBA DUAL DIODE 1SS306.
CONNECT EACH UNIT IN SERIES.
D5-D6 = 1N4148
* = 1% METAL FILM RESISTOR
10µs/DIV
+V
LT1431
COMP
FB
GND GND
5k
OUTPUT
ADJUST
7.5k*
AN118 F04
AN118 F03
Figure 3. Figure 1’s Output Noise is Just Discernable in
Monitoring Instrumentation’s 100µV Noise Floor
Figure 4. LT1431 Regulator Based Variant of Figure 1 Maintains
100µV Output Noise While Extending Input Supply Range to 32V.
Q1 May Require Heat Sinking at High Input Supply Voltages
an118fb
AN118-2
Application Note 118
DANGER! Lethal Potentials Present — See Text
250VOUT
1µF, 400V
= ZDT1048 DUAL
The previous resonant Royer examples utilize linear control of converter current to furnish harmonic free drive.
The trade off is decreased efficiency, particularly as input
voltage scales. Improved efficiency is possible by employing switched mode current drive to the Royer converter.
Unfortunately, such switched drive usually introduces
noise. As will be shown, this undesirable consequence
can be countered.
10k
L1 = CTX250-4, COILTRONICS
T1 = 210605R, COILTRONICS
1µF = WIMA MKS-4
0.22µF = WIMA MKP-2
D1-D4 = TOSHIBA DUAL DIODE
1SS306. CONNECT EACH
UNIT IN SERIES.
D5-D6 = 1N4148
* = 1% METAL FILM RESISTOR
1µF, 400V
D1-D4
10
1M
0.22µF
400V
10k
6
T1
5
5V
4
1
2
0.22µF
5V
x2
820Ω
3
L1
250µH
1N5817
D6
D5
5V
5V
SWITCH
VIN
VOUT
SD
GND
LTC3401
1M
FB
2.5k
OUTPUT
ADJUST
MODE
RT
VC
20k
Switched Current Source Based Resonant Royer
Converters
3.65k*
1µF
Figure 5 replaces the linearly operated current sink with a
switching regulator. The Royer converter and its loop are as
before; Figure 6’s transistor collector waveshape (trace A)
is similar to the other circuits. The high speed, switch
mode current sink drive (trace B) efficiently feeds L1. This
switched operation improves efficiency but degrades output
noise. Figure 7 shows switching regulator harmonic clearly
responsible for 3mV peak to peak output noise – about 30
times greater than the linearly operated circuits.
Careful examination of Figure 7 reveals almost no Royer
based residue. The noise is dominated by switching
regulator artifacts. Eliminating this switching regulator
originated noise while maintaining efficiency requires
special circuitry but is readily achievable.
AN118 F05
Figure 5. Replacing Linearly Operated Current Sink with
Switching Regulator Minimizes Heating Although Output Noise
Increases
A = 5V/DIV
A = 1mV/DIV
AC COUPLED
B = 5V/DIV
A = 20µs/DIV
B = 1µs/DIV
(TRIGGERS ASYNCRONOUS)
AN118 F06
Figure 6. Resonant Royer Collector Waveshape (Trace A) is
Similar to Previous Circuits. High Speed, Switched Mode Current
Sink Drive (Trace B) Efficiency Feeds L1
500ns/DIV
AN118 F07
Figure 7. Switching Regulator Harmonic
Results in 3mVP-P Output Noise
an118fb
AN118-3
Application Note 118
Low Noise Switching Regulator Driven Resonant
Royer Converters
Figure 8 examplifies the aforementioned “special circuitry”.
The resonant Royer converter and its loop are reminiscent of previous circuits. The fundamental difference is
the LT1534 switching regulator which utilizes controlled
transition times to retard high frequency harmonic while
maintaining efficiency. This approach blends switching and
linear current sink benefits4. Voltage and current transition rate, set by RV and RI respectively, is a compromise
between efficiency and noise reduction.
Figure 9’s Royer collector waveshape (trace A) is nearly
identical to the one produced by Figure 5’s circuit. Trace
B, depicting LT1534 controlled transition times, markedly
departs from its Figure 5 counterpart. These controlled
transition times dramatically reduce output noise (Figure
10) to 150µVP-P — a 20x improvement vs Figure 7’s
LTC3401 based results.
Note 4. As stated, this forum must suffer brevity to maintain focus. The
LT1534’s controlled transition time operation mandates further study. See
Reference 3.
DANGER! Lethal Potentials Present — See Text
250VOUT
1µF, 400V
= ZDT1048 DUAL
L2 = COILCRAFT B08T
L1 = CTX250-4, COILTRONICS
T1 = 210605R, COILTRONICS
1µF = WIMA MKS-4
0.22µF = WIMA MKP-2
D1-D4 = TOSHIBA DUAL DIODE
1SS306. CONNECT EACH
UNIT IN SERIES.
D5-D6 = 1N4148
* = 1% METAL FILM RESISTOR
1k
B = 5V/DIV
1µF, 400V
1M*
0.002µF
400V
D1-D4
10
A = 20µs/DIV
B = 10µs/DIV
(TRIGGERS ASYNCRONOUS)
100k
6
5V
2
1
0.22µF
5V
x2
4
820Ω
5V
3
1k
OUTPUT
ADJUST
L1
1N5817
AN118 F09
Figure 9. Resonant Royer Collector Waveshape
(Trace A) is Identical to Figure 5’s LT3401 Circuit;
LT1534 Current Sink’s Controlled Transition Times
(Trace B) Attenuate High Frequency Harmonic
T1
5
A = 5V/DIV
4.53k*
CT
+V
820pF
COL
FB
A = 100µV/DIV
AC COUPLED
LT1534
RT
16.9k*
GND VC
PGND
0.1µF
L2
28nH
RV
33k
RI
33k
AN118 F08
Figure 8. LT1534’s Controlled Transition Times Retard High
Frequency Harmonic and Maintain Low Heat Dissipation.
Approach Blends Switching and Linear Current Sink Benefits
10µs/DIV
AN118 F10
Figure 10. Switched Current Sinks Controlled Transition
Times Dramatically Lower Noise to 150µVP-P —A 20x
Improvement vs Figure 7’s LTC3401 Results
an118fb
AN118-4
Application Note 118
Figure 11 is essentially identical to Figure 8 except that
it produces a negative 1000V output. A1 provides low
impedance, inverting feedback to the LT1534. Figure 12a’s
output noise measures inside 1mV. As before, resonant
Royer ripple dominates the noise — no high frequency
content is detectable. It is worth noting that this noise
figure proportionally improves with increased filter capacitor values. For example, Figure 12b indicates only
100µV noise with filter capacitor values increased by 10x,
although capacitor physical size is large. The original values selected represent a reasonable compromise between
noise performance and physical size.
–1000VOUT
0.015µF, 2kV
DANGER! Lethal Potentials Present — See Text
10k
0.033µF, 2k
9M**
D1-D4
10
6
T1
5
1
0.22µF
820Ω
3
5V
5k
OUTPUT
ADJUST
= ZDT1048 DUAL
CT
8.06k*
1N5817
+V
820pF
COL
FB
LT1006
R1
+
5V
L1 = CTX150-4, COILTRONICS
T1 = 210609R, COILTRONICS
0.033µF/0.015µF = WIMA MKS-4
0.22µF = WIMA MKP-2
D1-D4 = 1N6529
L2 = COILCRAFT B08T
D5-D6 = 1N4148
* = 1% METAL FILM RESISTOR
** = IRC, CGH-1, 1%
2
–
5V
4
LT1534
RT
16.9k*
GND VC
+
PGND
15µF
L2
28nH
RV
24k
RI
39k
AN118 F11
Figure 11. Controlled Transition Time Switching Regulator Applied to a Negative Output,
1000V Converter. A1 Provides Low Impedance, Inverting Feedback to LT1534
A = 100µV/DIV
AC COUPLED
A = 500µV/DIV
AC COUPLED
10µs/DIV
AN118 F12a
Figure 12a. –1000V Converter Output Noise Measures
Inside 1mV (1PPM-0.0001%) in 100MHz Bandwidth.
Resonant Royer Related Ripple Dominates Residue—
No High Frequency Content is Detectable
10µs/DIV
AN118 F12b
Figure 12b. 10x Increase in Figure 11’s Filter Capacitor Values
Reduces Noise to 100µV. Penalty is Capacitor Physical Size
an118fb
AN118-5
Application Note 118
Controlled Transition Push-Pull Converters
Controlled transition techniques are also directly applicable
to push-pull architectures. Figure 13 uses a controlled
transition push-pull regulator in a simple loop to control
a 300V output converter. Symmetrical transformer drive
and controlled switching edge times promote low output
noise. The D1-D4 connected damper further minimizes
residual aberrations. In this case, inductors are used in
the output filter although appropriate resistor values could
be employed.
Figure 14 displays smooth transitions at the transformer
secondary outputs (trace A is T1 Pin 4, trace B, T1 Pin 7).
Absence of high frequency harmonic results in extremely
low noise. Figure 15’s fundamental related output residue approaches the 100µV measurement noise floor in
a 100MHz bandpass. This is spectacularly low noise
performance in any DC/DC converter and certainly in
one providing high voltage. Here, at 300V output, noise
represents less than 1 part in 3 million.
Figure 16 is similar except that output range is variable
from 0V to 300V. The LT1533 is replaced by an LT3439
which contains no control elements. It simply drives the
transformer with 50% duty cycle, controlled switching
transitions. Feedback control is enforced by A1-Q1-Q2
driving current into T1’s primary center tap. A1 compares
a resistively derived portion of the output with a user supplied control voltage. The values shown produce a 0V to
300V output in response to a 0V to 1V control voltage. An
RC network from Q2’s collector to A1’s positive input compensates the loop. Collector waveforms and output noise
signature are nearly identical to Figure 13. Output noise is
100µVP-P over the entire 0V to 300V output range.
DANGER! Lethal Potentials Present — See Text
0.01µF
L1
330µH
150Ω
L2
330µH
300VOUT
+
10µF
450V
D1-D4
A = 200V/DIV
+ 4.7µF
450V
B = 200V/DIV
4
7 (TIE 5-6)
1M*
T1
1
5V
680pF
+V
2
5V
COLA
3
COLB
CT
16.9k*
2µs/DIV
Figure 14. Transformer Secondary Outputs
Show No High Frequency Artifacts
FB
1k
OUTPUT ADJUST
LT1533
RT
3.83k*
0.1µF
VC
AN118 F14
GND PGND
L3
28nH
RV
33k
RI
43k
AN118 F13
* = 1% METAL FILM RESISTOR
L3 = COILCRAFT B08T
L1, L2 = COILCRAFT LPS5010-334MLB
D1-D4 = 1N6529
T1 = PICO 32195
Figure 13. A Push-Pull Drive, Controlled Transition,
300V Output Converter. Symmetrical Transformer
Drive and Slow Edges Promote Low Output Noise
A = 100µV/DIV
AC COUPLED
5µs/DIV
AN118 F15
Figure 15. Push-Pull Converter Related Residue Approaches
100µV Measurement Noise Floor. No Wideband Components
Appear in 100MHz Measurement Bandpass
an118fb
AN118-6
Application Note 118
Flyback Converters
voltage stress. Q1, operating as a “cascode” with the
LT1172’s internal switch, withstands L1’s high voltage
flyback events6.
Flyback converters, due to their abrupt, poorly controlled
energy delivery, are not usually associated with low noise
output. However, careful magnetic selection and layout
can provide surprisingly good performance, particularly
at low output current.
Diodes associated with Q1’s source terminal clamp L1
originated spikes arriving via Q1’s junction capacitance. The
high voltage is rectified and filtered, forming the circuits’
output. The ferrite bead, 100Ω and 300Ω resistors aid filter
Figure 17’s design provides 200V from a 5V input5. The
scheme is a basic inductor flyback boost regulator with
some important deviations. Q1, a high voltage device, has
been interposed between the LT1172 switching regulator
and the inductor. This permits the regulator to control
Q1’s high voltage switching without undergoing high
0.01µF
Note 5. LTC application note veterans, a weary crew, will recognize
material in this section from AN98 and AN113. The original circuits and
text have been modified as necessary to suit low noise operation. See
References.
Note 6. See References 13-17 for historical perspective and study on
cascodes.
L1
330µH
150Ω
+
+
10µF
400V
1µF
DANGER! Lethal Potentials Present — See Text
4
T1
7 (TIE 5-6)
1
2
10k
0.001µF
1M*
100k
–
3.32k*
2N3906
Q1
100µF
2.2µF
510k
5V
100Ω
680pF
COLA
CT
16.9k*
100k
LTC6240
D44
Q2
3
+
+
A1
5V
* = 1% METAL FILM RESISTOR
T1 = PICO 32195
5V
D1-D4 = 1N6529
L1 = COILCRAFT LP5010-334MLB
VOUT
0 TO 300V
4.7µF
400V
COLB
510k
LT3439
RT
+V
GND
RSL
27k
VCONTROL
0 TO 1V
PGND
AN118 F16
Figure 16. Full Range Adjustable Version of Figure 13. VCONTROL Directed A1 Sets T1 Drive Via Q1-Q2. 1M-3.32k Divider
Provides Feedback, Stabilized by A1’s Input Capacitors. Waveforms Are Similar to Figure 13. Output Noise is 100µVP-P
5V
+
1N5712
33µF
100k
1µF
1N5256B
30V 5%
15V
1N4702
Q1
IRF840
1N5819
300Ω
OUTPUT
200V
1µF = 2x –
0.47µF
250V
1M*
LT1172
VIN
FB
1µF
VC
= FERRITE BEAD FERRONICS 21-110J
* = 1% METAL FILM RESISTOR
L1 = 33µH, COILTRONICS UP2B
0.47µF = PANASONIC ECW-U2474KCV
GND
E2
BAS521
0.47µF
250V
SW
5V
+
DANGER! Lethal Potentials Present — See Text
L1
BAS521 100Ω
E1
6.19k*
1k
+
AN118 F17
1µF
Figure 17. 5V to 200V Output Converter. Cascoded Q1 Switches High Voltage, Allowing Low Voltage Regulator to Control Output. Diode
Clamps Protect Regulator from Transients; 100k Path Bootstraps Q1’s Gate Drive from L1’s Flyback Events. Output Connected 300ΩDiode Combination Provides Short-Circuit Protection. Ferrite Bead, 100Ω and 300Ω Resistors Minimize High Frequency Output Noise
an118fb
AN118-7
Application Note 118
efficiency7. Feedback to the regulator stabilizes the loop
and the VC pin network provides frequency compensation.
A 100k path from L1 bootstraps Q1’s gate drive to about
10V, ensuring saturation. The output connected diode
provides short-circuit protection by shutting down the
LT1172 if the output is accidentally grounded.
Figure 18’s traces A and C are LT1172 switch current and
voltage, respectively. Q1’s drain is trace B. Current ramp
termination results in a high voltage flyback event at Q1’s
drain. A safely attenuated version of the flyback appears
at the LT1172 switch. The sinosoidal signature, due to
inductor ring-off between conduction cycles, is harmless.
Figure 19, output noise, is composed of low frequency
ripple and wideband, flyback related spikes measuring
1mVP-P in a 100MHz bandpass.
Figure 20, contributed by Albert M. Wu of LTC, is a transformer coupled flyback circuit. The transformer secondary
provides voltage step-up referred to the flyback driven
primary. The 4.22M resistor supplies feedback to the regulator, closing a control loop. A 10k-0.68µF filter network
attenuates high frequency harmonic with minimal voltage
drop. Flyback related transients are clearly visible in Figure
21’s output noise although within 300µVP-P.
Note 7. Tutorial on ferrite beads appears in Appendix F.
DANGER! Lethal Potentials Present — See Text
T1
1:10:4
VIN
5V
7, 8
D1
4.7µH
A = 0.5A/DIV
5, 6
VIN
4
SYNC SS
C = 20V/DIV
464k
2µs/DIV
AN118 F18
VOUT = 350V
C3
0.68µF
400V
1N5817
C2
68nF
SW
C1
SHDN
GND
2.2µF
LT3580
RT
FB
B = 100V/DIV
10k
1
RFB 4.22M*
VC
10k
0.47µF
10nF
100pF
AN118 F20
Figure 18. Waveforms for 5V to 200V Converter Include LT1172
Switch Current and Voltage (Traces A and C, Respectively) and
Q1’s Drain Voltage (Trace B). Current Ramp Termination Results
in High Voltage Flyback Event at Q1 Drain. Safely Attenuated
Version Appears at LT1172 Switch. Sinosoidal Signature,
Due to Inductor Ring-Off Between Current Conduction Cycles,
is Harmless. All Traces Intensified Near Center Screen for
Photographic Clarity
C1: 2.2µF, 25V, X5R, 1206
C2: TDK C3225X7R2J683M
D1: VISHAY GSD2004S DUAL DIODE CONNECTED IN SERIES
T1: TDK LDT565630T-041
C3: WIMA MKS-4
* = IRC-CGH-1, 1%
Figure 20. 5V Powered Transformer Coupled
Flyback Converter Produces 350V Output
A = 200µV/DIV
AC COUPLED
1mV/DIV
AC COUPLED
5µs/DIV
AN118 F19
Figure 19. Figure 17’s Output Noise, Composed of Low
Frequency Ripple and Wideband, Flyback Related
Spikes, Measures 1mVP-P in 100MHz Bandpass
2µs/DIV
AN118 F21
Figure 21. High Speed Transients in Figure 20’s
Noise Signature are Within 300µVP-P
an118fb
AN118-8
Application Note 118
Figure 24’s post-regulator reduces Figure 22’s output
ripple and noise to only 2mV. A1 and the LT3468 are
identical to the previous circuit, except for the 15V zener
diode in series with the 10M-100k feedback divider. This
component causes C1’s voltage, and hence Q1’s collector,
to regulate 15V above the VPROGRAM input dictated point.
The VPROGRAM input is also routed to the A2-Q2-Q1 linear
post-regulator. A2’s 10M-100k feedback divider does not
include a zener, so the post-regulator follows the VPROGRAM
input with no offset. This arrangement forces 15V across
Q1 at all output voltages. This figure is high enough to
eliminate undesirable ripple and noise from the output
while keeping Q1 dissipation low.
5
4.7µF
T1
8
VIN
+
C1
13µF
330V
0V TO 300VOUT
5mA MAXIMUM
1
SW
LT3468
CHARGE
D1
4
D2
GND
10M*
DONE
A1
LT1006
33pF
+
Figure 23’s 250V DC output (Trace B) decays down about
2V until A1 (Trace A) goes high, enabling the LT3468 and
restoring the loop. This simple circuit works well, regulating over a programmable 0V to 300V range, although its
inherent hysteretic operation mandates the (unacceptable)
2V output ripple noted. Loop repetition rate varies with
input voltage, output set point and load but the ripple is
always present. The following circuit greatly reduces ripple
amplitude although complexity increases.
5V
–
Figure 22 employs the LT3468 photoflash capacitor charger
as a general purpose high voltage DC/DC converter. Normally, the LT3468 regulates its output at 300V by sensing
T1’s flyback pulse characteristic. This circuit allows the
LT3468 to regulate at lower voltages by truncating its
charge cycle before the output reaches 300V. A1 compares
a divided down portion of the output with the program
input voltage. When the program voltage (A1 + input) is
exceeded by the output derived potential (A1 – input) A1’s
output goes low, shutting down the LT3468. The feedback
capacitor provides AC hysteresis, sharpening A1’s output
to prevent chattering at the trip point. The LT3468 remains
shut down until the output voltage drops low enough to trip
A1’s output high, turning it back on. In this way, A1 duty
cycle modulates the LT3468, causing the output voltage
to stabilize at a point determined by the program input.
100k*
10k
10k
VPROGRAM INPUT
0V TO 3V = 0V TO 300VOUT
*1% METAL FILM RESISTOR
C1: RUBYCON 330FW13AK6325
AN118 F22
D1: TOSHIBA DUAL DIODE 1SS306,
CONNECT DIODES IN SERIES
D2: PANASONIC MA2Z720
T1: TDK LDT565630T-002
Figure 22. A Voltage Programmable 0V to 300V Output
Regulator. A1 Controls Regulator Output by Duty Cycle
Modulating LT3468/T1 DC/DC Converter Power Delivery
A = 5V/DIV
A = 1V/DIV
AC COUPLED
ON 250V DC
LEVEL
20ms/DIV
AN118 F23
Figure 23. Details of Figure 22’s Duty Cycle Modulated
Operation. High Voltage Output (Trace B) Ramps Down
Until A1 (Trace A) Goes High, Enabling LT3468/T1 to
Restore Output. Loop Repetition Rate Varies with Input
Voltage, Output Set Point and Load
Q3 and Q4 form a current limit, protecting Q1 from overload.
Excessive current through the 50Ω shunt turns Q3 on. Q3
drives Q4, shutting down the LT3468. Simultaneously, a
portion of Q3’s collector current turns Q2 on hard, shutting
off Q1. This loop dominates the normal regulation feedback,
protecting the circuit until the overload is removed.
an118fb
AN118-9
Application Note 118
DANGER! Lethal Potentials Present — See Text
Q3
D1
5V
5
4.7µF
T1
Q1
4
8
+
1
C1
13µF
330V
200k
1N4702
15V
LT3468 GND
CHARGE
DONE
†
0.01µF
0V TO 300VOUT
5mA MAXIMUM
10M*
10k
+
10k
Q2
D2
SW
1k
D3
0.68µF†
VIN
10k
50Ω
1k
A2
1/2 LT1013
0.1µF
100k**
–
10M*
100k
5V
A1
1/2 LT1013
33pF
+
Q4
2N3904
–
10k
100k**
10k
10k
100k
10k
AN118 F24
*1% METAL FILM RESISTOR
**0.1% METAL FILM RESISTOR
†WIMA MKS-4, 400V
C1: RUBYCON 330FW13AK6325
D1: TOSHIBA DUAL DIODE 1SS306,
CONNECT DIODES IN SERIES
D2: PANASONIC MA2Z720
D3: 1N4148
Q1, Q2: 2N6517
Q3: 2N6520
T1: TDK LDT565630T-002
VPROGRAM INPUT
0V TO 3V = 0V TO 300VOUT
Figure 24. Post-Regulation Reduces Figure 22’s 2V Output Ripple to 2mV. LT3468-Based DC/DC Converter, Similar
to Figure 22, Delivers High Voltage to Q1 Collector. A2, Q1, Q2 Form Tracking, High Voltage Linear Regulator.
Zener Sets Q1 VCE = 15V, Ensuring Tracking with Minimal Dissipation. Q3-Q4 Limit Short-Circuit Output Current
Figure 25 shows just how effective the post regulator
is. When A1 (trace A) goes high, Q1’s collector (trace B)
ramps up in response (note LT3468 switching artifacts
on ramps upward slope). When the A1-LT3468 loop is
satisfied, A1 goes low and Q1’s collector ramps down.
The output post-regulator (trace C), however, rejects the
ripple, showing only 2mV of noise. Slight trace blurring
derives from A1-LT3468 loop jitter.
Summary of Circuit Characteristics
Figure 26 summarizes the circuits presented with salient
characteristics noted. This chart is only a generalized
guideline and not an indicator of capabilities or limits.
There are too many variables and exceptions to accomodate the categorical statement a chart implies. The
interdependence of circuit parameters makes summarizing
or rating various approaches a hazardous exercise. There
is simply no intellectually responsible way to streamline
the selection and design process if optimum results are
desired. A meaningful choice must be the outcome of
laboratory-based experimentation. There are just too many
interdependent variables and surprises for a systematic,
AN118-10
A = 5V/DIV
B = 0.1V/DIV
AC COUPLED
C = 5mV/DIV
AC COUPLED
ON 200V DC
LEVEL
100µs/DIV
AN118 F25
Figure 25. Low Ripple Output (Trace C) is Apparent in PostRegulator’s Operation. Traces A and B are A1 Output and Q1’s
Collector, Respectively. Trace Blurring, Right of Photo Center,
Derives from Loop Jitter
theoretically based selection. Charts seek authority through
glib simplification and simplification is Disaster’s deputy.
Nonetheless, Figure 26, in all its appropriated glory, lists
input supply range, output voltage and current along with
comments for each circuit8.
Note 8. Readers detecting author ambivalence at Figure 26’s inclusion are
not hallucinating. Locally based marketeers champion such charts; the
writer is less enthusiastic.
an118fb
Application Note 118
FIGURE
NUMBER
SUPPLY RANGE
(1mA LOAD)
LT1635 - Linear Resonant Royer
1
2.7V to 12V
2mA at 250V
<100µV Wideband Noise. Easily Voltage Controlled.
Potential Dissipation Issue at High Supply Voltages.
LT1431 - Linear Resonant Royer
4
2.7V to 32V
2mA at 250V
<100µV Wideband Noise. Wide Supply Range.
Potential Dissipation Issue at High Supply Voltages.
LT3401 - Switched Resonant Royer
5
2.7V to 5V
3.5mA at 250V
3mV Wideband Noise. High Output Current, Better
Efficiency than Figures 1 and 4.
LT1534 - Switched Resonant Royer
8
2.7V to 15V
2mA at 250V
≈100µV Wideband Noise. Good Trade-Off Between
Figures 1, 4 and 5.
LT1534 - Swiched Resonant Royer
11
4.5V to 15V
1.2mA at –1000V
LT1533 Push-Pull
13
2.7V to 15V
2mA at 300V
LT3439 Push-Pull
16
4.5V to 6V
2mA at 0V to 300V
CIRCUIT TYPE
MAXIMUM OUTPUT
CURRENT AT TEST VOLTAGE COMMENTS
1mV Wideband Noise Reducable to 100µV. Negative
1000V Output Suits Photomultiplier Tubes.
≈100µV Wideband Noise.
Full Range Adjustable Version of Figure 13. ≈100µV
Wideband Noise.
LT1172 - Cascode Inductor Flyback
17
3.5V to 30V
2mA at 200V
VOUT Limit ≈200V. ≈1mV Wideband Noise.
LT3580 - XFMR Flyback
20
2.7V to 20V
4mA at 350V
300µV Wideband Noise. Wide Supply Range. High
Output Current. Small Transformer.
LT3468 - LT1006 XFMR Flyback
22
3.8V to 12V
5mA at 250V
1.5V Noise. Simple Voltage Control Input 0VIN to 3VIN
= 0VOUT – 300VOUT.
LT3468 - LT1013 XFMR Flyback
- Linear
24
3.8V to 12V
5mA at 250V
2mV Wideband Noise. Voltage Control Input
0VIN to 3VIN = 0VOUT to 300VOUT.
Figure 26. Summarized Characteristics of Techniques Presented. Applicable Circuit Depends on Application Specifics
Note: This application note was derived from a manuscript originally prepared for publication in EDN magazine.
REFERENCES
1. Williams, Jim, “A Fourth Generation of LCD Backlight
Technology,” Linear Technology Corporation, Application Note 65, November 1995, p. 32-34, 119.
6. Williams, Jim, “Minimizing Switching Residue in
Linear Regulator Outputs”. Linear Technology Corporation, Application Note 101, July 2005.
2. Bright, Pittman and Royer, “Transistors As On-Off
Switches in Saturable Core Circuits,” Electrical Manufacturing, December 1954. Available from Technomic
Publishing, Lancaster, PA.
7. Morrison, Ralph, “Grounding and Shielding Techniques in Instrumentation,” Wiley-Interscience,
1986.
3.
4.
Williams, Jim, “A Monolithic Switching Regulator with
100µV Output Noise,” Linear Technology Corporation,
Application Note 70, October 1997.
Baxendall, P.J., “Transistor Sine-Wave LC Oscillators,”
British Journal of IEEE, February 1960, Paper No.
2978E.
5. Williams, Jim, “Low Noise Varactor Biasing with
Switching Regulators,” Linear Technology Corporation, Application Note 85, August 2000, p. 4-6.
8. Fair-Rite Corporation, “Fair-Rite Soft Ferrites,” FairRite Corporation, 1998.
9. Sheehan, Dan, “Determine Noise of DC/DC Converters,” Electronic Design, September 27, 1973.
10. Ott, Henry W., “Noise Reduction Techniques in Electronic Systems,” Wiley Interscience, 1976.
11. Tektronix, Inc. “Type 1A7A Differential Amplifier
Instruction Manual,” “Check Overall Noise Level
Tangentially”, p. 5-36 and 5-37, 1968.
an118fb
AN118-11
Application Note 118
12. Witt, Jeff, “The LT1533 Heralds a New Class of Low Noise
Switching Regulators,” Linear Technology, Vol. VII, No.
3, August 1997, Linear Technology Corporation.
16. Williams, Jim, “Signal Sources, Conditioners and
Power Circuitry,” Linear Technology Corporation,
Application Note 98, November 2004, p. 20-21.
13. Williams, Jim, “Bias Voltage and Current Sense Circuits
for Avalanche Photodiodes,” Linear Technology Corporation, Application Note 92, November 2002, p.8.
17. Williams, Jim, “Power Conversion, Measurement
and Pulse Circuits,” Linear Technology Corporation,
Application Note 113, August 2007.
14. Williams, Jim, “Switching Regulators for Poets,” Appendix D, Linear Technology Corporation, Application
Note 25, September 1987.
18. Williams, Jim and Wu, Albert, “Simple Circuitry for
Cellular Telephone/Camera Flash Illumination,” Linear
Technology Corporation, Application Note 95, March
2004.
15. Hickman, R.W. and Hunt, F.V., “On Electronic Voltage
Stabilizers,” “Cascode,” Review of Scientific Instruments, January 1939, p. 6-21, 16.
19. LT3580 Data Sheet, Linear Technology Corporation.
APPENDIX A
FROM HV
RECTIFICER
OUT
FROM HV RECTIFICER
FROM HV
RECTIFICER
R
+VIN
OUT
+VIN
TO FB
TO FB
(a)
TO FB
(b)
FROM HV
RECTIFICER R
OUT
FROM HV
RECTIFICER
OUT
(c)
STRAY FLUX
C PARASITIC
R
OUT
FROM
HV
RECTIFIER
+VIN
+VIN
TO FB
L
OUT
LOSS TERMS
TO FB
LOSS TERMS
(d)
(e)
(f)
AN118 A1
Figure A1. Feedback Network Options. (a) Is Basic DC Feedback. (b) Adds AC Lead Network for Improved Dynamics. Diode Clamps
Protect Feedback Node from Capacitor’s Differentiated Response. (c)’s Low Ripple Two Section Filter Slows Loop Transmission but
Lead Network Provides Stability. Resistor R Sets DC Output Impedance. (d) Encloses R in DC Loop, Lowering Output Resistance.
Feedback Capacitor Supplies Leading Response. (e) Moves Feedback Capacitor to Filter Input, Further Extending (d)’s Leading
Response. (f), Replacing Filter Resistor (R) with Inductor, Lowers Output Resistance but Introduces Parasitic Shunt Capacitive Path
and Stray Flux Sensitivity
Feedback Considerations in High Voltage
DC/DC Converters
voltage induced overstress protection. Figure A1 lists
typical options.
A high voltage DC/DC converter feedback network is a
study in compromise. The appropriate choice is application dependent. Considerations include desired output
impedance, loop stability, transient response and high
(a) is basic DC feedback and requires no special commentary. (b) adds an AC lead network for improved
dynamics. Diode clamps protect the feedback node from
the capacitors differentiated response. (c)’s low ripple,
an118fb
AN118-12
Application Note 118
two section filter slows transient response but a lead
network provides stability. Resistor R, outside the loop,
sets DC output impedance. (d) encloses R within the
DC loop, lowering output resistance but delaying loop
transmission. A feedback capacitor supplies corrective
leading response. (e) moves the feedback capacitor to the
filter input, further extending (d)’s leading response. (f)
replaces filter resistor R with an inductor, lowering output
resistance but introducing parasitic shunt capacitance
which combines with capacitor loss terms to degrade
filtering. The inductor also approximates a transformer
secondary, vulnerable to stray flux pick-up with resulting
increased output noise1.
A common concern in any high voltage feedback network
is reliability. Components must be quite carefully chosen.
Voltage ratings should be conservative and strictly adhered
to. While component ratings are easily ascertained, more
subtle effects such as ill-suited board material and board
wash contaminants can be reliability hazards. Long term
electro-migration effects can have undesirable results.
Every potential unintended conductive path should be
considered as an error source and layout planned accordingly. Operating temperature, altitude, humidity and
condensation effects must be anticipated. In extreme cases,
it may be necessary to rout the board under components
operating at high voltage. Similarly, it is common practice
to use several units in series to minimize voltage across
the output connected feedback resistor. Contemporary
packaging requirements emphasize tightly packed layout
which may conflict with high voltage standoff requirements.
This tradeoff must be carefully reviewed or reliability will
suffer. The potentially deleterious (disastrous) effects of
environmental factors, layout and component choice over
time cannot be overstated. Clear thinking is needed to
avoid unpleasant surprises.
Note 1. See Appendix G.
Editor’s Note: Appendices B through E are thinly edited and modified versions of tutorials first
appearing in AN70. Although originally intended to address controlled transition applications
(e.g. LT1533, 4 and LT3439) the material is directly relevant and warrants inclusion here.
APPENDIX B
SPECIFYING AND MEASURING SOMETHING CALLED NOISE
Undesired output components in switching regulators are
commonly referred to as “noise.” The rapid, switched mode
power delivery that permits high efficiency conversion
also creates wideband harmonic energy. This undesirable
energy appears as radiated and conducted components, or
“noise.” Actually switching regulator output “noise” isn’t
really noise at all, but coherent, high frequency residue
directly related to the regulator’s switching. Unfortunately,
it is almost universal practice to refer to these parasitics
as “noise,” and this publication maintains this common,
albeit inaccurate, terminology.1
Measuring Noise
There are an almost uncountable number of ways to
specify noise in a switching regulator’s output. It is common industrial practice to specify peak-to-peak noise in
a 20MHz bandpass.2 Realistically, electronic systems are
readily upset by spectral energy beyond 20MHz, and this
specification restriction benefits no one.3 Considering all
this, it seems appropriate to specify peak-to-peak noise in
a verified 100MHz bandwidth. Reliable low level measurements in this bandpass require careful instrumentation
choice and connection practices.
Our study begins by selecting test instrumentation and
verifying its bandwidth and noise. This necessitates the
arrangement shown in Figure B1. Figure B2 diagrams
signal flow. The pulse generator supplies a subnanosecNote 1. Less genteelly, “If you can’t beat ‘em, join ‘em.”
Note 2. One DC/DC converter manufacturer specifies RMS noise in a
20MHz bandwidth. This is beyond deviousness and unworthy of comment.
Note 3. Except, of course, eager purveyors of power sources who specify
them in this manner.
an118fb
AN118-13
Figure B1. 100MHz Bandwidth Verification Test Setup.
Note Coaxial Connections for Wideband Signal Integrity
Application Note 118
an118fb
AN118-14
Application Note 118
ond rise time step to the attenuator, which produces a
<1mV version of the step. The amplifier takes 40dB of
gain (A = 100) and the oscilloscope displays the result.
The “front-to-back” cascaded bandwidth of this system
should be about 100MHz (tRISE = 3.5ns) and Figure B3
reveals this to be so. Figure B3’s trace shows 3.5ns rise
PULSE
GENERATOR
HP-215A
<1ns RISE TIME = 350MHz
ZIN
50Ω
ATTENUATOR
HP-355D
1000MHz
time and about 100µV of noise. The noise is limited by
the amplifier’s 50Ω noise floor.4
Note 4. Observed peak-to-peak noise is somewhat affected by the
oscilloscope’s “intensity” setting. Reference 11 describes a method for
normalizing the measurement.
ZIN
50Ω
<1mV
≈1ns RISE TIME
(350MHz)
AMPLIFIER
X40dB
HP-461A
50Ω
OSCILLOSCOPE
TEKTRONIX 454A
150MHz
(tr = 2.4ns)
150MHz
(tr = 2.4ns)
CASCADED BANDWIDTH ≈ 100MHz
(≈3.5ns RISE TIME)
AN118 B2
Figure B2. Subnanosecond Pulse Generator and Wideband
Attenuator Provide Fast Step to Verify Test Setup Bandwidth
100µV/DIV
100µV/DIV
2ns/DIV
AN118 B3
10µs/DIV
Figure B3. Oscilloscope Display Verifies Test Setup’s
100MHz (3.5ns Rise Time) Bandwidth. Baseline Noise
Derives from Amplifier’s 50Ω Input Noise Floor
100µV/DIV
AN118 B4
Figure B4. Output Switching Noise Is Just
Discernible in a 100MHz Bandpass
10mV/DIV
10µs/DIV
AN118 B5
Figure B5. 10MHz Band Limited Version of Preceding
Photo. All Switching Noise Information is Preserved,
Indicating Adequate Bandwidth
50µs/DIV
AN118 B6
Figure B6. Commercially Available Switching
Regulator’s Output Noise in a 1MHz Bandpass.
Unit Appears to Meet its 5mVP-P Noise Specification
an118fb
AN118-15
Application Note 118
Figure B4’s presentation of output noise shows barely visible switching artifacts (at vertical graticule lines 4, 6 and 8)
in the 100MHz bandpass. Fundamental ripple is seen more
clearly, although similarly noise floor dominated. Restricting measurement bandwidth to 10MHz (Figure B5) reduces
noise floor amplitude, although switching noise and ripple
amplitudes are preserved. This indicates that there is no
signal power beyond 10MHz. Further measurements as
bandwidth is successively reduced can determine the
highest frequency content present.
The importance of measurement bandwidth is further
illustrated by Figures B6 to B8. Figure B6 measures
a commercially available DC/DC converter in a 1MHz
bandpass. The unit appears to meet its claimed 5mVP-P
noise specification. In Figure B7, bandwidth is increased
10mV/DIV
to 10MHz. Spike amplitude enlarges to 6mVP-P, about
1mV outside the specification limit. Figure B8’s 50MHz
viewpoint brings an unpleasant surprise. Spikes measure
30mVP-P —six times the specified limit!5
Low Frequency Noise
Low frequency noise is rarely a concern, because it
almost never affects system operation. Low frequency
noise is shown in Figure B9. It is possible to reduce low
frequency noise by rolling off control loop bandwidth.
Figure B10 shows about a five times improvement when
this is done, even with greater measurement bandwidth.
A possible disadvantage is loss of loop bandwidth and
slower transient response.
Note 5. Caveat Emptor.
20mV/DIV
50µs/DIV
AN118 B7
Figure B7. Figure B6’s Regulator Noise in a 10MHz Bandpass.
6mVP-P Noise Exceeds Regulator’s Claimed 5mV Specification
500µV/DIV
50µs/DIV
AN118 B8
Figure B8. Wideband Observation of Figure B7 Shows
30mVP-P Noise—Six Times the Regulator’s Specification!
50µV/DIV
10ms/DIV
AN118 B9
Figure B9. 1Hz to 3kHz Noise Using Standard Frequency
Compensation. Almost All Noise Power is Below 1kHz
10ms/DIV
AN118 B10
Figure B10. Feedback Lead Network Decreases Low Frequency
Noise, Even as Measurement Bandwidth Expands to 100kHz
an118fb
AN118-16
Application Note 118
Preamplifier and Oscilloscope Selection
The low level measurements described require some form
of preamplification for the oscilloscope. Current generation
oscilloscopes rarely have greater than 2mV/DIV sensitivity,
although older instruments offer more capability. Figure
B11 lists representative preamplifiers and oscilloscope
plug-ins suitable for noise measurement. These units
feature wideband, low noise performance. It is particularly
significant that the majority of these instruments are no
longer produced. This is in keeping with current instrumentation trends, which emphasize digital signal acquisition
as opposed to analog measurement capability.
The monitoring oscilloscope should have adequate bandwidth and exceptional trace clarity. In the latter regard
high quality analog oscilloscopes are unmatched. The
exceptionally small spot size of these instruments is wellsuited to low level noise measurement.6 The digitizing
uncertainties and raster scan limitations of DSOs impose
display resolution penalties. Many DSO displays will not
even register the small levels of switching-based noise.
Note 6.In our work we have found Tektronix types 454, 454A, 547 and 556
excellent choices. Their pristine trace presentation is ideal for discerning
small signals of interest against a noise floor limited background.
INSTRUMENT
TYPE
MANUFACTURER
MODEL
NUMBER
–3dB
BANDWIDTH
MAXIMUM
SENSITIVITY/GAIN AVAILABILITY
Amplifier
Hewlett-Packard
461A
175MHz
Gain = 100
Secondary Market
50Ω Input, Stand-Alone. 100µVP-P
(≈20µV RMS) noise in 100MHz
bandwidth. Best of this group for noise
measurement described in text.
Differential Amplifier
Tektronix
1A5
50MHz
1mV/DIV
Secondary Market
Requires 500 Series Mainframe
Differential Amplifier
Tektronix
7A13
100MHz
1mV/DIV
Secondary Market
Requires 7000 Series Mainframe
COMMENTS
Differential Amplifier
Tektronix
11A33
150MHz
1mV/DIV
Secondary Market
Requires 11000 Series Mainframe
Differential Amplifier
Tektronix
P6046
100MHz
1mV/DIV
Secondary Market
Stand-Alone
Differential Amplifier
Preamble
1855
100MHz
Gain = 10
Current Production Stand-Alone, Settable Bandstops
Differential Amplifier
Tektronix
1A7/1A7A
1MHz
10µV/DIV
Secondary Market
Requires 500 Series Mainframe,
Settable Bandstops
Differential Amplifier
Tektronix
7A22
1MHz
10µV/DIV
Secondary Market
Requires 7000 Series Mainframe,
Settable Bandstops
Differential Amplifier
Tektronix
5A22
1MHz
10µV/DIV
Secondary Market
Requires 5000 Series Mainframe,
Settable Bandstops
Differential Amplifier
Tektronix
ADA-400A
1MHz
10µV/DIV
Current Production Stand-Alone with Optional Power
Supply, Settable Bandstops
Differential Amplifier
Preamble
1822
10MHz
Gain = 100
Current Production Stand-Alone, Settable Bandstops
Differential Amplifier
Stanford
Research
Systems
SR-560
1MHz
Gain = 50000
Current Production Stand-Alone, Settable Bandstops,
Battery or Line Operation
Differential Amplifier
Tektronix
AM-502
1MHz
Gain = 100000
Secondary Market
Requires TM-500 Series Power Supply
Figure B11. Some Applicable High Sensitivity, Low Noise Amplifiers. Trade-Offs Include Bandwidth, Sensitivity and
Availability. All Require Protective Input Network to Prevent Catastrophic Failure. See Figure B12 and Associated Text
an118fb
AN118-17
Application Note 118
Auxiliary Measurement Circuits
Figure B12 is the clamp circuit referred to in the preceding
figure caption. It must be employed with any of Figure
B12’s amplifiers to insure protection against catastrophic
overloading.7 The network is simply an AC coupled diode
clamp. The coupling capacitor specified withstands the
text examples high voltage outputs and the 10M resistors
bleed residual capacitor charge. Built into a small BNC
equipped enclosure, its output should be directly connected to the amplifier. 50Ω inputs may be directly driven;
high impedance input amplifiers should be shunted with
a coaxial 50Ω terminator.
BNC
INPUT
0.68µF, WIMA MKS-4, 1500V
10M
10M
B13’s battery powered, 1 MHz, 1mV square wave amplitude
calibrator facilitates “end-to-end” amplifier—oscilloscope
path gain verification. The 221k resistor associated area is
sensitive to variations in stray capacitance and is shielded
as per the schematic. A 4.5V reference stabilizes output
amplitude against battery voltage change and a peaking
trim optimizes front and trailing corner fidelity. Figure
B14 shows that the simple peaking network does not
quite achieve square corners, but 1mV pulse amplitude is
clearly delineated. Trace thickening in the waveform flats
indicates amplifier noise floor.
Note 7. Don’t say we didn’t warn you.
BNC
OUTPUT.
CONNECT DIRECTLY TO
50Ω INPUT AMPLIFIER.
NO CABLE
200µV/DIV
MUR-110’s
SHIELDED COAXIAL ENCLOSURE
AN118 B12
AN118 B14
500ns/DIV
Figure B12. Coaxially Fixtured Clamp Protects Figure B11’s Low
Noise Amplifiers From High Voltage Inputs. Resistors Insure
Capacitor Discharge
Figure B14. 1mV Amplitude Calibrator Output
Has Minor Corner Rounding but Pulse Flats
Indicate Desired Amplitude. Trace Thickening
Describes Amplifier Noise Floor
221k, 1%
SHIELD
4.5V
9V
+
10µF
LT1019-4.5
IN
OUT
GND
+
V+
22µF
LTC1799
R
GND
OUT
D
10pF
1k PEAKING
50pF
CONNECT
DIRECTLY TO
50Ω INPUT.
NO CABLE
100k
1%
4.5V
AN118 B13
Figure B13. Battery Powered, 1MHz, 1mV Square Wave Amplitude Calibrator Permits
Signal Path Gain Verification. Peaking Trim Optimizes Front and Trailing Corner Fidelity
an118fb
AN118-18
Application Note 118
APPENDIX C
PROBING AND CONNECTION TECHNIQUES FOR LOW LEVEL, WIDEBAND SIGNAL INTEGRITY
The most carefully prepared breadboard cannot fulfill its
mission if signal connections introduce distortion. Connections to the circuit are crucial for accurate information
extraction. The low level, wideband measurements demand
care in routing signals to test instrumentation.
Ground Loops
Figure C1 shows the effects of a ground loop between
pieces of line-powered test equipment. Small current flow
between test equipment’s nominally grounded chassis creates 60MHz modulation in the measured circuit output.
This problem can be avoided by grounding all line powered test equipment at the same outlet strip or otherwise
ensuring that all chassis are at the same ground potential.
100µV/DIV
Similarly, any test arrangement that permits circuit current
flow in chassis interconnects must be avoided.
Pickup
Figure C2 also shows 60Hz modulation of the noise measurement. In this case, a 4-inch voltmeter probe at the
feedback input is the culprit. Minimize the number of test
connections to the circuit and keep leads short.
Poor Probing Technique
Figure C3’s photograph shows a short ground strap affixed
to a scope probe. The probe connects to a point which
provides a trigger signal for the oscilloscope. Circuit output
noise is monitored on the oscilloscope via the coaxial cable
shown in the photo.
500µV/DIV
2ms/DIV
AN118 C1
Figure C1. Ground Loop Between Pieces of Test
Equipment Induces 60Hz Display Modulation
5ms/DIV
AN118 C2
Figure C2. 60Hz Pickup Due to Excessive
Probe Length at Feedback Node
an118fb
AN118-19
Figure C3. Poor Probing Technique. Trigger Probe Ground Lead Can
Cause Ground Loop-Induced Artifacts to Appear in Display
Application Note 118
an118fb
AN118-20
Application Note 118
Figure C4 shows results. A ground loop on the board
between the probe ground strap and the ground referred
cable shield causes apparent excessive ripple in the display.
Minimize the number of test connections to the circuit and
avoid ground loops.
Proper Coaxial Connection Path
In Figure C9, a coaxial cable transmits the noise signal to
the amplifier-oscilloscope combination. In theory, this affords the highest integrity cable signal transmission. Figure
C10’s trace shows this to be true. The former examples
aberrations and excessive noise have disappeared. The
switching residuals are now faintly outlined in the amplifier noise floor. Maintain coaxial connections in the noise
signal monitoring path.
Direct Connection Path
100µV/DIV
(INVERTED)
5µs/DIV
AN118 C4
Figure C4. Apparent Excessive Ripple Results from
Figure C3’s Probe Misuse. Ground Loop on Board
Introduces Serious Measurement Error
Violating Coaxial Signal Transmission—Felony Case
In Figure C5, the coaxial cable used to transmit the circuit output noise to the amplifier-oscilloscope has been
replaced with a probe. A short ground strap is employed
as the probe’s return. The error inducing trigger channel
probe in the previous case has been eliminated; the ’scope
is triggered by a noninvasive, isolated probe.1 Figure C6
shows excessive display noise due to breakup of the
coaxial signal environment. The probe’s ground strap
violates coaxial transmission and the signal is corrupted
by RF. Maintain coaxial connections in the noise signal
monitoring path.
Violating Coaxial Signal Transmission—Misdemeanor
Case
Figure C7’s probe connection also violates coaxial signal
flow, but to a less offensive extent. The probe’s ground
strap is eliminated, replaced by a tip grounding attachment.
Figure C8 shows better results over the preceding case,
although signal corruption is still evident. Maintain coaxial
connections in the noise signal monitoring path.
A good way to verify there are no cable-based errors is
to eliminate the cable. Figure C11’s approach eliminates
all cable between breadboard, amplifier and oscilloscope.
Figure C12’s presentation is indistinguishable from Figure
C10, indicating no cable-introduced infidelity. When results
seem optimal, design an experiment to test them. When
results seem poor, design an experiment to test them.
When results are as expected, design an experiment to
test them. When results are unexpected, design an experiment to test them.
Test Lead Connections
In theory, attaching a voltmeter lead to the regulator’s
output should not introduce noise. Figure C13’s increased
noise reading contradicts the theory. The regulator’s output
impedance, albeit low, is not zero, especially as frequency
scales up. The RF noise injected by the test lead works
against the finite output impedance, producing the 200µV
of noise indicated in the figure. If a voltmeter lead must be
connected to the output during testing, it should be done
through a 10kΩ-10µF filter. Such a network eliminates
Figure C13’s problem while introducing minimal error in
the monitoring DVM. Minimize the number of test lead
connections to the circuit while checking noise. Prevent
test leads from injecting RF into the test circuit.
Note 1. To be discussed. Read on.
an118fb
AN118-21
Application Note 118
Figure C5. Floating Trigger Probe Eliminates Ground Loop, but Output Probe
Ground Lead (Photo Upper Right) Violates Coaxial Signal Transmission
500µV/DIV
5µs/DIV
AN118 C6
Figure C6. Signal Corruption Due to
Figure C5’s Noncoaxial Probe Connection
an118fb
AN118-22
Application Note 118
Figure C7. Probe with Tip Grounding Attachment Approximates Coaxial Connection
100µV/DIV
5µs/DIV
AN118 C8
Figure C8. Probe with Tip Grounding Attachment
Improves Results. Some Corruption Is Still Evident
an118fb
AN118-23
Application Note 118
Figure C9. Coaxial Connection Theoretically Affords Highest Fidelity Signal Transmission
100µV/DIV
5µs/DIV
AN118 C10
Figure C10. Life Agrees with Theory. Coaxial Signal
Transmission Maintains Signal Integrity. Switching
Residuals Are Faintly Outlined in Amplifier Noise
an118fb
AN118-24
Application Note 118
Figure C11. Direct Connection to Equipment Eliminates Possible Cable-Termination
Parasitics, Providing Best Possible Signal Transmission
100µV/DIV
5µs/DIV
AN118 C12
Figure C12. Direct Connection to Equipment Provides
Identical Results to Cable-Termination Approach.
Cable and Termination Are Therefore Acceptable
an118fb
AN118-25
Application Note 118
200µV/DIV
AN118 C13
5µs/DIV
Figure C13. Voltmeter Lead Attached to Regulator Output
Introduces RF Pickup, Multiplying Apparent Noise Floor
Isolated Trigger Probe
The text associated with Figure C5 somewhat cryptically
alluded to an “isolated trigger probe.” Figure C14 reveals
this to be simply an RF choke terminated against ringing.
The choke picks up residual radiated field, generating
an isolated trigger signal. This arrangement furnishes
a ’scope trigger signal with essentially no measurement
corruption. The probe’s physical form appears in Figure
C15. For good results the termination should be adjusted
for minimum ringing while preserving the highest possible
amplitude output. Light compensatory damping produces
Figure C16’s output, which will cause poor ’scope triggering. Proper adjustment results in a more favorable
output (Figure C17), characterized by minimal ringing
and well-defined edges.
Trigger Probe Amplifier
The field around the switching magnetics is small and may
not be adequate to reliably trigger some oscilloscopes. In
such cases, Figure C18’s trigger probe amplifier is useful.
It uses an adaptive triggering scheme to compensate for
variations in probe output amplitude. A stable 5V trigger
output is maintained over a 50:1 probe output range. A1,
operating at a gain of 100, provides wideband AC gain.
The output of this stage biases a 2-way peak detector (Q1
through Q4). The maximum peak is stored in Q2’s emitter capacitor, while the minimum excursion is retained in
Q4’s emitter capacitor. The DC value of the midpoint of
A1’s output signal appears at the junction of the 500pF
capacitor and the 3MΩ units. This point always sits midway
between the signal’s excursions, regardless of absolute
amplitude. This signal-adaptive voltage is buffered by A2
to set the trigger voltage at the LT1394’s positive input.
The LT1394’s negative input is biased directly from A1’s
output. The LT1394’s output, the circuit’s trigger output, is
unaffected by >50:1 signal amplitude variations. An X100
analog output is available at A1.
Figure C19 shows the circuit’s digital output (trace B)
responding to the amplified probe signal at A1 (trace A).
Figure C20 is a typical noise testing setup. It includes the
breadboard, trigger probe, amplifier, oscilloscope and
coaxial components.
L1
PROBE
SHIELDED
CABLE
BNC CONNECTION
TO TERMINATION BOX
L1: J.W. MILLER #100267
TERMINATION BOX
1k DAMPING
ADJUST
4700pF
BNC
OUTPUT
AN70 FC14
Figure C14. Simple Trigger Probe Eliminates Board
Level Ground Loops. Termination Box Components
Damp L1’s Ringing Response
an118fb
AN118-26
Figure C15. The Trigger Probe and Termination Box.
Clip Lead Facilitates Mounting Probe, Is Electrically Neutral
Application Note 118
an118fb
AN118-27
Application Note 118
10mV/DIV
10mV/DIV
AN118 C16
10µs/DIV
10µs/DIV
Figure C16. Misadjusted Termination Causes Inadequate
Damping. Unstable Oscilloscope Triggering May Result
Figure C17. Properly Adjusted Termination Minimizes
Ringing with Small Amplitude Penalty
50Ω
3
2
5V
+
2k
Q2
3M
500pF
0.005µF
A1
LT1227
750Ω
6
4
–
5V
5
1
0.005µF
13
1k
Q3
14
15
10Ω
ANALOG BNC OUTPUT
TO ’SCOPE TRIGGER INPUT
5V
2k
Q1
12
3M
10
+
–
5V
A2
LT1006
Q4
11
470Ω
10µF
+
0.1µF
+
100µF
0.1µF
AN118 C17
2k
470Ω
0.1µF
+
–
LT1394
Q1, Q2, Q3, Q4 = CA3096 ARRAY: TIE SUBSTRATE (PIN 16) TO GROUND
= 1N4148
DIGITAL
TRIGGER
OUT BNC
TO ’SCOPE
AN118 C18
TRIGGER PROBE
AND TERMINATION BOX
(SEE FIGURE C14 FOR DETAILS)
Figure C18. Trigger Probe Amplifier Has Analog and Digital Outputs. Adaptive
Threshold Maintains Digital Output Over 50:1 Probe Signal Variations
A = 1V/DIV
AC COUPLED
B = 5V/DIV
10µs/DIV (UNCALIB)
AN118 C19
Figure C19. Trigger Probe Amplifier Analog (Trace A)
and Digital (Trace B) Outputs
an118fb
AN118-28
Figure C20. Typical Noise Test Setup Includes Trigger Probe,
Amplifier, Oscilloscope and Coaxial Components
Application Note 118
an118fb
AN118-29
Application Note 118
APPENDIX D
BREADBOARDING, NOISE MINIMIZATION AND LAYOUT CONSIDERATIONS
LT1533-based circuit’s low harmonic content allows their
noise performance to be less layout sensitive than other
switching regulators. However, some degree of prudence
is in order. As in all things, cavalierness is a direct route
to disappointment. Obtaining the absolute lowest noise
figure requires care, but performance below 500µV is
readily achieved. In general, lowest noise is obtained by
preventing mixing of ground currents in the return path.
Indiscriminate disposition of ground currents into a bus
or ground plane will cause such mixing, raising observed
output noise. The LT1533’s restricted edge rates mitigate
against corrupted ground path-induced problems, but
best noise performance occurs in a “single-point” ground
scheme. Single-point return schemes may be impractical
in production PC boards. In such cases, provide the lowest
possible impedance path to the power entry point from
the inductor associated with the LT1533’s power ground
pin. (Pin 16). Locate the output component ground returns
as close to the circuit load point as possible. Minimize
return current mixing between input and output sections
by restricting such mixing to the smallest possible common conductive area.
Noise Minimization
The LT1533’s controlled switching times allow extraordinarily low noise DC/DC conversion with surprisingly
little design effort. Wideband output noise well below
500µV is easily achieved. In most situations this level of
performance is entirely adequate. Applications requiring
the lowest possible output noise will benefit from special
attention to several areas.
Noise Tweaking
The slew time versus efficiency trace-off should be weighted
towards lowest noise to the extent tolerable. Typically, slew
times beyond 1.3µs result in “expensive” noise reduction in
terms of lost efficiency, but the benefit is available. The issue
is how much power is expendable to obtain incremental
decreases in output noise. Similarly, the layout techniques
previously discussed should be reviewed. Rigid adherence
to these guidelines will result in correspondingly lower
noise performance. The text’s breadboards were originally
constructed to provide the lowest possible noise levels,
and then systematically degraded to test layout sensitivity. This approach allows experimentation to determine
the best layout without expanding fanatical attention to
details that provide essentially no benefit.
The slow edge times greatly minimize radiated EMI, but
experimentation with the component’s physical orientation
can sometimes improve things. Look at the components
(yes, literally!) and try and imagine just what their residual
radiated field impinges on. In particular, the optional output
inductor may pick up field radiated by other magnetics,
resulting in increased output noise. Appropriate physical
layout will eliminate this effect, and experimentation is
useful. The EMI probe described in Appendix E is a useful
tool in this pursuit and highly recommended.
Capacitors
The filter capacitors used should have low parasitic impedance. Sanyo OS-CON types are excellent in this regard
and contributed to the performance levels quoted in the
text. Tantalum types are nearly as good. The input supply
bypass capacitor, which should be located directly at the
transformer center tap, needs similarly good characteristics. Aluminum electrolytics are not suitable for any
service in LT1533 circuits.
Damper Network
Some circuits may benefit from a small (e.g., 300Ω1000pF) damper network across the transformer secondary
if the absolutely lowest noise is needed. Extremely small
(20µV to 30µV) excursions can briefly appear during the
switching interval when no energy is coming through the
transformer. These events are so minuscule that they are
barely measurable in the noise floor, but the damper will
eliminate them.
an118fb
AN118-30
Application Note 118
Measurement Technique
Strictly speaking, measurement technique is not a way to
obtain lowest noise performance. Realistically, it is essential
that measurement technique be trustworthy. Uncountable
hours have been lost chasing “circuit problems” that in
reality are manifestations of poor measurement technique.
Please read Appendices B and C before pursuing solutions
to circuit noise that isn’t really there.1
Note 1. I do not wax pedantic here. My guilt in this offense runs deep.
APPENDIX E
APPLICATION NOTE E101: EMI “SNIFFER” PROBE
Bruce Carsten Associates, Inc.
6410 NW Sisters Place, Corvallis, Oregon 97330
541-745-3935
The EMI Sniffer Probe1 is used with an oscilloscope to
locate and identify magnetic field sources of electromagnetic interference (EMI) in electronic equipment. The
probe consists of a miniature 10 turn pickup coil located
in the end of a small shielded tube, with a BNC connector
provided for connection to a coaxial cable (Figure E1). The
Sniffer Probe output voltage is essentially proportional to
the rate of change of the ambient magnetic field, and thus
to the rate of change of nearby currents.
The principal advantages of the Sniffer Probe over simple
pickup loops are:
1. Spatial resolution of about a millimeter.
2. Relatively high sensitivity for a small coil.
3. A 50Ω source termination to minimize cable reflections
with unterminated scope inputs.
4. Faraday shielding to minimize sensitivity to electric
fields.
The EMI Sniffer Probe was developed to diagnose sources
of EMI in switch mode power converters, but it can also
be used in high speed logic systems and other electronic
equipment.
SOURCES OF EMI
Rapidly changing voltages and currents in electrical and
electronic equipment can easily result in radiated and conducted noise. Most EMI in switch mode power converters
is thus generated during switching transients when power
transistors are turned on or off.
Conventional scope probes can readily be used to see
dynamic voltages, which are the principal sources of
common mode conducted EMI. (High dV/dt can also feed
through poorly designed filters as normal mode voltage
spikes and may radiate fields from a circuit without a
conductive enclosure.)
Dynamic currents produce rapidly changing magnetic
fields which radiate far more easily than electric fields as
they are more difficult to shield. These changing magnetic
fields can also induce low impedance voltage transients in
other circuits, resulting in unexpected normal and common
mode conducted EMI.
These high dl/dt currents and resultant fields can not be
directly sensed by voltage probes, but are readily detected
and located with the Sniffer Probe. While current probes
can sense currents in discrete conductors and wires, they
are of little use with printed circuit traces or in detecting
dynamic magnetic fields.
PROBE RESPONSE CHARACTERISTICS
The Sniffer Probe is sensitive to magnetic fields only along
the probe axis. This directionality is useful in locating the
paths and sources of high dl/dt currents. The resolution
is usually sufficient to locate which trace on a printed
circuit board, or which lead on a component package, is
conducting the EMI generating current.
For “isolated” single conductors or PC traces, the Probe
response is greatest just to either side of the conductor
Note 1. The EMI Sniffer Probe is available from Bruce Carsten Associates
at the address noted in the title of this appendix.
an118fb
AN118-31
Application Note 118
© 1997, Bruce Carsten Associates, Inc.
*Approx. 160µ Wire, 1.5mm Coil Dia.
Figure E1. Construction of the EMI “Sniffer Probe” for Locating and Identifying Magnetic Field Sources of EMI
where the magnetic flux is along with probe axis. (Probe
response may be a little greater with the axis tilted towards
the center of the conductor.) As shown in Figure E2, there
is a sharp response null in the middle of the conductor,
with a 180° phase shift to either side and a decreasing
response with distance. The response will increase on the
inside of a bend where the flux lines are crowded together,
and is reduced on the outside of a bend where the flux
lines spread apart.
When the return current is in an adjacent parallel conductor, the Probe response is greatest between the two
conductors as shown in Figure E3. There will be a sharp
null and phase shift over each conductor, with a lower
peak response outside the conductor pair, again decreasing with distance.
The response to a trace with a return current on the opposite side of the board is similar to that of a single isolated
trace, except that the probe response may be greater with
the Probe axis tilted away from the trace. A “ground plane”
below a trace will have a similar effect, as there will be a
counter-flowing “image” current in the ground plane.
The Probe frequency response to a uniform magnetic
field is shown in Figure E4. Due to large variations in
field strength around a conductor, the Probe should be
considered as a qualitative indicator only, with no attempt made to “calibrate” it. The response fall-off near
300MHz is due to the pickup coil inductance driving the
coax cable impedance, and the mild resonant peaks (with
a 1MΩ scope termination) at multiples of 80MHz are due
to transmission line reflections.
an118fb
AN118-32
Application Note 118
PROBE
PROBE
PC TRACES
PC TRACE
PCB
PROBE VOLTAGE
PROBE VOLTAGE
PCB
Figure E2. Sniffer Probe Response to Current in a
Physically “Isolated” Conductor
Figure E3. Sniffer Probe Response with
Return Current in a Parallel Conductor
© 1997, Bruce Carsten Associates, Inc.
Figure E4. Typical EMI “Sniffer” Probe Frequency Response
Measured with 1.3m (51") of 50Ω Coax to Scope
Upper Traces: 1Meg Scope Input Impedance
Lower Traces: 50Ω Scope Input Impedance
an118fb
AN118-33
Application Note 118
PRINCIPLES OF PROBE USE
The Sniffer Probe is used with at least a 2-channel scope.
One channel is used to view the noise whose source is
to be located (which may also provide the scope trigger)
and the other channel is used for the Sniffer Probe. The
probe response nulls make it inadvisable to use this scope
channel for triggering.
A third scope trigger channel can be very useful, particularly if it is difficult to trigger on the noise. Transistor drive
waveforms (or their predecessors in the upstream logic)
are ideal for triggering; they are usually stable, and allow
immediate precursors of the noise to be viewed.
Start with the Probe at some distance from the circuit with
the Probe channel at maximum sensitivity. Move the probe
around the circuit, looking for “something happening” in
the circuit’s magnetic fields at the same time as the noise
problem. A precise “time domain” correlation between EMI
noise transients and internal circuit fields is fundamental
to the diagnostic approach.
As a candidate noise source is located, the Probe is moved
closer while the scope sensitivity is decreased to keep the
probe waveform on-screen. It should be possible to quickly
bring the probe down to the PC board trace (or wiring)
where the probe signal seems to be a maximum. This may
not be near the point of EMI generation, but it should be
near a PC trace or other conductor carrying the current
from the EMI source. This can be verified by moving the
probe back and forth in several directions; when the appropriate PC trace is crossed at roughly right angles, the
probe output will go through a sharp null over the trace,
with an evident phase reversal in probe voltage on each
side of the trace (as noted above).
This EMI “hot” trace can be followed (like a bloodhound
on the scent trail) to find all or much of the EMI generating current loop. If the trace is hidden on the back side
(or inside) of the board, mark its path with a felt pen and
locate the trace on disassembly, on another board or on
the artwork. From the current path and the timing of the
noise transient, the source of the problem usually becomes
almost self-evident.
Several not-uncommon problems (all of which have been
diagnosed with various versions of the Sniffer Probe) are
discussed here with suggested solutions or fixes.
TYPICAL DI/DT EMI PROBLEMS
Rectifier Reverse Recovery
Reverse recovery of rectifiers is the most common source
of dl/dt-related EMI in power converters; the charge
stored in P-N junction diodes during conduction causes
a momentary reverse current flow when the voltage reverses. This reverse current may stop very quickly (<1ns)
in diodes with a “snap” recovery (more likely in devices
with a PIV rating of less than 200V), or the reverse current
TYP. PROBE WAVEFORMS:
“SOFT”
RECOVERY
PROBE POINTS
“SNAP”
RECOVERY
Figure E5. Rectifier Reverse Recovery
Typical Fix: Tightly Coupled R-C Snubber
an118fb
AN118-34
Application Note 118
may decay more gradually with a “soft” recovery. Typical
Sniffer Probe waveforms for each type of recovery are
shown in Figure E5.
The “frequency halving” capacity is then connected in
series with the damping resistance and placed across the
diode, as tightly coupled as possible.
The sudden change in current creates a rapidly changing
magnetic field, which will both radiate external fields and
induce low impedance voltage spikes in other circuits. This
reverse recovery may “shock” parasitic L-C circuits into
ringing, which will result in oscillatory waveforms with
varying degrees of damping when the diode recovers. A
series R-C damper circuit in parallel with the diode is the
usual solution.
Snubber capacitors must have a high pulse current capability and low dielectric loss. Temperature stable (disc
or multilayer) ceramic, silvered mica and some plastic
filmfoil capacitors are suitable. Snubber resistors should
be noninductive; metal film, carbon film and carbon composition resistors are good, but wirewound resistors must
be avoided. The maximum snubber resistor dissipation
can be estimated from the product of the damper capacity,
switching frequency and the square of the peak snubber
capacitor voltage.
Output rectifiers generally carry the highest currents
and are thus the most prone to this problem, but this is
often recognized and they may be well-snubbed. It is not
uncommon for unsnubbed catch or clamp diodes to be
more of an EMI problem. (The fact that a diode in an RC-D snubber may need its own R-C snubber is not always
self-evident, for example).
The problem can usually be identified by placing the Sniffer
Probe near a rectifier lead. The signal will be strongest on
the inside of a lead bend in an axial package, or between the
anode and cathode leads in a TO-220, TO-247 or similar
type of package, as shown in Figure E5.
Using “softer” recovery diodes is a possible solution and
Schottky diodes are ideal in low voltage applications.
However, it must be recognized that a P-N diode with soft
recovery is also inherently lossy (while a “snap” recovery
is not), as the diode simultaneously develops a reverse
voltage while still conducting current: The fastest possible
diode (lowest recovered charge) with a moderately soft
recovery is usually the best choice. Sometimes a faster,
slightly “snappy” diode with a tightly coupled R-C snubber
works as well or better than a soft but excessively slow
recovery diode.
If significant ringing occurs, a “quick-and-dirty” R-C snubber design approach works fairly well: increasingly large
damper capacitors are placed across the diode until the
ringing frequency is halved. We know that the total ringing capacity is now quadrupled or that the original ringing
capacity is 1/3 of the added capacity. The damper resistance
required is about equal to the capacitive reactance of the
original ringing capacity at the original ringing frequency.
Snubbers on passive switches (diodes) or active switches
(transistors) should always be coupled as closely as
physically possible, with minimal loop inductance. This
minimizes the radiated field from the change in current
path from the switch to the snubber. It also minimizes
the turn-off voltage overshoot “required” to force the
current to change path through the switch-snubber loop
inductance.
Ringing in Clamp Zeners
A capacitor-to-capacitor ringing problem can occur when
a voltage clamping Zener or TransZorb® is placed across
the output of a converter for overvoltage protection (OVP).
Power Zeners have a large junction capacity, and this can
ring in series with the lead ESL and the output capacitors, with some of the ringing voltage showing up on the
output. This ringing current can be most easily detected
near the Zener leads, particularly on the inside of a bend
as shown in Figure E6.
R-C snubbers have not been found to work well in this
case as the ringing loop inductance is often as low or lower
than the obtainable parasitic inductance in the snubber.
Increasing the external loop inductance to allow damping
is not advisable as this would limit dynamic clamping
capability. In this case, it was found that a small ferrite
bead on one or both of the Zener leads dampened the
HF oscillations with minimal adverse side effects (a high
permeability ferrite bead quickly saturates as soon as the
Zener begins to conduct significant current).
an118fb
AN118-35
Application Note 118
PROBE POINTS
TYP. PROBE WAVEFORM:
100-500MHz
RINGING
I
Figure E6. Ringing Between Clamp Zener and Capacitor
Typical Fix: Small Ferrite Bead on Zener Lead(s)
+
Probe Point
(a)
IRING
Package
PCB Connections
Lead ESL
(b)
Rd
+
(c)
Rd
+
Figure E7. Ringing in Paralleled Dual Rectifiers
an118fb
AN118-36
Application Note 118
Paralleled Rectifiers
A less evident problem can occur when dual rectifier
diodes in a package are paralleled for increased current
capability, even with a tightly coupled R-C snubber. The
two diodes seldom recover at exactly the same time, which
can cause a very high frequency oscillation (hundreds of
MHz) to occur between the capacities of the two diodes
in series with the anode lead inductances, as shown in
Figure E7. This effect can really only be observed by
placing the probe between the two anode leads, as the
ringing current exists almost nowhere else (the ringing
is nearly “invisible” to a conventional voltage probe, like
many other EMI effects that can be easily found with a
magnetic field Sniffer Probe).
This “teeter-totter” oscillation has a voltage “null” about
where the R-C snubber is connected, so it provides little
or no damping (see Figure E7a). It is actually very difficult
to insert a suitable damping resistance into this circuit.
The easiest way to dampen the oscillation is to “slit” the
anode PC trace for an inch or so and place a damping
resistor at the anode leads as shown in Figure E7b. This
increases the inductance in series with the diode-diode
loop external to the package and leads, while having
minimal effect on the effective series inductance. Even
better damping is obtained by placing the resistor across
the anode leads at the entry point to the case, as shown
in Figure E7c, but this violates the mindset of many production engineers.
It is also preferable to split the original R-C damper into
two (2R) - (C/2) dampers, one on each side of the dual
Probe Point
rectifier (also shown in Figure E7c). In practice, it is always
preferable to use dual R-C dampers, one each side of the
diode; loop inductance is cut about in half, and the external
dl/dt field is reduced even further due to the oppositely
“handed” currents in the two snubber networks.
Paralleled Snubber or Damper Caps
A problem similar to that with the paralleled diodes occurs
when two or more low loss capacitors are paralleled and
driven with a sudden current change. There is a tendency
for a current to ring between the two capacitors in series
with their lead inductances (or ESL), as shown in Figure
E8a. This type of oscillation can usually be detected by
placing the Sniffer Probe between the leads of the paralleled capacitors. The ringing frequency is much lower than
with the paralleled diodes (due to the larger capacity), and
the effect may be benign if the capacitors are sufficiently
closer together.
If the resultant ringing is picked up externally, it can be
damped in a similar way as with the parallel diodes as
shown in Figure E8b. In either case, the dissipation in the
damping resistor tends to be relatively small.
Ringing in Transformer Shield Leads
The capacity of a transformer shield to other shields or
windings (CS in Figure E9) forms a series resonant circuit
with its “drain wire” inductance (LS) to the bypass point.
This resonant circuit is readily excited by typical square
wave voltages on windings, and a poorly damped oscillatory current may flow in the drain wire. The shield cur-
(a)
IRING
(b)
RD
Figure E8. Ringing in Paralleled “Snubber” Capacitors
an118fb
AN118-37
Application Note 118
VS
CS
CS
LS
LS
(1)
RD
SHIELD
PARASITICS
(2)
SHIELD RESONANCE DAMPING
10
SHIELD VOLTAGE FEEDTHROUGH
(1)
1
SHIELD RESONANCE
(2)
CAN BE DAMPED WITH
10 –1
A RESISTOR “RD” OR A
SMALL FERRITE BEAD:
10 –2
RD ≅
LS
CS
10 –3
10 –4
0.01
0.1
1
10
NORMALIZED FREQUENCY
F/Fres
Figure E9. Shield Effectiveness at High Frequencies is Limited
by Shield Capacity and Lead Inductance
an118fb
AN118-38
Application Note 118
rent may radiate noise into other circuits, and the shield
voltage will often show up as common mode conducted
noise. The shield voltage is very difficult to detect with a
voltage probe in most transformers, but the ringing shield
current can be observed by holding the Sniffer Probe near
the shield drain wire (Figure E10), or the shield current’s
return path in the circuit.
The first approach creates a “quadrupole” instead of a
dipole leakage field, which significantly reduces the distant
field intensity. It also reduces the eddy current losses in
any shorted strap electromagnetic shield used, which may
or may not be an important consideration.
This ringing can be dampened by placing a resistor RD in
series with the shield drain wire, whose value is approximately equal to the surge impedance of the resonant circuit,
which may be calculated from the formula in Figure E9.
External air gaps in an inductor, such as those in open
“bobbin core” inductors or with “E” cores spaced apart
(Figure E11b), can be a major source of external magnetic
fields when significant ripple or AC currents are present.
These fields can also be easily located with the Sniffer
Probe; response will be a maximum near an air gap or
near the end of an open inductor winding.
The shield capacitance (CS) can readily be measured with a
bridge (as the capacity from the shield to all facing shields
and/or windings), but LS is usually best calculated from
CS and the ringing frequency (as sensed by the Sniffer
Probe). This resistance is typically on the order of tens
of ohms.
One or more small ferrite beads can also be placed on
the drain wire instead to provide damping. This option
may be preferable as a late “fix” when the PC board has
already been laid out.
In either case, the damper losses are typically quite small.
The damper resistor has a moderately adverse impact on
shield effectiveness below the shield and drain wire resonant frequency; damper beads are superior in this respect
as their impedance is less at lower frequencies. The drain
wire connection should also be as short as possible to the
circuit bypass point, both to minimize EMI and to raise the
shield’s maximum effective (i.e., resonant) frequency.
Leakage Inductance Fields
Transformer leakage inductance fields emanate from
between primary and secondary windings. With a single
primary and secondary, a significant dipole field is created, which may be seen by placing the Sniffer Probe near
the winding ends as shown in Figure E11a. If this field is
generating EMI, there are two principal fixes:
1. Split the Primary or Secondary in two, to “sandwich”
the other winding, and/or:
2. Place a shorted copper strap “electromagnetic shield”
around the complete-core and winding assembly as
shown in Figue E12. Eddy currents in the shorted strap
largely cancel the external magnetic field.
External Air Gap Fields
“Open” inductor fields are not readily shielded and if they
present an EMI problem the inductor must usually be
redesigned to reduce external fields. The external field
around spaced E cores can be virtually eliminated by
placing all of the air gap in the center leg. Fields due to
a (possibly intentional) residual or minor outside air gap
can be minimized with the shorted strap electromagnetic
shield of Figure E12, if eddy current losses prove not to
be too high.
A less obvious problem may occur when inductors with
“open” cores are used as second stage filter chokes. The
minimal ripple current may not create a significant field,
but such an inductor can “pick up” external magnetic
fields and convert them to noise voltages or be an EMI
susceptibility problem.2
Poorly Bypassed High Speed Logic
Ideally, all high speed logic should have a tightly coupled
bypass capacitor for each IC and/or have power and ground
distribution planes in a multilayer PCB.
At the other extreme, I have seen one bypass capacitor
used at the power entrance to a logic board, with power
and ground led to the ICs from opposite sides of the board.
This created large spikes on the logic supply voltage and
produced significant electromagnetic fields around the
board.
Note 2. Ed Note. See Appendix D for additional commentary.
an118fb
AN118-39
Application Note 118
PROBE POINT (NEAR
SHIELD DRAIN PIN)
TYP. PROBE WAVEFORM:
10-100MHz
RINGING
Figure E10. Transformer Shield Ringing Typical Fix: 10Ω to 100Ω
Resistor (or Ferrite Bead in Drain Wire)
TRANSFORMER LEAKAGE
INDUCTANCE FIELD
INDUCTOR EXTERNAL
AIR GAP FIELD
PROBE
POINTS
(a)
TYPICAL FIXES:
SANDWICHED WINDINGS:
SHORTED STRAP SHIELD
(b)
TYPICAL FIX:
EXTERNAL AIR GAPS
Figure E11. Probe Voltages Resemble the Transformer
and Inductor Winding Waveforms
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AN118-40
Application Note 118
ELECTROMAGNETIC SHIELD
FORMED BY SHORTED COPPER
STRAP AROUND CORE AND WINDING
EM SHIELD ON AN INDUCTOR
WITH LARGE EXTERNAL CORE
AIR GAPS WILL HAVE HIGH
LOCALIZED EDDY CURRENT
LOSSES NEAR THE GAPS
Figure E12. A “Sandwiched” PRI-SEC Transformer Winding
Construction Reduces Electromagnetic Shield Eddy Current Losses
an118fb
AN118-41
Application Note 118
AC LINE OUTPUT
AC LINE INPUT
LINE IMPEDANCE
STABILIZATION
NETWORK
(LISN)
A
EQUIPMENT UNDER
EMI INVESTIGATION
NOISE
OUTPUT
(BNC)
EMI
“SNIFFER”
PROBE
LISN AC
LINE FILTER
INPUT
OUTPUT
(OPTIONAL)
SCOPE
B
CH 1
CH 2
EXT. TRIG.
CABLE LENGTH ≅ A + B
(OPTIONAL TRIGGER INPUT)
Figure E13. Using the Probe with a “LISN”
an118fb
AN118-42
Application Note 118
TO SCOPE
(50ΩTERM.)
2× SCALE
12.4Ω, 1/4W
M.F. RESISTOR
HI
COM
TO SIGNAL
GENERATOR
3/16" OD, 1/8" ID PLASTIC TUBE,
3/4" TO 1" LONG (AVAILABLE AT
MANY MODEL AND HOBBY SHOPS)
0.5"
20 TURNS #28 AWG WIRE WRAP WIRE
(OR SUB. #24 AWG MAGNET WIRE)
The Sniffer Probe Tip is centered inside the test coil where the Probe voltage
is greatest. The approximate flux density in the middle of a coil can be calculated
from the formula:
B = H = 1.257 N I / I
(CGS Units)
For the 1.27cm long, 20-turn test coil, the flux density is about 20 Gauss per amp.
At 1MHz, the Sniffer Probe voltage is 19mV P-P (±10%) per 100mA P-P for a 1M Ω
load impedance, and half that for a 50 Ω load.
Figure E14. EMI “Sniffer” Probe Test Coil
an118fb
AN118-43
Application Note 118
With a Sniffer Probe, I was able to show which pins of which
ICs had the larger current transients in synchronism with
the supply voltage transients. (The logic design engineers
were accusing the power supply vendor of creating the
noise. I found that the supplies were fairly quiet; it was
the poorly designed logic power distribution system that
was the problem.)
Probe Use with a “LISN”
A test setup using the Sniffer Probe with a Line Impedance
Stabilization Network (LISN) is shown in FIgure E13. The
optional “LISN AC LINE FILTER” reduces AC line voltage
feedthrough from a few 100mV to microvolt levels, simplifying EMI diagnosis when a suitable DC voltage source
is not available or cannot be used.
TESTING THE SNIFFER PROBE
The Sniffer Probe can be functionally tested with a jig
similar to that shown in Figure E14, which is used to test
probes in production.
CONCLUSION
The Sniffer Probe is a simple, but very fast and effective
means to locate dl/dt sources of EMI. These EMI sources
are very difficult to locate with conventional voltage or
current probes.
SUMMARY
A summarized procedure for using the EMI “Sniffer” Probe
appears in Figure E15.
1) Use a 2-channel scope, preferably one with an external trigger.
2) One scope channel is used for the Sniffer Probe, which is not to be used for
triggering.
3) The second channel is used to view the noise transient whose source is to be located,
which may also be used for triggering if practical.
4) More stable and reliable triggering is achieved with an “external trigger” (or a 3rd
channel) on a transistor drive waveform (or preceding logic transition), allowing
immediate precursors to the transient to be viewed. (Nearly all noise transients occur
during, or just after, a power transistor turn-on or turn-off.
5) Start with the Probe at some distance from the circuit with maximum sensitivity and
“sniff around” for something happening in precise sync with the noise transient. The
Probe waveform will not be identical to the noise transient, but will usually have a
strong resemblance.
6) Move the Probe closer to the suspected source while decreasing sensitivity. The
conductor carrying the responsible current is located by the sharp response null on
top of the conductor with inverted polarity on each side.
7) Trace out the noise current path as much as possible. Identify the current path on the
schematic.
8) The source of the noise transient is usually evident from the current path and the
timing information.
©1997, Bruce Carsten Associates, Inc.
Figure E15. EMI “Sniffer” Probe Procedure Outline
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AN118-44
Application Note 118
SNIFFER PROBE AMPLIFIER
probe’s uncalibrated, relative output means high frequency
termination aberrations are irrelevant. A simple film resistor, contained in the amplifier box, is adequate. Figure E17
shows the Sniffer Probe and the amplifier.
Figure E16 shows a 40MHz amplifier for the Sniffer Probe.
A gain of 200 allows an oscilloscope to display probe output
over a wide range of sensed inputs. The amplifier is built
into a small aluminum box. The probe should connect to
the amplifier via BNC cable, although the 50Ω termination does not have to be a high quality coaxial type. The
EMI “SNIFFER” PROBE
(SEE APPENDIX I
FOR DETAILS)
An alternate approach utilizes Appendix B’s (Figure B11)
HP-461A 50Ω amplifier.
15V
+
50Ω
A1
LT1223
–
+
1k
365Ω
A2
LT1223
+
–
1k
1µF
A3
LT1223
–
+
1k
365Ω
365Ω
USE RF LAYOUT TECHNIQUES.
SUPPLIES = ±15V, BYPASS EACH AMPLIFIER WITH
0.1µF CERAMIC CAPACITORS.
DIODE CLAMP SUPPLIES FOR REVERSE VOLTAGE
1k
50Ω
A4
LT1223
–
BNC OUTPUT
TO SCOPE
1k
–15V
365Ω
AN72 J16
Figure E16. 40MHz Amplifier for EMI Probe
an118fb
AN118-45
Figure E17. Sniffer Probe and Amplifier. Note All BNC-Based Signal
Transmission. ±15V Power Enters Box via Separate Cable
Application Note 118
an118fb
AN118-46
Application Note 118
APPENDIX F
About Ferrite Beads
60
0A
50
IMPEDENCE (Ω)
A ferrite bead enclosed conductor provides the highly desirable property of increasing impedance as frequency rises.
This effect is ideally suited to high frequency noise filtering of DC and low frequency signal carrying conductors.
The bead is essentially lossless within a linear regulator’s
passband. At higher frequencies the bead’s ferrite material
interacts with the conductors magnetic field, creating the
loss characteristic. Various ferrite materials and geometries
result in different loss factors versus frequency and power
level. Figure F1’s plot shows this. Impedance rises from
0.01Ω at DC to 50Ω at 100MHz. As DC current, and hence
constant magnetic field bias, rises, the ferrite becomes less
effective in offering loss. Note that beads can be “stacked”
in series along a conductor, proportionally increasing their
loss contribution. A wide variety of bead materials and
physical configurations are available to suit requirements
in standard and custom products.
0.1A
0.2A
40
0.5A
30
20
10
0
1
10
100
FREQUENCY (MHz)
DC = 0.01Ω
1000
AN118 F1
Figure F1. Impedance vs Frequency at Various DC Bias Currents
for a Surface Mounted Ferrite Bead (Fair-Rite 2518065007Y6).
Impedance is Essentially Zero at DC and Low Frequency, Rising
Above 50Ω Depending on Frequency and DC Current. Source:
Fair-Rite 2518065007Y6 Datasheet
APPENDIX G
Inductor Parasitics
Inductors can sometimes be used for high frequency
filtering instead of beads but parasitics must be kept in
mind. Advantages include wide availability and better effectiveness at lower frequencies, e.g., ≤100kHz. Figure G1
shows disadvantages are parasitic shunt capacitance
and potential susceptibility to stray switching regulator
radiation. Parasitic shunt capacitance allows unwanted
high frequency feedthrough. The inductors circuit board
position may allow stray magnetic fields to impinge its
PARASITIC
CAPACITANCE
USER
TERMINAL
STRAY
MAGNETIC
FIELD
winding, effectively turning it into a transformer secondary. The resulting observed spike and ripple related artifacts masquerade as conducted components, degrading
performance.
Figure G2 shows a form of inductance based filter constructed from PC board trace. Such extended length traces,
formed in spiral or serpentine patterns, look inductive at
high frequency. They can be surprisingly effective in some
circumstances, although introducing much less loss per
unit area than ferrite beads.
TERMINAL ACCESSABLE WITH PC VIA.
USER
TERMINAL
AN118 G2
AN118 G1
Figure G1. Some Parasitic Terms of an Inductor. Unwanted
Capacitance Permits High Frequency Feedthrough. Stray
Magnetic Field Induces Erroneous Inductor Current
Figure G2. Spiral and Serpentine PC Patterns are Sometimes
Used as High Frequency Filters, Although Less Effective Than
Ferrite Beads
an118fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN118-47
Application Note 118
an118fb
AN118-48
Linear Technology Corporation
LT 0615 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2008
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