LTC1628/LTC1628-PG - Linear Technology

LTC1628/LTC1628-PG - Linear Technology
LTC1628/LTC1628-PG
High Efficiency, 2-Phase
Synchronous Step-Down Switching Regulators
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FEATURES
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DESCRIPTIO
The LTC®1628/LTC1628-PG are high performance dual
step-down switching regulator controllers that drive all
N-channel synchronous power MOSFET stages. A constant frequency current mode architecture allows adjustment of the frequency up to 300kHz. Power loss and noise
due to the ESR of the input capacitors are minimized by
operating the two controller output stages out of phase.
Out-of-Phase Controllers Reduce Required Input
Capacitance and Power Supply Induced Noise
OPTI-LOOP® Compensation Minimizes COUT
±1% Output Voltage Accuracy
Dual N-Channel MOSFET Synchronous Drive
Power Good Output Voltage Monitor (LTC1628-PG)
DC Programmed Fixed Frequency 150kHz to 300kHz
Wide VIN Range: 3.5V to 36V Operation
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Soft-Start Current Ramping
Foldback Output Current Limiting
Latched Short-Circuit Shutdown with Defeat Option
Output Overvoltage Protection
Remote Output Voltage Sense
Low Shutdown IQ: 20µA
5V and 3.3V Standby Regulators
Selectable Constant Frequency or Burst Mode®
Operation
Available in 5mm × 5mm QFN and
28-Pin SSOP Packages
OPTI-LOOP compensation allows the transient response
to be optimized over a wide range of output capacitance and
ESR values. The precision 0.8V reference and power good
output indicator are compatible with future microprocessor generations, and a wide 3.5V to 30V (36V maximum)
input supply range encompasses all battery chemistries.
A RUN/SS pin for each controller provides both soft-start
and optional timed, short-circuit shutdown. Current
foldback limits MOSFET dissipation during short-circuit
conditions when overcurrent latchoff is disabled. Output
overvoltage protection circuitry latches on the bottom
MOSFET until VOUT returns to normal. The FCB mode pin
can select among Burst Mode, constant frequency mode
and continuous inductor current mode or regulate a
secondary winding. The LTC1628-PG includes a power
good output pin that replaces the FLTCPL, fault coupling
control pin of the LTC1628.
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APPLICATIO S
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Notebook and Palmtop Computers, PDAs
Battery Chargers
Portable Instruments
Battery-Operated Digital Devices
DC Power Distribution Systems
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TYPICAL APPLICATIO
, LTC and LT are registered trademarks of Linear Technology Corporation.
OPTI-LOOP and Burst Mode are registered trademarks of Linear Technology Corporation.
+
4.7µF
D3
VIN
M1
L1
6.3µH
CB1, 0.1µF
BOOST1
BG1
BOOST2
VOUT1
5V
5A
+
COUT1
47µF
6V
SP
R1
20k
1%
CC1
220pF
RC1
15k
L2
6.3µH
CB2, 0.1µF
M4
BG2
D2
PGND
SENSE1+
SENSE2 +
SENSE1–
VOSENSE1
ITH1
SENSE2 –
VOSENSE2
ITH2
1000pF
R2
105k
1%
VIN
5.2V TO 28V
CIN
22µF
50V
CERAMIC
SW2
LTC1628
SGND
RSENSE1
0.01Ω
M3
TG2
SW1
M2
D1
D4
INTVCC
TG1
1µF
CERAMIC
RSENSE2
0.01Ω
1000pF
RUN/SS1
CSS1
0.1µF
RUN/SS2
CSS2
0.1µF
CC2
220pF
RC2
15k
R3
20k
1%
M1, M2, M3, M4: FDS6680A
R4
63.4k
1%
VOUT2
3.3V
5A
COUT
56µF
6V
SP
+
1628 F01
Figure 1. High Efficiency Dual 5V/3.3V Step-Down Converter
1628fb
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LTC1628/LTC1628-PG
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ABSOLUTE
RATI GS (Note 1)
Input Supply Voltage (VIN).........................36V to – 0.3V
Top Side Driver Voltages
(BOOST1, BOOST2) ...................................42V to – 0.3V
Switch Voltage (SW1, SW2) .........................36V to – 5V
INTVCC, EXTVCC, RUN/SS1, RUN/SS2, (BOOST1-SW1),
(BOOST2-SW2), PGOOD .............................7V to – 0.3V
SENSE1+, SENSE2 +, SENSE1–,
SENSE2 – Voltages ........................ (1.1)INTVCC to – 0.3V
FREQSET, STBYMD, FCB,
FLTCPL Voltage ................................... INTVCC to – 0.3V
ITH1, ITH2, VOSENSE1, VOSENSE2 Voltages ...2.7V to – 0.3V
Peak Output Current <10µs (TG1, TG2, BG1, BG2) ... 3A
INTVCC Peak Output Current ................................ 50mA
Operating Temperature Range
LTC1628C/LTC1628C-PG ........................ 0°C to 85°C
LTC1628I/LTC1628I-PG ..................... – 40°C to 85°C
Junction Temperature (Note 2) ............................. 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)
(G Package Only) .................................................. 300°C
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PACKAGE/ORDER I FOR ATIO
4
25 BOOST1
FREQSET
5
24 VIN
STBYMD
6
23 BG1
FCB
7
22 EXTVCC
ITH1
8
21 INTVCC
SGND
9
20 PGND
3.3VOUT 10
LTC1628CUH
LTC1628IUH
32 31 30 29 28 27 26 25
VOSENSE1 1
24 BOOST1
FREQSET 2
23 VIN
STBYMD 3
22 BG1
FCB 4
21 EXTVCC
ITH1 5
20 INTVCC
SGND 6
19 PGND
3.3VOUT 7
19 BG2
18 BG2
UH PART
MARKING
17 BOOST2
ITH2 8
NC
SW2
15 RUN/SS2
TG2
SENSE2 + 14
RUN/SS2
16 TG2
SENSE2+
SENSE2 – 13
SENSE2–
9 10 11 12 13 14 15 16
17 SW2
NC
18 BOOST2
VOSENSE2 12
VOSENSE2
ITH2 11
SW1
VOSENSE1
LTC1628CG
LTC1628IG
LTC1628CG-PG
LTC1628IG-PG
TG1
26 SW1
FLTCPL
3
SENSE1
RUN/SS1
27 TG1
–
NC
28 FLTCPL*
2
SENSE1+
1
SENSE1 +
ORDER PART
NUMBER
TOP VIEW
SENSE1–
RUN/SS1
ORDER PART
NUMBER
NC
TOP VIEW
1628
1628I
UH PACKAGE
32-LEAD (5mm × 5mm) PLASTIC QFN
G PACKAGE
28-LEAD PLASTIC SSOP
θJA = 34°C/W
EXPOSED PAD IS SGND
(MUST BE SOLDERED TO PCB)
TJMAX = 125°C, θJA = 95°C/W
*PGOOD ON THE LTC1628-PG
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.792
0.800
0.808
V
Main Control Loops
VOSENSE1, 2
Regulated Feedback Voltage
(Note 3); ITH1, 2 Voltage = 1.2V
IVOSENSE1, 2
Feedback Current
(Note 3)
VREFLNREG
Reference Voltage Line Regulation
VIN = 3.6V to 30V (Note 3)
VLOADREG
Output Voltage Load Regulation
(Note 3)
Measured in Servo Loop; ∆ITH Voltage = 1.2V to 0.7V
Measured in Servo Loop; ∆ITH Voltage = 1.2V to 2.0V
●
●
●
–5
– 50
nA
0.002
0.02
%/V
0.1
– 0.1
0.5
– 0.5
%
%
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LTC1628/LTC1628-PG
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
gm1, 2
Transconductance Amplifier gm
ITH1, 2 = 1.2V; Sink/Source 5uA; (Note 3)
gmGBW1, 2
Transconductance Amplifier GBW
ITH1, 2 = 1.2V; (Note 3)
IQ
Input DC Supply Current
Normal Mode
Standby
Shutdown
(Note 4)
VIN = 15V; EXTVCC Tied to VOUT1; VOUT1 = 5V
VRUN/SS1, 2 = 0V, VSTBYMD > 2V
VRUN/SS1, 2 = 0V, VSTBYMD = Open;
MIN
TYP
MAX
1.3
UNITS
mmho
3
MHz
350
125
20
35
µA
µA
µA
0.76
0.800
0.84
V
– 0.30
– 0.18
– 0.1
µA
4.3
4.8
V
3.5
4
V
0.88
VFCB
Forced Continuous Threshold
IFCB
Forced Continuous Pin Current
VFCB = 0.85V
VBINHIBIT
Burst Inhibit (Constant Frequency)
Threshold
Measured at FCB pin
UVLO
Undervoltage Lockout
VIN Ramping Down
●
VOVL
Feedback Overvoltage Lockout
Measured at VOSENSE1, 2
●
0.84
0.86
ISENSE
Sense Pins Total Source Current
(Each Channel); VSENSE1–, 2 – = VSENSE1+, 2+ = 0V
– 85
– 60
VSTBYMD MS
Master Shutdown Threshold
VSTBYMD Ramping Down
0.4
VSTBYMD KA
Keep-Alive Power On-Threshold
VSTBYMD Ramping Up, RUNSS1, 2 = 0V
DFMAX
Maximum Duty Factor
In Dropout
99.4
%
IFLTCPL
VFLTCPL Input Current
LTC1628 Only
0.5V > VFLTCPL
INTVCC – 0.5V < VFLTCPL < INTVCC
–3
3
µA
µA
VFLTCPL
Fault Coupling Threshold;
LTC1628 Only
For FCB Signal and Individual Overcurrent
Faults to Affect Both Controllers
2
V
IRUN/SS1, 2
Soft-Start Charge Current
VRUN/SS1, 2 = 1.9V
0.5
1.2
µA
VRUN/SS1, VRUN/SS2 Rising
1.0
1.5
1.9
V
4.1
4.5
V
2
4
µA
VRUN/SS1, 2 ON RUN/SS Pin ON Threshold
●
98
VRUN/SS1, 2 LT RUN/SS Pin Latchoff Arming Threshold VRUN/SS1, VRUN/SS2 Rising from 3V
ISCL1, 2
RUN/SS Discharge Current
Soft Short Condition VOSENSE1, 2 = 0.5V;
VRUN/SS1, 2 = 4.5V
0.5
5
µA
88
85
mV
mV
(Note 5)
CLOAD = 3300pF
CLOAD = 3300pF
50
50
90
90
ns
ns
BG Transition Time:
Rise Time
Fall Time
(Note 5)
CLOAD = 3300pF
CLOAD = 3300pF
40
40
90
80
ns
ns
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
90
ns
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
90
ns
Minimum On-Time
Tested with a Square Wave (Note 6)
180
ns
VOSENSE1, 2 = 0.5V
Maximum Current Sense Threshold
VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V
●
VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V, LTC1628 Only
TG1, 2 tr
TG1, 2 tf
TG Transition Time:
Rise Time
Fall Time
BG1, 2 tr
BG1, 2 tf
tON(MIN)
V
75
75
Shutdown Latch Disable Current
VSENSE(MAX)
BG/TG t2D
V
2
1.6
ISDLHO
TG/BG t1D
0.6
1.5
V
µA
62
65
INTVCC Linear Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V, VEXTVCC = 4V
5.0
5.2
V
VLDO INT
INTVCC Load Regulation
ICC = 0 to 20mA, VEXTVCC = 4V
4.8
0.2
1.0
%
VLDO EXT
EXTVCC Voltage Drop
ICC = 20mA, VEXTVCC = 5V, LTC1628
120
240
mV
VLDO EXT-PG
EXTVCC Voltage Drop
ICC = 20mA, VEXTVCC = 5V, LTC1628-PG
80
160
mV
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, EXTVCC Ramping Positive
VLDOHYS
EXTVCC Hysteresis
●
4.5
4.7
V
0.2
V
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LTC1628/LTC1628-PG
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Oscillator
fOSC
Oscillator frequency
VFREQSET = Open (Note 7)
190
220
250
kHz
fLOW
Lowest Frequency
VFREQSET = 0V
120
140
160
kHz
fHIGH
Highest Frequency
VFREQSET = 2.4V
280
310
360
kHz
IFREQSET
FREQSET Input Current
VFREQSET = 0V
–2
–1
µA
3.3V Linear Regulator
V3.3OUT
3.3V Regulator Output Voltage
No Load
3.35
3.45
V
V3.3IL
3.3V Regulator Load Regulation
I3.3 = 0 to 10mA
0.5
2
%
V3.3VL
3.3V Regulator Line Regulation
6V < VIN < 30V
0.05
0.2
%
0.1
0.3
V
±1
µA
– 9.5
9.5
%
%
●
3.25
PGOOD Output (LTC1628-PG Only)
VPGL
PGOOD Voltage Low
IPGOOD = 2mA
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
VPG
PGOOD Trip Level, Either Controller
VOSENSE Respect to Set Output Voltage
VOSENSE Ramping Negative
VOSENSE Ramping Positive
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC1628G/LTC1628G-PG: TJ = TA + (PD • 95°C/W)
LTC1628CUH: TJ = TA + (PD • 34°C/W)
Note 3: The LTC1628/LTC1628-PG are tested in a feedback loop that
servos VITH1, 2 to a specified voltage and measures the resultant
VOSENSE1, 2.
–6
6
–7.5
7.5
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 5: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 6: The minimum on-time condition is specified for an inductor
peak-to-peak ripple current ≥ 40% of IMAX (see minimum on-time
considerations in the Applications Information section).
Note 7: VFREQSET pin internally tied to 1.19V reference through a large
resistance.
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Output Current
(Figure 13)
Efficiency vs Output Current
and Mode (Figure 13)
100
100
Burst Mode
OPERATION
70
60
50
40
30
FORCED
CONTINUOUS
MODE
CONSTANT
FREQUENCY
(BURST DISABLE)
20
80
0.1
0.01
1
OUTPUT CURRENT (A)
10
VIN = 20V
70
1628 G01
50
0.001
VOUT = 5V
IOUT = 3A
90
VIN = 10V
VIN = 15V
80
70
60
60
VIN = 15V
VOUT = 5V
10
0
0.001
100
VIN = 7V
90
EFFICIENCY (%)
EFFICIENCY (%)
80
VIN = 15V
VOUT = 5V
EFFICIENCY (%)
90
Efficiency vs Input Voltage
(Figure 13)
0.1
0.01
1
OUTPUT CURRENT (A)
50
10
1628 G02
5
25
15
INPUT VOLTAGE (V)
35
1628 G03
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LTC1628/LTC1628-PG
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TYPICAL PERFOR A CE CHARACTERISTICS
Supply Current vs Input Voltage
and Mode (Figure 13)
1000
600
BOTH
CONTROLLERS ON
400
200
STANDBY
5.05
INTVCC AND EXTVCC SWITCH VOLTAGE (V)
EXTVCC VOLTAGE DROP (mV)
250
800
SUPPLY CURRENT (µA)
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Voltage Drop
200
150
100
50
SHUTDOWN
0
0
0
5
20
15
10
25
INPUT VOLTAGE (V)
30
35
0
10
30
20
CURRENT (mA)
40
4.85
4.80
EXTVCC SWITCHOVER THRESHOLD
4.75
4.70
– 50 – 25
50
50
25
75
0
TEMPERATURE (°C)
100
125
1628 G06
Maximum Current Sense Threshold
vs Percent of Nominal Output
Voltage (Foldback)
75
80
ILOAD = 1mA
70
5.0
60
4.8
4.7
50
VSENSE (mV)
4.9
VSENSE (mV)
INTVCC VOLTAGE (V)
4.90
Maximum Current Sense Threshold
vs Duty Factor
Internal 5V LDO Line Reg
25
4.6
50
40
30
20
4.5
10
0
4.4
0
5
20
15
25
10
INPUT VOLTAGE (V)
30
0
35
20
40
60
DUTY FACTOR (%)
80
50
100
0
25
75
PERCENT ON NOMINAL OUTPUT VOLTAGE (%)
1628 G09
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
Maximum Current Sense Threshold
vs VRUN/SS (Soft-Start)
80
0
100
1628 G08
1628 G07
Current Sense Threshold
vs ITH Voltage
90
80
VSENSE(CM) = 1.6V
80
70
76
40
60
VSENSE (mV)
VSENSE (mV)
60
VSENSE (mV)
4.95
1628 G05
1628 G04
5.1
INTVCC VOLTAGE
5.00
72
68
50
40
30
20
10
20
0
64
–10
–20
0
0
1
2
3
4
5
6
VRUN/SS (V)
1628 G10
60
0
1
3
4
2
COMMON MODE VOLTAGE (V)
5
1628 G11
–30
0
0.5
1
1.5
VITH (V)
2
2.5
1628 G12
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LTC1628/LTC1628-PG
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TYPICAL PERFOR A CE CHARACTERISTICS
Load Regulation
FCB = 0V
VIN = 15V
FIGURE 1
SENSE Pins Total Source Current
100
VOSENSE = 0.7V
2.0
–0.2
50
ISENSE (µA)
–0.1
VITH (V)
NORMALIZED VOUT (%)
VITH vs VRUN/SS
2.5
0.0
1.5
1.0
–0.3
0
–50
0.5
–0.4
0
1
0
3
2
LOAD CURRENT (A)
4
5
0
2
1
3
4
5
–100
6
2
0
VRUN/SS (V)
4
1628 G14
1628 G13
Maximum Current Sense
Threshold vs Temperature
1628 G15
Dropout Voltage vs Output Current
(Figure 13)
80
4
6
VSENSE COMMON MODE VOLTAGE (V)
RUN/SS Current vs Temperature
1.8
VOUT = 5V
1.6
76
74
3
RUN/SS CURRENT (µA)
DROPOUT VOLTAGE (V)
VSENSE (mV)
78
2
RSENSE = 0.015Ω
1
72
1.4
1.2
1.0
0.8
0.6
0.4
RSENSE = 0.010Ω
0.2
70
–50 –25
0
50
25
0
75
TEMPERATURE (°C)
100
125
0
0.5
1.0 1.5 2.0 2.5 3.0
OUTPUT CURRENT (A)
3.5
4.0
0
–50
0
25
50
75
TEMPERATURE (°C)
1628 G18
1628 G17
Soft-Start Up (Figure 13)
100
125
1628 G25
Load Step (Figure 13)
VOUT
5V/DIV
–25
Load Step (Figure 13)
VOUT
200mV/DIV
VOUT
200mV/DIV
IOUT
2A/DIV
IOUT
2A/DIV
VRUN/SS
5V/DIV
IOUT
2A/DIV
VIN = 15V
VOUT = 5V
5ms/DIV
1628 G19
VIN = 15V
20µs/DIV
VOUT = 5V
LOAD STEP = 0A TO 3A
Burst Mode OPERATION
1628 G20
20µs/DIV
VIN = 15V
VOUT = 5V
LOAD STEP = 0A TO 3A
CONTINUOUS MODE
1628 G21
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LTC1628/LTC1628-PG
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TYPICAL PERFOR A CE CHARACTERISTICS
Input Source/Capacitor
Instantaneous Current (Figure 13)
IIN
2A/DIV
VIN
200mV/DIV
Constant Frequency (Burst Inhibit)
Operation (Figure 13)
Burst Mode Operation (Figure 13)
VOUT
20mV/DIV
VOUT
20mV/DIV
VSW1
10V/DIV
VSW2
10V/DIV
IOUT
0.5A/DIV
VIN = 15V
1µs/DIV
VOUT = 5V
IOUT5 = IOUT3.3 = 2A
VIN = 15V
VOUT = 5V
VFCB = OPEN
IOUT = 20mA
1628 G22
Current Sense Pin Input Current
vs Temperature
10µs/DIV
VIN = 15V
VOUT = 5V
VFCB = 5V
IOUT = 20mA
1628 G23
EXTVCC Switch Resistance
vs Temperature
35
10
1628 G24
350
31
29
27
50
25
0
75
TEMPERATURE (°C)
100
125
300
8
FREQUENCY (kHz)
EXTVCC SWITCH RESISTANCE (Ω)
VFREQSET = 5V
33
6
4
VFREQSET = OPEN
200
VFREQSET = 0V
150
100
50
0
–50 –25
50
25
0
75
TEMPERATURE (°C)
100
1628 G26
125
0
– 50 – 25
50
25
75
0
TEMPERATURE (°C)
1628 G27
Undervoltage Lockout
vs Temperature
100
125
1628 G28
Shutdown Latch Thresholds
vs Temperature
4.5
3.50
3.45
3.40
3.35
3.30
3.25
3.20
–50 –25
250
2
SHUTDOWN LATCH THRESHOLDS (V)
25
–50 –25
2µs/DIV
Oscillator Frequency
vs Temperature
VOUT = 5V
UNDERVOLTAGE LOCKOUT (V)
CURRENT SENSE INPUT CURRENT (µA)
IOUT
0.5A/DIV
50
25
75
0
TEMPERATURE (°C)
100
125
1628 G29
LATCH ARMING
4.0
3.5
3.0
LATCHOFF
THRESHOLD
2.5
2.0
1.5
1.0
0.5
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
1628 G30
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LTC1628/LTC1628-PG
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PI FU CTIO S
RUN/SS1, RUN/SS2: Combination of soft-start, run control inputs and short-circuit detection timers. A capacitor
to ground at each of these pins sets the ramp time to full
output current. Forcing either of these pins back below
1.0V causes the IC to shut down the circuitry required for
that particular controller. Latchoff overcurrent protection
is also invoked via this pin as described in the Applications
Information section.
SENSE1+, SENSE2+: The (+) Input to the Differential
Current Comparators. The Ith pin voltage and controlled
offsets between the SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip threshold.
SENSE1–, SENSE2–: The (–) Input to the Differential
Current Comparators.
VOSENSE1, VOSENSE2: Receives the remotely-sensed feedback voltage for each controller from an external resistive
divider across the output.
FREQSET: Frequency Control Input to the Oscillator. This
pin can be left open, tied to ground, tied to INTVCC or driven
by an external voltage source. This pin can also be used
with an external phase detector to build a true phaselocked loop.
STBYMD: Control pin that determines which circuitry remains active when the controllers are shut down and/or
provides a common control point to shut down both controllers. See the Operation section for details.
FCB: Forced Continuous Control Input. This input acts on
the first controller (or both controllers depending upon
the FLTCPL pin—see pin description), and is
normally used to regulate a secondary winding. Pulling
this pin below 0.8V will force continuous synchronous
operation for the first and optionally the second controller. Do not leave this pin floating.
ITH1, ITH2: Error Amplifier Output and Switching Regulator
Compensation Point. Each associated channels’ current
comparator trip point increases with this control voltage.
SGND: Small Signal Ground common to both controllers, must be routed separately from high current
grounds to the common (–) terminals of the COUT
capacitors.
3.3VOUT: Output of a linear regulator capable of supplying
10mA DC with peak currents as high as 50mA.
PGND: Driver Power Ground. Connects to the sources of
bottom (synchronous) N-channel MOSFETs, anodes of the
Schottky rectifiers and the (–) terminal(s) of CIN.
INTVCC: Output of the Internal 5V Linear Low Dropout
Regulator and the EXTVCC Switch. The driver and control
circuits are powered from this voltage source. Must be
decoupled to power ground with a minimum of 4.7µF tantalum or other low ESR capacitor. The INTVCC regulator
standby function is determined by the STBYMD pin.
EXTVCC: External Power Input to an Internal Switch Connected to INTVCC. This switch closes and supplies VCC
power, bypassing the internal low dropout regulator, whenever EXTVCC is higher than 4.7V. See EXTVCC connection
in Applications section. Do not exceed 7V on this pin.
BG1, BG2: High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing at these pins is
from ground to INTVCC.
VIN: Main Supply Pin. A bypass capacitor should be tied
between this pin and the signal ground pin.
BOOST1, BOOST2: Bootstrapped Supplies to the Top Side
Floating Drivers. Capacitors are connected between the
boost and switch pins and Schottky diodes are tied between the boost and INTVCC pins. Voltage swing at the
boost pins is from INTVCC to (VIN + INTVCC).
SW1, SW2: Switch Node Connections to Inductors. Voltage swing at these pins is from a Schottky diode (external)
voltage drop below ground to VIN.
TG1, TG2: High Current Gate Drives for Top N-Channel
MOSFETs. These are the outputs of floating drivers with a
voltage swing equal to INTVCC – 0.5V superimposed on
the switch node voltage SW.
FLTCPL: (LTC1628 Only) Fault Coupling Control Pin that
determines if fault/normal conditions on one controller
will act on the other controller. FLTCPL = INTVCC to couple
channels; FLTCPL = 0V to decouple.
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PGOOD: (LTC1628-PG Only) Open-Drain Logic Output.
PGOOD is pulled to ground when the voltage on either
VOSENSE pin is not within ±7.5% of its set point.
NC: These “No Connect” pins are not tied internally to
anything. On the PC layout, these pin landings should be
connected to the SGND plane under the IC.
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VIN
INTVCC
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
1.19V
BOOST
1M
FREQSET
DROP
OUT
DET
CLK1
OSCILLATOR
CLK2
FLTCPL
Q
R
Q
BOT
BOT
–
+
INTVCC
BINH
BG
COUT
PGND
B
+
4.5V
CIN
SW
SWITCH
LOGIC
+
+
D1
TOP ON
0.55V
CB
FCB
RUN/SS2
3V
0.18µA
R6
S
TG
TOP
RUN/SS1
MERGE LOGIC
VSEC
DB
–
SHDN
VOUT
RSENSE
FCB
+
–
FCB
I1
+
–
3.3VOUT
+
0.8V
VREF
–
++
SLOPE
COMP
VIN
+
30k SENSE
+
45k
EXTVCC
+
–
–
EA
+
5V
LDO
REG
OV
INTVCC
STBYMD
INTERNAL
SUPPLY
6V
VOSENSE
0.80V
SHDN
RST
4(VFB)
R2
R1
0.86V
ITH
1.2µA
SGND
VFB
+
–
+
CSEC
45k
2.4V
4.8V
DSEC
–
30k SENSE
VIN
5V
INTVCC
I2
–
3mV
0.86V
4(VFB)
–
–
+
R5
RUN
SOFT
START
CC
CC2
RC
RUN/SS
CSS
1628 FD/F02
Figure 2
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LTC1628/LTC1628-PG
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OPERATIO (Refer to Functional Diagram)
Main Control Loop
The LTC1628 uses a constant frequency, current mode
step-down architecture with the two controller channels
operating 180 degrees out of phase. During normal operation, each top MOSFET is turned on when the clock for that
channel sets the RS latch, and turned off when the main
current comparator, I1, resets the RS latch. The peak
inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH pin, which is the output of
each error amplifier EA. The VOSENSE pin receives the
voltage feedback signal, which is compared to the internal
reference voltage by the EA. When the load current increases, it causes a slight decrease in VOSENSE relative to
the 0.8V reference, which in turn causes the ITH voltage to
increase until the average inductor current matches the
new load current. After the top MOSFET has turned off, the
bottom MOSFET is turned on until either the inductor
current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle.
The top MOSFET drivers are biased from floating bootstrap capacitor CB, which normally is recharged during
each off cycle through an external diode when the top
MOSFET turns off. As VIN decreases to a voltage close to
VOUT, the loop may enter dropout and attempt to turn on
the top MOSFET continuously. The dropout detector detects this and forces the top MOSFET off for about 500ns
every tenth cycle to allow CB to recharge.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 1.2µA
current source to charge soft-start capacitor CSS. When
CSS reaches 1.5V, the main control loop is enabled with the
ITH voltage clamped at approximately 30% of its maximum
value. As CSS continues to charge, the ITH pin voltage is
gradually released allowing normal, full-current operation. When both RUN/SS1 and RUN/SS2 are low, all
LTC1628 controller functions are shut down, and the
STBYMD pin determines if the standby 5V and 3.3V
regulators are kept alive.
Low Current Operation
The FCB pin is a multifunction pin providing two functions: 1) to provide regulation for a secondary winding by
temporarily forcing continuous PWM operation on
controller 1 (or both controllers depending upon the
FLTCPL pin); and 2) select between two modes of low
current operation. When the FCB pin voltage is below
0.800V, the controller forces continuous PWM current
mode operation. In this mode, the top and bottom
MOSFETs are alternately turned on to maintain the output
voltage independent of direction of inductor current.
When the FCB pin is below VINTVCC␣ –␣ 2V but greater than
0.80V, the controller enters Burst Mode operation. Burst
Mode operation sets a minimum output current level
before inhibiting the top switch and turns off the synchronous MOSFET(s) when the inductor current goes negative. This combination of requirements will, at low currents, force the ITH pin below a voltage threshold that will
temporarily inhibit turn-on of both output MOSFETs until
the output voltage drops. There is 60mV of hysteresis in
the burst comparator B tied to the ITH pin. This hysteresis
produces output signals to the MOSFETs that turn them
on for several cycles, followed by a variable “sleep”
interval depending upon the load current. The resultant
output voltage ripple is held to a very small value by
having the hysteretic comparator after the error amplifier
gain block.
Constant Frequency Operation
When the FCB pin is tied to INTVCC, Burst Mode operation
is disabled and the forced minimum output current requirement is removed. This provides constant frequency,
discontinuous (preventing reverse inductor current) current operation over the widest possible output current
range. This constant frequency operation is not as efficient
as Burst Mode operation, but does provide a lower noise,
constant frequency operating mode down to approximately 1% of designed maximum output current.
Continuous Current (PWM) Operation
Tying the FCB pin to ground will force continuous current
operation. This is the least efficient operating mode, but
may be desirable in certain applications. The output can
source or sink current in this mode. When sinking current
while in forced continuous operation, current will be
forced back into the main power supply potentially boosting the input supply to dangerous voltage levels—
BEWARE!
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OPERATIO (Refer to Functional Diagram)
Frequency Setting
Fault Coupling Pin
The FREQSET pin provides frequency adjustment of the
internal oscillator from approximately 140kHz to 310kHz.
This input is nominally biased through an internal resistor
to the 1.19V reference, setting the oscillator frequency to
approximately 220kHz. This pin can be driven from an
external AC or DC signal source to control the instantaneous frequency of the oscillator.
The FLTCPL pin (LTC1628 only) controls two functions
that can operate individually (FLTCPL = 0V) or unilaterally
(FLTCPL = INTVCC) between the two controllers. When the
FLTCPL pin is grounded (internally tied default mode for
the LTC1628-PG), 1) the FCB input forces continuous
operation only on the first controller when the applied
voltage drops below 0.8V and 2) the short-circuit latchoff
function only latches off the controller having the shorted
output. When the FLTCPL pin is tied to INTVCC, 1) the FCB
input forces continuous operation on both controllers
when the applied voltage drops below 0.8V and 2) the
short-circuit latchoff function latches off both controllers
when either has a shorted output.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin.
When the EXTVCC pin is left open, an internal 5V low
dropout linear regulator supplies INTVCC power. If EXTVCC
is taken above 4.7V, the 5V regulator is turned off and an
internal switch is turned on connecting EXTVCC to INTVCC.
This allows the INTVCC power to be derived from a high
efficiency external source such as the output of the regulator itself or a secondary winding, as described in the
Applications Information.
Standby Mode Pin
The STBYMD pin is a three-state input that controls
common circuitry within the IC as follows: When the
STBYMD pin is held at ground, both controller RUN/SS
pins are pulled to ground providing a single control pin to
shut down both controllers. When the pin is left open, the
internal RUN/SS currents are enabled to charge the
RUN/SS capacitor(s), allowing the turn-on of either controller and activating necessary common internal biasing.
When the STBYMD pin is taken above 2V, both internal
linear regulators are turned on independent of the state on
the RUN/SS pins of the two switching regulator controllers, providing an output power source for “wake-up”
circuitry. Decouple the pin with a small capacitor (0.01µF)
to ground if the pin is not connected to a DC potential.
Output Overvoltage Protection
An overvoltage comparator, OV, guards against transient
overshoots (>7.5%) as well as other more serious conditions that may overvoltage the output. In this case, the top
MOSFET is turned off and the bottom MOSFET is turned on
until the overvoltage condition is cleared.
Power Good (PGOOD) Pin
The PGOOD pin (LTC1628-PG only) is connected to an
open drain of an internal MOSFET. The MOSFET turns on
and pulls the pin low when both the outputs are not within
±7.5% of their nominal output levels as determined by
their resistive feedback dividers. When both outputs meet
the ±7.5% requirement, the MOSFET is turned off within
10µs and the pin is allowed to be pulled up by an external
resistor to a source of up to 7V.
Foldback Current, Short-Circuit Detection
and Short-Circuit Latchoff
The RUN/SS capacitors are used initially to limit the inrush
current of each switching regulator. After the controller
has been started and been given adequate time to charge
up the output capacitors and provide full load current, the
RUN/SS capacitor is used in a short-circuit time-out
circuit. If the output voltage falls to less than 70% of its
nominal output voltage, the RUN/SS capacitor begins
discharging on the assumption that the output is in an
overcurrent and/or short-circuit condition. If the condition
lasts for a long enough period as determined by the size of
the RUN/SS capacitor, the controller (or both controllers
as determined by the FLTCPL pin, LTC1628 only) will be
shut down until the RUN/SS pin(s) voltage(s) are recycled.
This built-in latchoff can be overridden by providing a
>5µA pull-up at a compliance of 5V to the RUN/SS pin(s).
This current shortens the soft start period but also prevents net discharge of the RUN/SS capacitor(s) during an
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OPERATIO (Refer to Functional Diagram)
overcurrent and/or short-circuit condition. Foldback current limiting is also activated when the output voltage falls
below 70% of its nominal level whether or not the shortcircuit latchoff circuit is enabled. Even if a short is present
and the short-circuit latchoff is not enabled, a safe, low
output current is provided due to internal current foldback
and actual power wasted is low due to the efficient nature
of the current mode switching regulator.
With 2-phase operation, the two channels of the dualswitching regulator are operated 180 degrees out of
phase. This effectively interleaves the current pulses drawn
by the switches, greatly reducing the overlap time where
they add together. The result is a significant reduction in
total RMS input current, which in turn allows less expensive input capacitors to be used, reduces shielding requirements for EMI and improves real world operating
efficiency.
THEORY AND BENEFITS OF 2-PHASE OPERATION
Figure 3 compares the input waveforms for a representative single-phase dual switching regulator to the new
LTC1628 2-phase dual switching regulator. An actual
measurement of the RMS input current under these conditions shows that 2-phase operation dropped the input
current from 2.53ARMS to 1.55ARMS. While this is an
impressive reduction in itself, remember that the power
losses are proportional to IRMS2, meaning that the actual
power wasted is reduced by a factor of 2.66. The reduced
input ripple voltage also means less power is lost in the
input power path, which could include batteries, switches,
trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also
directly accrue as a result of the reduced RMS input
current and voltage.
The LTC1628 dual high efficiency DC/DC controller brings
the considerable benefits of 2-phase operation to portable
applications for the first time. Notebook computers, PDAs,
handheld terminals and automotive electronics will all
benefit from the lower input filtering requirement, reduced
electromagnetic interference (EMI) and increased efficiency associated with 2-phase operation.
Why the need for 2-phase operation? Up until the LTC1628,
constant-frequency dual switching regulators operated
both channels in phase (i.e., single-phase operation). This
means that both switches turned on at the same time,
causing current pulses of up to twice the amplitude of
those for one regulator to be drawn from the input capacitor and battery. These large amplitude current pulses
increased the total RMS current flowing from the input
capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.
Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 4 shows how
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
IIN(MEAS) = 2.53ARMS
(a)
DC236 F03a
IIN(MEAS) = 1.55ARMS
DC236 F03b
(b)
Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation
for Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The
Reduced Input Ripple with the LTC1628 2-Phase Regulator Allows Less Expensive
Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
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OPERATIO (Refer to Functional Diagram)
3.0
2.5
INPUT RMS CURRENT (A)
the input capacitor requirement to that for just one channel
operating at maximum current and 50% duty cycle.
SINGLE PHASE
DUAL CONTROLLER
the RMS input current varies for single-phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
A final question: If 2-phase operation offers such an
advantage over single-phase operation for dual switching
regulators, why hasn’t it been done before? The answer is
that, while simple in concept, it is hard to implement.
Constant-frequency current mode switching regulators
require an oscillator derived “slope compensation” signal
to allow stable operation of each regulator at over 50%
duty cycle. This signal is relatively easy to derive in singlephase dual switching regulators, but required the development of a new and proprietary technique to allow 2-phase
operation. In addition, isolation between the two channels
becomes more critical with 2-phase operation because
switch transitions in one channel could potentially disrupt
the operation of the other channel.
It can readily be seen that the advantages of 2-phase
operation are not just limited to a narrow operating range,
but in fact extend over a wide region. A good rule of thumb
for most applications is that 2-phase operation will reduce
The LTC1628 is proof that these hurdles have been surmounted. The new device offers unique advantages for the
ever-expanding number of high efficiency power supplies
required in portable electronics.
2.0
1.5
2-PHASE
DUAL CONTROLLER
1.0
0.5
0
VO1 = 5V/3A
VO2 = 3.3V/3A
0
10
20
30
INPUT VOLTAGE (V)
40
1628 F04
Figure 4. RMS Input Current Comparison
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APPLICATIO S I FOR ATIO
Figure 1 on the first page is a basic LTC1628 application
circuit. External component selection is driven by the
load requirement, and begins with the selection of RSENSE
and the inductor value. Next, the power MOSFETs and D1
are selected. Finally, CIN and COUT are selected. The
circuit shown in Figure 1 can be configured for operation
up to an input voltage of 28V (limited by the external
MOSFETs).
RSENSE Selection For Output Current
RSENSE is chosen based on the required output current.
The LTC1628 current comparator has a maximum threshold of 75mV/RSENSE and an input common mode range of
SGND to 1.1(INTVCC). The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current, ∆IL.
Allowing a margin for variations in the LTC1628 and
external component values yields:
RSENSE =
50mV
IMAX
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
the internal compensation required to meet stability criterion for buck regulators operating at greater than 50%
duty factor. A curve is provided to estimate this reducton
in peak output current level depending upon the operating
duty factor.
Selection of Operating Frequency
The LTC1628 uses a constant frequency architecture with
the frequency determined by an internal oscillator capacitor. This internal capacitor is charged by a fixed current
plus an additional current that is proportional to the
voltage applied to the FREQSET pin.
A graph for the voltage applied to the FREQSET pin vs
frequency is given in Figure 5. As the operating frequency
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FREQSET PIN VOLTAGE (V)
2.5
25% of the current limit determined by RSENSE. Lower
inductor values (higher ∆IL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
2.0
1.5
1.0
Inductor Core Selection
0.5
0
120
170
220
270
OPERATING FREQUENCY (kHz)
320
1628 F05
Figure 5. FREQSET Pin Voltage vs Frequency
is increased the gate charge losses will be higher, reducing
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 310kHz.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN:
∆IL =
 V

1
VOUT  1 – OUT 
( f)(L)
VIN 

Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron
cores, forcing the use of more expensive ferrite,
molypermalloy, or Kool Mµ® cores. Actual core loss is
independent of core size for a fixed inductor value, but it
is very dependent on inductance selected. As inductance
increases, core losses go down. Unfortunately, increased
inductance requires more turns of wire and therefore
copper losses will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more
difficult. However, designs for surface mount are available
that do not increase the height significantly.
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL=0.3(IMAX). Remember, the
maximum ∆IL occurs at the maximum input voltage.
Power MOSFET and D1 Selection
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average
inductor current required results in a peak current below
The peak-to-peak drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up (see
Two external power MOSFETs must be selected for each
controller with the LTC1628: One N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
Kool Mµ is a registered trademark of Magnetics, Inc.
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EXTVCC Pin Connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
The only exception is if low input voltage is expected
(VIN < 5V); then, sub-logic level threshold MOSFETs
(VGS(TH) < 3V) should be used. Pay close attention to the
BVDSS specification for the MOSFETs as well; most of the
logic level MOSFETs are limited to 30V or less.
The term (1+δ) is generally given for a MOSFET in the form
of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOSFET characteristics. The constant k = 1.7 can be used to
estimate the contributions of the two terms in the main
switch dissipation equation.
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1628 is operating in continuous mode the duty cycles
for the top and bottom MOSFETs are given by:
The Schottky diode D1 shown in Figure 1 conducts during
the dead-time between the conduction of the two power
MOSFETs. This prevents the body diode of the bottom
MOSFET from turning on, storing charge during the deadtime and requiring a reverse recovery period that could
cost as much as 3% in efficiency at high VIN. A 1A to 3A
Schottky is generally a good compromise for both regions
of operation due to the relatively small average current.
Larger diodes result in additional transition losses due to
their larger junction capacitance.
V
Main Switch Duty Cycle = OUT
VIN
V –V
Synchronous Switch Duty Cycle = IN OUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
( )( )
2
V
PMAIN = OUT IMAX 1 + δ RDS(ON) +
VIN
( ) (IMAX )(CRSS )(f)
k VIN
2
( )( )
2
V –V
PSYNC = IN OUT IMAX 1 + δ RDS(ON)
VIN
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V the
high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CRSS actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during a
short-circuit when the synchronous switch is on close to
100% of the period.
CIN and COUT Selection
The selection of CIN is simplified by the multiphase architecture and its impact on the worst-case RMS current
drawn through the input network (battery/fuse/capacitor).
It can be shown that the worst case RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula below to determine the maximum RMS current
requirement. Increasing the output current, drawn from
the other out-of-phase controller, will actually decrease
the input RMS ripple current from this maximum value
(see Figure 4). The out-of-phase technique typically reduces the input capacitor’s RMS ripple current by a factor
of 30% to 70% when compared to a single phase power
supply solution.
The type of input capacitor, value and ESR rating have
efficiency effects that need to be considered in the selection process. The capacitance value chosen should be
sufficient to store adequate charge to keep high peak
battery currents down. 20µF to 40µF is usually sufficient
for a 25W output supply operating at 200kHz. The ESR of
the capacitor is important for capacitor power dissipation
as well as overall battery efficiency. All of the power (RMS
ripple current • ESR) not only heats up the capacitor but
wastes power from the battery.
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Medium voltage (20V to 35V) ceramic, tantalum, OS-CON
and switcher-rated electrolytic capacitors can be used as
input capacitors, but each has drawbacks: ceramic voltage
coefficients are very high and may have audible piezoelectric effects; tantalums need to be surge-rated; OS-CONs
suffer from higher inductance, larger case size and limited
surface-mount applicability; electrolytics’ higher ESR and
dryout possibility require several to be used. Multiphase
systems allow the lowest amount of capacitance overall.
As little as one 22µF or two to three 10µF ceramic capacitors are an ideal choice in a 20W to 35W power supply due
to their extremely low ESR. Even though the capacitance
at 20V is substantially below their rating at zero-bias, very
low ESR loss makes ceramics an ideal candidate for
highest efficiency battery operated systems. Also consider parallel ceramic and high quality electrolytic capacitors as an effective means of achieving ESR and bulk
capacitance goals.
In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/VIN. To
prevent large voltage transients, a low ESR input capacitor
sized for the maximum RMS current of one channel must
be used. The maximum RMS capacitor current is given by:
CIN Re quiredIRMS ≈ IMAX
[V (V
OUT
IN − VOUT
)]
1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst case condition is commonly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the
capacitor, or to choose a capacitor rated at a higher
temperature than required. Several capacitors may also be
paralleled to meet size or height requirements in the
design. Always consult the manufacturer if there is any
question.
The benefit of the LTC1628 multiphase can be calculated
by using the equation above for the higher power controller and then calculating the loss that would have resulted
if both controller channels switch on at the same time. The
total RMS power lost is lower when both controllers are
operating due to the interleaving of current pulses through
the input capacitor’s ESR. This is why the input capacitor’s
requirement calculated above for the worst-case controller is adequate for the dual controller design. Remember
that input protection fuse resistance, battery resistance
and PC board trace resistance losses are also reduced due
to the reduced peak currents in a multiphase system. The
overall benefit of a multiphase design will only be fully
realized when the source impedance of the power supply/
battery is included in the efficiency testing. The drains of
the two top MOSFETS should be placed within 1cm of each
other and share a common CIN(s). Separating the drains
and CIN may produce undesirable voltage and current
resonances at VIN.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering.
The output ripple (∆VOUT) is determined by:

1 
∆VOUT ≈ ∆IL  ESR +

8 fCOUT 

Where f = operating frequency, COUT = output capacitance,
and ∆IL= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
with input voltage. With ∆IL = 0.3IOUT(MAX) the output
ripple will typically be less than 50mV at max VIN assuming:
COUT Recommended ESR < 2 RSENSE
and COUT > 1/(8fRSENSE)
The first condition relates to the ripple current into the ESR
of the output capacitance while the second term guarantees that the output capacitance does not significantly
discharge during the operating frequency period due to
ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging
term but can be compensated for by using capacitors of
very low ESR to maintain the ripple voltage at or below
50mV. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected.
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Manufacturers such as Nichicon, United Chemicon and
Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
recommended. Good bypassing is necessary to supply
the high transient currents required by the MOSFET gate
drivers and to prevent interaction between channels.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1628 to be
exceeded. The system supply current is normally dominated by the gate charge current. Additional external
loading of the INTVCC and 3.3V linear regulators also
needs to be taken into account for the power dissipation
calculations. The total INTVCC current can be supplied by
either the 5V internal linear regulator or by the EXTVCC
input pin. When the voltage applied to the EXTVCC pin is
less than 4.7V, all of the INTVCC current is supplied by the
internal 5V linear regulator. Power dissipation for the IC in
this case is highest: (VIN)(IINTVCC), and overall efficiency
is lowered. The gate charge current is dependent on
operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 2 of the
Electrical Characteristics. For example, the LTC1628 VIN
current is limited to less than 24mA from a 24V supply
when not using the EXTVCC pin as follows:
In surface mount applications multiple capacitors may
need to be used in parallel to meet the ESR, RMS current
handling and load step requirements of the application.
Aluminum electrolytic, dry tantalum and special polymer
capacitors are available in surface mount packages. Special polymer surface mount capacitors offer very low ESR
but have lower storage capacity per unit volume than other
capacitor types. These capacitors offer a very cost-effective output capacitor solution and are an ideal choice when
combined with a controller having high loop bandwidth.
Tantalum capacitors offer the highest capacitance density
and are often used as output capacitors for switching
regulators having controlled soft-start. Several excellent
surge-tested choices are the AVX TPS, AVX TPSV or the
KEMET T510 series of surface mount tantalums, available
in case heights ranging from 2mm to 4mm. Aluminum
electrolytic capacitors can be used in cost-driven applications providing that consideration is given to ripple current
ratings, temperature and long term reliability. A typical
application will require several to many aluminum electrolytic capacitors in parallel. A combination of the above
mentioned capacitors will often result in maximizing performance and minimizing overall cost. Other capacitor
types include Nichicon PL series, NEC Neocap, Pansonic
SP and Sprague 595D series. Consult manufacturers for
other specific recommendations.
Dissipation should be calculated to also include any added
current drawn from the internal 3.3V linear regulator. To
prevent maximum junction temperature from being exceeded, the input supply current must be checked operating in continuous mode at maximum VIN.
INTVCC Regulator
EXTVCC Connection
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. INTVCC powers
the drivers and internal circuitry within the LTC1628. The
INTVCC pin regulator can supply a peak current of 50mA
and must be bypassed to ground with a minimum of
4.7µF tantalum, 10µF special polymer, or low ESR type
electrolytic capacitor. A 1µF ceramic capacitor placed
directly adjacent to the INTVCC and PGND IC pins is highly
The LTC1628 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTVCC rises above 4.7V, the
internal regulator is turned off and the switch closes,
connecting the EXTVCC pin to the INTVCC pin thereby
supplying internal power. The switch remains closed as
long as the voltage applied to EXTVCC remains above 4.5V.
This allows the MOSFET driver and control power to be
TJ = 70°C + (24mA)(24V)(95°C/W) = 125°C
Use of the EXTVCC input pin reduces the junction temperature to:
TJ = 70°C + (24mA)(5V)(95°C/W) = 81°C
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derived from the output during normal operation (4.7V <
VOUT < 7V) and from the internal regulator when the output
is out of regulation (start-up, short-circuit). If more current is required through the EXTVCC switch than is specified, an external Schottky diode can be added between the
EXTVCC and INTVCC pins. Do not apply greater than 7V to
the EXTVCC pin and ensure that EXTVCC␣ <␣ VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of (Duty Cycle)/(Efficiency). For 5V regulators this
supply means connecting the EXTVCC pin directly to VOUT.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC Left Open (or Grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting in
an efficiency penalty of up to 10% at high input voltages.
2. EXTVCC Connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC Connected to an External supply. If an external
supply is available in the 5V to 7V range, it may be used to
power EXTVCC providing it is compatible with the MOSFET
gate drive requirements.
4. EXTVCC Connected to an Output-Derived Boost Network. For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage that has been boosted to greater
than 4.7V. This can be done with either the inductive boost
winding as shown in Figure 6a or the capacitive charge
pump shown in Figure 6b. The charge pump has the
advantage of simple magnetics.
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors CB connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the functional diagram is charged though
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on, the
driver places the CB voltage across the gate-source of the
desired MOSFET. This enhances the MOSFET and turns on
the topside switch. The switch node voltage, SW, rises to
VIN and the BOOST pin follows. With the topside MOSFET
on, the boost voltage is above the input supply: VBOOST =
VIN + VINTVCC. The value of the boost capacitor CB needs
to be 100 times that of the total input capacitance of the
topside MOSFET(s). The reverse breakdown of the external Schottky diode must be greater than VIN(MAX). When
adjusting the gate drive level, the final arbiter is the total
input current for the regulator. If a change is made and the
input current decreases, then the efficiency has improved.
If there is no change in input current, then there is no
change in efficiency.
VIN
OPTIONAL EXTVCC
CONNECTION
5V < VSEC < 7V
+
1µF
+
CIN
CIN
VSEC
VIN
BAT85
VIN
+
N-CH
LTC1628
+
VIN
1µF
BAT85
N-CH
LTC1628
TG1
0.22µF
BAT85
VN2222LL
TG1
RSENSE
RSENSE
VOUT
EXTVCC
SW
FCB
BG1
T1
1:N
R6
VOUT
EXTVCC
L1
SW
+
R5
+
COUT
N-CH
SGND
COUT
BG1
N-CH
PGND
PGND
1628 F06a
Figure 6a. Secondary Output Loop & EXTVCC Connection
1628 F06b
Figure 6b. Capacitive Charge Pump for EXTVCC
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Output Voltage
Soft-Start/Run Function
The LTC1628 output voltages are each set by an external
feedback resistive divider carefully placed across the
output capacitor. The resultant feedback signal is compared with the internal precision 0.800V voltage reference
by the error amplifier. The output voltage is given by the
equation:
The RUN/SS1 and RUN/SS2 pins are multipurpose pins
that provide a soft-start function and a means to shut
down the LTC1628. Soft-start reduces the input power
source’s surge currents by gradually increasing the
controller’s current limit (proportional to VITH). This pin
can also be used for power supply sequencing.
 R2 
VOUT = 0.8 V 1 + 
 R1
SENSE+/SENSE– Pins
The common mode input range of the current comparator
sense pins is from 0V to (1.1)INTVCC. Continuous linear
operation is guaranteed throughout this range allowing
output voltage setting from 0.8V to 7.7V, depending upon
the voltage applied to EXTVCC. A differential NPN input
stage is biased with internal resistors from an internal
2.4V source as shown in the Functional Diagram. This
requires that current either be sourced or sunk from the
SENSE pins depending on the output voltage. If the output
voltage is below 2.4V current will flow out of both SENSE
pins to the main output. The output can be easily preloaded
by the VOUT resistive divider to compensate for the current
comparator’s negative input bias current. The maximum
current flowing out of each pair of SENSE pins is:
ISENSE+ + ISENSE– = (2.4V – VOUT)/24k
Since VOSENSE is servoed to the 0.8V reference voltage, we
can choose R1 in Figure 2 to have a maximum value to
absorb this current.


0.8 V
R1(MAX ) = 24k 

 2.4V – VOUT 
for VOUT < 2.4V
Regulating an output voltage of 1.8V, the maximum value
of R1 should be 32K. Note that for an output voltage above
2.4V, R1 has no maximum value necessary to absorb the
sense currents; however, R1 is still bounded by the
VOSENSE feedback current.
An internal 1.2µA current source charges up the CSS
capacitor. When the voltage on RUN/SS1 (RUN/SS2)
reaches 1.5V, the particular controller is permitted to start
operating. As the voltage on RUN/SS increases from 1.5V
to 3.0V, the internal current limit is increased from 25mV/
RSENSE to 75mV/RSENSE. The output current limit ramps
up slowly, taking an additional 1.25s/µF to reach full
current. The output current thus ramps up slowly, reducing the starting surge current required from the input
power supply. If RUN/SS has been pulled all the way to
ground there is a delay before starting of approximately:
(
)
tDELAY =
1.5V
CSS = 1.25s / µF CSS
1.2µA
tIRAMP =
3V − 1.5V
CSS = 1.25s / µF CSS
1.2µA
(
)
By pulling both RUN/SS pins below 1V and/or pulling the
STBYMD pin below 0.2V, the LTC1628 is put into low
current shutdown (IQ = 20µA). The RUN/SS pins can be
driven directly from logic as shown in Figure 7. Diode D1
in Figure 7 reduces the start delay but allows CSS to ramp
up slowly providing the soft-start function. Each RUN/SS
pin has an internal 6V zener clamp (See Functional
Diagram).
VIN
3.3V OR 5V
INTVCC
RUN/SS
RSS*
D1
RSS*
RUN/SS
CSS
CSS
*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF
(a)
(b)
1628 F07
Figure 7. RUN/SS Pin Interfacing
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Fault Conditions: Overcurrent Latchoff
The RUN/SS pins also provide the ability to latch off the
controller(s) when an overcurrent condition is detected.
The RUN/SS capacitor, CSS, is used initially to turn on and
limit the inrush current. After the controller has been
started and been given adequate time to charge up the
output capacitor and provide full load current, the RUN/SS
capacitor is used for a short-circuit timer. If the regulator’s
output voltage falls to less than 70% of its nominal value
after CSS reaches 4.1V, CSS begins discharging on the
assumption that the output is in an overcurrent condition.
If the condition lasts for a long enough period as determined by the size of the CSS and the specified discharge
current, the controller will be shut down until the RUN/SS
pin voltage is recycled. If the overload occurs during startup, the time can be approximated by:
tLO1 ≈ [CSS (4.1 – 1.5 + 4.1 – 3.5)]/(1.2µA)
= 2.7 • 106 (CSS)
If the overload occurs after start-up the voltage on CSS will
begin discharging from the zener clamp voltage:
tLO2 ≈ [CSS (6 – 3.5)]/(1.2µA) = 2.1 • 106 (CSS)
The FLTCPL pin (LTC1628 only) determines whether an
overload on one channel will latch off only that channel
(FLTCPL = 0V) or both channels (FLTCPL = INTV CC). This
built-in overcurrent latchoff can be overridden by providing a pull-up resistor to the RUN/SS pin as shown in
Figure 7. This resistance shortens the soft-start period
and prevents the discharge of the RUN/SS capacitor
during an over current condition. Tying this pull-up resistor to VIN as in Figure 7a, defeats overcurrent latchoff.
Diode-connecting this pull-up resistor to INTVCC , as in
Figure 7b, eliminates any extra supply current during
controller shutdown while eliminating the INTV CC loading
from preventing controller start-up.
Why should you defeat overcurrent latchoff? During the
prototyping stage of a design, there may be a problem
with noise pickup or poor layout causing the protection
circuit to latch off. Defeating this feature will easily allow
troubleshooting of the circuit and PC layout. The internal
short-circuit and foldback current limiting still remains
active, thereby protecting the power supply system from
failure. After the design is complete, a decision can be
made whether to enable the latchoff feature.
The value of the soft-start capacitor CSS may need to be
scaled with output voltage, output capacitance and load
current characteristics. The minimum soft-start capacitance is given by:
CSS > (COUT )(VOUT) (10 – 4) (RSENSE)
The minimum recommended soft-start capacitor of
CSS = 0.1µF will be sufficient for most applications.
Fault Conditions: Current Limit and Current Foldback
The LTC1628 current comparator has a maximum sense
voltage of 75mV resulting in a maximum MOSFET current
of 75mV/RSENSE. The maximum value of current limit
generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the highest power
dissipation in the top MOSFET.
The LTC1628 includes current foldback to help further
limit load current when the output is shorted to ground.
The foldback circuit is active even when the overload
shutdown latch described above is overridden. If the
output falls below 70% of its nominal output level, then the
maximum sense voltage is progressively lowered from
75mV to 25mV. Under short-circuit conditions with very
low duty cycles, the LTC1628 will begin cycle skipping in
order to limit the short-circuit current. In this situation the
bottom MOSFET will be dissipating most of the power but
less than in normal operation. The short-circuit ripple
current is determined by the minimum on-time tON(MIN) of
the LTC1628 (less than 200ns), the input voltage and
inductor value:
∆IL(SC) = tON(MIN) (VIN/L)
The resulting short-circuit current is:
ISC =
25mV 1
+ ∆IL(SC)
RSENSE 2
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Fault Conditions: Overvoltage Protection (Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
much higher than nominal levels. The crowbar causes
huge currents to flow, that blow the fuse to protect against
a shorted top MOSFET if the short occurs while the
controller is operating.
A comparator monitors the output for overvoltage conditions. The comparator (OV) detects overvoltage faults
greater than 7.5% above the nominal output voltage.
When this condition is sensed, the top MOSFET is turned
off and the bottom MOSFET is turned on until the overvoltage condition is cleared. The output of this comparator is
only latched by the overvoltage condition itself and will
therefore allow a switching regulator system having a poor
PC layout to function while the design is being debugged.
The bottom MOSFET remains on continuously for as long
as the OV condition persists; if VOUT returns to a safe level,
normal operation automatically resumes. A shorted top
MOSFET will result in a high current condition which will
open the system fuse. The switching regulator will regulate properly with a leaky top MOSFET by altering the duty
cycle to accommodate the leakage.
The Standby Mode (STBYMD) Pin Function
The Standby Mode (STBYMD) pin provides several choices
for start-up and standby operational modes. If the pin is
pulled to ground, the RUN/SS pins for both controllers are
internally pulled to ground, preventing start-up and thereby
providing a single control pin for turning off both controllers at once. If the pin is left open or decoupled with a
capacitor to ground, the RUN/SS pins are each internally
provided with a starting current enabling external control
for turning on each controller independently. If the pin is
provided with a current of >3µA at a voltage greater than
2V, both internal linear regulators (INTVCC and 3.3V) will
be on even when both controllers are shut down. In this
mode, the onboard 3.3V and 5V linear regulators can
provide power to keep-alive functions such as a keyboard
controller. This pin can also be used as a latching “on” and/
or latching “off” power switch if so designed.
Frequency of Operation
The LTC1628 has an internal voltage controlled oscillator.
The frequency of this oscillator can be varied over a 2 to 1
range. The pin is internally self-biased at 1.19V, resulting
in a free-running frequency of approximately 220kHz. The
FREQSET pin can be grounded to lower this frequency to
approximately 140kHz or tied to the INTVCC pin to yield
approximately 310kHz. The FREQSET pin may be driven
with a voltage from 0 to INTVCC to fix or modulate the
oscillator frequency as shown in Figure 5.
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest time duration
that the LTC1628 is capable of turning on the top MOSFET.
It is determined by internal timing delays and the gate
charge required to turn on the top MOSFET. Low duty cycle
applications may approach this minimum on-time limit
and care should be taken to ensure that
tON(MIN) <
VOUT
VIN( f)
If the duty cycle falls below what can be accommodated by
the minimum on-time, the LTC1628 will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for the LTC1628 is generally less
than 200ns. However, as the peak sense voltage decreases
the minimum on-time gradually increases up to about
300ns. This is of particular concern in forced continuous
applications with low ripple current at light loads. If the
duty cycle drops below the minimum on-time limit in this
situation, a significant amount of cycle skipping can occur
with correspondingly larger current and voltage ripple.
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FCB Pin Operation
The FCB pin can be used to regulate a secondary winding
or as a logic level input. Continuous operation is forced
when the FCB pin drops below 0.8V. During continuous
mode, current flows continuously in the transformer primary. The secondary winding(s) draw current only when
the bottom, synchronous switch is on. When primary load
currents are low and/or the VIN/VOUT ratio is low, the
synchronous switch may not be on for a sufficient amount
of time to transfer power from the output capacitor to the
secondary load. Forced continuous operation will support
secondary windings providing there is sufficient synchronous switch duty factor. Thus, the FCB input pin removes
the requirement that power must be drawn from the
inductor primary in order to extract power from the
auxiliary windings. With the loop in continuous mode, the
auxiliary outputs may nominally be loaded without regard
to the primary output load.
The secondary output voltage VSEC is normally set as
shown in Figure 6a by the turns ratio N of the transformer:
VSEC ≅ (N + 1) VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
then VSEC will droop. An external resistive divider from
VSEC to the FCB pin sets a minimum voltage VSEC(MIN):
 R6 
VSEC(MIN) ≈ 0.8 V 1 + 
 R5 
If VSEC drops below this level, the FCB voltage forces
temporary continuous switching operation until VSEC is
again above its minimum.
In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.18µA
internal current source pulling the pin high. Include this
current when choosing resistor values R5 and R6.
The following table summarizes the possible states available on the FCB pin:
Table 1
FCB Pin
Condition
0V to 0.75V
Forced Continuous (Current Reversal
Allowed—Burst Inhibited)
0.85V < VFCB < 4.3V
Minimum Peak Current Induces
Burst Mode Operation
No Current Reversal Allowed
Feedback Resistors
Regulating a Secondary Winding
>4.8V
Burst Mode Operation Disabled
Constant Frequency Mode Enabled
No Current Reversal Allowed
No Minimum Peak Current
The FLTCPL pin determines whether only the first or both
controllers are temporarily forced into continuous mode
when the FCB pin falls below 0.8V. Tying the FLTCPL pin
to ground will send only the first controller into continuous
operation while tying the FLTCPL pin to INTVCC will send
both controllers into continuous operation.
Voltage Positioning
Voltage positioning can be used to minimize peak-to-peak
output voltage excursions under worst-case transient
loading conditions. The open-loop DC gain of the control
loop is reduced depending upon the maximum load step
specifications. Voltage positioning can easily be added to
the LTC1628 by loading the ITH pin with a resistive divider
having a Thevenin equivalent voltage source equal to the
midpoint operating voltage of the error amplifier, or 1.2V
(see Figure 8).
The resistive load reduces the DC loop gain while maintaining the linear control range of the error amplifier. The
maximum output voltage deviation can theoretically be
INTVCC
RT2
ITH
RT1
RC
LTC1628
CC
1628 F08
Figure 8. Active Voltage Positioning Applied to the LTC1628
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reduced to half or alternatively the amount of output
capacitance can be reduced for a particular application. A
complete explanation is included in Design Solutions 10.
(See www.linear.com.)
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1628 circuits: 1) LTC1628 VIN current (including loading on the 3.3V internal regulator), 2) INTVCC
regulator current, 3) I2R losses, 4) Topside MOSFET
transition losses.
1. The VIN current has two components: the first is the DC
supply current given in the Electrical Characteristics table,
which excludes MOSFET driver and control currents; the
second is the current drawn from the 3.3V linear regulator
output. VIN current typically results in a small (<0.1%) loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTVCC to
ground. The resulting dQ/dt is a current out of INTVCC that
is typically much larger than the control circuit current. In
continuous mode, IGATECHG =f(QT+QB), where QT and QB
are the gate charges of the topside and bottom side
MOSFETs.
Supplying INTVCC power through the EXTVCC switch input
from an output-derived source will scale the VIN current
required for the driver and control circuits by a factor of
(Duty Cycle)/(Efficiency). For example, in a 20V to 5V
application, 10mA of INTVCC current results in approximately 2.5mA of VIN current. This reduces the mid-current
loss from 10% or more (if the driver was powered directly
from VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor,
and input and output capacitor ESR. In continuous mode
the average output current flows through L and RSENSE,
but is “chopped” between the topside MOSFET and the
synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one
MOSFET can simply be summed with the resistances of L,
RSENSE and ESR to obtain I2R losses. For example, if each
RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE = 10mΩ and RESR
= 40mΩ (sum of both input and output capacitance
losses), then the total resistance is 130mΩ. This results in
losses ranging from 3% to 13% as the output current
increases from 1A to 5A for a 5V output, or a 4% to 20%
loss for a 3.3V output. Efficiency varies as the inverse
square of VOUT for the same external components and
output power level. The combined effects of increasingly
lower output voltages and higher currents required by
high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high input
voltages (typically 15V or greater). Transition losses can
be estimated from:
Transition Loss = (1.7) VIN2 IO(MAX) CRSS f
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses during
the design phase. The internal battery and fuse resistance
losses can be minimized by making sure that CIN has
adequate charge storage and very low ESR at the switching frequency. A 25W supply will typically require a
minimum of 20µF to 40µF of capacitance having a maximum of 20mΩ to 50mΩ of ESR. The LTC1628 2-phase
architecture typically halves this input capacitance requirement over competing solutions. Other losses including Schottky conduction losses during dead-time and
inductor core losses generally account for less than 2%
total additional loss.
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Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD (ESR), where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. OPTILOOP compensation allows the transient response to be
optimized over a wide range of output capacitance and
ESR values. The availability of the ITH pin not only allows
optimization of control loop behavior but also provides a
DC coupled and AC filtered closed loop response test
point. The DC step, rise time and settling at this test point
truly reflects the closed loop response. Assuming a predominantly second order system, phase margin and/or
damping factor can be estimated using the percentage of
overshoot seen at this pin. The bandwidth can also be
estimated by examining the rise time at the pin. The ITH
external components shown in the Figure 1 circuit will
provide an adequate starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80% of
full-load current having a rise time of 1µs to 10µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This is
why it is better to look at the ITH pin signal which is in the
feedback loop and is the filtered and compensated control
loop response. The gain of the loop will be increased by
increasing RC and the bandwidth of the loop will be
increased by decreasing CC. If RC is increased by the same
factor that CC is decreased, the zero frequency will be kept
the same, thereby keeping the phase shift the same in the
most critical frequency range of the feedback loop. The
output voltage settling behavior is related to the stability of
the closed-loop system and will demonstrate the actual
overall supply performance.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than1:50, the switch rise time
should be controlled so that the load rise time is limited to
approximately 25 • CLOAD. Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current to
about 200mA.
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Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging into
the supply from hell. The main power line in an automobile
is the source of a number of nasty potential transients,
including load-dump, reverse-battery, and double-battery.
Load-dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
50A IPK RATING
12V
just what it says, while double-battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 9 is the most straight forward
approach to protect a DC/DC converter from the ravages
of an automotive power line. The series diode prevents
current from flowing during reverse-battery, while the
transient suppressor clamps the input voltage during
load-dump. Note that the transient suppressor should not
conduct during double-battery operation, but must still
clamp the input voltage below breakdown of the converter.
Although the LTC1628 has a maximum input voltage of
36V, most applications will be limited to 30V by the
MOSFET BVDSS.
VIN
LTC1628
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
1628 F09
Figure 9. Automotive Application Protection
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LTC1628/LTC1628-PG
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Design Example
As a design example for one channel, assume VIN =
12V(nominal), VIN = 22V(max), VOUT = 1.8V, IMAX = 5A,
and f = 300kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQSET pin
to the INTVCC pin for 300kHz operation. The minimum
inductance for 30% ripple current is:
Choosing 1% resistors; R1 = 25.5k and R2 = 32.4k yields
an output voltage of 1.816V.
The power dissipation on the top side MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results
in; RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
()[
]
2
(0.042Ω) + 1.7(22V) (5A)(100pF)(300kHz)
PMAIN =
 V

V
∆IL = OUT  1 – OUT 
( f)(L) 
VIN 
A 4.7µH inductor will produce 23% ripple current and a
3.3µH will result in 33%. The peak inductor current will be
the maximum DC value plus one half the ripple current, or
5.84A, for the 3.3µH value. Increasing the ripple current
will also help ensure that the minimum on-time of 200ns
is not violated. The minimum on-time occurs at maximum
VIN:
tON(MIN) =
VOUT
VIN(MAX)f
=
1.8 V
= 273ns
22V(300kHz)
The RSENSE resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances:
50mV
RSENSE ≤
≈ 0.01Ω
5.84A
Since the output voltage is below 2.4V the output resistive
divider will need to be sized to not only set the output
voltage but also to absorb the SENSE pins specified input
current.
1.8 V 2
5 1 + (0.005)(50°C – 25°C)
22V
= 220mW
A short-circuit to ground will result in a folded back current
of:
ISC =
25mV 1  200ns(22V)
+ 
 = 3.2A
0.01Ω 2  3.3µH 
with a typical value of RDS(ON) and δ = (0.005/°C)(20) =
0.1. The resulting power dissipated in the bottom MOSFET
is:
( ) (1.1)(0.042Ω)
22V – 1.8 V
3.2A
22V
= 434mW
PSYNC =
2
which is less than under full-load conditions.
CIN is chosen for an RMS current rating of at least 3A at
temperature assuming only this channel is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
VORIPPLE = RESR(∆IL) = 0.02Ω(1.67A) = 33mVP–P


0.8 V
R1(MAX ) = 24k 

 2.4V – VOUT 

0.8 V 
= 24K
 = 32k
 2.4V – 1.8 V 
1628fb
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PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1628. These items are also illustrated graphically in
the layout diagram of Figure 10. The Figure 11 illustrates
the current waveforms present in the various branches of
the 2-phase synchronous regulators operating in the
continuous mode. Check the following in your layout:
1. Are the top N-channel MOSFETs M1 and M3 located
within 1cm of each other with a common drain connection
at CIN? Do not attempt to split the input decoupling for the
two channels as it can cause a large resonant loop.
FLTCPL
3
R2
4
R1
5
6
8
3.3V
10
11
12
R3
R4
13
14
SW1
VOSENSE1
BOOST1
FREQSET
VIN
STBYMD
BG1
FCB
EXTVCC
LTC1628
ITH1
SGND
3.3VOUT
ITH2
INTVCC
PGND
BG2
BOOST2
VOSENSE2
SW2
SENSE2 –
TG2
SENSE2 +
RUN/SS2
PGOOD
L1
RSENSE
VOUT1
26
25
CB1
M1
M2
D1
24
23
COUT1
RIN
22
CIN
CVIN
21
20
CINTVCC
GND
+
9
SENSE1 –
VPULL-UP
(<7V)
+
7
INTVCC
SENSE1 +
RUN/SS1
RPU
+
2
FLTCPL 28
(PGOOD)*
27
TG1
3. Do the LTC1628 VOSENSE pins resistive dividers connect to the (+) terminals of COUT? The resistive divider
must be connected between the (+) terminal of COUT and
+
1
2. Are the signal and power grounds kept separate? The
combined LTC1628 signal ground pin and the ground
return of CINTVCC must return to the combined COUT (–)
terminals. The path formed by the top N-channel MOSFET,
Schottky diode and the CIN capacitor should have short
leads and PC trace lengths. The output capacitor (–)
terminals should be connected as close as possible to the
(–) terminals of the input capacitor by placing the capacitors next to each other and away from the Schottky loop
described above.
VIN
COUT2
19
18
17
D2
CB2
M3
M4
RSENSE
16
15
VOUT2
L2
1628 F10
*PGOOD ON THE LTC1628-PG
Figure 10. LTC1628 Recommended Printed Circuit Layout Diagram
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LTC1628/LTC1628-PG
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SW1
L1
D1
RSENSE1
COUT1
VOUT1
+
RL1
VIN
RIN
CIN
+
SW2
BOLD LINES INDICATE
HIGH, SWITCHING
CURRENT LINES.
KEEP LINES TO A
MINIMUM LENGTH.
D2
L2
RSENSE2
COUT2
VOUT2
+
RL2
1628 F11
Figure 11. Branch Current Waveforms
signal ground and a small VOSENSE decoupling capacitor
should be as close as possible to the LTC1628 SGND pin.
The R2 and R4 connections should not be along the high
current input feeds from the input capacitor(s).
This capacitor carries the MOSFET drivers current peaks.
An additional 1µF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve noise
performance substantially.
4. Are the SENSE – and SENSE + leads routed together
with minimum PC trace spacing? The filter capacitor
between SENSE + and SENSE – should be as close as
possible to the IC. Ensure accurate current sensing with
Kelvin connections at the SENSE resistor.
6. Is the ITH pin filter capacitor connected close to the IC,
between ITH and the signal ground pins? This capacitor
helps to filter the effects of PCB noise and output voltage
ripple voltage from the compensaton loop
5. Is the INTVCC decoupling capacitor connected close to
the IC, between the INTVCC and the power ground pins?
7. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from the
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LTC1628/LTC1628-PG
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opposites channel’s voltage and current sensing feedback
pins. All of these nodes have very large and fast moving
signals and therefore should be kept on the “output side”
of the LTC1628 and occupy minimum PC trace area.
8. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on the
same side of the PC board as the input and output
capacitors with tie-ins for the bottom of the INTVCC
decoupling capacitor, the bottom of the voltage feedback
resistive divider and the SGND pin of the IC.
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use a
DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output
voltage as well. Check for proper performance over the
operating voltage and current range expected in the application. The frequency of operation should be maintained
over the input voltage range down to dropout and until the
output load drops below the low current operation threshold—typically 10% to 20% of the maximum designed
current level in Burst Mode operation.
The duty cycle percentage should be maintained from
cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic
rate can suggest noise pickup at the current or voltage
sensing inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC
layout if regulator bandwidth optimization is not required.
Only after each controller is checked for their individual
performance should both controllers be turned on at the
same time. A particularly difficult region of operation is
when one controller channel is nearing its current comparator trip point when the other channel is turning on its
top MOSFET. This occurs around 50% duty cycle on either
channel due to the phasing of the internal clocks and may
cause minor duty cycle jitter.
Short-circuit testing can be performed to verify proper
overcurrent latchoff, or 5µA can be provided to the RUN/
SS pin(s) by resistors from VIN to prevent the short-circuit
latchoff from occurring.
Reduce VIN from its nominal level to verify operation of the
regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher
output currents or only at higher input voltages. If problems coincide with high input voltages and low output
currents, look for capacitive coupling between the BOOST,
SW, TG, and possibly BG connections and the sensitive
voltage and current pins. The capacitor placed across the
current sensing pins needs to be placed immediately
adjacent to the pins of the IC. This capacitor helps to
minimize the effects of differential noise injection due to
high frequency capacitive coupling. If problems are encountered with high current output loading at lower input
voltages, look for inductive coupling between CIN, Schottky
and the top MOSFET components to the sensitive current
and voltage sensing traces. In addition, investigate common ground path voltage pickup between these components and the SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still be
maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
1628fb
29
LTC1628/LTC1628-PG
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TYPICAL APPLICATIO
59k
1M
FLTCPL
0.1µF
2
180pF
1000pF
3
105k, 1%
4
20k
1%
5
INTVCC
6
0.01µF
7
15k
1000pF 9
33pF
3.3V
10
11
15k
1000pF 12
20k
1%
13
63.4k
1%
180pF
1000pF
14
SW1
VOSENSE1
BOOST1
FREQSET
VIN
STBYMD
BG1
FCB
EXTVCC
LTC1628
ITH1
SGND
3.3VOUT
ITH2
INTVCC
PGND
BG2
BOOST2
VOSENSE2
SW2
SENSE2 –
TG2
SENSE2 +
RUN/SS2
+
0.015Ω
VOUT1
5V
3A; 4A PEAK
26
8
M1
0.1µF
25
M2
5
D1
MBRM
140T3
LT1121
3
ON/OFF
2
1
220k
24
23
10Ω
22
CMDSH-3TR
22µF
50V
150µF, 6.3V
PANASONIC SP
VOUT2
12V
120mA
+
100k
0.1µF
GND
21
20
19
1µF
10V
4.7µF
180µF, 4V
PANASONIC SP
CMDSH-3TR
18
17
1µF
25V
+
8
SENSE1 –
RUN/SS1
33µF
25V
T1, 1:1.8
10µH
+
33pF
SENSE1 +
FLTCPL* 28
(PGOOD)
27
TG1
MBRS1100T3
VPULL-UP
(<7V)
PGOOD
+
1
100k
0.1µF
M3
M4
D2
MBRM
140T3
VOUT2
3.3V
5A; 6A PEAK
16
15
L1
6.3µH
VIN
7V TO
28V
0.01Ω
0.1µF
1628 F12
VIN: 7V TO 28V
*PGOOD ON THE LTC1628-PG
VOUT: 5V, 3A/3.3V, 6A, 12V, 150mA
SWITCHING FREQUENCY = 300kHz
MI, M2, M3, M4: NDS8410A
L1: SUMIDA CEP123-6R3MC
T1: 10µH 1:1.8 — DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID
Figure 12. LTC1628 High Efficiency Low Noise 5V/3A, 3.3V/5A, 12/120mA Regulator
1628fb
30
LTC1628/LTC1628-PG
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PACKAGE DESCRIPTIO
(For purposes of clarity, drawings are not to scale)
G Package
28-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
10.07 – 10.33*
(0.397 – 0.407)
5.20 – 5.38**
(0.205 – 0.212)
1.73 – 1.99
(0.068 – 0.078)
28 27 26 25 24 23 22 21 20 19 18 17 16 15
0° – 8°
0.65
(0.0256)
BSC
0.55 – 0.95
(0.022 – 0.037)
0.13 – 0.22
(0.005 – 0.009)
NOTE: DIMENSIONS ARE IN MILLIMETERS
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.152mm (0.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.254mm (0.010") PER SIDE
7.65 – 7.90
(0.301 – 0.311)
0.05 – 0.21
(0.002 – 0.008)
0.25 – 0.38
(0.010 – 0.015)
1 2 3 4 5 6 7 8 9 10 11 12 13 14
G28 SSOP 1098
UH32 Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693)
0.57 ±0.05
5.35 ±0.05
4.20 ±0.05
3.45 ±0.05
(4 SIDES)
PACKAGE OUTLINE
0.23 ± 0.05
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
5.00 ± 0.10
(4 SIDES)
BOTTOM VIEW—EXPOSED PAD
R = 0.115
TYP
0.75 ± 0.05
0.00 – 0.05
0.40 ± 0.10
31 32
PIN 1
TOP MARK
1
2
3.45 ± 0.10
(4-SIDES)
(UH) QFN 0102
0.200 REF
NOTE:
1. DRAWING PROPOSED TO INCLUDE JEDEC PACKAGE OUTLINE
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
0.23 ± 0.05
0.50 BSC
1628fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC1628/LTC1628-PG
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TYPICAL APPLICATIO
FLTCPL
0.1µF
27pF
2
1000pF
105k
1%
3
4
20k
1%
5
INTVCC
6
0.01µF
7
8
220pF
9
3.3V
10
11
15k
220pF
12
20k
1%
27pF
13
63.4k
1%
1000pF
14
SW1
VOSENSE1
BOOST1
FREQSET
VIN
STBYMD
BG1
FCB
LTC1628
ITH1
SGND
EXTVCC
INTVCC
PGND
3.3VOUT
ITH2
BG2
BOOST2
VOSENSE2
SW2
SENSE2 –
TG2
SENSE2 +
RUN/SS2
L1
8µH
PGOOD
0.015Ω
VOUT1
5V
3A; 4A PEAK
26
M1A
0.1µF
25
M1B
D1
MBRM
140T3
24
47µF
6.3V
23
10Ω
22
CMDSH-3TR
22µF
50V
0.1µF
GND
21
20
1µF
10V
+
15k
33pF
SENSE1 –
RUN/SS1
VPULL-UP
(<7V)
+
33pF
SENSE1 +
FLTCPL* 28
(PGOOD)
27
TG1
+
1
4.7µF
56µF, 4V
19
CMDSH-3TR
18
17
0.1µF
M2A
M2B
D2
MBRM
140T3
VOUT2
3.3V
3A; 4A PEAK
16
15
L2
8µH
VIN
5.2V TO
28V
0.015Ω
0.1µF
1628 F13
VIN: 5.2V TO 28V
VOUT: 5V, 4A/3.3V, 4A
SWITCHING FREQUENCY = 300kHz
MI, M2: FDS8936A
L1, L2: 8µH SUMIDA CEP1238R0MC
OUTPUT CAPACITORS: PANASONIC SP SERIES
*PGOOD ON THE LTC1628-PG
Figure 13. LTC1628 5V/4A, 3.3V/4A Regulator
RELATED PARTS
PART NUMBER
LTC1159
LTC1438/LTC1439
LTC1538-AUX
LTC1539
LTC1530
LTC1625/LTC1775
LTC1629/LTC3729
DESCRIPTION
Synchronous Step-Down Switching Regulator
Dual Low Noise Synchronous Step-Down Switching Regulators
Dual Low Noise Synchronous Step-Down Switching Regulator
Dual Low Noise Synchronous Step-Down Switching Regulator
High Power Step-Down Synchronous DC/DC Controller in SO-8
No RSENSETM Current Mode Synchronous Step-Down Controller
20A to 200A PolyPhaseTM Synchronous Controller/LTC3729: 550kHz
COMMENTS
100% DC, Logic Level MOSFETs, VIN < 40V
POR, Auxiliary Regulator
Auxiliary Regulator, 5V Standby
5V Standby, POR, Low-Battery, Aux Regulator
High Efficiency 5V to 3.3V Conversion at Up to 15A
97% Efficiency, No Sense Resistor, 16-Pin SSOP
Expandable from 2-Phase to 12-Phase, All Surface Mount
Components, No Heat Sink, SSOP and QFN Packages
LTC1702
No RSENSE 2-Phase Dual Synchronous Step-Down Controller
550kHz, No Sense Resistor
LTC1703
No RSENSE 2-Phase Dual Synchronous Step-Down Controller
Mobile Pentium® III Processors, 550kHz,
with 5-Bit Mobile VID Control
VIN ≤ 7V
LT1709
2-Phase Synchronous Step-Down Switching Regulator
1.3V ≤ VOUT ≤ 3.5V, Current Mode Ensures
with 5-Bit VID
Accurate Current Sharing, 3.5V ≤ VIN ≤ 36V
LTC1735
Synchronous Step-Down Switching Regulator
Output Fault Protection, 16-Pin SSOP
LTC1736
Synchronous Controller with 5-Bit Mobile VID Control
Output Fault Protection, 24-Pin SSOP,
3.5V ≤ VIN ≤ 36V
LTC1929
2-Phase Synchronous Controller
Up to 42A, Uses All Surface Mount Components,
No Heat Sink, 3.5V ≤ VIN ≤ 36V
LTC3728
550kHz, Dual, 2-Phase Synchronous Step-Down Controller
Faster Switching Frequency Version of the LTC1628,
SSOP and QFN Packages, Small Inductors
Adaptive Power, No RSENSE and PolyPhase are trademarks of Linear Technology Corporation. Pentium is a registered trademark of Intel Corporation.
1628fb
32 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LT/TP 0302 1.5K REV B • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1998
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