Sound and Recording, Sixth Edition - Audio

Sound and Recording, Sixth Edition - Audio
Sound and Recording
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Sound and Recording
Sixth Edition
Francis Rumsey and Tim McCormick
AMSTERDAM • BOSTON • HEIDELBERG • LONDON • NEW YORK • OXFORD
PARIS • SAN DIEGO • SAN FRANCISCO • SINGAPORE • SYDNEY • TOKYO
Focal Press is an imprint of Elsevier
Focal Press is an imprint of Elsevier
Linacre House, Jordan Hill, Oxford OX2 8DP, UK
30 Corporate Drive, Suite 400, Burlington, MA 01803, USA
First edition 1992
Reprinted 1994
Second edition 1994
Reprinted 1995, 1996
Third edition 1997
Reprinted 1998 (twice), 1999, 2000, 2001
Fourth edition 2002
Reprinted 2003, 2004
Fifth edition 2006
Reprinted 2007, 2008
Sixth edition 2009
Copyright © 2009 Francis Rumsey & Tim McCormick. Published by Elsevier Ltd. All rights reserved.
The right of Francis Rumsey & Tim McCormick to be identified as the author of this work has been asserted in
accordance with the Copyright, Designs and Patents Act 1988
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British Library Cataloguing in Publication Data
Rumsey, Francis.
Sound and recording. — 6th ed.
1. Sound—Recording and reproducing.
I. Title II. McCormick, Tim, 1954621.3’893-dc22
Library of Congress Control Number: 2009929122
ISBN: 978-0-240-52163-3
For information on all Focal Press publications
visit our website at focalpress.com
Printed and bound in Great Britain
09 10 11 12
11 10 9 8 7 6 5 4 3 2 1
Contents
FACT FILE DIRECTORY ......................................................................... xi
PREFACE TO THE SECOND EDITION ..................................................... xv
PREFACE TO THE THIRD EDITION ...................................................... xvii
PREFACE TO THE FOURTH EDITION .................................................... xix
PREFACE TO THE FIFTH EDITION ........................................................ xxi
PREFACE TO THE SIXTH EDITION ...................................................... xxiii
CHAPTER 1
What is Sound? ................................................................ 1
A Vibrating Source ...................................................................... 1
Characteristics of a Sound Wave .................................................. 2
How Sound Travels in Air ............................................................ 4
Simple and Complex Sounds ....................................................... 5
Frequency Spectra of Repetitive Sounds ....................................... 6
Frequency Spectra of Non-repetitive Sounds ................................. 8
Phase ........................................................................................ 8
Sound in Electrical Form........................................................... 12
Displaying the Characteristics of a Sound Wave ........................... 15
The Decibel ............................................................................. 16
Sound Power and Sound Pressure .............................................. 16
Free and Reverberant Fields ...................................................... 19
Standing Waves ........................................................................ 23
Recommended Further Reading ................................................. 26
CHAPTER 2
Auditory Perception ....................................................... 29
The Hearing Mechanism .......................................................... 30
Frequency Perception .............................................................. 31
Loudness Perception ............................................................... 33
Practical Implications of Equal-loudness Contours ...................... 36
Spatial Perception ................................................................... 37
Recommended Further Reading ................................................ 46
Recommended Listening ........................................................... 46
CHAPTER 3
Microphones .................................................................. 47
The Moving-coil or Dynamic Microphone ..................................... 48
The Ribbon Microphone ............................................................ 51
The Capacitor or Condenser Microphone ..................................... 52
Directional Responses and Polar Diagrams .................................. 53
v
vi
Contents
Specialized Microphone Types ................................................... 59
Switchable Polar Patterns ......................................................... 61
Stereo Microphones .................................................................. 61
Microphone Performance .......................................................... 62
Microphone Powering Options.................................................... 66
Radio Microphones ................................................................... 69
Recommended Further Reading ................................................. 78
CHAPTER 4
Loudspeakers ................................................................. 79
The Moving-coil Loudspeaker..................................................... 80
Other Loudspeaker Types .......................................................... 81
Mounting and Loading Drive Units ............................................. 84
Complete Loudspeaker Systems ................................................. 88
Active Loudspeakers ................................................................. 90
Subwoofers .............................................................................. 91
Loudspeaker Performance ......................................................... 92
Setting up Loudspeakers ........................................................... 99
Thiele–Small Parameters and Enclosure
Volume Calculations ............................................................... 101
Digital Signal Processing in Loudspeakers ................................ 106
Recommended Further Reading ............................................... 106
CHAPTER 5
Mixers ......................................................................... 107
A Simple Six-channel Analog Mixer .......................................... 109
A Multitrack Mixer .................................................................. 115
Channel Grouping................................................................... 120
An Overview of Typical Mixer Facilities ..................................... 120
EQ Explained ......................................................................... 134
Stereo Line Input Modules ...................................................... 137
Dedicated Monitor Mixer ......................................................... 138
Technical Specifications.......................................................... 138
Metering Systems ................................................................... 142
Automation ............................................................................ 146
Digital Mixers ......................................................................... 155
Mixers with Integrated Control of Digital Workstations ................ 159
Introduction to Mixing Approaches ........................................... 161
Basic Operational Techniques .................................................. 163
Recommended Further Reading ............................................... 165
CHAPTER 6
Analog Recording ......................................................... 167
A Short History of Analog Recording ......................................... 168
Magnetic Tape ....................................................................... 170
Contents
The Magnetic Recording Process.............................................. 171
The Tape Recorder.................................................................. 176
Magnetic Recording Levels ...................................................... 178
What are Test Tapes For? ......................................................... 178
Tape Machine Alignment ......................................................... 180
Mechanical Transport Functions............................................... 184
The Compact Cassette ............................................................ 185
Recommended Further Reading ............................................... 187
CHAPTER 7
Noise Reduction ........................................................... 189
Why is Noise Reduction Required? ........................................... 190
Methods of Reducing Noise ..................................................... 190
Line-up of Noise Reduction Systems ........................................ 196
Single-ended Noise Reduction ................................................. 197
Recommended Further Reading ............................................... 199
CHAPTER 8
CHAPTER 9
Digital Audio Principles........................................................... 201
Digital and Analog Recording Contrasted .................................. 202
Binary for Beginners ............................................................... 203
The Digital Audio Signal Chain ................................................ 207
Analog-to-Digital Conversion .................................................... 209
D/A Conversion ....................................................................... 232
Direct Stream Digital (DSD)..................................................... 234
Changing the Resolution of an Audio Signal
(Requantization)..................................................................... 235
Introduction to Digital Signal Processing .................................. 238
Pitch Shifting and Time Stretching........................................... 247
Audio Data Reduction ............................................................. 248
Recommended Further Reading ............................................... 256
Digital Recording, Editing and Mastering Systems
257
Digital Tape Recording ............................................................ 258
Mass Storage-based Systems ................................................... 267
Audio Processing for Computer Workstations ............................. 276
Mass Storage-based Editing System Principles .......................... 280
Editing Software ..................................................................... 287
Mastering and Restoration ....................................................... 290
Preparing for and Understanding Release Media ........................ 294
Recommended Further Reading ............................................... 297
CHAPTER 10 Digital Audio Formats and Interchange ........................... 299
Audio File Formats for Digital Workstations ............................... 300
Consumer Optical Disk Formats ............................................... 315
Interconnecting Digital Audio Devices ...................................... 319
vii
viii
Contents
Recommended Further Reading ............................................... 340
Websites................................................................................ 340
CHAPTER 11 Power Amplifiers........................................................... 341
Domestic Power Amplifiers ...................................................... 342
Professional Amplifier Facilities ............................................... 344
Specifications ........................................................................ 345
Coupling ................................................................................ 351
CHAPTER 12 Lines and Interconnection ............................................. 353
Transformers .......................................................................... 354
Unbalanced Lines .................................................................. 356
Cable Effects with Unbalanced Lines ....................................... 358
Balanced Lines ...................................................................... 361
Working with Balanced Lines ................................................... 362
Star-quad Cable ..................................................................... 363
Electronic Balancing............................................................... 364
100 Volt Lines ....................................................................... 365
600 Ohms ............................................................................. 368
DI Boxes................................................................................ 370
Splitter Boxes ........................................................................ 373
Jackfields (Patchbays) ............................................................ 375
Distribution Amplifiers ............................................................ 379
CHAPTER 13 Plug-ins and Outboard Equipment .................................. 381
Plug-ins................................................................................. 381
Outboard Equipment .............................................................. 383
Connection of Outboard Devices .............................................. 394
Recommended Further Reading ............................................... 395
CHAPTER 14 MIDI and Synthetic Audio Control .................................. 397
Background ........................................................................... 399
What is MIDI? ........................................................................ 400
MIDI and Digital Audio Contrasted ........................................... 401
Basic Principles ..................................................................... 402
Interfacing a Computer to a MIDI System ................................. 406
How MIDI Control Works ......................................................... 410
MIDI Control of Sound Generators ............................................ 424
General MIDI ......................................................................... 434
Scalable Polyphonic MIDI (SPMIDI) ......................................... 437
RMID and XMF Files............................................................... 437
SAOL and SASL in MPEG 4 Structured Audio ........................... 439
Contents
MIDI over USB ....................................................................... 439
MIDI over IEEE 1394 ............................................................. 441
After MIDI? ............................................................................ 441
Sequencing Software .............................................................. 443
Recommended Further Reading ............................................... 452
Websites................................................................................ 452
CHAPTER 15 Synchronization ............................................................ 453
SMPTE/EBU Timecode ............................................................ 454
Recording Timecode ............................................................... 456
Machine Synchronizers ........................................................... 458
Digital Audio Synchronization .................................................. 461
MIDI and Synchronization ....................................................... 465
Synchronizing Audio/MIDI Computer Applications ..................... 469
Recommended Further Reading ............................................... 470
CHAPTER 16 Two-Channel Stereo ...................................................... 471
Principles of Loudspeaker Stereo ............................................. 473
Principles of Binaural or Headphone Stereo .............................. 482
Loudspeaker Stereo Over Headphones and Vice Versa ................ 484
Two-Channel Signal Formats .................................................... 487
Two-Channel Microphone Techniques ....................................... 489
Binaural Recording and ‘Dummy Head’ Techniques ................... 503
Spot microphones and Two-Channel Panning Laws .................... 505
Recommended Further Reading ............................................... 507
CHAPTER 17 Surround Sound ........................................................... 509
Three-Channel (3-0) Stereo ..................................................... 511
Four-Channel Surround (3-1 Stereo)......................................... 512
5.1-Channel Surround (3-2 Stereo) .......................................... 514
Other Multichannel Configurations ........................................... 520
Surround Sound Systems ........................................................ 522
Matrixed Surround Sound Systems ........................................... 522
Digital Surround Sound Formats .............................................. 528
Ambisonics ............................................................................ 534
Surround Sound Monitoring ..................................................... 540
Surround Sound Recording Techniques .................................... 545
Multichannel Panning Techniques ............................................ 557
Recommended Further Reading ............................................... 561
CHAPTER 18 Sound Quality .............................................................. 563
What is Sound Quality? ........................................................... 564
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Contents
Methods of Sound Quality Evaluation ....................................... 569
Aspects of Audio System Performance Affecting
Sound Quality ........................................................................ 575
Recommended Further Reading ............................................... 590
APPENDIX
Record Players.............................................................. 591
Pickup Mechanics .................................................................. 591
RIAA Equalization .................................................................. 595
Cartridge Types ...................................................................... 596
Connecting Leads ................................................................... 597
Arm Considerations ................................................................ 598
Laser Pickups ........................................................................ 599
Recommended Further Reading ............................................... 599
GLOSSARY OF TERMS....................................................................... 601
GENERAL FURTHER READING .......................................................... 609
INDEX .............................................................................................. 611
Fact File Directory
1.1
1.2
1.3
1.4
1.5
1.6
Ohm’s law................................................................................... 14
The decibel................................................................................. 17
The inverse-square law ................................................................. 18
Measuring SPLs .......................................................................... 20
Absorption, reflection and RT ....................................................... 22
Echoes and reflections ................................................................. 26
2.1
2.2
2.3
2.4
2.5
Critical bandwidth ....................................................................... 33
Equal-loudness contours .............................................................. 34
Masking ..................................................................................... 35
The precedence effect ................................................................. 39
Reflections affect spaciousness .................................................... 43
3.1
3.2
3.3
3.4
3.5
3.6
3.7
3.8
3.9
3.10
Electromagnetic transducers......................................................... 48
Dynamic microphone – principles ................................................. 49
Ribbon microphone – principles.................................................... 49
Capacitor microphone – principles ................................................ 50
Bass tip-up ................................................................................. 51
Sum and difference processing ..................................................... 63
Microphone sensitivity ................................................................. 64
Microphone noise specifications.................................................... 64
Phantom powering ....................................................................... 67
Frequency modulation ................................................................. 71
4.1
4.2
4.3
4.4
4.5
4.6
Electrostatic loudspeaker – principles............................................ 82
Transmission line system .............................................................. 85
Horn loudspeaker – principles....................................................... 86
A basic crossover network............................................................. 89
Loudspeaker sensitivity ................................................................ 94
Low-frequency Q ....................................................................... 103
5.1 Fader facts ............................................................................... 112
5.2 Pan control ............................................................................... 113
5.3 Pre-fade listen (PFL).................................................................. 114
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Fact File Directory
5.4
5.5
5.6
5.7
5.8
5.9
Audio groups............................................................................. 121
Control groups........................................................................... 122
Variable Q ................................................................................. 127
Common mode rejection ............................................................ 139
Clipping ................................................................................... 141
Metering, signal levels and distortion........................................... 143
6.1
6.2
6.3
6.4
6.5
6.6
A magnetic recording head ......................................................... 172
Replay head effects ................................................................... 175
Sync replay ............................................................................... 178
Magnetic reference levels ........................................................... 179
Bias adjustment ........................................................................ 181
Azimuth alignment .................................................................... 182
7.1 Pre-emphasis ............................................................................ 191
8.1
8.2
8.3
8.4
8.5
8.6
8.7
8.8
8.9
8.10
8.11
Analog and digital information .................................................... 204
Negative numbers ..................................................................... 206
Logical operations ..................................................................... 208
Sampling – frequency domain..................................................... 214
Audio sampling frequencies........................................................ 217
Parallel and serial representation ................................................ 222
Dynamic range and perception.................................................... 225
Quantizing resolutions ............................................................... 227
Types of dither .......................................................................... 228
Dynamic range enhancement ..................................................... 236
Crossfading .............................................................................. 241
9.1
9.2
9.3
9.4
9.5
9.6
9.7
9.8
Rotary and stationary heads ........................................................ 259
Data recovery ............................................................................ 261
Error handling ........................................................................... 263
RAID arrays .............................................................................. 269
Storage requirements of digital audio .......................................... 271
Peripheral interfaces .................................................................. 274
Audio processing latency ............................................................ 278
DVD discs and players ............................................................... 295
10.1
10.2
10.3
10.4
10.5
Broadcast wave format .............................................................. 307
Recordable DVD formats ............................................................ 316
Computer networks vs digital audio interfaces .............................. 321
Carrying data-reduced audio....................................................... 326
Extending a network .................................................................. 331
Fact File Directory
11.1 Amplifier classes ....................................................................... 343
11.2 Power bandwidth ...................................................................... 347
11.3 Slew rate .................................................................................. 348
12.1 The transformer ........................................................................ 355
12.2 Earth loops ............................................................................... 357
12.3 XLR-3 connectors ..................................................................... 363
13.1 Plug-in examples ...................................................................... 384
13.2 Compression and limiting .......................................................... 388
13.3 Simulating reflections ................................................................389
14.1
14.2
14.3
14.4
14.5
14.6
Midi hardware interface ............................................................. 404
MIDI connectors and cables ....................................................... 405
MIDI message format ................................................................ 411
Registered and non-registered parameter numbers ....................... 433
Standard MIDI files (SMF) ......................................................... 435
Downloadable sounds and SoundFonts ........................................ 438
15.1
15.2
15.3
15.4
Drop-frame timecode ................................................................. 455
Types of lock ............................................................................ 460
Synchronizer terminology ........................................................... 461
Relationships between video frame rates and
audio sampling rates .................................................................464
15.5 Quarter-frame MTC messages .....................................................470
16.1
16.2
16.3
16.4
16.5
16.6
16.7
Binaural versus ‘stereophonic’ localization ................................... 476
Stereo vector summation ........................................................... 479
The ‘Williams curves’................................................................. 481
Transaural stereo ....................................................................... 486
Stereo misalignment effects ....................................................... 489
Stereo width issues ................................................................... 491
End-fire and side-fire configurations ............................................ 497
17.1
17.2
17.3
17.4
17.5
17.6
Track allocations in 5.1 ............................................................. 517
Bass management in 5.1 ........................................................... 518
What is THX? ............................................................................ 526
Higher-order ambisonics ............................................................ 539
Loudspeaker mounting .............................................................. 541
Surround imaging ..................................................................... 546
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Fact File Directory
18.1
18.2
18.3
18.4
18.5
Effects of frequency response on sound quality ............................577
Effects of harmonic distortion on sound quality ............................580
Noise weighting curves ..............................................................583
Jitter........................................................................................584
Minimizing coding artifacts ........................................................588
A1.1 Stylus profile ............................................................................ 592
A1.2 Tracking weight ......................................................................... 594
Preface to the Second Edition
One of the greatest dangers in writing a book at an introductory level is
to sacrifice technical accuracy for the sake of simplicity. In writing Sound
and Recording: An Introduction we have gone to great lengths not to fall
into this trap, and have produced a comprehensive introduction to the field
of audio, intended principally for the newcomer to the subject, which is
both easy to understand and technically precise. We have written the book
that we would have valued when we first entered the industry, and as such
it represents a readable reference, packed with information. Many books
stop after a vague overview, just when the reader wants some clear facts
about a subject, or perhaps assume too much knowledge on the reader’s
behalf. Books by contributed authors often suffer from a lack of consistency
in style, coverage and technical level. Furthermore, there is a tendency for
books on audio to be either too technical for the beginner or, alternatively,
subjectively biased towards specific products or operations. There are also
quite a number of American books on sound recording which, although
good, tend to ignore European trends and practices. We hope that we have
steered a balanced course between these extremes, and have deliberately
avoided any attempt to dictate operational practice.
Sound and Recording: An Introduction is definitely biased towards an
understanding of ‘how it works’, as opposed to ‘how to work it’, although
technology is never discussed in an abstract manner but related to operational reality. Although we have included a basic introduction to acoustics and the nature of sound perception, this is not a book on acoustics or
musical acoustics (there are plenty of those around). It is concerned with
the principles of audio recording and reproduction, and has a distinct bias
towards the professional rather than the consumer end of the market. The
coverage of subject matter is broad, including chapters on digital audio,
timecode synchronization and MIDI, amongst other more conventional
subjects, and there is comprehensive coverage of commonly misunderstood
subjects such as the decibel, balanced lines, reference levels and metering
systems.
This second edition of the book has been published only two years after
the first, and the subject matter has not changed significantly enough in
the interim to warrant major modifications to the existing chapters. The
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Preface to the Second Edition
key difference between the second and first editions is the addition of a
long chapter on stereo recording and reproduction. This important topic
is covered in considerable detail, including historical developments, principles of stereo reproduction, surround sound and stereo microphone techniques. Virtually every recording or broadcast happening today is made in
stereo, and although surround sound has had a number of notable ‘flops’ in
the past it is likely to become considerably more important in the next ten
years. Stereo and surround sound are used extensively in film, video and
television production, and any new audio engineer should be familiar with
the principles.
Since this is an introductory book, it will be of greatest value to the student of sound recording or music technology, and to the person starting
out on a career in sound engineering or broadcasting. The technical level
has deliberately been kept reasonably low for this reason, and those who
find this frustrating probably do not need the book! Nonetheless, it is often
valuable for the seasoned audio engineer to go back to basics. Further reading suggestions have been made in order that the reader may go on to a
more in-depth coverage of the fields introduced here, and some of the references are considerably more technical than this book. Students will find
these suggestions valuable when planning a course of study.
Francis Rumsey
Tim McCormick
Preface to the Third Edition
Since the first edition of Sound and Recording some of the topics have
advanced quite considerably, particularly the areas dependent on digital and
computer technology. Consequently I have rewritten the chapters on digital recording and MIDI (Chapters 10 and 15), and have added a larger section on mixer automation in Chapter 7. Whereas the first edition of the
book was quite ‘analogue’, I think that there is now a more appropriate
balance between analogue and digital topics. Although analogue audio is by
no means dead (sound will remain analogue for ever!), most technological
developments are now digital.
I make no apologies for leaving in the chapter on record players,
although some readers have commented that they think it is a waste of
space. People still use record players, and there is a vast store of valuable
material on LP record. I see no problem with keeping a bit of history in the
book – you never know, it might come in useful one day when everyone has
forgotten (and some may never have known) what to do with vinyl discs. It
might even appease the faction of our industry that continues to insist that
vinyl records are the highest fidelity storage medium ever invented.
Francis Rumsey
Guildford
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Preface to the Fourth Edition
The fourth edition is published ten years after Sound and Recording was
first published, which is hard to believe. The book has been adopted widely
by students and tutors on audio courses around the world. In that time
audio technology and techniques have changed in some domains but not in
others. All the original principles still apply but the emphasis has gradually
changed from predominantly analogue to quite strongly digital, although
many studios still use analogue mixers and multitrack tape recorders for
a range of purposes and we do not feel that the death-knell of analogue
recording has yet been sounded. Readers of earlier editions will notice that
the chapter on record players has finally been reduced in size and relegated
to an appendix. While we continue to believe that information about the
LP should remain in the literature as the format lingers on, it is perhaps
time to remove it from the main part of the book.
In this edition a new chapter on surround sound has been added, complemented by a reworked chapter preceding it that is now called ‘two-channel
stereo’. Surround sound was touched upon in the previous edition but a
complete chapter reflects the increased activity in this field with the coming of new multichannel consumer replay formats.
The chapter on auditory perception has been reworked to include
greater detail on spatial perception and the digital audio chapter has been
updated to include DVD-A and SACD, with information about Direct
Stream Digital (DSD), the MiniDisc, computer-based editing systems and
their operation. Chapter 5 on loudspeakers now includes information about
distributed-mode loudspeakers (DML) and a substantial section on directivity and the various techniques used to control it. Finally a glossary of terms
has now been provided, with some additional material that supports the
main text.
Francis Rumsey
Tim McCormick
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Preface to the Fifth Edition
The fifth edition of Sound and Recording includes far greater detail on digital audio than the previous editions, reflecting the growing ‘all-digital’ trend
in audio equipment and techniques. In place of the previous single chapter
on the topic there are now three chapters (Chapters 8, 9 and 10) covering
principles, recording and editing systems, and applications. This provides
a depth of coverage of digital audio in the fifth edition that should enable
the reader to get a really detailed understanding of the principles of current
audio systems. We believe, however, that the detailed coverage of analogue
recording should remain in its current form, at least for this iteration of
the book. We have continued the trend, begun in previous new editions, of
going into topics in reasonable technical depth but without using unnecessary mathematics. It is intended that this will place Sound and Recording
slightly above the introductory level of the many broad-ranging textbooks
on recording techniques and audio, so that those who want to understand
how it works a bit better will find something to satisfy them here.
The chapter previously called ‘A guide to the audio signal chain’ has
been removed from this new edition, and parts of that material have now
found their way into other chapters, where appropriate. For example, the
part dealing with the history of analogue recording has been added to the
start of Chapter 6. Next, the material dealing with mixers has been combined into a single chapter (it is hard to remember why we ever divided
it into two) and now addresses both analogue and digital systems more
equally than before. Some small additions have been made to Chapters
12 and 13 and Chapter 14 has been completely revised and extended, now
being entitled ‘MIDI and synthetic audio control’.
Francis Rumsey
Tim McCormick
For information on all Focal Press publications visit our website at:
www.focalpress.com
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Preface to the Sixth Edition
When we first wrote this book it was our genuine intention to make it an
introduction to the topic of sound and recording that would be useful to
students starting out in the field. However, we readily admit that over the
years the technical level of the book has gradually risen in a number of
chapters, and that there are now many audio and music technology courses
that do not start out by covering the engineering aspects of the subject at
this level. For this reason, and recognizing that many courses use the book
as a somewhat more advanced text, we have finally allowed the book’s subtitle, ‘An Introduction’, to fall by the wayside.
In this edition we have overhauled many of the chapters, continuing the expansion and reorganization of the digital audio chapters to
include more recent details of pitch correction, file formats, interfaces
and Blu-Ray disk. The coverage of digital tape formats has been retained
in reduced form, partly for historical reasons. Chapters 6 and 7, covering
analog recording and noise reduction, have been shortened but it is felt
that they still justify inclusion given that such equipment is still in use in
the field. As fewer and fewer people in the industry continue to be familiar with such things as bias, replay equalization, azimuth and noise reduction line-up, we feel it is important that such information should continue
to be available in the literature while such technology persists. Likewise,
the appendix on record players survives, it being surprising how much this
equipment is still used.
The chapter on mixers has been thoroughly reworked and updated, as
it had become somewhat disorganized during its evolution through various editions, and the chapter on MIDI has been expanded to include more
information on sequencing principles. Chapter 15 on synchronization
has been revised to include substantially greater coverage of digital audio
synchronization topics, and the information about MIDI sync has also been
moved here. The section on digital plug-ins has been moved into the chapter on outboard equipment. A number of other additions have also been
made to the book, including an introduction to loudspeaker design parameters, further information on Class D amplifiers, and updated information
on wireless microphone frequencies.
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Preface to the Sixth Edition
Finally, a new chapter on sound quality has been added at the end of the
book, which incorporates some of the original appendix dealing with equipment specifications. This chapter introduces some of the main concepts
relating to the perception and evaluation of sound quality, giving examples
of relationships to simple aspects of audio equipment performance.
Francis Rumsey
Tim McCormick
January 2009
CHAPTER 1
What is Sound?
CH A P T E R C O N TE N T S
A Vibrating Source
Characteristics of a Sound Wave
How Sound Travels in Air
Simple and Complex Sounds
Frequency Spectra of Repetitive Sounds
Frequency Spectra of Non-repetitive Sounds
Phase
Sound in Electrical Form
Displaying the Characteristics of a Sound Wave
The Decibel
Sound Power and Sound Pressure
Free and Reverberant Fields
Standing Waves
1
2
4
5
6
8
8
12
15
16
16
19
23
A VIBRATING SOURCE
Sound is produced when an object (the source) vibrates and causes the air
around it to move. Consider the sphere shown in Figure 1.1. It is a pulsating sphere which could be imagined as something like a squash ball, and
it is pulsating regularly so that its size oscillates between being slightly
larger than normal and then slightly smaller than normal. As it pulsates it
will alternately compress and then rarefy the surrounding air, resulting in
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
1
2
CHAPTER 1: What is Sound?
FIGURE 1.1
(a) A simple sound source
can be imagined to be like
a pulsating sphere radiating
spherical waves. (b) The
longitudinal wave thus
created is a succession
of compressions and
rarefactions of the air.
(a)
Expanding sound field
Pulsating sphere
(b)
Rarefactions
Compressions
Direction of air
particle motion
Apparent direction of wave travel
a series of compressions and rarefactions traveling away from the sphere,
rather like a three-dimensional version of the ripples which travel away from
a stone dropped into a pond. These are known as longitudinal waves since
the air particles move in the same dimension as the direction of wave travel.
The alternative to longitudinal wave motion is transverse wave motion (see
Figure 1.2), such as is found in vibrating strings, where the motion of the
string is at right angles to the direction of apparent wave travel.
CHARACTERISTICS OF A SOUND WAVE
The rate at which the source oscillates is the frequency of the sound wave
it produces, and is quoted in hertz (Hz) or cycles per second (cps). 1000
hertz is termed 1 kilohertz (1 kHz). The amount of compression and rarefaction of the air which results from the sphere’s motion is the amplitude
of the sound wave, and is related to the loudness of the sound when it is
finally perceived by the ear (see Chapter 2). The distance between two adjacent peaks of compression or rarefaction as the wave travels through the
Characteristics of a Sound Wave
FIGURE 1.2
In a transverse wave the
motion of any point on the
wave is at right angles to
the apparent direction of
motion of the wave.
Motion of point
on string
Apparent direction of wave motion
FIGURE 1.3
A graphical representation
of a sinusoidal sound
waveform. The period of the
wave is represented by t,
and its frequency by 1/t.
t
Amplitude
+
0
Time
–
air is the wavelength of the sound wave, and is often represented by the
Greek letter lambda (λ). The wavelength depends on how fast the sound
wave travels, since a fast-traveling wave would result in a greater distance
between peaks than a slow-traveling wave, given a fixed time between compression peaks (i.e. a fixed frequency of oscillation of the source).
As shown in Figure 1.3, the sound wave’s characteristics can be represented on a graph, with amplitude plotted on the vertical axis and time
plotted on the horizontal axis. It will be seen that both positive and negative ranges are shown on the vertical axis: these represent compressions (⫹)
and rarefactions (⫺) of the air. This graph represents the waveform of the
sound. For a moment, a source vibrating in a very simple and regular manner is assumed, in so-called simple harmonic motion, the result of which is
a simple sound wave known as a sine wave. The simplest vibrating systems
oscillate in this way, such as a mass suspended from a spring, or a swinging
pendulum (see also ‘Phase’ below). It will be seen that the frequency (f) is
the inverse of the time between peaks or troughs of the wave (f ⫽ 1/t). So the
shorter the time between oscillations of the source, the higher the frequency.
The human ear is capable of perceiving sounds with frequencies between
3
4
CHAPTER 1: What is Sound?
approximately 20 Hz and 20 kHz (see ‘Frequency perception’, Chapter 2);
this is known as the audio frequency range or audio spectrum.
HOW SOUND TRAVELS IN AIR
Air is made up of gas molecules and has an elastic property (imagine putting
a thumb over the end of a bicycle pump and compressing the air inside –
the air is springy). Longitudinal sound waves travel in air in somewhat the
same fashion as a wave travels down a row of up-ended dominoes after the
first one is pushed over. The half-cycle of compression created by the vibrating source causes successive air particles to be moved in a knock-on effect,
and this is normally followed by a balancing rarefaction which causes a
similar motion of particles in the opposite direction.
It may be appreciated that the net effect of this is that individual air
particles do not actually travel – they oscillate about a fixed point – but
the result is that a wave is formed which appears to move away from the
source. The speed at which it moves away from the source depends on the
density and elasticity of the substance through which it passes, and in air
the speed is relatively slow compared with the speed at which sound travels
through most solids. In air the speed of sound is approximately 340 meters
per second (m s⫺1), although this depends on the temperature of the air. At
freezing point the speed is reduced to nearer 330 m s⫺1. In steel, to give an
example of a solid, the speed of sound is approximately 5100 m s⫺1.
The frequency and wavelength of a sound wave are related very simply
if the speed of the wave (usually denoted by the letter c) is known:
c ⫽ f λ or λ ⫽ c/f
To show some examples, the wavelength of sound in air at 20 Hz (the
low-frequency or LF end of the audio spectrum), assuming normal room
temperature, would be:
λ ⫽ 340 / 20 ⫽ 17 meters
whereas the wavelength of 20 kHz (at the high-frequency or HF end of the
audio spectrum) would be 1.7 cm. Thus it is apparent that the wavelength of
sound ranges from being very long in relation to most natural objects at low
frequencies, to quite short at high frequencies. This is important when considering how sound behaves when it encounters objects – whether the object
acts as a barrier or whether the sound bends around it (see Fact File 1.5).
Simple and Complex Sounds
SIMPLE AND COMPLEX SOUNDS
In the foregoing example, the sound had a simple waveform – it was a
sine wave or sinusoidal waveform – the type which might result from
a very simple vibrating system such as a weight suspended on a spring.
Sine waves have a very pure sound because they consist of energy at only
one frequency, and are often called pure tones. They are not heard very
commonly in real life (although they can be generated electrically) since
most sound sources do not vibrate in such a simple manner. A person
whistling or a recorder (a simple wind instrument) produces a sound
which approaches a sinusoidal waveform. Most real sounds are made
up of a combination of vibration patterns which result in a more complex waveform. The more complex the waveform, the more like noise the
sound becomes, and when the waveform has a highly random pattern the
sound is said to be noise (see ‘Frequency spectra of non-repetitive sounds’,
below).
The important characteristic of sounds which have a definite pitch is
that they are repetitive: that is, the waveform, no matter how complex,
repeats its pattern in the same way at regular intervals. All such waveforms
can be broken down into a series of components known as harmonics,
using a mathematical process called Fourier analysis (after the mathematician Joseph Fourier). Some examples of equivalent line spectra for different waveforms are given in Figure 1.4. This figure shows another way of
depicting the characteristics of the sound graphically – that is, by drawing a so-called line spectrum which shows frequency along the horizontal
axis and amplitude up the vertical axis. The line spectrum shows the relative strengths of different frequency components which make up a sound.
Where there is a line there is a frequency component. It will be noticed that
the more complex the waveform the more complex the corresponding line
spectrum.
For every waveform, such as that shown in Figure 1.3, there is a corresponding line spectrum: waveforms and line spectra are simply two different ways of showing the characteristics of the sound. Figure 1.3 is called
a time-domain plot, whilst the line spectrum is called a frequency-domain
plot. Unless otherwise stated, such frequency-domain graphs in this book
will cover the audio-frequency range, from 20 Hz at the lower end to 20 kHz
at the upper end.
In a reversal of the above breaking-down of waveforms into their component frequencies it is also possible to construct or synthesize waveforms
by adding together the relevant components.
5
CHAPTER 1: What is Sound?
+
Waveform
Line spectrum
Amplitude
(a)
–
f
(linear scale)
Frequency
(b)
+
Amplitude
FIGURE 1.4
Equivalent line spectra
for a selection of simple
waveforms. (a) The sine
wave consists of only
one component at the
fundamental frequency
f. (b) The sawtooth wave
consists of components
at the fundamental and
its integer multiples,
with amplitudes steadily
decreasing. (c) The
square wave consists
of components at
odd multiples of the
fundamental frequency.
–
f
(c)
+
2f
3f 4f
5f
(linear scale)
6f
7f
Frequency
Amplitude
6
–
f
3f
5f
(linear scale)
7f
Frequency
FREQUENCY SPECTRA OF REPETITIVE SOUNDS
As will be seen in Figure 1.4, the simple sine wave has a line spectrum
consisting of only one component at the frequency of the sine wave. This
is known as the fundamental frequency of oscillation. The other repetitive
waveforms, such as the square wave, have a fundamental frequency as well
as a number of additional components above the fundamental. These are
known as harmonics, but may also be referred to as overtones or partials.
Harmonics are frequency components of a sound which occur at integer
multiples of the fundamental frequency, that is at twice, three times, four
times and so on. Thus a sound with a fundamental of 100 Hz might also
contain harmonics at 200 Hz, 400 Hz and 600 Hz. The reason for the existence of these harmonics is that most simple vibrating sources are capable
Frequency Spectra of Repetitive Sounds
FIGURE 1.5
Modes of vibration of a
stretched string.
(a) Fundamental.
(b) Second harmonic.
(c) Third harmonic.
(a)
(b)
Antinode
Node
(c)
of vibrating in a number of harmonic modes at the same time. Consider a
stretched string, as shown in Figure 1.5. It may be made to vibrate in any
of a number of modes, corresponding to integer multiples of the fundamental frequency of vibration of the string (the concept of ‘standing waves’ is
introduced below). The fundamental corresponds to the mode in which the
string moves up and down as a whole, whereas the harmonics correspond
to modes in which the vibration pattern is divided into points of maximum
and minimum motion along the string (these are called antinodes and
nodes). It will be seen that the second mode involves two peaks of vibration, the third mode three peaks, and so on.
In accepted terminology, the fundamental is also the first harmonic, and
thus the next component is the second harmonic, and so on. Confusingly,
the second harmonic is also known as the first overtone. For the waveforms
shown in Figure 1.4, the fundamental has the highest amplitude, and the
amplitudes of the harmonics decrease with increasing frequency, but this
will not always be the case with real sounds since many waveforms have
line spectra which show the harmonics to be higher in amplitude than the
fundamental. It is also quite feasible for there to be harmonics missing in
the line spectrum, and this depends entirely on the waveform in question.
It is also possible for there to be overtones in the frequency spectrum of
a sound which are not related in a simple integer-multiple fashion to the
fundamental. These cannot correctly be termed harmonics, and they are
more correctly referred to as overtones or inharmonic partials. They tend
7
8
CHAPTER 1: What is Sound?
to arise in vibrating sources which have a complicated shape, and which
do not vibrate in simple harmonic motion but have a number of repetitive modes of vibration. Their patterns of oscillation are often unusual,
such as might be observed in a bell or a percussion instrument. It is still
possible for such sounds to have a recognizable pitch, but this depends on
the strength of the fundamental. In bells and other such sources, one often
hears the presence of several strong inharmonic overtones.
FREQUENCY SPECTRA OF NON-REPETITIVE SOUNDS
Non-repetitive waveforms do not have a recognizable pitch and sound
noise-like. Their frequency spectra are likely to consist of a collection of
components at unrelated frequencies, although some frequencies may be
more dominant than others. The analysis of such waves to show their frequency spectra is more complicated than with repetitive waves, but is still
possible using a mathematical technique called Fourier transformation, the
result of which is a frequency-domain plot of a time-domain waveform.
Single, short pulses can be shown to have continuous frequency spectra
which extend over quite a wide frequency range, and the shorter the pulse
the wider its frequency spectrum but usually the lower its total energy (see
Figure 1.6). Random waveforms will tend to sound like hiss, and a completely random waveform in which the frequency, amplitude and phase of
components are equally probable and constantly varying is called white
noise. A white noise signal’s spectrum is flat, when averaged over a period
of time, right across the audio-frequency range (and theoretically above it).
White noise has equal energy for a given bandwidth, whereas another type
of noise, known as pink noise, has equal energy per octave. For this reason
white noise sounds subjectively to have more high-frequency energy than
pink noise.
PHASE
Two waves of the same frequency are said to be ‘in phase’ when their compression (positive) and rarefaction (negative) half-cycles coincide exactly in
time and space (see Figure 1.7). If two in-phase signals of equal amplitude
are added together, or superimposed, they will sum to produce another signal of the same frequency but twice the amplitude. Signals are said to be
out of phase when the positive half-cycle of one coincides with the negative
half-cycle of the other. If these two signals are added together they will cancel each other out, and the result will be no signal.
Phase
(a)
Waveform
Amplitude
+
Continuous spectrum
–
(linear scale)
Frequency
(linear scale)
Frequency
+
Amplitude
(b)
–
Clearly, these are two extreme cases, and it is entirely possible to superimpose two sounds of the same frequency which are only partially in phase
with each other. The resultant wave in this case will be a partial addition or
partial cancelation, and the phase of the resulting wave will lie somewhere
between that of the two components (see Figure 1.7(c)).
Phase differences between signals can be the result of time delays
between them. If two identical signals start out at sources equidistant from
a listener at the same time as each other then they will be in phase by the
time they arrive at the listener. If one source is more distant than the other
then it will be delayed, and the phase relationship between the two will
depend upon the amount of delay (see Figure 1.8). A useful rule-of-thumb
is that sound travels about 30 cm (1 foot) per millisecond, so if the second source in the above example were 1 meter (just over 3 ft) more distant
than the first it would be delayed by just over 3 ms. The resulting phase
relationship between the two signals, it may be appreciated, would depend
on the frequency of the sound, since at a frequency of around 330 Hz the
3 ms delay would correspond to one wavelength and thus the delayed signal
would be in phase with the undelayed signal. If the delay had been half this
(1.5 ms) then the two signals would have been out of phase at 330 Hz.
Phase is often quoted as a number of degrees relative to some reference, and this must be related back to the nature of a sine wave. A diagram is the best way to illustrate this point, and looking at Figure 1.9 it
FIGURE 1.6
Frequency spectra of nonrepetitive waveforms.
(a) Pulse. (b) Noise.
9
10
CHAPTER 1: What is Sound?
(a)
+
–
+
}
+
equals
–
–
(b)
+
–
+
}
+
equals
–
–
(c)
+
–
+
}
+
equals
–
–
FIGURE 1.7 (a) When two identical in-phase waves are added together, the result is a wave of the
same frequency and phase but twice the amplitude. (b) Two identical out-of-phase waves add to give
nothing. (c) Two identical waves partially out of phase add to give a resultant wave with a phase and
amplitude which is the point-by-point sum of the two.
Phase
+
Speaker 2
Time
–
X
Delay of speaker 2 is
t2 – t1
t2
t1
t2
Wave leaves both speakers at
time X, and arrives at listener
at times t1 and t2 respectively
FIGURE 1.8
If the two loudspeakers in
the drawing emit the same
wave at the same time, the
phase difference between
the waves at the listener’s
ear will be directly related to
the delay t1 ⫺ t2.
Speaker 1
t1
α
180°
0°
Height of spot
90°
+
90°
270°
α
–
270°
will be seen that a sine wave may be considered as a graph of the vertical
position of a rotating spot on the outer rim of a disc (the amplitude of the
wave), plotted against time. The height of the spot rises and falls regularly
as the circle rotates at a constant speed. The sine wave is so called because
the spot’s height is directly proportional to the mathematical sine of the
angle of rotation of the disc, with zero degrees occurring at the origin of
the graph and at the point shown on the disc’s rotation in the diagram.
The vertical amplitude scale on the graph goes from minus one (maximum
negative amplitude) to plus one (maximum positive amplitude), passing
through zero at the halfway point. At 90° of rotation the amplitude of the
sine wave is maximum positive (the sine of 90° is 1), and at 180° it is zero
(sin 180° ⫽ 0). At 270° it is maximum negative (sin 270° ⫽ ⫺1), and at
FIGURE 1.9
The height of the spot
varies sinusoidally with
the angle of rotation of
the wheel. The phase
angle of a sine wave can
be understood in terms of
the number of degrees of
rotation of the wheel.
11
12
CHAPTER 1: What is Sound?
360° it is zero again. Thus in one cycle of the sine wave
the circle has passed through 360° of rotation.
It is now possible to go back to the phase relationship
between two waves of the same frequency. If each cycle is
90°
270°
0°
considered as corresponding to 360°, then one can say just
how many degrees one wave is ahead of or behind another
by comparing the 0° point on one wave with the 0° point
on the other (see Figure 1.10). In the example wave 1 is
90° out of phase with wave 2. It is important to realize
that phase is only a relevant concept in the case of continuous repetitive waveforms, and has little meaning in the
0°
90°
case of impulsive or transient sounds where time difference is the more relevant quantity. It can be deduced from
the foregoing discussion that (a) the higher the frequency,
FIGURE 1.10 The lower wave is 90° out of the greater the phase difference which would result from a
phase with the upper wave.
given time delay between two signals, and (b) it is possible
for there to be more than 360° of phase difference between two signals if
the delay is great enough to delay the second signal by more than one cycle.
In the latter case it becomes difficult to tell how many cycles of delay have
elapsed unless a discontinuity arises in the signal, since a phase difference
of 360° is indistinguishable from a phase difference of 0°.
SOUND IN ELECTRICAL FORM
Although the sound that one hears is due to compression and rarefaction
of the air, it is often necessary to convert sound into an electrical form in
order to perform operations on it such as amplification, recording and mixing. As detailed in Fact File 3.1 and Chapter 3, it is the job of the microphone to convert sound from an acoustical form into an electrical form.
The process of conversion will not be described here, but the result is
important because if it can be assumed for a moment that the microphone
is perfect then the resulting electrical waveform will be exactly the same
shape as the acoustical waveform which caused it.
The equivalent of the amplitude of the acoustical signal in electrical
terms is the voltage of the electrical signal. If the voltage at the output of a
microphone were to be measured whilst the microphone was picking up an
acoustical sine wave, one would measure a voltage which changed sinusoidally as well. Figure 1.11 shows this situation, and it may be seen that an
acoustical compression of the air corresponds to a positive-going voltage,
whilst an acoustical rarefaction of the air corresponds to a negative-going
Sound in Electrical Form
Acoustic
pressure
waves
Varying
electrical
voltage
Microphone
+
Compress
+
Positive
0
0
Rarefy
–
Negative
–
voltage. (This is the norm, although some sound reproduction systems
introduce an absolute phase reversal in the relationship between acoustical
phase and electrical phase, such that an acoustical compression becomes
equivalent to a negative voltage. Some people claim to be able to hear the
difference.)
The other important quantity in electrical terms is the current flowing
down the wire from the microphone. Current is the electrical equivalent of
the air particle motion discussed in ‘How sound travels in air ’, above. Just
as the acoustical sound wave was carried in the motion of the air particles,
so the electrical sound wave is carried in the motion of tiny charge carriers which reside in the metal of a wire (these are called electrons). When
the voltage is positive the current moves in one direction, and when it is
negative the current moves in the other direction. Since the voltage generated by a microphone is repeatedly alternating between positive and
negative, in sympathy with the sound wave’s compression and rarefaction
cycles, the current similarly changes direction each half cycle. Just as the
air particles in ‘Characteristics of a sound wave’, above, did not actually go
anywhere in the long term, so the electrons carrying the current do not go
anywhere either – they simply oscillate about a fixed point. This is known
as alternating current or AC.
A useful analogy to the above (both electrical and acoustical) exists
in plumbing. If one considers water in a pipe fed from a header tank, as
shown in Figure 1.12, the voltage is equivalent to the pressure of water
which results from the header tank, and the current is equivalent to the
rate of flow of water through the pipe. The only difference is that the diagram is concerned with a direct current situation in which the direction
of flow is not repeatedly changing. The quantity of resistance should be
introduced here, and is analogous to the diameter of the pipe. Resistance
impedes the flow of water through the pipe, as it does the flow of electrons
through a wire and the flow of acoustical sound energy through a substance.
FIGURE 1.11
A microphone converts
variations in acoustical
sound pressure into
variations in electrical
voltage. Normally, a
compression of the air
results in a positive voltage
and a rarefaction results in
a negative voltage.
13
14
CHAPTER 1: What is Sound?
FIGURE 1.12
There are parallels between
the flow of water in a pipe
and the flow of electricity
in a wire, as shown in this
drawing.
Water pressure
≡ Electrical voltage
Header tank
Diameter of pipe
≡ Electrical resistance
Outlet
pipe
Rate of flow
≡ Electrical current
FA C T F I L E 1 . 1 O H M’S LAW
Ohm’s law states that there is a fixed and simple relationship between the current flowing through a device (I), the
voltage across it (V ), and its resistance (R), as shown in
the diagram:
or:
There is also a relationship between the parameters
above and the power in watts (W) dissipated in a device:
W ⫽ I 2R ⫽ V 2 /R
V ⫽ IR
I ⫽ V /R
or:
R ⫽ V/I
V
Thus if the resistance of a device is known, and the
voltage dropped across it can be measured, then the current flow may be calculated, for example.
R
I
For a fixed voltage (or water pressure in this analogy), a high resistance
(narrow pipe) will result in a small current (a trickle of water), whilst a
low resistance (wide pipe) will result in a large current. The relationship
between voltage, current and resistance was established by Ohm, in the
form of Ohm’s law, as described in Fact File 1.1. There is also a relationship between power and voltage, current and resistance.
In AC systems, resistance is replaced by impedance, a complex term
which contains both resistance and reactance components. The reactance
part varies with the frequency of the signal; thus the impedance of an electrical device also varies with the frequency of a signal. Capacitors (basically
two conductive plates separated by an insulator) are electrical devices which
present a high impedance to low-frequency signals and a low impedance
to high-frequency signals. They will not pass direct current. Inductors
Displaying the Characteristics of a Sound Wave
(a)
(b)
FIGURE 1.13 (a) An oscilloscope displays the waveform of an electric signal by means of a
moving spot which is deflected up by a positive signal and down by a negative signal. (b) A spectrum
analyzer displays the frequency spectrum of an electrical waveform in the form of lines representing the
amplitudes of different spectral components of the signal.
(basically coils of wire) are electrical devices which present a high impedance to high-frequency signals and a low impedance to low-frequency
signals. Capacitance is measured in farads, inductance in henrys.
DISPLAYING THE CHARACTERISTICS OF A SOUND WAVE
Two devices can be introduced at this point which illustrate graphically the
various characteristics of sound signals so far described. It would be useful to (a) display the waveform of the sound, and (b) display the frequency
spectrum of the sound. In other words (a) the time-domain signal and (b)
the frequency-domain signal.
An oscilloscope is used for displaying the waveform of a sound, and a
spectrum analyzer is used for showing which frequencies are contained in
the signal and their amplitudes. Examples of such devices are pictured in
Figure 1.13. Both devices accept sound signals in electrical form and display their analyses of the sound on a screen. The oscilloscope displays a
moving spot which scans horizontally at one of a number of fixed speeds
from left to right and whose vertical deflection is controlled by the voltage
of the sound signal (up for positive, down for negative). In this way it plots
the waveform of the sound as it varies with time. Many oscilloscopes have
two inputs and can plot two waveforms at the same time, and this can be
useful for comparing the relative phases of two signals (see ‘Phase’, above).
The spectrum analyzer works in different ways depending on the
method of spectrum analysis. A real-time analyzer displays a constantly
updating line spectrum, similar to those depicted earlier in this chapter,
and shows the frequency components of the input signal on the horizontal
scale together with their amplitudes on the vertical scale.
15
16
CHAPTER 1: What is Sound?
THE DECIBEL
The unit of the decibel is used widely in sound engineering, often in preference to other units such as volts, watts, or other such absolute units,
since it is a convenient way of representing the ratio of one signal’s amplitude to another’s. It also results in numbers of a convenient size which
approximate more closely to one’s subjective impression of changes in
the amplitude of a signal, and it helps to compress the range of values
between the maximum and minimum sound levels encountered in real signals. For example, the range of sound intensities (see next section) which
can be handled by the human ear covers about 14 powers of ten, from
0.000 000 000 001 W m⫺2 to around 100 W m⫺2, but the equivalent range in
decibels is only from 0 to 140 dB.
Some examples of the use of the decibel are given in Fact File 1.2. The
relationship between the decibel and human sound perception is discussed
in more detail in Chapter 2. Operating levels in recording equipment are
discussed further in ‘Metering systems’, Chapter 5 and ‘Magnetic recording
levels’, Chapter 6.
Decibels are not only used to describe the ratio between two signals,
or the level of a signal above a reference, they are also used to describe the
voltage gain of a device. For example, a microphone amplifier may have a
gain of 60 dB, which is the equivalent of multiplying the input voltage by a
factor of 1000, as shown in the example below:
20 log 1000 / 1 ⫽ 60 dB
SOUND POWER AND SOUND PRESSURE
A simple sound source, such as the pulsating sphere used at the start of
this chapter, radiates sound power omnidirectionally – that is, equally in
all directions, rather like a three-dimensional version of the ripples moving
away from a stone dropped in a pond. The sound source generates a certain
amount of power, measured in watts, which is gradually distributed over
an increasingly large area as the wavefront travels further from the source;
thus the amount of power per square meter passing through the surface of
the imaginary sphere surrounding the source gets smaller with increasing
distance (see Fact File 1.3). For practical purposes the intensity of the direct
sound from a source drops by 6 dB for every doubling in distance from the
source (see Figure 1.14).
Sound Power and Sound Pressure
FA C T F I L E 1 . 2 T H E D E C IB E L
Basic Decibels
The decibel is based on the logarithm of the ratio between
two numbers. It describes how much larger or smaller one
value is than the other. It can also be used as an absolute
unit of measurement if the reference value is fixed and
known. Some standardized references have been established for decibel scales in different fields of sound engineering (see below).
The decibel is strictly ten times the logarithm to the
base ten of the ratio between the powers of two signals:
dB ⫽ 10 log10 (P1/P2 )
For example, the difference in decibels between
a signal with a power of 1 watt and one of 2 watts is
10 log(2/1) ⫽ 3 dB.
If the decibel is used to compare values other than
signal powers, the relationship to signal power must be
taken into account. Voltage has a square relationship to
power (from Ohm’s law: W ⫽ V 2/R); thus to compare two
voltages:
dB ⫽ 10 log(V12 /V22 ), or 10 log(V1 /V2 )2 ,
or 20 log(V1 /V2 )
For example, the difference in decibels between
a signal with a voltage of 1 volt and one of 2 volts is
20 log(2/1) ⫽ 6 dB. So a doubling in voltage gives rise to
an increase of 6 dB, and a doubling in power gives rise
to an increase of 3 dB. A similar relationship applies to
acoustical sound pressure (analogous to electrical voltage) and sound power (analogous to electrical power).
Decibels with a Reference
If a signal level is quoted in decibels, then a reference
must normally be given, otherwise the figure means nothing; e.g. ‘Signal level ⫽ 47 dB’ cannot have a meaning
unless one knows that the signal is 47 dB above a known
point. ‘ ⫹8 dB ref. 1 volt’ has a meaning since one now
knows that the level is 8 dB higher than 1 volt, and thus
one could calculate the voltage of the signal.
There are exceptions in practice, since in some fields
a reference level is accepted as implicit. Sound pressure
levels (SPLs) are an example, since the reference level is
defined worldwide as 2 ⫻ 10⫺5 N m⫺2 (20 μPa). Thus to
state ‘SPL ⫽ 77 dB’ is probably acceptable, although confusion can still arise due to misunderstandings over such
things as weighting curves (see Fact File 1.4). In sound
recording, 0 dB or ‘zero level’ is a nominal reference level
used for aligning equipment and setting recording levels,
often corresponding to 0.775 volts (0 dBu) although this
is subject to variations in studio centers in different locations. (Some studios use 4 dBu as their reference level, for
example.) ‘0 dB’ does not mean ‘no signal’, it means that
the signal concerned is at the same level as the reference.
Often a letter is placed after ‘dB’ to denote the reference standard in use (e.g. ‘dBm’), and a number of standard abbreviations are in use, some examples of which
are given below. Sometimes the suffix denotes a particular
frequency weighting characteristic used in the measurement of noise (e.g. ‘dBA’).
Abbrev.
dBV
dBu
dBv
dBm
dBA
Ref. level
1 volt
0.775 volt (Europe)
0.775 volt (USA)
1 milliwatt (see Chapter 12)
dB SPL, A-weighted response
Useful Decibel Ratios to Remember (Voltages or SPLs)
It is more common to deal in terms of voltage or SPL
ratios than power ratios in audio systems. Here are some
useful dB equivalents of different voltage or SPL relationships and multiplication factors:
dB
0
⫹3
⫹6
⫹20
⫹60
Multiplication factor
1
2
2
10
1000
17
18
CHAPTER 1: What is Sound?
FA C T F I L E 1 . 3 T H E IN V E R S E -S QUARE LAW
The law of decreasing power per unit area (intensity) of
a wavefront with increasing distance from the source is
known as the inverse-square law, because intensity drops
in proportion to the inverse square of the distance from
the source. Why is this? It is because the sound power
from a point source is spread over the surface area of a
sphere (S), which from elementary math is given by:
The sound intensity level (SIL) of this signal
in decibels can be calculated by comparing it with the
accepted reference level of 10⫺12 W m⫺2:
SIL(dB) ⫽ 10 log((5 ⫻ 10⫺4 ) ⫼ (10⫺12 )) ⫽ 87 dB
S ⫽ 4πr 2
where r is the distance from the source or the radius of
the sphere, as shown in the diagram.
If the original power of the source is W watts, then the
intensity, or power per unit area (I) at distance r, is:
r
Source
Power = W
I ⫽ W/4πr 2
For example, if the power of a source was 0.1 watt,
the intensity at 4 m distance would be:
1 m2
I ⫽ 0.1 ⫼ (4 ⫻ 3.14 ⫻ 16) ⫽ 0.0005 W m⫺2
Surface area of sphere = S
4m2
1m2
Source
r
r
FIGURE 1.14 The sound power which had passed through 1 m2 of space at distance r from the
source will pass through 4 m2 at distance 2r, and thus will have one quarter of the intensity.
The amount of acoustical power generated by real sound sources is
surprisingly small, compared with the number of watts of electrical power
involved in lighting a light bulb, for example. An acoustical source radiating 20 watts would produce a sound pressure level close to the threshold of
pain if a listener was close to the source. Most everyday sources generate
Free and Reverberant Fields
fractions of a watt of sound power, and this energy is eventually dissipated
into heat by absorption (see below). The amount of heat produced by the
dissipation of acoustic energy is relatively insignificant – the chances of
increasing the temperature of a room by shouting are slight, at least in the
physical sense.
Acoustical power is sometimes confused with the power output of an
amplifier used to drive a loudspeaker, and audio engineers will be familiar with power outputs from amplifiers of many hundreds of watts. It is
important to realize that loudspeakers are very inefficient devices – that
is, they only convert a small proportion of their electrical input power into
acoustical power. Thus, even if the input to a loudspeaker was to be, say,
100 watts electrically, the acoustical output power might only be perhaps
1 watt, suggesting a loudspeaker that is only 1% efficient. The remaining
power would be dissipated as heat in the voice coil.
Sound pressure is the effect of sound power on its surroundings. To
use a central heating analogy, sound power is analogous to the heat energy
generated by a radiator into a room, whilst sound pressure is analogous to
the temperature of the air in the room. The temperature is what a person
entering the room would feel, but the heat-generating radiator is the source
of power. Sound pressure level (SPL) is measured in newtons per square
meter (N m⫺2). A convenient reference level is set for sound pressure and
intensity measurements, this being referred to as 0 dB. This level of 0 dB
is approximately equivalent to the threshold of hearing (the quietest sound
perceivable by an average person) at a frequency of 1 kHz, and corresponds
to an SPL of 2 ⫻ 10⫺5 N m⫺2, which in turn is equivalent to an intensity of
approximately 10⫺12 W m⫺2 in the free field (see below).
Sound pressure levels are often quoted in dB (e.g. SPL ⫽ 63 dB means
that the SPL is 63 dB above 2 ⫻ 10⫺5 N m⫺2). The SPL in dB may not accurately represent the loudness of a sound, and thus a subjective unit of loudness has been derived from research data, called the phon. This is discussed
further in Chapter 2. Some methods of measuring sound pressure levels are
discussed in Fact File 1.4.
FREE AND REVERBERANT FIELDS
The free field in acoustic terms is an acoustical area in which there are no
reflections. Truly free fields are rarely encountered in reality, because there
are nearly always reflections of some kind, even if at a very low level. If
the reader can imagine the sensation of being suspended out of doors, way
above the ground, away from any buildings or other surfaces, then he or
19
20
CHAPTER 1: What is Sound?
FA C T F I L E 1 . 4 ME A S U R IN G S P Ls
Typically, a sound pressure level (SPL) meter is used to
measure the level of sound at a particular point. It is a
device that houses a high-quality omnidirectional (pressure)
microphone (see ‘Omnidirectional pattern’, Chapter 3) connected to amplifiers, filters and a meter (see diagram).
Weighting Filters
The microphone’s output voltage is proportional to the
SPL incident upon it, and the weighting filters may be
used to attenuate low and high frequencies according to
a standard curve such as the ‘A’-weighting curve, which
corresponds closely to the sensitivity of human hearing
at low levels (see Chapter 2). SPLs quoted simply in dB
are usually unweighted – in other words all frequencies
are treated equally – but SPLs quoted in dBA will have
been A-weighted and will correspond more closely to the
perceived loudness of the signal. A-weighting was originally designed to be valid up to a loudness of 55 phons,
since the ear’s frequency response becomes flatter at
higher levels; between 55 and 85 phons the ‘B’ curve was
intended to be used; above 85 phons the ‘C’ curve was
used. The ‘D’ curve was devised particularly for measuring aircraft engine noise at very high levels.
Now most standards suggest that the ‘A’ curve may
be used for measuring noise at any SPL, principally for
ease of comparability of measurements, but there is still
disagreement in the industry about the relative merits of
different curves. The ‘A’ curve attenuates low and high
frequencies and will therefore under-read quite substantially for signals at these frequencies. This is an advantage
Mic
in some circumstances and a disadvantage in others. The
‘C’ curve is recommended in the USA and Japan for aligning sound levels using noise signals in movie theaters, for
example. This only rolls off the very extremes of the audio
spectrum and is therefore quite close to an unweighted
reading. Some researchers have found that the ‘B’ curve
produces results that more closely relate measured sound
signal levels to subjective loudness of those signals.
Noise Criterion or Rating (NC or NR)
Noise levels are often measured in rooms by comparing
the level of the noise across the audible range with a standard set of curves called the noise criteria (NC) or noise
rating (NR) curves. These curves set out how much noise
is acceptable in each of a number of narrow frequency
bands for the noise to meet a certain criterion. The noise
criterion is then that of the nearest curve above which
none of the measured results rises. NC curves are used
principally in the USA, whereas NR curves are used principally in Europe. They allow considerably higher levels in
low-frequency bands than in middle- and high-frequency
bands, since the ear is less sensitive at low frequencies.
In order to measure the NC or NR of a location it is
necessary to connect the measuring microphone to a set
of filters or a spectrum analyzer which is capable of displaying the SPL in one octave or one-third octave bands.
Further Reading
British Standard 5969. Specification for sound level meters.
British Standard 6402. Sound exposure meters.
Amplifier
Amplifier
Weighting
filters
Meter
Free and Reverberant Fields
she will have an idea of the experience of a free-field condition. The result
is an acoustically ‘dead’ environment. Acoustic experiments are sometimes
performed in anechoic chambers, which are rooms specially treated so as
to produce almost no reflections at any frequency – the surfaces are totally
absorptive – and these attempt to create near free-field conditions.
In the free field all the sound energy from a source is radiated away
from the source and none is reflected; thus the inverse-square law (Fact File
1.3) entirely dictates the level of sound at any distance from the source. Of
course the source may be directional, in which case its directivity factor
must be taken into account. A source with a directivity factor of 2 on its
axis of maximum radiation radiates twice as much power in this direction
as it would have if it had been radiating omnidirectionally. The directivity index of a source is measured in dB, giving the above example a directivity index of 3 dB. If calculating the intensity at a given distance from a
directional source (as shown in Fact File 1.3), one must take into account
its directivity factor on the axis concerned by multiplying the power of the
source by the directivity factor before dividing by 4πr2.
In a room there is both direct and reflected sound. At a certain distance
from a source contained within a room the acoustic field is said to be diffuse or reverberant, since reflected sound energy predominates over direct
sound. A short time after the source has begun to generate sound a diffuse pattern of reflections will have built up throughout the room, and the
reflected sound energy will become roughly constant at any point in the
room. Close to the source the direct sound energy is still at quite a high
level, and thus the reflected sound makes a smaller contribution to the
total. This region is called the near field. (It is popular in sound recording to make use of so-called ‘near-field monitors’, which are loudspeakers
mounted quite close to the listener, such that the direct sound predominates over the effects of the room.)
The exact distance from a source at which a sound field becomes dominated by reverberant energy depends on the reverberation time of the room,
and this in turn depends on the amount of absorption in the room, and the
room’s volume (see Fact File 1.5). Figure 1.15 shows how the SPL changes
as distance increases from a source in three different rooms. Clearly, in
the acoustically ‘dead’ room, the conditions approach that of the free field
(with sound intensity dropping at close to the expected 6 dB per doubling
in distance), since the amount of reverberant energy is very small. The
critical distance at which the contribution from direct sound equals that
from reflected sound is further from the source than when the room is very
reverberant. In the reverberant room the sound pressure level does not
change much with distance from the source because reflected sound energy
21
CHAPTER 1: What is Sound?
FA C T F I L E 1. 5 A B S O R P T IO N , REFLECTI ON AND RT
Absorption
Reverberation Time
When a sound wave encounters a surface some of its
energy is absorbed and some reflected. The absorption
coefficient of a substance describes, on a scale from 0
to 1, how much energy is absorbed. An absorption coefficient of 1 indicates total absorption, whereas 0 represents
total reflection. The absorption coefficient of substances
varies with frequency.
The total amount of absorption present in a room can
be calculated by multiplying the absorption coefficient of
each surface by its area and then adding the products
together. All of the room’s surfaces must be taken into
account, as must people, chairs and other furnishings.
Tables of the performance of different substances are
available in acoustics references (see ‘Recommended further reading’). Porous materials tend to absorb high frequencies more effectively than low frequencies, whereas
resonant membrane- or panel-type absorbers tend to be
better at low frequencies. Highly tuned artificial absorbers
(Helmholtz absorbers) can be used to remove energy in
a room at specific frequencies. The trends in absorption
coefficient are shown in the diagram below.
W.C. Sabine developed a simple and fairly reliable formula for calculating the reverberation time (RT60) of a
room, assuming that absorptive material is distributed
evenly around the surfaces. It relates the volume of the
room (V) and its total absorption (A) to the time taken for
the sound pressure level to decay by 60 dB after a sound
source is turned off.
RT60 ⫽ (0.16V )/A seconds
In a large room where a considerable volume of air
is present, and where the distance between surfaces is
large, the absorption of the air becomes more important,
in which case an additional component must be added to
the above formula:
RT60 ⫽ (0.16V )/(A ⫹ xV ) seconds
where x is the absorption factor of air, given at various
temperatures and humidities in acoustics references.
The Sabine formula has been subject to modifications
by such people as Eyring, in an attempt to make it more
reliable in extreme cases of high absorption, and it should
be realized that it can only be a guide.
Reflection
The size of an object in relation to the wavelength of a
sound is important in determining whether the sound
wave will bend round it or be reflected by it. When an
object is large in relation to the wavelength the object
will act as a partial barrier to the sound, whereas when
it is small the sound will bend or diffract around it. Since
sound wavelengths in air range from approximately 18
meters at low frequencies to just over 1 cm at high frequencies, most commonly encountered objects will tend
to act as barriers to sound at high frequencies but will
have little effect at low frequencies.
1
Absorption coefficient
22
0
Helmholtz
Porous
Membrane
Frequency
Standing Waves
Rev. field level
0
Direct + rev. field level
Sound intensity level (dB)
–6
–12
–18
Reverberant room
–24
–30
–36
Average room
–42
–48
‘Dead’ or ‘dry’ room
Distance from source
predominates after only a short distance. This is important in room design,
since although a short reverberation time may be desirable in a recording
control room, for example, it has the disadvantage that the change in SPL
with distance from the speakers will be quite severe, requiring very highly
powered amplifiers and heavy-duty speakers to provide the necessary level.
A slightly longer reverberation time makes the room less disconcerting to
work in, and relieves the requirement on loudspeaker power.
STANDING WAVES
The wavelength of sound varies considerably over the audible frequency
range, as indicated in Fact File 1.5. At high frequencies, where the wavelength is small, it is appropriate to consider a sound wavefront rather like
light – as a ray. Similar rules apply, such as the angle of incidence of a
sound wave to a wall is the same as the angle of reflection. At low frequencies where the wavelength is comparable with the dimensions of the room
it is necessary to consider other factors, since the room behaves more as
a complex resonator, having certain frequencies at which strong pressure
peaks and dips are set up in various locations.
Standing waves or eigentones (sometimes also called room modes) may
be set up when half the wavelength of the sound or a multiple is equal to
one of the dimensions of the room (length, width or height). In such a case
(see Figure 1.16) the reflected wave from the two surfaces involved is in
FIGURE 1.15
As the distance from a
source increases direct
sound level drops but
reverberant sound level
remains roughly constant.
The resultant sound
level experienced at
different distances from
the source depends on
the reverberation time
of the room, since in a
reverberant room the level
of reflected sound is higher
than in a ‘dead’ room.
23
24
CHAPTER 1: What is Sound?
Wall
λ/2
Wall
Max. sound pressure
Min. sound pressure
FIGURE 1.16 When a standing wave is set up between two walls of a room there arise points of
maximum and minimum pressure. The first simple mode or eigentone occurs when half the wavelength
of the sound equals the distance between the boundaries, as illustrated, with pressure maxima at the
boundaries and a minimum in the center.
phase with the incident wave and a pattern of summations and cancelations
is set up, giving rise to points in the room at which the sound pressure is
very high, and other points where it is very low. For the first mode (pictured),
there is a peak at the two walls and a trough in the center of the room. It is
easy to experience such modes by generating a low-frequency sine tone into
a room from an oscillator connected to an amplifier and loudspeaker placed
in a corner. At selected low frequencies the room will resonate strongly and
the pressure peaks may be experienced by walking around the room. There
are always peaks towards the boundaries of the room, with troughs distributed at regular intervals between them. The positions of these depend on
whether the mode has been created between the walls or between the floor
and ceiling. The frequencies (f) at which the strongest modes will occur is
given by:
f ⫽ (c/ 2) ⫻ (n/d)
where c is the speed of sound, d is the dimension involved (distance
between walls or floor and ceiling), and n is the number of the mode.
A more complex formula can be used to predict the frequencies of all
the modes in a room, including those secondary modes formed by reflections between four and six surfaces (oblique and tangential modes).
The secondary modes typically have lower amplitudes than the primary
Standing Waves
modes (the axial modes) since they experience greater absorption. The formula is:
f ⫽ (c/ 2) (( p/L)2 ⫹ (q/W )2 ⫹ ( r/H)2 )
where p, q and r are the mode numbers for each dimension (1, 2, 3 …) and
L, W and H are the length, width and height of the room. For example, to
calculate the first axial mode involving only the length, make p ⫽ 1, q ⫽ 0
and r ⫽ 0. To calculate the first oblique mode involving all four walls, make
p ⫽ 1, q ⫽ 1, r ⫽ 0, and so on.
Some quick sums will show, for a given room, that the modes are widely
spaced at low frequencies and become more closely spaced at high frequencies. Above a certain frequency, there arise so many modes per octave that
it is hard to identify them separately. As a rule-of-thumb, modes tend only
to be particularly problematical up to about 200 Hz. The larger the room
the more closely spaced the modes. Rooms with more than one dimension
equal will experience so-called degenerate modes in which modes between
two dimensions occur at the same frequency, resulting in an even stronger
resonance at a particular frequency than otherwise. This is to be avoided.
Since low-frequency room modes cannot be avoided, except by introducing total absorption, the aim in room design is to reduce their effect
by adjusting the ratios between dimensions to achieve an even spacing.
A number of ‘ideal’ mode-spacing criteria have been developed by acousticians, but there is not the space to go into these in detail here. Larger
rooms are generally more pleasing than small rooms, since the mode spacing is closer at low frequencies, and individual modes tend not to stick out
so prominently, but room size has to be traded off against the target reverberation time. Making walls non-parallel does not prevent modes from
forming (since oblique and tangential modes are still possible); it simply
makes their frequencies more difficult to predict.
The practical difficulty with room modes results from the unevenness
in sound pressure throughout the room at mode frequencies. Thus a person sitting in one position might experience a very high level at a particular
frequency whilst other listeners might hear very little. A room with prominent LF modes will ‘boom’ at certain frequencies, and this is unpleasant
and undesirable for critical listening. The response of the room modifies
the perceived frequency response of a loudspeaker, for example, such that
even if the loudspeaker’s own frequency response may be acceptable it may
become unacceptable when modified by the resonant characteristics of the
room.
25
26
CHAPTER 1: What is Sound?
FA C T F I L E 1 . 6 E C H O E S A N D R EFLECTI ONS
Early Reflections
Early reflections are those echoes from nearby surfaces in
a room which arise within the first few milliseconds (up
to about 50 ms) of the direct sound arriving at a listener
from a source (see the diagram). It is these reflections
which give the listener the greatest clue as to the size of
a room, since the delay between the direct sound and the
first few reflections is related to the distance of the major
surfaces in the room from the listener. Artificial reverberation devices allow for the simulation of a number of early
reflections before the main body of reverberant sound
decays, and this gives different reverberation programs
the characteristic of different room sizes.
as a high ceiling or distant rear wall. Strong echoes are
usually annoying in critical listening situations and should
be suppressed by dispersion and absorption.
Flutter Echoes
A flutter echo is sometimes set up when two parallel
reflective surfaces face each other in a room, whilst the
other surfaces are absorbent. It is possible for a wavefront to become ‘trapped’ into bouncing back and forth
between these two surfaces until it decays, and this can
result in a ‘buzzing’ or ‘ringing’ effect on transients (at the
starts and ends of impulsive sounds such as hand claps).
Echoes
Echoes may be considered as discrete reflections of
sound arriving at the listener after about 50 ms from the
direct sound. These are perceived as separate arrivals,
whereas those up to around 50 ms are normally integrated
by the brain with the first arrival, not being perceived consciously as echoes. Such echoes are normally caused by
more distant surfaces which are strongly reflective, such
Early reflection
Source
Listener
Direct
Early reflection
Room modes are not the only results of reflections in enclosed spaces,
and some other examples are given in Fact File 1.6.
RECOMMENDED FURTHER READING
GENERAL ACOUSTICS
Alton Everest, F., 2000. The Master Handbook of Acoustics, fourth ed.
McGraw-Hill.
Benade, A.H., 1991. Fundamentals of Musical Acoustics. Oxford University Press.
Campbell, M., Greated, C., 2001. The Musician’s Guide to Acoustics. Oxford
University Press.
Eargle, J., 1995. Music, Sound, Technology, second ed. Van Nostrand Rheinhold.
Egan, M.D., 1988. Architectural Acoustics. McGraw-Hill.
Hall, D.E., 2001. Musical Acoustics, third ed. Brooks/Cole Publishing Co.
Recommended Further Reading
Howard, D., Angus, J., 2000. Acoustics and Psychoacoustics, second ed. Focal
Press.
Rettinger, M., 1988. Handbook of Architectural Acoustics and Noise Control. TAB
Books.
Rossing, T.D., 2001. The Science of Sound, third ed. Addison-Wesley.
27
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CHAPTER 2
Auditory Perception
CH A P T E R C O N TE N T S
The Hearing Mechanism
Frequency Perception
Loudness Perception
Practical Implications of Equal-loudness Contours
Spatial Perception
Sound source localization
Time-based cues
Amplitude and spectral cues
Effects of reflections
Interaction between hearing and other senses
Resolving conflicting cues
Distance and depth perception
Naturalness in spatial hearing
30
31
33
36
37
37
37
39
42
42
44
44
45
In this chapter the mechanisms by which sound is perceived will be introduced. The human ear often modifies the sounds presented to it before they
are presented to the brain, and the brain’s interpretation of what it receives
from the ears will vary depending on the information contained in the
nervous signals. An understanding of loudness perception is important
when considering such factors as the perceived frequency balance of a reproduced signal, and an understanding of directional perception is relevant to
the study of stereo recording techniques. Below, a number of aspects of the
hearing process will be related to the practical world of sound recording and
reproduction.
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
29
30
CHAPTER 2: Auditory Perception
THE HEARING MECHANISM
Although this is not intended to be a lesson in physiology, it is necessary to
investigate the basic components of the ear, and to look at how information
about sound signals is communicated to the brain. Figure 2.1 shows a diagram of the ear mechanism, not anatomically accurate but showing the key
mechanical components. The outer ear consists of the pinna (the visible
skin and bone structure) and the auditory canal, and is terminated by
the tympanic membrane or ‘ear drum’. The middle ear consists of a threebone lever structure which connects the tympanic membrane to the inner
ear via the oval window (another membrane). The inner ear is a fluid-filled
bony spiral device known as the cochlea, down the center of which runs a
flexible membrane known as the basilar membrane. The cochlea is shown
here as if ‘unwound’ into a straight chamber for the purposes of description. At the end of the basilar membrane, furthest from the middle ear,
there is a small gap called the helicotrema which allows fluid to pass from
the upper to the lower chamber. There are other components in the inner
ear, but those noted above are the most significant.
The ear drum is caused to vibrate in sympathy with the air in the auditory canal when excited by a sound wave, and these vibrations are transferred via the bones of the middle ear to the inner ear, being subject to a
multiplication of force of the order of 15:1 by the lever arrangement of the
bones. The lever arrangement, coupled with the difference in area between
the tympanic membrane and the oval window, helps to match the impedances of the outer and inner ears so as to ensure optimum transfer of
energy. Vibrations are thus transferred to the fluid in the inner ear in which
pressure waves are set up. The basilar membrane is not uniformly stiff
along its length (it is narrow and stiff at the oval window end and wider
and more flexible at the far end), and the fluid is relatively incompressible;
FIGURE 2.1
A simplified mechanical
diagram of the ear.
Outer ear
Pinna
Auditory canal
Tympanic
membrane
Middle
ear
Inner ear
Oval window
Helicotrema
Basilar membrane
Cochlea (fluid filled)
Eustachian tube
Frequency Perception
thus a high-speed pressure wave travels through the fluid and a pressure
difference is created across the basilar membrane.
FREQUENCY PERCEPTION
The motion of the basilar membrane depends considerably on the frequency of the sound wave, there being a peak of motion which moves closer
towards the oval window the higher the frequency (see Figure 2.2).
At low frequencies the membrane has been observed to move as a
whole, with the maximum amplitude of motion at the far end, whilst at
higher frequencies there arises a more well-defined peak. It is interesting
to note that for every octave (i.e. for every doubling in the frequency) the
position of this peak of maximum vibration moves a similar length up
the membrane, and this may explain the human preference for displaying
frequency-related information on a logarithmic frequency scale, which represents an increase in frequency by showing octaves as equal increments
along a frequency axis.
Frequency information is transmitted to the brain in two principal ways.
At low frequencies hair cells in the inner ear are stimulated by the vibrations of the basilar membrane, causing them to discharge small electrical
impulses along the auditory nerve fibers to the brain. These impulses are
found to be synchronous with the sound waveform, and thus the period of
the signal can be measured by the brain. Not all nerve fibers are capable of
discharging once per cycle of the sound waveform (in fact most have spontaneous firing rates of a maximum of 150 Hz with many being much lower
than this). Thus at all but the lowest frequencies the period information is
carried in a combination of nerve fiber outputs, with at least a few firing
on every cycle (see Figure 2.3). There is evidence to suggest that nerve
fibers may re-trigger faster if they are ‘kicked’ harder – that is, the louder
the sound the more regularly they may be made to fire. Also, whilst some
1500 Hz
10
500 Hz
20
50 Hz
30
Distance from oval window (mm)
FIGURE 2.2
The position of maximum
vibration on the basilar
membrane moves towards
the oval window as
frequency increases.
31
32
CHAPTER 2: Auditory Perception
FIGURE 2.3
Although each neurone
does not normally fire on
every cycle of the causatory
sound wave, the outputs of
a combination of neurones
firing on different cycles
represent the period of the
wave.
Original waveform
Neurone 1
Neurone 2
Neurone 3
Neurone 4
Neurone 5
Overall pattern
fibers will trigger with only a low level of stimulation, others will only fire
at high sound levels.
The upper frequency limit at which nerve fibers appear to cease firing
synchronously with the signal is around 4 kHz, and above this frequency
the brain relies increasingly on an assessment of the position of maximum
excitation of the membrane to decide on the pitch of the signal. There is
clearly an overlap region in the middle-frequency range, from about 200 Hz
upwards, over which the brain has both synchronous discharge information
and ‘position’ information on which to base its measurement of frequency.
It is interesting to note that one is much less able to determine the precise
musical pitch of a note when its frequency is above the synchronous discharge limit of 4 kHz.
The frequency selectivity of the ear has been likened to a set of filters,
and this concept is described in more detail in Fact File 2.1. It should be
noted that there is an unusual effect whereby the perceived pitch of a note
is related to the loudness of the sound, such that the pitch shifts slightly
with increasing sound level. This is sometimes noticed as loud sounds
decay, or when removing headphones, for example. The effect of ‘beats’
may also be noticed when two pure tones of very similar frequency are
sounded together, resulting in a pattern of addition and cancelation as they
come in and out of phase with each other. The so-called ‘beat frequency ’
is the difference frequency between the two signals, such that signals at
200 Hz and 201 Hz would result in a cyclic modulation of the overall level,
or beat, at 1 Hz. Combined signals slightly further apart in frequency result
in a ‘roughness’ which disappears once the frequencies of the two signals
are further than a critical band apart.
Loudness Perception
FA C T F I L E 2 . 1 C R IT IC A L B A N D W I DTH
The basilar membrane appears to act as a rough mechanical spectrum analyzer, providing a spectral analysis
of the incoming sound to an accuracy of between onefifth and one-third of an octave in the middle frequency
range (depending on which research data is accepted).
It acts rather like a bank of overlapping filters of a fixed
bandwidth. This analysis accuracy is known as the critical
bandwidth, which is the range of frequencies passed by
each notional filter.
The critical band concept is important in understanding hearing because it helps to explain why some signals
are ‘masked’ in the presence of others (see Fact File 2.3).
Fletcher, working in the 1940s, suggested that only signals
lying within the same critical band as the wanted signal
would be capable of masking it, although other work on
masking patterns seems to suggest that a signal may have
a masking effect on frequencies well above its own.
With complex signals, such as noise or speech, for
example, the total loudness of the signal depends to
some extent on the number of critical bands covered
by a signal. It can be demonstrated by a simple experiment that the loudness of a constant power signal does
not begin to increase until its bandwidth extends over
more than the relevant critical bandwidth, which appears
to support the previous claim. (A useful demonstration
of this phenomenon is to be found on the Compact Disc
entitled Auditory Demonstrations described at the end of
this chapter.)
Although the critical band concept helps to explain
the first level of frequency analysis in the hearing mechanism, it does not account for the fine frequency selectivity
of the ear which is much more precise than one-third of
an octave. One can detect changes in pitch of only a few
hertz, and in order to understand this it is necessary to
look at the ways in which the brain ‘sharpens’ the aural
tuning curves. For this the reader is referred to Moore
(2003), as detailed at the end of this chapter.
LOUDNESS PERCEPTION
The subjective quantity of ‘loudness’ is not directly related to the SPL of a
sound signal (see ‘Sound power and sound pressure’, Chapter 1). The ear
is not uniformly sensitive at all frequencies, and a set of curves has been
devised which represents the so-called equal-loudness contours of hearing (see Fact File 2.2). This is partially due to the resonances of the outer
ear which have a peak in the middle-frequency region, thus increasing the
effective SPL at the ear drum over this range.
The unit of loudness is the phon. If a sound is at the threshold of hearing (just perceivable) it is said to have a loudness of 0 phons, whereas if a
sound is at the threshold of pain it will probably have a loudness of around
140 phons. Thus the ear has a dynamic range of approximately 140 phons,
representing a range of sound pressures with a ratio of around 10 million to
one between the loudest and quietest sounds perceivable. As indicated in
Fact File 1.4, the ‘A’-weighting curve is often used when measuring sound
levels because it shapes the signal spectrum to represent more closely the
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CHAPTER 2: Auditory Perception
FA C T F I L E 2 . 2 E Q U A L-LO U D N ESS CONTOURS
Fletcher and Munson devised a set of curves to show the
sensitivity of the ear at different frequencies across the
audible range. They derived their results from tests on a
large number of subjects who were asked to adjust the
level of test tones until they appeared equally as loud as
a reference tone with a frequency of 1 kHz. The test tones
were spread across the audible spectrum. From these
results could be drawn curves of average ‘equal loudness’, indicating the SPL required at each frequency for a
sound to be perceived at a particular loudness level (see
diagram).
Loudness is measured in phons, the zero phon curve
being that curve which passes through 0 dB SPL at 1 kHz –
in other words, the threshold of hearing curve. All points
SPL (dB ref. 2 x 10 –5 Nm–2 )
34
along the 0 phon curve will sound equally loud, although
clearly a higher SPL is required at extremes of the spectrum
than in the middle. The so-called Fletcher–Munson curves
are not the only equal-loudness curves in existence –
Robinson and Dadson, amongst others, have published
revised curves based upon different test data. The shape
of the curves depends considerably on the type of sound
used in the test, since filtered noise produces slightly different results to sine tones.
It will be seen that the higher-level curves are flatter than the low-level curves, indicating that the ear’s
frequency response changes with signal level. This is
important when considering monitoring levels in sound
recording (see text).
130
120
110
100
90
80
70
60
50
40
30
20
10
0
–10
phons
100
80
60
40
20
0
20
100
1 kHz
10 kHz
Frequency Hz
subjective loudness of low-level signals. A noise level quoted in dBA is very
similar to a loudness level in phons.
To give an idea of the loudnesses of some common sounds, the background noise of a recording studio might be expected to measure at around
20 phons, a low-level conversation perhaps at around 50 phons, a busy office
at around 70 phons, shouted speech at around 90 phons, and a full symphony orchestra playing loudly at around 120 phons. These figures of course
depend on the distance from the sound source, but are given as a guide.
The loudness of a sound depends to a great extent on its nature. Broadband sounds tend to appear louder than narrow-band sounds, because they
Loudness Perception
FA C T F I L E 2 . 3 MA S K IN G
SPL (dB ref. 2 x 10 –5 Nm–2 )
Most people have experienced the phenomenon of masking, although it is often considered to be so obvious that
it does not need to be stated. As an example: it is necessary to raise your voice in order for someone to hear you
if you are in noisy surroundings. The background noise
has effectively raised the perception threshold so that a
sound must be louder before it can be heard. If one looks
at the masking effect of a pure tone, it will be seen that
it raises the hearing threshold considerably for frequencies which are the same as or higher than its own (see
diagram). Frequencies below the masking tone are less
affected. The range of frequencies masked by a tone
depends mostly on the area of the basilar membrane set
into motion by the tone, and the pattern of motion of this
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110
100
90
80
70
60
50
40
30
20
10
0
–10
20
membrane is more extended towards the HF end than
towards the LF end. If the required signal produces more
motion on the membrane than the masking tone produces at that point then it will be perceived.
The phenomenon of masking has many practical uses
in audio engineering. It is used widely in noise reduction
systems, since it allows the designer to assume that lowlevel noise which exists in the same frequency band as a
high-level music signal will be effectively masked by the
music signal. It is also used in digital audio data compression systems, since it allows the designer to use lower
resolution in some frequency bands where the increased
noise will be effectively masked by the wanted signal.
Masking tone
Normal threshold
Altered threshold
100
1 kHz
10 kHz
Frequency Hz
cover more critical bands (see Fact File 2.1), and distorted sounds appear
psychologically to be louder than undistorted sounds, perhaps because one
associates distortion with system overload. If two music signals are played
at identical levels to a listener, one with severe distortion and the other
without, the listener will judge the distorted signal to be louder.
A further factor of importance is that the threshold of hearing is raised
at a particular frequency in the presence of another sound at a similar
frequency. In other words, one sound may ‘mask’ another – a principle
described in more detail in Fact File 2.3.
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CHAPTER 2: Auditory Perception
In order to give the impression of a doubling in perceived loudness, an
increase of some 9–10 dB is required. Although 6 dB represents a doubling
of the actual sound pressure, the hearing mechanism appears to require
a greater increase than this for the signal to appear to be twice as loud.
Another subjective unit, rarely used in practice, is that of the sone: 1 sone
is arbitrarily aligned with 40 phons, and 2 sones is twice as loud as 1 sone,
representing approximately 49 phons; 3 sones is three times as loud, and
so on. Thus the sone is a true indication of the relative loudness of signals
on a linear scale, and sone values may be added together to arrive at the
total loudness of a signal in sones.
The ear is by no means a perfect transducer; in fact it introduces considerable distortions into sound signals due to its non-linearity. At high signal
levels, especially for low-frequency sounds, the amount of harmonic and intermodulation distortion (see Chapter 18) produced by the ear can be high.
PRACTICAL IMPLICATIONS OF EQUAL-LOUDNESS
CONTOURS
The non-linear frequency response of the ear presents the sound engineer with a number of problems. First, the perceived frequency balance of
a recording will depend on how loudly it is replayed, and thus a balance
made in the studio at one level may sound different when replayed in the
home at another. In practice, if a recording is replayed at a much lower
level than that at which it was balanced it will sound lacking in bass and
extreme treble – it will sound thin and lack warmth. Conversely, if a signal
is replayed at a higher level than that at which it was balanced it will have
an increased bass and treble response, sounding boomy and overbright.
A ‘loudness’ control is often provided on hi-fi amplifiers to boost low
and high frequencies for low-level listening, but this should be switched out
at higher levels. Rock-and-roll and heavy-metal music often sounds lacking
in bass when replayed at moderate sound levels because it is usually balanced at extremely high levels in the studio.
Some types of noise will sound louder than others, and hiss is usually
found to be most prominent due to its considerable energy content at middle–
high frequencies. Rumble and hum may be less noticeable because the ear is
less sensitive at low frequencies, and a low-frequency noise which causes large
deviations of the meters in a recording may not sound particularly loud in
reality. This does not mean, of course, that rumble and hum are acceptable.
Recordings equalized to give a strong mid-frequency content often sound
rather ‘harsh’, and listeners may complain of listening fatigue, since the ear
is particularly sensitive in the range between about 1 and 5 kHz.
Spatial Perception
SPATIAL PERCEPTION
Spatial perception principles are important when considering stereo sound
reproduction (see Chapters 16 and 17) and when designing PA rigs for large
auditoria, since an objective in both these cases is to give the illusion of
directionality and spaciousness.
Sound source localization
Most research into the mechanisms underlying directional sound perception
conclude that there are two primary mechanisms at work, the importance of
each depending on the nature of the sound signal and the conflicting environmental cues that may accompany discrete sources. These broad mechanisms involve the detection of timing or phase differences between the ears,
and of amplitude or spectral differences between the ears. The majority of
spatial perception is dependent on the listener having two ears, although
certain monoaural cues have been shown to exist – in other words it is
mainly the differences in signals received by the two ears that matter.
Time-based cues
A sound source located off the 0° (center front) axis will give rise to a time
difference between the signals arriving at the ears of the listener which
is related to its angle of incidence, as shown in Figure 2.4. This rises to
a maximum for sources at the side of the head, and enables the brain to
localize sources in the direction of the earlier ear. The maximum time delay
between the ears is of the order of 650 μs or 0.65 ms and is called the binaural delay. It is apparent that humans are capable of resolving direction down
to a resolution of a few degrees by this method. There is no obvious way
of distinguishing between front and rear sources or of detecting elevation
by this method, but one way of resolving this confusion is by taking into
account the effect of head movements. Front and rear sources at the same
angle of offset from center to one side, for example, will result in opposite
changes in time of arrival for a given direction of head turning.
Time difference cues are particularly registered at the starts and ends
of sounds (onsets and offsets) and seem to be primarily based on the lowfrequency content of the sound signal. They are useful for monitoring the
differences in onset and offset of the overall envelope of sound signals at
higher frequencies.
Timing differences can be expressed as phase differences when considering sinusoidal signals. The ear is sensitive to interaural phase differences
only at low frequencies and the sensitivity to phase begins to deteriorate
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CHAPTER 2: Auditory Perception
FIGURE 2.4
The interaural time
difference (ITD) for a
listener depends on the
angle of incidence of the
source, as this affects the
additional distance that the
sound wave has to travel
to the more distant ear. In
this model the ITD is given
by r(θ ⫹ sin θ)/c (where
c ⫽ 340 m/s, the speed of
sound, and θ is in radians).
Front center
Source
Paths to listener's
two ears
θ
r sinθ
rθ
θ
r
Left
ear
Right
ear
above about 1 kHz. At low frequencies the hair cells in the inner ear fire
regularly at specific points in the phase of the sound cycle, but at high frequencies this pattern becomes more random and not locked to any repeatable point in the cycle. Sound sources in the lateral plane give rise to phase
differences between the ears that depend on their angle of offset from the 0°
axis (center front). Because the distance between the ears is constant, the
phase difference will depend on the frequency and location of the source.
(Some sources also show a small difference in the time delay between
the ears at LF and HF.) Such a phase difference model of directional perception is only really relevant for continuous sine waves auditioned
in anechoic environments, which are rarely heard except in laboratories.
It also gives ambiguous information above about 700 Hz where the distance
between the ears is equal to half a wavelength of the sound, because it is
impossible to tell which ear is lagging and which is leading. Also there arise
Spatial Perception
FA C T F I L E 2 . 4 T H E P R E C E D E N CE EFFECT
The precedence effect is important for understanding
sound localization when two or more sources are emitting
essentially the same sound (e.g. a person speaking and
a loudspeaker in a different place emitting an amplified
version of their voice). It is primarily a feature of transient
sounds rather than continuous sounds. In such an example both ears hear both the person and the loudspeaker.
The brain tends to localize based on the interaural delay
arising from the earliest arriving wavefront, the source
appearing to come from a direction towards that of the
earliest arriving signal (within limits).
This effect operates over delays between the sources
that are somewhat greater than the interaural delay, of the
order of a few milliseconds. Similar sound arriving within
up to 50 ms of each other tend to be perceptually fused
together, such that one is not perceived as an echo of the
other. The time delay over which this fusing effect obtains
depends on the source, with clicks tending to separate
before complex sounds like music or speech. The timbre
and spatial qualities of this ‘fused sound’, though, may
be affected. One form of precedence effect is sometimes
referred to as the Haas effect after the Dutch scientist
who conducted some of the original experiments. It was
originally identified in experiments designed to determine
what would happen to the perception of speech in the presence of a single echo. Haas determined that the delayed
‘echo’ could be made substantially louder than the earlier sound before it was perceived to be equally loud, as
shown in the approximation below. The effect depends
considerably on the spatial separation of the two or more
sources involved. This has important implications for
recording techniques where time and intensity differences
between channels are used either separately or combined
to create spatial cues.
frequencies where the phase difference is zero. Phase differences can also be
confusing in reflective environments where room modes and other effects
of reflections may modify the phase cues present at the ears.
When two or more physically separated sources emit similar sounds the
precedence effect is important in determining the apparent source direction, as explained in Fact File 2.4.
Amplitude and spectral cues
The head’s size makes it an appreciable barrier to sound at high frequencies
but not at low frequencies. Furthermore, the unusual shape of the pinna
(the visible part of the outer ear) gives rise to reflections and resonances that
change the spectrum of the sound at the eardrum depending on the angle of
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CHAPTER 2: Auditory Perception
incidence of a sound wave. Reflections off the shoulders and body also modify the spectrum to some extent. A final amplitude cue that may be relevant
for spherical wave sources close to the head is the level difference due to the
extra distance traveled between the ears by off-center sources. For sources
at most normal distances from the head this level difference is minimal,
because the extra distance traveled is negligible compared with that already
traveled.
The sum of all of these effects is a unique head-related transfer function
or HRTF for every source position and angle of incidence, including different
elevations and front–back positions. Some examples of HRTFs at different
angles are shown in Figure 2.5. It will be seen that there are numerous spectral peaks and dips, particularly at high frequencies, and common features
have been found that characterize certain source positions. This, therefore,
is a unique form of directional encoding that the brain can learn. Typically,
sources to the rear give rise to a reduced high-frequency response in both
ears compared to those at the front, owing to the slightly forward-facing
shape of the pinna. Sources to one side result in an increased high-frequency
difference between the ears, owing to the shadowing effect of the head.
These HRTFs are superimposed on the natural spectra of the source
themselves. It is therefore hard to understand how the brain might use the
monoaural spectral characteristics of sounds to determine their positions
as it would be difficult to separate the timbral characteristics of sources
from those added by the HRTF. Monaural cues are likely to be more detectable with moving sources, because moving sources allow the brain to track
changes in the spectral characteristics that should be independent of a
source’s own spectrum. For lateralization it is most likely to be differences
in HRTFs between the ears that help the brain to localize sources, in conjunction with the associated interaural time delay. Monaural cues may be
more relevant for localization in the median plane where there are minimal
differences between the ears.
There are remarkable differences in HRTFs between individuals, although
common features can be found. Figure 2.6 shows just two HRTF curves measured by Begault for different subjects, illustrating the problem of generalization in this respect.
The so-called concha resonance (that created by the main cavity in the
center of the pinna) is believed to be responsible for creating a sense of
externalization – in other words a sense that the sound emanates from outside the head rather than within. Sound-reproducing systems that disturb
or distort this resonance, such as certain headphone types, tend to create
in-the-head localization as a result.
Spatial Perception
FIGURE 2.5 Monaural transfer functions of the left ear for several directions in the horizontal
plane, relative to sound incident from the front; anechoic chamber, 2 m loudspeaker distance, impulse
technique, 25 subjects, complex averaging (Blauert, 1997). (a) Level difference; (b) time difference.
(Courtesy of MIT Press.)
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CHAPTER 2: Auditory Perception
FIGURE 2.6
HRTFs of two subjects for
a source at 0° azimuth and
elevation. Note considerable
HF differences. (Begault,
1991.)
Effects of reflections
Reflections arising from sources in listening spaces affect spatial perception significantly, as discussed in Fact File 2.5. Reflections in the early time
period after direct sound (up to 50–80 ms) typically have the effect of broadening or deepening the spatial attributes of a source. They are unlikely to
be individually localizable. In the period up to about 20 ms they can cause
severe timbral coloration if they are at high levels. After 80 ms they tend
to contribute more to the sense of envelopment or spaciousness of the
environment.
Interaction between hearing and other senses
Some spatial cues are context dependent and may be strongly influenced
by the information presented by other senses, particularly vision. Learned
experience leads the brain to expect certain cues to imply certain spatial
conditions, and if this is contradicted then confusion may arise. For example, it is unusual to experience the sound of a plane flying along beneath
one, but the situation can occasionally arise when climbing mountains.
Generally one expects planes to fly above, and most people will look up or
duck when played loud binaural recordings of planes flying over, even if the
spectral cues do not imply this direction.
It is normal to rely quite heavily on the visual sense for information
about events within the visible field, and it is interesting to note that most
people, when played binaural recordings (see Chapter 16) of sound scenes
Spatial Perception
FA C T F I L E 2 . 5 R E FL E C T IO N S A FFECT SPACI OUSNESS
The subjective phenomenon of apparent or auditory source
width (ASW) has been studied for a number of years, particularly by psychoacousticians interested in the acoustics of concert halls. ASW relates to the issue of how large
a space a source appears to occupy from a sonic point
of view (ignoring vision for the moment), as shown below.
Individual source width should be distinguished from overall
‘sound stage width’ (in other words, the distance perceived
between the left and right limits of a stereophonic scene).
Early reflected energy in a space (up to about 80 ms)
appears to modify the ASW of a source by broadening it
somewhat, depending on the magnitude and time delay
of early reflections. Concert hall experiments seem to
show that subjects prefer larger amounts of ASW, but it
is not clear what is the optimum degree of ASW (presumably sources that appeared excessively large would be difficult to localize and unnatural).
Envelopment, spaciousness and sometimes ‘room
impression’ are typically spatial features of a reverberant environment rather than individual sources, and are
largely the result of late reflected sound (particularly lateral reflections after about 80 ms). Spaciousness is used
most often to describe the sense of open space or ‘room’
in which the subject is located, usually as a result of
some sound sources such as musical instruments playing in that space. It is also related to the sense of ‘externalization’ perceived – in other words whether the sound
appears to be outside the head rather than constrained
to a region close to or inside it. Envelopment is a similar
term and is used to describe the sense of immersivity and
involvement in a (reverberant) soundfield, with that sound
appearing to come from all around. It is regarded as a
positive quality that is experienced in good concert halls.
(b)
(a)
ASW
ASW
Source
Listener
Source
Listener
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CHAPTER 2: Auditory Perception
without accompanying visual information or any form of head tracking, localize the scene primarily behind them rather than in front. In fact
obtaining front images from any binaural system using headphones is surprisingly difficult. This may be because one is used to using the hearing
sense to localize things where they cannot be seen, and that if something
cannot be seen it is likely to be behind. In the absence of the ability to
move the head to resolve front–back conflicts the brain tends to assume a
rear sound image. So-called ‘reversals’ in binaural audio systems are consequently very common.
Resolving conflicting cues
In environments where different cues conflict in respect of the implied
location of sound sources, the hearing process appears to operate on a sort
of majority decision logic basis. In other words it evaluates the available
information and votes on the most likely situation, based on what it can
determine. Auditory perception has been likened to a hypothesis generation and testing process, whereby likely scenarios are constructed from the
available information and tested against subsequent experience (often over
a very short time interval). Context-dependent cues and those from other
senses are quite important here. Since there is a strong precedence effect
favoring the first-arriving wavefront, the direct sound in a reflective environment (which arrives at the listener first) will tend to affect localization
most, while subsequent reflections may be considered less important. Head
movements will also help to resolve some conflicts, as will visual cues.
Reflections from the nearest surfaces, though, particularly the floor, can
aid the localizing process in a subtle way. Moving sources also tend to provide more information than stationary ones, allowing the brain to measure
changes in the received information that may resolve some uncertainties.
Distance and depth perception
Apart from lateralization of sound sources, the ability to perceive distance
and depth of sound images is crucial to our subjective appreciation of sound
quality. Distance is a term specifically related to how far away an individual
source appears to be, whereas depth can describe the overall front–back distance of a scene and the sense of perspective created. Individual sources
may also appear to have depth.
A number of factors appear to contribute to distance perception,
depending on whether one is working in reflective or ‘dead’ environments.
Considering for a moment the simple differences between a sound source
Spatial Perception
close to a listener and the same source further away, the one further away
will have the following differences:
■
■
■
■
■
Quieter (extra distance traveled)
Less high-frequency content (air absorbtion)
More reverberant (in reflective environment)
Less difference between time of direct sound and first-floor reflection
Attenuated ground reflection
Numerous studies have shown that absolute distance perception, using
the auditory sense alone, is very unreliable in non-reflective environments,
although it is possible for listeners to be reasonably accurate in judging relative distances (since there is then a reference point with known distance
against which other sources can be compared). In reflective environments, on
the other hand, there is substantial additional information available to the
brain. The ratio of direct to reverberant sound is directly related to source distance. The reverberation time and the early reflection timing tells the brain a
lot about the size of the space and the distance to the surfaces, thereby giving
it boundaries beyond which sources could not reasonably be expected to lie.
Naturalness in spatial hearing
The majority of spatial cues received in reproduced sound environments
are similar to those received in natural environments, although their magnitudes and natures may be modified somewhat. There are, nonetheless,
occasional phenomena that might be considered as specifically associated
with reproduced sound, being rarely or never encountered in natural environments. The one that springs most readily to mind is the ‘out-of-phase’
phenomenon, in which two sound sources such as loudspeakers or headphones are oscillating exactly 180° out of phase with each other – usually
the result of a polarity inversion somewhere in the signal chain. This creates an uncomfortable sensation with a strong but rather unnatural sense
of spaciousness, and makes phantom sources hard to localize. The outof-phase sensation never arises in natural listening and many people find it
quite disorientating and uncomfortable. Its unfamiliarity makes it hard to
identify for naïve listeners, whereas for expert audio engineers its sound is
unmistakable. Naïve listeners may even quite like the effect, and extreme
phase effects have sometimes been used in low-end audio products to create a sense of extra stereo width.
Audio engineers also often refer to problems with spatial reproduction
as being ‘phasy ’ in quality. Usually this is a negative term that can imply
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CHAPTER 2: Auditory Perception
abnormal phase differences between the channels, or an unnatural degree
of phase difference that may be changing with time. Anomalies in signal
processing or microphone technique can create such effects and they are
unique to reproduced sound, so there is in effect no natural anchor or reference point against which to compare these experiences.
RECOMMENDED FURTHER READING
Blauert, J., 1997. Spatial Hearing, second ed. Translated by J.S. Allen. MIT Press.
Bregman, A., 1994. Auditory Scene Analysis: The Perceptual Organisation of
Sound. MIT Press.
Howard, D., Angus, J., 2000. Acoustics and Psychoacoustics, second ed. Focal
Press.
Moore, B.C.J., 2003. An Introduction to the Psychology of Hearing, fifth ed.
Academic Press.
RECOMMENDED LISTENING
Auditory Demonstrations (Compact Disc). Philips Cat. No. 1126-061. Available
from the Acoustical Society of America.
CHAPTER 3
Microphones
CH A P T E R C O N TE N T S
The Moving-coil or Dynamic Microphone
The Ribbon Microphone
The Capacitor or Condenser Microphone
Basic capacitor microphone
Electret designs
RF capacitor microphone
Directional Responses and Polar Diagrams
Omnidirectional pattern
Figure-eight or bidirectional pattern
Cardioid or unidirectional pattern
Hypercardioid pattern
Specialized Microphone Types
Rifle microphone
Parabolic microphone
Boundary or ‘pressure-zone’ microphone
Switchable Polar Patterns
Stereo Microphones
Microphone Performance
Microphone sensitivity in practice
Microphone noise in practice
Microphone Powering Options
Phantom power
A–B powering
Radio Microphones
Principles
Facilities
Licenses
Aerials
Aerial siting and connection
Diversity reception
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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CHAPTER 3: Microphones
A microphone is a transducer that converts acoustical sound energy into
electrical energy, based on the principle described in Fact File 3.1. It performs
the opposite function to a loudspeaker, which converts electrical energy
into acoustical energy. The three most common principles of operation are
the moving coil or ‘dynamic’, the ribbon, and the capacitor or condenser. The
principles of these are described in Fact Files 3.2–3.4.
THE MOVING-COIL OR DYNAMIC MICROPHONE
The moving-coil microphone is widely used in the sound reinforcement
industry, its robustness making it particularly suitable for hand-held vocal
FA C T F I L E 3. 1 E LE C T R O MA G N ETI C TRANSDUCERS
Electromagnetic transducers facilitate the conversion of
acoustic signals into electrical signals. They also act to
convert electrical signals back into acoustic sound waves.
The principle is very simple: if a wire can be made to
move in a magnetic field, perpendicular to the lines of flux
linking the poles of the magnet, then an electric current is
induced in the wire (see diagram). The direction of motion
governs the direction of current flow in the wire. If the wire
can be made to move back and forth then an alternating
current can be induced in the wire, related in frequency
and amplitude to the motion of the wire. Conversely, if a
current is made to flow through a wire that cuts the lines
of a magnetic field then the wire will move.
It is a short step from here to see how acoustic sound
signals may be converted into electrical signals and vice
versa. A simple moving-coil microphone, as illustrated in
Fact File 3.2, involves a wire moving in a magnetic field,
by means of a coil attached to a flexible diaphragm that
vibrates in sympathy with the sound wave. The output
of the microphone is an alternating electrical current,
whose frequency is the same as that of the sound wave
that caused the diaphragm to vibrate. The amplitude of
the electrical signal generated depends on the mechanical characteristics of the transducer, but is proportional to
the velocity of the coil.
Vibrating systems, such as transducer diaphragms,
with springiness (compliance) and mass, have a resonant
frequency (a natural frequency of free vibration). If the
driving force’s frequency is below this resonant frequency
then the motion of the system depends principally on its
stiffness; at resonance the motion is dependent principally on its damping (resistance); and above resonance
it is mass controlled. Damping is used in transducer
diaphragms to control the amplitude of the resonant
response peak, and to ensure a more even response
around resonance. Stiffness and mass control are used to
ensure as flat a frequency response as possible in the relevant frequency ranges. A similar, but reversed process,
occurs in a loudspeaker, where an alternating current
is fed into a coil attached to a diaphragm, there being a
similar magnet around the coil. This time the diaphragm
moves in sympathy with the frequency and magnitude of
the incoming electrical audio signal, causing compression
and rarefaction of the air.
Current in wire
Magnet
S
N
Motion of wire
Magnet
The Moving-Coil or Dynamic Microphone
FA C T F I L E 3 . 2 D Y N A MIC MIC R OPHONE – PRI NCI PLES
The moving-coil microphone functions like a moving-coil
speaker in reverse. As shown in the diagram, it consists of
a rigid diaphragm, typically 20–30 mm in diameter, which
is suspended in front of a magnet. A cylindrical former is
attached to the diaphragm on to which is wound a coil of
very fine-gauge wire. This sits in the gap of a strong permanent magnet. When the diaphragm is made to vibrate
by sound waves the coil in turn moves to and fro in the
magnet’s gap, and an alternating current flows in the coil,
producing the electrical output (see Fact File 3.1). Some
models have sufficient windings on the coil to produce a
high enough output to be fed directly to the output terminals, whereas other models use fewer windings, the lower
output then being fed to a step-up transformer in the
microphone casing and then to the output. The resonant
frequency of dynamic microphone diaphragms tends to
be in the middle frequency region.
The standard output impedance of professional microphones is 200 ohms. This value was chosen because it is
high enough to allow useful step-up ratios to be employed
in the output transformers, but low enough to allow a
microphone to drive long lines of 100 meters or so. It is
possible, though, to encounter dynamic microphones
with output impedances between 50 and 600 ohms.
Some moving-coil models have a transformer that can be
wired to give a high-level, high-impedance output suitable
for feeding into the lower-sensitivity inputs found on guitar amplifiers and some PA amplifiers. High-impedance
outputs can, however, only be used to drive cables of a
few meters in length, otherwise severe high-frequency
loss results. (This is dealt with fully in Chapter 12.)
Output leads
Magnet N
S
Diaphragm
S
S
Suspension
N
Coil
FA C T F I L E 3 . 3 R IB B O N MIC R O PHONE – PRI NCI PLES
The ribbon microphone consists of a long thin strip
of conductive metal foil, pleated to give it rigidity and
‘spring’, lightly tensioned between two end clamps, as
shown in the diagram. The opposing magnetic poles create a magnetic field across the ribbon such that when
it is excited by sound waves a current is induced into it
(see Fact File 3.1). The electrical output of the ribbon is
very small, and a transformer is built into the microphone
which steps up the output. The step-up ratio of a particular ribbon design is chosen so that the resulting output
impedance is the standard 200 ohms, this also giving an
electrical output level comparable with that of moving-coil
microphones. The resonant frequency of ribbon microphones is normally at the bottom of the audio spectrum.
Magnets
S
N
Corrugated
ribbon
Transformer
Output
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CHAPTER 3: Microphones
FA C T F I L E 3. 4 C A PA C IT O R MICROPHONE – PRI NCI PLES
The capacitor (or condenser) microphone operates on the
principle that if one plate of a capacitor is free to move
with respect to the other, then the capacitance (the ability to hold electrical charge) will vary. As shown in the
diagram, the capacitor consists of a flexible diaphragm
and a rigid back plate, separated by an insulator, the
diaphragm being free to move in sympathy with sound
waves incident upon it. The 48 volts DC phantom power
(see ‘Microphone powering options’, below) charges the
capacitor via a very high resistance. A DC blocking capacitor simply prevents the phantom power from entering the
head amplifier, allowing only audio signals to pass.
When sound waves move the diaphragm the capacitance varies, and thus the voltage across the capacitor
varies proportionally, since the high resistance only allows
very slow leakage of charge from the diaphragm (much
slower than the rate of change caused by audio frequencies). This voltage modulation is fed to the head amplifier
(via the blocking capacitor) which converts the very high
impedance output of the capacitor capsule to a much
lower impedance. The output transformer balances this
Insulator
signal (see ‘Balanced lines’, Chapter 12) and conveys it
to the microphone’s output terminals. The resonant frequency of a capacitor mic diaphragm is normally at the
upper end of the audio spectrum.
The head amplifier consists of a field-effect transistor
(FET) which has an almost infinitely high input impedance.
Other electronic components are also usually present
which perform tasks such as voltage regulation and output
stage duties. Earlier capacitor microphones had valves built
into the housing, and were somewhat more bulky affairs
than their modern counterparts. Additionally, extra wiring had to be incorporated in the mic leads to supply the
valves with HT (high-tension) and valve-heater voltages.
They were thus not particularly convenient to use, but such
is the quality of sound available from capacitor mics that
they quickly established themselves. Today, the capacitor
microphone is the standard top-quality type; other types
being used for relatively specialized applications. The electrical current requirement of capacitor microphones varies
from model to model, but generally lies between 0.5 mA
and 8 mA, drawn from the phantom power supply.
Very high resistance
Phantom power
Output
transformer
Diaphragm
DC blocking
capacitor
Head
amplifier
Output
Earthed back-plate
use. Wire-mesh bulbous wind shields are usually fitted to such models,
and contain foam material which attenuates wind noise and ‘p-blasting’
from the vocalist’s mouth. Built-in bass attenuation is also often provided
to compensate for the effect known as bass tip-up or proximity effect, a
phenomenon whereby sound sources at a distance of less than 50 cm or so
are reproduced with accentuated bass if the microphone has a directional
response (see Fact File 3.5). The frequency response of the moving-coil
The Ribbon Microphone
FA C T F I L E 3. 5 B A S S T IP -U P
Pressure-gradient microphones are susceptible to a phenomenon known as bass tip-up, meaning that if a sound
source is close to the mic (less than about a meter) the
low frequencies become unnaturally exaggerated. In normal operation, the driving force on a pressure-gradient
microphone is related almost totally to the phase difference of the sound wave between front and rear of the
diaphragm (caused by the extra distance traveled by the
wave). For a fixed path-length difference between front
and rear, therefore, the phase difference increases with
frequency. At LF the phase difference is small and at MF
to HF it is larger.
Close to a small source, where the microphone is in
a field of roughly spherical waves, sound pressure drops
as distance from the source increases (see Fact File 1.3).
Thus, in addition to the phase difference between front
and rear of the mic’s diaphragm, there is a pressure difference due to the natural level drop with distance from
the source. Since the driving force on the diaphragm
due to phase difference is small at LF, this pressure drop
makes a significant additional contribution, increasing the
overall output level at LF. At HF the phase difference is
larger, and thus the contribution made by pressure difference is smaller as a proportion of the total driving force.
At greater distances from the source, the sound field
approximates more closely to one of plane waves, and the
pressure drop over the front–back distance may be considered insignificant as a driving force on the diaphragm,
making the mic’s output related only to front–back phase
difference.
mic tends to show a resonant peak of several decibels in the upper-mid frequency or ‘presence’ range, at around 5 kHz or so, accompanied by a fairly
rapid fall-off in response above 8 or 10 kHz. This is due to the fact that
the moving mass of the coil–diaphragm structure is sufficient to impede
the diaphragm’s rapid movement necessary at high frequencies. The shortcomings have actually made the moving coil a good choice for vocalists
since the presence peak helps to lift the voice and improve intelligibility.
Its robustness has also meant that it is almost exclusively used as a bass
drum mic in the rock industry. Its sound quality is restricted by its slightly
uneven and limited frequency response, but it is extremely useful in applications such as vocals, drums, and the micing-up of guitar amplifiers.
One or two high-quality moving-coil mics have appeared with an
extended and somewhat smoother frequency response, and one way of
achieving this has been to use what are effectively two mic capsules in one
housing, one covering mid and high frequencies, one covering the bass.
THE RIBBON MICROPHONE
The ribbon microphone at its best is capable of very high-quality results. The
comparatively ‘floppy ’ suspension of the ribbon gives it a low-frequency resonance at around 40 Hz, below which its frequency response fairly quickly falls
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CHAPTER 3: Microphones
away. At the high-frequency end the frequency response remains smooth.
However, the moving mass of ribbon itself means that it has difficulty in
responding to very high frequencies, and there is generally a roll-off above
14 kHz or so. Reducing the size (therefore the mass) of ribbon reduces the
area for the sound waves to work upon and its electrical output becomes
unacceptably low. One manufacturer has adopted a ‘double-ribbon’ principle
which goes some way towards removing this dilemma. Two ribbons, each
half the length of a conventional ribbon are mounted one above the other
and are connected in series. They are thus analogous to a conventional ribbon that has been ‘clamped’ in the center. Each ribbon now has half the moving mass and thus a better top-end response. Both of them working together
still maintain the necessary output.
The ribbon mic is rather more delicate than the moving coil, and it is
better suited to applications where its smooth frequency response comes
into its own, such as the micing of acoustic instruments and classical ensembles. There are, however, some robust models which look like
moving-coil vocal mics and can be interchanged with them. Micing a rock
bass drum with one is still probably not a good idea, due to the very high
transient sound pressure levels involved.
THE CAPACITOR OR CONDENSER MICROPHONE
Basic capacitor microphone
The great advantage of the capacitor mic’s diaphragm over moving-coil and
ribbon types is that it is not attached to a coil and former, and it does not
need to be of a shape and size which makes it suitable for positioning along
the length of a magnetic field. It therefore consists of an extremely light disc,
typically 12–25 mm in diameter, frequently made from polyester coated with
an extremely thin vapor-deposited metal layer so as to render it conductive.
Sometimes the diaphragm itself is made of a metal such as titanium. The
resonant frequency of the diaphragm is typically in the 12–20 kHz range,
but the increased output here is rather less prominent than with moving
coils due to the diaphragm’s very light weight.
Occasionally capacitor microphones are capable of being switched to give
a line level output, this being simple to arrange since an amplifier is built into
the mic anyway. The high-level output gives the signal rather more immunity to interference when very long cables are employed, and it also removes
the need for microphone amplifiers at the mixer or tape recorder. Phantom
power does, however, still need to be provided (see ‘Phantom power ’, below).
Directional Responses and Polar Diagrams
Electret designs
A much later development was the so-called ‘electret’ or ‘electret condenser ’ principle. The need to polarize the diaphragm with 48 volts is
dispensed with by introducing a permanent electrostatic charge into it
during manufacture. In order to achieve this the diaphragm has to be of a
more substantial mass, and its audio performance is therefore closer to a
moving-coil than to a true capacitor type. The power for the head amplifier is supplied either by a small dry-cell battery in the stem of the mic or
by phantom power. The electret principle is particularly suited to applications where compact size and light weight are important, such as in small
portable cassette machines (all built-in mics are now electrets) and tie-clip
microphones which are ubiquitous in television work. They are also made
in vast quantities very cheaply.
Later on, the so-called ‘back electret’ technique was developed. Here,
the diaphragm is the same as that of a true capacitor type, the electrostatic
charge being induced into the rigid back plate instead. Top-quality examples
of back electrets are therefore just as good as conventional capacitor mics
with their 48 volts of polarizing voltage.
RF capacitor microphone
Still another variation on the theme is the RF (radio frequency) capacitor
mic, in which the capacitor formed by the diaphragm and back plate forms
part of a tuned circuit to generate a steady carrier frequency which is much
higher than the highest audio frequency. The sound waves move the diaphragm as before, and this now causes modulation of the tuned frequency.
This is then demodulated by a process similar to the process of FM radio
reception, and the resulting output is the required audio signal. (It must
be understood that the complete process is carried out within the housing
of the microphone and it does not in itself have anything to do with radio
microphone systems, as discussed in ‘Radio microphones’, below.)
DIRECTIONAL RESPONSES AND POLAR DIAGRAMS
Microphones are designed to have a specific directional response pattern,
described by a so-called ‘polar diagram’. The polar diagram is a form of twodimensional contour map, showing the magnitude of the microphone’s output at different angles of incidence of a sound wave. The distance of the polar
plot from the center of the graph (considered as the position of the microphone diaphragm) is usually calibrated in decibels, with a nominal 0 dB being
53
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CHAPTER 3: Microphones
marked for the response at zero degrees at 1 kHz.
The further the plot is from the center, the greater
the output of the microphone at that angle.
0°
0 dB
–6
–12
Omnidirectional pattern
Ideally, an omnidirectional or ‘omni’ microphone
picks up sound equally from all directions. The
omni polar response is shown in Figure 3.1, and
is achieved by leaving the microphone diaphragm
open at the front, but completely enclosing it at
the rear, so that it becomes a simple pressure
transducer, responding only to the change of air
pressure caused by the sound waves. This works
extremely well at low and mid frequencies, but
180°
at high frequencies the dimensions of the microFIGURE 3.1 Idealized polar diagram of an omnidirectional phone capsule itself begin to be comparable with
microphone.
the wavelength of the sound waves, and a shadowing effect causes high frequencies to be
0°
picked up rather less well to the rear and
sides of the mic. A pressure increase also
0 dB
results for high-frequency sounds from the
–6
front. Coupled with this is the possibility
–12
for cancelations to arise when a highfrequency wave, whose wavelength is comparable with the diaphragm diameter, is
incident from the side of the diaphragm.
270°
90°
In such a case positive and negative peaks
of the wave may result in opposing forces
on the diaphragm.
Figure 3.2 shows the polar response
plot
which can be expected from a real
20 Hz–2 kHz
omnidirectional
microphone with a cap3–6 kHz
above 8 kHz
sule half an inch (13 mm) in diameter. It
180°
is perfectly omnidirectional up to around
FIGURE 3.2 Typical polar diagram of an omnidirectional microphone 2 kHz, but then it begins to lose sensitivity
at a number of frequencies.
at the rear; at 3 kHz its sensitivity at 180°
will typically be 6 dB down compared with
lower frequencies. Above 8 kHz, the 180° response could be as much as
15 dB down, and the response at 90° and 270° could show perhaps a 10 dB
loss. As a consequence, sounds which are being picked up significantly off
270°
90°
Directional Responses and Polar Diagrams
axis from the microphone will be reproduced with considerable treble loss,
and will sound dull. It is at its best on axis and up to 45° either side of the
front of the microphone.
High-quality omnidirectional microphones are characterized by their
wide, smooth frequency response extending both to the lowest bass frequencies and the high treble with minimum resonances or coloration. This
is due to the fact that they are basically very simple in design, being just
a capsule which is open at the front and completely enclosed at the rear.
(In fact a very small opening is provided to the rear of the diaphragm in
order to compensate for overall changes in atmospheric pressure which
would otherwise distort the diaphragm.) The small tie-clip microphones
which one sees in television work are usually omnidirectional electret types
which are capable of very good performance. The smaller the dimensions
of the mic, the better the polar response at high frequencies, and mics such
as these have quarter-inch diaphragms which maintain a very good omnidirectional response right up to 10 kHz.
Omni microphones are usually the most immune to handling and wind
noise of all the polar patterns, since they are only sensitive to absolute
sound pressure. Patterns such as figure-eight (especially ribbons) and cardioid, described below, are much more susceptible to handling and wind noise
than omnis because they are sensitive to the large pressure difference created
across the capsule by low-frequency movements such as those caused by wind
or unwanted diaphragm motion. A pressure-gradient microphone’s mechanical impedance (the diaphragm’s resistance to motion) is always lower at LF
than that of a pressure (omni) microphone, and thus it is more susceptible to
unwanted LF disturbances.
Figure-eight or bidirectional pattern
The figure-eight or bidirectional polar response is shown in Figure 3.3.
Such a microphone has an output proportional to the mathematical cosine
of the angle of incidence. One can quickly draw a figure-eight plot on a
piece of graph paper, using a protractor and a set of cosine tables or pocket
calculator. Cos 0° ⫽ 1, showing a maximum response on the forward axis
(this will be termed the 0 dB reference point). Cos 90° ⫽ 0, so at 90° off axis
no sound is picked up. Cos 180° is ⫺1, so the output produced by a sound
which is picked up by the rear lobe of the microphone will be 180° out of
phase compared with an identical sound picked up by the front lobe. The
phase is indicated by the ⫹ and ⫺ signs on the polar diagram. At 45° off
axis, the output of the microphone is 3 dB down (cos 45° represents 0.707
or 1/2 times the maximum output) compared with the on-axis output.
55
56
CHAPTER 3: Microphones
Traditionally the ribbon microphone
has sported a figure-eight polar response,
0 dB
and the ribbon has been left completely
open both to the front and to the rear. Such
–6
a diaphragm operates on the pressure–12
+
gradient principle, responding to the difference in pressure between the front and
the rear of the microphone. Consider a
270°
90° sound reaching the mic from a direction
90° off axis to it. The sound pressure will
be of equal magnitude on both sides of the
diaphragm and so no movement will take
place, giving no output. When a sound
–
arrives from the 0° direction a phase difference arises between the front and rear
of the ribbon, due to the small additional
distance traveled by the wave. The result180°
ing difference in pressure produces moveFIGURE 3.3 Idealized polar diagram of a figure-eight microphone.
ment of the diaphragm and an output
results.
At very low frequencies, wavelengths are very long and therefore the
phase difference between front and rear of the mic is very small, causing a
gradual reduction in output as the frequency gets lower. In ribbon microphones this is compensated for by putting the low-frequency resonance
of the ribbon to good use, using it to prop up the bass response. Singlediaphragm capacitor mic designs which have a figure-eight polar response
do not have this option, since the diaphragm resonance is at a very high
frequency, and a gradual roll-off in the bass can be expected unless other
means such as electronic frequency correction in the microphone design
have been employed. Double-diaphragm switchable types which have a
figure-eight capability achieve this by combining a pair of back-to-back cardioids (see next section) that are mutually out of phase.
Like the omni, the figure-eight can give very clear uncolored reproduction. The polar response tends to be very uniform at all frequencies, except
for a slight narrowing above 10 kHz or so, but it is worth noting that a ribbon mic has a rather better polar response at high frequencies in the horizontal plane than in the vertical plane, due to the fact that the ribbon is
long and thin. A high-frequency sound coming from a direction somewhat
above the plane of the microphone will suffer partial cancelation, since at
frequencies where the wavelength begins to be comparable with the length
of the ribbon the wave arrives partially out of phase at the lower portion
0°
Directional Responses and Polar Diagrams
compared with the upper portion, therefore reducing the effective acoustical
drive of the ribbon compared with mid frequencies. Ribbon figure-eight
microphones should therefore be orientated either upright or upside-down
with their stems vertical so as to obtain the best polar response in the horizontal plane, vertical polar response usually being less important.
Although the figure-eight picks up sound equally to the front and to the
rear, it must be remembered that the rear pickup is out of phase with the
front, and so correct orientation of the mic is required.
Cardioid or unidirectional pattern
The cardioid pattern is described mathematically as 1 ⫹ cos θ, where θ is the
angle of incidence of the sound. Since the omni has a response of 1 (equal
all round) and the figure-eight has a response represented by cos θ, the cardioid may be considered theoretically as a product of these two responses.
Figure 3.4(a) illustrates its shape. Figure 3.4(b) shows an omni and a figureeight superimposed, and one can see that adding the two produces the cardioid shape: at 0°, both polar responses are of equal amplitude and phase, and
so they reinforce each other, giving a total output which is actually twice
that of either separately. At 180°, however, the two are of equal amplitude
but opposite phase, and so complete cancelation occurs and there is no output. At 90° there is no output from the figure-eight, but just the contribution from the omni, so the cardioid response is 6 dB down at 90°. It is 3 dB
down at 65° off axis.
One or two early microphone designs actually housed a figure-eight and
an omni together in the same casing, electrically combining their outputs
to give a resulting cardioid response. This gave a rather bulky mic, and also
the two diaphragms could not be placed close enough together to produce
a good cardioid response at higher frequencies due to the fact that at these
frequencies the wavelength of sound became comparable with the distance
between the diaphragms. The designs did, however, obtain a cardioid from
first principles.
The cardioid response is now obtained by leaving the diaphragm open at
the front, but introducing various acoustic labyrinths at the rear which cause
sound to reach the back of the diaphragm in various combinations of phase
and amplitude to produce a resultant cardioid response. This is difficult to
achieve at all frequencies simultaneously, and Figure 3.5 illustrates the polar
pattern of a typical cardioid mic with a three-quarter-inch diaphragm. As
can be seen, at mid frequencies the polar response is very good. At low frequencies it tends to degenerate towards omni, and at very high frequencies
it becomes rather more directional than is desirable. Sound arriving from,
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CHAPTER 3: Microphones
0°
0°
0 dB
0 dB
–6
–6
–12
–12
+
+
270°
90°
270°
90°
+
–
(a)
180°
(b)
180°
FIGURE 3.4 (a) Idealized polar diagram of a cardioid microphone. (b) A cardioid microphone can be seen to be the mathematical
equivalent of an omni and a figure-eight response added together.
0°
0 dB
–6
–12
270°
90°
180°
LF
MF
HF
FIGURE 3.5 Typical polar diagram of a cardioid microphone
at low, middle and high frequencies.
say, 45° off axis will be reproduced with treble
loss, and sounds arriving from the rear will not be
completely attenuated, the low frequencies being
picked up quite uniformly.
The above example is very typical of movingcoil cardioids, and they are in fact very useful for
vocalists due to the narrow pickup at high frequencies helping to exclude off-axis sounds, and
also the relative lack of pressure-gradient component at the bass end helping to combat bass
tip-up. High-quality capacitor cardioids with halfinch diaphragms achieve a rather more ideal cardioid response. Owing to the presence of acoustic
labyrinths, coloration of the sound is rather more
likely, and it is not unusual to find that a relatively cheap electret omni will sound better than
a fairly expensive cardioid.
Hypercardioid pattern
The hypercardioid, sometimes called ‘cottage loaf ’ because of its shape, is
shown in Figure 3.6. It is described mathematically by the formula 0.5 ⫹ cos θ,
i.e. it is a combination of an omni attenuated by 6 dB, and a figure-eight. Its
Specialized Microphone Types
response is in between the cardioid and figure-eight
patterns, having a relatively small rear lobe which
is out of phase with the front lobe. Its sensitivity
is 3 dB down at 55° off axis. Like the cardioid, the
polar response is obtained by introducing acoustic
labyrinths to the rear of the diaphragm. Because
of the large pressure-gradient component it too is
fairly susceptible to bass tip-up. Practical examples
of hypercardioid microphones tend to have polar
responses which are tolerably close to the ideal. The
hypercardioid has the highest direct-to-reverberant
ratio of the patterns described, which means that
the ratio between the level of on-axis sound and
the level of reflected sounds picked up from other
angles is very high, and so it is good for excluding
unwanted sounds such as excessive room ambience
or unwanted noise.
SPECIALIZED MICROPHONE TYPES
Rifle microphone
0°
0 dB
–6
–12
+
270°
90°
–
180°
FIGURE 3.6 Idealized polar diagram of a hypercardioid
microphone.
0°
0 dB
–6
The rifle microphone is so called because it con–12
sists of a long tube of around three-quarters of
an inch (1.9 cm) in diameter and perhaps 2 feet
(61 cm) in length, and looks rather like a rifle barrel. The design is effectively an ordinary cardioid 270°
90°
microphone to which has been attached a long
barrel along which slots are cut in such a way that
a sound arriving off axis enters the slots along the
length of the tube and thus various versions of the
sound arrive at the diaphragm at the bottom of
the tube in relative phases which tend to result in
cancelation. In this way, sounds arriving off axis
180°
are greatly attenuated compared with sounds arriving on axis. Figure 3.7 illustrates the characteris- FIGURE 3.7 Typical polar diagram of a highly directional
tic club-shaped polar response. It is an extremely microphone.
directional device, and is much used by news
sound crews where it can be pointed directly at a speaking subject, excluding
crowd noise. It is also used for wildlife recording, sports broadcasts, along the
front of theater stages in multiples, and in audience participation discussions
where a particular speaker can be picked out. For outside use it is normally
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CHAPTER 3: Microphones
Parabolic
reflector
Microphone
Incoming
sound
wavefront
completely enclosed in a long, fat wind
shield, looking like a very big cigar. Halflength versions are also available which
have a polar response midway between
a club shape and a hypercardioid. All
versions, however, tend to have a rather
wider pickup at low frequencies.
Parabolic microphone
An alternative method of achieving high
directionality is to use a parabolic dish, as
shown in Figure 3.8. The dish has a diameter usually of between 0.5 and 1 meter,
and a directional microphone is positioned
at its focal point. A large ‘catchment area’
FIGURE 3.8 A parabolic reflector is sometimes used to ‘focus’ the
is therefore created in which the sound is
incoming sound wavefront at the microphone position, thus making it highly
concentrated at the head of the mic. An
directional.
overall gain of around 15 dB is typical,
but at the lower frequencies where the
wavelength of sound becomes comparable with the diameter of the dish the
response falls away. Because this device actually concentrates the sound rather
than merely rejecting off-axis sounds, comparatively high outputs are achieved
from distant sound sources. They are very useful for capturing bird song, and
they are also sometimes employed around the boundaries of cricket pitches.
They are, however, rather cumbersome in a crowd, and can also produce a
rather colored sound.
Boundary or ‘pressure-zone’ microphone
The so-called boundary or pressure-zone microphone (PZM) consists basically of an omnidirectional microphone capsule mounted on a plate usually
of around 6 inches (15 cm) square or 6 inches in diameter such that the
capsule points directly at the plate and is around 2 or 3 millimeters away
from it. The plate is intended to be placed on a large flat surface such as a
wall or floor, and it can also be placed on the underside of a piano lid, for
instance. Its polar response is hemispherical. Because the mic capsule is a
simple omni, quite good-sounding versions are available with electret capsules fairly cheaply, and so if one wishes to experiment with this unusual
type of microphone one can do so without parting with a great deal of
money. It is important to remember though that despite its looks it is not a
contact mic – the plate itself does not transduce surface vibrations – and it
Stereo Microphones
should be used with the awareness that it is equivalent to an ordinary
omnidirectional microphone pointing at a flat surface, very close to
it. The frequency response of such a microphone is rarely as flat as
that of an ordinary omni, but it can be unobtrusive in use.
SWITCHABLE POLAR PATTERNS
The double-diaphragm capacitor microphone, such as the commercial
example shown in Figure 3.9, is a microphone in which two identical
diaphragms are employed, placed each side of a central rigid plate in FIGURE 3.9 A typical doublethe manner of a sandwich. Perforations in the central plate give both diaphragm condenser microphone
diaphragms an essentially cardioid response. When the polarizing volt- with switchable polar pattern: the
age on both diaphragms is the same, the electrically combined out- AKG C4141B-ULS. (Courtesy of AKG
put gives an omnidirectional response due to the combination of the Acoustics GmbH.)
back-to-back cardioids in phase. When the polarizing voltage of one
diaphragm is opposite to that of the other, and the potential of the rigid central plate is midway between the two, the combined output gives a figureeight response (back-to-back cardioids mutually out of phase). Intermediate
combinations give cardioid and hypercardioid polar responses. In this way
the microphone is given a switchable polar response which can be adjusted
either by switches on the microphone itself or via a remote control box.
Some microphones with switchable polar patterns achieve this by employing a conventional single diaphragm around which is placed appropriate
mechanical labyrinths which can be switched to give the various patterns.
Another method manufacturers have used is to make the capsule housing on the end of the microphone detachable, so that a cardioid capsule,
say, can be unscrewed and removed to be replaced with, say, an omni.
This also facilitates the use of extension tubes whereby a long thin pipe
of around a meter or so in length with suitably threaded terminations is
inserted between the main microphone body and the capsule. The body of
the microphone is mounted on a short floor stand and the thin tube now
brings the capsule up to the required height, giving a visually unobtrusive
form of microphone stand.
STEREO MICROPHONES
Stereo microphones, such as the example shown in Figure 3.10, are available in which two microphones are built into a single casing, one capsule
being rotatable with respect to the other so that the angle between the two
can be adjusted. Also, each capsule can be switched to give any desired polar
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CHAPTER 3: Microphones
response. One can therefore adjust the mic to give a pair
of figure-eight microphones angled at, say, 90°, or a pair
of cardioids at 120°, and so on. Some stereo mics, such
FIGURE 3.10 A typical stereo microphone: the as that pictured in Figure 3.11, are configured in a sumand-difference arrangement, instead of as a left–right pair,
Neumann SM69. (Courtesy of FWO Bauch Ltd.)
with a ‘sum’ capsule pointing forwards and a figure-eight
‘difference’ capsule facing sideways. The sum-and-difference or ‘middle
and side’ (M and S) signals are combined in a matrix box to produce
a left–right stereo signal by adding M and S to give the left channel
and subtracting M and S to give the right channel. This is discussed in
more detail in Fact File 3.6.
A sophisticated stereo microphone is the Soundfield Research
microphone. In this design, four ‘subcardioid’ capsules (i.e. between
omni and cardioid) are arranged in a tetrahedral array such that their
outputs can be combined in various ways to give four outputs, termed
‘B format’. The raw output from the four capsules is termed ‘A format’. The four B-format signals consist of a forward-facing figure-eight
(‘X’), a sideways-facing figure-eight (‘Y’), an up-and-down-facing figureeight (‘Z ’), and an omnidirectional output (‘W’). These are then appropriately combined to produce any configuration of stereo microphone
FIGURE 3.11 A typical ‘sum- output, each channel being fully adjustable from omni through carand-difference’ stereo microphone: dioid to figure-eight, the angles between the capsules also being fully
the Shure VP88. (Courtesy of HW adjustable. The tilt angle of the microphone, and also the ‘dominance’
International.)
(the front-to-back pickup ratio) can also be controlled. All of this is
achieved electronically by a remotely sited control unit. Additionally,
the raw B-format signals can be recorded on a four-channel tape
recorder, later to be replayed through the control unit where all of the
above parameters can be chosen after the recording session.
The ST250 is a second generation stereo microphone based on
soundfield principles, designed to be smaller and to be usable either
‘end-fire’ or ‘side-fire’ (see Figure 3.12). It can be electronically
inverted and polar patterns and capsule angles are variable remotely.
MICROPHONE PERFORMANCE
FIGURE 3.12 The SoundField
ST250 microphone is based on soundfield principles, and can be operated
either end- or side-fire, or upside-down,
using electrical matrixing of the capsules within the control unit. (Courtesy of SoundField Ltd.)
Professional microphones have a balanced low-impedance output usually via a three-pin XLR-type plug in their base. The impedance, which
is usually around 200 ohms but sometimes rather lower, enables long
microphone leads to be used. Also, the balanced configuration, discussed in ‘Balanced lines’, Chapter 12, gives considerable immunity
from interference. Other parameters which must be considered are
sensitivity (see Fact File 3.7) and noise (see Fact File 3.8).
Microphone Performance
FA C T F I L E 3 . 6 S U M A N D D IF F ERENCE PROCESSI NG
MS signals may be converted to conventional stereo very
easily, either using three channels on a mixer, or using an
electrical matrix. M is the mono sum of two conventional
stereo channels, and S is the difference between them.
Thus:
M ⫽ (L ⫹ R) ⫼ 2
S ⫽ (L ⫺ R) ⫼ 2
and
L ⫽ (M ⫹ S) ⫼ 2
R ⫽ (M ⫺ S) ⫼ 2
A pair of transformers may be used wired as shown
in the diagram to obtain either MS from LR, or vice versa.
Alternatively, a pair of summing amplifiers may be used,
with the M and S (or L and R) inputs to one being wired
in phase (so that they add) and to the other out of phase
(so that they subtract).
The mixer configuration shown in the diagram may
also be used. Here the M signal is panned centrally (feeding L and R outputs), whilst the S signal is panned left
(M ⫹ S ⫽ L). A post-fader insertion feed is taken from the
S channel to a third channel which is phase reversed to
give ⫺S. The gain of this channel is set at 0 dB and is
panned right (M ⫺ S ⫽ R). If the S fader is varied in level,
the width of the stereo image and the amount of rear
pickup can be varied.
M
L
S
R
L
Summing
amplifier
M
R
Inverting
amplifier
Summing
amplifier
S
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CHAPTER 3: Microphones
FA C T F I L E 3. 7 MIC R O P H O N E S ENSI TI VI TY
The sensitivity of a microphone is an indication of the
electrical output which will be obtained for a given acoustical sound pressure level (SPL). The standard SPL is
either 74 dB (⫽1 μB) or 94 dB (⫽1 pascal or 10 μB)
(μB ⫽ microbar). One level is simply ten times greater
than the other, so it is easy to make comparisons between
differently specified models. 74 dB is roughly the level of
moderately loud speech at a distance of 1 meter. 94 dB
is 20 dB or ten times higher than this, so a microphone
yielding 1 mV μB⫺1, will yield 10 mV in a soundfield
of 94 dB. Other ways of specifying sensitivity include
expressing the output as being so many decibels below a
certain voltage for a specified SPL. For example, a capacitor mic may have a sensitivity figure of ⫺60 dBV Pa⫺1
meaning that its output level is 60 dB below 1 volt for a
94 dB SPL, which is 1 mV (60 dB ⫽ times 1000).
Capacitor microphones are the most sensitive types,
giving values in the region of 5–15 mV Pa⫺1, i.e. a sound
pressure level of 94 dB will give between 5 and 15 millivolts of electrical output. The least sensitive microphones
are ribbons, having typical sensitivities of 1–2 mV Pa⫺1,
i.e. around 15 or 20 dB lower than capacitor types.
Moving coils are generally a little more sensitive than ribbons, values being typically 1.5–3 mV Pa⫺1.
FA C T F I L E 3 . 8 MIC R O P H O N E N OI SE SPECI FI CATI ONS
All microphones inherently generate some noise. The
common way of expressing capacitor microphone noise
is the ‘A’-weighted equivalent self-noise. A typical value of
‘A’-weighted self-noise of a high-quality capacitor microphone is around 18 dBA. This means that its output noise
voltage is equivalent to the microphone being placed in
a soundfield with a loudness of 18 dBA. A self-noise in
the region of 25 dBA from a microphone is rather poor,
and if it were to be used to record speech from a distance
of a couple of meters or so the hiss would be noticeable
on the recording. The very best capacitor microphones
achieve self-noise values of around 12 dBA.
When comparing specifications one must make sure
that the noise specification is being given in the same
units. Some manufacturers give a variety of figures, all
taken using different weighting systems and test meter
characteristics, but the ‘A’-weighted self-noise discussed
will normally be present amongst them. Also, a signalto-noise ratio is frequently quoted for a 94 dB reference
SPL, being 94 minus the self-noise, so a mic with a selfnoise of 18 dBA will have a signal-to-noise ratio of 76 dBA
for a 94 dB SPL, which is also a very common way of
specifying noise.
Microphone sensitivity in practice
The consequence of mics having different sensitivity values is that rather
more amplification is needed to bring ribbons and moving coils up to line
level than is the case with capacitors. For example, speech may yield, say,
0.15 mV from a ribbon. To amplify this up to line level (775 mV) requires a
gain of around ⫻5160 or 74 dB. This is a lot, and it taxes the noise performance of the equipment and will also cause considerable amplification of
any interference that manages to get into the microphone cables.
Microphone Performance
Consider now the same speech recording, made using a capacitor microphone of 1 mV μB⫺1 sensitivity. Now only ⫻775 or 57 dB of gain is needed
to bring this up to line level, which means that any interference will have
a rather better chance of being unnoticed, and also the noise performance
of the mixer will not be so severely taxed. This does not mean that highoutput capacitor microphones should always be used, but it illustrates that
high-quality mixers and microphone cabling are required to get the best out
of low-output mics.
Microphone noise in practice
The noise coming from a capacitor microphone is mainly caused by the head
amplifier. Since ribbons and moving coils are purely passive devices one might
think that they would therefore be noiseless. This is not the case, since a 200
ohm passive resistance at room temperature generates a noise output between
20 Hz and 20 kHz of 0.26 μV (μV ⫽ microvolts). Noise in passive microphones
is thus due to thermal excitation of the charge carriers in the microphone
ribbon or voice coil, and the output transformer windings. To see what this
means in equivalent self-noise terms so that ribbons and moving coils can be
compared with capacitors, one must relate this to sensitivity.
Take a moving coil with a sensitivity of 0.2 mV μB⫺1, which is 2 mV for
94 dB SPL. The noise is 0.26 μV or 0.000 26 mV. The signal-to-noise ratio is
given by dividing the sensitivity by the noise:
2 ⫼ 0.000 26 7600
and then expressing this in decibels:
dB ⫽ 20 log 7600 ⫽ 77 dB
This is an unweighted figure, and ‘A’ weighting will usually improve it
by a couple of decibels. However, the microphone amplifier into which the
mic needs to be plugged will add a bit of noise, so it is a good idea to leave
this figure as it is to give a fairly good comparison with the capacitor example. (Because the output level of capacitor mics is so much higher than that
of moving coils, the noise of a mixer’s microphone amplifier does not figure
in the noise discussion as far as these are concerned. The noise generated
by a capacitor mic is far higher than noise generated by good microphone
amplifiers and other types of microphone.)
A 200 ohm moving-coil mic with a sensitivity of 0.2 mV μB⫺1 thus has
a signal-to-noise ratio of about 77 dB, and therefore an equivalent self-noise
of 94 ⫺ 77 ⫽ 17 dB which is comparable with high-quality capacitor types,
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CHAPTER 3: Microphones
providing that high-quality microphone amplifiers are also used. A lowoutput 200 ohm ribbon microphone could have a sensitivity of 0.1 mV μB⫺1,
i.e. 6 dB less than the above moving-coil example. Because its 200 ohm
thermal noise is roughly the same, its equivalent self noise is therefore
6 dB worse, i.e. 23 dB. This would probably be just acceptable for recording
speech and classical music if an ultra-low-noise microphone amplifier were
to be used which did not add significantly to this figure.
The discussion of a few decibels here and there may seem a bit pedantic, but in fact self-noises in the low twenties are just on the borderline of
being acceptable if one wishes to record speech or the quieter types of classical music. Loud music, and mic positions close to the sound sources such
as is the practice with rock music, generate rather higher outputs from the
microphones and here noise is rarely a problem. But the high output levels
generated by close micing of drums, guitar amps and the like can lead to
overload in the microphone amplifiers. For example, if a high-output capacitor microphone is used to pick up a guitarist’s amplifier, outputs as high
as 150 mV or more can be generated. This would overload some fixed-gain
microphone input stages, and an in-line attenuator which reduces the level
by an appropriate amount such as 10–20 dB would have to be inserted at the
mixer or tape recorder end of the microphone line. Attenuators are available
built into a short cylindrical tube which carries an XLR-type plug at one end
and a socket at the other end. It is simply inserted between the mixer or
tape recorder input and the mic lead connector. It should not be connected
at the microphone end because it is best to leave the level of signal along the
length of the mic lead high to give it greater immunity from interference.
MICROPHONE POWERING OPTIONS
Phantom power
Consideration of capacitor microphones reveals the need for supplying
power to the electronics which are built into the casing, and also the need
for a polarizing voltage across the diaphragm of many capacitor types. It
would obviously be inconvenient and potentially troublesome to incorporate extra wires in the microphone cable to supply this power, and so
an ingenious method was devised whereby the existing wires in the cable
which carry the audio signal could also be used to carry the DC voltage
necessary for the operation of capacitor mics – hence the term ‘phantom
power ’, since it is invisibly carried over the audio wires. Furthermore, this
system does not preclude the connection of a microphone not requiring
power to a powered circuit. The principle is outlined in Fact File 3.9.
Microphone Powering Options
FA C T F I L E 3 . 9 P H A N T O M P O WERI NG
The diagram below illustrates the principle of phantom
powering. Arrows indicate the path of the phantom power
current. (Refer to Chapter 12 for details of the balanced
line system.) Here 48 volts DC are supplied to the capacitor microphone as follows: the voltage is applied to each
of the audio lines in the microphone cable via two equal
value resistors, 6800 (6k8) ohms being the standard
value. The current then travels along both audio lines and
into the microphone. The microphone’s output transformer
Capsule
Microphone
casing
secondary has either a ‘center tap’ – that is, a wire connected half-way along the transformer winding, as shown
in the diagram – or two resistors as in the arrangement
shown at the other end of the line. The current thus travels
towards the center of the winding from each end, and then
via the center tap to the electronic circuit and diaphragm
of the microphone. To complete the circuit, the return path
for the current is provided by the screening braid of the
microphone cable.
Head amp
To
mic
amp
Screen
+ 48 V
– DC
It will be appreciated that if, for instance, a ribbon microphone is connected to the line in place of a capacitor mic, no current will flow into the
microphone because there will be no center tap provided on the microphone’s output transformer. Therefore, it is perfectly safe to connect other
types of balanced microphone to this line. The two 6k8 resistors are necessary for the system because if they were replaced simply by two wires
directly connected to the audio lines, these wires would short-circuit the
lines together and so no audio signal would be able to pass. The phantom
power could be applied to a center tap of the input transformer, but if a
short circuit were to develop along the cabling between one of the audio
wires and the screen, potentially large currents could be drawn through
the transformer windings and the phantom power supply, blowing fuses
or burning out components. Two 6k8 resistors limit the current to around
14 mA, which should not cause serious problems. The 6k8 value was chosen so as to be high enough not to load the microphone unduly, but low
enough for there to be only a small DC voltage drop across them so that
the microphone still receives nearly the full 48 volts. This is known as the
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CHAPTER 3: Microphones
P48 standard. Two real-life examples will be chosen to investigate exactly
how much voltage drop occurs due to the resistors.
First, the current flows through both resistors equally and so the resistors
are effectively ‘in parallel’. Two equal-value resistors in parallel behave like a
single resistor of half the value, so the two 6k8 resistors can be regarded as a
single 3k4 resistor as far as the 48 V phantom power is concerned. Ohm’s law
(see Fact File 1.1) states that the voltage drop across a resistor is equal to its
resistance multiplied by the current passing through it. Now a Calrec 1050C
microphone draws 0.5 milliamps (⫽0.0005 amps) through the resistors, so
the voltage drop is 3400 ⫻ 0.0005 ⫽ 1.7 volts. Therefore the microphone
receives 48 ⫺ 1.7 volts, i.e. 46.3 volts. The Schoeps CMC-5 microphone
draws 4 mA so the voltage drop is 3400 ⫻ 0.004 ⫽ 13.6 volts. Therefore the
microphone receives 48 ⫺ 13.6 volts, i.e. 34.4 volts. The manufacturer normally takes this voltage drop into account in the design of the microphone,
although examples exist of mics which draw so much current that they load
down the phantom voltage of a mixer to a point where it is no longer adequate to power the mics. In such a case some mics become very noisy, some
will not work at all, and yet others may produce unusual noises or oscillation. A stand-alone dedicated power supply or internal battery supply may be
the solution in difficult cases.
The universal standard is 48 volts, but some capacitor microphones are
designed to operate on a range of voltages down to 9 volts, and this can
be advantageous, for instance, when using battery-powered equipment on
location, or out of doors away from a convenient source of mains power.
Figure 3.13 illustrates the situation with phantom powering when electronically balanced circuits are used, as opposed to transformers. Capacitors
are used to block the DC voltage from the power supply, but they present a
very low impedance to the audio signal.
A–B powering
Another form of powering for capacitor microphones which is sometimes
encountered is A–B powering. Figure 3.14 illustrates this system schematically. Here, the power is applied to one of the audio lines via a resistor and is
taken to the microphone electronics via another resistor at the microphone
end. The return path is provided by the other audio line as the arrows show.
The screen is not used for carrying any current. There is a capacitor at the
center of the winding of each transformer. A capacitor does not allow DC
to pass, and so these capacitors prevent the current from short-circuiting
via the transformer windings. The capacitors have a very low impedance at
audio frequencies, so as far as the audio signal is concerned they are not
there. The usual voltage used in this system is 12 volts.
Radio Microphones
6k8
Mic
amplifier
+
–
6k8
Screen
Microphone
casing
+ 48 V
– DC
Microphone
casing
DC–DC
convertor
Capacitor
Head amp
To mic
amplifier
Mic
diaphragm
Screen
+ 12 volts
– DC
FIGURE 3.14 A typical 12 volt A–B powering arrangement.
Although, like phantom power, the existing microphone lines are used
to carry the current, it is dangerous to connect another type of microphone
in place of the one illustrated. If, say, a ribbon microphone were to be connected, its output transformer would short-circuit the applied current.
Therefore 12 volt A–B powering should be switched off before connecting
any other type of microphone, and this is clearly a disadvantage compared
with the phantom powering approach. It is encountered most commonly in
location film sound recording equipment.
RADIO MICROPHONES
Radio microphones are widely used in film, broadcasting, theater and other
industries, and it is not difficult to think of circumstances in which freedom from trailing microphone cables can be a considerable advantage in all
of the above.
FIGURE 3.13
A typical 48 volt phantom
powering arrangement in
an electronically balanced
circuit.
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CHAPTER 3: Microphones
Principles
The radio microphone system consists of a microphone front end (which
is no different from an ordinary microphone); an FM (frequency modulation) transmitter, either built into the housing of the mic or housed in a
separate case into which the mic plugs; a short aerial via which the signal
is transmitted; and a receiver which is designed to receive the signal from a
particular transmitter. Only one specified transmission frequency is picked
up by a given receiver. The audio output of the receiver then feeds a mixer
or tape machine in the same manner as any orthodox microphone or line
level source would. The principle is illustrated in Figure 3.15.
The transmitter can be built into the stem of the microphone, or it can
be housed in a separate case, typically the size of a packet of cigarettes, into
which the microphone or other signal source is plugged. A small battery
which fits inside the casing of the transmitter provides the power, and this
can also supply power to those capacitor mics which are designed to operate at the typical 9 volts of the transmitter battery. The transmitter is of the
FM type (see Fact File 3.10), as this offers high-quality audio performance.
Frequently, two or more radio microphones need to be used. Each transmitter must transmit at a different frequency, and the spacing between
each adjacent frequency must not be too close otherwise they will interfere
with each other. In practice, channels with a minimum spacing of 0.2 MHz
are used. Although only one transmitter can be used at a given frequency,
any number of receivers can of course be used, as is the case with ordinary
radio reception.
Facilities
Transmitters are often fitted with facilities which enable the operator to set
the equipment up for optimum performance. A 1 kHz line-up tone is sometimes encountered which sends a continuous tone to the receiver to check
continuity. Input gain controls are useful, with an indication of peak input
level, so that the transmitter can be used with mics and line level sources
Aerials
Microphone
FM
transmitter
FM
receiver
Line out
FIGURE 3.15 A radio microphone incorporates an FM transmitter, resulting in no fixed link
between microphone and mixer.
Radio Microphones
FA C T F I L E 3 . 1 0 F R E Q U E N C Y MODULATI ON
In FM systems the transmitter radiates a high-frequency
radio wave (the carrier) whose frequency is modulated by
the amplitude of the audio signal. The positive-going part
of the audio waveform causes the carrier frequency to
deviate upwards, and the negative-going part causes it to
deviate downwards. At the receiver, the modulated carrier
is demodulated, converting variations in carrier frequency
back into variations in the amplitude of an audio signal.
Audio signals typically have a wide dynamic range,
and this affects the degree to which the carrier frequency
is modulated. The carrier deviation must be kept within
certain limits, and manufacturers specify the maximum
deviation permitted. The standard figure for a transmitter
with a carrier frequency of around 175 MHz is ⫾75 kHz,
meaning that the highest-level audio signal modulates the carrier frequency between 175.075 MHz and
174.925 MHz. The transmitter incorporates a limiter to
ensure that these limits are not exceeded.
of widely different output levels. It is important that the optimum setting
is found, as too great an input level may cause a limiter (see ‘The compressor/limiter ’, Chapter 13) to come into action much of the time, which can
cause compression and ‘pumping’ noises as the limiter operates. Too weak
a signal gives insufficient drive, and poor signal-to-noise ratios can result.
The receiver will have a signal strength indicator. This can be very useful for locating ‘dead spots’; transmitter positions which cause unacceptably low meter readings should be avoided, or the receiving aerial should be
moved to a position which gives better results. Another useful facility is an
indicator which tells the condition of the battery in the transmitter. When
the battery voltage falls below a certain level, the transmitter sends out an
inaudible warning signal to the receiver which will then indicate this condition. The operator then has a warning that the battery will soon fail, which
is often within 15 minutes of the indication.
Licenses
Transmitting equipment usually requires a license for its operation, and
governments normally rigidly control the frequency bands over which a
given user can operate. This ensures that local and network radio transmitters do not interfere with police, ambulance and fire brigade equipment,
etc. In the UK the frequency band for which radio mics do not have to be
licensed is between 173.8 MHz and 175 MHz. Each radio mic transmitter
needs to be spaced at least 0.2 MHz apart, and commonly used frequencies
are 173.8, 174.1, 174.5, 174.8, and 175.0 MHz. An additional requirement
is that the frequencies must be crystal controlled, which ensures that they
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CHAPTER 3: Microphones
cannot drift outside tightly specified limits. Maximum transmitter power
is limited to 10 milliwatts, which gives an effective radiated power (ERP)
at the aerial of 2 milliwatts which is very low, but adequate for the short
ranges over which radio mics are operated.
In recent years the VHF frequencies have been largely replaced by those
in the UHF band, and these have proved to be notably free from interference problems. Aerials are correspondingly shorter, and thus more convenient to wear. Frequencies are mainly in the 800 MHz band, and frequencies
for which a UK license is not required are between 863.1 and 864.9 MHz.
The Joint Frequency Management Group Ltd is the body which administers licenses in the UK, and their website www.jfmg.co.uk provides much
useful information on the subject. The situation in the USA is considerably more complicated, with both licensed and widespread unlicensed use
of wireless microphones, often in the ‘white space’ between other services
such as local television channels. There is increased competition for the
bandwidth previously used by wireless microphones, from services such as
commercial data networking and homeland security. The digital television
transition in 2009 is also likely to force licensed wireless microphones onto
frequencies below 698 MHz. However, this is accompanied by an FCC ruling that allows continued use of wireless microphones on unused television
band channels provided that they meet very strict technical criteria that
will limit their potential for interference.
Aerials
The dimensions of the transmitting aerial are related to the wavelength of
the transmitted frequency. The wavelength (λ) in an electrical conductor at
a frequency of 174.5 MHz is approximately 64 inches (160 cm). To translate this into a suitable aerial length, it is necessary to discuss the way in
which a signal resonates in a conductor. It is convenient to consider a simple dipole aerial, as shown in Figure 3.16. This consists of two conducting
rods, each a quarter of a wavelength long, fed by the transmitting signal as
shown. The center of the pair is the nodal point and exhibits a characteristic
impedance of about 70 ohms. For a radio mic, we need a total length of λ2,
i.e. 64/2 ⫽ 32 inches (80 cm).
A 32 inch dipole will therefore allow the standard range of radio mic
frequencies to resonate along its length to give efficient radiation, the precise length not being too critical. Consideration also has to be given to the
radiated polar response (this is not the same as the microphone’s polar
response). Figure 3.17 shows the polar response for a dipole. As can be
seen, it is a figure-eight with no radiation in the directions in which the
Radio Microphones
Total
le
ngth
= 0.9
5 λ/
2
Signal
Earth
λ/2
Nodal point at
center of aerial
FIGURE 3.16 A simple dipole aerial configuration.
two halves are pointing. Another factor is polarization of
the signal. Electromagnetic waves consist of an electric
wave plus a magnetic wave radiating at right angles to
each other, and so if a transmitting aerial is orientated vertically, the receiving aerial should also be orientated vertically. This is termed vertical polarization.
The radio mic transmitter therefore has a transmitting aerial of about 16 inches long: half of a dipole. The
other half is provided by the earth screen of the audio
input lead, and will be in practice rather longer than 16
inches. The first-mentioned half is therefore looked upon
as being the aerial proper, and it typically hangs vertically
downwards. The screened signal input cable will generally
be led upwards, but other practical requirements tend to
override its function as part of the aerial system.
Another type which is often used for hand-held radio
FIGURE 3.17 The dipole has a figure-eight
mics is the helical aerial. This is typically rather less than radiation pattern.
half the length of the 16 inch aerial, and has a diameter
of a centimeter or so. It protrudes from the base of the microphone. It consists of a tight coil of springy wire housed in a plastic insulator, and has the
advantage of being both smaller and reasonably tolerant of physical abuse.
Its radiating efficiency is, however, less good than the 16 inch length of wire.
At the receiver, a similar aerial is required. The helical aerial is very common here, and its short stubby form is very convenient for outside broadcast
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CHAPTER 3: Microphones
and film crews. A 16 inch length of metal tubing, rather like
a short car aerial, can be a bit unwieldy although it is a more
efficient receiver.
Other aerial configurations exist, offering higher gain and
Dipole
directionality. In the two-element aerial shown in Figure 3.18
the reflector is slightly larger than the dipole, and is spaced
behind it at a distance which causes reflection of signal back
on to it. It increases the gain, or strength of signal output,
by 3 dB. It also attenuates signals approaching from the rear
and sides. The three-element ‘Yagi’, named after its Japanese
Reflector
inventor and shown in Figure 3.19, uses the presence of a
FIGURE 3.18 A simple two-element aerial director and reflector to increase the gain of a conventional
incorporates a dipole and a reflector for greater dipole, and a greatly elongated rectangle called a folded dipole
directionality than a dipole.
is used, which itself has a characteristic impedance of about
300 ohms. The other elements are positioned such that
Direction of max. sensitivity
the final impedance is reduced to the standard 50 ohms.
The three-element Yagi is even more directional than the
dipole, and has increased gain. It can be useful in very difficult reception conditions, or where longer distances are
Director
involved such as receiving the signal from a transmitter
carried by a rock climber for running commentary! The
multi-element, high-gain, highly directional UHF television aerial is of course a familiar sight on our roof-tops.
Folded dipole
These aerials can also be used for transmitting, the
principles being exactly the same. Their increased directionality also helps to combat multipath problems. The
elements should be vertically orientated, because the transmitting aerial will normally be vertical, and the ‘direction
Reflector
of maximum sensitivity ’ arrows on the figures show the
FIGURE 3.19 The three-element ‘Yagi’
direction in which the aerials should be pointed.
configuration.
Another aerial type which has proved very useful where
a large number of radio mics are being used, for example in big musical productions, is the log-periodic type. It covers a considerably wider bandwidth
than the aerial types previously discussed. This type is rather like a dipole
but with an array of graduated dipole elements along a boom, the longer
ones to the rear, progressing down to shorter ones at the front. The appropriate pair then resonate according to the frequency of the transmission, and
by these means a single aerial (or, more usually, a pair in a diversity system)
can cover the wide band of frequencies over which the numerous transmitters operate. The presence of other elements behind and/or in front of a particular resonating pair gives the aerial a cardioid-like polar response, which
Direction of max. sensitivity
Radio Microphones
is useful for pointing the aerial to the area of desired coverage. Superficially
it resembles a Yagi but is fundamentally different in two respects. First, the
pairs of elements are somewhat different in length from their neighbors; and
second, each pair is an active dipole in its own right, the two halves being
insulated from each other. Usually such aerials are fabricated within a solid
flat plate rather like the paddle on the end of a rowing oar.
Another technique for improving the signal-to-noise ratio under difficult
reception conditions is noise reduction, which operates as follows. Inside the
transmitter there is an additional circuit which compresses the incoming
audio signal, thus reducing its overall dynamic range. At the receiver, a reciprocal circuit expands the audio signal, after reception and demodulation, and
as it pushes the lower-level audio signals back down to their correct level it
also therefore pushes the residual noise level down. Previously unacceptable
reception conditions will often yield usable results when such transmitters
and receivers are employed. It should be noted though that the system does
not increase signal strength, and all the problems of transmission and reception still apply. (Noise reduction systems are covered further in Chapter 7.)
Aerial siting and connection
It is frequently desirable to place the receiving aerial itself closer to the
transmitter than the receiver, in order to pick up a strong signal. To do
this an aerial is rigged at a convenient position close to the transmitter,
for example in the wings of a theater stage, or on the front of a balcony,
and then an aerial lead is run back to the receiver. A helical dipole aerial is
frequently employed. In such a situation, characteristic impedance must be
considered. As discussed in ‘Principles’, Chapter 12, when the wavelength
of the electrical signal in a conductor is similar to the length of the conductor, reflections can be set up at the receiving end unless the cable is properly
terminated. Therefore, impedance matching must be employed between
the aerial and the transmitter or receiver, and additionally the connecting
lead needs to have the correct characteristic impedance.
The standard value for radio microphone equipment is 50 ohms, and so
the aerial, the transmitter, the receiver, the aerial lead and the connectors
must all be rated at this value. This cannot be measured using a simple test
meter, but an aerial and cable can be tuned using an SWR (standing wave
ratio) meter to detect the level of the reflected signal. The aerial lead should
be a good-quality, low-loss type, otherwise the advantage of siting the aerial
closer to the transmitter will be wasted by signal loss along the cable. Poor
signal reception causes noisy performance, because the receiver has a builtin automatic gain control (AGC), which sets the amplification of the carrier
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CHAPTER 3: Microphones
frequency to an appropriate value. Weak signals simply require higher amplification and therefore higher noise levels result.
The use of several radio microphones calls for a complementary number
of receivers which all need an aerial feed. It is common practice to use just
one aerial which is plugged into the input of an aerial distribution amplifier. This distribution unit has several outputs which can be fed into each
receiver. It is not possible simply to connect an aerial to all the inputs in
parallel due to the impedance mismatch that this would cause.
Apart from obvious difficulties such as metallic structures between
transmitter and receiver, there are two phenomena which cause the reception of the radio signal to be less than perfect. The first phenomenon
is known as multi-path (see Figure 3.20). When the aerial transmits, the
signal reaches the receiving aerial by a number of routes. First, there is the
direct path from aerial to aerial. Additionally, signals bounce off the walls of
the building and reach the receiving aerial via a longer route. So the receiving aerial is faced with a number of signals of more or less random phase
and strength, and these will sometimes combine to cause severe signal
cancelation and consequently very poor reception. The movement of
the transmitter along with the person wearing it will alter the relationship between these multipath signals, and so ‘dead spots’ are sometimes
encountered where particular combinations of multipath signals cause
signal ‘drop-out’. The solution is to find out where these dead spots are by
trial and error, and re-siting the receiving aerial until they are minimized or
Secondary path
(due to reflection)
Microphone
transmitter
Primary path of RF signal
Receiver
Room boundary
FIGURE 3.20 Multipath distortion can arise between source and receiver due to reflections.
Radio Microphones
eliminated. It is generally good practice to site the aerial close to the transmitter so that the direct signal will be correspondingly stronger than many
of the signals arriving from the walls. Metal structures should be kept clear
of wherever possible due to their ability to reflect and screen RF signals.
Aerials can be rigged on metal bars, but at right angles to them, not parallel.
The other phenomenon is signal cancelation from other transmitters
when a number of channels are in use simultaneously. Because the transmitting frequencies of the radio mics will be quite close together, partial
cancelation of all the signals takes place. The received signals are therefore
weaker than for a single transmitter on its own. Again, siting the receiving
aerial close to the transmitters is a good idea. The ‘sharpness’ or ‘Q ’ of the
frequency tuning of the receivers plays a considerable part in obtaining good
reception in the presence of a number of signals. A receiver may give a good
performance when only one transmitter is in use, but a poor Q will vastly
reduce the reception quality when several are used. This should be checked
for when systems are being evaluated, and the testing of one channel on its
own will not of course show up these kinds of problems.
Diversity reception
A technique known as ‘spaced diversity ’ goes a good way towards combatting the above problems. In this system, two aerials feed two identical
receivers for each radio channel. A circuit continuously monitors the signal
strength being received by each receiver and automatically selects the one which is receiving the best signal (see Figure 3.21). When they are both receiving a
good signal, the outputs of the two are mixed together.
Signal strength
A crossfade is performed between the two as one RF
comparator
signal fades and the other becomes strong.
The two aerials are placed some distance apart, in
Receiver 1
Receiver 2
practice several meters gives good results, so that the
Control
multipath relationships between a given transmitter
Audio 2
position and each aerial will be somewhat different. A Audio 1
Mixer
dead spot for one aerial is therefore unlikely to coincide
with a dead spot for the other one. A good diversity
system overcomes many reception problems, and the
Best audio out
considerable increase in reliability of performance is
well worth the extra cost. The point at which diversity FIGURE 3.21 A diversity receiver incorporates
becomes desirable is when more than two radio micro- two aerials spaced apart and two receivers. The signal
phones are to be used, although good performance strength from each aerial is used to determine which
from four channels in a non-diversity installation is by output will have the higher quality.
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CHAPTER 3: Microphones
no means out of the question. Good radio microphones are very expensive,
a single channel of a quality example costing over a thousand pounds today.
Cheaper ones exist, but experience suggests that no radio microphone at all
is vastly preferable to a cheap one.
RECOMMENDED FURTHER READING
AES, 1979. Microphones: An Anthology. Audio Engineering Society.
Bartlett, B., 1991. Stereo Microphone Techniques. Focal Press.
Eargle, J., 2004. The Microphone Book, second ed.. Focal Press.
Gayford, M. (Ed.), 1994. Microphone Engineering Handbook. Focal Press.
See also ‘General further reading’ at the end of this book.
CHAPTER 4
Loudspeakers
CH A P T E R C O N TE N T S
The Moving-coil Loudspeaker
Other Loudspeaker Types
Mounting and Loading Drive Units
‘Infinite baffle’ systems
Bass reflex systems
Coupled cavity systems
Horn loading
Complete Loudspeaker systems
Two-way systems
Three-way systems
Active Loudspeakers
Subwoofers
Loudspeaker Performance
Impedance
Sensitivity
Distortion
Frequency response
Power handling
Directivity
Setting up Loudspeakers
Phase
Positioning
Thiele–Small Parameters and Enclosure Volume Calculations
Digital Signal Processing in Loudspeakers
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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91
92
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93
94
95
95
97
99
99
100
101
106
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CHAPTER 4: Loudspeakers
A loudspeaker is a transducer which converts electrical energy into acoustical
energy. A loudspeaker must therefore have a diaphragm of some sort which
is capable of being energized in such a way that it vibrates to produce sound
waves which are recognizably similar to the original sound from which the
energizing signal was derived. To ask a vibrating plastic loudspeaker cone to
reproduce the sound of, say, a violin is to ask a great deal, and it is easy to
take for granted how successful the best examples have become. Continuing
development and refinement of the loudspeaker has brought about a more
or less steady improvement in its general performance, but it is a sobering
thought that one very rarely mistakes a sound coming from a speaker for the
real sound itself, and that one nevertheless has to use these relatively imperfect devices to assess the results of one’s work. Additionally, it is easy to hear
significant differences between one model and another. Which is right? It is
important not to tailor a sound to suit a particular favorite model. There are
several principles by which loudspeakers can function, and the commonly
employed ones will be briefly discussed.
A word or two must be said about the loudspeaker enclosure. The box
can have as big an influence on the final sound of a speaker system as can
the drivers themselves. At first sight surprising, this fact can be more readily appreciated when one remembers that a speaker cone radiates virtually
the same amount of sound into the cabinet as out into the room. The same
amount of acoustical energy that is radiated is therefore also being concentrated in the cabinet, and the sound escaping through the walls and also
back out through the speaker cone has a considerable influence upon the
final sound of the system.
THE MOVING-COIL LOUDSPEAKER
The moving-coil principle is by far the most widely used, as it can be implemented in very cheap transistor radio speakers, PA (public address) systems,
and also top-quality studio monitors, plus all performance levels and applications in between. Figure 4.1 illustrates a cutaway view of a typical movingcoil loudspeaker. Such a device is also known as a drive unit or driver, as it
is the component of a complete speaker system which actually produces the
sound or ‘drives’ the air. Basically, the speaker consists of a powerful permanent magnet which has an annular gap to accommodate a coil of wire wound
around a cylindrical former. This former is attached to the cone or diaphragm
which is held in its rest position by a suspension system which usually consists of a compliant, corrugated, doped (impregnated) cloth material and a
compliant surround around the edge of the cone which can be made of a type
of rubber, doped fabric, or it can even be an extension of the cone itself, suitably treated to allow the required amount of movement of the cone.
Other Loudspeaker Types
The chassis usually consists either of
pressed steel or a casting, the latter being
Surround
particularly desirable where large heavy magChassis
nets are employed, since the very small clearance between the coil and the magnet gap
Cone or
demands a rigid structure to maintain the
diaphragm
alignment, and a pressed steel chassis can
Suspension
sometimes be distorted if the loudspeaker
is subject to rough handling as is inevitaMagnet
bly the case with portable PA systems and
the like. (A properly designed pressed steel
chassis should not be overlooked though.)
The cone itself can in principle be made of
almost any material, common choices being Magnet gap
paper pulp (as used in many PA speaker
Coil
Input
cones for its light weight, giving good effiterminals
ciency), plastics of various types (as used in
many hi-fi speaker cones due to the greater
consistency achievable than with paper pulp,
and the potentially lower coloration of the
sound, usually at the expense of increased FIGURE 4.1 Cross-section through a typical moving-coil loudspeaker.
weight and therefore lower efficiency which
is not crucially important in a domestic loudspeaker), and sometimes metal foil.
The principle of operation is based on the principle of electromagnetic
transducers described in Fact File 3.1, and is the exact reverse of the process involved in the moving-coil microphone (see Fact File 3.2). The cone
vibration sets up sound waves in the air which are an acoustic analog of
the electrical input signal. Thus in principle the moving-coil speaker is a
very crude and simple device, but the results obtained today are incomparably superior to the original 1920s Kellog and Rice design. It is, however, a
great tribute to those pioneers that the principle of operation of what is still
today’s most widely used type of speaker is still theirs.
OTHER LOUDSPEAKER TYPES
The electrostatic loudspeaker first became commercially viable in the
1950s, and is described in Fact File 4.1. The electrostatic principle is far
less commonly employed than is the moving coil, since it is difficult and
expensive to manufacture and will not produce the sound levels available
from moving-coil speakers. The sound quality of the best examples, such
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CHAPTER 4: Loudspeakers
as the Quad ESL 63 pictured in Figure 4.2, is, however, rarely equaled by
other types of speaker.
Another technique in producing a panel-type speaker membrane has been to employ a light film on which is attached a
series of conductive strips which serve as the equivalent of the
coil of a moving-coil cone speaker. The panel is housed within
a system of strong permanent magnets, and the drive signal is
applied to the conductive strips. Gaps in the magnets allow the
sound to radiate. Such systems tend to be large and expensive
like the electrostatic models, but again very high-quality results
are possible. In order to get adequate bass response and output
level from such panel speakers the diaphragm needs to be of
considerable area.
The ribbon loudspeaker principle has sometimes been
employed in high-frequency applications (‘tweeters’) and has
FIGURE
4.2 The Quad ESL63 recently also been employed in large full-range models. Figure 4.3
electrostatic loudspeaker. (Courtesy of Quad illustrates the principle. A light corrugated aluminum ribbon,
clamped at each end, is placed between two magnetic poles, one
Electroacoustics Ltd.)
FA C T F I L E 4. 1 E L E C T R O S TATIC LOUDSPEAKER – PRI NCI PLES
The electrostatic loudspeaker’s drive unit consists of a
large, flat diaphragm of extremely light weight, placed
between two rigid plates. The diagram shows a side view.
There are parallels between this loudspeaker and the
capacitor microphone described in Chapter 3.
The diaphragm has a very high resistance, and a DC
polarizing voltage in the kilovolt (kV) range is applied to
the center tap of the secondary of the input transformer,
and charges the capacitor formed by the narrow gap
between the diaphragm and the plates. The input signal
appears (via the transformer) across the two rigid plates
and thus modulates the electrostatic field. The diaphragm, being the other plate of the capacitor, thus experiences a force which alters according to the input signal.
Being free to move within certain limits with respect to the
two rigid plates, it thus vibrates to produce the sound.
There is no cabinet as such to house the speaker, and
sound radiates through the holes of both plates. Sound
therefore emerges equally from the rear and the front of
the speaker, but not from the sides. Its polar response is
therefore a figure-eight, similar to a figure-eight microphone with the rear lobe being out of phase with the front
lobe.
Rigid
perforated
plates
Diaphragm
Polarizing
voltage
Step-up
transformer
Input
signal
Other Loudspeaker Types
north, one south. The input signal is applied, via a step-down transformer, to
each end of the ribbon. The alternating nature of the signal causes an alternating magnetic field around the ribbon, which behaves like a single turn of
a coil in a moving-coil speaker. The magnets each side thus cause the ribbon to vibrate, producing sound waves. The impedance of the ribbon is often
extremely low, and an amplifier cannot drive it directly. A transformer is therefore used which steps up the impedance of the ribbon. The ribbon itself produces a very low acoustic output and often has a horn in front of it to improve
its acoustical matching with the air, giving a higher output for a given electrical input. Some ribbons are, however, very long –
Magnets
half a meter or more – and drive the air directly.
A recent panel type of speaker is the so-called
‘distributed mode loudspeaker ’ (DML), developed
S
by the NXT company following the UK’s Defence
N
Evaluation and Research Agency’s discovery that
certain lightweight composite panels used in military aircraft could act as efficient sound radiators
(Figure 4.4). Its operating principle is the antithesis
Corrugated
of conventional wisdom: whereas it is normal pracribbon
tice to strive for ‘pistonic’ motion of a cone driver
or panel, the complete area of the radiating surface
moving backwards and forwards as a whole with
Transformer
progressively smaller areas of the surface moving as
frequency increases, the DML panel is deliberately
Input
made very flexible so that a multiplicity of bending modes or resonances, equally distributed in fre- FIGURE 4.3 A ribbon loudspeaker mechanism.
quency, are set up across its surface. This creates a
large number of small radiating areas which are virtually independent of each other, giving an uncorrelated set of signals but summing to give a resultant output. The panel is driven not across its
whole area but usually at a strategically placed point by a movingcoil transducer. Because of the essentially random-phase nature
of the radiating areas, the panel is claimed not to suffer from the
higher-frequency beaming effects of conventional panels, and also
there is not the global 180° out-of-phase radiation from the rear.
Further research into DML materials has brought the promise of integrated audio-visual panels, a single screen radiating
both sound and vision simultaneously.
There are a few other types of speaker in use, but these are
sufficiently uncommon for descriptions not to be merited in this FIGURE 4.4 DML loudspeaker. (Courtesy of New Transducers Ltd.)
brief outline of basic principles.
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CHAPTER 4: Loudspeakers
MOUNTING AND LOADING DRIVE UNITS
‘Infinite baffle’ systems
The moving-coil speaker radiates sound equally in front of and to the
rear of the diaphragm or cone. As the cone moves forward it produces a
compression of the air in front of it but a rarefaction behind it, and vice
versa. The acoustical waveforms are therefore 180° out of phase with each
other and when they meet in the surrounding air they tend to cancel out,
particularly at lower frequencies where diffraction around the cone occurs.
A cabinet is therefore employed in which the drive unit sits, which has the
job of preventing the sound radiated from the rear of the cone from reaching the open air. The simplest form of cabinet is the sealed box (commonly,
but wrongly, known as the ‘infinite baffle’) which will usually have some
sound-absorbing material inside it such as plastic foam or fiber wadding.
A true ‘infinite baffle’ would be a very large flat piece of sheet material
with a circular hole cut in the middle into which the drive unit would be
mounted. Diffraction around the baffle would then only occur at frequencies below that where the wavelength approached the size of the baffle, and
thus cancelation of the two mutually out-of-phase signals would not occur
over most of the range, but for this to be effective at the lowest frequencies
the baffle would have to measure at least 3 or 4 meters square. The only
practical means of employing this type of loading is to mount the speaker
in the dividing wall between two rooms, but this is rarely encountered for
obvious reasons.
Bass reflex systems
Drive unit
Cabinet
Port
FIGURE 4.5
A ported bass reflex cabinet
construction.
Another form of loading is the bass reflex system, as shown in Figure 4.5.
A tunnel, or port, is mounted in one of the walls of the cabinet, and the
various parameters of cabinet internal volume, speaker cone weight, speaker
cone suspension compliance, port dimensions, and thus mass of air inside
the port are chosen so that at a specified low frequency the air inside the
port will resonate, which reduces the movement of the speaker cone at that
frequency. The port thus produces low-frequency output of its own, acting
in combination with the driver. In this manner increased low-frequency
output, increased efficiency, or a combination of the two can be achieved.
However, it is worth remembering that at frequencies lower than the resonant frequency the driver is acoustically unloaded because the port now
behaves simply as an open window. If extremely low frequencies from, say,
mishandled microphones or record player arms reach the speaker they will
Mounting and Loading Drive Units
cause considerable excursion of the speaker cone which can cause damage.
The air inside a closed box system, however, provides a mechanical supporting ‘spring’ right down to the lowest frequencies.
A device known as an auxiliary bass radiator (ABR) is occasionally used
as an alternative to a reflex port, and takes the form of a further bass unit
without its own magnet and coil. It is thus undriven electrically. Its cone
mass acts in the same manner as the air plug in a reflex port, but has the
advantage that mid-range frequencies are not emitted, resulting in lower
coloration.
A further form of bass loading is described in Fact File 4.2.
Coupled cavity systems
A form of bass loading that has found much favor in small domestic surround sound subwoofers is the coupled cavity, although the technique has
been in use even in large sound reinforcement subwoofers for many years.
The simplest arrangement of the loading is shown in Figure 4.6a. The drive
unit looks into a second or ‘coupled’ enclosure which is fitted with a port.
Whereas reflex ports are tuned to a specific low frequency, here the port is
tuned to a frequency above the pass band of the system, e.g. above 120 Hz
or so, and the port therefore radiates all sound below that. Above the tuned
frequency, it exhibits a 12 dB/octave roll-off. However, secondary resonances
FA C T F I L E 4 . 2 T R A N S MIS S IO N LI NE SYSTEM
A form of bass loading is the acoustic labyrinth or ‘transmission line’, as shown in the diagram. A large cabinet
houses a folded tunnel the length of which is chosen
so that resonance occurs at a specified low frequency.
Above that frequency, the tunnel, which is filled or partially filled with acoustically absorbent material, gradually
absorbs the rear-radiated sound energy along its length.
At resonance, the opening, together with the air inside
the tunnel, behaves like the port of a bass reflex design.
An advantage of this type of loading is the very good bass
extension achievable, but a large cabinet is required for
its proper functioning.
Drive
unit
Port
Labyrinth
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CHAPTER 4: Loudspeakers
in the system require that a low pass filter must still be employed. The
increased loading on the driver – it now drives an air cavity coupled to the
air plug in the port – produces greater efficiency. Figures
4.6(b) and (c) show other arrangements. In (b) two drivers
look into a common central cavity. In (c), the two drivers
are also reflex loaded.
(a)
Horn loading
(b)
(c)
FIGURE 4.6 Coupled cavity loading.
Horn loading is a technique commonly employed in large
PA loudspeaker systems, as described in Fact File 4.3. Here,
a horn is placed in front of the speaker diaphragm. The socalled ‘long-throw ’ horn tends to beam the sound over an
included angle of perhaps 90° horizontally and 40° vertically. The acoustical energy is therefore concentrated principally in the forward direction, and this is one reason for
the horn’s high efficiency. The sound is beamed forwards
towards the rear of the hall with relatively little sound
reaching the side walls. The ‘constant directivity ’ horn
FA C T F I L E 4 . 3 H O R N L O U D S P EAKER – PRI NCI PLES
A horn is an acoustic transformer, that is it helps to match
the air impedance at the throat of the horn (the throat is
where the speaker drive unit is) with the air impedance
at the mouth. Improved acoustic efficiency is therefore achieved, and for a given electrical input a horn
can increase the acoustical output of a driver by 10 dB
or more compared with the driver mounted in a conventional cabinet. A horn functions over a relatively limited
frequency range, and therefore relatively small horns are
used for the high frequencies, larger ones for upper mid
frequencies, and so on. This is very worthwhile where
high sound levels need to be generated in large halls,
rock concerts and open-air events.
Each design of horn has a natural lower cut-off frequency which is the frequency below which it ceases to
load the driver acoustically. Very large horns indeed are
needed to reproduce low frequencies, and one technique
has been to fold the horn up by building it into a more
conventional-looking cabinet. The horn principle is rarely
employed at bass frequencies due to the necessarily large
size. It is, however, frequently employed at mid and high
frequencies, but the higher coloration of the sound it produces tends to rule it out for hi-fi and studio monitoring
use other than at high frequencies if high sound levels
are required. Horns tend to be more directional than conventional speakers, and this has further advantages in PA
applications.
Horn
Driver
Mouth
Rear
chamber
Throat
Mounting and Loading Drive Units
aims to achieve a consistent spread of sound throughout the whole of its
working frequency range, and this is usually achieved at the expense of an
uneven frequency response. Special equalization is therefore often applied
to compensate for this.
The long-throw horn does not do much for those members of an audience who are close to the stage between the speaker stacks, and an acoustic
lens is often employed, which, as its name suggests, diffracts the sound,
such that the higher frequencies are spread out over a wider angle to give
good coverage at the front. Figure 4.7 shows a typical acoustic lens. It consists of a number of metal plates which are shaped and positioned with
respect to each other in such a manner as to cause outward diffraction of
the high frequencies. The downward slope of the plates is incidental to the
design requirements and it is not incorporated to project the sound downwards. Because the available acoustic output is spread out over a wider area
than is the case with the long-throw horn, the on-axis sensitivity tends to
be lower.
The high efficiency of the horn has also been much exploited in
those PA applications which do not require high sound quality, and their
use for outdoor events such as fêtes, football matches and the like, as well
as on railway station platforms, will have been noticed. Often, a contrivance known as a re-entrant horn is used, as shown in Figure 4.8. It can
be seen that the horn has been effectively cut in half, and the half which
carries the driver is turned around and placed inside the bell of the other.
Quite a long horn is therefore accommodated in a compact structure,
and this method of construction is particularly applicable to hand-held
loudhailers.
The high-frequency horn is driven not by a cone speaker but by a ‘compression driver ’ which consists of a dome-shaped diaphragm usually with
a diameter of 1 or 2 inches (2.5 or 5 cm). It resembles a hi-fi dome tweeter
but with a flange or thread in front of the dome for fixing on to the horn.
The compression driver can easily be damaged if it is driven by frequencies
below the cut-off frequency of the horn it is looking into.
FIGURE 4.7
An example of an acoustic
lens.
Front view
Side view
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CHAPTER 4: Loudspeakers
COMPLETE LOUDSPEAKER SYSTEMS
Two-way systems
It is a fact of life that no single drive unit can adequately
reproduce the complete frequency spectrum from, say, 30 Hz
to 20 kHz. Bass frequencies require large drivers with relatively high cone excursions so that adequate areas of air can
be set in motion. Conversely, the same cone could not be
expected to vibrate at 15 kHz–15 000 times a second to reproduce very high frequencies. A double bass is much larger than
FIGURE 4.8 A re-entrant horn.
a flute, and the strings of a piano which produce the low notes
are much fatter and longer than those for the high notes.
Input
The most widely used technique for reproducing virtually
terminal
the whole frequency spectrum is the so-called two-way speaker
N
system, which is employed at many quality levels from fairly
cheap audio packages to very high-quality studio monitors.
Dome diaphragm
It consists of a bass/mid driver which handles frequencies
Magnet S
up to around 3 kHz, and a high-frequency unit or ‘tweeter ’
which reproduces frequencies from 3 kHz to 20 kHz or more.
Figure 4.9 shows a cutaway view of a tweeter. Typically of
N
Input
around 1 inch (2.5 cm) in diameter, the dome is attached to a
terminal
coil in the same way that a cone is in a bass/mid driver. The
Coil
dome can be made of various materials, ‘soft’ or ‘hard’, and
FIGURE 4.9 Cross-section through a
metal domes are also frequently employed. A bass/mid driver
typical dome tweeter.
cannot adequately reproduce high frequencies as has been said.
Similarly, such a small dome tweeter would actually be damaged if bass frequencies were fed to it; thus a crossover network is required to feed each
drive unit with frequencies in the correct range, as described in Fact File 4.4.
In a basic system the woofer would typically be of around 8 inches
(20 cm) in diameter for a medium-sized domestic speaker, mounted in a cabinet having several cubic feet internal volume. Tweeters are usually sealed
at the rear, and therefore they are simply mounted in an appropriate hole
cut in the front baffle of the enclosure. This type of speaker is commonly
encountered at the cheaper end of the price range, but its simplicity makes
it well worth study since it nevertheless incorporates the basic features of
many much more costly designs. The latter differ in that they make use of
more advanced and sophisticated drive units, higher-quality cabinet materials and constructional techniques, and a rather more sophisticated crossover which usually incorporates both inductors and capacitors in the treble
and bass sections as well as resistors which together give much steeper filter
Driver
Complete Loudspeaker Systems
FA C T F I L E 4 . 4 A B A S IC C R O S S OVER NETW ORK
A frequency-dividing network or ‘crossover’ is fitted into
the speaker enclosure which divides the incoming signal into high frequencies (above about 3 kHz) and lower
frequencies, sending the latter to the bass/mid unit or
‘woofer’ and the former to the tweeter. A simple example
of the principle involved is illustrated in the diagram. In
practical designs additional account should be taken of
the fact that speaker drive units are not pure resistances.
The tweeter is fed by a capacitor. A capacitor has an
impedance which is inversely proportional to frequency,
that is at high frequencies its impedance is very low and
at low frequencies its impedance is relatively high. The
typical impedance of a tweeter is 8 ohms, and so for
signals below the example of 3 kHz (the ‘crossover frequency’) a value of capacitor is chosen which exhibits
an impedance of 8 ohms also at 3 kHz, and due to the
nature of the voltage/current phase relationship of the signal across a capacitor the power delivered to the tweeter
is attenuated by 3 dB at that frequency. It then falls at a
rate of 6 dB per octave thereafter (i.e. the tweeter’s output is 9 dB down at 1.5 kHz, 15 dB down at 750 Hz and
so on) thus protecting the tweeter from lower frequencies.
The formula which contains the value of the capacitor for
the chosen 3 kHz frequency is:
The capacitor value will more conveniently be
expressed in microfarads (millionths of a farad) and so
the final formula becomes:
C ⫽ 159 155 ⫼ (8 ohms ⫻ 3000 Hz) ⫽ 6.7 μF
Turning now to the woofer, it will be seen that an
inductor is placed in series with it. An inductor has an
impedance which rises with frequency; therefore, a value
is chosen that gives an impedance value similar to that
of the woofer at the chosen crossover frequency. Again,
the typical impedance of a woofer is 8 ohms. The formula
which contains the value of the inductor is:
f ⫽ R/(2πL )
where L ⫽ inductance in henrys, R ⫽ speaker resistance,
f ⫽ crossover frequency. The millihenry (one-thousandth
of a henry, mH) is more appropriate, so this gives:
L ⫽ 8000 ⫼ (2π ⫻ 3000) ⫽ 0.42 mH
Tweeter
Input
Capacitor
Inductor
f ⫽ 1/(2πRC )
where R is the resistance of the tweeter, and C is the
value of the capacitor in farads.
slopes than our 6 dB/octave example. Also, the overall frequency response
can be adjusted by the crossover to take account of, say, a woofer which gives
more acoustic output in the mid range than in the bass: some attenuation
of the mid range can give a flatter and better-balanced frequency response.
Three-way systems
Numerous three-way loudspeaker systems have also appeared where a
separate mid-range driver is incorporated along with additional crossover
components to restrict the frequencies feeding it to the mid range, for
Woofer
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CHAPTER 4: Loudspeakers
example between 400 Hz and 4 kHz. It is an attractive technique due to the
fact that the important mid frequencies where much of the detail of music
and speech resides are reproduced by a dedicated driver designed specially
for that job. But the increased cost and complexity does not always bring
about a proportional advance in sound quality.
ACTIVE LOUDSPEAKERS
So far, only ‘passive’ loudspeakers have been discussed, so named because
simple passive components – resistors, capacitors and inductors – are used
to divide the frequency range between the various drivers. ‘Active’ loudspeakers are also encountered, in which the frequency range is divided by active
electronic circuitry at line level, after which each frequency band is sent
to a separate power amplifier and thence to the appropriate speaker drive
unit. The expense and complexity of active systems has tended to restrict
the active technique to high-powered professional PA applications where
four-, five- and even six-way systems are employed, and to professional
studio monitoring speakers, such as the Rogers LS5/8 system pictured in
Figure 4.10. Active speakers are still comparatively rare in domestic audio.
Each driver has its own power amplifier, which of course immediately
increases the cost and complexity of the speaker system, but the advantages
include: lower distortion (due to the fact that the signal is now being split at
line level, where only a volt or so at negligible current is involved, as compared with the tens of volts and several amps that passive crossovers have
to deal with); greater system-design flexibility due to the fact that almost
any combination of speaker components can be used because their differing sensitivities, impedances and power requirements
can be compensated for by adjusting the gains of the
separate power amplifiers or electronic crossover outputs; better control of final frequency response, since
it is far easier to incorporate precise compensating circuitry into an electronic crossover design than is the
case with a passive crossover; better clarity of sound
and firmer bass simply due to the lack of passive components between power amplifiers and drivers; and an
improvement in power amplifier performance due to
the fact that each amplifier now handles a relatively
restricted band of frequencies.
In active systems amplifiers can be better matched
FIGURE 4.10 Rogers LS5/8 high-quality active studio
to loudspeakers, and the system can be designed as
loudspeaker. (Courtesy of Swisstone Electronics Ltd.)
Subwoofers
a whole, without the problems which arise when an unpredictable load is
attached to a power amplifier. In passive systems, the designer has little
or no control over which type of loudspeaker is connected to which type
of amplifier, and thus the design of each is usually a compromise between
adaptability and performance. Some active speakers have the electronics
built into the speaker cabinet which simplifies installation.
Electronic equalization has also been used to extract a level of bass performance from relatively small sized enclosures which would not normally
be expected to extend to very low frequencies. For example, looking at
Figure 4.13(a), the response of this speaker can be seen to be about 6 dB down
at 55 Hz compared with the mid frequencies, with a roll-off of about 12 dB
per octave. Applying 6 dB of boost at 55 Hz with an appropriate filter shape
would extend the bass response of the speaker markedly. However, a 6 dB
increase corresponds to a four times increase in input power at low frequencies, causing a large increase in speaker cone excursion. For these reasons,
such a technique can only be implemented if special high powered long throw
bass drivers are employed, designed specifically for this kind of application.
SUBWOOFERS
Good bass response from a loudspeaker requires a large internal cabinet
volume so that the resonant frequency of the system can be correspondingly low, the response of a given speaker normally falling away below this
resonant point. This implies the use of two large enclosures which are
likely to be visually obtrusive in a living room, for instance. A way around
this problem is to incorporate a so-called ‘subwoofer ’ system. A separate
speaker cabinet is employed which handles only the deep bass frequencies,
and it is usually driven by its own power amplifier. The signal to drive the
power amp comes from an electronic crossover which subtracts the low
bass frequencies from the feed to the main stereo amplifier and speakers,
and sends the mono sum of the deep bass to the subwoofer system.
Freed from the need to reproduce deep bass, the main stereo speakers
can now be small high-quality systems; the subwoofer can be positioned
anywhere in the room according to the manufacturers of such systems
since it only radiates frequencies below around 100 Hz or so, where sources
tend to radiate only omnidirectionally anyway. Degradation of the stereo
image has sometimes been noted when the subwoofer is a long way from
the stereo pair, and a position close to one of these is probably a good idea.
Subwoofers are also employed in concert and theater sound systems. It
is difficult to achieve both high efficiency and a good bass response at the
same time from a speaker intended for public address use, and quite large
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FIGURE 4.11
Impedance plot of a
typical two-way sealed-box
domestic loudspeaker.
32
Impedance Ω
92
16
8
4
50
100
1 kHz
10 kHz
Frequency Hz
and loud examples often have little output below 70 Hz or so. Subwoofer
systems, if properly integrated into the system as a whole, can make a large
difference to the weight and scale of live sound.
LOUDSPEAKER PERFORMANCE
Impedance
The great majority of loudspeaker drive units and systems are labelled
‘Impedance ⫽ 8 ohms’. This is, however, a nominal figure, the impedance
in practice varying widely with frequency (see ‘Sound in electrical form’,
Chapter 1). A speaker system may indeed have an 8 ohm impedance at,
say, 150 Hz, but at 50 Hz it may well be 30 ohms, and at 10 kHz it could be
4 ohms. Figure 4.11 shows the impedance plot of a typical two-way, sealed
box, domestic hi-fi speaker.
The steep rise in impedance at a certain low frequency is indicative of
the low-frequency resonance of the system. Other undulations are indicative of the reactive nature of the speaker due to capacitive and inductive
elements in the crossover components and the drive units themselves.
Also, the driver/box interface has an effect, the most obvious place being at
the already-mentioned LF resonant frequency.
Figure 4.12 shows an impedance plot of a bass reflex design. Here we
see the characteristic ‘double hump’ at the bass end. The high peak at about
70 Hz is the bass driver/cabinet resonance point. The trough at about 40 Hz
is the resonant frequency of the bass reflex port where maximum LF sound
energy is radiated from the port itself and minimum energy is radiated from
Loudspeaker Performance
FIGURE 4.12
Impedance plot of a typical
bass reflex design.
Impedance Ω
32
16
8
4
50
100
1 kHz
10 kHz
Frequency Hz
the bass driver. The low peak at about 20 Hz is virtually equal to the free-air
resonance of the bass driver itself because at very low frequencies the driver
is acoustically unloaded by the cabinet due to the presence of the port opening. A transmission-line design exhibits a similar impedance characteristic.
The DC resistance of an 8 ohm driver or speaker system tends to
lie around 7 ohms, and this simple measurement is a good guide if the
impedance of an unlabeled speaker is to be estimated. Other impedances
encountered include 15 ohm and 4 ohm models. The 4 ohm speakers
are harder to drive because for a given amplifier output voltage they draw
twice as much current. The 15 ohm speaker is an easy load, but its higher
impedance means that less current is drawn from the amplifier and so
the power (volts ⫻ amps) driving the speaker will be correspondingly less.
So a power amplifier may not be able to deliver its full rated power into
this higher impedance. Thus 8 ohms has become virtually standard, and
competently designed amplifiers can normally be expected to drive competently designed speakers. Higher-powered professional power amplifiers can
also be expected to drive two 8 ohm speakers in parallel, giving a resultant
nominal impedance of 4 ohms.
Sensitivity
A loudspeaker’s sensitivity is a measure of how efficiently it converts
electrical sound energy into acoustical sound energy. The principles are
described in Fact File 4.5. Loudspeakers are very inefficient devices indeed.
A typical high-quality domestic speaker system has an efficiency of less
than 1%, and therefore if 20 watts is fed into it the resulting acoustic
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CHAPTER 4: Loudspeakers
output will be less than 0.2 acoustical watts. Almost all of the rest of the
power is dissipated as heat in the voice coils of the drivers. Horn-loaded
systems can achieve a much better efficiency, figures of around 10% being
typical. An efficiency figure is not in itself a very helpful thing to know,
parameters such as sensitivity and power handling being much more useful. But it is as well to be aware that most of the power fed into a speaker
has to be dissipated as heat, and prolonged high-level drive causes high
voice-coil temperatures.
It has been suggested that sensitivity is not an indication of quality. In
fact, it is often found that lower-sensitivity models tend to produce a better sound. This is because refinements in sound quality usually come at
the expense of reduced acoustical output for a given input, and PA speaker
designers generally have to sacrifice absolute sound quality in order to
achieve the high sensitivity and sound output levels necessary for the
intended purpose.
Distortion
Distortion in loudspeaker systems is generally an order of magnitude or
more higher than in other audio equipment. Much of it tends to be second-harmonic distortion whereby the loudspeaker will add frequencies an
octave above the legitimate input signal. This is especially manifest at low
frequencies where speaker diaphragms have to move comparatively large
distances to reproduce them. When output levels of greater than 90 dB
for domestic systems and 105 dB or so for high-sensitivity systems are
FA C T F I L E 4 . 5 L O U D S P E A K E R SENSI TI VI TY
Sensitivity is defined as the acoustic sound output for a
given voltage input. The standard conditions are an input
of 2.83 volts (corresponding to 1 watt into 8 ohms) and
an acoustic SPL measurement at a distance of 1 meter
in front of the speaker. The input signal is pink noise
which contains equal sound energy per octave (see
‘Frequency spectra of non-repetitive sounds’, Chapter 1).
A single frequency may correspond with a peak or dip in
the speaker’s response, leading to an inaccurate overall
assessment. For example, a domestic speaker may have
a quoted sensitivity of 86 dB W⫺1, that is 1 watt of input
will produce 86 dB output at 1 meter.
Sensitivities of various speakers differ quite widely and
this is not an indication of the sound quality. A high-level
professional monitor speaker may have a sensitivity of
98 dB W⫺1 suggesting that it will be very much louder than
its domestic cousin, and this will indeed be the case. Highfrequency PA horns sometimes achieve a value of 118 dB
for just 1 watt input. Sensitivity is thus a useful guide when
considering which types of speaker to choose for a given
application. A small speaker having a quoted sensitivity of
84 dB W⫺1 and 40 watts power handling will not fill a large
hall with sound. The high sound level capability of large
professional models will be wasted in a living room.
Loudspeaker Performance
being produced, low-frequency distortion of around 10% is quite common,
this consisting mainly of second-harmonic and partly of third-harmonic
distortion.
At mid and high frequencies distortion is generally below 1%, this being
confined to relatively narrow bands of frequencies which correspond to
areas such as crossover frequencies or driver resonances. Fortunately, distortion of this magnitude in a speaker does not indicate impending damage,
and it is just that these transducers are inherently non-linear to this extent.
Much of the distortion is at low frequencies where the ear is comparatively
insensitive to it, and also the predominantly second-harmonic character
is subjectively innocuous to the ear. Distortion levels of 10–15% are fairly
common in the throats of high-frequency horns.
Frequency response
The frequency response of a speaker also indicates how linear it is. Ideally,
a speaker would respond equally well to all frequencies, producing a
smooth ‘flat’ output response to an input signal sweeping from the lowest to the highest frequencies at a constant amplitude. In practice, only
the largest speakers produce a significant output down to 20 Hz or so, but
even the smallest speaker systems can respond to 20 kHz. The ‘flatness’
of the response, i.e. how evenly a speaker responds to all frequencies, is
a rather different matter. High-quality systems achieve a response that is
within 6 dB of the 1 kHz level from 80 Hz to 20 kHz, and such a frequency
response might look like Figure 4.13(a). Figure 4.13(b) is an example of a
rather lower-quality speaker which has a considerably more ragged response
and an earlier bass roll-off.
The frequency response can be measured using a variety of different methods, some manufacturers taking readings under the most favorable conditions to hide inadequacies. Others simply quote something like
‘ ⫾ 3 dB from 100 Hz to 15 kHz’. This does at least give a fairly good idea
of the smoothness of the response. These specifications do not, however,
tell you how a system will sound, and they must be used only as a guide.
They tell nothing of coloration levels, or the ability to reproduce good
stereo depth, or the smoothness of the treble, or the ‘tightness’ of the bass.
Power handling
Power handling is the number of watts a speaker can handle before unacceptable amounts of distortion ensue. It goes hand in hand with sensitivity in determining the maximum sound level a speaker can deliver. For
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CHAPTER 4: Loudspeakers
FIGURE 4.13
Typical loudspeaker
frequency response plots.
(a) A high-quality unit. (b)
A lower-quality unit.
(a)
6 dB
100
1 kHz
10 kHz
100
1 kHz
10 kHz
(b)
6 dB
example, a domestic speaker may be rated at 30 watts and have a sensitivity of 86 dB W⫺1. The decibel increase of 30 watts over 1 watt is given by:
dB increase ⫽ 10 log 30 ⫽ 15 dB
Therefore, the maximum output level of this speaker is 86 ⫹ 5 ⫽ 101 dB
at 1 meter for 30 watts input. This is loud, and quite adequate for domestic
use. Consider now a PA speaker with a quoted sensitivity of 99 dB W⫺1. 30
watts input now produces 99 ⫹ 15 ⫽ 114 dB, some 13 dB more than with
the previous example for the same power input. To get 114 dB out of the
86 dB W⫺1 speaker one would need to drive it with no less than 500 watts,
which would of course be way beyond its capabilities. This dramatically
demonstrates the need to be aware of the implications of sensitivity and
power handling.
Loudspeaker Performance
A 30 watt speaker can, however, safely be driven even by a 500 watt amplifier providing that sensible precautions are taken with respect to how hard the
amplifier is driven. Occasional peaks of more than 30 watts will be quite happily tolerated; it is sustained high-level drive which will damage a speaker. It
is perfectly all right to drive a high power speaker with a low-power amplifier,
but care must be taken that the latter is not overdriven otherwise the harsh
distortion products can easily damage high-frequency horns and tweeters even
though the speaker system may have quoted power handling well in excess of
the amplifier. The golden rule is to listen carefully. If the sound is clean and
unstressed, all will be well.
Directivity
Directivity, or dispersion, describes the angle of coverage of a loudspeaker’s
output. Very low frequencies radiated from a speaker are effectively omnidirectional, because the wavelength of the sound is large compared with the dimensions of the speaker and its enclosure, and efficient diffraction of sound around
the latter is the result. As the frequency increases, wavelengths become comparable to the dimensions of the speaker’s front surface, diffraction is curtailed,
and the speaker’s output is predominantly in the forwards direction. At still
higher frequencies, an even narrower dispersion angle results as a further effect
comes into play: off-axis phase cancelation. If one listens, say, 30° off-axis from
the front of a speaker, a given upper frequency (with a short wavelength) arrives
which has been radiated both from the closest side of the speaker cone to the
listener and from the furthest side of the cone, and these two sound sources will
not therefore be in phase with each other because of the different distances they
are away from one another. Phase cancelation therefore occurs, perceived output level falls, and the effect becomes more severe as frequencies increase. The
phenomenon is mitigated by designing for progressively smaller radiating areas
of the speaker cone to be utilized as the frequency increases, finally crossing
over to a tweeter of very small dimensions. By these means, fairly even dispersion of sound, at least in the mid and lower treble regions, can be maintained.
Various other methods have been used to control directivity (the acoustic
lens has been covered) and one or two will be described. Low frequencies,
which are normally omnidirectional, have been given a cardioid-like dispersion pattern by mounting large speaker drivers on essentially open baffles
which by themselves give a figure-of-eight polar response, the output falling
with falling frequency. To these was added a considerable amount of absorbent material to the rear, and together with appropriate bass boost to flatten
the frequency response of the speakers, predominantly forward-radiation of
low frequencies was achieved. A more elegant technique has been to mount
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essentially open-baffle speakers (the rear radiation therefore being 180° out of
phase with the front producing a figure-of-eight polar pattern, and with bass
boost applied to flatten the frequency response) adjacent to closed-box omnidirectional speakers. Their combined acoustical outputs thereby produce a
cardioid dispersion pattern, useful for throwing low frequencies forwards into
an auditorium rather than across a stage where low-frequency feedback with
microphones can be a problem.
A more sophisticated technique has been to use a conventional forward facing subwoofer, adding to it another which is behind, and facing
rearward.
Using DSP processing to give appropriate delay and phase change with
frequency, the rear-facing driver’s output can be made to cancel the sound
which reaches the rear of the enclosure from the front-facing driver. A very
consistent cardioid response can thereby be achieved over the limited range
of frequencies across which the subwoofer operates.
Another fascinating technique, introduced by Philips in 1983, is the
Bessel Array. It was developed to counteract the beaming effects of multiplespeaker systems. Essentially it makes use of Bessel coefficients to specify
phase relationships and output level requirements from each of a horizontal
row of speakers necessary to obtain an overall dispersion pattern from the
row which is the same as one speaker on its own. Normally, path-length
differences between off-axis listeners and the various speaker drivers result
in phase cancelations and consequent loss of level, particularly in the upper
frequency range. For a horizontal five-speaker row, labeled A, B, C, D and
E, the Bessel function gives:
A:B:C:D:E ⫽ 12
: :2: ⫺ 21
:
In other words, speakers A and E are required to draw
half the current of speakers B, C and D; and speaker D
must be connected out of phase. A practical implementation would be to connect speakers A and E in series, with
speakers B, C and D each connected straight across the
system’s input terminals but with D wired out of phase.
The speaker drivers are mounted side by side very close
together to give good results across the frequency range.
For a seven-speaker row, the Bessel function gives:
A:B:C:D:E:F:G ⫽ 12
: :2:0: ⫺ 2:2: ⫺ 1
FIGURE 4.14 A Bessel Array.
Speaker D can therefore be omitted, but a space in
the row must be left in its position so as to preserve the
correct distance relationships between the others.
Setting up Loudspeakers
Both horizontal and vertical rows of speakers can be combined into a
square arrangement so that an array of, for example, 25 speakers, together
having potentially very high power handling and output level capability, can,
however, give the same dispersion characteristics of one speaker on its own.
The amplitude and phase relationships necessary in such an array are given
by the numbers in the circles representing the speakers in Figure 4.14.
It is worth mentioning that the same technique can also be applied
to microphones, offering potential for a high output, very low noise array
whilst still maintaining a good polar response.
A highly directional speaker incorporating a parabolic reflector of about
1.3 meters in diameter has been developed by the Meyer loudspeaker company as their type SB-1. Designed to work between 500 Hz and 15 kHz, the
system comprises an outrigger supporting a small horn and compression
driver at the focus of the dish which fires into it, and a small hole at the
dish’s center admits sound from a 12-inch cone driver. Claimed dispersion
(⫺6 dB points) is 10° vertical and 10° horizontal, and maximum peak output at 100 meters distance is 110 dB.
SETTING UP LOUDSPEAKERS
Phase
Phase is a very important consideration when wiring up speakers. A positivegoing voltage will cause a speaker cone to move in a certain direction, which
is usually forwards, although at least two American and two British manufacturers have unfortunately adopted the opposite convention. It is essential
that both speakers of a stereo pair, or all of the speakers of a particular type
in a complete sound rig, are ‘in phase’, that is all the cones are moving in
the same direction at any one time when an identical signal is applied. If
two stereo speakers are wired up out of phase, this produces vague ‘swimming’ sound images in stereo, and cancelation of bass frequencies. This can
easily be demonstrated by temporarily connecting one speaker in opposite
phase and then listening to a mono signal source – speech from the radio is a
good test. The voice will seem to come from nowhere in particular, and small
movements of the head produce sudden large shifts in apparent sound source
location. Now reconnect the speakers in phase and the voice will come from
a definite position in between the speakers. It will also be quite stable when
you move a few feet to the left or to the right.
Occasionally it is not possible to check the phase of an unknown
speaker by listening. An alternative method is to connect a 1.5 V battery
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across the input terminals and watch which way the cone of the bass driver
moves. If it moves forwards, then the positive terminal of the battery corresponds to the positive input terminal of the speaker. If it moves backwards as the battery is connected, then the positive terminal of the battery
is touching the negative input terminal of the speaker. The terminals can
then be labeled ⫹ and ⫺.
Positioning
Loudspeaker positioning has a significant effect upon the performance. In
smaller spaces such as control rooms and living rooms the speakers are
likely to be positioned close to the walls, and ‘room gain’ comes into effect
whereby the low frequencies are reinforced. This happens because at these
frequencies the speaker is virtually omnidirectional, i.e. it radiates sound
equally in all directions. The rear- and side-radiated sound is therefore
reflected off the walls and back into the room to add more bass power. As
we move higher in frequency, a point is reached whereby the wavelength
of lower mid frequencies starts to become comparable with the distance
between the speaker and a nearby wall. At half wavelengths the reflected
sound is out of phase with the original sound from the speaker and some
cancelation of sound is caused. Additionally, high-frequency ‘splash’ is
often caused by nearby hard surfaces, this often being the case in control
rooms where large consoles, tape machines, outboard processing gear, etc.
can be in close proximity to the speakers. Phantom stereo images can thus
be generated which distort the perspective of the legitimate sound. A loudspeaker which has an encouragingly flat frequency response can therefore
often sound far from neutral in a real listening environment. It is therefore
essential to give consideration to loudspeaker placement, and a position
such that the speakers are at head height when viewed from the listening
position (high-frequency dispersion is much narrower than at lower frequencies, and therefore a speaker should be listened to on axis) and also
away from room boundaries will give the most tonally accurate sound.
Some speakers, however, are designed to give their best when mounted
directly against a wall, the gain in bass response from such a position being
allowed for in the design. A number of professional studio monitors are
designed to be let into a wall such that their drivers are then level with the
wall’s surface. The manufacturers’ instructions should be heeded, in conjunction with experimentation and listening tests. Speech is a good test signal. Male speech is good for revealing boominess in a speaker, and female
speech reveals treble splash from hard-surfaced objects nearby. Electronic
music is probably the least helpful since it has no real-life reference by
Thiele – Small Parameters and Enclosure Volume Calculations
which to assess the reproduced sound. It is worth emphasizing that the
speaker is the means by which the results of previous endeavor are judged,
and that time spent in both choosing and siting is time well spent.
Speakers are of course used in audio-visual work, and one frequently
finds that it is desirable to place a speaker next to a video monitor screen.
But the magnetic field from the magnets can affect the picture quality by
pulling the internal electron beams off course. Some speakers are specially
magnetically screened so as to avoid this.
Loudspeaker positioning issues affecting two-channel stereo and surround
sound reproduction are covered in greater detail in Chapters 16 and 17.
THIELE–SMALL PARAMETERS AND ENCLOSURE
VOLUME CALCULATIONS
Low-frequency performance of a driver/box combination is one of the few
areas of loudspeaker design where the performance of the practical system
closely resembles the theoretical design aims. This is because at low frequencies the speaker cone acts as a pure piston, and wavelengths are long,
minimizing the effects of enclosure dimensions and objects close to the
speaker. Nearby boundaries, e.g. walls and the floor, have a significant effect
at very low frequencies, but these are predictable and easily allowed for.
It was A.N. Thiele and Richard Small, working largely independently of
each other in Australia mainly during the 1960s, who modeled driver and
enclosure behavior in terms of simple electrical circuits. They substituted,
for instance, the DC resistance of the coil with a resistor, its inductance with
an inductor of the same value, and the places where the speaker’s impedance rose with decreasing frequency could be represented by a capacitor of
an appropriate value in the circuit model. The latter could also represent the
‘stiffness’ of the air enclosed in the box. Electrical formulae could then be
applied to these models to predict the behavior of particular drive units in
various sizes and types of enclosure. A series of ‘Thiele–Small’ parameters
are therefore associated with a particular drive unit and they enable systems
to be designed, the low frequency performance of which can be predicted
with considerable accuracy, something which had previously been a largely
empirical affair before their work. A host of parameters are specified by the
manufacturers, and include such things as magnet flux density, sensitivity,
diaphragm moving mass, mechanical resistance of suspension, force factor,
equivalent volume of suspension compliance, mechanical Q, electrical Q,
and free air resonance. The list is a comprehensive description of the drive
unit in question, but fortunately only three need to be considered when
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designing an enclosure for the target low-frequency performance. These are
the free-air resonance, represented by the symbol fo; the equivalent air volume of the suspension compliance, VAS; and the total Q of the driver, QT.
They will be considered in turn.
fo is the free-air resonance of the driver, determined by taking an impedance plot and noting the frequency at which a large, narrow peak in the
impedance takes place. Such a rise can be seen in Figure 4.12 at about 20 Hz.
VAS can be explained as follows. Imagine a bass driver with an infinitely
compliant suspension, its cone being capable of being moved to and fro
with the fingers with no effort. Now mount the driver in a closed box of
say 70 liters internal volume. Push the cone with the fingers again, and the
spring of the enclosed air now supports the cone, and one feels an impedance as one pushes against that spring. The suspension of the drive unit is
thus specified as an equivalent air volume. The enclosure volume of both
closed-box and reflex systems is always smaller than the drive unit’s VAS in
order that the air stiffness loads the cone of the speaker adequately, as its
own suspension system is insufficient to control low frequency excursion
alone.
QT, the total Q of the driver, is the average between the electrical Q, QE,
and the mechanical Q, QM. Briefly, QE is determined using a formula containing total moving mass of the diaphragm, the resistance of the coil, and
the Bl factor (flux density multiplied by a length of coil wire immersed in
the magnetic field, indicating the force with which the coil pushes the cone
for a given input level). QM is determined by calculating the Q of the lowfrequency peak in the impedance plot, dividing the center peak frequency by
the bandwidth at the ⫺3 dB points each side of this. QT is always quoted,
so one does not need to calculate it from the other parameters.
Before moving on to discuss how the above is used in calculations, system Q must be looked at. This is explained in Fact File 4.6.
For the following discussions, a Q of 0.7 will be assumed. Different Q
values can be substituted by the reader to explore the effect this has on
enclosure volume.
For a closed box (‘infinite baffle’) system, a three-stage process is
involved. The following formula is used first:
QTC /QT ⫽ x ⫽ f3 / f0
where QTC is the chosen system Q (assumed to be 0.7), and f3 is the resonant frequency of the system. x is the ratio between the two quantities. For
example, if the driver’s QT is 0.35, then x is 2. If the driver’s fo is 25 Hz,
Thiele – Small Parameters and Enclosure Volume Calculations
FA C T F I L E 4 . 6 L O W-FR E Q U E N C Y Q
The graph shows a family of curves for various possible
low-frequency alignments. A system Q of 0.7 for a closed
box is usually the target figure for medium-sized enclosures
in both the domestic and the sound reinforcement contexts. The roll-off is a smooth 12 dB per octave, and there
is no emphasis at any frequency. A Q of 0.7 means that the
response is 3 dB down compared with the flat part of the
frequency response above it. (Refer to Q in the ‘Glossary
of terms’ for a discussion of how Q value relates to dB of
attenuation at the point of resonance.) A Q of 0.6 has an
earlier but gentler bass roll-off, and the frequency response
is about 4.5 dB down at resonance. Q values above 0.7
progressively overemphasize the response at resonance,
producing an undesirable ‘hump’. Large enclosures for
domestic use have to be designed with room boundary
gain taken into consideration. A model with an impressively extended response down to very low frequencies in
an anechoic chamber will often sound ‘boomy’ in a room
because the low frequencies, which are omni-directional,
reflect off the rear wall and floor to reinforce the primary
sound from the speaker, adding to its output. A Q value of
0.6 or even 0.5 is therefore best chosen for large domestic
enclosures so that the total combined response in a real
listening environment is more even and natural.
For very small speakers, a Q value of greater that 0.7,
say 0.9 or slightly more, can be chosen which gives slight
bass emphasis and helps give the impression of a ‘fuller’
sound from a small enclosure. Such an alignment is used
judiciously so as to avoid overemphasizing the upper bass
frequencies.
Q = 1.4
+6
+3
dB
0
−3
−6
−9
−12
−15
Q=1
Q = .7
Q = .5
f3
then f3 is 50 Hz. Therefore, such a driver in a box giving a Q of 0.7 will
have a resonant frequency of 50 Hz.
The next stage is to calculate the box volume required to achieve this performance. For this, x needs first to be converted into α, the compliance ratio.
This is the ratio between the drive unit’s VAS and the box volume. The latter
is always smaller than the former. This is done using the simple formula:
x2 ⫺ 1 ⫽ α
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In the present example, this gives an α of 3. To calculate box size, the
following formula is used:
α ⫽
VAS
VB
where VB is the box volume. If the driver has a VAS of 80 liters, this then
gives a box size of about 27 liters. These results are quite typical of a
medium-sized domestic speaker system.
The bass reflex design is rather more complex. This consists of a box
with a small port or ‘tunnel’ mounted in one of its walls, the air plug in
the port together with the internal air volume resonating at a particular
low frequency. At this frequency, maximum sound output is obtained from
the port, and the drive unit’s cone movement is reduced compared with a
closed box system. First, we will look at the formula for the enclosure:
fc ⫽
344.8R
2 (πVb [ L ⫻ 1.7])
where R is the port radius (assuming a circular port), L is the port length,
and Vb is the enclosure volume. All dimensions are in meters; Vb is in cubic
meters. fc is the resonant frequency of the system. 344.8 is the speed of
sound in meters per second at normal temperatures and pressures. The port
can in principle have any cross-sectional shape, and more than one port can
be used. It is the total cross-sectional area combined with the total length of
the port or ports which are required for the calculation. Note that nowhere
does the drive unit in question appear in the calculation. The box with port
is a resonant system alone, and the design must be combined with drive
unit calculations assuming a closed box system with a target Q rather lower
than is customary for a closed box, usually much nearer to 0.5 so that the
driver has a slow low-frequency roll-off as the output from the port rises,
producing a smooth transition. The reflex enclosure volume is therefore
typically in the order of about 80% of the drive unit’s VAS. Looking again at
Figure 4.12, we see two low-frequency peaks in the impedance with a trough
in between. The lowest one is the free-air resonance of the driver (altered
slightly by air loading), the trough is the reflex port resonant frequency, and
the upper peak is the box/driver resonant frequency. The designer must
ensure that the design arrived at with the particular chosen driver ensures
these conditions. One does not, for instance, design for a port resonant frequency of 30 Hz when the drive unit’s free-air resonance is 40 Hz.
Thiele – Small Parameters and Enclosure Volume Calculations
The design procedure is normally to calculate for a closed-box system
with a chosen drive unit, the target Q being closer to 0.5 than to 0.7. (Reflex
systems are larger than closed-box systems for a given driver.) One notes the
driver/box resonant frequency, and then chooses port dimensions which give
a port resonance which is midway between the driver/box resonance and the
driver’s free-air resonance. Final dimensions will be chosen during prototyping for optimum subjective results in a real listening environment.
Above port resonance, the output from the port falls at the rate of 12 dB/
octave, and the design aim is to give a smooth transition between the port’s
output and the driver’s output. The port gives a useful degree of bass extension. Below resonance, the speaker cone simply pumps air in and out of the
port, and the latter’s output is therefore 180° out of phase with the former’s, producing a rapid 24 dB/octave roll-off in the response. Furthermore,
the drive unit is not acoustically loaded below resonance, and particularly
in sound reinforcement use where very high powers are involved care must
be taken to curtail very low-frequency drive to reflex enclosures, otherwise
excessive cone excursions combined with large currents drawn from the
amplifiers can cause poor performance and premature failure.
The abrupt 24 dB/octave roll-off of the reflex design means that it interfaces with room boundaries less successfully than does a closed-box system
with its more gradual roll-off. However, some reflex designs are deliberately
‘de-tuned’ to give a less rapid fall in response, helping to avoid a ‘boomy ’
bass quality when the speaker is placed close to walls.
Drive units with QT values of 0.2 and below are well suited to bass
reflex use. Drivers with QT values of 0.3 and above are better suited to
closed-box designs.
In the above discussions, no mention has been made of sound absorbent
material in the box, either in the form of foam lining or wadding filling the
volume of the enclosure. This should not be necessary for low-frequency
considerations, and it is usually included to absorb mid frequency energy
from the rear of the speaker cone to prevent it from re-emerging back
through the cone and the enclosure walls, coloring the sound. However, the
presence particularly of volume-filling material has the effect of reducing
the speed of sound in the enclosure, making it apparently larger, sometimes
by as much as 15% for some types of filling. This must be taken into consideration in the final design. An over-dense filling tends to behave in a
manner more like a solid mass, and the box volume is apparently reduced.
There is no reason in principle to include sound absorbent material in a
box intended purely for low-frequency use, unless one wishes to increase its
apparent acoustic volume for economic or space-saving reasons.
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DIGITAL SIGNAL PROCESSING IN LOUDSPEAKERS
Digital signal processing (DSP) is used increasingly in loudspeakers to compensate for a range of linear and non-linear distortion processes that typically
arise. DSP can also be used in crossover design and for controlling the spatial radiation characteristics of loudspeakers or loudspeaker arrays. With the
help of such technology it may be possible to get better performance out of
smaller loudspeaker units by using electronics to counteract physical inadequacies. Some such processes can make use of psychoacoustical phenomena,
such as a means of extending the perceived bass response without actually
reproducing the relevant low frequencies, and it may also be possible to
modify the way in which the loudspeaker interacts with the listening room.
Finally, there are various ways by which it may be possible to engineer an
‘all-digital’ signal chain, even using digital forms of representation right up
to the point where the binary data is converted into an acoustical waveform.
RECOMMENDED FURTHER READING
Borwick, J. (Ed.) 2001. Loudspeaker and Headphone Handbook. Focal Press.
Colloms, M., 2005. High Performance Loudspeakers, sixth ed. Wiley.
Eargle, J., 2003. Loudspeaker Handbook. Kluwer Academic Publishers.
CHAPTER 5
Mixers
CH A P T E R C O N TE N T S
A Simple Six-channel Analog Mixer
Overview
Input channels
Output section
Miscellaneous features
A Multitrack Mixer
Overview
In-line and split configurations
Further aspects of the in-line design
Channel Grouping
An Overview of Typical Mixer Facilities
Input section
Routing section
Dynamics section
Equalizer section
Channel and mix controls
Auxiliary sends
Master control section
Effects returns
Patchbay or jackfield
EQ Explained
Principal EQ bands
Filters
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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C H A P TE R C O N TENTS
Stereo Line Input Modules
Dedicated Monitor Mixer
Technical Specifications
Input noise
Output noise
Impedance
Frequency response
Distortion
Crosstalk
Metering Systems
Mechanical metering
Problems with mechanical meters
Electronic bargraph metering
Relationship between different metering standards
Meter take-off point
Automation
Background
Fader automation
Grouping automated faders
Mute automation
Storing the automation data
Integrating machine control
Retrofitting automation
Total automation systems
Dynamic and static systems
Digital Mixers
Audio handling
Assignable control surfaces
Digital mixers – a case study
Digitally controlled analog mixers
Mixers with Integrated Control of Digital Workstations
Introduction to Mixing Approaches
Basic Operational Techniques
Level setting
Using auxiliary sends
Using audio groups
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A Simple Six-channel Analog Mixer
This chapter describes the principles and basic operation of audio mixers.
It begins with a description of a simple system and moves on to consider
the facilities of large-scale multitrack systems. Because many design and
layout concepts of analog mixers find a place in more recent digital mixers,
these aspects are covered here in a fairly generic way. Those features found
more commonly only in digital systems are described towards the end of
the chapter.
In its simplest form an audio mixer combines several incoming signals
into a single output signal. This cannot be achieved simply by connecting all the incoming signals in parallel and then feeding them into a single
input because they may influence each other. The signals need to be isolated from each other. Individual control of at least the level of each signal
is also required.
In practice, mixers do more than simply mix. They can provide phantom
power for capacitor microphones (see ‘The capacitor or condenser microphone’, Chapter 3); pan control (whereby each signal can be placed in any
desired position in a stereo image); filtering and equalization; routing facilities;
and monitoring facilities, whereby one of a number of sources can be routed
to loudspeakers for listening, often without affecting the mixer’s main output.
A SIMPLE SIX-CHANNEL ANALOG MIXER
Overview
By way of example, a simple six-channel analog mixer will be considered,
having six inputs and two outputs (for stereo). Figure 5.1 illustrates such
a notional six-into-two mixer with basic facilities. It also illustrates the
back panel. The inputs illustrated are via XLR-type three-pin latching connectors, and are of a balanced configuration. Separate inputs are provided
for microphone and line level signals, although it is possible to encounter
systems which simply use one socket switchable to be either mic or line.
Many cheap mixers have unbalanced inputs via quarter-inch jack sockets,
or even ‘phono’ sockets such as are found on hi-fi amplifiers. Some mixers employ balanced XLR inputs for microphones, but unbalanced jack
or phono inputs for line level signals, since the higher-level line signal is
less susceptible to noise and interference, and will probably have traveled a
shorter distance.
On some larger mixers a relatively small number of multipin connectors are provided, and multicore cables link these to a large jackfield which
consists of rows of jack sockets mounted in a rack, each being individually
labeled. All inputs and outputs will appear on this jackfield, and patch cords
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CHAPTER 5: Mixers
LINE/
MIC
GAIN
SLATE
HF
MIC
LEVEL
LF
PAN
MONITOR
LEVEL
OUTPUT/PFL
OUTPUTS
L
R
PFL
+10
LEVEL
0
–10
PHONES
–20
–40
48 V
MAIN OUTPUTS
MICS
MONITOR O/Ps
LINE
POWER
FIGURE 5.1 Front panel and rear connectors of a typical simple six-channel mixer.
of a meter or so in length with GPO-type jack plugs at each end enable
the inputs and outputs to be interfaced with other equipment and tie-lines
in any appropriate combination. (The jackfield is more fully described in
‘Patchfield or jackfield’, below, and ‘Jackfields (patchbays)’, Chapter 12.)
A Simple Six-channel Analog Mixer
The outputs are also on three-pin XLR-type connectors. The convention
for these audio connections is that inputs have sockets or holes, outputs
have pins. This means that the pins of the connectors ‘point’ in the direction
of the signal, and therefore one should never be confused as to which connectors are inputs and which are outputs. The microphone inputs also
have a switch each for supplying 48 V phantom power to the microphones
if required. Sometimes this is found on the input module itself, or sometimes on the power supply, switching 48 V for all the inputs at once.
The term ‘bus’ is frequently used to describe a signal path within the
mixer to which a number of signals can be attached and thus combined.
For instance, routing some input channels to the ‘stereo bus’ conveys those
channels to the stereo output in the manner of a bus journey in the conventional everyday sense. A bus is therefore a mixing path to which signals
can be attached.
Input channels
All the input channels in this example are identical, and so only one will
be described. The first control in the signal chain is input gain or sensitivity. This control adjusts the degree of amplification provided by the input
amplifier, and is often labeled in decibels, either in detented steps or continuously variable. Inputs are normally switchable between mic and line.
In ‘mic’ position, depending on the output level of the microphone connected to the channel (see ‘Microphone performance’, Chapter 3), the input
gain is adjusted to raise the signal to a suitable line level, and up to 80 dB
or so of gain is usually available here (see ‘Miscellaneous features’, below).
In ‘line’ position little amplification is used and the gain control normally
provides adjustment either side of unity gain (0 dB), perhaps ⫾20 dB either
way, allowing the connection of high-level signals from such devices as CD
players, tape machines and musical keyboards.
The equalization or EQ section which follows (see ‘Equalizer section’,
below) has only two bands in this example – treble and bass – and these
provide boost and cut of around ⫾12 dB over broad low and high-frequency
bands (e.g. centered on 100 Hz and 10 kHz). This section can be used like the
tone controls on a hi-fi amplifier to adjust the spectral balance of the signal.
The fader controls the overall level of the channel, usually offering a small
amount of gain (up to 12 dB) and infinite attenuation. The law of the fader
is specially designed for audio purposes (see Fact File 5.1). The pan control
divides the mono input signal between left and right mixer outputs, in order
to position the signal in a virtual stereo sound stage (see Fact File 5.2).
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FA C T F I L E 5. 1 FA D E R FA C TS
Fader law
Channel and output faders, and also rotary level controls,
can have one of two laws: linear or logarithmic (the latter
sometimes also termed ‘audio taper’). A linear law means
that a control will alter the level of a signal (or the degree
of cut and boost in a tone control circuit) in a linear fashion: that is, a control setting midway between maximum
and minimum will attenuate a signal by half its voltage,
i.e. ⫺6 dB. But this is not a very good law for an audio
level control because a 6 dB drop in level does not produce a subjective halving of loudness. Additionally, the
rest of the scaling (⫺10 dB, ⫺20 dB, ⫺30 dB and so on)
has to be accommodated within the lower half of the control’s travel, so the top half gives control over a mere 6 dB,
the bottom half all the rest.
For level control, therefore, the logarithmic or ‘log’
law is used whereby a non-linear voltage relationship is
employed in order to produce an approximately even spacing when the control is calibrated in decibels, since the
decibel scale is logarithmic. A log fader will therefore attenuate a signal by 10 dB at a point approximately a quarter
of the way down from the top of its travel. Equal dB increments will then be fairly evenly spaced below this point. A
rotary log pot (‘pot’ is short for potentiometer) will have its
maximum level usually set at the 5 o’clock position and,
the ⫺10 dB point will be around the 2 o’clock position.
An even subjective attenuation of volume level is therefore produced by the log law as the control is gradually
turned down. A linear law causes very little to happen
subjectively until one reaches the lowest quarter of the
range, at which point most of the effect takes place.
The linear law is, however, used where a symmetrical
effect is required about the central position; for example,
the cut and boost control of a tone control section will
have a central zero position about which the signal is cut
and boosted to an equal extent either side of this.
Electrical quality
There are two types of electrical track in use in analog
faders, along which a conductive ‘wiper’ runs as the fader
is moved to vary its resistance. One type of track consists
of a carbon element, and is cheap to manufacture. The
quality of such carbon tracks is, however, not very consistent and the ‘feel’ of the fader is often scrapy or grainy,
and as it is moved the sound tends to jump from one
level to another in a series of tiny stages rather than in
a continuous manner. The carbon track wears out rather
quickly, and can become unreliable.
The second type employs a conductive plastic track.
Here, an electrically conductive material is diffused into a
strip of plastic in a controlled manner to give the desired
resistance value and law (linear or log). Much more
expensive than the carbon track, the conductive plastic
track gives smooth, continuous operation and maintains
this standard over a long period of time. It is the only serious choice for professional-quality equipment.
Output section
The two main output faders (left and right) control the overall level of all
the channel signals which have been summed on the left and right mix
buses, as shown in the block diagram (Figure 5.2). The outputs of these
faders (often called the group outputs) feed the main output connectors
on the rear panel, and an internal feed is taken from the main outputs to
the monitor selector. The monitor selector on this simple example can be
A Simple Six-channel Analog Mixer
FA C T F I L E 5 . 2 PA N C O N T R O L
the left and right outputs of that channel, since two identical signals summed together will give a rise in level of
6 dB. A pot which gives a 6 dB drop in the center results
in no level rise for centrally panned signals in the mono
sum. Unfortunately, the 3 dB drop works best for stereo
reproduction, resulting in no perceived level rise for centrally panned signals.
Only about 18 dB of level difference is actually
required between left and right channels to give the
impression that a source is either fully left or fully right
in a loudspeaker stereo image, but most pan-pots are
designed to provide full attenuation of one channel when
rotated fully towards the other. This allows for the two
buses between which signals are panned to be treated
independently, such as when a pan control is used to
route a signal either to odd or even channels of a multitrack bus (see ‘Routing section’, below).
0
–3dB
–6
Gain (dB)
The pan control on a mixer is used for positioning a signal somewhere between left and right in the stereo mix
image. It does this by splitting a single signal from the
output of a fader into two signals (left and right), setting
the position in the image by varying the level difference
between left and right channels. It is thus not the same as
the balance control on a stereo amplifier, which takes in a
stereo signal and simply varies the relative levels between
the two channels. A typical pan-pot law would look similar
to that shown in the diagram, and ensures a roughly constant perceived level of sound as the source is panned
from left to right in stereo. The output of the pan-pot usually feeds the left and right channels of the stereo mix
bus (the two main summation lines which combine the
outputs of all channels on the mixer), although on mixers
with more than two mix buses the pan-pot’s output may
be switched to pan between any pair of buses, or perhaps
simply between odd and even groups (see Fact File 5.4).
On some older consoles, four-way routing is provided
to a quadraphonic mix bus, with a left–right pot and a
front–back pot. These are rare now. Many stereo panpots use a dual-gang variable resistor which follows a law
giving a 4.5 dB level drop to each channel when panned
centrally, compared with the level sent to either channel at the extremes. The 4.5 dB figure is a compromise
between the ⫺3 dB and ⫺6 dB laws. Pan-pots which only
drop the level by 3 dB in the center cause a rise in level of
any centrally panned signal if a mono sum is derived from
–12
–18
R channel
–24
Left
L channel
Center
switched to route either the main outputs or the PFL bus (see Fact File 5.3)
to the loudspeakers. The monitor gain control adjusts the loudspeaker output level without affecting the main line output level, but of course any
changes made to the main fader gain will affect the monitor output.
The slate facility on this example allows for a small microphone
mounted in the mixer to be routed to the main outputs, so that comments from the engineer (such as take numbers) can be recorded on a tape
machine connected to the main outputs. A rotary control adjusts the slate
level.
Right
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Input channel
FIGURE 5.2
Block diagram of a typical
signal path from channel
input to main output on a
simple mixer.
Master section
Mix bus
L
R
To PFL
To monitor
selector
Left output
fader
EQ
Input
amplifier
Fader
Pan
control
Main
O/P
Right output
fader
FA C T F I L E 5. 3 P R E -FA D E LIS T EN ( PFL)
Pre-fade listen, or PFL, is a facility which enables a signal
to be monitored without routing it to the main outputs of
the mixer. It also provides a means for listening to a signal
in isolation in order to adjust its level or EQ.
Normally, a separate mono mixing bus runs the
length of the console picking up PFL outputs from each
channel. A PFL switch on each channel routes the signal from before the fader of that channel to the PFL bus
(see diagram), sometimes at the same time as activating
internal logic which switches the mixer’s monitor outputs
to monitor the PFL bus. If no such logic exists, the mixer’s monitor selector will allow for the selection of PFL, in
which position the monitors will reproduce any channel
currently with its PFL button pressed. On some broadcast and live consoles a separate small PFL loudspeaker
is provided on the mixer itself, or perhaps on a separate
output, in order that selected sources can be checked
without affecting the main monitors.
Sometimes PFL is selected by ‘overpressing’ the
channel fader concerned at the bottom of its travel (i.e.
pushing it further down). This activates a microswitch
which performs the same functions as above. PFL has
great advantages in live work and broadcasting, since it
allows the engineer to listen to sources before they are
faded up (and thus routed to the main outputs which
would be carrying the live program). It can also be used
in studio recording to isolate sources from all the others
without cutting all the other channels, in order to adjust
equalization and other processing with greater ease.
PFL
bus
PFL
Channel
fader
Main
mix
bus
A Multitrack Mixer
Miscellaneous features
Professional-quality microphones have an output
impedance of around 200 ohms, and the balanced
microphone inputs will have an input impedance of between 1000 and 2000 ohms (‘2 kΩ‘,
k ⫽ thousand). The outputs should have an impedance of around 200 ohms or lower. The headphone
output impedance will typically be 100 ohms or so.
Small mixers usually have a separate power supply
which plugs into the mains. This typically contains a mains transformer, rectifiers and regulating
circuitry, and it supplies the mixer with relatively
low DC voltages. The main advantage of a separate
power supply is that the mains transformer can
be sited well away from the mixer, since the alter- FIGURE 5.3 A compact stereo mixer: the Seemix
nating 50 Hz mains field around the former can ‘Seeport’. (Courtesy of Seemix Sound AS.)
be induced into the audio circuits. This manifests
itself as ‘mains hum’ which is only really effectively dealt with by increasing
the distance between the mixer and the transformer. Large mixers usually
have separate rack-mounting power supplies.
The above-described mixer is very simple, offering few facilities, but it
provides a good basis for the understanding of more complex models. A
typical commercial example of a compact mixer is shown in Figure 5.3.
A MULTITRACK MIXER
Overview
The stereo mixer outlined in the previous section forms only half the story
in a multitrack recording environment. Conventionally, popular music
recording involves at least two distinct stages: the ‘track-laying’ phase and
the ‘mixdown’ phase. In the former, musical tracks are layed down on a
multitrack recorder in stages, with backing tracks and rhythm tracks being
recorded first, followed by lead tracks and vocals. In the mixdown phase,
all the previously recorded tracks are played back through the mixer and
combined into a stereo or surround mix to form the finished product which
goes to be made into a commercial release. Since the widespread adoption
of electronic instruments and MIDI-controlled equipment (see Chapter 14),
MIDI-sequenced sound sources are often played directly into the mix in the
second stage.
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FIGURE 5.4
In multitrack recording two
signal paths are needed –
one from mic or line input
to the multitrack recorder,
and one returning from the
recorder to contribute to a
‘monitor’ mix.
Mic
or
line
Source
To
MT
Input
Processing
Multitrack
replay
Fader
Pan
Routing
To
stereo mix
For these reasons, as well as requiring mixdown signal paths from many
inputs to a stereo bus the mixer also requires signal paths for routing many
input signals to a multitrack recorder. Often it will be necessary to perform
both of these functions simultaneously – that is, recording microphone
signals to multitrack whilst also mixing the return from multitrack into
stereo, so that the engineer and producer can hear what the finished result
will sound like, and so that any musicians who may be overdubbing additional tracks can be given a mixed feed of any previously recorded tracks in
headphones. The latter is known as the monitor mix and this often forms
the basis for the stereo mixdown when the tracklaying job is finished.
So there are two signal paths in this case: one from the microphone or
line source to the multitrack recorder, and one from the multitrack recorder
back to the stereo mix, as shown in Figure 5.4. The path from the microphone input which usually feeds the multitrack machine will be termed the
channel path, whilst the path from the line input or tape return which usually feeds the stereo mix will be termed the monitor path.
It is likely that some basic signal processing such as equalization will
be required in the feed to the multitrack recorder (see below), but the more
comprehensive signal processing features are usually applied in the mixdown path. The situation used to be somewhat different in the American
market where there was a greater tendency to record on multitrack
‘wet’, that is with all effects and EQ, rather than applying the effects on
mixdown.
In-line and split configurations
As can be seen from Figure 5.4, there are two complete signal paths, two
faders, two sets of EQ, and so on. This takes up space, and there are two
A Multitrack Mixer
From mics
To multitrack
Input modules
Channel faders
From multitrack
Master
section
Stereo mix
Monitor
mixer
Monitor faders
ways of arranging this physically, one known as the split-monitoring, or
European-style console, the other as the in-line console. The split console is
the more obvious of the two, and its physical layout is shown in Figure 5.5.
It contains the input channels on one side (usually the left), a master control section in the middle, and the monitor mixer on the other side. So it
really is two consoles in one frame. It is necessary to have as many monitor
channels as there are tracks on the tape, and these channels are likely to
need some signal processing. The monitor mixer is used during track laying for mixing a stereo version of the material that is being recorded, so
that everyone can hear a rough mix of what the end result will sound like.
On mixdown every input to the console can be routed to the stereo mix bus
so as to increase the number of inputs for outboard effects, etc. and so that
the comprehensive facilities provided perhaps only on the left side of the
console are available for the multitrack returns.
This layout has advantages in that it is easily assimilated in operation, and it makes the channel module less cluttered than the in-line
design (described below), but it can make the console very large when a
lot of tracks are involved. It can also increase the build cost of the console because of the near doubling in facilities and metalwork required,
and it lacks flexibility, especially when switching over from track laying to
remixing.
The in-line layout involves the incorporation of the monitor paths from
the right-hand side of the split console (the monitor section) into the left
side, rather as if the console were sawn in half and the right side merged
with the left, as shown in Figure 5.6. In this process a complete monitor
FIGURE 5.5
A typical ‘split’ or
‘European-style’ multitrack
mixer has input modules
on one side and monitor
modules on the other: two
separate mixers in effect.
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FIGURE 5.6
A typical ‘in-line’ mixer
incorporates two signal
paths in one module,
providing two faders per
module (one per path).
This has the effect of
reducing the size of the
mixer for a given number of
channels, when compared
with a split design.
From
multitrack
To multitrack
From
mics
Stereo mix
In-line modules
Master
section
(Additional
in-line
modules)
Small faders
Large faders
FIGURE 5.7
The in-line design allows for
sound processing facilities
such as EQ and dynamics
to be shared or switched
between the signal paths.
Small fader
Channel input
Routing
Processing
Monitor
mix
Monitor input
Large fader
signal path is fitted into the module of the same-numbered channel path,
making it no more than a matter of a few switches to enable facilities to be
shared between the two paths. In such a design each module will contain
two faders (one for each signal path), but usually only one EQ section, one
set of auxiliary sends (see below), one dynamics control section, and so on,
with switches to swap facilities between paths. (A simple example showing only the switching needed to swap one block of processing is shown in
Figure 5.7.) Usually this means that it is not possible to have EQ in both the
multitrack recording path and the stereo mix path, but some more recent
designs have made it possible to split the equalizer so that some frequencyband controls are in the channel path whilst others are in the monitor
path. The band ranges are then made to overlap considerably which makes
the arrangement quite flexible.
A Multitrack Mixer
Further aspects of the in-line design
It has already been stated that there will be two main faders associated with
each channel module in an in-line console: one to control the gain of each
signal path. Sometimes the small fader is not a linear slider but a rotary
knob. It is not uniformly agreed as to whether the large fader at the bottom
of the channel module should normally control the monitor level of the
like-numbered tape track or whether it should control the channel output
level to multitrack tape. Convention originally had it that American consoles made the large fader the monitor fader in normal operation, while
British consoles tended to make it the channel fader. Normally their functions can be swapped over, depending on whether one is mixing down or
track laying, either globally (for the whole console), in which case the fader
swap will probably happen automatically when switching the console from
‘recording’ to ‘remix’ mode, or on individual channels, in which case the
operation is usually performed using a control labeled something like ‘fader
flip’, ‘fader reverse’ or ‘changeover ’. The process of fader swapping is mostly
used for convenience, since more precise control can be exercised over a
large fader near the operator than over a small fader which is further away,
so the large fader is assigned to the function that is being used most in the
current operation. This is coupled with the fact that in an automated console it is almost invariably the large fader that is automated, and the automation is required most in the mixdown process.
Confusion can arise when operating in-line mixers, such as when a
microphone signal is fed into, say, mic input 1 and is routed to track 13 on
the tape. In such a case the operator will control the monitor level of that
track (and therefore the level of that microphone’s signal in the stereo mix)
on monitor fader 13, whilst the channel fader on module 1 will control the
multitrack record level for that mic signal.
If a 24 track recorder is in use with the mixer, then monitor faders
higher than number 24 will not normally carry a tape return, but will be
free for other sources. More than one microphone signal can be routed
to each track on the tape (as each multitrack output on the mixer has its
own mix bus), so there will be a number of level controls that affect each
source’s level in the monitor mix, each of which has a different purpose:
■
MIC LEVEL TRIM – adjusts the gain of the microphone pre-amplifier
at the channel input. Usually located at the top of the module.
■
CHANNEL FADER – comes next in the chain and controls the
individual level of the mic (or line) signal connected to that module’s
input before it goes to tape. Located on the same-numbered module
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CHAPTER 5: Mixers
as the input. (May be switched to be either the large or small fader,
depending on configuration.)
■
BUS TRIM or TRACK SUBGROUP – will affect the overall level of
all signals routed to a particular track. Usually located with the track
routing buttons at the top of the module. Sometimes a channel fader
can be made to act as a subgroup master.
MONITOR FADER – located in the return path from
the multitrack recorder to the stereo mix. Does not affect the
recorded level on the multitrack tape, but affects the level of
this track in the mix. (May be switched to be either the large or
small fader, depending on configuration.)
■
A typical in-line multitrack mixer is shown in the photograph
in Figure 5.8.
CHANNEL GROUPING
Grouping is a term that refers to the simultaneous control of more
than one signal at a time. It usually means that one fader controls the levels of a number of slave channels. Two types of channel grouping are currently common: audio grouping and ‘control’
grouping. The latter is often called VCA grouping, but there are
other means of control grouping that are not quite the same as
the direct VCA control method. The two approaches have very
different results, although initially they may appear to be similar
FIGURE 5.8 A typical in-line mixer: because one fader appears to control a number of signal levels.
the Soundcraft ‘Sapphyre’. (Courtesy of
The primary reason for using group faders of any kind is in order
Soundcraft Electronics Ltd.)
to reduce the number of faders that the engineer has to handle at
a time. This can be done in a situation where a number of channels are carrying audio signals that can be faded up and down together. These
signals do not all have to be at the same initial level, and indeed one is still free
to adjust levels individually within a group. A collection of channels carrying
drum sounds, or carrying an orchestral string section, would be examples of
suitable groups. The two approaches are described in Fact Files 5.4 and 5.5.
AN OVERVIEW OF TYPICAL MIXER FACILITIES
Most mixing consoles provide a degree of sound signal processing on board,
as well as routing to external processing devices. The very least of these
facilities is some form of equalization (a means of controlling the gain at
An Overview of Typical Mixer Facilities
FA C T F I L E 5 . 4 A U D IO G R O U P S
Audio groups are so called because they create a single
audio output which is the sum of a number of channels.
A single fader controls the level of the summed signal,
and there will be a group output from the console which
is effectively a mix of the audio signals in that group, as
shown in the diagram. The audio signals from each input
to the group are fed via equal-value resistors to the input
of a summing or virtual-earth amplifier.
The stereo mix outputs from an in-line console are
effectively audio groups, one for the left, one for the
right, as they constitute a sum of all the signals routed
to the stereo output and include overall level control. In
the same way, the multitrack routing buses on an in-line
console are also audio groups, as they are sums of all the
channels routed to their respective tracks. More obviously,
some smaller or older consoles will have routing buttons
on each channel module for, say, four audio group destinations, these being really the only way of routing channels to the main outputs.
The master faders for audio groups will often be in
the form of four or eight faders in the central section of
the console. They may be arranged such that one may
Channel outputs
pan a channel between odd and even groups, and it
would be common for two of these groups (an odd and
an even one) to be used as the stereo output in mixdown.
It is also common for perhaps eight audio group faders
to be used as ‘subgroups’, themselves having routing
to the stereo mix, so that channel signals can be made
more easily manageable by routing them to a subgroup
(or panning between two subgroups) and thence to the
main mix via a single level control (the subgroup fader),
as shown in the diagram. (Only four subgroups are shown
in the diagram, without pan controls. Subgroups 1 and 3
feed the left mix bus, and 2 and 4 feed the right mix bus.
Sometimes subgroup outputs can be panned between left
and right main outputs.)
Subgroup
faders
1
2
Channel outputs
Subgroup
fader
Summing
amplifier
Group
output
Mix bus
L
R
Left output
fader
3
Main
O/P
4
Right output
fader
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CHAPTER 5: Mixers
FA C T F I L E 5 . 5 C O N T R O L G R O UPS
Control grouping differs from audio grouping primarily
because it does not give rise to a single summed audio
output for the group: the levels of the faders in the group
are controlled from one fader, but their outputs remain
separate. Such grouping can be imagined as similar in its
effect to a large hand moving many faders at the same
time, each fader maintaining its level in relation to the
others.
The most common way of achieving control grouping is to use VCAs (voltage-controlled amplifiers), whose
gain can be controlled by a DC voltage applied to a control pin. In the VCA fader, audio is not passed through
the fader itself but is routed through a VCA, whose gain
is controlled by a DC voltage derived from the fader position, as shown in the diagram. So the fader now carries
DC instead of audio, and the audio level is controlled indirectly. A more recent alternative to the VCA is the DCA, or
digitally controlled attenuator, whose gain is controlled by
a binary value instead of a DC voltage. This can be easier
to implement in digitally controlled mixers.
Indirect gain control opens up all sorts of new possibilities. The gain of the channel could be controlled externally from a variety of sources, either by combining the
voltage from an external controller in an appropriate way
with the fader’s voltage so that it would still be possible
to set the relative level of the channel, or by breaking the
direct connection between the DC fader and the VCA so
that an automation system could intervene, as discussed
in ‘Automation’, below. It becomes possible to see that
group faders could be DC controls which could be connected to a number of channel VCAs such that their gains
would go up and down together. Further to this, a channel
VCA could be assigned to any of the available groups
simply by selecting the appropriate DC path: this is often
achieved by means of thumbwheel switches on each
fader, as shown in the diagram.
Normally, there are dedicated VCA group master faders in a non-automated system. They usually reside in the
central section of a mixer and will control the overall levels of any channel faders assigned to them by the thumbwheels by the faders. In such a system, the channel
audio outputs would normally be routed to the main mix
directly, the grouping affecting the levels of the individual
channels in this mix.
In an automated system grouping may be achieved
via the automation processor which will allow any fader to
be designated as the group master for a particular group.
This is possible because the automation processor reads
the levels of all the faders, and can use the position of
the designated master to modify the data sent back to the
other faders in the group (see ‘Automation’, below).
DC voltages
from group
faders
Channel
fader
V
Thumbwheel switch
Resistive
network
DC
CV in
VCA
Audio in
Out
various frequencies), and there are few consoles which do not include this.
As well as signal processing, there will be a number of switches that make
changes to the signal path or operational mode of the console. These may
operate on individual channels, or they may function globally (affecting the
whole console at once). The following section is a guide to the facilities
commonly found on multitrack consoles. Figure 5.9 shows the typical location of these sections on an in-line console module.
An Overview of Typical Mixer Facilities
Input section
■
■
■
■
■
Input gain control
Sets the microphone or line input amplifier gain to
match the level of the incoming signal. This control
is often a coarse control in 10 dB steps, sometimes
accompanied by a fine trim. Opinion varies as to
whether this control should be in detented steps or
continuous. Detented steps of 5 or 10 dB make for
easy reset of the control to an exact gain setting,
and precise gain matching of channels.
Phantom power
Many professional mics require 48 volts phantom
powering (see ‘Microphone powering options’,
Chapter 3). There is sometimes a switch on the
module to turn it on or off, although most balanced
mics which do not use phantom power will not be
damaged if it is accidentally left on. Occasionally
this switch is on the rear of the console, by the mic
input socket, or it may be in a central assignable
switch panel. Other methods exist: for example,
one console requires that the mic gain control is
pulled out to turn on the phantom power.
MIC/LINE switch
Switches between the channel’s mic input and line
input. The line input could be the playback output
from a tape machine, or another line level signal
such as a synth or effects device.
PAD
Usually used for attenuating the mic input signal
by something like 20 dB, for situations when the
mic is in a field of high sound pressure. If the mic is
in front of a kick drum, for example, its output may
be so high as to cause the mic input to clip. Also,
capacitor mics tend to produce a higher output level
than dynamic mics, requiring that the pad be used
on some occasions.
Phase reverse or ‘φ’
Sometimes located after the mic input for reversing
the phase of the signal, to compensate for a reversed
Channel
routing
section
Bus trim and
channel pan
controls
Input selection
Pad and phase
Input section
Input gain
Dynamics
section
Auxiliary
sends
Equalizer
Small fader
(rotary)
Monitor pan
Channel
and mix
controls
Monitor routing
Fader swap and PFL
Bounce and SOLO
12
Monitor mute
FIGURE 5.9 Typical layout of controls on an inline mixer module (for description see text).
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directional mic, a mis-wired lead, or to create an effect. This is often
left until later in the signal path.
■
HPF/LPF
Filters can sometimes be switched in at the input stage, which will
usually just be basic high- and low-pass filters which are either in
or out, with no frequency adjustment. These can be used to filter
out unwanted rumble or perhaps hiss from noisy signals. Filtering
rumble at this stage can be an advantage because it saves clipping
later in the chain.
Routing section
■
Track routing switches
The number of routing switches depends on the console: some will
have 24, some 32 and some 48. The switches route the channel path
signal to the multitrack machine, and it is possible to route a signal
to more than one track. The track assignment is often arranged as
pairs of tracks, so that odd and even tracks can be assigned together,
with a pan-pot used to pan between them as a stereo pair, e.g. tracks
3 and 4 could be a stereo pair for background vocals, and each
background vocal mic could be routed to 3 and 4, panned to the
relevant place in the image. In an assignable console these controls
may be removed to a central assignable routing section.
■
It is common for there to be fewer routing switches than there
are tracks, so as to save space, resulting in a number of means of
assigning tracks. Examples are rotary knobs to select the track,
one button per pair of tracks with ‘odd/even/both’ switch, and a
‘shift’ function to select tracks higher than a certain number. The
multitrack routing may be used to route signals to effects devices
during mixdown, when the track outputs are not being used for
recording. In this case one would patch into the track output on the
patchfield (see below) and take the relevant signal to an effects input
somewhere else on the patchfield. In order to route monitor path
signals to the track routing buses it may be necessary to use a switch
which links the output of the monitor fader to the track assignment
matrix.
In theater sound mixers it is common for output routing to be
changed very frequently, and thus routing switches may be located
close to the channel fader, rather than at the top of the module as
in a music mixer. On some recent mixers, track routing is carried
An Overview of Typical Mixer Facilities
out on a matrix which resides in the central section above the main
faders. This removes unnecessary clutter from the channel modules
and reduces the total number of switches required. It may also allow
the storing of routing configurations in memory for later recall.
■
Mix routing switches
Sometimes there is a facility for routing the channel path output
signal to the main monitor mix, or to one of perhaps four output
groups, and these switches will often be located along with the track
routing.
■
Channel pan
Used for panning channel signals between odd and even tracks of the
multitrack, in conjunction with the routing switches.
■
Bus trim
Used for trimming the overall level of the send to multitrack for a
particular bus. It will normally trim the level sent to the track which
corresponds to the number of the module.
■
Odd/Even/Both
Occasionally found when fewer routing buttons are used than there are
tracks. When one routing button is for a pair of tracks, this switch will
determine whether the signal is sent to the odd channel only, the even
channel only, or to both (in which case the pan control is operative).
■
DIRECT
Used for routing the channel output directly to the corresponding
track on the multitrack machine without going via the summing
buses. This can reduce the noise level from the console since the
summing procedure used for combining a number of channel outputs
to a track bus can add noise. If a channel is routed directly to a track,
no other signals can be routed to that track.
Dynamics section
Some advanced consoles incorporate dynamics control on every module,
so that each signal can be treated without resorting to external devices.
The functions available on the best designs rival the best external devices,
incorporating compressor and expander sections which can act as limiters
and gates respectively if required. One system allows the EQ to be placed
in the side-chain of the dynamics unit, providing frequency-sensitive limiting, among other things, and it is usually possible to link the action of one
channel’s dynamics to the next in order to ‘gang’ stereo channels so that
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the image does not shift when one channel has a sudden change in level
while the other does not.
When dynamics are used on stereo signals it is important that left
and right channels have the same settings, otherwise the image may be
affected. If dynamics control is not available on every module, it is sometimes offered on the central section with inputs and outputs on the patchbay. Dynamics control will not be covered further here, but is discussed in
more detail in ‘The compressor/limiter ’, below.
Equalizer section
The EQ section is usually split into three or four sections, each operating
on a different frequency band. As each band tends to have similar functions
these will be described in general. The principles of EQ are described in
greater detail in ‘EQ explained’, below.
■
HF, MID 1, MID 2, LF
A high-frequency band, two mid-frequency bands, and a lowfrequency band are often provided. If the EQ is parametric these
bands will allow continuous variation of frequency (over a certain
range), ‘Q ’, and boost/cut. If it is not parametric, then there may be
a few switched frequencies for the mid band, and perhaps a fixed
frequency for the LF and HF bands.
■
Peaking/shelving or BELL
Often provided on the upper and lower bands for determining
whether the filter will provide boost/cut over a fixed band (whose
width will be determined by the Q), or whether it will act as a shelf,
with the response rising or rolling off above or below a certain
frequency (see ‘EQ explained’, below).
■
Q
The Q of a filter is defined as its center frequency divided by its
bandwidth (the distance between frequencies where the output of
the filter is 3 dB lower than the peak output). In practice this affects
the ‘sharpness’ of the filter peak or notch, high Q giving the sharpest
response, and low Q giving a very broad response. Low Q would be
used when boost or cut over a relatively wide range of frequencies is
required, while high Q is used to boost or cut one specific region (see
Fact File 5.6).
■
Frequency control
Sets the center frequency of a peaking filter, or the turnover frequency
of a shelf.
An Overview of Typical Mixer Facilities
FA C T F I L E 5 . 6 VA R IA B L E Q
Some EQ sections provide an additional control whereby
the Q of the filter can be adjusted. This type of EQ section is termed a parametric EQ since all parameters,
cut/boost, frequency, and Q can be adjusted. The diagram below illustrates the effect of varying the Q of an EQ
section. High Q settings affect very narrow bands of frequencies, low Q settings affect wider bands. The low Q
settings sound ‘warmer’ because they have gentle slopes
and therefore have a more gradual and natural effect on
the sound. High Q slopes are good for a rather more overt
emphasis of a particular narrow band, which of course
can be just as useful in the appropriate situation. Some
EQ sections are labeled parametric even though the Q is
not variable. This is a misuse of the term, and it is wise
to check whether or not an EQ section is truly parametric
even though it may be labeled as such.
■
Boost/cut
Determines the amount of boost or cut applied to the selected band,
usually up to a maximum of around ⫾15 dB.
■
HPF/LPF
Sometimes the high- and low-pass filters are located here instead
of at the input, or perhaps in addition. They normally have a fixed
frequency turnover point and a fixed roll-off of either 12 or 18 dB per
octave. Often these will operate even if the EQ is switched out.
■
CHANNEL
The American convention is for the main equalizer to reside
normally in the monitor path, but it can be switched so that it is
in the channel path. Normally the whole EQ block is switched at
once, but on some recent models a section of the EQ can be switched
separately. This would be used to equalize the signal which is being
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CHAPTER 5: Mixers
recorded on multitrack tape. If the EQ is in the monitor path then
it will only affect the replayed signal. The traditional European
convention is for EQ to reside normally in the channel path, so as to
allow recording with EQ.
■
IN/OUT
Switches the EQ in or out of circuit. Equalization circuits can
introduce noise and phase distortion, so they are best switched out
when not required.
Channel and mix controls
■
Pan
This control is a continuous rotary knob, and is used to place the
signal of that channel in any desired position in the stereo picture.
See Fact File 5.2.
■
Fader reverse
Swaps the faders between mix and channel paths, so that the large
fader can be made to control either the mix level or the channel level.
Some systems defeat any fader automation when the large fader is
put in the channel path. Fader reverse can often be switched globally,
and may occur when the console mode is changed from recording to
mixdown.
■
Line/Tape or Bus/Tape
Switches the source of the input to the monitor path between the line
output of the same-numbered channel and the return from multitrack
tape. Again it is possible that this may be switched globally. In ‘line’
or ‘bus’ mode the monitor paths are effectively ‘listening to’ the line
output of the console’s track assignment buses, while in ‘tape’ mode
the monitor paths are listening to the off-tape signal (unless the
tape machine’s monitoring is switched to monitor the line input of
the tape machine, in which case ‘line’ and ‘tape’ will effectively be
the same thing!). If a problem is suspected with the tape machine,
switching to monitor ‘line’ will bypass the tape machine entirely and
allow the operator to check if the console is actually sending anything.
■
Broadcast, or ‘mic to mix’, or ‘simulcast’
Used for routing the mic signal to both the channel and monitor
paths simultaneously, so that a multitrack recording can be
made while a stereo mix is being recorded or broadcasted. The
configuration means that any alterations made to the channel path
An Overview of Typical Mixer Facilities
will not affect the stereo mix, which is important when the mix
output is live (see Figure 5.10).
■
BUS or ‘monitor-to-bus’
Routes the output of the monitor fader to the input of the channel
path (or the channel fader) so that the channel path can be used as
a post-fader effects send to any one of the multitrack buses (used
in this case as aux sends), as shown in Figure 5.11. If a BUS TRIM
FIGURE 5.10
A ‘broadcast mode’ switch
in an in-line console allows
the microphone input to be
routed to both signal paths,
such that a live stereo mix
may be made independent
of any changes to
multitrack recording levels.
1
Small fader
P
A
N
Bounce path
Multitrack
routing
24
BUS or BOUNCE switch
DUMP switch
L
Large fader
P
A
N
Monitor mix
R
FIGURE 5.11
Signal routings for ‘bounce’,
‘bus’ and ‘dump’ modes
(see text).
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control is provided on each multitrack output this can be used as the
master effects-send level control.
■
DUMP
Incorporated (rarely) on some consoles to route the stereo panned
mix output of a track (i.e. after the monitor path pan-pot) to the
multitrack assignment switches. In this way, the mixed version
of a group of tracks can be ‘bounced down’ to two tracks on the
multitrack, panned and level-set as in the monitor mix (see
Figure 5.11).
■
BOUNCE
A facility for routing the output of the monitor fader to the
multitrack assignment matrix, before the pan control, in order
that tracks can be ‘bounced down’ so as to free tracks for more
recording by mixing a group of tracks on to a lower number of tracks.
BOUNCE is like a mono version of DUMP (see Figure 5.11).
■
MUTE or CUT
Cuts the selected track from the mix. There may be two of these
switches, one for cutting the channel signal from the multitrack
send, the other for cutting the mix signal from the mix.
■
PFL
See Fact File 5.3.
■
AFL
After fade listen is similar to PFL, except that it is taken from after
the fader. This is sometimes referred to as SOLO, which routes a
panned version of the track to the main monitors, cutting everything
else. These functions are useful for isolating signals when setting up
and spotting faults. On many consoles the AFL bus will be stereo.
Solo functions are useful when applying effects and EQ, in order that
one may hear the isolated sound and treat it individually without
hearing the rest of the mix. Often a light is provided to show that a
solo mode is selected, because there are times when nothing can be
heard from the loudspeakers due to a solo button being down with
no signal on that track. A solo safe control may be provided centrally,
which prevents this feature from being activated.
■
In-place solo
On some consoles, solo functions as an ‘in-place’ solo, which means
that it actually changes the mix output, muting all tracks which
are not solo’ed and picking out all the solo’ed tracks. This may be
An Overview of Typical Mixer Facilities
preferable to AFL as it reproduces the exact contribution of each
channel to the mix, at the presently set master mix level. Automation
systems often allow the solo functions to be automated in groups, so
that a whole section can be isolated in the mix. In certain designs,
the function of the automated mute button on the monitor fader may
be reversed so that it becomes solo.
Auxiliary sends
The number of aux(iliary) sends depends on the console, but there can be
up to ten on an ordinary console, and sometimes more on assignable models. Aux sends are ‘take-off points’ for signals from either the channel or
mix paths, and they appear as outputs from the console which can be used
for foldback to musicians, effects sends, cues, and so on. Each module will
be able to send to auxiliaries, and each numbered auxiliary output is made
up of all the signals routed to that aux send. So they are really additional
mix buses. Each aux will have a master gain control, usually in the center
of the console for adjusting the overall gain of the signal sent from the console, and may have basic EQ. Aux sends are often a combination of mono
and stereo buses. Mono sends are usually used as routes to effects, while
stereo sends may have one level control and a pan control per channel for
mixing a foldback source.
■
Aux sends 1–n
Controls for the level of each individual channel in the numbered aux
mix.
■
Pre/post
Determines whether the send is taken off before or after the fader.
If it is before then the send will still be live even when the fader is
down. Generally, ‘cue’ feeds will be pre-fade, so that a mix can be
sent to foldback which is independent of the monitor mix. Effects
sends will normally be taken post-fade, in order that the effect follows
a track’s mix level.
■
Mix/channel
Determines whether the send is taken from the mix or channel
paths. It will often be sensible to take the send from the channel path
when effects are to be recorded on to multitrack rather than on to the
mix. This function has been labeled ‘WET ’ on some designs.
■
MUTE
Cuts the numbered send from the aux mix.
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Master control section
The master control section usually resides in the middle of the console,
or near the right-hand end. It will contain some or all of the following
facilities:
■
Monitor selection
A set of switches for selecting the source to be monitored. These will
include recording machines, aux sends, the main stereo mix, and
perhaps some miscellaneous external sources like CD players, cassette
machines, etc. They only select the signal going to the loudspeakers,
not the mix outputs. This may be duplicated to some extent for a set of
additional studio loudspeakers, which will have a separate gain control.
■
DIM
Reduces the level sent to the monitor loudspeakers by a considerable
amount (usually around 40 dB), for quick silencing of the room.
■
MONO
Sums the left and right outputs to the monitors into mono so that
mono compatibility can be checked.
■
Monitor phase reverse
Phase reverses one channel of the monitoring so that a quick check
on suspected phase reversals can be made.
■
TAPE/LINE
Usually a global facility for switching the inputs to the mix path
between the tape returns and the console track outputs. Can be
reversed individually on modules.
■
FADER REVERSE
Global swapping of small and large faders between mix and channel
paths.
■
Record/Overdub/Mixdown
Usually globally configures mic/line input switching, large and small
faders and auxiliary sends depending on mode of operation. (Can be
overridden on individual channels.)
■
Auxiliary level controls
Master controls for setting the overall level of each aux send output.
■
Foldback and Talkback
There is often a facility for selecting which signals are routed to
the stereo foldback which the musicians hear on their headphones.
An Overview of Typical Mixer Facilities
Sometimes this is as comprehensive as a cue mixer which allows
mixing of aux sends in various amounts to various stereo cues, while
often it is more a matter of selecting whether foldback consists of the
stereo mix, or one of the aux sends. Foldback level is controllable,
and it is sometimes possible to route left and right foldback signals
from different sources. Talkback is usually achieved using a small
microphone built into the console, which can be routed to a
number of destinations. These destinations will often be aux sends,
multitrack buses, mix bus, studio loudspeakers and foldback.
■
Oscillator
Built-in sine-wave oscillators vary in quality and sophistication, some
providing only one or two fixed frequencies, while others allow the
generation of a whole range. If the built-in oscillator is good it can
be used for lining up the tape machine, as it normally can be routed
to the mix bus or the multitrack outputs. The absolute minimum
requirement is for accurate 1 kHz and 10 kHz tones, the 10 kHz being
particularly important for setting the bias of an analog tape machine.
The oscillator will have an output level control.
■
Slate
Provides a feed from the console talkback mic to the stereo output,
often superimposing a low-frequency tone (around 50 Hz) so that the
slate points can be heard when winding a tape at high speed. Slate
would be used for recording take information on to tape.
■
Master faders
There may be either one stereo fader or left and right faders to control
the overall mix output level. Often the group master faders will reside
in this section.
Effects returns
Effects returns are used as extra inputs to the mixer, supplied specifically for inputs from external devices such as reverberation units. These
are often located in the central section of the console and may be laid out
like reduced-facility input channels. Returns sometimes have EQ, perhaps
more basic than on channels, and they may have aux sends. Normally they
will feed the mix, although sometimes facilities are provided to feed one
or more returns to the multitrack via assignment switches. A small fader
or rotary level control is provided, as well as a pan-pot for a mono return.
Occasionally, automated faders may be assigned to the return channels so
as to allow automated control of their levels in the mix.
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HF
Patchbay or jackfield
0
–4
+4
–8
+8
–12
dB
Hi-MID
+12
0
–4
+4
–8
+8
–12
dB
5
+12
7
3
9
11
1
kHz
Lo-MID
0
–4
+4
–8
+8
–12
dB
700
+12
1k
400
1k5
2k5
200
Hz
LF
0
–4
+4
–8
+8
–12
dB
+12
Filters
10k
50
8k
5k
EQ EXPLAINED
12k
65
15k
80
35
20
Hz
HF
Most large consoles employ a built-in jackfield or patchbay for routing signals in ways which the console switching does not allow, and for sending
signals to and from external devices. Just about every input and output
on every module in the console comes up on the patchbay, allowing signals to be cross-connected in virtually any configuration. The jackfield is
usually arranged in horizontal rows, each row having an equal number of
jacks. Vertically, it tries to follow the signal path of the console as closely
as possible, so the mic inputs are at the top and the multitrack outputs are
nearer the bottom. In between these there are often insert points which
allow the engineer to ‘break into’ the signal path, often before or after the
EQ, to insert an effects device, compressor, or other external signal processor. Insert points usually consist of two rows, one which physically breaks
the signal chain when a jack is inserted, and one which does not. Normally
it is the lower row which breaks the chain, and should be used as inputs.
The upper row is used as an output or send. Normaling is usually applied
at insert points, which means that unless a jack is inserted the signal will
flow directly from the upper row to the lower.
At the bottom of the patchfield will be all the master inputs and outputs, playback returns, perhaps some parallel jacks, and sometimes some
spare rows for connection of one’s own devices. Some consoles bring the
microphone signals up to the patchbay, but there are some manufacturers who would rather not do this unless absolutely necessary as it is more
likely to introduce noise, and phantom power may be present on the jackfield. Jackfields are covered in further detail in ‘Jackfields (patchbays)’,
Chapter 12.
100
18k
LF
EQ
FIGURE 5.12
Typical layout of an EQ
section.
The tone control or EQ (⫽equalization) section provides mid-frequency
controls in addition to bass and treble. A typical comprehensive EQ section
may have first an HF (high-frequency) control similar to a treble control but
operating only at the highest frequencies. Next would come a hi-mid control, affecting frequencies from around 1 kHz to 10 kHz, the center frequency
being adjusted by a separate control. Lo-mid controls would come next,
similar to the hi-mid but operating over a range of say 200 Hz to 2 kHz.
Then would come an LF (low-frequency) control. Additionally, high- and
low-frequency filters can be provided. The complete EQ section looks something like that shown in Figure 5.12. An EQ section takes up quite a bit
of space, and so it is quite common for dual concentric or even assignable
EQ Explained
controls (see below) to be used. For instance, the cut/boost controls of the
hi- and lo-mid sections can be surrounded by annular skirts which select
the frequency.
Principal EQ bands
The HF section affects the highest frequencies and provides up to 12 dB of
boost or cut. This type of curve is called a shelving curve because it gently
boosts or cuts the frequency range towards a shelf where the level remains
relatively constant (see Figure 5.13(a)). Next comes the hi-mid section. Two
dB gain
LF shelf curve
HF shelf curve
12
9
6
3
0
–3
–6
–9
–12
100
1k
FIGURE 5.13 (a)
Typical HF and LF shelf EQ
characteristics shown at
maximum settings.
10 k
Frequency Hz
FIGURE 5.13 (b)
Typical MF peaking filter
characteristic.
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CHAPTER 5: Mixers
FIGURE 5.13 (c)
MF peaking filter
characteristics at 1, 5 and
10 kHz.
FIGURE 5.13 (d)
High-pass filters
with various turnover
frequencies.
dB gain
136
12
9
6
3
0
–3
–6
–9
–12
100
1 kHz
10 kHz
Frequency Hz
controls are provided here, one to give cut or boost, the other to select the
desired center frequency. The latter is commonly referred to as a ‘swept
mid’ because one can sweep the setting across the frequency range.
Figure 5.13(b) shows the result produced when the frequency setting is
at the 1 kHz position, termed the center frequency. Maximum boost and
cut affects this frequency the most, and the slopes of the curve are considerably steeper than those of the previous shelving curves. This is often
referred to as a ‘bell’ curve due to the upper portion’s resemblance to the
shape of a bell. It has a fairly high ‘Q ’, that is its sides are steep. Q is
defined as:
Q ⫽ center frequency ⫼ bandwidth
Stereo Line Input Modules
where the bandwidth is the spacing in hertz between the two points at
which the response of the filter is 3 dB lower than that at the center frequency. In the example shown the center frequency is 1 kHz and the bandwidth is 400 Hz, giving Q ⫽ 2.5.
MF EQ controls are often used to hunt for trouble-spots; if a particular instrument (or microphone) has an emphasis in its spectrum somewhere which does not sound very nice, some mid cut can be introduced,
and the frequency control can be used to search for the precise area in the
frequency spectrum where the trouble lies. Similarly, a dull sound can be
given a lift in an appropriate part of the spectrum which will bring it to life
in the overall mix. Figure 5.13(c) shows the maximum cut and boost curves
obtained with the frequency selector at either of the three settings of 1, 5
and 10 kHz. The high Q of the filters enables relatively narrow bands to be
affected. Q may be varied in some cases, as described in Fact File 5.6.
The lo-mid section is the same as the hi-mid section except that it covers a lower band of frequencies. Note though that the highest frequency
setting overlaps the lowest setting of the hi-mid section. This is quite
common, and ensures that no ‘gaps’ in the frequency spectrum are left
uncovered.
Filters
High- and low-cut filters provide fixed attenuation slopes at various frequencies. Figure 5.13(d) shows the responses at LF settings of 80, 65, 50,
35 and 20 Hz. The slopes are somewhat steeper than is the case with the
HF and LF shelving curves, and slope rates of 18 or 24 dB per octave are
typical. This enables just the lowest, or highest, frequencies to be rapidly
attenuated with minimal effect on the mid band. Very low traffic rumble
could be removed by selecting the 20 or 35 Hz setting. More serious lowfrequency noise may require the use of one of the higher turnover frequencies. High-frequency hiss from, say, a noisy guitar amplifier or air escaping
from a pipe organ bellows can be dealt with by selecting the turnover frequency of the HF section which attenuates just sufficient HF noise without
unduly curtailing the HF content of the wanted sound.
STEREO LINE INPUT MODULES
In broadcast situations it is common to require a number of inputs to be
dedicated to stereo line level sources, such as CD players, electronic musical instruments, etc. Such modules are sometimes offered as an option for
multitrack consoles, acting as replacements for conventional I/O modules
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and allowing two signals to be faded up and down together with one fader.
Often the EQ on such modules is more limited, but the module may provide
for the selection of more than one stereo source, and routing to the main
mix as well as the multitrack. It is common to require that stereo modules
always reside in special slots on the console, as they may require special wiring. Such modules also used to provide facilities for handling LP turntable
outputs, offering RIAA equalization (see ‘RIAA equalization’, Appendix 1).
Stereo microphone inputs can also be provided, with the option for MS
(middle and side) format signals as well as AB (conventional left and right)
format (see ‘Stereo microphones’, Chapter 3). A means of control over
stereo width can be offered on such modules.
DEDICATED MONITOR MIXER
A dedicated monitor mixer is often used in live sound reinforcement work
to provide a separate monitor mix for each musician, in order that each
artist may specify his or her precise monitoring requirements. A comprehensive design will have, say, 24 inputs containing similar facilities to a
conventional mixer, except that below the EQ section there will be a row of
rotary or short-throw faders which individually send the signal from that
channel to the group outputs, in any combination of relative levels. Each
group output will then provide a separate monitor mix to be fed to headphones or amplifier racks.
TECHNICAL SPECIFICATIONS
This section contains some guidance concerning the meanings and commonly encountered values of technical specifications for mixers.
Input noise
The output from a microphone is in the millivolt range, and so needs considerable amplification to bring it up to line level. Amplification of the signal also brings with it amplification of the microphone’s own noise output
(discussed in ‘Microphone noise in practice’, Chapter 3), which one can do
nothing about, as well as amplification of the mixer’s own input noise. The
latter must therefore be as low as possible so as not to compromise the
noise performance unduly. A 200 ohm source resistance on its own generates 0.26 μV of noise (20 kHz bandwidth). Referred to the standard line
level of 775 mV (0 dBu) this is ⫺129.6 dBu. A microphone amplifier will
Technical Specifications
add its own noise to this, and so manufacturers quote an ‘equivalent input
noise’ (EIN) value which should be measured with a 200 ohm source resistance across the input.
An amplifier with a noise contribution equal to that of the 200 ohm
resistor will degrade the theoretically ‘perfect’ noise level by 3 dB, and so
the quoted equivalent input noise will be ⫺129.6 ⫹ 3 ⫽ ⫺126.6 dBm.
(Because noise contributions from various sources sum according to their
power content, not their voltage levels, dBm is traditionally used to express
input noise level.) This value is quite respectable, and good-quality mixers
should not be noisier than this. Values of around ⫺128 dBm are sometimes
encountered, which are excellent, indicating that the input resistance is
generating more noise than the amplifier. Make sure that the EIN is quoted
with a 200 ohm source, and a bandwidth up to 20 kHz, unweighted. A 150
ohm source, sometimes specified, will give an apparently better EIN simply
because this resistor is itself quieter than a 200 ohm one, resistor noise
being proportional to ohmic value. Also, weighting gives a flattering result,
so one always has to check the measuring conditions. Make sure that
EIN is quoted in dBm or dBu. Some manufacturers quote EIN in dBV (i.e.
ref. 1 volt) which gives a result that looks 2.2 dB better. An input should
have high common mode rejection as well as low noise, as discussed in
Fact File 5.7.
Output noise
The output residual noise of a mixer, with all faders at minimum, should
be at most ⫺90 dBu. There is no point in having a very quiet microphone
FA C T F I L E 5 . 7 C O MMO N MO D E REJ ECTI ON
As discussed in ‘Balanced lines’, Chapter 12, common
mode rejection is the ability of a balanced input to reject
interference which can be induced into the signal lines. A
microphone input should have a CMRR (common mode
rejection ratio) of 70 dB or more; i.e. it should attenuate
the interference by 70 dB. But look at how this measurement is made. It is relatively easy to achieve 70 dB at, say,
500 Hz, but rejection is needed most at high frequencies
– between 5 and 20 kHz – and so a quoted CMRR of
‘70 dB at 15 kHz’ or ‘70 dB between 100 Hz and 10 kHz’
should be sought. Line level CMRR can be allowed to be
rather lower since the signal voltage level is a lot higher
than in microphone cabling. CMRRs of as low as 30 dB at
10 kHz are deemed to be adequate.
Common mode rejection is a property of a balanced
input, and so it is not applicable to a balanced output.
However, output balance is sometimes quoted which
gives an indication of how closely the two legs of a balanced output are matched. If the two legs were to be
combined in antiphase total cancelation would ideally be
achieved. In practice, around 70 dB of attenuation should
be looked for.
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amplifier if a noisy output stage ruins it. With all channels routed to the
output, and all faders at the ‘zero’ position, output noise (or ‘mixing’ noise)
should be at least ⫺80 dBu with the channel inputs switched to ‘line’ and
set for unity gain. Switching these to ‘mic’ inevitably increases noise levels because this increases the gain of the input amplifier. It underlines the
reason why all unused channels should be unrouted from the mix buses,
or their faders brought down to a minimum. Digital mixers with ‘scene’
memories tend to be programmed by copying a particular scene to another
vacant scene, then modifying it for the new requirements. When doing this,
one needs to ensure that all unwanted inputs and routing from the copied
scene are removed so as to maintain the cleanest possible signal. Make sure
that the aux outputs have a similarly good output noise level.
Impedance
A microphone input should have a minimum impedance of 1 kΩ. A lower
value than this degrades the performance of many microphones. A line
level input should have a minimum impedance of 10 kΩ. Whether it is balanced or unbalanced should be clearly stated, and consideration of the type
of line level equipment that the mixer will be partnered with will determine the importance of balanced line inputs. All outputs should have a low
impedance, below 200 ohms, balanced. Check that the aux outputs are also
of very low impedance, as sometimes they are not. If insert points are provided on the input channels and/or outputs, these also should have very
low output and high input impedances.
Frequency response
A frequency response that is within 0.2 dB between 20 Hz and 20 kHz for all
combinations of input and output is desirable. The performance of audio
transformers varies slightly with different source and load impedances,
and a specification should state the range of loads between which a ‘flat’
frequency response will be obtained. Above 20 kHz, and probably below
15 Hz or so, the frequency response should fall away so that unwanted outof-band frequencies are not amplified, for example radio-frequency breakthrough or subsonic interference.
Distortion
With an analog mixer, distortion should be quoted at maximum gain
through the mixer and a healthy output level of, say, 10 dBu or more. This
Technical Specifications
will produce a typical worst case, and should normally be less than 0.1%
THD (total harmonic distortion). The distortion of the low-gain line level
inputs to outputs can be expected to be lower: around 0.01%. The outputs
should be loaded with a fairly low impedance which will require more current from the output stages than a high impedance will, this helping to
reveal any shortcomings. A typical value is 600 ohms.
Clipping and overload margins are discussed in Fact File 5.8.
Crosstalk
In analog mixers, a signal flowing along one path may induce a small signal
in another, and this is termed ‘crosstalk’. Crosstalk from adjacent channels should be well below the level of the legitimate output signal, and a
figure of ⫺80 dB or more should be looked for at 1 kHz. Crosstalk performance tends to deteriorate at high frequencies due to capacitive coupling in
wiring harnesses, for instance, but a crosstalk of at least ⫺60 dB at 15 kHz
should still be sought. Similarly, very low-frequency crosstalk often deteriorates due to the power supply source impedance rising here, and a figure of
⫺50 dB at 20 Hz is reasonable.
Ensure that crosstalk between all combinations of input and output is
of a similarly good level. Sometimes crosstalk between channel auxiliaries
is rather poorer than that between the main outputs.
FA C T F I L E 5 . 8 C LIP P IN G
A good mixer will be designed to provide a maximum
electrical output level of at least 20 dBu. Many will provide 24 dBu. Above this electrical level clipping will occur,
where the top and bottom of the audio waveform are
chopped off, producing sudden and excessive distortion (see diagram). Since the nominal reference level of
0 dBu usually corresponds to a meter indication of PPM
4 or ⫺4 VU, it is very difficult to clip the output stages of
a mixer. The maximum meter indication on a PPM would
correspond in this case to an electrical output of around
12 dBu, and thus one would have to be severely bending
the meter needles to cause clipping.
Clipping, though, may occur at other points in the
signal chain, especially when large amounts of EQ boost
have been added. If, say, 12 dB of boost has been applied
on a channel, and the fader is set well above the 0 dB
mark, clipping on the mix bus may occur, depending on
overload margins here. Large amounts of EQ boost should
not normally be used without a corresponding overall gain
reduction of the channel for this reason.
An input pad or attenuator is often provided to prevent the clipping of mic inputs in the presence of highlevel signals (see ‘Input section’, above).
Max. +ve output voltage
Max. –ve output voltage
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METERING SYSTEMS
Metering systems are provided on audio mixers to indicate the levels of
audio signals entering and leaving the mixer. Careful use of metering is
vital for optimizing noise and distortion, and to the recording of the correct
audio level on tape. In this section the merits of different metering systems
are examined.
Mechanical metering
Two primary types of mechanical meters have been used: the VU (volume
unit) meter (Figure 5.14) and the PPM (peak program meter), as shown in
Figure 5.15. These are very different to each other, the only real similarity
being that they both have swinging needles. The British, or BBC-type, PPM
is distinctive in styling in that it is black with numbers ranging from 1 to
7 equally spaced across its scale, there being a 4 dB level difference between
each gradation, except between 1 and 2 where there is usually a 6 dB change in level. The EBU PPM (also shown in
Figure 5.15) has a scale calibrated in decibels. The VU, on
–3 –2 –1 0
–5
1
–7
the other hand, is usually white or cream, with a scale run2
–10
ning from ⫺20 dB up to 3 dB, ranged around a zero point
3
–20
which is usually the studio’s electrical reference level.
Originally the VU meter was associated with a variable
VU
attenuator which could vary the electrical alignment level
for 0 VU up to ⫹24 dBu, although it is common for this to
be fixed these days at 0 VU ⫽ ⫹4 dBu.
It is important to know how meter readings relate to the
FIGURE 5.14 Typical VU meter scale.
line-up standard in use in a particular environment, and
FIGURE 5.15
(Left) BBC-type peak
programme meter (PPM).
(Right) European-type
PPM.
Metering Systems
to understand that these standards may vary between establishments and
areas of work. Fact File 5.9 discusses the relationship between meter indication and signal levels, considering practical issues such as the onset of
distortion.
Problems with mechanical meters
PPMs respond well to signal peaks, that is they have a fast rise-time,
whereas VUs are quite the opposite: they have a very slow rise-time. This
means that VUs do not give a true representation of the peak level going
on to tape, especially in cases when a signal with a high transient content,
such as a harpsichord, is being recorded, often showing as much as 10–
15 dB lower than a peak-reading meter. This can result in overmodulation
of the recording, especially with digital recorders where the system is very
sensitive to peak overload. None the less, many people are used to working
with VUs, and have learned to interpret them. They are good for measuring
continuous signals such as tones, but their value for monitoring program
material is dubious in the age of digital recording.
VUs have no control over the fall-time of the needle, which is much the
same as the rise-time, whereas PPMs are engineered to have a fast rise-time
FA C T F I L E 5 . 9 ME T E R IN G , S IG NAL LEVELS AND DI STORTI ON
Within a studio there is usually a ‘reference level’ and a
‘peak recording level’. In the broadcast domain these
are usually referred to as ‘alignment level’ and ‘permitted maximum level (PML)’ as shown in Figure 5.17. The
reference or alignment level usually relates to the level at
which a 1 kHz line-up tone should play back on the console’s meters. In analog mixers this may correspond to
PPM 4 on a BBC-type PPM or ‘Test’ on a European PPM.
Electrically PPM 4 usually corresponds to a level of 0 dBu,
although the German and Nordic metering standards traditionally had it as ⫺3 dBu. In the digital domain, line-up
level usually corresponds to either ⫺20 dBFS (SMPTE) or
⫺18 dBFS (EBU), depending on the area of the world and
standard concerned. A relationship is therefore established
between meter reading and signal level in each domain.
In digital audio systems, where compatibility with
other systems is not an issue, it is possible to peak close
to 0 dBFS without incurring increases in distortion, and
many recording engineers use all this ‘headroom’ in order
to maximize dynamic range. Digital systems clip hard at
0 dBFS whereas analog tape tends to give rise to gradually
increasing distortion and level compression as levels rise.
Because of the typical relationship between analog and
digital levels, peak digital level can correspond to electrical levels as high as ⫹18 to ⫹22 dBu, which can be
inconvenient in mixed format systems sharing the same
meters. In broadcasting it is normal to peak no more than
8⫺9 dB above line-up level (PPM 6 in the UK) as higher
levels than this can have serious effects on analog transmitter distortion. Limiters are normally used in broadcasting systems, which start to take effect rapidly above this
level. Because of the standardized relationship between
analog and digital levels in broadcasting, the PML is lower
than digital peak level, leading to a degree of undermodulation of the digital system that is considered acceptable
in the interests of compatibility.
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and a longer fall-time, which tends to be more subjectively useful. The
PPM was designed to indicate peaks that would cause audible distortion, but does not measure the absolute peak level of a signal. Mechanical
meters take up a lot of space on a console, and it can be impossible to find
space for one meter per channel in the case of a multitrack console. In this
case there are often only meters on the main outputs, and perhaps some
auxiliary outputs, these being complemented on more expensive consoles
by electronic bargraph meters, usually consisting of LED or liquid crystal
displays, or some form of ‘plasma’ display.
Electronic bargraph metering
+12
+6
+3
0
–3
0
–5
–3
–15
–6
–35
–10
–50
Peak
–20
VU
FIGURE 5.16
Typical peak-reading
bargraph meter with
optional VU scale.
Unlike mechanical meters, electronic bargraphs have no mechanical inertia to overcome, so they can effectively have an infinitely fast rise-time
although this may not be the ideal in practice. Cheaper bargraphs are made
out of a row of LEDs (light emitting diodes), and the resolution accuracy
depends on the number of LEDs used. This type of display is sometimes
adequate, but unless there are a lot of gradations it is difficult to use them
for line-up purposes. Plasma and liquid crystal displays look almost continuous from top to bottom, and do not tend to have the glare of LEDs, being
more comfortable to work with for any length of time. Such displays often
cover a dynamic range far greater than any mechanical meter, perhaps from
⫺50 dB up to ⫹12 dB, and so can be very useful in showing the presence of
signals which would not show up on a mechanical PPM. Such a meter is
illustrated in Figure 5.16.
There may be a facility provided to switch the peak response of these
meters from PEAK to VU mode, where they will imitate the scale and
ballistic response of a VU meter. On more up-market designs it may be
possible to use the multitrack bargraphs as a spectrum analyzer display,
indicating perhaps a one-third octave frequency-band analysis of the signal fed to it. Occasionally, bargraph displays incorporate a peak-hold facility. A major advantage of these vertical bargraphs is that they take up very
little horizontal space on a meter bridge and can thus be used for providing one meter for every channel of the console: useful for monitoring the
record levels on a multitrack tape machine. In this case, the feed to the
meter is usually taken off at the input to the monitor path of an in-line
module.
Miscellaneous meters may also be provided on the aux send outputs for
giving some indication of the level being sent to auxiliary devices such as
effects. These are commonly smaller than the main meters, or may consist
of LED bargraphs with lower resolution. A phase meter or correlation meter
Metering Systems
is another option, this usually being connected between the left and right
main monitor outputs to indicate the degree of phase correlation between
these signals. This can be either mechanical or electronic. In broadcast
environments, sum and difference (or M and S) meters may be provided
to show the level of the mono-compatible and stereo difference signals in
stereo broadcasting. These often reside alongside a stereo meter for left and
right output levels.
Relationship between different metering standards
Figure 5.17 shows a number of common meter scales and the relationship
between them. The relationship between meter indication and electrical
level varies depending on the type of meter and the part of the world concerned. As introduced in Fact File 5.9, there is a further relationship to be
concerned with, this being that between the electrical output level of the
mixer and the recording or transmitted level, and between the analog and
digital domains.
0
–2
–4
+5
7
–6
+12
+12
Broadcast
permitted
maximum level
(PML)
–8
6
0
+8
+3
+6
5
+4
4
TEST
0
–5
–3
TEST
–10
–20
–7
–6
Alignment
level
–18
–5
–4
3
–12
–10
–8
2
–20
–12
–30
–12
1
–20
BBC
PPM
EBU
PPM
IRT/
DIN
PPM
Nordic
PPM
VU
EBU
digital
PPM
FIGURE 5.17
Graphical comparison of
commonly encountered
meter scalings and
electrical levels in dBu.
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Meter take-off point
Output level meter-driving circuits should normally be connected directly
across the outputs so that they register the real output levels of the mixer.
This may seem self-evident but there are certain models in which this is
not the case, the meter circuit taking its drive from a place in the circuit
just before the output amplifiers. In such configurations, if a faulty lead or
piece of equipment connected to the mixer places, say, a short-circuit across
the output the meter will nevertheless read normal levels, and lack of signal reaching a destination will be attributed to other causes. The schematic
circuit diagrams of the mixer can be consulted to ascertain whether such
an arrangement has been employed. If it is not clear, a steady test tone can
be sent to the mixer’s output, giving a high meter reading. Then a shortcircuit can be deliberately applied across the output (the output amplifier
will not normally be harmed by several seconds of short-circuit) and the
meter watched. If the indicated level drastically reduces then the meter is
correctly registering the real output. If it stays high then the meter is taking
its feed from elsewhere.
AUTOMATION
Background
The original and still most common form of mixer automation is a means
of storing fader positions dynamically against time for reiteration at a later
point in time, synchronous with recorded material. The aim of automation has been to assist an engineer in mixdown when the number of faders
that need to be handled at once becomes too great for one person. Fader
automation has resulted in engineers being able to concentrate on subareas of a mix at each pass, gradually building up the finished product and
refining it.
MCI first introduced VCA (voltage controlled amplifier) automation for
their JH500 series of mixing consoles in the mid-1970s, and this was soon
followed by imitations with various changes from other manufacturers.
Moving fader automation systems, such as Neve’s NECAM, were introduced slightly later and tended to be more expensive than VCA systems.
During the mid-1980s, largely because of the falling cost of microprocessor hardware, console automation enjoyed further advances resulting in
developments such as snapshot storage, total dynamic automation, retrofit automation packages, and MIDI-based automation. The rise of digital
mixers and digitally controlled analog mixers with integral automation has
Automation
continued the trend towards total automation of most mixer controls as a
standard feature of many new products.
In the following sections a number of different approaches to console
automation will be presented and discussed.
Fader automation
There are two common means of memorizing and controlling the gain of
a channel: one which stores the positions of the fader and uses this data to
control the gain of a VCA or digitally controlled attenuator (DCA), the other
which also stores fader movements but uses this information actually to
drive the fader’s position using a motor. The former is cheaper to implement
than the latter, but is not so ergonomically satisfactory because the fader’s
physical position may not always correspond to the gain of the channel.
It is possible to combine elements of the two approaches in order that
gain control can be performed by a VCA but with the fader being moved
mechanically to display the gain. This allows for rapid changes in level
which might be impossible using physical fader movements, and also
allows for dynamic gain offsets of a stored mix whilst retaining the previous gain profile (see below). In the following discussion the term ‘VCA
faders’ may be taken to refer to any approach where indirect gain control
of the channel is employed, and many of the concepts apply also to DCA
implementations.
With VCA faders it is possible to break the connection between a fader
and the corresponding means of level control, as was described in Fact File
5.5. It is across this breakpoint that an automation system will normally be
connected. The automation processor then reads a digital value corresponding to the position of the fader and can return a value to control the gain of
the channel (see Figure 5.18).
The information sent back to the VCA would depend on the operational
mode of the system at the time, and might or might not correspond directly
to the fader position. Common operational modes are:
■
WRITE: channel level corresponds directly to the fader position
■
READ: channel level controlled by data derived from a previously
stored mix
■
UPDATE: channel level controlled by a combination of previously
stored mix data and current fader position
■
GROUP: channel level controlled by a combination of the channel
fader’s position and that of a group master
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CHAPTER 5: Mixers
FIGURE 5.18
Fader position is encoded so that
it can be read by an automation
computer. Data returned from the
computer is used to control a VCA
through which the audio signal flows.
Fader position data
Fader position data
A/D
convertor
V
D/A
convertor
Voltage returned
from automation processor
Voltage proportional
to position of fader
DC fader
Control voltage
Out
Audio in
VCA
In a VCA implementation the fader position is measured by an analogto-digital convertor (see Chapter 8), which turns the DC value from the
fader into a binary number (usually eight or ten bits) which the microprocessor can read. An eight bit value suggests that the fader’s position can
be represented by one of 256 discrete values, which is usually enough to
give the impression of continuous movements, although professional systems tend to use ten bit representation for more precise control (1024
steps). The automation computer ‘scans’ the faders many times a second
and reads their values. Each fader has a unique address and the information obtained from each address is stored in a different temporary memory
location by the computer. A generalized block diagram of a typical system
is shown in Figure 5.19.
The disadvantage of such a system is that it is not easy to see what the
level of the channel is. During a read or update pass the automation computer is in control of the channel gain, rather than the fader. The fader can
be half way to the bottom of its travel whilst the gain of the VCA is near
the top. Sometimes a mixer’s bargraph meters can be used to display the
value of the DC control voltage which is being fed from the automation to
the VCA, and a switch is sometimes provided to change their function to
this mode. Alternatively a separate display is provided for the automation
computer, indicating fader position with one marker and channel gain with
another.
Automation
From faders
(DC levels)
To fader VCAs
(DC levels)
MPX
From
switches
To
switches
Demux.
Addr.
decode
Addr.
decode
A/D
Switch buffering
and addressing
D/A
Address
Data
Control
CPU
RAM
ROM
Disk
Display
User
IF
Timecode
IF
Such faders are commonly provided with ‘null’ LEDs: little lights on the
fader package which point in the direction that the fader must be moved to
make its position correspond to the stored level. When the lights go out (or
when they are both on), the fader position is correct. This can sometimes
be necessary when modifying a section of the mix by writing over the original data. If the data fed from the automation is different to the position of
the fader, then when the mode is switched from read to write there will be
a jump in level as the fader position takes over from the stored data. The
null lights allow the user to move the fader towards the position dictated by
the stored data, and most systems only switch from read to write when the
null point is crossed, to ensure a smooth transition. The same procedure is
followed when coming out of rewrite mode, although it can be bypassed in
favor of a sudden jump in level.
Update mode involves using the relative position of the fader to modify
the stored data. In this mode, the fader’s absolute position is not important because the system assumes that its starting position is a point of
unity gain, thereafter adding the changes in the fader’s position to the
FIGURE 5.19
Generalized block diagram
of a mixer automation
system handling switches
and fader positions. The
fader interfaces incorporate
a multiplexer (MPX) and
demultiplexer (Demux) to
allow one convertor to be
shared between a number
of faders. RAM is used
for temporary mix data
storage; ROM may hold the
operating software program.
The CPU is the controlling
microprocessor.
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CHAPTER 5: Mixers
stored data. So if a channel was placed in update mode and the fader
moved up by 3 dB, the overall level of the updated passage would be
increased by 3 dB (see Figure 5.20). For fine changes in gain the fader can be
preset near the top of its range before entering update mode, whereas larger
changes can be introduced nearer the bottom (because of the gain law of
typical faders).
Some systems make these modes relatively invisible, anticipating which
mode is most appropriate in certain situations. For example, WRITE mode
is required for the first pass of a new mix, where the absolute fader positions are stored, whereas subsequent passes might require all the faders to
be in UPDATE.
A moving fader system works in a similar fashion, except that the data
which is returned to the fader is used to set the position of a drive mechanism which physically moves the fader to the position in which it was
when the mix was written. This has the advantage that the fader is its own
means of visual feedback from the automation system and will always represent the gain of the channel.
If the fader were to be permanently driven, there would be a problem
when both the engineer and the automation system wanted to control the
gain. Clutches or other forms of control are employed to remove the danger
of a fight between fader and engineer in such a situation, and the fader is
usually made touch-sensitive to detect the presence of a hand on it.
Such faders are, in effect, permanently in update mode, as they can at
any time be touched and the channel gain modified, but there is usually
some form of relative mode which can be used for offsetting a complete
FIGURE 5.20
Graphical illustration of
stages involved in entering
and leaving an UPDATE
or RELATIVE mode on an
automated VCA fader.
Re-enter
Cancel
UPDATE READ mode
Initiate
UPDATE
3 dB
Channel gain (dB)
150
Updated mix gain profile
Original mix gain profile
Raise gain
by 3 dB
Time
Return fader
to null point
Automation
section by a certain amount. The problem with relative offsets and moving
faders is that if there is a sudden change in the stored mix data while the
engineer is holding the fader, it will not be executed. The engineer must
let go for the system to take control again. This is where a combination of
moving fader and VCA-type control comes into its own.
Grouping automated faders
Conventional control grouping (Fact File 5.5) is normally achieved by using
dedicated master faders. In an automated console it may be possible to do
things differently. The automation computer has access to data representing the positions of all the main faders on the console, so it may allow any
fader to be designated a group master for a group of faders assigned to it. It
can do this by allowing the user to set up a fader as a group master (either
by pressing a button on the fader panel, or from a central control panel). It
will then use the level from this fader to modify the data sent back to all
the other faders in that group, taking into account their individual positions as well. This idea means that a master fader can reside physically
within the group of faders to which it applies, although this may not always
be the most desirable way of working.
Sometimes the computer will store automation data relating to groups
in terms of the motions of the individual channels in the group, without
storing the fact that a certain fader was the master, whereas other systems
will store the data from the master fader, remembering the fact that it was
a master originally.
Mute automation
Mutes are easier to automate than faders because they only have two states.
Mute switches associated with each fader are also scanned by the automation computer, although only a single bit of data is required to represent
the state of each switch. A simple electronic switch can be used to effect
the mute, and in analog designs this often takes the form of a FET (field
effect transistor) in the signal path, which has very high attenuation in its
‘closed’ position (see Figure 5.21). Alternatively, some more basic systems
effect mutes by a sudden change in channel gain, pulling the fader down to
maximum attenuation.
Storing the automation data
Early systems converted the data representing the fader positions and mute
switches into a modulated serial data stream which could be recorded
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alongside the audio to which it related on a multitrack tape.
In order to allow updates of the data, at least two tracks
Audio out
were required: one to play back the old data, and one to
record the updated data, these usually being the two outside tracks of the tape (1 and 24 in the case of a 24 track
machine). This was limiting, in that only the two most
recent mixes were ever available for comparison (unless
Mute switch
more tracks were set aside for automation), whole tracks
had to be mixed at a time (because otherwise the updated
track would be incomplete), and at least two audio tracks
–V
were lost on the tape. Yet it meant that the mix data was
FIGURE 5.21
always available alongside the music, eliminating the possiTypical implementation of a FET mute switch.
bility of losing a disk with the mix data stored separately.
More recent systems use computer hardware to store mix data, in RAM
and on disks. Data is synchronized to the audio by recording time-code
alongside audio which uniquely identifies any point in time, this being read
by the automation system and used to relate recording position to stored
data. This method gives almost limitless flexibility in the modification of a
mix, allowing one to store many versions, of which sections can be joined
together ‘off-line’ (that is, without the recorder running) or on-line, to form
the finished product. The finished mix can be dumped to a disk for more
permanent storage, and this disk could contain a number of versions of
the mix.
It is becoming quite common for some automation systems to use MIDI
or MIDI over Ethernet (ipMIDI) for the transmission of automation data.
A basic automation computer associated with the mixer converts control
positions into MIDI information and transmits/receives it using a device
known as a UART which generates and decodes serial data at the appropriate rate for the MIDI standard, as shown in Figure 5.22. MIDI data can
then be stored on a conventional sequencer or using dedicated software, as
described in Chapter 14.
Audio
in
FET
Integrating machine control
Control of recording machines is a common feature of modern mixers. It may
only involve transport remotes being mounted in the center panel somewhere,
or it may involve a totally integrated autolocator/synchronizer associated with
the rest of the automation system. On top-flight desks, controls are provided on the channel modules for putting the relevant tape track into recordready mode, coupled with the record function of the transport remotes. This
requires careful interfacing between the console and the recording machine,
Automation
but means that it is not necessary to work
with a separate recording machine remote
unit by the console.
It is very useful to be able to address the
automation in terms of the mix in progress: in other words, ‘go back to the second
chorus’, should mean something to the system, even if abbreviated. The alternative is
to have to address the system in terms of
timecode locations. Often, keys are provided which allow the engineer to return to
various points in the mix, both from a mix
data point-of-view and from the recording
machines’ point-of-view, so that the automation system locates the recorder to the
position described in the command, ready
to play.
Mixer control interface
CPU
RAM
ROM
UART
Serial MIDI
data
FIGURE 5.22 A UART is used to route MIDI data to and from the
automation computer.
Retrofitting automation
Automation can sometimes be retrofitted into existing analog consoles
that do not have any automation. These systems usually control only the
faders and the mutes, as anything else requires considerable modification
of the console’s electronics, but the relatively low price of some systems
makes them attractive, even on a modest budget. Fitting normally involves
a modification or replacement of the fader package, to incorporate VCAs in
consoles which don’t have them, or to break into the control path between
fader and VCA in systems which do. This job can normally be achieved in
a day. It is also possible to retrofit moving fader automation.
A separate control panel may be provided, with buttons to control the
modes of operation, as well as some form of display to show things like
VCA gains, editing modes, and set-up data. The faders will be interfaced
to a processor rack which would reside either in a remote bay, or under
the console, and this will normally contain a disk drive to store the final
mixes. Alternatively a standard desktop computer will be used as the control interface.
Total automation systems
SSL originally coined the term ‘Total Recall’ for its system, which was a
means of telling the operator where the controls should be and leaving him
or her to reset them him/herself. This saved an enormous amount of time
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in the resetting of the console in between sessions, because it saved having
to write down the positions of every knob and button.
True Total Reset is quite a different proposition and requires an interface between the automation system and every control on the console,
with some means of measuring the position of the control, some means of
resetting it, and some means of displaying what is going on. A number of
options exist, for example one could:
■
■
■
■
motorize all the rotary pots
make all the pots continuously rotating and provide a display
make the pots into up/down-type incrementers with display
provide assignable controls with larger displays
Of these, the first is impractical in most cases due to the space that
motorized pots would take up, the reliability problem and the cost,
although it does solve the problem of display. The second would work, but
again there is the problem that a continuously rotating pot would not have
a pointer because it would merely be a means of incrementing the level
from wherever it was at the time, so extra display would be required and
this takes up space. None the less some ingenious solutions have been
developed, including incorporating the display in the head of rotary controls (see Figure 5.23). The third is not ergonomically very desirable, as the
human prefers analog interfaces rather than digital ones, and there is no
room on a conventional console for all the controls to be of this type with
their associated displays. Most of the designs which have implemented
total automation have adopted a version of the fourth option: that is to use
fewer controls than there are functions, and to provide larger displays.
The concept of total automation is inherent in the principles of an
assignable mixing console, as described below in the section on digital mixers. In such a console, few of the controls carry audio directly as they are
only interfaces to the control system, so one knob may control
the HF EQ for any channel to which it is assigned, for example.
Because of this indirect control, usually via a microprocessor,
it is relatively easy to implement a means of storing the switch
closures and settings in memory for reiteration at a later date.
FIGURE 5.23 Two possible options for
positional display with continuously rotating
knobs in automated systems. (a) Lights
around the rim of the knob itself; (b) lights
around the knob’s base.
Dynamic and static systems
Many analog assignable consoles use the modern equivalent
of a VCA: the digitally controlled attenuator, also to control
the levels of various functions such as EQ, aux sends, and so
on. Full dynamic automation requires regular scanning of all
Digital Mixers
controls so as to ensure smooth operation, and a considerable amount of
data is generated this way. Static systems exist which do not aim to store
the continuous changes of all the functions, but they will store ‘snapshots’
of the positions of controls which can be recalled either manually or with
respect to timecode. This can often be performed quite regularly (many
times a second) and in these cases we approach the dynamic situation, but
in others the reset may take a second or two which precludes the use of it
during mixing. Changes must be silent to be useful during mixing.
Other snapshot systems merely store the settings of switch positions,
without storing the variable controls, and this uses much less processing
time and memory. Automated routing is of particular use in theater work
where sound effects may need to be routed to a complex combination of
destinations. A static memory of the required information is employed so
that a single command from the operator will reset all the routing ready for
the next set of sound cues.
DIGITAL MIXERS
Much of what has been said in this chapter applies equally to both analog and digital mixers, at least in conceptual and operational terms. To
complete the picture some features specific to digital mixers will now be
described, although the principles of digital audio processing are explained
in more detail in Chapter 8. Digital mixing has now reached the point
where it can be implemented cost effectively, and there are a number of
reasonably priced digital mixers with full automation. At the other end of
the scale companies are manufacturing large-scale studio mixers with an
emphasis on ultra-high sound quality and an ergonomically appropriate
control interface. At the low cost end of the scale, digital mixers are implemented within computer-based workstations and represented graphically on
the computer display. Faders and other controls are moved using a mouse.
Audio handling
In a digital mixer incoming analog signals are converted to the digital
domain as early as possible so that all the functions are performed entirely
in the digital domain, with as much as 32-bit internal processing resolution to cope with extremes of signal level, EQ settings and other effects.
The advantage of this is that once the signal is in the digital domain it is
inherently more robust than its analog counterpart: it is virtually immune
from crosstalk, and is unaffected by lead capacitance, electromagnetic fields
from mains wiring, additional circuit distortion and noise, and other forms
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of interference. Digital inputs and outputs can be provided to connect
recording devices and other digital equipment without conversion to analog. Inputs can be a mixture of analog and digital (the latter configurable
via plug-in modules for Tascam, ADAT, Yamaha and AES/EBU formats, for
example) with digital and analog main and monitoring outputs. Functions
such as gain, EQ, delay, phase, routing, and effects such as echo, reverb,
compression and limiting, can all be carried out in the digital domain precisely and repeatably using digital signal processing.
Inputs and outputs, digital and analog, are often provided by a series
of outboard rack-mounting units which incorporate the D/A and A/D
convertors, microphone amplifiers and phantom power, and these can be
positioned where needed, the gain still being adjusted from the mixer. In a
recording studio one rack could be by the control surface itself, one or more
in the recording studio area, and one by the recording machines. In a theater, units would be placed next to power amplifiers, and in off-stage areas
where musicians play. These units are connected in a daisy-chain loop to
the main control surface via coaxial BNC cabling, MADI interface, or proprietary fiber optic links, the latter being preferable for longer distances.
Assignable control surfaces
Operationally a digital mixer can remain similar to its analog counterpart,
although commercial examples have tended at least partially to use assignable control surface designs as they can be easier to implement in digital
mixers. With assignable designs many of the controls of the traditional
console such as pan, EQ, aux send and group assign are present only as
single assignable sections or multi-function controls, many facilities can be
packed into a unit of modest dimensions and quite modest cost.
A fully assignable digital mixing console is ergonomically quite different
from its analog counterpart. Typically, the control surface consists of many
input channel faders and/or rotary knobs, each channel having ‘active’ and
‘select’ buttons. Much smaller areas of the control surface are given over
to single sections of EQ, routing (aux and group) and processing: these sections are automatically assigned to one particular channel when its ‘select’
button is active before adjustments can take place. Thus many processes
which affect the signals are not continuously on view or at the fingertips of
the operator as is the case with the traditional analog desk. The assignable
design is therefore more suitable to recording work (particularly post-session
mixdowns) where desk states can be built up gradually and saved to scene
memories, rather than to live performance and primary recording work where
continuous visual indication of and access to controls remains desirable.
Digital Mixers
Facilities such as channel delay, effects processing, moving fader automation and fader ganging, scene memories offering total recall of all settings, MIDI (including memory load and dump via separate MIDI data
filers), and timecode interface are typically offered, and a display screen
shows the status of all controls: either in simple global formats for the
whole console for parameters such as routing, channel delay, scene memory
details and the like, or in much greater detail for each individual channel.
Metering can also be shown. Cursors facilitate both navigation around the
screen displays and adjustments of the various parameters.
Digital mixers – a case study
The difficulty and expense of implementing
true ‘total recall’ of an analog mixer, that is
automated resetting of all surface controls, has
already been discussed. Digital mixers can incorporate such a feature routinely, and the Digico
D5T illustrated in Figure 5.24 is a typical example of a console in which all set-up parameters
including such things as input gain and phantom power switching are recallable in seconds
when a particular project or show is loaded into
the mixer or recalled from its memory store.
Such mixers are essentially versions of computer mixing systems but with a hardware control surface to provide a more traditional mode
of hands-on operation, still an essential feature
for live mixing work and many types of recording and broadcast session. Ergonomically, the
mixer combines traditional ‘analog’ facilities of
channel faders, aux send and EQ knobs, ‘VCA’
and group faders, with a considerable degree
of assignability using large touch-sensitive
screens and several selectable ‘layers’ across
which banks of inputs can be displayed and
accessed (Figure 5.25). A master output screen
can display a variety of things such as group
outputs, automation parameters, scene memory information, matrix settings and the like.
A console such as this can offer 96 input channels, 20 aux sends, 24 group sends, and in a
FIGURE 5.24 Digico D5T. (Courtesy of the RSC.)
FIGURE 5.25 Detail of input channels display. (Digico D5T,
courtesy of the RSC.)
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theater version a 32-output matrix section. It is not difficult to appreciate
the huge size and cost of an equivalent analog console. A QWERTY key pad
facilitates the labeling of all sections.
Typically, adjacent to each bank of channel faders will be a row of buttons enabling access to different control layers. Layer 1 could be input
channels 1 to 8, the accompanying screen display showing such things as
input gain, phantom power, routing, aux send levels, and EQ. Touching the
appropriate area of the display expands that area for ease of viewing and
adjustment, e.g. touching the EQ section of a channel displays the settings
in much more detail and assigns the EQ controls adjacent to the screen to
that channel. Layer 2 could display channels 25 to 30 (channels 9 to 24
being provided by adjacent banks of faders). Layer 3 of all the fader banks
could give fader control of all the matrix outputs, or all the group outputs, or all the aux master outputs, or a combination. All of these things
are chosen by the operator and set up to his or her requirements. The top
layer would normally be assigned to inputs which need to be continuously
on view, e.g. musicians’ microphones and vocal mics, radio mics, and DI
inputs. The lower layers would be assigned to things such as CD players,
sampler and other replay machine outputs, and probably to some of the
effects returns. These inputs do not normally need to be accessed quickly.
Other features such as digital delay and EQ on inputs and outputs (the latter particularly useful in live sound work), compressors and limiters, and
internal effects processors, are routinely available. This reduces the number of outboard effects processors needed. The settings for these are all programmable and recordable along with the rest of the console settings.
Two main observations can be made regarding the operation of such
consoles compared with their analog counterparts. First, a good deal of
initial setting up, assigning and labeling needs to be carried out before a
session can begin. Input and output channels need to be assigned to appropriate sockets on the outboard units around the building; the various layers have to be assigned to inputs/outputs/auxs/VCAs as appropriate,
and labeled; and a series of scene memories has to be created in anticipation of what will be required for the show or recording session. Second,
the operation of the console often requires a two-stage thinking process.
Although channel faders and some other facilities for a particular layer will
be instantly available for adjustment, many other facilities will need to be
accessed either on a different layer or by touching an area of a screen before
adjustments can be made. Additionally, adjustments need to be stored in a
scene memory. Normally, storing changes such as input gain, EQ, and aux
send levels in a particular scene will automatically store those changes to
the other scene memories. Channel fader adjustments will be stored only
Mixers with Integrated Control of Digital Workstations
to that particular scene. Just what adjustments are stored to the present
scene, and which ones are automatically stored to a bank of scenes, can be
chosen by the operator. The complete project then needs to be stored to the
mixer’s hard disk drive, and preferably also to an external backup. This all
needs an operator who is familiar with that particular console and its software quirks. Digital consoles necessarily have many common features, but
manufacturers have their own proprietary ways of doing things. The typical analog console, in contrast, will be fairly familiar to a user after ten or
15 minutes.
Digitally controlled analog mixers
A digitally controlled analog console will be
looked at briefly next. The Midas Heritage 3000
shown in Figure 5.26 is a good example of such
a mixer. Its control surface is analog, and the
signals remain in the analog domain throughout. Digital control gives such things as mute
and mute group automation, VCA assign, and
virtual fader automation (a row of LEDs adjacent to each fader displays the audio level of the
fader regardless of its physical position; moving
the fader to the top lit LED gives the operator
manual control). Scene memories can thus be
programmed into the desk giving the appropriate fader positions and channel mutes, these
being of great value in the live mixing situations
for which such consoles are designed. Other FIGURE 5.26 The Midas Heritage 3000. (Courtesy of Klark
consoles also provide automation of EQ in/out, Teknik.)
insert, aux send enable, group assign, and moving fader automation, albeit at a somewhat higher cost, and such consoles
undoubtedly prove their worth in the live sound market where visiting and
freelance sound engineers need to become quickly familiar with a console
which does not have too much automation. Such consoles are likely to be
in use until well into the second decade of the present century.
MIXERS WITH INTEGRATED CONTROL OF
DIGITAL WORKSTATIONS
Integrated control of digital audio workstations is now a growing feature
of either analog or digital mixing consoles. In the case of some designs the
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mixing console has become little more than
a sophisticated control surface, enabling the
functions of a workstation to be adjusted using
more conventional controls. This is offered as
an alternative to using a computer display and
mouse, which can be inconvenient when trying
to handle complex mixes. In such cases most of
the audio processing is handled by the workstation hardware, either using desktop computer
FIGURE 5.27 A typical workstation-integrated mixer control
processing power or dedicated signal processing
surface from the Digidesign ICON series (D-Control ES). (Courtesy
cards. Some audio handling can be included,
of Digidesign.)
such as monitoring and studio communication
control. The mixer control surface is connected
to the workstation using a dedicated interface
such as MIDI or Ethernet. Such control surfaces often include remote control facilities for
the workstation transport and editing functions.
An example from the Digidesign ICON series is
shown in Figure 5.27.
An alternative is to employ an external
analog mixer that has dedicated workstation
control facilities, such as the SSL Matrix, pictured in Figure 5.28. In this case the mixer
handles more of the audio processing itself
and can be used as an adjunct or alternative
to the onboard digital processing of the workstation, with comprehensive routing to and
FIGURE 5.28 Analog mixer with integrated workstation from conventional analog outboard equipcontrol: SSL Matrix. (Courtesy of Solid-State Logic.)
ment. It enables analog mixing of audio channels outside the workstation, which has
become a popular way of working for some. The control interface to the
workstation is by means of MIDI over Ethernet (ipMIDI), and USB to
carry the equivalent of computer keyboard commands. The control protocol can be made to conform either to Mackie’s HUI (Human User
Interface) or MCU (Mackie Control Universal), or to a special configuration of MIDI controllers (see Chapter 14), and the mixer’s digitally controlled attenuators can be remote controlled using MIDI commands.
The latter facility enables the mixer to be automated using MIDI tracks
on the workstation. A typical configuration of this system is shown in
Figure 5.29.
Introduction to Mixing Approaches
FIGURE 5.29
A typical system
configuration of the SSL
Matrix console showing
interconnections to studio
equipment and workstation.
DAW audio interface
DAW channel I/O
DAW FX I/O
Line inputs
Monitor LS and
headphones
Insert
sends and
returns
External
monitor
inputs
Ethernet
Soft-key and
DAW control plug-in control
Transport
control
DAW control interface
INTRODUCTION TO MIXING APPROACHES
This section provides an introduction to basic mixer operation and level
setting.
Acoustic sources will be picked up by microphones and fed into the mic
inputs of a mixer (which incorporates amplifiers to raise the low-voltage
output from microphones), whilst other sources usually produce so-called
‘line level’ outputs, which can be connected to the mixer without extra
amplification. In the mixer, sources are combined in proportions controlled
by the engineer and recorded. In ‘straight-to-stereo’ (or surround) techniques, such as a classical music recording, microphone sources are often
mixed ‘live’ without recording to a multitrack medium, creating a session
master which is the collection of original recordings, often consisting of a
number of takes of the musical material. The balance between the sources
must be correct at this stage, and often only a small number of carefully
positioned microphones are used. The session master recordings will then
proceed to the editing stage where takes are assembled in an artistically satisfactory manner, under the control of the producer, to create a final master
which will be transmitted or made into a commercial release. This final
master could be made into a number of production masters which will be
used to make different release formats. In the case of ‘straight-to-stereo’
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mixing the console used may be a simpler affair than that used for multitrack recording, since the mixer’s job is to take multiple inputs and combine them to a single stereo output, perhaps including processing such as
equalization. This method of production is clearly cheaper and less time
consuming than multitrack recording, but requires skill to achieve a usable
balance quickly. It also limits flexibility in post-production. Occasionally,
classical music is recorded in a multitrack form, especially in the case of
complex operas or large-force orchestral music with a choir and soloists,
where to get a correct balance at the time of the session could be costly
and time consuming. In such a case, the production process becomes more
similar to the pop recording situation described below.
‘Pop’ music is rarely recorded live, except at live events such as concerts,
but is created in the recording studio. Acoustic and electrical sources are fed
into a mixer and recorded on to a multitrack medium, often a few tracks at
a time, gradually building up a montage of sounds. The resulting recording
then contains a collection of individual sources on multiple tracks which
must subsequently be mixed into the final release format. Individual songs
or titles are recorded in separate places on the tape, to be compiled later. It
is not so common these days to record multitrack pop titles in ‘takes’ for
later editing, as with classical music, since mixer automation allows the
engineer to work on a song in sections for automatic execution in sequence
by a computer. In any case, multitrack machines have comprehensive
‘drop-in’ facilities for recording short inserted sections on individual tracks
without introducing clicks, and a pop-music master is usually built up by
laying down backing tracks for a complete song (drums, keyboards, rhythm
guitars, etc.) after which lead lines are overdubbed using drop-in facilities.
Occasionally multitrack recordings are edited or compiled (‘comped’) early
on during a recording session to assemble an acceptable backing track from
a number of takes, after which further layers are added. Considerable use
may be made of computer-sequenced electronic instruments, under MIDI
control, often in conjunction with multitrack disk recording. The computer
controlling the electronic instruments is synchronized to the recording
machine using time code and the outputs of the instruments are fed to the
mixer to be combined with the non-sequenced sources.
Once the session is completed, the multitrack recording is mixed
down. This is often done somewhere differently from the original session, and involves feeding the outputs of each track into individual inputs
of the mixer, treating each track as if it were an original source. The balance between the tracks, and the positioning of the tracks in the stereo
image, can then be carried out at leisure (within the budget constraints of
the project!), often without all the musicians present, under control of the
Basic Operational Techniques
producer. During the mixdown, further post-production takes place such
as the addition of effects from outboard equipment to enhance the mix.
An automation system is often used to memorize fader and mute movements on the console, since the large number of channels involved in modern recording makes it difficult if not impossible for the engineer to mix a
whole song correctly in one go. Following mixdown, the master that results
will be edited very basically, in order to compile titles in the correct order
for the production master. The compiled tape will then be mastered for the
various distribution media.
BASIC OPERATIONAL TECHNIQUES
Level setting
If one is using a microphone to record speech or classical music then normally a fairly high input gain setting will be required. If the microphone is
placed up against a guitar amplifier then the mic’s output will be high and
a much lower input gain setting can be used. There are essentially three
ways of setting the gain control to the optimum position. First, using PFL
or prefade listen (see Fact File 5.3).
PFL is pressed, or the fader overpressed (i.e. pressed beyond the bottom of its travel against a sprung microswitch), on the input module concerned and the level read on either a separate PFL meter or with the main
meters switched to monitor the PFL bus. The channel input gain should
be adjusted to give a meter reading of, say, PPM 5, or 0 VU on older analog
desks, and a meter reading of perhaps 6–10 dB below maximum on a digital
desk. This gain-setting procedure must be carried out at a realistic input
level from the source. It is frequently the case during rehearsals that vocalists and guitarists will produce a level that is rather lower than that which
they will use when they actually begin to play.
The pan control should be set next (see Fact File 5.2) to place the source
in the stereo image. The main output faders will normally be set to 0 dB
on their calibration, which is usually at the top. The channel faders can
then be set to give both the desired subjective sound balance and appropriate output meter readings.
The second way of setting the gain is a good way in its own right, and it
has to be used if PFL facilities are not provided. First of all both the channel
fader and the output faders need to be positioned to the 0 dB point. This
will be either at the top of the faders’ travels or at a position about a quarter
of the way down from the top of their travel. If no 0 dB position is indicated
then the latter position should be set. After the pan control and faders have
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CHAPTER 5: Mixers
been positioned, the input gain may then be adjusted to give the desired
reading on the output level meters. When several incoming signals need
to be balanced the gain controls should all be positioned to give both the
desired sound balance between them and the appropriate meter readings –
normally PPM 6 or just over 0 VU during the loudest passages.
These two gain-setting methods differ in that with the former method
the channel fader positions will show a correspondence to the subjective
contribution each channel is making towards the overall mix, whereas the
latter method places all the channel faders at roughly the same level.
The third way is similar to the second way, but one channel at a time is
set up, placing channel and output faders at 0 dB and adjusting the gain for a
peak meter reading. That channel fader is then turned completely down and
the next channel is set up in a similar way. When all the channels which are
to be used have been set up, the channel faders can then be advanced to give
both the desired subjective balance and peak meter readings.
Use of the EQ controls often necessitates the resetting of the channel’s
gain. For example, if a particular instrument requires a bit of bass boost,
applying this will also increase the level of signal and so the gain will often
need to be reduced a little to compensate. Applying bass or treble cut will
sometimes require a small gain increase.
Using auxiliary sends
Aux facilities were described in ‘Auxiliary sends’, above. The auxiliaries
are configured either ‘pre-fade’ or ‘post-fade’. Pre-fade aux sends are useful
for providing a monitor mix for musicians, since this balance will be unaffected by movements of the faders which control the main mix. The engineer then retains the freedom to experiment in the control room without
disturbing the continuity of feed to the musicians.
Post-fade sends are affected by the channel fader position. These are
used to send signals to effects devices and other destinations where it is
desirable to have the aux level under the overall control of the channel
fader. For example, the engineer may wish to add a little echo to a voice.
Aux 2, set to post-fade, is used to send the signal to an echo device, probably positioning the aux 2 control around the number 6 position and the
aux 2 master at maximum. The output of the echo device is returned to
another input channel or an echo return channel, and this fader can be
adjusted to set the amount of echo. The level of echo will then rise and fall
with the fader setting for the voice.
The post-fade aux could also be used simply as an additional output to drive separate amplifiers and speakers in another part of a hall, for
example.
Recommended Further Reading
Using audio groups
The group outputs (see ‘Channel grouping’, above) or multitrack routing
buses (see ‘Routing section’, above) can be used for overall control of various separate groups of instruments, depending on whether mixing down or
track laying. For example, a drum kit may have eight microphones on it.
These eight input channels can be routed to groups 1 and 2 with appropriate stereo pan settings. Groups 1 and 2 would then be routed to stereo outputs left and right respectively. Overall control of the drum kit level is now
achieved simply by moving group faders 1 and 2.
RECOMMENDED FURTHER READING
See ‘General further reading’ at the end of this book.
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CHAPTER 6
Analog Recording
CH A P T E R C O N TE N T S
A Short History of Analog Recording
Early recording machines
Electrical recording
Later developments
Magnetic Tape
Structure
Open-reel tape
The Magnetic Recording Process
Introduction
Equalization
The Tape Recorder
Studio recorder
The multitrack machine
Magnetic Recording Levels
What are Test Tapes For?
Tape Machine Alignment
Head inspection and demagnetization
Replay alignment
Record alignment
Mechanical Transport Functions
The Compact Cassette
Background
Alignments
Multitrack cassette recorders
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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CHAPTER 6: Analog Recording
Successive editions of this book have quite naturally seen the emphasis
placed more and more on digital topics and away from analog. Yet even the
gramophone record has survived for rather more years than many would
have predicted, and the analog open reel tape recorder, particularly in multitrack form in rock recording studios where some engineers and artists value
its distortion characteristics, still enjoys a significant amount of use. An
analog recording chapter continues to be justified in this edition therefore,
albeit in a shortened form.
A SHORT HISTORY OF ANALOG RECORDING
Early recording machines
When Edison and Berliner first developed recording machines in the last
years of the nineteenth century they involved little or no electrical apparatus.
Certainly the recording and reproduction process itself was completely
mechanical or ‘acoustic’, the system making use of a small horn terminated
in a stretched, flexible diaphragm attached to a stylus which cut a groove
of varying depth into the malleable tin foil on Edison’s ‘phonograph’ cylinder or of varying lateral deviation in the wax on Berliner’s ‘gramphone’ disc
(see Figure 6.1). On replay, the undulations of the groove caused the stylus
and diaphragm to vibrate, thus causing the air
in the horn to move in sympathy, thus reproCylinder covered with foil
ducing the sound – albeit with a very limited
Stylus
frequency range and very distorted.
Cylinders for the phonograph could be
Horn pickup
recorded by the user, but they were difficult to
Diaphragm
duplicate for mass production, whereas discs
for the gramophone were normally replay only,
but they could be duplicated readily for mass
production. For this reason discs fairly quickly
won the day as the mass-market prerecorded
music medium. There was no such thing as
magnetic recording tape at the time, so recordings were made directly onto a master disc,
lasting for the duration of the side of the disc –
a maximum of around 4 minutes – with no
FIGURE 6.1 The earliest phonograph used a rotating foilcovered cylinder and a stylus attached to a flexible diaphragm. The possibility for editing. Recordings containing
recordist spoke or sang into the horn causing the stylus to vibrate, errors were either remade or they were passed
thus inscribing a modulated groove into the surface of the soft foil. On with mistakes intact. A long item of music
replay the modulated groove would cause the stylus and diaphragm would be recorded in short sections with gaps
to vibrate, resulting in a sound wave being emitted from the horn.
to change the disc, and possibilities arose for
A Short History of Analog Recording
discontinuities between the sections as well as variations in pitch and tempo.
Owing to the deficiencies of the acoustic recording process, instruments had
to be grouped quite tightly around the pick-up horn in order for them to be
heard on the recording, and often louder instruments were substituted for
quieter ones (the double bass was replaced by the tuba, for example) in order
to correct for the poor frequency balance. It is perhaps partly because of this
that much of the recorded music of the time consisted of vocal soloists and
small ensembles, since these were easier to record than large orchestras.
Electrical recording
During the 1920s, when broadcasting was in its infancy, electrical recording
became more widely used, based on the principles of electromagnetic transduction (see Chapter 3). The possibility for a microphone to be connected
remotely to a recording machine meant that microphones could be positioned
in more suitable places, connected by wires to a complementary transducer at
the other end of the wire, which drove the stylus to cut the disc. Even more
usefully, the outputs of microphones could be mixed together before being
fed to the disc cutter, allowing greater flexibility in the balance. Basic variable
resistors could be inserted into the signal chain in order to control the levels
from each microphone, and valve amplifiers would be used to increase the
electrical level so that it would be suitable to drive the cutting stylus.
The sound quality of electrical recordings shows a marked improvement over acoustic recordings, with a wider frequency range and a greater
dynamic range. Experimental work took place both in Europe and the USA
on stereo recording and reproduction, but it was not to be until much later
that stereo took its place as a common consumer format, nearly all records
and broadcasts being in mono at that time.
Later developments
During the 1930s work progressed on the development of magnetic recording
equipment, and examples of experimental wire recorders and tape recorders
began to appear, based on the principle of using a current flowing through a
coil to create a magnetic field which would in turn magnetize a moving metal
wire or tape coated with magnetic material. The 1940s, during wartime, saw
the introduction of the first AC-biased tape recorders, which brought with
them good sound quality and the possibility for editing. Tape itself, though,
was first made of paper coated with metal oxide which tended to deteriorate rather quickly, and only later of plastics which proved longer lasting and
easier to handle. In the 1950s the microgroove LP record appeared, with
markedly lower surface noise and improved frequency response, having a
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CHAPTER 6: Analog Recording
playing time of around 25 minutes per side. This was an ideal medium for
distribution of commercial stereo recordings, which began to appear in the
late 1950s, although it was not until the 1960s that stereo really took hold.
In the early 1960s the first multitrack tape recorders appeared, the Beatles
making use of an early four-track recorder for their ‘Sergeant Pepper’s Lonely
Hearts Club Band’ album. The machine offered the unprecedented flexibility
of allowing sources to be recorded separately, and the results in the stereo
mix are panned very crudely to left and right in somewhat ‘gimmicky ’ stereo.
Mixing equipment in the 1950s and 1960s was often quite basic, compared
with today’s sophisticated consoles, and rotary faders were the norm. There
simply was not the quantity of tracks involved as exists today.
MAGNETIC TAPE
Structure
Magnetic tape consists of a length of plastic material which is given a surface coating capable of retaining magnetic flux rather in the manner that,
say, an iron rod is capable of being magnetized. The earliest recorders
actually used a length of iron wire as the recording medium. In practice
all modern tape has a polyester base which was chosen, after various trials with other formulations which proved either too brittle (they snapped
easily) or too plastic (they stretched), for its good strength and dimensional
stability. It is used throughout the tape industry from the dictation microcassette to the 2 inch (5 cm) multitrack variety. The coating is of a metal
oxide, or metal alloy particles.
Open-reel tape
Open-reel quarter-inch tape intended for analog recorders has been available in a variety of thicknesses. Standard Play tape has an overall thickness
of 50 microns (micrometers), and a playing time (at 15 inches (38 cm) per
second) of 33 minutes is obtained from a 10 inch (25 cm) reel. Long Play
tape has an overall thickness of 35 microns giving a corresponding 48 minutes of playing time, which is very useful for live recording work. In the
past ‘Double Play ’ and even ‘Triple Play ’ thicknesses have been available,
these being aimed at the domestic open-reel market. These formulations
are prone to snapping or stretching, as well as offering slightly poorer sound
quality, and should not really be considered for professional use.
Standard Play tape is almost always ‘back coated’. A rough coating is
applied to the back of the tape during manufacture which produces neater
and more even winding on a tape machine, by providing a certain amount of
The Magnetic Recording Process
friction between layers which holds the tape in place. Also, the rough surface
helps prevent air being trapped between layers during fast spooling which
can contribute to uneven winding. Long Play tape is also available with a
back coating, but as often as not it will be absent. It is worth noting that the
flanges of a tape spool should only serve to protect the tape from damage. The
‘pancake’ of tape on the spool should not touch these flanges. Metal spools are
better than plastic spools because they are more rigid and they do not warp.
Professional open-reel tape can be purchased either on spools or in ‘pancake’
form on hubs without flanges. The latter is of course cheaper, but considerable
care is needed in its handling so that spillage of the unprotected tape does not
occur. Such pancakes are either spooled onto empty reels before use, or they
can be placed on top of a special reel with only a lower flange. Professional tape
machines are invariably operated with their decks horizontal. Half inch, 1 inch
and 2 inch tape intended for multitrack recorders always comes on spools, is
always of Standard Play thickness, and is always back coated.
THE MAGNETIC RECORDING PROCESS
Introduction
Since tape is magnetic, the recording process must convert an electrical
audio signal into a magnetic form. On replay the recorded magnetic signal must be converted back into electrical form. The process is outlined
in Fact File 6.1. Normally a professional tape recorder has three heads, as
shown in Figure 6.2, in the order erase–record–replay. This allows for the
tape to be first erased, then re-recorded, and then monitored by the third
head. The structure of the three heads is similar, but the gap of the replay
head is normally smaller than that of the record head. It is possible to use
the same head for both purposes, but usually with a compromise in performance. Such a two-head arrangement is often found in cheaper cassette
machines which do not allow off-tape monitoring whilst recording. A simplified block diagram of a typical tape recorder is shown in Figure 6.3.
Erase
Record
Direction of tape travel
Replay
FIGURE 6.2
Order of heads on a
professional analog tape
recorder.
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CHAPTER 6: Analog Recording
FA C T F I L E 6 . 1 A MA G N E TIC R ECORDI NG HEAD
When an electrical current flows through a coil of wire a
magnetic field is created. If the current only flows in one
direction (DC) the electromagnet thus formed will have a
north pole at one end and a south pole at the other (see
diagram). The audio signal to be recorded onto tape is
alternating current (AC), and when this is passed through
a similar coil the result is an alternating magnetic field
whose direction changes according to the amplitude and
phase of the audio signal.
Magnetic flux is rather like the magnetic equivalent
of electrical current, in that it flows from one pole of the
magnet to the other in invisible ‘lines of flux’. For sound
recording it is desirable that the tape is magnetized with a
pattern of flux representing the sound signal. A recording
head is used which is basically an electromagnet with a
small gap in it. The tape passes across the gap, as shown
in the diagram. The electrical audio signal is applied
across the coil and an alternating magnetic field is created across the gap. Since the gap is filled with a nonmagnetic material it appears as a very high ‘resistance’ to
magnetic flux, but the tape represents a very low resistance in comparison and thus the flux flows across the
gap via the tape, leaving it magnetized.
On replay, the magnetized tape moves across the
head gap of a similar or identical head to that used during
Erase and
bias oscillator
recording, but this time the magnetic flux on the tape
flows through the head and thus induces a current in the
coil, providing an electrical output.
+
Current
N
S
AC input
Magnetic
material
Record
EQ
Output
amplifier
Replay EQ
Bias
trap
Flux path
Magnetic coating
Erase
head
Record
head
Output
Replay
pre-amp
Coil
Gap
Input
Record
amplifier
–
DC input
Replay
head
Tape
Tape backing
FIGURE 6.3 Simplified
block diagram of a typical
analog tape recorder. The bias
trap is a filter which prevents
the HF bias signal feeding
back into an earlier stage.
The Magnetic Recording Process
The magnetization characteristics of tape are by no means linear, and
therefore a high-frequency signal known as bias is added to the audio signal at the record head, generally a sine wave of between 100 and 200 kHz,
which biases the tape towards a more linear part of its operating range.
Without bias the tape retains very little magnetization and distortion is
excessive. The bias signal is of too high a frequency to be retained by the
tape, so does not appear on the output during replay. Different types of
tape require different levels of bias for optimum recording conditions to be
achieved, and this will be discussed in bias requirements, below.
Equalization
Short-circuit flux (dB)
‘Pre-equalization’ is applied to the audio signal before recording. This equalization is set in such a way that the replayed short-circuit flux in an ideal
head follows a standard frequency response curve (see Figure 6.4). A number of standards exist for different tape speeds, whose time constants are
the same as those quoted for replay EQ in Table 6.1. Although the replayed
flux level must conform to these curves, the electrical pre-EQ may be very
different, since this depends on the individual head and tape characteristics.
Replay equalization (see Figure 6.5) is used to ensure that a flat response is
available at the tape machine’s output. It compensates for losses incurred
in the magnetic recording/replay process, the rising output of the replay
head with frequency, the recorded flux characteristic, and the fall-off in HF
FIGURE 6.4
Examples of standardized
recording characteristics
for short-circuit flux. (NB:
this is not equivalent to
the electrical equalization
required in the record
chain, but represents the
resulting flux level replayed
from tape, measured using
an ideal head).
0
–10
–20
100
1 kHz
10 kHz
Frequency Hz
Equalization
standard
17.5 μs (AES, 30 ips)
35 μs (IEC, 15 ips)
3180+50 μs (NAB, 7.5 and 15 ips)
70 μs (IEC, 7.5 ips)
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CHAPTER 6: Analog Recording
Table 6.1
Replay equalization time constants
Time constants (μs)
Tape speed
FIGURE 6.5
Examples of replay
equalization required to
correct for the recording
characteristic (see Figure
6.5), replay-head losses,
and the rising output of the
replay head with frequency.
ips (cm/s)
Standard
30 (76)
15 (38)
15 (38)
7.5 (19)
7.5 (19)
3.75 (9.5)
1.875 (4.75)
1.875 (4.75)
AES/IEC
IEC/CCIR
NAB
IEC/CCIR
NAB
All
DIN (Type I)
DIN (Type II or IV)
Replay EQ characteristic (dB)
174
HF
17.5
35
50
70
50
90
120
70
LF
–
–
3180
–
3180
3180
3180
3180
0
–10
–20
–30
–40
–50
100
1 kHz
10 kHz
Frequency Hz
Equalization
standard
17.5 μs (AES, 30 ips)
35 μs (IEC, 15 ips)
3180+50 μs (NAB, 7.5 and 15 ips)
70 μs (IEC, 7.5 ips)
Likely additional HF EQ (70 μs only shown)
response where the recorded wavelength approaches the head gap width
(see Fact File 6.2). Table 6.1 shows the time constants corresponding to the
turnover frequencies of replay equalizers at a number of tape speeds. Again
a number of standards exist. Time constant (normally quoted in microseconds) is the product of resistance and capacitance (RC) in the equivalent
equalizing filter, and the turnover frequency corresponding to a particular
time constant can be calculated using:
f ⫽ 1/(2πRC)
The Magnetic Recording Process
FA C T F I L E 6 . 2 R E P L AY H E A D E FFECTS
through the head, or when flux takes a ‘short-circuit’ path
through the head. This results in low-frequency ‘head
bumps’ or ‘woodles’ in the frequency response. The diagram below summarizes these effects on the output of the
replay head.
Gap
Recorded wavelength
Extinction
frequency
Output level
The output level of the replay head coil is proportional
to the rate of change of flux, and thus the output level
increases by 6 dB per octave as frequency rises (assuming a constant flux recording). Replay equalization is used
to correct for this slope.
At high frequencies the recorded wavelength on tape
is very short (in other words the distance between magnetic flux reversals is very short). The higher the tape
speed, the longer the recorded wavelength. At a certain
high frequency the recorded wavelength will equal the
replay-head gap width (see diagram) and the net flux in
the head will be zero, thus no current will be induced. The
result of this is that there is an upper cut-off frequency on
replay (the extinction frequency), which is engineered to
be as high as possible.
Gap effects are noticeable below the cut-off frequency, resulting in a gradual roll-off in the frequency
response as the wavelength approaches the gap length.
Clearly, at low tape speeds (in which case the recorded
wavelength is short) the cut-off frequency will be lower
than at high tape speeds for a given gap width.
At low frequencies, the recorded wavelength
approaches the dimensions of the length of tape in contact with the head, and various additive and cancelation
effects occur when not all of the flux from the tape passes
6 dB/octave
LF bumps
Frequency
The LF time constant of 3180 μs was introduced in the American NAB
standard to reduce hum in early tape recorders, and has remained. HF time
constants resulting in low turnover frequencies tend to result in greater
replay noise, since HF is boosted over a wider band on replay, thus amplifying tape noise considerably. This is mainly why Type I cassette tapes
(120 μs EQ) sound noisier than Type II tapes (70 μs EQ). Most professional
tape recorders have switchable EQ to allow the replay of NAB- and IEC/
CCIR-recorded tapes. EQ switches automatically with tape speed in most
machines.
Additional adjustable HF and LF EQ is provided on many tape machines,
so that the recorder’s frequency response may be optimized for a variety of
operational conditions, bias levels and tape types.
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THE TAPE RECORDER
Studio recorder
Professional open-reel recorders fall into two categories: console mounted
and portable. The stereo console recorder, intended for permanent or
semi-permanent installation in a recording studio, outside broadcast truck
or whatever generally sports rather few facilities, but has balanced inputs
and outputs at line level (no microphone inputs), transport controls, editing modes, possibly a headphone socket, a tape counter (often in real time
rather than in arbitrary numbers or revs), tape speed selector, reel size selector, and probably (though not always) a pair of level meters. It is deliberately
simple because its job is to accept a signal, store it as faithfully as possible,
and then reproduce it on call. It is also robustly built, stays aligned for long
periods without the need for frequent adjustment, and will be expected to
perform reliably for long periods. A typical example is pictured in Figure 6.6.
The inputs of such a machine will be capable of accepting high electrical
levels – up to at least ⫹20 dBu or around 8 volts – so that
there is virtually no possibility of electrical input overload.
The input impedance will be at least 10 kΩ. The outputs
will be capable of driving impedances down to 600 ohms,
and will have a source impedance of below 100 ohms. A
facility will be provided for connecting a remote control
unit so that the transport can be controlled from the mixing console, for instance.
Its semi-professional counterpart will be capable at
its best of a performance that is a little inferior, and in
addition to being smaller and lighter will sport rather
more facilities such as microphone inputs and various
alternative input and output options. Headphone outlets
will be provided along with record-level meters, source/
tape monitor switching, variable output level, and perhaps ‘sound on sound’-type facilities for simple overdub
work. The semi-professional machine will not usually be as robustly constructed, this being of particular
concern for machines which are to be transported since
rough treatment can easily send a chassis askew, causing misalignment of the tape transport system which
FIGURE 6.6 A typical professional open-reel will be virtually impossible to correct. Some chassis are
two-track analog tape recorder: the Studer A807-TC. constructed of pressed steel which is not very rigid. A
(Courtesy of FWO Bauch Ltd.)
casting is much better.
The Tape Recorder
The professional portable tape machine, unlike
its console equivalent, needs to offer a wide range
of facilities since it will be required to provide such
things as balanced outputs and inputs, both at line
and microphone level, phantom and A–B mic powering, metering, battery operation which allows
usefully long recording times, the facility to record
timecode and pilot tone for use in TV and film
work, illumination of the important controls and
meters, and possibly even basic mixing facilities. It
FIGURE 6.7 A typical professional portable two-track
must be robust to stand up to professional field use,
and small enough to be carried easily. Nevertheless, recorder: the Nagra IV-S. (Courtesy of Nagra Kudelski
it should also be capable of accepting professional 10 (GB) Ltd.)
inch (25 cm) reels, and adaptors are usually available
to facilitate this. A lot has to be provided in a small
package, and the miniaturization necessary does not come cheap. The
audio performance of such machines is at least as good as that of a studio
recorder. A typical commercial example is pictured in Figure 6.7.
The multitrack machine
Multitrack machines come in a variety of track configurations and quality
levels. The professional multitrack machine tends to be quite massively engineered and is designed to give consistent, reliable performance on a par with
the stereo mastering machine. The transport needs to be particularly fine so
that consistent performance across the tracks is achieved. A full reel of 2 inch
tape is quite heavy, and powerful spooling motors and brakes are required to
keep it under control. Apart from the increased number of tracks, multitrack
machines are basically the same as their stereo counterparts and manufacturers tend to offer a range of track configurations within a given model type.
Alignment of course takes a lot longer, and computer control of this is most
welcome when one considers that 24 tracks implies 168 separate adjustments!
A useful feature to have on a multitrack recorder is an automatic repeat
function or autolocate. The real-time counter can be programmed so that
the machine will repeat a section of the tape over and over again within the
specified start and end points to facilitate mixdown rehearsals. Multitrack
recorders will be equipped with a number of unique features which are vital
during recording sessions. For example, sync replay (see Fact File 6.3), gapless, noiseless punch-in (allowing any track to be dropped into record at
any point without introducing a gap or a click) and spot erasure (allowing a
track to be erased manually over a very small portion of tape).
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FA C T F I L E 6 . 3 S YN C R E P L AY
The overdubbing process used widely in multitrack
recording requires musicians to listen to existing tracks
on the tape whilst recording others. If replay was to
come from the replay head and the new recording was
to be made onto the record head, a recorded delay would
arise between old and new material due to the distance
between the heads. Sync replay allows the record head
to be used as a replay head on the tracks which are not
currently recording, thus maintaining synchronization.
The sound quality coming off the record head (called
~ +10 dB
0 dB
the sync head in this mode) is not always as good as that
coming off the replay head, because the gap is larger, but
it is adequate for a cue feed. Often separate EQ is provided for sync replay to optimize this. Mixdown should
always be performed from the replay head.
Some manufacturers have optimized their head technology such that record and replay heads are exactly the
same, and thus there is no difference between true replay
and sync replay.
MOL (1 kHz, 3% distortion)
Reference level (320 nWb m –1)
MAGNETIC RECORDING
LEVELS
It has already been said that the equivalent of electrical current in magnetic
terms is magnetic flux, and it is necessary
to understand the relationship between
electrical levels and magnetic recording
levels on tape. The performance of an
Typical noise floor (CCIR 468-2 wtd.) analog tape recorder depends very much
~ –50 dB
on the magnetic level recorded on the
FIGURE 6.8 The available dynamic range on an analogue tape lies
tape, since at high levels one encounbetween the noise floor and the MOL. Precise figures depend on tape and
ters distortion and saturation, whilst at
tape machine.
low levels there is noise (see Figure 6.8).
A window exists, between the noise and the distortion, in which the audio
signal must be recorded, and the recording level must be controlled to lie
optimally within this region. For this reason the relationship between the
electrical input level to the tape machine and the flux level on tape must be
established so that the engineer knows what meter indication on a mixer
corresponds to what magnetic flux level. Once a relationship has been set up
it is possible largely to forget about magnetic flux levels and concentrate on
the meters. Fact File 6.4 discusses magnetic flux reference levels.
Available
dynamic
range
WHAT ARE TEST TAPES FOR?
A test tape is a reference standard recording containing pre-recorded tones
at a guaranteed magnetic flux level. A test tape is the only starting point for
What Are Test Tapes For?
FA C T F I L E 6 . 4 MA G N E T IC R E FERENCE LEVELS
Magnetic flux density on tape is measured in nanowebers per meter (nWb m⫺1), the weber being the unit of
magnetic flux. Modern tapes have a number of important
specifications, probably the most significant being maximum output level (MOL), HF saturation point and noise
level. (These parameters are also discussed in Appendix
1.) The MOL is the flux level at which third-harmonic distortion reaches 3% of the fundamental’s level, measured
at 1 kHz (or 5% and 315 Hz for cassettes), and can be
considered as a sensible peak recording level unless
excessive distortion is required for some reason. The MOL
for a modern high-quality tape lies at a magnetic level of
around 1000 nWb m⫺1, or even slightly higher in some
cases, and thus it is wise to align a tape machine such
that this magnetic level corresponds fairly closely to the
peak level indication on a mixer’s meters.
A common reference level in electrical terms is
0 dBu, which often lines up with PPM 4 or ⫺4 VU on a
mixer’s meter. This must be aligned to correspond to a
recognized magnetic reference level on the tape, such as
320 nWb m⫺1. Peak recording level, in this case, would
normally be around 8 dBu if the maximum allowed PPM
indication was to be 6, as is conventional. This would
in turn correspond to a magnetic recording level of
804 nWb m⫺1, which is close to the MOL of the tape and
would probably result in around 2% distortion.
There are a number of accepted magnetic reference
levels in use worldwide, the principal ones being 200,
250 and 320 nWb m⫺1. There is 4 dB between 200 and
320 nWb m⫺1, and thus a 320 nWb m⫺1 test tape should
replay 4 dB higher in level on a meter than a 200 nWb m⫺1
test tape. American test tapes often use 200 nWb m⫺1
(so-called NAB level), whilst German tapes often use
250 nWb m⫺1 (sometimes called DIN level). Other European
tapes tend to use 320 nWb m⫺1 (sometimes called IEC
level). Test tapes are discussed further in the main text.
There is currently a likelihood in recording studios
that analog tapes are being under-recorded, since the
performance characteristics of modern tapes are now
good enough to allow higher peak recording levels than
before. A studio which aligned PPM 4 to equal 0 dBu, in
turn to correspond to only 200 nWb m⫺1 on tape, would
possibly be leaving 4–6 dB of headroom unused on the
tape, sacrificing valuable signal-to-noise ratio.
aligning a tape machine, since otherwise there is no way of knowing what
magnetic level will end up on the tape during recording. During alignment,
the test tape is replayed, and a 1 kHz tone at the specified magnetic flux
level (say 320 nWb m⫺1) produces a certain electrical level at the machine’s
output. The output level would then be adjusted for the desired electrical
level, according to the studio’s standard (say 0 dBu), to read at a standard
meter indication (say PPM 4). It is then absolutely clear that if the output level of the tape machine is 0 dBu then the magnetic level on tape is
320 nWb m⫺1. After this relationship has been set up it is then possible to
record a signal on tape at a known magnetic level – for example, a 1 kHz
tone at 0 dBu could be fed to the input of the tape machine, and the input
level adjusted until the output read 0 dBu also. The 1 kHz tone would then
be recording at a flux level of 320 nWb m⫺1.
Test tapes also contain tones at other frequencies for such purposes as
azimuth alignment of heads and for frequency response calibration of replay
EQ (see below). A test tape with the required magnetic reference level should
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be used, and it should also conform to the correct EQ standard (NAB or
CCIR, see ‘Equalization’, above). Tapes are available at all speeds, standards
and widths, with most being recorded across the full width of the tape.
TAPE MACHINE ALIGNMENT
Head inspection and demagnetization
Heads and tape guides must be periodically inspected for wear. Flats on
guides and head surfaces should be looked for; sometimes it is possible to
rotate a guide so that a fresh portion contacts the tape. Badly worn guides
and heads cause sharp angles to contact the tape which can damage the
oxide layer. Heads have been made of several materials. Mu-metal heads
have good electromagnetic properties, but are not particularly hard wearing.
Ferrite heads wear extremely slowly and their gaps can be machined to tight
tolerances. The gap edges can, however, be rather brittle and require careful
handling. Permalloy heads last a long time and give a good overall performance, and are often chosen. Head wear is revealed by the presence of a flat
area on the surface which contacts the tape. Slight wear does not necessarily
indicate that head replacement is required, and if performance is found to be
satisfactory during alignment with a test tape then no action need be taken.
Replay-head wear is often signified by exceptionally good high-frequency
response, requiring replay EQ to be reduced to the lower limit of its range. This
seems odd but is because the replay gap on many designs gets slightly narrower as the head wears down, and is at its narrowest just before it collapses!
Heads should be cleaned regularly using isopropyl alcohol and a cotton
bud. They should also be demagnetized fairly regularly, since heads can gradually become slightly permanently magnetized, especially on older machines,
resulting in increased noise and a type of ‘bubbling’ modulation noise in the
background on recordings. A demagnetizer is a strong AC electromagnet
which should be switched on well away from the tape machine, keeping it
clear of anything else magnetic or metal. This device will erase a tape if placed
near one! Once turned on the demagger should be drawn smoothly and slowly
along the tape path (without a tape present), across the guides and heads, and
drawn away gently on the far side. Only then should it be turned off.
Replay alignment
Replay alignment should be carried out before record alignment, as
explained above. The method for setting replay and record levels has
already been covered in the previous section. HF tones for azimuth
adjustment normally follow (see Fact File 6.5). The test tape will contain
a sequence of tones for replay frequency response alignment, often at 10 or
Tape Machine Alignment
FA C T F I L E 6 . 5 B IA S A D JU S TMENT
Bias level affects the performance of the recording process
and the correct level of bias is a compromise between output level, distortion, noise level and other factors. The graph
below shows a typical tape’s performance with increasing
bias, and it can be seen that output level increases up to a
point, after which it falls off. Distortion and noise go down
as bias increases, but unfortunately the point of minimum
noise and distortion is not quite the same as the point of
maximum output level. Typically the optimum compromise
between all the factors, offering the best dynamic range, is
where the bias level is set just slightly higher than the point
giving peak output. In order to set bias, a 10 kHz tone is
recorded at, say, 10 dB below reference level, whilst bias
is gradually increased from the minimum. The output level
from the tape machine gradually rises to a peak and then
begins to drop off as bias continues to increase. Optimum
bias is set for a number of decibels of fall-off in level after
this peak – the so-called ‘overbias’ amount.
The optimum bias point depends on tape speed
and formulation, but is typically around 3 dB of overbias
at a speed of 15 ips (38 cm s⫺1). At 7.5 ips the overbias
increases to 6 dB and at 30 ips it is only around 1.5 dB.
If bias is adjusted at 1 kHz there is much less change of
output level with variation in bias, and thus only between
0.5 and 0.75 dB of overbias is required at 15 ips. This is
difficult to read on most meters.
Peak output
10 kHz
output
level
Overbias
Distortion
Bias level
20 dB below reference level so that tape saturation is avoided at frequency
extremes, starting with a 1 kHz reference followed by, say, 31.5 Hz, 63 Hz,
125 Hz, 250 Hz, 500 Hz, 2 kHz, 4 kHz, 8 kHz and 16 kHz. Spoken identification of each section is provided. As the tape runs, the replay equalization
is adjusted so as to achieve the flattest frequency response. Often both LF
and HF replay adjustment is provided, sometimes just HF, but normally
one should only adjust HF response on replay, since LF can suffer from the
head bumps described in Fact File 6.2 and a peak or dip of response may
coincide with a frequency on the test tape, leading to potential misalignment. Also full-track test tapes can cause ‘fringing’ at LF, whereby flux from
the guard band leaks on to adjacent tracks. (Although it seems strange,
replay LF EQ is normally adjusted during recording, to obtain the flattest
record–replay response.)
Record alignment
The frequency response of the machine during recording is considerably
affected by bias adjustment, and therefore bias is aligned first before record
equalization. The effects and alignment of bias are described in Fact File 6.5.
Optimum
bias
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FA C T F I L E 6 . 6 A ZIMU TH A L IG NM ENT
Azimuth
Height
Azimuth describes the orientation of the head gap with
respect to the tape. The gap should be exactly perpendicular to the edge of the tape otherwise two consequences
follow. First, high frequencies are not efficiently recorded
or replayed because the head gap becomes effectively
wider as far as the tape is concerned, as shown in the
diagram (B is wider than A). Second, the relative phase
between tracks is changed.
The high-frequency tone on a test tape (8, 10, or
16 kHz) can be used with the outputs of both channels
combined, adjusting replay azimuth so as to give maximum output level which indicates that both channels are
in phase. Alternatively, the two channels can be displayed
separately on a double-beam oscilloscope, one wave being
positioned above the other on the screen, where it can easily be seen if phase errors are present. Azimuth is adjusted
until the two sine waves are in step. It is advisable to begin
with a lower-frequency tone than 8 kHz if a large azimuth
error is suspected, since there is a danger of ending up
with tracks a multiple of 360° out of phase otherwise.
In multitrack machines a process of trial and error is
required to find a pair of tracks which most closely represents the best phase alignment between all the tracks.
Head manufacturing tolerances result in gaps which
are not perfectly aligned on all tracks. Cheap multitrack
machines display rather wider phase errors between
various tracks than do expensive ones.
Azimuth of the replay head is normally adjusted
regularly, especially when replaying tapes made on other
machines which may have been recorded with a different
azimuth. Record-head azimuth is not modified unless
there is reason to believe that it may have changed.
Absolute height of the head should be such that the
center of the face of the head corresponds with the
center of the tape. Height can be adjusted using a
test tape that is not recorded across the full width of the
tape but with two discrete tracks. The correct height
gives both equal output level from both channels and
minimum crosstalk between them. It is also possible to
buy tapes which are only recorded in the guard band,
allowing the user to adjust height for minimum breakthrough onto the audio tracks. It can also sometimes be
adjusted visually.
Azimuth
Correct
Zenith
Zenith is the vertical orientation of the head with respect
to the surface of the tape. The head should neither lean
forwards towards the tape, nor lean backwards, otherwise
uneven wrap of the tape across the surface of the head
results causing inconsistent tape-to-head contact and
uneven head wear. Zenith is not normally adjusted unless
the head has been changed or there is reason to believe
that the zenith has changed.
Wrap
Wrap is the centrality of the head gap in the area of tape
in contact with the head. The gap should be exactly
in the center of that portion, so that the degree of
approach and recede contact of the tape with respect to
the gap is exactly equal. Uneven frequency response can
be caused if this is not the case. Wrap can be adjusted
by painting the head surface with a removable dye and
running the tape across it. The tape will remove the dye
over the contact area, and adjustments can be made
accordingly.
Height
Wrap
Zenith
Front view
Top view
Side view
Incorrect
A
B
Front view
Tape Machine Alignment
It is wise to set a roughly correct input level before adjusting bias, by sending a 1 kHz tone at reference level to the tape machine and adjusting the
input gain until it replays at the same level.
After bias levels have been set, record azimuth can be adjusted if
necessary (see Fact File 6.5) by recording an HF tone and monitoring the
now correctly aligned replay output. It may also be necessary to go back
and check the 1 kHz record level if large changes have been made to bias.
Record equalization can now be aligned. Normally only HF EQ is available on record. A 1 kHz tone is recorded at between 10 and 20 dB below
reference level and the meter gain adjusted so that this can be seen easily
on replay. Spot frequencies are then recorded to check the machine’s frequency response, normally only at the extremes of the range. A 5 kHz tone,
followed by tones at 10 kHz and 15 kHz can be recorded and monitored
off tape. The HF EQ is adjusted for the flattest possible response. The LF
replay EQ (see above) can similarly be adjusted, sweeping the oscillator over
a range of frequencies from, say, 40 Hz to 150 Hz, and adjusting for the best
compromise between the upper and lower limits of the ‘head bumps’.
Some machines have a built-in computer which will automatically align
it to any tape. The tape is loaded and the command given, and the machine
itself runs the tape adjusting bias, level and EQ as it goes. This takes literally seconds. Several settings can be stored in its memory so that a change of
tape type can be accompanied simply by telling the machine which type is to
be used, and it will automatically set its bias and EQ to the previously stored
values. This is of particular value when aligning multitrack machines!
Once the tape machine has been correctly aligned for record and replay,
a series of tones should be recorded at the beginning of every tape made
on the machine. This allows the replay response of any machine which
might subsequently be used for replaying the tape to be adjusted so as to
replay the tape with a flat frequency response. The minimum requirement
should be a tone at 1 kHz at reference level, followed by tones at HF and
LF (say 10 kHz and 63 Hz) at either reference level (if the tape can cope) or
at ⫺10 dB. The levels and frequencies of these tones must be marked on
the tape box (e.g. ‘Tones @ 1 kHz, 320 nWb m⫺1 (⫽0 dB); 10 kHz and 63 Hz
@ ⫺10 dB). Designations on the tape box such as ‘1 kHz @ 0 VU’ mean
almost nothing, since 0 VU is not a magnetic level. What the engineer
means in this case is that he/she sent a tone from his/her desk to the tape
machine, measuring 0 VU on the meters, but this gives no indication of the
magnetic level that resulted on the tape. Noted on the box should also be
an indication of where peak recording level lies in relation to the 1 kHz reference level (e.g. ‘peak recording level @ 8 dB above 320 nWb m⫺1), in order
that the replay chain can be set up to accommodate the likely signal peaks.
In broadcasting, for example, it is most important to know where the peak
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signal level will be, since this must be set to peak at PPM 6 on a program
meter, corresponding to maximum transmitter modulation.
When this tape comes to be replayed, the engineer will adjust the replay
level and EQ controls of the relevant machine, along with replay azimuth,
to ensure that the recorded magnetic reference level replays at his or her
studio’s electrical reference level, and to ensure a flat response. This is the
only way of ensuring that a tape made on one machine replays correctly on
another day or on another machine.
MECHANICAL TRANSPORT FUNCTIONS
Properly, mechanical alignment of the tape transport should be looked at
before electrical alignment, because the electromagnetic performance is
affected by it, but the converse is not the case. Mechanical alignment should
be required far less frequently than electrical adjustments, and sometimes it
also requires rather specialized tools. Because most mechanical alignments
are fairly specialized, and because they differ with each tape machine, detailed
techniques will not be covered further here. The manual for a machine normally details the necessary procedures. Looking at the diagram in Figure 6.9,
it can be seen that the tape unwinds from the reel on the left, passes through
various guides on its way to the head block, and then through various further guides and onto the take-up reel on the right. Some tape guides may be
loaded with floppy springs which give on the instant of start-up, then slowly
swing back in order to control the tension of the tape as the machine starts.
The capstan is the shaft of a motor which pokes up through the deck of the
machine by a couple of centimeters or so
(more of course for multitrack machines
with their increased tape widths) and lies
fairly close to the tape when the tape is
at rest, on the right-hand side of the head
block. A large rubber wheel will be located
close to the capstan but on the opposite side
Supply reel
Take-up reel
of the tape. This is called the pinch roller
or pinch wheel. The capstan motor rotates
at a constant and carefully controlled speed,
Guide
and its speed of rotation defines the speed
Capstan
at which the tape runs. When record or play
Head block
is selected the pinch roller rapidly moves
towards the capstan, firmly sandwiching
Pinch roller
the tape in between the two. The rotation
FIGURE 6.9 Typical layout of mechanical components on the of the capstan now controls the speed of
deckplate of an analog open-reel recorder.
tape travel across the heads.
The Compact Cassette
The take-up reel is controlled by a motor which applies a low anticlockwise torque so that the tape is wound on to it. The supply reel on
the left is also controlled by a motor, which now applies a low clockwise
torque, attempting to drag the tape back in the opposite direction, and this
‘back tension’ keeps the tape in firm contact with the heads. Different reel
sizes require different degrees of back tension for optimum spooling, and a
reel size switch will usually be provided although this is sometimes automatic. One or two transports have been designed without pinch rollers,
an enlarged diameter capstan on its own providing speed control. The reel
motors need to be rather more finely controlled during record and replay
so as to avoid tape slippage across the capstan. Even capstanless transports
have appeared, the tape speed being governed entirely by the reel motors.
When fast wind or rewind is selected the tape is lifted away from the
heads by tape lifters, whilst spooling motors apply an appropriately high
torque to the reel which is to take up the tape and a low reverse torque to
the supply reel to control back tension. The tape is kept away from the
heads so that its rapid movement does not cause excessive heating and
wear of the tape heads. Also, very high-level, high-frequency energy is
induced into the playback head if the tape is in contact with it which can
easily damage speakers, particularly tweeters and HF horns. Nevertheless, a
facility for moving the tape into contact with the heads during fast spooling
is provided so that a particular point in the tape can be listened for.
Motion sensing and logic control is an important feature of a modern
open-reel machine. Because the transport controls are electronically governed on modern machines, one can go straight from, say, rewind to play,
leaving the machine itself to store the command and bring the tape safely
to a halt before allowing the pinch wheel to approach the capstan. Motion
sensing can be implemented by a number of means, often either by sensing
the speed of the reel motors using tachometers, or by counting pulses from
a roller guide.
The tape counter is usually driven by a rotating roller between the head
block and that reel. Slight slippage can be expected, this being cumulative
over a complete reel of tape, but remarkably accurate real-time counters are
nevertheless to be found.
THE COMPACT CASSETTE
Background
The Compact Cassette was invented by Philips, and was launched in 1963.
It was originally intended as a convenient low-quality format suitable for
office dictation machines and the like. It was envisaged that domestic tape
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recording would be open-reel, and a boom in this area was predicted. Prerecorded open-reel tapes were launched. The expected boom never really
materialized, however, and the sheer convenience of the cassette medium
meant that it began to make inroads into the domestic environment. The
format consists of tape one-eighth of an inch wide (3 mm), quarter track,
running at a speed of 1.875 ips. Such drastically reduced dimensions and
running speed compared with open-reel returned a poor level of audio performance, and if it was to be used for reasonable-quality music reproduction considerable development was needed.
Leaving the standards which had been set for the Compact Cassette
unaltered, tape and machine manufacturers worked hard to develop the format, and the level of performance now available from this medium is quite
impressive given its humble beginnings.
Alignments
Cassette test tapes are available, enabling frequency response and azimuth
checks. Thorough cleaning and demagnetization of the machine should be
carried out before one is used. Small azimuth adjustments can bring particularly worthwhile improvements in cassette performance, especially
when replaying tapes recorded on another machine. Azimuth can simply be
adjusted for maximum subjective HF response using the little spring-loaded
screw on one side of the record/replay head. Some machines incorporate
computer systems similar to those found in certain professional open-reel
models which automatically align the machine for a particular type of tape.
Settings for several types can be stored in the computer’s memory.
Automatic replay azimuth adjustment is also possible. The two channels of the stereo output are filtered, converted into square waves and then
fed to a comparator. Phase differences produce an output control voltage
and this drives a small low-speed motor which adjusts the azimuth setting
of the replay head. When azimuth is correct no control voltage is produced
and the azimuth is left alone. The system is continuously active throughout the replay process and it is designed to extract the best performance
from pre-recorded musicassettes and recordings made on other machines.
Multitrack cassette recorders
In the late 1970s the Japanese TEAC company introduced a machine called
the Portastudio. It was a four-channel multitrack cassette recorder with
mixing facilities and multiple inputs built in. The tape ran at twice normal
speed, 3.75 ips, and the four tracks were recorded across the full width of
the tape. Each track could be recorded on separately, sync facilities were
Recommended Further Reading
provided, and ‘bounce down’ could be achieved in the manner of a professional multitrack machine whereby signals recorded on, say, tracks 1, 2
and 3 could be mixed and recorded onto the fourth track, freeing the other
three tracks for further use. The final four-track tape could then be mixed
down into stereo, these stereo outputs being fed to a conventional cassette
recorder (or even open-reel).
One mixer company even offered an eight-track cassette-based system
which incorporated a mixing section that offered facilities such as multiband EQ and auxiliary sends.
RECOMMENDED FURTHER READING
Jorgensen, F., 1995. The Complete Handbook of Magnetic Recording, fourth ed.
McGraw-Hill.
See also ‘General further reading’ at the end of this book.
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CHAPTER 7
Noise Reduction
CH A P T E R C O N TE N T S
Why is Noise Reduction Required?
Methods of Reducing Noise
Variable pre-emphasis
Dolby noise reduction systems
dbx
Telcom c4
Line-up of Noise Reduction Systems
Single-ended Noise Reduction
General systems
Noise gates
Digital noise extraction
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Much of the material contained in this chapter could be regarded as historical given the widespread adoption of digital recording and processing, but
the justification given at the beginning of the last chapter for its continued
inclusion requires that its partnering chapter, also in reduced form, should
hold a place in this edition. Noise reduction techniques have been applied
to analog tape machines of all formats, radio microphones, radio transmission and reception, land lines, satellite relays, gramophone records, and
even some digital tape machines. The general principles of operation will
be outlined, followed by a discussion of particular well-known examples.
Detailed descriptions of some individual systems are referred to in the
‘Further reading’ list at the end of this chapter.
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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FIGURE 7.1
Graphical representation
of a companding noise
reduction process.
Original
signal
Dynamic range
compressed
Noise added
Dynamic range
expanded
Noise
reduced
Programme
restored to
original
level
In
Noise reduction
encoder
Out
Recording or
transmission
process
Noise reduction
decoder
WHY IS NOISE REDUCTION REQUIRED?
A noise reduction system, used correctly, reduces the level of unwanted signals introduced in a recording–replay or transmission–reception process (see
Figure 7.1). Noise such as hiss, hum, and interference may be introduced,
as well as, say, print-through in analog recording, due to imperfections in
the storage or transmission process. In communications, a signal sent along
a land line may be prone to interference from various sources, and will
therefore emerge with some of this interference signal mixed with it.
High-quality radio microphone systems routinely incorporate noise
reduction in the form of variable ratio compression and expansion, and this
can make previously unacceptably noisy reception conditions (weak signal
strength coming into the receiver requiring high RF gain with its attendant
high noise level) usable.
METHODS OF REDUCING NOISE
Variable pre-emphasis
Pre-emphasis (see Fact File 7.1) is a very straightforward solution to the problem of noise reduction, but is not a panacea. Many sound sources, including music, have a falling energy content at high frequencies, so lower-level
HF signals can be boosted to an extent without too much risk of saturating the tape. But tape tends to saturate more easily at HF than at LF (see
the previous chapter), so high levels of distortion and compression would
result if too much pre-emphasis were applied at the recording stage. What
is needed is a circuit which senses the level of the signal on a continuous basis, controlling the degree of pre-emphasis so as to be non-existent
at high signal levels but considerable at low signal levels (see Figure 7.2).
Methods of Reducing Noise
FA C T F I L E 7 . 1 P R E -E MP H A S IS
One approach to the problem of reducing the apparent
level of noise could be to precondition the incoming signal
in some way so as to raise it further above the noise. Hiss
is most annoying at high frequencies, so one could boost
HF on recording. On replay, HF signals would therefore be
reproduced with unnatural emphasis, but if the same region
is now attenuated to bring the signal down to its original level
any hiss in the same band will also be attenuated by a corresponding amount, and so a degree of noise reduction can
be achieved without affecting the overall frequency balance
of the signal. This is known as pre-emphasis (on record) and
de-emphasis (on replay), as shown in the diagram.
Pre-emphasis
De-emphasis
100
Variable
filter
Variable
filter
Subtract
Add
Output
Input
Output level (dB)
1 kHz
0
–20
–40
50
100
1 kHz
Frequency Hz
0 dB
Original
signal level
–20 dB
–40 dB
10 kHz
10
kHz
FIGURE 7.2
A simple complementary
noise reduction system
could boost high
frequencies at low signal
levels during encoding,
and cut them on decoding
(encoding characteristic
shown).
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This can be achieved by incorporating a filter into a side-chain which passes
only high-frequency, low-level signals, adding this component into the unpreemphasized signal. On replay, a reciprocal de-emphasis circuit could then be
used. The lack of noise reduction at high signal levels does not matter, since
high-level signals have a masking effect on low-level noise (see Fact File 2.3).
Such a process may be called a compansion process, in other words a
process which compresses the dynamic range of a signal during recording and expands it on replay. The variable HF emphasis described above is
an example of selective compansion, acting only on a certain band of frequencies. It is most important to notice that the decoding stage is an exact
mirror image of the encoding process, and that it is not possible to use one
without the other. Recordings not encoded by a noise reduction system
cannot simply be passed through a decoder to reduce their noise. Similarly,
encoded tapes sound unusual unless properly decoded, normally sounding
overbright and with fluctuations in HF level.
Dolby noise reduction systems
The above process is used as the basis for the Dolby B noise reduction system, found in most cassette decks. Specifically, the threshold below which
noise reduction comes into play is around 20 dB below a standard magnetic
reference level known as ‘Dolby level’ (200 nWb m⫺1). The maximum HF
boost of the Dolby B system is 10 dB above 8 kHz, and therefore a maximum of 10 dB of noise reduction is provided. A high-quality cassette deck,
without noise reduction, using a good ferric tape, will yield a signal-tonoise ratio of about 50 dB ref. Dolby level. When Dolby B noise reduction
is switched in, the 10 dB improvement brings this up to 60 dB (which is
more adequate for good-quality music and speech recording). The quoted
improvement is seen when noise is measured according to the CCIR 468-2
weighting curve (see ‘Dynamic range and signal-to-noise ratio’, Appendix 1)
and will not be so great when measured unweighted.
Dolby B became widely incorporated into cassette players in the early
1970s, but by the end of the 1970s competition from other companies
offering greater levels of noise reduction prompted Dolby to introduce
Dolby C, which gives 20 dB of noise reduction. The system acts down to
a lower frequency than Dolby B (100 Hz), and incorporates additional circuitry (known as ‘anti-saturation’) which reduces HF tape squashing when
high levels of signal are present. Most of the noise reduction action takes
place between 1 kHz and 10 kHz, and less action is taken on frequencies
above 10 kHz (where noise is less noticeable) in order to desensitize the
system to HF response errors from such factors as azimuth misalignment
Methods of Reducing Noise
which would otherwise be exaggerated (this is known as ‘spectral skewing’). Dolby C, with its greater compression/expansion ratio compared with
Dolby B, will exaggerate tape machine response errors to a correspondingly
greater degree, and undecoded Dolby C tapes will sound extremely bright.
Dolby A was introduced in 1965, and is a professional noise reduction
system. In essence there is a similarity to the processes described above,
but in the Dolby A encoder the noise reduction process is divided into four
separate frequency bands, as shown in Figure 7.3. A low-level ‘differential’
component is produced for each band, and the differential side-chain output is then recombined with the main signal. The differential component’s
contribution to the total signal depends on the input level, having maximum effect below ⫺40 dB ref. Dolby level (see Figures 7.4(a) and (b)).
The band splitting means that each band acts independently, such that
a high-level signal in one band does not cause a lessening of noise reduction effort in another low-level band, thus maintaining maximum effectiveness with a wide range of program material. The two upper bands are high
pass and overlap, offering noise reduction of 10 dB up to around 5 kHz,
rising to 15 dB at the upper end of the spectrum.
Filters
9 kHz
HP
Compressors
3 kHz
HP
Compressors
+
Input
80 Hz–
3 kHz
Compressors
80 Hz
LP
Compressors
+
Output
To rec
FIGURE 7.3 In the Dolby A system a low-level ‘differential’ signal is added to the main signal
during encoding. This differential signal is produced in a side-chain which operates independently on
four frequency bands. The differential signal is later subtracted during decoding.
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CHAPTER 7: Noise Reduction
(a)
(b)
Input (dB)
Input (dB)
0 (Dolby level)
–40
0
–40
Encoding
characteristic
Output (dB)
Output (dB)
FIGURE 7.4
(a) Differential signal
component produced in
a Dolby A side-chain. (b)
Input level plotted against
output level of Dolby A unit
after adding or subtracting
differential component.
–40
–40
Decoding
characteristic
The decoder is the mirror image of the encoder, except that the differential signal produced by the side-chain is now subtracted from the main
signal, restoring the signal to its original state and reducing the noise introduced between encoding and decoding.
The late 1980s saw the introduction of Dolby SR – Spectral Recording –
which gives greater noise reduction of around 25 dB. It has been successful
in helping to prolong the useful life of analog tape machines, both stereo
mastering and multitrack, in the face of the coming of digital tape recorders. Dolby SR differs from Dolby A in that whereas the latter leaves the
signal alone until it drops below a certain threshold, the former seeks to
maintain full noise reduction (i.e. maximum signal boost during recording) across the whole frequency spectrum until the incoming signal rises
above the threshold level. The band of frequencies where this happens
is then subject to appropriately less boost. This is rather like looking at
the same process from opposite directions, but the SR system attempts to
place a comparably high recording level on the tape across the whole frequency spectrum in order that the dynamic range of the tape is always used
optimally.
This is achieved by ten fixed and sliding-band filters with gentle slopes.
The fixed-band filters can vary in gain. The sliding-band filters can be
adjusted to cover different frequency ranges. It is therefore a fairly complex
multiband system, requiring analysis of the incoming signal to determine
its energy at various frequencies. Spectral skewing and anti-saturation are
also incorporated (see ‘Dolby C ’, above). Dolby SR is a particularly inaudible
noise reduction system, more tolerant of level mismatches and replay speed
changes than previous systems. A simplified ‘S’-type version was introduced for the cassette medium, and is also used on some semi-professional
multitrack recorders.
Methods of Reducing Noise
The Dolby process, being level dependent, requires that the reproduced
signal level on decoding is exactly the same with respect to Dolby level as
on encoding, otherwise frequency response errors will result: for instance,
a given level of treble boost applied during encoding must be cut by exactly
the same amount during replay.
dbx
dbx is another commonly encountered system. It offers around 30 dB of
noise reduction and differs from the various Dolby systems as follows. dbx
globally compresses the incoming signal across the whole of the frequency
spectrum, and in addition gives pre-emphasis at high frequencies (treble
boost). It is not level dependent, and seeks to compress an incoming signal with, say, a 90 dB dynamic range into one with a 60 dB dynamic range
which will now fit into the dynamic range capabilities of the analog tape
recorder. On replay, a reciprocal amount of expansion is applied together
with treble de-emphasis.
Owing to the two factors of high compansion ratios and treble pre- and
de-emphasis, frequency response errors can be considerably exaggerated.
Therefore, dbx type 1 is offered which may be used with professional equipment and type 2 is to be used with domestic equipment such as cassette
decks where the noise reduction at high frequencies is relaxed somewhat
so as not to exaggerate response errors unduly. The degree of compression/
expansion is fixed, that is it does not depend on the level of the incoming signal. There is also no division of noise reduction between frequency
bands. These factors sometimes produce audible modulation of background
hiss with critical program material such as wide dynamic range classical
music, and audible ‘pumping’ noises can sometimes be heard. The system does, however, offer impressive levels of noise reduction, particularly
welcome with the cassette medium, and does not require accurate level
alignment.
Telcom c4
The ANT telcom c4 noise reduction system arrived somewhat later than
did Dolby and dbx, in 1978. Capitalizing on the experience gained by those
two systems, the telcom c4 offers a maximum noise reduction of around
30 dB, is level dependent like Dolby, and also splits the frequency spectrum
up into four bands which are then treated separately. The makers claim
that the c4 system is less affected by record/replay-level errors than is Dolby
A. The system works well in operation, and side-effects are minimal.
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There is another system offered by the company, called ‘hi-com’, which
is a cheaper, simpler version intended for home studio setups and domestic
cassette decks.
LINE-UP OF NOISE REDUCTION SYSTEMS
In order to ensure unity gain through the system on recording and replay,
with correct tracking of a Dolby decoder, it is important to align the noise
reduction signal chain. Many methods are recommended, some more rigorous than others, but in a normal studio operation for everyday alignment,
the following process should be satisfactory. It should be done after the tape
machine has been aligned (this having been done with the NR unit bypassed).
For a Dolby A encoder, a 1 kHz tone should be generated from the mixer
at ⫹4 dBu (usually PPM 5), and fed to the input of the NR unit. The unit
should be in ‘NR out’ mode, and set to ‘record’. The input level of the NR
unit should normally be adjusted so that this tone reads on the ‘NAB’ level
mark on the meter (see Figure 7.5). The output of the unit should then be
adjusted until its electrical level is also ⫹4 dBu. (If the tape machine has
meters then the level can be read here, provided that these meters are reliable and the line-up is known.)
It is customary to record a passage of ‘Dolby tone’ (in the case of Dolby A)
or Dolby Noise (in the case of Dolby SR) at the beginning of a Dolby-encoded
tape, along with the other line-up tones (see ‘Record alignment’, Chapter 6).
During record line-up, the Dolby tone is generated by the Dolby unit itself,
and consists of a frequency-modulated 700 Hz tone at the Dolby’s internal
line-up reference level, which is easily recognized and distinguished from
other line-up tones which may be present on a tape. Once the output level of
the record Dolby has been set then the Dolby tone button on the relevant unit
should be pressed, and the tone recorded at the start of the tape.
To align the replay Dolby (set to ‘NR out’,
‘replay ’ mode), the recorded Dolby tone should
Red
be replayed and the input level adjusted so that
Green
the tone reads at the NAB mark on the internal
NAB
DIN
meter. The output level should then be adjusted for
Green
⫹4 dBu, or so that the mixer’s meter reads PPM 5
Red
when switched to monitor the tape machine replay.
FIGURE 7.5 Dolby level is indicated on Dolby units using
For operation, the record and replay units
either a mechanical meter (shown left), or using red and green should be switched to ‘NR in’.
LEDs (shown right). The meter is normally aligned to the ‘18.5
Dolby SR uses pink noise instead of Dolby tone
NAB’ mark or set such that the two green LEDs are on together.
to distinguish tapes recorded with this system, and
Single-Ended Noise Reduction
it is useful because it allows for line-up of the replay Dolby in cases where
accurate level metering is not available. Since level misalignment will result
in response errors the effects will be audible on a band of pink noise. A facility is provided for automatic switching between internally generated pink
noise and off-tape noise, allowing the user to adjust replay-level alignment
until there appears to be no audible difference between the spectra of the
two. In normal circumstances Dolby SR systems should be aligned in a similar way to Dolby A, except that a noise band is recorded on the tape instead
of a tone. Most systems use LED meters to indicate the correct level, having
four LEDs as shown in Figure 7.5.
SINGLE-ENDED NOISE REDUCTION
General systems
Several companies offer so-called ‘single-ended’ noise reduction systems,
and these are intended to ‘clean up’ an existing noisy recording or signal.
They operate by sensing the level of the incoming signal, and as the level
falls below a certain threshold the circuit begins to roll off the treble progressively, thereby reducing the level of hiss. The wanted signal, being low in
level, in theory suffers less from this treble reduction than would a high-level
signal due to the change in response of the ear with level (see Fact File 2.2).
High-level signals are left unprocessed. The system is in fact rather similar
to the Dolby B decoding process, but of course the proper reciprocal Dolby B
encoding is absent. The input level controls of such systems must be carefully adjusted so as to bring in the effect of the treble roll-off at the appropriate threshold for the particular signal being processed so that a suitable
compromise can be achieved between degree of hiss reduction and degree
of treble loss during quieter passages. Such single-ended systems should be
judiciously used – they are not intended to be left permanently in circuit
– and value judgments must always be made as to whether the processed
signal is in fact an improvement over the unprocessed one.
If a single-ended system is to be used on a stereo program, units which
are capable of being electronically ‘ganged’ must be employed so that exactly
the same degree of treble cut is applied to each channel; otherwise varying
frequency balance between channels will cause stereo images to wander.
Noise gates
The noise gate can be looked upon as another single-ended noise reduction
system. It operates as follows. A threshold control is provided which can
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be adjusted such that the output of the unit is muted (the gate is ‘closed’)
when the signal level falls below the threshold. During periods when signal level is very low (possibly consisting of tape or guitar amplifier noise
only) or absent the unit shuts down. A very fast attack time is employed so
that the sudden appearance of signal opens up the output without audible
clipping of the initial transient. The time lapse before the gate closes, after
the signal has dropped below the chosen threshold level, can also be varied. The close threshold is engineered to be lower than the open threshold (known as hysteresis) so that a signal level which is on the borderline
does not confuse the unit as to whether it should be open or closed, which
would cause ‘gate flapping’.
Such units are useful when, for instance, a noisy electric guitar setup
is being recorded. During passages when the guitarist is not playing the
output shuts down so that the noise is removed from the mix. They are
sometimes also used in a similar manner during multitrack mixdown
where they mute outputs of the tape machine during the times when the
tape is unmodulated, thus removing the noise contribution from those
tracks.
Noise gates can also be used as effects in themselves, and the ‘gated
snare drum’ is a common effect on pop records. The snare drum is given
a heavy degree of gated reverb, and a high threshold level is set on the gate
so that around half a second or so after the drum is hit the heavy ‘foggy ’
reverb is abruptly cut off. Drum machines can mimic this effect, as can
some effects processors.
Digital noise extraction
Extremely sophisticated single-ended computer-based noise reduction systems have been developed. A given noisy recording will normally have a
short period somewhere in which only the noise is present without any program, for instance the run-in groove of an old 78 rpm shellac disc recording
provides a sample of that record’s characteristic noise. This noise is analyzed by a computer and can subsequently be recognized as an unwanted
constituent of the signal, and then extracted electronically from it. Sudden
discontinuities in the program caused by scratches and the like can be recognized as such and removed. The gap is filled by new material which is
made to be similar to that which exists either side of the gap. Not all of
these processes are currently ‘real time’, and it may take several times longer than the program’s duration for the process to be carried out, but as the
speed of digital signal processing increases more operations become possible in real time.
Recommended Further Reading
RECOMMENDED FURTHER READING
Dolby, R., 1967. An audio noise reduction system. J. Audio Eng. Soc. 15, 383–388.
Dolby, R., 1970. A noise reduction system for consumer tape applications.
Presented at the 39th AES Convention. J. Audio Eng. Soc. (Abstracts) 18, 704.
Dolby, R., 1983. A 20 dB audio noise reduction system for consumer applications.
J. Audio Eng. Soc. 31, 98–113.
Dolby, R., 1986. The spectral recording process. Presented at the 81st AES
Convention. Preprint 2413 (C-6). Audio Engineering Society.
See also ‘General further reading’ at the end of this book.
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CHAPTER 8
Digital Audio Principles
CH A P T E R C O N TE N T S
Digital and Analog Recording Contrasted
Binary for Beginners
The Digital Audio Signal Chain
Analog-to-Digital Conversion
A basic example
Introduction to audio A/D conversion
Audio sampling
Filtering and aliasing
Sampling frequency and sound quality
Quantizing
Quantizing resolution and sound quality
Use of dither
Oversampling in A/D conversion
Noise shaping in A/D conversion
D/A Conversion
A basic D/A convertor
Oversampling in D/A conversion
Direct Stream Digital (DSD)
Changing the Resolution of an Audio Signal (Requantization)
Introduction to Digital Signal Processing
Gain changing (level control)
Mixing
Digital filters and equalization
Digital reverberation and other effects
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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C H A P TE R C O N TENTS
Dynamics processing
Sample rate conversion
Pitch Shifting and Time Stretching
Audio Data Reduction
Why reduce the data rate?
Lossless and lossy coding
MPEG – an example of lossy coding
Other data-reduced formats
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This chapter contains an introduction to the main principles of digital
audio, described in a relatively non-mathematical way. Further reading
recommendations at the end of this chapter are given for those who want
to study the subject in more depth. Subsequent chapters deal with digital
recording and editing systems and with digital audio applications.
DIGITAL AND ANALOG RECORDING CONTRASTED
In analog recording, as described in the previous chapters, sound is
recorded by converting continuous variations in sound pressure into continuous variations in electrical voltage, using a microphone. This varying
voltage is then converted into a varying pattern of magnetization on a tape,
or, alternatively, into a pattern of light and dark areas on an optical-film
soundtrack, or a groove of varying deviation on an LP.
Because the physical characteristics of analog recordings relate closely
to the sound waveform, replaying them is a relatively simple matter.
Variations in the recorded signal can be converted directly into variations
in sound pressure using a suitable collection of transducers and amplifiers. The replay system, however, is unable to tell the difference between
wanted signals and unwanted signals. Unwanted signals might be distortions, noise and other forms of interference introduced by the recording
process. For example, a record player cannot distinguish between the stylus
movement it experiences because of a scratch on a record (unwanted) and
that caused by a loud transient in the music (wanted). Imperfections in the
recording medium are reproduced as clicks, crackles and other noises.
Binary for Beginners
Digital recording, on the other hand, converts the electrical waveform
from a microphone into a series of binary numbers, each of which represents the amplitude of the signal at a unique point in time, recording these
numbers in a coded form which allows the system to detect whether the
replayed signal is correct or not. A reproducing device is then able to distinguish between the wanted and the unwanted signals introduced above,
and is thus able to reject all but the wanted original information in most
cases. Digital audio can be engineered to be more tolerant of a poor recording channel than analog audio. Distortions and imperfections in the storage or transmission process need not affect the sound quality of the signal
provided that they remain within the design limits of the system and that
timing and data errors are corrected. These issues are given further coverage in Fact File 8.1.
Digital audio has made it possible for sound engineers to take advantage of developments in the computer industry, and this is particularly
beneficial because the size of that industry results in mass production (and
therefore cost savings) on a scale not possible for audio products alone.
Today it is common for sound to be recorded, processed and edited on relatively low cost desktop computer equipment, and this is a trend likely to
continue.
BINARY FOR BEGINNERS
First, we introduce the basics of binary number systems, because nearly all
digital audio systems are based on this.
In the decimal number system each digit of a number represents a
power of ten. In a binary system each digit or bit represents a power of two
(see Figure 8.1). It is possible to calculate the decimal equivalent of a binary
integer (whole number) by using the method shown. Negative numbers
need special treatment, as described in Fact File 8.2. A number made up of
more than 1 bit is called a binary ‘word’, and an 8 bit word is called a ‘byte’
(from ‘by eight’). Four bits is called a ‘nibble’. The more bits there are in a
word the larger the number of states it can represent, with 8 bits allowing
256 (28) states and 16 bits allowing 65 536 (216). The bit with the lowest
weight (20) is called the least significant bit or LSB and that with the greatest weight is called the most significant bit or MSB. The term kilobyte or
Kbyte is used to mean 1024 or 210 bytes and the term megabyte or Mbyte
represents 1024 Kbytes.
Electrically it is possible to represent a binary word in either serial or
parallel form. In serial communication only one connection need be used
203
FACT FILE 8.1 ANALOG AND DIGITAL INFORMATION
Analog information is made up of a continuum of values,
which at any instant may have any value between the limits of the system. For example, a rotating knob may have
one of an infinite number of positions – it is therefore
an analog controller (see the diagram below). A simple
switch, on the other hand, can be considered as a digital controller, since it has only two positions – off or on. It
cannot take any value in between. The brightness of light
that we perceive with our eyes is analog information and
as the sun goes down the brightness falls gradually and
smoothly, whereas a household light without a dimmer
may be either on or off – its state is binary (that is it has
only two possible states).
(a)
(a)
+V
Variable
resistor
Voltage
between
0 and +V
depending
on position
Output
0
(b)
+V
Continuously-variable
position
Switch
O/P
Voltage is
0 or +V
depending
on position
(b)
ON
0
OFF
Electrically, analog information may be represented
as a varying voltage or current. If a rotary knob is used
to control a variable resistor connected to a voltage supply, its position will affect the output voltage as shown
below. This, like the knob’s position, may occupy any
value between the limits – in this case anywhere between
zero volts and ⫹V. The switch could be used to control a
similar voltage supply and in this case the output voltage
could only be either zero volts or ⫹V. In other words the
electrical information that resulted would be binary. The
high (⫹V) state could be said to correspond to a binary
one and the low state to binary zero (although in many
real cases it is actually the other way around).
Binary information is inherently more resilient to noise
and interference than analog information, as shown in
the diagram below. If noise is added to an analog signal
it becomes very difficult to tell what is the wanted signal
and what is the unwanted noise, as there is no means
of distinguishing between the two. If noise is added to a
binary signal it is possible to extract the important information at a later stage. By comparing the signal amplitude with a fixed decision point it is possible for a receiver
to treat everything above the decision point as ‘high’ and
everything below it as ‘low’. For any noise or interference
to influence the state of a digital signal it must be at least
large enough in amplitude to cause a high level to be
interpreted as ‘low’, or vice versa.
The timing of digital signals may also be corrected
to some extent, giving digital signals another advantage
over analog ones. This is because digital information has
a discrete time structure in which the intended sample
instants are known. If the timing of bits in a digital message
becomes unstable, such as after having been passed over a
long cable with its associated signal distortions, resulting in
timing ‘jitter’, the signal may be reclocked at a stable rate.
(a)
+
=
+
=
(b)
High
Low
Binary for Beginners
One bit
(a)
A binary
word or ‘byte’
(b)
0 1 1 1 0 0 1 0
0 1 1 1 0 0 1 0
Binary
Decimal weights
128 64
Decimal equivalent
of the binary number
(c)
32
16
8
4
2
1
FIGURE 8.1
(a) A binary number (word or
‘byte’) consists of bits. (b) Each
bit represents a power of two.
(c) Binary numbers can be
represented electrically in pulse
code modulation (PCM) by a
string of high and low voltages.
0 + 64 + 32 + 16 + 0 + 0 + 2 + 0 = 114
High = 1
Low = 0
0
1
1
1
0
0
1
0
and the word is clocked out one bit at a time using a device known as a
shift register. The shift register is previously loaded with the word in parallel form (see Figure 8.2). The rate at which the serial data is transferred
depends on the rate of the clock. In parallel communication each bit of the
word is transferred over a separate connection.
Because binary numbers can become fairly unwieldy when they get long,
various forms of shorthand are used to make them more manageable. The
most common of these is hexadecimal. The hexadecimal system represents
decimal values from 0 to 15 using the 16 symbols 0–9 and A–F, according
to Table 8.1. Each hexadecimal digit corresponds to 4 bits or one nibble
of the binary word. An example showing how a long binary word may be
written in hexadecimal (hex) is shown in Figure 8.3 – it is simply a matter
of breaking the word up into 4 bit chunks and converting each chunk to
hex. Similarly, a hex word can be converted to binary by using the reverse
process.
Logical operations can be carried out on binary numbers, which enables
various forms of mathematics to be done in binary form, as introduced in
Fact File 8.3.
Fixed-point binary numbers are often used in digital audio systems to
represent sample values. These are usually integer values represented by
a number of bytes (2 bytes for 16 bit samples, 3 bytes for 24 bit samples,
etc.). In some applications it is necessary to represent numbers with a very
large range, or in a fractional form. Here floating-point representation may
be used. A typical floating-point binary number might consist of 32 bits,
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CHAPTER 8: Digital Audio Principles
FA C T F I L E 8. 2 N E G ATIVE N U MBERS
Negative integers are usually represented in a form known
as ‘two’s complement’. Negative values are represented
by taking the positive equivalent, inverting all the bits and
adding a one. Thus to obtain the 4 bit binary equivalent
of decimal minus five (⫺510) in binary two’s complement
form:
10
⫺5
510 ⫽ 01012
⫽ 1010 ⫹ 0001 ⫽ 10112
The carry bit that may result from adding the two
MSBs is ignored.
An example is shown here of 4 bit, two’s complement
numbers arranged in a circular fashion. It will be seen
that the binary value changes from all zeros to all ones
as it crosses the zero point and that the maximum positive value is 0111 whilst the maximum negative value is
1000, so the values wrap around from maximum positive
to maximum negative.
0100
Two’s complement numbers have the advantage that the
MSB represents the sign (1 ⫽ negative, 0 ⫽ positive) and
that arithmetic may be performed on positive and negative numbers giving the correct result:
0101
0011
0010
0110
0111
e.g. (in decimal): 5
⫹ (⫺3)
⫽2
or (in binary): 0101
⫹ 1101
⫽ 0010
Positive
values
Max +
0001
0000
0
Max –
1000
1001
Negative
values
–1
1010
1111
1110
1011
1101
1100
Parallel input
0
1
1
1
0
0
1
0
Serial output
Clock (bit rate)
FIGURE 8.2 A shift register is used to convert a parallel binary word into a serial format. The
clock is used to shift the bits one at a time out of the register, and its frequency determines the bit rate.
The data may be clocked out of the register either MSB or LSB first, depending on the device and its
configuration.
arranged as 4 bytes, as shown in Figure 8.4. Three bytes are used to represent the mantissa and 1 byte the exponent (although the choice of number
of bits for the exponent and mantissa are open to variance depending on
the application). The mantissa is the main part of the numerical value and
the exponent determines the power of two to which the mantissa must be
The Digital Audio Signal Chain
Table 8.1
Hexadecimal and decimal equivalents to binary numbers
Binary
Hexadecimal
Decimal
0000
0001
0010
0011
0100
0101
0110
0111
1000
1001
1010
1011
1100
1101
1110
1111
0
1
2
3
4
5
6
7
8
9
A
B
C
D
E
F
0
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
0010111110111110
2
F
B
E
FIGURE 8.3 This 16 bit binary number may be represented in hexadecimal as shown, by breaking
it up into 4 bit nibbles and representing each nibble as a hex digit.
raised. The MSB of the exponent is used to represent its sign and the same
for the mantissa.
It is normally more straightforward to perform arithmetic processing
operations on fixed-point numbers than on floating-point numbers, but signal processing devices are available in both forms.
THE DIGITAL AUDIO SIGNAL CHAIN
Figure 8.5 shows the signal chain involved in a typical digital recording or
broadcasting system. First, the analog audio signal (a time-varying electrical voltage) is passed through an analog-to-digital (A/D) convertor where it
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FA C T F I L E 8 . 3 LO G IC A L O P E R ATI ONS
Most of the apparently complicated processing operations that occur within a computer are actually just a fast
sequence of simple logical operations. The apparent power
of the computer and its ability to perform complex tasks
are really due to the speed with which simple operations
are performed.
The basic family of logical operations is shown here in
the form of a truth table next to the electrical symbol that
A B C
AND
0 0 0
A
C
B
represents each ‘logic gate’. The AND operation gives an
output only when both its inputs are true; the OR operation gives an output when either of its inputs are true; and
the XOR (exclusive OR) gives an output only when one of
its inputs is true. The inverter or NOT gate gives an output
which is the opposite of its input and this is often symbolized using a small circle on inputs or outputs of devices to
indicate inversion.
1 0 0
0 1 0
1 1 1
A B C
OR
0 0 0
A
C
B
Inverter (NOT)
1 0 1
0 1 1
1 1 1
A B C
EXOR
0 0 0
A
B
C
1 0 1
0 1 1
1 1 0
FIGURE 8.4
An example of floating-point
number representation in a
binary system.
1 byte
Exponent
MSB=sign of exp.
3 bytes
Mantissa
Mantissa
Mantissa
MSB=sign of mantissa
is transformed from a continuously varying voltage into a series of ‘samples’,
which are ‘snapshots’ of the analog signal taken many thousand times per
second. Each sample is represented by a number. If the system uses some
form of data reduction (see below) this will be carried out here, after A/D
Analog-to-Digital Conversion
conversion and before channel coding. The resulting sequence of audio data
is coded into a form that makes it suitable for recording or broadcasting (a
process known as coding or channel coding), and the signal is then recorded
or transmitted. Upon replay or reception the signal is decoded and subjected
to error correction, and it is this latter process which works out what damage
has been done to the signal since it was coded. The channel coding and error
detection/correction processes are usually integral to the recording or transmission system and modern disk-based recording systems often rely on the
built-in processes of generic computer mass storage systems to deal with this.
After decoding, any errors in timing or value of the samples are corrected if
possible and the result is fed to a digital-to-analog (D/A) convertor, which
turns the numerical data back into a time-continuous analog audio signal.
In the following sections each of the main processes involved in this
chain will be explained, followed by a discussion of the implementation of
this technology in real audio systems.
ANALOG-TO-DIGITAL CONVERSION
A basic example
In order to convert analog information into digital information it is necessary to measure its amplitude at specific points in time (called ‘sampling’)
and to assign a binary digital value to each measurement (called ‘quantizing’). A simple example of the process can be taken from control technology in which it is wished to convert the position of a rotary knob into a
digital control signal that could be used by a computer. This concept can be
extended to the conversion of audio signals.
The diagram in Figure 8.6 shows such a rotary knob against a fixed
scale running from 0 to 9. The position of the control should be measured
or ‘sampled’ at regular intervals to register changes. The rate at which
switches and analog controls are sampled depends on how important it is
that they are updated regularly. Some older audio mixing consoles sampled
the positions of automated controls once per television frame (40 ms in
Europe), whereas some modern digital mixers sample controls as often as
once per audio sample period (roughly 20 μs). Clearly the more regularly a
control’s position is sampled the more data will be produced, since there
will be one binary value per sample. A smooth representation of changing
control movements is ensured by regular sampling.
To quantize the position of the knob it is necessary to determine which
point of the scale it is nearest at each sampling instant and assign a binary
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CHAPTER 8: Digital Audio Principles
In
A/D
conversion
Channel
coding
Store/
Transmit
Decoding
D/A
conversion
Out
FIGURE 8.5 Block diagram of the typical digital recording or broadcasting signal chain.
number that is equivalent to its position. Unless the pointer is at exactly
one of the increments the quantizing process involves a degree of error. The
maximum error is plus or minus half of an increment, because once the
pointer is more than halfway between one increment and the next it should
be quantized to the next.
Introduction to audio A/D conversion
The process of A/D conversion is of paramount importance in determining
the inherent sound quality of a digital audio signal. The technical quality of
the audio signal, once converted, can never be made any better, only worse.
Some applications deal with audio purely in the digital domain, in which
case A/D conversion is not an issue, but most operations involve the acquisition of audio material from the analog world at one time or another. The
quality of convertors varies very widely in digital audio workstations and
their peripherals because the price range of such workstations is also great.
Some stand-alone professional convertors can easily cost as much as the
complete digital audio hardware and software for a desktop computer. One
can find audio A/D convertors built into many multimedia desktop computers now, but these are often rather low performance devices when compared with the best available. As will be seen below, the sampling rate and
the number of bits per sample are the main determinants of the quality of a
digital audio signal, but the design of the convertors determines how closely
the sound quality approaches the theoretical limits.
Despite the above, it must be admitted that to the undiscerning ear one
16 bit convertor sounds very much like another and that there is a law of
diminishing returns when one compares the increased cost of good convertors with the perceivable improvement in quality. Convertors are very
much like wine in this respect.
Audio sampling
An analog audio signal is a time-continuous electrical waveform and the
A/D convertor’s task is to turn this signal into a time-discrete sequence
Analog-to-Digital Conversion
4
5
3
6
2
7
8
1
0
9
FIGURE 8.6 A rotary knob’s position could be measured against a numbered scale such as the
decimal scale shown. Quantizing the knob’s position would involve deciding which of the limited number
of values (0–9) most closely represented the true position.
of binary numbers. The sampling process employed in an A/D convertor involves the measurement or ‘sampling’ of the amplitude of the audio
waveform at regular intervals in time (see Figure 8.7). From this diagram it
will be clear that the sample pulses represent the instantaneous amplitudes
of the audio signal at each point in time. The samples can be considered
as instantaneous ‘still frames’ of the audio signal which together and in
sequence form a representation of the continuous waveform, rather like the
still frames that make up a movie film give the impression of a continuously moving picture when played in quick succession.
In order to represent the fine detail of the signal it is necessary to take
a large number of these samples per second. The mathematical sampling
theorem proposed by Shannon indicates that at least two samples must be
taken per audio cycle if the necessary information about the signal is to be
conveyed. This means that the sampling frequency must be at least twice
as high as the highest audio frequency to be handled by the system (this is
known as the Nyquist criterion).
Another way of visualizing the sampling process is to consider it in terms
of modulation, as shown in Figure 8.8. The continuous audio waveform is
used to modulate a regular chain of pulses. The frequency of these pulses is
the sampling frequency. Before modulation all these pulses have the same
amplitude (height), but after modulation the amplitude of the pulses is modified according to the instantaneous amplitude of the audio signal at that
point in time. This process is known as pulse amplitude modulation (PAM).
Fact File 8.4 describes a frequency domain view of this process.
Filtering and aliasing
It can be seen from Figure 8.9 that if too few samples are taken per cycle
of the audio signal then the samples may be interpreted as representing a
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CHAPTER 8: Digital Audio Principles
FIGURE 8.7
An arbitrary audio signal
is sampled at regular
intervals of time t to create
short sample pulses whose
amplitudes represent the
instantaneous amplitude
of the audio signal at each
point in time.
Sample pulses
Audio waveform
+
Amplitude
212
Time
0
–
t = sample period
wave other than that originally sampled. This is one way of understanding
the phenomenon known as aliasing. An ‘alias’ is an unwanted representation of the original signal that arises when the sampled signal is reconstructed during D/A conversion.
It is relatively easy to see why the sampling frequency must be at least
twice the highest baseband audio frequency from Figure 8.10. It can be
seen that an extension of the baseband above the Nyquist frequency results
in the lower sideband of the first spectral repetition overlapping the upper
end of the baseband and appearing within the audible range that would be
reconstructed by a D/A convertor. Two further examples are shown to illustrate the point – the first in which a baseband tone has a low enough frequency for the sampled sidebands to lie above the audio frequency range,
and the second in which a much higher frequency tone causes the lower
sampled sideband to fall well within the baseband, forming an alias of the
original tone that would be perceived as an unwanted component in the
reconstructed audio signal.
The aliasing phenomenon can be seen in the case of the well-known
‘spoked-wheel’ effect on films, since moving pictures are also an example
of a sampled signal. In film, still pictures (image samples) are normally
taken at a rate of 24 per second. If a rotating wheel with a marker on it is
filmed it will appear to move round in a forward direction as long as the
rate of rotation is much slower than the rate of the still photographs, but as
its rotation rate increases it will appear to slow down, stop, and then appear
to start moving backwards. The virtual impression of backwards motion
gets faster as the rate of rotation of the wheel gets faster and this backwards motion is the aliased result of sampling at too low a rate. Clearly the
Analog-to-Digital Conversion
Constant-amplitude
pulses (frequency = 1/t)
modulates
Amplitude
Amplitude
Audio
waveform
Time
Time
t = sample period
Amplitude
produces
Pulse-amplitude
modulation
Time
wheel is not really rotating backwards, it just appears to be. Perhaps ideally one would arrange to filter out moving objects that were rotating faster
than half the frame rate of the film, but this is hard to achieve in practice
and visible aliasing does not seem to be as annoying subjectively as audible
aliasing.
If audio signals are allowed to alias in digital recording one hears the
audible equivalent of the backwards-rotating wheel – that is, sound components in the audible spectrum that were not there in the first place, moving
downwards in frequency as the original frequency of the signal increases. In
basic convertors, therefore, it is necessary to filter the baseband audio signal
before the sampling process, as shown in Figure 8.11, so as to remove any
components having a frequency higher than half the sampling frequency. It
is therefore clear that in practice the choice of sampling frequency governs
the high frequency limit of a digital audio system.
In real systems, and because filters are not perfect, the sampling frequency is usually made higher than twice the highest audio frequency to be
FIGURE 8.8
In pulse amplitude
modulation, the
instantaneous amplitude
of the sample pulses is
modulated by the audio
signal amplitude (positive
only values shown).
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CHAPTER 8: Digital Audio Principles
FA C T F I L E 8 . 4 S A MP L IN G – F R EQUENCY DOM AI N
Before modulation the audio signal has a frequency spectrum extending over the normal audio range, known as the
baseband spectrum (upper diagram). The shape of the
waveform and its equivalent spectrum is not significant in
this diagram – it is just an artist’s impression of a complex audio signal such as music. The sampling pulses,
before modulation, have a line spectrum at multiples of
the sampling frequency, which is much higher than the
highest audio frequency (middle diagram). The frequency
spectrum of the pulse-amplitude-modulated (PAM) signal
is as shown in the lower diagram. In addition to the ‘baseband’ audio signal (the original audio spectrum before
sampling) there are now a number of additional images of
this spectrum, each centered on multiples of the sampling
frequency. Sidebands have been produced either side of
the sampling frequency and its multiples, as a result of the
amplitude modulation, and these extend above and below
the sampling frequency and its multiples to the extent of
the base bandwidth. In other words these sidebands are
pairs of mirror images of the audio baseband.
Frequency domain
Time domain
Amplitude
Amplitude
Audio
waveform
Time
Frequency
Amplitude
Amplitude
Constant amplitude
pulses
Time
fs
2fs
3fs
Pulse amplitude
modulation
Time
Amplitude
Nyquist frequency
Amplitude
214
0.5fs
fs
2fs
3fs
Analog-to-Digital Conversion
FIGURE 8.9
In example (a) many
samples are taken per cycle
of the wave. In example
(b) less than two samples
are taken per cycle, making
it possible for another
lower-frequency wave to
be reconstructed from the
samples. This is one way
of viewing the problem of
aliasing.
(a)
(b)
(a)
fn
Alias
region
fn
fs
fs
2fs
Sampled
spectrum
Audio
baseband
(c)
fs
1 kHz
(b)
fs
29
(d)
2fs
31
59
61
Alias
Orig.
13 17
43 47
FIGURE 8.10 Aliasing viewed in the frequency domain. In (a) the audio baseband extends up
to half the sampling frequency (the Nyquist frequency fn) and no aliasing occurs. In (b) the audio
baseband extends above the Nyquist frequency and consequently overlaps the lower sideband of the
first spectral repetition, giving rise to aliased components in the shaded region. In (c) a tone at 1 kHz
is sampled at a sampling frequency of 30 kHz, creating sidebands at 29 and 31 kHz (and at 59 and
61 kHz, etc.). These are well above the normal audio frequency range, and will not be audible. In (d) a
tone at 17 kHz is sampled at 30 kHz, putting the first lower sideband at 13 kHz – well within the normal
audio range. The 13 kHz sideband is said to be an alias of the original wave.
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CHAPTER 8: Digital Audio Principles
FIGURE 8.11
In simple A/D convertors
an analog anti-aliasing filter
is used prior to conversion,
which removes input
signals with a frequency
above the Nyquist limit.
Analog
input
Anti-aliasing
filter
A/D
convertor
To coder
Filter response
Output
level
Audio band
Nyquist
limit
Sampling
frequency
f
represented, allowing for the filter to roll off more gently. The filters incorporated into both D/A and A/D convertors have a pronounced effect on
sound quality, since they determine the linearity of the frequency response
within the audio band, the slope with which it rolls off at high frequency
and the phase linearity of the system. In a non-oversampling convertor,
the filter must reject all signals above half the sampling frequency with
an attenuation of at least 80 dB. Steep filters tend to have an erratic phase
response at high frequencies and may exhibit ‘ringing’ due to the high ‘Q ’ of
the filter. Steep filters also have the added disadvantage that they are complicated to produce. Although filter effects are unavoidable to some extent,
manufacturers have made considerable improvements to analog antialiasing and reconstruction filters and these may be retro-fitted to many
existing systems with poor filters. A positive effect is normally noticed on
sound quality.
The process of oversampling and the use of higher sampling frequencies
(see below) have helped to ease the problems of such filtering. Here the first
repetition of the baseband is shifted to a much higher frequency, allowing
the use of a shallower anti-aliasing filter and consequently fewer audible
side effects.
Sampling frequency and sound quality
The choice of sampling frequency determines the maximum audio bandwidth available. There is a strong argument for choosing a sampling frequency no higher than is strictly necessary, in other words not much higher
Analog-to-Digital Conversion
FA C T F I L E 8 . 5 A U D IO S A MP LING FREQUENCI ES
The table shows commonly encountered sampling frequencies and their applications.
Frequency (kHz)
Application
8
16
22.05
Telephony (speech quality). ITU-T G711 standard
Used in some telephony applications. ITU-T G722 data reduction
Half the CD frequency is 22.05 kHz. Used in some older computer applications. The original
Apple Macintosh audio sampling frequency was 22 254.5454 … Hz
Used in some broadcast coding systems, e.g. NICAM. DAT long play mode. AES-5
secondary rate
A slight modification of the 44.1 kHz frequency used in some older equipment to
synchronize digital audio with the NTSC television frame rate of 29.97 frames per second.
Such ‘pull-down’ rates are sometimes still encountered in video sync situations
CD sampling frequency. AES-5 secondary rate
Occasionally encountered when 48 kHz equipment is used in NTSC video operations.
Another ‘pull-down’ rate, ideally to be avoided
AES-5 primary rate for professional applications. Basic rate for Blu-Ray disk (which no
longer specifies 44.1 kHz as an option)
Twice the CD sampling frequency. Optional for DVD-Audio
AES-5-1998 secondary rate for high bandwidth applications. Optional for DVD-Video, DVDAudio and Blu-Ray disks
Four times the basic standard rates. Optional in DVD-Audio. 192 kHz is the highest sampling
frequency allowed on Blu-Ray audio disks
DSD sampling frequency. A highly oversampled rate used in 1 bit PCM systems such as
SuperAudio CD
32
44.056
44.1
47.952
48
88.2
96
176.4 and 192
2.8224 MHz
than twice the highest audio frequency to be represented. This often starts
arguments over what is the highest useful audio frequency and this is an
area over which heated debates have raged. Conventional wisdom has it that
the audio frequency band extends up to 20 kHz, implying the need for a sampling frequency of just over 40 kHz for high quality audio work. There are in
fact two standard sampling frequencies between 40 and 50 kHz: the Compact
Disc rate of 44.1 kHz and the so-called ‘professional’ rate of 48 kHz. These
are both allowed in the original AES-5 standard of 1984, which sets down
preferred sampling frequencies for digital audio equipment. Fact File 8.5
shows commonly encountered sampling frequencies.
The 48 kHz rate was originally specified for professional use because it left
a certain amount of leeway for downward varispeed in tape recorders. When
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CHAPTER 8: Digital Audio Principles
many digital recorders are varispeeded, the sampling frequency changes proportionately and the result is a shifting of the first spectral repetition of the
audio baseband. If the sampling frequency is reduced too far aliased components may become audible. Most professional digital tape recorders allowed
for only around ⫾12.5% of varispeed for this reason. It is possible now,
though, to avoid such problems using digital low-pass filters whose cut-off frequency varies with the sampling frequency, or by using digital signal processing to vary the pitch of audio without varying the output sampling frequency.
The 44.1 kHz frequency had been established earlier on for the consumer
Compact Disc and is very widely used in the industry. In fact in many ways
it has become the sampling rate of choice for most professional recordings.
It allows for full use of the 20 kHz audio band and oversampling convertors
allow for the use of shallow analog anti-aliasing filters which avoid phase
problems at high audio frequencies. It also generates 10% less data per second than the 48 kHz rate, making it economical from a storage point of view.
A rate of 32 kHz is used in some broadcasting applications, such as
NICAM 728 stereo TV transmissions, and in some radio distribution systems. Television and FM radio sound bandwidth is limited to 15 kHz and
a considerable economy of transmission bandwidth is achieved by the use
of this lower sampling rate. The majority of important audio information
lies below 15 kHz in any case and little is lost by removing the top 5 kHz
of the audio band. Some professional audio applications offer this rate as
an option, but it is not common. It is used for the long play mode of some
DAT machines, for example.
Arguments for the standardization of higher sampling rates have become
stronger in recent years, quoting evidence from sources claiming that information above 20 kHz is important for higher sound quality, or at least that
the avoidance of steep filtering must be a good thing. The DVD standards,
for example, incorporate such sampling frequencies as standard features.
AES-5-1998 (a revision of the AES standard on sampling frequencies)
now allows 96 kHz as an optional rate for applications in which the audio
bandwidth exceeds 20 kHz or where relaxation of the anti-alias filtering
region is desired. Doubling the sampling frequency leads to a doubling in
the overall data rate of a digital audio system and a consequent halving
in storage time per megabyte. It also means that any signal processing algorithms need to process twice the amount of data and alter their algorithms
accordingly. It follows that these higher sampling rates should be used only
after careful consideration of the merits.
Low sampling frequencies such as those below 30 kHz are sometimes
encountered for lower quality sound applications such as the storage and
transmission of speech, the generation of computer sound effects and so
Analog-to-Digital Conversion
forth. Multimedia applications may need to support these rates because
such applications often involve the incorporation of sounds of different
qualities. There are also low sampling frequency options for data reduction
codecs, as discussed below.
Quantizing
After sampling, the modulated pulse chain is quantized. In quantizing a
sampled audio signal the range of sample amplitudes is mapped onto
a scale of stepped binary values, as shown in Figure 8.12. The quantizer
determines which of a fixed number of quantizing intervals (of size Q) each
sample lies within and then assigns it a value that represents the mid-point
of that interval. This is done in order that each sample amplitude can be
represented by a unique binary number in pulse code modulation (PCM).
(PCM is the designation for the form of modulation in which signals are
represented as a sequence of sampled and quantized binary data words.) In
linear quantizing each quantizing step represents an equal increment of signal voltage and most high quality audio systems use linear quantizing.
Quantizing error is an inevitable side effect in the process of A/D conversion and the degree of error depends on the quantizing scale used.
Considering binary quantization, a 4 bit scale offers 16 possible steps, an
8 bit scale offers 256 steps, and a 16 bit scale 65 536. The more bits, the
more accurate the process of quantization. The quantizing error magnitude
will be a maximum of plus or minus half the amplitude of one quantizing step and a greater number of bits per sample will therefore result in a
smaller error (see Figure 8.13), provided that the analog voltage range represented remains the same.
Figure 8.14 shows the binary number range covered by digital audio signals at different resolutions using the usual two’s complement hexadecimal
representation. It will be seen that the maximum positive sample value of
a 16 bit signal is &7FFF, whilst the maximum negative value is &8000.
The sample value changes from all zeros (&0000) to all ones (&FFFF) as
it crosses the zero point. The maximum digital signal level is normally
termed 0 dBFS (FS ⫽ full scale).
The quantized output of an A/D convertor can be represented in either
serial or parallel form, as shown in Fact File 8.6.
Quantizing resolution and sound quality
The quantizing error may be considered as an unwanted signal added to
the wanted signal, as shown in Figure 8.15. Unwanted signals tend to be
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CHAPTER 8: Digital Audio Principles
FIGURE 8.12
When a signal is quantized,
each sample is mapped
to the closest quantizing
interval Q, and given the
binary value assigned to
that interval. (Example of
a 3 bit quantizer shown.)
On D/A conversion each
binary value is assumed to
represent the voltage at the
mid point of the quantizing
interval.
Binary values of quantizing intervals (3 bit quantizer)
011
Q
010
001
time
000
111
110
101
100
011
3 bit quantizing scale
220
Max. error 0.5Q
Q
010
Error
Mid points of
quantizing
intervals
001
time
000
(a)
FIGURE 8.13(a) In (a) a 3 bit scale is used and only a small number of quantizing intervals
covers the analog voltage range, making the maximum quantizing error quite large. The second
sample in this picture will be assigned the value 010, for example the corresponding voltage of which
is somewhat higher than that of the sample. During D/A conversion the binary sample values from (a)
would be turned into pulses with the amplitudes shown in
Voltages equivalent to these binary numbers
Analog-to-Digital Conversion
011
010
001
time
000
(b)
0111
4 bit quantizing scale
0110
Q
0101
Max. error
0.5Q
0100
0011
Mid points of
quantizing
intervals
Error
0010
0001
time
0000
(c)
FIGURE 8.13 (b, c) (Continued))(b), where many samples have been forced to the same level
owing to quantizing. In (c) the 4 bit scale means that a larger number of intervals are used to cover the
same range and the quantizing error is reduced. (Expanded positive range only shown for clarity.)
221
222
CHAPTER 8: Digital Audio Principles
Max. +ve signal voltage
(a)
(b)
(c)
7F
7FFF
7FFFF
Positive values
00
FF
Zero volts
0000
FFFF
00000
FFFFF
Negative values
Max. –ve signal voltage
80
8000
80000
FIGURE 8.14 Binary number ranges (in hexadecimal) related to analog voltage ranges for
different convertor resolutions, assuming two’s complement representation of negative values. (a) 8 bit
quantizer, (b) 16 bit quantizer, (c) 20 bit quantizer.
FA C T F I L E 8 . 6 PA R A L L E L A N D SERI AL REPRESENTATI ON
Electrically it is possible to represent the quantized binary
signal in either serial or parallel form. When each bit of
the audio sample is carried on a separate wire, the signal is said to be in a parallel format, so a 16 bit convertor
would have 16 single bit outputs. If the data is transmitted
down a single wire or channel, one bit after the other, the
data is said to be in serial format. In serial communication
the binary word is clocked out one bit at a time using a
device known as a shift register. The shift register is previously loaded with the word in parallel form as shown in
the diagram. The rate at which the serial data is transferred depends on the rate of the clock.
Serial form is most useful for transmission over
interconnects or transmission links that might cover
substantial distances or where the bulk and cost of the
interconnect limits the number of paths available. Parallel
form tends to be used internally, within high speed digital
systems, although serial forms are increasingly used here
as well. Most digital audio interfaces are serial, for example, although the Tascam TDIF interface uses a parallel
representation of the audio data.
Parallel input
0
1
1
1
0
0
1
0
Serial output
Clock (bit rate)
Analog-to-Digital Conversion
classified either as distortion or noise, depending on their characteristics,
and the nature of the quantizing error signal depends very much upon
the level and nature of the related audio signal. Here are a few examples, the
illustrations for which have been prepared in the digital domain for clarity,
using 16 bit sample resolution.
First, consider a very low level sine wave signal, sampled then quantized, having a level only just sufficient to turn the least significant bit of
the quantizer on and off at its peak (see Figure 8.16(a)). Such a signal would
have a quantizing error that was periodic, and strongly correlated with the
signal, resulting in harmonic distortion. Figure 8.16(b) shows the frequency
spectrum, analyzed in the digital domain of such a signal, showing clearly
the distortion products (predominantly odd harmonics) in addition to the
original fundamental. Once the signal falls below the level at which it
just turns on the LSB there is no modulation. The audible result, therefore, of fading such a signal down to silence is that of an increasingly distorted signal suddenly disappearing. A higher level sine wave signal would
cross more quantizing intervals and result in more non-zero sample values.
As signal level rises the quantizing error, still with a maximum value of
⫾0.5Q, becomes increasingly small as a proportion of the total signal level
and the error gradually loses its correlation with the signal.
Consider now a music signal of reasonably high level. Such a signal has
widely varying amplitude and spectral characteristics and consequently the
quantizing error is likely to have a more random nature. In other words
it will be more noise-like than distortion-like, hence the term quantizing
noise that is often used to describe the audible effect of quantizing error. An
analysis of the power of the quantizing error, assuming that it has a noiselike nature, shows that it has an r.m.s. amplitude of Q/12, where Q is the
voltage increment represented by one quantizing interval. Consequently the
signal-to-noise ratio of an ideal n bit quantized signal can be shown to be:
6.02 n ⫹ 1.76 dB
This implies a theoretical S/N ratio that approximates to just over 6 dB
per bit. So a 16 bit convertor might be expected to exhibit an S/N ratio
of around 98 dB, and an 8 bit convertor around 50 dB. This assumes an
undithered convertor, which is not the normal case, as described below. If
a convertor is undithered there will only be quantizing noise when a signal
is present, but there will be no quiescent noise floor in the absence of a
signal. Issues of dynamic range with relation to human hearing are discussed further in Fact File 8.7.
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CHAPTER 8: Digital Audio Principles
FIGURE 8.15
Quantizing error depicted
as an unwanted signal
added to the original
sample values. Here the
error is highly correlated
with the signal and will
appear as distortion.
(Courtesy of Allen
Mornington West.)
FIGURE 8.16
(a) A 1 kHz sine wave at
very low level (amplitude
⫾1 LSB) just turns the
least significant bit of
the quantizer on and off.
Analyzed in the digital
domain with sample values
shown in hex on the
vertical axis and time in
ms on the horizontal axis.
(b) Frequency spectrum
of this quantized sine
wave, showing distortion
products.
5 bit 2’s complement
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
0
1
0
0
1
1
0
0
1
0
1
0
1
1
0
0
0
0 10
1
8
0
6
1
4
0
2
1
0
0
1 –2
1 –4
1 –6
0 –8
1 –10
0
–50
1
0
Time
0
50
100
150
200
250
300
1 kHz sine wave, ampl. 1 LSB, no dither
(a)
000400
Hex
000300
000200
000100
000000
FFFF00
FFFE00
FFFD00
SCOPE
FFFC00
ms
0.00
0.36
0.73
1.09
1.45
2.18
SCOPE
2.54
2.90
16537
19293 FFT 22050
1.81
1 kHz sine wave, ampl. 1 LSB, no dither
(b)
0.00
dBFS
–17.50
–35.00
–52.50
–70.00
–87.50
–105.00
–122.50
FFT
–140.00
0 Hz 2756
5512
8268
11025
13781
Analog-to-Digital Conversion
FA C T F I L E 8 . 7 D YN A MIC R A N G E AND PERCEPTI ON
It is possible with digital audio to approach the limits of
human hearing in terms of sound quality. In other words,
the unwanted artefacts of the process can be controlled so
as to be close to or below the thresholds of perception. It is
also true, though, that badly engineered digital audio can
sound poor and that the term ‘digital’ does not automatically imply high quality. The choice of sampling parameters
and noise shaping methods, as well as more subtle aspects
of convertor design, affect the frequency response, distortion and perceived dynamic range of digital audio signals.
The human ear’s capabilities should be regarded as
the standard against which the quality of digital systems
is measured, since it could be argued that the only distortions and noises that matter are those that can be heard.
Work carried out by Louis Fielder and Elizabeth Cohen
attempted to establish the dynamic range requirements
for high quality digital audio systems by investigating
the extremes of sound pressure available from acoustic
sources and comparing these with the perceivable noise
floors in real acoustic environments. Using psychoacoustic theory, Fielder was able to show what was likely to
be heard at different frequencies in terms of noise and
distortion, and where the limiting elements might be in a
typical recording chain. He determined a dynamic range
requirement of 122 dB for natural reproduction. Taking
into account microphone performance and the limitations
of consumer loudspeakers, this requirement dropped to
115 dB for consumer systems.
The dynamic range of a digital audio system is limited at high signal
levels by the point at which the quantizing range of the convertor has been
‘used up’ (in other words, when there are no more bits available to represent a higher level signal). At this point the waveform will be hard clipped
(see Figure 8.17) and will become very distorted. This point will normally
be set to occur at a certain electrical input voltage, such as ⫹24 dBu in
some professional systems. (The effect is very different from that encountered in analog tape recorders which tend to produce gradually more distortion as the recording level increases. Digital recorders remain relatively
undistorted as the recording level rises until the overload point is reached,
at which point very bad distortion occurs.)
The number of bits per sample therefore dictates the signal-to-noise ratio
of a linear PCM digital audio system. Fact File 8.8 summarizes the applications for different quantizing resolutions. For many years 16 bit linear PCM
was considered the norm for high quality audio applications. This is the CD
standard and is capable of offering a good S/N ratio range of over 90 dB. For
most purposes this is adequate, but it fails to reach the psychoacoustic ideal
of 122 dB for subjectively noise-free reproduction in professional systems. To
achieve such a performance requires a convertor resolution of around 21 bits,
which is achievable with today’s convertor technology, depending on how the
specification is interpreted. So-called 24 bit convertors are indeed available
today, but their audio performance is strongly dependent upon the stability
225
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CHAPTER 8: Digital Audio Principles
FIGURE 8.17
Signals exceeding peak
level in a digital system
are hard-clipped, since no
more digits are available
to represent the sample
value.
0 dB FS (full scale) signals above this level are clipped
Large signal
Zero-crossing level
Range of
convertor
Maximum negative signal level
of the timing clock, electrical environment, analog stages, grounding and
other issues.
For professional recording purposes one may need a certain amount of
‘headroom’ – in other words some unused dynamic range above the normal peak recording level which can be used in unforeseen circumstances
such as when a signal overshoots its expected level. This can be particularly necessary in live recording situations where one is never quite sure
what is going to happen with recording levels. This is another reason why
many professionals feel that a resolution of greater than 16 bits is desirable
for original recording. Twenty and 24 bit recording formats are becoming
increasingly popular for this reason, with mastering engineers then optimizing the finished recording for 16 bit media (such as CD) using noiseshaped requantizing processes.
Use of dither
The use of dither in A/D conversion, as well as in conversion between one
sample resolution and another, is now widely accepted as correct. It has
the effect of linearizing a normal convertor (in other words it effectively
makes each quantizing interval the same size) and turns quantizing distortion into a random, noise-like signal at all times. This is desirable for
a number of reasons. First, because white noise at a very low level is less
subjectively annoying than distortion; second, because it allows signals to
be faded smoothly down without the sudden disappearance noted above;
and third, because it often allows signals to be reconstructed even when
their level is below the noise floor of the system. Undithered audio signals
begin to sound ‘grainy ’ and distorted as the signal level falls. Quiescent hiss
will disappear if dither is switched off, making a system seem quieter, but a
small amount of continuous hiss is considered preferable to low level distortion. The resolution of modern high resolution convertors is such that
the noise floor is normally inaudible in any case.
Analog-to-Digital Conversion
FA C T F I L E 8 . 8 Q U A N TIZIN G R ESOLUTI ONS
The table shows some commonly encountered quantizing resolutions and their applications.
Bits per
Approx. dynamic)
Sample
range with dither (dB)
8
44
12
14
16
68
80
92
20
24
116
140
Application
Low–moderate quality for older PC internal sound generation. Some older
multimedia applications.
Usually in the form of unsigned binary numbers
Older Akai samplers, e.g. S900
Original EIAJ format PCM adaptors, such as Sony PCM-100
CD standard. DAT standard. Commonly used high quality resolution for
consumer media, some professional recorders and multimedia PCs. Usually
two’s complement (signed) binary numbers
High quality professional audio recording and mastering applications
Maximum resolution of most recent professional recording systems,
also of AES 3 digital interface. Dynamic range exceeds psychoacoustic
requirements. Hard to convert accurately at this resolution.
Dithering a convertor involves the addition of a very low-level signal to
the audio whose amplitude depends upon the type of dither employed (see
Fact File 8.9). The dither signal is usually noise, but may also be a waveform at half the sampling frequency or a combination of the two. A signal
that has not been correctly dithered during the A/D conversion process cannot thereafter be dithered with the same effect, because the signal will have
been irrevocably distorted. How then does dither perform the seemingly
remarkable task of removing quantizing distortion?
It was stated above that the distortion was a result of the correlation
between the signal and the quantizing error, making the error periodic and
subjectively annoying. Adding noise, which is a random signal, to the audio
has the effect of randomizing the quantizing error and making it noise-like
as well (shown in Figure 8.18(a) and (b)). If the noise has an amplitude similar in level to the LSB (in other words, one quantizing step) then a signal
lying exactly at the decision point between one quantizing interval and
the next may be quantized either upwards or downwards, depending on the
instantaneous level of the dither noise added to it. Over time this random
effect is averaged, leading to a noise-like quantizing error and a fixed noise
floor in the system.
Dither is also used in digital processing devices such as mixers, but
in such cases it is introduced in the digital domain as a random number
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CHAPTER 8: Digital Audio Principles
FA C T F I L E 8 . 9 T Y P E S O F D ITH ER
Research has shown that certain dither signals are more
suitable than others for high quality audio work. Dither
noise is often characterized in terms of its probability distribution, which is a statistical method of showing the likelihood of the signal having a certain amplitude. A simple
graph is used to indicate the shape of the distribution.
The probability is the vertical axis and the amplitude in
terms of quantizing steps is the horizontal axis.
Logical probability distributions can be understood
simply by thinking of the way in which dice fall when
thrown (see the diagram). A single throw has a rectangular probability distribution function (RPDF), as shown in
(a), because there is an equal chance of the throw being
between 1 and 6. The total value of a pair of dice, on the
other hand, has a roughly triangular probability distribution function (TPDF), as shown in (b), with the peak
grouped on values from 6 to 8, because there are more
combinations that make these totals than there are combinations making 2 or 12. Going back to digital electronics,
one could liken the dice to random number generators
and see that RPDF dither could be created using a single
random number generator, and that TPDF dither could be
created by adding the outputs of two RPDF generators.
RPDF dither has equal likelihood that the amplitude
of the noise will fall anywhere between zero and maximum, whereas TPDF dither has greater likelihood that
the amplitude will be zero than that it will be maximum.
Although RPDF and TPDF dither can have the effect of
linearizing a digital audio system and removing distortion,
RPDF dither tends to result in noise modulation at low
signal levels. The most suitable dither noise is found to
be TPDF with a peak-to-peak amplitude of 2Q. If RPDF
dither is used it should have a peak-to-peak amplitude of
1Q. Analog white noise has Gaussian probability, whose
shape is like a normal distribution curve. With Gaussian
noise, the optimum r.m.s. amplitude for the dither signal
is 0.5Q, at which level noise modulation is minimized but
not altogether absent. Dither at this level has the effect
of reducing the undithered dynamic range by about 6 dB,
making the dithered dynamic range of an ideal 16 bit
convertor around 92 dB.
Probability
(a)
1
2
3
4
5
6
Value
(b)
Probability
228
1
6–8
13
Analog-to-Digital Conversion
(a)
Amplitude
12
10
8
6
4
2
0
–2
–4
–6
–8
–10
–12
–50
Time
0
50
(b)
100
150
200
250
300
Amplitude
10
8
6
4
2
0
–2
–4
Time
–6
–8
–10
–50
0
50
100
150
200
250
300
sequence (the digital equivalent of white noise). In this context it is used
to remove low-level distortion in signals whose gains have been altered and
to optimize the conversion from high resolution to lower resolution during
post-production.
Oversampling in A/D conversion
Oversampling involves sampling audio at a higher frequency than
strictly necessary to satisfy the Nyquist criterion. Normally, though, this
high rate is reduced to a lower rate in a subsequent digital filtering process, in order that no more storage space is required than for conventionally sampled audio. It works by trading off quantizing resolution against
sampling rate, based on the principle that the information carrying capacity of a channel is related to the product of these two factors. Samples at a
high rate with low resolution can be converted into samples at a lower rate
with higher resolution, with no overall loss of information. Oversampling
FIGURE 8.18
(a) Dither noise added to
a sine wave signal prior to
quantization. (b) Postquantization the error signal
is now random and noiselike. (Courtesy of Allen
Mornington West.)
229
230
CHAPTER 8: Digital Audio Principles
has now become so popular that it is the norm in most high quality audio
convertors.
Although oversampling A/D convertors often quote very high sampling
rates of up to 128 times the basic rates of 44.1 or 48 kHz, the actual rate at
the digital output of the convertor is reduced to a basic rate or a small multiple thereof (e.g. 48, 96 or 192 kHz). Samples acquired at the high rate are
quantized to only a few bits’ resolution and then digitally filtered to reduce
the sampling rate, as shown in Figure 8.19. The digital low-pass filter limits the bandwidth of the signal to half the basic sampling frequency in order
to avoid aliasing, and this is coupled with ‘decimation’. Decimation reduces
the sampling rate by dropping samples from the oversampled stream.
A result of the low-pass filtering operation is to increase the word length
of the samples very considerably. This is not simply an arbitrary extension
of the word length, but an accurate calculation of the correct value of each
sample, based on the values of surrounding samples. Although oversampling convertors quantize samples initially at a low resolution, the output of the decimator consists of samples at a lower rate with more bits of
resolution. The sample resolution can then be shortened as necessary (see
‘Requantizing’, below) to produce the desired word length.
Oversampling brings with it a number of benefits and is the key to
improved sound quality at both the A/D and D/A ends of a system. Because
the initial sampling rate is well above the audio range (often tens or hundreds of times the nominal rate) the spectral repetitions resulting from PAM
are a long way from the upper end of the audio band (see Figure 8.20). The
analog anti-aliasing filter used in conventional convertors is replaced by
a digital decimation filter. Such filters can be made to have a linear phase
response if required, resulting in higher sound quality. If oversampling is
also used in D/A conversion the analog reconstruction filter can have a shallower roll-off. This can have the effect of improving phase linearity within
the audio band, which is known to improve audio quality. In oversampled
D/A conversion, basic rate audio is up-sampled to a higher rate before conversion and reconstruction filtering. Oversampling also makes it possible to
introduce so-called ‘noise shaping’ into the conversion process, which allows
quantizing noise to be shifted out of the most audible parts of the spectrum.
Oversampling without subsequent decimation is a fundamental principle of Sony’s Direct Stream Digital system, described below.
Noise shaping in A/D conversion
Noise shaping is a means by which noise within the most audible parts of
the audio frequency range is reduced at the expense of increased noise at
Analog-to-Digital Conversion
Small number
of bits per sample
Sample
Audio in
Quantize
Larger number
of bits per sample
Digital LPF
and
decimate
Rate = Multiple of fs
Rate = fs
Amplitude
(a)
Potential alias region
Audio
band
Amplitude
(b)
fs
Frequency
fs
2fs
Multiple
of fs
Digital LPF
Audio
band
FIGURE 8.19
Block diagram of
oversampling A/D
conversion process.
3fs
4fs
other frequencies, using a process that ‘shapes’ the spectral energy of the
quantizing noise. It is possible because of the high sampling frequencies
used in oversampling convertors. A high sampling frequency extends the
frequency range over which quantizing noise is spread, putting much of it
outside the audio band.
Quantizing noise energy extends over the whole baseband, up to the
Nyquist frequency. Oversampling spreads the quantizing noise energy
over a wider spectrum, because in oversampled convertors the Nyquist frequency is well above the upper limit of the audio band. This has the effect
of reducing the in-band noise by around 3 dB per octave of oversampling (in
other words, a system oversampling at twice the Nyquist rate would see the
noise power within the audio band reduced by 3 dB).
FIGURE 8.20
(a) Oversampling in A/D
conversion initially creates
spectral repetitions that lie
a long way from the top
of the audio baseband.
The dotted line shows the
theoretical extension of the
baseband and the potential
for aliasing, but the audio
signal only occupies the
bottom part of this band.
(b) Decimation and digital
low-pass filtering limits
the baseband to half
the sampling frequency,
thereby eliminating any
aliasing effects, and creates
a conventional collection
of spectral repetitions at
multiples of the sampling
frequency.
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CHAPTER 8: Digital Audio Principles
FIGURE 8.21
Block diagram of a noise
shaping delta–sigma A/D
convertor.
Audio
signal
+
Add
Noise-shaping
filter/integrator
A/D
convertor
Decimator/
low-pass filter
–
D/A
convertor
In oversampled noise-shaping A/D conversion an integrator (low-pass
filter) is introduced before the quantizer, and a D/A convertor is incorporated into a negative feedback loop, as shown in Figure 8.21. This is the
so-called ‘sigma–delta convertor ’. Without going too deeply into the principles of such convertors, the result is that the quantizing noise (introduced
after the integrator) is given a rising frequency response at the input to the
decimator, whilst the input signal is passed with a flat response. There
are clear parallels between such a circuit and analog negative-feedback
circuits.
Without noise shaping, the energy spectrum of quantizing noise is flat
up to the Nyquist frequency, but with first-order noise shaping this energy
spectrum is made non-flat, as shown in Figure 8.22. With second-order
noise shaping the in-band reduction in noise is even greater, such that the
in-band noise is well below that achieved without noise shaping.
D/A CONVERSION
A basic D/A convertor
The basic D/A conversion process is shown in Figure 8.23. Audio sample
words are converted back into a staircase-like chain of voltage levels corresponding to the sample values. This is achieved in simple convertors by
using the states of bits to turn current sources on or off, making up the
required pulse amplitude by the combination of outputs of each of these
sources. This staircase is then ‘resampled’ to reduce the width of the pulses
before they are passed through a low-pass reconstruction filter whose cutoff frequency is half the sampling frequency. The effect of the reconstruction filter is to join up the sample points to make a smooth waveform.
Resampling is necessary to avoid any discontinuities in signal amplitude
at sample boundaries and because otherwise the averaging effect of the filter would result in a reduction in the amplitude of high-frequency audio
signals (the so-called ‘aperture effect’). Aperture effect may be reduced by
D/A Conversion
Audio band
Power
Power
Audio band
Quantizing noise
Quantizing noise
fn
Frequency
Frequency
(a)
(b)
Power
Audio band
Frequency
fn
FIGURE 8.22
Frequency spectra of quantizing
noise. In a non-oversampled
convertor, as shown in (a), the
quantizing noise is constrained to
lie within the audio band. In an
oversampling convertor, as shown
in (b), the quantizing noise power
is spread over a much wider range,
thus reducing its energy in the
audio band. (c) With noise shaping
the noise power within the audio
band is reduced still further, at the
expense of increased noise outside
that band.
fn
Quantizing noise power
no noise shaping
with 1st order filter
with 2nd order filter
Time
D/A convertor
output
FIGURE 8.23
Processes involved in D/A
conversion (positive sample
values only shown).
Amplitude
Amplitude
Amplitude
(c)
Time
Time
is resampled
to reduce pulse width
and is low-pass
filtered
to reconstruct original
waveform
limiting the width of the sample pulses to perhaps one-eighth of the sample
period. Equalization may be required to correct for aperture effect.
Oversampling in D/A conversion
Oversampling may be used in D/A conversion, as well as in A/D conversion. In the D/A case additional samples must be created in between the
233
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CHAPTER 8: Digital Audio Principles
Nyquist rate samples in order that conversion can be performed at a higher
sampling rate. These are produced by sample rate conversion of the PCM
data. These samples are then converted back to analog at the higher rate,
again avoiding the need for steep analog filters. Noise shaping may also be
introduced at the D/A stage, depending on the design of the convertor, to
reduce the subjective level of the noise.
A number of advanced D/A convertor designs exist which involve oversampling at a high rate, creating samples with only a few bits of resolution.
The extreme version of this approach involves very high rate conversion of
single bit samples (so-called ‘bit stream conversion’), with noise shaping to
optimize the noise spectrum of the signal. The theory of these convertors is
outside the scope of this book.
DIRECT STREAM DIGITAL (DSD)
Direct Stream Digital (DSD) is Sony’s proprietary name for its 1 bit digital
audio coding system that uses a very high sampling frequency (2.8224 MHz
as a rule). This system is used for audio representation on the consumer
Super Audio CD (SACD) and in various items of professional equipment
used for producing SACD material. It is not directly compatible with conventional PCM systems although DSD signals can be down-sampled and
converted to multibit PCM if required.
DSD signals are the result of delta–sigma conversion of the analog signal, a technique used at the front end of some oversampling convertors
described above. As shown in Figure 8.24, a delta–sigma convertor employs
a comparator and a feedback loop containing a low-pass filter that effectively quantizes the difference between the current sample and the accumulated value of previous samples. If it is higher then a ‘1’ results, if it is lower
a ‘0’ results. This creates a 1 bit output that simply alternates between
one and zero in a pattern that depends on the original signal waveform, as
shown in Figure 8.24. Conversion to analog can be as simple a matter as
passing the bit stream through a low-pass filter, but is usually somewhat
more sophisticated, involving noise shaping and higher order filtering.
Although one would expect 1 bit signals to have an appalling signal-tonoise ratio, the exceptionally high sampling frequency spreads the noise over
a very wide frequency range leading to lower noise within the audio band.
Additionally, high-order noise shaping is used to reduce the noise in the audio
band at the expense of that at much higher (inaudible) frequencies, as discussed earlier. A dynamic range of around 120 dB is therefore claimed, as well
as a frequency response extending smoothly to over 100 kHz.
Changing the Resolution of an Audio Signal
FIGURE 8.24 Direct Stream Digital bitstream generation. (a) Typical binary representation of a
sine wave. (b) Pulse density modulation. (c) DSD signal chain.
CHANGING THE RESOLUTION OF AN AUDIO SIGNAL
(REQUANTIZATION)
There may be points in an audio production when the need arises to change
the resolution of a signal. A common example of this in high quality audio
is when mastering 16 bit consumer products from 20 or 24 bit recordings,
but it also occurs within signal processors of all types because sample word
lengths may vary at different stages. It is important that this operation is
performed correctly because incorrect requantization results in unpleasant
distortion, just like undithered quantization in A/D conversion. Dynamic
range enhancement can also be employed when requantizing for consumer
media, as shown in Fact File 8.10.
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CHAPTER 8: Digital Audio Principles
FA C T F I L E 8. 1 0 D YN A MIC R A N GE ENHANCEM ENT
Noise spectrum (dB)
It is possible to maximize the subjective dynamic range diagram, although many other shapes are in use. Some
of digital audio signals during the process of requantiza- approaches allow the mastering engineer to choose from
tion. This is particularly useful when mastering high reso- a number of ‘shapes’ of noise until one is found that is
lution recordings for CD because the reduction to 16 bit subjectively the most pleasing for the type of music conword lengths would normally result in increased quan- cerned, whereas others stick to one theoretically derived
tizing noise. It is in fact possible to retain most of the ‘correct’ shape.
dynamic range of a higher resolution recording, even though it
(a) 40
is being transferred to a 16 bit
medium. This remarkable feat
30
is achieved by a noise-shaping
Shape A
process similar to that described
Shape B
20
Shape C
earlier.
During requantization digital
10
filtering is employed to shape
the spectrum of the quantizing
0
noise so that as much of it as
possible is shifted into the least
–10
audible parts of the spectrum.
–20
This usually involves moving the
noise away from the 4 kHz region
–30
where the ear is most sensitive
10000
15000
20000
0
5000
and increasing it at the highFrequency (Hz)
frequency end of the spectrum.
The result is often quite high
(b) 40
levels of noise at high frequency,
Shape D
but still lying below the audibil30
Shape E
ity threshold. In this way CDs
can be made to sound almost
20
as if they had the dynamic range
of 20 bit recordings. Some typi10
cal weighting curves used in a
commercial mastering processor
0
from Meridian are shown in the
Noise spectrum (dB)
236
–10
–20
–30
0
5000
10000
Frequency (Hz)
15000
20000
Changing the Resolution of an Audio Signal
FIGURE 8.25
Truncation of audio
samples results in
distortion. (a) Shows the
spectrum of a 1 kHz signal
generated and analyzed at
20 bit resolution. In (b) the
signal has been truncated
to 16 bit resolution and
the distortion products are
clearly noticeable.
1 kHz sine wave, –90 dBFS, 20-bit original
(a) 0.00
dBFS
–17.50
–35.00
–52.50
–70.00
–87.50
–105.00
–122.50
FFT
–140.00
0
Hz 2756
5512
8268
11025
13781
16537
19293 FFT 22050
1 kHz sine wave, –90 dBFS, 20-bit truncated to 16-bit
(b) 0.00
dBFS
–17.50
–35.00
–52.50
–70.00
–87.50
–105.00
–122.50
FFT
–140.00
0
Hz 2756
5512
8268
11025
13781
16537
19293 FFT 22050
If the length of audio samples needs to be reduced then the worst possible solution is simply to remove unwanted LSBs. Taking the example of a
20 bit signal being reduced to 16 bits, one should not simply remove the 4
LSBs and expect everything to be all right. By removing the LSBs one would
be creating a similar effect to not using dither in A/D conversion – in other
words one would introduce low-level distortion components. Low-level
237
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CHAPTER 8: Digital Audio Principles
signals would sound grainy and would not fade smoothly into noise. Figure
8.25 shows a 1 kHz signal at a level of ⫺90 dBFS that originally began life
at 20 bit resolution but has been truncated to 16 bits. The harmonic distortion is clearly visible.
The correct approach is to redither the signal for the target resolution
by adding dither noise in the digital domain. This digital dither should be
at an appropriate level for the new resolution and the LSB of the new sample should then be rounded up or down depending on the total value of
the LSBs to be discarded, as shown in Figure 8.26. It is worrying to note
how many low cost digital audio applications fail to perform this operation
satisfactorily, leading to complaints about sound quality. Many professional
quality audio workstations allow for audio to be stored and output at a variety of resolutions and may make dither user-selectable. They also allow the
level of the audio signal to be changed in order that maximum use may be
made of the available bits. It is normally important, for example when mastering a CD from a 20 bit recording, to ensure that the highest level signal
on the original recording is adjusted during mastering so that it peaks close
to the maximum level before requantizing and redithering at 16 bit resolution. In this way as much as possible of the original low-level information
is preserved and quantizing noise is minimized. This applies in any requantizing operation, not just CD mastering. A number of applications are
available that automatically scale the audio signal so that its level is optimized in this way, allowing the user to set a peak signal value up to which
the highest level samples will be scaled. Since some overload detectors
on digital meters and CD mastering systems look for repeated samples at
maximum level to detect clipping, it is perhaps wise to set peak levels so
that they lie just below full modulation. This will ensure that master tapes
are not rejected for a suspected recording fault by duplication plants and
subsequent users do not complain of ‘over ’ levels.
INTRODUCTION TO DIGITAL SIGNAL PROCESSING
Just as processing operations like equalization, fading and compression
can be performed in the analog domain, so they can in the digital domain.
Indeed it is often possible to achieve certain operations in the digital
domain with fewer side effects such as phase distortion. It is possible to
perform operations in the digital domain that are either very difficult or
impossible in the analog domain. High quality, authentic-sounding artificial
reverberation is one such example, in which the reflection characteristics
of different halls and rooms can be accurately simulated. Digital signal
Introduction to Digital Signal Processing
Peak level
New res. noise floor
Original res. noise floor
Original signal
dynamic range
Optimize
dynamic
range
Add dither
at correct level
for new res.
Requantize
to new res.
processing (DSP) involves the high speed manipulation of the binary data
representing audio samples. It may involve changing the values and timing
order of samples and it may involve the combining of two or more streams
of audio data. DSP can affect the sound quality of digital audio in that it
can add noise or distortion, although one must assume that the aim of
good design is to minimize any such degradation in quality.
In the sections that follow an introduction will be given to some of
the main applications of DSP in audio workstations without delving into the
mathematical principles involved. In some cases the description is an oversimplification of the process, but the aim has been to illustrate concepts
not to tackle the detailed design considerations involved.
Gain changing (level control)
It is relatively easy to change the level of an audio signal in the digital
domain. It is most easy to shift its gain by 6 dB since this involves shifting
the whole sample word either one step to the left or right (see Figure 8.27).
Effectively the original value has been multiplied or divided by a factor of
two. More precise gain control is obtained by multiplying the audio sample
value by some other factor representing the increase or decrease in gain.
The number of bits in the multiplication factor determines the accuracy of
gain adjustment. The result of multiplying two binary numbers together is
to create a new sample word which may have many more bits than the original and it is common to find that digital mixers have internal structures
capable of handling 32 bit words, even though their inputs and outputs
FIGURE 8.26
The correct order of events
when requantizing an audio
signal at a lower resolution
is shown here.
239
240
CHAPTER 8: Digital Audio Principles
FIGURE 8.27
The gain of a sample may
be changed by 6 dB simply
by shifting all the bits one
step to the left or right.
MSB
0
LSB
0
1
1
1
1
1
1
0
+6 dB
=
MSB
0
0
1
0
1
0
Original 8 bit sample
LSB
0
New sample with higher level
may handle only 20. Because of this, redithering is usually employed in
mixers at points where the sample resolution has to be shortened, such as
at any digital outputs or conversion stages, in order to preserve sound quality as described above.
The values used for multiplication in a digital gain control may be
derived from any user control such as a fader, rotary knob or on-screen representation, or they may be derived from stored values in an automation
system. A simple ‘old-fashioned’ way of deriving a digital value from an
‘analog’ fader is to connect the fader to a fixed voltage supply and connect
the fader wiper to an A/D convertor, although it is quite common now to
find controls capable of providing a direct binary output relating to their
position. The ‘law ’ of the fader (the way in which its gain is related to its
physical position) can be determined by creating a suitable look-up table
of values in memory which are then used as multiplication factors corresponding to each physical fader position.
Mixing
Mixing is the summation of independent data streams representing the different audio channels. Time coincident samples from each input channel are
summed to produce a single output channel sample. Clearly it is possible to
have many mix ‘buses’ by having a number of separate summing operations
for different output channels. The result of summing a lot of signals may be to
increase the overall level considerably and the architecture of the mixer must
allow enough headroom for this possibility. In the same way as an analog
mixer, the gain structure within a digital mixer must be such that there is an
appropriate dynamic range window for the signals at each point in the chain,
also allowing for operations such as equalization that change the signal level.
Crossfading is a combination of gain changing and mixing, as described
in Fact File 8.11.
Digital filters and equalization
Digital filtering is something of a ‘catch-all’ term, and is often used
to describe DSP operations that do not at first sight appear to be filtering.
Introduction to Digital Signal Processing
FA C T F I L E 8 . 1 1 C R O S S FA D IN G
Crossfading is employed widely in audio workstations
at points where one section of sound is to be joined to
another (edit points). It avoids the abrupt change of waveform that might otherwise result in an audible click and
allows one sound to take over smoothly from the other.
The process is illustrated conceptually here. It involves
two signals each undergoing an automated fade (binary
multiplication), one downwards and the other upwards,
followed by an addition of the two signals. By controlling the rates and coefficients involved in the fades one
can create different styles of crossfade for different
purposes.
Decreasing coefficients
Amplitude
Time
Outgoing
samples
Sum
Multipliers
Adder
Incoming
samples
Amplitude
Time
Increasing coefficients
A digital filter is essentially a process that involves the time delay, multiplication and recombination of audio samples in all sorts of configurations,
from the simplest to the most complex. Using digital filters one can create low- and high-pass filters, peaking and shelving filters, echo and reverberation effects, and even adaptive filters that adjust their characteristics to
affect different parts of the signal.
To understand the basic principle of digital filters it helps to think
about how one might emulate a certain analog filtering process digitally.
Filter responses can be modeled in two main ways – one by looking at their
frequency domain response and the other by looking at their time
domain response. (There is another approach involving the so-called z-plane
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CHAPTER 8: Digital Audio Principles
FIGURE 8.28
Examples of (a) the
frequency response of a
simple filter, and (b) the
equivalent time domain
impulse response.
Amplitude
Amplitude
Time
Frequency
(a)
(b)
transform, but this is not covered here.) The frequency domain response
shows how the amplitude of the filter’s output varies with frequency,
whereas the time domain response is usually represented in terms of an
impulse response (see Figure 8.28). An impulse response shows how the filter’s output responds to stimulation at the input by a single short impulse.
Every frequency response has a corresponding impulse (time) response
because the two are directly related. If you change the way a filter responds
in time you also change the way it responds in frequency. A mathematical process known as the Fourier transform is often used as a means of
transforming a time domain response into its equivalent frequency domain
response. They are simply two ways of looking at the same thing.
Digital audio is time discrete because it is sampled. Each sample represents the amplitude of the sound wave at a certain point in time. It is
therefore normal to create certain filtering characteristics digitally by operating on the audio samples in the time domain. In fact if it were desired
to emulate a certain analog filter characteristic digitally one would theoretically need only to measure its impulse response and model this in the
digital domain. The digital version would then have the same frequency
response as the analog version, and one can even envisage the possibility
for favorite analog filters to be recreated for the digital workstation. The
question, though, is how to create a particular impulse response characteristic digitally, and how to combine this with the audio data.
As mentioned earlier, all digital filters involve delay, multiplication
and recombination of audio samples, and it is the arrangement of these
elements that gives a filter its impulse response. A simple filter model is
the finite impulse response (FIR) filter, or transversal filter, shown in Figure
8.29. As can be seen, this filter consists of a tapped delay line with each
tap being multiplied by a certain coefficient before being summed with
the outputs of the other taps. Each delay stage is normally a one sample
period delay. An impulse arriving at the input would result in a number of
Introduction to Digital Signal Processing
In
Delay
1/fs
N
Delay
1/fs
X
N
Delay
1/fs
X
N
Delay
1/fs
X
N
Delay
1/fs
X
N
X
+
Out
Output
Input
1/fs
Time
separate versions of the impulse being summed at the output, each with a
different amplitude. It is called a finite impulse response filter because a
single impulse at the input results in a finite output sequence determined
by the number of taps. The more taps there are the more intricate the filter’s response can be made, although a simple low-pass filter only requires
a few taps.
The other main type is the infinite impulse response (IIR) filter, which
is also known as a recursive filter because there is a degree of feedback
between the output and the input (see Figure 8.30). The response of such
a filter to a single impulse is an infinite output sequence, because of the
feedback. IIR filters are often used in audio equipment because they involve
fewer elements for most variable equalizers than equivalent FIR filters, and
they are useful in effects devices. They are unfortunately not phase linear,
though, whereas FIR filters can be made phase linear.
Digital reverberation and other effects
It can probably be seen that the IIR filter described in the previous section
forms the basis for certain digital effects, such as reverberation. The impulse
response of a typical room looks something like Figure 8.31, that is an
FIGURE 8.29
A simple FIR filter
(transversal filter).
N ⫽ multiplication
coefficient for each tap.
Response shown below
indicates successive
outputs’ samples,
multiplied by decreasing
coefficients.
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CHAPTER 8: Digital Audio Principles
FIGURE 8.30
A simple IIR filter (recursive
filter). The output impulses
continue indefinitely but
become very small. N in
this case is about 0.8.
A similar response to
the previous FIR filter is
achieved but with fewer
stages.
Delay
1/fs
+
X
N
Output
Input
Time
Direct sound
Early reflections
Level
244
Diffuse decay
Time
FIGURE 8.31 The impulse response of a typical reflective room.
initial direct arrival of sound from the source, followed by a series of early
reflections, followed by a diffuse ‘tail’ of densely packed reflections decaying
gradually to almost nothing. Using a number of IIR filters, perhaps together
with a few FIR filters, one could create a suitable pattern of delayed and
attenuated versions of the original impulse to simulate the decay pattern of
a room. By modifying the delays and amplitudes of the early reflections and
the nature of the diffuse tail one could simulate different rooms.
The design of convincing reverberation algorithms is a skilled task, and
the difference between crude approaches and good ones is very noticeable.
Some audio workstations offer limited reverberation effects built into the
basic software package, but these often sound rather poor because of
Introduction to Digital Signal Processing
Delay
RMS detection
and time constants
Attack
time
Recovery
time
Log
X
Threshold and
slope control
Threshold
Antilog
Slope
the limited DSP power available (often processed on the computer’s own
CPU) and the crude algorithms involved. More convincing reverberation
processors are available which exist either as stand-alone devices or as
optional plug-ins for the workstation, having access to more DSP capacity
and tailor-made software.
Other simple effects can be introduced without much DSP capacity, such
as double-tracking and phasing/flanging effects. These often only involve
very simple delaying and recombination processes. Pitch shifting can also
be implemented digitally, and this involves processes similar to sample rate
conversion, as described below. High quality pitch shifting requires quite
considerable horsepower because of the number of calculations required.
Dynamics processing
Digital dynamics processing involves gain control that depends on the
instantaneous level of the audio signal. A simple block diagram of such a
device is shown in Figure 8.32. A side chain produces coefficients corresponding to the instantaneous gain change required, which are then used to
multiply the delayed audio samples. First, the r.m.s. level of the signal must
be determined, after which it needs to be converted to a logarithmic value
in order to determine the level change in decibels. Only samples above a
certain threshold level will be affected, so a constant factor must be added
to the values obtained, after which they are multiplied by a factor to represent the compression slope. The coefficient values are then antilogged to
produce linear coefficients by which the audio samples can be multiplied.
Sample rate conversion
Sample rate conversion is necessary whenever audio is to be transferred
between systems operating at different rates. The aim is to convert the
FIGURE 8.32
A simple digital dynamics
processing operation.
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CHAPTER 8: Digital Audio Principles
audio to the new rate without any change in pitch or addition of distortion
or noise. These days sample rate conversion can be a very high quality process, although it is never an entirely transparent process because it involves
modifying the sample values and timings. As with requantizing algorithms,
it is fairly common to encounter poorly implemented sample rate conversion on low cost digital audio workstations, often depending very much on
the specific software application rather than the hardware involved.
The easiest way to convert from one rate to another is by passing
through the analog domain and resampling at the new rate, but this may
introduce a small amount of extra noise. The most basic form of digital
rate conversion involves the translation of samples at one fixed rate to a
new fixed rate, related by a simple fractional ratio. Fractional-ratio conversion involves the mathematical calculation of samples at the new rate
based on the values of samples at the old rate. Digital filtering is used to
calculate the amplitudes of the new samples such that they are correct
based on the impulse response of original samples, after low-pass filtering
with an upper limit of the Nyquist frequency of the original sampling rate.
A clock rate common to both sample rates is used to control the interpolation process. Using this method, some output samples will coincide with
input samples, but only a limited number of possibilities exist for the interval between input and output samples.
If the input and output sampling rates have a variable or non-simple
relationship the above does not hold true, since output samples may
be required at any interval in between input samples. This requires an
interpolator with many more clock phases than for fractional-ratio conversion, the intention being to pick a clock phase that most closely corresponds to the desired output sample instant at which to calculate the
necessary coefficient. There will clearly be an error, which may be made
smaller by increasing the number of possible interpolator phases. The audible result of the timing error is equivalent to the effects of jitter on an audio
signal (see above), and should be minimized in design so that the effects of
sample rate conversion are below the noise floor of the signal resolution in
hand. If the input sampling rate is continuously varied (as it might be in
variable-speed searching or cueing) the position of interpolated samples
in relation to original samples must vary also. This requires real-time calculation of the filter phase.
Many workstations now include sample rate conversion as either a standard or optional feature, so that audio material recorded and edited at one
rate can be reproduced at another. It is important to ensure that the quality
of the sample rate conversion is high enough not to affect the sound quality of your recordings, and it should only be used if it cannot be avoided.
Pitch Shifting and Time Stretching
Poorly implemented applications sometimes omit to use correct low-pass
filtering to avoid aliasing, or incorporate very basic digital filters, resulting
in poor sound quality after rate conversion.
Sample rate conversion is also useful as a means of synchronizing an
external digital source to a standard sampling frequency reference, when it
is outside the range receivable by a workstation.
PITCH SHIFTING AND TIME STRETCHING
Pitch alteration is now a common feature of many digital audio processing systems. It can be used either to modify the musical pitch of notes in
order to create alternative versions, including harmony lines, or it can be
used to ‘correct’ the pitch of musical lines such as vocals that were not sung
in tune. The most basic way of doing this is to alter the sampling frequency
of the audio signal (slowing down the sampling frequency without doing
any sophisticated rate conversion will cause the perceived pitch to drop),
but this also changes the speed and duration, and the resulting sample data
is then at a non-standard sampling frequency. Resampling the original signal in the digital domain at a different frequency, followed by replay at the
original sampling frequency, is an alternative, but the speed will be similarly affected. Modern pitch alteration algorithms are usually much more
sophisticated than this and can alter the pitch without altering the speed.
Time stretching is a related application, and involves altering the duration
of a clip without altering its pitch. Pitch correction algorithms attempt to
identify the fundamental pitch of individual notes in a phrase and quantize
them to a fixed pitch scale in order to force melodies into tune. This can be
done with varying degrees of severity and musicality, leading to results anywhere on a range from crude effect to subtle correction of tuning.
Both effects can be achieved by transforming a signal into the frequency
domain, modifying it and resynthesizing it in the time domain. Techniques
based on phase vocoding or spectral modeling are sometimes used. Both
approaches succeed to some extent in enabling pitch and time information to be analyzed and modified independently, although with varying side
effects depending on the content of the signal and the parameters of the
processing. The signal is transformed to the frequency domain in overlapping blocks using a short time Fourier transform (STFT). It then becomes
possible to modify the signal in the frequency domain, for example by scaling certain spectral components, before performing an inverse transform to
return it to the time domain with modified pitch. Alternatively the original spectral components can be resynthesized with a new time scale at the
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CHAPTER 8: Digital Audio Principles
stage of the inverse transform in order to change the duration. In the time
domain, time stretch processing typically involves identifying the fundamental period of the wave and extracting or adding individual cycles with
crossfades, to shorten or lengthen the clip. It may also involve removing or
adding samples in silent gaps between notes or phrases. (The latter is particularly used in algorithms that attempt to fit overdubbed speech dynamically to a guide track, for movie sound applications.)
In general, both time and pitch-shifting only work successfully over a
limited range of 30% or so either way, although occasionally they can be
made to work over wider ranges depending on the nature of the signal and
the sophistication of the algorithm. One problem with simple pitch shifting
is the well-known ‘Pinky and Perky ’ effect that can be noticed when shifting
musical sounds (particularly voices) too far from their original frequency,
making them sound unnatural. This is because real musical sounds and
human voices have so-called formants, which are peaks and troughs in
the spectrum that are due to influences like resonances in the instrument
or vocal tract. These give a voice its unique character, for example. When
the pitch is shifted the formant structure can become shifted too, so that
the peaks and troughs are no longer in the right place in the frequency
spectrum for the voice in question. Some sophisticated pitch shifting algorithms therefore employ advanced signal processing methods that can identify the so-called spectral envelope of an instrument or voice (its pattern of
peaks and troughs in the frequency spectrum), and attempt to retain this
even when the pitch is shifted. In this way a voice can be made to sound
like the original singer even when shifted over quite a wide range.
AUDIO DATA REDUCTION
Conventional PCM audio has a high data rate, and there are many applications for which it would be an advantage to have a lower data rate without
much (or any) loss of sound quality. Sixteen bit linear PCM at a sampling
rate of 44.1 kHz (‘CD quality digital audio’) results in a data rate of about
700 kbit/s. For multimedia applications, broadcasting, communications and
some consumer purposes (e.g. streaming over the Internet) the data rate
may be reduced to a fraction of this with minimal effect on the perceived
sound quality. At very low rates the effect on sound quality is traded off
with the bit rate required. Simple techniques for reducing the data rate,
such as reducing the sampling rate or number of bits per sample would
have a very noticeable effect on sound quality, so most modern low bit rate
coding works by exploiting the phenomenon of auditory masking to ‘hide’
Audio Data Reduction
the increased noise resulting from bit rate reduction in parts of the audio
spectrum where it will hopefully be inaudible. There are a number of types
of low bit rate coding used in audio systems, working on similar principles,
and used for applications such as consumer disk and tape systems (e.g.
Sony ATRAC), digital cinema sound (e.g. Dolby Digital, Sony SDDS, DTS)
and multimedia applications (e.g. MPEG).
Why reduce the data rate?
Nothing is inherently wrong with linear PCM from a sound quality point of
view, indeed it is probably the best thing to use. The problem is simply that
the data rate is too high for a number of applications. Two channels of linear PCM require a rate of around 1.4 Mbit/s, whereas applications such as
Digital Audio Broadcasting (DAB) or Digital Radio need it to be more like
128 kbit/s (or perhaps lower for some applications) in order to fit sufficient
channels into the radio frequency spectrum – in other words more than ten
times less data per second. Some Internet streaming applications need it to
be even lower than this, with rates down in the low tens of kilobits per second for mobile communications.
The efficiency of mass storage media and data networks is related to
their data transfer rates. The more data can be moved per second, the
more audio channels may be handled simultaneously; the faster a disk can
be copied, the faster a sound file can be transmitted across the world. In
reducing the data rate that each audio channel demands, one also reduces
the requirement for such high specifications from storage media and networks, or alternatively one can obtain greater functionality from the same
specification. A network connection capable of handling eight channels of
linear PCM simultaneously could be made to handle, say, 48 channels of
data-reduced audio, without unduly affecting sound quality.
Although this sounds like magic and makes it seem as if there is no
point in continuing to use linear PCM, it must be appreciated that the data
reduction is achieved by throwing away data from the original audio signal.
The more data is thrown away the more likely it is that unwanted audible
effects will be noticed. The design aim of most of these systems is to try
to retain as much as possible of the sound quality whilst throwing away as
much data as possible, so it follows that one should always use the least
data reduction necessary, where there is a choice.
Lossless and lossy coding
There is an important distinction to be made between the type of data
reduction used in some computer applications and the approach used in
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CHAPTER 8: Digital Audio Principles
Original
data
Lossless
encoding
Storage or
transmission
Decoding
Original
data
Decoding
Approximation
to original data
(a)
Original
data
Lossy
encoding
Storage or
transmission
(b)
FIGURE 8.33 (a) In lossless coding the original data is reconstructed perfectly upon decoding,
resulting in no loss of information. (b) In lossy coding the decoded information is not the same as that
originally coded, but the coder is designed so that the effects of the process are minimal.
many audio coders. The distinction is really between ‘lossless’ coding and
coding which involves some loss of information (see Figure 8.33). It is
quite common to use data compression on computer files in order to fit
more information onto a given disk or tape, but such compression is usually lossless in that the original data is reconstructed bit for bit when the
file is decompressed. A number of tape backup devices for computers have
a compression facility for increasing the apparent capacity of the medium,
for example. Methods are used which exploit redundancy in the information, such as coding a string of 80 zeros by replacing them with a short
message stating the value of the following data and the number of bytes
involved. This is particularly relevant in single-frame bit-mapped picture
files where there may be considerable runs of black or white in each line of
a scan, where nothing in the image is changing. One may expect files compressed using off-the-shelf PC data compression applications to be reduced
to perhaps 25–50% of their original size, but it must be remembered that
they are often dealing with static data, and do not have to work in real
time. Also, it is not normally acceptable for decompressed computer data
to be anything but the original data.
It is possible to use lossless coding on audio signals. Lossless coding
allows the original PCM data to be reconstructed perfectly by the decoder
and is therefore ‘noiseless’ since there is no effect on audio quality. The data
reduction obtained using these methods ranges from nothing to about 2.5:1
and is variable depending on the program material. This is because audio
signals have an unpredictable content, do not make use of a standard limited
character set, and do not spend long periods of time in one binary state or
Audio Data Reduction
the other. Although it is possible to perform this reduction in real time,
the coding gains are not sufficient for many applications. Nonetheless, a
halving in the average audio data rate is certainly a useful saving. A form of
lossless data reduction known as Direct Stream Transfer (DST) can be used
for Super Audio CD in order to fit the required multichannel audio data
into the space available. A similar system was designed for DVD-Audio,
called MLP (Meridian Lossless Packing), which evolved into Dolby TrueHD,
used as the lossless coding scheme for the Blu-Ray disc.
‘Noisy ’ or lossy coding methods make possible a far greater degree of
data reduction, but require the designer and user to arrive at a compromise
between the degree of data reduction and potential effects on sound quality. Here data reduction is achieved by coding the signal less accurately
than in the original PCM format (using fewer bits per sample), thereby
increasing quantizing noise, but with the intention that increases in noise
will be ‘masked’ (made inaudible) by the signal. The original data is not
reconstructed perfectly on decoding. The success of such techniques therefore relies on being able to model the characteristics of the human hearing
process in order to predict the masking effect of the signal at any point in
time – hence the common term ‘perceptual coding’ for this approach. Using
detailed psychoacoustic models it is possible to code high quality audio at
rates under 100 kbit/s per channel with minimal effects on audio quality.
Higher data rates, such as 192 kbit/s, can be used to obtain an audio quality
that is demonstrably indistinguishable from the original PCM.
MPEG – an example of lossy coding
The following is a very brief overview of how one approach works, based
on the technology involved in the MPEG (Moving Pictures Expert Group)
standards.
As shown in Figure 8.34, the incoming digital audio signal is filtered
into a number of narrow frequency bands. Parallel to this a computer model
of the human hearing process (an auditory model) analyzes a short portion
of the audio signal (a few milliseconds). This analysis is used to determine
what parts of the audio spectrum will be masked, and to what degree, during that short time period. In bands where there is a strong signal, quantizing noise can be allowed to rise considerably without it being heard,
because one signal is very efficient at masking another lower level signal
in the same band as itself (see Figure 8.35). Provided that the noise is kept
below the masking threshold in each band it should be inaudible.
Blocks of audio samples in each narrow band are scaled (low-level
signals are amplified so that they use more of the most significant bits of the
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CHAPTER 8: Digital Audio Principles
FIGURE 8.34
Generalized block diagram
of a psychoacoustic low bit
rate coder.
Perceptual
model
Quantization
and scale
factors
Band
splitting
PCM
audio
FIGURE 8.35
Quantizing noise lying
under the masking
threshold will normally be
inaudible.
Framing
Data-reduced
bit stream
Masking tone
Normal threshold
Altered threshold
Noise
20
100
1 kHz
10 kHz
Frequency Hz
range) and the scaled samples are then reduced in resolution (requantized)
by reducing the number of bits available to represent each sample – a process that results in increased quantizing noise. The output of the auditory
model is used to control the requantizing process so that the sound quality remains as high as possible for a given bit rate. The greatest number of
bits is allocated to frequency bands where noise would be most audible, and
the fewest to those bands where the noise would be effectively masked
by the signal. Control information is sent along with the blocks of bit ratereduced samples to allow them to be reconstructed at the correct level and
resolution upon decoding.
The above process is repeated every few milliseconds, so that the masking model is constantly being updated to take account of changes in the
audio signal. Carefully implemented, such a process can result in a reduction of the original data rate to anything from about one-quarter to less
than one-tenth. A decoder uses the control information transmitted with
the bit rate-reduced samples to restore the samples to their correct level
and can determine how many bits were allocated to each frequency band by
Audio Data Reduction
Data reduced
frames
Frame
unpacking
Reconstruction
of subband
samples
PCM audio at
Reconstruction original rate
of PCM audio
Table 8.2
MPEG-1 layers
Layer
Complexity
Min. delay
Bit rate range
Target
1
2
3
Low
Moderate
High
19 ms
35 ms
59 ms
32–448 kbit/s
32–384 kbit/s*•
32–320 kbit/s
192 kbit/s
128 kbit/s
64 kbit/s
•*
In Layer 2, bit rates of 224 kbit/s and above are for stereo modes only
the encoder, reconstructing linear PCM samples and then recombining the
frequency bands to form a single output (see Figure 8.36). A decoder can be
much less complex, and therefore cheaper, than an encoder, because it does
not need to contain the auditory model.
A standard known as MPEG-1, published by the International Standards
Organization (ISO 11172-3), defines a number of ‘layers’ of complexity for
low bit rate audio coders as shown in Table 8.2. Each of the layers can be
operated at any of the bit rates within the ranges shown (although some
of the higher rates are intended for stereo modes) and the user must make
appropriate decisions about what sound quality is appropriate for each
application. The lower the data rate, the lower the sound quality that
will be obtained. At high data rates the encoding–decoding process has
been judged by many to be audibly ‘transparent’ – in other words listeners
cannot detect that the coded and decoded signal is different from the original input. The target bit rates were for ‘transparent’ coding.
‘MP3’ will be for many people the name associated with downloading
music files from the Internet. The term MP3 has caused some confusion;
it is short for MPEG-1 Layer 3, but MP3 has virtually become a generic
term for the system used for receiving compressed audio from the Internet.
There is also MPEG-2 which can handle multichannel surround, and further developments in this and later systems will be briefly touched upon.
MPEG-2 BC (Backwards Compatible with MPEG-1) additionally supports sampling frequencies from 16 kHz to 22.05 kHz and 24 kHz at bit
rates from 32 to 256 kbit/s for Layer 1. For Layers 2 and 3, bit rates are from
8 to 160 kbit/s. Developments, intended to supersede MPEG-2 BC, have
included MPEG-2 AAC (Advanced Audio Coding). This defines a standard
for multichannel coding of up to 48 channels, with sampling rates from
FIGURE 8.36
Generalized block diagram
of an MPEG-Audio decoder.
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CHAPTER 8: Digital Audio Principles
8 kHz to 96 kHz. It also incorporates a Modified Discrete Cosine transform
system as used in the MiniDisc coding format (ATRAC). MPEG-2AAC was
not, however, designed to be backwards compatible with MPEG-1.
MPEG-4 ‘natural audio coding’ is based on the standards outlined for
MPEG-2 AAC; it includes further coding techniques for reducing transmission bandwidth and it can scale the bit rate according to the complexity
of the decoder. This is used in Apple’s iPod, for example. There are also
intermediate levels of parametric representation in MPEG-4 such as used
in speech coding, whereby speed and pitch of basic signals can be altered
over time. One has access to a variety of methods of representing sound
at different levels of abstraction and complexity, all the way from natural
audio coding (lowest level of abstraction), through parametric coding systems based on speech synthesis and low-level parameter modification, to
fully synthetic audio objects.
When audio signals are described in the form of ‘objects’ and ‘scenes’,
it requires that they be rendered or synthesized by a suitable decoder.
Structured Audio (SA) in MPEG-4 enables synthetic sound sources to be
represented and controlled at very low bit rates (less than 1 kbit/s). An SA
decoder can synthesize music and sound effects. SAOL (Structured Audio
Orchestra Language), as used in MPEG-4, was developed at MIT and is an
evolution of CSound (a synthesis language used widely in the electroacoustic music and academic communities). It enables ‘instruments’ and ‘scores’
to be downloaded. The instruments define the parameters of a number of
sound sources that are to be rendered by synthesis (e.g. FM, wavetable,
granular, additive) and the ‘score’ is a list of control information that governs what those instruments play and when (represented in the SASL or
Structured Audio Score Language format). This is rather like a more refined
version of the established MIDI control protocol, and indeed MIDI can be
used if required for basic music performance control. This is discussed further in Chapter 14.
Sound scenes, as distinct from sound objects, are usually made up of
two elements – that is the sound objects and the environment within which
they are located. Both elements are integrated within one part of MPEG-4.
This part of MPEG-4 uses so-called BIFS (Binary Format for Scenes) for
describing the composition of scenes (both visual and audio). The objects
are known as nodes and are based on VRML (virtual reality modeling language). So-called Audio BIFS can be post-processed and represents parametric descriptions of sound objects. Advanced Audio BIFS also enables virtual
environments to be described in the form of perceptual room acoustics
parameters, including positioning and directivity of sound objects. MPEG4 audio scene description distinguishes between physical and perceptual
Audio Data Reduction
representation of scenes, rather like the low- and high-level description
information mentioned above.
Other data-reduced formats
Dolby Digital or AC-3 encoding was developed as a means of delivering
5.1-channel surround to cinemas or the home without the need for analog
matrix encoding. The AC-3 coding algorithm can be used for a wide range
of different audio signal configurations and bit rates from 32 kbit/s for a
single mono channel up to 640 kbit/s for surround signals. It is used widely
for the distribution of digital sound tracks on 35 mm movie films, the data
being stored optically in the space between the sprocket holes on the film.
It is sufficient to say here that the process involves a number of techniques by which the data representing audio from the source channels is
transformed into the frequency domain and requantized to a lower resolution, relying on the masking characteristics of the human hearing process to hide the increased quantizing noise that results from this process.
A common bit pool is used so that channels requiring higher data rates
than others can trade their bit rate requirements provided that the overall
total bit rate does not exceed the constant rate specified.
Aside from the representation of surround sound in a compact digital
form, Dolby Digital includes a variety of operational features that enhance
system flexibility and help adapt replay to a variety of consumer situations.
These include dialog normalization (‘dialnorm’) and the option to include
dynamic range control information alongside the audio data for use in environments where background noise prevents the full dynamic range of the
source material being heard. Downmix control information can also be carried alongside the audio data in order that a two-channel version of the
surround sound material can be reconstructed in the decoder. As a rule,
Dolby Digital data is stored or transmitted with the highest number of
channels needed for the end product to be represented and any compatible downmixes are created in the decoder. This differs from some other
systems where a two-channel downmix is carried alongside the surround
information.
The DTS (Digital Theater Systems) ‘Coherent Acoustics’ system is
another digital signal coding format that can be used to deliver surround
sound in consumer or professional applications, using low bit rate coding techniques to reduce the data rate of the audio information. The
DTS system can accommodate a wide range of bit rates from 32 kbit/s up
to 4.096 Mbit/s (somewhat higher than Dolby Digital), with up to eight
source channels and with sampling rates up to 192 kHz. Variable bit rate
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and lossless coding are also optional. Downmixing and dynamic range control options are provided in the system. Because the maximum data rate is
typically somewhat higher than that of Dolby Digital or MPEG, a greater
margin can be engineered between the signal and any artefacts of low bit
rate coding, leading to potentially higher sound quality. Such judgments,
though, are obviously up to the individual and it is impossible to make
blanket statements about comparative sound quality between systems.
SDDS stands for Sony Dynamic Digital Sound, and was the third of the
main competing formats for digital film sound. Using Sony’s ATRAC data
reduction system (also used on MiniDiscs), it too encodes audio data with
a substantial saving in bit rate compared with the original PCM (about 5:1
compression).
RECOMMENDED FURTHER READING
Bosi, M., Goldberg, R., 2003. Introduction to Digital Audio Coding and Standards.
Kluwer Academic Publishers.
Pohlmann, K., 2005. Principles of Digital Audio. McGraw-Hill Professional.
Rumsey, F., 2004. Desktop Audio Technology. Focal Press.
Watkinson, J., 2001. The Art of Digital Audio. Focal Press.
Zölzer, U., 2008. Digital Audio Signal Processing. Wiley Blackwell.
CHAPTER 9
Digital Recording, Editing and
Mastering Systems
CH A P T E R C O N TE N T S
Digital Tape Recording
Background to digital tape recording
Channel coding for dedicated tape formats
Error correction
Digital tape formats
Editing digital tape recordings
Mass Storage-based Systems
Magnetic hard disks
Optical discs
Memory cards
Recording audio on to mass storage media
Media formatting
Audio Processing for Computer Workstations
Introduction
DSP cards
Host-based audio processing
Integrated sound cards
Mass Storage-based Editing System Principles
Introduction
Sound files and sound segments
Edit point handling
Crossfading
Editing modes
Simulation of ‘reel-rocking’
Editing Software
Mastering and Restoration
Software
Level control in mastering
Preparing for and Understanding Release Media
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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This chapter describes digital audio recording systems and the principles of
digital audio editing and mastering.
DIGITAL TAPE RECORDING
Although it is still possible to find examples of dedicated digital tape recording formats in use, they have largely been superseded by recording systems
that use computer mass storage media. The economies of scale of the
computer industry have made data storage relatively cheap and there is no
longer a strong jusitification for systems dedicated to audio purposes. Tape
has a relatively slow access time, because it is a linear storage medium.
However, a dedicated tape format can easily be interchanged between
recorders, provided that another machine operating to the same standard
can be found. Computer mass storage media, on the other hand, come in
a very wide variety of sizes and formats, and there are numerous levels at
which compatibility must exist between systems before interchange can
take place. This matter is discussed in the next chapter.
Background to digital tape recording
When commercial digital audio recording systems were first introduced
in the 1970s and early 1980s it was necessary to employ recorders with
sufficient bandwidth for the high data rates involved (a machine capable of
handling bandwidths of a few megahertz was required). Analog audio tape
recorders were out of the question because their bandwidths extended only
up to around 35 kHz at best, so video tape recorders (VTRs) were often utilized because of their wide recording bandwidth. PCM adaptors converted
digital audio data into a waveform which resembled a television waveform,
suitable for recording on to a VTR. The Denon company of Japan developed such a system in partnership with the NHK broadcasting organization
and they released the world’s first PCM recording onto LP in 1971. In the
early 1980s, devices such as Sony’s PCM-F1 became available at modest
prices, allowing 16 bit, 44.1 kHz digital audio to be recorded on to a consumer VTR, resulting in widespread proliferation of stereo digital recording. Dedicated open-reel digital recorders using stationary heads were also
developed (see Fact File 9.1). High-density tape formulations were then
manufactured for digital use, and this, combined with new channel codes
(see below), improvements in error correction and better head design, led
to the use of a relatively low number of tracks per channel, or even singletrack recording of a given digital signal, combined with playing speeds of
15 or 30 inches per second. Dedicated rotary-head systems, not based on a
VTR, were also developed – the R-DAT format being the most well known.
Digital Tape Recording
FA C T F I L E 9 . 1 R O TA RY A N D S TATI ONARY HEADS
There are two fundamental mechanisms for the recording of digital audio on tape, one which uses a relatively
low linear tape speed and a quickly rotating head, and
one which uses a fast linear tape speed and a stationary head. In the rotary-head system the head either
describes tracks almost perpendicular to the direction
of tape travel, or it describes tracks which are almost in
the same plane as the tape travel. The former is known
as transverse scanning, and the latter is known as helical scanning, as shown in (a). Transverse scanning uses
more tape when compared with helical scanning. It is not
common for digital tape recording to use the transverse
scanning method. The reason for using a rotary head is to
achieve a high head-to-tape speed, since it is this which
governs the available bandwidth. Rotary-head recordings
cannot easily be splice-edited because of the track pattern, but they can be electronically edited using at least
two machines.
Stationary heads allow the design of tape machines
that are very similar in many respects to analog transports. With stationary-head recording it is possible to
record a number of narrow tracks in parallel across the
width of the tape, as shown in (b). Tape speed can be
traded off against the number of parallel tracks used for
each audio channel, since the required data rate can be
made up by a combination of recordings made on sepa-
rate tracks. This approach was used in the DASH format,
where the tape speed could be 30 ips (76 cm s⫺1) using
one track per channel, 15 ips using two tracks per channel, or 7.5 ips using four tracks per channel.
(a)
Typical track pattern
(b)
Head is stationary
Tape travel
Track pattern
Digital recording tape is thinner (27.5 microns) than that used for analog recordings; long playing times can be accommodated on a reel, but also
thin tape contacts the machine’s heads more intimately than does standard 50 micron thickness tape which tends to be stiffer. Intimate contact is
essential for reliable recording and replay of such a densely packed and high
bandwidth signal.
Channel coding for dedicated tape formats
Since ‘raw ’ binary data is normally unsuitable for recording directly by dedicated digital recording systems, a ‘channel code’ is used which matches the
data to the characteristics of the recording system, uses storage space efficiently, and makes the data easy to recover on replay. A wide range of channel codes exists, each with characteristics designed for a specific purpose.
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The channel code converts a pattern of binary data into a different pattern
of transitions in the recording or transmission medium. It is another stage
of modulation, in effect. Thus the pattern of bumps in the optical surface of
a CD bears little resemblance to the original audio data, and the pattern of
magnetic flux transitions on a DAT cassette would be similarly different.
Given the correct code book, one could work out what audio data was represented by a given pattern from either of these systems.
Many channel codes are designed for a low DC content (in other words,
the data is coded so as to spend, on average, half of the time in one state
and half in the other) in cases where signals must be coupled by transformers (see ‘Transformers’, Chapter 12), and others may be designed for narrow
bandwidth or a limited high-frequency content. Certain codes are designed
specifically for very high-density recording, and may have a low clock content with the possibility for long runs in one binary state or the other without a transition. Channel coding involves the incorporation of the data to
be recorded with a clock signal, such that there is a sufficient clock content
to allow the data and clock to be recovered on replay (see Fact File 9.2).
Channel codes vary as to their robustness in the face of distortion, noise
and timing errors in the recording channel.
Some examples of channel codes used in audio systems are shown in
Figure 9.1. FM is the simplest, being an example of binary frequency modulation. It is otherwise known as ‘bi-phase mark’, one of the Manchester
codes, and is the channel code used by SMPTE/EBU timecode (see Chapter
15). MFM and Miller-squared are more efficient in terms of recording density. MFM is more efficient than FM because it eliminates the transitions
between successive ones, only leaving them between successive zeros.
Miller-squared eliminates the DC content present in MFM by removing the
transition for the last one in an even number of successive ones.
Group codes, such as that used in the Compact Disc and R-DAT, involve
the coding of patterns of bits from the original audio data into new codes
with more suitable characteristics, using a look-up table or ‘code book’ to
keep track of the relationship between recorded and original codes. This has
clear parallels with coding as used in intelligence operations, in which the
recipient of a message requires the code book to be able to understand the
message. CD uses a method known as 8-to-14 modulation, in which 16 bit
audio sample words are each split into two 8 bit words, after which a code
book is used to generate a new 14 bit word for each of the 256 possible combinations of 8 bits. Since there are many more words possible with 14 bits
than with 8, it is possible to choose those which have appropriate characteristics for the CD recording channel. In this case, it is those words which
have no more than 11 consecutive bits in the same state, and no less than
Digital Tape Recording
FA C T F I L E 9 . 2 D ATA R E C O V E RY
Channel-coded data must be decoded on replay, but first
the audio data must be separated from the clock information which was combined with it before recording. This
process is known as data and sync separation, as shown
in (a).
It is normal to use a phase-locked loop for the purpose
of regenerating the clock signal from the replayed data, as
shown in (b), this being based around a voltage-controlled
oscillator (VCO) which runs at some multiple of the offtape clock frequency. A phase comparator compares the
relative phases of the divided VCO output and the clock
data off tape, producing a voltage proportional to the error
which controls the frequency of the VCO. With suitable
damping, the phase-locked oscillator will ‘flywheel’ over
short losses or irregularities of the off-tape clock.
Recorded data is usually interspersed with synchronizing patterns in order to give the PLL in the data separator a
regular reference in the absence of regular clock data from
the encoded audio signal, since many channel codes have
long runs without a transition. Even if the off-tape data and
clock have timing irregularities, such as might manifest
themselves as ‘wow’ and ‘flutter’ in analog reproducers (see
Chapter 18), these can be removed in digital systems. The
erratic data (from tape or disk, for example) is written into
a short-term solid state memory (RAM) and read out again
a fraction of a second later under control of a crystal clock
(which has an exceptionally stable frequency), as shown in
(c). Provided that the average rate of input to the buffer is
the same as the average rate of output, and the buffer is of
sufficient size to soak up short-term irregularities in timing,
the buffer will not overflow or become empty.
Clock
(a) Channelcoded data
(c) Erratic input
data
Data
and
clock
separator
Phase
comparator
Erratic write
clock
Control voltage
Voltagecontrolled
oscillator
Clock at n x f
Filter
÷n
Clock at f
Stable output
data
RAM buffer
Data
101101
(b) Channelcoded signal
with clock at
frequency f
Delay
Clock divider
three. This limits the bandwidth of the recorded data, and makes it suitable
for the optical pick-up process, whilst retaining the necessary clock content.
Error correction
There are two stages to the error correction process used in digital tape
recording systems. First, the error must be detected, and then it must be
corrected. If it cannot be corrected then it must be concealed. In order for
Stable read
clock
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the error to be detected it is necessary to build in certain protection mechanisms.
Two principal types of error exist: the burst error and the
FM
random error. Burst errors result in the loss of many successive
samples and may be due to major momentary signal loss, such
MFM
as might occur at a tape drop-out or at an instant of impulsive
interference such as an electrical spike induced in a cable or piece
2
Miller
of dirt on the surface of a CD. Burst error correction capability is
usually quoted as the number of consecutive samples which may
FIGURE 9.1 Examples of three
be corrected perfectly. Random errors result in the loss of single
channel codes used in digital recording.
Miller-squared is the most efficient of those
samples in randomly located positions, and are more likely to be
shown since it involves the smallest number the result of noise or poor signal quality. Random error rates are
of transitions for the given data sequence.
normally quoted as an average rate, for example 1 in 106. Error
correction systems must be able to cope with the occurrence of
Original sample order
both burst and random errors in close proximity.
1 2 3 4 5 6 7 8 9 10 11 12 13
Audio data is normally interleaved before recording, which
means that the order of samples is shuffled (as shown conceptuInterleaved sample order
ally in Figure 9.2). Samples that had been adjacent in real time
3 7 13 9 4 10 1 5 11 8 2 6 12
are now separated from each other on the tape. The benefit of
this is that a burst error, which destroys consecutive samples
Burst error destroys three samples
on tape, will result in a collection of single-sample errors in
3 7 13 9 4
11 8 2 6 12
between good samples when the data is deinterleaved, allowing for the error to be concealed. A common process, associated
Consequent random errors in
with interleaving, is the separation of odd and even samples by a
de-interleaved data
delay. The greater the interleave delay, the longer the burst error
2 3 4
6 7 8 9
11 12 13
that can be handled. A common example of this is found in the
DASH tape format (an open-reel digital recording format), and
FIGURE 9.2 Interleaving is used in
involves delaying odd samples so that they are separated from
digital recording and broadcasting systems
to rearrange the original order of samples for even samples by 2448 samples, as well as reordering groups of
storage or transmission. This can have the
odd and even samples within themselves.
effect of converting burst errors into random
Redundant data is also added before recording. Redundancy,
errors when the samples are deinterleaved.
in simple terms, involves the recording of data in more than one
form or place. A simple example of the use of redundancy is found in the
twin-DASH format, in which all audio data is recorded twice. On a second
pair of tracks (handling the duplicated data), the odd–even sequence of data is
reversed to become even–odd. First, this results in double protection against
errors, and second, it allows for perfect correction at a splice, since two burst
errors will be produced by the splice, one in each set of tracks. Because of the
reversed odd–even order in the second set of tracks, uncorrupted odd data
can be used from one set of tracks, and uncorrupted even data from the other
set, obviating the need for interpolation (see Fact File 9.3).
Data
1
0
1
1
0
1
Digital Tape Recording
FA C T F I L E 9 . 3 E R R O R H A N D L ING
True Correction
Up to a certain random error rate or burst error duration
an error correction system will be able to reconstitute
erroneous samples perfectly. Such corrected samples are
indistinguishable from the originals, and sound quality will
not be affected. Such errors are often signaled by green
lights showing ‘CRC’ failure or ‘Parity’ failure.
Interpolation
When the error rate exceeds the limits for perfect correction, an error correction system may move to a process
involving interpolation between good samples to arrive at
a value for a missing sample (as shown in the diagram).
The interpolated value is the mathematical average of the
foregoing and succeeding samples, which may or may
not be correct. This process is also known as concealment or averaging, and the audible effect is not unpleasant, although it will result in a temporary reduction in
audio bandwidth. Interpolation is usually signaled by an
orange indicator to show that the error condition is fairly
serious. In most cases the duration of such concealment
is very short, but prolonged bouts of concealment should
be viewed warily, since sound quality will be affected. This
will usually point to a problem such as dirty heads or a
misaligned transport, and action should be taken.
a few samples before muting. Hold is normally indicated
by a red light.
Mute
When an error correction system is completely overwhelmed it will usually effect a mute on the audio output
of the system. The duration of this mute may be varied by
the user in some systems. The alternative to muting is to
hear the output, regardless of the error. Depending on the
severity of the error, it may sound like a small ‘spit’, click,
or even a more severe breakup of the sound. In some
cases this may be preferable to muting.
Original sample amplitudes
Sample missing due to error
Hold
In extreme cases, where even interpolation is impossible (when there are not two good samples either side of
the bad one), a system may ‘hold’. In other words, it will
repeat the last correct sample value. The audible effect
of this will not be marked in isolated cases, but is still a
severe condition. Most systems will not hold for more than
Mathematically-interpolated value
inserted
Cyclic redundancy check (CRC) codes, calculated from the original
data and recorded along with that data, are used in many systems to detect
the presence and position of errors on replay. Complex mathematical procedures are also used to form codewords from audio data which allow
for both burst and random errors to be corrected perfectly up to a given
limit. Reed–Solomon encoding is another powerful system which is used to
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protect digital recordings against errors, but it is beyond the scope of this
book to cover these codes in detail.
Digital tape formats
There have been a number of commercial recording formats over the last
20 years, and only a brief summary will be given here of the most common.
Sony’s PCM-1610 and PCM-1630 adaptors dominated the CD-mastering
market for a number of years, although by today’s standards they used a fairly
basic recording format and relied on 60 Hz/525 line U-matic cassette VTRs
(Figure 9.3). The system operated at a sampling rate of 44.1 kHz and used
16 bit quantization, being designed specifically for the making of tapes to
be turned into CDs. Recordings made in this format could be electronically
edited using the Sony DAE3000 editing system, and the playing time of tapes
ran up to 75 minutes using a tape specially developed for digital audio use.
The R-DAT or DAT format was a small stereo, rotary-head, cassettebased format offering a range of sampling rates and recording times, including the professional rates of 44.1 and 48 kHz. Originally, consumer machines
operated at 48 kHz to avoid the possibility for digital copying of CDs, but
professional versions became available which
would record at either 44.1 or 48 kHz. Consumer
machines would record at 44.1 kHz, but usually
only via the digital inputs. DAT was a 16 bit format, but had a non-linearly encoded long-play
mode as well, sampled at 32 kHz. Truly professional designs offering editing facilities, external sync and IEC-standard timecode were also
developed. The format became exceptionally
popular with professionals owing to its low cost,
high performance, portability and convenience.
Various non-standard modifications were introFIGURE 9.3 Sony DMR-4000 digital master recorder.
duced, including a 96 kHz sampling rate machine
(Courtesy of Sony Broadcast and Professional Europe.)
and adaptors enabling the storage of 20 bit audio
on such a high sampling rate machine (sacrificing the high sampling rate for more bits). The
IEC timecode standard for R-DAT was devised
in 1990. It allowed for SMPTE/EBU timecode of
any frame rate to be converted into the internal
DAT ‘running-time’ code, and then converted
back into any SMPTE/EBU frame rate on replay.
FIGURE 9.4 Sony PCM-7030 professional DAT machine.
A typical machine is pictured in Figure 9.4.
(Courtesy of Sony Broadcast and Professional Europe.)
Digital Tape Recording
The Nagra-D recorder (Figure 9.5) was designed as a digital replacement
for the world-famous Nagra analog recorders, and as such was intended for
professional use in field recording and studios. The format was designed
to have considerable commonality with the audio format used in D1- and
D2-format digital VTRs, having rotary heads, although it used open reels
for operational convenience. Allowing for 20–24 bits of audio resolution,
the Nagra-D format was appropriate for use with high-resolution convertors. The error correction and recording density used in this format were
designed to make recordings exceptionally robust, and recording time could
be up to 6 hours on a 7 inch (18 cm) reel, in two-track mode. The format
was also designed for operation in a four-track mode at twice the stereo
tape speed, such that in stereo the tape travels at 4.75 cm s⫺1, and in four
track at 9.525 cm s⫺1.
The DASH (Digital Audio Stationary Head) format consisted of a whole
family of open-reel stationary-head recording formats from two tracks
up to 48 tracks. DASH-format machines operated at 44.1 kHz or 48 kHz
rates (and sometimes optionally at 44.056 kHz), and they allowed varispeed ⫾12.5%. They were designed to allow gapless punch-in and punchout, splice editing, electronic editing and easy synchronization. Multitrack
DASH machines (an example is shown in Figure 9.6) gained wide acceptance in studios, but the stereo machines did not. Later developments
resulted in DASH multitracks capable of storing 24 bit audio instead of the
original 16 bits.
Subsequently budget modular multitrack formats were introduced. Most
of these were based on eight-track cassettes using rotary head transports
borrowed from consumer video technology.
The most widely used were the DA-88 format
(based on Hi-8 cassettes) and the ADAT format
(based on VHS cassettes). These offered most of
the features of open-reel machines and a number of them could be synchronized to expand
the channel capacity. An example is shown in
Figure 9.7.
Editing digital tape recordings
Razor blade cut-and-splice editing was possible
on open-reel digital formats, and the analog cue
tracks were monitored during these operations.
A 90° butt joint was used for the splice editing of digital tape. The discontinuity in the
FIGURE 9.5 Nagra-D open-reel digital tape recorder.
(Courtesy of Sound PR.)
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FIGURE 9.6 An open-reel digital multitrack recorder: the
Sony PCM-3348. (Courtesy of Sony Broadcast and Professional
Europe.)
FIGURE 9.7 A modular digital multitrack machine, Sony
PCM-800. (Courtesy of Sony Broadcast and Professional Europe.)
data stream caused by the splice would cause
complete momentary drop-out of the digital signal if no further action were taken, so circuits
were incorporated that sensed the splice and
performed an electronic crossfade from one side
of the splice to the other, with error concealment to minimize the audibility of the splice. It
was normally advised that a 0.5 mm gap should
be left at the splice so that its presence would
easily be detected by the crossfade circuitry. The
thin tape could easily be damaged during the
cut-and-splice edit procedure and this method
failed to gain an enthusiastic following, despite
its having been the norm in the analog world.
Electronic editing was far more desirable, and
was the usual method.
Electronic editing normally required the
use of two machines plus a control unit, as
shown in the example in Figure 9.8. A technique was employed whereby a finished master
tape was assembled from source takes on player
machines. This was a relatively slow process,
as it involved real-time copying of audio from
one machine to another, and modifications to
Controller
Player
(original recorded material)
Recorder
(master tape)
Selected takes copied from player to
recorder in appropriate order
FIGURE 9.8 In electronic tape copy editing selected takes are copied in sequence from player to
recorder with appropriate crossfades at joins.
Mass Storage-Based Systems
the finished master were difficult. The digital editor could often store several seconds of program in its memory and this could be replayed at normal
speed or under the control of a search knob which enabled very slow to-andfro searches to be performed in the manner of rock and roll editing on an
analog machine. Edits could be rehearsed prior to execution. When satisfactory edit points had been determined the two machines were synchronized
using timecode, and the record machine switched to drop in the new section
of the recording from the replay machine at the chosen moment. Here a
crossfade was introduced between old and new material to smooth the join.
The original source tape was left unaltered.
MASS STORAGE-BASED SYSTEMS
Once audio is in a digital form it can be handled by a computer, like any
other data. The only real difference is that audio requires a high sustained
data rate, substantial processing power and large amounts of storage
compared with more basic data such as text. The following is an introduction to some of the technology associated with computer-based audio
workstations and audio recording using computer mass storage media such
as hard disks. More detail will be found in Desktop Audio Technology,
as detailed in the ‘Recommended further reading’ list. The MIDI-based
aspects of such systems are covered in Chapter 14.
Magnetic hard disks
Magnetic hard disk drives are probably the most common form of mass
storage. They have the advantage of being random-access systems – in
other words any data can be accessed at random and with only a short
delay. There exist both removable and fixed media disk drives, but in
almost all cases the fixed media drives have a higher performance than
removable media drives. This is because the design tolerances can be made
much finer when the drive does not have to cope with removable media,
allowing higher data storage densities to be achieved. Some disk drives have
completely removable drive cartridges containing the surfaces and mechanism, enabling hard disk drives to be swapped between systems for easy
project management (an example is shown in Figure 9.9).
The general structure of a hard disk drive is shown in Figure 9.10. It consists of a motor connected to a drive mechanism that causes one or more
disk surfaces to rotate at anything from a few hundred to many thousands
of revolutions per minute. This rotation may either remain constant or may
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stop and start, and it may either be at a constant rate or a variable rate, depending on the
drive. One or more heads are mounted on a
positioning mechanism which can move the
head across the surface of the disk to access
FIGURE 9.9 A typical removable disk drive system allowing
particular points, under the control of hardware
multiple drives to be inserted or removed from the chassis at will.
and software called a disk controller. The heads
Frame housing multiple removable drives. (Courtesy of Glyph
Technologies Inc.)
read data from and write data to the disk surface
by whatever means the drive employs.
The disk surface is normally divided up
Disk surface
Common spindle
into tracks and sectors, not physically but by
means of ‘soft’ formatting (see Figure 9.11).
Low-level formatting places logical markers,
which indicate block boundaries, amongst
other processes. On most hard disks the tracks
are arranged as a series of concentric rings, but
with some optical discs there is a continuous
spiral track.
Disk drives look after their own channel
coding, error detection and correction so there
is no need for system designers to devise dediRead/write head
cated audio processes for disk-based recordHead positioner
FIGURE 9.10 The general mechanical structure of a disk drive. ing systems. The formatted capacity of a disk
drive is all available for the storage of ‘raw ’
audio data, with no additional overhead required
for redundancy and error checking codes. ‘Bad
blocks’ are mapped out during the formatting of a
disk, and not used for data storage. If a disk drive
detects an error when reading a block of data it will
attempt to read it again. If this fails then an error is
normally generated and the file cannot be accessed,
requiring the user to resort to one of the many file
recovery packages on the market. Disk-based audio
systems do not resort to error interpolation or sample hold operations, unlike tape recorders. Replay is
normally either correct or not possible.
Sector
RAID arrays enable disk drives to be
combined in various ways as described in Fact
File 9.4.
FIGURE 9.11 Disk formatting divides the storage area
into tracks and sectors.
Mass Storage-Based Systems
FA C T F I L E 9 . 4 R A ID A R R AYS
Hard disk drives can be combined in various ways to
improve either data integrity or data throughput. RAID stands
for Redundant Array of Inexpensive Disks, and is a means of
linking ordinary disk drives under one controller so that they
form an array of data storage space. A RAID array can be
treated as a single volume by a host computer. There are a
number of levels of RAID array, each of which is designed
for a slightly different purpose, as summarized in the table.
RAID
level
Features
0
Data blocks split alternately between a pair of disks, but no redundancy so actually less reliable than a
single disk. Transfer rate is higher than a single disk. Can improve access times by intelligent controller
positioning of heads so that next block is ready more quickly
Offers disk mirroring. Data from one disk is automatically duplicated on another. A form of real-time backup
Uses bit interleaving to spread the bits of each data word across the disks, so that, say, eight disks each
hold one bit of each word, with additional disks carrying error protection data. Non-synchronous head
positioning.
Slow to read data, and designed for mainframe computers
Similar to level 2, but synchronizes heads on all drives, and ensures that only one drive is used for error
protection data. Allows high-speed data transfer, because of multiple disks in parallel. Cannot perform
simultaneous read and write operations
Writes whole blocks sequentially to each drive in turn, using one dedicated error protection drive. Allows
multiple read operations but only single write operations
As level 4 but splits error protection between drives, avoiding the need for a dedicated check drive. Allows
multiple simultaneous reads and writes
As level 5 but incorporates RAM caches for higher performance
1
2
3
4
5
6
Optical discs
There are a number of families of optical disc drive that have differing operational and technical characteristics, although they share the universal benefit of removable media. They are all written and read using a laser, which
is a highly focused beam of coherent light, although the method by which
the data is actually stored varies from type to type. Optical discs are sometimes enclosed in a plastic cartridge that protects the disc from damage,
dust and fingerprints, and they have the advantage that the pickup never
touches the disc surface making them immune from the ‘head crashes’ that
can affect magnetic hard disks.
Compatibility between different optical discs and drives is something of
a minefield because the method of formatting and the read/write mechanism may differ. The most obvious differences lie in the erasable or nonerasable nature of the discs and the method by which data is written to
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and read from the disc, but there are also physical sizes and the presence or
lack of a cartridge to consider. Drives tend to split into two distinct families from a compatibility point of view: those that handle CD/DVD formats
and those that handle magneto-optical (M-O) and other cartridge-type ISO
standard disc formats. The latter may be considered more suitable for ‘professional purposes’ whereas the former are often encountered in consumer
equipment.
WORM discs (for example, the cartridges that were used quite widely
for archiving in the late 1980s and 1990s) may only be written once by the
user, after which the recording is permanent (a CD-R is therefore a type of
WORM disc). Other types of optical discs can be written numerous times,
either requiring pre-erasure or using direct overwrite methods (where new
data is simply written on top of old, erasing it in the process). The read/
write process of most current rewritable discs is typically ‘phase change’ or
‘magneto-optical’. The CD-RW is an example of a rewritable disc that now
uses direct overwrite principles.
The speed of some optical drives approaches that of a slow hard disk,
which makes it possible to use them as an alternative form of primary storage, capable of servicing a number of audio channels.
Memory cards
Increasing use is also made in audio systems of small flash memory cards,
particularly in portable recorders. These cards are capable of storing many
gigabytes of data on a solid state chip with fast access time, and they have
no moving parts which makes them relatively robust. Additionally they
have the benefit of being removable, which makes them suitable for transfer of some projects between systems, although the capacity and speed
limitations still make disks the medium of choice for large professional
projects. Such memory cards come in a variety of formats such as Compact
Flash (CF), Secure Digital (SD) and Memory Stick, and card readers can be
purchased that will read multiple types. There is a limit to the number of
times such devices can be rewritten, which is likely to be lower than that
for a typical magnetic disk drive.
Recording audio on to mass storage media
Mass storage media need to offer at least a minimum level of performance
capable of handling the data rates and capacities associated with digital audio, as described in Fact File 9.5. The discontinuous ‘bursty ’ nature
of recording onto such media usually requires the use of a buffer RAM
(Random Access Memory) during replay, which accepts this interrupted
Mass Storage-Based Systems
FA C T F I L E 9 . 5 S TO R A G E R E Q U I REM ENTS OF DI GI TAL AUDI O
The table shows the data rates required to support a single
channel of digital audio at various resolutions. Media to be
used as primary storage would need to be able to sustain
data transfer at a number of times these rates to be useful for multimedia workstations. The table also shows the
number of megabytes of storage required per minute of
audio, showing that the capacity needed for audio purposes
is considerably greater than that required for text or simple
graphics applications. Storage requirements increase pro
rata with the number of audio channels to be handled.
Storage systems may use removable media but many
have fixed media. It is advantageous to have removable
media for audio purposes because it allows different jobs
to be kept on different media and exchanged at will, but
unfortunately the highest performance is still obtainable from storage systems with fixed media. Although
the performance of removable media drives is improving
all the time, fixed media drives have so far retained their
advantage.
Sampling rate
Resolution
Bit rate
Capacity/min
Capacity/hour
kHz
bits
kbit/s
Mbytes/min
Mbytes/hour
96
88.1
48
48
44.1
44.1
32
22.05
11
16
16
20
16
16
8
16
8
8
1536
1410
960
768
706
353
512
176
88
11.0
10.1
6.9
5.5
5.0
2.5
3.7
1.3
0.6
659
605
412
330
303
151
220
76
38
Data rates and capacities for linear PCM
data stream and stores it for a short time before releasing it as a continuous stream. It performs the opposite function during recording, as shown
in Figure 9.12. Several things cause a delay in the retrieval of information
from disks: the time it takes for the head positioner to move across a disk,
the time it takes for the required data in a particular track to come around
to the pickup head, and the transfer of the data from the disk via the buffer RAM to the outside world, as shown in Figure 9.13. Total delay, or data
access time, is in practice several milliseconds. The instantaneous rate at
which the system can accept or give out data is called the transfer rate and
varies with the storage device.
Sound is stored in named data files on the disk, the files consisting of
a number of blocks of data stored either separately or together. A directory
stored on the disk keeps track of where the blocks of each file are stored
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Delay
Store
Record buffer
Audio
in
Replay buffer
Audio
out
Continuous data
Intermittent data
RAM
FIGURE 9.12 RAM buffering is used to convert burst data flow to continuous data flow, and
vice versa.
Direction of rotation
Rotational latency
Required block
Head positioner
Seek latency
FIGURE 9.13 The delays involved in accessing a block of data stored on a disk.
so that they can be retrieved in correct sequence. Each file normally corresponds to a single recording of a single channel of audio, although some
stereo file formats exist.
Multiple channels are handled by accessing multiple files from the disk
in a time-shared manner, with synchronization between the tracks being
performed subsequently in RAM. The storage capacity of a disk can be
divided between channels in whatever proportion is appropriate, and it is
not necessary to pre-allocate storage space to particular audio channels. For
example, a 360 Mbyte disk will store about 60 minutes of mono audio at
professional rates. This could be subdivided to give 30 minutes of stereo,
Mass Storage-Based Systems
15 minutes of four track, etc., or the proportions could be shared unequally.
A feature of the disk system is that unused storage capacity is not necessarily ‘wasted’ as can be the case with a tape system. During recording of a
multitrack tape there will often be sections on each track with no information recorded, but that space cannot be allocated elsewhere. On a disk
these gaps do not occupy storage space and can be used for additional space
on other channels at other times.
The number of audio channels that can be recorded or replayed simultaneously depends on the performance of the storage device, interface,
drivers and host computer. Slow systems may only be capable of handling
a few channels whereas faster systems with multiple disk drives may be
capable of expansion up to a virtually unlimited number of channels.
Some systems are modular, allowing for expansion of storage and other
audio processing facilities as means allow, with all modules communicating over a high-speed data bus, as shown in Figure 9.14. Increasingly external disks are connected using high-speed serial interfaces such as Firewire
SCSI IF
Disk drives
High speed bus
Disk I/O card 2
Increase storage capacity
Disk I/O card n
Signal processing
Increase channel capacity
Disk I/O card 1
SCSI chain
FIGURE 9.14
Arrangement of multiple
disks in a typical modular
system, showing how a
number of disks can be
attached to a single SCSI
chain to increase storage
capacity. Additional IO
cards can be added to
increase data throughput
for additional audio
channels.
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(IEEE 1394) (see Fact File 9.6), and as desktop computers get faster and
more capable there is no longer a strong need to have dedicated cards for
connecting audio-only disk drives. These days one or more of the host
computer’s internal or external disks is usually employed, although it is
FA C T F I L E 9 . 6 P E R IP H E R A L IN TERFACES
A variety of different physical interfaces can be used for
interconnecting storage devices and host workstations.
Some are internal buses only designed to operate over
limited lengths of cable and some are external interfaces
that can be connected over several meters. The interfaces
can be broadly divided into serial and parallel types, the
serial types tending to be used for external connections
owing to their size and ease of use. The disk interface
can be slower than the drive attached to it in some cases,
making it into a bottleneck in some applications. There
is no point having a super-fast disk drive if the interface
cannot handle data at that rate.
SCSI
For many years the most commonly used interface for
connecting mass storage media to host computers was
SCSI (the Small Computer Systems Interface), pronounced ‘scuzzy’. It is still used quite widely for very high
performance applications but EIDE interfaces and drives
are now capable of very good performance that can be
adequate for many purposes. SCSI is a high-speed parallel interface found on many computer systems, originally
allowing up to seven peripheral devices to be connected
to a host on a single bus. SCSI has grown through a
number of improvements and revisions, the latest being
Ultra160 SCSI, capable of addressing 16 devices at a
maximum data rate of 160 Mbyte/sec. A new generation
of Serial Attached SCSI (SAS) interfaces is also beginning
to become available, which retains many of the features
of SCSI but uses a serial format.
ATA/IDE
The ATA and IDE family of interfaces has evolved through
the years as the primary internal interface for connecting
disk drives to PC system buses. It is cheap and ubiquitous. Although drives with such interfaces were not considered adequate for audio purposes in the past, many
people are now using them with the on-board audio processing of modern computers as they are cheap and the
performance is adequate for many needs. Recent flavors
of this interface family include Ultra ATA/66 and Ultra
ATA/100 that use a 40-pin, 80 conductor connector and
deliver data rates up to either 66 or 100 Mbyte/sec. ATAPI
(ATA Packet Interface) is a variant used for storage media
such as CD drives.
Serial ATA is a relatively recent development designed
to enable disk drives to be interfaced serially, thereby
reducing the physical complexity of the interface. High
data transfer rates are planned, eventually up to
600 Mbyte/sec. It is intended primarily for internal connection of disks within host workstations, rather than as
an external interface like USB or Firewire.
PCMCIA
PCMCIA is a standard expansion port for notebook computers and other small-size computer products. A number of storage media and other peripherals are available
in PCMCIA format, and these include flash memory cards,
modem interfaces and super-small hard disk drives. The
standard is of greatest use in portable and mobile applications where limited space is available for peripheral storage.
Firewire and USB
Firewire and USB are both serial interfaces for connecting
external peripherals. They both enable disk drives to be
connected in a very simple manner, with high transfer rates
(many hundreds of megabits per second), although USB
1.0 devices are limited to 12 Mbit/s. A key feature of these
interfaces is that they can be ‘hot plugged’ (in other words
devices can be connected and disconnected with the power
on). The interfaces also supply basic power that enables
some simple devices to be powered from the host device.
Interconnection cables can usually be run up to between 5
and 10 meters, depending on the cable and the data rate.
Mass Storage-Based Systems
often recommended that this is not the same disk as used for system software in order to avoid conflicts of demand betweeen system housekeeping
and audio needs.
Media formatting
The process of formatting a storage device erases all of the information in
the volume. (It may not actually do this, but it rewrites the directory and
volume map information to make it seem as if the disk is empty again.)
Effectively the volume then becomes virgin territory again and data can be
written anywhere.
When a disk is formatted at a low level the sector headers are written and the bad blocks mapped out. A map is kept of the locations of
bad blocks so that they may be avoided in subsequent storage operations.
Low-level formatting can take quite a long time as every block has to be
addressed. During a high-level format the disk may be subdivided into a
number of ‘partitions’. Each of these partitions can behave as an entirely
independent ‘volume’ of information, as if it were a separate disk drive (see
Figure 9.15). It may even be possible to format each partition in a different
way, such that a different filing system may be used for each partition. Each
volume then has a directory created, which is an area of storage set aside to
contain information about the contents of the disk. The directory indicates
the locations of the files, their sizes, and various other vital statistics.
Disk store
Total available storage capacity
Volume A
Volume A Header info
(size, block size,
interleave ratio,
filing system, etc.)
Volume B
Volume C
Ditto
Ditto
Directory
Subdirectories
Files
FIGURE 9.15 A disk may be divided up into a number of different partitions, each acting as an
independent volume of information.
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The most common general purpose filing systems in audio workstations are HFS (Hierarchical Filing System) or HFS (for Mac OS), FAT 32
(for Windows PCs) and NTFS (for Windows NT and 2000). The Unix
operating system is used on some multi-user systems and high-powered
workstations and also has its own filing system. These were not designed
principally with real-time requirements such as audio and video replay in
mind but they have the advantage that disks formatted for a widely used
filing system will be more easily interchangeable than those using proprietary systems. Further information about audio file formats and interchange
is provided in the next chapter.
When an erasable volume like a hard disk has been used for some time
there will be a lot of files on the disk, and probably a lot of small spaces
where old files have been erased. New files must be stored in the available
space and this may involve splitting them up over the remaining smaller
areas. This is known as disk fragmentation, and it seriously affects the
overall performance of the drive. The reason is clear to see from Figure
9.16. More head seeks are required to access the blocks of a file than if they
had been stored contiguously, and this slows down the average transfer rate
considerably. It may come to a point where the drive is unable to supply
data fast enough for the purpose.
There are only two solutions to this problem: one is to reformat the disk
completely (which may be difficult, if one is in the middle of a project), the
other is to optimize or consolidate the storage space. Various software utilities exist for this purpose, whose job is to consolidate all the little areas of
free space into fewer larger areas. They do this by juggling the blocks of files
between disk areas and temporary RAM – a process that often takes a number
of hours. Power failure during such an optimization process can result in total
corruption of the drive, because the job is not completed and files may be only
half moved, so it is advisable to back up the drive before doing this. It has
been known for some such utilities to make the files unusable by some audio
editing packages, because the software may have relied on certain files being
in certain physical places, so it is wise to check first with the manufacturer.
AUDIO PROCESSING FOR COMPUTER WORKSTATIONS
Introduction
A lot of audio processing now takes place within the workstation, usually
relying either on the host computer’s processing power (using the CPU to
perform signal processing operations) or on one or more DSP (digital signal
processing) cards attached to the workstation’s expansion bus. Professional
Audio Processing for Computer Workstations
systems usually use external A/D and D/A converBlock 3
(a)
Block 2
tors, connected to a ‘core’ card attached to the comBlock 1
puter’s expansion bus. This is because it is often
difficult to obtain the highest technical performance
from convertors mounted on internal sound cards,
owing to the relatively ‘noisy ’ electrical environment
inside most computers. Furthermore, the number of
Pickup positioner
channels required may not fit onto an internal card.
As more and more audio work takes place entirely in
Block 1
the digital domain, though, the need for analog con(b)
Block 3
vertors decreases. Digital interfaces are also often
provided on external ‘breakout boxes’, partly for convenience and partly because of physical size of the
connectors. Compact connectors such as the optical Block 2
connector used for the ADAT eight-channel interface or the two-channel SPDIF phono connector are
Pickup positioner
accommodated on some cards, but multiple AES/EBU
connectors cannot be.
It is also becoming increasingly common for sub- FIGURE 9.16 At (a) a file is stored in three
stantial audio processing power to exist on integrated contiguous blocks and these can be read sequentially
sound cards that contain digital interfaces and possi- without moving the head. At (b) the file is fragmented and
bly A/D and D/A convertors. These cards are typically is distributed over three remote blocks, involving movement
used for consumer or semi-professional applications of the head to read it. The latter read operation will take
on desktop computers, although many now have very more time.
impressive features and can be used for advanced operations. Such cards are
now available in ‘full duplex’ configurations that enable audio to be received
by the card from the outside world, processed and/or stored, then routed
back to an external device. Full duplex operation usually allows recording
and replay simultaneously.
Sound cards and DSP cards are commonly connected to the workstation using the PCI (peripheral component interface) expansion bus. Older
ISA (PC) buses or NuBus (Mac) slots did not have the same data throughput capabilities and performance was therefore somewhat limited. PCI or
the more recent PCI Express bus can be extended to an external expansion chassis that enables a larger number of cards to be connected than
allowed for within the host computer. Sufficient processing power can now
be installed for the workstation to become the audio processing ‘heart’ of a
larger studio system, as opposed to using an external mixing console and
effects units. The higher the sampling frequency, the more DSP operations will be required per second, so it is worth bearing in mind that going
up to, say, 96 kHz sampling frequency for a project will require double the
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processing power and twice the storage space of 48 kHz. The same is true
of increasing the number of channels to which processing is applied.
The issue of latency is important in the choice of digital audio hardware
and software, as discussed in Fact File 9.7.
DSP cards
DSP cards can be added to widely used workstation packages such as
Digidesign’s ProTools. These so-called ‘DSP Farms’ or ‘Mix Farms’ are
expansion cards that connect to the PCI bus of the workstation and take
on much of the ‘number crunching’ work involved in effects processing and
mixing. ‘Plug-in’ processing software is becoming an extremely popular and
cost-effective way of implementing effects processing within the workstation, and this is discussed further in Chapter 13. ProTools plug-ins usually
rely either on DSP Farms or on host-based processing (see the next section)
to handle this load.
Digidesign’s TDM (Time Division Multiplex) architecture is a useful
example of the way in which audio processing can be handled within the
workstation. Here the processing tasks are shared between DSP cards, each
card being able to handle a certain number of operations per second. If the
system runs out of ‘horse power ’ it is possible to add further DSP cards
to share the load. Audio is routed and mixed at 24 bit resolution, and a
common audio bus links the card that is connected on a separate multiway
ribbon cable.
FA C T F I L E 9 . 7 A U D IO P R O C E S SI NG LATENCY
Latency is the delay incurred in executing audio operations
between input and output of a system. The lower the better is the rule, particularly when operating systems in ‘full
duplex’ mode, because processed sound may be routed
back to musicians (for foldback purposes) or may be combined with undelayed sound at some point. The management of latency is a software issue and some systems have
sophisticated approaches to ensuring that all supposedly
synchronous audio reaches the output at the same time no
matter what processing it has encountered on the way.
Minimum latency achievable is both a hardware and
a software issue. The poorest systems can give rise to
tens or even hundreds of milliseconds between input and
output whereas the best reduce this to a few milliseconds.
Audio I/O that connects directly to an audio processing
card can help to reduce latency, otherwise the communication required between host and various cards can add
to the delay. Some real-time audio processing software
also implements special routines to minimize and manage critical delays and this is often what distinguishes
professional systems from cheaper ones. The audio driver
software or ‘middleware’ that communicates between
applications and sound cards influences latency considerably. One example of such middleware intended for low
latency audio signal routing in computers is Steinberg’s
ASIO (Audio Stream Input Output).
Audio Processing for Computer Workstations
Host-based audio processing
An alternative to using dedicated DSP cards is to use the now substantial
processing capacity of a typical desktop workstation. The success of such
‘host-based processing’ obviously depends on the number of tasks that the
workstation is required to undertake and this capacity may vary with time
and context. It is, however, quite possible to use the host’s own CPU to run
DSP ‘plug-ins’ for implementing equalization, mixing and limited effects,
provided it is fast enough. The ‘multi-core’ (e.g. quad-core) processor architectures of some modern computers enables the division of processing
power between applications, and in some cases one can allocate a specific
number of processor cores to an audio application, leaving, say, one or two
for system tasks and other applications. This ensures the greatest degree of
hardware independence between processing tasks, and avoids conflicts of
demand at times of peak processor load.
The software architecture required to run plug-in operations on the host
CPU is naturally slightly different to that used on dedicated DSP cards, so
it is usually necessary to specify whether the plug-in is to run on the host
or on a dedicated resource such as Digidesign’s TDM cards. A number of
applications are now appearing, however, that enable the integration of
host-based (or ‘native’) plug-ins and dedicated DSP such as TDM-bus cards.
Audio processing that runs on the host may be subject to greater latency
(input to output delay) than when using dedicated signal processing, and
it obviously takes up processing power that could be used for running the
user interface or other software. It is nonetheless a cost-effective option for
many users that do not have high expectations of a system and it may be
possible to expand the system to include dedicated DSP in the future.
Integrated sound cards
Integrated sound cards typically contain all the components necessary to
handle audio for basic purposes within a desktop computer and may be able
to operate in full duplex mode (in and out at the same time). They typically incorporate convertors, DSP, a digital interface, FM and/or wavetable
synthesis engines. Optionally, they may also include some sort of I/O
daughter board that can be connected to a break-out audio interface,
increasing the number of possible connectors and the options for external
analog conversion. Such cards also tend to sport MIDI/joystick interfaces.
A typical example of this type of card is the ‘SoundBlaster ’ series from
Creative Labs.
Any analog audio connections are normally unbalanced and the convertors may be of only limited quality compared with the best external devices.
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For professional purposes it is advisable to use high-quality external convertors and balanced analog audio connections.
MASS STORAGE-BASED EDITING SYSTEM PRINCIPLES
Introduction
The random access nature of mass storage media led to the coining of the
term non-linear editing for the process of audio editing. With non-linear
editing the editor may preview a number of possible masters in their entirety
before deciding which should be the final one. Even after this, it is a simple
matter to modify the edit list to update the master. Edits may also be previewed and experimented with in order to determine the most appropriate
location and processing. Crossfades may be modified and adjustments made
to equalization and levels, all in the digital domain. Non-linear editing has
also come to feature very widely in post-production for video and film.
Non-linear editing is truly non-destructive in that the edited master only exists as a series of instructions to replay certain parts of sound
files at certain times, with specified signal processing overlaid, as shown
in Figure 9.17. The original sound files remain intact at all times, and a
single sound file can be used as many times as desired in different locations
and on different tracks without the need for copying the audio data. Editing
FIGURE 9.17
Instructions from an edit
decision list (EDL) are
used to control the replay
of sound file segments
from disk, which may
be subjected to further
processing (also under EDL
control) before arriving at
the audio outputs.
EDL
Host computer
running editing software
Commands
derived from
EDL
Signal
processing
Sound files on disk
Disk I/O and
memory
Audio
outputs
Edited
audio
Mass Storage-Based Editing System Principles
may involve the simple joining of sections, or it may involve more complex
operations such as long crossfades between one album track and the next,
or gain offsets between one section and another. All these things are possible without affecting the original source material.
Sound files and sound segments
In the case of music editing sound files might be session takes, anything
from a few bars to a whole movement, while in picture dubbing they might
contain a phrase of dialog or a sound effect. Specific segments of these
sound files can be defined while editing, in order to get rid of unwanted
material or to select useful extracts. The terminology varies but such identified parts of sound files are usually termed either ‘clips’ or ‘segments’.
Rather than creating a copy of the segment or clip and storing it as a separate sound file, it is normal simply to store it as a ‘soft’ entity – in other
words as simply commands in an edit list or project file that identify the
start and end addresses of the segment concerned and the sound file to
which it relates. It may be given a name by the operator and subsequently
used as if it were a sound file in its own right. An almost unlimited number of these segments can be created from original sound files, without the
need for any additional audio storage space.
Edit point handling
Edit points can be simple butt joins or crossfades. A butt join is very simple
because it involves straightforward switching from the replay of one sound
segment to another. Since replay involves temporary storage of the sound
file blocks in RAM (see above) it is a relatively simple matter to ensure
that both outgoing and incoming files in the region of the edit are available in RAM simultaneously (in different address areas). Up until the edit,
blocks of the outgoing file are read from the disk into RAM and thence to
the audio outputs. As the edit point is reached a switch occurs between
outgoing and incoming material by instituting a jump in the memory read
address corresponding to the start of the incoming material. Replay then
continues by reading subsequent blocks from the incoming sound file. It is
normally possible to position edits to single sample accuracy, making the
timing resolution as fine as a number of tens of microseconds if required.
The problem with butt joins is that they are quite unsubtle. Audible
clicks and bumps may result because of the discontinuity in the waveform
that may result, as shown in Figure 9.18. It is normal, therefore, to use at
least a short crossfade at edit points to hide the effect of the join. This is
what happens when analog tape is spliced, because the traditional angled
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CHAPTER 9: Digital Recording, Editing and Mastering Systems
cut has the same effect as a short crossfade
(of between 5 and 20 ms depending on the
tape speed and angle of cut). Most workstations have considerable flexibility with crossPoor join
owing to
fades and are not limited to short durations.
discontinuity
It is now common to use crossfades of many
(b)
shapes and durations (e.g. linear, root cosine,
equal power) for different creative purposes.
This, coupled with the ability to preview edits
Good join at
and fine-tune their locations, has made it poszero crossing
sible to put edits in places previously considFIGURE 9.18 (a) A bad butt edit results in a waveform
ered impossible.
discontinuity. (b) Butt edits can be made to work if there is minimal
The locations of edit points are kept in an
discontinuity.
edit decision list (EDL) which contains information about the segments and files to be replayed at each time, the in and
the out points of each section and details of the crossfade time and shape at
each edit point. It may also contain additional information such as signal
processing operations to be performed (gain changes, EQ, etc.).
(a)
Crossfading
Crossfading is similar to butt joining, except that it requires access to data
from both incoming and outgoing files for the duration of the crossfade.
The crossfade calculation involves simple signal processing, during which
the values of outgoing samples are multiplied by gradually decreasing coefficients whilst the values of incoming samples are multiplied by gradually
increasing coefficients. Time coincident samples of the two files are then
added together to produce output samples, as described in the previous
chapter. The duration and shape of the crossfade can be adjusted by altering the coefficients involved and the rate at which the process is executed.
Crossfades are either performed in real time, as the edit point passes,
or pre-calculated and written to disk as a file. Real-time crossfades can be
varied at any time and are simply stored as commands in the EDL, indicating the nature of the fade to be executed. The process is similar to that for
the butt edit, except that as the edit point approaches samples from both
incoming and outgoing segments are loaded into RAM in order that there
is an overlap in time. During the crossfade it is necessary to continue to
load samples from both incoming and outgoing segments into their respective areas of RAM, and for these to be routed to the crossfade processor, as
shown in Figure 9.19. The resulting samples are then available for routing to the output. Alternatively the crossfade can be calculated in non-real
Mass Storage-Based Editing System Principles
Replay
schedule
control
X
Solid-state
memory
Store
Crossfade
processor
Y
Output
buffer
X and Y
blocks
Audio
output
XXXX
X
XXXXYYYYY
Y
YYYYY
FIGURE 9.19 Conceptual diagram of the sequence of operations which occur during a crossfade. X and Y are the incoming and
outgoing sound segments.
Time
Butt
join
Output
File X
Butt
join
Crossfade
file
File Y
FIGURE 9.20 Replay of a precalculated crossfade file at an edit point between files X and Y.
time. This incurs a short delay while the system works out the sums, after
which a new sound file is stored which contains only the crossfade. Replay
of the edit then involves playing the outgoing segment up to the beginning
of the crossfade, then the crossfade file, then the incoming segment from
after the crossfade, as shown in Figure 9.20. Load on the disk drive is no
higher than normal in this case.
The shape of the crossfade can usually be changed to suit different
operational purposes. Standard linear fades (those where the gain changes
uniformly with time) are not always the most suitable for music editing,
especially when the crossfade is longer than about ten millseconds. The
result may be a momentary drop in the resulting level in the center of the
crossfade that is due to the way in which the sound levels from the two
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files add together. If there is a random phase difference between the signals, as there will often be in music, the rise in level resulting from adding
the two signals will normally be around 3 dB, but the linear crossfade is
6 dB down in its center resulting in an overall level drop of around 3 dB (see
Figure 9.21). Exponential crossfades and other such shapes may be more
suitable for these purposes, because they have a smaller level drop in the
center. It may even be possible to design customized crossfade laws. It is
often possible to alter the offset of the start and end of the fade from the
actual edit point and to have a faster fade-up than fade-down.
Many systems also allow automated gain changes to be introduced as
well as fades, so that level differences across edit points may be corrected.
Figure 9.22 shows a crossfade profile which has a higher level after the edit
point than before it, and different slopes for the in- and out-fades. A lot of
the difficulties that editors encounter in making edits work can be solved
using a combination of these facilities.
Editing modes
(a)
Amplitude
Summed output level
over crossfade
Outgoing take
Incoming take
Time
Summed output level
over crossfade
(b)
Amplitude
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Outgoing take
Incoming take
Time
FIGURE 9.21 Summation of levels at a crossfade. (a) A linear
crossfade can result in a level drop if the incoming and outgoing material
are non-coherent. (b) An exponential fade, or other similar laws, can help to
make the level more constant across the edit.
During the editing process the operator will load appropriate sound files
and audition them, both on their own
and in a sequence with other files. The
exact method of assembling the edited
sequence depends very much on the user
interface, but it is common to present
the user with a visual analogy of moving tape, allowing files to be ‘cut and
spliced’ or ‘copied and pasted’ into appropriate locations along the virtual tape.
These files, or edited clips of them, are
then played out at the timecode locations
corresponding to their positions on this
‘virtual tape’. It is also quite common
to display a representation of the audio
waveform that allows the editor to see
as well as hear the signal around the edit
point (see Figure 9.23).
In non-linear systems the tape-based
approach is often simulated, allowing
the user to roughly locate an edit point
while playing the virtual tape followed by
Mass Storage-Based Editing System Principles
Gain
a fine trim using simulated reel-rocking or a detailed view of the waveform.
Some software presents source and destination streams as well, in further
simulation of the tape approach. It is also possible to insert or change sections in the middle of a finished master, provided that the EDL and source
files are still available. To take an example, assume that an edited opera has
been completed and that the producer now wishes to change a take somewhere in the middle (see Figure 9.24). The replacement take is unlikely to
be exactly the same length but it is possible
E
simply to shuffle all of the following mate- +6 dB
rial along or back slightly to accommodate it,
A
0 dB
this being only a matter of changing the EDL
rather than modifying the stored music in
C
any way. The files are then simply played out
B
D
at slightly different times than in the first
version of the edit.
It is also normal to allow edited segments
to be fixed in time if desired, so that they are
not shuffled forwards or backwards when
–∞
Time
other segments are inserted. This ‘anchoring’
of segments is often used in picture dubbing FIGURE 9.22 The system may allow the user to program a gain
when certain sound effects and dialog have profile around an edit point, defining the starting gain (A), the fadedown time (B), the fade-up time (D), the point below unity at which
to remain locked to the picture.
the two files cross over (C) and the final gain (E).
FIGURE 9.23 Example from SADiE editing system showing the ‘trim editor’ in which is displayed
a detailed view of the audio waveform around the edit point, together with information about the
crossfade.
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(a)
To be replaced
Take M
Take N
Take O
Take M
Take P is just
too long for space
left by N
Take O
Take P
(replaces N)
(b)
To be replaced
Take M
Take M
Take N
Gap left by N may be
widened to accept P
Take O
Take O
Take P
(replaces N)
FIGURE 9.24 Replacing a take in the middle of an edited program. (a) Tape-based copy editing
results in a gap of fixed size, which may not match the new take length. (b) Non-linear editing allows
the gap size to be adjusted to match the new take.
Simulation of ‘reel-rocking’
It is common to simulate the effect of analog tape ‘reel-rocking’ in nonlinear editors, providing the user with the sonic impression that reels of
analog tape are being ‘rocked’ back and forth as they are in analog tape editing when fine-searching edit points. Editors are used to the sound of tape
moving in this way, and are skilled at locating edit points when listening to
such a sound.
The simulation of variable speed replay in both directions (forwards
and backwards) is usually controlled by a wheel or sideways movement of
Editing Software
a mouse which moves the ‘tape’ in either direction around the current play
location. The magnitude and direction of this movement is used to control
the rate at which samples are read from the disk file, via the buffer, and this
replaces the fixed sampling rate clock as the controller of the replay rate.
Systems differ very greatly as to the sound quality achieved in this mode,
because it is in fact quite a difficult task to provide convincing simulation.
So poor have been many attempts that many editors do not use the feature,
preferring to judge edit points accurately ‘on the fly ’, followed by trimming
or nudging them either way if they are not successful the first time. Good
simulation requires very fast, responsive action and an ergonomically suitable control. A mouse is very unsuitable for the purpose. It also requires a
certain amount of DSP to filter the signal correctly, in order to avoid the
aliasing that can be caused by varying the sampling rate.
EDITING SOFTWARE
It is increasingly common for MIDI (see Chapter 14) and digital audio editing to be integrated within one software package, particularly for pop music
recording and other multitrack productions where control of electronic
sound sources is integrated with recorded natural sounds. Such applications used to be called sequencers but this is less common now that MIDI
sequencing is only one of many tasks that are possible. Although most
sequencers contain some form of audio editing these days, there are some
software applications more specifically targeted at high-quality audio editing and production. These have tended to come from a professional audio
background rather than a MIDI sequencing background, although it is
admitted that the two fields have met in the middle now and it is increasingly hard to distinguish a MIDI sequencer with added audio features from
an audio editor with added MIDI features.
Audio applications such as those described here are used in contexts
where MIDI is not particularly important and where fine control over editing crossfades, dithering, mixing, mastering and post-production functions
are required. Here the editor needs tools for such things as: previewing
and trimming edits, such as might be necessary in classical music postproduction; PQ editing CD masters; preparing surround sound DVD material for encoding; MLP or AC-3 encoding of audio material; editing of DSD
material for SuperAudio CD. The following example, based on the SADiE
audio editing system, demonstrates some of the practical concepts.
SADiE workstations run on the PC platform and most utilize an external audio interface. Recent Series 5 systems, however, can be constructed
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as an integrated rack-mounted unit containing audio interfaces and a
Pentium PC. Both PCM and DSD signal processing options are available
and the system makes provision for lossless encoding for DVD-Audio and
Blu-Ray, as well as SACD mastering and encoding. A typical user interface
for SADiE is shown in Figure 9.25. It is possible to see transport controls,
the mixer interface and the playlist display. The main part of the screen
is occupied by a horizontal display of recording tracks or ‘streams’, and
these are analogous to the tracks of a multitrack tape recorder. A record
icon associated with each stream is used to arm it ready for recording. As
recording proceeds, the empty streams are filled from left to right across
the screen in real time, led by a vertical moving cursor. These streams
can be displayed either as solid continuous blocks or as waveforms, the
latter being the usual mode when editing is undertaken. After recording,
extra streams can be recorded if required simply by disarming the record
icons of the streams already used and arming the record icons of empty
FIGURE 9.25 SADiE editor displays, showing mixer, playlist, transport controls and project
elements.
Editing Software
streams below them, making it possible to build up a large number of
‘virtual’ tracks as required. The maximum number that can be replayed
simultaneously depends upon the memory and DSP capacity of the system
used. A basic two-input/four-output might allow up to eight streams to be
replayed (depending on the amount of DSP being used for other tasks),
and a fully equipped system can allow at least 32 simultaneous streams of
program material to be recorded and replayed, i.e. it is a complete multitrack recording machine.
Replay involves either using the transport control display or clicking the
mouse at a desired position on a time-bar towards the top of the screen,
this positioning the moving cursor (which is analogous to a tape head)
where one wishes replay to begin. Editing is performed by means of a razorblade icon, which will make the cut where the moving cursor is positioned.
Alternatively, an edit icon can be loaded to the mouse’s cursor for positioning anywhere on any individual stream to make a cut.
Audio can be arranged in the playlist by the normal processes of placing,
dragging, copying and pasting, and there is a range of options for slipping
material left or right in the list to accommodate new material (this ensures
that all previous edits remain attached in the right way when the list is
slipped backwards or forwards in time). Audio to be edited in detail can be
viewed in the trim window (shown earlier in Figure 9.23) which shows a
detailed waveform display, allowing edits to be previewed either to or from
the edit point, or across the edit, using the play controls in the top righthand corner (this is particularly useful for music editing). The crossfade
region is clearly visible, with different colors and shadings used to indicate
the ‘live’ audio streams before and after the edit. There are many stages
of undo and redo so that nothing need be permanent at this stage. When
a satisfactory edit is achieved, it can be written back to the main display
where it will be incorporated. Scrub and jog actions for locating edit points
are also possible. A useful ‘lock to time’ icon is provided which can be activated to prevent horizontal movement of the streams so that they cannot
be accidentally moved out of sync with each other during editing.
The mixer section can be thought of in conventional terms, and indeed
some systems offer physical plug-in interfaces with moving fader automation
for those who prefer them. As well as mouse control of such things as fader,
pan, solo and mute, processing such as EQ, filters, aux send and compression
can be selected from an effects ‘rack’, and each can be dragged across and
dropped in above a fader where it will become incorporated into that channel.
Third party ‘plug-in’ software is also available for many systems to enhance
the signal processing features, including CEDAR audio restoration software,
as described below. The latest software allows for the use of DirectX plug-ins
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for audio processing. Automation of faders and
other processing is also possible.
MASTERING AND RESTORATION
Software
(a)
(b)
FIGURE 9.26 CEDAR Retouch display for SADiE, showing
frequency (vertical) against time (horizontal) and amplitude (color/
density). Problem areas of the spectrographic display can be
highlighted and a new signal synthesized using information from
the surrounding region. (a) Harmonics of an interfering signal
can be clearly seen. (b) A short-term spike crosses most of the
frequency range.
Some software applications are designed specifically for the mastering and restoration markets.
These products are designed either to enable
‘fine tuning’ of master recordings prior to commercial release, involving subtle compression,
equalization and gain adjustment (mastering),
or to enable the ‘cleaning up’ of old recordings
that have hiss, crackle and clicks (restoration).
CEDAR applications or plug-ins are good
examples of the restoration group. Sophisticated
controls are provided for the adjustment of
dehissing and decrackling parameters, which
often require considerable skill to master.
Recently the company has introduced advanced
visualization tools that enable restoration engineers to ‘touch up’ audio material using an interface not dissimilar to that used for photo editing
on computers. Audio anomalies (unwanted content) can be seen in the time and frequency
domains, highlighted and interpolated based on
information either side of the anomaly. A typical
display from its Retouch product for the SADiE
platform is shown in Figure 9.26.
CEDAR’s
restoration
algorithms
are
typically divided into ‘decrackle’, ‘declick’,
‘dethump’ and ‘denoise’, each depending on the
nature of the anomaly to be corrected. Some
typical user interfaces for controlling these processes are shown in Figure 9.27.
Mastering software usually incorporates
advanced dynamics control such as the TC
Works Master X series, based on its Finalizer
products, a user interface of which is pictured
in Figure 9.28. Here compressor curves and
Mastering and Restoration
FIGURE 9.27
CEDAR restoration plug-ins
for SADiE, showing
(a) declick and (b) denoise
processes.
(a)
(b)
frequency dependency of dynamics can be adjusted and metered. The display also allows the user to view the number of samples at peak level to
watch for digital overloads that might be problematic.
Level control in mastering
Level control, it might be argued, is less crucial than it used to be in the
days when a recording engineer struggled to optimize a recording’s dynamic
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FIGURE 9.28 TC Works Master X mastering dynamics plug-in interface.
(a)
(b)
Clipping level (0 dBFS)
3% distortion ceiling
Reference level
Effective dynamic range
Effective dynamic range
Tape noise floor
Dither noise floor
FIGURE 9.29 Comparison of analog and digital dynamic range. (a) Analog tape has increasing distortion
as the recording level increases, with an effective maximum output level at 3% third harmonic distortion. (b)
Modern high-resolution digital systems have wider dynamic range with a noise floor fixed by dither noise
and a maximum recording level at which clipping occurs. The linearity of digital systems does not normally
become poorer as signal level increases, until 0 dBFS is reached. This makes level control a somewhat
less important issue at the initial recording stage, provided sufficient headroom is allowed for peaks.
Mastering and Restoration
range between the noise floor and the distortion ceiling (see Figure 9.29).
However, there are still artistic and technical considerations.
The dynamic range of a typical digital audio system can now be well over
100 dB and there is room for the operator to allow a reasonable degree of
‘headroom’ between the peak audio signal level and the maximum allowable
level. Meters are provided to enable the signal level to be observed, and they
are usually calibrated in dB, with zero at the top and negative dBs below this.
The full dynamic range is not always shown, and there may be a peak bar
that can hold the maximum level permanently or temporarily. As explained
in Chapter 8, 0 dBFS (full scale) is the point at which all of the bits available
to represent the signal have been used. Above this level the signal clips and
the effect of this is quite objectionable, except on very short transients where
it may not be noticed. It follows that signals should never be allowed to clip.
There is a tendency in modern audio production to want to master
everything so that it sounds as loud as possible, and to ensure that the signal peaks as close to 0 dBFS as possible. This level maximizing or normalizing process can be done automatically in most packages, the software
searching the audio track for its highest level sample and then adjusting the
overall gain so that this just reaches 0 dBFS. In this way the recording can be
made to use all the bits available, which can be useful if it is to be released
on a relatively low-resolution consumer medium where noise might be more
of a problem. (It is important to make sure that correct redithering is used
when altering the level and requantizing, as explained in Chapter 8.) This
does not, of course, take into account any production decisions that might
be involved in adjusting the overall levels of individual tracks on an album
or other compilation, where relative levels should be adjusted according to
the nature of the individual items, their loudness and the producer’s intent.
A little-known but important fact is that even if the signal is maximized in
the automatic fashion, so that the highest sample value just does not clip, subsequent analog electronics in the signal chain may still do so. Some equipment
is designed in such a way that the maximum digital signal level is aligned to
coincide with the clipping voltage of the analog electronics in a D/A convertor.
In fact, owing to the response of the reconstruction filter in the D/A convertor
(which reconstructs an analog waveform from the PAM pulse train) intersample signal peaks can be created that slightly exceed the analog level corresponding to 0 dBFS, thereby clipping the analog side of the convertor. For this reason
it is recommended that digital-side signals are maximized so that they peak
a few dB below 0 dBFS, in order to avoid the distortion that might otherwise
result on the analog side. Some mastering software provides detailed analysis
of the signal showing exactly how many samples occur in sequence at peak
level, which can be a useful warning of potential or previous clipping.
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PREPARING FOR AND UNDERSTANDING
RELEASE MEDIA
Consumer release formats such as CD, DVD (see Chapter 10), and MP3
(see Chapter 8) usually require some form of mastering and pre-release
preparation. This can range from subtle tweaks to the sound quality and
relative levels on tracks to PQ encoding, DVD authoring, data encoding
and the addition of graphics, video and text. Some of these have already
been mentioned in other places in this book.
PQ encoding for CD mastering can often be done in some of the application packages designed for audio editing, such as SADiE and Pyramix.
In this case it may involve little more than marking the starts and ends
of the tracks in the playlist and allowing the software to work out the relevant frame advances and Red Book requirements for the assembly of the
PQ code that will either be written to a CD-R or included in the DDP file
for sending to the pressing plant. The CD only comes at one resolution
and sampling frequency (16 bit, 44.1 kHz) making release preparation a relatively straightforward matter.
DVD mastering is considerably more complicated than CD and requires
advanced authoring software that can deal with all the different options
possible on this multi-faceted release format. A number of different combinations of players and discs are possible, as explained in Fact File 9.8,
although the DVD-Audio format has not been particularly successful commercially. DVD-Video allows for 48 or 96 kHz sampling frequency and 16,
20 or 24 bit PCM encoding. A two-channel downmix must be available
on the disc in linear PCM form (for basic compatibility), but most discs
also include Dolby Digital or possibly DTS surround audio. Dolby Digital
encoding usually involves the preparation of a file or files containing the
compressed data, and a range of settings have to be made during this process, such as the bit rate, dialog normalization level, rear channel phase
shift and so on. A typical control screen is shown in Figure 9.30. Then of
course there are the pictures, but they are not the topic of this book.
Playing time depends on the way in which producers decide to use the
space available on the disc, and this requires the juggling of the available
bit budget. DVD-Audio can store at least 74 minutes of stereo audio even
at the highest sample rate and resolution (192/24). Other modes are possible, with up to six channels of audio playing for at least 74 minutes, using
combinations of sample frequency and resolution, together with MLP. Sixchannel audio can only operate at the two lower sample rates of either class
(44.1/88.2 or 48/96).
DVD masters are usually transferred to the pressing plant on DLT
tapes, using the Disc Description Protocol (DDP), or on DVD-R(A) discs as
Preparing for and Understanding Release Media
FA C T F I L E 9 . 8 D V D D IS C S A N D PLAYERS
There are at least three DVD player types in existence
(audio, universal and video), although only the DVD-Video
format has gained a substantial market presence. There
are also two types of DVD-Audio disc, one containing only
audio objects and the other (the DVD-AudioV) capable
of holding video objects as well. The video objects on a
DVD-AudioV are just the same as DVD-Video objects and
therefore can contain video clips, Dolby AC-3 compressed
audio and other information. In addition, there is the standard DVD-Video disc.
DVD-AudioV discs should play back in audio players
and universal players. Any video objects on an AudioV disc
should play back on video-only players. The requirement
for video objects on DVD-AudioV discs to contain PCM
audio was dropped at the last moment so that such
objects could only contain AC-3 audio if desired. This
means that an audio disc could contain a multichannel
AC-3 audio stream in a video object, enabling it to be
played in a video player. This is a good way of ensuring
that a multichannel audio disc plays back in as many different types of player as possible, but requires that the
content producer makes sure to include the AC-3 video
object in addition to MLP or PCM audio objects. The video
object can also contain a DTS audio bitstream if desired.
Figure courtesy of Bike Suzuki (DVD-Audio Forum).
Audio player
DVD-Audio disc
DVD
12 25:00
Universal player
Video player
DVD-AudioV disc
DVD-Video disc
FIGURE 9.30
Screen display of Dolby
Digital encoding software
options.
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FIGURE 9.31 Example of SACD text authoring screen from SADiE.
a disc image with a special CMF (cutting master format) header in the disc
lead-in area containing the DDP data.
SACD Authoring software enables the text information to be added, as
shown in Figure 9.31. SACD masters are normally submitted to the pressing plant on AIT format data tapes.
Sony and Philips have paid considerable attention to copy protection
and anti-piracy measures on the disc itself. Comprehensive visible and
invisible watermarking are standard features of the SACD. Using a process
known as PSP (Pit Signal Processing) the width of the pits cut into the disc
surface is modulated in such a fashion as to create a visible image on the
surface of the CD layer, if desired by the originator. This provides a visible
means of authentication. The invisible watermark is a mandatory feature
of the SACD layer and is used to authenticate the disc before it will play
on an SACD player. The watermark is needed to decode the data on the
disc. Discs without this watermark will simply be rejected by the player.
It is apparently not possible to copy this watermark by any known means.
Encryption of digital music content is also optional, at the request of software providers.
MP3, as already explained elsewhere, is actually MPEG-1, Layer 3
encoded audio, stored in a data file, usually for distribution to consumers
Recommended Further Reading
either on the Internet or on other release media. Consumer disc players are
increasingly capable of replaying MP3 files from CDs, for example. MP3
mastering requires that the two-channel audio signal is MPEG-encoded,
using one of the many MP3 encoders available. Some mastering software
now includes MP3 encoding as an option, as well as other data-reduced formats such as AAC.
Some of the choices to be made in this process concern the data rate
and audio bandwidth to be encoded, as this affects the sound quality. The
lowest bit rates (e.g. below 64 kbit/s) will tend to sound noticeably poorer
than the higher ones, particularly if full audio bandwidth is retained. For
this reason some encoders limit the bandwidth or halve the sampling frequency for very low bit rate encoding, because this tends to minimize the
unpleasant side-effects of MPEG encoding. It is also possible to select joint
stereo coding mode, as this will improve the technical quality somewhat at
low bit rates, possibly at the expense of stereo imaging accuracy. As mentioned above, at very low bit rates some audio processing may be required
to make sound quality acceptable when squeezed down such a small pipe.
Commercial tools for interactive authoring and MPEG-4 encoding are
only just beginning to appear at the time of writing. Such tools enable
audio scenes to be described and data encoded in a scalable fashion so that
they can be rendered at the consumer replay end of the chain, according to
the processing power available.
RECOMMENDED FURTHER READING
Katz, B., 2007. Mastering Audio: The Art and Science. Focal Press.
Leider, C., 2004. Digital Audio Workstation: Mixing, Recording and Mastering Your
MAC or PC. McGraw-Hill Professional.
Rumsey, F., 2004. Desktop Audio Technology. Focal Press.
Watkinson, J., 2001. The Art of Digital Audio. Focal Press.
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CHAPTER 10
Digital Audio Formats and Interchange
CH A P T E R C O N TE N T S
Audio File Formats for Digital Workstations
File formats in general
Sound Designer formats
AIFF and AIFF-C formats
RIFF WAVE format
MPEG audio file formats
DSD-IFF file format
Edit decision list (EDL) files and project interchange
AES-31 format
MXF – the media exchange format
AAF – the advanced authoring format
Disk pre-mastering formats
Consumer Optical Disk Formats
Compact discs and drives
DVD
Super Audio CD (SACD)
Blu-Ray disk
Interconnecting Digital Audio Devices
Introduction
Dedicated audio interface formats
The AES/EBU interface (AES-3)
Standard consumer interface (IEC 60958-3)
Proprietary digital interfaces
Data networks and computer interconnects
Audio network requirements
Storage area networks
Protocols for the internet
Wireless networks
Audio over firewire (IEEE 1394)
Audio over Universal Serial Bus (USB)
AES-47: audio over ATM
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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This chapter provides further details about the main formats in which digital audio data is stored and moved between systems. This includes coverage
of audio file formats, consumer disk standards, digital interfaces and networked audio interchange, concentrating mainly on those issues of importance for professional applications.
AUDIO FILE FORMATS FOR DIGITAL WORKSTATIONS
There used to be almost as many file formats for audio as there are days in
the year. In the computer games field, for example, this is still true to some
extent. For a long time the specific file storage strategy used for disk-based
digital audio was the key to success in digital workstation design, because
disk drives were relatively slow and needed clever strategies to ensure that
they were capable of handling a sufficiently large number of audio channels. Manufacturers also worked very much in isolation and the size of the
market was relatively small, leading to virtually every workstation or piece
of software using a different file format for audio and edit list information.
There are still advantages in the use of filing structures specially designed
for real-time applications such as audio and video editing, because one may
obtain better performance from a disk drive in this way, but the need is not as
great as it used to be. Interchange is becoming at least as important as, if not
more important than, ultimate transfer speed and the majority of hard disk
drives available today are capable of replaying many channels of audio in real
time without needing to use a dedicated storage strategy. As the use of networked workstations grows, the need for files to be transferred between systems also grows and either by international standardization or by sheer force
of market dominance certain file formats are becoming the accepted means
by which data is exchanged. This is not to say that we will only be left with
one or two formats, but that systems will have to be able to read and write
files in the common formats if users are to be able to share work with others.
The recent growth in the importance of metadata (data about data), and
the representation of audio, video and metadata as ‘objects’, has led to the
development of interchange methods that are based on object-oriented concepts and project ‘packages’ as opposed to using simple text files and separate media files. There is increasing integration between audio and other
media in multimedia authoring and some of the file formats mentioned
below are closely related to international efforts in multimedia file exchange.
It is not proposed to attempt to describe all of the file formats in existence, because that would be a relatively pointless exercise and would not
make for interesting reading. It is nonetheless useful to have a look at some
Audio File Formats for Digital Workstations
examples taken from the most commonly encountered file formats, particularly those used for high-quality audio by desktop and multimedia systems,
since these are amongst the most widely used in the world and are often
handled by audio workstations even if not their native format. It is not proposed to investigate the large number of specialized file formats developed
principally for computer music on various platforms, nor the files used for
internal sounds and games on many computers.
File formats in general
A data file is simply a series of data bytes formed into blocks and stored
either contiguously or in fragmented form. Files themselves are largely
independent of the operating system and filing structure of the host computer, because a file can be transferred to another platform and still exist
as an identical series of data blocks. It is the filing system that is often
the platform- or operating-system-dependent entity. There are sometimes
features of data files that relate directly to the operating system and filing
system that created them, these being fairly fundamental features, but they
do not normally prevent such files being translated by other platforms.
For example, there are two approaches to byte ordering: the so-called
little-endian order in which the least significant byte comes first or at the
lowest memory address, and the big-endian format in which the most significant byte comes first or at the highest memory address. These originally
related to the byte ordering used in data processing by the two most common microprocessor families and thereby to the two most common operating systems used in desktop audio workstations. Motorola processors, as
originally used in the Apple Macintosh, deal in big-endian byte ordering,
and Intel processors, as used in MS-DOS machines, deal in little-endian
byte ordering. It is relatively easy to interpret files either way around but it
is necessary to know that there is a need to do so if one is writing software.
Second, some Macintosh files may have two parts – a resource fork
and a data fork – whereas Windows files only have one part. High-level
‘resources’ are stored in the resource fork (used in some audio files for storing information about the file, such as signal processing to be applied, display information and so forth) whilst the raw data content of the file is
stored in the data fork (used in audio applications for audio sample data).
The resource fork is not always there, but may be. The resource fork can
get lost when transferring such files between machines or to servers, unless
Mac-specific protocols are used (e.g. MacBinary or BinHex).
Some data files include a ‘header ’, that is a number of bytes at the start of
the file containing information about the data that follows. In audio systems
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this may include the sampling rate and resolution of the file. Audio replay
would normally be started immediately after the header. On the other hand,
some files are simply raw data, usually in cases where the format is fixed.
ASCII text files are a well-known example of raw data files – they simply begin with the first character of the text. More recently file structures
have been developed that are really ‘containers’ for lots of smaller files, or
data objects, each with its own descriptors and data. The RIFF structure,
described below is an early example of the concept of a ‘chunk-based’ file
structure. Apple’s Bento container structure, used in OMFI, and the container structure of AAF are more advanced examples of such an approach.
The audio data in most common high-quality audio formats is stored in
two’s complement form (see Chapter 8) and the majority of files are used
for 16 or 24 bit data, thus employing either 2 or 3 bytes per audio sample.
Eight bit files use 1 byte per sample.
Sound Designer formats
Sound Designer files originate with the Californian company Digidesign,
manufacturer of probably the world’s most widely used digital audio hardware for desktop computers. Many systems handle Sound Designer files
because they were used widely for such purposes as the distribution of
sound effects on CD-ROM and for other short music sample files.
The Sound Designer I format (SD I) is for mono sounds and it is recommended principally for use in storing short sounds. It originated on the
Macintosh, so numerical data is stored in big-endian byte order but it has
no resource fork. The data fork contains a header of 1336 bytes, followed
by the audio data bytes. The header contains information about how the
sample should be displayed in Sound Designer editing software, including
data describing vertical and horizontal scaling. It also contains details of
‘loop points’ for the file (these are principally for use with audio/MIDI sampling packages where portions of the sound are repeatedly cycled through
while a key is held down, in order to sustain a note). The header contains
information on the sample rate, sample period, number of bits per sample,
quantization method (e.g. ‘linear ’, expressed as an ASCII string describing
the method) and size of RAM buffer to be used. The audio data is normally
either 8 or 16 bit, and always MS byte followed by LS byte of each sample.
Sound Designer II has been one of the most commonly used formats for
audio workstations and has greater flexibility than SD I. Again it originated
as a Mac file and unlike SD I it has a separate resource fork which contains the file’s ‘vital statistics’. The data fork contains only the audio data
bytes in two’s complement form, either 8 or 16 bits per sample. SD II files
Audio File Formats for Digital Workstations
First byte of file
A1
B1
C1
D1
A2
MS byte
B2
C2
LS byte
D2
A3
B3
C3
D3
...
...
...
Dn
A1 = first sample of channel A
B1 = first sample of channel B
etc.
16 bit audio sample
FIGURE 10.1 Sound Designer II files allow samples for multiple audio channels to be interleaved.
Four channel, 16 bit example shown.
can contain audio samples for more than one channel, in which case the
samples are interleaved, as shown in Figure 10.1, on a sample by sample
basis (i.e. all the bytes for one channel sample followed by all the bytes for
the next, etc.). It is unusual to find more than stereo data contained in SD
II files and it is recommended that multichannel recordings are made using
separate files for each channel.
AIFF and AIFF-C formats
The AIFF format is widely used as an audio interchange standard, because
it conforms to the EA IFF 85 standard for interchange format files used
for various other types of information such as graphical images. AIFF is
an Apple standard format for audio data and is encountered widely on
Macintosh-based audio workstations and some Silicon Graphics systems.
Audio information can be stored at a number of resolutions and for any
number of channels if required, and the related AIFF-C (file type ‘AIFC ’)
format allows also for compressed audio data. It consists only of a data fork,
with no resource fork, making it easier to transport to other platforms.
All IFF-type files are typically made up of ‘chunks’ of data as shown
in Figure 10.2. A chunk consists of a header and a number of data bytes
to follow. The simplest AIFF files contain a ‘common chunk’, which is
equivalent to the header data in other audio files, and a ‘sound data’ chunk
containing the audio sample data. These are contained overall by a ‘form’
chunk as shown in Figure 10.3. AIFC files must also contain a ‘version
chunk’ before the common chunk to allow for future changes to AIFC.
RIFF WAVE format
The RIFF WAVE (often called WAV) format is the Microsoft equivalent
of Apple’s AIFF. It has a similar structure, again conforming to the IFF
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CHAPTER 10: Digital Audio Formats and Interchange
Header
8 bytes
304
Chunk ID
Chunk data size
Data
4 byte ASCII type identifier
Size in bytes of following data
(32 bit signed integer)
Data content of chunk
FIGURE 10.2 General format of an IFF file chunk.
Form chunk format
AIFF file format
4
'FORM'
'FORM'
4
Size
Size
4
Form type
'AIFF'
Bytes
'COMM'
Size
–4
Size
Chunks
Data
'SSND'
Size
Sound
data
FIGURE 10.3 General format of an AIFF file.
pattern, but with numbers stored in little-endian rather than big-endian
form. It is used widely for sound file storage and interchange on PC workstations, and for multimedia applications involving sound. Within WAVE
files it is possible to include information about a number of cue points, and
a playlist to indicate the order in which the cues are to be replayed. WAVE
files use the file extension ‘.wav ’.
Audio File Formats for Digital Workstations
RIFF
W
A
V
E
f
i
l
e
FORMAT
DATA
(Audio)
"RIFF"
4
Size (of file –8)
4
"WAVE"
4
"fmt_"
4
Length (&00000010)
4
12
24
Format information
16
FIGURE 10.4 Diagrammatic representation of a simple RIFF WAVE file, showing the three
principal chunks. Additional chunks may be contained within the overall structure, for example a ‘bext’
chunk for the Broadcast WAVE file.
A basic WAV file consists of three principal chunks, as shown in Figure
10.4, the RIFF chunk, the FORMAT chunk and the DATA chunk. The
RIFF chunk contains 12 bytes, the first four of which are the ASCII characters ‘RIFF ’, the next four indicating the number of bytes in the remainder
of the file (after the first eight) and the last four of which are the ASCII
characters ‘WAVE’. The format chunk contains information about the format of the sound file, including the number of audio channels, sampling
rate and bits per sample, as shown in Table 10.1.
The audio data chunk contains a sequence of bytes of audio sample
data, divided as shown in the FORMAT chunk. Unusually, if there are only
8 bits per sample or fewer each value is unsigned and ranges between 0
and 255 (decimal), whereas if the resolution is higher than this the data is
signed and ranges both positively and negatively around zero. Audio samples are interleaved by channel in time order, so that if the file contains two
channels a sample for the left channel is followed immediately by the associated sample for the right channel. The same is true of multiple channels
(one sample for time-coincident sample periods on each channel is inserted
at a time, starting with the lowest numbered channel), although basic WAV
files were nearly always just mono or two channel.
The RIFF WAVE format is extensible and can have additional chunks to
define enhanced functionality such as surround sound and other forms of
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Table 10.1
Contents of FORMAT chunk in a basic WAVE PCM file
Byte
ID
Contents
0–3
4–7
ckID
nChunkSize
8–9
wFormatTag
10–11
nChannels
12–15
16–19
20–21
nSamplesPerSec
nAvgBytesPerSec
nBlockAlign
22–23
nBitsPerSample
‘fmt_’ (ASCII characters)
Length of FORMAT chunk (binary, hex value:
&00000010)
Audio data format (e.g. &0001 ⫽ WAVE format
PCM) Other formats are allowed, for example IEEE
floating point and MPEG format (&0050 ⫽ MPEG 1)
Number of channels (e.g. &0001 ⫽ mono,
&0002 ⫽ stereo)
Sample rate (binary, in Hz)
Bytes per second
Bytes per sample: e.g. &0001 ⫽ 8 bit mono;
&0002 ⫽ 8 bit stereo or 16 bit mono; &0004 ⫽ 16
bit stereo
Bits per sample
coding. This is known as ‘WAVE-format extensible’ (see http://www.microsoft.com/hwdev/tech/audio/multichaud.asp). Chunks can include data relating to cue points, labels and associated data, for example. The Broadcast
WAVE format is one example of an enhanced WAVE file (see Fact File 10.1),
which is used widely in professional applications for interchange purposes.
MPEG audio file formats
It is possible to store MPEG-compressed audio in AIFF-C or WAVE files,
with the compression type noted in the appropriate header field. There are
also older MS-DOS file extensions used to denote MPEG audio files, notably .MPA (MPEG Audio) or .ABS (Audio Bit Stream). However, owing to the
ubiquity of the so-called ‘MP3’ format (MPEG 1, Layer 3) for audio distribution on the Internet, MPEG audio files are increasingly denoted with the
extension ‘.MP3’. Such files are relatively simple, being really no more than
MPEG audio frame data in sequence, each frame being preceded by a frame
header.
DSD-IFF file format
The DSD-IFF file format is based on a similar structure to other IFF-type
files, described above, except that it is modified slightly to allow for the large
file sizes that may be encountered with the high-resolution Direct Stream
Digital format used for SuperAudio CD. Specifically the container FORM
Audio File Formats for Digital Workstations
FA C T F I L E 1 0 . 1 B R O A D C A S T WAVE FORM AT
The Broadcast WAVE format, described in EBU Tech.
3285, was standardized by the European Broadcasting
Union (EBU) because of a need to ensure compatibility
of sound files and accompanying information when transferred between workstations. It is based on the RIFF WAVE
format described above, but contains an additional chunk
that is specific to the format (the ‘broadcast_audio_extension’ chunk, ID ⫽ ‘bext’) and also limits some aspects of
the WAVE format. Version 0 was published in 1997 and
Version 1 in 2001, the only difference being the addition
of an SMPTE UMID (Unique Material Identifier) in version
1 (this is a form of metadata). Such files currently only
contain either PCM or MPEG-format audio data.
Broadcast WAVE files contain at least three chunks:
the broadcast_audio_extension chunk, the format
chunk and the audio data chunk. The broadcast extension chunk contains the data shown in the table below.
Optionally files may also contain further chunks for specialized purposes and may contain chunks relating to
MPEG audio data (the ‘fact’ and ‘mpeg_audio_extension’
chunks). MPEG applications of the format are described
in EBU Tech. 3285, Supplement 1 and the audio data
chunk containing the MPEG data normally conforms to
the MP3 frame format.
A multichannel extension chunk defines the channel ordering, surround format, downmix coefficients for
creating a two-channel mix, and some descriptive information. There are also chunks defined for metadata
describing the audio contained within the file, such as
the ‘quality chunk’ (ckID ⫽ ‘qlty’), which together with
the coding history contained in the ‘bext’ chunk make
up the so-called ‘capturing report’. These are described
in Supplement 2 to EBU Tech. 3285. Finally there is a
chunk describing the peak audio level within a file, which
can aid automatic program level setting and program
interchange.
BWF files can be either mono, two-channel or multichannel (sometimes called polyfiles, or BWF-P), and utilities exist for separating polyfiles into individual mono files
which some applications require.
Broadcast audio extension chunk format
Data
Size (bytes)
Description
ckID
ckSize
Description
Originator
OriginatorReference
OriginationDate
OriginationTime
TimeReferenceLow
TimeReferenceHigh
Version
UMID
4
4
256
32
32
10
8
4
4
2
64
Reserved
190
Chunk ID ⫽ ‘bext’
Size of chunk
Description of the sound clip
Name of the originator
Unique identifier of the originator (issued by the EBU)
‘yyyy-mm-dd’
‘hh-mm-ss’
Low byte of the first sample count since midnight
High byte of the first sample count since midnight
BWF version number, e.g. &0001 is Version 1
UMID according to SMPTE 330M. If only a 32 byte UMID then
the second half should be padded with zeros
Reserved for extensions. Set to zero in Version 1
CodingHistory
Unrestricted
A series of ASCII strings, each terminated by CR/LF (carriage
return, line feed) describing each stage of the audio coding
history, according to EBU R-98
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chunk is labeled ‘FRM8’ and this identifies all local chunks that follow as
having ‘length’ indications that are 8 bytes long rather than the normal 4.
In other words, rather than a 4 byte chunk ID followed by a 4 byte length
indication, these files have a 4 byte ID followed by an 8 byte length indication. This allows for the definition of chunks with a length greater than 2
Gbytes, which may be needed for mastering SuperAudio CDs. There are
also various optional chunks that can be used for exchanging more detailed
information and comments such as might be used in project interchange.
Further details of this file format, and an excellent guide to the use of
DSD-IFF in project applications, can be found in the DSD-IFF specification, as described in the ‘Recommended further reading’ at the end of this
chapter.
Edit decision list (EDL) files and project interchange
EDL formats were historically proprietary but the need for open interchange
of project data has increased the use of standardized EDL structures and
‘packaged’ project formats to make projects transportable between systems
from different manufacturers. There is an old and widely used format for
EDLs in the video world that is known as the CMX-compatible form. CMX
is a well-known manufacturer of video editing equipment and most editing systems will read CMX EDLs for the sake of compatibility. These can
be used for basic audio purposes, and indeed a number of workstations can
read CMX EDL files for the purpose of auto-conforming audio edits to video
edits performed on a separate system. The CMX list defines the cut points
between source material and the various transition effects at joins, and it
can be translated reasonably well for the purpose of defining audio cut points
and their timecode locations, provided video frame accuracy is adequate.
Project interchange can involve the transfer of edit list, mixing, effects
and audio data. Many of these are proprietary, such as the Digidesign
ProTools session format. Software, such as SSL Pro-Convert, can be
obtained for audio and video workstations that translates EDLs or projects between a number of different systems to make interchange easier,
although it is clear that this process is not always problem-free and good
planning of in-house processes is vital. The OMFI (Open Media Framework
Interchange) structure, originally developed by Avid, was one early attempt
at an open project interchange format and contained a format for interchanging edit list data. Other options include XML-tagged formats that
identify different items in the edit list in a text-based form. AES-31 is now
gaining considerable popularity among workstation software manufacturers
Audio File Formats for Digital Workstations
as a simple means of exchanging audio editing projects between systems,
and is described in more detail below.
AES-31 format
AES-31 is an international standard designed to enable straightforward
interchange of audio files and projects between systems. Audio editing
packages are increasingly offering AES-31 as a simple interchange format
for edit lists. In Part 1 the standard specifies a disk format that is compatible with the FAT32 file system, a widely used structure for the formatting
of computer hard disks. Part 2 describes the use of the Broadcast WAVE
audio file format. Part 3 describes simple project interchange, including a
format for the communication of edit lists using ASCII text that can be
parsed by a computer as well as read by a human. The basis of this is the
edit decision markup language (EDML). It is not necessary to use all the
parts of AES-31 to make a satisfactory interchange of elements. For example, one could exchange an edit list according to part 3 without using a disk
based on part 1. Adherence to all the parts would mean that one could take
a removable disk from one system, containing sound files and a project file,
and the project would be readable directly by the receiving device.
EDML documents are limited to a 7 bit ASCII character set in which
white space delimits fields within records. Standard carriage return (CR)
and line-feed (LF) characters can be included to aid the readability of lists
but they are ignored by software that might parse the list. An event location
is described by a combination of time code value and sample count information. The time code value is represented in ASCII using conventional
hours, minutes, seconds and frames (e.g. HH:MM:SS:FF) and the optional
sample count is a four figure number denoting the number of samples
after the start of the frame concerned at which the event actually occurs.
This enables sample-accurate edit points to be specified. It is slightly more
complicated than this because the ASCII delimiters between the time code
fields are changed to indicate various parameters:
HH:MM delimiter ⫽ Frame count and timebase indicator (see Table
10.2)
MM:SS delimiter ⫽ Film frame indicator (if not applicable, use the
previous delimiter)
SS:FF delimiter ⫽ Video field and timecode type (see Table 10.3)
The delimiter before the sample count value is used to indicate the
audio sampling frequency, including all the pull-up and pull-down options
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Table 10.2
Frame count and timebase indicator coding in AES-31
Timebase
Frame count
30
25
24
Table 10.3
Unknown
?
!
#
1.000
|
.
⫽
1.001
:
/
-
Video field and timecode type indicator in AES-31
Video Field
Counting mode
PAL
NTSC non-drop-frame
NTSC drop-frame
Field 1
.
.
,
Field 2
:
:
;
(e.g. fs times 1/1.001). There are too many of these possibilities to list here
and the interested reader is referred to the standard for further information.
This is an example of a time code and (after the slash denoting 48 kHz
sampling frequency) optional sample count value:
14:57:24.03 / 0175
The Audio xDecision List (ADL) is contained between two ASCII keyword tags ⬍ADL⬎ and ⬍/ADL⬎. It in turn contains a number of sections, each contained within other keyword tags such as ⬍VERSION ⬎,
⬍PROJECT⬎, ⬍SYSTEM⬎ and ⬍SEQUENCE⬎. The edit points themselves are contained in the ⬍EVENT_LIST⬎ section. Each event begins
with the ASCII keyword ‘‘(Entry)’’, which serves to delimit events in the
list, followed by an entry number (32 bit integer, incrementing through
the list) and an entry type keyword to describe the nature of the event (e.g.
‘‘(Cut)’’). Each different event type then has a number of bytes following
that define the event more specifically. The following is an example of a
simple cut edit, as suggested by the standard:
(Entry) 0010 (Cut) F ‘‘FILE://VOL/DIR/FILE’’ 1 1 03:00:00;00/0000
01:00:00:00/0000 01:00:10:00/0000 _
This sequence essentially describes a cut edit, entry number 0010, the
source of which is the file (F) with the path shown, using channel 1 of the
Audio File Formats for Digital Workstations
source file (or just a mono file), placed on track 1 of the destination timeline, starting at timecode three hours in the source file, placed to begin at
one hour in the destination timeline (the ‘in point’) and to end ten seconds
later (the ‘out point’). Some workstation software packages store a timecode
value along with each sound file to indicate the nominal start time of the
original recording (e.g. BWF files contain a timestamp in the ‘bext’ chunk),
otherwise each sound file is assumed to start at time zero.
It is assumed that default crossfades will be handled by the workstation software itself. Most workstations introduce a basic short crossfade at
each edit point to avoid clicks, but this can be modified by ‘event modifier ’
information in the ADL. Such modifiers can be used to adjust the shape
and duration of a fade in or fade out at an edit point. There is also the
option to point at a rendered crossfade file for the edit point, as described
in Chapter 9.
MXF – the media exchange format
MXF was developed by the Pro-MPEG forum as a means of exchanging
audio, video and metadata between devices, primarily in television operations. It is based on the modern concept of media objects that are split into
‘essence’ and ‘metadata’. Essence files are the raw material (i.e. audio and
video) and the metadata describes things about the essence (such as where
to put it, where it came from and how to process it).
MXF files attempt to present the material in a ‘streaming’ format, that
is one that can be played out in real time, but they can also be exchanged
in conventional file transfer operations. As such they are normally considered to be finished program material, rather than material that is to be
processed somewhere downstream, designed for playout in broadcasting
environments. The bit stream is also said to be compatible with recording
on digital videotape devices.
AAF – the advanced authoring format
AAF is an authoring format for multimedia data that is supported by numerous vendors, including Avid which has adopted it as a migration path from
OMFI. Parts of OMFI 2.0 form the basis for parts of AAF and there are also
close similarities between AAF and MXF (described in the previous section).
Like the formats to which it has similarities, AAF is an object-oriented
format that combines essence and metadata within a container structure.
Unlike MXF it is designed for project interchange such that elements
within the project can be modified, post-processed and resynchronized. It
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is not, therefore, directly suitable as a streaming format but can easily be
converted to MXF for streaming if necessary.
Rather like OMFI it is designed to enable complex relationships to be
described between content elements, to map these elements onto a timeline,
to describe the processing of effects, synchronize streams of essence, retain
historical metadata and refer to external essence (essence not contained
within the AAF package itself). It has three essential parts: the AAF Object
Specification (which defines a container for essence and metadata, the logical
contents of objects and rules for relationships between them); the AAF LowLevel Container Specification (which defines a disk filing structure for the
data, based on Microsoft’s Structured Storage); and the AAF SDK Reference
Implementation (which is a software development kit that enables applications to deal with AAF files). The Object Specification is extensible in that it
allows new object classes to be defined for future development purposes.
The basic object hierarchy is illustrated in Figure 10.5, using an example of a typical audio post-production scenario. ‘Packages’ of metadata are
defined that describe either compositions, essence or physical media. Some
package types are very ‘close’ to the source material (they are at a lower level
in the object hierarchy, so to speak) – for example, a ‘file source package’
FIGURE 10.5
Graphical conceptualization
of some metadata package
relationships in AAF –
a simple audio postproduction example.
Time
Segment
Video slot
Segment
Audio 1 slot
Composition
package
Audio 2 slot
Audio 3 slot
Seg
Audio 4 slot
Source clip segment
refers to....
Slot 1
File source package
00:13:00:00
00:12:00:00
refers to....
Sound file on disk (essence)
Audio File Formats for Digital Workstations
might describe a particular sound file stored on disk. The metadata package, however, would not be the file itself, but it would describe its name and
where to find it. Higher-level packages would refer to these lower-level packages in order to put together a complex program. A composition package
is one that effectively describes how to assemble source clips to make up a
finished program. Some composition packages describe effects that require
a number of elements of essence to be combined or processed in some way.
Packages can have a number of ‘slots’. These are a bit like tracks in more
conventional terminology, each slot describing only one kind of essence
(e.g. audio, video, graphics). Slots can be static (not time-dependent),
timeline (running against a timing reference) or event-based (one-shot,
triggered events). Slots have segments that can be source clips, sequences,
effects or fillers. A source clip segment can refer to a particular part of a slot
in a separate essence package (so it could refer to a short portion of a sound
file that is described in an essence package, for example).
Disk pre-mastering formats
The original tape format for submitting CD masters to pressing plants was
Sony’s audio-dedicated PCM 1610/1630 format on U-matic video tape.
This is now ‘old technology ’ has been replaced by alternatives based on
more recent data storage media and file storage protocols. These include
the PMCD (pre-master CD), CD-R, Exabyte and DLT tape formats. DVD
mastering also requires high-capacity media for transferring the many
gigabytes of information to mastering houses so that glass masters can be
created.
The Disk Description Protocol (DDP) developed by Doug Carson and
Associates is now widely used for describing disk masters. Version 1 of the
DDP layed down the basic data structure but said little about higher-level
issues involved in interchange, making it more than a little complicated
for manufacturers to ensure that DDP masters from one system would be
readable on another. Version 2 addressed some of these issues.
DDP is a protocol for describing the contents of a disk, which is not
medium specific. That said it is common to interchange CD masters with
DDP data on 8 mm Exabyte data cartridges and DVD masters are typically
transferred on DLT Type III or IV compact tapes or on DVD-R(A) format
disks with CMF (cutting master format) DDP headers. DDP files can be
supplied separately to the audio data if necessary. DDP can be used for
interchanging the data for a number of different disk formats, such as CDROM, CD-DA, CD-I and CD-ROM-XA, DVD-Video and -Audio, and the
protocol is really extremely simple. It consists of a number of ‘streams’ of
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data, each of which carries different information to describe the contents of
the disk. These streams may be either a series of packets of data transferred
over a network, files on a disk or tape, or raw blocks of data independent of
any filing system. The DDP protocol simply maps its data into whatever
block or packet size is used by the medium concerned, provided that the
block or packet size is at least 128 bytes. Either a standard computer filing structure can be used, in which case each stream is contained within a
named file, or the storage medium is used ‘raw ’ with each stream starting
at a designated sector or block address.
The ANSI tape labeling specification is used to label the tapes used for
DDP transfers. This allows the names and locations of the various streams
to be identified. The principal streams included in a DDP transfer for CD
mastering are as follows:
1. DDP ID stream or ‘DDPID ’ file. 128 bytes long, describing the type
and level of DDP information, various ‘vital statistics’ about the
other DDP files and their location on the medium (in the case of
physically addressed media), and a user text field (not transferred to
the CD).
2. DDP Map stream or ‘DDPMS’ file. This is a stream of 128 byte data
packets which together give a map of the CD contents, showing
what types of CD data are to be recorded in each part of the CD,
how long the streams are, what types of subcode are included, and so
forth. Pointers are included to the relevant text, subcode and main
streams (or files) for each part of the CD.
3. Text stream. An optional stream containing text to describe the
titling information for volumes, tracks or index points (not currently
stored in CD formats), or for other text comments. If stored as a file,
its name is indicated in the appropriate map packet.
4. Subcode stream. Optionally contains information about the
subcode data to be included within a part of the disk, particularly for
CD-DA. If stored as a file, its name is indicated in the appropriate
map packet.
5. Main stream. Contains the main data to be stored on a part of the
CD, treated simply as a stream of bytes, irrespective of the block or
packet size used. More than one of these files can be used in cases
of mixed-mode disks, but there is normally only one in the case of
a conventional audio CD. If stored as a file, its name is indicated in
the appropriate map packet.
Consumer Optical Disk Formats
CONSUMER OPTICAL DISK FORMATS
Compact discs and drives
The CD is not immediately suitable for real-time audio editing and production, partly because of its relatively slow access time compared with hard
disks, but can be seen to have considerable value for the storage and transfer of sound material that does not require real-time editing. Broadcasters
use them for sound effects libraries, and studios and mastering facilities
use them for providing customers and record companies with ‘acetates’ or
test pressings of a new recording. They have also become quite popular as a
means of transferring finished masters to a CD pressing plant in the form
of the PMCD (pre-master CD). They are ideal as a means of ‘proofing’ CDROMs and other CD formats, and can be used as low-cost backup storage
for computer data.
Compact Discs (CDs) are familiar to most people as a consumer readonly optical disc for audio (CD-DA) or data (CD-ROM) storage. Standard
audio CDs (CD-DA) conform to the Red Book standard published by
Philips. The CD-ROM standard (Yellow Book) divides the CD into a structure with 2048 byte sectors, adds an extra layer of error protection, and
makes it useful for general purpose data storage including the distribution
of sound and video in the form of computer data files. It is possible to find
discs with mixed modes, containing sections in CD-ROM format and sections in CD-Audio format. The CD Plus is one such example.
CD-R is the recordable CD, and may be used for recording CD-Audio
format or other CD formats using a suitable drive and software. The Orange
Book, Part 2, contains information on the additional features of CD-R, such
as the area in the center of the disc where data specific to CD-R recordings
is stored. Audio CDs recorded to the Orange Book standard can be ‘fixed’
to give them a standard Red Book table of contents (TOC), allowing them
to be replayed on any conventional CD player. Once fixed into this form,
the CD-R may not subsequently be added to or changed, but prior to this
there is a certain amount of flexibility, as discussed below. CD-RW discs are
erasable and work on phase-change principles, requiring a drive compatible
with this technology, being described in the Orange Book, Part 3.
The degree of reflectivity of CD-RW discs is much lower than that of
typical CD-R and CD-ROM. This means that some early drives and players may have difficulties reading them. However, the ‘multi-read’ specification developed by the OSTA (Optical Storage Technology Association)
describes a drive that should read all types of CD, so recent drives should
have no difficulties here.
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DVD
DVD was the natural successor to CD, being a higher density optical disc
format aimed at the consumer market, having the same diameter as CD
and many similar physical features. It uses a different laser wavelength to
CD (635–650 nm as opposed to 780 nm) so multi-standard drives need to
be able to accommodate both. Data storage capacity depends on the number of sides and layers to the disc, but ranges from 4.7 Gbytes (single-layer,
single-sided) up to about 18 Gbytes (double-layer, double-sided). The data
transfer rate at ‘one times’ speed is just over 11 Mbit/s.
DVD can be used as a general purpose data storage medium. Like CD,
there are numerous different variants on the recordable DVD, partly owing
to competition between the numerous different ‘factions’ in the DVD consortium. These include DVD-R, DVD-RAM, DVD-RW and DVD ⫹ RW, all
of which are based on similar principles but have slightly different features,
leading to a compatibility minefield (see Fact File 10.2). The ‘DVD Multi’
guidelines produced by the DVD Forum are an attempt to foster greater
FA C T F I L E 1 0 . 2 R E C O R D A B L E DVD FORM ATS
Recordable DVD type
Description
DVD-R (A and G)
DVD equivalent of CD-R. One-time recordable in sequential manner, replayable
on virtually any DVD-ROM drive. Supports ‘incremental writing’ or ‘disc at once’
recording. Capacity either 3.95 (early discs) or 4.7 Gbyte per side. ‘Authoring’ (A)
version (recording laser wavelength ⫽ 635 nm) can be used for pre-mastering DVDs
for pressing, including DDP data for disc mastering (see Chapter 6). ‘General’ (G)
version (recording laser wavelength ⫽ 650 nm) intended for consumer use, having
various ‘content protection’ features that prevent encrypted commercial releases
from being cloned
Sectored format, rather more like a hard disk in data structure when compared with
DVD-R. Uses phase-change (PD-type) principles allowing direct overwrite. Version 2
discs allow 4.7 Gbyte per side (reduced to about 4.2 Gbyte after formatting). Type 1
cartridges are sealed and Type 2 allow the disc to be removed. Double-sided discs
only come in sealed cartridges. Can be rewritten about 100 000 times. The recent
Type 3 is a bare disc that can be placed in an open cartridge for recording
Pioneer development, similar to CD-RW in structure, involving sequential writing.
Does not involve a cartridge. Can be rewritten about 1000 times. 4.7 Gbyte per side
Non-DVD-Forum alternative to DVD-RAM (and not compatible), allowing direct
overwrite. No cartridge. Data can be written in either CLV (for video recording) or
CAV (for random access storage) modes. There is also a write-once version known as
DVD⫹R
DVD-RAM
DVD-RW
DVD ⫹ RW
Consumer Optical Disk Formats
compatibility between DVD drives and discs, and many drives are now
available that will read and write most of the DVD formats.
Writeable DVDs are a useful option for backup of large projects, particularly DVD-RAM because of its many-times overwriting capacity and its
hard disk-like behavior. It is possible that a format like DVD-RAM could
be used as primary storage in a multitrack recording/editing system, as it
has sufficient performance for a limited number of channels and it has the
great advantage of being removable. Indeed one company has used it in a
multichannel location recorder. However, it is likely that hard disks will
retain the performance edge for the foreseeable future.
DVD-Video is the format originally defined for consumer distribution of
movies with surround sound, typically incorporating MPEG-2 video encoding and Dolby Digital surround sound encoding. It also allows for up to
eight channels of 48 or 96 kHz linear PCM audio, at up to 24 bit resolution. DVD-Audio was intended for very high-quality multichannel audio
reproduction and allowed for linear PCM sampling rates up to 192 kHz,
with numerous configurations of audio channels for different surround
modes, and optional lossless data reduction (MLP). However, it has not
been widely adopted in the commercial music industry.
DVD-Audio had a number of options for choosing the sampling frequencies and resolutions of different channel groups, it being possible to use a
different resolution on the front channels from that used on the rear, for
example. The format was more versatile in respect of sampling frequency
than DVD-Video, having also accommodated multiples of the CD sample
frequency of 44.1 kHz as options (the DVD-Video format allows only for
multiples of 48 kHz). Bit resolution could be 16, 20 or 24 bits per channel,
and again this could be divided unequally between the channels, according
to the channel group split described below.
Meridian Lossless Packing (MLP) was licensed through Dolby
Laboratories for DVD-A and is a lossless coding technique designed to
reduce the data rate of audio signals without compromising sound quality.
It has both a variable bit rate mode and a fixed bit rate mode. The variable mode delivers the optimum compression for storing audio in computer
data files, but the fixed mode is important for DVD applications where one
must be able to guarantee a certain reduction in peak bit rate.
Super Audio CD (SACD)
Version 1.0 of the SACD specification is described in the ‘Scarlet Book’,
available from Philips licensing department. SACD uses DSD (Direct
Stream Digital) as a means of representing audio signals, as described
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in Chapter 8, so requires audio to be
sourced in or converted to this form.
Multichannel area
SACD aims to provide a playing time of
at least 74 minutes for both two channel and six channel balances. The disc
2-channel area
is divided into two regions, one for twochannel audio, the other for multichannel, as shown in Figure 10.6. A lossless
data packing method known as Direct
File system
Stream Transfer (DST) can be used to
Master TOC
achieve roughly 2:1 data reduction of the
signal stored on disc so as to enable highquality multichannel audio on the same
disc as the two channel mix. SACD has
only achieved a relatively modest market penetration compared with formats
such as CD and DVD-Video, but is used
by some specialized high-quality record
labels. A considerable number of recordings have nonetheless been issued in the
FIGURE 10.6 Different regions of a Super Audio CD, showing
SACD format.
separate two-channel and multichannel regions.
SACDs can be manufactured as single- or dual-layer discs, with the option of
the second layer being a Red Book CD layer (the so-called ‘hybrid disc’ that
will also play on a normal CD player). SACDs, not being a formal part of the
DVD hierarchy of standards (although using some of the optical disc technology), do not have the same options for DVD-Video objects as DVD-Audio.
The disc is designed first and foremost as a super-high-quality audio medium.
Nonetheless there is provision for additional data in a separate area of the
disc. The content and capacity of this is not specified but could be video clips,
text or graphics, for example.
Extra data area
Blu-Ray disk
The Blu-Ray disk is a higher density optical disk format than DVD, which
uses a shorter wavelength blue-violet laser (wavelength 405 nm) to achieve
a high packing density of data on the disk surface. Single-layer disks offer
25 Gbytes of storage and dual-layer disks offer 50 Gbytes, and the basic
transfer rate is also higher than DVD at around 36 Mbit/s although a higher
rate of 54 Mbit/s is required for HD movie replay, which is achieved by
using at least 1.5 times playback speed. Like DVD, a range of read-only,
Interconnecting Digital Audio Devices
writeable and rewriteable formats is possible. There is an audio-only version of the player specification, known as BD-Audio, which does not have
to be able to decode video, making possible a high-resolution surround playback format that might offer an alternative to DVD-Audio or SACD. Audioonly transfer rates of the disk vary depending on the format concerned.
As far as audio formats are concerned, Linear PCM, Dolby Digital and
DTS Digital Surround are mandatory in Blu-Ray players and recorders, but
it is up to individual studios to decide what formats to include on their disk
releases. Alternative optional audio formats include higher-resolution versions of Dolby and DTS formats, known as Dolby Digital Plus and DTS-HD
respectively, as well as losslessly encoded versions known as Dolby TrueHD
and DTS-HD Master Audio. High sampling frequencies (up to 192 kHz) are
possible on Blu-Ray, as are audio sample resolutions of 16, 20 or 24 bits. The
standard limits audio reproduction to six channels of 192 kHz, 24 bit uncompressed digital audio, which gives rise to a data transfer rate of 27.7 Mbit/s.
INTERCONNECTING DIGITAL AUDIO DEVICES
Introduction
In the case of analog interconnection between devices, replayed digital
audio is converted to the analog domain by the replay machine’s D/A convertors, routed to the recording machine via a conventional audio cable and
then reconverted to the digital domain by the recording machine’s A/D convertors. The audio is subject to any gain changes that might be introduced
by level differences between output and input, or by the record gain control
of the recorder and the replay gain control of the player. Analog domain
copying is necessary if any analog processing of the signal is to happen
in between one device and another, such as gain correction, equalization,
or the addition of effects such as reverberation. Most of these operations,
though, are now possible in the digital domain.
An analog domain copy cannot be said to be a perfect copy or a clone
of the original master, because the data values will not be exactly the same
(owing to slight differences in recording level, differences between convertors, the addition of noise, and so on). For a clone it is necessary to make
a true digital copy. This can either involve a file copying process, perhaps over a network using a workstation, or a digital interface or network
infrastructure may be used for the streamed interconnection of recording
systems and other audio devices such as mixers and effects units.
Professional digital audio systems, and some consumer systems, have
digital interfaces conforming to one of the standard protocols and allow for
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a number of channels of digital audio data to be transferred between devices
with no loss of sound quality. Any number of generations of digital copies
may be made without affecting the sound quality of the latest generation,
provided that errors have been fully corrected. (This assumes that the audio
is in a linear PCM format and has not been subject to low bit rate decoding
and re-encoding.) The digital outputs of a recording device are taken from
a point in the signal chain after error correction, which results in the copy
being error corrected. The copy does not suffer from any errors that existed
in the master, provided that those errors were correctable. This process
takes place in real time, requiring the operator to put the receiving device
into record mode such that it simply stores the incoming stream of audio
data. Any accompanying metadata may or may not be recorded (often most
of it is not).
Making a copy of a recording or transferring audio data between devices
using any of the digital interface standards involves the connection of appropriate cables between player and recorder, and the switching of the receiver’s input to ‘digital’ as opposed to ‘analog’, since this sets it to accept a
signal from the digital input as opposed to the A/D convertor. It is necessary
for both machines to be operating at the same sampling frequency (unless
a sampling frequency convertor is used) and may require the recorder to be
switched to ‘external sync’ mode, so that it can lock its sampling frequency
to that of the player. (Some devices such as effects may lock themselves to
an incoming digital signal as a matter of course.) Alternatively (and preferably) a common reference (e.g. word clock) signal may be used to synchronize
all devices that are to be interconnected digitally. If one of these methods
of ensuring a common sampling frequency is not used then either audio
will not be decoded at all by the receiver, or regular clicks will be audible at
a rate corresponding to the difference between the two sampling frequencies (at which point samples are either skipped or repeated owing to the
‘sample slippage’ that is occurring between the two machines). A receiver
should be capable of at least the same quantizing resolution (number of
bits per sample) as the source device, otherwise audio resolution will be
lost. If there is a difference in resolution between the systems it is advisable to use a processor in between the machines that optimally dithers the
signal for the new resolution, or alternatively to use redithering options
on the source machine to prepare the signal for its new resolution (see
Chapter 8).
Increasingly generic computer data interconnects are used to transfer
audio as explained in Fact File 10.3.
Interconnecting Digital Audio Devices
FA C T F I L E 1 0 . 3 C O MP U T E R N E TW ORKS VS DI GI TAL
AU D I O I N T E R FA C E S
Dedicated ‘streaming’ interfaces, as employed in broadcasting, production and post-production environments,
are the digital audio equivalent of analog signal cables,
down which signals for one or more channels are carried
in real time from one point to another, possibly with some
auxiliary information (metadata) attached. An example
is the AES-3 interface, described in the main text. Such
an audio interface uses a data format dedicated to audio
purposes, whereas a computer data network may carry
numerous types of information.
Dedicated interfaces are normally unidirectional,
point-to-point connections, and should be distinguished
from computer data interconnects and networks that are
often bidirectional and carry data in a packet format for
numerous sources and destinations. With dedicated interfaces sources may be connected to destinations using a
routing matrix or by patching individual connections, very
much as with analog signals. Audio data are transmitted
in an unbroken stream, there is no handshaking process
involved in the data transfer, and erroneous data are not
retransmitted because there is no mechanism for requesting its retransmission. The data rate of a dedicated audio
interface is usually directly related to the audio sampling
frequency, word length and number of channels of the
audio data to be transmitted, ensuring that the interface is
always capable of serving the specified number of channels. If a channel is unused for some reason its capacity
is not normally available for assigning to other purposes
(such as higher-speed transfer of another channel, for
example).
There is an increasing trend towards employing standard computer interconnects and networks to transfer
audio information, as opposed to using dedicated audio
interfaces. Such computer networks are typically used for
a variety of purposes in general data communications and
they may need to be adapted for audio applications that
require sample-accurate real-time transfer. The increasing ubiquity of computer systems in audio environments
makes it inevitable that generic data communication
technology will gradually take the place of dedicated interfaces. It also makes sense economically to take advantage
of the ‘mass market’ features of the computer industry.
Computer networks are typically general purpose data
carriers that may have asynchronous features and may
not always have the inherent quality-of-service (QoS) features that are required for ‘streaming’ applications. They
also normally use an addressing structure that enables
packets of data to be carried from one of a number of
sources to one of a number of destinations and such
packets will share the connection in a more or less controlled way. Data transport protocols such as TCP/IP are
often used as a universal means of managing the transfer
of data from place to place, adding overheads in terms of
data rate, delay and error handling that may work against
the efficient transfer of audio. Such networks may be
designed primarily for file transfer applications where the
time taken to transfer the file is not a crucial factor – ‘as
fast as possible’ will do. This has required some special
techniques to be developed for carrying real-time data
such as audio information.
Desktop computers and consumer equipment are
also increasingly equipped with general purpose serial
data interfaces such as USB (Universal Serial Bus) and
Firewire (IEEE 1394). These are examples of personal
area network (PAN) technology, allowing a number of
devices to be interconnected within a limited range around
the user. These have a high enough data rate to carry a
number of channels of audio data over relatively short distances, either over copper or optical fiber. Audio protocols
also exist for these as discussed in the main text.
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Dedicated audio interface formats
There are a number of types of digital interface, some of which are international standards and others of which are manufacturer specific. They all
carry digital audio for one or more channels with at least 16 bit resolution
and will operate at the standard sampling rates of 44.1 and 48 kHz, as well
as at 32 kHz if necessary, some having a degree of latitude for varispeed.
Some interface standards have been adapted to handle higher sampling
frequencies such as 88.2 and 96 kHz. The interfaces vary as to how many
physical interconnections are required. Some require one link per channel
plus a synchronization signal, whilst others carry all the audio information
plus synchronization information over one cable.
The most common interfaces are described below in outline. It is common for subtle incompatibilities to arise between devices, even when
interconnected with a standard interface, owing to the different ways in
which non-audio information is implemented. This can result in anything
from minor operational problems to total non-communication and the
causes and remedies are unfortunately far too detailed to go into here.
The reader is referred to The Digital Interface Handbook by Rumsey and
Watkinson, as well as to the standards themselves, if a greater understanding of the intricacies of digital audio interfaces is required.
The AES/EBU interface (AES-3)
The AES-3 interface, described almost identically in AES-3-1992, IEC 60958
and EBU Tech. 3250E among others, allows for two channels of digital audio
(A and B) to be transferred serially over one balanced interface, using drivers
and receivers similar to those used in the RS422 data transmission standard, with an output voltage of between 2 and 7 volts as shown in Figure
10.7. The interface allows two channels of audio to be transferred over distances up to 100 m, but longer distances may be covered using combinations
FIGURE 10.7
Recommended electrical
circuit for use with the
standard two-channel
interface.
Driver
Cable
Receiver
3
3
2
2
1
1
EQ
Interconnecting Digital Audio Devices
of appropriate cabling, equalization and termination. Standard XLR-3 connectors are used, often labeled DI (for digital in) and DO (for digital out).
Each audio sample is contained within a ‘subframe’ (see Figure 10.8),
and each subframe begins with one of three synchronizing patterns to identify the sample as either the A or B channel, or to mark the start of a new
channel status block (see Figure 10.9). These synchronizing patterns violate the rules of bi-phase mark coding (see below) and are easily identified
by a decoder. One frame (containing two audio samples) is normally transmitted in the time period of one audio sample, so the data rate varies with
the sampling frequency. (Note, though, that the recently introduced ‘single-channel-double-sampling-frequency ’ mode of the interface allows two
samples for one channel to be transmitted within a single frame in order to
allow the transport of audio at 88.2 or 96 kHz sampling frequency.)
1 Frame = 64 bits; Duration = 1 sample period = 20.8 μs @ 48 kHz sampling rate
Subframe A = 32 bits
4
4
Subframe B = 32 bits
20
4
4
Audio sample data
Sync Aux
4
20
4
Audio sample data
Sync Aux
LSB
LSB
VUCP
VUCP
1 bit
period
FIGURE 10.9
Three different preambles
(X, Y and Z) are used to
synchronize a receiver at
the starts of subframes.
X (M)
Y (W)
Z (B)
X
Channel
A
Y
Channel
B
32
bit periods
Frame 191
FIGURE 10.8
Format of the standard twochannel interface frame.
Z
Channel
A
Subframe
Channel
B
Y
X
Channel
A
Y
Subframe
Frame 0
Start of new channel
status block
Frame 1
Channel
B
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CHAPTER 10: Digital Audio Formats and Interchange
Byte
0
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
Basic control data
Mode and user bit management
Use of aux bits, alignment and audio word length
Multichannel mode and channel number
Sampling frequency and sync reference
Reserved
Source identification (4 bytes of 7 bit ASCII, no
parity)
Destination identification (4 bytes of 7 bit ASCII, no
parity)
Local sample address code (32 bit binary)
Time-of-day sample address code (32 bit binary)
Channel status reliability flags
CRCC
FIGURE 10.10 Overview of the professional channel status
block.
0
1
1
0
1
Original
NRZ data
Bi-phase mark
coded data
Additional data is carried within the subframe in the form of 4 bits of auxiliary data
(which may either be used for additional
audio resolution or for other purposes such as
low-quality speech), a validity bit (V), a user
bit (U), a channel status bit (C) and a parity
bit (P), making 32 bits per subframe and 64
bits per frame. Channel status bits are aggregated at the receiver to form a 24 byte word
every 192 frames, and each bit of this word
has a specific function relating to interface
operation, an overview of which is shown in
Figure 10.10. Examples of bit usage in this
word are the signaling of sampling frequency
and pre-emphasis, as well as the carrying of
a sample address ‘timecode’ and labeling of
source and destination. Bit 1 of the first byte
signifies whether the interface is operating
according to the professional (set to 1) or consumer (set to 0) specification.
Bi-phase mark coding, the same channel
code as used for SMPTE/EBU timecode, is
used in order to ensure that the data is selfclocking, of limited bandwidth, DC free, and
polarity independent, as shown in Figure
10.11. The interface has to accommodate
a wide range of cable types and a nominal
110 ohm characteristic impedance is recommended. Originally (AES-3-1985) up to four
receivers with a nominal input impedance of
250 ohms could be connected across a single
professional interface cable, but a later modification to the standard recommended the use
of a single receiver per transmitter, having a
nominal input impedance of 110 ohms.
FIGURE 10.11 An example of the bi-phase mark channel code.
Standard consumer interface (IEC 60958-3)
The most common consumer interface (historically related to SPDIF – the
Sony/Philips digital interface) is very similar to the AES-3 interface, but uses
Interconnecting Digital Audio Devices
Driver
Cable
Receiver
unbalanced electrical interconnection over a coaxial cable having
a characteristic impedance of 75 ohms, as shown in Figure
10.12. It can be found on many items of semi-professional or
consumer digital audio equipment, such as CD players, DVD
players and DAT machines, and is also widely used on computer
sound cards because of the small physical size of the connectors. It usually terminates in an RCA phono connector, although
some equipment makes use of optical fiber interconnects (TOSlink) carrying the same data. Format convertors are available
for converting consumer format signals to the professional format, and vice versa, and for converting between electrical and
optical formats. Both the professional (AES-3 equivalent) and
consumer interfaces are capable of carrying data-reduced stereo
and surround audio signals such as MPEG and Dolby Digital as
described in Fact File 10.4.
The data format of subframes is the same as that used in the
professional interface, but the channel status implementation is
almost completely different, as shown in Figure 10.13. The second byte of channel status in the consumer interface has been
set aside for the indication of ‘category codes’, these being set to
define the type of consumer usage. Examples of defined categories
are (00000000) for the General category, (10000000) for Compact
Disc and (11000000) for a DAT machine. Once the category has
been defined, the receiver is expected to interpret certain bits of
the channel status word in a particular way, depending on the
category. For example, in CD usage, the four control bits from the
CD’s ‘Q’ channel subcode are inserted into the first four control
bits of the channel status block (bits 1–4). Copy protection can be
FIGURE 10.12
The consumer electrical
interface. (transformer and
capacitor are optional but
may improve the electrical
characteristics of the
interface.)
Byte
0
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
Basic control and mode data
Category code
Source and channel number
Sampling rate and clock accuracy
Depends on application
Default to binary 0
16
17
18
19
20
21
22
23
FIGURE 10.13 Overview of the
consumer channel status block.
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FACT FILE 10.4 CARRYING DATA-REDUCED AUDIO
The increased use of data-reduced multichannel audio
has resulted in methods by which such data can be carried over standard two-channel interfaces, for either professional or consumer purposes. This makes use of the
‘non-audio’ or ‘other uses’ mode of the interface, indicated in the second bit of channel status, which tells
conventional PCM audio decoders that the information
is some other form of data that should not be converted
directly to analog audio. Because data-reduced audio
has a much lower rate than the PCM audio from which
it was derived, a number of audio channels can be carried in a data stream that occupies no more space than
two channels of conventional PCM. These applications of
the interface are described in SMPTE 337M (concerned
with professional applications) and IEC 61937, although
the two are not identical. SMPTE 338M and 339M specify data types to be used with this standard. The SMPTE
standard packs the compressed audio data into 16, 20 or
24 bits of the audio part of the AES-3 subframe and can
use the two subframes independently (e.g. one for PCM
audio and the other for data-reduced audio), whereas the
IEC standard only uses 16 bits and treats both subframes
the same way.
Consumer use of this mode is evident on DVD players,
for example, for connecting them to home cinema decoders. Here the Dolby Digital or DTS-encoded surround sound
is not decoded in the player but in the attached receiver/
decoder. IEC 61937 has parts, either pending or published,
dealing with a range of different codecs including ATRAC,
Dolby AC-3, DTS and MPEG (various flavors). An ordinary
PCM convertor trying to decode such a signal would simply reproduce it as a loud, rather unpleasant noise, which is
not advised and does not normally happen if the second bit
of channel status is correctly observed. Professional applications of the mode vary, but are likely to be increasingly
encountered in conjunction with Dolby E data reduction – a
relatively recent development involving mild data reduction
for professional multichannel applications in which users
wish to continue making use of existing AES-3-compatible equipment (e.g. VTRs, switchers and routers). Dolby
E enables 5.1-channel surround audio to be carried over
conventional two-channel interfaces and through AES-3transparent equipment at a typical rate of about 1.92 Mbit/s
(depending on how many bits of the audio subframe are
employed). It is designed so that it can be switched or edited
at video frame boundaries without disturbing the audio.
implemented in consumer-interfaced equipment, according to the Serial Copy
Management System (SCMS).
The user bits of the consumer interface are often used to carry information derived from the subcode of recordings, such as track identification
and cue point data. This can be used when copying CDs and DAT tapes,
for example, to ensure that track start ID markers are copied along with
the audio data. This information is not normally carried over AES/EBU
interfaces.
Proprietary digital interfaces
Tascam’s interfaces became popular owing to the widespread use of the
company’s DA-88 multitrack recorder and derivatives. The primary TDIF1 interface uses a 25-pin D-sub connector to carry eight channels of audio
information in two directions (in and out of the device), sampling frequency
Interconnecting Digital Audio Devices
Word length
Parity
User
Odd audio (24 bits)
Even audio (24 bits)
MSB
Emphasis
L/R
switch
signal
and pre-emphasis information (on separate wires, two for fs and one for
emphasis) and a synchronizing signal. The interface is unbalanced and uses
CMOS voltage levels. Each data connection carries two channels of audio
data, odd channel and MSB first, as shown in Figure 10.14. As can be seen,
the audio data can be up to 24 bits long, followed by 2 bits to signal the
word length, 1 bit to signal emphasis and 1 bit for parity. There are also 4
user bits per channel that are not usually used.
The Alesis ADAT multichannel optical digital interface, commonly
referred to as the ‘light pipe’ interface or simply ‘ADAT Optical’, is a serial,
self-clocking, optical interface that carries eight channels of audio information. It is described in US Patent 5,297,181: ‘Method and apparatus for providing a digital audio interface protocol’. The interface is capable of carrying
up to 24 bits of digital audio data for each channel and the eight channels
of data are combined into one serial frame that is transmitted at the sampling frequency. The data is encoded in NRZI format for transmission, with
forced ones inserted every 5 bits (except during the sync pattern) to provide clock content. This can be used to synchronize the sampling clock of a
receiving device if required, although some devices require the use of a separate 9-pin ADAT sync cable for synchronization. The sampling frequency
is normally limited to 48 kHz with varispeed up to 50.4 kHz and TOSLINK
optical connectors are typically employed (Toshiba TOCP172 or equivalent).
In order to operate at 96 kHz sampling frequency some implementations
use a ‘double-speed’ mode in which two channels are used to transmit one
channel’s audio data (naturally halving the number of channels handled by
one serial interface). Although 5 m lengths of optical fiber are the maximum
recommended, longer distances may be covered if all the components of the
interface are of good quality and clean. Experimentation is required.
As shown in Figure 10.15 the frame consists of an 11 bit sync pattern
consisting of 10 zeros followed by a forced one. This is followed by 4 user
bits (not normally used and set to zero), the first forced one, then the first
FIGURE 10.14
Basic Format of TDIF data
and LRsync signal.
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Sync
User
Chan. 1
Chan. 8
MSB
FIGURE 10.15 Basic format of ADAT data.
FIGURE 10.16
Direct Stream Digital
interface data is either
transmitted ‘raw’, as shown
at (a), or phase modulated,
as in the SDIF-3 format
shown at (b).
(a)
DSD-raw
Bit cell transitions
(b)
SDIF-3
audio channel sample (with forced ones every 5 bits), the second audio
channel sample, and so on.
SDIF is the original Sony interface for digital audio, most commonly
encountered in SDIF-2 format on BNC connectors, along with a word clock
signal. However, this is not often used these days. SDIF-3 is Sony’s interface for high-resolution DSD data (see Chapter 8), although some early
DSD equipment used a data format known as ‘DSD-raw ’, which was simply
a stream of DSD samples in non-return-to-zero (NRZ) form, as shown in
Figure 10.16(a). (The latter is essentially the same as SDIF-2.) In SDIF-3 data
is carried over 75 ohm unbalanced coaxial cables, terminating in BNC connectors. The bit rate is twice the DSD sampling frequency (or 5.6448 Mbit/s
at the sampling frequency given above) because phase modulation is used
for data transmission as shown in Figure 10.16(b). A separate word clock at
44.1 kHz is used for synchronization purposes. It is also possible to encounter a DSD clock signal connection at the 64 times 44.1 kHz (2.8224 MHz).
Sony also developed a multichannel interface for DSD signals, capable of carrying 24 channels over a single physical link. The transmission
method is based on the same technology as used for the Ethernet 100BASETX (100 Mbit/s) twisted-pair physical layer (PHY), but it is used in this
application to create a point-to-point audio interface. Category 5 cabling is
used, as for Ethernet, consisting of eight conductors. Two pairs are used for
bi-directional audio data and the other two pairs for clock signals, one in
each direction.
Twenty-four channels of DSD audio require a total bit rate of 67.7 Mbit/s,
leaving an appreciable spare capacity for additional data. In the MAC-DSD
Interconnecting Digital Audio Devices
interface this is used for error correction (parity) data, frame header and
auxiliary information. Data is formed into frames that can contain
Ethernet MAC headers and optional network addresses for compatibility
with network systems. Audio data within the frame is formed into 352 32
bit blocks, 24 bits of each being individual channel samples, six of which
are parity bits and two of which are auxiliary bits.
More recently Sony introduced ‘SuperMAC ’ which is capable of handling either DSD or PCM audio with very low latency (delay), typically
less than 50 μs, over Cat-5 Ethernet cables using the 100BASE-TX physical
layer. The number of channels carried depends on the sampling frequency.
Twenty-four bidirectional DSD channels can be handled, or 48 PCM channels at 44.1/48 kHz, reducing proportionately as the sampling frequency
increases. In conventional PCM mode the interface is transparent to AES-3
data including user and channel status information. Up to 5 Mbit/s of
Ethernet control information can be carried in addition. A means of interchange based on this was standardized by the AES as AES-50. ‘HyperMAC ’
runs even faster, carrying up to 384 audio channels on gigabit Ethernet
Cat-6 cable or optical fiber, together with 100 Mbit/s Ethernet control data.
Recently Sony sold this networking technology to Klark Teknik.
The advantage of these interfaces is that audio data thus formatted can
be carried over the physical drivers and cables common to Ethernet networks, carrying a lot of audio at high speed. In a sense these interfaces bridge
the conceptual gap between dedicated audio interfaces and generic computer
networks, as they use some of the hardware and the physical layer of a computer network to transfer audio in a convenient form. They do not, however,
employ all the higher layers of computer network protocols as mentioned in
the next section. This means that the networking protocol overhead is relatively low, minimal buffering is required and latency can be kept to a minimum. Dedicated routing equipment is, however, required. One of the main
applications so far has been in a Midas router for live performance mixing.
Data networks and computer interconnects
A network carries data either on wire or optical fiber, and is normally
shared between a number of devices and users. The sharing is achieved
by containing the data in packets of a limited number of bytes (usually
between 64 and 1518), each with an address attached. The packets usually share a common physical link, normally a high-speed serial bus of
some kind, being multiplexed in time either using a regular slot structure
synchronized to a system clock (isochronous transfer) or in an asynchronous fashion whereby the time interval between packets may be varied or
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transmission may not be regular, as shown in Figure 10.17. The length of
packets may not be constant, depending on the requirements of different
protocols sharing the same network. Packets for a particular file transfer
between two devices may not be contiguous and may be transferred eratically, depending on what other traffic is sharing the same physical link.
Figure 10.18 shows some common physical layouts for local area networks (LANs). LANs are networks that operate within a limited area, such
Time slot
(a)
A1
B1
C1
(b)
A1
A2
A3
A2
B2
B1
C2
B2
A3
B3
C1
FIGURE 10.17 Packets for different destinations (A, B and C) multiplexed onto a common serial
bus. (a) Time division multiplexed into a regular time slot structure. (b) Asynchronous transfer showing
variable time gaps and packet lengths between transfers for different destinations.
FIGURE 10.18
Two examples of computer
network topologies. (a)
Devices connected by
spurs to a common hub,
and (b) devices connected
to a common ‘backbone’.
The former is now by far
the most common, typically
using CAT 5 cabling.
(a)
Network
spur
Network
hub
(b)
Network backbone
Interconnecting Digital Audio Devices
as an office building or studio center, within which it is common for every
device to ‘see’ the same data, each picking off that which is addressed to it
and ignoring the rest. Routers and bridges can be used to break up complex
LANs into subnets. WANs (wide area networks) and MANs (metropolitan
area networks) are larger entities that link LANs within communities or
regions. PANs (personal area networks) are typically limited to a range of a
few tens of meters around the user (e.g. Firewire, USB, Bluetooth). Wireless
versions of these network types are increasingly common. Different parts of
a network can be interconnected or extended as explained in Fact File 10.5.
Network communication is divided into a number of ‘layers’, each relating to an aspect of the communication protocol and interfacing correctly
with the layers either side. The ISO seven-layer model for open systems
interconnection (OSI) shows the number of levels at which compatibility
between systems needs to exist before seamless interchange of data can
be achieved (Figure 10.19). It shows that communication begins when the
application is passed down through various stages to the layer most people
understand – the physical layer, or the piece of wire over which the information is carried. Layers 3, 4 and 5 can be grouped under the broad heading of ‘protocol’, determining the way in which data packets are formatted
7 Application layer
6 Presentation layer
5 Session layer
4 Transport layer
3 Network layer
2 Data link layer
1 Physical layer
FIGURE 10.19
The ISO model for Open
Systems Interconnection is
arranged in seven layers, as
shown here.
FA C T F I L E 10 . 5 E XT E N D IN G A NETW ORK
It is common to need to extend a network to a wider area
or to more machines. As the number of devices increases
so does the traffic, and there comes a point when it is
necessary to divide a network into zones, separated by
‘repeaters’, ‘bridges’ or ‘routers’. Some of these devices
allow network traffic to be contained within zones, only
communicating between the zones when necessary. This
is vital in large interconnected networks because otherwise data placed anywhere on the network would be
present at every other point on the network, and overload
could quickly occur.
A repeater is a device that links two separate segments
of a network so that they can talk to each other, whereas a
bridge isolates the two segments in normal use, only transferring data across the bridge when it has a destination
address on the other side. A router is very selective in that it
examines data packets and decides whether or not to pass
them depending on a number of factors. A router can be
programmed only to pass certain protocols and only certain source and destination addresses. It therefore acts as
something of a network policeman and can be used as a
first level of ensuring security of a network from unwanted
external access. Routers can also operate between different
standards of network, such as between FDDI and Ethernet,
and ensure that packets of data are transferred over the
most time-/cost-effective route.
One could also use some form of router to link a local
network to another that was quite some distance away,
forming a wide area network (WAN). Data can be routed
either over dialed data links such as ISDN, in which the
time is charged according to usage just like a telephone
call, or over leased circuits. The choice would depend on
the degree of usage and the relative costs. The Internet
provides a means by which LANs are easily interconnected, although the data rate available will depend on
the route, the service provider and the current traffic.
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and transferred. There is a strong similarity here with the exchange of
data on physical media, as discussed earlier, where a range of compatibility
layers from the physical to the application determine whether or not one
device can read another’s disks.
Audio network requirements
The principal application of computer networks in audio systems is in the
transfer of audio data files between workstations, or between workstations
and a central ‘server ’ which stores shared files. The device requesting the
transfer is known as the ‘client’ and the device providing the data is known
as the ‘server ’. When a file is transferred in this way a byte-for-byte copy
is reconstructed on the client machine, with the file name and any other
header data intact. There are considerable advantages in being able to perform this operation at speeds in excess of real time for operations in which
real-time feeds of audio are not the aim. For example, in a news editing
environment a user might wish to upload a news story file from a remote
disk drive in order to incorporate it into a report, this being needed as fast
as the system is capable of transferring it. Alternatively, the editor might
need access to remotely stored files, such as sound files on another person’s
system, in order to work on them separately. In audio post-production for
films or video there might be a central store of sound effects, accessible by
everyone on the network, or it might be desired to pass on a completed portion of a project to the next stage in the post-production process.
Wired Ethernet is fast enough to transfer audio data files faster than
real time, depending on network loading and speed. For satisfactory operation it is advisable to use 100 Mbit/s or even 1 Gbit/s Ethernet as opposed
to the basic 10 Mbit/s version. Switched Ethernet architectures allow the
bandwidth to be more effectively utilized, by creating switched connections
between specific source and destination devices. Approaches using FDDI
or ATM are appropriate for handling large numbers of sound file transfers
simultaneously at high speed. Unlike a real-time audio interface, the speed
of transfer of a sound file over a packet-switched network (when using conventional file transfer protocols) depends on how much traffic is currently
using it. If there is a lot of traffic then the file may be transferred more
slowly than if the network is quiet (very much like motor traffic on roads).
The file might be transferred erratically as traffic volume varies, with the
file arriving at its destination in ‘spurts’. There therefore arises the need
for network communication protocols designed specifically for the transfer
of real-time data, which serve the function of reserving a proportion of the
Interconnecting Digital Audio Devices
network bandwidth for a given period of time. This is known as engineering a certain ‘quality of service’.
Without real-time protocols the computer network can not be relied
upon for transferring audio where an unbroken audio output is to be reconstructed at the destination from the data concerned. The faster the network
the more likely it is that one would be able to transfer a file fast enough to
feed an unbroken audio output, but this should not be taken for granted.
Even the highest speed networks can be filled up with traffic! This may
seem unnecessarily careful until one considers an application in which a
disk drive elsewhere on the network is being used as the source for replay
by a local workstation, as illustrated in Figure 10.20. Here it must be possible to ensure guaranteed access to the remote disk at a rate adequate for
real-time transfer, otherwise gaps will be heard in the replayed audio.
Storage area networks
An alternative setup involving the sharing of common storage by a number
of workstations is the Storage Area Network (SAN). This employs a networking technology known as Fibre Channel that can run at speeds of 4 Gbit/s,
and can also employ fiber optic links to allow long connections between
shared storage and remote workstations. RAID arrays (see Chapter 9) are
typically employed with SANs, and special software such as Apple’s XSAN is
needed to enable multiple users to access the files on such common storage.
Audio out
On-air
station
Central
server
Disk drive
High-speed network
Data path for packets
Edit station
1
Studio
recording
Edit station
2
Edit station
3
FIGURE 10.20
In this example of a
networked system a remote
disk is accessed over the
network to provide data
for real-time audio playout
from a workstation used
for on-air broadcasting.
Continuity of data flow to
the on-air workstation is
of paramount importance
here.
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Protocols for the internet
The Internet is now established as a universal means for worldwide communication. Although real-time protocols and quality of service do not sit
easily with the idea of a free-for-all networking structure, there is growing
evidence of applications that allow real-time audio and video information
to be streamed with reasonable quality. The RealAudio format, for example, developed by Real Networks, is designed for coding audio in streaming
media applications, achieving respectable quality at the higher data rates.
People are also increasingly using the Internet for transferring multimedia
projects between sites using FTP (file transfer protocol).
The Internet is a collection of interlinked networks with bridges and routers in various locations, which originally developed amongst the academic and
research community. The bandwidth (data rate) available on the Internet varies
from place to place, and depends on the route over which data is transferred.
In this sense there is no easy way to guarantee a certain bandwidth, nor a certain ‘time slot’, and when there is a lot of traffic it simply takes a long time for
data transfers to take place. Users access the Internet through a service provider (ISP), originally using a telephone line and a modem, or ISDN, or these
days more likely an ADSL connection. The most intensive users will probably
opt for high-speed leased lines giving permanent access to the Internet.
The common protocol for communication on the Internet is called
TCP/IP (Transmission Control Protocol/Internet Protocol). This provides a
connection-oriented approach to data transfer, allowing for verification of
packet integrity, packet order and retransmission in the case of packet loss.
At a more detailed level, as part of the TCP/IP structure, there are highlevel protocols for transferring data in different ways. There is a file transfer
protocol (FTP) used for downloading files from remote sites, a simple mail
transfer protocol (SMTP) and a post office protocol (POP) for transferring
email, and a hypertext transfer protocol (HTTP) used for interlinking sites
on the world wide web (WWW). The WWW is a collection of file servers
connected to the Internet, each with its own unique IP address (the method
by which devices connected to the Internet are identified), upon which may
be stored text, graphics, sounds and other data.
UDP (user datagram protocol) is a relatively low-level connectionless protocol that is useful for streaming audio over the Internet. Being connectionless, it does not require any handshaking between transmitter and receiver, so
the overheads are very low and packets can simply be streamed from a transmitter without worrying about whether or not the receiver gets them. If packets are missed by the receiver, or received in the wrong order, there is little to
be done about it except mute or replay distorted audio, but UDP can be efficient when bandwidth is low and quality of service is not the primary issue.
Interconnecting Digital Audio Devices
Various real-time protocols have also been developed for use on the
Internet, such as RTP (real-time transport protocol). Here packets are timestamped and may be reassembled in the correct order and synchronized with a
receiver clock. RTP does not guarantee quality of service or reserve bandwidth
but this can be handled by a protocol known as RSVP (reservation protocol).
RTSP is the real-time streaming protocol that manages more sophisticated
functionality for streaming media servers and players, such as stream control
(play, stop, fast-forward, etc.) and multicast (streaming to numerous receivers).
Wireless networks
Increasing use is made of wireless networks these days, the primary advantage
being the lack of need for a physical connection between devices. There are
various IEEE 802 standards for wireless networking, including 802.11 which
covers wireless Ethernet or ‘Wi-Fi’. These typically operate on either the
2.4 GHz or 5 GHz radio frequency bands, at relatively low power, and use various interference reduction and avoidance mechanisms to enable networks to
coexist with other services. It should, however, be recognized that wireless networks will never be as reliable as wired networks owing to the differing conditions under which they operate, and that any critical applications in which
real-time streaming is required would do well to stick to wired networks where
the chances of experiencing drop-outs owing to interference or RF fading are
almost non-existent. They are, however, extremely convenient for mobile
applications and when people move around with computing devices, enabling
reasonably high data rates to be achieved with the latest technology.
Bluetooth is one example of a wireless personal area network (WPAN)
designed to operate over limited range at data rates of up to 1 Mbit/s. Within
this there is the capacity for a number of channels of voice quality audio at
data rates of 64 kbit/s and asynchronous channels up to 723 kbit/s. Taking
into account the overhead for communication and error protection, the
actual data rate achievable for audio communication is usually only sufficient to transfer data-reduced audio for a few channels at a time.
Audio over firewire (IEEE 1394)
Firewire is an international standard serial data interface specified in IEEE
1394-1995. One of its key applications has been as a replacement for SCSI
(Small Computer Systems Interface) for connecting disk drives and other
peripherals to computers. It is extremely fast, running at rates of 100, 200
and 400 Mbit/s in its original form, with higher rates appearing all the time
up to 3.2 Gbit/s. It is intended for optical fiber or copper interconnection, the
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copper 100 Mbit/s (S100) version being limited to 4.5 m between hops (a hop
is the distance between two adjacent devices). The S100 version has a maximum realistic data capacity of 65 Mbit/s, a maximum of 16 hops between
nodes and no more than 63 nodes on up to 1024 separate buses. On the copper version there are three twisted pairs – data, strobe and power – and the
interface operates in half duplex mode, which means that communications
in two directions are possible, but only one direction at a time. The ‘direction’ is determined by the current transmitter which will have arbitrated for
access to the bus. Connections are ‘hot pluggable’ with auto-reconfiguration –
in other words one can connect and disconnect devices without turning off
the power and the remaining system will reconfigure itself accordingly. It is
also relatively cheap to implement. A recent implementation, 1394c, allows
the use of gigabit Ethernet connectors, which may improve the reliability
and usefulness of the interface in professional applications.
Firewire combines features of network and point-to-point interfaces,
offering both asynchronous and isochronous communication modes, so
guaranteed latency and bandwidth are available if needed for time-critical
applications. Communications are established between logical addresses,
and the end point of an isochronous stream is called a ‘plug’. Logical connections between devices can be specified as either ‘broadcast’ or ‘pointto-point’. In the broadcast case either the transmitting or receiving plug is
defined, but not both, and broadcast connections are unprotected in that
any device can start and stop it. A primary advantage for audio applications
is that point-to-point connections are protected – only the device that initiated a transfer can interfere with that connection, so once established the
data rate is guaranteed for as long as the link remains intact. The interface
can be used for real-time multichannel audio interconnections, file transfer,
MIDI and machine control, carrying digital video, carrying any other computer data and connecting peripherals (e.g. disk drives).
Originating partly in Yamaha’s ‘m-LAN’ protocol, the 1394 Audio and
Music Data Transmission Protocol is now also available as an IEC PAS
component of the IEC 61883 standard (a PAS is a publically available specification that is not strictly defined as a standard but is made available for
information purposes by organizations operating under given procedures). It
offers a versatile means of transporting digital audio and MIDI control data.
Audio over Universal Serial Bus (USB)
The Universal Serial Bus is not the same as IEEE 1394, but it has some
similar implications for desktop multimedia systems, including audio
peripherals. USB has been jointly supported by a number of manufacturers
Interconnecting Digital Audio Devices
including Microsoft, Digital, IBM, NEC, Intel and Compaq. Version 1.0 of
the copper interface runs at a lower speed than 1394 (typically either 1.5 or
12 Mbit/s) and is designed to act as a low-cost connection for multiple input
devices to computers such as joysticks, keyboards, scanners and so on. USB
2.0 runs at a higher rate up to 480 Mbit/s and is supposed to be backwardscompatible with 1.0.
USB 1.0 supports up to 127 devices for both isochronous and asynchronous communication and can carry data over distances of up to 5 m per hop
(similar to 1394). A hub structure is required for multiple connections to the
host connector. Like 1394 it is hot pluggable and reconfigures the addressing
structure automatically, so when new devices are connected to a USB setup
the host device assigns a unique address. Limited power is available over the
interface and some devices are capable of being powered solely using this
source – known as ‘bus-powered’ devices – which can be useful for field operation of, say, a simple A/D convertor with a laptop computer.
Data transmissions are grouped into frames of 1 ms duration in USB
1.0 but a ‘micro-frame’ of 1/8th of 1 ms was also defined in USB 2.0. A
start-of-frame packet indicates the beginning of a cycle and the bus clock is
normally at 1 kHz if such packets are transmitted every millisecond. So the
USB frame rate is substantially slower than the typical audio sampling rate.
The transport structure and different layers of the network protocol will not
be described in detail as they are long and complex and can be found in the
USB 2.0 specification. However, it is important to be aware that transactions are set up between sources and destinations over so-called ‘pipes’ and
that numerous ‘interfaces’ can be defined and run over a single USB cable,
only dependent on the available bandwidth.
The way in which audio is handled on USB is well defined and somewhat more clearly explained than the 1394 audio/music protocol. It defines
three types of communication: audio control, audio streaming and MIDI
streaming. We are concerned primarily with audio streaming applications.
Audio data transmissions fall into one of three types. Type 1 transmissions
consist of channel-ordered PCM samples in consecutive subframes, whilst
Type 2 transmissions typically contain non-PCM audio data that does not
preserve a particular channel order in the bitstream, such as certain types
of multichannel data-reduced audio stream. Type 3 transmissions are a
hybrid of the two such that non-PCM data is packed into pseudo-stereo
data words in order that clock recovery can be made easier.
Audio samples are transferred in subframes, each of which can be
1–4 bytes long (up to 24 bits resolution). An audio frame consists of one or
more subframes, each of which represents a sample of different channels in
the cluster (see below). As with 1394, a USB packet can contain a number
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of frames in succession, each containing a cluster of subframes. Frames are
described by a format descriptor header that contains a number of bytes
describing the audio data type, number of channels, subframe size, as well
as information about the sampling frequency and the way it is controlled
(for Type 1 data). An example of a simple audio frame would be one containing only two subframes of 24 bit resolution for stereo audio.
Audio of a number of different types can be transferred in Type 1 transmissions, including PCM audio (two’s complement, fixed point), PCM-8
format (compatible with original 8 bit WAV, unsigned, fixed point), IEEE
floating point, A-law and μ-law (companded audio corresponding to relatively old telephony standards). Type 2 transmissions typically contain
data-reduced audio signals such as MPEG or AC-3 streams. Here the
data stream contains an encoded representation of a number of channels
of audio, formed into encoded audio frames that relate to a large number of original audio samples. An MPEG encoded frame, for example, will
typically be longer than a USB packet (a typical MPEG frame might be
8 or 24 ms long), so it is broken up into smaller packets for transmission
over USB rather like the way it is streamed over the IEC 60958 interface
described in Fact File 10.4. The primary rule is that no USB packet should
contain data for more than one encoded audio frame, so a new encoded
frame should always be started in a new packet. The format descriptor for
Type 2 is similar to Type 1 except that it replaces subframe size and number of channels indication with maximum bit rate and number of audio
samples per encoded frame. Currently only MPEG and AC-3 audio are
defined for Type 2.
Audio data for closely related synchronous channels can be clustered
for USB transmission in Type 1 format. Up to 254 streams can be clustered and there are 12 defined spatial positions for reproduction, to simplify the relationship between channels and the loudspeaker locations to
which they relate. (This is something of a simplification of the potentially
complicated formatting of spatial audio signals and assumes that channels
are tied to loudspeaker locations, but it is potentially useful. It is related
to the channel ordering of samples within a WAVE format extensible file,
described earlier.) The first six defined streams follow the internationally
standardized order of surround sound channels for 5.1 surround, that is
left, right, centre, LFE (low frequency effects), left surround, right surround
(see Chapter 17). Subsequent streams are allocated to other loudspeaker
locations around a notional listener. Not all the spatial location streams
have to be present but they are supposed to be presented in the defined
order. Clusters are defined in a descriptor field that includes ‘bNrChannels’
Interconnecting Digital Audio Devices
Table 10.4
Channel identification in USB audio cluster descriptor
Data bit
Spatial location
D0
D1
D2
D3
D4
D5
D6
D7
D8
D9
D10
D11
D12…15
Left Front (L)
Right Front (R)
Center Front (C)
Low Frequency Enhancement (LFE)
Left Surround (LS)
Right Surround (RS)
Left of Center (LC)
Right of Center (RC)
Surround (S)
Side Left (SL)
Side Right (SR)
Top (T)
Reserved
(specifying how many logical audio channels are present in the cluster) and
‘wChannelConfig’ (a bit field that indicates which spatial locations are present in the cluster). If the relevant bit is set then the relevant location is
present in the cluster. The bit allocations are shown in Table 10.4.
AES-47: audio over ATM
AES-47 defines a method by which linear PCM data, either conforming to
AES-3 format or not, can be transferred over ATM (Asynchronous Transfer
Mode) networks. There are various arguments for doing this, not least
being the increasing use of ATM-based networks for data communications
within the broadcasting industry and the need to route audio signals over
longer distances than possible using standard digital interfaces. There is
also a need for low latency, guaranteed bandwidth and switched circuits,
all of which are features of ATM. Essentially an ATM connection is established in a similar way to making a telephone call. A SETUP message is
sent at the start of a new ‘call’ that describes the nature of the data to be
transmitted and defines its vital statistics. The AES-47 standard describes a
specific professional audio implementation of this procedure that includes
information about the audio signal and the structure of audio frames in the
SETUP at the beginning of the call.
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RECOMMENDED FURTHER READING
1394 Trade Assocation, 2001. TA Document 2001003: Audio and Music Data
Transmission Protocol 2.0.
AES, 2002. AES 47-2002: Transmission of Digital Audio over Asynchronous
Transfer mode Networks.
IEC, 1998. IEC/PAS 61883-6. Consumer Audio/Video Equipment – Digital Interface –
Part 6: Audio and Music Data Transmission Protocol.
IEEE, 1995. IEEE 1394: Standard for a High Performance Serial Bus.
Page, M., et al., 2002. Multichannel audio connection for direct stream digital.
Presented at AES 113th Convention, Los Angeles, 5–8 October.
Rumsey, F., Watkinson, J., 2003. The Digital Interface Handbook, third ed.. Focal Press.
Rumsey, F., 2004. Desktop Audio Technology. Focal Press.
WEBSITES
Audio Engineering Society standards. ⬍www.aes.org/⬎
IEEE 1394. ⬍www.1394ta.org/⬎
Universal Serial Bus. ⬍www.usb.org/⬎
CHAPTER 11
Power Amplifiers
CH A P T E R C O N TE N T S
Domestic Power Amplifiers
Professional Amplifier Facilities
Specifications
Sensitivity
Power output
Frequency response
Distortion
Crosstalk
Signal-to-noise ratio
Impedance
Damping factor
Phase response
Coupling
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Power amplifiers are uneventful devices. They are usually big and heavy,
take up a lot of rack space, and feature very little (or sometimes nothing)
beyond input and output sockets. Because one tends to ignore them, it is all
the more important that they are chosen and used with due care. Coming
in a variety of shapes, sizes and ‘generations’, they are all required to do
the ostensibly simple job of providing voltage amplification – converting line
levels of up to a volt or so into several tens of volts, with output currents
in the ampere range to develop the necessary power across the loudspeaker
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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terminals. Given these few requirements, it is perhaps surprising how many
designs there are on the market.
DOMESTIC POWER AMPLIFIERS
The domestic power amplifier, at its best, is designed for maximum fidelity
in the true sense of that word, and this will usually mean that other considerations such as long-term overload protection and complete stability into any
type of speaker load are not always given the type of priority which is essential in the professional field. A professional power amp may well be asked to
drive a pair of 6 ohm speakers in parallel on the other end of 30 meters of
cable, at near to maximum output level for hours on end if used in a rock PA
rig. This demands large power supplies and heavy transformers, with plenty
of heat sink area (the black fins usually found on the outer casing) to keep it
from overheating. Cooling fans are frequently employed which will often run
at different speeds depending on the temperature of the amplifier.
The domestic amplifier is unlikely to be operated at high output levels
for a significant length of time, and the power supplies are often therefore
designed to deliver high currents for short periods to take care of short, loud
passages. A power supply big enough to supply high currents for lengthy
periods is probably wasted in a domestic amplifier. Also, the thermal inertia of the transformer and the heat sinks means that unacceptable rises in
temperature are unlikely. Although there are one or two domestic speakers
which are notoriously difficult to drive due to various combinations of low
impedance, low efficiency (leading to high power demand), and wide phase
swings (current and voltage being out of step with each other due to crossover components and driver behavior in a particular speaker enclosure), the
majority of domestic hi-fi speakers are a comfortable load for an amplifier,
and usually the speaker leads will be less than 10 meters in length.
It is unlikely that the amplifier will be driven into a short-circuit due
to faulty speaker lines for any length of time (silence gives an immediate warning), which is not the case with a professional amplifier which
may well be one of many, driving a whole array of speakers. A shortcircuit developing soon after a show has begun may cause the amplifier to
be driven hard into this condition for the whole evening. Protection circuitry needs to be incorporated into the design to allow the professional
amplifier to cope with this without overheating or catastrophically failing
which can affect other amplifiers in the same part of the rig.
Several ‘classes’ of amplifier design have appeared over the years, these
being labels identifying the type of output stage topology employed to drive
the speaker. These are outlined in Fact File 11.1.
Domestic Power Amplifiers
FA C T F I L E 1 1 . 1 A MP LIFIE R C LASSES
Class A
The output stage draws a constant high current from the
power supply regardless of whether there is an audio signal
present or not. Low-current class A stages are used widely
in audio circuits. The steady bias current as it is known is
employed because transistors are non-linear devices, particularly when operated at very low currents. A steady current
is therefore passed through them which biases them into the
area of their working range at which they are most linear.
The constant bias current makes class A amplification inefficient due to heat generation, but there is the
advantage that the output transistors are at a constant
steady temperature. Class A is capable of very high sound
quality, and several highly specified upmarket domestic
class A power amplifiers exist.
Class B
No current flows through the output transistors when no
audio signal is present. The driving signal itself biases
the transistors into conduction to drive the speakers. The
technique is therefore extremely efficient because the
current drawn from the power supply is entirely dependent upon the level of drive signal, and so it is particularly
attractive in battery-operated equipment. The disadvantage is that at low signal levels the output transistors
operate in a non-linear region. It is usual for pairs (or multiples) of transistors to provide the output current of a power
amplifier. Each of the pair handles opposite halves of the
output waveform (positive and negative with respect to zero)
and therefore as the output swings through zero from positive to negative and vice versa the signal suffers so-called
‘crossover distortion’. The result is relatively low sound quality, but class B can be used in applications which do not
require high sound quality such as telephone systems, handheld security transceivers, paging systems and the like.
Class A–B
In this design a relatively low constant bias current flows
through the output transistors to give a low-power class
A amplifier. As the input drive signal is increased, the
output transistors are biased into appropriately highercurrent conduction in order to deliver higher power to
the speakers. This part of the operation is the class B
part, i.e. it depends on input drive signal level. But the
low-level class A component keeps the transistors biased
into a linear part of their operating range so that crossover
distortion is largely avoided. The majority of high-quality
amplifiers operate on this principle.
Other Classes
Class C drives a narrow band of frequencies into a resonant load, and is appropriate to radio-frequency (RF) work
where an amplifier is required to drive a single frequency
into an appropriately tuned aerial.
Class D uses ‘pulse width modulation’. It has seen
increasing use since the late 1980s, although the technique appeared as far back as the 1960s in Sinclair
designs. In a conventional output stage, the voltage across
the transistors varies in proportion to the input signal voltage, and on average they spend much of their time under
the conditions of moderate voltage and moderate output
current which together require them to dissipate power in
the form of heat via the amplifier’s heat sinks. With class
D, however, the output transistors are driven by an ultrasonic square wave whose mark-to-space ratio is varied by
the audio signal. The transistors are therefore either in a
state of minimum voltage across them combined with
maximum current output (full power), or maximum voltage
across them combined with virtually no current output.
Power dissipation in the transistors is therefore minimal,
they run cool, and they are therefore much more efficient. Output low-pass filtering is required to remove the
ultrasonic square wave component, leaving just the audio
waveform. Combined with efficient switched-mode power
supplies, the technique can offer high-powered compact
power amplifier designs which do not need fan cooling.
The D designation simply means that it was the fourth output topology to emerge, but sometimes it is erroneously
termed ‘Digital’. There are indeed analogies with digital
processing in two respects: the output transistors are
always either fully on or fully off (a ‘1’ or a ‘0’), and the output is low-pass filtered for the same reasons as for digitalto-analog conversion, to remove ultrasonic components.
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FA C T F I L E 1 1 . 1 (C O N TIN U E D )
A variation on class D, called class T (from the Tripath
company), has recently been seen. Here, the ultrasonic
frequency is continuously varied in accordance with the
amplitude of the audio signal. The frequency is about
1.2 MHz at low signal levels, falling to around 200 kHz
for very high signal levels; a greater overall efficiency is
claimed as a result. Classes E and F were concerned with
increasing efficiency, and currently no commercial models conform to these particular categories.
Class G incorporates several different voltage rails
which progressively come into action as the drive signal
voltage is increased. This technique can give very good
efficiency because for much of the time only the lowervoltage, low-current supplies are in operation. Such designs
can be rather smaller than their conventional class A–B
counterparts of comparable output power rating. Class H is
a variation on class G in that the power supply voltage rails
are made to track the input signal continuously, maintaining just enough headroom to accommodate the amplifier’s
requirements for the necessary output voltage swing.
Since the early 1980s the MOSFET (Metal Oxide
Semiconductor Field-Effect Transistor) has been widely
employed for the output stages of power amplifiers.
MOSFET techniques claim lower distortion, better thermal
tracking (i.e. good linearity over a wide range of operating
temperatures), simpler output stage design, and greater
tolerance of adverse loudspeaker loads without the need
for elaborate protection circuitry.
PROFESSIONAL AMPLIFIER FACILITIES
The most straightforward power amplifiers have input sockets and output
terminals, and nothing else. Single-channel models are frequently encountered, and in the professional field these are often desirable because if one
channel of a stereo power amplifier develops a fault then the other channel
also has to be shut down, thus losing a perfectly good circuit. The singlechannel power amplifier is thus a good idea when multi-speaker arrays are
in use such as in rock PA systems and theater sound.
Other facilities found on power amplifiers include input level controls,
output level meters, overload indicators, thermal shutdown (the mains
feed is automatically disconnected if the amplifier rises above a certain
temperature), earth-lift facility to circumvent earth loops, and ‘bridging’
switch. This last facility, applicable to a stereo power amplifier, is a facility
sometimes provided whereby the two channels of the amp can be bridged
together to form a single-channel higher-powered one, the speaker(s) now
being connected across the two positive output terminals with the negative
terminals left unused. Only one of the input sockets is now used to drive it.
Cooling fans are often incorporated into an amplifier design. Such a forcecooled design can be physically smaller than its convection-cooled counterpart, but fans tend to be noisy. Anything other than a genuinely silent fan is
unacceptable in a studio or broadcast control room, or indeed in theater work,
and such models will need to be housed in a separate well-ventilated room.
Ventilation of course needs to be a consideration with all power amplifiers.
Specifications
SPECIFICATIONS
Power amplifier specifications include sensitivity, maximum output power
into a given load, power bandwidth, frequency response, slew rate, distortion, crosstalk between channels, signal-to-noise ratio, input impedance,
output impedance, damping factor, phase response, and DC offset. Quite
surprising differences in sound quality can be heard between certain models, and steady-state measurements do not, unfortunately, always tell a user
what he or she can expect to hear.
Sensitivity
Sensitivity is a measurement of how much voltage input is required to produce the amplifier’s maximum rated output. For example, a model may be
specified ‘150 watts into 8 ohms, input sensitivity 775 mV ⫽ 0 dBu’. This
means that an input voltage of 775 mV will cause the amplifier to deliver
150 watts into an 8 ohm load. Speakers exhibit impedances which vary considerably with frequency, so this is always a nominal specification when real
speakers are being driven. Consideration of sensitivity is important because
the equipment which is to drive the amp must not be allowed to deliver
a greater voltage to the amplifier than its specification states, otherwise
the amplifier will be overloaded causing ‘clipping’ of the output waveform
(a squaring-off of the tops and bottoms of the waveform resulting in severe
distortion). This manifests itself as a ‘breaking-up’ of the sound on musical
peaks, and will often quickly damage tweeters and high-frequency horns.
Many amplifiers have input level controls so that if, for instance, the
peak output level of the mixer which drives the amplifier is normally say
‘PPM 6’ – about 2 volts – then the amp’s input levels can be turned down
to prevent overload. In the given example, 2 volts is 8 B higher than 775 mV
(PPM 4 ⫽ 0 dBu) and so the input level control should be reduced by 8 dB to
allow for this. If a dB calibration is not provided on the level control, and
many are not particularly accurate anyway, a reasonable guide is that, compared with its maximum position of about ‘5 o’clock’, reducing the level
to about 2 o’clock will reduce the sensitivity by about 10 dB, or by a factor of three. In this position, the power amplifier with an input sensitivity
of 775 mV will now require 0.775 ⫻ 3, or about 2 volts, to develop its full
output.
If input level controls are not provided, one can build a simple resistive attenuator which reduces the voltage being fed to the amplifier’s input.
Two examples are shown in Figure 11.1. It is best to place such attenuators close to the power amp input in order to keep signal levels high while
they are traveling down the connecting leads. In both cases the 3k3 resistor
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(a)
Signal
6k8
From
mixer
Signal
3k3
To
amplifier
Earth
(b)
Earth
3k3
+
+
From
mixer
3k3
To
amplifier
3k3
–
Earth
–
Earth
FIGURE 11.1 (a) An unbalanced resistive attenuator. (b) A balanced resistive attenuator.
which is in parallel with the amplifier’s input can be increased in value for
less attenuation, and decreased in value for greater attenuation. With care,
the resistors can be built into connecting plugs, the latter then needing to
be clearly labeled.
Power output
A manufacturer will state the maximum power a particular model can provide into a given load, e.g. ‘200 watts into 8 ohms’, often with ‘both channels driven’ written after it. This last means that both channels of a stereo
amplifier can deliver this simultaneously. When one channel only is being
driven, the maximum output is often a bit higher, say 225 watts, because
the power supply is less heavily taxed. Thus 200 watts into 8 ohms means
that the amplifier is capable of delivering 40 volts into this load, with a current of 5 amps. If the load is now reduced to 4 ohms then the same amplifier should produce 400 watts. A theoretically perfect amplifier should then
double its output when the impedance it drives is halved. In practice, this is
beyond the great majority of power amplifiers and the 4 ohm specification
Specifications
of the above example may be more like 320 watts, but this only around
1 dB below the theoretically perfect value. A 2 ohm load is very punishing
for an amplifier, and should be avoided even though a manufacturer sometimes claims a model is capable of, say, 800 watts of short-term peaks into
2 ohms. This at least tells us that the amp should be able to drive 4 ohm
loads without any trouble.
Because 200 watts is only 3 dB higher than 100 watts, then, other
things being equal, the exact wattage of an amplifier is less important than
factors such as its ability to drive difficult reactive loads for long periods.
Often, ‘RMS’ will be seen after the wattage rating. This stands for rootmean-square, and defines the raw ‘heating’ power of an amplifier, rather
than its peak output. All amplifiers should be specified RMS so that they
can easily be compared. The RMS value is 0.707 times the instantaneous
peak capability, and it is unlikely that one would encounter a professional
amplifier with just a peak power rating.
Power bandwidth is not the same as power rating, as discussed in Fact
File 11.2.
Frequency response
Frequency response, unlike power bandwidth, is simply a measure of the
limits within which an amplifier responds equally to all frequencies when
delivering a very low power. The frequency response is usually measured
with the amplifier delivering 1 watt into 8 ohms. A specification such as
‘20 Hz – 20 kHz ⫾ 0.5 dB’ should be looked for, meaning that the response is
virtually flat across the whole of the audible band. Additionally, the ⫺3 dB
FA C T F I L E 1 1 . 2 P O WE R B A N D WI DTH
Power bandwidth is a definition of the frequency response
limits within which an amplifier can sustain its specified
output. Specifically, a 3 dB drop of output power is allowed
in defining a particular amplifier’s power bandwidth. For
example, a 200 watt amplifier may have a power bandwidth of 10 Hz to 30 kHz, meaning that it can supply 200
watts – 3 dB (⫽100 watts) at 10 Hz and 30 kHz, compared with the full 200 watts at mid frequencies. Such an
amplifier would be expected to deliver the full 200 watts
at all frequencies between about 30 Hz and 20 kHz, and
this should also be looked for in the specification. Often,
though, the power rating of an amplifier is much more
impressive when measured using single sine-wave tones
than with broad-band signals, since the amplifier may be
more efficient at a single frequency.
Power bandwidth can indicate whether a given amplifier is capable of driving a subwoofer at high levels in a PA
rig, as it will be called upon to deliver much of its power
at frequencies below 100 Hz or so. The driving of highfrequency horns also needs good high-frequency power
bandwidth so that the amplifier never clips the high frequencies, which easily damages horns as has been said.
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FA C T F I L E 1 1 . 3 S L E W R AT E
Slew rate is a measure of the ability of an amplifier to
respond accurately to high-level transients. For instance,
the leading edge of a transient may demand that the output of an amplifier swings from 0 to 120 watts in a fraction
of a millisecond. The slew rate is defined in V μs⫺1 (volts
per microsecond) and a power amplifier which is capable
of 200 watts output will usually have a slew rate of at least
30 V μs⫺1. Higher-powered models require a greater slew
rate simply because their maximum output voltage swing
is greater. A 400 watt model might be required to swing
57 volts into 8 ohms as compared with the 200 watt model’s 40, so its slew rate needs to be at least:
30 ⫻ 57 ⫼ 40) ⫽ 43 V μs⫺1
In practice, modern power amplifiers achieve slew
rates comfortably above these figures.
An absolute minimum can be estimated by considering the highest frequency of interest, 20 kHz, then doubling it for safety, 40 kHz, and considering how fast a
given amplifier must respond to reproduce this accurately
at full output. A sine wave of 40 kHz reaches its positivegoing peak in 6.25 μs, as shown in the diagram. A 200
watt model delivers a peak voltage swing of 56.56 volts
peak to peak (1.414 times the RMS voltage). It may
seem then that it could therefore be required to swing
from 0 V to 28.28 V in 6.25 μs, thus requiring a slew rate
of 28.28 ÷ 6.25, or 4.35 V μs⫺1. But the actual slew rate
requirement is rather higher because the initial portion
of the sine wave rises steeply, tailing off towards its maximum level.
Musical waveforms come in all shapes and sizes of
course, including near-square waves with their almost
vertical leading edges, so a minimum slew rate of around
eight times this (i.e. 30 V μs⫺1) might be considered as
necessary. It should be remembered, though, that the harmonics of an HF square wave are well outside the audible
spectrum, and thus slew rate distortion of such waves at
HF is unlikely to be audible. Extremely high slew rates of
several hundred volts per microsecond are sometimes
encountered. These are achieved in part by a wide frequency response and ‘fast’ output transistors, which are
not always as stable into difficult speaker loads as are their
‘ordinary’ counterparts. Excessive slew rates are therefore
to be viewed with skepticism.
Actual slew rate needed
+
40 kHz sine wave
0
6.25 μs
25 μs
–
points are usually also stated, e.g. ‘ ⫺ 3 dB at 12 Hz and 40 kHz’, indicating
that the response falls away smoothly below and above the audio range.
This is desirable as it gives a degree of protection for the amp and speakers
against subsonic disturbances and RF interference.
Distortion
Distortion should be 0.1% THD or less across the audio band, even close to
maximum-rated output. It often rises slightly at very high frequencies, but
this is of no consequence. Transient distortion, or transient intermodulation
Specifications
distortion (TID), is also a useful specification. It is usually assessed by feeding both a 19 kHz and a 20 kHz sine wave into the amplifier and measuring the relative level of 1 kHz difference tone. The 1 kHz level should be at
least 70 dB down, indicating a well-behaved amplifier in this respect. The
test should be carried out with the amplifier delivering at least two-thirds
of its rated power into 8 ohms. Slew rate distortion is also important (see
Fact File 11.3).
Crosstalk
Crosstalk figures of around ⫺70 dB at mid frequencies should be a reasonable minimum, degrading to around ⫺50 dB at 20 kHz, and by perhaps the
same amount at 25 Hz or so. ‘Dynamic crosstalk’ is sometimes specified,
this manifesting itself mainly at low frequencies because the power supply
works hardest when it is called upon to deliver high currents during highlevel, low-frequency drive. Current demand by one channel can modulate
the power supply voltage rails, which gets into the other channel. A number of amplifiers have completely separate power supplies for each channel,
which eliminates such crosstalk, or at least separate secondary windings on
the mains transformer plus two sets of rectifiers and reservoir capacitors
which is almost as good.
Signal-to-noise ratio
Signal-to-noise ratio is a measure of the output residual noise voltage
expressed as a decibel ratio between that and the maximum output voltage,
when the input is short-circuited. Noise should never be a problem with
a modern power amplifier and signal-to-noise ratios of at least 100 dB are
common. High-powered models (200 watts upwards) should have signalto-noise ratios correspondingly greater (e.g. 110 dB or so) in order that the
output residual noise remains below audibility.
Impedance
The input impedance of an amplifier ought to be at least 10 kΩ, so that if a
mixer is required to drive, say, ten amplifiers in parallel, as is often the case
with PA rigs, the total load will be 10 k ÷ 10, or 1 k, which is still a comfortable load for the mixer. Because speakers are of very low impedance, and
because their impedance varies greatly with frequency, the amplifier’s output impedance must not be greater than a fraction of an ohm, and a value
of 0.1 ohms or less is needed. A power amplifier needs to be a virtually
perfect ‘voltage source’, its output voltage remaining substantially constant
with different load impedances.
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The output impedance does, however, rise a little at frequency extremes.
At LF, the output impedance of the power supply rises and therefore so does
the amplifier’s. It is common practice to place a low-valued inductor of a
couple of microhenrys in series with a power amp’s output which raises its
output impedance a little at HF, this being to protect the amp against particularly reactive speakers or excessively capacitive cables, which can provoke HF oscillation.
Damping factor
Damping factor is a numerical indication of how well an amplifier can ‘control’ a speaker. There is a tendency for speaker cones and diaphragms to go
on vibrating a little after the driving signal has stopped, and a very low output impedance virtually short-circuits the speaker terminals which ‘damps’
this. Damping factor is the ratio between the amplifier’s output impedance and the speaker’s rated impedance, so a damping factor of ‘100 into 8
ohms’ means that the output impedance of the amplifier is 8 ÷ 100 ohms,
or 0.08 ohms. One hundred is quite a good figure (the higher the better,
but a number greater than 200 could imply that the amplifier is insufficiently well protected from reactive loads and the like), but it is better if
a frequency is given. Damping factor is most useful at low frequencies
because it is the bass cones which vibrate with greatest excursion, requiring
the tightest control. A damping factor of ‘100 at 40 Hz’ is therefore a more
useful specification than ‘100 at 1 kHz’.
Phase response
Phase response is a measurement of how well the frequency extremes keep
in step with mid frequencies. At very low and very high frequencies, 15°
phase leads or phase lags are common, meaning that in the case of phase
lag, there is a small delay of the signal compared with mid frequencies, and
phase lead means the opposite. At 20 Hz and 20 kHz, the phase lag or phase
lead should not be greater than 15°, otherwise this may imply a degree of
instability when difficult loads are being driven, particularly if HF phase
errors are present.
The absolute phase of a power amplifier is simply a statement of
whether the output is in phase with the input. The amplifier should be nonphase-inverting overall. One or two models do phase invert, and this causes
difficulties when such models are mixed with non-inverting ones in multispeaker arrays when phase cancelations between adjacent speakers, and
incorrect phase relationships between stereo pairs and the like, crop up. The
cause of these problems is not usually apparent and can waste much time.
Coupling
COUPLING
The vast majority of power amplifier output stages are ‘direct coupled’,
that is the output power transistors are connected to the speakers with
nothing in between beyond perhaps a very low-valued resistor and a small
inductor. The DC voltage operating points of the circuit must therefore be
chosen such that no DC voltage appears across the output terminals of the
amplifier. In practice this is achieved by using ‘split’ voltage rails of opposite polarity (e.g. ⫾46 volts DC) between which the symmetrical output
stage ‘hangs’, the output being the mid-point of the voltage rails (i.e. 0 V).
Small errors are always present, and so ‘DC offsets’ are produced which
means that several millivolts of DC voltage will always be present across
the output terminals. This DC flows through the speaker, causing its cone
to deflect either forwards or backwards a little from its rest position. As low
a DC offset as possible must therefore be achieved, and a value of ⫾40 mV
is an acceptable maximum. Values of 15 mV or less are quite common.
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CHAPTER 12
Lines and Interconnection
CH A P T E R C O N TE N T S
Transformers
Transformers and impedances
Limitations of transformers
Unbalanced Lines
Cable Effects with Unbalanced Lines
Cable resistance
Cable and transformer inductance
Cable capacitance
Balanced Lines
Working with Balanced Lines
Star-quad Cable
Electronic Balancing
100 Volt Lines
Principles
Working with 100 volt lines
600 Ohms
Principles
Problems with 600 ohm equipment
DI Boxes
Overview
Passive DI boxes
Active DI boxes
Splitter Boxes
Jackfields (patchbays)
Overview
Patch cords
Normaling
Other jackfield facilities
Distribution Amplifiers
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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This chapter is concerned with the interconnection of analog audio signals,
and the solving of problems concerned with analog interfacing. It is not
intended to cover digital interfacing systems here, since this subject is adequately covered in Chapter 10. The proper interconnection of analog audio
signals, and an understanding of the principles of balanced and unbalanced
lines, is vital to the maintenance of high quality in an audio system, and
will remain important for many years notwithstanding the growing usage
of digital systems.
TRANSFORMERS
Mains transformers are widely used throughout the electrical and electronics
industries, usually to convert the 240 V AC mains voltage to a rather lower
voltage. Audio transformers are widely used in audio equipment for balancing and isolating purposes, and whereas mains transformers are required
only to work at 50 Hz, audio transformers must give a satisfactory performance over the complete audio spectrum. Fortunately most audio transformers are only called upon to handle a few volts at negligible power, so
they are generally much smaller than their mains counterparts. The principles of transformer operation are outlined in Fact File 12.1.
Transformers and impedances
Consider Figure 12.1(a). The turns ratio is 1:2, so the square of the
turns ratio (used to calculate the impedance across the secondary) is 1:4,
and therefore the impedance across the secondary will be found to be
10 ⫻ 4 ⫽ 40 k. Another example is shown in Figure 12.1(b). The turns ratio
is 1:4. 0.7 volts is applied across the primary and gives 2.8 volts across the
secondary. The square of the turns ratio is 1:16, so the impedance across
the secondary is 2 k ⫻ 16 ⫽ 32 k. The transformer also works backwards,
as shown in Figure 12.1(c). A 20 k resistor is now placed across the secondary. The square of the turns ratio is 1:16, and therefore the impedance
across the primary is 20 k ÷ 16 ⫽ 1 k25.
Consider now a microphone transformer that is loaded with an impedance on both sides, as shown in Figure 12.2. The transformer presents the
2 k impedance of the mixer to the microphone, and the 200 ohm impedance
of the microphone to the mixer. With a step-up ratio of 1:4 the square of
the turns ratio would be 1:16. The microphone would be presented with an
impedance of 2 k ÷ 16 ⫽ 125 ohms, whereas the mixer would be presented
with an impedance of 200 ⫻ 16 ⫽ 3200 ohms. In this particular case a 1:4
step-up transformer is unsuitable because microphones like to work into an
Transformers
FA C T F I L E 1 2 . 1 T H E T R A N S FO RM ER
From the diagrams it can be seen that the transformer
consists of a laminated core (i.e. a number of thin sheets
of metal ‘laminated’ together to form a single thick
core) around which is wound a ‘primary’ winding and a
‘secondary’ winding. If an alternating current is passed
through the primary winding, magnetic flux flows in the
core (in a similar fashion to the principle of the tape head;
see Fact File 6.1), and thus through the secondary winding. Flux changes in the secondary winding cause a current to be induced in it. The voltage across the secondary
winding compared with that across the primary is proportional to the ratio between the number of turns on each
coil. For example, if the primary and secondary windings
each have the same number of turns, then 1 volt across
the primary will also appear as 1 volt across the secondary. If the secondary has twice the number of turns as the
primary then twice the voltage will appear across it. The
transformer also works in reverse – voltage applied to
the secondary will be induced into the primary in proportion to the turns ratio.
The current flowing through the secondary is in inverse
proportion to the turns ratio, such that equal power exists
on the primary and secondary sides of the transformer (it
is not magic – the increased voltage across the secondary of a step-up transformer is traded off against reduced
current!).
It is important to remember that the principle of operation of the transformer depends on AC in the windings
(a)
10k
(b)
1:2
?
0.7 V 2k
inducing an alternating field into the core (i.e. it is the
change in direction of magnetic flux which induces a
current in the secondary, not simply the presence of
constant flux). A DC signal, therefore, is not passed by a
transformer.
Impedances are proportional to the square of the
turns ratio, as discussed in the main text. A transformer
will ‘reflect’ the impedances between which it works. In the
case of a 1:1 transformer the impedance across the secondary is equal to the impedance across the primary, but
in the case of a 1:2 transformer the impedance seen across
the secondary would be four times that across the primary.
Metal core
Primary
winding
Core
?
Secondary
Primary
(c)
1:4
?
Secondary
winding
1:4
20k
FIGURE 12.1 Examples of transformer circuits. (a) What is the impedance across the secondary?
(b) What are the impedance and voltage across the secondary? (c) What is the impedance across the
primary?
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Mic output
impedance
200
ohms
Mixer input
2k impedance
FIGURE 12.2 The input impedance of the mixer
is seen by the microphone, modified by the turns ratio of
the transformer, and vice versa.
impedance five times or more their own impedance,
so 125 ohms is far too low. Similarly, electronic inputs
work best when driven by an impedance considerably
lower than their own, so 3200 ohms is far too high.
Limitations of transformers
Earlier it was mentioned that an audio transformer
must be able to handle the complete audio range. At
very high and very low frequencies this is not easy to achieve, and it is
usual to find that distortion rises at low frequencies, and also to a lesser
extent at very high frequencies. The frequency response falls away at the
frequency extremes, and an average transformer may well be 3 dB down
at 20 Hz and 20 kHz compared with mid frequencies. Good (⫽expensive)
transformers have a much better performance than this. All transformers
are designed to work within certain limits of voltage and current, and if too
high a voltage is applied a rapid increase in distortion results.
The frequency response and distortion performance is affected by the
impedances between which the transformer works, and any particular model
will be designed to give its optimum performance when used for its intended
application. For example, a microphone transformer is designed to handle
voltages in the millivolt range up to around 800 mV or so. The primary
winding will be terminated with about 200 ohms, and the secondary will be
terminated with around 1–2 k (or rather more if a step-up ratio is present). A
line level transformer on the other hand must handle voltages up to 8 volts
or so, and will probably be driven by a source impedance of below 100 ohms,
and will feed an impedance of 10 k or more. Such differing parameters as
these require specialized designs. There is no ‘universal’ transformer.
Transformers are sensitive to electromagnetic fields, and so their siting
must be given consideration. Place an audio transformer next to a mains
transformer and hum will be induced into it, and thus into the rest of the
audio circuit. Most audio transformers are built into metal screening cans
which considerably reduce their susceptibility to radio-frequency interference and the like.
UNBALANCED LINES
‘Unbalanced’ in this context does not mean unstable or faulty. The unbalanced audio line is to be found in virtually all domestic audio equipment,
much semi-professional and some professional audio equipment as
Unbalanced Lines
well. It consists of a ‘send’ and ‘return’ path for the audio signal, the
return path being an outer screening braid which encloses the send wire
and screens it from electromagnetic interference, shown in Figure 12.3.
The screening effect considerably reduces interference such as hum, RF,
and other induction, without eliminating it entirely. If the unbalanced
line is used to carry an audio signal over tens of meters, the cumulative effect of interference may be unacceptable. Earth loops can also be
formed (see Fact File 12.2). Unbalanced lines are normally terminated
in connectors such as phono plugs, DIN plugs and quarter-inch ‘A-gauge’
jack plugs.
An improved means of unbalanced interconnection is shown in Figure
12.4. The connecting lead now has two wires inside the outer screen. One
is used as the signal wire, and instead of the return being provided by the
outer screen, it is provided by the second inner wire. The screening braid is
connected to earth at one end only, and so it merely provides an interference screen without affecting the audio signal.
Output
Input
FIGURE 12.3
Simple unbalanced
interconnection.
Signal wire
Earth
Screen wire (enclosing signal wire along its length)
Earth
FA C T F I L E 1 2 . 2 E A RT H LO O P S
It is possible to wire cables such that the screening braid
of a line is connected to earth at both ends. In many
pieces of audio equipment the earth side of the audio circuit is connected to the mains earth. When two or more
pieces of equipment are connected together this creates
multiple paths to the mains earth, and low-level mains
currents can circulate around the screening braids of the
connecting leads if the earths are at even slightly different potentials. This induces 50 Hz mains hum into the
inner conductor. A common remedy for this problem is
to disconnect the earth wires in the mains plugs on all
the interconnected pieces of equipment except one, the
remaining connection providing the earth for all the other
pieces of equipment via the audio screening braids. This,
though, is potentially dangerous, since if a piece of equipment develops a fault and the mains plug with the earth
connection is unplugged, then the rest of the system is
now unearthed and the fault could in serious cases place
a mains voltage on the metal parts of the equipment. A lot
of units are now ‘double insulated’, so that internal mains
wiring cannot place mains voltage on the metal chassis.
The mains lead is just two core, live and neutral.
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Output
FIGURE 12.4
Alternative unbalanced
interconnection.
Input
Signal wire
Return wire
Earth
Screen (connected at one end only)
Earth
CABLE EFFECTS WITH UNBALANCED LINES
Cable resistance
‘Loop’ resistance is the total resistance of both the send and return paths for
the signal, and generally, as long as the loop resistance of a cable is a couple
of orders of magnitude (i.e. a factor of 100) lower than the input impedance
of the equipment it is feeding, it can be ignored. For example, the output
impedance of a tape recorder might be 200 ohms. The input impedance
of the amplifier it would be connected to would normally be 10 k or more.
The DC resistance of a few meters of connecting cable would only be a
fraction of an ohm and so would not need to be considered. But what about
100 meters of microphone cable? The input impedance of a microphone
amplifier would normally be at least 1000 ohms. Two orders of magnitude
lower than this is 10 ohms. Even 100 meters of mic lead will have a lower
resistance than this unless very thin cheap wire is used, and so again the
DC resistance of microphone cables can be ignored.
Speaker cables do need to be watched, because the input impedance
of loudspeakers is of the order of 8 ohms. Wiring manufacturers quote the
value of DC resistance per unit length (usually 1 meter) of cable, and a typical cable suitable for speaker use would be of 6 amp rating and about 12
milliohms (0.012 ohms) resistance per meter. Consider a 5 meter length of
speaker cable. Its total loop resistance then would be 10 meters multiplied by
0.012 ohms ⫽ 0.12 ohms. This is a bit too high to meet the criterion stated
above, an 8 ohm speaker requiring a cable of around 0.08 ohm loop resistance.
In practice, though, this would probably be perfectly adequate, since there are
many other factors which will affect sound quality. Nevertheless, it does illustrate that quite heavy cables are required to feed speakers, otherwise too much
power will be wasted in the cable itself before the signal reaches the speaker.
If the same cable as above were used for a 40 meter feed to a remote 8
ohm loudspeaker, the loop resistance would be nearly 1 ohm and nearly oneeighth of the amplifier power would be dissipated in heat in the cable. The
moral here is to use the shortest length of cable as is practicable, or if long
runs are required use the 100 volt line system (see ‘100 volt lines’, below).
Cable Effects with Unbalanced Lines
Cable and transformer inductance
The effect of cable inductance (see ‘Sound in electrical form’, Chapter 1)
becomes more serious at high frequencies, but at audio frequencies it is insignificant even over long runs of cable. Conversely, inductance is extremely
important in transformers. The coils on the transformer cores consist
of a large number of turns of wire, and the electromagnetic field of each
turn works against the fields of the other turns. The metallic core greatly
enhances this effect. Therefore, the inductance of each transformer coil is
very high and presents a high impedance to an audio signal. For a given
frequency, the higher the inductance the higher the impedance in ohms.
Cable capacitance
The closer the conductors in a cable are together, the greater the capacitance
(see ‘Sound in electrical form’, Chapter 1). The surface area of the conductors is also important. Capacitance is the opposite of inductance in that, for
a given frequency, the greater the capacitance the lower is the impedance
in ohms. In a screened cable the screening braid entirely encloses the inner
conductor and so the surface area of the braid, as seen by this inner conductor, is quite large. Since large surface area implies high capacitance,
screened cable has a much higher capacitance than ordinary mains wiring,
for example. So when an audio signal looks into a connecting cable it sees
the capacitance between the conductors and therefore a rather less-thaninfinite impedance between them, especially at high frequencies. A small
amount of the signal can therefore be conducted to earth via the screen.
In the diagram in Figure 12.5 there are two resistors of equal value.
A voltage V1 is applied across the two. Because the value of the resistors is
the same, V1 is divided exactly in half, and V2 will be found to be exactly half
the value of V1. If the lower resistor were to be increased in value to
400 ohms then twice the voltage would appear across it than across the
upper resistor. The ratio of the resistors equals the ratio of the voltages
200 ohms
across them.
V1
Consider a 200 ohm microphone looking into a mic lead, as
shown in Figure 12.6(a). C is the capacitance between the screening
braid and the inner core of the cable. The equivalent of this circuit
200 ohms
V2
is shown in Figure 12.6(b). Manufacturers quote the capacitance of
cables in picofarads (pF) per unit length. A typical value for screened
cable is 200 pF (0.0002 μF) per meter. A simple formula exists for
determining the frequency at which 3 dB of signal is lost for a given
FIGURE 12.5 The voltage
capacitance and source resistance:
f ⫽ 159 155/RC
V2 across the output is half the
input voltage (V1).
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CHAPTER 12: Lines and Interconnection
FIGURE 12.6
(a) A microphone with
a 200 ohm output
impedance is connected
to an amplifier. (b) Lead
capacitance conducts high
frequencies to ground more
than low frequencies, and
thus the cable introduces
HF roll-off. V2 is lower at HF
than at LF.
(a)
Amplifier
Microphone
200 ohms
Signal
C
Screen
(b)
200 ohms
V1
Lead
capacitance
V2
where f ⫽ frequency in hertz (Hz), R ⫽ resistance in ohms, and
C ⫽ capacitance in microfarads (μF).
To calculate the capacitance which will cause a 3 dB loss at 40 kHz, putting it safely out of the way of the audio band, the formula must be rearranged to give the maximum value of acceptable capacitance:
C ⫽ 159 155/Rf
Thus, if R ⫽ 200 (mic impedance), f ⫽ 40 000:
C ⫽ 159 155 ⫼ (200 ⫻ 40 000) ⫽ 0.02 μF
So a maximum value of 0.02 μF of lead capacitance is acceptable for a mic
lead. Typical lead capacitance was quoted as 0.0002 μF per meter, so 100 meters
will give 0.02 μF, which is the calculated acceptable value. Therefore one could
safely use up to 100 meters of typical mic cable with a standard 200 ohm
microphone without incurring significant signal loss at high frequencies.
The principle applies equally to other audio circuits, and one more
example will be worked out. A certain tape recorder has an output impedance of 1 k. How long a cable can it safely drive? From the above formula:
C ⫽ 159 155 ⫼ (1000 ⫻ 40 000) ⫽ 0.004 μF
In this case, assuming the same cable capacitance, the maximum safe
cable length is 0.004 ÷ 0.0002 ⫽ 20 meters. In practice, modern audio
equipment generally has a low enough source impedance to drive long
Balanced Lines
leads, but it is always wise to check up on this in the manufacturer’s specification. Probably of greater concern will be the need to avoid long runs of
unbalanced cable due to interference problems.
BALANCED LINES
The balanced line is better at rejecting interference than the unbalanced
line, and improvements upon the performance of the unbalanced line in
this respect can be 80 dB or more for high-quality microphone lines.
As shown in Figure 12.7, the connecting cable consists of a pair of inner
conductors enclosed by a screening braid. At each end of the line is a ‘balancing’ transformer. The output amplifier feeds the primary of the output
transformer and its voltage appears across the secondary. The send and
return paths for the audio signal are provided by the two inner conductors,
and the screen does not form part of the audio circuit. If an interference
signal breaks through the screen it is induced equally into both signal lines.
At the secondary transformer’s primary the induced interference current,
flowing in the same direction in both legs of the balanced line, cancels
out, thus rejecting the interference signal. Two identical signals, flowing in
opposite directions, cancel out where they collide.
Such an interfering signal is called a ‘common mode’ signal because
it is equal and common to both audio lines. The rejection of this in the
transformer is termed ‘common mode rejection’ (CMR). A common mode
rejection ratio (CMRR) of at least 80 dB may be feasible. Meanwhile, the
legitimate audio signal flows through the primary of the transformer
as before, because the signal appears at each end of the coil with equal
strength but opposite phase. Such a signal is called a ‘differential signal’,
and the balanced input is also termed a ‘differential input’ because it
accepts differential mode signals but rejects common mode signals.
So balanced lines are used for professional audio connections because of
their greatly superior rejection of interference, and this is particularly useful
when sending just a few precious millivolts from a microphone down many
meters of cable to an amplifier.
Output
Input
Interference
1:1
Audio signal
Interference
1:1
FIGURE 12.7
A balanced interconnection
using transformers.
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WORKING WITH BALANCED LINES
In order to avoid earth loops (see Fact File 12.2) with the balanced line, the
earth screen is often connected at one end only, as shown in Figure 12.8(a),
and still acts as a screen for the balanced audio lines. There is now no earth
link between the two pieces of equipment, and so both can be safely earthed
at the mains without causing an earth loop. The transformers have ‘isolated’
the two pieces of equipment from each other. The one potential danger with
this is that the connecting lead with its earth disconnected in the plug at
one end may later be used as a microphone cable. The lack of earth continuity between microphone and amplifier will cause inadequate screening of
the microphone, and will also prevent a phantom power circuit being made
(see ‘Microphone powering options’, Chapter 3), so such cables and tie-lines
should be marked ‘earth off ’ at the plug without the earth connection.
Unfortunately, not all pieces of audio equipment have balanced inputs
and outputs, and one may be faced with the problem of interfacing a
FIGURE 12.8
(a) Balanced output to
balanced input with screen
connected to earth only
at output. (b) Balanced
output to unbalanced input.
(c) Unbalanced output to
balanced input.
(a)
(b)
(c)
Balanced output
Balanced input
Balanced output
Unbalanced input
Unbalanced output
Balanced input
Star-Quad Cable
balanced output with an unbalanced input, and an unbalanced output with
a balanced input. A solution is shown in Figure 12.8(b), where the output
transformer is connected to the signal and earth of the unbalanced input to
give signal continuity. Because the input is unbalanced, there is no common
mode rejection and the line is as susceptible to interference as an ordinary
unbalanced line is. But notice that the screen is connected at one end only,
so at least one can avoid an earth loop.
Figure 12.8(c) illustrates an unbalanced output feeding a balanced input.
The signal and earth from the output feed the primary of the input transformer. Again the screen is not connected at one end so earth loops are avoided.
Common mode rejection of interference at the input is again lost, because one
side of the transformer primary is connected to earth. A better solution is to
use a balancing transformer as close to the unbalanced output as possible, preferably before sending the signal over any length of cable. In the longer term it
would be a good idea to fit balancing transformers inside unbalanced equipment with associated three-pin XLR-type sockets (see Fact File 12.3) if space
inside the casing will allow. (Wait until the guarantee has elapsed first!)
STAR-QUAD CABLE
Two audio lines can never occupy exactly the same physical space, and any
interference induced into a balanced line may be slightly stronger in one
line than in the other. This imbalance is seen by the transformer as a small
differential signal which it will pass on, so a small amount of the unwanted
FA C T F I L E 1 2 . 3 X L R -3 C O N N E CTORS
The most common balanced connector in professional
audio is the XLR-3. This connector has three pins (as
shown in the diagram), carrying respectively:
Pin 1 Screen
Pin 2 Signal (Live or ‘hot’)
Pin 3 Signal (Return or ‘cold’)
It is easy to remember this configuration, since X–
L–R stands for Xternal, Live, Return. Unfortunately, an
American convention still hangs on in some equipment
which reverses the roles of pins 2 and 3, making pin
2 return and pin 3 live (or ‘hot’). The result of this is an
apparent absolute phase reversal in signals from devices
using this convention when compared with an identical
signal leaving a standard device. Modern American equipment mostly uses the European convention, and American
manufacturers have now agreed to standardize on this
approach.
3
2
1
Viewed from end of
male pins
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CHAPTER 12: Lines and Interconnection
signal will still get through. To help combat this, the two
audio lines are twisted together during manufacture so as to
Outer sleeve
present, on average, an equal face to the interference along
both lines. A further step has been taken in the form of a
cable called ‘star-quad’. Here, four audio lines are incorpoScreening braid
rated inside the screen, as shown in Figure 12.9.
It is connected as follows. The screen is connected
FIGURE 12.9 Four conductors are used in
as usual. The four inner cores are connected in pairs
star-quad cable.
such that two of the opposite wires (top and bottom
in the figure) are connected together and used as one line, and the other
two opposite wires are used as the other. All four are twisted together
along the length of the cable during manufacture. This configuration
ensures that for a given length of cable, both audio lines are exposed to an
interference signal as equally as possible so that any interference is induced
as equally as possible. So the input transformer sees the interference as a
virtually perfect common mode signal, and efficiently rejects it. This may
seem like taking things to extremes, but star-quad is in fact quite widely
used for microphone cables. When multicore cables are used which contain
many separate audio lines in a single thick cable, the balanced system gives
good immunity from crosstalk, due to the fact that a signal in a particular
wire will be induced equally into the audio pair of the adjacent line, and
will therefore be a common mode signal. Star-quad multicores give even
lower values of crosstalk.
Four inner
conductors
ELECTRONIC BALANCING
Much audio equipment uses an electronically balanced arrangement
instead of a transformer, and it is schematically represented as in Figure
12.10. The transformers have been replaced by a differential amplifier. The
differential amplifier is designed to respond only to differential signals, as is
the case with the transformer, and has one positive and one negative input.
Electronically balanced and transformer-balanced equipment can of course
be freely intermixed. Reasons for dispensing with transformers include
lower cost (a good transformer is rather more expensive than the electronic
components which replace it), smaller size (transformers take up at least
a few cubic centimeters, the alternative electronics rather less), less susceptibility to electromagnetic interference, and rather less sensitivity to the
impedances between which they work with respect to distortion, frequency
response and the like.
Good electronic balancing circuitry is, however, tricky to design, and the
use of high-quality transformers in expensive audio equipment may well
100 Volt Lines
Differential output
+
–
Balanced input
Differential amplifier
+
–
be a safer bet than electronic balancing of unknown performance. The best
electronic balancing is usually capable of equal CMR performance to the
transformer. Critics of transformer balancing cite factors such as the lowfrequency distortion performance of a transformer, and its inability to pass
extremely low frequencies, whereas critics of electronic balancing cite the
better CMR available from a transformer when compared with a differential amplifier, and the fact that only the transformer provides true isolation
between devices. Broadcasters often prefer to use transformer balancing
because signals are transferred over very long distances and isolation is
required, whereas recording studios often prefer electronic balancing, claiming that the sound quality is better.
100 VOLT LINES
Principles
In ‘Cable resistance’, above, it was suggested that the resistance of even quite
thick cables was still sufficient to cause signal loss in loudspeaker interconnection unless short runs were employed. But long speaker lines are frequently
unavoidable, examples being: backstage paging and show relay speakers in
theaters; wall-mounted speakers in lecture theaters and halls; paging speakers
in supermarkets and factories; and open-air ‘tannoy ’ horns at fairgrounds and
fêtes. All these require long speaker runs, or alternatively a separate power
amplifier sited close to each speaker, each amplifier being driven from the line
output of a mixer or microphone amplifier. The latter solution will in most
cases be considered an unnecessarily expensive and complicated solution.
So the ‘100 volt line’ was developed so that long speaker cable runs could be
employed without too much signal loss along them.
The problem in normal speaker connection is that the speaker cable has
a resistance comparable with, or even greater than, the speaker’s impedance over longer runs. It was shown in ‘limitations of transformers’, above,
that a transformer reflects impedance according to the square of the turns
ratio. Suppose a transformer with a turns ratio of 5:1 is connected to the
input of an 8 ohm speaker, as shown in Figure 12.11. The square of
FIGURE 12.10
An electronically balanced
interconnection using
differential amplifiers.
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CHAPTER 12: Lines and Interconnection
the turns ratio is 25:1 so the impedance across the
primary of the transformer is 25 ⫻ 8 ⫽ 200 ohms.
8 ohm
Now, the effective impedance of the speaker is much
Speaker cable
speaker
greater than the cable resistance, so most of the voltage will now reach the primary of the transformer
FIGURE 12.11 Transformer coupling to a
and thence to the secondary and the speaker itself.
loudspeaker as used in 100 volt line systems.
But the transformer also transforms voltage, and the
voltage across the secondary will only be a fifth of that
across the primary. To produce 20 volts across the speaker then, one must
apply 100 volts to the primary.
In a 100 volt line system, as shown in Figure 12.12, a 50 W power
amplifier drives a transformer with a step-up ratio of 1:5. Because the output impedance of a power amplifier is designed to be extremely low, the
impedance across the secondary is also low enough to be ignored. The
20 volts, 2.5 amp output of the 50 watt amplifier is stepped up to 100 volts.
The current is correspondingly stepped down to 0.5 amps (see Fact File
12.1), so that the total power remains the same. Along the speaker line
there is a much higher voltage than before, and a much lower current. The
voltage drop across the cable resistance is proportional to the current flowing through it, so this reduction in current means that there is a much
smaller voltage drop due to the line. At the speaker end, a transformer
restores the voltage to 20 volts and the current to 2.5 amps, and so the
original 50 watts is delivered to the speaker.
A 50 watt amplifier has been used in the discussion. Any wattage of
amplifier can be used, the transformer being chosen so that the step-up
ratio gives the standard 100 volts output when the amplifier is delivering
its maximum power output. For example, an amplifier rated at 100 watts
into 8 ohms produces about 28 V. The step-up ratio of the line transformer
would then have to be 28:100, or 1:3.6, to give the standard 100 volt output when the amplifier is being fully driven.
Returning to the loudspeaker end of the circuit. What if the speaker is
only rated at 10 watts? The full 100 watts of the above amplifier would
burn it out very quickly, and so a step-down ratio of the speaker transformer is chosen so that it receives only 10 watts. As 10 watts across
8 ohms is equivalent to around 9 volts, the required speaker transformer
would have a step-down ratio of 100:9, or approximately 11:1.
5:1
Working with 100 volt lines
Speaker line transformers usually have a range of terminals labeled such
that the primary side has a choice of wattage settings (e.g. 30 W, 20 W, 10 W,
100 Volt Lines
FIGURE 12.12
Voltage/current relationships
in an example of 100 volt
line operation.
2 W) and the secondary gives a choice of speaker impedance, usually 15
ohms, 8 ohms and 4 ohms. This choice means that a number of speaker
systems can be connected along the line (a transformer being required for
each speaker enclosure), the wattage setting being appropriate to the speaker’s coverage. For example, a paging system in the backstage area of a theater could be required to feed paging to six dressing rooms, a large toilet, and
a fairly noisy green room. The dressing rooms are small and quiet, and so
small speakers rated at 10 watts are employed with line transformers wired
for 2 watts output. The large toilet requires greater power from the speaker,
so one could use a 10 W speaker with a line transformer wired for 10 W output. The noisy green room could have a rather larger 20 W speaker with a
line transformer wired for 20 W output. In this way, each speaker is supplied
only with the wattage required to make it loud enough to be clearly heard in
that particular room. A 20 W speaker in a small dressing room would be far
too loud, and a 2 W speaker in a larger, noisier room would be inadequate.
As a string of loudspeakers is added to the system, one must be careful
that the total wattage of the speakers does not exceed the output wattage of
the power amplifier, or the latter will be overloaded. In the example, the six
dressing rooms were allocated 2 W each, total 12 W. The toilet was allocated
10 W, the green room 20 W. The total is therefore 42 W, and a 50 W amplifier and line transformer would be adequate. In practice, a 100 W amplifier
would be chosen to allow for both a good safety margin and plenty of power
in hand if extra speakers need to be connected at a later date.
From the foregoing, it might well be asked why the 100 volt line system is not automatically used in all speaker systems. One reason is that
100 volts is high enough to give an electric shock and so is potentially dangerous in the domestic environment and other places where members of
the public could interfere with an inadequately installed system. Second,
the ultimate sound quality is compromised by the presence of transformers
in the speaker lines – they are harder to design than the microphone and
line-level transformers already discussed, because they have to handle high
voltages as well as several amps – and whereas they still give a perfectly
adequate and extremely useful performance in paging and background
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CHAPTER 12: Lines and Interconnection
music applications, they are not therefore used in high-quality PA systems
or hi-fi and studio monitor speakers.
600 OHMS
One frequently sees 600 ohms mentioned in the specifications of mixers,
microphone amplifiers, and other equipment with line-level outputs. Why
is 600 ohms so special? The short answer is: it is not.
Principles
As has been shown, the output impedances of audio devices are low, typically 200 ohms for microphones and the same value or rather less for linelevel outputs. The input impedances of devices are much higher, at least
1–2 k for microphone inputs and 10 k or more for line inputs. This is to
ensure that virtually the whole of the output voltage of a piece of equipment appears across the input it is feeding. Also, lower input impedances
draw a higher current for a given voltage, the obvious example of this being
an 8 ohm loudspeaker which draws several amps from a power amplifier.
So a high input impedance means that only very small currents need to be
supplied by the outputs, and one can look upon microphone and line-level
signals as being purely voltage without considering current.
This works fine, unless you are a telephone company that needs to send
its signals along miles of cable. Over these distances, a hitherto unmentioned parameter comes into play which would cause signal loss if not dealt
with, namely the wavelength of the signal in the cable. Now the audio signal is transmitted along a line at close to the speed of light (186 000 miles
per second or 3 ⫻ 108 meters per second). The shortest signal wavelength
will occur at the upper limit of the audio spectrum, and will be around
9.3 miles (14.9 km) at 20 kHz.
When a cable is long enough to accommodate a whole wavelength or
more, the signal can be reflected back along the line and cause some cancelation of the primary signal. Even when the cable run is somewhat less
than a wavelength some reflection and cancelation still occurs. To stop this
from happening the cable must be terminated correctly, to form a so-called
‘transmission line’, and input and output impedances are chosen to be
equal. The value of 600 ohms was chosen many decades ago as the standard value for telecommunications, and therefore the ‘600 ohm balanced
line’ is used to send audio signals along lines which need to be longer than
a mile or so. It was chosen because comparatively little current needs to
be supplied to drive this impedance, but it is not high enough to allow
600 Ohms
much interference, as it is much easier for interference to affect a highimpedance circuit than a low one. Thus, professional equipment began to
appear which boasted ‘600 ohms’ to make it compatible with these lines.
Unfortunately, many people did not bother to find out, or never understood,
why 600 ohms was sometimes needed, and assumed that this was a professional audio standard per se, rather than a telecommunications standard. It
was used widely in broadcasting, which has parallels with telecommunications, and may still be found in many cases involving older equipment.
The 600 ohm standard also gave rise to the standard reference level unit
of 0 dBm, which corresponds to 1 mW of power dissipated in a resistance
of 600 ohms. The corresponding voltage across the 600 ohm resistance at
0 dBm is 0.775 volts, and this leads some people still to confuse dBm with
dBu, but 0 dBu refers simply to 0.775 volts with no reference to power or
impedance: dBu is much more appropriate in modern equipment, since,
as indicated above, the current flowing in most interfaces is negligible and
impedances vary; dBm should only correctly be used in 600 ohm systems,
unless an alternative impedance is quoted (e.g. dBm (75 ohms) is sometimes used in video equipment where 75 ohm termination is common).
Problems with 600 ohm equipment
A 600 ohm output impedance is too high for normal applications. With 600
ohms, 200 pF per meter cable, and an acceptable 3 dB loss at 40 kHz, the
maximum cable length would be only around 33 m, which is inadequate for
many installations. (This HF loss does not occur with the properly terminated 600 ohm system because the cable assumes the properties of a transmission line.) Furthermore, consider a situation where a mixer with a 600
ohm output impedance is required to drive five power amplifiers, each with
an input impedance of 10 k. Five lots of 10 k in parallel produce an effective
impedance of 2 k, as shown in Figure 12.13(a). Effectively then 600 ohms is
driving 2 k, as shown in Figure 12.13(b). If V1 is 1 volt then V2 (from Ohm’s
law) is only 0.77 volts. Almost a quarter of the audio signal has been lost,
and only a maximum of 33 meters of cable can be driven anyway.
Despite this, there are still one or two manufacturers who use 600 ohm
impedances in order to appear ‘professional’. It actually renders their equipment less suitable for professional use, as has been shown. One specification that is frequently encountered for a line output is something like:
‘capable of delivering ⫹20 dBu into 600 ohms’. Here ⫹20 dBu is 7.75 volts,
and 600 ohms is quite a low impedance, thus drawing more current from
the source for a given voltage than, say, 10 kΩ does. The above specification
is therefore useful, because it tells the user that the equipment can deliver
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FIGURE 12.13
(a) Considerable signal
loss can result if a 600
ohm output is connected
to a number of 10 k inputs
in parallel. (b) Electrical
equivalent.
(a)
Mixer output
600 ohms
5 power amplifiers
10k
10k
10k
10k
10k
= 2k
(b)
600 ohms
V1
2k
V2
7.75 volts even into 600 ohms, and can therefore safely drive, say, a stack
of power amplifiers, and/or a number of tape recorders, etc., without being
overloaded. A domestic cassette recorder, for instance, despite having a low
output impedance, could not be expected to do this. Its specification may
well state ‘2 volts into 10 kΩ or greater ’. This is fine for domestic applications, but one should do a quick calculation or two before asking it to drive
all the power amplifiers in the building at once.
DI BOXES
Overview
A frequent requirement is the need to interface equipment that has basically non-standard unbalanced outputs with the standard balanced inputs
of mixers, either at line level or microphone level. An electric guitar, for
example, has an unbalanced output of fairly high impedance – around
10 kΩ or so. The standard output socket is the ‘mono’ quarter-inch jack,
and output voltage levels of around a volt or so (with the guitar’s volume
controls set to maximum) can be expected. Plugging the guitar directly into
the mic or line-level input of a mixer is unsatisfactory for several reasons:
the input impedance of the mixer will be too low for the guitar, which likes
DI Boxes
to drive impedances of 500 kΩ or more; the guitar output is unbalanced so
the interference-rejecting properties of the mixer’s balanced input will be
lost; the high output impedance of the guitar renders it incapable of driving long studio tie-lines; and the guitarist will frequently wish to plug the
instrument into an amplifier as well as the mixer, and simply using the
same guitar output to feed both via a splitter lead electrically connects
the amplifier to the studio equipment which causes severe interference and
low-frequency hum problems. Similar problems are encountered with other
instruments such as synthesizers, electric pianos, and pickup systems for
acoustic instruments.
To connect such an instrument with the mixer, a special interfacing unit
known as a DI box (DI ⫽ direct injection) is therefore employed. This unit
will convert the instrument’s output to a low-impedance balanced signal,
and also reduce its output level to the millivolt range suitable for feeding
a microphone input. In addition to the input jack socket, it will also have
an output jack socket so that the instrument’s unprocessed signal can be
passed to an amplifier as well. The low-impedance balanced output appears
on a standard three-pin XLR panel-mounted plug which can now be looked
upon as the output of a microphone. An earth-lift switch is also provided
which isolates the earth of the input and output jack sockets from the XLR
output, to trap earth loop problems.
Passive DI boxes
The simplest DI boxes contain just a transformer, and are termed ‘passive’
because they require no power supply. Figure 12.14 shows the circuit. The
transformer in this case has a 20:1 step-down ratio, converting the fairly
high output of the instrument to a lower output suitable for feeding microphone lines. Impedance is converted according to the square of the turns
ratio (400:1), so a typical guitar output impedance of 15 kΩ will be stepped
Input
jack
socket
20:1
XLR output
2
1
3
Link
output
jack
socket
Earth lift switch
Input earth connected
to metal case
FIGURE 12.14
A simple passive directinjection box.
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CHAPTER 12: Lines and Interconnection
down to about 40 ohms which is comfortably low enough to drive long
microphone lines. But the guitar itself likes to look into a high impedance.
If the mixer’s microphone input impedance is 2 kΩ, the transformer will
step this up to 800 kΩ which is adequately high for the guitar. The ‘link
output jack socket’ is used to connect the guitar to an amplifier if required.
Note the configuration of the input jack socket: the make-and-break contact normally short-circuits the input which gives the box immunity from
interference, and also very low noise when an instrument is not plugged in.
Insertion of the jack plug opens this contact, removing the short-circuit. The
transformer isolates the instrument from phantom power on the microphone line.
This type of DI box design has the advantages of being cheap, simple, and
requiring no power source – there are no internal batteries to forget to change.
On the other hand, its input and output impedances are entirely dependent
on the reflected impedances each side of the transformer. Unusually low
microphone input impedances will give insufficiently high impedances for
many guitars. Also, instruments with passive volume controls can exhibit
output impedances as high as 200 kΩ with the control turned down a few
numbers from maximum, and this will cause too high an impedance at the
output of the DI box for driving long lines. The fixed turns ratio of the transformer is not equally suited to the wide variety of instruments the DI box
will encounter, although several units have additional switches which alter
the transformer tapping giving different degrees of attenuation.
Active DI boxes
The active DI box replaces the transformer with an electronic circuit which
presents a constant very high impedance to the instrument and provides
a constant low-impedance output. Additionally, the presence of electronics provides scope for including other features such as several switched
attenuation values (say ⫺20 dB, ⫺40 dB, ⫺60 dB), high and low filters
and the like. The box is powered either by internal batteries, or preferably
by the phantom power on the microphone line. If batteries are used, the
box should include an indication of battery status; a ‘test’ switch is often
included which lights an LED when the battery is good. Alternatively, an
LED comes on as a warning when the voltage of the battery drops below
a certain level. The make-and-break contacts of the input jack socket are
often configured so that insertion of the jack plug automatically switches
the unit on. One should be mindful of this because if the jack plug is left
plugged into the unit overnight, for instance, this will waste battery power.
Usually the current consumption of the DI box is just a few milliamps, so
Splitter Boxes
the battery will last for perhaps a hundred hours. Some guitar and keyboard
amplifiers offer a separate balanced output on an XLR socket labeled ‘DI’ or
‘studio’ which is intended to replace the DI box, and it is often convenient
to use this instead.
DI boxes are generally small and light, and they spend much of their
time on the floor being kicked around and trodden on by musicians and
sound engineers. Therefore, rugged metal (not plastic) boxes should be
used and any switches, LEDs, etc. should be mounted such that they are
recessed or shrouded for protection. Switches should not be easily moved
by trailing guitar leads and feet. The DI box can also be used for interfacing
domestic hi-fi equipment such as cassette recorders and radio tuners with
balanced microphone inputs.
SPLITTER BOXES
The recording or broadcasting of live events calls for the outputs of microphones and instruments to be fed to at least two destinations, namely the
PA mixer and the mixer in the mobile recording or outside broadcast van.
The PA engineer can then balance the sound for the live audience, and the
recording/broadcast balancer can independently control the mix for these
differing requirements. It is possible of course to use two completely separate sets of microphones for this, but when one considers that there may be
as many as ten microphones on a drum kit alone, and a vocalist would find
the handling of two mics strapped together with two trailing leads rather
unacceptable, a single set of microphones plugged into splitter boxes is the
obvious way to do it. Ten or 15 years ago recording studios and broadcasting companies would have frowned on this because the quality of some of
the microphones then being used for PA was insufficient for their needs;
but today PA mics tend to be every bit as good as those found in studios,
and indeed many of the same models are common to both environments.
A microphone cannot be plugged into two microphone inputs of two
separate mixers directly because on the one hand this will electrically connect one mixer with the other causing ground loop and interference problems, not to mention the fact that one phantom power circuit will be
driving directly into the other phantom power circuit; and on the other
hand the impedance seen by the microphone will now be the parallel result
of the two mixers resulting in impedances of as low as 500 ohms which is
too low for many microphones. A splitter box is therefore used which isolates the two mixers from each other and maintains a suitable impedance
for the microphone. A splitter box will often contain a transformer with
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one primary winding for the microphone and two separate secondary windings giving the two outputs, as shown in Figure 12.15.
The diagram requires a bit of explanation. First, phantom power must
be conveyed to the microphone. In this case, output 2 provides it via the
center tap of its winding which conveys the power to the center tap of the
primary. The earth screen, on pin 1 of the input and output XLR sockets,
is connected between the input and output 2 only, to provide screening of
the microphone and its lead and also the phantom power return path. Note
that pin 1 of output 1 is left unconnected so that earth loops cannot be
created between the two outputs.
The turns ratio of the transformer must be considered next. The
1:0.7:0.7 indicates that each secondary coil has only 0.7 times the windings
of the primary, and therefore the output voltage of the secondaries will each
be 0.7 times the microphone output, which is about 3 dB down. There is
therefore a 3 dB insertion loss in the splitter transformer. The reason for this
is that the impedance as seen by the microphone must not be allowed to go
too low. If there were the same number of turns on each coil, the microphone would be driving the impedance across each output directly in parallel. Two mixers with input impedances of 1 kΩ would therefore together
load the microphone with 500 ohms, which is too low. But the illustrated
turns ratio means that each 1 kΩ impedance is stepped up by a factor of
1:(0.7)2 ⫽ 1:0.5, so each mixer input appears as 2 kΩ to the microphone,
giving a resultant parallel impedance of 1 kΩ, equal to the value of each
mixer on its own which is fine. The 3 dB loss of signal is accompanied by an
effective halving of the microphone impedance as seen by each mixer, again
due to the transformer’s impedance conversion according to the square of
the turns ratio, so there need not be a signal-to-noise ratio penalty.
FIGURE 12.15
A simple passive
splitter box.
Microphone input
1:0.7:0.7
1
2
1
Output 1
2
3
3
2
1
Output 2
3
Earth connected
to metal case
Jackfields (Patchbays)
Because of the simple nature of the splitter box, a high-quality transformer and a metal case with the necessary input and output sockets are all
that is really needed. Active electronic units are also available which eliminate the insertion loss and can even provide extra gain if required. The
advantages of an active splitter box over its transformer counterpart are,
however, of far less importance than, say, the advantages that an active DI
box has over its passive counterpart.
JACKFIELDS (PATCHBAYS)
Overview
A jackfield (or patchbay) provides a versatile and comprehensive means of
interconnecting equipment and tie-lines in a non-permanent manner such
that various source and destination configurations can quickly and easily
be set up to cater for any requirements that may arise.
For example, a large mixing console may have microphone inputs, line
inputs, main outputs, group outputs, auxiliary outputs, and insert send
and returns for all input channels and all outputs. A jackfield, which
usually consists of banks of 19 inch (48 cm) wide rack-mounting modules filled with rows of quarter-inch ‘GPO ’ (Telecom)-type balanced jack
sockets, is used as the termination point for all of the above facilities so
that any individual input or output can be separately accessed. There are
usually 24 jack sockets to a row, but sometimes 20 or 28 are
encountered. Multicore cables connect the mixer to the jackfield, multipin connectors normally being employed at the
mixer end. At the jackfield end multipin connectors can again
be used, but as often as not the multicores will be hard wired
to the jack socket terminals themselves. The layout of a mixer
jackfield was discussed in ‘Patchfield or jackfield’, Chapter 5.
In addition to the mixer’s jackfield there will be other jackfields either elsewhere in the rack or in adjacent racks which provide connection points for the other equipment and tie-lines. In
a recording studio control room there will be such things as multitrack inputs and outputs, mic and line tie-lines linking control
room to studio, outboard processor inputs and outputs, and tielines to the other control rooms and studios within a studio complex. A broadcasting studio will have similar arrangements, and
there may also be tie-lines linking the studio with nearby concert
halls or transmitter distribution facilities. A theater jackfield will
FIGURE 12.16 A typical busy
in addition carry tie-lines leading to various destinations around
jackfield. (Courtesy of the RSC.)
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the auditorium, back stage, in the wings, in the orchestra pit, and almost
anywhere else. There is no such thing as too many tie-lines in a theater.
Patch cords
Patch cords (screened leads of around 1 meter in length terminated at each
end with a ‘B’-gauge jack plug (small tip) giving tip/ring/sleeve connections)
are used to link two appropriate sockets. The tip is live (corresponding to
pin 2 on an XLR), the ring is return (pin 3 on an XLR), and the sleeve is
earth (pin 1 on an XLR), providing balanced interconnection. The wire or
‘cord’ of a normal patch cord is colored red. Yellow cords should indicate
that the patch cord reverses the phase of the signal (i.e. tip at one end is
connected to ring of the other end) but this convention is not followed rigidly, leading to potential confusion. Green cords indicate that the earth is
left unconnected at one end, and such cords are employed if an earth loop
occurs when two separately powered pieces of equipment are connected
together via the jackfield.
A large-tipped ‘A’-gauge jack plug, found on the end of most stereo headphone leads and often called a stereo jack (although it was originally used as
a single-line balanced jack before stereo came into vogue), may sometimes
be inserted into a GPO-type socket. This works, but the spring-loaded tip
connector is positioned so as to make secure contact with the small ‘B’-type
tip and the large ‘A’ tip can sometimes bend the contact. When the correct
type of jack plug is later inserted there will be intermittent or no contact
between the tip and the socket’s tip connector. The insertion of large tipped
jack plugs should therefore be discouraged.
Normaling
Normally, jackfield insertion points will be unused. Therefore the insertion
send socket must be connected to the insertion return socket so that signal continuity is achieved. When an outboard unit is to be patched in, the
insertion send socket is used to feed the unit’s input. The unit’s output
is fed to the insertion return socket on the jackfield. This means that the
send signal must be disconnected from the return socket and replaced by
the return from the processor. To effect this requirement, extra make-andbreak ‘normaling’ contacts on the jack socket are employed. Figure 12.17
shows how this is done. The signal is taken from the top jack socket to the
bottom jack socket via the black triangle make-and-break contactors on the
bottom socket. There is signal continuity. If a jack plug is now inserted into
the bottom socket the contacts will be moved away from the make-andbreak triangles, disconnecting the upper socket’s signal from it. Signal from
Jackfields (Patchbays)
Insertion
send
socket
FIGURE 12.17
Normaling at jackfield
insertion points.
From insertion
point output
Common
earth
Insertion
return
socket
To insertion
point input
that jack plug now feeds the return socket. The make-and-break contacts
on the upper jack socket are left unused, and so insertion of a jack plug into
this socket alone has no effect on signal continuity. The send socket therefore simply provides an output signal to feed the processor. This arrangement is commonly termed ‘half normaling’ because only the lower socket
in Figure 12.17 uses the make-and-break contacts.
Sometimes these make-and-break contacts are wired so that the signal is
also interrupted if a jack plug is inserted into the send socket alone. This
can be useful if, for instance, an alternative destination is required for, say,
a group output. Insertion of a jack plug into the send socket will automatically mute the group’s output and allow its signal to be patched in elsewhere,
without disturbing the original patching arrangement. Such a wiring scheme
is, however, rather less often encountered. It is termed ‘full normaling’.
In addition to providing insert points, normaling can also be used to
connect the group outputs of a mixer to the inputs of a multitrack recorder
or a set of power amplifiers. The recorder or amplifier inputs will have
associated sockets on the jackfield, and the mixer’s group output sockets
will be normaled to these as described. If
need be, the input sockets can be overplugged
in order to drive the inputs with alterna1
2
3
4
5
6
tive signals, automatically disconnecting the
Matrix outputs
mixer’s outputs in the process. Figure 12.18
illustrates a small section of a mixer’s jackfield, and how it could be labeled. The upper
row gives access to the mixer’s matrix outN
N
N
N
N
N
6
5
1
2
4
3
puts. Patch cords inserted here can convey
AMP input tie-lines
the signals to other sockets where they are
FIGURE 12.18 Part of a mixer’s jackfield.
required, for example to processor inputs,
7
8
N
N
7
8
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CHAPTER 12: Lines and Interconnection
or for foldback feeds. The lower row is connected to another jackfield in
a room which houses racks of power amplifiers. It will be seen that an N
in a circle is written above each of the numbers; this indicates that these
are normaled to the sockets above. Thus, with no patch cords inserted, the
upper row is by default connected to the lower row, and the matrix outputs
automatically drive the amp input tie-lines and thence the power amplifiers. This would be the routine mode of operation, and the two rows are
half normaled together. If, however, it is desired that an amp room tie-line
needs to be fed from another device, say, for example, a digital delay unit is
required to be inserted between matrix output 1 and tie-line 1, the digital
delay’s output is plugged into tie-line 1 socket, breaking the normaling to
the socket above. Matrix output 1 socket is then patched to drive the digital
delay’s input.
If an N in a circle also appears beneath the upper sockets, this would
indicate full normaling. Inserting a jack plug into an upper socket so labeled
would then disconnect that matrix output from the amp input tie-line.
Other jackfield facilities
Other useful facilities in a jackfield include multiple rows of interconnected
jacks, or ‘mults’. These consist of, say, six or eight or however many adjacent sockets which are wired together in parallel so that a mixer output
can be patched into one of the sockets, the signal now appearing on all of
the sockets in the chain which can then be used to feed a number of power
amplifier inputs in parallel, or several tape machines. The disadvantage of
this poor man’s distribution amplifier is that if there is a short-circuit or an
interference signal on any of the inputs it is feeding, this will affect all of
the sockets on the mult. There is no isolation between them.
Outboard equipment will generally be equipped with XLR-type input
and output connectors (or sometimes ‘mono’ jack sockets). It is useful
therefore to have a panel of both male and female XLR sockets near to
where such outboard gear is usually placed, these being wired to a row of
sockets on the jackfield to facilitate connection between the mixer or tielines and these units.
Special additional make-and-break contacts can be included in the jack
socket which are operated in the usual manner by jack plug insertion but
have no contact with any part of the audio signal. These can be used to
activate warning indicators to tell operators that a certain piece of equipment is in use, for example.
Since most if not all of the interconnections in a rig pass through the
jackfield it is essential that the contacts are of good quality, giving reliable
Distribution Amplifiers
service. Palladium metal plating is employed which is tough, offering good
resistance to wear and oxidation. This should always be looked for when
jackfield is being ordered. Gold or silver plating is not used because it would
quickly wear away in the face of professional use. The latter also tarnishes
rather easily.
There is a miniature version known as the Bantam jack. This type is
frequently employed in the control surface areas of mixers to give convenient access to the patching. Very high-density jackfields can be assembled,
which has implications for the wiring arrangements on the back. Several
earlier examples of Bantam-type jackfields were unreliable and unsuited to
professional use. Later examples are rather better, and of course palladium
contacts should always be specified.
Electronically controlled ‘jackfields’ dispense with patch cords altogether. Such systems consist of a digitally controlled ‘stage box’ which carries a number of input and output sockets into which the mixer inputs and
outputs and any other tie-lines, processor and tape machine inputs and
outputs are plugged. The unit is controlled by a keypad, and a VDU displays the state of the patch. Information can also be entered identifying
each input and output by name according to the particular plug-up arrangement of the stage box. Any output can be routed to any input, and an output can be switched to drive any number of inputs as required. Various
patches can be stored in the system’s memory and recalled; rapid repatching is therefore possible, and this facility can be used in conjunction with
timecode to effect automatic repatches at certain chosen points on a tape
during mixdown for instance. MIDI control is also a possibility.
DISTRIBUTION AMPLIFIERS
A distribution amplifier is an amplifier used for distributing one input to a
number of outputs, with independent level control and isolation for each
output. It is used widely in broadcast centers and other locations where signals must be split off and routed to a number of independent locations.
This approach is preferable to simple parallel connections, since each output is unaffected by connections made to the others, preventing one from
loading down or interfering with the others.
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CHAPTER 13
Plug-ins and Outboard Equipment
CH A P T E R C O N TE N T S
Plug-ins
Outboard Equipment
The graphic equalizer
The compressor/limiter
Echo and reverb devices
Digital reverb
Multi-effects processors
Frequency shifter
Digital delay
Miscellaneous devices
Connection of Outboard Devices
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PLUG-INS
Plug-ins are now one of the fastest-moving areas of audio development, providing audio signal processing and effects that run either on a workstation’s CPU
or on dedicated DSP. (The hardware aspects of this were described in Chapter 9.)
Audio data can be routed from a sequencer or other audio application, via an
API (application programming interface) to another software module called a
‘plug-in’ that does something to the audio and then returns it to the source
application. In this sense it is rather like inserting an effect into an audio
signal path, but done in software rather than using physical patch cords and
rack-mounted effects units. Plug-ins can be written for the host processor in a
language such as C⫹, using the software development toolkits (SDK) provided
by the relevant parties. Plug-in processing introduces a delay that depends on
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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the amount of processing and the type of plug-in architecture used. Clearly
low latency architectures are highly desirable for most applications.
Many plug-ins are versions of previously external audio devices that
have been modeled in DSP, in order to bring favorite EQs or reverbs into
the workstation environment. The sound quality of these depends on the
quality of the software modeling that has been done. Some host-based
(native) plug-ins do not have as good a quality as dedicated DSP plug-ins as
they may have been ‘cut to fit’ the processing power available, but as hosts
become ever more powerful the quality of native plug-ins increases.
A number of proprietary architectures have been developed for plug-ins,
including Microsoft’s DirectX, Steinberg’s VST, Digidesign’s TDM, Mark of
the Unicorn’s MAS, TC Works’ PowerCore and EMagic’s host-based plug-in
format. Apple’s OS X Audio Units are a feature built into the OS that
manages plug-ins without the need for third-party middleware solutions.
It is usually necessary to specify for which system any software plug-in
is intended, as the architectures are not compatible. As OS-based plug-in
architectures for audio become more widely used, the need for proprietary
approaches may diminish.
Digidesign, for example, has had four different plug-in approaches that
are used variously in its products, as shown in Table 13.1.
DirectX is a suite of multimedia extensions developed by Microsoft
for the Windows platform. It includes an element called DirectShow that
deals with real-time streaming of media data, together with the insertion
of so-called ‘filters’ at different points. DirectX audio plug-ins work under
DirectShow and are compatible with a wide range of Windows-based audio
software. They operate at 32 bit resolution, using floating-point arithmetic
Table 13.1
Digidesign Plug-in Formats
Plug-in Architecture
Description
TDM
Uses dedicated DSP cards for signal
processing. Does not affect the host
CPU load and processing power can be
expanded as required
Uses the host processor for TDM plug-ins,
instead of dedicated DSP
Uses host processor for plug-ins. Not as
versatile as HTDM
Non-real-time processing that uses the host
CPU to perform operations such as timestretching that require the audio file to be
rewritten
HTDM (Host TDM)
RTAS (Real Time Audio Suite)
AudioSuite
Outboard Equipment
and can run in real time or can render audio files in non-real time. They do
not require dedicated signal processing hardware, running on the host CPU,
and the number of concurrent plug-ins depends on CPU power and available memory. DirectX plug-ins are also scalable – in other words they can
adapt to the processing resource available. They have the advantage of being
compatible with the very wide range of DirectX-compatible software in the
general computing marketplace. DXi, for example, is a software synthesizer
plug-in architecture developed by Cakewalk, running under DirectX.
One example of a proprietary approach used quite widely is VST,
Steinberg’s Virtual Studio Technology plug-in architecture. It runs on multiple platforms and works in a similar way to DirectX plug-ins. On Windows
machines it operates as a DLL (dynamic link library) resource, whereas on
Macs it runs as a raw Code resource. It can also run on BeOS and SGI systems, as a Library function. VST incorporates both virtual effects and virtual
instruments such as samplers and synthesizers. There is a cross-platform
GUI development tool that enables the appearance of the user interface to
be ported between platforms without the need to rewrite it each time.
Plug-ins cover all the usual traditional outboard functions including
graphic equalization, compressor/limiting, reverb and multi-effects processing, and a variety of ‘vintage’ examples mimic old guitar amplifiers and analog processors. Examples other than these more traditional types include
plug-ins which will essentially transform one instrument into another; and
others which are designed to create new and unfamiliar sounds. Fact file
13.1 gives some examples. Outboard equipment, the ‘hardware’ equivalent of the plug-in, which of course preceded it historically, is still much in
use in studios and particularly in the live sound environment. Sometimes
a ‘half-way house’ is encountered where a mixer will incorporate electronics from another manufacturer’s effects processor with front panel controls
similar to the original unit. This offers useful on-board processing from
elsewhere which will normally be a familiar and respected device.
OUTBOARD EQUIPMENT
The graphic equalizer
The graphic equalizer, pictured in Figure 13.1, consists of a row of faders (or
sometimes rotary controls), each of which can cut and boost a relatively narrow band of frequencies. Simple four- or five-band devices exist which are
aimed at the electronic music market, these being multiband tone controls.
They perform the useful function of expanding existing simple tone controls
on guitars and amplifiers, and several amplifiers actually incorporate them.
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FA C T F I L E 1 3 . 1 P LU G -IN E XA MPLES
Three examples of plug-in user interfaces are shown
below: a reverb device, a program for creating the sound
of another instrument from an existing one, and an
effects processor. The Applied Acoustic Systems Strum
Acoustics GS-1 acoustic guitar synth, taking its input signal from a keyboard, mimics a variety of nylon- and steelstrung acoustic guitars; a voicing module automatically
arranges notes of the chords played on the keyboard as a
guitar player would play them on a fret board. Strumming
and picking actions are also created, helping to give an
authentic guitar sound. The Platinum Enigma is an example of a plug-in which processes incoming signals using
flanging, phase, delay, filtering and modulation to create
a variety of sounds which at the extreme are not recognizably related to the original signal.
The quality of such plug-ins is now getting to the point
where it is on a par with the sound quality achievable on
external devices, depending primarily on the amount of
DSP available.
Outboard Equipment
The professional rack-mounting graphic
equalizer will have at least ten frequency
bands, spaced at octave or one-third-octave
intervals. The ISO (International Standards
Organization) center frequencies for octave
bands are 31 Hz, 63 Hz, 125 Hz, 250 Hz,
500 Hz, 1 kHz, 2 kHz, 4 kHz, 8 kHz, and
16 kHz. Each fader can cut or boost its
band by typically 12 dB or more. Figure
13.2 shows two possible types of filter
FIGURE 13.1 A typical two-channel graphic equalizer. (Courtesy of
Klark-Teknik Research Ltd.)
FIGURE 13.2 Two types of filter action shown with various degrees of boost and cut. (a) Typical
graphic equalizer with Q dependent upon degree of boost/cut. (b) Constant Q filter action.
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action. The 1 kHz fader is chosen, and three levels of cut and boost are illustrated. Maximum cut and boost of both types produces very similar Q (see
Fact File 5.6). A high Q result is obtained by both types when maximum cut
or boost is applied. The action of the first type is rather gentler when less cut
or boost is applied, and the Q varies according to the degree of deviation from
the fader’s central position.
Many graphic equalizers conform to this type of action, and it has the disadvantage that a relatively broad band of frequencies is affected when moderate
degrees of boost or cut are applied. The second type maintains a tight control
of frequency bandwidth throughout the cut and boost range, and such filters
are termed constant Q, the Q remaining virtually the same throughout the
fader’s travel. This is particularly important in the closely spaced one-thirdoctave graphic equalizer which has 30 separate bands, so that adjacent bands
do not interact with each other too much. The ISO center frequencies for 30
bands are 25 Hz, 31 Hz, 40 Hz, 50 Hz, 63 Hz, 80 Hz, 100 Hz, 125 Hz, 180 Hz,
200 Hz, 250 Hz, 315 Hz, 400 Hz, 500 Hz, 630 Hz, 800 Hz, 1 kHz, 1k25 Hz,
1k8 Hz, 2 kHz, 2k5 Hz, 3k15 Hz, 4 kHz, 5 kHz, 6k3 Hz, 8 kHz, 10 kHz,
12k5 Hz, 16 kHz, and 20 kHz. The value of using standard center frequencies is that complementary equipment such as spectrum analyzers which will
often be used in conjunction with graphic equalizers have their scales centered
on the same frequencies.
Even with tight constant Q filters, the conventional analog graphic
equalizer still suffers from adjacent filter interaction. If, say, 12 dB of boost
is applied to one frequency and 12 dB of cut applied to the next, the result
will be more like a 6 dB boost and cut, the response merging in between
to produce an ill-defined Q value. Such extreme settings are, however,
unlikely. The digital graphic equalizer applies cut and boost in the digital
domain, and extreme settings of adjacent bands can be successfully accomplished without interaction if required.
Some graphic equalizers are single channel, some are stereo. All will
have an overall level control, a bypass switch, and many also sport separate
steep-cut LF filters. A useful facility is an overload indicator – usually an
LED which flashes just before the signal is clipped – which indicates signal
clipping anywhere along the circuit path within the unit. Large degrees of
boost can sometimes provoke this. Some feature frequency cut only, these
being useful as notch filters for getting rid of feedback frequencies in PA/
microphone combinations. Some can be switched between cut/boost, or
cut only. It is quite possible that the graphic equalizer will be asked to drive
very long lines, if it is placed between mixer outputs and power amplifiers
for example, and so it must be capable of doing this. The ‘⫹20 dBu into
Outboard Equipment
600 ohms’ specification should be looked for as is the case with mixers. It
will be more usual though to patch the graphic equalizer into mixer output
inserts, so that the mixer’s output level meters display the effect on level the
graphic equalizer is having. Signal-to-noise ratio should be at least 100 dB.
The graphic equalizer can be used purely as a creative tool, providing
tone control to taste. It will frequently be used to provide overall frequency
balance correction for PA rigs. It has formerly been used to equalize control
room speakers, but poor results are frequently obtained due to the fact that
a spectrum analyzer’s microphone samples the complete room frequency
response whereas the perceived frequency balance is a complex combination of direct and reflected sound arriving at different times. The graphic
equalizer can also change the phase response of signals, and there has been
a trend away from their use in the control room for monitor EQ, adjustments being made to the control room acoustics instead.
The parametric equalizer was fully described in ‘Equalizer section’,
Chapter 5.
The compressor/limiter
The compressor/limiter (see Fact File 13.2) is used in applications such
as dynamics control and as a guard against signal clipping. Such a device
is pictured in Figure 13.3. The three main variable parameters are attack,
release and threshold. The attack time, in microseconds (μs) and milliseconds (ms), is the time taken for a limiter to react to a signal. A very fast
attack time of 10 μs can be used to avoid signal clipping, any high-level transients being rapidly brought under control. A fast release time will rapidly
restore the gain so that only very short-duration peaks will be truncated.
A ducking effect can be produced by using rapid attack plus a release of
around 200–300 ms. A threshold level is chosen which causes the limiting
to come in at a moderate signal level so that peaks are pushed down before
the gain is quickly reinstated. Such a ducking effect is ugly on speech, but is
useful for overhead cymbal mics for example.
A slow release time of several seconds, coupled with a moderate threshold, will compress the signal dynamics into a narrower window, allowing a higher mean signal level to be produced. Such a technique is often
used in vocals to obtain consistent vocal level from a singer. AM radio is
compressed in this way so as to squeeze wide dynamic range material into
this narrow dynamic range medium. It is also used on FM radio to a lesser
extent, although very bad examples of its application are frequently heard
on pop stations. An oppressive, raspy sound is the result, and in the pauses
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FA C T F I L E 1 3 . 2 C O MP R E S S IO N AND LI M I TI NG
A compressor is a device whose output level can be made
to change at a different rate to input level. For example,
a compressor with a ratio of 2:1 will give an output level
that changes by only half as much as the input level above
a certain threshold (see diagram). For example, if the
input level were to change by 6 dB the output level would
change by only 3 dB. Other compression ratios are available such as 3:1, 5:1, etc. At the higher ratios, the output
level changes only a very small amount with changes in
input level, which makes the device useful for reducing the
dynamic range of a signal. The threshold of a compressor
determines the signal level above which action occurs.
A limiter is a compressor with a very high compression ratio. A limiter is used to ensure that signal level does
not rise above a given threshold. A ‘soft’ limiter has an
action which comes in only gently above the threshold,
rather than acting as a brick wall, whereas a ‘hard’ limiter
has the effect almost of clipping anything which exceeds
the threshold.
Output level (dB)
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Uncompressed
(1:1)
2:1
Threshold
Limiting (>20:1)
Input level (dB)
in between items or speech one hears the
system gain creeping back up, causing
pumping noises. Background noise rapidly ducks back down when the presenter
again speaks. The effect is like listening to
FIGURE 13.3 A typical compressor/limiter. (Courtesy of Drawmer a tape with noise reduction encoding, the
reciprocal decoding being absent.
Distribution Ltd.)
Many units offer separate limiting and
compressing sections, their attack, release and threshold controls in each
section having values appropriate to the two applications. Some also include
gates (see ‘Noise gates’, Chapter 7) with variable threshold and ratio, rather
like an upside-down limiter, such that below a certain level the level drops
faster than would be expected. ‘Gain make-up’ is often available to compensate for the overall level-reducing effect of compression. Meters may indicate
the amount of level reduction occurring.
Echo and reverb devices
Before the advent of electronic reverb and echo processing, somewhat more
basic, ‘physical’ means were used to generate the effects. The echo chamber was literally that, a fairly large reverberant room being equipped with a
speaker and at least two spaced microphones. Signal was sent to the speaker
Outboard Equipment
FA C T F I L E 1 3 . 3 S IMU L AT IN G R EFLECTI ONS
from a certain large room and a smaller, more reflective
room. Early reflections, though, are dictated only by the
distance of the surfaces.
Direct sound
Early reflections
Level
Pre-delay in a reverb device is a means of delaying the
first reflection to simulate the effect of a large room with
distant surfaces. Early reflections may then be programmed to simulate the first few reflections from the
surfaces as the reverberant field builds up, followed by
the general decay of reverberant energy in the room as
random reflections lose their energy (see diagram).
Pre-delay and early reflections have an important
effect on one’s perception of the size of a room, and it is
these first few milliseconds which provide the brain with
one of its main clues as to room size. Reverberation time
(RT) alone is not a good guide to room size, since the RT
is affected both by room volume and absorption (see Fact
Files 1.5 and 1.6); thus the same RT could be obtained
Diffuse decay
Time
and the reverb generated in the room was picked up by the two microphones
which constituted the ‘stereo return’. The echo plate was a large thin resonant
plate of several meters in area suspended in a frame. A driving transducer
excited vibrations in the plate, and these were picked up by several transducers placed in various positions on its surface. Some quite high-quality reverb
effects were possible. The spring reverb, made popular by the Hammond
organ company many decades ago, consists literally of a series of coil springs
about the diameter of a pencil and of varying lengths (about 10–30 cm)
and tensions depending on the model. A driving transducer excites torsional
vibrations in one end of the springs, and these are picked up by transducers at the other end. Quite a pleasing effect can be obtained, and it is still
popular for guitar amplifiers. The tape echo consisted of a short tape loop
together with a record head followed by several replay heads spaced a few
centimeters apart, then lastly an erase head. The output levels from the
replay heads could be adjusted as could the speed of the tape, generating
a variety of repeat-echo effects. Control of the erase head could generate a
huge build-up of multi-echoes.
When one hears real reverb, one hears ‘pre-delay ’ in effects processing parlance: a sound from the source travels to the room boundaries and
then back to the listener, so there is a delay of several tens of milliseconds
between hearing the direct sound and hearing the reverberation. This plays
a large part in generating realistic reverb effects, and Fact File 13.3 explains
the requirements in more detail.
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Digital reverb
The present-day digital reverb, such as that pictured in Figure 13.4,
can be quite a sophisticated device. Research into path lengths,
boundary and atmospheric absorption, and the physical volume and
dimensions of real halls, have been taken into account when algorithms have been designed. Typical front panel controls will include
selection of internal pre-programmed effects, labeled as ‘large hall’,
‘medium hall’, ‘cathedral’, ‘church’, etc., and parameters such as
degree of pre-delay, decay time, frequency balance of delay, dry-towet ratio (how much direct untreated sound appears with the effect
signal on the output), stereo width, and relative phase between the
stereo outputs can often be additionally altered by the user. A small
display gives information about the various parameters.
FIGURE 13.4 The Lexicon 960-L
Memory stores generally contain a volatile and a non-volatile
Multichannel effects system. (Courtesy of
section. The non-volatile section contains factory preset effects,
Stirling/Pure distribution.)
and although the parameters can be varied to taste the alterations
cannot be stored in that memory. Settings can be stored in the volatile section, and it is usual to adjust an internal preset to taste and then transfer
and store this in the volatile section. For example, a unit may contain 100
presets. The first 50 are non-volatile, and cannot be permanently altered.
The last 50 can store settings arrived at by the user by transferring existing
settings to this section. The method of doing this varies between models,
but is usually a simple two- or three-button procedure, for example by pressing 5, 7 and ‘store’. This means that a setting in the non-volatile memory
which has been adjusted to taste will be stored in memory 57. Additional
adjustments can then be made later if required.
Several units provide a lock facility so that stored effects can be made
safe against accidental overwriting. An internal battery backup protects the
memory contents when the unit is switched off. Various unique settings
can be stored in the memories, although it is surprising how a particular
model will frequently stamp its own character on the sound however it is
altered. This can be both good or bad of course, and operators may have
a preference for a particular system. Certain processors sound a bit like
clangy spring reverbs whatever the settings. Some sound dull due to limited
bandwidth, the frequency response extending only up to around 12 kHz or
so. This must be carefully looked for in the specification. Sometimes the
bandwidth reduces with increasing reverberation or echo decay times. Such
a shortcoming is sometimes hard to find in a unit’s manual.
In some of the above devices it should be noted that the input is normally mono and the output stereo. In this way ‘stereo space’ can be added
to a mono signal, there being a degree of decorrelation between the outputs.
Outboard Equipment
A reverb device may have stereo inputs, so that the source
can be assumed to be other than a point.
Multi-effects processors
Digital multi-effects processors such as that shown in Figure
13.5 can offer a great variety of features. Parametric equalization is available, offering variations in Q, frequency and degree
of cut and boost. Short memory capacity can store a sample,
the unit being able to process this and reproduce it according to the incoming signal’s command. MIDI interfacing (see
Chapter 14) has become popular for the selection of effects
FIGURE 13.5 The tc Electronics System
under remote control, as has the RS 232 computer interface,
6000 multi-effects processor. (Courtesy of tc
and a USB port or RAM card slot is sometimes encountered Electronics.)
for loading and storing information. Some now have Firewire
or USB ports for digital audio streaming, or an alternative realtime digital interface (see Chapter 10). Repeat echo, autopan, phase, modulation, flange, high and low filters, straight signal delay, pitch change, gating,
and added harmony may all be available in the presets, various multifunction
nudge buttons being provided for overall control. Many units are only capable
of offering one type of effect at a time. Several have software update options so
that a basic unit can be purchased and updates later incorporated internally to
provide, say, longer delay times, higher sample storage capacity, and new types
of effect as funds allow and as the manufacturer develops them. This helps to
keep obsolescence at bay in an area which is always rapidly changing.
Frequency shifter
The frequency shifter shifts an incoming signal by a few hertz. It is used for
acoustic feedback control in PA work, and operates as follows. Feedback is
caused by sound from a speaker re-entering a microphone to be reamplified
and reproduced again by the speaker, forming a positive feedback loop which
builds up to a continuous loud howling noise at a particular frequency.
The frequency shifter is placed in the signal path such that the frequencies
reproduced by the speakers are displaced by several hertz compared with the
sound entering the microphone, preventing additive effects when the sound
is recycled, so the positive feedback loop is broken. The very small frequency
shift has minimal effect on the perceived pitch of the primary sound.
Digital delay
The digital delay line is incorporated into any substantial sound reinforcement installation as a matter of course. Consider a typical SR setup which
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may consist of main speakers each side of the stage (or against the proscenium arch or ‘prosc’ in a theater); front-fill speakers across the front of
the stage; additional speakers covering extreme right- and left-hand sides
under galleries; a line of speakers under a gallery covering the rear of the
stalls; and a flown cluster covering an upper balcony, these latter speakers
being rigged somewhat forward of the stage. Arrival times of the sounds
from the various speakers to a particular location in the auditorium will of
course vary due to the different physical path lengths, and the sound can
be quite blurred as a result. Comb-filtering effects – abrupt attenuation of
sound at certain frequencies as sound from one speaker is canceled by antiphase sound coming from another due to the different path lengths – can
also be encountered at some locations, and a slow walk from one side of an
auditorium to the other whilst listening to pink noise will quite often make
these problems apparent.
Digital delay lines are patched in between the mixer’s outputs and the
power amplifiers to alleviate the problem (digital mixers often incorporate
delays on their outputs for this sort of application) and are set up as follows.
First, the sound of sharp clicks, rather like those of a large ticking clock –
about one tick per second – is played through the main speakers each side
of the stage with the other speakers switched off. Whilst standing close to
the front of the stage, the front-fill speakers can then be added in. The clicks
from these will reach the listener sooner than those from the main speakers, and the clicks will sound blurred, or perhaps even a double click will
be heard each time. Adding a delay of perhaps 10 ms or so to the front-fills
will bring the sound back into sharp focus, the exact value of delay needed
depending upon the distance between the listener and the two sets of speakers. The front-fills are then switched off, and the listener moves to the side,
under the balcony, and the speakers covering these areas are switched on.
Again, indistinct clicks will be heard, and a delay to the side speakers of perhaps 25 ms will be required here to bring the sound into focus. For the line
of speakers covering the rear stalls, a delay of perhaps 50 ms will be found to
be needed. A rule of thumb when setting initial values is that sound travels
just over a foot per millisecond, and so if, say, the line of rear stalls speakers is about 50 feet in front of the main stage speakers, an initial setting of
50–55 ms can be set. Many delay devices will also display the delay in terms
of feet or meters, which is very useful. Moving up to the balcony, the flown
cluster can now be switched on. A delay of perhaps 120 ms will be needed
here to time-align the flown cluster with the main speakers each side of the
stage. As well as giving a much cleaner sound, the use of delays in this manner also has the effect of making the speakers forward of the stage ‘disappear ’, and the sound appears to come just from the stage itself.
Outboard Equipment
As air temperature rises, sound travels faster, and delay settings obtained
in a fairly cold auditorium during the day may well be a little high for the
evening concert when the air is somewhat warmer. Some digital delay
devices have an input for a temperature sensor, and if this is used the delay
settings will automatically adjust to compensate for temperature changes.
Several computer-based systems are available which in conjunction with
measuring microphones placed in the auditorium will display required delay
settings for the various speaker locations when special test tones are played
through the system. Additionally, using pink noise to drive the speakers the
EQ curve requirements for flat frequency responses can be displayed, and
parametric equalization can be used to mirror the display curves. A word
of caution concerning EQ settings – air absorbs high frequencies to a rather
greater extent than low frequencies, and sounds coming from a distance
naturally sound duller. At a distance of 50 meters, assuming a 20°C temperature and 20% relative humidity, there is about a 10 dB loss at 8 kHz,
and about a 35 dB loss at 16 kHz. Measuring microphones placed some
distance from flown speakers will therefore register treble loss as a natural
consequence of air absorption, and one must add treble boost with caution.
An unnaturally bright sound can result, and excessive power can be fed to
high-frequency horns.
Miscellaneous devices
The Aphex company of America introduced a unit called the Aural Exciter
in the 1970s, and for a time the mechanisms by which it achieved its effect
were shrouded in a certain amount of mystery. The unit made a signal
‘sparkle’, enhancing its overall presence and life, and it was usually applied
to individual sounds in a mix such as solo instruments and voices, but
sometimes also to the complete stereo signal. Such devices succeed entirely
by their subjective effect, and several companies later produced similar units. They achieve their psycho-acoustic effect by techniques such as
comb filtering, selective boosting of certain frequencies, and by introducing
relatively narrow-band phase shifts between stereo channels.
Effects such as these go back a long way, and in many cases will be long
obsolete and unavailable. But the value of such things as old valve (tube)
compressors and limiters, distortion devices, tape and plate echoes, room
simulators and other vintage-sounding devices such as certain guitar amplifiers is reflected in the market for ‘plug-ins’ for computer-based digital workstations: computer programs which have been developed to simulate the
sounds of such old and well-loved devices. These things succeed purely on
their own subjective terms, and are very much part of the creative process.
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Outboard
processor
From
channel
input
Send
Ret.
Pre-fade
insertion
point
To
channel
output
Fader
Plug-ins do of course also provide up-to-the-minute effects and processing.
The de-esser cleans up closely miced vocals.
Sibilant sounds can produce a rasping quality,
and the de-esser dynamically filters the high-level,
high-frequency component of the sound to produce
a more natural vocal quality.
FIGURE 13.6 Outboard processors such as compressors
are normally patched in at an insertion point of the required
mixer channel.
Outboard
processor
To echo
return input
CONNECTION OF OUTBOARD
DEVICES
A distinction needs to be made between processors which need to interrupt a signal for
Aux send
treatment, and those which basically add
(via level control)
something to an existing signal. Graphic and
From
To
parametric equalizers, compressors, de-essers
channel
channel
and gates need to be placed in the signal
input
output
path. One would not normally wish to mix,
Fader
say, an uncompressed signal with its comFIGURE 13.7 Reverberation and echo devices are usually fed
pressed version, or an ungated with the gated
from a post-fader auxiliary send, and brought back to a dedicated echo
sound. Such processors will generally be
return or another channel input.
patched in via the mixer’s channel insertion
send and returns (see Figure 13.6), or patched in ahead of the incoming signal or immediately after an output. Devices such as echo, reverb, chorus,
flange are generally used to add something to an existing signal and usually
a channel aux send will be used to drive them. Their outputs will be brought
back to additional input channels and these signals mixed to taste with the
existing dry signal (see Figure 13.7).
Sometimes just the effects signal will be required, in which case either
the aux send will be switched to pre-fade and that channel’s fader closed, or
the channel will simply be de-routed from the outputs. The channel is then
used merely to send the signal to the effects unit via the aux. The returns
will often contain a degree of dry signal anyway, the ratio of dry to effect
being adjusted on the processor.
MIDI control for selecting a program has already been mentioned.
Additionally, MIDI can be used in a musical way with some devices. For
instance, a ‘harmonizer ’ device, designed to add harmony to a vocal or
instrumental line, is normally set to add appropriate diatonic harmonies to
the incoming line in the appropriate key with the desired number of voices
above and/or below it. Results are thereafter in the hands of the machine.
Recommended Further Reading
Alternatively, a MIDI keyboard can be used to control the device so that
the harmonizer adds the notes which are being held down on the keyboard.
Composition of the harmonies and voice lines is then under the control of
the musician. This can be used in recording for adding harmonies to an
existing line, or in a live situation where a keyboard player plays along with
a soloist to generate the required harmonies.
RECOMMENDED FURTHER READING
White, P., 2003. Basic Effects and Processors. New Amsterdam Books.
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CHAPTER 14
MIDI and Synthetic Audio Control
CH A P T E R C O N TE N T S
Background
What is MIDI?
MIDI and Digital Audio Contrasted
Basic Principles
The interface
Simple interconnection
Interfacing a Computer to a MIDI System
Adding MIDI ports
Drivers and audio I/O software
How MIDI Control Works
MIDI channels
Channel and system messages contrasted
Note on and note off messages
Velocity information
Running status
Polyphonic key pressure (aftertouch)
Control change
Channel modes
Program change
Channel aftertouch
Pitch bend wheel
System exclusive
Universal system exclusive messages
Tune request
Active sensing
Reset
Sound and Recording
Copyright © 2009 Elsevier Ltd. All rights reserved.
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C H A P TE R C O N TENTS
MIDI Control of Sound Generators
MIDI note assignment in synthesizers and samplers
Polyphony, voice and note assignment
MIDI functions of sound generators
MIDI data buffers and latency
Handling of velocity and aftertouch data
Handling of controller messages
Voice selection
General MIDI
Scalable Polyphonic MIDI (SPMIDI)
RMID and XMF files
SAOL and SASL in MPEG 4 structured audio
MIDI over USB
MIDI over IEEE 1394
After MIDI?
Sequencing Software
Introduction
Tracks, channels, instruments and environments
Input and output filters
Timing resolution
Displaying, manipulating and editing information
Quantization of rhythm
Automation and non-note MIDI events
MIDI mixing and external control
Synchronization
Synchronized digital video
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MIDI is the Music Instrument Digital Interface, a control protocol and interface standard for electronic musical instruments which has also been used
widely in other music and audio products. Although it is relatively dated by
modern standards it is still used extensively, which is a testament to its simplicity and success. Even if the MIDI hardware interface is used less these
days, either because more synthesis, sampling and processing takes place
using software within the workstation, or because other data interfaces such
as USB, Firewire and Ethernet are becoming popular, the original protocol for
communicating events and other control information is still widely encountered. A lot of software that runs on computers uses MIDI as a basis for controlling the generation of sounds and external devices.
Background
399
Synthetic audio is used increasingly in audio workstations and mobile
devices as a very efficient means of audio representation, because it only
requires control information and sound object descriptions to be transmitted. Standards such as MPEG-4 Structured Audio enable synthetic audio
to be used as an alternative or an addition to natural audio coding and this
can be seen as a natural evolution of the MIDI concept in interactive multimedia applications.
BACKGROUND
Electronic musical instruments existed widely before MIDI was developed
in the early 1980s, but no universal means existed of controlling them
remotely. Many older musical instruments used analog voltage control,
rather than being controlled by a microprocessor, and thus used a variety of
analog remote interfaces (if indeed any facility of this kind was provided at
all). Such interfaces commonly took the form of one port for timing information, such as might be required by a sequencer or drum machine, and
another for pitch and key triggering information, as shown in Figure 14.1.
The latter, commonly referred to as ‘CV and gate’, consisted of a DC (direct
current) control line carrying a variable control voltage (CV) which was proportional to the pitch of the note, and a separate line to carry a trigger pulse.
A common increment for the CV was 1 volt per octave (although this was
by no means the only approach) and notes on a synthesizer could be triggered remotely by setting the CV to the correct pitch and sending a ‘note on’
trigger pulse which would initiate a new cycle of the synthesizer’s envelope
generator. Such an interface would deal with only
one note at a time, but many older synths
Pitch
Tempo
CV
were only monophonic in any case (that is, they
Start
were only capable of generating a single voice).
Note trigger
Stop
Instruments with onboard sequencers would
Continue
need a timing reference in order that they could
be run in synchronization with other such
devices, and this commonly took the form of a
square pulse train at a rate related to the current
musical tempo, often connected to the device
using a DIN-type connector, along with trigElectronic musical instrument
ger lines for starting and stopping a sequence’s
FIGURE 14.1 Prior to MIDI control, electronic musical
execution. There was no universal agreement
instruments tended to use a DC remote interface for pitch and note
over the rate of this external clock, and frequentriggering. A second interface handled a clock signal to control
cies measured in pulses per musical quarter note
tempo and trigger pulses to control the execution of a stored
sequence.
(ppqn), such as 24 ppqn and 48 ppqn, were used
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by different manufacturers. A number of conversion boxes were available
which divided or multiplied clock signals in order that devices from different manufacturers could be made to work together.
As microprocessor control began to be more widely used in musical
instruments a number of incompatible digital control interfaces sprang up,
promoted by the large synthesizer manufacturers, some serial and some
parallel. Needless to say the plethora of non-standardized approaches to
remote control made it difficult to construct an integrated system, especially when integrating equipment from different manufacturers. Owing
to collaboration between the major parties in America and Japan, the way
became clear for agreement over a common hardware interface and command protocol, resulting in the specification of the MIDI standard in late
1982/early 1983. This interface grew out of an amalgamation of a proposed
universal interface called USI (the Universal Synthesizer Interface) which
was intended mainly for note on and off commands, and a Japanese specification which was rather more complex and which proposed an extensive
protocol to cover other operations as well. Since MIDI’s introduction, the
use of older remote interfaces has died away very quickly, but there remain
available a number of specialized interfaces which may be used to interconnect non-MIDI equipment to MIDI systems by converting the digital MIDI
commands into the type of analog information described above.
The standard has been subject to a number of addenda, extending the
functionality of MIDI far beyond the original. The original specification was
called the MIDI 1.0 specification, to which has been added such addenda as
the MIDI Sample Dump protocol, MIDI Files, General MIDI (1 and 2), MIDI
TimeCode, MIDI Show Control, MIDI Machine Control and Downloadable
Sounds. A new ‘HD ’ (High Definition) version of the standard is planned for
release in 2009, which is expected to include support for more channels and
controllers, as well as greater controller resolution using single messages. It
is aimed to make this compatible with existing hardware and software. The
MIDI Manufacturer’s Association (MMA) is now the primary association
governing formal extensions to the standard, liaising closely with a Japanese
association called AMEI (Association of Musical Electronics Industry).
WHAT IS MIDI?
MIDI is a digital remote control interface for music systems, but has come
to relate to a wide range of standards and specifications to ensure interoperability between electronic music systems. MIDI-controlled equipment is
normally based on microprocessor control, with the MIDI interface forming an I/O port. It is a measure of the popularity of MIDI as a means of
MIDI and Digital Audio Contrasted
control that it has now been adopted in many other audio and visual systems, including the automation of mixing consoles, the control of studio
outboard equipment, lighting equipment and other machinery. Although
many of its standard commands are music related, it is possible either
to adapt music commands to non-musical purposes or to use command
sequences designed especially for alternative methods of control.
The adoption of a serial standard for MIDI was dictated largely by economic and practical considerations, as it was intended that it should be
possible for the interface to be installed on relatively cheap items of equipment and that it should be available to as wide a range of users as possible.
A parallel system might have been more professionally satisfactory, but
would have involved a considerable manufacturing cost overhead per MIDI
device, as well as parallel cabling between devices, which would have been
more expensive and bulky than serial interconnection. The simplicity and
ease of installation of MIDI systems was largely responsible for its rapid
proliferation as an international standard.
Unlike its analog predecessors, MIDI integrates timing and system control commands with pitch and note triggering commands, such that everything may be carried in the same format over the same piece of wire. MIDI
makes it possible to control musical instruments polyphonically in pseudo
real time: that is, the speed of transmission is such that delays in the transfer of performance commands are not audible in the majority of cases. It
is also possible to address a number of separate receiving devices within a
single MIDI data stream, and this allows a controlling device to determine
the destination of a command.
MIDI AND DIGITAL AUDIO CONTRASTED
For many the distinction between MIDI and digital audio may be a clear
one, but those new to the subject often confuse the two. Any confusion is
often due to both MIDI and digital audio equipment appearing to perform
the same task – that is the recording of multiple channels of music using
digital equipment – and is not helped by the way in which some manufacturers refer to MIDI sequencing as digital recording.
Digital audio involves a process whereby an audio waveform (such as
the line output of a musical instrument) is sampled regularly and then converted into a series of binary words that represent the sound waveform, as
described in Chapter 8. A digital audio recorder stores this sequence of data
and can replay it by passing the original data through a digital-to-analog
convertor that turns the data back into a sound waveform, as shown in
Figure 14.2. A multitrack recorder has a number of independent channels
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FIGURE 14.2
(a) Digital audio recording
and (b) MIDI recording
contrasted. In (a) the sound
waveform itself is converted
into digital data and stored,
whereas in (b) only control
information is stored, and
a MIDI-controlled sound
generator is required during
replay.
(a)
Sound source
Analog
waveform
Data
Line out
A/D
convertor
Data
D/A
convertor
Store
(b)
Control data
Control data
MIDI out
MIDI in
Line
out
Store
Line
out
that work in the same way, allowing a sound recording to be built up in
layers. MIDI, on the other hand, handles digital information that controls
the generation of sound. MIDI data does not represent the sound waveform
itself. When a multitrack music recording is made using a MIDI sequencer
(described later) this control data is stored, and can be replayed by transmitting the original data to a collection of MIDI-controlled musical instruments. It is the instruments that actually reproduce the recording.
A digital audio recording, then, allows any sound to be stored and
replayed without the need for additional hardware. It is useful for recording
acoustic sounds such as voices, where MIDI is not a great deal of help here.
A MIDI recording is almost useless without a collection of sound generators.
An interesting advantage of the MIDI recording is that, since the stored data
represents event information describing a piece of music, it is possible to
change the music by changing the event data. MIDI recordings also consume
a lot less memory space than digital audio recordings. It is also possible to
transmit a MIDI recording to a different collection of instruments from those
used during the original recording, thus resulting in a different sound. It is
now common for MIDI and digital audio recording to be integrated in one
software package, allowing the two to be edited and manipulated in parallel.
BASIC PRINCIPLES
The interface
The MIDI standard specifies a unidirectional serial interface (see Fact File
8.6, Chapter 8) running at 31.25 kbit/s ⫾1%. The rate was defined at a
time when the clock speeds of microprocessors were typically much slower
Basic Principles
Idle state
320 μs
Data word
LSB
MSB
1
0
Start bit
One clock period
Stop bit
than they are today, this rate being a convenient division of the typical 1
or 2 MHz master clock rate. The rate had to be slow enough to be carried
without excessive losses over simple cables and interface hardware, but fast
enough to allow musical information to be transferred from one instrument to another without noticeable delays. Control messages are sent as
groups of bytes. Each byte is preceded by one start bit and followed by one
stop bit per byte in order to synchronize reception of the data which is
transmitted asynchronously, as shown in Figure 14.3. The addition of start
and stop bits means that each 8 bit word actually takes ten bit periods to
transmit (lasting a total of 320 μs). Standard MIDI messages typically consist of one, two or three bytes, although there are longer messages for some
purposes.
The hardware interface is shown in Fact File 14.1. In the MIDI specification, the opto-isolator is defined as having a rise time of no more
than 2 μs. The rise time affects the speed with which the device reacts to
a change in its input and if slow will tend to distort the leading edge of
data bit cells. The same also applies in practice to fall times. Rise-time distortion results in timing instability of the data, since it alters the time at
which a data edge crosses the decision point between one and zero. If the
rise time is excessively slow the data value may be corrupted since the output of the device will not have risen to its full value before the next data bit
arrives. If a large number of MIDI devices are wired in series (that is from
THRU to IN a number of times) the data will be forced to pass through
a number of opto-isolators and thus will suffer the combined effects of a
number of stages of rise-time distortion. Whether or not this will be sufficient to result in data detection errors at the final receiver will depend
to some extent on the quality of the opto-isolators concerned, as well as
on other losses that the signal may have suffered on its travels. It follows
that the better the specification of the opto-isolator, the more stages of
FIGURE 14.3
A MIDI message consists
of a number of bytes, each
transmitted serially and
asynchronously by a UART
in this format, with a start
and stop bit to synchronize
the receiving UART. The
total period of a MIDI data
byte, including start and
stop bits, is 320 μs.
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FA C T F I L E 1 4 . 1 MID I H A R D WA RE I NTERFACE
Most equipment using MIDI has three interface connectors: IN, OUT, and THRU. The OUT connector carries
data that the device itself has generated. The IN connector receives data from other devices and the THRU connector is a direct throughput of the data that is present
at the IN. As can be seen from the hardware interface
diagram, it is simply a buffered feed of the input data,
and it has not been processed in any way. A few cheaper
devices do not have THRU connectors, but it is possible
to obtain ‘MIDI THRU boxes’ which provide a number
of ‘THRUs’ from one input. Occasionally, devices without a THRU socket allow the OUT socket to be switched
between OUT and THRU functions.
The interface incorporates an opto-isolator between the
MIDI IN (that is the receiving socket) and the device’s
microprocessor system. This is to ensure that there is
no direct electrical link between devices and helps to
reduce the effects of any problems which might occur if
one instrument in a system were to develop an electrical fault. An opto-isolator is an encapsulated device in
which a light-emitting diode (LED) can be turned on or off
depending on the voltage applied across its terminals, illuminating a photo-transistor which consequently conducts
or not, depending on the state of the LED. Thus the data
is transferred optically, rather than electrically.
+5V
To UART
From UART
Buffers
Buffers
Opto-isolator
220
220
+5V
+5V
220
220
220
5 2 4
5 2 4
5 2 4
IN
THRU
OUT
device cascading will be possible before unacceptable distortion is introduced. The delay in data passed between IN and THRU is only a matter
of microseconds, so this contributes little to any audible delays perceived
in the musical outputs of some instruments in a large system. The bulk of
any perceived delay will be due to other factors like processing delay, buffer
delays and traffic.
Basic Principles
FA C T F I L E 1 4 . 2 MID I C O N N E C TORS AND CABLES
The connectors used for MIDI interfaces are 5-pin DIN
types. The specification also allows for the use of XLRtype connectors (such as those used for balanced audio
signals in professional equipment), but these are rarely
encountered in practice. Only three of the pins of a
5-pin DIN plug are actually used in most equipment (the
three innermost pins). In the cable, pin 5 at one end
should be connected to pin 5 at the other, and likewise
pin 4 to pin 4, and pin 2 to pin 2. Unless any hi-fi DIN
cables to be used follow this convention they will not
work. Professional microphone cable terminated in DIN
connectors may be used as a higher-quality solution,
because domestic cables will not always be a shielded
twisted pair and thus are more susceptible to external
interference, as well as radiating more themselves which
could interfere with adjacent audio signals. A 5 mA current loop is created between a MIDI OUT or THRU and
a MIDI IN, when connected with the appropriate cable,
and data bits are signaled by the turning on and off of
this current by the sending device. This principle is
shown in the diagram.
The cable should be a shielded twisted pair with the
shield connected to pin 2 of the connector at both ends,
although within the receiver itself, as can be seen from the
diagram above, the MIDI IN does not have pin 2 connected
to earth. This is to avoid earth loops and makes it possible to
use a cable either way round. If two devices are connected
together whose earths are at slightly different potentials, a
current is caused to flow down any earth wire connecting
them. This can induce interference into the data wires, possibly corrupting the data, and can also result in interference
such as hum on audio circuits. It is recommended that no
more than 15 m of cable is used for a single cable run in a
simple MIDI system and investigation of typical cables indicates that corruption of data does indeed ensue after longer
distances, although this is gradual and depends on the electromagnetic interference conditions, the quality of cable and
the equipment in use. Longer distances may be accommodated with the use of buffer or ‘booster’ boxes that compensate for some of the cable losses and retransmit the data.
It is also possible to extend a MIDI system by using a data
network with an appropriate interface.
From UART
+5V
4
OUT
or
THRU
5
5
IN
MIDI cable
The specification of cables and connectors is described in Fact File 14.2.
This form of hardware interface is increasingly referred to as ‘MIDI-DIN’
to distinguish it from other means of transferring MIDI data.
Implementations of MIDI that work over other hardware interfaces
such as Ethernet (using Internet Protocol/UDP), USB and Firewire (IEEE
4
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CHAPTER 14: MIDI and Synthetic Audio Control
1394) have also been introduced, sometimes in proprietary form. The latter two are described briefly later in the chapter.
Simple interconnection
In the simplest MIDI system, one instrument could be connected to
another as shown in Figure 14.4. Here, instrument 1 sends information
relating to actions performed on its own controls (notes pressed, pedals
pressed, etc.) to instrument 2, which imitates these actions as far as it is
able. This type of arrangement can be used for ‘doubling-up’ sounds, ‘layering’ or ‘stacking’, such that a composite sound can be made up from two
synthesizers’ outputs. (The audio outputs of the two instruments would
have to be mixed together for this effect to be heard.) Larger MIDI systems could be built up by further ‘daisy-chaining’ of instruments, such that
instruments further down the chain all received information generated by
the first (see Figure 14.5), although this is not a very satisfactory way of
building a large MIDI system. In large systems some form of central routing helps to avoid MIDI ‘traffic jams’ and simplifies interconnection.
INTERFACING A COMPUTER TO A MIDI SYSTEM
Adding MIDI ports
In order to use a workstation as a central controller for a MIDI system it
must have at least one MIDI interface, consisting of at least an IN and an
MIDI cable
MIDI OUT
MIDI IN
Instrument 1
Instrument 2
FIGURE 14.4 The simplest form of MIDI interconnection involves connecting two instruments
together as shown.
OUT
Instrument 1
IN
THRU
Instrument 2
IN
THRU
Instrument 3
IN
Instrument 4
FIGURE 14.5 Further instruments can be added using THRU ports as shown, in order that
messages from instrument 1 may be transmitted to all the other instruments.
Interfacing a Computer to a MIDI System
OUT port. (THRU is not strictly necessary in most cases.) Unless the computer has a built-in interface, as found on the old Atari machines, some
form of third-party hardware interface must be added and there are many
ranging from simple single ports to complex multiple port products.
A typical single port MIDI interface can be connected either to one of
the spare I/O ports of the computer (a serial or USB port, for example), or
can be installed as an expansion slot card (perhaps as part of an integrated
sound card). Depending on which port it is connected to, some processing may be required within the MIDI interface to convert the MIDI data
stream to and from the relevant interface protocol. Older PCs had serial
interfaces that would operate at a high enough data rate for MIDI, but were
not normally able to operate at precisely the 31.25 kbaud required. External
interfaces were able to transpose the data stream from a higher serial data
rate (often 38.4 kbaud) down to the MIDI rate using intermediate buffering
and flow control. Some PCs and soundcards also had the so-called ‘MIDI/
Joystick port’ that conformed to the old Roland MPU-401 interface standard. Adaptor cables were available that provided MIDI IN and OUT connectors from this port. Some older PC interfaces also attach to the parallel
port. The majority of recent MIDI interfaces are connected either to USB
or Firewire ports of host workstations.
Multiport interfaces have become widely used in MIDI systems where
more than 16 MIDI channels are required, and they are also useful as
a means of limiting the amount of data sent or received through any
one MIDI port. (A single port can become ‘overloaded’ with MIDI data
if serving a large number of devices, resulting in data delays.) Multiport
interfaces are normally more than just a parallel distribution of a single
MIDI data stream, typically handling a number of independent MIDI
data streams that can be separately addressed by the operating system
drivers or sequencer software. USB and Firewire MIDI protocols allow a
particular stream or ‘cable’ to be identified so that each stream controlling 16 MIDI channels can be routed to a particular physical port or
instrument.
EMagic’s Unitor8 interface is pictured in Figure 14.6. It has RS-232 and
-422 serial ports as well as a USB port to link with the host workstation.
There are eight MIDI ports with two on the front panel for easy connection of ‘guest’ devices or controllers that are not installed at the back. This
device also has VITC and LTC timecode ports in order that synchronization information can be relayed to and from the computer. A multi-device
MIDI system is pictured in Figure 14.7, showing a number of multi-timbral
sound generators connected to separate MIDI ports and a timecode
connection to an external video tape recorder for use in synchronized
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CHAPTER 14: MIDI and Synthetic Audio Control
FIGURE 14.6
(a) Front and (b) back
panels of the Emagic Unitor
8 interface, showing USB
port, RS-422 port, RS-232
port, LTC and VITC ports
and multiple MIDI ports.
(a)
(b)
Multiport MIDI
interface
FIGURE 14.7
A typical multi-machine
MIDI system interfaced to
a computer via a multiport
interface connected by a
high-speed link (e.g. USB).
High speed I/O
Multi-timbral
expander
Multi-timbral
expander
Controller
Multi-timbral
expander
Timecode
Video or audio recorder
Drum
machine
Sampler
Master keyboard
Effects 1
Effects 2
Automated mixer
Audio
routing
matrix
Interfacing a Computer to a MIDI System
post-production. As more of these functions are now being provided within
the workstation (e.g. synthesis, video, mixing) the number of devices connected in this way will reduce.
Drivers and audio I/O software
Most audio and MIDI hardware requires ‘driver ’ software of some sort
to enable the operating system (OS) to ‘see’ the hardware and use it correctly. There are also sound manager or multimedia extensions that form
part of the operating system of the workstation in question, designed to
route audio to and from hardware in the absence of dedicated solutions.
The standard multimedia extensions of the OS that basic audio software
used in older systems to communicate with sound cards could result in
high latency and might also be limited to only two channels and 48 kHz
sampling frequency. Dedicated low latency approaches were therefore developed as an alternative, allowing higher sampling frequencies, full audio resolution, sample-accurate synchronization and multiple channels. Examples
of these are Steinberg’s ASIO (Audio Stream Input Output) and E-Magic’s
EASI. These are software extensions behaving as ‘hardware abstraction
layers’ (HALs) that replace the OS standard sound manager and enable
applications to communicate more effectively with I/O hardware. ASIO,
for example, handles a range of sampling frequencies and bit depths, as
well as multiple channel I/O, and many sound cards and applications are
ASIO-compatible.
As high-quality audio begins to feature more prominently in general
purpose desktop computers, audio architectures and OS audio provision
improve to keep step. OS native audio provision may now take the place
of what third-party extensions have provided in the past. For example,
Apple’s OS X Core Audio standard is designed to provide a low latency
HAL between applications and audio hardware, enabling multichannel audio data to be communicated to and from sound cards and external interfaces such as USB and Firewire. Core Audio handles audio in 32
bit floating-point form for high-resolution signal processing, as well as
enabling sample accurate timing information to be communicated alongside audio data. Microsoft has also done something similar for Windows
systems, with the Windows Driver Model (WDM) audio drivers that also
include options for multichannel audio, high resolutions and sampling
frequencies. DirectSound is the Microsoft equivalent of Apple’s OS X
Core Audio.
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Core MIDI and DirectMusic do a similar thing for MIDI data. Whereas
previously it would have been necessary to install a third-party MIDI
HAL such as OMS (Opcode’s Open Music System) or MIDI Manager to
route MIDI data to and from multiport interfaces and applications, these
features are now included within the operating system’s multimedia
extensions.
HOW MIDI CONTROL WORKS
MIDI channels
MIDI messages are made up of a number of bytes as explained in Fact File
14.3. Each part of the message has a specific purpose, and one of these is
to define the receiving channel to which the message refers. In this way,
a controlling device can make data device specific – in other words it can
define which receiving instrument will act on the data sent. This is most
important in large systems that use a computer sequencer as a master controller, when a large amount of information will be present on the MIDI
data bus, not all of which is intended for every instrument. If a device is set
in software to receive on a specific channel or on a number of channels it
will act only on information which is ‘tagged’ with its own channel numbers. Everything else it will usually ignore. There are 16 basic MIDI channels and instruments can usually be set to receive on any specific channel
or channels (omni off mode), or to receive on all channels (omni on mode).
The latter mode is useful as a means of determining whether anything at
all is being received by the device.
Later it will be seen that the limit of 16 MIDI channels can be overcome easily by using multiport MIDI interfaces connected to a computer.
In such cases it is important not to confuse the MIDI data channel with
the physical port to which a device may be connected, since each physical
port will be capable of transmitting on all 16 data channels.
Channel and system messages contrasted
Two primary classes of message exist: those that relate to specific MIDI
channels and those that relate to the system as a whole. One should bear
in mind that it is possible for an instrument to be receiving in ‘omni on’
mode, in which case it will ignore the channel label and attempt to respond
to anything that it receives.
Channel messages start with status bytes in the range &8n to &En
(they start at hexadecimal eight because the MSB must be a one for a status
411
FA C T F I L E 14 . 3 MID I ME S S A G E FORM AT
There are two basic types of MIDI message byte: the
status byte and the data byte. The first byte in a MIDI
message is normally a status byte. Standard MIDI messages can be up to three bytes long, but not all messages
require three bytes, and there are some fairly common
exceptions to the rule which are described below. The
standard has been extended and refined over the years
and the following is only an introduction to the basic messages. The prefix ‘&’ will be used to indicate hexadecimal
values (see Table 8.1); individual MIDI message bytes
will be delineated using square brackets, e.g. [&45], and
channel numbers will be denoted using ‘n’ to indicate that
the value may be anything from &0 to &F (channels 1 to
16). The table shows the format and content of MIDI messages under each of the statuses.
Status bytes always begin with a binary one to distinguish them from data bytes, which always begin with
a zero. Because the most significant bit (MSB) of each
byte is reserved to denote the type (status or data) there
are only seven active bits per byte which allows 27 (that
is 128) possible values. As shown in the figure below,
the first half of the status byte denotes the message
type and the second half denotes the channel number.
Because four bits of the status byte are set aside to
indicate the channel number, this allows for 24 (or 16)
possible channels. There are only three bits to denote
the message type, because the first bit must always be
a one. This theoretically allows for eight message types,
but there are some special cases in the form of system
messages (see below).
8 bits
1 s s s n n n n
Status
0 x x x x x x x
Data 1
0 y y y y y y y
Data 2
The MMA has defined Approved Protocols (APs) and
Recommended Practices (RPs). An AP is a part of the
standard MIDI specification and is used when the standard is further defined or when a previously undefined
command is defined, whereas an RP is used to describe
an optional new MIDI application that is not a mandatory
or binding part of the standard. Not all MIDI devices will
have all the following commands implemented, since it is
not mandatory for a device conforming to the MIDI standard to implement every possibility.
Message
Status
Data 1
Data 2
Note off
Note on
Polyphonic aftertouch
Control change
Program change
Channel aftertouch
Pitch wheel
&8n
&9n
&An
&Bn
&Cn
&Dn
&En
Note number
Note number
Note number
Controller number
Program number
Pressure
LSbyte
Velocity
Velocity
Pressure
Data
–
–
MSbyte
&F0
&F7
Manufacturer ID
–
Data, (Data), (Data)
&F1
&F2
&F3
&F6
Data
LSbyte
Song number
–
–
MSbyte
–
&F8
&FA
&FB
&FC
&FE
&FF
–
–
–
–
–
–
–
–
–
–
–
–
System exclusive
System exclusive start
End of SysEx
System common
Quarter frame
Song pointer
Song select
Tune request
System real time
Timing clock
Start
Continue
Stop
Active sensing
Reset
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CHAPTER 14: MIDI and Synthetic Audio Control
byte). System messages all begin with &F, and do not contain a channel
number. Instead the least significant nibble of the system status byte is
used for further identification of the system message, such that there is
room for 16 possible system messages running from &F0 to &FF. System
messages are themselves split into three groups: system common, system
exclusive and system real time. The common messages may apply to any
device on the MIDI bus, depending only on the device’s ability to handle
the message. The exclusive messages apply to whichever manufacturers’
devices are specified later in the message (see below) and the real-time messages are intended for devices which are to be synchronized to the prevailing musical tempo. (Some of the so-called real-time messages do not really
seem to deserve this appellation, as discussed below.) The status byte &F1
is used for MIDI TimeCode.
MIDI channel numbers are usually referred to as ‘channels one to 16’,
but the binary numbers representing these run from zero to 15 (&0 to &F),
as 15 is the largest decimal number which can be represented with four
bits. Thus the note on message for channel 5 is actually &94 (nine for note
on, and four for channel 5).
Note on and note off messages
Much of the musical information sent over a typical MIDI interface will
consist of these two message types. As indicated by the titles, the note on
message turns on a musical note, and the note off message turns it off.
Note on takes the general format:
[&8n ] [Note number ] [Velocity ]
and note off takes the form:
[&9n ] [Note number ] [Velocity ]
A MIDI instrument will generate note on messages at its MIDI OUT
corresponding to whatever notes are pressed on the keyboard, on whatever
channel the instrument is set to transmit. Also, any note which has been
turned on must subsequently be turned off in order for it to stop sounding,
thus if one instrument receives a note on message from another and then
loses the MIDI connection for any reason, the note will continue sounding
ad infinitum. This situation can occur if a MIDI cable is pulled out during
transmission.
How MIDI Control Works
Table 14.1
MIDI note numbers related to the musical scale
Musical note
MIDI note number
C–2
C–1
C0
C1
C2
C3 (middle C)
C4
C5
C6
C7
C8
G8
0
12
24
36
48
60 (Yamaha convention)
72
84
96
108
120
127
MIDI note numbers relate directly to the western musical chromatic
scale and the format of the message allows for 128 note numbers which
cover a range of a little over ten octaves – adequate for the full range of
most musical material. This quantization of the pitch scale is geared very
much towards keyboard instruments, being less suitable for other instruments and cultures where the definition of pitches is not so black and
white. Nonetheless, means have been developed of adapting control to situations where unconventional tunings are required. Note numbers normally
relate to the musical scale as shown in Table 14.1, although there is a certain degree of confusion here. Yamaha established the use of C3 for middle
C, whereas others have used C4. Some software allows the user to decide
which convention will be used for display purposes.
Velocity information
Note messages are associated with a velocity byte that is used to represent the speed at which a key was pressed or released. The former will
correspond to the force exerted on the key as it is depressed: in other
words, ‘how hard you hit it’ (called ‘note on velocity ’). It is used to control
parameters such as the volume or timbre of the note at the audio output
of an instrument and can be applied internally to scale the effect of one
or more of the envelope generators in a synthesizer. This velocity value
has 128 possible states, but not all MIDI instruments are able to generate
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CHAPTER 14: MIDI and Synthetic Audio Control
or interpret the velocity byte, in which case they will set it to a value half
way between the limits, i.e. 6410. Some instruments may act on velocity
information even if they are unable to generate it themselves. It is recommended that a logarithmic rather than linear relationship should be established between the velocity value and the parameter which it controls,
since this corresponds more closely to the way in which musicians expect
an instrument to respond, although some instruments allow customized
mapping of velocity values to parameters. The note on, velocity zero value
is reserved for the special purpose of turning a note off, for reasons that
will become clear under ‘Running status’, below. If an instrument sees a
note number with a velocity of zero, its software should interpret this as
a note off message.
Note off velocity (or ‘release velocity ’) is not widely used, as it relates to
the speed at which a note is released, which is not a parameter that affects
the sound of many normal keyboard instruments. Nonetheless it is available for special effects if a manufacturer decides to implement it.
Running status
Running status is an accepted method of reducing the amount of data
transmitted. It involves the assumption that once a status byte has been
asserted by a controller there is no need to reiterate this status for each
subsequent message of that status, so long as the status has not changed in
between. Thus a string of notes on messages could be sent with the note on
status only sent at the start of the series of note data, for example:
[&9n ] [Data] [Velocity ] [Data] [Velocity ] [Data] [Velocity ]
For a long string of notes this could reduce the amount of data sent by
nearly one-third. But in most music each note on is almost always followed
quickly by a note off for the same note number, so this method would
clearly break down as the status would be changing from note on to note
off very regularly, thus eliminating most of the advantage gained by running status. This is the reason for the adoption of note on, velocity zero as
equivalent to a note off message, because it allows a string of what appears
to be note on messages, but which is, in fact, both note on and note off.
Running status is not used at all times for a string of same-status messages and will often only be called upon by an instrument’s software when
the rate of data exceeds a certain point. Indeed, an examination of the data
from a typical synthesizer indicates that running status is not used during
a large amount of ordinary playing.
How MIDI Control Works
Polyphonic key pressure (aftertouch)
The key pressure messages are sometimes called ‘aftertouch’ by keyboard
manufacturers. Aftertouch is perhaps a slightly misleading term as it does
not make clear what aspect of touch is referred to, and many people have
confused it with note off velocity. This message refers to the amount of
pressure placed on a key at the bottom of its travel, and it is used to instigate effects based on how much the player leans onto the key after depressing it. It is often applied to performance parameters such as vibrato.
The polyphonic key pressure message is not widely used, as it transmits
a separate value for every key on the keyboard and thus requires a separate
sensor for every key. This can be expensive to implement and is beyond the
scope of many keyboards, so most manufacturers have resorted to the use
of the channel pressure message (see below). The message takes the general
format:
[&An ] [Note number ] [Pressure ]
Implementing polyphonic key pressure messages involves the transmission of a considerable amount of data that might be unnecessary, as
the message will be sent for every note in a chord every time the pressure
changes. As most people do not maintain a constant pressure on the bottom of a key whilst playing, many redundant messages might be sent per
note. A technique known as ‘controller thinning’ may be used by a device
to limit the rate at which such messages are transmitted and this may be
implemented either before transmission or at a later stage using a computer.
Alternatively this data may be filtered out altogether if it is not required.
Control change
As well as note information, a MIDI device may be capable of transmitting control information that corresponds to the various switches, control
wheels and pedals associated with it. These come under the control change
message group and should be distinguished from program change messages.
The controller messages have proliferated enormously since the early days
of MIDI and not all devices will implement all of them. The control change
message takes the general form:
[&Bn ] [Controller number ] [Data]
so a number of controllers may be addressed using the same type of status
byte by changing the controller number.
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Although the original MIDI standard did not lay down any hard and
fast rules for the assignment of physical control devices to logical controller numbers, there is now common agreement amongst manufacturers
that certain controller numbers will be used for certain purposes. These
are assigned by the MMA. There are two distinct kinds of controller: the
switch type and the analog type. The analog controller is any continuously
variable wheel, lever, slider or pedal that might have any one of a number
of positions and these are often known as continuous controllers. There
are 128 controller numbers available and these are grouped as shown in
Table 14.2. Table 14.3 shows a more detailed breakdown of some of these,
as found in the majority of MIDI-controlled musical instruments, although
the full list is regularly updated by the MMA. The control change messages
have become fairly complex and interested users are referred to the relevant
standards.
The first 64 controller numbers (that is up to &3F) relate to only 32
physical controllers (the continuous controllers). This is to allow for greater
resolution in the quantization of position than would be feasible with the
seven bits that are offered by a single data byte. Seven bits would only allow
128 possible positions of an analog controller to be represented and this
might not be adequate in some cases. For this reason the first 32 controllers
handle the most significant byte (MSbyte) of the controller data, whilst the
second 32 handle the least significant byte (LSbyte). In this way, controller
numbers &06 and &38 both represent the data entry slider, for example.
Together, the data values can make up a 14 bit number (because the first
bit of each data word has to be a zero), which allows the quantization of
a control’s position to be one part in 214 (16 38410). Clearly, not all controllers will require this resolution, but it is available if needed. Only the
LSbyte would be needed for small movements of a control. If a system opts
not to use the extra resolution offered by the second byte, it should send
only the MSbyte for coarse control. In practice this is all that is transmitted
on many devices.
Table 14.2
MIDI controller classifications
Controller number (hex)
Function
&00–1F
&20–3F
&40–65
&66–77
&78–7F
14 bit controllers, MSbyte
14 bit controllers, LSbyte
7 bit controllers or switches
Originally undefined
Channel mode control
Table 14.3
MIDI controller functions
Controller number (hex)
Function
00
01
02
03
04
05
06
07
08
09
0A
0B
0C
0D
0E–0F
10–13
14–1F
20–3F
40
41
42
43
44
45
46–4F
50–53
54
55–5A
5B–5F
60
61
62
63
64
65
66–77
78
79
7A
7B
7C
7D
7E
7F
Bank select
Modulation wheel
Breath controller
Undefined
Foot controller
Portamento time
Data entry slider
Main volume
Balance
Undefined
Pan
Expression controller
Effect control 1
Effect control 2
Undefined
General purpose controllers 1–4
Undefined
LSbyte for 14 bit controllers (same function order as 00–1F)
Sustain pedal
Portamento on/off
Sostenuto pedal
Soft pedal
Legato footswitch
Hold 2
Sound controllers
General purpose controllers 5–8
Portamento control
Undefined
Effects depth 1–5
Data increment
Data decrement
NRPC LSbyte (non-registered parameter controller)
NRPC MSbyte
RPC LSbyte (registered parameter controller)
RPC MSbyte
Undefined
All sounds off
Reset all controllers
Local on/off
All notes off
Omni receive mode off
Omni receive mode on
Mono receive mode
Poly receive mode
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On/off switches can be represented easily in binary form (0 for OFF, 1
for ON), and it would be possible to use just a single bit for this purpose,
but, in order to conform to the standard format of the message, switch
states are normally represented by data values between &00 and &3F for
OFF and &40 and &7F for ON. In other words switches are now considered as seven bit continuous controllers. In older systems it may be found
that only &00 ⫽ OFF and &7F ⫽ ON.
The data increment and decrement buttons that are present on many
devices are assigned to two specific controller numbers (&60 and &61)
and an extension to the standard defines four controllers (&62 to &65)
that effectively expand the scope of the control change messages. These
are the registered and non-registered parameter controllers (RPCs and
NRPCs).
The ‘all notes off ’ command (frequently abbreviated to ‘ANO ’) was
designed to be transmitted to devices as a means of silencing them, but
it does not necessarily have this effect in practice. What actually happens
varies between instruments, especially if the sustain pedal is held down or
notes are still being pressed manually by a player. All notes off is supposed
to put all note generators into the release phase of their envelopes, and
clearly the result of this will depend on what a sound is programmed to do
at this point. The exception should be notes which are being played whilst
the sustain pedal is held down, which should only be released when that
pedal is released. ‘All sounds off ’ was designed to overcome the problems
with ‘all notes off ’, by turning sounds off as quickly as possible. ‘Reset all
controllers’ is designed to reset all controllers to their default state, in order
to return a device to its ‘standard’ setting.
Channel modes
Although grouped with the controllers, under the same status, the channel
mode messages differ somewhat in that they set the mode of operation of the
instrument receiving on that particular channel.
‘Local on/off ’ is used to make or break the link between an instrument’s
keyboard and its own sound generators. Effectively there is a switch between
the output of the keyboard and the control input to the sound generators
which allows the instrument to play its own sound generators in normal
operation when the switch is closed (see Figure 14.8). If the switch is opened,
the link is broken and the output from the keyboard feeds the MIDI OUT
whilst the sound generators are controlled from the MIDI IN. In this mode
the instrument acts as two separate devices: a keyboard without any sound,
How MIDI Control Works
and a sound generator without a keyboard. This
configuration can be useful when the instrument
Sound generators
in use is the master keyboard for a large sequencer
system, where it may not always be desired that
everything played on the master keyboard results in
sound from the instrument itself.
MIDI IN
Local on/off
‘Omni off ’ ensures that the instrument will only
switch
MIDI OUT
act on data tagged with its own channel number(s),
as set by the instrument’s controls. ‘Omni on’ sets
the instrument to receive on all of the MIDI channels. In other words, the instrument will ignore the
channel number in the status byte and will attempt
Keyboard
to act on any data that may arrive, whatever its
FIGURE 14.8 The ‘local off’ switch disconnects a
channel. Devices should power up in this mode
keyboard from its associated sound generators in order
according to the original specification, but more
that the two parts may be treated independently in a MIDI
recent devices will tend to power up in the mode in
system.
which they were left. Mono mode sets the instrument such that it will only reproduce one note at a time, as opposed to ‘Poly ’
(phonic) in which a number of notes may be sounded together.
In older devices the mono mode came into its own as a means of operating an instrument in a ‘multi-timbral’ fashion, whereby MIDI information on each channel controlled a separate monophonic musical voice.
This used to be one of the only ways of getting a device to generate more
than one type of voice at a time. The data byte that accompanies the mono
mode message specifies how many voices are to be assigned to adjacent
MIDI channels, starting with the basic receive channel. For example, if the
data byte is set to 4, then four voices will be assigned to adjacent MIDI
channels, starting from the basic channel which is the one on which the
instrument has been set to receive in normal operation. Exceptionally, if
the data byte is set to zero, all 16 voices (if they exist) are assigned each to
one of the 16 MIDI channels. In this way, a single multi-timbral instrument can act as 16 monophonic instruments, although on cheaper systems
all of these voices may be combined to one audio output.
Mono mode tends to be used mostly on MIDI guitar synthesizers
because each string can then have its own channel and each can control
its own set of pitch bend and other parameters. The mode also has the
advantage that it is possible to play in a truly legato fashion – that is with
a smooth takeover between the notes of a melody – because the arrival of
a second note message acts simply to change the pitch if the first one is
still being held down, rather than retriggering the start of a note envelope.
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The legato switch controller allows a similar type of playing in polyphonic
modes by allowing new note messages only to change the pitch.
In poly mode the instrument will sound as many notes as it is able to at
the same time. Instruments differ as to the action taken when the number
of simultaneous notes is exceeded: some will release the first note played
in favor of the new note, whereas others will refuse to play the new note.
Some may be able to route excess note messages to their MIDI OUT ports
so that they can be played by a chained device. The more intelligent of
them may look to see if the same note already exists in the notes currently
sounding and only accept a new note if is not already sounding. Even more
intelligently, some devices may release the quietest note (that with the lowest velocity value), or the note furthest through its velocity envelope, to
make way for a later arrival. It is also common to run a device in poly mode
on more than one receive channel, provided that the software can handle
the reception of multiple polyphonic channels. A multi-timbral sound generator may well have this facility, commonly referred to as ‘multi’ mode,
making it act as if it were a number of separate instruments each receiving
on a separate channel. In multi mode a device may be able to dynamically
assign its polyphony between the channels and voices in order that the user
does not need to assign a fixed polyphony to each voice.
Program change
The program change message is used most commonly to change the ‘patch’ of
an instrument or other device. A patch is a stored configuration of the device,
describing the setup of the tone generators in a synthesizer and the way in
which they are interconnected. Program change is channel specific and there
is only a single data byte associated with it, specifying to which of 128 possible
stored programs the receiving device should switch. On non-musical devices
such as effects units, the program change message is often used to switch
between different effects and the different effects programs may be mapped to
specific program change numbers. The message takes the general form:
&[Cn ] [Program number ]
If a program change message is sent to a musical device it will usually result
in a change of voice, as long as this facility is enabled. Exactly which voice
corresponds to which program change number depends on the manufacturer. It is quite common for some manufacturers to implement this
function in such a way that a data value of zero gives voice number one.
How MIDI Control Works
This results in a permanent offset between the program change number
and the voice number, which should be taken into account in any software.
On some instruments, voices may be split into a number of ‘banks’ of 8, 16
or 32, and higher banks can be selected over MIDI by setting the program
change number to a value which is 8, 16 or 32 higher than the lowest bank
number. For example, bank 1, voice 2, might be selected by program change
&01, whereas bank 2, voice 2, would probably be selected in this case by
program change &11, where there were 16 voices per bank.
There are also a number of other approaches used in commercial sound
modules. Where more than 128 voices need to be addressed remotely, the
more recent ‘bank select’ command may be implemented.
Channel aftertouch
Most instruments use a single sensor, often in the form of a pressuresensitive conductive plastic bar running the length of the keyboard, to
detect the pressure applied to keys at the bottom of their travel. In the case
of channel aftertouch, one message is sent for the entire instrument and
this will correspond to an approximate total of the pressure over the range
of the keyboard, the strongest influence being from the key pressed the
hardest. (Some manufacturers have split the pressure detector into upper
and lower keyboard regions, and some use ‘intelligent’ zoning.) The message takes the general form:
&[Dn ] [Pressure value ]
There is only one data byte, so there are 128 possible values and, as with
the polyphonic version, many messages may be sent as the pressure is varied at the bottom of a key’s travel. Controller ‘thinning’ may be used to
reduce the quantity of these messages, as described above.
Pitch bend wheel
The pitch wheel message has a status byte of its own, and carries information about the movement of the sprung-return control wheel on many
keyboards which modifies the pitch of any note(s) played. It uses two data
bytes in order to give 14 bits of resolution, in much the same way as the
continuous controllers, except that the pitch wheel message carries both
bytes together. Fourteen data bits are required so that the pitch appears to
change smoothly, rather than in steps (as it might with only seven bits).
The pitch bend message is channel specific so ought to be sent separately
for each individual channel. This becomes important when using a single
multi-timbral device in mono mode (see above), as one must ensure that
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a pitch bend message only affects the notes on the intended channel. The
message takes the general form:
&[En ] [LSbyte ] [MSbyte ]
The value of the pitch bend controller should be halfway between the lower
and upper range limits when it is at rest in its sprung central position, thus
allowing bending both down and up. This corresponds to a hex value of
&2000, transmitted as &[En] [00] [40]. The range of pitch controlled by
the bend message is set on the receiving device itself, or using the RPC designated for this purpose (see ‘Control change’, above).
System exclusive
A system exclusive message is one that is unique to a particular manufacturer and often a particular instrument. The only thing that is defined
about such messages is how they are to start and finish, with the exception of the use of system exclusive messages for universal information, as
discussed elsewhere. System exclusive messages generated by a device will
naturally be produced at the MIDI OUT, not at the THRU, so a deliberate
connection must be made between the transmitting device and the receiving device before data transfer may take place.
Occasionally it is necessary to make a return link from the OUT of
the receiver to the IN of the transmitter so that two-way communication
is possible and so that the receiver can control the flow of data to some
extent by telling the transmitter when it is ready to receive and when it has
received correctly (a form of handshaking).
The message takes the general form:
&[F0] [ident.] [data] [data]…[F7]
where [ident.] identifies the relevant manufacturer ID, a number defining which manufacturer’s message is to follow. Originally, manufacturer
IDs were a single byte but the number of IDs has been extended by setting
aside the [00] value of the ID to indicate that two further bytes of ID follow.
Manufacturer IDs are therefore either one or three bytes long. A full list of
manufacturer IDs is available from the MMA.
Data of virtually any sort can follow the ID. It can be used for a variety of miscellaneous purposes that have not been defined in the MIDI standard and the message can have virtually any length that the manufacturer
requires. It is often split into packets of a manageable size in order not to
cause receiver memory buffers to overflow. Exceptions are data bytes that
How MIDI Control Works
look like other MIDI status bytes (except real-time messages), as they will
naturally be interpreted as such by any receiver, which might terminate
reception of the system exclusive message. The message should be terminated with &F7, although this is not always observed, in which case the
receiving device should ‘time-out’ after a given period, or terminate the system exclusive message on receipt of the next status byte. It is recommended
that some form of error checking (typically a checksum) is employed for long
system exclusive data dumps, and many systems employ means of detecting
whether the data has been received accurately, asking for retries of sections
of the message in the event of failure, via a return link to the transmitter.
Examples of applications for such messages can be seen in the form of
sample data dumps (from a sampler to a computer and back for editing purposes), although this is painfully slow, and voice data dumps (from a synthesizer to a computer for storage and editing of user-programmed voices).
There are now an enormous number of uses of system exclusive messages,
both in the universal categories and in the manufacturer categories.
Universal system exclusive messages
The three highest numbered IDs within the system exclusive message
have been set aside to denote special modes. These are the ‘universal noncommercial’ messages (ID: &7D), the ‘universal non-real-time’ messages
(ID: &7E) and the ‘universal real-time’ messages (ID: &7F). Universal sysex
messages are often used for controlling device parameters that were not originally specified in the MIDI standard and that now need addressing in most
devices. Examples are things like ‘chorus modulation depth’, ‘reverb type’
and ‘master fine tuning’.
Universal non-commercial messages are set aside for educational
and research purposes and should not be used in commercial products.
Universal non-real-time messages are used for universal system exclusive
events which are not time critical and universal real-time messages deal
with time critical events (thus being given a higher priority). The two latter
types of message normally take the general form of:
&[F0] [ID] [dev. ID] [sub-ID 1] [sub-ID 2] [data]……[F7]
Device ID used to be referred to as ‘channel number ’, but this did not
really make sense since a whole byte allows for the addressing of 128 channels and this does not correspond to the normal 16 channels of MIDI. The
term ‘device ID ’ is now used widely by software as a means of defining one
of a number of physical devices in a large MIDI system, rather than defining a MIDI channel number. It should be noted, though, that it is allowable
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for a device to have more than one ID if this seems appropriate. Modern
MIDI devices will normally allow their device ID to be set either over MIDI
or from the front panel. The use of &7F in this position signifies that the
message applies to all devices as opposed to just one.
The sub-IDs are used to identify first the category or application of the message (sub-ID #1) and secondly the type of message within that category (sub-ID
#2). For some reason, the original MIDI sample dump messages do not use the
sub-ID #2, although some recent additions to the sample dump do.
Tune request
Older analog synthesizers tended to drift somewhat in pitch over the time
that they were turned on. The tune request is a request for these synthesizers to retune themselves to a fixed reference. (It is advisable not to transmit
pitch bend or note on messages to instruments during a tune-up because of
the unpredictable behavior of some products under these conditions.)
Active sensing
Active sensing messages are single status bytes sent roughly three times per
second by a controlling device when there is no other activity on the bus.
It acts as a means of reassuring the receiving devices that the controller has
not disappeared. Not all devices transmit active sensing information, and
a receiver’s software should be able to detect the presence or lack of it. If a
receiver has come to expect active sensing bytes then it will generally act by
turning off all notes if these bytes disappear for any reason. This can be a useful function when a MIDI cable has been pulled out during a transmission, as
it ensures that notes will not be left sounding for very long. If a receiver has
not seen active sensing bytes since last turned on, it should assume that they
are not being used.
Reset
This message resets all devices on the bus to their power-on state. The
process may take some time and some devices mute their audio outputs,
which can result in clicks, therefore the message should be used with care.
MIDI CONTROL OF SOUND GENERATORS
MIDI note assignment in synthesizers and samplers
Many of the replay and signal processing aspects of synthesis and sampling
now overlap so that it is more difficult to distinguish between the two. In
basic terms a sampler is a device that stores short clips of sound data in RAM,
MIDI Control of Sound Generators
enabling them to be replayed subsequently at different pitches, possibly
looped and processed. A synthesizer is a device that enables signals to be
artificially generated and modified to create novel sounds. Wavetable synthesis is based on a similar principle to sampling, though, and stored samples can form the basis for synthesis. A sound generator can often generate
a number of different sounds at the same time. It is possible that these
sounds could be entirely unrelated (perhaps a single drum, an animal noise
and a piano note), or that they might have some relationship to each other
(perhaps a number of drums in a kit, or a selection of notes from a grand
piano). The method by which sounds or samples are assigned to MIDI
notes and channels is defined by the replay program.
The most common approach when assigning note numbers to samples
is to program the sampler with the range of MIDI note numbers over which
a certain sample should be sounded. Akai, one of the most popular sampler
manufacturers, called these ‘keygroups’. It may be that this ‘range’ is only
one note, in which case the sample in question would be triggered only on
receipt of that note number, but in the case of a range of notes the sample
would be played on receipt of any note in the range. In the latter case transposition would be required, depending on the relationship between the note
number received and the original note number given to the sample (see
above). A couple of examples highlight the difference in approach, as shown
in Figure 14.9. In the first example, illustrating a possible approach to note
assignment for a collection of drum kit sounds, most samples are assigned
to only one note number, although it is possible for tuned drum sounds
such as tom-toms to be assigned over a range in order to give the impression of ‘tuned toms’. Each MIDI note message received would replay the
particular percussion sound assigned to that note number in this example.
In the second example, illustrating a suggested approach to note assignment for an organ, notes were originally sampled every musical fifth across
the organ’s note range. The replay program has been designed so that each of
these samples is assigned to a note range of a fifth, centered on the original
pitch of each sample, resulting in a maximum transposition of a third up or
down. Ideally, of course, every note would have been sampled and assigned
to an individual note number on replay, but this requires very large amounts
of memory and painstaking sample acquisition on the first place.
In further pursuit of sonic accuracy, some devices provide the facility for
introducing a crossfade between note ranges. This is used where an abrupt
change in the sound at the boundary between two note ranges might be undesirable, allowing the takeover from one sample to another to be more gradual. For example, in the organ scenario introduced above, the timbre could
change noticeably when playing musical passages that crossed between two
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CHAPTER 14: MIDI and Synthetic Audio Control
Cow bell
Rattle
Cymbal
Tom-toms
Brushed cymbal
Timpani
Hi-hat
(a)
Snare
FIGURE 14.9
(a) Percussion samples
are often assigned to one
note per sample, except
for tuned percussion which
sometimes covers a range
of notes. (b) Organ samples
could be transposed over
a range of notes, centered
on the original pitch of the
sample.
Kick drum
426
(b)
E2
E2
C3
C3
G3
G3
E4
E4
note ranges because replay would switch from the upper limit of transposition
of one sample to the lower limit of the next (or vice versa). In this case the
ranges for the different samples are made to overlap (as illustrated in Figure
14.10). In the overlap range the system mixes a proportion of the two samples together to form the output. The exact proportion depends on the range
of overlap and the note’s position within this range. Very accurate tuning of
the original samples is needed in order to avoid beats when using positional
crossfades. Clearly this approach would be of less value when each note was
assigned to a completely different sound, as in the drum kit example.
Crossfades based on note velocity allow two or more samples to be
assigned to one note or range of notes. This requires at least a ‘loud sample’ and a ‘soft sample’ to be stored for each original sound and some systems may accommodate four or more to be assigned over the velocity
range. The terminology may vary, but the principle is that a velocity value
is set at which the replay switches from one stored sample to another, as
many instruments sound quite different when they are loud to when
MIDI Control of Sound Generators
E2
G3
C3
E2
E4
FIGURE 14.10
Overlapped sample ranges
can be crossfaded in order
that a gradual shift in
timbre takes place over the
region of takeover between
one range and the next.
G3
C3
E4
Crossfade
region
they are soft (it is more than just the volume that
changes: it is the timbre also). If a simple switching
point is set, then the change from one sample to the
other will be abrupt as the velocity crosses either side
of the relevant value. This can be illustrated by storing
two completely different sounds as the loud and soft
samples, in which case the output changes from one to
the other at the switching point. A more subtle effect
is achieved by using velocity crossfading, in which the
proportion of loud and soft samples varies depending on the received note velocity value. At low velocity
values the proportion of the soft sample in the output
would be greatest and at high values the output content
would be almost entirely made up of the loud sample
(see Figure 14.11).
Polyphony, voice and note assignment
Crossfade
region
7F
Max. velocity
Loud sample
range
Crossfade region
Crossfade
region
50
4F
Switch point (example)
Soft sample
range
00
Min. velocity
FIGURE 14.11 Illustration of velocity switch
and velocity crossfade between two stored samples
(‘soft’ and ‘loud’) over the range of MIDI note velocity
values.
Modern sound modules (synthesizers and samplers)
tend to be multi-note polyphonic. When the polyphony of a
device is exceeded the device should follow a predefined set of rules to determine what to do with the extra notes. Typically a sound module will either
release the ‘oldest’ notes first, or possibly release the quietest. Alternatively,
new notes that exceed the polyphony will simply not be sounded until others
are released. Rules for this are defined in some of the recent General MIDI
specifications (see below), and composers may now even be able to exercise
some control over what happens in devices with limited polyphony.
It is important to distinguish between the degree of polyphony offered by
a device and the number of simultaneous voices it can generate. Sometimes
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these may be traded off against each other in multi-timbral devices, by allocating a certain number of notes to each voice, with the total adding up to
the total polyphony. Either 16 notes could be allocated to one voice or four
notes to each of four voices, for example. Dynamic allocation is often used
to distribute the polyphony around the voices depending on demand and
this is a particular feature of General MIDI sound modules.
A multi-timbral sound generator is one that is capable of generating
more than one voice at a time, independent of polyphony considerations.
A voice is a particular sound type, such as ‘grand piano’ or ‘accordion’. This
capability is now the norm for modern sound modules. Older synthesizers used to be able to generate only one or two voices at a time, possibly
allowing a keyboard split, and could sometimes make use of MIDI channel
mode 4 (monophonic, omni off) to allow multiple monophonic voices to be
generated under MIDI control. They tended only to receive polyphonically
on one MIDI channel at a time. More recent systems are capable of receiving on all 16 MIDI channels simultaneously, with each channel controlling
an entirely independent polyphonic voice.
MIDI functions of sound generators
The MIDI implementation for a particular sound generator should be
described in the manual that accompanies it. A MIDI implementation chart
indicates which message types are received and transmitted, together with
any comments relating to limitations or unusual features. Functions such as
note off velocity and polyphonic aftertouch, for example, are quite rare. It is
quite common for a device to be able to accept certain data and act upon it,
even if it cannot generate such data from its own controllers. The note range
available under MIDI control compared with that available from a device’s
keyboard is a good example of this, since many devices will respond to note
data over a full ten octave range yet still have only a limited (or no) keyboard.
This approach can be used by a manufacturer who wishes to make a cheaper
synthesizer that omits the expensive physical sensors for such things as
velocity and aftertouch, whilst retaining these functions in software for use
under MIDI control. Devices conforming to the General MIDI specification
described below must conform to certain basic guidelines concerning their
MIDI implementation and the structure of their sound generators.
MIDI data buffers and latency
All MIDI-controlled equipment uses some form of data buffering for received
MIDI messages. Such buffering acts as a temporary store for messages that
have arrived but have not yet been processed and allows for a certain prioritization in the handling of received messages. Cheaper devices tend to have
MIDI Control of Sound Generators
relatively small MIDI input buffers and these can overflow easily unless care
is taken in the filtering and distribution of MIDI data around a large system (usually accomplished by a MIDI router or multiport inferface). When a
buffer overflows it will normally result in an error message displayed on the
front panel of the device, indicating that some MIDI data is likely to have
been lost. More advanced equipment can store more MIDI data in its input
buffer, although this is not necessarily desirable because many messages that
are transmitted over MIDI are intended for ‘real-time’ execution and one
would not wish them to be delayed in a temporary buffer. Such buffer delay
is one potential cause of latency in MIDI systems. A more useful solution
would be to speed up the rate at which incoming messages are processed.
Handling of velocity and aftertouch data
Sound generators able to respond to note on velocity will use the value of
this byte to control assigned functions within the sound generators. It is
common for the user to be able to program the device such that the velocity value affects certain parameters to a greater or lesser extent. For example, it might be decided that the ‘brightness’ of the sound should increase
with greater key velocity, in which case it would be necessary to program
the device so that the envelope generator that affected the brightness was
subject to control by the velocity value. This would usuall