Design of Integrated, Low Power, Radio Receivers in

Design of Integrated, Low Power, Radio Receivers in
Design of Integrated, Low Power, Radio
Receivers in BiCMOS Technologies
by
William B. Kuhn
Dissertation submitted to the faculty of the
Virginia Polytechnic Institute and State University
in partial fulfillment of the requirements for the degree of
DOCTOR OF PHILOSOPHY
in
Electrical Engineering
William
c
B. Kuhn and VPI & SU 1995
APPROVED:
Aicha Elshabini-Riad, Co-chairman
F. William Stephenson, Co-chairman
Peter M. Athanas
Charles W. Bostian
Lee W. Johnson
Timothy Pratt
December, 1995
Blacksburg, Virginia
Design of Integrated, Low Power, Radio Receivers in
BiCMOS Technologies
by
William B. Kuhn
Committee Co-chairmen:
Aicha Elshabini-Riad
Bradley Department of Electrical Engineering
and
F. William Stephenson
Bradley Department of Electrical Engineering
(ABSTRACT)
Despite increasing levels of integration in modern electronic products, radio receiver
designs continue to rely on discrete LC, ceramic, and electro-acoustic devices for the realization of RF and IF bandpass filtering. Although considerable research has been directed
at developing suitable switched-capacitor and Gm-C based replacements for these filters,
the resulting designs have yet to see substantial commercial application.
A critical problem faced by existing active filter implementations is found to be the
power consumption required to simultaneously achieve narrow fractional bandwidths and
acceptable dynamic range. This power consumption, which can reach several hundred milliwatts, is incompatible with portable wireless product design. Additional problems include
the complexity of tuning control circuits required to achieve small fractional bandwidths,
and difficulties in extending filter designs to higher frequencies. These problems are examined in depth, and performace bounds and new implementation techniques are considered.
A detailed study of active filters reveals that their dynamic range limitations are fundamentally the result of regenerative gain associated with the realization of high-Q poles.
Thus, some form of energy storage and exchange mechanism is shown to be required to
decrease the regeneration needed. This leads to an investigation of on-chip LC filtering. It
is shown that on-chip spiral inductors can be designed to resonate with both intentional
and parasitic capacitances, forming stable tuned circuits operating from 100 MHz to over
1 GHz. Although the Q of the inductors employed is typically small (Q < 10), negative
resistance circuits can be used to increase the effective Q to arbitrarily high values. Hence,
very small fractional bandwidths (< 2%) can be obtained. Moreover, even a small inductor
Q is shown to provide significant increases in dynamic range over that achievable in fully
active filter designs.
Important practical considerations surrounding the implementation of Q-enhanced LC
filters in silicon CMOS processes are then investigated, including realizing the necessary onchip spiral inductors and Q-enhancement circuits, predicting frequency and Q tolerances
and temperature stability, and developing real-time frequency and Q tuning mechanisms.
These issues are studied in depth and two prototype filters designed to validate theoretical
predictions are reported. Performance levels achieved by these prototypes indicate that Qenhanced filtering offers a viable approach to solving the on-chip bandpass filtering problem.
These filters can therefore be expected to play an important role in the development of future
integrated receiver products.
iii
ACKNOWLEDGEMENTS
Many individuals have contributed either directly, or indirectly, to this research, and to
the education I have received over the years which made it possible. First and foremost, I
am indebted to my chairmen and committee for overseeing the work, for reading through
many progress reports, for helpful guidance in preparation of research papers, and for
reviewing and editing the dissertation itself. Their time, encouragement, and advice are
greatly appreciated.
In particular, I would like to thank Professor Elshabini-Riad for her encouragement,
patience, and excellent insight into how to get things done, and Professor Stephenson for
his support of the work and of my education, and for always being available, even when
the duties of being Dean of the College of Engineering demanded all of his time. Also,
many of those on my committee have been my teachers, and I would like to thank them
for providing many helpful and engaging lectures and for their devotion to their chosen
profession. Indeed, it is the faculty of a university that makes education possible, and I am
deeply indebted to all of the teachers I have had over the years.
Special thanks are also due Professor Bill Davis for many enjoyable discussions, and for
providing access to laboratory equipment critical to the evaluation of prototype hardware
created during this research. In addition, I would like to thank Loretta Estes for always
knowing how the department works, Virginia McWhorter for assigning me to teach the right
courses during the summers, Brian Fox and Dan Fague for their insights into “real-world”
issues, Tony Franks for constructing hardware used in testing, the faculty and students of
MPRG for allowing me to sit in on their meetings and keep abreast of the broader world of
iv
research in wireless telecommunications, and all the members of the Virginia Tech Amateur
Radio Association for the opportunity for relaxation away from the rigors of text books.
Thanks also go to MOSIS for fabrication of the prototype ICs, to Motorola and Tandy
Corporation for allowing schematics and block diagrams in Chapter 2 to be used, and to
the IEE and IEEE for permission to reprint portions of conference and journal papers
published in the course of this work.
My greatest appreciation however, goes to my wife Anne, and to my family. Thanks go
to Anne for standing by me over 10+ years, and for encouraging me to leave my “lucrative”
job at Georgia Tech and realize my dream of returning to the student life. Thanks go to
my family for their love and guidance, for supporting me through college, and for their
continuing support throughout the years.
Finally, I would like to offer my deep appreciation to the Via Family for the Bradley
Endowment. The Bradley Fellowship has provided living expenses and tuition for the last
three years, allowing me to pursue full-time study and research in my chosen area. Moreover,
as Anne will attest, the Fellowship was the key factor in our decision to leave the hustle of
metro-Atlanta and make our new home in the mountains of Virginia.
v
TABLE OF CONTENTS
1 Introduction
1
1.1
Scope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1
1.2
Problem Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1
1.3
Dissertation Outline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
2
2 Background and Literature Review
2.1
2.2
2.3
2.4
6
Commercial Integrated Circuits . . . . . . . . . . . . . . . . . . . . . . . . .
8
2.1.1
Receiver Building Blocks . . . . . . . . . . . . . . . . . . . . . . . .
9
2.1.2
Cellular Telephone Design Example . . . . . . . . . . . . . . . . . .
12
2.1.3
Cellular/PCS Front End, Power Amplifier, and Transverter ICs . . .
16
2.1.4
Cellular/PCS Chip Sets . . . . . . . . . . . . . . . . . . . . . . . . .
18
2.1.5
Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
21
Integrated Bandpass Filter Research . . . . . . . . . . . . . . . . . . . . . .
22
2.2.1
Filter Performance Benchmarks . . . . . . . . . . . . . . . . . . . . .
22
On-Chip Filter Alternatives . . . . . . . . . . . . . . . . . . . . . . . . . . .
24
2.3.1
Digital Filters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
25
2.3.2
Analog Passive Filters . . . . . . . . . . . . . . . . . . . . . . . . . .
26
2.3.3
Analog Active Filters . . . . . . . . . . . . . . . . . . . . . . . . . .
30
2.3.4
Summary of Alternatives . . . . . . . . . . . . . . . . . . . . . . . .
42
Integrated Synthesizers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
44
2.4.1
On-Chip Local Oscillators and VCOs . . . . . . . . . . . . . . . . .
45
2.4.2
Reported Implementations . . . . . . . . . . . . . . . . . . . . . . . .
47
2.4.3
Direct Digital Synthesis . . . . . . . . . . . . . . . . . . . . . . . . .
49
vi
CONTENTS
2.5
Integrated Receivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
50
2.5.1
Classical Superheterodyne Architectures . . . . . . . . . . . . . . . .
51
2.5.2
Ultra-Low IF Architectures . . . . . . . . . . . . . . . . . . . . . . .
54
2.5.3
Direct Conversion Architectures . . . . . . . . . . . . . . . . . . . .
55
3 System Requirements
58
3.1
The Spectrum Environment . . . . . . . . . . . . . . . . . . . . . . . . . . .
59
3.2
Commercial Wireless Services . . . . . . . . . . . . . . . . . . . . . . . . . .
61
3.3
Receiver Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
64
3.3.1
Sensitivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
64
3.3.2
Dynamic Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
67
3.3.3
Selectivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
84
3.3.4
Fidelity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
90
4 Alternative Receiver Architectures
4.1
93
Early Receiver Architectures . . . . . . . . . . . . . . . . . . . . . . . . . .
94
4.1.1
Tuned RF Receivers . . . . . . . . . . . . . . . . . . . . . . . . . . .
94
4.1.2
Regenerative Receivers . . . . . . . . . . . . . . . . . . . . . . . . . .
96
4.1.3
Super-Regenerative Receivers . . . . . . . . . . . . . . . . . . . . . .
98
Superheterodyne Receivers . . . . . . . . . . . . . . . . . . . . . . . . . . .
100
4.2.1
Modern Single Conversion Implementations . . . . . . . . . . . . . .
101
4.2.2
Multiple Conversion Implementations . . . . . . . . . . . . . . . . .
102
4.2.3
Up Conversion Implementations . . . . . . . . . . . . . . . . . . . .
104
4.2.4
Designs with Ultra-Low IFs . . . . . . . . . . . . . . . . . . . . . . .
106
4.2.5
Designs with Image Rejection Mixers . . . . . . . . . . . . . . . . . .
108
4.2.6
Designs with Selective Demodulators . . . . . . . . . . . . . . . . . .
109
4.3
Direct Conversion Receivers . . . . . . . . . . . . . . . . . . . . . . . . . . .
110
4.4
Digital Receivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
113
4.5
Ideal Low-Power Receivers . . . . . . . . . . . . . . . . . . . . . . . . . . . .
115
4.2
vii
CONTENTS
5 Integrated Bandpass Filter Design Options
5.1
5.2
Continuous-Time Active Filters . . . . . . . . . . . . . . . . . . . . . . . . .
120
5.1.1
Bandpass Filter Structures . . . . . . . . . . . . . . . . . . . . . . .
122
5.1.2
Gm-C Filter Design . . . . . . . . . . . . . . . . . . . . . . . . . . .
125
5.1.3
Q-enhanced LC Filter Design . . . . . . . . . . . . . . . . . . . . . .
131
Electro-Acoustic Filters . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
134
6 Active Filter Dynamic Range
6.1
119
136
Dynamic Range of Gm-C Filters . . . . . . . . . . . . . . . . . . . . . . . .
138
6.1.1
Optimum Dynamic Range . . . . . . . . . . . . . . . . . . . . . . . .
141
6.1.2
Dynamic Range Versus Power Consumption and Bandwidth . . . . .
142
6.2
Dynamic Range of Q-Enhanced LC Filters . . . . . . . . . . . . . . . . . . .
146
6.3
Relationship of Filter Dynamic Range to Receiver Dynamic Range . . . . .
149
6.4
Blocking and Spurious Free Dynamic Range . . . . . . . . . . . . . . . . . .
150
6.5
Limitations on Filter Q . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
154
7 Tuning Techniques for Active Filters
159
7.1
The Need for Filter Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . .
159
7.2
Tuning Control Sytem Designs . . . . . . . . . . . . . . . . . . . . . . . . .
165
7.2.1
Master-Slave Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . .
166
7.2.2
Self-Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
168
7.2.3
Correlated Tuning Loop . . . . . . . . . . . . . . . . . . . . . . . . .
169
7.2.4
Adaptive Filter Tuning . . . . . . . . . . . . . . . . . . . . . . . . .
171
7.2.5
Orthogonal Reference Tuning . . . . . . . . . . . . . . . . . . . . . .
172
7.2.6
KIS Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
186
8 Q-Enhanced LC Filter Design
8.1
192
On-Chip Spiral Inductors . . . . . . . . . . . . . . . . . . . . . . . . . . . .
193
8.1.1
194
Inductance Value . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
viii
CONTENTS
8.2
8.3
8.4
8.1.2
Inductor Q . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
195
8.1.3
Modeling of Inductor Q and Self-Resonant Frequency . . . . . . . .
197
Circuit Design Considerations . . . . . . . . . . . . . . . . . . . . . . . . . .
198
8.2.1
Component Value Selection . . . . . . . . . . . . . . . . . . . . . . .
200
8.2.2
Balanced Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
204
8.2.3
Center-Tapped Spiral Inductors . . . . . . . . . . . . . . . . . . . . .
206
Filter Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
211
8.3.1
Frequency Tolerance and Temperature Coefficient . . . . . . . . . .
211
8.3.2
Q Tolerance and Temperature Coefficient . . . . . . . . . . . . . . .
212
8.3.3
Frequency and Q Adjustment . . . . . . . . . . . . . . . . . . . . . .
213
Higher-Order Filter Design . . . . . . . . . . . . . . . . . . . . . . . . . . .
215
9 Q-Enhanced LC Filter Implementations
9.1
9.2
220
Ultra-High Q VHF Receiver Front-End . . . . . . . . . . . . . . . . . . . . .
221
9.1.1
Design and Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . .
222
9.1.2
Experimental Results . . . . . . . . . . . . . . . . . . . . . . . . . .
227
Cellular and PCS Receiver IF Filter . . . . . . . . . . . . . . . . . . . . . .
233
9.2.1
Design and Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . .
234
9.2.2
Experimental Results . . . . . . . . . . . . . . . . . . . . . . . . . .
240
10 Integrated Receiver Design
248
10.1 Integration of Local Oscillators . . . . . . . . . . . . . . . . . . . . . . . . .
248
10.1.1 Phase Noise in Oscillators . . . . . . . . . . . . . . . . . . . . . . . .
249
10.1.2 Phase Noise Effects on Receiver Operation . . . . . . . . . . . . . .
253
10.2 Receiver Design Example . . . . . . . . . . . . . . . . . . . . . . . . . . . .
257
10.2.1 DECT System Requirements . . . . . . . . . . . . . . . . . . . . . .
258
10.2.2 Dual Conversion Superheterodyne Receiver Design . . . . . . . . . .
260
10.2.3 Direct Conversion Receiver Design . . . . . . . . . . . . . . . . . . .
268
10.2.4 Single Conversion Receiver Design . . . . . . . . . . . . . . . . . . .
270
ix
CONTENTS
11 Conclusions and Future Directions
272
References
280
Receiver Architectures and System Requirements . . . . . . . . . . . . . . . . . .
280
Commercial Products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
282
Integrated Receivers/Tranceivers . . . . . . . . . . . . . . . . . . . . . . . . . . .
283
Integrated Active Filters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
284
Filter Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
287
Filter Dynamic Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
288
Spiral Inductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
289
Frequency Synthesis and Oscillators . . . . . . . . . . . . . . . . . . . . . . . . .
291
Electroacoustic Devices, Filters, and Oscillators . . . . . . . . . . . . . . . . . . .
291
General References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
293
Curriculum Vitae
295
x
LIST OF FIGURES
2.1
MC3363 Based 49 MHz Portable Phone. . . . . . . . . . . . . . . . . . . . .
11
2.2
Cellular Phone Block Diagram. . . . . . . . . . . . . . . . . . . . . . . . . .
14
2.3
National Semiconductor DECT Chip Set. . . . . . . . . . . . . . . . . . . .
18
2.4
HP and MOSAIC Microsystems DECT Chip Set. . . . . . . . . . . . . . . .
20
2.5
On-Chip Bandpass Filter Alternatives. . . . . . . . . . . . . . . . . . . . . .
24
2.6
Q-enhanced LC Bandpass Filter. . . . . . . . . . . . . . . . . . . . . . . . .
41
2.7
Phase Noise Comparison of On-Chip Oscillators. . . . . . . . . . . . . . . .
49
2.8
Single-Chip Paging Receiver. . . . . . . . . . . . . . . . . . . . . . . . . . .
57
3.1
Frequency assignments in the VHF to L-band spectrum. . . . . . . . . . . .
59
3.2
Radio frequency environment from 80 MHz to 180 MHz. . . . . . . . . . . .
60
3.3
Superheterodyne receiver architecture. . . . . . . . . . . . . . . . . . . . . .
65
3.4
Received signal levels versus distance from transmitter. . . . . . . . . . . . .
69
3.5
Representative receiver hardware design. . . . . . . . . . . . . . . . . . . . .
72
3.6
Simple nonlinear system model. . . . . . . . . . . . . . . . . . . . . . . . . .
76
3.7
Intermodulation products within a receiver front-end. . . . . . . . . . . . .
78
3.8
Graphical determination of dynamic range. . . . . . . . . . . . . . . . . . .
80
3.9
IF channel select filter attenuation versus frequency. . . . . . . . . . . . . .
86
3.10 Preselect filter attenuation versus frequency. . . . . . . . . . . . . . . . . . .
88
4.1
Tuned RF receiver architecture. . . . . . . . . . . . . . . . . . . . . . . . . .
95
4.2
Early tuned-RF receiver circuit. [British patent no. 147,147] . . . . . . . . .
95
4.3
Regenerative receiver architecture. . . . . . . . . . . . . . . . . . . . . . . .
96
4.4
Super-regenerative receiver architecture. . . . . . . . . . . . . . . . . . . . .
99
xi
LIST OF FIGURES
4.5
Armstrong’s original superheterodyne receiver. . . . . . . . . . . . . . . . .
100
4.6
Modern superheterodyne receiver. . . . . . . . . . . . . . . . . . . . . . . .
101
4.7
Single conversion superheterodyne with tracking preselect. . . . . . . . . . .
103
4.8
Multiple conversion superheterodyne. . . . . . . . . . . . . . . . . . . . . . .
104
4.9
Up-conversion supetheterodyne. . . . . . . . . . . . . . . . . . . . . . . . . .
105
4.10 Ultra-low IF superheterodyne. . . . . . . . . . . . . . . . . . . . . . . . . . .
107
4.11 Image rejection mixer. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
108
4.12 Selective AM demodulator. . . . . . . . . . . . . . . . . . . . . . . . . . . .
110
4.13 Direct conversion receiver architecture. . . . . . . . . . . . . . . . . . . . . .
111
4.14 Simplified digital receiver architecture. . . . . . . . . . . . . . . . . . . . . .
113
4.15 Simplified direct-conversion digital receiver architecture. . . . . . . . . . . .
114
4.16 Ideal receiver architecture. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
117
5.1
Lowpass to bandpass filter transformation. . . . . . . . . . . . . . . . . . . .
122
5.2
Flow diagram representation of filter structure. . . . . . . . . . . . . . . . .
123
5.3
Coupled resonator bandpass filter structure. . . . . . . . . . . . . . . . . . .
124
5.4
Simplified transconductance amplifier. . . . . . . . . . . . . . . . . . . . . .
126
5.5
Practical differential transconductance amplifier. . . . . . . . . . . . . . . .
126
5.6
Gm-C based bandpass filter. . . . . . . . . . . . . . . . . . . . . . . . . . . .
127
5.7
Flowgraph of Gm-C based bandpass filter. . . . . . . . . . . . . . . . . . . .
128
5.8
RLC equivalents in Gm-C resonator design. . . . . . . . . . . . . . . . . . .
129
5.9
RLC resonator equivalent. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
129
5.10 Gm-C filter flowgraph with currents and voltages labeled. . . . . . . . . . .
130
5.11 Equivalent RLC filter flowgraph. . . . . . . . . . . . . . . . . . . . . . . . .
131
5.12 Q-enhanced RLC resonator circuit design. . . . . . . . . . . . . . . . . . . .
132
5.13 Q-enhanced RLC resonator flow diagram. . . . . . . . . . . . . . . . . . . .
132
6.1
Noise models for circuit components. . . . . . . . . . . . . . . . . . . . . . .
137
6.2
Gm-C flowgraph with noise sources included. . . . . . . . . . . . . . . . . .
138
xii
LIST OF FIGURES
6.3
Gm-C based coupled resonator filters. . . . . . . . . . . . . . . . . . . . . .
142
6.4
Q-enhanced resonator flow diagram with noise sources. . . . . . . . . . . . .
147
6.5
Transfer functions to internal nodes in coupled resonator filter. . . . . . . .
154
6.6
Receiver front-ends employing a) passive filter, and b) active filter. . . . . .
155
7.1
Use of cascode design to raise circuit output resistance. . . . . . . . . . . .
160
7.2
Effects of excess integrator phase on resonator design. . . . . . . . . . . . .
162
7.3
Control of frequency and Q in Gm-C resonators. . . . . . . . . . . . . . . .
163
7.4
Control of frequency and Q in Q-enhanced LC resonators. . . . . . . . . . .
165
7.5
Controlling the effective inductance value in Q-enhanced LC resonators. . .
165
7.6
Master-slave frequency tuning. . . . . . . . . . . . . . . . . . . . . . . . . .
166
7.7
Self-tuning technique. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
168
7.8
Correlated Tuning Loop technique. . . . . . . . . . . . . . . . . . . . . . . .
170
7.9
Adaptive Filter Tuning. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
171
7.10 SS-ORT Technique Simplified Block Diagram. . . . . . . . . . . . . . . . . .
173
7.11 SS-ORT Control System. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
175
7.12 Frequency Control Loop Block Diagram. . . . . . . . . . . . . . . . . . . . .
177
7.13 Tunable active filter used for ORT technique prototype. . . . . . . . . . . .
182
7.14 Tunable active filter used for ORT technique prototype. . . . . . . . . . . .
183
7.15 Prototype ORT technique control system circuitry. . . . . . . . . . . . . . .
184
7.16 Prototype ORT technique PRBS generator circuitry. . . . . . . . . . . . . .
185
7.17 Self-tuning simplifications in direct conversion receivers. . . . . . . . . . . .
189
7.18 Self-tuning simplifications in superheterodyne receivers. . . . . . . . . . . .
189
7.19 Orthogonal reference tuning simplifications. . . . . . . . . . . . . . . . . . .
191
8.1
Spiral inductor layout. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
194
8.2
Simplified spiral inductor model. . . . . . . . . . . . . . . . . . . . . . . . .
195
8.3
Detailed circuit model of spiral inductor. . . . . . . . . . . . . . . . . . . . .
197
8.4
Parallel mode Q-enhancement. . . . . . . . . . . . . . . . . . . . . . . . . .
199
xiii
LIST OF FIGURES
8.5
Parallel mode Q-enhanced filter. . . . . . . . . . . . . . . . . . . . . . . . .
199
8.6
Source and load resistances seen by filter. . . . . . . . . . . . . . . . . . . .
201
8.7
Q-enhanced filter circuit design. . . . . . . . . . . . . . . . . . . . . . . . . .
205
8.8
Center-tapped spiral inductor geometry. . . . . . . . . . . . . . . . . . . . .
206
8.9
Center-tapped versus traditional spiral performance. . . . . . . . . . . . . .
208
8.10 Representative balanced filter circuit designs. . . . . . . . . . . . . . . . . .
209
8.11 Frequency fine tuning using capacitively loaded gyrator. . . . . . . . . . . .
214
8.12 Coupled resonator filter architecture. . . . . . . . . . . . . . . . . . . . . . .
215
8.13 Circuit models of magnetic coupling. . . . . . . . . . . . . . . . . . . . . . .
217
8.14 Calculated transfer functions for coupled resonators. . . . . . . . . . . . . .
218
8.15 Simulated coupling coefficients versus inductor separations. . . . . . . . . .
219
9.1
Block diagram of ultra-high Q VHF receiver front-end IC. . . . . . . . . . .
221
9.2
VHF receiver architecture block diagram. . . . . . . . . . . . . . . . . . . .
222
9.3
Input transconductor and negative resistance designs. . . . . . . . . . . . .
223
9.4
Mixer circuit design. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
224
9.5
VHF front-end IC photograph. . . . . . . . . . . . . . . . . . . . . . . . . .
226
9.6
Test circuits for VHF front-end IC. . . . . . . . . . . . . . . . . . . . . . . .
228
9.7
Test circuits for VHF front-end IC. . . . . . . . . . . . . . . . . . . . . . . .
229
9.8
Measured transfer functions of Q-enhanced filter. . . . . . . . . . . . . . . .
230
9.9
Measured Q factor versus control voltage. . . . . . . . . . . . . . . . . . . .
230
9.10 Block diagram of IF filter second-order section. . . . . . . . . . . . . . . . .
234
9.11 Core circuits of IF filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
235
9.12 Switched tuning capacitors and coupling neutralization circuits. . . . . . . .
238
9.13 Amplitude detection circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . .
239
9.14 Photograph of chip layout. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
240
9.15 Test circuits for IF filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
241
9.16 Test circuits for IF filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
242
xiv
LIST OF FIGURES
9.17 Measured |S21| of second-order section at different Q enhancements. . . . .
243
9.18 Measured blocking and spurious-free dynamic range. . . . . . . . . . . . . .
244
9.19 Master/slave Q tuning from 20o C to 50oC. . . . . . . . . . . . . . . . . . . .
245
9.20 Measured fourth-order response. . . . . . . . . . . . . . . . . . . . . . . . .
246
10.1 Effects of phase noise on received signals. . . . . . . . . . . . . . . . . . . .
254
10.2 Dual conversion integrated receiver design. . . . . . . . . . . . . . . . . . . .
261
10.3 Direct conversion integrated receiver design. . . . . . . . . . . . . . . . . . .
269
10.4 Single conversion integrated receiver. . . . . . . . . . . . . . . . . . . . . . .
270
xv
LIST OF TABLES
2.1
Building Block ICs for Radio Receivers. . . . . . . . . . . . . . . . . . . . .
10
2.2
Specifications for 1991 Cellular Phone. . . . . . . . . . . . . . . . . . . . . .
15
2.3
Cellular/PCS Front Ends and Transverters. . . . . . . . . . . . . . . . . . .
17
2.4
Cellular/PCS RF Chip Sets. . . . . . . . . . . . . . . . . . . . . . . . . . . .
19
2.5
Cellular/PCS Baseband Chip Sets. . . . . . . . . . . . . . . . . . . . . . . .
19
2.6
Discrete Filter Performance Benchmarks. . . . . . . . . . . . . . . . . . . .
23
2.7
On-Chip Spiral Inductors in Silicon Processes. . . . . . . . . . . . . . . . . .
27
2.8
On-Chip Electro-Acoustic RF Bandpass Filters. . . . . . . . . . . . . . . . .
28
2.9
Switched-Capacitor RF Bandpass Filters. . . . . . . . . . . . . . . . . . . .
32
2.10 Continuous-Time Gm-C RF Bandpass Filters. . . . . . . . . . . . . . . . . .
34
2.11 Summary of On-Chip RF Bandpass Filter Alternatives. . . . . . . . . . . .
43
2.12 Reported Work in On-Chip Oscillators. . . . . . . . . . . . . . . . . . . . .
48
2.13 Reported Work in Integrated Receivers. . . . . . . . . . . . . . . . . . . . .
51
3.1
Example wireless services and system parameters . . . . . . . . . . . . . . .
62
3.2
Example BDR performance. . . . . . . . . . . . . . . . . . . . . . . . . . . .
74
3.3
Example SFDR performance. . . . . . . . . . . . . . . . . . . . . . . . . . .
83
3.4
Example IF filter complements. . . . . . . . . . . . . . . . . . . . . . . . . .
87
5.1
Filter implementation alternatives. . . . . . . . . . . . . . . . . . . . . . . .
120
6.1
Approximate upper bounds on active filter Q when used in radio receivers.
157
8.1
Simulated inductance, self-resonant frequency, and Q. . . . . . . . . . . . .
198
8.2
Comparison of traditional and center-tapped spiral performance. . . . . . .
210
xvi
9.1
Measured performance of VHF front-end IC. . . . . . . . . . . . . . . . . .
231
9.2
Blocking dynamic range versus frequency offset. . . . . . . . . . . . . . . . .
232
9.3
Spurious-free dynamic range versus frequency offset. . . . . . . . . . . . . .
233
9.4
Measured performance of IF filter second-order section. . . . . . . . . . . .
247
10.1 DECT System Requirements. . . . . . . . . . . . . . . . . . . . . . . . . . .
259
10.2 RF Preselect Filter and LO Performance. . . . . . . . . . . . . . . . . . . .
265
10.3 First IF Filter and LO Performance. . . . . . . . . . . . . . . . . . . . . . .
266
10.4 Second IF Filter Peformance. . . . . . . . . . . . . . . . . . . . . . . . . . .
267
10.5 Overall Receiver Performance. . . . . . . . . . . . . . . . . . . . . . . . . . .
268
xvii
Chapter 1
Introduction
1.1
Scope
The goal of this research is to identify and address problems involved in building complete
radios in integrated circuit form. To contain the scope of the effort, the research is focused
primarily on receiver design in the VHF to L-band spectrum, and on low cost, low power,
silicon IC technologies. The selected frequency spectrum encompasses a wide range of
commercially important radio services including FM and television broadcast, pagers, lowearth-orbit navigation and communication satellites, portable and cellular telephones, and
the rapidly evolving Personal Communications Services (PCS) industry as a whole.
1.2
Problem Overview
At the present time, various manufacturers offer a wide range of IC building blocks for receiver design, including RF amplifiers, mixers, synthesizers, and IF amplifiers/demodulators.
However, completely integrated, high-performance receivers are essentially non-existent.
One of the main challenges facing complete integration of receiver hardware has been a lack
of suitable on-chip RF and IF filtering. Thus, a major portion of this dissertation is focused
on the problem of realizing high performance, integrated bandpass filters. An additional
obstacle to full integration is frequency synthesis. Although many “complete” synthesizer
IC offerings can be found, a closer examination reveals that the VCO and tuned circuits for
1
the synthesizer are not included on-chip. This problem is also considered and is shown to
be closely related to the realization of bandpass filters.
Issues surrounding receiver integration are addressed in this dissertation from both a systems and a circuits perspective. The problem faced is ultimately a circuits problem, but the
degree to which any given circuit approach is successful must be assessed in the larger context of the host receiver system and its intended function and environment. This introduces
an additional dimension to the research – a study of system requirements and an assessment of alternative receiver architectures. Indeed, a review of work on integrated receiver
design over the past two decades shows that alternative architectures have been extensively
applied. These architectures can, to some extent, reduce the need for high-performance RF
bandpass filters. However, dynamic range, image rejection, and power consumption considerations make the total elimination of RF bandpass filtering impractical in many wireless
services. This conclusion, addressed in Chapter 4, is a major reason for the emphasis on
integrated RF filtering in this dissertation.
1.3
Dissertation Outline
Chapter 1 provides a statement of scope, an introduction to the problem addressed, and an
outline of the structure of the dissertation.
The study of integrated receiver design begins in Chapter 2 with a survey of the state-of-theart in both research and commercial products. Commercial integrated circuits developed
over the past three decades are reviewed and cellular and PCS chip sets currently being
introduced are discussed. This material is followed by a comprehensive survey of the literature illustrating the technical approaches taken and the levels of performance achieved
by researchers working on integrated filters, integrated synthesizers, and fully integrated
receivers. Published work by the author is included in this chapter to provide a continuity
2
of discussion, and to serve as a preview of more detailed discussions in subsequent chapters.
Having examined current technology and research efforts in Chapter 2, Chapter 3 turns
to the issue of receiver system requirements. The purpose of this chapter is to identify
the level of technical performance that integrated receiver technology must provide to be
commercially viable. The subject of receiver system performance is introduced through
an examination of the spectrum environment in which receivers are required to operate.
Commercial wireless services which can benefit from low-power, integrated receiver technology are then identified, and system parameters that affect receiver design are summarized for each service. Finally, receiver performance measures including sensitivity, dynamic
range, selectivity, and fidelity are quantified, and system level performance requirements are
mapped to requirements on integrated filters.
Chapter 4 continues the investigation of system level issues by examining receiver architecture alternatives. A historical overview of receiver development is presented first, showing
the origins of the superheterodyne receiver and the implicit assumptions about amplifier and
filter technology on which this architecture is based. Variants of the superheterodyne architecture are then investigated in detail, with emphasis on identifying fundamental bounds
and practical limits to the performance that can be achieved. Finally an “ideal” low-power
receiver architecture is derived, and practical considerations regarding its viability are examined.
Throughout the study of receiver architectures in Chapter 4, the importance of high dynamic
range bandpass filtering is illustrated, setting the stage for the study of these filters in subsequent chapters. Chapter 5 thus returns to the investigation of circuit level filter design issues
begun in Chapter 2. The major categories of on-chip filter implementation technologies are
reviewed and implementation issues are discussed. Although on-chip electro-acoustic filters
are found to show some promise, their construction requires fabrication technology that
is beyond the scope of this research. Hence, discussion of these filters is left primarily to
3
the work cited in the references. Within the remaining filter technologies, Gm-C and Qenhanced LC designs are identified as the most viable for implementation at IF and RF
frequencies.
The two main problems with these filters, dynamic range limitations and the need for realtime tuning are then studied in Chapters 6 and 7 respectively. In the discussion of dynamic
range, it is demonstrated that Q-enhanced LC filters possess a significant advantage over
Gm-C and other fully active filter designs operating at comparable power levels. In the
discussion of filter tuning, the sensitivities of filter center frequency and Q to manufacturing tolerances and temperature effects are investigated and the need for real-time tuning
is demonstrated. The major control system designs that have been proposed in the literature are then reviewed and a new “orthogonal reference tuning” technique is presented.
The concept of “keep-it-simple” (KIS) tuning, wherein minimum complexity and cost are
considered important goals, concludes the chapter.
Based on the advantages of Q-enhanced LC filters identified in Chapters 5 - 7, Q-enhanced
filter design is selected as the technology of choice. Chapter 8 provides a detailed study
of factors affecting the realization of these devices in integrated circuit form. Particular
emphasis is placed on the performance of on-chip inductors on which the filters are based,
and on the issues of Q enhancement, frequency and Q stability, and higher-order filter
design.
Chapter 9 describes the design and performance of two prototype integrated circuits fabricated in the course of this research. These ICs were developed to demonstrate the viability
of this new area of filter design, and to validate theoretical performance predictions in earlier chapters. The first IC includes a Q-enhanced, second-order filter which relies on off-chip
inductors and varactor tuning diodes. This chip was designed primarily to validate dynamic
range predictions and to investigate filter tuning requirements. However, coupled with an
on-chip mixer, the device provides the basis for an ultra-high-Q, regenerative, VHF receiver
4
front-end. Dynamic range performance of this filter, which draws less than 1.5 mA and
provides a gain of over 30 dB, is shown to be suitable for demanding applications such as
low-earth-orbit satellite receivers. The second IC is a two pole (fourth-order) Q-enhanced
filter with on-chip inductors and tuning capability. This filter, which operates at a frequency of 200 MHz and a selectivity Q of 100, is targeted at the first IF section of modern
cordless and cellular phones. The filter draws less than 6 mA when operating from a 3
V supply, and consumes less than 3.3 mm2 of IC area, making it a viable alternative to
off-chip electro-acoustic filters currently used in these products.
Having demonstrated the viability of on-chip bandpass filters using Q-enhanced LC technology, Chapter 10 presents a preliminary study of a fully integrated receiver design. The
remaining problem of integrating local oscillators is investigated, followed by the development and performance analysis of a receiver architecture suitable for use in future wireless
products. Finally, Chapter 11 concludes the dissertation with a summary of results and a
list of key research areas for further investigation.
5
Chapter 2
Background and Literature Review
This chapter surveys reported work in the area of integrated receiver design. The survey is
organized into the following main sections:
• Commodity ICs
• Integrated bandpass filters
• Integrated synthesizers
• Integrated receivers
Throughout this chapter, published work by the author is cited together with discussions of
work by previous researchers. This is done where appropriate to provide a more complete
treatment, and to help introduce work covered in subsequent chapters.
The first section of this chapter presents a sampling of commercially available, integrated
circuit receiver building blocks. These blocks include RF amplifiers, mixers, IF and AGC
amplifiers, frequency synthesizers, and demodulators – plus devices that incorporate several
of these functions in a single chip. Absent from this list are bandpass filters and local
oscillator tuned circuits. A look at modern receiver designs in this section shows that these
functions are currently implemented off-chip in the form of discrete LC, ceramic, crystal,
or surface acoustic wave (SAW) filters and resonators.
6
Section 2.2 reviews the extensive body of research on integrated filter design conducted
over the past three decades. Emphasis is placed on bandpass filters suitable for use in
radio receivers. Performance figures of reported designs are tabulated and discussed. The
important problems of power consumption, filter tuning, and dynamic range are introduced
and the technologies available for implementing on-chip filters in different frequency ranges
are considered.
Section 2.3 reviews research into fully integrated frequency synthesizers, with concentration
on high quality oscillator designs. Here the most important problems are found to be phase
noise and spurious signal levels. VCOs for phase locked loop architectures, as well as direct
digital synthesis (DDS) are considered.
Finally, Section 2.4 discusses implementations of partially or fully integrated receivers. In
the commercial sector, application specific integrated circuits (ASICs) have been designed
for selected markets with varying levels of integration. Reported designs for broadcast
receivers and pagers are reviewed. In these radio services, consumer demand is sufficiently
high to justify designing for a fixed frequency band and information bandwidth. Thus, onchip bandpass filtering becomes practical from an economic viewpoint. However, we find
that for technical reasons these ASIC designs depart from the classical superheterodyne
receiver architecture in order to avoid or minimize the demand for RF and IF bandpass
filtering. The extensive work on integrated bandpass filtering discussed in Section 2.2
is thus seen to have found limited use in commercial impementations. The reasons and
possible solutions to this technology transfer problem will be looked at in depth throughout
this dissertation.
7
2.1
Commercial Integrated Circuits
One approach to assessing the state of the art in receiver integration is to conduct a study
of commercial wireless products and the IC components from which they are built. Such a
study gives an indication of what is feasible with current manufacturing technology, subject
to business and market constraints. In this section, we present an overview of commercial
IC devices offered at the time of this writing (1Q95), with emphasis on those ICs applicable
to the design of high volume products, including radio and television broadcast receivers,
pagers, portable and cellular telephones, and the evolving PCS digital wireless market.
Commercial ICs can be divided into two broad classifications:
• Commodity ICs
• Application Specific ICs (ASICs)
This section looks exclusively at commodity ICs, defined here as devices offered for sale
by IC manufacturers to manufacturers of wireless products and systems. These ICs are
usually intended for wide market penetration, and consequently often adopt a “building
block” approach to system design. Information on these devices is widely available in data
books and in trade journal advertisements and new product announcements.
In contrast to commodity ICs, ASICs are devices targeted at a single application and as
such, may allow a higher level of integration. Information on ACICs is more difficult to
find, but is often available in research papers. These devices will be covered in Section 2.4
where integrated receiver research is reviewed.
8
2.1.1
Receiver Building Blocks
A summary of commodity ICs introduced over the past three decades is presented in Table
2.1.
Beginning in the late 1960s and early 1970s, IC technology had advanced sufficiently that
relatively large sections of products such as televisions could be economically integrated
onto a single chip. Receiver ICs, including the MC1350 IF amp, CA3089 IF subsystem,
CA3065 sound circuit, and MC1310 FM stereo demodulator represented the first steps
toward reducing the part count, size, and cost of these products.
By the late 1970s and early 1980s, so-called “single-chip” broadcast receiver ICs such as the
National Semiconductor LM1868 were being produced. The LM1868 provides all the active
circuitry necessary for an AM receiver, plus most of the active circuits needed for an FM
receiver. However, the RF amplifier, local oscillator (LO), and mixer for the FM receiver
section must be implemented with off-chip discrete transistors – providing higher gain and
better noise figure performance than that obtainable with on-chip devices at the time the
IC was introduced. Beyond these discrete transistors and their bias circuitry, additional offchip components required include bandpass filters, LO tank circuits, the FM demodulator
phase shift network, and various coupling and bypass capacitors.
Toward the middle of the 1980s, significant new markets began to open up largely due to
growth in use of personal pagers and in portable and cellular telephones. The Signetics
NE/SA615 FM IF subsystem, targeted at these markets, represents a modern derivative of
the earlier generation CA3089 IF subsystem. Although the RF amplifier, tuned circuits,
and filters remained off-chip in this device, the NE/SA615 integrated numerous functions
including a low noise mixer with capability extending to 500 MHz, a local oscillator transistor, an IF amplifier/limiter, a quadrature detector, muting, and a high dynamic range
received signal strength indicator (RSSI). In addition, the operating voltage and current
9
Table 2.1: Building Block ICs for Radio Receivers.
Manufacturer
Part #
Function
Applications
Features
Motorola
MC1350
IF Amp
90 MHz bandwidth, AGC with 60 dB
range, 50 dB gain at 45 MHz, 12V
RCA
CA3065
Sound Circuit
Television,
general
purpose
Television
RCA/Harris
CA3089
FM IF
Motorola
MC1310
Stereo Demod
Signetics
NE/SA615
FM IF
VHF/UHF receivers, cellular IF, general
purpose
National
Semiconductor
LM1868
AM/FM Receiver
AM/FM
broadcast
Motorola
MC3363
VHF FM Receiver
49
MHz portable
phone, pagers,
VHF radios
Motorola
MC4044
Plessey
SP4653
Phase/Frequency
Detector
Prescaler
Motorola
MC44802
PLL Synthesizer
PLL
synthesizers
PLL
synthesizers
Television
tuners
National
Semiconductor
LMX1501A
PLL Synthesizer
Cellular/PCS
Harris
HSP45102
DDS Synthesizer
Frequency
hopped PCS
FM broadcast,
FM
2-way, general
purpose
FM Broadcast
10
4.5 MHz subcarrier IF amp/limiter, detector, audio amp, and volume control,
12V
10.7 MHz IF amp/limiter, detector, signal level detector, audio amp, muting/squelch, AFC amp, 8.5 - 16V
Baseband L-R subcarrier demodulation, L/R audio matrixing, stereo indicator, 8 - 14V
500 MHz downconversion mixer, local
oscillator, IF amp/limiter, quadrature
detector, muting, and RSSI. 500 MHz
mixer RF bandwidth with 5 dB NF at
45 MHz and 18 dB gain, 25 MHz IF
bandwidth with 102 dB gain, 90 dB
RSSI range, 4.5 - 8V
AM: RF/AGC amp, mixer, LO, IF
amp, detector; FM: IF amp/limiter,
quadrature detector, muting, meter
drive; audio preamp and power amp,
3 - 15V
Dual conversion receiver including 450
MHz RF amplifier transistor, first
mixer and LO with tuning diode, second mixer and LO, IF amp/limiter,
quadrature detector, muting, FSK
comparator, RSSI, 2 - 7V at 3.6mA excluding RF transistor
Linear phase/frequency detector, lock
detection, charge pump, 5V
1.1 GHz divide-by-256, 10 mV input
sensitivity, 5V
1.3 GHz divide-by-8 prescaler, programmable
15bit divider, programmable reference divider, crystal reference oscillator, tristate phase/frequency detector, tuner
bankswitching drivers, serial programming, 5V, 60mA
1.1 GHz synthesizer similar to above,
but with dual-modulus prescaler, 2.7 5.5V, 6mA
40 MHz clock, up to 20 MHz output
with 0.009 Hz resolution, 5V, 99mA
consumption of the device were lower than that of the CA3089, reflecting growing market
demand for smaller and lighter weight portable equipment.
One of the highest levels of integration in commodity ICs available today is represented
by the Motorola MC3363 VHF FM receiver introduced around 1990. Shown in a 49 MHz
portable phone application circuit in Figure 2.1, this device is similar in many respects
to the NE/SA615, but adds several key components including an RF amplifier transistor,
an additional mixer, a varactor tuning diode, and a data slicer (comparator) for FSK bit
decisions. However, in common with all other commodity ICs, preselection/image filtering,
channel select filtering, quadrature detection phase shifter, and tuned circuits for the local
oscillators must still be implemented off-chip with discrete components.
Figure 2.1: MC3363 Based 49 MHz Portable Phone. Copyright of Motorola. Used by
permission.
The lower half of Table 2.1 lists several ICs used in the tuning function of radio receivers.
11
In early receivers, this tuning was provided by manually tuned variable capacitors, or for
the case of television receivers, by mechanical switching of inductors. Today, most receivers
(except very low cost AM/FM broadcast units) rely on digitally programmed frequency synthesizers. Modern tuners offer crystal controlled frequency stability, free from adjustment,
plus enhanced ease of use and additional features including search/scan.
The first frequency synthesizers were constructed from a set of phase locked loop (PLL)
building blocks such as the Plessey SP4653 prescaler and Motorola MC4044 phase detector, together with suitable TTL or CMOS programmable dividers. In modern receivers,
these functions have been integrated into “single-chip” PLL synthesizer ICs such as the
Motorola MC44802 for television tuners and the National Semiconductor LMX1501A for
cellular radios. These devices offer most functions needed for designing a digitally tuned
LO controlled by keyboard entry or a receiver’s microprocessor. However, like the receiver
ICs discussed previously, the tuned circuits required for the PLL’s VCO remain off-chip.
An interesting alternative to PLL synthesizers is offered by the relatively new direct digital
synthesis (DDS) ICs. An example device is listed at the end of Table 2.1. DDS devices
do not require tuned circuits (except for a crystal reference) and would therefore appear
to bypass the need for on-chip VCOs in integrated receivers. Unfortunately however, the
output frequencies of DDS devices are comparatively low and must be multiplied or mixed
up to VHF through L-band for use as a receiver’s first LO, usually involving the use of
auxiliary PLL synthesizers and tuned circuits. In addition, current devices are relatively
expensive ($20 or more) and consume substantial power (e.g. > 100 mW).
2.1.2
Cellular Telephone Design Example
Modern hand-held cellular telephones represent one of the most sophisticated and challenging areas of wireless circuit design today. Hence, it is worthwhile studying the architecture
and level of integration currently being achieved in these products. This section overviews
12
the design of a 1991 vintage cellular phone marketed by the Radio ShackT M division of
the Tandy Corporation in the United States.1 The phone operates in the North American
Advanced Mobile Phone System (AMPS) service at 800 MHz, providing synthesized tuning
of 832, 30 kHz wide channels. Frequency division duplex is employed with a 45 MHz split
between transmit and receive subbands. Voice modulation is analog FM with ± 12 kHz
deviation, and control data are sent using 20 k baud Manchester encoded FSK. The phone
is approximately 7 x 2.2 x 1.4 inches in size and sold for around $430 (without service
activation) in 1994. It contains two PC boards, each measuring 6.8 x 2 inches. One board
performs the analog RF/IF functions, while the other provides microprocessor control, user
interface, and baseband audio processing. The phone is powered by six NiCad batteries
delivering a nominal voltage of 7.2V and a current capacity of 700 mAh. Performance
specifications for this phone are summarized in Table 2.2 [34], and a block diagram of the
RF/IF board is shown in Figure 2.2.
The RF/IF PC board connects the phone’s antenna to a duplexer formed by two ceramic
bandpass filters, one centered on the receive subband and one on the transmit subband.
Following the duplexer, receive signals are amplified by a discrete transistor LNA and fed
through a second ceramic filter to achieve the required image frequency rejection levels.
The first down conversion is provided by a double balanced passive mixer, converting the
received signal to a 45 MHz IF where it is then amplified by a second discrete transistor LNA
and filtered by a 4-pole (8th order) ladder filter composed of two cascaded monolithic crystal
filters (MCFs). This filter narrows the receive bandwidth sufficiently to enable the second
IF to be implemented at a comparatively low 455 kHz. A Signetics SA615 is employed to
provide the second down conversion mixer, 455 kHz IF amplification, limiting, quadrature
detection, and signal strength monitoring. An off-chip 44.45 MHz crystal provides the
local oscillator tank circuit and an off-chip 455 kHz ceramic BPF is used for final channel
selection. Baseband audio is delivered to the adjacent PC board for FSK data decisions
1
Radio Shack is a registered trademark of the Tandy Corporation.
13
Figure 2.2: Cellular Phone Block Diagram. Copyright 1991 Tandy Corporation. Reprinted
by written permission, from [34].
14
Table 2.2: Specifications for 1991 Cellular Phone.
Parameter
Batteries
Minimum Voltage
Current Consumption
Receive
Transmit
Temperature
Number of Channels
Frequency Stability
FM Deviation
Sensitivity
RSSI Dynamic Range
Selectivity
Adjacent channel
Alternate channel
Image Rejection
Receiver Intermodulation
Value
7.2 V nominal @ 700 mAh
6.5 V
50 mA (360 mW)
550 mA (4 W)
-30o to + 65o C
832
± 2.5 ppm
± 12 kHz
-116 dBm (12 dB SINAD)
> 60 dB
>
>
>
<
15
16 dB
65 dB
60 dB
-26 dB
and voice band filtering and amplification.
Tuning is performed on the RF/IF board using two MB1502 synthesizer ICs (one for the
transmit frequency and one for the receive frequency), with discrete component loop filters,
VCOs, and buffer amplifiers. The FM transmit signal is formed by phase modulating
the transmitter synthesizer VCO, and then amplifying the signal with a series of seven
discrete transistors. The final two transistors operate in class C and harmonic suppression
is provided by microstrip impedance matching networks and by the transmit portion of the
antenna duplexer.
In all, the RF/IF circuit board contains 4 ICs and 344 discrete components - representing
a relatively low level of integration. Thus, the phone’s small size is due primarily to the use
of fine geometry PC board construction (10 mil traces), and of low profile surface mount
components.
2.1.3
Cellular/PCS Front End, Power Amplifier, and Transverter ICs
In the years since the cellular phone in the previous section was designed, several new
ICs have become available, enabling further size and weight reductions in newer products.
These ICs, summarized in Table 2.3, provide integrated solutions to the design of receiver
front ends (LNAs and mixers), transmitter power amplifiers and, in the case of newer time
division duplex (TDD) services such as DECT, transmit/receive (T/R) switching.
The products listed in Table 2.3 span both the existing cellular phone frequencies and the
newly allocated spectra from 900 MHz to 2.4 GHz. Several products, such as the NEC
UPC2757T downconverter and Siemens CMY90 front end ICs offer broadband operation
across this full spectrum, but at the cost of requiring external matching networks. Most
newer products are targeted at particular spectrum segments (e.g 900 MHz, 1.8 GHz, or 2.4
GHz) and provide on-chip 50 Ohm impedance matching for reduced external part count.
16
Table 2.3: Cellular/PCS Front Ends and Transverters.
Manufacturer
Part #
Function
Technology
Features
NEC
UPC2757T
Downconverter
Silicon
Siemens
CMY90
Front End
GaAs
Triquint
TQ9203J
Front End
GaAs
Motorola
Ref: [44]
Front End
GaAs
Philips
SA620
Front End
Silicon
Motorola
MRFIC2401
Front End
GaAs
AT&T
Ref: [41]
Transverter
GaAs
0.1 - 2 GHz mixer with 10 - 300 MHz IF, local
oscillator, IF amp, 13 dB gain, 3V, 5.6mA
0.4 - 3 GHz LNA, mixer, IF amp, 4.5 dB NF with
15 dB gain at 900 MHz, 3V, 2.5mA
0.8 - 1 GHz, 2 LNAs with switch for antenna
diversity, mixer with 45 - 200 MHz IF, 2.7 dB
NF with 20 dB gain, -10 dBm IIP3, 5V, 10.5mA
0.7 - 0.9 GHz, RF amp, mixer, IF amp, 50 Ohm
matching at RF, LO ports, 3.6V, 2.7mA
1.2 GHz, LNA, mixer, VCO, gain switching,
14.5/-4.5 dB gain with -18/+1.5 dBm IIP3, 50
Ohm RF port, 2.7 - 5.5V, 10mA
2.0 - 2.8 GHz, LNA, mixer, 3dB NF with 21 dB
gain, -15 dBm IIP3, 50 Ohm RF/LO ports, 5V,
10mA
2.4 GHz RF to 915 MHz IF half-duplex conversion, T/R switch, LNA, mixer, and LPF
in receive path; mixer, amp with power control/detect, harmonic suppression filtering, T/R
switch in transmit path; shared VCO with onchip tank circuit – excluding varcap; 0.25W output; ± 5V
Impedance matching is provided by on-chip LC networks incorporating spiral inductors.
A typical example described in [46], employs an L-type network to convert 600 Ohms to
50 Ohms with a loaded Q of 1.7. This low-Q circuit provides good matching across the
desired band, but introduces a bandpass response restricting the IC to the targeted spectrum
segment (the 900 MHz ISM band in the case of this IC). On the plus side, the bandpass
response can be helpful in reducing transmitter harmonic and spurious outputs, and in
receivers it provides attenuation to out-of-band signals. However the Q is not sufficient in
any of the devices to serve as preselection/image filters and off-chip filters are still required.
This fact is clearly illustrated in the next section when modern “chip sets” for cellular and
PCS wireless products are reviewed.
17
2.1.4
Cellular/PCS Chip Sets
The state of the art in cellular and PCS RF integration is illustrated by chip set offerings currently being introduced. These chip sets are targeted at emerging digital wireless
markets such as Digital European Cordless Telecommunications (DECT), Groupe Special
Mobile (GSM) and its 1.8 GHz derivative (DCS1800), North American Digital Cellular
(NADC) IS-54, and Code Division Multiple Access (CDMA) systems such as IS-95. Table
2.4 summarizes ICs targeted at the analog portion (both transmitter and receiver) of products in these markets, while Table 2.5 summarizes some of the baseband processing chips
becoming available.
The National Semiconductor DECT chip set illustrated in Figure 2.3 is representative of
the level of integration being achieved [28]. The chip set includes the LMX2215/16 receiver
LNA and downconversion mixer, the LMX2240 IF amplifier/limiter/detector/RSSI, the
LMX2410 baseband data processor, and the LMX2320 synthesizer IC and is expected to
sell for under $30 in quantity.
LMX2240
LMX2215/16
LNA
BPF
LMX2410
BPF
LPF
d
BPF
RSSI
LMX2320
T/R
PLL
Baseband
Processor
To / From
Microprocessor
LPF
T/R
PA
Figure 2.3: National Semiconductor DECT Chip Set. After Eccles [28].
An alternative chip set targeted at DECT but not yet commercially available is shown in
Figure 2.4 [39]. This chip set, developed by Hewlett Packard and MOSAIC Microsystems
adopts a different tranceiver architecture and partitioning of functions, but represents a
similar level of integration. Although the T/R switch, LNA, and PA are not part of the
set, these functions are available from other manufacturers as seen in the previous section.
18
Table 2.4: Cellular/PCS RF Chip Sets.
Manufacturer
Application
Technology
Partitioning
Philips
FM Cellular
Silicon
National
Semiconductor
DECT
Silicon
SA620 – Front end (LNA, mixer, VCO)
SA615 – IF (mixer, amp/limiter, detector, RSSI)
SA7025DK – Synthesizer (prescaler, dividers,
phase detector)
LMX2215/16 – Front end (LNA, mixer)
HP & MOSAIC
Microsystems
DECT
Silicon
Philips
GSM/DCS1800
Silicon
RF Micro Devices
CDMA/FM Cellular
GaAs
Qualcomm
LMX2240 – IF (amp/limiter, detector, RSSI)
LMX2320 – Synthesizer (prescaler, dividers,
phase detector)
LMX2410 – Baseband processor (demod comparator with threshold D/A, RSSI A/D, digital
FIR gaussian filter and D/A for transmit data)
Transverter (T/R mixers, VCO, frequency doubler, prescaler)
IF (LO, mixer, IF amp/limiter, detector, RSSI,
data detector)
SA1620 – Transverter (LNAs and mixer with 60
dB gain control in receive path; IF level control,
SSB mixer, and PA driver in transmit path)
SA1638 – IF (I/Q mixers for transmit and receive,
fixed frequency synthesizer with quadrature LO
output)
UMA1019 – Synthesizer (prescaler, dividers,
phase detector)
RF9906 – Front End (LNA, mixer, IF amp)
RF9907 – Receive IF (IF amp with 90 dB AGC
range)
RF9909 – Transmit IF (IF amp with 84 dB gain
adjust)
RF9908 – Up Converter (Mixer, buffer amps)
BBE – Mod/demod (Fixed frequency synthesizer with VCO, mixers for I/Q modulation/demodulation, D/As and A/Ds)
Silicon
Table 2.5: Cellular/PCS Baseband Chip Sets.
Manufacturer
Application
Partitioning
Philips
GSM/DCS1800
Qualcomm
CDMA/FM Cellular
Texas Instruments
NADC (IS-54)
PCD5072 – ADCs for I/Q detection, DSP, audio D/A and amp for
receive; DSP, GMSK modulator, I/Q A/Ds for transmit; AGC and
AFC D/As
PCF5083 – Speech coding/decoding, channel coding/decoding, encryption/decryption, burst buffering, TDMA timing
MSM2 – CDMA and FM demods, viterbi decoder, voice coding, data
interleaving/deinterleaving
— – Audio A/D and D/As
TCM4300 – Data demodulation, timing, and power management
TMS320IS54B – Voice and channel coding/decoding
TVL320AC3X – Audio A/D and D/As and amplifiers
19
In addition, the CMOS synthesizer shown in Figure 2.4 is readily available elsewhere.
Figure 2.4: HP and MOSAIC Microsystems DECT Chip Set. Copyright 1993 IEEE.
Reprinted with permission, from [39].
From these examples, we note that the following functions must still be implemented off-chip
to form a complete tranceiver product:
• Bandpass filters for RF preselection/image filtering
• Bandpass filters for channel selection at first and second IFs
• Tank circuits for first and second LOs
• Quadrature detection phase shift network
• Lowpass data filters
Examination of chip sets for other cellular and PCS markets listed in Table 2.4 and of
20
associated tranceiver designs [25] [38] [31], yields the same conclusions – as of 1995, virtually
all analog portions of a tranceiver are available in a small number of ICs, with the exception
of high frequency filters and tank circuits.
2.1.5
Discussion
Given current market pressures to reduce the size and cost of wireless products, it is reasonable to ask why these key components have so far not been integrated with the remainder of
the receiver functions. For wireless commodity ICs, the reasons may be either economically
or technically based.
From an economic viewpoint, including filters and tuned circuits on-chip can have a negative
effect on the breadth of the market that a commodity IC can penetrate. For example, if
a front-end chip targeted at the cellular market is restricted to the US band of 869 894 MHz by an on-chip preselection/image filter, that chip becomes unsuitable for use in
other markets such as Europe where GSM cellular is implemented at 935 - 960 MHz. The
commodity IC manufacturer is then forced to weigh the costs associated with producing
multiple versions of the chip against the benefits of higher integration – smaller end product
size, plus capture of a portion of the filter vendor’s market.
However, for other cases, such as IF filters in broadcast receivers, where the operating frequency and bandwidth are standardized, we would expect integration to be fully warranted
from an IC manufacturer’s perspective. Yet, broadcast receiver commodity ICs still rely on
off-chip ceramic IF filters purchased from separate vendors. In this case it appears clear
that technical reasons must be at least partly involved. These technical considerations
are investigated in the following sections where research into on-chip bandpass filters and
oscillators is reviewed.
21
2.2
Integrated Bandpass Filter Research
Research into bandpass filters for radio applications dates to the very beginning of electrical
engineering. Research on integrated RF bandpass filters however, is comparatively recent,
with work beginning around the 1960’s [59] [79]. In this section we attempt to organize
this area of research into a coherent picture and to assess why these filters have so far
seen little or no commercial applications. We begin with a look at the competition – offchip ceramic, crystal, SAW, and LC filters. Following this, we look at on-chip filtering
alternatives including:
• Digital filters,
• Analog passive filters (including electro-acoustic and LC types),
• Analog active filters (including switched-capacitor, and Gm-C), and
• Q-enhanced LC filters.
The important topics of filter tuning and dynamic range are also considered.
2.2.1
Filter Performance Benchmarks
In order for an integrated bandpass filter to see commercial application, it must be both
technically and economically competitive with the discrete filters which it is to replace.
Table 2.6 lists performance data for some discrete ceramic, crystal, SAW, and LC filters
available today. These filters span the range of IF and RF frequencies from 262 kHz to 914
MHz. All of the filters listed except three (the 881 and 914 MHz filters) are intended for
use as channel select filters at standardized IF frequencies. The 881 and 914 MHz filters are
RF preselect/image filters intended for use at the front end of cellular and cordless phones.
22
Table 2.6: Discrete Filter Performance Benchmarks.
Part #
Type
Application
Bandwidth
Shape Factor
AM Broadcast
IF
AM Broadcast
IF
Pager IF
262 kHz
Freq
6 kHz (2.3%)
455 kHz
6 kHz (1.3%)
450 kHz
6 kHz (1.3%)
Television
Sound IF
FM Broadcast
IF
Cellular Phone
IF
DECT IF
4.5 MHz
120
kHz
(2.7%)
230
kHz
(2.1%)
25
kHz
(0.2%)
1.1 MHz (1%)
-16 dB @ ± 9
kHz
-16 dB @ ± 9
kHz
-40 dB @ ±
12.5 kHz
-20 dB @ ± 270
kHz
-20 dB @ ± 290
kHz
-40 dB @ ± 25
kHz
-20 dB @ ± 1.5
MHz
-20 dB @ ± 78
MHz
TOKO
HCFM8-262B
TOKO CFMR455B
MuRata
SFP450F
MuRata
SFE4.5MBF
MuRata
SFE10.7MS2-Z
ECS ECS-10.715B
Siemens B4535
Ceramic
MuRata
LFC3001B0881B025
Toko
6DFA-881E-11
Toko
6DFA-914A-14
LC
Cellular RF
881 MHz
25
MHz
(2.8%)
Dielectric
Cellular RF
881 MHz
Dielectric
Cordless Phone
RF
914 MHz
25
MHz
(2.8%)
1 MHz (0.1%)
Ceramic
Ceramic
Ceramic
Ceramic
MCF
SAW
10.7 MHz
10.7 MHz
110 MHz
-20 dB @ ± 78
MHz
-24 dB @ ± 45
MHz
IL
Price
6 dB
$1
6 dB
$1
6 dB
–
6 dB
–
6 dB
$0.30
2.5 dB
$3
–
$3
3.5 dB
–
1.8 dB
–
2.2 dB
–
Some general conclusions that can be drawn from this table include:
• Fractional bandwidths are small (1% to 3%, with 0.1% possible),
• Shape factors are moderate (16 to 20 dB attenuation at 2 to 3 times the nominal
bandwidth), but can be higher for cascaded designs such as the SFP450F,
• Insertion loss is moderate (1.5 to 6 dB)
• Cost is low ($0.30 to $3) when purchased in high quantities
The small fractional bandwidth of these filters minimizes the number of IFs required in a
receiver, which helps to minimize overall receiver cost. The moderate shape factors provide
acceptable selectivity for some applications, although two of these filters will frequently
be used in cascade to achieve a better alternate channel selectivity. (Note however that
adjacent channel selectivity may still be relatively low as seen in the specifications for the
23
cellular phone in Table 2.2.) The relatively low insertion loss of these filters minimizes
the amount of RF/IF amplification required preceding the filter, thereby improving the
receiver’s noise figure and intermodulation dynamic range. Finally, the cost of the filters
helps to explain their nearly universal use in consumer products. This cost also hints at
one reason integrated filters have not yet replaced these discrete devices. At $1 per filter,
only a very small fraction of chip area can be used by an on-chip filter if it is to be cost
competitive.
2.3
On-Chip Filter Alternatives
A wide range of technologies exists through which filters may conceivably be implemented
on-chip. These technologies, ranging from digital signal processors (DSP) to active analog
designs and electro-acoustic implementations, are summarized in Figure 2.5 and examined
in the following sections.
On-Chip Bandpass Filters
Digital Filters
Analog Filters
Passive
LC
Electro-Acoustic
Active
Switched
Capacitor
Gm-C
Q-Enhanced
LC
Figure 2.5: On-Chip Bandpass Filter Alternatives.
24
2.3.1
Digital Filters
In theory, DSP could be used to implement any of the filters shown in Table 2.6. In practice,
however, current technology rules out this possibility for all but perhaps the lowest frequency
( < 1 MHz ) devices. Practical problems with the DSP filtering approach include:
• the need for analog anti-aliasing filters,
• chip area requirements,
• electromagnetic compatibility with low level analog signals,
• requirements for a high resolution, high speed analog to digital converter (ADC), and
• power consumption at high frequencies.
If we neglect all of these problems except the last, we still find that the DSP solution is
generally unsuitable for all but the lowest frequency applications. For integrated receiver
designs within the scope of this dissertation, low power consumption is a requirement. Based
on data in Tables 2.1, 2.3, and 2.4, we conclude that any solution drawing more than a few
milliamps will not be greeted favorably by designers of portable equipment.
In a DSP solution, the two major power consumers will be the ADC and the DSP computational circuitry, both of which increase consumption with frequency of operation. Vittoz
[182] has developed bounds on the power consumption of DSP circuitry (excluding the
ADC) for lowpass filters as a function of dynamic range requirements. For a 1 um, 3V
digital process and 60 - 80 dB dynamic range (corresponding to 10 - 13 bit resolution), he
computes a power consumption on the order of 1 nJ per pole, per Hz of corner frequency.
To apply these results to bandpass filters, we must use the center frequency of the filter.
As an example, a 10 MHz, fourth-order IF filter requires (1nJ/pole)(4poles)(107Hz) = 40
mW. In a 3 V process, this corresponds to 13 mA. Cascading two such filters for improved
25
alternate channel rejection raises the figure to 26 mA at 10 MHz. Hence, even without considering power consumption of the ADC, the need for an anti-alias filter, electromagnetic
compatibility with small signals, or chip area requirements, the DSP solution is not suitable
at frequencies of 10 MHz and above.
While it can be argued that power consumption will decrease for DSP circuits as smaller
geometries become available, these improvements will be gradual and will not change the
general conclusions reached here in the near future [182]. Therefore, in the remainder of this
dissertation we will exclude digital solutions and look exclusively at the remaining analog
circuit options in Figure 2.5.
2.3.2
Analog Passive Filters
Within certain constraints, it is possible to implement completely passive on-chip filters.
As shown in Figure 2.5, the two basic technologies are the same as those used to realize the
most popular off-chip filter types – LC filters and electroacoustic filters. Both types have
been realized in silicon IC processes [167] [60] [131], although with performance well below
that currently possible off-chip.
Passive LC Filters
On-chip capacitors can be fabricated in one or more forms in all IC processes. On-chip
inductors can also be fabricated through the use of planar spiral geometries formed in one
or more of the process’ metal layers. Such inductors are employed routinely in the design of
GaAs monolithic microwave integrated circuits (MMICs) operating at several GHz. More
recently, spiral inductors have begun to see commercial application in silicon processes at
lower frequencies, where they are employed in on-chip impedance matching networks [46].
Their use in filter design at frequecies below 2 GHz has been investigated by Nguyen and
26
Meyer [131] for lowpass filters, and by Chang, Abidi, and Gaitan [122] for bandpass filters.
Negus, et. al. [46] has also reported their use in silicon for both bandpass filtering and
impedance matching functions.
A significant problem with all on-chip LC filters realized to date is low Q, as illustrated in
Table 2.7.2 In silicon processes operating at 2 GHz and below, the quality factor of on-chip
spiral inductors is generally 10 or less [131] [77], while in-circuit Q can be significantly lower.
For matching networks or lowpass filter design, these quality factors are often acceptable –
provided the impedance ratio or filter order is sufficiently low. However, for bandpass filters
intended for use in superheterodyne receivers, they are far too low to achieve the desired
fractional bandwidths. An in-circuit Q of 5 implies a bandwidth of 20% – nearly an order
of magnitude away from than that offered by the discrete filters seen previously in Table
2.6.
Table 2.7: On-Chip Spiral Inductors in Silicon Processes.
Ref
[131]
[131]
[122]
[122]
Inductance
1.9 nH
9.7 nH
100 nH
100 nH
Resonance
9.7 GHz
2.5 GHz
800 MHz
3 GHz*
In-circuit Q
8 @ 4.1 GHz
3 @ 900 MHz
1.3 @ 400 MHz
3 @ 800 MHz*
Turns
4
9
20
20
Dimension
115 um
230 um
440 um
440 um
Reasons for low spiral inductor quality factors are discussed in Chapter 8 together with
possibilities for process modifications that could be used to provide some improvements.
However, improvements sufficient to realize narrow bandwidth passive LC filters are unlikely
because of the substantial process modifications needed. Realization of on-chip, high-Q, LC
bandpass filters therefore requires the addition of active circuitry to compensate for coil
losses [57]. These techniques are considered in Section 2.3.3 below and in Chapters 5 - 9
later in this dissertation.
2
The performance of inductors marked with an asterisk (*) has been enhanced by removal of the underlying substrate using a post-fabrication etch.
27
Passive Electro-Acoustic Filters
A handfull of researchers have investigated the possibility of implementing electro-acoustic
filters on-chip as shown in Table 2.8 [60] [167] [158] [163] [165]. Such filters have been integrated with active circuitry in silicon IC processes, but require process additions/modifications.
To date, these filters have also required relatively large chip areas. Both of these factors are
likely to limit their ability to compete with low cost off-chip designs. Nevertheless, their
performance is reasonably good and they offer the advantage of being completely passive
– consuming no power and yielding excellent dynamic range. These performance features
may allow them to compete in some applications in the future, if and when their fabrication
reaches a suitable level of maturity.
Table 2.8: On-Chip Electro-Acoustic RF Bandpass Filters.
Ref
[163]
[159]
[165]
[165]
[167]
Year
1990
1992
1989
1989
1989
Type
BAW
BAW
SAW
SAW
SAW
Freq
1.1 GHz
1.8 GHz
283 MHz
283 MHz
160 MHz
Bandwidth
30 MHz (3%)
180 MHz (10%)
18.7 MHz (6.6%)
500 kHz (0.18%)
180 kHz (0.1%)
IL
1.1 dB
3.6 dB
–
–
–
Size
58 mm2
–
≈10 mm2
≈10 mm2
10 mm2
Like their discrete counterparts, on-chip electro-acoustic filters can be divided into two
major subcategories:
• Bulk acoustic wave, and
• Surface acoustic wave devices.
Bulk acoustic wave (BAW) devices are usually referred to in the literature as either thin
film resonators (TFRs) or film bulk acoustic resonators (FBARs) to emphasize their construction and crystal resonator-like behavior. In [163], a FBAR filter operating at 1.1 GHz
28
is described. This filter was constructed in four layers consisting of a sputtered SiO2 base
for support and temperature compensation, followed by a 0.25 um layer of oriented gold
onto which a piezoelectric layer of ZnO was sputtered. The final layer consisted of evaporated aluminum, with a 0.2 to 0.5 um thickness. The fabrication procedure is stated to be
compatible with the fabrication of active devices, although this was not demonstrated in the
reported examples. The filter consisted of two resonators together with shunt inductors to
resonate out the FBARs’ parallel plate capacitances. The resulting 2-pole response showed
very low insertion loss (1.1 dB) and a 3% fractional bandwidth. Such a device could be
used as a front end preselect/image filter for cellular or PCS receivers. The techniques are
reported to be suitable for use from 1 to 5 GHz. However, for a 1 GHz filter, the chip size
is relatively large (58mm2) – primarily due to the use of the on-chip spiral inductors.
A more complex stacked crystal filter (SCF) construction, not requiring spiral inductors has
also been reported [159]. This device yielded a 10% fractional bandwidth, one pole response
at 1.8 GHz. To the author’s knowledge, no MCF structures have yet been implemented,
although their possibility has been considered [158].
On-chip surface acoustic wave (SAW) filters have been reported in [60], [167], and [165].
These devices were constructed together with RF active circuitry in a modified BIFET
process. As shown in Table 2.8, on-chip SAW filters include both wideband transversal
designs (6.6% fractional bandwidth), and narrowband resonators (0.2% fractional bandwidth). These filters, operating at 283 MHz were fabricated using a ZnO piezoelectric
deposited on top of aluminum interdigital transducers, resting on a field oxide SiO2 base.
The surface acoustic wave propagates at the boundary of the SiO2 and ZnO and shows
good performance, provided the thickness of the SiO2 and ZnO are selected properly [165].
For example, the dynamic range for a SAW delay line constructed in this form was found to
be approximately 130 dB. The additional SAW filter entries in Table 2.8 listed as operating
at 160 MHz are preliminary extrapolations of this work targeted at use in an up-conversion
FM broadcast receiver [167]. The size of these filters (21mm2 and 9mm2) is significant, but
29
not impractical.
Based on these reported results and projections, integrated electro-acoustic filters appear
to offer good performance in moderate chip areas and could become practical as RF and IF
filters in PCS receivers with further development. However, due to lack of access to suitable
fabrication facilities, this option is not pursued in this dissertation. The interested reader
is referred to the references for more information.
2.3.3
Analog Active Filters
The vast majority of research in on-chip filtering has dealt with the design of active filters – a
mature subject on which hundreds of papers have been published and numerous textbooks
have been written. Here we will concentrate on a subset of this work – the design of
integrated bandpass filters targeted at radio receiver applications. As we will see, dynamic
range, a significant area of concern in all active filter applications, is particularly troublesome
in high-Q bandpass designs. In addition, fabrication tolerances and temperature drift also
present significant technical challenges.
Implementation approaches for active filters can be grouped into three main categories,
which are considered in the following subsections:
• Switched-capacitor filters,
• Continuous-time Gm-C filters, and
• Q-enhanced LC filters.
30
Switched-Capacitor Filters
At low frequencies (below about 10 MHz), switched-capacitor (SC) designs can provide
precision filtering in the face of wide fabrication tolerances. By simulating a resistor’s
current-voltage relationship with charge sharing via capacitors and FET switches, “RC”
time constants become dependent on capacitor ratios and clock rates alone. Since capacitor
ratios can be held to tolerances as tight as 0.1% to 0.5% on a chip [91], very accurate
responses can be achieved. The primary disadvantages of SC implementations are the need
for fast settling amplifiers, and the need for anti-alias filtering at the input and reconstructive
smoothing at the output. The former limits the frequency range over which the SC filter
can operate, while the latter adds complexity and requires some degree of oversampling,
beyond the Nyquist rate.
Some examples of reported SC bandpass filter implementations targeted at radio receiver
design are shown in Table 2.9. The 260 kHz filter in [56] is a good example of early work
in this area. This 3-pole (6th order) bandpass filter provides a fractional bandwidth of
2.5%, a dynamic range of 70 dB, and a power dissipation of 70 mW. With a 4 MHz clock,
anti-aliasing can be provided with a simple low order, lowpass filter.
The remaining implementations listed in Table 2.9 illustrate both the possibilities and the
practical problems of implementing SC filters at higher frequencies. These filters show
reduced dynamic range, higher power dissipation, or both. Moreover, except for the GaAs
implementation, the clock to center frequency ratio has been lowered to approximately 4:1
in order to reduce amplifier settling time problems, requiring more aggressive anti-alias
filtering. The GaAs 20 MHz filter achieves a respectable dynamic range (for its bandwidth)
and good clock to center frequency ratio, but its power consumption is unsuitable for a
portable radio receiver application. In addition, at 8.6 mm2 chip area for a 1-pole (2 nd
order) design, its cost may also be a problem.
31
Table 2.9: Switched-Capacitor RF Bandpass Filters.
Ref
Year
Freq
Order
Power
DR
[56]
[85]
1983
1986
260 kHz
3.1 MHz
Bandwidth
6.5 kHz (2.5%)
56 kHz (1.8%)
6th
6th
70 mW
45 mW
70 dB
51 dB
[86]
1988
10.7 MHz
430 kHz (4%)
6th
500 mW
42 dB
[61]
1991
20 MHz
1.25
(6.3%)
2nd
440 mW
65 dB
MHz
Process
4 um CMOS
1.75
um
CMOS
2.25
um
CMOS
0.5 um GaAs
Size
Voltage
–
2mm2
±5V
5V
2mm2
10V
8.6mm2
–
Continuous-Time Gm-C Filters
The problems associated with implementing SC filters at frequencies of several MHz or
higher have led researchers to the design of continuous-time (CT) active filters. While
CT filters actually predate SC filter design, implementation of CT filters in completely
monolithic form is comparatively new.
One significant difference between discrete active filter design and the design of on-chip active filters can be seen in the building blocks employed. In discrete designs, these building
blocks usually include resistors, capacitors, and operational amplifiers. The resulting filters
are referred to as RC active filters. In the design of most modern integrated CT filters,
suitable resistors are often not available, and MOSFETs biased in the resistive region are
often used in their place. The resulting filters are then referred to as MOSFET-C filters
[91]. In addition, the desire to operate at high frequencies and with smaller chip areas
often rules out the use of operational amplifiers. Simpler active filters with fewer internal nodes and associated parasitic poles can be created using transconductance amplifiers,
implemented with a small number of FETs. These Gm or operational transconductance
amplifiers (OTAs) can be used to form gain stages and integrators. The names given to the
resulting filter designs include Gm-C filters [94], OTA-C filters [94], transconductance-C
filters [75], and transconductance - amplifier - capacitor (TAC) filters [68]. For uniformity
and simplicity, we shall refer to all of these as Gm-C filters in this dissertation.
32
Today the dicipline of on-chip Gm-C filter design is well established. Several excellent review
articles have appeared on the subject in leading journals [81] [91] [94], as well as books [63]
[72] [84] and collections of reprint articles [93]. Nevertheless, commercial applications of GmC filters to date have been limited primarily to lowpass filters. An example is the AD896
from Analog Devices, designed for use in read channels of disk drives. The only potential
applications of bandpass designs known to the author include a 5.5 MHz filter with a 5%
fractional bandwidth designed at Philips as part of a television chroma, luminance, and
sound separator IC [71], and a 55 kHz, 10% fractional bandwidth design used at Sony in an
AM receiver [48]. However, it is not clear from the available literature if these filters have
made it into marketed products.
Possible reasons for this limited commercialization of integrated bandpass filters can be
seen by examining Table 2.10. This table summarizes the performance of bandpass filters
reported from 1968 to 1994 that have been targeted at receiver design. Conclusions that
can be drawn from these data include:
• Fractional bandwidths of many implementations are often excessive (up to 20%),
• Power dissipations are often high (100 mW or above),
• Chip area consumption is moderate to high (4mm2 or above), and
• Dynamic range is low to moderate (40 to 70 dB).
Further research overviewed in the section on dynamic range below reveals that all of these
problems are related. In fact, there are fundamental limitations to the dynamic range
that can be achieved for a given power consumpation and filter fractional bandwidth. In
the opinion of the author, these dynamic range limitations are perhaps the single most
important reason why these filters have so far not seen significant commercial use.
Another significant problem with on-chip CT filters is frequency and Q accuracy. Achieving
33
Table 2.10: Continuous-Time Gm-C RF Bandpass Filters.
Ref
Year
Freq
Order
Power
DR
Process
Size
Voltage
[79]
[71]
[64]
1968
1980
1984
650 kHz
5.5 MHz
500 kHz
Bandwidth
13 kHz (2%)
280 kHz (5%)
96 kHz (19%)
2nd
2nd
6th
100 mW*
–
55 mW
65 dB
–
60 dB
–
11mm2
4mm2
+12, −6V
12V
10V
[55]
[65]
[75]
1986
1987
1988
10.7 MHz
455 kHz
4 MHz
2 MHz (19%)
7 kHz (1.5%)
800 kHz (20%)
6th
8th
8th
650 mW
–
900 mW
60 dB
40 dB
75 dB*
16mm2
–
23mm2
±5V
5V
±5V
[98]
1989
12.5 MHz
250 kHz (2%)
4th
360 mW
60 dB
7.8mm2
±6V
[66]
[66]
[68]
1989
1989
1989
10.7 MHz
40 MHz
95 MHz
1.1 MHz (10%)
3.2 MHz (8%)
8 MHz (8%)
4th
4th
2nd
80 mW
80 mW
–
50 dB
49 dB
–
4mm2
2mm2
–
6V
6V
–
[78]
1990
1.0 MHz
300 kHz (30%)
6th
25 mW
45 dB
4mm2
5V
[69]
1990
200 kHz
20 kHz (10%)
18th
1.2 mW
54 dB
2.5mm2
4V
[83]
1992
10.7 MHz
300 kHz (3%)
4th
220 mW
68 dB
6mm2
±2.5V
[54]
1994
455 kHz
27 kHz (6%)
18th
33 mW
64 dB
Bipolar
Bipolar
6
um
CMOS
Bipolar
–
3
um
CMOS
3
um
CMOS
Bipolar
Bipolar
3
um
CMOS
1.75 um
CMOS
3
um
CMOS
1.5 um
CMOS
1.2 um
CMOS
17mm2
5V
full integration requires tightly controlled or trimmed on-chip resistor and capacitor tolerances together with precision temperature compensation. The alternative is to employ some
form of on-chip tuning mechanism. The former adds considerably to the cost of design and
fabrication, and is practical only at relatively large fractional bandwidths The latter adds
complexity to the design, but has nevertheless become a necessary part of the on-chip Gm-C
filter design dicipline [106]. In the subsections below, these two problems are investigated
in greater depth.
Tuning Techniques Tuning of an on-chip Gm-C filter’s frequency and Q is required due
to fabrication tolerances and temperature drift of capacitances and transistor gains. To
compensate for these factors, all of the filter implementations in Table 2.10 which have
included tuning circuits on-chip employ some variant of a technique originally introduced
by Tan and Gray [89]. This technique is now known as master/slave tuning [106].
34
In the simplest implementations of the master/slave technique, the “master” is an on-chip
tunable oscillator whose component values are related by known ratios to those used in
the “slave”, which is the filter being tuned. Both master and slave receive the same tuning
control voltages and/or currents so that changes in the master’s frequency and Q are tracked
by the slave. The desired filter tuning is then effected by phase locking the master oscillator
to a known reference frequency supplied to the chip from an external source. In a radio
receiver, such a reference might be available in the form of either a master local oscillator
for frequency synthesis, or a clock for the control microprocessor.
With proper design, matching and tracking for both capacitors and transistor gains in Gm-C
filters can be held to as tight as 0.1% to 0.5% [91], which is adaquate for realizing selectivity
Q values as high as 50-100 and pole frequencies as accurate as about 1%. This implies a
practical lower limit to bandpass filter fractional bandwidths of about 2% – a conclusion
supported by filter realizations listed in Table 2.10.
If smaller fractional bandwidths are desired, some form of “self-tuning” technique is needed.
A handfull of researchers have addressed this problem, offering control systems with varying degrees of complexity and performance. In 1981, Tsividis [108] proposed a technique
involving switching between two filters, allowing the filter used in the signal processing to
be periodically taken off-line and retuned. Tuning off-line can be relatively easy and can
provide arbitrary precision. For example, for a second-order filter, a sinusoid at the filter’s
center frequency can be fed to the filter. The amplitude and phase of the sinusoid at the
filter output can then be used to sense and adjust the filter’s Q and center frequency respectively. Later, Brooks and VanPeteghem [100] introduced an approach that allows the filter
involved in the signal processing path to be tuned without interruption. Their Correlated
Tuning Loop (CTL) technique performs a narrowband autocorrelation on the filter’s input
signal and a narrowband cross correlation between the filter’s input and output signals to
derive tuning control signals. More recently, Kozma, Johns, and Sedra [103] [104] presented
a tuning technique based on adaptive filtering and described an approach incorporating two
35
filters which could allow tuning the filter used in the signal processing path without interruption. In this approach, filter A continuously processes the input signal while filter B is
first tuned with a known reference as descibed above, and then connected to the input signal
so that its output can serve as a reference for adjusting filter A. With this arrangement,
filter B could be periodically tuned as in [108], and no interruption of the signal processing
by filter A is needed.
In a paper by the author [105], a technique called orthogonal reference tuning (ORT) was
developed and prototyped. In this approach, a known reference signal is employed and
is present in the filter passband together with the RF/IF signal being processed. This is
permissible provided the reference signal can be made orthogonal, or nearly orthogonal, to
the processed signal so that the two can be separated at the filter output. This is shown to
be possible for certain types of modulation, including wideband FM and digital signaling
such as BPSK and spread spectrum. The details of this technique are presented in Chapter
7 later in this dissertation.
It is interesting to note that to date none of these approaches have been implemented onchip. This fact is most likely due partly to their complexity, and partly to the inherent
dynamic range limitations of high-Q filters for which they are most applicable. These
dynamic range limitations are investigated in some depth in the following section.
Dynamic Range Limitations Dynamic range (DR) is an important performance parameter in radio receiver design. For receivers incorporating active filters, the DR of the
filter employed will upper bound the receiver DR and is therefore of paramount importance
in the filter’s design.
Dynamic range is typically defined as the maximum output signal power Psat prior to some
level of gain compression and/or signal distortion, divided by the integrated output noise
power PN in the bandwidth of interest B. In filter design, the filter noise bandwidth is
36
often used, which for high order filters is approximately equal to the filter’s nominal (e.g.
3 dB) bandwidth. Thus,
DR =
Psat
V2
= sat
PN
VN2
(2.1)
where Vsat and VN are the saturation and integrated noise voltages measured at the filter
output.
All filters, whether active or passive possess a finite DR. For example, in a passive electroacoustic filter, PN is determined by thermal noise, and can be computed by
PN = kT B
(2.2)
where k is Boltzmann’s constant, and T is the physical temperature in Kelvin (e.g. K =
290). The limit on maximum signal power is determined by non-linearities in stress and
strain relationships in the piezoelectric material, and although it may be very large, it is
still finite. As an example, the DR of an on-chip SAW delay line is given in [167] as 130 dB.
This figure is considerably higher than the instantaneous DR of many receivers.3 Hence,
the issue of DR in passive filters is often ignored.
In active filters however, DR limitations can be much more severe. One obvious limitation is
that imposed by saturation of the active circuits employed. For most filters, the maximum
output signal power will be some fraction of the total power consumed by the circuit. A less
obvious, but equally important factor in active filters is an increase in the noise floor over
that given in (2.2) due to shot and thermal noise in the devices used, and the amplification
3
In receivers such as televisions, where linear modulation is used, the overall DR may be 100 dB or
higher due to the use of AGC, but the instantaneous DR is usually less. For FM receivers where limiting
is employed, the circuits prior to the channel select filter must still be linear and their DR is usually below
100 dB.
37
of this noise in certain areas of the circuit.
In a 1983 paper on SC filters, Choi [56] made the observation that noise in high-Q SC filters
is greater than that of low-Q filters due to the inherent gains associated with the filter
resonators. However, no quantitative treatment was given. Khorramabadi [64] quantified
this effect for the case of continuous-time filters, giving the following expression for the noise
at the output of a 6th order Gm-C ladder filter
VN2 =
3kT Q
Cintg
(2.3)
where Q is the filter’s selectivity Q and Cintg is the capacitance associated with the integrators used. Applying the definition of DR in (2.1), the DR of this filter becomes
DR =
2
Vsat
Cintg
3kT Q
(2.4)
in which the dependence of DR on Q is clearly evident.
The issue of fundamental limits on DR for arbitrary Gm-C filter designs was investigated
in detail by Groenewold [115] [116]. In a 1992 paper, he showed that for a high-Q biquad,
the optimum DR that can be achieved is given by
DRopt =
2
Ctotal
Vsat
4kT F Q
(2.5)
where Ctotal is the total capacitance in the filter’s integrators, and F is a noise factor term
(F > 1) used to account for possible excess noise contributions by non-ideal devices. In the
same paper, an approximate upper bound on optimum DR for a general high-Q BPF was
derived as
38
DRopt ≤
2
Ctotal
Vsat
2πkT F Q
(2.6)
a result in general agreement with that predicted by Khorramabadi for a particular 6th
order filter. Similar relationships for the dependence of DR on filter Q for high-Q designs
have also been published by Abidi [110].
The results published by Groenewold have been extended by the author in [119] to show the
dependence of DR on filter power consumption Pdiss and filter bandwidth B – parameters
more appropriate to system level receiver design studies. Under these constraints, it was
shown that the optimum DR is inversely proportional to the square of the filter Q
DRopt =
ηPdiss
4πkT F BQ2
(2.7)
where η is an efficiency factor relating the power consumed by the filter to the maximum
output signal power. In the same paper, the DR of several hypothetical low power Gm-C
filters were compared to DR measurements reported for commercial radio receivers. From
these comparisons it was concluded that Q values up to approximately 100 in Gm-C bandpass filters can theoretically yield acceptable DR at acceptable power consumption, if an
efficiency of 0.1 and a noise factor of 2 can be achieved. This result is studied in detail
in Chapters 5 and 6. A comparison of this result with DR figures listed in Table 2.10 on
a normalized, per Hz basis suggests that efficiencies and/or noise figures of realizations to
date are about one order of magnitude worse than this, indicating where some room for
improvement might be found.
A detailed study of Groenewold’s work reveals that the mechanism responsible for limiting
DR in high-Q bandpass filters is essentially the regenerative gain associated with the realization of high-Q complex poles – exactly the reason cited by Choi [56] for SC filters. This
implies that the only way to improve DR performance in an active filter (for a fixed power
39
consumption and bandwidth) is to decrease the regenerative gain. In turn, this suggests
a need for resonance circuits which possess energy storage and exchange mechanisms (see
Chapters 5 and 6).
This line of reasoning by the author led to an exploration of the dynamic range of Qenhanced LC filters. In [119] it was shown that a 2nd order bandpass LC filter whose
Q has been increased by active negative resistance circuitry does indeed provide marked
improvements in DR over Gm-C filters. The DR for such a filter is found to be:
DRopt =
ηPdiss
Q2
4πkT F BQ2 o
(2.8)
where Qo is the base Q of the LC circuit prior to Q enhancement. Thus, even though
the Q of on-chip inductors may be limited to 10 or less, an on-chip Q-enhanced LC filter
should be able to realize small fractional bandwidths while providing a 20 dB improvement
in DR over an Gm-C realization. Alternatively, for a given DR, the Q-enhanced filter with
a coil Q of 10 requires two orders of magnitude less power than a Gm-C filter operating at
similar efficiency. This feature is one of the primary reasons for the emphasis on Q-enhanced
filtering in this dissertation.
Q-Enhanced LC Filters
Q-enhanced (QE) LC filtering is an emerging research area on which few papers have so far
been published [57] [76] [119]. One possible realization of an Q-enhanced LC filter which
has been used by the author is illustrated in Figure 2.6
In this realization, a MOSFET differential input amplifier Q1A, Q1B provides current drive
to an LC tank circuit composed of L1A, L1B, and C1. Stray capacitances CSA and CSB
represent both the inductor turn-to-substrate capacitance as well as transistor gate and
40
Vdd
L1a
L1b
C1
Out+
Out-
In+
In-
Q1a
Q1b
Q2a Q2b
Csa
Csb
Ii
Input Amp
Q3a Q3b
Iq
Negative Resistance
Io
Output Buffer
Figure 2.6: Q-enhanced LC Bandpass Filter.
drain loading, modifying the tank circuit’s resonance frequency. By itself, the resulting
modified tank circuit’s differential impedance seen between the drains of Q1A and Q1B
has low Q due largely to the spiral inductor series resistance losses. These losses, combined
with smaller losses from the tank circuit capacitances, can be represented (near the resonant
frequency) by a suitable valued resistance in parallel with C1. The negative resistance circuit
composed of cross-coupled differential amplifier Q2A, Q2B cancels most of this resistance,
producing a Q-enhanced resonance. This circuit is tuned by varying the tail current Iq to
effect a desired Q. Finally, output buffer Q3A, Q3B serves to isolate the filter load from the
tank circuit and provide a lower impedance drive capability.
While the concept of increasing the Q of a tank circuit dates to the early days of radio
design in the form of regenerative receivers [2] [20], the use of this technique for high-order,
precision filtering has so far received little attention. One possible explanation for this is
the applicable frequency range of on-chip inductors. According to [77], this range is limited
to about 1 GHz and above. In fact, work by the author indicates that this is a soft limit
which depends on fabrication process characteristics, and on what constitutes acceptable
41
Q for a coil – factors considered in depth in Chapters 6 - 8. With the advent of new
frequency spectrum allocations at 900 MHz, 1.8 GHz, and 2.4 GHz for cellular and PCS,
there is now increased motivation for exploring this area of active filter design. Such designs
could function either as RF preselection/image filters or as IF channel selection filters in
cellular/PCS receivers.
The earliest reference to Q-enhanced bandpass filtering in the 1 - 2 GHz range known to
the author is a paper by Duncan, et. al. in 1993 [57]. This work described a “series
mode” compensation of inductor losses to raise the Q of 5 nH inductors from 5 to 20 at
1.6 GHz. Automatic frequency or Q tuning was not addressed, and only simulated results
were reported. In [93], a “parallel mode” approach was descibed in which an active negative
resistance circuit was placed in parallel with the inductance as in Figure 2.6. Later [77],
results were reported for a frequency scaled, board-level prototype using the parallel mode
approach. An approach to frequency tuning without requiring a varicap diode was also
discussed [76] [77].
Q-enhanced LC filters possess a number of attractive features besides the dynamic range advantage discussed in the previous section. For example, they should exhibit lower sensitivity
to active components [57], which can make Q tuning easier. In addition, stray capacitances
such as CSA and CSB in Figure 2.6 which present significant problems in Gm-C filter design
[65] are easily tuned out in Q-enhanced filters by adjusting the value of the inductors or the
resonanting capacitance C1. These issues are explored more fully in Chapters 5 and 8.
2.3.4
Summary of Alternatives
The options discussed in the preceding sections for implementing on-chip, low power, RF
bandpass filters are summarized in Table 2.11.
Digital filters operating at acceptable power levels have been shown to be viable only at
42
Table 2.11: Summary of On-Chip RF Bandpass Filter Alternatives.
Filter Type
Advantages
Digital
Precision,
Power consumption, chip
dynamic range, area, aliasing, ADC reprogrammability quirements,
external
clock requirement
Dynamic range, Quality factor, chip area
stability
Dynamic range, Process
modifications,
stability
chip area
Precision
Dynamic range at high
Q, aliasing, external clock
requirement
Frequency
of Dynamic range at high Q,
operation
tuning requirement
Dynamic range, chip
area,
tuning
stability
requirement
Passive LC
Electro
acoustic
Switched
capacitor
Gm-C
Q-enhanced
LC
Disadvantages
Frequency
Applications
< 10 MHz
Low IF filtering, baseband filtering and signal processing.
> 100 MHz
Power amp harmonic suppression, low Q RF preselection
RF preselection, IF filtering
> 100 MHz
< 10 MHz
Low frequency, moderate Q IF
filtering, baseband filtering
< 100 MHz
Moderate Q IF filtering, baseband filtering
RF preselection, IF filtering
> 100 MHz
relatively low frequecies, limiting their application to low IF and baseband processing.
Passive LC circuits are limited to operating at high frequencies due to the size of on-chip
inductors, and to the design of low Q, large fractional bandwidth filters due to inductor
losses. Nevertheless, passive LC circuits are used routinely in GaAs ICs and are beginning to
be applied in silicon ICs both as impedance matching networks and as harmonic suppression
filters in power amplifiers.
On-chip electro-acoustic filters, either in the form of bulk acoustic wave (BAW) or surface
acoustic wave (SAW) devices, offer potentially good performance for high frequency applications with both moderate (e.g 10%) and narrow (< 1%) fractional bandwidths. Economic
acceptance of these devices will hinge on the availability of modified fabrication facilities
and on what constitutes acceptable chip areas.
Switched-capacitor (SC) filters, which are used extensively in baseband signal processing
applications, have also been demonstrated at IF frequencies. Their frequency range is
limited to less than about 10 MHz in low cost silicon processes. Moreover, high-Q filters are
fundamentally limited in dynamic range – a property shared with continuous-time active
43
filters. For low power applications, this limits SC filter realizations to moderate fractional
bandwidths with pole Qs of 50 to 100.
Continuous-time on-chip active filters designed for high frequency operation employ Gm-C
architectures rather than the more traditional operational amplifier based designs used in
discrete RC filters. This allows them to work at higher frequencies than SC filters. Although
most reported designs have been implemented at or below about 10 MHz, operation to 95
MHz has been demonstrated, and finer geometry processes could double or quadruple this
range in the future. However, in common with SC designs, fundamental dynamic range
problems will limit their application to moderate fractional bandwidths (pole Qs of 50 to
100) in low power designs.
Finally, for frequencies above 100 MHz, Q-enhanced LC filtering offers a potentially good
alternative to electro-acoustic filtering – without the need for fabrication process modifications. Dynamic range of Q-enhanced LC filters is improved over that of SC and Gm-C
designs by a factor of Q2o , where Qo is the Q of the LC resonator employed before Q enhancement. Commercial viability of this new area of research will depend on the dynamic
range values achieved in practice, on the development of suitable methods for frequency
and Q tuning, and on chip area requirements of the inductors employed.
2.4
Integrated Synthesizers
In Section 2.1.1 we saw that frequency synthesizer ICs as well as receiver ICs with which
they were designed to operate both lacked one key component – on-chip tuned circuits. In
this section we examine some of the technical reasons why complete synthesizers, including
oscillator resonant circuits, are not currently implemented on-chip. A brief look at the
direct digital synthesis alternative to traditional PLL synthesizers is also included.
44
2.4.1
On-Chip Local Oscillators and VCOs
The reasons for oscillator resonant circuits remaining off-chip have much in common with
the problems associated with on-chip filtering. From an economic viewpoint, restricting
an IC to a particular frequency range limits its market size, just as on-chip filters would.
Nevertheless, just as for filters, there are several markets which are large enough to warrant
fixed frequency band products, with broadcast radio and television as well as cellular and
PCS among them. Thus the reasons for their limited use must be largely technical.
A look at commercial offerings shows that various forms of integrated oscillators have been
available for many years. For example, ring oscillators and Gm-C oscillators such as those
employed in master/slave tuning of Gm-C filters can be easily built. These circuits can be
made tunable, and can operate at frequencies up to at least several hundred MHz [141] [96],
satisfying two important requirements for use in tuners. A third requirement however, that
of spectral purity, is more problematic.
In radio receivers, phase noise and spurious outputs of local oscillators can often be critical
to receiver performance [147]. Phase noise and spurious outputs can be a limiting factor in
receiver dynamic range, as well as a contributor to poor signal to noise ratios and bit error
rates of demodulated signals. In addition, spurious outputs can result in spurious signals
being received which are not actually present in the environment.
One additional technical problem in implementing on-chip oscillators is on-chip signal coupling. Placing oscillator circuits on-chip with a digital synthesizer IC can be expected to
produce substantial spurious outputs due to the presence of high frequency harmonics of
digital switching waveforms. This problem may be solvable however, by careful choice of
frequencies and divider ratios. An alternative is to partition the system design into one
digital IC and one analog IC, physically separating the digital synthesizer circuits from
the analog oscillator. However, even if this is done, the problem of oscillator phase noise
45
remains.
Oscillator Phase Noise
Theoretical formulas for predicting oscillator phase noise were developed by Leeson and
Robins in the early 1960s [142] [147]. For an LC based oscillator, Robins has derived the
ratio of the one-sided phase noise power Nop in a 1 Hz bandwidth to the oscillator output
power Posc as a function of frequency offset ∆f from the nominal oscillator frequency fo as
Nop
kT F
fo 2
=
(
)
2
Posc
8Q Posc ∆f
(2.9)
where k is Boltzmann’s constant, T is temperature in Kelvin, F is a noise factor of the
oscillator active circuits, and Q is the loaded quality factor of the LC resonant circuit. This
formula is derived by treating the oscillator as a linear filter with sufficient Q enhancement
that it amplifies the thermal noise to the desired output power level of the oscillator.
1
f
noise of active devices is not included in the derivation, so that (2.9) applies only at large
frequency offsets. The linear assumption, which can be achieved with a detector and AGC
circuit if necessary, is a relatively good one for high quality oscillators. This simple formula
provides significant insight into the spectral purity problem and shows fundamental limits to
achieving good phase noise performance. Good phase noise requires a large resonant circuit
Q, with performance improving as Q2 – exactly the same behavior noted for Q-enhanced
LC filters.
Since (2.9) was derived for LC oscillators, it is not directly applicable to active circuits without inductors. We may expect nevertheless that the phase noise of inductorless oscillators
would be on the order of that predicted by (2.9) for the case of Q = 1. This expectation is
supported by the following result for the phase noise of a Gm-C based oscillator published,
but not derived, in [149]:
46
Nop = kT R(1 + A + 2Q)(
ωo 2
)
∆ω
(2.10)
In (2.10), Q is now defined as Xc /R – a figure of merit for the active inductor and capacitor
employed. Xc is the tank circuit capacitive reactance, R is the series resistance of the tank
circuit, and A is a noise factor term which is taken to be 1 for an ideal OTA.
The value of Nop in (2.10) is strictly increasing with R and is minimum for R = 0, corresponding to use of an ideal active inductor, and lossless capacitor. Under this condition,
we can rewrite (2.10) in the form of (2.9) to allow a comparison:
Nop
2kT fo 2
=
(
)
Posc
Posc ∆f
(2.11)
This result supports our earlier conclusions. The phase noise of oscillators using active
inductors is significantly and fundamentally inferior to that of LC oscillators – just as the
dynamic range of Gm-C filters was found to be inferior to that of Q-enhanced LC filters.
The important issue of what level of phase noise is acceptable will be considered in detail
in Chapter 10. The answer to this question will be found to depend on the radio system in
which the oscillator is employed.
2.4.2
Reported Implementations
Table 2.12 summarizes some of the work in on-chip oscillators reported to date and the
phase noise performance obtained.
In [96], Voorman, et. al. report an oscillator constructed with a Gm-C architecture operating
at 215 MHz, which is converted to 430 MHz using an on-chip frequency doubler. Although
few details are given, a spectrum analyzer display shows a phase noise at 100 kHz offset of
approximately -90 dBc/Hz.
47
Table 2.12: Reported Work in On-Chip Oscillators.
Reference
[96]
[144]
Type
Gm-C
LC
fo
430 MHz
1.8 GHz
Q
5*
Power
70 mW @ 5 V
[149]
[150]
LC
TFR
1.1 GHz
300 MHz
21
-
1 mW @ 1 V
-
Phase Noise
-90 dBc @ 100 kHz
-67 dBc @ 20 kHz
-88 dBc @ 100 kHz
-75 dBc @ 10 kHz
-110 dBc @ 1 kHz
In 1992, Nguyen [144] et. al. describe the design of an integrated LC-based VCO operating
at 1.8 GHz and possessing a 5% tuning range. As an alternative to the use of varactors
for tuning, a cross-coupled, dual tank circuit Colpitts configuration is used. The inductors
employed are square spirals with outer dimensions of approximately 200 um. One has 9
turns, giving approximately 7 nH inductance, while the other has 7 turns, giving 5 nH of
inductance. The circuit operated at 5V and consumed 70 mW of power. The measured
phase noise is -67 dBc at 20 kHz offset and -88 dBc at 100 kHz offset.
As an alternative to on-chip spiral inductors which are known to be low in Q, Steyaert and
Craninckx [149] investigated the use of bondwires to create inductors. A 1.1 GHz oscillator
was constructed which consumed 1 mW of power from a supply as low as 1 V. Even at
this reduced power, the phase noise is improved over Nguyen’s result, thanks to the higher
inductor Q.
In [150] and [151], Burns, et. al. report results for an oscillator at 300 MHz based on a
thin film bulk acoustic wave resonator. These resonators provide Qs in the range of 1000 or
above, which accounts for the significant improvement in phase noise over other work listed
in the table. A phase noise of -110 dBc/Hz was reported at an offset of only 1 kHz.
To put all of these results into some perspective, they are shown plotted together with the
phase noise of a cellular phone VCO in Figure 2.7. The curve for the cellular phone VCO is
taken from the National Semiconductor LMX1501A synthesizer data sheet and represents
48
a highly optimized discrete VCO design. To provide a meaningful comparison, all phase
noise values plotted in Figure 2.7 have been normalized to a 1 GHz operating frequency
according to the frequency scaling behavior predicted in (2.9) and 2.11.
-70
x
LMX1501A
x LC
Phase Noise in dBc
x LC
-80
x Gm-C
-90
-100
x
x
x TFR
-110
x
-120
-130
10K
1K
100K
Frequency Offset in Hz
Figure 2.7: Phase Noise Comparison of On-Chip Oscillators.
Finally, we note that there appears to be at least one example of an on-chip synthesizer
being produced in a commercial chip. In [29], a product announcement for an RF IC for a
global positioning system (GPS) receiver shows a block diagram with an on-chip oscillator
together with PLL synthesizer divider circuits. However, no description of, or technical
information on, the oscillator was provided.
2.4.3
Direct Digital Synthesis
One possible alternative to use of on-chip VCOs is direct digital synthesis (DDS) technology
[143] [33]. Direct digital synthesis has several attractive features including:
• Phase noise is equivalent or better than that of the crystal reference clock signal used,
49
• Frequency resolution can be very small ( 1 Hz), and
• Switching time is nearly instantaneous.
The biggest problems facing these devices are power consumption, spurious frequency generation, and chip area. As an example, a DDS operating at a 150 MHz clock rate has been
reported in [145]. Built in a 1.25 um CMOS process, it consumes approximately 1 W of
power and 25 mm2 of chip area. Spurious levels are -67 dBc when using an external 12 bit
digital to analog converter (DAC). A more recent, lower power DDS has been reported in
[140]. Implemented in a 1 um CMOS process, it includes an on-chip 10 bit DAC, dissipates
40 mW at 40 MHz clock frequency, and consumes approximately 15 mm2 of chip real estate.
Perhaps the biggest problem with DDS devices in radio receiver applications is their relatively low frequency of operation. The synthesized sinusoid of a DDS is limited to less than
half of the clock rate by the Nyquist criterion. This requires that the synthesized signal
either be frequency multiplied to the desired final local oscillator operating frequency (e.g.
1 GHz), or that it be mixed with an auxiliary oscillator. Frequency multiplication by a
factor N results in an increase in both phase noise and spurious levels by 20log(N ) [147]
[143], resulting in poor performance for large N. The other alternative, frequency mixing,
requires a high purity oscillator operating near the desired final frequency together with an
image rejection mixer, a bandpass filter, or both. Thus, the generation of the desired LO
signal may actually be complicated by the use of the DDS in comparison with a simpler
VCO and PLL synthesizer approach. The advantages of DDS are primarily found only
when fine frequency resolution and/or fast tuning is required.
2.5
Integrated Receivers
In Section 2.1, we looked at commodity ICs targeted at radio design and discovered that
despite high levels of integration in many devices and names such as “single-chip receiver”
50
coined by their manufacturers, two key components – bandpass filters and LO tuned circuits
– were not included. In this section, we investigate research and development efforts directed
toward complete integration of receivers from antenna input to baseband (audio or digital)
output. A summary of this work is provided in Table 2.13 and discussed below.
Table 2.13: Reported Work in Integrated Receivers.
Ref
Application
Architecture
[85]
[167]
Narrowband
FM
FM Broadcast
[43]
FM Broadcast
[48]
AM/FM
Broadcast
[51]
[52]
Paging
Paging
Dual Conversion Superhet
Up Conversion
Superhet
Low
IF
w/ Modulation
Compression
Low IF w/ Image
Reject
Mixer
Zero IF
Zero IF
2.5.1
Sensitivity
Selectivity
Size
Voltage
Current
5 mV
31 dB
7.7 mm2
5V
20 mA
1 uV
–
> 9mm2
–
–
1 uV
35 dB
4.5 mm2
3-18 V
8 mA
1 uV
40 dB
12 mm2
0.9-4.5 V
16 mA
–
-126 dBm
–
70 dB
5 mm2
18 mm2
1.8 V
2-3.5 V
2.5 mA
2.7 mA
Classical Superheterodyne Architectures
Reported work on integration of classical superheterodyne receivers is limited, and to date no
complete single-chip designs have been realized in hardware. In Section 2.2, we postulated
that the availability of suitable bandpass filters is largely to blame for this – a conclusion
supported by work reported in [42], [50], and [167] which is reviewed below.
Downconversion Designs
In 1986, Song, et. al. [50] employed two switched-capacitor bandpass filters in the design of
a CMOS dual conversion superheterodyne narrowband RF “receiver”. The design consisted
of an RF down conversion mixer followed by a 3 MHz first IF bandpass filter with a 6th
order response and a bandwidth of 55 kHz. Continuous-time anti-alias filters are included
at input and output ports of this filter. A second mixer downconverts the first IF filter
51
output to a 72 kHz second IF where a 6th order switched capacitor bandpass filter narrows
the channel bandwidth to 24 kHz.
Demodulation of narrowband FM is provided on-chip through the digital equivalent of a
one-shot FM detector. Following demodulation, a 3.75 kHz switched capacitor, 7th order
lowpass filter is used to smooth the output signal. A single reference clock together with
appropriate dividers provides all clocking for the SC filters as well as the LO signal for the
second mixer. The RF front end and first mixer LO are not included on-chip. Aside from
these omissions, the largest problems with this design are the basic receiver architecture
and the performance achieved.
According to the authors, the low first IF (3 MHz) was selected to avoid problems with
available on-chip bandpass filters which are difficult to implement at higher frequency (especially for SC designs), and are limited in Q. Although the chip was stated as being targeted
at use with RF inputs in the range of 50 - 100 MHz, this low IF would make this difficult in
some applications. If operated at 50 MHz, a 4th order preselect filter with a 1% fractional
bandwidth would be required to provide adaquet image rejection. At 100 MHz, the fractional bandwidth needed would be 0.5%. Moreover, if the service band of interest exceeds
500 kHz, this filter would need either a higher order response to provide sharper rolloff, or
would have to be tunable.
In addition to these problems, the 1.8% distortion achieved is somewhat high and the 31 dB
alternate channel selectivity is probably too low to be practical - especially when competing
with other receivers which routinely achieve figures of 40 - 70 dB or higher.
Upconversion Designs
One approach to dealing with the image problems of superheterodyne receivers is to use an
up-conversion architecture in which the first IF is higher in frequency than the desired RF
52
signals. In an up conversion design, requirements on RF preselection/image filtering are
relaxed substantially, although other requirements become more severe. For example, the
up-conversion is usually followed by a down conversion to a low second IF for amplification
and detection. The ratio of the first to second IF frequencies may be as high as 100,
requiring a very narrow first IF filter implemented as a ceramic or SAW design. However,
this narrow filter can now be implemented at a single frequency (assuming receiver tuning
is done using the first mixer LO), regardless of how wide the service band of interest is.
These features together with the problems of implementing on-chip high-Q active bandpass
filters led researchers at Delft University to explore the possibility of building up conversion
receivers with on-chip SAW filtering [167] [120] [165] [42]. In [167], P. T. M. van Zeijl et. al.
explored the feasibility of an integrated up conversion FM broadcast receiver with a first
IF at 160 MHz. The integration of RF front-end circuitry (amplifiers and mixers) together
with SAW filters was demonstrated, although a filter with the required bandwidth was not
realized in this effort. In addition, other required circuits such as the synthesizer, second
IF, and demodulator were omitted. General conclusions drawn from this work include:
• Integration of active circuits with SAW filters is viable if “minor” fabrication process
modifications are made,
• filter area is relatively large and a resonator filter design should be used to meet the
very narrow fractional bandwidth required (0.1% in this case),
• dynamic range performance is excellent, with both the electrical circuits and a SAW
delay line exhibiting over 100 dB in a bandwidth of 180 kHz,
• the image frequency at 400 MHz can be attenuated by 50 dB using a simple antenna
matching network / filter circuit, and
• SAW filter frequency tolerance and temperature sensitivity are potential problems
(300 ppm accuracy and 3 ppm / o K temperature stability required).
53
Related work by Eikenbroek [42] and van der Plas [49] explored the feasibility of an up
conversion receiver for AM broadcast and shortwave (150 kHz - 30 MHz) using on-chip
SAW filtering. Recognizing the frequency accuracy and stability problems of the necessarily
narrowband SAW filters, they investigated the use of a selective AM detector, rather than
the use of narrowband active filtering at a 2nd IF. No selectivity figures were reported for
the proposed detector, but concerns were raised about its required complexity.
2.5.2
Ultra-Low IF Architectures
For other researchers, problems with implementing suitable on-chip bandpass filters led to
departures from the classical superheterodyne receiver. An excellent review of this work is
provided by Abidi in [40].
One of the earliest such departures was reported by Kasperkovitz [43] at Philips Research
Laboratories. In this work, an FM broadcast receiver was designed and implemented using
an ultra low IF frequency of 70 kHz. This choice places the image frequency in the gap
halfway between the desired station and the adjacent channel. Distortion problems resulting
from large deviations (∆f > 70kHz) were avoided by employing AFC feedback to compress
the peak modulation deviations to ± 15 kHz. Selectivity prior to demodulation is provided
by a 4th order continuous-time filter with a cutoff of 100 kHz. This filter gives 38 dB of
attenuation to an adjacent channel signal displaced 300 kHz away (European FM). Adjacent
channel selectivity, although not quoted, should be in the neighorhood of 35 dB after capture
ratio is subtracted from the 38 dB attenuation. Sensitivity of the receiver was quoted at
approximately 1 uV.
The Philips receiver, which was marketed as the TDA 7000, was perhaps the first IC to
qualify as a true “radio-on-a-chip”. External components were limited primarily to the
antenna, manually tuned tank circuit, batteries, and speaker power amplifier. However,
the performance was rather limited as seen by figures quoted in Table 2.13. In fact, the
54
device data sheet provides little information in the form of specifications beyond sensitivity
and maximum signal level. Adjacent and alternate channel selectivity and intermodulation
dynamic range figures are not given – possibly indicating that the device performs poorly
in these areas.
Approximately a decade later, work at SONY on developing a single-chip AM/FM stereo
receiver resulted in a non-traditional architecture similar in many ways to that of the Philips
receiver. The SONY receiver is a dual conversion superheterodyne with a 30 MHz first IF
and an ultra-low second IF. Images from the first down conversion are attenuated by a low
complexity off-chip antenna matching/filter network. Interestingly, no bandpass filtering is
provided at the 30 MHz first IF. Instead, this IF is used solely to simplify the implementation
of an image-rejection (IR) second mixer which downconverts to 150 kHz for FM and to 55
kHz for AM. An off-chip adjustment for mixer balance allows the IR mixer to attenuate
images by approximately 40 dB.
As in the Philips design, final channel selectivity is provided in FM mode by an on-chip
high order lowpass filter. In AM, selectivity is provided by an on-chip 55 kHz, 6th order,
continuous time bandpass filter with a Q of 10. Overall receiver performance is improved
over that of the earlier Philips design as seen in Table 2.13, although at the price of higher
current consumption, larger chip area, and increased off-chip component count. Selectivity
and image rejection are nevertheless quite low relative to more traditional superheterodyne
receiver designs implemented with off-chip ceramic filters.
2.5.3
Direct Conversion Architectures
In addition to these designs for FM broadcast, single-chip receiver ICs have been implemented for the radio paging market [51] [52]. Here too, alternatives to traditional superheterodyne architectures have been adopted in order to avoid on-chip bandpass filters. In
[51], Vance describes a zero IF (direct conversion) architecture which is ideally suited to the
55
narrowband FSK modulation used in pagers. The method employs in-phase and quadrature
phase (I and Q channel) mixers to generate two baseband waveforms. On-chip lowpass filters follow the mixers to provide channel selectivity, after which the signals in each channel
are amplified, limited, and fed to a flip flop detector.
The flip flop detector represents a perfect solution to the demodulation of FSK signals in
a direct conversion receiver. The flip flop is clocked by the square wave output from the
I channel limiter and latches the value of the Q channel limiter output on each I channel
rising edge. Because of the quadrature downconversion and the binary FSK signal, the Q
channel output is either 90 degrees leading or lagging from the I channel, depending on
the FSK bit polarity. Thus the data latched will either be zeros or ones, depending on the
modulation bit.
Details of the circuit implementation and performance measurements are not given by
Vance, but a similar design by Wilson, et. al. [52] reported in 1991 gives some indication
of the level of integration and performance achievable. Their design, shown in Figure 2.8,
employs a ferrite core loop antenna resonated with an off-chip tunable capacitor. This tuned
antenna is connected to an RF amplifier. An LC tuned circuit load employed with the RF
amplifier as well as the 90 degree phase splitter required for I/Q mixing are implemented
off-chip. The RF amplifier active circuits together with all remaining circuitry (except for
the crystal resonator for the LO) are implemented on-chip.
Lowpass filters consisting of a third-order Sallen-Key stage followed by a 7th order gyratorcapacitor stage provide channel selectivity, and also account for approximately half of the 18
mm2 chip area. The filter cutoff frequency is one-time tuned by an external resistor which
controls bias currents and transistor gains. As seen in Table 2.13, this high order on-chip
filtering results in good adjacent channel selectivity, although at the expense of substantial
chip real-estate. The receiver also achieves a respectable blocking dynamic range (BDR) of
82 dB at 1 MHz offset, while operating at 2.7 mA from a 2 V supply. However, it must be
56
noted that the pager IC employs the equivalent of 4th order preselection filtering (tuned
antenna plus tuned load RF amplifier) implemented off-chip which may be responsible for
part of its BDR achievement.
Figure 2.8: Single-Chip Paging Receiver. Copyright 1991 IEEE. Reprinted by permission,
from [52].
As a final example of work in highly integrated receivers, we mention the on-going Wireless
Tranceiver Project at the University of California at Los Angeles [40]. In this project, the
UCLA team is designing and implementing a two chip, direct conversion, frequency hopped,
FSK tranceiver for use in the 900 MHz ISM band. At the time of this writing, little
performance information is available outside of that reported for individual components
such as the DDS and RF amplifier chips referred to earlier in this chapter [122] [140].
57
Chapter 3
System Requirements
Commercial acceptance of integrated radio receivers will depend, to a large degree, on
the performance that can be achieved. Thus, before studying circuit level approaches to
integration, it is important to understand the system level performance requirements that
must be met. In this chapter, identification of performance requirements begins with a brief
study of the radio frequency spectrum in which modern receivers are required to operate.
An overview of important commercial wireless services that could benefit from integrated
receiver technology is then provided. Finally, the critical performance measures that most
directly affect the design of integrated receiver hardware are examined.
In Chapter 2, RF and IF bandpass filters were found to be the primary components hindering complete integration of radio receivers. Thus, discussion of receiver performance will
be heavily weighted toward requirements that have a significant bearing on filter design.
Throughout this chapter, a superheterodyne architecture will be assumed. This selection
is reasonable since nearly all present day receivers use some variant of the superheterodyne
approach, and because performance measures applicable to superheterodyne receivers are
shared to a large degree by alternative receiver designs. The issue of alternative receiver
architectures and their impact on filter design will be examined in detail in Chapter 4.
58
3.1
The Spectrum Environment
The useful radio frequency spectrum extends from ultra low frequencies in the neighborhood
of 10 KHz to millimeter wave frequencies of 30 GHz and above. Within this spectrum, a
large number of radio services exist, ranging from maritime, mobile, aeronautical, and
satellite communications, to radio and television broadcast. Recently, commercial interest
has focused on the use of VHF to L-band frequencies to provide personal, low power, pointto-point communications. Applications for this technology include radio paging, portable
and cellular telephones, and a wide range of future personal communications services such
as wireless local area networks.
These new systems reside in regions of spectrum previously allocated for other uses, and
must cooperate with a variety of existing services, as illustrated in Figure 3.1. This problem
is complicated by the fact that portable radio receivers typically employ omni-directional
antennas and are therefore susceptable to signals arriving from any source emitting radio
25 MHz
100 MHz
250 MHz
GPS
PCS
ISM
GPS
Cellular
Unlicensed/CT1
Television
Business/Govt
FM Broadcast
Aeronautical
Space
Amateur
Paging
Television
CB
Amateur
Land Mobile
Cordless Phone
Amateur
energy in their direction.
1GHz
2.5 GHz
Figure 3.1: Frequency assignments in the VHF to L-band spectrum.
To illustrate some of the specific challenges involved in modern receiver design, Figure 3.2
shows a spectrum plot of signals present in the 80 - 180 MHz frequency band. This view was
taken using a spectrum analyzer connected to a 12 inch dipole antenna at approximately
59
20 foot elevation in rural Blacksburg, Virginia. Signal densities in many metro areas are as
much as an order of magnitude higher.
Figure 3.2: Radio frequency environment from 80 MHz to 180 MHz.
The cluster of signals to the left of center in Figure 3.2 is the FM broadcast band. Within this
band, stations geographically close to the receiver appear at substantially higher amplitudes
than those farther away. In the spectrum plot shown, signal level variations of 40 dB can
be seen, whereas in some cases, variations of 80 dB or higher must be accommodated by
the receiver hardware. In addition, up to 100 stations may exist at any one time within the
FM broadcast service, each 200 KHz wide, and separated from other signals by as little as
200 KHz.1 The same receiver must therefore select, amplify, and demodulate the desired
signal in the presence of a wide range of interfering stations on nearby frequencies.
Finally, interference from neighboring services must be considered. This problem is best
1
Where possible the Federal Communications Commission attempts to allocate station frequency assignments at 400 KHz intervals or greater to minimize adjacent channel interference problems. However, in large
metro areas, stations can often be found spaced by the minimum 200 KHz interval.
60
illustrated by referring to the right side of Figure 3.2. The large signals in this portion
of the spectrum are from pager service transmitters located approximately two miles from
the receiver site. Frequencies just 15 MHz below these signals (at about 140 MHz) are
allocated to other services, including weak signal satellite applications [9]. Signals in these
services may originate from low-earth-orbit (LEO) satellites, and enter terrestrial handheld
receivers at levels as low as -120 dBm (far below the spectrum analyzer’s noise floor seen
in Figure 3.2). This level is over 80 dB below that of the paging service signals located just
15 MHz away.
3.2
Commercial Wireless Services
In addition to challenges presented by the radio spectrum environment, receiver designs
must take into account a wide range of system design parameters including operating frequency, signal bandwidth, and modulation type. These parameters can have a significant
impact on the design of integrated receivers and are therefore reviewed in this section.
A sampling of wireless services within the VHF through L-band spectrum which can benefit
from integrated, low power receiver technology is illustrated in Table 3.1. The first column
in this table shows the type of service, while the remaining columns show the system parameters that define the physical layer of the communication link. The listed parameters
are those that most directly affect requirements for integrated receiver and filter designs.
Column two indicates the frequency at which front-end preselect and image filters would
be required to operate. The range of frequencies shown also defines the required preselect
filter bandwidth, which varies from 1% to 20% of the RF center frequency depending on
the service in question.2 In cases where two ranges are shown, one range is for the base
2
Preselect filters wider than the service band can be used, but will result in potential problems with
out-of-band service interference. Preselect filters narrower than the service band are also possible, but must
track the local oscillator as the receiver is tuned. This issue is investigated in Section 3 and in Chapter 4.
61
Table 3.1: Example wireless services and system parameters
Service
Broadcast
FM
AM
Land Mobile
Public Safety
Public Safety
Paging
POCSAG
Cordless Phone
CT1
CT1/900
CT2
CT3/DCT900
Cellular Phone
AMPS
IS54
IS95
PDC/JDC
GSM
DCS1800
PCS
DECT
PHP/PHS
Satellite
ORBcomm
Band
Channel BW
Modulation
Duplex Method
Multiplex Method
88 - 108 MHz
200 KHz
None
N/A
0.54 - 1.7 MHz
10 KHz
FM w/DSB
subcarrier
AM
None
N/A
150 - 174 MHz
450 - 512 MHz
12.5 - 25 KHz
12.5 - 25 KHz
NBFM
NBFM
TDD (Simplex)
TDD (Simplex)
FDM
FDM
150 - 174 MHz
12.5 - 25 KHz
FSK
None
FDM/TDM
46.61 - 47 MHz
49.67 - 50.00 MHz
900 - 928 MHz
< 20 KHz
FM
FDD
FDMA
Various
CDMA/FDMA
864.1 - 868.1 MHz
864.1 - 868.1 MHz
100 KHz
1 MHz
Spread
Spectrum
GMSK
MSK
TDD
TDD
FDMA
FDMA/TDMA
30 KHz
FM
FDD
FDMA
30 KHz
DQPSK
FDD
FDMA/TDMA
Spread
Spectrum
FDD
CDMA
25 KHz
DQPSK
FDD
FDMA/TDMA
200 KHz
GMSK
FDD
FDMA/TDMA
200 KHz
GMSK
FDD
FDMA/TDMA
1882 - 1897 MHz
1895 - 1918 MHz
1.728 MHz
300 KHz
GMSK
DQPSK
TDD
TDD
FDMA/TDMA
FDMA/TDMA
137 - 138 MHz
≤9 KHz
PSK
TDD
FDMA/TDMA
869
825
869
825
869
-
894
849
894
849
894
MHz
MHz
MHz
MHz
MHz
825
810
940
890
935
1710
1805
-
849 MHz
826 MHz
956 MHz
915 MHz
960 MHz
1785 MHz
1880 MHz
Various
≤1.7 MHz
62
to portable link, while the other is for the portable to base link. These systems employ
full duplex (simultaneous transmission and reception), and typically must share a single
antenna between the receiver and the transmitter through the use of duplex filters.
The third column provides a rough indication of the bandwidth that the final IF filter in the
receiver is required to have. In addition, this parameter, combined with the RF frequency
gives an indication of the accuracy required in the receiver’s local oscillator. For example,
systems such as AMPS cellular have very stringent requirements on frequency accuracy
due to the use of a 30 KHz bandwidth when operating at an RF frequency of 900 MHz,
whereas systems such as DECT and IS95 have more relaxed requirements due to the use of
bandwidths in the MHz range.
The fourth column specifies the modulation method used. Besides determining the demodulator type, this feature influences the manner in which signal level variations are handled.
For example, AM and DQPSK may require automatic gain control (AGC), whereas FM,
FSK, and MSK/GMSK modulations can be decoded with a simpler limiting IF amplifier
receiver design.
Column five specifies the duplex method used. In a frequency-division duplex (FDD) system, a duplex filter is needed to simultaneously use the antenna for transmit and receive.
This is an important issue for integration of radio hardware since duplex filters must handle
substantial tranmitted power and therefore may not permit the use of active filter technology. The alternative time-division duplex (TDD) approach requires only an RF switch, and
therefore is more integration friendly.
Finally, column six specifies the multiplex method used. This aspect of the system architecture has important implications for filter tuning approaches discussed in Chapter 7.
Systems which employ some form of time division multiplexing (TDM) may allow filters to
be taken off-line during timeslots that are not of interest to the receiver, allowing simplified
tuning approaches to be developed.
63
3.3
Receiver Performance
The fundamental tasks a receiver is required to perform include:
• selection of a desired signal from a potentially dense spectral environment;
• amplification of the signal to a level suitable for demodulation; and
• demodulation of the signal to recover the transmitted information.
The ability of the receiver to carry out these tasks is typically quantified through a set of
performance measures which fall into the following general categories:
• Sensitivity;
• Dynamic Range;
• Selectivity; and
• Fidelity.
These performance measures are examined in depth in the following sections and will play a
central role in the assessment of proposed techniques for integrated receiver design throughout this dissertation.
3.3.1
Sensitivity
The sensitivity of a receiver is a measure of its ability to amplify and demodulate weak
signals. Sensitivity is often expressed in terms of the minimum signal level at the antenna
which produces some acceptable level of fidelity in the demodulator output. This measure of
64
Preselect
Filter
IF
Filter
Image
Filter
Demod
Audio/
Data
Figure 3.3: Superheterodyne receiver architecture.
sensitivity depends on the particular signaling method and demodulator design employed,
as well as on the receiver’s RF and IF circuits. This dissertation, however, is primarily
concerned with the development of techniques for integrating filters, since circuits for amplifiers and demodulators are well established. Hence, a measure of sensitivity independent
of the signaling method and receiver demodulator implementation is more appropriate. For
this reason, receiver noise figure will be the primary measure of sensitivity used.
The noise figure F of an arbitrary system can be defined as:
F =
Si /Ni
So /No
(3.1)
where S and N represent signal and noise powers and the subscripts i and o represent the
input and output of the system in question. For radio receivers, these locations are typically
defined as the antenna port and the demodulator input, respectively.
The importance of noise figure in the design of integrated bandpass filters can be understood
by referring to the receiver shown previously in Figure 3.3. In theory, it may be possible to
replace the preselect filter, low noise amplifier (LNA), and image filter in this design with a
single fully integrated active filter. If such a replacement is made however, the noise figure
65
of the filter must compare favorably with the circuit blocks it replaces to avoid degrading
the overall receiver sensitivity. For typical receivers with omni-directional antennas, this
places a requirement of 3 to 15 dB on the filter implementation, depending on the system
in which the receiver is used.
For the case of integrated IF filters, the noise figure requirements are relaxed. Here, the
impact of the filter’s noise figure on that of the overall receiver is reduced by the gain in the
receiver’s front-end. To quantify this statement, recall that the overall receiver noise figure
F can be related to the noise figure of individual portions of the receiver by [172]:
F = F1 +
F2 − 1
G1
(3.2)
where F1 and G1 represent the noise figure and gain of the front-end receiver circuits
preceding the IF filter, respectively, and F2 is the noise figure of the filter itself. Thus,
given a noise figure for the front-end and a maximum acceptable noise figure for the overall
receiver, the maximum allowed noise figure for the filter can be found from the relation:
F2 =
F
− 1 G1 F1 + 1
F1
(3.3)
For a typical receiver design, the gain G1 used is based on tradeoffs between noise figure and
dynamic range performance, and lies in the range of 10 to 10,000 (10 to 40 dB). Assuming
that the allowed degradation in receiver noise figure from the filter is 3 dB ( FF1 = 2), then
the above expression can be simplified to:
F2 ≈ G1F1
(3.4)
Expressed in dB, this equation states that the noise figure of the filter should be the sum
of the receiver’s front-end gain and noise figure. Thus, for a receiver with a 3 dB front-end
66
noise figure, a 30 dB front-end gain, and an overall noise figure target of 6 dB for the receiver
as a whole, the noise figure of the filter should be 33 dB.
While a lower noise figure in the filter will decrease the impact on the receiver as a whole, care
must be taken to avoid lowering the filter’s already limited dynamic range. For example,
analysis of Equation (3.3) indicates that the degradation in the above example can be
lowered to 1 dB by cutting the filter noise figure to 27 dB. However, this will cause a loss
of 6 dB to the effective dynamic range of the filter.
3.3.2
Dynamic Range
In Section 3.1, it was shown that receivers may be required to cope with signal level variations of up to 80 dB or more in some applications. The lowest signal level that can be
received is determined by the thermal noise floor in the environment together with the
receiver’s noise figure, while the highest signal allowed is a function of saturation and distortion effects in the receiver’s active circuits. In a traditional receiver, this overall dynamic
range is limited by the performance of amplifiers and/or mixers. In a receiver employing
active bandpass filters, the filters’ dynamic range may become the limiting factor.
In the published literature, the dynamic range of active filters is generally quoted as a
simple ratio of maximum in-band signal level to the total in-band filter noise. In receiver
design, however, several different definitions of dynamic range exist, and the most important
dynamic range measures are those that deal with interfering signals outside the signal passband. Thus, a more complete specification of filter dynamic range performance is needed.
In the subsections which follow, each of the three main definitions of receiver dynamic range
(total, blocking, and spurious-free) are reviewed, and the implications of these performance
parameters for the design of integrated bandpass filters are discussed. Detailed relationships
between the definitions presented below and the classic definition of filter dynamic range
67
will be undertaken in Chapter 6.
Total Dynamic Range
The total dynamic range of a receiver DRtot may be defined (in dB) as
DRtot ≡ Pmax − Pmin
(3.5)
where Pmax is the power above which unacceptable distortion in the demodulated signal
occurs, and Pmin is the minimum usable input signal power for satisfactory reception.
Typical values of DRtot found in modern receivers range from 40 dB to well over 80 dB
depending on the type of radio service in which the receiver is used, and on the power
consumption and cost of the receiver hardware.
The value of DRtot required in a receiver depends primarily on how close the receiver is
allowed to come to the transmitter and on how large the effective radiated power is at the
transmitter site. For FM broadcast receivers, the ratio of maximum to minimum power
encountered can reach 100 dB or more under worst case conditions, while for satellite
ground station receivers, values as low as 20 dB may be found. The value quoted for
the FM broadcast case is based on receiver design data published in references [43] and
[48], while the value for satellite reception is easily derived from the carrier to noise ratios
required at the demodulator input (e.g. 13 dB) added to a typical link margin of 7 dB [179].
For the important case of cellular telephone receivers, extensive studies of signal levels
versus distance from base station transmitters have been performed, and dynamic range requirements can be derived directly from propagation and transmitter power considerations.
The following analysis is based on information in reference [11] and illustrates some of the
factors involved.
68
Received Power (dBm)
-20
-40
-60
-80
-100
-120
0.01
0.03
0.1
0.3
1
Distance (miles)
3
10
Figure 3.4: Received signal levels versus distance from transmitter.
Figure 3.4 shows a typical cellular radio system signal level profile plotted for transmitter
to receiver separations R of 0.01 miles (53 ft) to 10 miles. This plot was constructed based
on a transmitter antenna with a directivity gain Gt = 4 (6 dB), mounted at a height of 30
m (100 ft), and driven with a power level (per user) of Pt = 10 watts. Based on this power
and directivity, the power density P in the region between 0.1 and 1 miles can be estimated
as [11]:
P ≈
Pt Gt
4πR2.4
(3.6)
and the received power Pr can then be found from:
Pr = P Ae ≈ P
λ2
4π
(3.7)
where Ae is the effective aperture of the receiving antenna, λ is the free-space wavelength
at the operating frequency, and a propagation constant of 24 dB per decade (close to that
of free-space) has been assumed. At larger distances from the transmitter (i.e. > 1 mile),
the signal falls at a faster rate of approximately 40 dB per decade due to ground reflection
69
and shadowing considerations, while at very close distances (< 0.1 miles), the signal falls
below the 24 dB per decade asymptote predicted by Equations (3.6) and (3.7) due to large
elevation angles which place the angle to the receiver outside the main lobe of the transmit
antenna [11]. Finally, given the overall signal level variation of 60 dB seen in this plot,
a total dynamic range requirement of approximately 90 dB can be derived by adding a
variance of 20 dB to account for multipath fading, and a bias of 10 dB to provide an
adequate carrier-to-noise level at the receiver’s demodulator input.
In order to translate these requirements to requirements on active filters used within a
receiver, additional factors must be taken into account. For example, in systems employing modulation such as AM or PSK, automatic gain control (AGC) amplifiers are often
used within the receiver to normalize the signal level presented to the demodulator circuits.
Since the AGC is typically placed before the IF channel selection filters, the total dynamic
range requirement on an active IF filter can be significantly smaller than that of the overall receiver. Similar considerations also apply to the RF preselect filter when switchable
attenuators are used at the antenna input port.
For example, an IF filter with a 50 dB total dynamic range can be used in a receiver
system with a 60 dB AGC and still meet a total dynamic range requirement exceeding 100
dB. Meeting a more relaxed 80 dB requirement can be achieved with a single 30 dB step
attenuator placed at any location ahead of the filter, while meeting the total dynamic range
requirement in the satellite receiver would require no AGC or attenuator at all.
Blocking Dynamic Range
The most difficult dynamic range requirements in any receiver design are those which deal
not with signal variations in the desired signal, but rather with the reception of a weak
desired signal in the presence of one or more large interfering signals outside the desired
signal channel. Such signals may reside in other channels of the service band, or in other
70
portions of the radio spectrum.
Under the condition that the desired signal is weak (e.g. near the noise floor), the full
sensitivity, and hence, the full gain of the receiver may be required. Thus, AGC and
attenuator circuits, if present, are not active and large interfering signals will be amplified
to very high levels, resulting in saturation of circuits and reduction of receiver gain. If the
gain is reduced to the point that the desired signal is no longer received, the signal is said
to be “blocked”. Thus, the blocking dynamic range (BDR) of a receiver can be estimated3
(in dB) as:
BDR ≈ P1dB − Pmin
(3.8)
where P1dB is the power of an interfering signal at the antenna that results in 1 dB gain
compression and Pmin is the minimum usable signal power defined earlier.
The BDR value required in a receiver depends on the environment in which the receiver
operates, and ideally can be found based on considerations similar to those discussed in
the preceding section. However, the problem is complicated here by the presence of a large
and variable number of potential interferers, and the existence and physical proximity of
transmitters in other service bands which may be difficult to fully characterize. Thus, an
alternative method of establishing requirements is preferable.
One useful method is to benchmark the performance of existing receivers. In fact, the
allocation of frequencies and the design of wireless services are typically based on this
approach [4] [7] [14]. An alternative approach is to study the theoretical and practical
limits to performance through an analysis of receiver design techniques. Both of these
3
This approximation assumes that noise measured at the system output is not subject to compression.
An exact analysis of the densitization problem requires a detailed characterization of circuit non-linearities,
noise mixing processes, and the system in which the circuits are used [176]. However, the given expression
is approximately valid for many situations and produces acceptable results for this discussion.
71
methods are considered in the following discussion.
The BDR of a receiver is a function of several parameters, including:
• the 1 dB compression point and noise figure of individual active circuits preceding the
IF channel select filter,
• the gain distribution preceding the IF channel select filter, and
• the frequency offset of the interfering signal causing blocking.
G = -2 dB
G = 21 dB
G = -6 dB
G = 21 dB
G = -15 dB
IF
Filter
Image
Filter
Preselect
Filter
B = 25 MHz
G = -2 dB
Pco = 0 dBm
Pco = 0 dBm
Demod
B = 25 KHz
Figure 3.5: Representative receiver hardware design.
To observe the effects of these parameters on dynamic range performance, consider the
low-power receiver design shown in Figure 3.5, in which two amplifiers, two passive filters,
and one passive mixer precede the IF channel select filter. Each amplifier is assumed to
have a compression point referred to its output Pco of 0 dBm, and a power gain of 21 dB.
The compression point for the receiver as a whole can then be found from the IF amplifier
output compression point minus the net gain preceding it, giving:
P1dB = Pco − G
72
(3.9)
or -32 dBm. To find the blocking dynamic range from Equation (3.9), the receiver noise
floor Pmin must also be known. This power may be found (in dBm) from the relation:
Pmin = 10log(kTaB) + N F + C/Nmin + 30
(3.10)
where k is Boltzmann’s constant, Ta is the antenna noise temperature, B is the signal or IF
bandwidth, N F is the receiver noise figure, and C/Nmin is the minimum required carrier
to noise ratio needed at the demodulator input. (The final factor of 30 dB converts the
power in dBW to dBm.) Assuming an antenna noise temperature of 290 K, a bandwidth of
30 KHz, a noise figure of 6 dB, and a minimum carrier to noise ratio of 10 dB, Pmin found
from Equation (3.10) is -113 dBm. Hence, the blocking dynamic range for this receiver is
BDR ≈ −32 − (−113) = 81 dB.
This value of BDR will hold for any interfering signal within the 20 MHz passband defined
by the preselect and image filters, since the receiver amplifies and translates all signals
within this bandwidth to IF. Outside this service band, BDR will generally be higher due
to attenuation of interfering signals by preselect and image filters.
The in-band BDR of this design can be improved either through an increase in the IF
amplifier compression point, or by lowering the gain ahead of the IF filter. However, both
options come at a price. Raising Pco of the amplifiers employed will generally require a
circuit design which consumes more power, and therefore may be an unacceptable option
for portable receiver operation. Alternatively, lowering the gain will increase the receiver’s
noise figure, with diminishing returns on dynamic range performance, and possibly unacceptable lowering of the receiver’s sensitivity. Thus, the art of receiver design includes
trade-offs between noise figure and dynamic range performance. Moreover, these observations demonstrate that there are practical as well as theoretical upper bounds on the
dynamic range that can be achieved within a receiver design at a given power consumption
and a specified bandwidth.
73
To validate these statements, Table 3.2 shows the measured dynamic range of three representative commercial receivers [52] [118] [111]. Since the bandwidth of the receivers affects
the value of Pmin , and hence BDR, dynamic range values are shown normalized to a constant
25 KHz bandwidth in the third column. The BDR performance of the paging receiver is in
good agreement with the example receiver calculation above. In the other two cases, BDR
is larger due to the higher power consumptions typically used in mobile and base station
receiver designs. Amplifiers and mixers in these radios often operate at power consumptions
as high as 1 W, making output compression points up to +20 dBm possible. In addition,
the higher cost and larger size of these units may permit the use of lower loss filters and
hence, lower front-end gain.
Table 3.2: Example BDR performance.
Receiver Type
VHF Pager
VHF Mobile
HF Base
BDR / BW
≤ 82 dB / 20 KHz
108 dB / 2.4 KHz
115 dB / 2.4 KHz
BDR in 25 KHz BW
≤ 83 dB
98 dB
105 dB
The implications of this discussion for the design of integrated receivers and active filters
are significant. Receiver BDR performance is seen to be dependent on power consumption,
bandwidth, and gain of circuits preceding the IF filter. Similarly, in Chapter 2, the dynamic
range of active filters was found to depend on power consumption, bandwidth, and filter Q.
Thus, active filters may offer acceptable dynamic range performance in receiver applications
despite their inherent dynamic range limitations if the filter Q is limited to moderate values.
These factors will be examined in detail in Chapter 6 after expressions for the dynamic range
of active filters have been derived.
BDR at Frequency Offsets In classic superheterodyne receiver architectures, BDR is
relatively constant for any interfering signals within the service band, but increases for
74
out-of-band signals due to the attenuation offered by the receiver’s preselect filter. For
integrated active filters, however, BDR may or may not be constant with frequency. In
fact, it will be shown in Chapter 6 that the BDR of appropriately designed filters increases
at a rate of 6 dB per octave outside the filter passband. In general, this may be less than
the improvement offered by passive filters, which may use a multipole design.
Thus, a receiver with an integrated active preselect filter may not be able to meet the
performance of existing receivers and it becomes necessary to investigate how much BDR
is actually required. This is a complex issue since it depends on the entire RF spectrum
the receiver will encounter during use. Factors such as how close the receiver may come
to transmitter installations in other services, what effective aperture and impedance level
the receiver’s antenna presents at the interfering signal’s operating frequency, and what the
active preselect filter’s input impedance is at this frequency will all play a role in making this
decision, and should be carefully assessed before committing to a fully integrated preselect
filter design.
Spurious-Free Dynamic Range
In addition to the blocking problems discussed in the preceding section, strong interfering
signals may generate spurious signals within the receiver’s front-end circuits. These signals,
which are products of non-linear intermodulation distortion in active circuits, may interfere
with the reception of a desired signal in certain situations. A measure of the receiver’s
immunity to this problem is the receiver’s spurious-free dynamic range (SFDR) discussed
in this section.
Nonlinearities responsible for spurious signal generation within a receiver are the same as
those responsible for blocking. However, since the mechanism involved is more complex
than that of simple gain compression, some review of concepts and notation is needed
before proceeding further.
75
To understand the production of spurious signals within a receiver, consider a system h
(such as a small signal amplifier) with input xi and output xo as illustrated in Figure 3.6.
For simplicity, we will assume that h is memoryless, so that the time dependence of xi
and xo may be ignored in determining the system’s nonlinear behavior. For notational
convenience, we will further assume that DC bias in the input is removed.
xi (t)
h
xo(t)
Figure 3.6: Simple nonlinear system model.
If the input xi to this system is sufficiently small, then a simple Taylor series may be used
to provide an approximation of xo of the following form:
xo ≈ h(0) +
∂h
1 ∂2h 2 1 ∂3h 3
xi +
x +
x + ...
∂xi
2! ∂x2i i 3! ∂x3i i
(3.11)
or
xo ≈ a0 + a1 xi + a2 x2i + a3 x3i + ...
(3.12)
Intermodulation distortion is produced when two or more signals at different frequencies
are present at the system input. Following established convention, consider the case of two
sinusoidal signals, at frequencies ω1 and ω2 , where ω1 and ω2 are “closely spaced”. That is,
where |ω1 − ω2 | |ω1 + ω2 |, and:
xi (t) = V1 cos(ω1t) + V2cos(ω2 t)
76
(3.13)
Inserting Equation (3.13) into Equation (3.12), expanding, and collecting terms, xo (t) may
then be written in the form:
xo (t)
V22
V12
+
= a0 + a2
2
2
3
3
3V1
3V2
2V12 V2
2V22 V1
+ a1 V1 + a3
+
cos(ω1 t) + a1 V2 + a3
+
cos(ω2 t)
4
4
4
4
2
V1 V2
V 2 V1
cos((2ω1 − ω2 )t) + 2 cos((2ω2 − ω1 )t)
+a3
4
4
+...
(3.14)
The first two terms of this result comprise the DC output of the system plus a DC offset
which depends on the second-order (quadratic) non-linearity and the input signal amplitudes. In general, this DC shift is unimportant and can be ignored. The next two terms
contain the desired output plus additional responses that account for the gain compression.
This gain compression behavior is responsible for determining the blocking dynamic range.
Finally, the last term contains the intermodulation products of interest in the following
discussion. Additional terms such as harmonics of the input frequencies are also produced,
but are not shown since they can be easily removed by filtering.
The importance of the intermodulation products can be understood by assuming that ω1
and ω2 are frequencies located one and two channel spacings above the frequency ωo to
which the receiver in Figure 3.7 is tuned.4 Under this condition, the intermodulation
product frequency (2ω1 − ω2 ) falls exactly at ωo . Thus, two strong interferers at ω1 and ω2
passing through the receiver’s amplifier and mixer circuits produce a spurious signal that
falls within the receiver’s passband and cannot be removed by any form of filtering. If the
spurious product is of sufficient amplitude, it may dominate the desired signal and prevent
4
This choice of frequencies is far from unique. Many other channel spacings will produce the effects
described, especially when more than two signals are considered.
77
successful reception. Alternatively, if no desired signal is currently present, the spurious
product may be falsely interpreted by the receiver as a valid signal. Whether or not either
of these conditions occur will depend on the amplitudes V1 and V2 of the interferers and on
Preselect
Filter
ωο
ω1
ω2
ωο
ω1
ω2
the nonlinearity term a3.
IF
Filter
Image
Filter
Demod
Figure 3.7: Intermodulation products within a receiver front-end.
For simplicity, assume that V1 = V2 = Vi . Then, the amplitudes of each of the expected
outputs at ω1 and ω2 from the interferers due to the gain coefficient a1 of the system
(ignoring gain compression effects) is:
Vo1 = a1Vi
(3.15)
and the amplitudes of the intermodulation product resulting from the a3 term in the nonlinearity expansion is of the form:
Vo3 = a3
Vi3
4
(3.16)
If the input and output impedances of the circuit are specified, these relationships can be
written more simply in terms of power in dB as:
78
Po1 = Pi + G
(3.17)
Po3 = 3Pi + constant
(3.18)
and
where Pi is the input power, G is the small signal power gain, and Po1 and Po3 are the
output power of the first-order (expected) and third-order (spurious) signals, respectively.
These expressions are plotted in Figure 3.8 together with the effects of gain compression
and higher-order terms of the series expansion not considered above. This figure provides a
convenient illustration of the important performance parameters related to receiver dynamic
range, including [112]:
G
The small signal amplifier gain
Pci
The gain compression point referred to the input
Pco
The gain compression point referred to the output
IIP3
The third-order intercept point referred to the input
OIP3
The third-order intercept point referred to the output
M DS
The minimum discernible signal, (noise floor referred to the input),
and
Pi3
The input power at which intermodulation products rise above the
noise floor.
If the circuit noise floor (in some specified bandwidth) is known, both the blocking dynamic
range and the spurious-free dynamic range can be found in terms of these parameters
through relatively straightforward graphical constructions.
In Figure 3.8, the noise floor referred to the output is represented by the upper border of
79
Po (dBm)
Intercept Point
OIP3
P
o3
P
o1
Pco
G
Pi3
MDS
Pci IIP
3
P i (dBm)
SFDR
BDR
Output Noise Floor
Figure 3.8: Graphical determination of dynamic range.
80
the shaded region. The MDS or noise floor referred to the input is then found as the input
power at which the Po1 curve hits the noise floor, and the blocking dynamic range can be
found as the distance between this point and the 1 dB compression point Pci as shown
below the x axis, giving:
BDR = Pci − M DS
(3.19)
The spurious free dynamic range (SFDR) is defined as the input power Pi3 above which
third-order intermodulation products rise above the noise floor, divided by the MDS. Expressed in dB, the SFDR is of the form:
SF DR = Pi3 − M DS
(3.20)
The power Pi3 can be found in terms of the third-order input intercept point IIP3 and MDS
through simple geometric considerations. Based on the slopes shown in Figure 3.8, for the
Po1 and Po3 curves, Pi3 is given by:
1
Pi3 = IIP3 − (IIP3 − M DS)
3
(3.21)
Combining this result with equation (3.20), the following expression for SFDR results.
2
SF DR = (IIP3 − M DS)
3
(3.22)
Thus, from equations (3.19) and (3.22), both the BDR and SFDR figures of merit can be
found, provided the 1 dB compression point and third-order intercept points are known.
These quantities are often specified by manufacturers or measured for new designs. In
81
[119], it is suggested that these two points may also be found in terms of each other. This
conclusion is based on empirical evidence showing that IIP3 and Pci can be related by:
IIP3 = Pci + ∆
(3.23)
where ∆ lies in the range of 5 to 15 dB, depending on details of circuit implementation.
In [176], Meyer and Wong have subsequently noted that a ∆ value of 9 dB can be derived
directly from Equation (3.14) if the Taylor series expansion is truncated after the cubic
term. Combining Equations (3.19), (3.22), and (3.23), and using ∆ = 9 dB, yields:
2
SF DR ≈ (BDR + 9)
3
(3.24)
Based on Equation (3.24), all of the conclusions concerning BDR performance of receivers
arrived at in the preceding section are seen to apply to SFDR performance as well. In
particular:
• There are upper bounds on the spurious-free dynamic range that a receiver can
achieve. As in the case of BDR, these limits depend on the power consumption
of the active circuits employed and on the gain preceding the IF channel select filter.
• Increasing the receiver’s SFDR performance requires either an increase in amplifier
compression points, which requires higher power consumption, or a reduction in gain,
which sacrifices the overall noise figure, raising the receiver’s noise floor together with
the the input compression point and produces diminishing results.
• The SFDR of a superheterodyne receiver is relatively constant within the passband
defined by the preselect filter, and increases outside this passband by an amount
determined by the attenuation characteristic of upstream filters.
82
SFDR requirements for signals which fall within the preselect filter’s passband can be established either through an analysis of receiver hardware, or through benchmark measurements
in commercial receivers. For the case of the receiver design shown previously in Figure 3.5,
for which a BDR estimate of 81 dB was found, Equation (3.24) yields a SFDR of 60 dB.
This value agrees well with values assumed by studies such as those used in the Federal
Aviation Administration’s software model used to allocate operating frequencies [14].
As an additional validation of the theory, measured results cited in [113] [114] are given
in Table 3.3. These data values, taken from Amateur Radio receivers operating in the 144
- 148 MHz service band, are for FM receivers operating with a bandwidth of 12.5 KHz.
Values shown in the table have been normalized to a 25 KHz bandwidth by subtracting 2
dB from the reported values.
Table 3.3: Example SFDR performance.
Receiver Type
Mobile
Mobile
Mobile
Handheld
Handheld
Handheld
In-Band SFDR
67 dB
68 dB
71 dB
58 dB
60 dB
64 dB
SFDR at 10 MHz Offset
84 dB
94 dB
85 dB
76 dB
83 dB
74 dB
As in the case of BDR measurements, mobile receivers, which consume higher powers than
handheld units, generally provide the highest performance. For handheld units, the SFDR
typically falls in the range of 60 - 65 dB (for 25 KHz bandwidth), validating the general
conclusions above.
Out-of-band SFDR measurements are also shown in the table for a 10 MHz offset from
the center of the service band giving an indication of the level of performance that may be
acceptable. These values were measured with the receiver tuned to 146 MHz and interferers
placed at 156 MHz and 166 MHz to simulate the effects of large adjacent service band paging
83
signals. As expected, the values are higher than the in-band measurements. However, the
improvement shown is suggestive of only a 1-pole to 2-pole roll-off in the receiver’s preselect
filter. Hence, the concerns raised in the preceding section relative to using active filters in
preselect applications may not represent a serious problem.
3.3.3
Selectivity
The selectivity of a receiver is a measure of its ability to amplify and demodulate signals at a
desired frequency while rejecting signals at other frequencies in the environment. While the
blocking and intermodulation characteristics of receivers discussed in the previous section
can limit the selectivity obtainable, selectivity is primarily a function of IF channel select
filter design.
For the classic superheterodyne architecture, which provides the basis for most modern
receiver designs, primary measures of receiver selectivity include:
• adjacent/alternate channel selectivity;
• image rejection; and
• spurious response rejection.
Adjacent and Alternate Channel Selectivity
To determine channel selectivity requirements of a receiver, the distinction between adjacent
and alternate channels must first be understood. The term “adjacent channel selectivity”
is often used in the literature to refer to any channel in the receiver’s service band which
is near the channel being received. In this dissertation, however, a more precise definition
will be assumed in which an adjacent channel is either the channel immediately above or
84
immediately below the one to which the receiver is tuned. The term “alternate channel”
will then be used to refer to signals either two channels above or below the channel tuned.
Channels at larger distances will not be specifically referenced.
For practical as well as economic reasons, radio system designers often specify that two
signals in the same geographic area may not reside in adjacent channels. This system
constraint, which has been adopted by broadcast services and by first generation cellular
services, relaxes performance demands on both receiver IF filtering and on transmitter
spectral purity. An alternative approach that achieves the same result is to constrain the
signal bandwidth to some fraction of the channel spacing so that “guard bands” occur
on either side of the transmitted signal. In systems employing this method of spectrum
management, signals within the same geographic area may be assigned to adjacent channels,
but the spectrum utilization remains comparable to that above due to decreased usage of
the channel bandwidth. A third option is to rely on the fact that the spectral energy
in a transmitted signal employing certain types of modulation and program material is
statistically concentrated at the channel center, so that guard bands are implicitly present.
In the discussions below, this third case will be treated as equivalent to the second in which
guard bands are explicitly defined.
These issues are illustrated in Figure 3.9 in which the attenuation of an IF channel select
filter versus frequency offset from the filter center is plotted. For the case where guard
bands are not used, the filter bandwidth is generally equal to the channel spacing as shown
by the lower attenuation curve. Assuming a Butterworth-type filter rolloff5, the mean
attenuation of this filter and hence the approximate receiver selectivity at adjacent and
alternate channels is:
5
The rolloff of other all-pole filter types commonly used in receivers is similar. Somewhat higher attenuations can be obtained in Chebyshev designs or in filters employing zeros to produce elliptic responses.
However, the phase and group delay characteristics of these filters often prevent their use in receiver IF
applications.
85
AT T NAdjCh = 6N dB
(3.25)
AT T NAltCh = 12N dB
(3.26)
where N is the number of filter poles.6
Sy
ste
m
Sy
wi
th
ste
m
ou
tg
wi
ua
th
rd
gu
ba
nd
ard
s
ban
ds
ATTN
f
chan -2
chan -1
chan 0
chan 1
chan 2
Figure 3.9: IF channel select filter attenuation versus frequency.
Equation (3.25) implies that a selectivity of 50 dB to adjacent channel signals would require
a filter with at least 8 poles. In fact, significantly more than 8 poles would be needed if the
adjacent channel signal energy is evenly spread across the channel bandwidth. In contrast,
50 dB alternate channel selectivity requires a filter with a more realistic complement of 4
to 5 poles.
For the case where guard bands are present in the system architecture (either explicitly
or implicitly), the IF filter bandwidth can be reduced as illustrated. This allows good
selectivity in adjacent channels to be achieved with a similar number of poles as that in the
case without guard bands above.
6
Following established convention for bandpass filter designs, an N-pole filter is defined here as one of
order 2N. That is, the number of poles refers to the order of an equivalent lowpass prototype filter from
which the bandpass filter can be derived.
86
Requirements for the number of poles used in receivers incorporating on-chip IF channel
select filters can be determined through the considerations discussed above if specifications
for adjacent or alternate channel selectivity are given. For example, the AMPS cellular
phone discussed in Chapter 2 lists specifications of > 16 dB adjacent channel selectivity and
> 65 dB alternate channel selectivity, implying a requirement of 5 to 6 poles. Alternatively,
filter requirements may be determined directly by examining the hardware design of existing
products. Examples of filter complements for several radio receiver designs are summarized
in Table 3.4 indicating that 4 pole filtering is acceptable in many applications, while 6 pole
designs may be found in more expensive products and in more demanding applications.
Table 3.4: Example IF filter complements.
Receiver Type
Portable FM Broadcast
Hi-Fi FM Broadcast
Cordless Phone
DECT
FM Cellular
Filter Poles
2-4
4-6
4
3 - 4 (SAW)
6
Image Rejection
The image rejection of a receiver is the attenuation offered by the receiver’s preselect and
image filters to signals residing at the image frequency fim . The frequency fim is a function
of the IF frequency fIF used, and can be found from:
fim = fRF ± 2fIF
(3.27)
depending on whether the LO operates above or below the RF frequency.
The image rejection offered by a preselect filter in a given receiver design can be found
87
by examining Figure 3.10. The attenuation curve of a preselect filter with bandwidth B
is shown under conditions where the receiver is tuned to the center of the filter passband
and to the band edge. The former case is applicable to tracking preselect filters, whereas
the latter yields worst-case image rejection for the more common case of receivers with
fixed-tuned preselect designs.
ATTN
f
fRF
fimage
B
Figure 3.10: Preselect filter attenuation versus frequency.
From these attenuation curves, the image rejection offered by an N-pole filter design to a
signal at the image frequency can be found from:7
AT T N ≈ 6N log2
2fIF
B/2
(3.28)
This equation can be used to determine the preselect filter order and IF frequency required
to yield a specified requirement. For example, an AMPS analog cellular phone requires
B = 25 MHz in a fixed-tuned preselect receiver architecture. Using the typical choice of a
45 MHz IF, the number of poles required to provide 60 dB worst case image rejection is 4.
Alternatively, for a DECT phone operating with B = 20 MHz and fIF = 110 MHz, a 60
dB image rejection can be achieved with only 3 poles. Finally, a manually tuned, low-cost
FM broadcast receiver typically employs an IF of 10.7 MHz and a tracking preselect filter
7
The attenuation for the fixed-tuned filter with the receiver tuned to the band edge will be slightly less,
but the given equation is sufficiently accurate for this discussion provided that fI F ≈ 2B or greater.
88
with B = 2 MHz. Assuming a somewhat relaxed image rejection requirement of 50 dB, the
number of poles required in this design is 2.
The level of image rejection required in any given receiver will depend on the location of
the image frequencies in the RF spectrum, and on the expected proximity of the receiver
to transmitters operating on those frequencies. These issues must be carefully considered
when choosing IF and LO frequencies during the receiver design process. For example, an
image rejection requirement of 50 dB or less for the FM broadcast receiver discussed above
may be permissible since the image frequency with a low-side injection LO design falls into
the Aviation band at 110 - 130 MHz. Outside of the immediate vicinity of major airports,
these frequencies tend to be sparsely populated and contain only low power signals.
Spurious Response
In early receiver designs, mixer circuits often created frequency conversions other than
those desired, leading to spurious responses on a wide range of possible frequencies. This
problem is less severe in modern receiver implementations due to the use of improved mixer
technology and IF frequencies which are either well below or well above that of the RF
signal. However, spurious response problems can still arise in multiple-conversion receiver
architectures.8
In multiple-conversion designs, images resulting from conversion from one IF to the next
create spurious responses, which typically fall at frequencies within the receiver’s own service
band. Hence, the spurious response attenuation requirements are of the same order of
magnitude as those for alternate channel rejection (e.g. ≥ 60 dB).
As an example, consider the AMPS cellular phone discussed in Chapter 2, in which a dual
8
Spurious responses may also arise when complex synthesizer designs are used for tuning. Only the
problems associated with filters are covered here.
89
conversion architecture was employed with a first IF at 45 MHz and a second IF at 455
KHz. The mixer used in the down conversion from 45 MHz to 455 KHz will produce an
image at frequencies either 910 KHz above or below the desired signal, and the extent to
which this response is attenuated will depend on the bandwidth and order of the first IF
filter.
The filter order required to meet a given spurious specification depends on the ratio of the
second IF frequency f2IF to the first IF filter bandwidth B1IF , in a manner similar to that
for the preselect filter considered earlier.
AT T Nspur ≈ 6N log2
2f2IF
B1IF /2
(3.29)
The actual number of filter poles needed for an early IF filter depends heavily on tradeoffs
made in the overall receiver architecture. This issue will be considered in more depth in
Chapter 4. However, as an example calculation, consider a receiver in which a first IF filter
bandwidth of 2% (selectivity Q of 50) is used. Assume further that the second IF frequency
is 10% of the first IF frequency to allow the selectivity Q of the final channel selection filter
to be decreased by a factor of 10 relative to a single-conversion design. Then, to provide 60
dB of spurious response rejection, Equation (3.29) shows that the first IF filter requires N
= 3 poles.
3.3.4
Fidelity
Discussions of selectivity in the previous section have addressed filter requirements needed
to achieve certain levels of stopband rejection. However, no requirements have yet been
determined for filter in-band response. In this final section, in-band response characteristics
are briefly considered and related to the issue of receiver fidelity.
90
Fidelity is a measure of a receiver’s ability to accurately demodulate the information contained in the desired radio signal. In an analog FM receiver, fidelity may be measured in
terms of the total harmonic distortion present in the recovered audio waveform, while in
a digital system, fidelity is typically expressed in terms of symbol or bit error rate. These
factors depend on both the filtering used, and the demodulator design employed.
Assuming the demodulator is ideal, the receiver’s fidelity can be related to the IF channel
select filter’s in-band response.9 A perfect filter would provide a flat amplitude response
and a linear phase over the signal bandwidth producing only scaling and delay. Real filters
however, possess neither of these characteristics and an assessment of amplitude and phase
deviations on fidelity is needed. Using appropriate analysis techniques, the effects of various
nonideal amplitude and phase response features can be determined, and the results can then
be used to specify filter performance based on fidelity requirements. This analysis is complex
and outside the scope of this discussion. A simpler alternative approach to determining
filter requirements is thus desired. As in the case of dynamic range, benchmarks relative to
existing receiver designs can serve this purpose.
In Chapter 2, the most common filter type used in receiver channel select filtering is the
MCF design. These filters are members of a broad class of recursive filters with similar response characteristics known as coupled-resonators. Thus, any filter with in-band responses
matching or approximating that of a coupled-resonator design should prove acceptable.
In some newer PCS products operating at higher frequencies and larger bandwidths, surface
acoustic wave filters may be used. A survey of manufacturer data shows that the response
of these filters varies considerably depending on the type of SAW construction (transversal
or resonator), and on the filter cost. However, despite their potential for highly linear phase,
many of these filters exhibit responses that are similar to those of simple recursive Butter9
The effect of preselect and image reject filters is generally negligible because the gain and phase variations
over the signal bandwidth are minimal.
91
worth forms. Thus, replacements for these filters should provide acceptable performance
provided their amplitude and phase responses are at least as good as that of Butterworth
designs.
92
Chapter 4
Alternative Receiver Architectures
In the last chapter, a superheterodyne receiver architecture was assumed in discussing
system and filter requirements. Although this choice is generally reasonable, it contains
an implicit assumption that receiver channel selectivity is best achieved with fixed-tuned,
high-order, bandpass filters operating at relatively low frequencies. In light of the difficulties
in integrating bandpass filters identified in Chapter 2 and changes in RF device technology
which occur over time, this assumption may no longer be valid. Indeed, there is renewed
interest today in alternative receiver designs which can decrease, or if possible, eliminate
the need for on-chip bandpass filters altogether. These architectural alternatives are the
subject of this chapter.
The study of alternative receiver architectures begins with an overview of early receiver
designs, providing an important historical context. Next, the superheterodyne architecture
and its many variations are examined in detail. One of these variations, the design of
superheterodyne receivers with ultra-low IFs, then leads to the concept of direct conversion receivers in which received signals are converted directly to baseband and all channel
selection filtering is done with lowpass filters. Finally, an “ideal” receiver architecture is
proposed in which channel selection filters are moved to the receiver’s front-end to provide
good performance with minimum possible power consumption.
93
4.1
Early Receiver Architectures
The earliest radio receiver architecture was developed by H.R. Hertz in 1887 [24]. Hertz’s
receiver was designed purely as a means to validate the existence of radio waves predicted
by Maxwell 23 years earlier. The receiver operated at a frequency of 31 MHz and consisted
of a simple loop of wire broken by a microscopic gap. This arrangement produced a weak
spark when radio energy was created in Hertz’s laboratory through the use of a powerful
spark gap transmitter.
Commercial development of radio receivers began approximately 10 years later in 1896 when
Marconi demonstrated reception of signals over distances of up to 9 miles. Marconi’s system
used a spark gap transmitter similar to Hertz’s, and a receiver consisting of a battery,
coherer, and bell. These “demodulator” circuits were connected directly to an antenna
constructed with two metal plates, one fixed to a tree and one buried in the ground.
The concept of tuned radio receivers using LC circuits was developed at about the same
time by Sir Oliver Lodge, and patented in 1897. Later in 1899, Marconi patented similar
technology, giving the Marconi Wireless Telegraph and Signal Company a near monopoly
on radio design, and setting the stage for rapid progress in subsequent years.
4.1.1
Tuned RF Receivers
Lodge’s, Marconi’s, and Hertz’s earlier receivers can all be classified as “Tuned-RF” architectures in which selectivity and detection are provided by circuits operating at the same
frequency as the transmitted signal. The general concept is illustrated in Figure 4.1.
With the invention of the triode vacuum tube by de Forest, and the development of radio
telephony (AM voice transmission) by Fessenden and Alexandersen at the turn of the century, more sophisticated tuned-RF receivers emerged to handle an increasingly congested
94
Bandpass
Filter
Detector
Figure 4.1: Tuned RF receiver architecture.
RF spectrum [24]. These receivers used amplifiers and cascaded stages of tuned circuits
(essentially high order filters) to provide both improved sensitivity and increased selectivity. Thus, within only a few years, tuned-RF receivers reached a relatively high level of
sophistication as illustrated in the design of Figure 4.2 which was patented in 1913.
Figure 4.2: Early tuned-RF receiver circuit. [British patent no. 147,147]
Nevertheless, tuned-RF receivers required a fair amount of patience and skill to manually
adjust the filtering circuits used. This fact, combined with the cost of using three or more
95
tubes and the difficulties of extending the designs to higher frequencies, motivated the
development of a variety of alternative approaches discussed in the following sections.
4.1.2
Regenerative Receivers
At about the same time that multi-stage tuned-RF receivers were being built, de Forest,
von Strauss, Meissner, and others discovered that coupling between the plate coils and grid
coils of a tuned amplifier could result in the phenomonon of “reaction” or regeneration [24].
This effect provided a wealth of benefits including an increase in amplification, a reduction
in the number of tubes needed within the receiver, and high selectivity without the need for
multiple, independently tuned circuits. A block diagram of a regenerative receiver is shown
in Figure 4.3 in which the simplicity of the design is readily apparent.
Bandpass
Filter
Detector
Figure 4.3: Regenerative receiver architecture.
Although the effects of regeneration were not well understood at the time, they may be
derived quite easily using modern feedback theory. To quantify these effects, assume that
the filter shown in Figure 4.3 is a basic second-order bandpass design with transfer function
H(s) given by the relation:
H(s) =
s ωQo
s2 + s ωQo + ωo2
(4.1)
where ωo is the center frequency and Q is the quality factor. With the numerator shown,
96
the midband gain of H(s) is 1.0, while the 3 dB bandwidth B (in Hz) is related to the Q
through the well known expression:
B=
2πωo
Q
(4.2)
If the amplifier in Figure 4.3 has a gain of A (A < 1), then from feedback theory, the closed
loop gain HCL (s) of the regenerative circuit in the dashed box is of the form:
HCL (s) =
AH(s)
1 − AH(s)
(4.3)
Substituting Equation (4.1) into Equation (4.3) then yields:
HCL (s) =
s ωQo A
s2 + s ωQo (1 − A) + ωo2
(4.4)
for which the midband gain, effective Q, and bandwidth are:
|HCL (ωo )| =
(4.5)
Q
1−A
(4.6)
2πωo
= B(1 − A)
QCL
(4.7)
QCL =
BCL =
A
1−A
As an example, assuming A = 0.99, the gain is increased by a factor of 100 (40 dB), the Q
is multiplied by 100, and the bandwidth is reduced by 100.
97
Despite these useful effects, regenerative receivers have a number of limitations. One serious
limitation is the need for critical manual adjustment during use to achieve precisely the value
of A desired. If the regeneration is advanced too far, the receiver will break into oscillation
and reception of the desired signal will be lost.
Although oscillation represented a problem for regenerative receivers intended for AM reception, it was recognized by de Forest and others as a means of producing sine waves
electronically, and was exploited to create heterodyne detectors for receiving continuouswave (CW) telegraph signals. A more elegant solution to the reception of these signals
however, was invented by Round [24]. Round showed that if a regenerative amplifier was
operated at exactly the point of entering oscillation, it could function simultaneously as
a regenerative amplifier, heterodyning oscillator, and audio detector. This special case of
regenerative receiver design known as an “oscillating detector” was popular for many years,
especially within the amateur community because of its excellent performance and low cost.
4.1.3
Super-Regenerative Receivers
Regenerative receivers were manufactured until at least the 1960’s because of their ability
to minimize the number of active devices used, and hence to minimize cost. However, their
reliance on critical manual adjustment of the regeneration control was a major drawback to
their design. In an early attempt to address this problem, Bolitho and Armstrong developed
the super-regenerative architecture shown in Figure 4.4 in 1919 and 1922, respectively [24].
The principle of super-regenerative design was to employ regeneration with a feedback
gain of slightly greater than one. Although this assured that oscillations would build up,
a “quenching” circuit was used to periodically damp the oscillations as shown. With this
setup, the presence of an input signal caused oscillations to build up at a rate that increased
with increasing signal amplitude. Hence, the average magnitude of oscillation energy provided an indication of both the presence and strength of the received signal.
98
Bandpass
Filter
Detector
Quench
Oscillator f
Q
t
1
fQ
Figure 4.4: Super-regenerative receiver architecture.
Super-regeneration may be viewed as a simple, yet somewhat crude (by today’s standards)
method of automatically controlling the regeneration setting, thereby eliminating the need
for manual adjustment. However, while the need for manual control is eliminated, the superregenerative architecture retains many other limitations of the basic regenerative concept
including:
• Bandpass filtering is limited to a single-pole rolloff;
• Oscillations which occur at the received frequency radiate through the antenna, interfering with other receivers;
• Selectivity decreases for large input signals due to limited dynamic range; and
• Bandwidth is dependent on gain and therefore not matched to the signal being received.
These problems were overlooked in early regenerative and super-regenerative receivers because of the cost benefits afforded by reductions in the number of active devices required.
However, with the advent of integrated circuits, this advantage disappeared, and the superheterodyne receiver became the architecture of choice in virtually all receiver designs.
99
4.2
Superheterodyne Receivers
To
Low Frequency
Receiver
Preselect
Filter
Figure 4.5: Armstrong’s original superheterodyne receiver.
The concept of heterodyning an incoming signal to convert it to a lower frequency was
developed by Armstrong and others in 1918. Armstrong’s original design, shown in Figure
4.5, was intended to allow low frequency radiotelephone receivers to be adapted for use at
newer HF frequencies being used in Europe. However, it was quickly recognized that the
basic approach offered many additional benefits, including:
• The low-frequency receiver (typically a high quality tuned-RF design) could be adjusted once, and thereafter all tuning could be done by varying the heterodyne oscillator.
• Amplification could be provided primarily at a lower frequency where high gains were
easier to achieve.
• Amplification was split between two frequencies, so that the risk of unwanted regenerative feedback could be reduced.
• Narrow, high-order filtering was more easily achieved in the low frequency receiver
than at the actual incoming RF frequency being received.
100
Eventually, the separate tuned-RF receiver was replaced by the dedicated IF section of the
modern superheterodyne design, in which pre-tuned fixed-frequency filters are employed.
The result became the well-known architecture used today with high quality channel-select
filtering and no adjustments aside from volume and tuning controls.
4.2.1
Modern Single Conversion Implementations
The essential components of a modern single conversion superheterodyne receiver architecture are shown in Figure 4.6 and briefly reviewed in the following paragraphs.
In the superheterodyne design of Figure 4.6, the preselect filter serves both to attenuate
signals outside the service band and to provide partial rejection of the image frequency
created by the heterodyning process. This filter must have low insertion loss to minimize
degradation of the receiver’s noise figure.1
Optional AGC
IF
Filter
Image
Filter
Preselect
Filter
Demod
LNA
Figure 4.6: Modern superheterodyne receiver.
After preselection, a low noise amplifier (LNA) provides moderate gain to overcome losses
in passive circuits up to the input of the first IF amplifier. This gain is needed to provide
1
In some designs (such as satellite receivers with directional dish antennas), this filter may be placed after
the LNA to achieve the best possible noise performance. However, in most terrestrial receiver applications,
this option is unacceptable since it leaves the LNA exposed to all signals throughout the RF spectrum.
101
good noise figure performance, but must be minimized to avoid degrading dynamic range
as discussed in Chapter 3.
Following the LNA, an image filter is provided. The purpose of this filter is two-fold. First,
it supplements out-of-band attenuation provided by the preselect filter, further increasing
image rejection performance and improving BDR and SFDR performance relative to outof-band interferers. Its primary purpose however is to improve noise figure performance by
attenuating noise at the image frequency generated in the LNA and translated to IF in the
mixing process.
Next, the mixer and LO convert the signal to the selected IF frequency where additional
moderate levels of gain may be provided. Thereafter, the IF channel select filter passes only
the signal being received. IF amplification can then be used as needed to increase the signal
to an amplitude that can be conveniently processed by demodulator circuits to recover
the transmitted information. In analog FM and digital FSK systems, the post-filter IF
amplifier subsystem is typically designed to provide very high gains, intentionally forcing
the amplifiers into amplitude limiting. Intermodulation products are (ideally) no longer
important at this stage in the receiver since all interfering signals have been attenuated by
the IF channel select filter. In other designs intended for use with AM and digital phase
modulation, automatic gain control is usually employed so that amplitude information is
retained and minimal phase distortion occurs.
4.2.2
Multiple Conversion Implementations
The design of superheterodyne receivers involves many trade-offs including selection of
IF and LO frequencies to meet image rejection and spurious response objectives, and to
minimize the complexity of required bandpass filter components. If the service band of
interest is sufficiently narrow relative to the band center frequency, then the single conversion
design described in the preceding section can be implemented without requiring an excessive
102
number of filter poles. However, if the service band is wide, either a tracking preselect filter
or a multiple conversion design may be needed.
The tracking design illustrated in Figure 4.7 employs preselection and image filters with
bandwidths smaller than the width of the service band of interest. Hence, the number of
poles required to achieve a given image rejection performance can be reduced while still
converting the signal to a frequency sufficiently low to avoid an excessively narrow IF filter
fractional bandwidth. The price paid in this case is the need for a narrow bandwidth,
tunable filter, and associated tracking between the filter and the local oscillator.
Optional AGC
IF
Filter
Image
Filter
Preselect
Filter
Demod
Filter Tuning
Figure 4.7: Single conversion superheterodyne with tracking preselect.
The alternative multiple conversion design shown in Figure 4.8 converts the incoming signal
first to an IF at a relatively high frequency, and then to a lower second IF. This approach
minimizes the number of poles needed in the preselect/image filters and allows narrow
channel bandwidths to be handled with lower Q filter designs. The price paid in this design
is additional circuitry, and the associated power consumption needed to maintain good
dynamic range performance in all active devices ahead of the final channel select filter.
The tracking preselect design has historically been difficult to implement due to problems
in realizing high quality tunable RF filters. However, in considering receiver integration
using active filters, this architecture may prove to be desirable. Integrated active filters
necessarily include tuning to address manufacturing tolerances and temperature drift, and
103
Preselect
Filter
Image
Filter
IF1
IF2
IF
Filter
IF
Filter
LO1
Demod
LO2
Figure 4.8: Multiple conversion superheterodyne.
this tuning can be used to advantage in this application. Nevertheless, the degree to which
the filter can be narrowed will depend on the resulting Q values needed. This issue is
considered in Chapter 6, where relationships between Q and dynamic range are derived in
detail.
Integration of a multiple conversion design is also a potentially attractive alternative when
faced with the quality factor limitations of on-chip active filters. In this case, the increased
first LO frequency and the low second IF frequency simultaneously decrease the Q and
number of poles needed in both the preselect and channel select filters. However, the need
for an additional local oscillator and the additional power consumed must be carefully
considered.
4.2.3
Up Conversion Implementations
Historically, one of the main goals of the superheterodyne architecture was to convert the
incoming RF signal to the lowest frequency possible, allowing use of less expensive amplification and filtering components. However, with the development of semiconductors
operating at several hundred MHz and above, and of quartz and surface acoustic wave
filters with fractional bandwidths well under 1%, new architectural options have become
104
available. Around the late 1970’s and early 1980’s, these options began to be exploited
heavily in the form of up-conversion receiver designs.
For very wide service bands (e.g. 0.1 to 30 MHz), achieving good image rejection with down
conversion superheterodyne designs requires elaborate tracking filters and band switching
mechanisms. Image rejection requirements can be met more easily however, if the incoming
signal is first upconverted to an IF frequency above the desired RF frequency as shown in
Figure 4.9. Here, both the LO frequency and the IF frequency are relatively high, placing
the image frequency well above the highest frequency in the service band. Hence, preselect
and image filters can be implemented with simpler lowpass filter designs.
Lowpass
Filter
fIF2 < fIF1
fIF1 > f RF
f RF
IF
Filter
Lowpass
Filter
LO1
IF
Filter
LO2
Demod
Optional Second IF
Figure 4.9: Up-conversion supetheterodyne.
Up-conversion architectures are especially useful in applications such as AM and FM broadcast receivers where the service band is broad and the RF frequency is “low” relative to
the performance of active devices. However, one of the drawbacks of this architecture is
the need for narrow IF filter fractional bandwidths if full channel selectivity is desired at
the up-converted frequency, or if subsequent down conversion to a low frequency is desired.
In receiver designs with discrete filters, quartz and surface acoustic wave (SAW) filters are
often required, leading to increased product cost.
Work on partially and fully integrated AM/FM broadcast receivers has been done at Delft
University of Technology in conjunction with Philips Laboratories [45] [47] [53]. In [45]
105
and [47], a partially integrated solution for the AM portion of a car radio is described in
which a 10.7 MHz IF is adopted and an off-chip crystal MCF is used to provide the small
fractional bandwidth needed. In [53], the design of an up conversion FM broadcast receiver
is described and the use of an on-chip SAW filter operating at 160 MHz is investigated. To
achieve acceptable channel selectivity at the 160 MHz IF, the IF filter in this application
requires a selectivity Q of 800 (160 MHz / 200 KHz), leading the investigators to conclude
that a coupled resonator SAW design with physical length of at least 6 mm was required.
In addition, the temperature stability needed to maintain the filter centered on the IF
frequency was cited as a significant problem to be solved.
Integrated active LC filters could also serve as a first IF in up-conversion designs. However,
the Q values needed are generally beyond those that can be realized with adequate dynamic
range, so that one or more subsequent down conversions would be required. Thus, this
architecture does not appear to offer a good solution to the receiver integration problem.
4.2.4
Designs with Ultra-Low IFs
Problems with image responses can be reduced in some special situations without actually
attenuating the image frequency at all. The key to achieving this result is to carefully
select the RF to LO frequency relationship so that the image frequency falls within an
unused portion of the RF spectrum. This concept led to the development of the ultra-low
IF integrated receiver technology described briefly in Chapter 2 and detailed below.
The ultra-low IF architecture is illustrated in Figure 4.10, where the RF frequency fRF is
related to the LO frequency fLO through the channel bandwidth B:
fLO = fRF + KB
106
(4.8)
Lowpass
Filter
Demod
Figure 4.10: Ultra-low IF superheterodyne.
Assuming that the constant K is selected as K = 12 , the RF signal will be down converted
to an IF frequency of fIF =
B
2,
and channel selectivity can be provided with a lowpass filter
with cutoff frequency fc = B. Assuming also that operating frequencies in the system are
assigned so that adjacent channel signals are not occupied, the image response, which falls
at the center of the adjacent channel, will contain no energy and will not interfere with
reception of the desired signal. This basic approach can also be used in systems in which
the adjacent channel is occupied, provided that guard bands are present. For example, if
guard bands account for half of the channel bandwidth, then K =
1
4
lowpass filter would be designed with a cutoff frequency of fc =
B
2.
could be used, and the
This basic approach
was pioneered at Philips in the early 1980’s for use in constructing a fully integrated FM
broadcast receiver in which no bandpass filters were required [43].
The performance of the ultra-low IF receiver architecture rests on the degree to which
the image frequency is occupied. In turn, this issue depends heavily on system level design
considerations and on the ability of transmitters to avoid contaminating neighboring regions
of the frequency spectrum. One additional problem is the resulting demand placed on the
lowpass filters needed for final channel selection. A careful study of the imaging problem
shows that the number of poles required in the lowpass filter is more than twice that needed
in an equivalent traditional receiver using bandpass filters with the same final selectivity.
107
This feature results from the fact that signals in alternate channels are converted into the
region of baseband spectrum immediately above the bandwidth defined by the lowpass filter.
4.2.5
Designs with Image Rejection Mixers
Demands on filter requirements can also be reduced through the use of image rejection
mixers such as shown in Figure 4.11. In this approach, the use of quadrature mixing
attenuates the signal at the image frequency. With precise 90o phase shifts, a theoretically
infinite attenuation results, whereas in practice, the performance is limited to about 30 to
40 dB depending on whether or not trimming is used [48].
90 o
90 o
Figure 4.11: Image rejection mixer.
Nevertheless, 30 to 40 dB of attenuation can significantly decrease the number of poles
needed in bandpass filters when converting the signal to a subsequent IF stage. For example,
if an image rejection of 60 dB is required in the receiver, and 30 dB is provided by the mixer,
then the required number of poles in the filter is reduced by half.
Drawbacks of the image rejection approach include difficulties in achieving the necessary
phase shift accuracies, and the extra power consumption involved in the use of two mixers.
In addition, if an image reject mixer is considered for use at the receiver’s front-end, special
108
consideration must be given to the dynamic range issues discussed in Chapter 3. Use of
a reduced number of poles in the RF preselection filter will decrease the attenuation of
out-of-band signals at large frequency offsets, lowering the blocking and intermodulation
dynamic range performance relative to signals at these frequencies. However, the benchmark
receiver performance measurements shown in Table 3.3 suggest that this problem may not
be significant provided that one to two poles of preselection filtering is retained.
4.2.6
Designs with Selective Demodulators
A final technique that can be applied to minimize the need for on-chip bandpass filtering
in superheterodyne receivers is the implementation of selective demodulators. In theory, it
may be possible to reduce the requirements on channel select IF filters while obtaining good
receiver selectivity if the demodulator itself is designed to be sensitive to signals only within
the desired signal’s bandwidth. This concept was studied by Nauta [45], and later by van
der Plas [49] at Delft with mixed results. Their work addressed the subject of synchronous
detection in AM receivers using the demodulator shown in Figure 4.12.
In this design, a narrow bandwidth PLL recovers the carrier of the AM signal in the presence
of one or more interfering signals on adjacent frequencies. The recovered carrier is then
mixed with the IF signal to translate the AM spectrum to baseband. A lowpass filter
provides the necessary channel selectivity by eliminating adjacent channel interferers which
have been translated by the mixer to frequencies above the desired signal’s audio bandwidth.
Through careful design of PLL and AGC circuits, van der Plas achieved a 50 dB selectivity
using this approach. However, he cites difficulties including the need for precision receiver
tuning to prevent false PLL lockup, and a relatively large amount of circuitry for carrier
regeneration.
Similar techniques can be developed for other modulation types. For example, with digital
phase modulated systems, demodulation often involves carrier regeneration and translation
109
Phase Locked Loop
Lowpass
Filter
IF input
Lowpass
Filter
Audio
Output
Synchronous Detector
Figure 4.12: Selective AM demodulator.
directly to baseband. Lowpass filters can then be used to remove interferers as before.
However, as in van der Plas’ work, the implementation of carrier recovery circuits capable
of discriminating against large amplitude interfering signals presents significant challenges.
In the case of frequency modulated systems, the three demodulator types commonly used in
traditional receivers include frequency discriminators, one-shot multivibrators, and PLLs,
none of which is well suited to selective demodulation. A fourth type, however, shows
considerable promise for the special case of digital FM systems. The essential concept is
equivalent to the use of the direct conversion receiver architecture described in the next
section. The interested reader is referred to the references for additional details [52] [51]
[40].
4.3
Direct Conversion Receivers
Of all the alternative architectures developed to date, the direct conversion or “zero-IF”
design shown in Figure 4.13 offers the most promise for eliminating on-chip bandpass filters
110
while simultaneously retaining high performance operation.
Lowpass
Filter
Preselect
Filter
I
Baseband
Lowpass
Filter
Q
90 o
Figure 4.13: Direct conversion receiver architecture.
In a direct conversion architecture, the received signal is passed through a preselect filter
whose sole responsibility is to improve dynamic range performance by rejecting potentially
large out-of-band interferers. The portion of the spectrum passed by the preselect filter
is then amplified and mixed with a local oscillator equal to the frequency of the desired
signal. This converts the signal directly to baseband where final channel select filtering is
performed with integration-friendly, lowpass filter designs.
Since the center of the “IF” is at zero frequency, no image response is produced, and no image
filtering is required. Signals above and below the desired signal frequency are translated into
baseband frequencies above that of the desired signal, and are removed by lowpass filtering.
The concept of direct conversion thus shares many of the features of selective demodulators
dicussed in the preceding section. Indeed, the direct conversion receiver architecture can be
viewed as a selective demodulator operating directly at the RF frequency.
The integration-friendly features of the direct conversion design have led several researchers
to embrace this architecture as a foundation on which to build fully integrated receiver
designs [52] [40]. However, before these receivers can achieve widespread acceptance, several
111
important practical problems must be addressed.
First, reception of signals in which transmitted information is carried in both sidebands (or
equivalently in both dimensions of a sinusoidal basis vector) requires the implementation
of in-phase/quadrature (I/Q) channel conversion as shown. The problem of maintaining a
precision 90o phase shift over the range of LO frequencies to which the receiver is tuned is
complicated by the fact that the circuits must operate directly at the RF frequency.
Second, the transmitted information may be difficult to extract once it has been separated
into I and Q parts. This is especially true for analog FM, but may not be a serious problem
for newer digital modulation types.
Third, LO to RF port isolation of the mixers must be carefully considered to prevent
radiation and interference to nearby receivers operating at the same frequency. This problem
is reduced through the use of an LNA placed between the antenna and mixers. However,
operation at very high frequencies (900 MHz to 2 GHz) limits the isolation achievable in
practice, making this problem a potentially significant concern.
Fourth, DC offsets occur at the I/Q outputs due to LO to RF feedthrough in the mixers.
These effects must be removed with highpass filtering, making reception of low frequency
baseband information difficult.
Finally, translation of the entire service band to baseband places significant dynamic range
requirements on the mixers as well as on the low pass filters employed. This problem
is fundamentally the same as that in traditional superheterodyne receivers. Hence, all the
tradeoffs between noise figure, dynamic range, and power consumption discussed in Chapter
3 carry over to the direct conversion architecture.
The advantages and disadvantages of direct conversion receiver designs relative to those of
receivers based on the traditional superheterodyne architecture must therefore be decided
based on practical rather than theoretical issues. To date, the radio hardware industry
112
has favored the classic superheterodyne as more robust and lower risk in the design of
board-level receiver solutions. Whether or not this decision changes within the domain of
integrated receiver design will depend on risk and performance tradeoffs associated with
solving the problems listed above, versus solving the problems associated with on-chip
bandpass filtering addressed in this dissertation.
4.4
Digital Receivers
Periodically, the concept of an “all digital” receiver design is proposed as a solution to
analog signal processing limitations and apparent complexity. Thus, it is prudent to consider
whether such an approach could minimize or eliminate the requirement for on-chip filtering.
Two simplified block diagrams of digital receivers are shown in Figures 4.14 and 4.15. The
receiver in Figure 4.14 is a superheterodyne-like design in which the sampling frequency is
above the desired RF frequency and a “digital down-converter” is used to mix the signal
to a lower IF, or directly to baseband. The other design shown in Figure 4.15 is a directconversion architecture in which the RF is sampled at the frequency of the desired signal
using a quadrature approach.
LPF
Sampler
f clock
Decimation
Filters
A/D
D i gi t a l
LO
Lowpass
Channel
Select
Filters
I
Q
Digital Circuits
Figure 4.14: Simplified digital receiver architecture.
On the surface, these designs have significant appeal. Assuming for the moment that the RF
113
Sampler
A/D
LPF
Sampler
Decimation
Filters
A/D
Lowpass
Channel
Select
Filters
I
Q
Digital LO and
Phase Shift
Digital Circuits
Figure 4.15: Simplified direct-conversion digital receiver architecture.
can be directly sampled, the digital local oscillator can produce high precision quadrature
waveforms and the benefits of digital signal processing can be applied to realize channel
selection filters with very high selectivity. In addition, adaptive algorithms can be used to
combat problems such as multipath distortions in the received signal.
Unfortunately, the assumption that the receiver can sample directly at the RF frequency is
fundamentally flawed. In modern cellular and PCS systems, sampling directly at RF entails
sampling rates of 900 MHz and above. Further, since the RF front-end is open to the full
spectrum environment (no narrowband preselect filtering has yet been assumed), very high
dynamic range would be required, implying an analog-to-digital (A/D) converter with 10
bits or more. Attempting to sample to this precision at GHz frequencies is a very difficult
task, implying high cost and high power consumptions.2 Moreover, even if the problems of
GHz rate sampling and A/D conversion are put aside, the data rate and word widths alone
will be too large to allow processing within acceptable power budgets for portable operation
(see Chapter 2 and reference [182]).
In an attempt to circumvent this obvious flaw, most designers have adopted either a conventional receiver front-end, or the use of “sub-sampling” techniques. The former requires all
2
Even with 10 bits, some form of AGC will generally be required and the spurious-free dynamic range is
unlikely to be sufficiently high to handle all the interfering signals present in the environment.
114
the analog filtering implied by analog receiver designs considered in previous sections, while
the latter introduces aliasing which must be eliminated by very narrow preselect filtering
prior to sampling.
For example, assume that a sub-sampled, direct-conversion digital receiver architecture is
being considered in which the RF frequency is 2 GHz and the sampling frequency is 20
MHz. Such a design will alias any signals spaced by integer multiples of 20 MHz from
the desired signal into the baseband processing circuits where they will interfere with, or
prevent reception. To solve this problem, a preselect filter with a bandwidth B 20 MHz
and an attenuation on the order of 60 dB or more would need to be added. To achieve
this level of attenuation with a manageable 3-pole bandpass filter design would require a
selectivity Q of 500. Thus, rather than simplifying the design of integrated receivers, this
approach actually increases the demands on bandpass filter designs.
To reduce the demands on preselection filtering, one might consider increasing the sampling
rate. However, at higher rates, the power consumption required in the digital lowpass filters
which follow would exceed allowable limits as discussed in Chapter 2. Thus, the use of an
analog front-end offers the only viable alternative for low-power, high performance design,
and the “digital receiver” concept is more properly seen as a digital demodulator operating
at a final, low IF frequency.
4.5
Ideal Low-Power Receivers
Having surveyed existing receiver architectures and assessed their practical problems, one
may wish to consider what shape a receiver might assume if only physical laws, rather
than technology limitations determined its ultimate performance. In this final section, the
design of an ideal receiver targeted at future low-power, portable, wireless applications is
considered. Such a design must provide all the basic receiver functions identified in Chapter
115
3, including:
• Selecting one signal from a dense spectral environment;
• Amplifying this signal to a level suitable for demodulation; and
• Recovering the transmitted information.
In addition, it should be able to receive signals of any power level (provided they are above
the noise floor), and should be immune to the problems of spurious responses created by
arbitrarily large interferers.
To avoid violating physical laws in attempting to design such a device, the following assumptions are made:
• The noise figure of passive circuits is equal to their insertion loss.
• The noise figure of active circuits used is greater than or equal to one (0 dB).
• The efficiency of active circuits is less than or equal to one.
As additional constraints, it is assumed that the demodulator used is not frequency selective, and that it requires some amplification of the signal at the antenna. Otherwise the
demodulator and the receiver would be the same and the “solution” would be contained in
itself.
Based on these assumptions, a little thought shows that the ideal receiver must assume
a form similar to that shown in Figure 4.16 in which all selectivity is moved to the RF
input. If this were not the case, then large amplitude interferers would reach downstream
circuits, causing saturation and intermodulation distortion products resulting in blocking
and spurious responses. The only alternative would be to increase the power consumption
116
of the amplifiers or of the mixer drive circuits to a level at least as high as that of the
interferer, which would violate the assumption of low power operation.
Optional Down Conversion
Optional
Variable
Attenuator
Preselect
Filter
Image
Filter
Demod
Filter Tuning
Figure 4.16: Ideal receiver architecture.
With the solution shown, several requirements on the preselect filter must be met:
• It must be high order to attenuate undesired signals.
• It must be tunable so that more than a single station in the service band can be
received.
• It must be linear to avoid blocking and spurious responses at high frequencies.
• It must have zero (or negative) insertion loss to avoid degrading the receiver’s noise
figure performance.
Assuming these requirements are satisfied, downstream circuits can then be implemented
with relatively low dynamic range circuits, and very low power. With interfering signals
removed by the front-end filter, the dynamic range needed is reduced to that required to
establish a specified C/N ratio at the demodulator input. For many modulation types, this
can be as low as 15 to 20 dB.
For satellite receivers, where the strength of the desired signal does not vary significantly,
the overall receiver dynamic range can be as low as 20 dB, as discussed in Chapter 3. For
117
terrestrial systems, signal levels at the antenna may vary by as much as 100 dB depending
on the distance to the transmitter. However, this range is easily compressed to that of
the required C/N for demodulation using suitable attenuators, AGC amplifiers, or limiting
mechanisms (for FM modulations). Hence, an ideal receiver designed for these situations
could also be implemented with very low dynamic range circuits.
The practicality of such an ideal, low-power receiver is of course open to serious question.3
Although it is possible to create high quality, tunable, passive filters with narrow bandwidths
using technologies such as YIG resonators, such filters are currently too large and too
expensive for consumer products. The real value of examining this architecture lies instead,
in contrasting it with existing receiver design to illustrate their basic limitations.
In the past, receiver power consumption was a less serious issue than it is today, and traditional superheterodyne receivers served well. With current increases in spectral occupancy
and increasing emphasis on reducing power consumption in portable wireless devices, however, low power circuits with good dynamic range have become a necessity. These two
conflicting needs can only be addressed through designs which move selectivity closer to
the antenna. Hence, while the direct conversion architecture discussed in the previous section shows potential for solving some of the on-chip bandpass filtering problem, a need for
bandpass filtering at the front end remains. This conclusion, coupled with the many practical problems associated with realizing the direct conversion receiver architecture, is largely
responsible for the emphasis on integrated bandpass filters in the following chapters.
3
Perhaps the closest consumer market product approximating the ideal receiver design of Figure 4.16 is
a fixed-tuned paging receiver produced by Motorola [154] in which a 4-pole crystal filter was used at the RF
frequency and the signal was converted directly to a 35 KHz IF for amplification, limiting, and demodulation.
118
Chapter 5
Integrated Bandpass Filter Design Options
The subject of filter design is a major field of study within the discipline of Electrical
Engineering. An exhaustive treatment of bandpass filter design options is therefore well beyond the scope of this dissertation. Fortunately, however, technological constraints imposed
by silicon IC processes and by the specialization to low power, high frequency, integrated
implementations permits the range of study to be narrowed significantly.
As shown in Chapter 2, digital filtering is not a viable option at center frequencies above
about 1 MHz, due to problems of excessive power consumption. This power consumption
is determined by device parasitic capacitance and cannot be reduced by alternative filter
structures or computational algorithms. Although the development of submicron processes
will help in the coming years, the upper limit is unlikely to exceed a few MHz in the
immediate future. Thus, the following discussion is restricted to analog methods. This
decision narrows implementation alternatives to the general class of linear time invarient
(LTI) filters, within which filter implementation options may be grouped, as shown in Table
5.1.
Additional observations made in Chapter 2 rule out the use of switched-capacitor and delay
line filters for all but extremely low IF frequencies (e.g. < 1 MHz) when implemented in
silicon processes at low power. In this case the main problems are the clock rates and
circuit time constants needed to achieve sufficient settling between samples, and the clock
noise found in practical realizations. Hence, implementation alternatives for high frequency
119
Table 5.1: Filter implementation alternatives.
Recursive
Transversal
Continuous-Time
Active RC, Gm-C, etc.
Bulk Acoustic Wave
Surface Acoustic Wave
Surface Acoustic Wave
Discrete-Time
Switched-capacitor
CCD delay line
designs are essentially constrained to the continuous-time filter options listed in the first
column.
Within the domain of continuous-time filters, designs may be divided into recursive structures and transversal structures. Continuous-time recursive filters have been widely studied
and are easily constructed in integrated circuits using active filter design methods. Traditionally, these methods have involved only transistors (or opamps), resistors, and capacitors.
Two additional methods are less well known, but also viable options. The first, Q-enhanced
LC filters, adds inductors to the list of on-chip elements, extending the capabilities to VHF
frequencies and above. The second, electro-acoustic filters, adds piezo-electric components
to the list of on-chip elements, raising the possibility of implementing on-chip, bulk and
surface acoustic wave devices.
This chapter will concentrate on continuous-time active filter design techniques and recursive
filter structures. The subject of electro-acoustic filters is briefly discussed in Section 5.2,
but is left mainly to the references for the reasons cited in Chapter 2.
5.1
Continuous-Time Active Filters
In the most general case, continuous-time active filters may be defined through a system of
first-order differential equations:
120
d
X(t) = AX(t) + Bu(t)
dt
y(t) = CX(t) + du(t)
(5.1)
where
X(t)
is an n x 1 state vector
A
is an n x n coefficient matrix
B
is an n x 1 coefficient vector
u(t)
is the system input
y(t)
is the system output,
C
is a 1 x n coefficient vector, and
d
is a scalar.
Taking Laplace transforms and solving for the frequency domain transfer function H(s),
yields the familiar ratio of polynomials in s:
H(s) ≡
y(s)
an sn + an−1 sn−1 + ... + ao
=
u(s)
bn sn + bn−1 sn−1 + ... + bo
(5.2)
Through suitable choices of coefficients ai and bi, desired filter responses can be synthesized.
For example, the class of all-pole lowpass responses can be obtained by setting ao = 1 and
ai = 0 for i > 1, yielding:
HLP (s) =
ao
bn sn + bn−1 sn−1 + ... + bo
(5.3)
The familiar filter forms such as Butterworth, Chebyshev, and Bessel can then be obtained
by proper selection of bi [187].
121
To obtain a bandpass filter design, a common approach is to begin with Equation (5.3) and
then apply the transformation:
s→
s2 + ωo2
Bs
(5.4)
where ωo is the desired center frequency and B is the desired filter bandwidth. This transformation yields a transfer function of the form:
HBP (s) =
a0n sn
b02ns2n + b02n−1 s2n−1 + ... + b0o
(5.5)
defining an all-pole bandpass design with filter response characteristics essentially identical
to the lowpass prototype, but shifted in frequency and scaled in bandwidth as illustrated
in Figure 5.1.
|H| dB
B
0
-20 N log (ω)
− ωo
-20 N log ω B/2
|
ωο
1.0
|
log(ω)
Figure 5.1: Lowpass to bandpass filter transformation.
5.1.1
Bandpass Filter Structures
A practical realization of a filter requires a filter structure. This structure can be defined
graphically in the form of a signal flow diagram. For example, the structure shown in Figure
5.2 implements the familiar second-order bandpass transfer function:
122
HBP (s) =
sωo
s2 + s ωQo + ωo2
(5.6)
with center frequency of ωo , and quality factor of Q.
1
ωο
ωο
1
y(s)
s
s
u(s)
-1
Q
-1
Figure 5.2: Flow diagram representation of filter structure.
A state-space description in the form of Equation (5.1) implies a particular filter structure.
However, a desired transfer function expressed in the form of Equation (5.2) has many
possible state space representations, and hence, can be realized by more than one structure.
The selection of a particular structure must therefore be based on lower-level implementation
issues. In active filter design, these issues include sensitivity to component variations,
availability of component values, component matching considerations, and filter dynamic
range.
Of all possible structures, those that have been found to have the most desirable properties
include:
• State Variable
• Cascaded Biquad
• Follow-the-Leader, and
123
• Multiple Feedback
For the special case of high-Q bandpass active filter design, structures built around secondorder resonators of the form expressed in Equation (5.6) have received the most attention
[64] [68] [69] [75] [83] [90] [99]. Of these choices, the cascaded biquad structure has been
implemented in some realizations [75] [83], and at least one reseacher has considered the
follow-the-leader structure [68]. However, the simple coupled-resonator structure shown in
Figure 5.3 has generally been preferred because of the following desirable features:
• It can be derived from low sensitivity ladder structures [64].
• It provides good equalization of gains across all nodes at all frequencies within the
filter passband, and hence, has good dynamic range performance [68].
• Higher-order filters can be implemented using a set of identical second-order resonator
sections [69].
k
s
k
u(s)
1
H(s)
H(s)
s
k
s
H(s)
y(s)
k
s
Figure 5.3: Coupled resonator bandpass filter structure.
Of these features, the ability to employ identical resonators represents perhaps the most
significant advantage over competing approaches. This property allows a single second-order
filter section to be designed and then replicated as needed to create higher-order responses,
taking optimum advantage of the matching and temperature tracking features of monolithic
IC fabrication processes.
124
Another important advantage of the coupled-resonator structure is its similarity to existing
discrete RF and IF bandpass filter designs. Discrete filters ranging from monolithic crystal
filters, to crystal ladders, tuned transformers, helical, dielectric resonator, and interdigital
stripline designs are all based on the coupled resonator concept. Hence, they share similar
gain and phase response characteristics, allowing on-chip filters to achieve comparable inband responses simply by providing an equivalent number of resonator sections (equivalent
number of poles).
5.1.2
Gm-C Filter Design
Having selected a filter structure, the next step in the design process is to realize a circuitlevel implementation. In classic RC active filter design, circuit elements used consist of resistors, capacitors, and operational amplifiers. However, as noted in Chapter 2, the frequency
of operation of these designs, as well as that of their on-chip “MOSFET-C” counterparts,
is limited by the operational amplifiers employed.
To solve this problem, designers have moved to circuit implementations based on simple
transconductance amplifiers such as that shown in Figure 5.4. These amplifiers can be used
to perform multiplication by constants through appropriate choices of transistor geometries
(W/L ratios), and can be used to implement integrations by loading the outputs with capacitors. Thus, filters based on these techniques are referred to as transconductor amplifier
- capacitor (TAC), operational transconductance amplifier - capacitor (OTA-C), or simply
transconductor - capacitor (Gm-C) designs.
In practice, transconductors are typically implemented in differential form using some varient of the configuration shown in Figure 5.5. Such differential transconductors have several
attractive features including [82] [96]:
• simplification of bias circuit design;
125
io
io
vi
vi
g mvi
io
vi
r ds
g mvi
Figure 5.4: Simplified transconductance amplifier.
io
io
vi
I bias
Figure 5.5: Practical differential transconductance amplifier.
• rejection of power supply noise and crosstalk from other circuits present on the same
die;
• ability to create both inverted and non-inverted output signals; and
• ability to process signal voltages twice that possible in single-ended circuits with the
same fidelity.
Although these features are interesting and important design considerations, they are not
essential to the concepts covered in this chapter. Thus, the reader is directed to the cited
references for additional low-level implementation details. In the discussions which follow,
the simplified, single-ended form of Figure 5.4 will be assumed, with the understanding that
similar results can be obtained with fully differential designs.
Using the Gm-C design approach, bandpass resonators can be constructed as shown in
Figure 5.6. In this representation, symbols used in [75] have been adopted for the transcon126
ductors, and the ground reference for the transconductors is not explicitly shown.1 Taking
as states the voltages VC1 and VC2 on capacitors C1 and C2, this circuit can be redrawn in
flow diagram form as shown in Figure 5.7, which can be shown to be structurally equivalent
to that of Figure 5.2. Thus, if the output is taken to be the voltage VC1 , the transfer function is that of a second-order bandpass filter, and the center frequency, Q, and midband
gain can be shown to be:
r
gm1 gm2
C1 C2
s
C1 gm1 gm2
Q =
2
C2 gmQ
ωo =
VC1 V i
=
ωo
g mi
(5.8)
gmi
gmQ
(5.9)
VC 2
VC 1
vi
(5.7)
C
gm2
1
C
vi
gmvi
vi
gmvi
2
gmQ
gm1
Figure 5.6: Gm-C based bandpass filter.
The degrees of freedom in selecting component values can be reduced by setting gm1 =
gm2 = gm and C1 = C2 = C, yielding:
ωo =
1
gm
C
(5.10)
The transconductor gmQ , which appears to be short circuited, actually draws current from the node
common to its input and output, simulating a grounded resistor.
127
vi
g mi
1
g m2
sC1
sC2
I C1
VC1
-gmQ
VC2
-g m1
Figure 5.7: Flowgraph of Gm-C based bandpass filter.
Q =
VC1 V i
=
ωo
gm
gmQ
gmi
gmQ
(5.11)
(5.12)
In addition to giving simpler formulas, this choice yields a circuit in which the gain from the
input to each of the capacitor node voltages is equal — a necessary condition for maximizing
the dynamic range [115].
An examination of Equations (5.10) - (5.12) produces two important observations related
to filter tuning [75]. First, the resonator’s center frequency is controlled by a ratio of
transconductances and capacitances, which vary independently in IC fabrication processes.
Thus, accurate frequencies cannot be realized, and some form of frequency tuning becomes
a necessary part of the filter design process. Second, the filter’s Q and midband gain
are controlled by ratios of transconductances, which can be accurately achieved, making
tuning of these parameters theoretically unnecessary. In practice, however, Q tuning may
be required in designs operating at very high frequencies, depending on the circuit Q.
The need for Q tuning arises from practical limitations of transconductors such as drainsource conductances and drain/gate capacitances. These problems will be examined in
Chapter 7 where tuning control systems designed to correct both frequency and Q errors
will be described. A more fundamental problem, however, is that of limited dynamic range.
128
Qualitatively, dynamic range limitations in active filters arise from the need to recharge the
circuit capacitances on each new cycle of the signal being processed. As will be shown in
Chapter 6, the circuits used to perform this task consume significant power, and generate
substantial noise in the process. These problems can be minimized if energy storage elements
can be incorporated in the filter design, a feature offered by the approaches described in
the following sections.
RLC Circuit Equivalence
The circuit of Figure 5.5 is equivalent to, and can be replaced by a classic, parallel RLC
resonator excited by a current Ii = gmi Vi . To illustrate this fact, the Gm-C circuit is redrawn
in Figure 5.8 with the R, L, and C component equivalences indicated. The corresponding
RLC circuit is shown in Figure 5.9.
gm1
VC 1
vi
g mi
VC 2
C
1
gm2
R
C
C
gmQ
2
L
Figure 5.8: RLC equivalents in Gm-C resonator design.
VC 1
vi
g mi
C
R
L
Figure 5.9: RLC resonator equivalent.
The equivalence of R to the gmQ transconductance connected in negative feedback can be
129
seen by examining the current-voltage relationship of each. A voltage V applied to the
transconductor results in a current I = gmQ V , yielding an effective resistance of
R=
1
(5.13)
gmQ
With a little additional work, it can also be shown that the inductance L is simulated
through the capacitively-loaded gyrator consisting of C2 , gm1 , and gm2 , yielding:
L=
C2
gm1 gm2
(5.14)
Alternatively, this equivalence can be seen through a flow graph analysis by redrawing
Figure 5.7 as shown in Figure 5.10. In this modified representation, capacitor currents have
been shown explicitly, and the right-most node has been labeled IL to indicate the simulated
inductor current. For comparison, the flow graph of the RLC resonator circuit of Figure
5.9 is shown in Figure 5.11. An examination of Figures 5.10 and 5.11 then leads directly to
(5.13) and (5.14).
vi
1
g mi
I C1
-gmQ
1
g m2
sC1
g m1
sC2
VC2
I C2
VC1
L
-1
Figure 5.10: Gm-C filter flowgraph with currents and voltages labeled.
130
IL
vi
g mi
I C1
1
1
sC
sL
IL
VC
-1
R
-1
Figure 5.11: Equivalent RLC filter flowgraph.
5.1.3
Q-enhanced LC Filter Design
At frequencies above about 100 MHz, high-Q, Gm-C filters become difficult to implement
due to parasitic capacitances in circuit elements and interconnects. In RLC designs, however, these parasitics can be absorbed into the tuned circuits, permitting operation at frequencies approaching the limits of the transistors employed. In addition, RLC designs can
theoretically implement bandpass functions without the need for active circuits. However,
as noted in Chapter 2, current IC technology limits achievable Q values of on-chip inductors
to 10 and below, making integration of purely passive high-Q designs impossible.
Nevertheless, integrated RLC filters with Q > 10 can be realized if active circuits are
admitted into the design process. These active circuits can be used to cancel losses in
on-chip inductors, thereby enhancing the effective inductor Q.
To illustrate the fundamentals of the Q-enhanced filter design approach, consider the modified RLC resonator circuit shown in Figure 5.12, and its flowgraph equivalent in Figure
5.13.
Here, a lossy inductor is modeled near the circuit’s resonant frequency as an ideal inductor
in parallel with a loss resistance Rp . Connected in parallel with Rp is an active negative
resistance implemented with a transconductor gmn arranged in a positive feedback config-
131
VC
vi
g mi
C
Rp
gmn
L
Figure 5.12: Q-enhanced RLC resonator circuit design.
g mn
vi
g mi
IC
1
1
sC
sL
VC
-1
Rp
IL
-1
Figure 5.13: Q-enhanced RLC resonator flow diagram.
132
uration. The parallel combination of Rp and gmn yields an effective resistance value Ref f
given by:
Ref f = Rp|| −
1
gmn
=
Rp
1 − gmn Rp
(5.15)
Based on the well-known relationships between inductor quality factor Qo and the value of
Rp, and between the parallel resistance, reactance, and Q of a resonant circuit, the effective
Q of this resonator can be found from:
Qef f =
1
Qo
1 − gmn Rp
(5.16)
and can be made arbitrarily large by a suitable choice of gmn .
With the introduction of the transconductance gmn , this circuit becomes an active filter,
and like the Gm-C design, the transconductor will consume power and add noise to the
circuit. However, both the power consumption and the added noise will be significantly
less than in the Gm-C case due to the energy storage and exchange that takes place in and
between the (lossy) LC resonant circuit elements. These considerations will be addressed
in detail in Chapter 6, where quantitative expressions for the DR of both types of filters
will be derived and compared.
An additional problem with the Q-enhancement approach is the difficulty of achieving
accurate and stable results at large effective Q values. In general, circuit parameters which
control Qo (metal sheet resistance and substrate losses) will be uncorrelated with those
determining gmn , and some form of Q tuning will be necessary. In addition, frequency
tuning will typically be required in all but the lowest Q designs due to variations in oxide
thicknesses between different fabrication runs. Both of these issues can be addressed through
the automatic tuning techniques discussed in Chapter 7, and the detailed circuit design
133
techniques discussed in Chapters 8 and 9.
5.2
Electro-Acoustic Filters
Before proceeding to a detailed discussion of dynamic range in the following chapter, it
should be emphasized that the advantages offered by energy storage and exchange are not
unique to LC resonators. These advantages can conceivably be realized by any technology
in which these mechanisms exist.
One such technology which has been shown to be compatible with IC design, is the field of
electro-acoustics [153] [167] [165]. Through the use of piezo-electric materials added to the
fabrication process, devices can be constructed in which an electric field impressed across two
capacitive electrodes develops localized stress and strain fields within the material. These
fields then exchange energy in the same manner as electric and magnetic fields, leading to
propagation of energy away from the source. The source electrodes, or electrodes placed at
some other location within the material can then be used to couple acoustic energy back
out of the acoustic fields and into the electric circuit. The velocity of propagation through
the material is on the order of 3000 m/s, approximately 5 orders of magnitude lower than
that of electro-magnetic energy in free space. Thus, filters operating at frequencies of 100
MHz or above can be constructed in areas of a few mm2 or less, and become compatible
with dimensions found in low-cost integrated circuits.
As discussed in Chapter 2, on-chip electro-acoustic filters in the form of both bulk acoustic wave (BAW) and surface acoustic wave (SAW) designs have been studied by various
researchers over the past several years, and show considerable promise for use at VHF frequencies and above. Being passive, the resulting designs have significant advantages over
both Gm-C and Q-enhanced LC filters in terms of both power consumption and potential
dynamic range. To date however, these filters have not reached a suitable level of maturity
134
for commercialization, and several significant problems remain. Since experimental work
in this area requires special fabrication technology not available at the author’s institution,
these filters will not be addressed further in this dissertation. Developments in the coming
years will determine what role, if any, electro-acoustic filtering will play in the design and
production of on-chip filters for low-cost commercial wireless products.
135
Chapter 6
Active Filter Dynamic Range
The dynamic range of an active filter may be defined as in Chapters 2 and 3 as:
DR =
Pmax
Pnoise
(6.1)
where Pmax is the maximum allowable signal level and Pnoise is the noise floor measured in
some defined bandwidth. Since Pmax and Pnoise must be taken at the same circuit node,
an equivalent definition of dynamic range may be written in terms of voltages as:
DR =
2
Vmax
2
Vnoise
(6.2)
where Vmax is the maximum allowable signal voltage and Vnoise is the total RMS noise voltage within the specified bandwidth. For active circuits, both Pmax and Vmax are functions
of circuit biasing and power consumption and are often known constraints in the design
process. Under this condition, DR is determined by the noise introducted by the circuit.
In order to compute either Pnoise or Vnoise , noise models for the circuit elements must be
available. Models for capacitors, inductors, resistors, and transconductors which will be
used in the discussions to follow are shown in Figure 6.1.
Ideal capacitors and inductors do not dissipate energy, and therefore generate no noise.
Thus, the noise models for ideal capacitors and inductors are simply the elements themselves
136
io
C
L
vi
in
g mvi
in
Figure 6.1: Noise models for circuit components.
with no noise sources present. Lossy capacitors and inductors generate noise equal to that
of a resistor which gives equivalent energy losses. Thus, if an inductor is modeled as in
Figure 5.12 with a parallel loss resistance Rp , the noise of the inductor is simply that of
the resistor used to model the losses. For a resistance of value R, the one-sided RMS noise
current density is [171] [177]:
r
in (f ) =
1
R
4kT
√
A/ Hz
(6.3)
where k is Boltzmann’s constant, and T is temperature in degrees Kelvin.1
Finally, for a transconductor, the RMS noise current density may be approximated as [73]:
in (f ) =
p
4kT F gm
(6.4)
where gm is the transconductance value and F is a noise figure which typically ranges
between 1 and 2, and accounts for variations in the actual transconductor implementation.
This model is approximately valid for transconductors created with FET or BJT devices,
as well as those using source or emitter degeneration [65].
1
Since the noise generated by the resistor is uncorrelated with other elements in the circuit, the noise
current source is shown without a polarity indication.
137
Given these models, the noise level at any node or branch in a circuit may be found by
computing the noise due to each source independently, and taking the RMS value of the individual contributions. This process will be used in the following sections to find the dynamic
range of the Gm-C and Q-enhanced LC resonators discussed in the previous chapter.
6.1
Dynamic Range of Gm-C Filters
The dynamic range of a Gm-C filter may be found by augmenting the flowgraph of Figure
5.10 to show the noise currents injected into the circuit by each transconductor used. The
resulting flowgraph is shown in Figure 6.2.
Ing
g mi
vi
I n gm
mi
1
1
I C1
1
Ing
-gmQ
1
gm
sC1
I n gm
1
I C2
VC1
gm
sC2
VC2
IL
mQ
-1
Figure 6.2: Gm-C flowgraph with noise sources included.
To simplify the following discussion, and to maximize the dynamic range [115], both the
capacitors C1 and C2 and the transconductors gm1 and gm2 are assumed to be of equal
value so that the resonant frequency ωo , Q, and midband gain are given by Equations
(5.10), (5.11), and (5.12), respectively.
The dynamic range of this circuit will be evaluated using Equation (6.2) by examining the
signal and noise voltages at the output node VC1 . Assuming Vmax on C1 is known, the
dynamic range then depends on Vnoise at this node alone.
138
To find Vnoise , assume that the resonant frequency, Q, and capacitance are specified. Equations (5.10) and (5.11) can then be solved to find gm and gmQ . Assume also that gmi is
set equal to gmQ to give a midband gain of unity. The actual choice of gmi does not significantly affect the circuit’s dynamic range provided gmi gm . Under this restriction, the
noise produced by transconductor gmi will be significantly smaller than that produced by
transconductor gm feeding the same node, and may be neglected. Similarly, if Q is large,
Equation (5.11) implies that gmQ gm , so that the noise contribution of gmQ may be
neglected as well. The noise produced at the output node VC1 may then be found based
solely on the noise contributions of the transconductors gm feeding the capacitors.
An analysis of the closed-loop system shown in Figure 6.2 shows that the gain from both
noise currents Ingm to the current IC1 entering the integrating capacitor is:
IC1
s2
= 2 ωo
Ingm
s + Q s + ωo2
(6.5)
which has magnitudes of 0, Q, and 1 at frequencies of 0, ωo , and ∞, respectively. Thus,
at resonance, both noise currents are amplified by the effective quality factor of the circuit,
and when added on a power basis, yield a mean squared noise density at node IC1 of:
In2 (fo ) = 2(4kT F gm )Q2
(6.6)
To translate this noise current to the noise voltage at the output node VC1 , Equation (6.6)
is multiplied by the square magnitude of the capacitor’s impedance, yielding:
Vn2(fo ) = 2(4kT F gm)Q2
1
ωo C
2
(6.7)
Equation (6.6) gives the noise voltage power spectral density (PSD) at the center frequency
of the filter only. The total mean squared noise voltage at node VC1 is found by integrating
139
the PSD across all frequencies. Since the noise spectrum is shaped by the filter transfer
function, the total mean square noise voltage may be found from:
Z
2
Vnoise
=
0
∞
Vn2 (fo )|H(f )|2df
(6.8)
where H(f ) is a second-order resonance curve with center frequency fo , quality factor Q,
and midband gain of 1. Since Vn2 (fo ) is a constant with respect to the integration, it
can be removed from the integrand. The remaining integral is then the definition of the
noise bandwidth Bn of the filter, which for a high-Q second-order resonator is given by the
well-known result:
Bn =
π fo
2Q
(6.9)
Combining equations (6.7) – (6.9) with the expression for ωo given in equation (5.10), then
yields:
2
Vnoise
= 2kT F
Q
C
(6.10)
2
, which is typical of equations found in the literature, shows a
This expression for Vnoise
scaling of noise by the quality factor of the filter if capacitance is held constant. To complete
the evaluation of dynamic range, Equation (6.10) can be combined with Equation (6.2) to
give
DR =
2
Vmax
C
2kT F Q
suggesting that dynamic range scales inversely as the filter Q.
140
(6.11)
6.1.1
Optimum Dynamic Range
Equation (6.11) gives the dynamic range for the particular Gm-C second-order resonator of
Figures 5.5. However, it is not clear whether other possible filter structures could achieve
a better result. In addition, the dynamic range of higher-order filters has not yet been
addressed. These issues were investigated by Groenewold in his Ph.D work at Delft University in 1992 and formed the basis of his disseration [116]. By returning to the general
state space description of a linear filter in (5.1), Groenewold showed that the best possible
dynamic range for a high-Q second-order active filter biquad is given by (6.11).2 Thus, we
can conclude that the filter in Figure 5.6 is in fact an optimal design.
In addition to this result for a high-Q resonator, Groenewold derived the following bound
for the dynamic range of higher-order bandpass filters [115]:
DR ≤
2
Vmax
C
2πkT F Q
(6.12)
This result, which differs from that in (6.11) only by a small constant, indicates that the
optimum dynamic range for a high-order filter is essentially limited to that of a simple
resonator design. To understand how such a dynamic range might be achieved in practice,
consider the coupled resonator structure of Figure 6.3.
As shown, coupling can be achieved either with capacitors, or with small transconductors
whose values are selected to give a voltage gain of approximately 1 from one resonator to
the next. In either case, the coupling circuits introduce negligible additional noise into
the circuit compared with that of the gyrator transconductors used in the resonator core.
Moreover, since the interstage gain is 1, the noise injected into any given resonator from
neighboring resonators will be at most equal in magnitude to the noise generated within
2
Groenwold’s result differs from the result here by a factor of 2. This discrepancy arises from differing
definitions used for C, which in [115] is defined as the total capacitance of the two integrators combined.
141
g
vi
vo
g mi
vo
g mi
Resonator
m1
vi
Resonator
gm2
Resonator
Capacitive Coupling
Resonator
Active Coupling
Figure 6.3: Gm-C based coupled resonator filters.
the resonator itself. Thus, the dynamic range of the overall high-order filter is seen to be
close to that of the core resonator on which it is built.3
6.1.2
Dynamic Range Versus Power Consumption and Bandwidth
Dynamic range expressions such as (6.11) are common in the literature, and have led to the
widespread notion that the dynamic range of active bandpass filters is inversely proportional
to their quality factor. This conclusion, while correct if C is taken as a constant, is at best
misleading. As shown below, dynamic range actually varies inversely as the square of the
filter Q under conditions of constant power consumption and constant bandwidth.
Fundamentally, all active circuits have a limited dynamic range. Their maximum signal
handling capability is limited by their power consumption, while their minimum signal
handling capability is limited by thermal and shot noise, and by the bandwidth in which
the noise is measured. Yet in (6.11), neither the power consumption nor the filter bandwidth appear explicitly, preventing a direct comparison of filter and system dynamic range
performance.
For example, in an amplifier or active mixer circuit used within a radio receiver, the maximum output signal is limited to some fraction η of the DC power PDC consumed, while the
3
Similar reasoning can also be applied to the cascaded biquad filter structure with similar results.
142
noise at the output is proportional to the thermal noise kT B, the noise figure F , and the
circuit gain G. Thus, the dynamic range may be written in the form:
DR =
ηPDC
kT BF G
(6.13)
In Chapter 3, these concepts were used to illustrate that radio receiver blocking and spurious
free dynamic range are fundamentally limited by power consumption and gain of active
circuits that precede the IF channel select filters. If the dynamic range of an integrated
filter is to be compared with these dynamic range performance measures, Equation (6.11)
must be recast in a form similar to that of (6.13).
The effect of power consumption on filter dynamic range was briefly addressed in Groenewold’s dissertation, resulting in an expression showing that dynamic range is inversely
proportional to the product of center frequency and Q [116].4
DR ∼
PDC
fo Q
(6.14)
However, the important issue of bandwidth was ignored. This oversight is easily corrected
by replacing the frequency of operation with the product of bandwidth and Q:
fo = BQ
(6.15)
yielding a result that is inversely proportional to the square of the filter Q.
DR ∼
PDC
BQ2
(6.16)
4
Similar conclusions have been reached by others (c.f. [110] [94]), all leading to the conclusion that DR
is inversely proportional to Q.
143
A full expression for filter dynamic range showing this dependence can be derived using
(6.11) together with basic circuit analysis concepts. For the filter of Figure 5.5, the maximum signal current imax that can be delivered to the integrating capacitors is limited by
the transconductor’s bias current Ibias .5
imax ≤ Ibias
(6.17)
Similarly, the maximum signal voltage on the capacitors vmax must be less than the supply
voltage VDC .
vmax ≤ VDC
(6.18)
Thus, the product of vmax and imax is less than the DC power consumed, as expected:
vmax imax ≤ PDC
(6.19)
Equation (6.19) can be rewritten as an equality if a suitable efficiency factor η is defined to
account for the ratio of vmax to VDC , the ratio of imax to Ibias , and the ratio of Ibias to the
total filter current consumption IDC .
PDC
=
η ≡
1
vmax imax
η
vmax imax Ibias
VDC Ibias IDC
(6.20)
(6.21)
5
This assumes class A operation which is typically used in small signal circuit design. However, other
modes of operation can be taken into account with a suitable definition for efficiency η.
144
Finally, for a filter with a given capacitance C, vmax and imax are related by the capacitive
susceptance ωo C, giving:
imax = vmaxωo C
(6.22)
Thus, combining Equations (6.15), (6.20), and (6.22) with Equation (6.11) yields:
DR =
ηPDC
4πkT F BQ2
(6.23)
which can now be compared directly with the performance of the amplifier circuit expressed
in (6.13).
The significance of the dynamic range degradation with Q is perhaps best illustrated by a
numerical example. Suppose that a filter with a Q of 50 and a bandwidth of 25 KHz is
needed in a low power handheld receiver. In Chapter 3, the dynamic range of such a receiver
was investigated and found to be on the order of 90 dB.6 This level of performance was based
on amplifiers with output compression points of 0 dBm, implying a power consumption on
the order of 20 mW, assuming 10% efficiency.
Using (6.23), the minimum power consumption (per pole) of a Gm-C filter satisfying this
dynamic range requirement can now be found. If an efficiency of η = 0.1 and a noise figure
of F = 2 are assumed, the minimum power needed is 63 mW per pole. Thus, a 3-pole
design would require 189 mW. In practice, obtaining an efficiency of 0.1 in a Gm-C circuit
may be difficult due to MOSFET linearity problems. Repeating the calculation with a more
realistic value of η = 0.01, a power consumption of 630 mW per pole, or nearly 2 watts
for a 3-pole design is found. Clearly, this is unacceptable in a low power, battery operated
receiver design.
6
The actual result was 81 dB, but included a 10 dB C/N bias not accounted for by (6.23).
145
6.2
Dynamic Range of Q-Enhanced LC Filters
The dynamic range problems of Gm-C filters can be traced to two primary causes:
• the power required to charge the capacitors in each new cycle, and
• the excess noise generated by transconductors operating at large currents.
In Chapter 5, it was postulated that energy storage and exchange in Q-enhanced filters
could help reduce both factors, resulting in substantial improvements in dynamic range
performance. In this section, these improvements will be demonstrated quantitatively.
The dynamic range of Q-enhanced LC filters filters could be derived directly using an
approach similar to that taken for the Gm-C case, and then compared with (6.23). However,
the fundamental mechanisms responsible are better illustrated if the derivation is leveraged
off previous results. Thus, in the following analysis, the noise sources present in the flow
diagrams of the two circuits will be compared first. This leads to the observation that the
noise in the Q-enhanced LC filter is reduced over that of the Gm-C design by a factor of
Qo , where Qo is the base Q of the passive (un-enhanced) LC resonator on which the filter
is built. Then, the currents required in the Q-enhanced LC circuit are examined and shown
to be a factor of Qo less than those in the Gm-C design. Finally, these results are combined
to show a total improvement of Q2o when power consumption is held constant.
The Q-enhanced LC resonator flow diagram of Figure 5.13 is redrawn in Figure 6.4 with the
noise souces added. Of the three noise sources shown, the source associated with gmi can
be neglected for reasons identical to those stated in the Gm-C filter case. An examination
of Equation (5.16) then shows that for Q Qo ,
gmn ≈
146
1
Rp
(6.24)
Ing
g mn
mn
1
g mi
vi
Ing
IC
1
1
mi
In R
1
1
sC
sL
IL
VC
-1
Rp
-1
p
Figure 6.4: Q-enhanced resonator flow diagram with noise sources.
Hence, by (6.3) and (6.4), the noise current power spectral density from the loss resistance
and the transconductor are approximately equal and can be estimated as:
i2nR p (fo ) ≈ 4kT gmn
(6.25)
i2ng m (fo ) ≈ 4kT F gmn
(6.26)
As in the Gm-C filter case, these two sources are multiplied by the factor Q to find the
noise current entering the capacitor, and when converted to the voltage on the capacitor,
yield:
vn2 (fo )
≈ (4kT (F + 1)gmn )Q
2
1
ωo C
2
(6.27)
However, in the Q-enhanced LC filter case, gmn is given by:
gmn ≈
1
1
=
Rp
Qo ωo1C
147
(6.28)
Combining this with (5.10) for the Gm-C filter’s resonant frequency ωo yields
gmn ≈
1
gm
Qo
(6.29)
where gm is the Gm-C filter’s transconductance value.
Thus, the noise originating from the loss resistance and the negative resistance circuit
tranconductance used in the Q-enhanced LC filter is a factor of Qo less than that from the
two transconductances used in the Gm-C filter. The desired result then follows immediately
from analogy with (6.11):
DR ≈
2 C
Vmax
Qo
kT (F + 1)Q
(6.30)
Dynamic Range Versus Power Consumption and Bandwidth
To complete the comparison of the two filters, an expression in the form of (6.23) must now
be derived. In the Gm-C filter, transconductors drive the circuit capacitances directly and
must supply all current necessary to charge the capacitors to the designed voltage vmax ,
resulting in the current imax given by (6.22). However, in the Q-enhanced LC filter, the
current imax needed from the transconductor is only that required to make up for losses in
the inductor. Thus,
imax =
vmax
vmax
1
=
=
vmax ωo C
1
Rp
Qo
Qo ωo C
(6.31)
which is a factor of Qo less than in (6.22). Thus, the Q-enhanced LC filter achieves a power
reduction of Qo at the same time the dynamic range is improved.
148
To cast this result in a form that can be compared directly with (6.23), (6.31) can be
combined with (6.20) and (6.30) to give:
DR ≈
ηPDC
Q2
2πkT (F + 1)BQ2 o
(6.32)
This is a significant improvement, even for the case of low quality factor inductors. For
example, if Qo = 5, the dynamic range improvement (assuming other factors are equal) is
over 14 dB. Alternatively, for the same dynamic range as a Gm-C filter, the Q-enhanced
LC filter requires over an order of magnitude less power.
Returning to the numerical example used for the Gm-C filter, a 90 dB dynamic range Qenhanced LC filter with B = 25 KHz, Q = 50, and Qo = 5 requires only 1.9 mW per pole
if η = 0.1 and F = 2 are assumed. Even for the more practical case of η = 0.01, the power
is 19 mW per pole, and is still compatible with the 20 mW per amplifier consumption of
the example receiver design.
6.3
Relationship of Filter Dynamic Range to Receiver Dynamic Range
Applications for integrated bandpass filters within radio receivers fall into two major categories: preselect filters, and channel select filters. In the preceding sections, the bandwidth
B used in deriving (6.23) and (6.32) was the 3 dB bandwidth of the filter resonator. For
channel select filters, this bandwidth can be assumed to be approximately equal to the
width of the radio channel, and hence the noise bandwidth seen by the receiver’s demodulator. Thus, (6.23) and (6.32) can be used directly to upper bound the instantaneous in-band
dynamic range of a receiver in which the filter is used.7 However, for preselect filters, the
filter bandwidth is generally wider than the radio channel, and (6.23) and (6.32) require
7
The term instantaneous is used here to emphasize that the total dynamic range of the receiver can be
higher due to the use of AGC or attenuators as discussed in Chapter 3.
149
modification.
For preselect filters, the noise actually seen by the receiver’s demodulator is reduced by the
IF channel select filter according to
PnIF
BnIF
=
Pnf ilt
Bnf ilt
(6.33)
where PnIF is the noise power measured in the IF filter’s noise bandwidth BnIF , and Pnf ilt
is the noise power measured in the preselect filter’s noise bandwidth Bnf ilt . Thus, the
dynamic range measured at the receiver’s detector will be a factor of
Bnf ilt
BnIF
larger than that
suggested by (6.23) and (6.32).
Combining (6.33) with the dynamic range expressions in (6.23) and (6.32), and with the
resonator noise bandwidth in (6.9) yields the following results for the instantaneous dynamic
range of active preselect filters:
ηPDC
8kT F BnIF Q2
(6.34)
ηPDC
Q2
4kT (F + 1)BnIF Q2 o
(6.35)
DRGm−C ≈
DRQE−LC ≈
6.4
Blocking and Spurious Free Dynamic Range
Expressions (6.23), (6.32) and (6.34), (6.35) provide upper bounds on the instantaneous
total dynamic range of a receiver employing Gm-C and Q-enhanced LC filters. However, as
noted in Chapter 3, the total dynamic range of a receiver can be increased through AGC
and switchable attenuator circuits, and is therefore not a major concern. The most critical
dynamic range measures are, instead, those that determine the receiver’s ability to function
150
relative to out-of-band interferers. Thus, the blocking dynamic range and spurious free
dynamic range of the filters are critical performance concerns.
If the efficiency used in evaluating (6.23), (6.32), (6.34), and (6.35) is defined as the ratio
of the 1 dB compression signal power to the DC power consumed, then these expressions
can be used to estimate the filter’s BDR relative to in-band signals. The in-band SFDR
can then be estimated through (3.24), provided the BDR is properly converted to dB.
For preselect filters, these equations will give reasonable bounds on receiver performance
relative to signals that fall within the service band. However, for signals which fall outside
the service band, or for the case of IF filters where interferers are necessarily outside the
passband of the filter, further study is needed.
In the following analysis, assume that the filter in question is a single second-order resonator
of the form shown in Figure 5.5 or Figure 5.12, and that the input transconductor gmi is
chosen to provide a voltage gain greater than 1. Under this condition, signal compression
will be produced by nonlinearities at the filter’s output before being produced in the input
transconductor’s voltage to current transfer characteristic.
The output compression point will be set by the filter’s power consumption, and will be
essentially the same for signals inside or outside the filter passband. However, since the
gain of the filter is a function of frequency, the input compression point will be larger for
signals outside the passband than for those within the passband and the effective BDR will
be higher than predicted by (6.23), (6.32), (6.34), and (6.35).
These effects may be most easily quantified by adopting the terminology of Chapter 3 in
which powers and gains are written in dBm and dB respectively. Let the compression point
and the noise floor referred to the input (in dBm) be written as Pci and M DS, respectively.
Then the BDR may be expressed in dB as:
151
BDR = Pci − M DS
(6.36)
If the filter has a power gain G in dB, then the output compression point Pco may be written
(in dBm) as
Pco = Pci + G
(6.37)
which may be combined with (6.36), to yield:
BDR = Pco − G − M DS
(6.38)
In this equation, both Pco and M DS are fixed, but the gain term G varies with frequency
f . For signals outside the filter passband, G may be approximated as:
f − fo B/2 G(f ) = Go − 20log (6.39)
where Go is the midband gain, fo is the center frequency, and B is the 3 dB bandwidth.
Inserting (6.39) into (6.38), and combining with (6.36) and (6.37), the BDR may then be
written as:
f − fo BDR(f ) = BDR(fo ) + 20log B/2 (6.40)
Thus, BDR is found to improve with frequency offset as anticipated. For a preselect filter,
this improvement occurs only for signals outside the service band. However, for an IF
channel select design, the improvement occurs for signals inside the service band and can
be significant even at relatively small frequency offsets. For example, evaluation of equation
152
(6.40) yields 6 and 12 dB increases for the cases of adjacent and alternate channel interferers
respectively.
Variations in SFDR with frequency offset can be derived similarly, and are found to be
f − fo +2
SF DR(f ) = SF DR(fo ) + 20log B/2 (6.41)
where the additional factor of 2 dB results from the fact that the second tone in the two
tone signal used in the SFDR measurement is at a frequency of 2(f − fo ), and is therefore
attenuated more than the tone at f − fo .
The ultimate dynamic range performance at large offsets is determined by nonlinearities in
the input transconductor’s voltage to current transfer function and can be very high, even
in low power designs. For example, with a MOSFET-based transconductor biased at 1 V
above threshold, and a 1 KΩ source impedance, the 1 dB compression point referred to the
filter input will be on the order of 1 mW or 0 dBm. If the input signal noise floor is -120
dBm, the ultimate BDR at large signal offsets will approach 120 dB.
Higher-Order Filters
Equations (6.40) and (6.41) were derived for the case of a simple second-order resonator.
To extend these results to higher-order filter designs, the structure of the higher-order filter
must be considered. The improvement will depend on how the gain from the filter input to
each internal node within the filter varies with frequency. The limiting factor will be the
node at which saturation first occurs.
For example, Figure 6.5 shows transfer functions from the input of a fourth-order (2-pole)
coupled resonator filter to the capacitor node in each of the filter’s two resonator sections.
The effect of coupling is a broadening of the first resonator’s bandwidth and the creation of
153
the desired two-pole response at the second resonator’s output. However, the filter rolloff to
the first resonator is still that of a single-pole transfer function. Thus, the equations derived
above for the simple resonator remain valid in this higher-order design provided suitable
adjustments are made for the bandwidth B.
40
Gain (dB)
30
20
10
0
0.95
1
1.05
Frequency
Figure 6.5: Transfer functions to internal nodes in coupled resonator filter.
Although it is possible to design filters in which the response to all internal nodes is N-pole,
preliminary analysis of these filter structures suggests that they possess a significantly degraded initial in-band dynamic range. Thus, they would possess an overall inferior dynamic
range performance relative to that of the coupled resonator structure assumed here.
6.5
Limitations on Filter Q
The dynamic range of Gm-C filters has been extensively studied in the literature and is
described as being inversely proportional to Q. However, as shown in this chapter, the
dynamic range of these filters is actually inversely proportional to the square of the filter Q
when power consumption and bandwidth are held constant.
154
To understand this result and the limits it places on Q values of active filters used in radio
receivers, consider the case of a receiver front-end design such as that shown in Figure 6.6a
in which a passive preselect filter is employed. Although the passive filter may have very
high dynamic range by itself, downstream active circuits within the receiver will limit the
maximum signal levels allowed, and thus the dynamic range of the system as a whole.
G = 0 dB
G = 20 dB
G = 20 dB
Passive
Filter
Active
Filter
NF = 6 dB
NF = 6 dB
(b)
(a)
Figure 6.6: Receiver front-ends employing a) passive filter, and b) active filter.
The dynamic range of this system when used in a receiver with a signal or IF bandwidth
BnIF , can be written in the form:
DR =
Pco
kT F GBnIF
(6.42)
where Pco is the amplifier’s output compression point, F and G are the amplifier’s noise
figure and gain, respectively, and k and T have their usual meanings. For simplicity, the
insertion loss of the filter is assumed to be zero.
In general, Pco will be limited by the amplifier’s power consumption PDC and efficiency η,
so that (6.42) may be rewritten as:
DR =
ηPDC
kT F GBnIF
155
(6.43)
Hence, in this system dynamic range is seen to be independent of filter Q. For example, a
system employing a 1 MHz filter centered at 10 MHz (Q1 = 10) will have the same dynamic
range as a system employing a 1 MHz filter centered at 100 MHz (Q2 = 100). In an active
filter implementation, however, the dynamic range of the 100 MHz filter will be a factor of
(Q2/Q1 )2 = 100 less than that of the 10 MHz filter due to the higher Q needed to obtain
the smaller fractional bandwidth.
This degradation comes about because the only way to raise the Q of the active filter without
increasing its power consumption is to hold the capacitive reactances constant. Thus, to
raise the filter Q, the transconductance gmQ in the Gm-C filter design must be decreased,
or the equivalent parallel resistance in the Q-enhanced LC filters must be increased. In
either case, the principal noise sources in the circuit remain constant, but the gain of the
filter from the noise sources to the filter output rises, increasing the filter’s noise floor and
lowering the dynamic range. Thus, degradation in dynamic range can be viewed as the
result of increases in regenerative gain needed to realize the desired Q.
However, a moderate amount of gain can be tolerated and is needed in a receiver front-end
to maintain good noise performance as discussed in Chapter 3. Thus, filters with Q > 1
may be acceptable, despite their inherent dynamic range degradations, if Q is limited to
moderate values. In the receiver implementation of Figure 6.6a, the gain that degrades
receiver dynamic range is that of the amplifier, whereas in the active filter implementation
in Figure 6.6b, the gain is an integral component of the Q enhancement process and can be
provided by the filter itself.
The in-band dynamic range of these two systems can be compared by taking the ratio of
Equations (6.42) and (6.34) with appropriate subscripts added to distinguish the terms in
each expression. For the case of a Gm-C active preselect filter, the following expression
results:
156
DRpassive
ηpassive FGm−C PDCpassive 8Q2
=
DRGm−C
ηGm−C Fpassive PDCGm−C G
(6.44)
By specifying that the powers for the two systems should be equal, and by making the
simplifying assumption that similar efficiencies and noise figures can be achieved in the
two cases, (6.44) can then be solved for the maximum allowable filter Q that will prevent
in-band dynamic range degradation in the overall receiver system:
s
QmaxGm−C ≈
√
G
= 0.35 G
8
(6.45)
Repeating this analysis for the case of Q-enhanced LC filters, yields the result:
s
QmaxQE−LC ≈
√
G
Qo = 0.35 GQo
8
(6.46)
To determine numerical values for Q, the gain G must be specified. Depending on system
performance requirements and cost factors, typical receiver designs may use gains ranging
from 10 dB to 40 dB within the front-end and down-converter circuits preceding the IF
filter. Thus, it is instructive to tabulate Qmax values for cases with G varying between
these values. The results are shown in Table 6.1.
Table 6.1: Approximate upper bounds on active filter Q when used in radio receivers.
System Gain
10
20
30
40
dB
dB
dB
dB
QmaxGm−C
QmaxQE−LC
Qo = 2.5 Qo = 5 Qo = 10
1.1
3.5
11
35
2.8
8.8
28
88
5.5
18
55
175
11
35
110
350
In using Table 6.1, it is important to remember that the derivation of (6.45) and (6.46)
157
assumed similar noise figures and efficiencies could be achieved in the two systems. Moreover, the power consumption of only a single resonator was accounted for, and the case of
out-of-band dynamic range performance was not considered. Thus, these results should be
used only as a preliminay guide.
Equations similar to (6.45) and (6.46) can be derived for the case of IF filters. However, in
this case, higher Q values can be tolerated because the signals causing blocking and intermodulation distortion are necessarily outside the filter passband. The increase in allowable
Q will depend on what interference signals are expected. If the system design precludes
the existance of large adjacent channel signals, then the worst case interferers will reside in
alternate radio channels and an improvement of 6 to 12 dB can be expected depending on
the details of the filter design. This can be used to buy an additional factor of 2 to 4 in Q,
or to make up for the increased power consumption of a high-order filter design.
158
Chapter 7
Tuning Techniques for Active Filters
In addition to dynamic range limitations discussed in Chapter 6, continuous-time integrated
active filters suffer from significant center frequency and quality factor errors, especially
when operating at high Q (e.g. Q > 10). These problems generally require the incorporation
of some form of frequency and/or Q tuning control system into the filter design.
In this chapter, the major tuning control systems that have been proposed in the literature
are reviewed and a new class of techniques referred to as orthogonal reference tuning (ORT)
is investigated in detail. In addition, several simplifications of control system designs are
proposed. Before proceeding with these discussions, however, some introductory remarks
are presented to illustrate the need for tuning control.
7.1
The Need for Filter Tuning
The need for tuning of active, continuous-time filters was introduced briefly in Chapter 5.
In the case of a Gm-C design, the filter’s center frequency ωo and quality factor Q were
derived in terms of the transconductances and capacitances used, and were found to be:
ωo =
Q =
159
gm
C
gm
gmQ
(7.1)
(7.2)
Since transconductances and capacitances in integrated circuits have large tolerances (typically ±10% or higher [97]), and their deviations are largely uncorrelated, equation (7.1)
suggests that initial frequency errors of ±20% or more can be expected. In contrast, matching and tracking of like components on an integrated circuit can be as tight as ±0.1% [175]
[97]. Thus, (7.2) suggests that Q accuracies as good as ±0.2% may be feasible. In practice, however, the Q of high frequency Gm-C filters is a more complex function of circuit
parameters than suggested by (7.2), and further discussion is needed.
The derivation of (7.2) was based on the assumption that ideal transconductors and integrators can be created. At low frequencies (e.g. < 1 MHz), this assumption is valid since
transconductors with very high output resistances can be realized with the cascode design
approach illustrated in Figure 7.1. At low frequencies, cascoded transconductors can be
loaded with capacitances to form integrators with precise 90o phase shifts at the filter’s
natural frequency ωo . As long as ωo is sufficiently low relative to that of the parsitic pole
ωp introduced at the internal node between the two transistors, the existence of parasitics
can be ignored in the design.
io
io
vbias
vi
g mvi
r ds (2+ g mr ds )
vi
Figure 7.1: Use of cascode design to raise circuit output resistance.
The required separation between ωo and ωp , however, is a function of the filter Q. In high
Q designs, transfer function distortions and stability problems can occur even when ωp is
several orders of magnitude above ωo [99]. To see this, consider the modified resonator flow
diagram shown in Figure 7.2 in which the effect of the pole at ωp in each transconductor
has been represented by small phase errors added to the nominal −90o integrator phase.
160
The effect of the pole can be modeled as a multiplication of the effective transconductor
value by:
P (s) =
ωp
s + ωp
(7.3)
and for ωo ωp , the phase error introduced can be approximated at ω = ωo by:
θ=6
ωp
ωo
≈−
jωo + ωp
ωp
radians
(7.4)
From the derivation shown in Figure 7.2, the main effect of excess integrator phase is seen
to be an enhancement of the resonator Q, and the effective Q may be found in terms of the
ratio of ωo and ωp as:
Qef f =
1
Q
1
Q
=
1 − 2 ωωop Q
+ 2θ
(7.5)
Evaluation of (7.5) shows that a filter with Q = 100 will experience a 10% Q increase
for ωp = 2000ωo , while for ωp = 200ωo , the same filter would oscillate. These two cases
correspond to filters with center frequencies of 1 MHz and 10 MHz implemented in a process
with fT > 2 GHz, clearly illustrating the practical problems associated with extending GmC bandpass filter designs to higher frequencies.
Various approaches exist, of course, for correcting the Q errors predicted by (7.5). The
simplest is to increase the intentional damping term
1
Q
by increasing the transconductance
gmQ , while a more robust approach described in [99] introduces a pole-zero pair to correct
effects over a wider frequency range. Both approaches, however, involve the use of circuits
which do not match or track the ratio
ωo
ωp
accurately enough to avoid the use of post-
fabrication Q tuning.
161
ωο
u(s)
ωο
eθ
j
s
1
s
e
jθ
-1
Q
-1
s = j ωο ,
ωο
u(s)
1
j ωο
|θ| << 1
ωο
(1+j θ)
j ωο
(1+j θ)
-1
Q
-1
u(s)
1
θ
θ
-j
-j
-1
Q
-1
θ
Q
u(s)
1
<< 1 , |θ| << 1
-j
-j
-1
Q
−2θ
-1
Figure 7.2: Effects of excess integrator phase on resonator design.
162
One additional option is to avoid the use of the cascade circuit altogether, so that internal
nodes and associated parasitic poles are eliminated [68]. Unfortunately, the output resistance of the basic single-transistor transconductor is then too low to realize high-Q filter
designs. While intentional excess phase lag could be introduced, or negative resistance circuits could be added to raise the effective Q, the Q tuning problem re-appears in either case.
Thus, in general, high frequency, high-Q, Gm-C filters require tuning of both frequency and
Q to realize accurate (and stable) responses [75].
The requirement for filter tuning implies the need for circuits that allow frequency and Q
to be varied through current or voltage control. This need can be satisfied by a variety of
design techniques, but is often addressed through use of current programmable transconductors such as the differential pair shown earlier in Figure 5.5 [64]. By implementing the
transconductances gm and gmQ with these designs, frequency and Q can be independently
set as shown in Figure 7.3.
gm
VC 1
vi
gmi
C
gmQ
C
1
Q Control
gm
VC 2
2
f Control
Figure 7.3: Control of frequency and Q in Gm-C resonators.
Similar tuning considerations apply to the case of Q-enhanced LC filter design, although
the circuit parameters defining resonant frequency and Q are somewhat different. This
subject has not been treated in significant depth in the literature and a full discussion in
this dissertation is deferred to the next chapter. However, the essential points are outlined
in the paragraphs below.
163
The resonant frequency ωo of an ideal Q-enhanced LC resonator is given by the well-known
expression:
1
ωo = √
LC
(7.6)
while the effective Q was found in Chapter 5 to be of the form:
Qef f =
1
Qo
1 − gmn Rp
(7.7)
The inductance L in (7.6) is determined largely by the inductor geometry and can be
accurately controlled. However, the capacitance C may vary by ±10% or more, so that
frequency deviations of ≥ 5% can be expected. Thus, tuning of frequency becomes a
necessity for all but the lowest Q resonator designs.
In the case of quality factor, the need for tuning may be even more pronounced. Parameters
that determine gmn are uncorrelated with those that determine the inductor loss resistance
Rp, so that the denominator in (7.7) becomes difficult to control when gmn Rp → 1. Thus,
at large Q enhancements (
Qef f
Qo
1), tuning of Q is required.
The need for tunable frequency and Q circuits in Q-enhanced LC filters can be satisfied
though techniques such as those shown in Figure 7.4. Here, the control of filter Q is provided
in a manner essentially equivalent to that used in the Gm-C case. However, control of
frequency requires some means of varying circuit capacitance or circuit inductance. This
requirement can be satisfied with varactor diodes (if supported in the process), or through
switchable capacitances. Alternatively, the effective inductance value can be varied through
the technique shown in Figure 7.5 [76]. The advantage of this technique is that continuous
tuning can be provided without the need for varactor diodes. However, since additional
active circuitry and noise is introduced, care must be taken to avoid degrading the filter’s
164
already limited dynamic range.
VC
vi
g mi
gmn
Rp
C
Q Control
L
f Control
Figure 7.4: Control of frequency and Q in Q-enhanced LC resonators.
VC
vi
g mi
gmn
C
Rp
L
K iL
iL
Figure 7.5: Controlling the effective inductance value in Q-enhanced LC resonators.
7.2
Tuning Control Sytem Designs
Having demonstrated the need for filter tuning, this section turns to the question of how
to design frequency and Q control systems. Ideally, one-time tuning could be performed
using simple trimming techniques following fabrication, and no control system would be
required. Unfortunately, cost and technical problems associated with trimming, combined
with strong temperature and voltage dependences found in Gm-C and Q-enhanced circuits
rule out this option for all but the lowest frequency and lowest Q implementations. Thus,
165
control system designs are needed in which tuning can be performed while the filter is in
use.
Unlike the fundamental dynamic range limitations discussed in the previous chapter, the
design of tuning control systems is an engineering problem, which in theory can be solved.
In practice, however, control systems proposed in the literature are sometimes complex and
may involve substantial amounts of additional circuitry, over and above that required by
the filter itself. The issue of complexity is often overlooked, but is an important facet of
design, and will play an important part in discussions in the following sections.
7.2.1
Master-Slave Tuning
Master-slave tuning, was the first technique developed for real-time tuning of active continuoustime filters [89], and continues to be the dominant approach used today. In the classic
master-slave architecture shown in Figure 7.6, an on-chip oscillator (the master) is implemented using circuits similar to those employed within the filter to be tuned (the slave).
Both circuits receive the same frequency control input, which is derived by phase locking
the master oscillator to an external reference. Thus, when the master oscillator’s frequency
is set, the passband of the filter is properly tuned.
Filter
In
External
Reference
Filter
Out
Slave Filter
Phase/Freq
Detector
Loop
Filter
F Control
Master
Oscillator
Figure 7.6: Master-slave frequency tuning.
166
The accuracy of this technique depends on the degree of matching and tracking between
master and slave circuits. With careful design, capacitances as well as transconductances
can be matched to closer than 1% [97], implying frequency accuracies down to 2% or better.
Although not indicated in Figure 7.6, the master-slave approach can also be used to provide
control of filter Q [106]. One approach is to construct the master oscillator in the form of
a second-order resonator structurally identical to that used within the filter to be tuned,
but with a lower value of gmQ (or gmn ). The value of gmQ (gmn ) for both master and
slave are made tunable as shown in Figure 7.3, and share a common Q control input. By
adding a control loop to precisely set the amplitude of the master circuit’s oscillation, the
exact setting for Q → ∞ is then obtained, and with a fixed offset between master and slave
transconductances designed into the filter, a finite and known filter Q is achieved.
Several variations of the master-slave technique have been reported in the literature (c.f.
[75] and [83]. However, all implementations share the following essential features:
• Tuning relies on matching and tracking of on-chip components, limiting the frequency
accuracies and Q values that can be obtained.
• Substantial additional circuitry is required, adding to chip real-estate usage, power
consumption, and cost.
• An external tuning reference signal must be supplied, introducing the need for careful
design to minimize reference signal feedthrough, and potentially adding to the cost of
the system in which the filter is used.
Although none of these limitations is severe enough to prevent use of the master-slave technique in low to moderate Q filter designs (Q < 50), higher Q designs require an alternative
approach. These approaches, discussed in the following sections, are designed to eliminate
the reliance on component matching, but in some cases can help to reduce or eliminate
other limitations of the master-slave approach as well.
167
7.2.2
Self-Tuning
Self-tuning [108] was the first technique proposed for achieving higher Q values than those
possible with the master-slave approach. The essential concept behind this technique is
illustrated in Figure 7.7.
Filter
In
Filter
Out
Filter
Freq
Q
Tuning
Control
Figure 7.7: Self-tuning technique.
During use, the filter is periodically taken off-line and tuned directly, either by passing
a known reference signal through it, or by some other suitable means. Since the filter is
directly tuned, no ratios of on-chip components are involved, and the accuracy of tuning is
theoretically unlimited. However, in a practical implementation, the following issues must
be addressed:
• Processing of the desired signal is interrupted while the filter is being tuned.
• Suitable tuning control circuits must be designed to measure and adjust the filter
frequency and Q.
• The filter must show good short-term stability to maintain its frequency and Q settings
between tuning updates.
A solution to the first problem was offered by Tsividis in [108]. This solution consisted of
switching between two filters, with one performing the desired signal processing task while
168
the other undergoes tuning updates. Whether or not the resulting transients associated
with filter switching are acceptable will depend on the design of the system in which the
filter is used.
The second problem, involving the design of the required tuning control circuits was not
discussed in the original paper. However, such circuits could be built using a variety of
approaches. One possibility is to employ an external reference signal with a frequency
equal to that of the desired filter center frequency. Assuming that only a single resonator
in the filter is tuned at a time, this signal could be passed through the resonator and its
phase and amplitude measured at the resonator’s output to determine center frequency and
Q errors, respectively. Appropriate control updates could then be applied to force the filter
back to the proper frequency and Q values. Details of this approach are presented in the
discussion of the orthogonal reference technique later in this chapter.
The third problem, the need for short-term frequency and Q stability to hold the filter
parameters constant between tuning updates, will depend on the circuit design. Although
this issue is not addressed in the literature, tests conducted by the author and described
in [105] show that this requirement can generally be met for filters with Q values up to at
least several thousand.
7.2.3
Correlated Tuning Loop
An interesting alternative technique which avoids both the matching and tracking limitations of the master-slave design and the filter switching problems associated with the
self-tuning approach is illustrated in Figure 7.8 [100]. This technique, referred to as the
Correlated Tuning Loop (CTL), uses the signal being processed to characterize the filter’s
response. Assuming that this signal contains energy in the neighborhood of the filter center
frequency, narrowband in-phase and quadrature samples of this energy can be measured at
both the filter input and output with the circuits shown. These measurements can then be
169
used to determine the phase and amplitude shift introduced by the filter, and to update the
filter’s frequency and Q settings appropriately.
Filter
In
Filter
Out
Second-Order
Filter
Freq
External
Reference
Q
I/Q
Mixers
I/Q
Mixers
Q
I
Q
I
LPFs
LPFs
I
External
Reference
Q
I
Q
Control Algorithm
Figure 7.8: Correlated Tuning Loop technique.
Although the CTL technique is theoretically sound, and has been demonstrated in hardware
[109], it contains at least three significant limitations:
• It assumes certain spectral properties are always present within the input signal.
• It requires significant extra circuitry (4 mixers plus a quadrature reference).
• It requires measurements to be made directly at the filter input.
The last two points are of particular concern in radio receiver applications. Essentially
what is required are two auxiliary direct-conversion receivers. Aside for the complexity and
power consumption of these circuits, the need for measurements at the input of the filter
implies that at least one of these receivers must have very high dynamic range - possibly
greater than that of the filter itself. Thus, the practicality of the CTL technique is open to
serious question in this application.
170
7.2.4
Adaptive Filter Tuning
A potentially attractive alternative to the CTL technique which eliminates the need for
measurements at the filter’s input has been proposed by Kozma, Johns, and Sedra in [103].
This technique uses adaptive filter theory to adjust filter parameters based on comparisons
with a known model.
Filter
In
Main Filter
Freq
Filter
Out
Q
Adaptive
Control
Algorithm
Reference
Filter
Freq
Q
Tuning
Control
Figure 7.9: Adaptive Filter Tuning.
In the configuration shown in Figure 7.9, the main filter can be tuned directly without
interrupting the signal processing task. The reference filter shown is used as a model, and
is periodically adjusted using the self-tuning technique described previously. Since the selftuning technique does not rely on matching, the reference filter can be adjusted to arbitrary
accuracy. The adaptive control algorithm then compares the output of this filter with that
of the main filter, adjusting the latter to make the two outputs match. Hence, the main
filter is tuned to the same level of accuracy achieved by self-tuning of the reference filter
while continuously processing the desired signal.
171
The price paid for this achievement, however, is a relatively high level of complexity, adding
to power consumption and cost. In addition, this approach, like the CTL design discussed
in the previous section, assumes certain spectral properties are present in the input signal to
allow it to be tuned. This latter limitation is removed by the orthogonal reference technique
described in the following section.
7.2.5
Orthogonal Reference Tuning
Copyright 1994 IEEE. Reprinted with permission.
1
In this section, an alternative approach to tuning high-Q filters is proposed and investigated
in detail. The new technique, referred to orthogonal reference tuning (ORT), uses only a
single instance of the filter, requires no filter switching, and does not rely on the input
signal and its characterization for tuning. Instead, a known reference signal is employed
and is present in the filter passband together with the RF/IF signal being processed. This
is permissible provided the reference signal can be made orthogonal, or nearly orthogonal,
to the processed signal so that the two signals can be separated at the filter output to the
degree necessary to meet overall system design goals.
The extent to which orthogonality can be achieved depends on the signal type being processed, and different methods will be appropriate for different applications. For example,
with a receiver designed for use with digital biphase shift-keyed (BPSK) modulation [23],
essentially complete orthogonality can be achieved using a quadrature phase signal as a
tuning reference. Another approach offering complete orthogonality is possible in communication systems employing time division multiple access (TDMA). In TDMA systems, time
orthogonality could be achieved by applying the tuning reference signal to the filter during
1
Portions of this subsection are extracted from the following paper by the author: W. B. Kuhn,
A. Elshabini-Riad, and F. W. Stephenson, “A New Tuning Technique for Implementing Very High Q,
Continuous-Time, Bandpass Filters in Radio Receiver Applications”, IEEE Int. Symp. on Circuits and
Systems, 1994, pp 5-257 - 5-260.
172
time slots which are not of interest to the receiver. In this case, the concept of an orthogonal
reference reduces to the self-tuning technique discussed earlier.
A more general approach which achieves approximate orthogonality is illustrated in Figure
7.10. In this design, referred to as spread-spectrum orthogonal reference tuning (SS-ORT),
the reference signal is modulated with a direct-sequence spread-spectrum pseudo-random
bit sequence (PRBS) [23].
Automatic
Gain
Control
Tunable
Bandpass
Filter
From
IF Amplifier
Q
Spreading
Sequence
Generator
To
Demodulator
Freq
Control
Loop Filters
Lowpass
Filters
Reference
Oscillator
Reference
Recovery
Correlators
Figure 7.10: SS-ORT Technique Simplified Block Diagram.
The SS-ORT technique is well suited to use with two of the most popular signal formats
found in radio receiver applications - wideband frequency modulation (WBFM) and low
dimension digital phase modulations such as QPSK. In the following discussion, assume
that the filter being tuned is a simple second-order bandpass section. Higher order filters
can be implemented as cascade designs with straightforward extensions that tune individual
second-order sections.
During operation, the receiver automatic gain control (AGC) shown in Figure 7.10 maintains
the desired IF input signal at a constant level as seen by the filter. A reference oscillator at
the filter’s center frequency is modulated with a PRBS waveform and then summed with this
173
IF input signal so that both are present in the filter simultaneously. The amplitude of this
reference waveform is held to a fraction of the IF signal’s amplitude to minimize interference
in the demodulation process. This fraction is described by a carrier-to-reference power ratio
(C/R) at the filter output, which is maintained at a level of approximately 20 dB.
At the filter output, the reference must be recovered from the combined signal and then
used to determine control inputs for adjusting the filter’s frequency and Q. Using established
methods from the area of spread-spectrum systems, recovery can be achieved by correlating
the output of the filter with the reference signal applied at the filter’s input, and then
lowpass filtering the result. In the process, the higher amplitude IF signal is spread, so
that only a fraction of its power passes through the lowpass and control loop filters. If this
fraction is sufficiently small, accurate tuning information can be obtained.
Reference Recovery Analysis
A quantitative analysis of the reference recovery problem can be made with the help of
Figure 7.11. The input signal to the bandpass filter is a summation of a spread reference
waveform r(t) and an IF input signal waveform c(t).
To simplify the analysis, the IF input signal will be represented as a simple sinusoid. The
two signals may then be expressed as:
√
2Rs(t)cos(2πfot)
√
c(t) =
2Ccos(2πfot + θc )
r(t) =
(7.8)
(7.9)
where R and C represent the signal powers, fo represents the frequency, s(t) is a pseudorandom sequence waveform with discrete amplitudes of +1 and -1 and bit duration Tb =
1
Rb ,
and θc is a relative phase angle. These two signals are passed through the second-order
174
Frequency Control
ve (t)
K
f
s
rF (t)
c(t) + r(t)
Bandpass
y(t)
- 90
o
Filter
ve (t)
rQ (t)
Reference
KQ
s
+
Q Set
Q Control
Figure 7.11: SS-ORT Control System.
bandpass filter producing an output that may be approximated by:
y(t) =
√
Q(t) √
[ 2Rs(t)cos(2πfot + θF (t)) + 2Ccos(2πfot + θc + θF (t))]
Qo
(7.10)
where Qo is the nominal filter Q, Q(t) is the actual Q at time t, and θF (t) is a phase offset
created by an error in the filter center frequency. This approximation is valid for small
frequency errors provided the reference signal is narrowband with respect to the filter’s 3
dB bandwidth Bo . For a small frequency error of ∆F (t), θF (t) is calculated from the filter
transfer function phase angle as:
θF (t) =
dθ
2
∆F (t)
∆F (t) =
dF
Bo
(7.11)
As shown in Figure 7.11, the filter output y(t) given in (7.11) is then correlated with scaled
and phase-shifted versions of the reference waveform. The response of the control system to
disturbances in the filter Q and frequency, and the magnitude of tuning errors introduced
175
by the presence of the IF input signal c(t) may be determined by correlating y(t) with the
reference signals rQ(t) and rF (t) for the respective control loops:
√
2s(t)cos(2πfo t)
√
rF (t) =
2s(t)sin(2πfo t)
rQ(t) =
(7.12)
(7.13)
The result for the frequency control loop is derived below. The Q control loop may be
analyzed similarly.
Using superposition, we examine the control system under the influence of the reference
signal r(t) first and then add in the effects of c(t). With c(t) set to zero, the output of the
correlator in the frequency control loop averaged over any large number of RF cycles by the
lowpass filter is found from (7.10), (7.11), and (7.13) as:
√
√ 2
ve (t) ≈ − Rsin(θF (t)) ≈ − R ∆F (t)
Bo
(7.14)
The resulting control system response may be evaluated with the help of Figure 7.12 in
which Fd represents an internal disturbance in the filter frequency, Fc represents the frequency correction applied by the control system, and Fe is the frequency error of the filter
(equivalent to ∆F (t) in (7.14)). The error voltage at the output of the discriminator and
the frequency control voltage are represented as ve and vc respectively, and Kd , Kf , and
Kc are gain values relating the various voltages and frequency quantities, with Kd given by
(7.14). The closed-loop response of this system is found to be:
H(f ) =
Fc (f )
1
=
Fd (f )
1 + j BfCL
where BCL is the 3 dB bandwidth of the control loop given by BCL =
176
(7.15)
Kd Kf Kc
2π
Fd (s)
Fe (s)
+
-
Kd
ve (s)
KF
vc (s)
s
Kc
Fc (s)
Figure 7.12: Frequency Control Loop Block Diagram.
In general one would like to set BCL in (7.15) as large as possible for best correction of
frequency disturbances Fd . However, as shown below, BCL must be kept relatively small
to minimize errors in the control system caused by the IF signal c(t) which until now has
been ignored in the analysis. These errors will be evaluated first by determining a frequency
disturbance value Fd that would produce an output from the discriminator equivalent to
that produced by the presence of c(t) in the filter. Equation (7.15) will then be applied to
find the resulting frequency correction signal Fc which in this case is a frequency control
error produced by c(t).
The worst case (maximum) output from the frequency discriminator is obtained when the
second term of y(t) in (7.10) is in phase with the reference signal rF (t). Under this condition:
ve (t) =
√
Cs(t)
(7.16)
Using (7.14), the equivalent frequency disturbance is:
s
Bo
Fd (t) = −
2
C
s(t)
R
(7.17)
To determine the resulting control system output for this input, the waveform s(t) may be
approximated as a stationary random process with power spectral density (PSD):
177
f
1
SFd (f ) =
sinc2
Rb
Rb
(7.18)
resulting in a PSD at the control system output equal to:
SFc (f ) =
Bo2 C
f
|H(f )|2
sinc2
4Rb R
Rb
(7.19)
If the cutoff frequency of H(f ) is low in comparison with the bit rate Rb of s(t), then the
sinc2 term may be approximated as one, and the mean square magnitude of Fc is found to
be:
Z
Fc2
=
=
∞
SFc (f )df
−∞
Bo2 C
Z
4Rb R
∞
−∞
|H(f )|2df
(7.20)
Recognizing the integral of |H(f )|2 as the noise bandwidth BN of the control system response, the root mean square (RMS) magnitude of frequency error caused by the presence
of the carrier c(t) may then be written:
q
∆FRM S =
The quantity
Rb
BN
Bo
Fc2 =
2
s
C BN
R Rb
(7.21)
is the processing gain Gp of this spread-spectrum system. In terms of this
gain, the normalized RMS error of filter frequency tuning may be written as:
∆FRM S
1
=
Bo
2
s
C 1
R GPF
(7.22)
The normalized RMS error of the filter Q can be derived similarly and is found to be:
178
∆QRM S
=
Qo
s
C 1
R GPQ
(7.23)
Equations (7.22) and (7.23) can be solved for the required processing gain for a specified
RMS variation in frequency and Q. For example, to achieve a 10 percent RMS variation in
Q with C/R=20 dB, the required processing gain is 40 dB. This corresponds to a control
loop noise bandwidth 10,000 times smaller than the PRBS modulation rate. Clearly, the
control loop bandwidth is relatively low, and good short term stability of frequency and Q
in the filter is required.
PRBS Modulation Rate
From the preceding discussion, it is clear that the PRBS modulation rate should be selected
as large as possible to maximize processing gain and control loop bandwidth. However, if
the rate is made too high, the reference signal spreading will be excessive relative to the
filter bandwidth, and group delay introduced by the filter will complicate the correlation
process required for reference recovery.
The group delay of the filter depends both on the filter bandwidth and on which control
loop is being analyzed. The worst case is found to be the delay for the frequency control
loop. For low frequency components of the reference modulation waveform this delay is
given by:
τg =
2
πBo
(7.24)
where Bo is the 3dB bandwidth of the filter begin tuned. For proper correlation, this should
be held to a fraction of the PRBS period. Hence, the constraint on the PRBS rate may be
stated as:
179
RP RBS π
Bo
2
(7.25)
In the prototype implementation described below, however, the relaxed constraint:
RP RBS < Bo
(7.26)
has been found to give satisfactory results.
Effects on the Demodulation Process
To a first-order approximation, the reference signal used in the SS-ORT technique may be
treated as additive white Gaussian noise in estimating the effects on signal-to-noise (S/N)
or bit error rate (BER) at the demodulator output. This approximation is a good one if the
PRBS rate is on the order of the filter bandwidth. The demodulator output S/N or BER
can then be found from established formulas or graphs.
Demodulation theory for wideband FM systems predicts a thresholding effect at carrier-tonoise (C/N) values near 10 dB and substantial S/N improvement factors above threshold
(depending on the modulation index) [179]. For example, S/N at the demodulator output
can be on the order of 50 dB for a C/N value of only 15 dB at the demodulator input. Digital
modulations such as QPSK also exhibit a similar thresholding behavior in the neighborhood
of 10 - 15 dB C/N depending on BER requirements, and at a C/N ratio of 20 dB, the
demodulator output is virtually error free. Hence, a C/R ratio of 20 dB in the SS-ORT
technique should yield similarly error-free performance.
180
Effects on Receiver Sensitivity
Since the reference signal behaves approximately as additive white Gaussian noise, the noise
floor of the receiver will be raised and the sensitivity of the receiver will be reduced by the
presence of the reference signal. However, the effect of the reference on receiver sensitivity
will be small provided that the C/R used is at least several dB higher than the demodulator
threshold. For example, if the demodulator threshold is 10 dB and C/R is set to 20 dB,
then the effective C/N seen by the demodulator is found to be 9.6 dB. Thus, the receiver
sensitivity is decreased by only 0.4 dB.
Experimental Verification
A board-level prototype of the SS-ORT technique was built to verify the technique’s viability
and to check the analytical predictions discussed above. The tuned filter shown in Figures
7.13 and 7.14 was designed to implement a simple second-order bandpass response with a
nominal center frequency of 10.7 MHz and a nominal Q value of 100. These values allow
the filter to be placed in the IF section of a commercial FM broadcast receiver, providing
a realistic demodulator and the ability to assess S/N at the demodulator output through
critical listening tests in addition to more traditional laboratory measurements.
The filter was tested first under manual control to check short term stability. Both frequency
and Q were found to be stable to within less than 5 percent of the nominal bandwidth and
quality factor, respectively, over periods of several seconds at Q values as high as 1000.
These tests confirmed that the small control loop bandwidth used (BN = 1.6 Hz) would be
sufficient to tune the filter transfer function.
The filter was then placed in the control loop implemented with the circuits shown in Figures
7.15 and 7.16 and locked at a frequency of 10.7 MHz and a Q of 100 using a PRBS rate of
100 KHz. C/R was adjusted to 20 dB and swept frequency measurements were performed
181
Figure 7.13: Tunable active filter used for ORT technique prototype.
182
FOUT
F
FIN
Q
+12
VB
R23
10K
R25
430
Q3
2N3904
R24
1K
33p
C15
R26
100K
C16
Q5
2N3904
VB
R29
10K
+12
0.1
C17
0.01
R31
1K
R30
100
R28
430
R32
10K
VB
33p
C18
D2
MV2109
0.1
Q4
2N3904
0.1
D1
MV2109
C19
R27
1K
C14
R33
100K
1000p
C20
Q6
2N3904
R35
620
R34
100
VB
VB
R39
10K
R38
10K
VB
R37
430
Q7
2N3904
R36
1K
Q8
2N5486
R42
1.2K
R41
4.7K
+12
33p
C22
D3
MV2109
0.1
C21
R40
1.5K
R43
100K
0.1
C24
C23
0.01
Figure 7.14: Tunable active filter used for ORT technique prototype.
183
J3
AGC
J2
REFIN
J1
SIGIN
CW
CCW
R3
3.3K
R2
75
R1
75
R5
6.8K
VR1
10K
+12
22K
R4
0.01
C1
C2
0.01
+
-
0.1
C3
6
4
AGC
3
G1
IC1
MC1350
AGC
5
AGC
7
G2
V+
2
OO+
+12
8
1
R6
200
J4
F
2K
R8
2K
R7
C4
0.01
R10
3.3K
J5
PWR
0.01
C6
0.01
C5
R9
CW
CCW
+12
4.7K
VR2
10K
R12
10K
2.2U
C9
0.1
0.1
+12
C8
C7
R11
10K
L1
100U
FIN
+12
FOUT
F
0.01
C10
+12
R14
47K
R13
47K
J6
Q
R16
4.7K
Q1
2N3904
CW
CCW
R19
3.3K
R15
1K
+12
Q2
2N3904
VR3
10K
+12
0.01
C12
C11
R18
10K
R17
5.6K
0.01
C13
2.2U
75
R22
75
R20
75
R21
Q
J9
MON
J8
REF
J7
SIG
Figure 7.15: Prototype ORT technique control system circuitry.
184
J3
REFIN
J2
PRNSEQ
J1
PWR
R1
82
0.01
C2
R2
2K
0.01
C3
C30
R0
7
8
1
OSCE VCC
4
IN
OUT
IC2
NE602 OUT 5
2 IN
OSCB GND
6
3
0.01
+8
1.0U
0.1
0.1
C5
C4
OUT
COM
C1
IN
IC1
78L08
0.1
C6
+8
+12
0.1
C7
VR1
10K
CW
100K
R6
100K
R4
47p
C9
2.7M
R5
0.01
C10
Q1
2N3904
CCW
+12
C8
0.01
R3
3.9K
22p
C11
0.47
C13
+8
3
+
2 -
2
C14
33p
C31
0.47
IC3
LF353
R8
100K
1000p
C12
R7
82K
1
1
5-60p
VC1
D1
1N4148
1
2
0.01
C16
L1
3U
39K
R9
D2
1N4148
1
+8
0.01
C18
0.01
R90
C23
0.1
C20
0.1
0.01
C19
7
8
OSCE VCC
4
IN
OUT
IC5
NE602 OUT 5
2 IN
OSCB GND
6
3
1
7
8
OSCE VCC
4
IN
OUT
IC4
NE602 OUT 5
2 IN
OSCB GND
C15
6
3
J4
Q
1000p
C17
+8
10p
C22
100K
R10
100K
R11
0.01
C21
+8
5
0.47
+
6 -
C25
R14
4.7K
R13
470
Q2
2N3904
R12
820
C28
IC3
LF353
0.47
0.1
C24
5-60p
VC2
7
0.01
C29
1000p
C27
1000p
C26
R16
1.5K
R90
R0
J6
F
J5
REFOUT
Figure 7.16: Prototype ORT technique PRBS generator circuitry.
185
CCW
CW
+5
1
2
3
3900p
C2
VR1
10K
R3
10K
0.1
C1
R1
1M
+5
DIS
THR
CTR
R2
10K
LM556
IC1
TRG
OP
RST
CTR
THR
DIS
0.001
C3
11
12
13
6
5
4
LM556
IC1
RST
OP
TRG
10
9
8
1
2
IC2
+5
74C86
3
J1
PWR
6
5
4
3
2
1
QD
QC
QB
QA
B
A
QH
QE
QF
QG
CLR
CK
74C164
IC4
8
9
10
11
12
13
9
8
IC2
5
6
74C86
IC2
74C86
0.1
0.1
4
C5
C4
OUT
COM
IC3
78L05
IN
10
6
5
4
3
2
1
+5
QD
QC
QB
QA
B
A
QE
QF
QG
QH
CLR
CK
74C164
IC5
8
9
10
11
12
13
R4
10K
J2
PRNSEQ
to check the filter response. Stability of the control loops and variations in frequency and
Q caused by finite processing gain were also assessed. The frequency and Q of the filter in
lock were found to be correct to within the accuracy of the 90 degree phase shift setting
and the Q-adjustment setting provided on the control board. Both loops remained stable
at all input signal frequencies inside and outside the filter bandwidth, and the expected
variations of frequency and Q were observed at zero beat. With a PRBS rate of 100 KHz
(Gp = 48 dB), RMS variations on the order of a few percent were observed as predicted by
Equations (7.12) and (7.12).
With the PRBS rate set to 10 KHz, the control system was also tested at a filter Q value
of 400 (the limit of the adjustment range in the control loop circuitry). Swept frequency
measurements showed proper filter operation and good stability.
Effects of the reference on demodulator performance were measured by placing the locked
filter into the IF section of a commercial FM broadcast receiver. The PRBS rate was set
to 100 KHz and the filter Q was set to 100. With the receiver’s multiplex switch set to
monaural mode the expected high S/N was observed in the audio output. S/N improvement
threshold was observed at the predicted value of 10 dB C/R and at a C/R value of 20 dB
the presence of the reference was virtually undetectable in listening tests. Measurements
performed with the receiver set to stereo mode, however showed clearly perceptible noise.
This behavior is expected since the stereo modulation format for commercial broadcast
uses essentially narrowband FM modulation for the L-R channel information. Hence, the
S/N improvement is substantially reduced and the reference signal injects noise into the
demodulation process.
7.2.6
KIS Tuning
All of the tuning techniques discussed in this chapter have been found to have important
technical and practical limitations, and in some cases, relatively high levels of complexity.
186
Complexity was clearly evident in the SS-ORT technique described in the previous section,
but is also a potential problem in other techniques. In this final section, the issue of
complexity is examined more carefully to determine which techniques, if any, are suitable
for use in commercial products.
The importance of simplicity should not be overlooked in any design endeavor. Intuitively,
simple techniques are often attractive because they possess a certain level of elegance and
robustness. However, from a purely pragmatic viewpoint, simplicity helps to keep development cost and time within allowable limits, and to minimize overall project risk. In the
design of any real system or device, these issues often become deciding factors in what is
implemented and what is not. Thus, the concept of Keep-It-Simple (KIS) tuning is proposed and examined in this section. Where possible, new approaches designed to reduce
complexity and supporting circuitry are also suggested.
Master-slave tuning
The basic master-slave technique discussed in Section 7.2.1 qualifies as a KIS technique
in many ways. Typically, only a simple second-order master oscillator is needed, and the
design and layout of this oscillator can be copied from that of a resonator section used
within the filter being tuned. Design of the remaining amplitude and phase-locked-loop
circuits may require additional time, but is straightforward and involves relatively low risk.
However, the master-slave tuning approach does require significant chip real-estate and
power consumption, and can suffer from problems with clock signal feedthrough into the
signal processing circuits. In addition, for filters operating at very high Q, matching and
tracking may be insufficient to realize a desired level of accuracy. Thus, additional techniques should be considered before committing to this approach.
187
Self-Tuning
Self-tuning offers the potential for very high levels of tuning accuracy, but may be difficult
to implement in systems which require continuous processing of the input signal. In newer
digital radio designs however, where continuous reception is not necessarily required, simplifications may be found by carefully integrating filter tuning requirements into the overall
system design.
For example, in systems employing time division multiple access or time division duplex,
it may be possible to tune the filter during timeslots which are not of interest to the host
receiver. In this case, only a single copy of the filter would be needed as shown in Figure 7.7,
and any potential problems associated with reference feedhrough could be eliminated by
disabling the tuning reference signal when tuning is not in progress. Both of these features
represent significant advantages over the master-slave technique, apart from the ability to
obtain theoretically unlimited tuning accuracy.
The need for an external reference and tuning control loop circuitry in the self-tuning
approach is still a potential problem. However, simplifications in this area may also be
found if the filter design is considered together with that of the host system. One interesting
possibility arises in the case of direct conversion receivers as illustrated in Figure 7.17. In
this design, the preselect filter is self-tuned using the receiver’s existing local oscillator as a
reference, and the receiver’s demodulator circuits and microprocessor are used to measure
and adjust the filter’s frequency and Q. Thus, on-chip circuitry required to support tuning is
reduced to a switch or adder at the filter input, a simple passive attenuator, and any digital
control ports, registers, and D/As needed on the filter die to accept tuning commands and
data and provide suitable control voltages or currents to frequency and Q circuits.
One final possibility for simplification of the self-tuning technique is illustrated in Figure
7.18. This approach could be used in traditional superheterodyne receivers as well as direct-
188
Lowpass
Filter
F,Q
I
Active
Preselect Filter
Lowpass
Filter
Q
A/D, Data
Detection,
and Control
Attenuator
90 o
Figure 7.17: Self-tuning simplifications in direct conversion receivers.
conversion designs.
Filter
Input
Second-Order
BPF Section
Freq
Filter
Output
Q
MUX
From uP
To uP
N
Figure 7.18: Self-tuning simplifications in superheterodyne receivers.
As before, filter frequency and Q control inputs are derived from the host receiver’s microprocessor. However, measurement of frequency and Q is now performed using an amplitude
detector and frequency prescaler incorporated into the filter die. Periodically, the filter is
taken off-line and the Q is increased until the amplitude detector indicates that oscillation
is present. The frequency of oscillation is then measured by the microprocessor’s built-in
189
counter-timer and used to derive frequency correction signals to be fed to the filter. Finally the microprocessor restores the filter to a finite Q by applying a preset offset from
the Q control setting needed to achieve oscillation. With this approach, no reference signal is needed, and on-chip supporting circuitry is reduced to a simple amplitude detector,
frequency prescaler, and associated circuits for accepting tuning control updates2 .
Additional Techniques
Of the remaining tuning techniques, both the correlated tuning loop and the adaptive filter
approach are relatively complex. The spread-spectrum orthogonal reference technique is
also potentially complex, but simplifications may be found in a few special cases.
The need for generating and correlating with a spreading code can be eliminated when
the technique is used within a direct-conversion receiver designed to receive direct-sequence
spread-spectrum signals as shown in Figure 7.19. In this design, no spreading waveform
is needed since the signal itself is spread and therefore approximately orthogonal to the
LO sinusoid. As in the technique illustrated in Figure 7.17, supporting circuitry can be
minimized by the use of the receiver’s existing LO, detector, and data processing circuits.
One additional simplification of the orthogonal reference technique may be possible in the
case of receivers designed for use with satellites. In this case, the potentially complex
AGC function may be unnecessary since the signal being received is relatively constant in
amplitude. At most, a simple AGC with a 10 to 20 dB range would be required.
2
The frequency reference in this case is implicit in the microprocessor’s counter/timer.
190
Lowpass
Filter
F,Q
Variable
Attenuator
I
Active
Preselect Filter
Lowpass
Filter
Q
Attenuator
90 o
Figure 7.19: Orthogonal reference tuning simplifications.
191
A/D, Data
Detection,
and Control
Chapter 8
Q-Enhanced LC Filter Design
In previous chapters, the primary considerations for selecting an on-chip filter implementation technology were identified as operating frequency, Q, dynamic range, and filter tuning.
The need to operate at high frequencies with low power consumption ruled out digital
as well as switched-capacitor options, while non-standard IC processing required to form
on-chip electro-acoustic designs placed this technology outside the scope of the current research. A detailed comparison between Gm-C and Q-enhanced LC filters then led to the
conclusion that the latter technology is the only viable option. Q-enhanced filters provide
an improvement of Q2o in dynamic range over comparable Gm-C designs, making on-chip
bandpass filters with Q values as high as 50 or above practical in low-power receiver designs.
Moreover, the frequency of operation for these filters extends from about 100 MHz to well
into the microwave range, due to the ability to resonate out inevitable circuit parasitic capacitances. Thus, this technology is ideally suited to the design of both RF preselect filters,
and first-IF filters in modern cellular and PCS products.
Q-enhanced LC filtering is a relatively new area of research within the discipline of active
filter design, and to date, only a few investigators have considered this option [57] [77] [94].1
Indeed, at the time of this writing, no fully integrated Q-enhanced LC filters implemented
in silicon have been reported, and circuit-level design techniques are largely unexplored.
Therefore, this chapter will concentrate on detailed circuit-level design issues, complement1
Portions of this chapter have been submitted to the IEEE for possible publication. Copyright may be
transferred from the author to the IEEE without notice.
192
ing and extending the high-level issues presented in Chapters 2 through 7. Further detail
on designing these filters will then be provided in Chapter 9, where two prototype ICs are
described and their measured performance is reported.
The most critical circuits in any Q-enhanced LC design are the on-chip inductors employed.
The quality factors in these devices must be carefully optimized to provide the best possible dynamic range performance, at the lowest possible power consumption. Thus, the first
section in this chapter investigates the design and characterization of spiral inductors and
presents techniques for estimating a spiral’s inductance, quality factor, and self-resonant
frequency. The following section then turns to the issue of Q-enhancement, the implementation of differential circuits, and the design of basic second-order resonator filters. Next,
the accuracy and temperature stability of frequency and Q are studied and methods for
varying frequency and Q to achieve a desired response are considered. Finally, the design
of higher-order transfer functions using magnetically coupled resonators is discussed.
8.1
On-Chip Spiral Inductors
Inductors can be implemented in standard IC processes using planar spiral geometries. The
design and characterization of these devices has been studied for many years within the field
of microwave and millimeter wave integrated circuits (MMICs), where usable inductances
can be fabricated in acceptable chip areas. However, their use in silicon IC processes is
comparatively new, and the properties of spiral inductors in silicon are sufficiently different
from those in GaAs MMIC devices that new modeling techniques are needed.
In a typical two-metal silicon IC process, spiral inductors can be fabricated using techniques
similar to those used in MMIC devices. A typical geometry is illustrated in Figure 8.1
[131]. Top layer metallization usually provides the lowest resistivity and capacitance to the
underlying substrate, and is therefore used for the spiral turns, while the lower metallization
193
layer is used for connection to the spiral center. The inductance values of spirals constructed
in silicon technology are similar to those of spirals operating in free space. This is in contrast
to spirals implemented in GaAs technology where the presence of a high conductivity ground
plane a short distance under the turns lowers the inter-turn magnetic coupling and hence
the overall inductance value [133].
Metal 2
D
Metal 1
0.5um
SiO2
1.5 um
Silicon Substrate
> 200 um
w
(a)
(b)
Figure 8.1: Spiral inductor layout. a) Top view. b) Side view.
8.1.1
Inductance Value
For the square spiral illustrated in Figure 8.1, operating in free space, Bryan and Dill have
suggested that the inductance value can be approximated by [121], [123]:
L ≈ αDN β
(8.1)
where L is the inductance in nH, D is the outer dimension in µm, N is the number of turns,
and α and β are empirically determined constants. Dill [123] provides values of 8.5 × 10−4
and 1.67 for α and β, respectively, for PC board and hybrid spirals. However, the author
has found that values of 8.9 × 10−4 and 1.8 provide a better fit to measurements reported
194
in [131] for spirals implemented in silicon technology. For improved accuracy, numerical
techniques such as those employed by Grover [125] and Greenhouse [124] may also be used.
8.1.2
Inductor Q
A complete characterization of a spiral inductor for circuit design purposes generally involves developing an equivalent circuit to model finite inductor Q and self resonance effects,
in addition to the low-frequency inductance value. A simplified equivalent circuit is shown
in Figure 8.2 and will be used initially to illustrate important concepts. More accurate modeling usually requires either more complex lumped circuit models such as those described
later in this section [128], or the use of two-port S parameters taken from analytic models,
EM field simulations, or prototype measurements [136], [130].
L
Rs
C
C
R sub
Figure 8.2: Simplified spiral inductor model.
Finite inductor Q results primarily from series resistance in the inductor turns (modeled
as Rs in Figure 8.2) [131]. Additional losses result from currents conducted through the
turn-to-substrate capacitances (C) and the underlying semiconducting substrate resistance
(Rsub ). The dominant losses resulting from series resistance can be estimated directly
from the outer dimension D, trace width W , number of turns N , and metallization sheet
resistance rs as:
Rs ≈
N 4D
rs
2 W
195
(8.2)
where Rs is in Ohms, D and W are in µm, and rs is in Ohms/2.
If the metallization thickness is less than the skin depth (2.1 µm at 2 GHz for AlSiCu
metallization), this resistance can be treated as approximately constant with frequency.
Assuming that Rs is the primary inductor loss mechanism, an upper bound on the inductor
quality factor Qo can then be estimated at a frequency f in Hz from Equations (8.1) and
(8.2) as:
Qo ≤
απf N β−1W
2πf L
≈ 10−9
Rs
rs
(8.3)
Letting W scale with the outer dimension D as:
W =
D
4N
(8.4)
to provide a gap between traces equal to the trace width, and noting that (β − 1) ≈ 1, the
following general observations can be made:
• Qo increases linearly with frequency when skin effect can be ignored.
• For a given frequency and number of turns, Qo increases approximately linearly with
W , and hence with outer dimension D.
• For a given frequency and outer dimension D, W decreases with increasing N , so that
Qo is relatively independent of N , and hence of L.
Values of Qo achievable in practice depend on the operating frequency of the filter, the
resistivity of the metallization layers in the target process, the chip real estate that can
be allocated to the spiral inductor, and additional losses from the underlying substrate
and connected circuitry. Achievable values of Qo also depend indirectly on the spiral’s
196
self resonant frequency, which upper bounds the outer dimensions and inductance values
possible at a given operating frequency.
8.1.3
Modeling of Inductor Q and Self-Resonant Frequency
An accurate prediction of self-resonance and Qo requires modeling distributed turn-tosubstrate capacitances as well as substrate resistances between turns. These difficulties
have driven most designers to fabrication and measurement of prototypes, or to the use
of 3-D field simulations. An alternative approach involves simulating the detailed lumped
element model shown in Figure 8.3 using SPICE [128].
K1 n
K 12
C1
R1
L1
R sub12
C2
R2
K2 n
L2
R sub23
C3
Cn
Rn
Ln
Cn+1
R sub n n+1
Figure 8.3: Detailed circuit model of spiral inductor.
The accuracy of this approach rests on the ability to predict circuit element values from
spiral geometry and process parameters. For the design described in this paper, values for
self inductances of individual turns Li and coupling coefficients between turns Kij were
obtained using a finite element calculation of flux densities, while values for per-turn capacitance and resistance elements were computed directly from geometry and IC process data.
The resulting model was validated against data published in [131] and [122], and then run
to generate the data shown in Table 8.1, based on typical CMOS process parameters (rs
= 0.025 Ω/2, metal-to-substrate capacitance CA = 0.02 fF/µm2 , and substrate resistivity
rsub = 25 Ω/2 (doped active to maximize Q)). For all simulations, the trace width W was
197
set to W =
D
4N ,
giving a turn pitch of 2W , and skin effect and substrate eddy currents were
assumed negligible.
Table 8.1: Simulated inductance L, self-resonant frequency fR , and quality factor Q for
grounded spiral inductors.
D (µm)
330
330
330
1000
1000
1000
8.2
N
4
8
16
12
24
48
L (nH)
3.5
11
39
71
250
980
fR (MHz)
3900
2300
1200
300
160
65
Qo at fR
11
8.1
4.5
3.4
1.8
<1
Circuit Design Considerations
From Table 8.1, it is clear that achieveable Qo values are quite low, especially at frequencies in the VHF range. Therefore, if a high selectivity filter is desired, some form of Q
enhancement is needed. Two possible solutions to this problem are discussed in references
[57] and [93]. In [57], the effective Q of a lossy inductor is raised through the insertion of an
active negative resistance placed in series with the inductor. Difficulties with this approach
include the need to measure current within the inductor, and the potential degradation to
dynamic range that results from the insertion of current sensing resistors. Alternatively, a
series-to-parallel impedance transformation can be performed on the lossy inductor and a
negative resistance can be added, yielding the “parallel mode” Q-enhanced resonator shown
in Figure 8.4 [93], where:
Rp = (1 + Q2o )Rs
and:
198
(8.5)
Lp =
1 + Q2o
Q2o
!
L
(8.6)
In this approach, the negative resistance has been implemented as a transconductor with
positive feedback to cancel losses represented by Rp.
L
C
Lp
C
gm
Rp
Rs
Figure 8.4: Parallel mode Q-enhancement.
To form a complete Q-enhanced second-order filter, input and output transconductors may
be added to this circuit, as shown in Figure 8.5. In this configuration, the input transconductor provides a current source drive to the parallel RLC impedance, creating a second-order
bandpass response, while the output transconductor isolates the RLC impedance from the
load.
g mn
g mi
Vi
C
Lp
g mo
Io
Rp
Figure 8.5: Parallel mode Q-enhanced filter.
199
As shown in Chapter 5, the effective parallel resistance Req of the resonator, and the effective
quality factor Qenh of the resulting filter can be found from:
Req = Rp k
−1
1
=
Rp
gmn
1 − gmn Rp
(8.7)
and:
Qenh =
Req
1
=
Qo
XLp
1 − gmn Rp
(8.8)
and can be set to any desired value above Qo by controlling gmn using techniques such
as those described in Chapter 7. To proceed with the design beyond this point, however,
values for the circuit elements of Figure 8.5 must be chosen.
8.2.1
Component Value Selection
In general, the filter will be excited by a source with resistance Rs as shown in Figure 8.6,
and depending on the application, an input load resistor Ri may also be used.2
3
Thus, the
circuit parameters to be considered in the design process include Rs , Ri, gmi , C, Lp, Rp,
gmn , gmo , and RL .
Rs and RL are typically set by the system design in which the filter will be used. However,
the remaining values must be selected to satisfy various design constraints such as achieving:
• the desired operating frequency fo , and filter Q,
2
If the filter interfaces to external (off-chip) circuitry, Ri may be needed to provide good SWR performance. Otherwise, Ri will typically be omitted.
3
Real valued source and load impedances are assumed here for simplicity, but similar results follow for
complex impedances.
200
Rs
V
V
V
i
Filter
Ri
s
o
RL
Figure 8.6: Source and load resistances seen by filter.
• a desired value of gain,
• a desired noise figure, and
• the maximum possible dynamic range.
Considering the design of the RLC resonator, note that once the inductor is designed, Rp
is determined by Equation 8.5 and C and gmn can be picked to yield the desired frequency
and Q, respectively. However, the only constraint for designing the inductor is to maximize
Qo in order to maximize dynamic range. Since Qo is determined primarily by the inductor’s
trace resistivity and outer dimension, and is relatively independent of the number of turns
used, the actual inductance value must be chosen based on other considerations.
To constrain the inductance value, the impedance of the RLC resonator must be related to
the filter’s input and output transconductances, and the input, source, and load resistances.
These factors combine to determine the overall filter gain and noise figure, as well as the
extent to which dynamic range improves at large frequency offsets.
An interesting and useful case occurs if Rs and RL are known4 and gmi , gmn , and gmo are
4
These impedances are typically determined by the operating voltage and power level of the source and
load circuitry, and therefore set the overall system dynamic range.
201
chosen according to:
gmi = gmo = gmn ≈
1
RL
(8.9)
Under this condition, the inductive reactance XL and hence the inductance Lp are fixed by
the required condition between gmn and Rp , and can be found from:
XL =
Rp
1
=
gm
Qo
Qo n
(8.10)
where Qo is assumed known from the process parameters and outer dimension D of the
spiral used.
With these values selected, the voltage gain from the resonator core to the load is found to
be unity and the gain from the filter input voltage Vi to the resonator core voltage Vc is:
Vc
= gmi Req
Vi
(8.11)
which, when combined with (8.8) under the condition Qenh Qo , can be reduced to the
form:
Vc
Qenh
≈
Vi
Qo
(8.12)
Thus, the filter provides a substantial gain equal to the Q enhancement, and a reasonably
low noise figure may be expected.
An approximate expression for the filter’s noise figure when Ri → ∞ can be found by
writing the noise figure in terms of voltages as:
202
F
=
V 2 /V 2
Si /Ni
≡ s2 n2s
So /No
Vc /Vnc
=
1 Vn2c
A2v Vn2s
(8.13)
where Vns and Vnc are the noise voltages at the filter input and within the resonator core,
respectively, and Av is the voltage gain given by (8.11) (since Ri is assumed infinite).
Combining (8.13) with (8.11) and with the following approximations for the noise voltages:
Vns =
Vnc
p
4kT RsB
v
u
u
≈ t4kT
gmi + gmn
1
+
Rp
!
BR2eq + A2v Vn2s
(8.14)
yields:
F ≈1+
1
Rp + gmn
2 R
gm
s
i
gmi +
(8.15)
which, with the assumed relationship in (8.9), reduces to:
F ≈1+
3
gmi Rs
(8.16)
A similar relationship can be found under the matched input condition Ri = Rs , giving:
F ≈1+
12
gmi Rs
(8.17)
Both (8.16) and (8.17) indicate that the filter noise figure depends on the relationship
between the input transconductance and the source resistance, and can be minimized by
203
making the product gmi Rs large.5 Under the condition Rs = RL , (8.9) implies that gmi Rs ≈
1 and noise figures of 6 and 11 dB result for the two cases of Ri = ∞ and Ri = Rs ,
respectively. These values may be acceptable for filters intended for use in RF preselection,
but may actually be too low for IF filters, as discussed in Chapter 3. For IF designs, a larger
noise figure can be obtained either by lowering gmi , or by lowering Rs . If gmi is lowered,
the noise injected into the resonant circuit from the input transconductance will be less
and slight improvements (e.g. 1 - 2 dB) in in-band dynamic range will result. However, a
better choice is to lower Rs while maintaining the relationship in (8.9). This choice retains
the voltage gain in the filter, but effectively lowers the voltage produced by the source.
The advantage offered by this approach is that the ultimate out-of-band dynamic range
improvement described in Chapter 6 is increased since the input is generally voltage limited
and a lower input impedance implies a larger input power.
8.2.2
Balanced Circuits
Having selected values for the essential components in Figure 8.5, the task of circuit-level
design and layout can begin. A critical decision at this stage is whether to adopt singleended or differential circuit design techniques. For Q-enhanced filters which operate at high
frequencies, several considerations favor the use of differential design approaches, including:
• Negative resistances can be implemented by cross-coupling gate and drain circuits,
avoiding problems of intermediate nodes and associated parasitic poles.
• Coupling to power supply noise and to other circuits on the same die can be significantly reduced.
5
This product cannot be increased arbitrarily because of problems with gate-drain feedback capacitance.
At high frequencies, this feedback results in modifications of the expected input impedance and potential
stability problems if gmi Rs 1.
204
• Second-order non-linearities are canceled, leading to better dynamic range and noise
performance.
• Higher voltage swings can be supported.
A simple balanced circuit design for the second-order filter of Figure 8.5 was briefly presented in Chapter 2 and is repeated in Figure 8.7 for convenience. In this design, the
primary resonance is between the series connected inductors L1A and L1B , and capacitor
C1. However, to determine the value of C1 needed to achieve a given frequency, stray capacitances CSA and CSB must be taken into account. These capacitances represent both
the effects of self-resonance in the inductor and the parasitic capacitances associated with
all interconnects and devices attached to these nodes.
Vdd
L
L
1A
1B
C1
Out+
Out-
In+
In-
M1A
M1B
M2A M2B
C
SA
M3A M3B
C
SB
Ii
Iq
Io
Figure 8.7: Q-enhanced filter circuit design.
In an alternative design, C1 could be eliminated and capacitances of 2 C1 could be paralleled
with CSA and CSB to achieve the same frequency of operation. This technique has the
advantage that the capacitances are larger and can be more accurately realized at high
frequencies. In either case, however, two inductors are required for a fully balanced design,
adding significantly to the overall area consumption and the die size. This problem can
be solved if an approach similar to that used to build transmission-line transformers is
205
adopted, in which two sprials are inter-wound as illustrated in Figure 8.8, to form a single,
center-tapped spiral inductor.
Figure 8.8: Center-tapped spiral inductor geometry.
8.2.3
Center-Tapped Spiral Inductors
Copyright 1995 IEE. Reprinted with permission.
6
To compare the performance of the proposed center-tapped spiral geometry with that of
the traditional spiral shown previously in Figure 8.1, simulations were performed and then
checked with a board-level, scaled prototype. These investigations, described below, revealed that the center-tapped geometry consumes only 63% of the area required by two
traditional sprials, and delivers a common-mode rejection ratio approaching 40 dB. Moreover, the self-resonance frequency is twice that of a traditional grounded spiral of the same
value, enabling higher reactances and Q values to be obtained.
6
Portions of this subsection are extracted from the following paper by the author: W. B. Kuhn, A.
Elshabini-Riad, and F. W. Stephenson, “Centre-tapped Spiral Inductors for Monolithic Bandpass Filters,”
Electron. Lett., vol. 31, no. 8, pp. 625-626, 13 April 1995.
206
Performance Comparison
First, the resonance frequency and Q of a traditional spiral were investigated using the
circuit of Figure 8.3 with substrate losses excluded. A resonance curve simulated with
SPICE for a 100 nH, 28 turn inductor with outer dimension D = 450µm, trace width
w = 4µm, area capacitance C = 0.016f F/µm2, trace resistivity rs = 0.026Ω/2, and r = 4
is shown in curve a of Figure 8.9. This curve was generated with a current source applied
to one end of the spiral with the opposite end grounded - the appropriate condition for
circuits reported in the literature. If the source is applied across the inductor with neither
end grounded, the self-resonant frequency is found to be approximately doubled, contrary to
√
the 2 increase predicted by the simplified model assumed in [131]. At the grounded spiral’s
resonant frequency of 560 MHz, the Q calculated from the simulated peak impedance is 2.3,
in general agreement with the value predicted from (8.3).
A model for the center-tapped spiral of Figure 8.8 was then derived from the model of
Figure 8.3 by modifying the connections between turns, and grounding the center tap node.
Since the connections between turns are the only substantial change, the total inductance,
the total series resistance, and the Q of the center-tapped spiral can be predicted to be
approximately equal to that of a traditional spiral with the same total number of turns.
To determine the self-resonant frequency, a SPICE simulation was performed for a centertapped inductor with the same dimensions and total number of turns as the traditional spiral
above. The results, shown in curve b of Figure 8.9 indicate a factor of two improvement
over the traditional grounded spiral. Hence, the center-tapped spiral behaves essentially as
a floating inductor with respect to resonance. To determine the balance of the center-tapped
spiral, additional simulations were performed with the two ends of the inductor driven inphase. The result shown in curve c of Figure 8.9, indicates a common mode rejection ratio
(CMRR) on the order of 40 dB.
207
Figure 8.9: Center-tapped versus traditional spiral performance. (Simulated inductor voltage produced by unity amplitude current source inputs for a) grounded traditional spiral,
b) center-tapped spiral, and c) center-tapped spiral driven with in-phase currents.)
208
Size Comparison
Representative circuits employing the traditional spiral and the center-tapped spiral geometries are shown in Figure 8.10. For equal load impedance to be presented to the differential
amplifier transistors, the circuit of Figure 8.10a requires two inductors, each with half the
inductance of the single center-tapped inductor in the circuit of Figure 8.10b. To achieve
the same electrical performance from the two circuits, however, one must also guarantee
that the Q value of the inductors in the circuits are equal. Evaluations of (8.3) shows
that Q scales roughly linearly with D, and for any given D, is comparatively independent
of L and N (assuming w scales proportional to D and inversely proportional to N). Each
of the two inductors employed in Figure 8.10a must therefore be approximately equal in
size to the single inductor in Figure 8.10b. An exact analysis based on (8.3) shows that
the center-tapped spiral based circuit requires 63% of the chip area used by the traditional
spiral based design.
Vdd
Vdd
L1
L
C
L
1A
1B
C
C
SA
C
SA
C1
In+
a
M1A
SB
SB
C1
M1B
In-
In+
b
I bias
M1A
M1B
In-
I bias
Figure 8.10: Representative balanced filter circuit designs using a) two traditional spirals,
and b) a single center-tapped spiral.
209
Experimental Verification
To validate the simulation results, dimensionally scaled prototypes of center-tapped and
traditional spiral inductors were constructed. The prototypes were built using a scale factor
of 400 with 1/32” thick double-sided epoxy glass printed circuit board (r = 4.7). Trace
widths were scaled by the ratio of the board thickness to the thickness of a typical field
oxide layer between substrate and top-level metal (2 µm), resulting in w = 62.5 mil and
D = 7.0”. To approximate the ground plane behavior of the substrate in a typical silicon
IC process, 8 radial, 62.5 mil wide cuts were made in the ground plane, extending from
the spiral center to 0.25” beyond the outer trace, with one cut extending completely to
the board edge. These cuts provide low resistance between inter-turn capacitances, but
high resistance to major eddy current loops – properties characteristic of spiral inductors
constructed over a moderate resistivity substrate.
Simulation predictions and measured results for inductance L and self-resonant frequency fR
are compared in Table 8.2. As expected, the overall inductance values for the two coils are
essentially equal and the self-resonant frequency of the center-tapped spiral is approximately
twice that of the grounded traditional spiral. To validate the coil balance simulation, the
center-tapped spiral was used as the inductive load in a bipolar transistor amplifier circuit
similar to the FET circuit of Figure 8.10b. A CMRR of 36 dB was measured at 1 MHz,
and 37 dB at the in-circuit resonance frequency of 2.14 MHz.
Table 8.2: Comparison of traditional and center-tapped spiral performance.
Traditional spiral (grounded)
Traditional spiral (floating)
Center-tapped spiral
Simulated
L(µH) fR (M Hz)
47.1
1.33
47.1
2.54
47.1
2.69
210
Measured
L(µH) fR (M Hz)
47.9
1.10
46.4
2.42
46.5
2.38
8.3
Filter Tuning
For high-Q filter implementations (Q ≥ 50), fractional bandwidths will be small compared
with the deviations in center frequency expected due to fabrication tolerances. In addition,
the Q enhancement required will be large, leading to problems in obtaining accurate Q
values. These issues, together with circuits that can be used to vary the filter’s frequency
and Q are considered in the following paragraphs.
8.3.1
Frequency Tolerance and Temperature Coefficient
The center frequency fo of a Q-enhanced filter is a function of the spiral inductance and
of the capacitance of inductor turns and connected circuitry. The inductance value is
primarily controlled by the spiral’s dimensions and is therefore relatively unaffected by
process variations. Capacitances, however, may vary considerably. For example, field oxide
thicknesses, and therefore inductor turn-to-substrate capacitances may vary by ±10% or
more. Since the sensitivity of fo to C is 0.5, a ±10% capacitance tolerance implies a
manufacturing tolerance for frequency of ±5%. Recalling that the bandwidth B of a secondorder response is given by B =
fo
Q,
the maximum filter Q possible without an initial trim is
therefore found to be on the order of 10 or less, depending on the filter precision required.
The temperature coefficient of fo depends on variations in oxide and circuit capacitances.
For oxide capacitances, the temperature coefficient of capacitance can be as low as ±100
ppm/o C [97]. If capacitance variations of connected circuitry can be neglected, this results
in a 0.35% frequency shift for a temperature change of 0 − 70oC, implying a maximum
filter pole Q on the order of 100 without real-time tuning. In practice, however, this level
of stability may be difficult to achieve. For example, in a parallel mode Q enhancement
design, the effective inductance Lp is a function of Qo , making frequency also dependent
on trace series resistance tolerances and temperature coefficients. The sensitivity of fo to
211
Rs can be found using the well known expression for fo in conjunction with equation (8.6),
yielding:
fo
SR
=−
s
1
1 + Q2o
(8.18)
For AlSiCu metallization, the temperature coefficient of resistivity, and hence of Rs is on
the order of 4000 ppm/o C [185]. Thus, for the case of a relatively low Q inductor with
Qo = 2.5, the expected frequency drift due to Rs alone is -550 ppm/o C.
8.3.2
Q Tolerance and Temperature Coefficient
Effects of manufacturing tolerance and temperature coefficients on Qenh are even more
severe than those for center frequency fo , especially for large Q enhancements ( QQenh
1).
o
To determine these effects, variations in both the negative resistance circuit’s gm value and
the spiral’s series resistance must be considered. Using Equations (8.5) and (8.8) to solve
for the sensitivities of Qenh to gm and Rs for a parallel mode filter yields:
SgQmenh =
Qenh
−1
Qo
Qenh
SR
≈−
s
Qenh
Qo
(8.19)
(8.20)
Manufacturing tolerances on gm are functions of both FET channel mobility µ and threshold
voltage VT , with values of ±15% and ±0.2V being typical [180]. Manufacturing tolerances
on Rs can be even broader since metallization thicknesses are not tightly controlled in a
typical CMOS process. Deviations as high as ±25% are possible. Thus, even for low Q
enhancements, at least a one-time trim will be required.
212
Temperature coefficients for gm and Rs are also significant. As stated, the temperature
coefficient of Rs is on the order of +4000 ppm/o C. For gm , the temperature coefficient is
determined by variations in both µ and VT . Assuming operation at low gate-source voltage
Vgs so that:
gm ≈ µCox
W
(Vgs − VT )
L
(8.21)
one can find a first-order estimate of these effects. Sze [180] shows data indicating coefficients
for µ and VT on the order of -5000 ppm/o C and -3 mV/o C, respectively. Since the effects
of µ and VT on gm are opposite in sign, some first order temperature compensation of the
combined effects may be possible by careful design. However, even if a reduction to ±1000
ppm/o C is achieved, and the Q enhancement is limited to 10, variations in Qenh of up to
60% can result for a 70o C temperature change, and some form of real time tuning becomes
necessary.
8.3.3
Frequency and Q Adjustment
Both one-time tuning and real-time tuning require mechanisms for adjusting frequency and
Q. For Q adjustment, the negative resistance circuit can be implemented with tunable
transconductors. A popular architecture for balanced circuits is a differential pair such as
shown in Figure 5.5, whose gain can be varied through bias current adjustments. This circuit
has the advantages of very high frequency operation and provision for positive feedback
by cross-coupling transistor gates and drains. Other tunable transconductor circuits can
be borrowed from the design of Gm-C filters to the extent that they function at high
frequencies. For example, linearized differential pairs have been proposed in the literature
[68]. However, since increases in supported signal voltages can be offset by increased noise
injected into the LC circuit, limiting potential dynamic range improvements and degrading
213
the filter’s noise figure, the excess noise behavior of these techniques should be carefully
assessed.
For frequency adjustment, variable capacitances can be provided either in the form of
varactor diodes (if supported in the process), or by switching capacitances in and out of the
circuit [70]. In either case, the Q of the tuning capacitances must be large enough to avoid
significant degradation of the base LC circuit Q. An interesting alternative proposed in [77]
is to vary the effective inductance value through tunable reactance multiplication circuits. If
this approach is adopted, care must be taken to avoid degradation of the circuit’s dynamic
range due to the additional active circuitry. An alternative approach which obtains similar
results, but does not require the insertion of a resistor to detect inductor current is suggested
in Figure 8.11. Here, a small valued inductor is simulated with a gyrator, and paralleled
with the RLC resonator to provide fine adjustment of frequency.
VC
vi
gmn
g mi
Rp
C
L
Freq
Q
Figure 8.11: Frequency fine tuning using capacitively loaded gyrator.
An analysis of the noise behavior of these two approaches using the techniques of Chapter 6
suggests that significant dynamic range degradation occurs if the tuning range is on the order
of
1
Qo
or higher. Thus, for Qo ≥ 5, it may not be possible to implement a sufficient tuning
range with these techniques alone. In this case, some combination of switchable capacitances
to achieve coarse control (e.g. to within 1% with 3 to 4 bits of digital adjustment), combined
with fine tuning over a ±1% range could be used.
214
8.4
Higher-Order Filter Design
In the previous sections, a single resonator was used within the filter design, providing only
a single-pole response. To obtain higher-order filter designs, such filters could be cascaded,
or a coupled-resonator design similar to that considered in Chapter 5 could be adopted. The
coupled-resonator option has the advantage that identical resonator sections can be used,
simplifying the tuning process. In the following discussion, considerations for the design
of coupled-resonator higher-order Q-enhanced LC filters are presented using a fourth-order
design as an example. The approaches discussed can be extended to higher-order filters
provided that magnetic coupling from non-adjacent resonators can be suitably controlled.
A fourth-order coupled-resonator design is shown in Figure 8.12 in single-ended form. For
differential circuits, the same considerations apply, with the components shown representing
one half-circuit of the differential design. In this simplified equivalent circuit, gmi and gmo
provide input and output buffering to the two resonant circuits which are assumed identical
and magnetically coupled. Resistors R1 and R2 represent the combination of equivalent
inductor parallel loss resistance and the negative resistance used for Q enhancement.
V1
Vi
C1
I1
R1
k
L2
L1
I2
R2
V2
Vo
C2
g mi
g mo
Figure 8.12: Coupled resonator filter architecture.
The response of this circuit can be derived in several ways. The approach taken below is
intended to provide some insight into the coupling coefficient’s effects on transfer function
215
formation. This approach can also help in analyzing effects of oscillator signal feedthrough
due to magnetic coupling when one of the tuned circuits is used as the master in a master/slave tuning architecture.
The voltage V2 in Figure 8.12 can be written in terms of currents I1 and I2 in inductors L1
and L2 as:
V2 = sL2 I2 + sM I1
(8.22)
√
where M is related to the magnetic coupling coefficient k by M = k L1L2 . If the two
resonant circuits are assumed identical, one may set L1 = L2 = L to obtain:
M = kL
(8.23)
From (8.22), the parallel resonant circuit composed of L2 , C2 , and R2 can be modeled by
the equivalent circuit shown in Figure 8.13a, where Vk is found from Equations (8.22) and
(8.23) to be given by:
Vk = skLI1
(8.24)
Vk = kV1
(8.25)
or, since I1 = V1/sL:
Thus, from Figure 8.13a, V2 is seen to be k times V1, magnified by the effects of series
resonance (since R2 XC2 ). At the resonant frequency:
216
L2
Vk
V2
V2
I2
R2
Vk
s L2
C2
+
_
(a)
R2
L2
C2
(b)
Figure 8.13: Circuit models of magnetic coupling.
|V2| ≈ kQ|V1|
and if k is on the order of
1
Q
(8.26)
or above, essentially full coupling results.
To find an exact expression for V1 and V2 in Figure 8.12, the Norton equivalent circuit in
Figure 8.13b may be used to write:
V2 =
kV1
Z(s)
sL2
(8.27)
where Z(s) is the impedance of the parallel resonant LCR circuit. Similarly, the expression
for V1 in Figure 8.12 may be written as:
V1 = gmi Vi Z(s) +
kV2
Z(s)
sL1
(8.28)
Solving Equations (8.27) and (8.28) for the transfer functions from Vi to V1 and V2 with
L1 = L2 = L, yields:
217


V1
1


= gmi Z(s) 
2 
Vi
k
1 − sL
Z(s)
(8.29)
V2
k
V1
=
Z(s)
Vi
sL
Vi
(8.30)
Thus, the response from Vi to V1 is second-order bandpass, modified by an additional term
that depends on the coupling coefficient and the Q value. The response from Vi to V2
is then found by cascading that to V1 with an additional second-order section, yielding a
fourth-order response. These transfer functions are plotted in Figure 8.14 for the cases of
k<
1
Q
(under coupling), k =
1
Q
(critical coupling), and k >
1
Q
(over coupling), a normalized
frequency of 1 Hz, and a Q of 100.
40
30
| V2 / Vi | (dB)
| V1 / Vi | (dB)
40
20
k = 0.02
k = 0.01
k = 0.005
10
0
k = 0.02
k = 0.01
k = 0.005
30
20
10
0
0.95
1
1.05
0.95
Frequency
1
1.05
Frequency
Figure 8.14: Calculated transfer functions for coupled resonators with Q = 100, and fo = 1.
In practice, Q is determined by the desired filter transfer function, while the coupling
coefficient k is a function primarily of inductor placement on the die. To determine expected
coupling values, simulations were performed using a finite element model of flux linkages
for two identical spirals in free space. Results are shown in Figure 8.15 for the cases of
218
lateral and diagonal displacements. These results were found to be relatively independent
of absolute inductor size or number of turns.
0.06
S
0.05
D
Lateral
Separation
|K|
0.04
S
0.03
Diagonal
Separation
0.02
0.01
0
0.01 0.03 0.1 0.33
1
3
S/D
Figure 8.15: Simulated coupling coefficients versus inductor separations.
If the available chip area is insufficient to provide adequate separation for a desired k, some
method of coupling neutralization will be required. One solution to this problem is suggested
in Figure 8.13. Since the Norton current source value is
kV1
sL ,
the effect of coupling is to
inject a current proportional to the integral of the voltage present in the opposite resonant
circuit. Hence, the coupling can be reduced by injecting a current of the opposite sign using
an integrating transconductance.
Measurements conducted on the prototype filter described in the following chapter together
with subsequent simulations indicate that when Qo is low (e.g. < 10), the use of a pure
integration results in an asymetric response characteristic. For the case of low Qo , the solution to this problem is to employ a lossy integrator providing both in-phase and quadrature
neutralization. This can be observed by repeating the above analysis with a series RL circuit in place of the ideal inductors, and using this RL model to find the value of the Norton
source.
219
Chapter 9
Q-Enhanced LC Filter Implementations
This chapter describes two Q-enhanced LC filters designed and fabricated during the course
of this research.1 The main purpose of these circuits was to demonstrate the overall feasibility of Q-enhanced filter technology and to validate theoretical performance predictions
of earlier chapters. However, each filter was also designed with an application in mind, and
with additional development, could potentially be used within commercial products.
The first design, described in Section 9.1, is a simple 100 MHz second-order filter with offchip inductors, developed early in the research. The purpose of this design was to prototype
essential components including negative resistance transconductors, and to validate performance predictions including dynamic range, noise figure, and short-term tuning stability.
Since the inductors in this filter were off-chip, relatively high Qo values could be realized
(e.g. Q ≈ 100), and ultra-high Q values could be achieved through Q enhancement (e.g.
Q = 10, 000). Thus, the application envisioned is a narrowband VHF receiver front-end
approximating the ideal receiver described in Chapter 4.
The second design, described in Section 9.2, is a more elaborate fourth-order filter with
fully integrated inductors. This filter was designed to validate several issues including the
performance of the newly developed center-tapped spiral inductor, the use of magnetic
coupling and coupling compensation circuits for realizing higher-order filter designs, and
1
Portions of this chapter have been submitted to the IEEE for possible publication. Copyright may be
transferred from the author to the IEEE without notice.
220
the potential viability of KIS tuning control systems such as those proposed in Chapter 7.
With an operating frequency of 200 MHz and a nominal Q of 100, this design was targeted
at use within the IF section of a modern cellular or PCS radio receiver.
Both filters were fabricated through the MOSIS service in a 2 µm, 2-poly, 2-metal, N-well
BiCMOS process, and were packaged in standard 0.6” wide, 40 pin, ceramic DIP carriers.
Hence, the frequency of both designs was confined to the VHF-UHF range. Based on the
performance observed with this technology, scaling of the techniques demonstrated here to
finer-line processes coupled with the use of modern surface-mount RF packaging should
allow similar filters to be constructed at operating frequencies of 900 and 1800 MHz.
9.1
Ultra-High Q VHF Receiver Front-End
A block diagram of circuits implemented on this IC is shown in Figure 9.1. The design
includes a differential transconductor at the input, driving an off-chip LC tank circuit
connected to pins P1 and P2. Negative resistance circuits on the IC provide Q enhancement
to raise the effective Q of the tank circuit to values determined by the control input Vq . A
mixer then buffers the signal at pins P1 and P2 and converts this signal to an IF frequency
for further processing.
Vf
Off-Chip
Tank Circuit
P1
RFin
RFin
Vbias
Vdd
Differential
Amplifier
P2
Mixer
IFout
IFout
Negative
Resistance
Vq
LO
Rbias
Figure 9.1: Block diagram of ultra-high Q VHF receiver front-end IC.
221
Use of a mixer, rather than a simple transconductor at the filter output allows the design
to be used as a receiver front-end as shown in Figure 9.2. Moreover, operation of the filter
at ultra-high Q values (e.g. 10,000) allows the preselection bandwidth to be narrowed to
that of the signal being received (assumed to be 10 KHz in the application shown), thereby
approximating the design of the ideal receiver of Chapter 4.
Front-End IC
VHF
BPF
Freq, Q
1 MHz
IF
BPF
To
Demod
LO
Figure 9.2: VHF receiver architecture block diagram.
In the envisioned application shown in Figure 9.2, VHF signals from the antenna are delivered directly to the filter, which is tuned in conjunction with the LO signal using the
receiver’s control microprocessor to select different RF channels within the service band.
Since the filter operates at a small fractional bandwidth, a simple second-order filter topology can provide good image rejection, allowing the mixer to convert VHF signals to a low IF
in a single step. Final channel selection can then be implemented at this IF using conventional fixed-tuned switched-capacitor or gm-C filter designs. Dynamic range requirements
on the mixer and subsequent circuits are reduced significantly by the narrow preselection,
decreasing the probability of producing spurious (intermod) responses and thereby allowing
these circuits to be implemented at very low power.
9.1.1
Design and Layout
Circuits used to implement the input transconductor, negative resistance, and mixer blocks
of Figure 9.2 are shown in Figures 9.3, and 9.4.
222
Figure 9.3: Input transconductor and negative resistance designs.
223
VCQ
VBIAS
IN-
IN+
M4
3/100
VDD/2
4p
C1
P2
M1
9/2
M3
48/2
M2
9/2
P1
M5
3/100
VDD/2
M8
3/20
M7
24/2
M6
12/2
4p
C2
M9
48/2
VDD
M10
12/2
4p
C3
M12
3/20
M11
24/2
VDD/2
Figure 9.4: Mixer circuit design.
224
MBIAS2
LO2
VDD/2
M14
3/100
M13
8/4
M15
24/2
P1
M17
3/100
VDD/2
M16
15/2
Q1
1x1
M18
15/2
4p
C4
M19
48/2
Q2
1x1
M20
15/2
M21
15/2
4p
C5
P2
M23
3/100
M22
8/4
VDD/2
IF2
IF1
A principle goal in the design of these circuits was to provide good dynamic range performance at very low power. Thus, nominal operating currents for the three circuit blocks
were selected as 500 µA, 500 µA, and 1 mA, respectively, and signal processing circuits
were designed to support maximum possible voltage swings.
For example, for the differential transconductor of Figure 9.3, a bias level (ggs − vT ) of 1.5
volts was chosen to allow large out-of-band signals to be present without saturating the
input transconductor’s I/V characteristic. This choice fixed the transconductor β at:
β=
2ID
= 220µA/V 2
(vgs − vT )2
(9.1)
and the (half-circuit) transconductance at:
gmi = β(vgs − vT ) = 333µA/V
(9.2)
To provide both good dynamic range and good voltage gain, the condition gmn = gmi was
then adopted as discussed in Chapter 8 to establish the value of Rp at 3000 Ω, and hence,
the inductive reactance (assuming Qo = 100) as:
XL =
Rp
= 30Ω
Qo
(9.3)
Finally, to provide good noise figure performance, the condition gmi Rs ≈ 1 was adopted,
setting the source resistance at 3 KΩ for the half-circuit, or 6 KΩ for the differential input.
Although this is substantially higher than the 50 Ω terminal impedance of typical antennas and test equipment, a simple L-type matching network will transform between these
impedance levels with a Q of 11.
The nominal 500 µA bias current for the input transconductor is set by an off-chip control
225
voltage, providing some flexibility in testing. Similar externally-controlled biasing is used
in the negative resistance and mixer circuits, for similar reasons. Within the negative
resistance circuit of Figure 9.3, the bias voltage Vq establishes the tail current and thus the
transconductance of cross-coupled transconductor M6, M10, thereby controlling the filter
Q. The additional transistors M7, and M11 act as source followers to increase the allowable
voltage swing at pins P1, P2 to approximately 1 V RMS.
Finally, the mixer shown in Figure 9.4 is implemented as a Gilbert cell in which the lower
bipolar differential pair switches the bias current between the two upper MOS transconductor pairs on alternate half-cycles. This somewhat non-standard configuration was adopted
to allow the upper transconductors to be implemented with FETs, minimizing loading and
pulling of the resonant circuit Q and frequency. Bipolar transistors were then used below
to allow full switching to be obtained with minimum possible LO voltages.
Figure 9.5: VHF front-end IC photograph.
The layout for the circuit was constrained by the need to share space between these circuits
and an un-related digital design on the same die. Nevertheless, the relatively small amount
of circuitry described here was easily accommodated in the available area as shown in the
photograph of Figure 9.5. The only critical layout consideration was the placement of the
226
input, output, and tank circuit pins which were separated and paired with power and ground
pins to the extent possible to maximize signal isolation.
9.1.2
Experimental Results
The IC was tested using the circuits in Figures 9.6 and 9.7 to provide the required bias
voltages, tuning controls, and I/O matching networks. Test circuits were constructed on
double-sided, glass-epoxy PC board with one side serving as a ground plane. All filter
tuning was performed manually with potentiometer controls.
The tank circuit inductors were constructed as air-wound coils with a (calculated) value
of 30 nH. The remaining inductors used in the I/O matching networks were commercially
acquired, ferrite-core, adjustable units. The two output pins (IF+ and IF-) and their
associated matching networks were used to provide outputs at the RF and IF frequencies,
respectively. During the majority of tests, the mixer circuit was bypassed by grounding pin
29 through R10, reducing the mixer to a simple transconductor output stage and providing
a direct RF output. This allowed the filter to be tested independently of the mixer’s
performance.
Experimental results obtained are summarized in Figures 9.8 and 9.9, and Tables 9.1 – 9.3.
Figures 9.8 and 9.9 show the effects of the Q control input. The base Q of the external tank
circuit was found to be approximately 70 with the Q control voltage at zero volts, and a Q
of 10,000 was achieved at 1.73V, corresponding to approximately 650 µA of transconductor
current. The manual settability of the filter Q was found to be excellent, with Q values of up
to 100,000 possible with careful adjustment.2 In addition, no hysteresis effects were noted
when adjusting the Q control to enter and leave oscillation, and good orthogonality was
observed between frequency and Q control inputs – important considerations for automatic
2
The primary limitation here was the potentiometer itself, and the small rotation angles needed between
Q = 100,000 and oscillation.
227
Figure 9.6: Test circuits for VHF front-end IC.
228
J1
RFIN
J2
LO
VDD/2
47K
R1
2p
C1
0.01U
C3
L1
350n
0.01U
C2
R2
6.2K
R3
1.0K
VCF
29
28
27
26
11
7
R4
47K
LO2
LO1
REF-
REF+
RF-
RF+
12
VB
P1
33
VCQ
VCQ
14
P2
C5
VDD
R5
10K
VSEN DAT T2 RST CLK T1
C1
C2 VSS VSS
13
17
19
20
21
22
23
24
10
25
U1
VT100
0.01U
39
40 5
34 1
6
2
3
4
8
9
15
V/2 V/2 DP3 DP2 DP1 DP0 BSY RDY AP0 TST VDD VDD
VDD/2
RB1 RB2
16
18
32
MV2103
D1
D2
MV2103
10p
L3
30n
L2
30n
C4
VD
VSS
VSS
IF-
IF+
VDD
VDD
35
31
37
36
38
30
VDD
VDD
VDD
27p
C6
R7
6.2K
R6
6.2K
C7
0.01U
C9
L4
350n
0.01U
L5
8.2U
2p
C8
J4
IFOUT
J3
RFOUT
Figure 9.7: Test circuits for VHF front-end IC.
229
J6
Q
J5
F
VR2
10K
VR1
100K
R8
39K
CW
R9
27K
CCW
VDD
CW
CCW
VDD
0.01U
C11
0.01U
C10
+
5
+
6 -
3
2 -
J7
+15V
IC1
LM358
IC1
LM358
2
7
1
1N4148
D3
1
100
R14
100
R12
C13
VCQ
R13
10K
0.01U
TP5
0.01U
VCF
R11
10K
C12
TP4
470
R10
1.0
C14
VR3
1k
R15
1200
CW
CCW
OUT
COM
IC2
LM317L
IN
330
R16
VDD
VDD
1.0
C15
R20
100K
R19
33K
R18
100K
R17
100K
C17
0.01U
C18
0.01U
3
+
2 -
0.01U
C16
5
+
6 IC3
LM358
IC3
LM358
VDD
1
7
R21
TP2
100
100
R25
R22
10K
C20
0.01U
R24
10K
VD
VDD/2
TP3
R23
10K
0.01U
C19
TP6
control system design.
| S21 ! (dB)
30
Q = 10,000
Q = 3,000
Q = 1,000
Q = 300
20
10
0
Q = 100
-10
-20
101.5
101.7
101.9 102.1 102.3
Frequency (MHz)
102.5
Figure 9.8: Measured transfer functions of Q-enhanced filter.
100000
100000
1000
Q
Q
10000
10000
100
10
0.7 0.9 1.1 1.3 1.5 1.7 1.9
Control Voltage (V)
1000
1.7
1.71 1.72 1.73 1.74 1.75
Control Voltage (V)
Figure 9.9: Measured Q factor versus control voltage.
Short-term stability of both frequency and Q were found to be excellent as shown in Table
9.1, indicating that the filter is suitable for use with either the self-tuning or orthogonal
reference tuning techniques and their simplifications discussed in Chapter 7.
In-band dynamic range and noise figure goals for the design, derived from expressions
developed in Chapters 6 and 8 were 71 dB and 11 dB, respectively. These targets were
computed based on an assumed maximum capacitor voltage of 1 V RMS, and a matched
input condition. As shown in Table I, neither goal was met. The measured in-band dynamic
range of 56 dB is attributed to a substantially lower voltage within the tank circuit than
230
Table 9.1: Measured performance of VHF front-end IC.
Performance Measure
Minimum Q (Base Q of tuned circuit)
Maximum Q (Manually controlled)
Q stability
Frequency tuning range
Frequency Stability
Minimum discernable signal
Noise figure
Gain (mixer bypassed)
Gain (to IF)
Blocking dynamic range
In-band
1 MHz offset
Spurious free dynamic range
In-band
1 MHz offset
Current Consumption (VDD = 8V )
Filter
Mixer
231
Value
70
105
30% / hour
88 - 102 MHz
5 KHz / hour
-117 dBm
15 dB
31 dB
22 dB
56 dB
102 dB
42 dB
87 dB
1.3 mA
1.0 mA
that assumed (vmax ≈ 200mV versus 1 V). Subsequent simulations and study showed this
problem to be related to non-linearities in the negative resistance transconductor circuit –
a problem common in transconductors used in Gm-C filter designs as well [64]. At large Q
enhancements, a drop in transconductor gain as small as 1% results in substantial lowering
of the filter Q, and hence lowering of the filter gain (compression) and widening of the
bandwidth. This problem could be corrected to some extent in future implementations by
employing more linear transconductors within the negative resistance block. In conjunction
with this method, designing for lower impedance levels could provide somewhat higher
efficiencies.
The higher than expected noise figure was attributed to problems in the input matching
network implementation. PC board trace capacitances, combined with the matching network inductor’s self-resonance effects, raised the effective inductance value and required the
substitution of a lower-valued device. In turn, the loss resistance associated with this new
inductor resulted in a lower effective value of Rs seen by the input transconductor. This
problem could be corrected in future implementations by designing for lower impedance
levels throughout the circuits.
Table 9.2: Blocking dynamic range versus frequency offset.
∆f
B/2
Measured BDR (dB)
Calculated BDR (dB)
0
20
100
200
1000
56
81
96
102
111
–
82
96
102
116
Finally, Tables 9.2 and 9.3 compare the dynamic range performance with the predicted
behavior discussed in Chapters 3 and 6. The relationship between measured in-band dynamic range and spurious-free dynamic range agrees closely with that predicted by Equation
(3.24), while the improvement with frequency offset is in excellent agreement with (6.40)
232
Table 9.3: Spurious-free dynamic range versus frequency offset.
∆f
B/2
Measured SFDR (dB)
Calculated SFDR (dB)
0
2
4
20
100
200
42
49
55
70
83
87
–
50
56
70
84
90
and (6.41).3
Despite the IC’s somewhat limited in-band dynamic range performance, the ultimate dynamic range at large frequency offsets is excellent. BDR and SFDR measurements of 102
and 87 dB respectively were measured at only 1 MHz offset. Thus, the performance of the
circuit makes it an excellent candidate for use within receivers designs for VHF satellites
operating under the harsh interference conditions described in Chapter 3.
9.2
Cellular and PCS Receiver IF Filter
The second IC fabricated in the course of this research was a fourth-order (2-pole) coupledresonator design with on-chip inductors and partial on-chip frequency and Q tuning. The
nominal operating frequency and selectivity Q are 200 MHz and 100 respectively, and the
application envisioned for this design is a first IF filter in a receiver intended for use in new
PCS services such as DECT (see Chapters 4 and 10).
3
Values at the largest offsets diverge from those calculated due to expected saturation effects within the
input transconductor. These saturation effects are not accounted for by the equations, but can be predicted
as discussed in Chapter 6.
233
9.2.1
Design and Layout
The prototype filter was implemented using two identical copies of the circuitry shown in
the block diagram of Figure 9.10. With this architecture, the circuits can be operated in
a variety of modes. For example, with one section disabled, a single second-order filter
can be evaluated. Alternatively, with both sections enabled, a coupled resonator 4th order
response is obtained. Finally, a master/slave tuned second-order filter can be created by
using one section for filtering and configuring the other section as an oscillator.
Vdd
From
Other
Section
Input
Rectout
g
gm
g
m
Output
Obias
Ibias
Fcntl
m
3
Switch Array
1
s
Qcntl
Kcntl
To
Other
Section
Figure 9.10: Block diagram of IF filter second-order section.
As indicated in the block diagram, balanced circuitry was adopted throughout the design
to provide good power supply rejection and signal isolation, while simplifying the implementation of negative resistance circuitry. Circuitry used to implement the input buffer
amplifier, spiral inductor load, controlled negative resistance, and output buffer amplifier
are shown in Figure 9.11.
The on-chip, center-tapped inductor L1 consists of 26 turns with an outer dimension of 850
234
Figure 9.11: Core circuits of IF filter.
235
IBIAS
IN+
4/100
VDD
3/10
VDD
500f
72/2
COILVDD
72/2
144/2
250n
IN-
QCNTL
3/10
VDD
4/100
VDD
500f
144/2
288/2
144/2
OBIAS
4/100
VDD
500f
72/2
144/2
72/2
COIL-
COIL+
OUT-
OUT+
µm, yielding a series equivalent circuit inductance value of approximately 250 nH, and a
simulated self-resonant frequency of 270 MHz. Trace width and turn pitch are 12 µm and
16 µm, respectively, and nominal process parameters are: metal2 to substrate capacitance
= 0.016 fF/µm2 , metal resistivity = 0.03Ω/2, and substrate resistivity = 2500Ω/2. The
substrate resistivity is that of an N well region, which was used in place of a lower resistance
active region to achieve thicker oxide separation and a higher self-resonant frequency.4
At 200 MHz, the equivalent parallel inductance is found from Equation 8.6 to be 300 nH,
for an effective parallel reactance of 377 Ω, and a parallel loss resistance of 830 Ω.5 This
loss resistance is in parallel with the voltage-controlled negative resistance transconductor
consisting of M3A, M3B, and M4.
Despite the problem of limited voltage swings encountered in the first IC design, a simple
cross-coupled differential design was adopted for the negative resistance function. The
reasons for this decision were three-fold. First, by accepting a small voltage swing, the
source followers previously used to provide DC voltage offsets between gate and drain nodes
could be eliminated, allowing operation of the filter down to 3 V and below. Second,
difficulties in simulating designs with transconductors such as those described in [68] left
too many unknowns and the overall risk of adopting a new design approach was judged too
high. Finally, the 830 Ω impedance level established by the inductor design used, meant
that reasonable efficiency could be obtained (signal currents are a higher fraction of the
transconductor bias current than for the IC described in the previous section).
The negative resistance circuit was designed to provide the required gmn value of
1
Rp
=
4
Based on these values, the simulated quality factor Qo at self resonance, including effects of substrate
losses, but excluding connected circuitry is 3.0. When operating at the nominal frequency of 200 MHz, a
Qo value of 2.2 is therefore expected.
5
Inductor self-resonance and inductor Q have strong effects on filter operating frequency and Q, and must
be carefully considered. An initial design and fabrication of the chip had to be repeated because simplified
calculations were used. The second chip fabricated is described here and used more accurate simulations
including substrate losses, inter-turn sidewall and fringing capacitances, and turn-to-turn crossover trace
capacitances.
236
1.2mA/V with a current of 1 mA, implying the transistor W/L values shown in Figure
9.11. Similar, but slightly smaller transistor sizes were then used in the input and output
transconductors as shown. The selected values yield a nominal filter gain on the order of 0
dB with 50 Ω source and load impedances when operating at bias currents of approximately
500 µA each.
Frequency tuning is implemented with three copies of the capacitor switching circuit of
Figure 9.12a, providing 8 steps of approximately 1.2 MHz each. Four to five bits of control
would be provided in practice for more range and finer resolution, but was not implemented
in the prototype due to pinout limitations.
Each copy of the capacitor switching circuit consists of a pair of pulldown transistors M7A
and M7B which ground tuning capacitors C1A and C1B when on. Since these pulldown
transistors must be large to minimize Q degradation, drain capacitance is significant when
the transistors are off and must be taken into account. When M7A and M7B are off, the
additional circuitry shown biases the switching transistor drains to approximately 2V to
lower the junction capacitance and minimize non-linearities for large signal swings. Further
drain junction capacitance reduction is achieved by implementing M7A, M7B with arrays of
square transistors, each consisting of an annular gate surrounding a minimum size central
drain diffusion and contact, a geometry adopted throughout the filter design where possible
to limit the effect of parasitic capacitance on filter operation.
Coupling neutralization is performed by the integrating transconductor shown in Figure
9.12b. The RC filters provide approximately 75o of phase shift, with the remaining 15o
provided by source followers M8A, M8B and transconductor M9A, M9B.
Finally, an amplitude detection circuit is implemented as shown in Figure 9.13 with full
wave rectifier M11A - M12B, smoothing filter C2, and DC amplifier M13, M14. The rectifier
threshold is set by R1 in conjunction with resistor programmable current source M15, M16.
237
Figure 9.12: Switched tuning capacitors and coupling neutralization circuits.
238
4/2
3/100
VDD
3/100
4/2
VDD
FCNTL
4/2
3/30
500f
4/2
8/2
PADVDD
4/2
3/4
VDD
COIL-
COIL+
96/2
280f
96/2
280f
4/2
3/4
VDD
3/30
500f
4/2
24/2
4/6
VDD
COIL-
COIL+
KCNTL
VDD
10K
4/100
96/2
48/2
VDD
200f
500f
96/2
48/2
96/2
200f
10K
96/2
48/2
VDD
COIL2+
COIL2-
Figure 9.13: Amplitude detection circuit.
239
24/2
THRESH
5K
COIL-
24/2
VDD
COIL+
24/2
4/8
VDD
6/2
24/2
VDD
4/2
5p
4/2
24/2
3/100
6/2
VDD
4/2
4/8
VDD
RECTOUT
A photograph of the fabricated circuitry is shown in Figure 9.14. To provide minimum
possible inductor coupling between the two identical second-order sections implemented on
the die, the inductors were oriented diagonally opposite to each other. Chip area not used
by the circuits discussed above was used to provide on-chip decoupling and supply bypass
capacitors and to implement test structures. Total chip area for both second-order sections,
excluding pads, is 3.3 mm2 .
Figure 9.14: Photograph of chip layout.
9.2.2
Experimental Results
The IC was tested using the circuits shown in Figures 9.15 and 9.16 to provide I/O terminations and frequency and Q adjustments.
Figure 9.17 shows measured gain |S21| of a single second-order section at varying values
of Q enhancement with the opposite section disabled and powered down. The base Q of
240
Figure 9.15: Test circuits for IF filter.
241
J1
INA
R1
51
0.01U
C1
0.01U
C3
OBIASB
IBIASB
RECTBIAS
0.01U
C2
OBIASA
IBIASA
N/C
27
35
34
33
32
1
31
39
7
15
14
13
12
21
11
19
VQ2B
GNDB
INB
INB
GNDB
F1A
F1A
OBIASB
IBIASB
8
RECTBIASB
VQ2A
GNDA
INA
INA
GNDA
OBIASA
IBIASA
F2A
F2A
9
20
10
F3A
F3A
F1B
28
VQB
F1B
U1
VT200
VCOUPLA
17
RECTBIASA
VQA
VCOUPL
VQA
37
F2B
29
F2B
VQB
40
30
F3B
F3B
RECTOUTB
VDDPADB
OUTB
OUTB
VDDPADB
VDDCOILB
VDDB
RECTOUTA
VDDPADA
OUTA
OUTA
VDDPADA
VDDCOILA
VDDA
VCOUPLB
38
5
4
3
2
26
36
18
25
24
23
22
6
16
VDDB
N/C
VDDA
0.01U
C5
RECTOUT
R3
33
R2
33
C7
0.01U
R5
100
R4
100
C6
0.01U
C4
0.01U
C8
0.01U
C13
0.01U
C11
0.01U
C10
0.01U
C9
0.01U
C12
0.01U
J3
OUTB
J2
OUTA
Figure 9.16: Test circuits for IF filter.
242
J4
VDD
VR2
10K
CCW
CW
R7
10K
VR1
10K
VR4
1M
CCW
CW
R6
10K
CCW
CW
VQ
VR3
1K
VBIAS
RECTOUT
CCW
CW
R8
1K
1
2
3
4
5
6
7
8
SW1
16
15
14
13
12
11
10
9
R9
470K
R10
470K
R11
470K
OBIASA
VDDA
F1A
F2A
F3A
IBIASA
VQA
RECTBIAS
VCOUPLE
1
2
3
4
5
6
7
8
SW2
16
15
14
13
12
11
10
9
R12
470K
R13
470K
R14
470K
OBIASB
VDDB
F1B
F2B
F3B
IBIASB
VQB
the resonant circuit was estimated from the lowest curve as 2.3, in general agreement with
results predicted through simulation. For minimum and maximum Q conditions, current
consumptions were measured at 2.54 and 3.88 mA, respectively, with a 5V supply, and 1.81
and 2.94 mA, respectively, with a 3V supply.
| S21 | (dB)
0
Q = 100
Q = 30
Q = 10
Q= 3
Q = 2.3
-10
-20
-30
-40
-50
100
150
200
250
300
Frequency (MHz)
Figure 9.17: Measured |S21 | of second-order section at different Q enhancements.
Measured blocking dynamic range (BDR) and spurious free dynamic range (SFDR) are
plotted in Figure 9.18 with the second-order section set to a Q of 100 (3 dB bandwidth
of 2 MHz). The dynamic range figures shown are in general agreement with predictions
and are respectable for filters of this bandwidth. Overall receiver dynamic range would be
higher if the received signal were subsequently downconverted to a lower IF with a narrower
bandwidth prior to demodulation. For example, the increase in BDR would be 18 dB for a
receiver with a 30 KHz signal bandwidth.
A significant limitation on dynamic range was found to be gain compression in the resonant
circuits, and associated broadening of the filter bandwidth. These factors were expected
based on the results of the first chip design effort. The 1 dB output compression point was
found to be -37 dBm, approximately 10 dB lower than the maximum signal output power.
243
This behavior can be traced to non-linearities in the negative resistance circuitry, indicating
that dynamic range performance may be improved through better transconductor design
in this area. Additional improvements in dynamic range are possible if a lower resistivity
metallization can be used in the spiral inductor implementation, as seen through Equation
Dynamic Range (dB)
(8.3) in Chapter 8.
90
BDR
70
SFDR
50
30
0
1
2
4
8
16
32
64
128
Frequency Offset [(f - fo)/B]
Figure 9.18: Measured blocking and spurious-free dynamic range.
Temperature coefficients of fo and Qo were found to be -735 ppm/o C and -15%/o C, respectively, in agreement with predictions in Chapter 8. To investigate the possibility of
applying master-slave tuning to compensate for the relatively large Q temperature drift,
the opposite second-order section was powered up to serve as a master oscillator and its
amplitude detector output was connected to the Q control input of both sections. To provide a lower finite Q in the slave section and to reduce oscillator feedthrough, the master
oscillator was operated at maximum frequency, while the slave filter was operated at minimum frequency. Filter response curves measured at 20 to 50o C are plotted in Figure 9.19,
where the oscillator feedthrough is clearly visible.
Frequency tuning was not included in these tests, but could be provided by a suitable
244
|S21| (dB)
0
50 C
40 C
30 C
20 C
-10
-20
-30
185
190
195
200
205
Frequency (MHz)
Figure 9.19: Master/slave Q tuning from 20oC to 50oC.
control arrangement designed to monitor and adjust the master oscillator’s frequency. In a
typical application, this function could be implemented in software through a host receiver’s
microprocessor and associated counter/timer, thereby limiting additional on-chip circuitry
to a simple frequency prescaler as described in Chapter 7. Q stability and frequency tracking
between master and slave seen in Figure 9.19 suggest that Q values up to 100 or higher
could be supported with this approach.
Oscillator output power was adjusted to -32 dBm for these tests by resistor programming of
the amplitude detector. At this level, the oscillator signal feedthrough in the filter output
was -47 dBm, which is relatively large compared with the -37 dBm output compression
point. Single-sided phase noise of the master oscillator in a 1 Hz bandwidth was measured
as -84 dBc at 10 KHz offset and -105 dBc at 100 KHz offset. These are acceptable levels to
allow the master oscillator to double as a local oscillator for downconversion to a subsequent
IF, as will be shown in Chapter 10.
Finally, the chip was configured as a fourth-order filter by applying equal frequency and
Q control signals to both sections using manual adjustments. To maintain Q balance, the
input and output buffers of both sections were enabled. The resulting filter response is
245
0
0
-10
-10
| S21 | (dB)
| S21 | (dB)
shown in Figure 9.20a.
-20
-30
-40
-50
188
-20
-30
-40
193
198
203
208
Frequency (MHz)
-50
185
190
195
200
205
Frequency (MHz)
(a)
(b)
Figure 9.20: Measured fourth-order response. a) Without neutralization. b) With neutralization.
The assymetry seen in the passband is due to the presence of in-phase as well as quadrature coupling between the two coupled resonators, as discussed in Section 9.2.1. Since no
in-phase neutralization circuitry was included in the prototype filter, an alternative technique consisting of offsetting the frequency of the second resonator by 2.5 MHz was used to
equalize the passband ripple. The coupling adjustment control was then applied to achieve
the response shown in Figure 9.20b. Future designs will incorporate both in-phase and
quadrature neutralization to provide a more ideal response characteristic. Simulations indicate that demands on stability of the coupling neutralization circuitry are not excessive.
Variations as high as 10% result in less than 1 dB of passband assymetry if a Qo of 5 can
be achieved.
Tuning of frequency and Q for the fourth-order filter was not implemented in the prototype,
but could be provided using either a Master/Slave approach, or the Self-Tuning approach
reviewed earlier in Chapter 7. Measurements made on the prototype filter indicate that the
short term stability of both fo and Q are sufficient to support the latter technique. Short
246
Table 9.4: Measured performance of IF filter second-order section.
Performance Measure
Frequency tuning range
Q tuning range
Nominal 3 dB bandwidth
Nominal gain (50Ω I/O)
Ultimate rejection
Blocking dynamic range
In-band
4 MHz offset (alternate channel)
32 MHz offset
Spurious free dynamic range
In-band
4 MHz offset (alternate channel)
32 MHz offset
Temperature coefficients (open loop)
Frequency
Q
Supply voltage
Supply current (3 V)
Chip area (excluding pads)
Value
194 - 203 MHz
2.3 - ∞
2 MHz
-5 dB
40 dB
47 dB
54 dB
72 dB
37 dB
49 dB
63 dB
-735 ppm/o C
-15%/o C
3-5V
2.94 mA
1.7 mm2
term frequency drift was measured as less than 10 KHz in 5 seconds and less than 50 KHz
in 5 minutes at room temperature. Short term drift for Q was negligible over periods of
several minutes at Q values as high as 200.
247
Chapter 10
Integrated Receiver Design
In this final chapter, the issue of integrating local oscillators is briefly considered, completing
the analysis of receiver components which have so far resisted integration in commercial
products. System level design considerations and circuit-level design techniques developed
throughout this dissertation are then combined in developing a hypothetical, fully integrated
receiver. The intent of this receiver design exercise is not to embark on a full-scale receiver
development, but rather to show how the complete theory developed thus far can be used
in future commercial products. Thus, only a top-level design is presented, and circuit level
implementation is left to future work.
10.1
Integration of Local Oscillators
In addition to bandpass filters, a fully integrated receiver will typically need to include integrated local oscillators. Oscillators, like filters, can be implmented in integrated form using
either Gm-C, or LC based active circuits. Indeed, such oscillators are routinely used within
integrated filters to serve in master-slave tuning. However, before using these techniques to
implement receiver LOs, the issue of oscillator spectral purity must be addressed.
In the first subsection below, the phase noise performance of LOs will be examined and phase
noise estimates will be found through simple extensions of the dynamic range expressions
for filters developed in Chapter 6. Following this derivation, the effects of phase noise on
248
receiver performance are considered, and expressions for determining required phase noise
performance are developed.
10.1.1
Phase Noise in Oscillators
An oscillator may be created from a simple second-order resonator (filter) by letting Q → ∞,
and most practical oscillator designs can be reduced to this form. A critical study of
oscillator operation shows that the actual Q of the oscillator is large, but finite. Thus, the
spectrum of the oscillator is that of filtered noise, rather than the single-frequency spectral
line normally assumed in a simplified analysis.
A study of oscillator noise based on these concepts was carried out as early as 1964 by
Robins [147] who showed that the ratio of phase noise in a 1 Hz bandwidth Nop to oscillator
power Posc may be written in the form:
Nop (∆f )
F kT 1
=
Posc
Posc 8Q2
fo
∆f
2
(10.1)
where ∆f is the frequency offset from the oscillator center frequency fo at which the noise
is measured, F is a noise figure for the active circuits used, Q is the quality factor of the
resonator on which the oscillator is based, and k and T have their usual meanings. This
expression is approximately valid for ∆f greater than some lower limit in the range of 10
Hz to 100 KHz, depending on the actual oscillator construction. At frequencies below this
limit, the noise spectrum varies at rates of (∆f )−3 or higher due to 1/f noise in the active
components used in the realization [147].
Although Robin’s derivation is straightforward and could be reviewed here in more depth
to illustrate the factors and assumptions involved, similar results can be obtained with less
effort by leveraging the dynamic range expressions for Gm-C and Q-enhanced LC filters
249
developed in Chapter 6. This analysis, detailed below, provides expressions directly relevant
to the cases of oscillators constructed using Gm-C and LC based on-chip resonators.
Phase Noise in Gm-C and LC Oscillators
Consider the dynamic range of a Gm-C, second-order resonator derived in Chapter 6 and
repeated in Equation (10.2) below for convenience:
DR ≡
2
Vmax
ηPDC
=
2
4πkT BF BQ2
Vnoise
(10.2)
This resonator will become an oscillator when Q is raised to the point where Vnoise = Vmax,
or equivalently, when DR = 1. Substituting this value for DR and the expression for Q in
terms of center frequency and bandwidth into (10.2), the 3 dB bandwidth of the oscillator’s
noise spectrum can be found from:
BGm−C =
4πkT F fo2
ηPDC
(10.3)
The denominator ηPDC in this expression is simply the oscillator signal power Posc within
the resonator core:
ηPDC ≡ Posc = vmax imax
(10.4)
so that (10.3) may be rewritten in the form:
BGm−C =
4πkT F fo2
Posc
250
(10.5)
Evaluation of (10.5) for low power, high frequency operation (e.g. Posc = 100µW, fo = 100
MHz), gives a bandwidth BGm−C = 5 Hz. While this result gives some indication of the
oscillator’s purity, one is generally more interested in finding an expression similar to that
of (10.1) from which the noise density as a function of frequency offset can be evaluated.
To derive such a result, note that the oscillator power can be written in terms of the noise
density No at the center frequency (∆f = 0) as:
π
Posc ≡ No BN = No B
2
(10.6)
where BN is the resonator’s noise bandwidth. At frequencies outside the 3 dB bandwidth
B, the noise is shaped by the second-order resonator’s response characteristic to give:
NoGm−C (∆f ) = No
B/2
∆f
2
(10.7)
Thus, combining (10.5) - (10.7), the noise density at any frequency outside the oscillator’s
bandwidth may be written as:
NoGm−C (∆f ) = 2kT F
fo
∆f
2
(10.8)
or, dividing by the oscillator power to give a normalized expression similar to the form in
Equation (10.1):
NoGm−C (∆f )
2kT F
=
Posc
Posc
fo
∆f
2
(10.9)
The noise in this expression contains half its power in amplitude noise and half its power
in phase noise. Hence, an expression for the phase noise alone may be found by dividing
(10.9) by 2 to give:
251
NopGm−C (∆f )
kT F
=
Posc
Posc
fo
∆f
2
(10.10)
Repeating the above analysis starting with the dynamic range expression for the Q-enhanced
LC filter developed in Chapter 6, yields similar results, except for an improvement of Q2o :
1 kT (F + 1)
NopLC (∆f )
= 2
Posc
Qo 2Posc
fo
∆f
2
(10.11)
Example Phase Noise Calculations
As an example of numerical values, consider the oscillator used in the Q-enhanced filter
prototype of Section 9.2. The oscillator power required in Equation (10.11) is that within
the resonator core, and can be found for the case of the prototype filter design discussed in
Section 9.2 from:
Posc =
2
Vdif
fRM S
Rp
(10.12)
where the resistance Rp is known from the design to be approximatly 830 Ω. The voltage
Vdif fRM S however, was not specified in Chapter 9 and must be found from the measured
oscillator output power of -32 dBm by solving for the output voltage across the 50 Ω load
and dividing by the output buffer gain. At -32 dBm, the voltage across the 50 Ω load
is 5.62 mV, and at a bias current of 500 µA, the gain of the output stage (differential to
single-ended) is found to be 0.015. Thus, the value of Vdif fRM S is found to be 375 mV, and
from (10.12), the oscillator power is found 170 µW.
The expected phase noise in a 1 Hz bandwidth at a 10 KHz frequency offset can now be
found from (10.11) to be 450 fW/Hz (with F = 2 and Qo = 2.3), or, dividing by Posc :
252
NopLC (∆f )
= 2.7x10−9 ≡ −86dBc
Posc
(10.13)
This result compares favorably with the measured value of - 84 dBc reported in Section 9.2.
At a frequency offset of 100 KHz, the expected noise level is computed to be -106 dBc, and
is within 1 dB of the measured result of -105 dBc.
10.1.2
Phase Noise Effects on Receiver Operation
Oscillator phase noise results in two main degradations in radio receiver designs:
• Phase (or frequency) deviations are added to the received signal, degrading the demodulator output signal-to-noise ratio or bit-error-rate, and
• Phase noise mixed with strong adjacent channel interferers translates noise into the
IF passband, limiting the receiver’s selectivity and dynamic range.
These effects are illustrated in Figure 10.1 and are discussed quantitatively in the following
sections.
Phase/Frequency Deviations Added to Received Signal
As illustrated in Figure 10.1, the noise of the local oscillator is directly impressed onto the
received signal as a result of the mixing process. Whether or not this noise is significant
depends on the modulation type and bandwidth used in the radio system, and must be
determined by comparing the total RMS phase (or frequency) deviations induced, to the
phase or frequency deviations expected in the desired signal.
From [147], the RMS phase deviation resulting from the LO phase noise can be found from:
253
dB
dB
f
f
RF In
IF Out
LO
dB
f
Figure 10.1: Effects of phase noise on received signals.
sZ
∆ΦRM S =
f2
f1
2Nop(f )
df
Posc
(10.14)
where f1 and f2 define the range of frequencies over which the noise is significant. For
example, assuming QPSK modulation and a matched filter in the receiver’s IF, f2 will be
approximately Rs /2, where Rs is the symbol rate. The remaining frequency f1 depends on
whether coherent or differential demodulation is employed, and on the particular demodulator implementation used (e.g. the carrier recovery loop bandwidth).
Based on (10.10) and (10.11), the phase noise of an oscillator may be written in the form:
Nop(f ) = Nop (fA )
fA
f
2
(10.15)
where fA is an arbitrary reference frequency at which the phase noise is known. Substituting
this result into (10.14) and assuming coherent demodulation and a carrier recovery loop with
254
bandwidth f1 = Rs /10, Equation (10.14) can be solved to give:
s
∆ΦRM S =
16
Nop(Rs )
Rs
Posc
(10.16)
As an example, a QPSK signal with Rs = 500 Ks/s mixed with the oscillator in Section 9.2
where:
−105dB
Nop(500KHz)
= 10 10
Posc
100KHz
500KHz
2
= 1.27x10−12
(10.17)
would, according to (10.16) have an RMS phase deviation of 3.2 milli radians added to it by
the LO. Since this noise is significantly below the 1.6 radian minimum phase deviation for a
QPSK signal, the phase noise of the oscillator is not a significant problem in this example.
In the case of FM or FSK modulated systems, the RMS frequency noise in the demodulated
output is the parameter of interest. From [147], the frequency noise induced by the LO can
be found from:
sZ
f2
∆FRM S =
f1
2Nop(f ) 2
f df
Posc
(10.18)
Solving this expression with (10.15) then gives:
s
∆FRM S =
2
Nop(f2 ) 2
f (f2 − f1 )
Posc 2
(10.19)
and for the case of a FSK system with a 500 Kb/s data rate, f2 = 500 KHz, and f1 = 0
Hz, the oscillator of Section 9.2 would induce a frequency deviation of ∆FRM S = 400 Hz.
Again, this is well below the peak modulation of such a system and oscillator phase noise
influence on demodulation performance can be ignored.
255
Phase Noise Limitations on Selectivity and Dynamic Range
The second and most severe effect of LO phase noise on receiver performance is the translation of noise at large frequency offsets into the receiver’s passband. This problem occurs
when large adjacent channel signals are present as shown in Figure 10.1, and can limit the
receiver’s selectivity and dynamic range performance.
To find the magnitude of these effects, suppose that the ratio of the interferer signal power
I to the desired signal power S in Figure 10.1 is I/S with I/S 1. Let the interferer be
displaced from the desired signal by a distance ∆f , and let the receiver’s IF bandwidth be
BIF . Then, if we assume for simplicity that the LO phase noise in a range BIF /2 either
side of the frequency offset ∆f is approximately constant and the mixer’s gain is 1.0, the
noise power Nout translated into the IF passband by the mixing process will be:
Nout = I
Nop(∆f )
BIF
Posc
(10.20)
and the ratio of the desired signal power Sout to the added noise will be
Sout
=
Nout
1
I Nop (∆f )
S Posc BIF
(10.21)
For the case of a sufficiently large interferer and a sufficiently noisy LO, this ratio will fall
below the minimum carrier-to-noise ratio (C/N )min required at the demodulator input and
reception will be lost. Solving (10.21) for the required phase noise performance to keep
Sout /Nout ≥ (C/N )min, gives:
Nop (∆f )
≤
Posc
1
I
S (C/N )minBIF
256
(10.22)
Suppose, for example that I/S represents a desired adjacent or alternate channel selectivity
performance level of 50 dB, and that the receiver’s IF bandwidth is 1 MHz. Then, for a
demodulator with (C/N )min = 15 dB, the required phase noise is found from (10.22) to be
-125 dBc, which is comparable to the noise of the oscillator in Section 9.2 evaluated at a
1 MHz offset. Note that in this case, a Gm-C oscillator operating at a comparable power
level would not provide suitable performance since its phase noise is higher by a factor of
Q2o . Moreover, neither oscillator would meet the 50 dB requirement if it was operating at
2
2 GHz, since the phase noise at the same offset would be a factor of ( 2000
200 ) higher at this
frequency. For operation at frequencies in the GHz range, the only solutions would be to
employ an LC oscillator with a higher inductor Q, to increase the oscillator power, or to
relax the performance requirement.1
10.2
Receiver Design Example
With the issue of LO purity put to rest, this section turns to the main thrust of this final
chapter — a top-level design for a fully integrated receiver. The design considered will be
a Digital European Cordless Telephone (DECT) receiver with power consumption and cost
targets at or below those of current DECT chip sets discussed in [39] and [28] and overviewed
in Chapter 2. The DECT service was selected for this exercise for several reasons, including:
• There is widespread commercial interest in this standard.
• The standard operates in the newly allocated 1800 MHz portion of the radio spectrum
and employs wideband modulation typical of many newer digital wireless services.
• DECT employs time-division duplexing (TDD), which is more integration-friendly
than frequency-division duplexing system architectures.
1
The subject of frequency synthesis has not been covered here, but was discussed briefly in Chapter 2.
There it was concluded that LOs operating at the RF frequency are needed since multiplication of lower
frequency signals to RF by a factor N results in a factor of N 2 increase in phase noise and spurious products.
257
The final point is particularly important because the use of TDD eliminates the need for
antenna duplex filters, which would be required to handle the full transmitted power, and
therefore cannot be integrated using active filter techniques.2
In the first subsection below, the DECT standard is reviewed, and requirements for DECT
receivers are examined. These requirements set the minimum performance that a fully integrated DECT receiver must have to receive type-approval and to compete with existing implementations constructed from commercially available chip sets and off-chip ceramic/SAW
filters. In the next subsection, a design for a fully integrated, dual-conversion receiver is
proposed and the performance of the filters and local oscillators employed are computed
using techniques developed in the preceding section and in earlier chapters. Finally, the
last two sections examine possible approachs to improve the receiver’s performance and
reduce the power consumption by employing direct-conversion techniques and a simplified
single-conversion architecture.
10.2.1
DECT System Requirements
The DECT standard defines a wireless architecture for implementing next-generation cordless phones, as well as a variety of digital personal communications services. As shown
in Table 10.1, DECT products operate at approximately 1.9 GHz and employ frequency
division multiple access (FDMA) and time division multiple access (TDMA) techniques to
allow high capacity cellular-like systems to be built. Time division duplexing (TDD) is used
to simulate simultaneous transmission and reception of data, allowing bulky and expensive
duplex filters to be replaced with smaller and more integration-friendly RF switches.
DECT specifies the use of Gaussian frequency-shift keying (GFSK) modulation with a bit
2
Some form of “roofing” filter may still be needed at the antenna in TDD systems for the transmitter
side of the product. However, such a filter would be required only to attenuate harmonics and therefore may
be constructed with on-chip LC lowpass filter designs such as those described in [131]
258
Table 10.1: DECT System Requirements.
Parameter
RF Frequency
Modulation
Data Rate
Access/Duplex Method
Number of RF Channels
Channel Spacing
Peak Deviation
Timeslots
Frame Length
Frequency Accuracy
Sensitivity (BER = 10−3 )
Noise Figure (Eb/No = 14 dB)
Noise Floor
Input Compression
Dynamic Range
Input Intercept
Spurious Free Dynamic Range
Selectivity (IF filter attenuation)
Adjacent Channel
Alternate Channel
Current Consumption
Die Size
259
Requirement
1881 - 1898 MHz
GFSK (BT = 0.5)
1.152 Mb/s
FDMA/TDMA/TDD
10
1.728 MHz
288 KHz
12 / frame
5 ms TX, 5 ms RX
50 KHz (26 ppm)
≤ −83 dBm
≤ 16 dB
≤ −97 dBm
≥ −36 dBm
≥ 61 dB
≥ −27 dBm
≥ 47 dB
≥ 30 dB
≥ 45 dB
≤ 50 mA
≤ 10 mm2
rate of 1.152 Mb/s and a bandwidth-time product of 0.5. Thus, most designs will employ
channel select filters with a bandwidth in the neighborhood of 1.1 MHz. The defined
channel spacing of 1.728 MHz then provides the equivalent of guard bands, permitting use
of adjacent as well as alternate channels in many applications.
Unlike many older wireless systems, requirements for DECT receivers are specified in the
DECT standard and type-approval documents. A recent article by Madsen and Fague
[13] provides an excellent summary of these requirements and was used as the basis for
many of the performance parameters listed in Table 10.1, including sensitivity, noise figure,
and dynamic range. Although no data was provided in the article directly for channel
selectivity, a Murata SAFC110.6MA50T SAW filter was suggested, and reference to this and
other similar filters’ data sheets provided the adjacent and alternate channel performance
specifications shown. Finally, the current consumption and die size values shown, although
not strictly DECT requirements, are important factors in the design of commercial products
and play a critical role in assessing receiver integration technology. The values listed for
these parameters were estimated from the DECT chipset described in [39].3
10.2.2
Dual Conversion Superheterodyne Receiver Design
A block diagram for a future, fully integrated DECT receiver based on the techniques
developed in this dissertation is proposed in Figure 10.2. This design adopts a conservative
dual conversion superheterodyne architecture to provide the required channel selectivity of
1.1 MHz while limiting the Q of all filters to a maximum of 200. Both the preselect and
first IF filters are Q-enhanced designs which could be implemented using the techniques
discussed in Chapters 8 and 9, while the lower-Q second IF filter employs traditional Gm-C
filter techniques. The Q-enhanced designs have companion oscillators used for master-slave
3
Although the 50 mA current consumption shown may appear high, these receivers implement powerdown modes to allow the average consumption to be reduced.
260
tuning of filter Q as demonstrated in Section 9.2, and which double as local oscillators
for down conversion to the next IF. In both cases, oscillator and filter frequency tuning
is performed through the simplified approach discussed in Chapter 7 wherein a frequency
prescaler is provided on-chip and used in conjunction with the host receiver’s microprocessor
to form a frequency locked loop. Finally, note that no amplifier circuits are employed. In this
design, the Q-enhanced filters provide all the front-end gain needed to achieve the required
noise figure performance. Hence, the power normally consumed by amplifiers in discrete
filter based receivers can be used within the integrated filters to achieve a comparable total
receiver power consumption.
fo = 1889 MHz
B = 20 MHz
1882 - 1897 MHz
Step
Attenuator
From
Transmitter
fc = 300 MHz
2 Pole
Preselect
Filter
fo = 200 MHz
B = 1 MHz
1 Pole
IF Filter
LPF
IR Mixer
fo = 10 MHz
B = 1.1 MHz
3 Pole
IF Filter
LPF
200 MHz
To Limiter/RSSI
and Demod
10 MHz
fLO= 2082 - 2097 MHz
-1 dB
+26 dB
0 dB
Gain
0, -20 dB
1 dB
NF
8 dB
10 dB
0, 20 dB
1, 21 dB
Cumulative NF
1 dB
9, 29 dB
-1,-21 dB
Cumulative Gain
-1 dB
+25,+5 dB
fc = 15 MHz
fLO= 210 MHz
-2 dB
2 dB
+12 dB
32 dB
+23,+3 dB
0 dB
8 dB
12, 32 dB
+35,+15 dB
-2 dB
2 dB
0 dB
39 dB
+33, +13 dB
13, 33 dB
+33, +13 dB
Figure 10.2: Dual conversion integrated receiver design.
As shown below, the weakest link in this design is the first IF filter which operates at
the maximum selectivity Q of 200, and therefore has the lowest dynamic range. This
filter employs a design similar to that of the prototype IF filter of Section 9.2. Since
the inductor Q is relatively low for Q-enhanced LC filters operating at these frequecies,
the in-band dynamic range possible in this design with reasonable chip areas and power
consumptions is less than that required in the overall receiver. This difficulty is handled by
adding a 20 dB step attenuator immediately in front of the receiver’s preselect filter. The
attenuator could be implemented with simple MOSFET switches and would be controlled
by the host receiver’s microprocessor in response to the received signal strength indicator
(RSSI) level measured in the IF limiter/amplifier chain (not shown). This design allows the
261
receiver to handle the required variations in the level of the desired signal due to changes in
distance from receiver to transmitter. Dynamic range improvements with frequency offset
described in Chapter 6 then provide the necessary blocking and spurious-free dynamic range
performance relative to adjacent and alternate channel signals.
The image rejection performance of the first IF filter is improved by using an image rejection (IF) mixer in the down conversion to the second IF. Without this mixer, substantial
additional chip area and power consumption would be required to implement a multiplepole filter design, and potential problems of providing tight control over the filter passband
centering and shape would have to be addressed. However, with a simple, untrimmed image
reject mixer, the number of filter poles can be reduced to one while maintaining good image
(spurious) response performance as discussed in Chapter 4. Even if the mixer image rejection is limited to only 20 dB, the additional 32 dB provided by the filter brings the total
spurious response attenuation to over 50 dB. Trying to achieve a full 50 dB with only an
image reject mixer is generally impractical, even if trimming is permitted, due to requirements to set and hold amplitude and phase balances of better than 1 % and 1o respectively.
Thus, the second IF filter is required in this design, despite its limitations.
Finally, a 3-pole filter in the second IF supplies the majority of channel selectivity. Since
this filter operates at a relatively low frequency of 10 MHz, and a low selectivity Q of 9.1,
established Gm-C techniques can be applied to achieve accurate response and good dynamic
range while keeping power consumption to manageable levels.
Detailed Peformance Estimates
A complete circuit-level design for the proposed receiver is a significant engineering effort
and is therefore outside the scope of this dissertation. However, the performance of the
receiver can be predicted using the equations developed in previous chapters and by extrapolating from the prototype performance of Section 9.2. These techniques have been used to
262
generate the performance estimates shown in Tables 10.2 - 10.4 and to arrive at the overall
gain and noise figure allocations shown previously in the receiver block diagram of Figure
10.2. The resulting data was then combined to determine the overall receiver performance
estimates shown in Table 10.5 where the design is compared with the requirements of Table
10.1 and with the performance of the DECT receivers reported in [39] and [28].
Table 10.2 summarizes the performance of the RF preselect filter and local oscillator. The
preselect filter assumed is a 2-pole coupled resonator design similar to that of the prototype
filter of Section 9.2, but scaled in frequency to 1900 MHz. Thus, this filter assumes the
use of an advanced, fine-geometry IC process with gate lengths in the neighborhood of 0.5
µm. Apart from this potentially aggressive scaling of operating frequency, the design is
otherwise easier to implement than that of the prototype filter due to the smaller inductors
and higher Qo values possible at higher frequencies. Based on inductor Q simulations
discussed in Chapter 8, a Qo value of 8 is assumed for an inductor outer dimension of 500
µm. The required Q enhancement is therefore substantially less than that in the prototype
filter design, leading to better dynamic range and lower temperature drifts. With smaller
inductors, the coupling can be set in this filter with inductor spacing alone, and lower
oscillator feedthrough can be achieved in acceptable chip areas. In addition, the higher
base Q minimizes the passband asymetry problems experienced in the prototype design, so
that all coupling neutralization circuitry can be eliminated.
The LO associated with this filter is designed to operate at 2082 to 2097 MHz, resulting in
a 200 MHz first IF. Frequency tuning is provided as in the prototype design with switched
tuning capacitors, but with 5 to 6 bits of control to allow a resolution down to approximately
2 MHz, while still maintaining the necessary range to tune out manufacturing tolerances
and temperature related drift. Although this resolution is not fine enough to address the
50 KHz tuning increment required for accurate demodulation, the operating frequency of
the LO can be measured to within 25 KHz by the host microprocessor with an N = 128
prescaler ratio and a 5 ms counter gate time. The IF frequency is then known to this level
263
of accuracy and the ± 1 MHz error can be corrected by suitable control of the first IF filter’s
center frequency and LO.
The design and performance of the first IF filter and LO is summarized in Table 10.3. Except
for minor modifications and additions, this design is essentially equivalent to the prototype
filter configured to provide master-slave tuning of filter Q. To allow operation at a higher
Q (Q = 200 versus Q = 100), the power consumption of the filter has been increased by a
factor of 2, and to provide finer frequency control, a digital-to-analog converter controlled,
simulated inductance is assumed to be used in both the filter and oscillator. Finally, the
coupling neutralization circuits provided in the prototype implementation are eliminated
since only a 1-pole filter is used.
An interesting feature of the first IF filter is the relatively high noise figure used. A noise
figure of 32 dB is specified for the reasons described in Chapter 3. This value is equal to
the (decibel) sum of the gain and noise figure of upstream circuits, guaranteeing that the
noise floor produced by these circuits will be equal in magnitude to the noise floor within
the filter. Although this choice increases the effective noise figure of the overall receiver
by 3 dB, it guarantees minimal degradation of the first IF filter’s already limited dynamic
range.
The final filter used is described in Table 10.4. As previously mentioned, this filter can
be implemented using Gm-C techniques due to the low selectivity Q of 9.1 that results
from the 1.1 MHz bandwidth operating at a second IF frequency of 10.0 MHz. The low
Q value allows a higher dynamic range than that of the first IF filter, despite the Gm-C
technique’s inherent limitations. In turn, a noise figure of 39 dB, 6 dB less than the sum
of the upstream circuits’s gain and noise figures is specified. While this noise figure means
that the incoming noise will be 6 dB higher than the noise generated within the filter and
that the filter’s effective dynamic range will therefore be cut by 6 dB, it reduces the impact
of the filter on the overall receiver noise figure to only 1 dB.
264
Table 10.2: RF Preselect Filter and LO Performance.
Parameter
Value
Preselect Filter
Type
Tuning
Q
Frequency
Center Frequency
Bandwidth
Selectivity Q
Number of Poles
Image Rejection
Inductor Size
Resonator Base Q
Q Enhancement
Supply Voltage
DC Current Consumption
Resonator core
Total
Resonator 1 dB Compression
Dynamic Range (BIF = 1.1 MHz)
Spurious Free Dynamic Range
Gain
Noise Figure
Local Oscillator
Frequency
Resolution
Accuracy (N=128 prescaler, 5 ms count)
DC Current
Oscillator Power
Inductor Size
Resonator Base Q
Phase Noise (∆f = 1.728 MHz)
FM Noise (∆f = 0 − 550 KHz)
Noise Limited Adjacent Channel
Selectivity (at 15 dB C/N)
265
Q-Enhanced LC
Master-Slave
µP Assisted FLL
1890 MHz
20 MHz
95
2
64 dB
500 µm
8
11.9
3-5V
2 mA
≤ 10 mA
200 µW (-7 dBm)
73 dB
54 dB
26 dB
8 dB
2082 - 2097 MHz
2 MHz
25 KHz
5 mA
350 µW
500 µm
8
-124 dBc
1.1 KHz
48 dB
Table 10.3: First IF Filter and LO Performance.
Parameter
Value
Preselect Filter
Type
Tuning
Q
Frequency
Center Frequency
Bandwidth
Selectivity Q
Number of Poles
Image (Spurious) Rejection
Filter
Filter + IR Mixer
Inductor Size
Resonator Base Q
Q Enhancement
Supply Voltage
DC Current Consumption
Resonator core
Total
Resonator 1 dB Compression
Dynamic Range
In-Band
Adjacent Channel
Spurious Free Dynamic Range
In-Band
Adjacent Channel
Gain
Noise Figure
Local Oscillator
Frequency
Resolution
Switched Capacitors
Gyrator Simulated Inductor
Accuracy (N=16 prescaler, 5 ms count)
DC Current
Oscillator Power
Inductor Size
Resonator Base Q
Phase Noise (∆f = 1.728 MHz)
FM Noise (∆f = 0 − 550 KHz)
Noise Limited Adjacent Channel
Selectivity (at 15 dB C/N)
266
Q-Enhanced LC
Master-Slave
µP Assisted FLL
199 - 201 MHz
1 MHz
200
1
32 dB
≥ 52 dB
850 µm
2.5
80
3-5V
2 mA
≤ 6 mA
60 µW (-12 dBm)
52 dB
63 dB
41
54
12
32
dB
dB
dB
dB
209 - 211 MHz
1 MHz
25 KHz
3.2 KHz
6 mA
350 µW
850 µm
2.5
-133 dBc
144 Hz
57 dB
Table 10.4: Second IF Filter Peformance.
Parameter
Type
Tuning
Q
Frequency
Center Frequency
Bandwidth
Selectivity Q
Number of Poles
Supply Voltage
DC Current Consumption
Resonator core
Total
Resonator 1 dB Compression
Dynamic Range
In-Band
Adjacent Channel
Spurious Free Dynamic Range
In-Band
Adjacent Channel
Gain
Noise Figure
267
Value
Gm-C
Master-Slave
Master-Slave
10.00 MHz
1.1 MHz
9.1
3
3-5V
1 mA
≤ 8 mA
100 µW (-10 dBm)
72 dB
78 dB
54 dB
62 dB
0 dB
39 dB
Table 10.5: Overall Receiver Performance.
Parameter
Requirement
Noise Figure
Sensitivity
Input Noise Floor
Dynamic Range (In-Band)
Spurious Free Dynamic Range (Adj Ch)
Selectivity (IF Filter Attn.)
Adjacent Channel
Alternate Channel
Current Consumption
≤ 16
≤ −83
≤ −97
≥ 61
≥ 46
Die Size
dB
dBm
dBm
dB
dB
30 dB
45 dB
≤ 50 mA
≤ 10 mm2
Proposed Design
Design in [13]
Design in [39]
13 dB
-86 dBm
-100 dBm
68 dB
50 dB
12.6 dB
-87 dBm
-101 dBm
67 dB
52 dB
8 dB
-91 dBm
-105 dBm
75 dB
55 dB
40 dB
64 dB
35 mA
+ mixer/demod
2.2 mm2
+ active circuits
-
50 mA
-
< 8.5 mm2
Based on the performance of these individual components, and the gain and noise distributions shown in Figure 10.2, the overall performance of the proposed receiver is summarized
in Table 10.5. As indicated by comparison with the required performance and the performance of the receivers described in the references,4 the proposed integrated receiver design is
competitive in all areas. The integrated receiver meets or exceeds performance requirements
and compares favorably with designs based on commercial chip sets and ceramic/SAW offchip filters. Thus, with a suitable engineering effort, full integration of DECT receivers is
indeed feasible. Moreover, the relatively small chip area consumed by the required inductors
indicates that the cost of production can be kept low, and by eliminating external filters,
overall system cost can be reduced at the same time product size is decreased.
10.2.3
Direct Conversion Receiver Design
The design of the previous section can be improved (in theory) if the dynamic range bottleneck at the first IF filter is eliminated by the use of a direct conversion architecture
such as shown in Figure 10.3. In this implementation, the preselect filter is retained to
4
Values for the design described in [39] are computed based on stated performance of the chip set combined
with a 12 dB gain, 4 dB noise figure GaAs preamp.
268
provide good out-of-band rejection while achieving a front-end gain in the neighborhood
of 30 dB. However, all channel selectivity is now provided by integration-friendly lowpass
filter designs, and the power consumption of the first IF filter and LO, and of the second IF
filter are eliminated. In addition, since the preselect filter performance shown previously in
Table 10.2 meets the dynamic range requirements of the DECT specification, it may also
be possible to eliminate the step atteunator from the design.
fc = 550 KHz
fo = 1889 MHz
B = 20 MHz
1882 - 1897 MHz
Optional
Step
Attenuator
From
Transmitter
2 Pole
Preselect
Filter
+45
o
-45
o
4 Pole
LPF
I
4 Pole
LPF
Q
To Limiter/RSSI
and Demod
fc = 550 KHz
f LO= 1882 - 1897 MHz
Figure 10.3: Direct conversion integrated receiver design.
Unfortunately, closer consideration indicates that this design is probably not feasible. The
most severe problem is coupling of the LO oscillator to the filter circuits. Even if the
on-chip coils can be spaced by several inductor diameters to lower the coupling coefficient
to 0.001, the 350 µW oscillator signal will induce a signal in the filter on the order of 30
µW, or -15 dBm. This signal will create DC offsets in the mixer outputs several orders of
magnitude higher than the desired signal level. Moreover, the reverse isolation of the input
transconductor used in the filter design may not be sufficient to limit radiation of the LO
signal to acceptable limits. Additional problems with this design include the LO tuning
resolution needed to achieve accuracies on the order of 50 KHz or less, and the the need to
maintain accurate mixer phase shifts and amplitude balance over the full RF tuning range.
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10.2.4
Single Conversion Receiver Design
All of the problems of the direct conversion design listed above can be reduced if the
receiver architecture is modified as shown in Figure 10.4. Strictly speaking, this design
is a single-conversion receiver architecture with a non-coherent quadrature demodulator.
However, it can also be viewed as a fixed-tuned direct conversion receiver operating at 200
MHz preceeded by a block down converter. This latter interpretation allows the design to
be contrasted with the direct conversion design to see how the problems of the previous
receiver are addressed.
fc = 550 KHz
fo = 1889 MHz
B = 20 MHz
1882 - 1897 MHz
fc = 300 MHz
Optional
Step
Attenuator
From
Transmitter
2 Pole
Preselect
Filter
LPF
200 MHz
+45
fLO= 2082 - 2097 MHz
o
-45
o
4 Pole
LPF
I
4 Pole
LPF
Q
To Limiter/RSSI
and Demod
fc = 550 KHz
200 MHz
LO
Figure 10.4: Single conversion integrated receiver.
In the new design, the LO operates at a frequency offset from the frequency to which
the filter is tuned, reducing oscillator feedthrough by 25 to 30 dB (depending on filter
bandwidth). Hence, the oscillator signal coupled into the filter circuits will now be at a
level on the order of -40 dBm or less. Since two LOs are used, one at the RF and one in
the 200 MHz IF, the techniques used in the dual conversion architecture can be applied to
solve the tuning resolution problem. Coarse steps in the range of 2 MHz can be used in the
preselect filter, while fine tuning to within 50 KHz can be achieved using the 200 MHz LO.
Unlike the earlier dual conversion design however, the new architecture does not employ
IF filtering, so there is no inductor coupling problem within the 200 MHz direct conversion
receiver block. Moreover, the power consumption is reduced over the dual conversion design
270
by approximately 16 mA through the elimination of the first and second IF filters. While
the lowpass filters now used to provide channel selection will consume power, their pole
Q values will be low (Q < 5), so that they can provide good dynamic range at very low
currents.
Finally, we note that the quadrature mixing should be simplified over that of the previous
direct conversion design by operation at a low and relatively fixed 200 MHz IF. While the
addition of the 200 MHz LO will consume power not used in the direct conversion design,
the total power of the single conversion receiver may actually be lower since only a single 2
GHz mixer is now required.
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Chapter 11
Conclusions and Future Directions
Worldwide allocations of radio frequency spectrum for cellular and PCS services in recent
years have placed new emphasis on the development of small, low cost wireless products. As
shown in Chapter 2, RF device manufacturers have responded by introducing chip sets with
increasing levels of integration. However, to date, virtually all high performance wireless
products continue to rely on discrete LC, crystal, ceramic, or SAW devices for the realization
of RF preselection filters, IF channel selection filters, and local oscillator tuned circuits.
During the past decade, considerable research has been directed at developing suitable
switched-capacitor and Gm-C based on-chip replacements for discrete filters now in use.
Although this work has produced several potentially viable designs at IF frequencies from
455 KHz to 10.7 MHz, these designs have yet to see substantial commercial application. A
critical problem faced by switched-capacitor and Gm-C filters was found to be the power
consumption required to simultaneously achieve narrow fractional bandwidths and acceptable dynamic range. This power consumption, which can reach several hundred milliwatts,
is incompatible with portable wireless product design.
These problems have caused some researchers to look for alternatives to the classic superheterodyne architecure as a means to integrate receiver hardware. Thus, the issues of
receiver system requirements and alternative receiver designs were investigated in Chapters
3 and 4 respectively. In the superheterodyne architecture on which virtually all present-day
receiver designs are based, RF and IF bandpass filters with small fractional bandwidths
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(e.g ≤ 2 %) are needed to suppress image responses, to provide required channel selectivity,
and to achieve good dynamic range in the presence of large amplitude inteferers on nearby
frequencies and within other service bands. Alternative receiver designs such as direct
conversion (or “zero-IF”) architectures have no image frequency and can theoretically provide all channel selectivity using integration-friendly lowpass filters. Unfortunately, direct
conversion receivers with performance matching that of present-day superheterodynes are
difficult to implement in practice. Problems include LO to RF coupling, resulting DC offsets and receiver radiation, and the need for precision quadrature mixers operating directly
at RF. Moreover, as shown in Chapters 3 and 4, direct conversion designs possess the same
fundamental design tradeoffs faced by superheterodyne receivers in the critical areas of dynamic range, gain, and noise figure. Thus, even if the practical problems of implementing
the direct conversion architecture can be overcome, narrowband preselect filters will still be
required, and good dynamic range must still be maintained in all circuits preceeding the
final channel-select filters.
Potential technologies for implementing on-chip bandpass filters at both RF and IF frequencies were reviewed in Chapter 5, and energy storage in resonant circuits was identified
as a critical factor for achieving good dynamic range at low power. On-chip electro-acoustic
filters possess this property and offer the potential of purely passive filtering, but require
IC fabrication process modifications, and have not yet reached a suitable level of maturity.
On-chip LC filters also possess this property and are fully compatibility with current IC
processing techniques. Thus, filter design using integrated inductors was chosen as a potentially viable solution to the on-chip bandpass filtering problem and studied in depth in
later chapters.
A study of the literature on integrated inductors in silicon processes showed that a few
passive on-chip filters had been reported, but that no high-Q bandpass designs existed. A
principle problem was found to be the low Q values of spiral inductor designs. With Q
values of less than 10 at frequencies of 1 GHz and below, purely passive high-Q bandpass
273
filters cannot be realized. However, a few researchers have noted that by adding active
circuits, inductor losses can be compensated for, and arbitrarily high Q values can be
achieved. Proposals for constructing such Q-enhanced LC filters had appeared in circuits
and systems conferences, but no fully integrated realizations had been reported.
In Chapters 5 and 6, the dynamic range performance of Q-enhanced LC filter designs
was investigated and found to be limited by the active circuits used. However, a detailed
study of this problem revealed that even a small inductor Q results in significant dynamic
range and power consumption advantages over Gm-C and other fully active designs. For
example, an inductor with Qo = 5 results in a dynamic range advantage of Q2o = 25 or
14 dB. Alternatively, for a given dynamic range requirement and receiver bandwidth, the
Q-enhanced filter design consumes over an order of magnitude less power.
The issue of how much dynamic range is required was then considered to determine if these
advantages are sufficient to allow integrated, low-power receivers to be built. Specification
of dynamic range in receivers involves at least three separate definitions of performance. In
Chapter 3, blocking and spurious-free dynamic range were identified as the most important
characteristics since the total dynamic range needed to compensate for varying distances
between transmitter and receiver can be increased through the addition of attenuator and
AGC circuits. However, the dynamic range of filters is nearly always quoted for in-band
signals only, using the filter’s noise bandwidth to define the total integrated noise power.
Thus, to allow comparisons with receiver performance to be made, the relationship of this
measure of performance to blocking and spurious-free dynamic range was investigated in
considerable depth.
Blocking and spurious-free dynamic range were shown in Chapter 6 to depend on the receiver’s IF bandwidth, rather than on the bandwidth of the filter itself. For IF filters, these
bandwidths are approximately equal, but for RF preselect filter designs, the preselect bandwidth is many times that of the signal and the dynamic range when used within a receiver
274
is higher than that of the filter alone. Moreover, blocking and spurious-free dynamic range
are fundamentally defined relative to signals which fall outside the filter passband, and for
a properly designed active filter, this fact was shown to result in dynamic range values
significantly above that of the filter’s in-band performance. With these factors in hand, the
dynamic range of Gm-C and Q-enhanced LC filters was then compared with the dynamic
range of traditionally designed receiver circuits and expressions were developed to determine fundamental limits on the Q values allowed within receivers having restricted power
consumption budgets. This study revealed that the Q of Gm-C filters must be limited to
less than 30, while the Q of LC based designs can be as high as 100 or more depending on
the base inductor Q that can be achieved.
An additional consideration in on-chip filter design is the accuracy and temperature stability
of the transfer functions that can be realized. Thus, Chapter 7 considered the need for
tuning in integrated filters, and surveyed the design of control systems which can be used
to achieve a desired level of tuning precision. In Q-enhanced filters, as in more traditional
Gm-C filter designs, both frequency and Q tuning were found to be needed during use and
the basic problem of controlling the filter transfer function was shown to be the same in
both cases. Thus, control systems developed for Gm-C filters can be adapted to the tuning
of Q-enhanced filters as well. A review of these control system designs showed that the
common master/slave technique offers a simple and robust solution for filters with Q values
up to about 50, but that other techniques may provide better performance in particular
situations. A new orthogonal-reference-tuning (ORT) technique was proposed and examined
in detail as a method to tune high-Q designs to very high levels of precision. The concept of
keep-it-simple (KIS) was then investigated and several simplifications of existing techniques
were proposed.
Having established bounds on filter performance and the theoretical viability of the Qenhanced filter approach, detailed design considerations for implementing these filters were
investigated in Chapter 8. On-chip inductor Q was found to be primarily dependent on
275
frequency of operation and on inductor size, and relatively independent of the actual inductance value. Significant additional losses however, can occur due to the underlying
substrate, making detailed modeling of inductors critical to the successful design of high
Q filters. A lumped circuit simulation model was used to determine representative performance achievable in standard CMOS processes with inductor outer dimensions ranging
from 330 µm to 1000 µm. Q values from 2 to 8 were found depending on frequency of
operation, inductor size, and other properties of the fabrication process.
Techniques for Q enhancement were then investigated and criteria for selecting values of
essential components were shown to depend on the terminating impedances, filter gain,
and noise figure desired. A study of balanced filter design led to the developement of a
novel center-tapped sprial inductor geometry which allows higher self-resonant frequencies
and Q values to be achieved. In turn, this development allows better dynamic range and
lower power operation to be realized, and allows the usable frequency range of operation
of Q-enhanced filters to be extended down to about 200 MHz — far lower than the 1 GHz
suggested in the literature.
The full viability of Q-enhanced filtering was then demonstrated in Chapter 9 through
the design, fabrication, and testing of two prototype ICs. In addition to serving as a
vehicle to work out device-level implementation issues for these filters, these prototypes were
constructed to validate dynamic range performance predictions and to assess the viability
of various tuning options considered in previous chapters.
The first IC, which operated at a nominal frequency of 100 MHz, employed off-chip inductors
and tuning capacitors and served primarily as an early prototype of the proposed technique.
Starting from an LC tank circuit Q of 70, the filter exhibited excellent short-term tuning
stability and good dynamic range when operating at Q values up to 10,000. A gain of 31 dB
and an in-band dynamic range of 56 dB were achieved at a current consumption of less than
2 mA (excluding the on-chip mixer). Dynamic range increases with frequency offset were
276
found to be in excellent agreement with predictions in previous chapters, with blocking and
spurious-free dynamic range performance reaching 100 and 85 dB respectively at 1 MHz
offset from the filter center.
The second IC was a fourth-order, fully integrated design, with two on-chip center-tapped
spiral inductors, each with an outer dimension of 850 µm. This IC, which consumed approximately 3.3 mm2 of chip area, operated at a nominal frequency of 200 MHz with a
selectivity Q of 100, and consumed less than 6 mA from a 3 V supply, making it potentially
suitable for use as a first IF filter in modern wideband cellular and PCS products. Base
Q of the resonant circuit at 200 MHz was measured at 2.3, and when enhanced to the
nominal Q of 100, provided respectable in-band dynamic range, and good dynamic range
at frequency offsets. Experiments with filter tuning showed excellent agreement with theory and demonstrated the viability of using the master/slave technique to control filter Q.
Short-term, open-loop stability of both frequency and Q were also excellent (frequency drift
of less than 50 KHz in 5 minutes, and negligable Q drift at Q values up to 200) indicating
that the simplified tuning techniques proposed in Chapter 7 can be applied in the design
of these filters. Although the prototype IC was constructed for use at 200 MHz, this choice
was primarily dictated by the available 2 µm IC process and by the low frequency, 40 pin,
DIP packaging used. Thus, scaling of the demonstrated design techniques to finer-line fabrication processes and the use of RF compatible packaging is expected to allow extension
to RF preselect applications as well.
Finally, Chapter 10 investigated performance requirements and performance bounds for onchip local oscillators — the remaining obstacle to full receiver integration. This problem was
found to be related to, but less severe than that of filter dynamic range, at least for newer
wideband wireless system designs. A top-level design for a hypothetical, fully integrated
receiver was then proposed and evaluated to investigate the possibility of constructing a
complete “receiver-on-a-chip”. This study, combined with results obtained from the filter
prototypes confirms the theoretical as well as practical viability of the technology. Using
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the theory developed throughout this disseration, a detailed set of performance data for the
three filters and two local oscillators used in the design was generated. The performance
of these components was then used to determine overall receiver performance, which was
shown to not only meet the required specifications, but to compete well with two different
designs based on current technology chipsets employing off-chip ceramic and SAW filters.
Nevertheless, several practical issues remain to be addressed before this new technology can
be used to its full potential and adopted for routine commercial receiver designs. First,
the design and characterization of spiral inductors should be examined in more depth. Onchip inductor Q is critical to the optimization of dynamic range and minimization of power
consumption, and to the simplification of filter tuning. Techniques should therefore be
investigated for realizing higher Q inductors in newer, multi-layer metal processes. Higher
inductance values, and hence, higher quality factors may also be achievable if thin-film
ferrites can be incorporated into the fabrication process. In addition, design tools should be
developed to accurately predict the inductance values, self-resonant frequencies, and quality
factors that result, so that filter designs can be created with minimal iteration.
Second, the design of improved linearity transconductors should be investigated. A simple,
long-tailed differential pair was used in the prototype designs to assure working devices, at
the expense of limited voltage swings at large Q enhancements. Several linearized transconductors have been proposed in the literature for use in Gm-C filters, and may be adapted
to the design of Q-enhanced filters to the extent that they operate at high frequencies. The
effects of these techniques on dynamic range, and on filter stability when operating at high
Q values however, must be assessed.
Third, additional work is needed to determine the best and least costly tuning techniques,
and to refine the design of tuning circuits. In particular, a more detailed design is needed to
determine what, if any, practical problems may arise in implementing the simplified tuning
techniques proposed in Chapter 7 for use in TDMA/TDD wireless systems. In addition,
278
techniques for implementing coupling neutralization should be studied in more detail.
Finally, the practical tradeoffs discussed in Chapters 4 and 10 with respect to superheterodyne and direct conversion receiver architectures should be assessed. These tradeoffs are
perhaps best studied through a full circuit-level design effort with subsequent fabrication
and testing. Such a detailed engineering effort is needed to determine which receiver architecture will ultimately win approval in the design of completely integrated, low power,
radio receivers, and to prepare the technology for transfer from the laboratory to commercial
product development.
279
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Vita
Bill Kuhn was born in Newport News, Virginia in 1956. His interests in electronics and
radio date to the early 1970s when he obtained amateur radio and commercial FCC licenses
and worked in the areas of television/radio repair and electronic marine navigation during
High School and College. He received the B.S. degree in Electrical Engineering in 1979 from
Virginia Tech, and from 1979 to 1981 he worked at Ford Aerospace and Communications
Corporation where he developed satellite ground receiver equipment, including frequency
synthesis and digital bit synchronizer circuitry. He obtained the M.S. degree in Electrical
Engineering in 1982 from the Georgia Institute of Technology and was employed during the
summer of 1982 at the NASA Ames Research Center in the area of error correction coding
technology assessment. From 1983 to 1993 he was employed at the Georgia Tech Research
Institute where he worked as a systems engineer in the area of electronic countermeasures
and as a task leader and software engineer in the area of analog and mixed-signal circuit
simulation and model development. He received the Bradley Fellowship from Virginia Tech
in 1993 where he is currently pursuing the Ph.D. degree. He holds two U.S. patents and
has published papers in the areas of computer music, computer aided engineering, and
integrated RF electronics. His current research interests are focused on the problem of
integrating complete wireless products in low-cost, low-power ICs.
295
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