U-131 APPLICATION NOTE Simple Switchmode Lead
Simple Switchmode Lead-Acid Battery Charger
John A. O’Connor
Lead-acid batteries are finding considerable use as both primary and backup power sources. For complete
battery utilization, the charger circuit must charge the battery to full capacity, while minimizing over-charging
for extended battery life. Since battery capacity varies with temperature, the charger must vary the amount
of charge with temperature to realize maximum capacity and life. Simple, low cost circuits are currently
available for small, low power requirements, while more complex solutions are affordable only on larger more
expensive systems. Often the greatest challenge is in designing mid-size, mid-price systems, where obtaining
optimum performance at moderate cost and complexity may be nearly impossible without dedicated integrated
circuits. This paper describes a compact lead-acid battery charger, which achieves high efficiency at low cost
by utilizing switchmode power circuitry, and provides high charging accuracy by employing a dedicated control
IC. The circuit described can be easily adapted to lower or higher power applications.
Lead-Acid Basics
Lead-acid battery chargers typically have two tasks
to accomplish. The first is to restore capacity, often
as quickly as practical. The second is to maintain
capacity by compensating for self discharge. In both
instances optimum operation requires accurate
sensing of battery voltage and temperature.
When a typical lead-acid cell is charged, lead sulfate
is converted to lead on the battery’s negative plate
and lead dioxide on the positive plate. Over-charge
reactions begin when the majority of lead sulfate has
been converted, typically resulting in the generation
of hydrogen and oxygen gas. At moderate charge
rates most of the hydrogen and oxygen will
recombine in sealed batteries. In unsealed batteries
however, dehydration will occur.
The onset of over-charge can be detected by
monitoring battery voltage. Figure 1 shows battery
voltage verses percent of previous discharge
capacity returned at various charge rates. Over
charge reactions are indicated by the sharp rise in
cell voltage. The point at which over-charge
reactions begin is dependent on charge rate, and as
charge rate is increased, the percentage of returned
capacity at the onset of over-charge diminishes. For
over-charge to coincide with 100% return of
capacity, the charge rate must typically be less than
C/100 (1/100 amps of its amp-hour capacity). At high
charge rates, controlled over-charging is typically
Figure 1. Over-charge reactions begin earlier (indicated by
the sharp rise in cell voltage) when charge rate is increased.
(Reprinted with the permission of Gates Energy Products,
employed with sealed batteries to return full capacity
as quickly as possible.
To maintain capacity on a fully charged battery, a
constant voltage is applied. The voltage must be
high enough to compensate for self discharge, yet
not too high as to cause excessive over-charging.
While simply maintaining a fixed output voltage is a
relatively simple function, the battery’s temperature
coefficient of -3.9mV/degree C per cell adds
complication. If battery temperature is not
compensated for, loss of capacity will occur below
the nominal design temperature, and over-charging
with degradation in life will occur at elevated
Charging Algorithm
To satisfy the aforementioned requirements and
thus provide maximum battery capacity and life, a
charging algorithm which breaks the charging cycle
down into four states is employed. The charging
algorithm is illustrated by the charger state diagram
shown in figure 2. Assuming a fully discharged
battery, the charger sequences through the states
as follows:
1. Trickle-charge If the battery voltage is below a
predetermined threshold, indicative of a very
deep discharge or one or more shorted cells, a
small trickle current is applied to bring the
battery voltage up to a level corresponding to
near zero capacity (typically 1.7V/[email protected] 25
degrees C). Trickle charging at low battery
voltages prevents the charger from delivering
high currents into a short as well as reducing
excessive out-gassing when a shorted cell is
present. Note that as battery voltage increases,
detection of a shorted cell becomes more
2. Bulk-charge Once the trickle-charge threshold
is exceeded the charger transitions into the
bulk-charge state. During this time full current is
delivered to the battery and the majority of its
capacity is restored.
3. Over-charge Controlled over charging follows
bulk-charging to restore full capacity in a
minimum amount of time. The over-charge
voltage is dependent on the bulk-charge rate as
illustrated by figure 1. Note that on unsealed
batteries minimal over-charging should be
Figure 2. The charging algorithm is broken down into four
employed to minimize out-gassing and
subsequent dehydration. Initially overcharge
current is the same as bulk-charge current. As
the over-charge voltage is approached, the
charge current diminishes. Over-charge is
terminated when the current reduces to a low
value, typically one-tenth the bulk charge rate.
4. Float-Charge To maintain full capacity a fixed
voltage is applied to the battery. The charger
will deliver whatever current is necessary to
sustain the float voltage and compensate for
leakage current. When a load is applied to the
battery, the charger will supply the majority of
the current up to the bulk-charge current level.
It will remain in the float state until the battery
voltage drops to 90% of the float voltage, at
which point operation will revert to the bulk
charge state.
Charger Circuit Design
There are many possible circuit configurations which
will provide the necessary control and output
charging current. For efficient operation, particularly
at higher output currents, switching power circuitry
is preferred. To minimize cost as well as complexity
each IC used must provide as much functionality as
possible. A circuit topology was chosen which
utilizes two special purpose ICs and a general
purpose op-amp to provide all of the control
functions, while a discrete MOSFET output stage
handles the power. The circuit design is modular to
simplify modification for different application
The charger circuit can be divided into three basic
blocks. The first is the voltage loop control and state
control logic which executes the control algorithm
while providing temperature compensation. The
second is the switchmode controller which regulates
the current to the battery as commanded by the
voltage loop control and state control logic. The third
is the output power stage which is sized to efficiently
deliver the charging current.
Voltage Loop Control and State Control Logic
Initially designed for charging small lead-acid
batteries using a linear pass transistor for current
control, the UC3906 directly implements the voltage
loop control and state control logic while providing
the appropriate temperature compensation. The
block diagram of the UC3906 is shown in figure 3.
Battery voltage is monitored with a resistor divider
string. This network establishes the float voltage, the
over-charge voltage, and the trickle-charge
threshold voltage by comparing to the precision
temperature compensated reference. Since
temperature is monitored on chip it is critical that the
battery and the UC3906 are in close proximity, and
Figure 3. UC3906 Lead-Acid Battery Charger block diagram
that self-heating or heating from other components
is minimized.
The differential current sense comparator is used to
terminate over-charging and transition to the float
state. The voltage amplifier provides gain and
compensation for the voltage loop. The UC3906 is
covered in detail in reference [3].
Switchmode Current Source
The charging algorithm places great demands on
the current loop. during bulk charge full current must
be supplied, yet during the float state the current
draw may be only a few milliamps. This equates to
a dynamic range in excess of 60 dB which can be
very difficult to achieve with common peak current
mode techniques. The wide dynamic range also
requires operation with both continuous and
discontinuous inductor current, potentially adding
complication to voltage loop stabilization. Although
load resistors can be employed to reduce the
required dynamic range, their use can significantly
degrade efficiency, particularly while in the float
state. Note that a high value load resistor (10 k) is
employed to assure operation down to zero output
current and to provide a discharge path for the output
capacitor. Additionally, to provide precise bulk and
trickle-charge current levels the closed current loop
transconductance must be accurate. Average
current feedback will circumvent these potential
problems, and is the key to a successful
implementation of the switching current source for
this application.
Figure 4 shows the basic implementation of average
current feedback. While slightly more complicated
than typical peak current mode control schemes,
average current feedback offers several critical
performance enhancements. The high gain of the
error amplifier at lower frequencies provides high
closed current loop accuracy and accommodates
the large output stage nonlinearity which occurs
when the inductor current becomes discontinuous.
Good switching spike noise immunity is inherent with
this technique permitting stable operation at narrow
duty cycles.
A UC3823 PWM controller shown in figure 5 was
chosen for the current loop control circuit for several
reasons. First and most importantly it is capable of
operating linearly from very small duty cycles to near
Figure 4. Average Current Feedback Loop
100% duty cycle. Secondly the error amplifier
bandwidth and configuration are well suited to the
average current loop’s requirements. Additionally,
the output driver affords a simple interface to most
discrete output power stages.
A separate op-amp configured as a differential
amplifier senses the output current and level shifts
the signal to the appropriate voltage. The offset and
common mode rejection of this amplifier are the
major source of current loop error.
Output Power Stage
To simplify development a simple buck regulator
output stage was used. For further simplicity the
high-side switch is implemented using a direct
coupled P-channel MOSFET. A switched current
sink provides gate charge, turning the MOSFET on
while a zener diode limits the gate to source voltage
to 12 volts. A second emitter switched current sink
drives a PNP which removes gate charge, turning
the MOSFET off. Undoubtedly this output stage is
suitable for many applications, although higher
power capability and efficiency can be achieved
using N-channel devices. A relatively low value
output inductor was chosen to minimize size and
cost since operation in the discontinuous current
mode is of no concern with average current
feedback. Output ripple voltage is also not critical so
the output capacitor was selected for ripple current
capability. High frequency ringing caused by circuit
parasitics is damped with a small RC snubber across
the catch rectifier. A rectifier in series with the output
Figure 5. UC3823 High speed PWM Controller Block Diagram
prevents the battery from back driving the charger
when input power is disconnected.
Complete Charger Circuit
A complete schematic for the switch-mode charger
is shown in figure 6. Control circuit power is supplied
from an emitter follower off a zener shunt regulator.
The PWM frequency is set to 100 kHz as a
reasonable compromise between output filter
component size and switching loss. Output current
is sensed in the battery return lead to minimize
common mode voltage errors. This arrangement
also allows direct current sensing for pulse by pulse
current limiting adding further protection during
abnormal conditions. The differential amplifier is set
to a gain of 5 with the output signal referenced to the
UC3823s 5.1 V reference.
The current feedback signal is summed with the
current command signal at the error amplifier’s
inverting input. To accommodate worst case offset
in both the error amplifier and the differential
amplifier and allow zero output current, the
non-inverting input of the error amplifier is biased
130 mV below the 5.1 V reference. Trickle bias is
accomplished by injecting a small current into the
differential amplifier’s negative op-amp input, thus
causing a proportional output current to balance the
loop. Additionally, a 100 pF capacitor across the
PWM comparator inputs enhances noise immunity,
particularly at low duty cycles.
For maximum control and float voltage accuracy, the
UC3906s ground is connected to the battery’s
negative terminal, thereby rejecting the current
sense resistors voltage drop. The internal emitter
follower output transistor interfaces to the current
source as illustrated in figure 7. The voltage amplifier
drives the output current command signal. The
current command signal is limited by clamping the
voltage amplifier output through a diode to 4.2 V. The
clamp also prevents the emitter follower from
Figure 6. Switchmode Lead-Acid Battery charger schematic
frequency is set such that the amplified inductor
current down-slope is less than the oscillator-ramp
up-slope as seen by the PWM comparator. By
setting the two slopes equal under worst case
conditions (at maximum output voltage) maximum
closed loop bandwidth is achieved without
subharmonic oscillation.
Placing a zero below the minimum loop crossover
frequency significantly boosts low frequency gain
while a pole placed above the maximum crossover
frequency enhances noise immunity. Note that since
loop response is not particularly critical for battery
charging, conservative compensation with plenty of
phase margin is normally employed.
Figure 7. The UC3906’s output transistor provides the
interface to the switchmode current source.
saturating which would cause a large difference
between collector and emitter currents due to
excessive base drive.
Battery voltage is sensed by the resistor divider
string, with the values shown for a typical 24 V (12
cell) application. Other battery voltages are easily
accommodated by simply changing the divider
values using the procedure presented in the
UC3906 data sheet, although changes in input
voltage may require modification of the output circuit
and the control circuit power supply. The resistor
divider establishes all of the state transitions with the
exception of over-charge terminate, which is
determined by detecting when the output current
has tapered off to approximately one-tenth the bulk
charge level. This is accomplished by the UC3906s
current sense comparator which senses the
appropriately scaled signal from the differential
amplifier output.
Current and Voltage Loop Compensation
The charger circuit implements a two loop control
system with the current loop operating inside the
voltage loop. During trickle-charge, bulk-charge and
the beginning of over-charge the voltage loop is
saturated and the current loop is essentially driven
from a fixed reference.
With continuous inductor current the control to
output gain of the current loop shown in figure 4
exhibits a single pole response from the output
inductor. The error amplifier gain at the switching
When inductor current becomes discontinuous, the
power circuit gain suddenly drops, requiring large
duty cycle changes to significantly effect output
current. The single pole characteristic of continuous
inductor current with its 90 degree phase lag
disappears. The current loop becomes more stable,
but less responsive. Fortunately the high gain of the
error amplifier easily provides the large duty cycle
changes necessary to accommodate changes in
output current, thereby maintaining good average
current regulation.
The block diagram of the voltage loop is shown in
figure 8. With an inner transconductance loop the
control to output gain of the voltage loop exhibits a
single pole response from the output capacitor and
equivalent load resistance. While it may initially
appear that a simple fixed gain on the voltage
amplifier would provide suitable loop compensation,
further examination shows a severe drop in voltage
gain at high loads, which would drastically reduce
DC accuracy. A zero is placed in the voltage
amplifier’s transfer function to boost low frequency
gain and therefore restore DC accuracy.
The current loop’s single pole response above its
crossover frequency cancels the output stage zero
resulting from the output capacitor’s capacitance
and ESR. Note again that since wide bandwidth is
not required for battery charging, the voltage loop
crossover frequency is well below both the current
loop’s pole and the output capacitor’s zero. Low
leakage capacitors must be used for the
compensation network to maintain high DC gain
since the voltage amplifier is a transconductance
type. Loop stabilization is covered extensively in
references [1] and [2].
Charger Performance Summary
The charger circuit properly executes the charging
algorithm, exhibiting stable operation regardless of
battery conditions including an open circuit load. The
circuit was tested with 6, 12 and 24 V batteries by
modifying only the battery voltage sense divider. As
would be expected, circuit efficiency was best at high
battery voltage, approaching 85% while
bulk-charging a 24 V battery with a 40 V input supply
An analysis of circuit losses indicates several areas
where efficiency could be improved. Any accuracy
and offset improvement in the differential amplifier
will allow a corresponding decrease in current sense
resistor value and hence dissipation, while
maintaining the same overall current loop accuracy.
Replacing the output blocking rectifier with a
Schottky would save a few watts if the Schottky’s
leakage could be tolerated. Further improvement
could be made in that area by using a relay to
disconnect the charger when input power is
removed. A more conservative inductor design with
less resistance would save a little over one watt. As
expected, the greatest losses occur in the output
switch. A lower on resistance FET and a higher peak
current gate drive to reduce switching losses could
save more than 5 watts. Incorporating a few of these
improvements will easily increase circuit efficiency
to greater than 90%.
Alternate Circuit Configurations
While the charger circuit as designed may be
suitable for many applications, a few modifications
should satisfy the majority of additional
requirements. Higher voltage batteries can be
charged by designing a higher voltage output stage.
N-channel MOSFETs are preferable for cost and
efficiency reasons, but are more difficult to drive than
P-channels. Fortunately, the remainder of the circuit
will require minimal modification.
Some applications may require both the battery and
charger to share a common ground and thus prohibit
current sensing in the batteries negative return. The
differential amplifier can sense current at the
inductor output if tighter tolerance resistors to
improve CMRR are used. While this simple
modification renders a suitable signal for closing the
current loop, another current sense signal
referenced to ground must be developed for pulse
by pulse current limiting. This signal is most easily
derived by using a PNP level shift transistor,
connecting the base to the 5.1 V reference and the
emitter through a resistor to the differential amplifier
At higher battery voltages it may be desirable to float
with a current rather than a voltage. Varying
self-discharge rates of individual cells in high voltage
batteries causes inevitable differences in cell charge
levels. By employing a float current and applying a
small continuous overcharge, variation of charge
between cells is minimized. Precise output at float
current levels places great demands on current loop
accuracy, and will add unnecessary expense to the
current sensing circuitry. A more cost effective
alternative is to use a fixed linear current source
which should be small and inexpensive considering
the very low output current.
Thus far the input supply has not been addressed
and is assumed to be from a voltage required
elsewhere in the system or from a typical line
frequency transformer, rectify bridge and filter
capacitor. This may represent more than half the
cost of the charger, and is certainly the majority of
its size and weight. An obvious alternative is to
replace the buck output section with a transformer
coupled output, taking advantage of the switching
control circuit already present. Buck derived circuits
such as forward, half-bridge and full-bridge easily
interface with the existing design, however resonant
and flyback circuits are also applicable. A small (0.75
W) auxiliary supply will be required to power the
control circuitry since the modulator will output zero
at times, prohibiting the use of a bootstrap winding
commonly used on switching power supplies. This
Figure 8. Voltage Control Loop block diagram
approach is particularly cost effective for
stand-alone applications, allowing the design of a
compact, light weight, high performance charger.
A practical switchmode lead acid battery charger
circuit has been presented which incorporates all of
the features necessary to assure long battery life
with rapid charging capability. By utilizing special
function ICs, component count is minimized,
reducing system cost and complexity. With the
circuit as presented, or with its many possible
variations, designers need no longer compromise
charging performance and battery life to achieve a
cost effective system.
1. L. Dixon, “Average Current Mode Control of
Switching Power Supplies”, Unitrode Power
Supply Design Seminar, SEM700, Topic 5
2. L. Dixon, “Closing the Feedback Loop”,
Unitrode Power Supply Design Seminar,
SEM700, Section C
3. R. Valley, “Improved Charging Methods for
Lead-Acid Batteries Using the UC3906”,
Unitrode Linear Integrated Circuits Data and
Applications Handbook, IC600
4. L. Woffard, “New Pulse Width Modulator Chip
Controls IMHz Switchers”, Unitrode Linear
Integrated Circuits Data and Applications
Handbook, IC600
TEL. (603) 424-2410 l FAX (603) 424-3460
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