User manual | H DELAY l

United States Patent [191
[11]
[45]
Powell
[54] SOLID STATE ELECTRICAL MUSICAL
4,581,587
Apr. 8, 1986
instrument signal input, a preampli?er, and circuitry in
INSTRUMENT AMPLIFIER
[76] Inventor:
Patent Number:
Date of Patent:
association with the preampli?er and preferably in a
feedback loop of the preampli?er, for adjusting the
frequency compensation of the preampli?er. The ampli
?er may also include distortion circuitry, again prefera
Brent L. Powell, 310 W. 9th St., St.
Anthony, Id. 83445
[21] Appl. No.: 607,934
[22] Filed:
May 7,1984
bly located in a feedback loop of the preampli?er,
[51]
[52]
Int. Cl.4 ......................... .. H03F 1/34; H03F 3/08
US. Cl. .................................... .. 330/107; 330/59;
maintains the relative even and odd harmonic content
[58]
Field of Search ............... .. 330/59, 107, 109, 149,
[56]
330/294, 302, 308; 328/167; 307/521, 522, 271
References Cited
which compresses the peaks of the input signal thereby
giving a distorted output of the preampli?er which
of the input. The ampli?er may also include a variable
band reject ?lter operated by a sweep rate generator
which varies the light output of a light emitting diode
which in turn varies the resistance of photoresistors in
330/109; 330/294; 330/302
the variable band reject ?lter. If signal delay circuitry is
included to create a reverberation effect, the ampli?er
U.S. PATENT DOCUMENTS
4,107,622 8/ 1978
4,151,477 4/1979
4,179,669 12/1979
also includes a high pass ?lter in parallel with the delay
circuitry to increase the high frequency content of the
output signal. In addition, a modi?ed power ampli?er
may be included with an integrated circuit operational
ampli?er input D.C. coupled to the rest of the circuit.
Toyomaki ..................... .. 330/294 X
Yokoyama
..... .. 330/294 X
Dodson et a1. ................. .. 330/59 X
Primary Examiner-Eugene R. LaRoche
Assistant Examiner-Steven J. Mottola
Attorney, Agent, or Firm-Mallinckrodt & Mallinckrodt
[57]
ABSTRACT
A solid state electrical instrument ampli?er includes an
20 Claims, 8 Drawing Figures
l7
ms'rnuusnr
INPUT
wrm
VARIABLE
POWER
TONE
COMPENSATION
AND DI$TORTION
FREAMP
VOLUME /‘"
AND
CONTROL
TE
uznemi 1 ON
POWER
SUPPLY
AMP
HIGH PASS
FILTER
H DELAY l
CIRCUITRY
AMPLIFlED
SIGNAL
OUT TO
SPEAKER
1
4,581,587
2
the signal from the instrument pickup. The initial pre
SOLID STATE ELECTRICAL MUSICAL
ampli?cation stage is where the frequency compensa
INSTRUMENT AMPLIFIER
tion takes place. Thus, the tone controls modify the
compensated signal and do not act to modify the fre
BACKGROUND OF THE INVENTION
5 quency compensation of the initial pre-ampli?cation
stage. Further, the tone controls are generally tied in
1. Field
with the volume controls so that at high volume levels,
The invention is in the ?eld of ampli?ers for musical
the tone controls are almost nonfunctional.
instruments, particularly instruments such as electric
guitars and electric basses.
Frequency compensation is not as important where a
piezoelectric pick-up or a dynamic microphone pick-up
2. State of the Art
With electrical instruments such as electric guitars
and electric basses, the characteristics of the sound
are used. These pick-ups are generally substantially
produced are not as dependent upon the instrument
ampli?cation. With these pick-ups, the tone controls, if
they provide a :10 db tone compensation, will usually
be suf?cient to give any desired frequency compensa
linear in their output, so require a substantially linear
itself as upon the instrument’s pickup and ampli?er. The
particular pickup used determines the relative ampli
tude of various frequency signals sent from the instru
ment to the ampli?er. The signal processing done by the
ampli?er determines the characteristics of the ampli?ed
signal, and hence its audio characteristics when the
tion. However, as pointed out above, while such tone
controls may operate satisfactorily at lower volume
levels, such tone controls may not provide the desired
compensation at higher volume levels (those generally
electrical signal is converted to an audio signal by a
used with electric instruments). Thus, even with piezo
electric pick-ups, some frequency compensation is usu
ally required or desired.
speaker.
There are currently many instrument ampli?ers on
the market and each produces a somewhat different
In addition to frequency compensation in ampli?ers,
instrument sound. The difference may be small between
it
is sometimes desired to introduce changes into the
25
some ampli?ers and large between others.
signal to give it a special, modi?ed sound. One popular
It is not unusual today for a versatile performer to
modi?cation is commonly referred to as “distortion”. In
have several instruments and several ampli?ers and use
present ampli?ers, this is generally achieved by clipping
one or another depending upon the desired type of
the sign wave music signal to form a wave similar to a
music to be played along with its characteristic sound.
square wave. This produces what is commonly called
For example, a performer playing rock music generally
“fuzz tone”. The problem with this procedure is that a
square wave emphasizes the odd order harmonics and
desires a hard, driving sound which is characterized by
an emphasis on treble volume. On the other hand, a
performer playing jazz desires a more mellow sound
which is characterized by an emphasis on bass volume,
but still producing sharp clear sounds. A performer
playing country music will generally prefer an in-be
cancels the even order harmonics in the signal. The
cancelling of the even order harmonics produces a
harsh and unpleasant sound. It has been found that the
most desirable “distortion” signal is one that keeps both
tween sound.
the even and odd harmonics in about the same propor
and one that results in the largest difference in the sound
tion as in the undistored signal.
Another effect often incorporated into an ampli?er is
produced by the ampli?er, is the frequency response of
the “phase shifting” effect, which produces an ethereal
One of the principal differences between ampli?ers,
sweeping, swishing sound that seems to surround the
listener. This can also produce the effect of a rotating
speaker or vibrato. To create this effect, the frequency
of a particular sound is ?rst summed to an original fre
nonlinear, meaning that signals of different frequency
produce different amplitude or strength signals. The 45 quency and then is gradually subtracted or nulled.
the ampli?er. Most electric guitars and electric basses
use a wound magnetic type of pick-up to convert string
vibrations into electrical signals. These pick-ups are
lower frequency vibrations produce the strongest sig
While the currently known circuitry for achieving this
nal, and the strength of the electrical signal decreases as
the frequency increases. To make up for this difference,
it is necessary that the ampli?er apply what is com
effect is generally satisfactory, room remains for im
provement in the circuitry both from the standpoint of
simplicity and operation.
A still further effect usually incorporated into ampli
monly called frequency compensation. In order that all
?ers is the so-called “rever ” effect, which results from
the combination of a signal with a similar, but delayed
signal. This produces an echo type of sound. The nor
frequencies of an instrument signal are included in the
audio signal produced by the speaker in substantially
the strength they are produced by the instrument, it is
necessary to compensate for the lower amplitude high
frequency signals produced by a magnetic pick-up by
55
mal delay lines used in musical instrument ampli?ers
substantially reduce signals above six kHz thereby re
adjusting the ampli?er so that it ampli?es the high fre
ducing substantially the higher frequency signals in the
quency signals to a greater degree than the low fre
combined signal. No effective way to compensate for
quency signals. The amount of frequency compensa
this reduction in high frequency signals when using a
reverb unit is currently available.
tion, where it occurs, and what frequencies are compen
sated to what degree are usually designed into an ampli 60 It would thus be desirable to have an instrument
?er and are the principal reasons for the substantial
ampli?er where the frequency compensation of the
differences in the sound of the output from ampli?er to
ampli?er.
initial pre-ampli?er could be adjusted by the user to
adjust for differences in pick-ups being used and to give
Most ampli?ers have tone controls which modify the
an adjustable desired type of sound to a particular in
frequency response of the ampli?er to some extent upon 65 strument. It would also be desirable to have an ampli?er
adjustment of the controls by the user. These controls
where the tone control would operate over its full range
generally take the form of either high pass or low pass
filters in the circuit after the initial pre-ampli?cation of
regardless of the volume setting, and where the distor
tion circuit would not unduly cancell even harmonics of
3
4,581,587
the signal. Further, it would be desirable to have an
ampli?er with improved “phase shifting” circuitry and
impedance and may be D.C. coupled to the rest of the
circuitry. This makes it possible to have a completely
high frequency compensation for use in conjunction
with “reverb” circuitry.
D.C. coupled ampli?er.
THE DRAWINGS
In the accompanying drawings, which illustrate an
embodiment of the invention constituting the best mode
SUMMARY OF THE INVENTION
According to the invention, a solid state electrical
instrument ampli?er includes an instrument signal input
presently contemplated for carrying out the invention
in actual practice:
with means for the musician to adjust the frequency O
FIG. 1, is a block diagram of an ampli?er for electri
compensation of the preampli?er, and signal output
cal musical instruments which is built according to the
means such as a standard phono jack, a preampli?er
means, such as terminals to connect to a power ampli
invention;
?er. The ampli?er will generally include the power
FIG. 2, a circuit diagram, partially in block form,
showing the circuitry of the preamp with frequency
compensation and distortion block of FIG. 1;
FIG. 3, a circuit diagram showing the circuitry of
FIG. 2 with the circuitry of the variable frequency
compensation block of FIG. 2, for ease of explanation,
not showing the distortion circuitry;
FIG. 4, a circuit diagram showing the circuitry of
FIG. 2 with the circuitry of the distortion circuitry
block, but, for ease of explanation not showing the
ampli?er and may include such additional items as dis
tortion means for distorting the signal, band reject ?lter
means for creating ethereal sweeping, swishing sounds,
signal delay means for creating reverberation effects,
5
and volume and tone controls.
In a preferred embodiment of the invention, the pre
ampli?er takes the form of an ampli?er having negative 20
feedback, with the frequency compensation means lo
cated in the negative feedback loop. Such a preampli?er
variable frequency compensation circuitry;
may take the form of an integrated circuit operational
ampli?er connected as a noninverting ampli?er. A feed
back resistor is connected between the output of the
FIG. 5, a circuit diagram of the preamp with fre
quency compensation and distortion, preamp, variable
band reject ?lter, and sweep rate generator blocks of
operational ampli?er and its inverting input. A variable
resistance-capacitance network is connected between
the inverting input and ground as part of the negative
feedback loop. Adjustment of the resistance-capaci
FIG. 1;
FIG. 6, a circuit diagram of the volume and tone
control, high pass ?lter, delay circuitry, and summing
tance network varies the frequency compensation of the
amp blocks of FIG. ll;
FIG. 7, a circuit diagram of the power amp block of
FIG. 1; and
preampli?er. By providing the frequency compensation
means in the feedback loop of the preampli?er, the
frequency compensation may be varied by the musician
FIG. 8, a graph showing the frequency vs. gain char
acteristics of the circuitry of FIG. 3.
over a wide range without substantially changing the
input or output impedence of the preampli?er. This
DETAILED DESCRIPTION OF THE
ILLUSTRATED EMBODIMENT
FIG. 1 shows a block diagram of the overall circuitry
of an ampli?er incorporating all of the features of the
current invention. Thus, the signal from an electrical
means that the frequency compensation can be varied
over a wide range without changing other characteris
tics of the ampli?er.
A preferred embodiment of the invention also in
cludes distortion circuitry located in the preampli?er
negative feedback loop. The distortion circuitry oper
instrument such as an electric guitar or an electric bass
ates to compress the peaks of the input signal and pro
vide a distored output signal of the preampli?er which
is connected to a preampli?er 110 which has variable
frequency compensation and adjustable distortion. The
still substantially maintains the same relative even and
odd harmonic content of the input signal. Such circuitry
may take the form of two parallel diodes with opposite
orientations connected in series with a resistor, the en
tire diode resistor combination being connected in par
allel with the feedback resistor between the output and
inverting input of an operational ampli?er.
The ampli?er of the invention may also include band
reject ?lter means connected between the preampli?er
and the signal output means wherein the reject band of
the ?lter is determined by the resistance of two photore
45
output of the preampli?er is a signal which has been
compensated to any desired degree for the nonlinearity
of the instrument pickup and also to emphasize treble or
bass frequencies. Further, if desired, the signal has been
distorted to a desired degree by compressing it to pro
vide an effect similar to that commonly known as “fuzz
tone”, but much more pleasing because even order
harmonics are not cancelled from or substantially re
duced in the signal. The output of the preampli?er is
connected directly to the volume and tone controls 11
and also to a variable band reject ?lter 12 which can be
sistors. A sweep rate generator, such as a triangle wave 55 operated, if desired, to produce the special effect of an
ethereal sweeping, swishing sound that seems to sur
round the listener, or a vibrato sound. The effect is
generator varies the intensity of the illumination of at
least one light emitting diode which is located to vary
the illumination on, and thus the resistance of, the pho
toresistors of the band reject ?lter.
The ampli?er may also include delay circuitry to
delay a portion of the signal from the preampli?er and
then add it to the nondelayed signal to create a “reverb”
effect. With such delay circuitry, a high pass ?lter is
included in parallel with the delay circuitry to increase
the high frequency content of the output signal.
The ampli?er may also include a power ampli?er
having an integrated circuit operational ampli?er as an
input stage so that the power ampli?er has a high input
produced by sweeping the rejection frequency of the
?lter over a range of frequencies and by varying the
sweep frequency of the ?lter. The sweep rate of the
?lter is controlled by the sweep rate generator 12. The
signal from the variable band reject ?lter is summed at
the volume and tone controls with the signal directly
from the preampli?er to give the desired effect.
The volume and tone controls 11 are standard and the
resulting output signal is then divided with the signal
going directly to a summing ampli?er 14, to a high pass
?lter 15 for further treble tone control, and to delay
5
4,581,587
circuitry 16 where it is delayed when desired to create
a “reverb” effect. The signals, if any, from the high pass
?lter and the delay circuitry are summed with the signal
6
passes to ground. This reduces the amount of negative
feedback to the operational ampli?er and increases the
overall gain of the ampli?er. As the signal frequency
decreases, the capacitive resistance of the R-C circuit
increases thereby causing an increase of negative feed
as it comes from the tone and volume controls in the
summing ampli?er 14.
The output of the summing ampli?er is further ampli
?ed in the power ampli?er 17 and supplied to the speak
back to the ampli?er and a decrease in the overall gain.
Thus, high frequencies have greater gain than low fre
quencies, the speci?c gain characteristics being deter
ers.
A second preampli?er 18 is provided for input signals
mined by the speci?c R-C circuit used.
from sources not needing variable frequency compensa 0
In the circuitry of the invention, the circuit functions
tion or added distortion and the output of that preampli
as with a ?xed R-C circuit, but the R-C circuit is vari
?er is connected to the output of preampli?er l0 and is
able. Thus, by changing the adjustment of variable resis
connected to the remaining blocks of the circuitry as
tor VR1, the value of the R-C circuit is changed. With
indicated above.
variable resistor VR1 adjusted so that the wiper is at
A power supply 19 supplies power to the circuitry
one limit of its travel (the far left in FIG. 3), the effec
while power supply 20 supplies power to the power
tive impedance of the R-C circuit is the parallel combi
ampli?er.
nation of the series connection of capacitor C1 and
Where a power ampli?er is included in the circuitry,
resistor R3 in one parallel branch and variable resistor
the signal output means will generally take the form of
VR1, capacitor C2 and resistor R4 in the other parallel
terminals for connection of audio speakers. If a power 20 branch. With the wiper of variable resistor VR1 at the
ampli?er is not included, the signal output means will
other limit of its travel (the far right in FIG. 3), the
generally take the form of terminals for the connection
effective impedance of the R-C circuit is the parallel
of a separate power ampli?er or of other equipment
combination of the series connection of variable resistor
such as recording equipment.
VR1, capacitor C1 and resistor R3 in one parallel
FIG. 2 shows a block diagram of the preampli?er 25 branch and capacitor C2 and resistor R3 in the other
with frequency compensation and distortion as indi
parallel branch. These two positions of VR1 determine
cated in block 10 in FIG. 1. This shows the preampli?er
the two extremes of frequency compensation, with set
as an integrated circuit operational ampli?er IC1. The
tings of variable resistor VR1 between its extremes
frequency compensation is achieved by variable fre
giving intermediate settings of frequency compensation.
quency compensation circuitry 21 connected between 30 The graph of FIG. 8 shows variations in frequency
ground and the inverting input of operational ampli?er
compensation for a typical circuit of FIG. 3, the line 30
IC1 as part of the feedback loop. The distortion is ob
shows the gain vs. frequency characteristics at one
tained by distortion circuitry 22 connected in the feed
extreme setting of variable resistor VR1 and the line 31
back loop of IC1 in parallel with normal feedback resis
shows the gain vs. frequency characteristics at the other
tor R1. If only variable frequency compensation or 35 extreme setting. The area between the lines show the
distortion is needed in connection with the preampli?er,
range of intermediate settings. With this arrangement
only the desired circuit need be used. Both circuits need
for varying frequency compensation, the input and
not be used together, and the distortion circuit will
output impedance of the overall preampli?er circuit
generally be switched in or out of the circuitry as de
remains substantially constant as the frequency equal
sired. An output isolation resistor R2 is connected in the 40 ization is changed. The output of IC1 is connected
output of IC1 as shown.
through output resistor R2 and variable resistor VR2 to
Since the variable frequency compensation circuitry
ground. Variable resistor VR2 is shown for explanation
and the distortion circuitry are separate, for ease of
illustration and explanation, the preampli?er circuitry
incorporating only variable frequency compensation is
shown in FIG. 3, while the preampli?er circuitry incor
porating only distortion circuitry is shown in FIG. 4.
The combined preampli?er circuitry is shown in FIG.
5, along with additional circuitry of the invention.
Referring to FIG. 3, the input, which is the instru
ment signal input from an electrical instrument such as
a guitar, bass, or synthesizer, is connected to the nonin
verting input of operational ampli?er IC1 with IC1
connected in normal fashion as a noninverting ampli?er.
purposes and represents the volume control of the am
45
pli?er and any other resistances between the output of
IC1 and ground. Thus, the DC bias path for the invert
ing input of IC1 is set by resistors R1 and R2 and vari
able resistor VR2, however, in terms of bias, VR2 is of
substantially constant resistance. In the actual circuitry
of FIGS. 5 and 6, the resistance represented by VR2
will vary to some degree but for purposes of explana
tion of the frequency compensation and distortion cir
cuitry, it is shown as substantially constant. Thus, the
DC bias on the inverting input of IC1 remains substan
tially constant regardless of the setting of the frequency
compensation. The DC bias for the noninverting input
With such ampli?ers, a feedback loop is provided from
the output of IC1 through resistor R1 back to the in
verting input of IC1. A resistance is also provided be
tween the inverting input of IC1 and ground. Here the
strument pickup is connected to the input, the signal
resistance is made up of the combination of variable
source shunts resistor R5 and the resulting DC bias
of IC1 is set by resistor R5 when the instrument input is
open, but when a signal source such as a magnetic in
resistor VR1, resistors R3 and R3 and capacitors C1 and 60 impedance very closely matches the value of R1 to
C2. While it is common to provide an R-C circuit be
provide a very low output offset voltage of typically
tween the inverting input of an operational ampli?er
less than 20 mv. Under this condition, DC coupling of
connected as a noninverting ampli?er and ground to
other subsequent circuitry to this preampli?er stage is
provide frequency compensation in the gain of the am
possible. Thus, as explained above, the bias condition on
pli?er, such circuit always uses a ?xed R-C circuit. In 65 IC1 remains substantially constant regardless of the
such instance, as the frequency of the input and output
setting of frequency equalization and the offset voltage
signal increases, the capacitive resistance of the R-C
of IC1 is substantially constant. In this regard, however,
circuit is reduced so that more of the feedback signal
it has been found that it is important to keep the value of
7
4,581,587
8
the variable impedance created by the combination of
resistors R1 and R2 and variable resistor VR2. R5 is a
variable resistor VR1, resistors R3 and R4 and capaci
tors C1 and C2 less than about one-tenth the resistance
value of R1. When greater than about one-tenth the
bias resistor for the noninverting input to IC1 when the
input is open. When connected to an instrument, the
DC bias for the noninverting input will generally be
resistance of R1, the offset voltage IC1 will begin to
through the instrument pickup.
vary.
The gain of the ampli?er is provided by the following
equation:
The distortion circuit is connected in parallel‘ with
feedback resistor R1 and is made up of diodes D1 and
D2, variable resistor VR3, and switch SW1. With
switch SW1 open, the distortion circuitry is not in the
10 preampli?er circuitry and has no effect on the preampli
gain =
?er operation. With switch SW1 closed, and variable
resistor VR3 set at its maximum resistance, the voltage
across the parallel diodes D1 and D2 is of insuf?cient
where R’s represent the resistance values of the indi
magnitude to cause forward conduction of the diodes.
cated resistors, VR1 represents the resistance value of
However as the resistance of VR3 is reduced, the volt
VR1, and XCl and XC2 represent the AC resistance
age drop across the parallel diode combination begins to
values of the indicated capacitors at a particular fre
increase until forward conduction occurs. The amount
quency. Further, the above equation is for variable
of negative feedback is limited by the resistance of resis
resistor VR1 with its wiper at the extreme left in FIG.
tor R1 which is chosen so that the voltage drop between
3 so that all of the resistance of VR1 is in sereis with C1
the output of IC1 and the inverting input of IC1 is never
and R3. For the other extreme setting of VR1, VR1 has
large enough to bring the forward conductance of the
to be moved from the association with R3 and XC1 to
parallel diodes to their minimum resistance. The diodes
similar association with R4 and XC2. Thus, at the other
can be controlled from full off to 90% conduction by
extreme:
varying variable resistance VR3. As indicated, the
25
value of resistor R1 is selected to achieve a voltage drop
Rl
gain =
It has been found that for use with magnetic or piezo
electric guitar or bass pickups or for use with music
synthesizers, the following component values for the
. circuitry of FIG. 3 give excellent results: R1 - 68K
ohms, R2 - 2.2K ohms, R3 - 10k ohms, R4 - 1.8K ohms,
R5 - 1.5 meg. ohms, VR1 - 25K ohms linear taper poten
which will not allow the diodes to be turned fully on.
The effect of full conduction would result in an output
from the preampli?er of a modi?ed square wave which
cancells even order harmonics and results in a signal
containing predominantly odd order harmonics. The
optimum value of resistor R1 will limit the on resistance
of the diodes to approximately 70% of their full on
resistance. By using the nonlinear portion of the for
ward resistance of the diodes, i.e. the area between
tiometer, VR2 - 100k ohms linear taper potentiometer, 35
about 0.45 volts and 0.55 volts, a voltage averaging
C1 - 0.68 microfarads, C2 - 0.033 microfarads, and IC1
circuit is achieved. With resistor R1 in parallel with the
.. a Texas Instruments TL094CN linear operational am
distortion circuit, the net output of IC1 is close in form
. pli?er or a National Semiconductor LM 348N opera
to a compressed sine wave. The peaks of the output
tional ampli?er.
signal are compressed, but not clipped. This results in an
Using the above gain equation and component values
output signal having substantially equally reduced even
for an arbitrary condition of input signal of 100 mv at
and odd order harmonics and thus still has substantially
500 Hz, the gain at one extreme of VR1 works out to
16.2 db while at the other extreme of VR1 works out to
the same balance of even and odd order harmonics in
The particular circuitry shown is designed for input
linear taper potentiometer, R6 - 3.9k ohms, C3 - 0.033
microfarads, D1 and D2 - 1N4002 diodes.
the distorted signal as in the original signal. This gives a
15.5 db. In similar fashion, gain calculatons can be made
much more pleasing sound than a clipped signal which
45
at various frequencies and the results shown in a graph
tends to cancell even order harmonics. The amount and
as FIG. 8. Line 30 in FIG. 8 represents the frequency
intensity of the distortion effect can be adjusted by
vs. gain curve when VR1 is set so that it is in series with
adjusting variable resistor VR3.
C1 and R3 and line 31 represents the frequency vs. gain
For the circuitry shown, the following component
curve when VR1 is set so that it is in series with C2 and
50 values have been found satisfactory: VR3 - 50k ohms
R4.
frequencies between 20 Hz and 20 kHz and an average
input signal amplitude of about 100 mv. Above 20 kHz,
the circuitry may become unstable and oscillate. There
fore, if it is to be used at frequencies greater than 20 kHz
additional circuitry may be needed to reduce gain above
that frequency.
FIG. 4 shows IC1 connected as a noninverting ampli
FIGS. 5, 6 and 7 constitute a circuit diagram of a
presently preferred embodiment of a guitar ampli?er
which includes the features of the invention as shown in
the block diagram of FIG. 1. The power supplies are
not shown since their contruction and operation will be
obvious to one skilled in the art.
Referring to FIG. 5, the preampli?er with frequency
?er in standard fashion without the frequency equaliza
tion circuitry of FIG. 3, but with the distortion circuitry 60 compensation and distortion is shown as block 10 with
the frequency compensation circuitry and the distortion
of the invention. Again, the instrument input is con
circuitry included in blocks 21 and 22 within block 10.
nected to the noninverting input of IC1. Feedback from
The frequency compensation circuitry and the distor
the output of IC1 is connected through resistor R1 to
tion circuitry is as shown in FIGS. 3 and 4, respectively.
the inverting input of IC1. Resistor R6 and capacitor C3
are connected in normal manner between the inverting 65 Here both the frequency compensation circuitry and
input of IC1 and ground, here taking the place of the
the distortion circuitry is combined, but the operation of
variable frequency compensation circuitry. Again the
each circuit is as previously described. Thus, the opera
DC bias for the inverting input to IC1 is provided by
tion of the preampli?er 10 produces an output signal
9
4,581,587
which has been frequency compensated to the desired
10
extent and, if desired, distorted to a desired extent. The
instrument input to preampli?er 10 is shown as a stan
low frequency oscillator capable of driving a light emit
ting element. The photo resistive cells are manufactured
using calcium sul?de and therefore their peak response
dard phone plug 32 where the lead wire from the instru
is to light frequency of 560 nanometers wave length.
ment is connected in standard fashion to the ampli?er
In operation, the signal from the preampli?ers pass
through resistor R13 to the noninverting input of opera
tional ampli?er IC3. The output of IC3 is fed through
resistor R14 to the noninverting input of operational
ampli?er IC6 as a high pass signal. Simultaneously, the
output of IC3 passes through photocell 33 to the invert
circuitry.
A second preampli?er 18 is provided as an auxiliary
input to the ampli?er to be used when variable fre
quency compensation or distortion as provided by pre
ampli?er 10 is not desired. For example, preampli?er 18
may be used with a microphone.
ing input of operational ampli?er IC4, which operates
The input signal to preampli?er 18 is connected
through phone jack 33, and isolation capacitor C4 to the
as an integrator to give a band pass output which is fed
back to the noninverting input of IC3. This band pass
noninverting input of operational ampli?er IC2 which is
signal is inverted with respect to the original signal and
connected similarly to 1C1 in a standard noninverting
hence nulls to a large extent the input signals of those
ampli?er con?guration. Resistor R7 establishes the DC
bias on the noninverting input to IC2. Resistor R8 is the ‘
frequencies. The output of IC4, the band pass signal, is
fed through photo cell 34 to the inverting input of ICS.
feedback resistor connected between the output of IC2
and its inverting input, with resistors R9 and R10 and
the inverting input of IC3. This negative feedback can
capacitor C5 connected between the inverting input
IC5 acts as a low pass ?lter with its output connected to
20
cells the low frequency component of the input signal to
and ground in standard fashion to provide the desired
gain and frequency response. Resistor R11 is an output
isolation resistor similar to R4 in preampli?er 10. The
DC bias on the inverting input of IC2 is set by a path
through variable resistor VR2 which is the volume
control shown in FIG. 6, and resistors R11 and R8, as
IC3. The output of ICS, the low pass signal, passes
through resistor R18 and is summed with the output
from IC3, the high pass signal, and both are connected
to the noninverting input of operational ampli?er IC6.
The output signal from IC6 is connected through
capacitor C7 and resistor R19, acting as a signal loss
well as resistors R9 and R10 which also provide a DC
resistor, to the wiper of variable resistor VR4. Variable
resistor VR4 is connected between the output of the
path from the inverting input to ground.
Satisfactory component values for the preampli?er 18
preampli?ers and ground. The output signal from the
variable band reject ?lter is mixed with the signal di
R10 - 33k ohms, R11 - 2.2k ohms, C4 -0.068 microfar
rectly from the preampli?er in any proportional amount
ads, C5 - 0.068 microfarads, and IC2 a Texas Instru
or depth by adjustment of the wiper of VR4. Resistor
ments TLO94CN linear operational ampli?er.
R20 is a bias resistor for IC6, and switch SW2 is prefera
The output of preampli?ers 10 and 18 are connected
bly a normally closed, foot operated switch positioned
together and pass through resistor R12 to the volume 35 to be operated by a performer when operation of the
and tone controls, block 11, through what has been
band reject ?lter is desired. With switch SW2 closed,
indicated in FIGS. 5 and 6 as connection A. Resistor
the output of the ?lter is grounded and no signal appears
R12 may have a value of 3.3k ohms. The output of
on the wiper of VR4 to be mixed with the preampli?er
preampli?ers 10 and 18 are also connected as inputs to
outputs. When switch SW2 is opened, the ?lter pro
variable band rejection ?lter 11. The variable band 40 duces an output which is mixed through VR4 with the
are R7 - l meg. ohms, R8 - 150k ohms, R9 - 10k ohms, 30
rejection ?lter is designed to pass all but a desired fre
quency and to provide an output signal similar to the
input, but with a selected frequency band substantially
preampli?er signals.
The following component values have been found
satisfactory for the band rejection ?lter circuitry as
attenuated. To accomplish this, the variable band rejec
shown: R13 through R16 - 15k ohms, R17 - 270k ohms,
tion ?lter provides a high pass ?lter and a low pass ?lter
R18 - 3.3k ohms, R19 - 4.7k ohms, R20 - 47k ohms, C5
with the rejected band between.
and C6 - 0.01 microfarad, C7 - 0.47 microfarad, VR4 -
In the circuitry shown, operational ampli?er IC3
forms a high pass ?lter, operational ampli?er IC4 forms
a band pass ?lter which determines the frequencies
rejected, and operational ampli?er ICS forms a low pass
?lter. Operational ampli?er IC6 is an impedance isola
tor and voltage follower for summing the high pass and
low pass signals. Resistors R13, R14, R15, and R16 must
be equal in value and provide gain control. Capacitors
25k ohms linear taper potentiometer. IC3 through IC6
may be National Semiconductor LM 348N’s while the
photocells may be Radio Shack number 276-1 l6’s.
With the photocells used, and a proper variable light
source, the range of the ?lter extends over eight oc
taves.
'
The photo cells are used to change the center fre
quency of the notch ?lter in a sweeping mode. This is
C5 and C6 in conjunction with the resistance of the 55 accomplished by modulating light to the photocells to
photo resistive cells 33 and 34, set the center frequency
change their resistance. While various methods of mod
of the rejected band or notch. Capacitors C5 and C6
ulating light to the photocells can be used, it is presently
must be equal in value and photo resistive cells 33 and
preferred to use a triangle wave generator to cause
34 must have equal resistance values. The “Q” or depth
varying illumination of two light emitting diodes which
of the center frequency or notch is set by the ratio of the 60 in turn, illuminate the photocells to vary their resis
resistance of resistor R17 divided by the resistance of
tance.
resistor R13. With the high pass and low pass signals
A triangle wave generator is shown in FIG. 5 as the
summed by IC6, and the band pass nulled by IC3, the
sweep rate generator enclosed by box 13. The inverting
result is a band rejection or notch ?lter. In order to take
input of operational ampli?er 1C7 is connected to a
full advantage of the notch ?lter, it is necessary to vary 65 voltage divider made up of resistors R21 and R22. At
the center frequency or notch. This can be done by
the instant of startup, using a split power supply of :18
varying the resistance of resistances 33 and 34 equally.
volts dc, the offset voltage saturates 1C7 to give a posi
To change the resistance of the photocells requires a
tive output voltage which is connected to the inverting
111
4,581,587
input of IC8 through variable resistor VRS and resistor
R23. A capacitor C8 is connected as the feedback loop
between the inverting input of IC8 and its output. With
before reaching the wiper, the signal is attenuated and
of lesser amplitude and thus the volume is reduced.
Capacitor C9 will pass the higher frequencies unaf~
a positive output on IC7, a current I flows through
variable resistor VRS and resistor R23 to charge capaci
fected by the volume control VR6. It should be noted
that VR6 is equivalent to VR2 as shown in FIGS. 3 and
4.
The signal from VR6 is connected to three tone con
trol branches. The bass control is made up of resistors
tor C8. IC8 and capacitor C8 act as an integrator. The
output of IC8 is connected through resistor R24 to the
noninverting input of IC7, which also receives a feed
back signal from the ouput of IC7 through resistor R25.
IC8 generates a negative going ramp with a period of
rate of I/(VR5)(R23)(C8) volts/second until the output
of IC8 equals the negative saturation point of IC7. IC7
then clamps to the negative state and provides a nega
12
entering at A must pass through some resistance of VR6
10
R27, R28, and R29, capacitor C10, and variable resistor
VR7. Capacitor C10 will pass the higher frequency
signals around variable resistor VR7 so that when the
wiper of VR7 is set with maximum resistance to ground,
the bass signals will be strongest. When the wiper of
VR7 is set with minimum resistance to ground, the base
signals will be attenuated by the resistance of VR7
tive current I to the inverting input of IC8. IC8 now
generates a positive going ramp with a rate of
I/(VR5)(R23)(C8) volts/second until the output of IC8
equals the positive saturation point of IC7 where IC7
again changes output state and the cycle repeats.
while the treble signals will have been passed by capaci
Frequency of the triangle wave is determined by the
RC time constant which is the combination of variable
tor C10 so the bass signals will have been attenuated in
relation to the treble signals. Capacitor C10 is chosen so
that all but the desired bass tones are passed. With an
resistor VRS, resistor R23 and capacitor C8, and the
positive and negative saturation voltages of IC7. Ampli
instrument ampli?er a satisfactory value for C10 is 0.047
microfarad. The remaining components in the bass con
tude of the waveform is determined by the ratio of
resistor R25 to resistor R24, and the saturation voltages
trol may be R27 - 10k ohms, VR7 - 100k ohms linear
taper potentiometer, R28 - lk ohms, and R29 - 6.8k
of IC7. The output reference center voltage with re
spect to ground is set by resistors R21 and R22. The
ohms. Resistor R29 is an isolation resistor for the output
of the bass control.
output waveform is symmetrical about the positive and
negative peaks with respect to ground. Resistor R26
The midrange control is made up of resistors R30,
R31, and R32, variable resistor VR8, and capacitors
sets the load current magnitude and the associated volt
C11 and C12. This branch operates similarly to the bass
age drop across light emitting diodes LEDl and LED2. 30 tone control branch, but the capacitor C11 is of a
The frequency of the generator is regulated by variable
smaller value than capacitor C10 in the bass control so
resistor VRS. Resistor R23 is used as the upper RC
that the signals passed by capacitor C11 around variable
1 element to limit the upper frequency so the output
resistor VR8 are higher in frequency than the signals
waveform is not distorted on its positive peaks.
passed by capacitor C10 around variable resistor VR7.
The following component values have been found
Thus, the signal passing through VR8 and supplied to
satisfactory R21 - 22k ohms, R22 - 39k ohms, R23 - 3.3k
ohms, R24 - 27k ohms, R25 - 100k ohms, R26 » 910
the wiper of VR8 contain not only the low frequency
bass signals, but also higher frequency midrange signals.
ohms, VRS - 500k ohms linear taper potentiometer, and
The bass signals, however, are blocked by capacitor
C8 - 22 microfarads. The light emitting diodes provide
C12 so that the output from the wiper of VR7 is limited
green light with a wavelength of 560 nanometers and 40 to midrange frequencies. Component values to give a
are made by Monsonto Chemical as well as others. The
satisfactory midrange tone control are: R30 - 3.9k ohms,
light emitting diodes should be placed within 10 mm of
R31 =- 5.6k ohms, R32 - 4.7k ohms, VR8 - 100k ohms
the photocells. IC7 and IC8 are National Semiconduc
tor LMl458N’s. It should be noted that while two light
linear taper potentiometer, C11 - 0.0047 microfarads,
emitting diodes are shown, a single light emitting diode
could be used if placed so that it illuminates both photo
cells simultaneously.
and C12 - 0.33 microfarads. R32 is an isolation resistor
45
for the output.
The treble control is made up of capacitors C13 and
C14, resistor R33 and variable resistor VR9. The high
With the values given, the period rate of the wave
form is a maximum of 33 seconds (0.03 Hz) to a mini
mum of 0.083 seconds (12 Hz) with an output wave
frequencies are passed by capacitors C13 and C14 and
amplitude of +0.8 volts DC. to +4.8 volts D.C.
across VR9. Satisfactory component values are C13 0.015 microfarads, C14 - 0.047 microfarads, R33 ~ 12k
As indicated, the signal directly from the preampli?
ers 10 and 18, and the signal added in from the variable
band reject ?lter are connected to the volume and tone
are taken off of variable resistor VR9. The lower fre
quencies are blocked by capacitor C13, so do not appear
ohms, VR9 - 100k ohms linear taper potentiometer.
The signals from the three tone controls are com
controls, block 11, FIG. 6, at the connection marked A 55 bined and simultaneously sent directly to summing am
in FIGS. 5 and 6. With VR4, the musician can adjust the
pli?er 14 through loss and isolation resistor R34, to high
relative amount of phase shifted signal to normal signal
pass ?lter 15, and to delay circuitry 16. The delay cir
he desires, and switch the phase shifted signal on or off
cuitry is used to create a reverb effect when desired by
with switch SW2.
delaying a portion of the signal which is then added to
The signal at point A, FIG. 6, is connected to ground 60 the signal passing through resistor R34. The delay cir
through variable resistor VR6. This variable resistor is
cuitry uses a standard delay spring reverb unit which
the main volume control with the output signal coming
substantially attenuates signals above six kHz. For this
from the wiper of VR6. Thus, when the wiper of VR6
reason, when the reverb is used, the higher frequencies
is set with maximum resistance between it and ground,
of the output signal have been substantially reduced.
the incoming signal at A passes directly to the wiper 65 The high pass ?lter is provided speci?cally to compen
before passing through any of VR6. The signal is not
sate for this loss of high frequency signals through the
attenuated to any extent by VR6 and volume is maxi
reverb unit. It does so by emphasizing the treble tones in
mum. As the wiper of VR6 is moves so that the signal
the non-delayed signal.
13
4,581,587
The high pass ?lter is connected in the circuitry by
closing switch SW3 which places capacitor C15 in par
taper potentiometer. 1C9 and IC10 may be National
Semiconductor LM 348N’s.
The combination of the main signal and the reverb
signal is ampli?ed by IC11 connected in normal fashion
as a noninverting ampli?er and serving the purpose of a
summing ampli?er. Resistor R45 is a bias resistor. The
series connection of variable resistor VR11 and resistor
R46 provide a variable feedback resistance for IC11.
allel with resistor R34 to form a high pass ?lter with the
high frequency signals bypassing resistor R34. Switch
SW3 allows the ?lter to be used if and when desired. It
has been found that with R34 having a value of 150k
ohms and capacitor C15 having a value of 0.0047 micro
farads, ?lter action begins at 23 Hz and steadly increases
to 20 kHz. The ultimate slope of +6 db per octive
occurs at about 2,258 Hz. This ?lter action occurs re
0
gardless of the volume level of the signal.
The signal entering the delay circuitry of block 16 is
connected to the noninverting input of operational am
R47 completes the feedback loop to ground. By varying
the resistance of VR11, the gain of IC11 may be varied
since the gain of IC11 is given by (R46+VR11)/R47.
The output of IC11 is connected across variable resis
tor VR12 which forms another volume control for the
pli?er 1C9 which is connected in normal manner as a
noninverting ampli?er. Resistor R35 is a bias resistor for
the noninverting input while resistor R36 is the feed
back resistor. Resistor R37 and capacitor C16 complete
the feedback loop to ground and determine the fre
ampli?er. The position of the wiper of VR12 will deter
mine the amplitude of the signal connected through the
wiper to the remaining circuitry. Capacitor C18 pro
vides an additional high pass ?lter connected in normal
manner with respect to volume control VR12. The
quency response of IC9. The purpose of 1C9 is to in
crease the signal strength prior to the delay spring
which substantially attenuates the signal.
14
ohms, R44 - 150 k ohms, and VR10 - 100k ohms linear
20
signal passing through capacitor C18 is summed with
The input signal to 1C9 will generally be in the range
of about ?ve volts peak-to-peak with the output of 1C9
being about 15 volts peak-to-peak. The output of 1C9 is
connected to a standard delay spring reverb unit 35 25
summing ampli?er 14. This output signal passes through
such as a No. 900-0000751 reverb unit made by O.C.
Electronics. In the reverb unit, the electrical signal is
converted to mechanical vibrations which travel along
the spring and are then converted back to electrical
to its passage through resistor R48, is connected to the
noninverting input of IC12 which is connected as a
buffer ampli?er to provide an output at terminal C
signals. When converted back to electrical signals, the
signal has been delayed by about 200-300 milliseconds.
needed. This can go to an additional power ampli?er, to
recorders, etc. Resistor R49 is a bias resistor as is resis
The signal has also been attenuated about one thousand
times and is now about 2 millivolts peak-to-peak. The
signal is now connected to the noninverting input of
tor R50. Variable resistor VR13 in the output of IC12
provides an additional volume control.
operational ampli?er IC10 connected in normal fashion
as a noninverting ampli?er where the signal is ampli?ed
buffer ampli?ers are R45 - l megaohm, R46 - 120k
ohms, R47 - 51k ohms, VR11 - 2 megaohms linear taper
the signal on the wiper of VR12 and forms the output of
isolation resistor R48 to terminal B which is the direct
connection to terminal B of the power ampli?er shown
in FIG. 7. The signal from the summing ampli?er, prior
which can be used anytime a preampli?er signal is
Satisfactory component values for the summing and
to make up for the loss in the reverb unit. The output
potentiometer, VR12 - 100k ohms linear taper potenti
signal from IC10 is again about 15 volts peak-to-peak.
ometer, C18 - 0.0022 microfarads, R48 - b 2.2k ohms,
Resistor R38 is a bias resistor and R39 is a feedback
R49 - 470k ohms, R50 - 470k ohms, and VR13 - 5k ohms
resistor. Resistors R40 and R41 and capacitor C17 com 40 linear taper potentiometer. IC11 and IC12 may be Na
plete the feedback loop to ground and determine the
tional Semiconductor LM 348N’s.
frequency response of IC10.
The signal from the preampli?er at terminal B is
The delayed and ampli?ed signal from IC10 passes
connected to the power ampli?er of FIG. 7, a modi?ed
through isolation resistors R42 and R44 to variable
quasi-complimentary Class B power ampli?er. The
resistor VR10. Variable resistor VR10 sets the reverb 45 signal enters the power ampli?er through the nonin
depth, i.e., the strength of the reverb signal which is
added to the main signal. The reverb signal passes
through isolation resistor R44 where it joins the main
signal at the noninverting input to operational ampli?er
National Semiconductor LM 343H. Resistor R51 is a
bias resistor while resistor R52 is a feedback resistor
preferably foot operated so when opened by the foot of
collector of transistor Q1. Capacitor C23 helps prevent
verting input of operational ampli?er IC13 such as a
from the output of the entire power ampli?er. Resistor
IC11. With the wiper of variable resistor VR10 set with
R53 completes the feedback loop to ground. Resistors
maximum resistance to ground, the maximum reverb
R54'through R59, along with diodes D3 through D8
signal will be added to the main signal. With the wiper
provide the bias for the bases of transistors Q1 and Q2.
of variable resistor VR10 set with minimum resistance
Capacitors C20 and C21 are ?lter capacitors while ca
to ground, the reverb signal is essentially grounded so
pacitor C22 helps prevent oscillation of the ampli?er.
no reverb signal is added to the main signal.
55
Transistor Q1 is connected in series between resistors
A normally closed switch SW4 is connected to
R60 and R61 between the negative power supply
ground between resistors R42 and R43 and when
—VCC and the positive terminal of speakers 36. The
closed, acts to ground the reverb signal. The switch is
bases of transistors Q3 and Q4 are connected to the
a performer causes the delayed signal to be added to the 60 oscillation of the ampli?er. Resistors R62 and R63 are
main signal. The reverb depth will have been previ
ously set so activation of the foot switch adds the de
the load resistors for transisters Q3 and Q4 respectively.
Transistor Q2 is connected in series with resistor R6
sired reverb signal.
Satisfactory component values for the delay circuitry
between the positive power supply +VCC and the
positive terminal of speakers 36. The bases of transistors
are R35 - 390k ohms, R36 - 470k ohms, R37 - 56k ohms, 65
Q5 and Q6 are connected to the emitter of transistor Q2.
Resistors R65 and R66 are the load resistors for transis
tors Q5 and Q6 respectively. Capacitor C24 is a ?lter
capacitor for the negative power supply while capacitor ,
capacitor C16 - 0.033 microfarads, R38 - l megaohm,
R39 - l.5 megaohms, R40 ~22k ohms, R41 - 33k ohms,
C17 - 0.068 microfarads, R42 - 2.2k ohms, R43 - 150k
l5
4,581,587
C26 is a ?lter capacitor for the positive power supply.
The series connection of resistor R67 and capacitor C28
is connected across the speakers to ground.
The following component values have been found
I claim:
l. A solid state electrical instrument ampli?er com
prising an instrument signal input means; a preampli?er
having a feedback loop; distortion means connected in
satisfactory for the power ampli?er: RSI-100k ohms,
parallel with the feedback loop and having nonlinear
R52 - 2 meg. ohms, R53 - 100k ohms, R54 and R55 - 10
resistances versus applied voltage characteristics over
at least a portion of its operating range immediately
ohms, R56 through R59 ~ 2.7k ohms, R60 - 100 ohms, R
below a preset voltage; means in association with said
61 ~ 2.2 ohms, R62 and R63 - 0.51 ohms, R64 ~ 100 ohms,
R65 and R66 - 0.51 ohms, R67 - 22 ohms, C20 and C21
- 50 microfarads, C22 - 0.0022 microfarads, C23 - 0.033
distortion means for limiting voltages applied to such
0 distortion means to below said preset voltage and caus
ing said distortion means to operate at least partially in
said nonlinear portion of its operating range so that said
microfarads, C24 and C26 - 0.068 microfarads, C28 - 0.1
microfarads, D3 through D8 - 1N4003, Q1 - TIP-41C,
Q2 - TIP-42C, and Q3 through Q6 - 2N6339. ICll3 is a
preampli?er compresses the peaks of the input signal
National Semiconductor type LM-343H. The power
supply for the power ampli?er, block 20, FIG. ll, should
and provides a distorted output signal which substan
supply DC. voltage of +VCC of about +37.5 volts
content of the input signal; and signal output means.
2. A solid state electrical instrument ampli?er accord
tially preserves the relative even and odd harmonic
and -VCC of about —37.5 volts.
The operation of the power ampli?er is basically the
ing to claim 1, wherein the distortion means is a parallel
connection of two diodes, each diode having an oppo
site orientation.
3. A solid state electrical instrument ampli?er accord
same as a quasicomplimentary Class B power ampli?er
except that the usual transistor differential input stage
has been replaced with an operational ampi?er so that
the input signal is introduced at the zero D.C. point.
This eliminates the normal input stage transistors from
ing to claim 2, wherein the preampli?er is an integrated
circuit operational ampli?er connected as a noninvert
ing ampli?er, and wherein the feedback loop is a nega
tive feedback loop which includes a feedback resistor
the circuitry and eliminates the need for factory adjust
ment of the input transistor bias. With the modi?ed
circuitry as shown, bias is simple and not as critical with
respect to the transistors as is normally the case.
connected between the operational ampli?er output and
the operational ampli?er inverting input.
Further, providing an input operational ampli?er
4. A solid state electrical instrument ampli?er accord
ing to claim 3, wherein the means for limiting voltages
allows direct D.C. coupling of the ampli?er to the rest
of the circuit and gives the power ampli?er stage a high
input impedence, a high slew rate of 2.5 volts per micro
second which provides a power bandwidth of 25K Hz,
and gives short circuit protection due to short circuit of
the load or short circuits caused by a power supply.
applied to the distortion means is a variable resistor
connected in series with the distortion means.
5. A solid state electrical instrument ampli?er accord
ing to claim ll, wherein there is additionally included a
switch in series with the distortion means so that the
It should be realized that although not shown, each of 35 distortion means can be selectively connected in paral
lel with the feedback loop.
both positive and negative bias voltages. These are
6. A solid state distortion circuit for use with an elec
supplied by a power supply, shown as block 19 in FIG.
trical instrument ampli?er, comprising an instrument
1, but not shown otherwise, which provides +18 volts
signal input means; a preampli?er having a feedback
and - 18 volts to the integrated circuits and to the ter 40 loop; distortion means connected in parallel with the
~ minals marked +V and —V in the sweep rate generator
feedback loop and having nonlinear resistance versus
the integrated circuit operational ampli?ers require
13 of FIG. 5.
applied voltage characteristics over at least a portion of
From the above description of the circuitry, it will be
its operating range immediately below a preset voltage;
realized that except for the standard tone controls, the
means in association with said distortion means for lim
signal modi?cation takes place in feedback loops or by 45 iting voltages applied to such distortion means to below
auxilary circuitry so that ampli?er input and output
said preset voltage and causing said distortion means to
impedences are not substantially affected by the signal
operate at least partially in said nonlinear portion of its
modi?cation and the adjustment of that modi?cation.
operating range so that said preampli?er compresses the
Further, satisfactory tone modi?cation can be accom
peaks of the input signal and provides a distorted output
plished merely through varying the frequency compen
sation of the circuitry without the normal tone controls.
signal which substantially preserves the relative even
and odd harmonic content of the input signal; and signal
Such normal tone controls are not necessary, but are
output means.
provided merely for added adjustment and for perform
7. A solid state distortion circuit according to claim 6,
wherein the distortion means is a parallel connection of
ers who are accustomed to having such adjustments and
might object to their not being present.
Further, it will be realized that various individual
features of the circuitry may be used in a musical instru
ment ampli?er without including other features of the
circuitry.
55
two diodes, each diode having an opposite orientation.
8. A solid state distortion circuit according to claim 7,
wherein the preampli?er is an integrated circuit opera
tional ampli?er connected as a noninverting ampli?er,
and wherein the feedback loop is a negative feedback
Whereas this invention is here illustrated and de 60 loop which includes a feedback resistor connected be
scribed with speci?c reference to an embodiment
tween the operational ampli?er output and the opera
thereof presently contemplated as the best mode of
tional ampli?er inverting input.
carrying out such invention in actual practice, it is to be
9. A solid state distortion circuit according to claim 8,
understood that various changes may be made in adapt
wherein the means for limiting voltages applied to the
ing the invention to different embodiments without 65 distortion means is a variable resistor connected in se
departing from the broader inventive concepts dis
closed herein and comprehended by the claims that
follow.
ries with the distortion means.
10. A solid state distortion circuit according to claim
6, wherein there is additionally included a switch in
17
4,581,587
18
series with the distortion means so that the distortion
means can be selectively connected in parallel with the
the preampli?er in selectable proportions; and signal
feedback loop.
15. A solid state variable band rejection ?lter accord
ing to claim 14, wherein the sweep rate generator is a
triangle wave generator and wherein the triangle waves
cause a constant variation in the light output of the at
least one light source.
16. A solid state variable band rejection ?lter accord
ing to claim 15, wherein the at least one light source is
output means.
11. A solid state electrical instrument ampli?er com
prising an instrument signal input means; a preampli?er;
a variable band reject ?lter for attenuating a variable
frequency band of the signal from the preampli?er, the
reject band of the ?lter being controlled by controlling
the resistance of two photoresistors in such ?lter; a
sweep rate generator in association with at least one
light source so that the output of the sweep rate genera
at least one light emitting diode.
17. A solid state electrical instrument ampli?er com
prising an instrument signal input means; a preampli?er;
delay means for delaying a portion of the output signal
tor is represented by a variable light output of the light
source and said light source being physically arranged
in association with the photoresistors so that light from
from the preampli?er; summing means for summing the
the at least one light source falls upon the photoresistors
delayed signal from the delay circuitry with the nonde
to control the reject band; means for combining the
signal from the variable band reject ?lter with the signal
tion effect, high pass ?lter circuitry electrically con
from the preampli?er in selectable propotions; and sig
nected in parallel with the delay means to pass the rela
layed signal from the preampli?er to create a reverbera
nal outputs means.
tively high frequencies and attenuate the relatively low
12. A solid state electrical instrument ampli?er ac 20 frequencies in the nondelayed signal to thereby increase
cording to claim 11, wherein the sweep rate generator is
a triangle wave generator and wherein the triangle
the relatively high frequencies in the summed signal.
waves cause a constant variation in the light output of
the at least one light source.
cording to claim 17, wherein the high pass circuitry
18. A solid state electrical instrument ampli?er ac
includes a resistor connected in parallel with a capacitor
13. A solid state electrical instrument ampli?er ac 25 both connected in parallel with the delay means.
cording to claim 12, wherein the at least one light
19. Solid state reverberation circuitry for use with an
source is at least one light emitting diode.
electrical instrument ampli?er, comprising an instru
14. A solid state variable band rejection ?lter for use
ment signal input means; a preampli?er; delay means for
with an electrical instrument ampli?er, comprising an
delaying a portion of the output signal from the pream
instrument signal input means; a preampli?er; a variable 30 pli?er; summing means for summing the delayed signal
i band reject ?lter for attenuating a variable frequency
from the delay circuitry with the nondelayed signal
band of the signal from the preampli?er, the reject band
from the preampli?er to create a reverberation effect,
of the ?lter being controlled by controlling the resis
high pass ?lter circuitry electrically connected in paral
tance of two photoresistors in such ?lter; a sweep rate
generator in association with at least one light source so 35
that the output of the sweep rate generator is repre
lel with the delay means to pass the relatively high
frequencies and attenuate the relatively low frequencies
in the nondelayed signal to thereby increase the rela
sented by a variable light output of the light source and
tively high frequencies in the summed signal.
said light source being physically arranged in associa
20. Solid state reverberation. circuitry according to
tion with the photoresistors so that light from the at
claim 19, wherein the high pass circuitry includes a
least one light source falls upon the photoresistors to 40 resistor connected in parallel with a capacitor both
control the reject band; means for combining the signal
connected in parallel with the delay means.
from the variable band reject ?lter with the signal from
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