Agilent Ultra-Wideband Communication RF Measurements

Agilent Ultra-Wideband Communication RF Measurements
Ultra-Wideband Communication
RF Measurements
Application Note 1488
This application note is written for people who need to understand the
configuration and testing of ultra-wideband (UWB) devices, and some of the
issues surrounding their use. A broad range of topics is addressed in this
paper, including practical test techniques. Further details on many of them
may found in the references in Appendix B.
The basic concepts behind UWB signals are not new, but the radios are
becoming more sophisticated. The signals are split into three main groups,
depending on the signal generation technique: baseband-pulsed, pulsemodulated RF, and orthogonal frequency division multiplexing (OFDM).
Pulsed signals have been used in air and ground-penetrating RADAR
systems of various forms for many years. Ultra-wideband OFDM involves
adapting standard OFDM principles to meet the regulatory requirements of
an underlay technology.
RADAR and position location in the form of radio frequency identification
(RFID) tags are good applications of UWB, but it is the application to short
range, very high speed data transfer that has recently triggered increased
interest, and is the main focus of this application note. Communications
applications like streaming video can make use of the latest mixed signal IC
technology to provide viably-priced consumer devices.
Spectrum allocation is the key to new radio development. In 2002 the FCC
in the United States allocated 3.1 to 10.6 GHz for use with unlicensed UWB
signals as an underlay technology. It has stimulated many proposals to meet
the specific requirements of the ruling. The IEEE 802.15.3a Working Group
is one of the bodies looking to develop a standard that can be used generally
by the industry for high-speed communication. Similar to Bluetooth™, the
Multi-Band OFDM Alliance Special Interest Group has been established to
promote an open OFDM standard. Other groups in Europe, Japan, and
Asia are also showing interest, but do not currently have definite spectrum
allocation rules with which to work.
There are alternative approaches to very high-speed wireless data transfer.
One example, known as mmWave, uses conventional modulation of carriers
above 20 GHz. This application note does not directly deal with this, but
some of the measurement techniques will be applicable.
It is not only the RF transmission that has to be addressed to make a radio;
the digital signal paths to and from the radio also need suitable hardware
interfaces and a software medium to work. Industry groups such as JEDEC
are tackling the hardware interface definition, while the IEEE 802.15.3 standard describes a medium access control that is suitable for the very high
throughputs being sought. It continues to be enhanced.
In working with UWB devices, it is important to understand what you are
trying to achieve, before making assumptions about what will be the correct
measurements and equipment. Table 1 lists the basic options.
Table 1.
Understanding and developing the radio design or a module
Testing for spectrum regulation purposes
Checking for interoperability between different vendors’ designs
Testing the effect of interference on other systems
Sections of interest
6, 7, 8
4, 7
Table of Contents
1. Basic Concepts Behind UWB Radio
Definition of UWB ............................................................................................5
Spectrum occupancy and channel capacity ................................................5
Frequencies, power levels, and applications ...............................................6
IEEE802.15.3a (alternate PHYsical layer) section criteria ...............................7
Signal generation and modulation.................................................................8
Baseband pulsed ............................................................................................8
Pulse modulated RF ........................................................................................11
Orthogonal frequency division multiplexing (OFDM) .....................................13
TDMA and packet structures .........................................................................15
Notes on MB-ODFM .......................................................................................15
Notes on pulse modulated RF DS-CDMA .......................................................16
2. Circuit and Device Simulation .........................................................................17
3. RF (PHYsical) Layer Test...................................................................................18
4. Interference Testing for Non-UWB Devices ...................................................19
Device test configuration ................................................................................20
RF signal coupling ............................................................................................21
Generating the interference signal................................................................22
5. Component and Network Measurements.......................................................25
Component impedance and reflection measurements ..............................25
Antenna and channel response measurements .........................................26
Use of equalizer characteristics .....................................................................28
Multi-path reflection and wavelets ................................................................29
Differential network analysis .........................................................................29
Delta (additive) EVM ........................................................................................31
6. Transmitter Measurements ..............................................................................33
Test conditions and measurement setup .....................................................33
Equivalent isotropic radiated power (EIRP) ....................................................34
Interoperability testing....................................................................................34
Hardware probing ...........................................................................................35
Measurement triggering .................................................................................36
Use of captured time records .........................................................................39
Test modes ......................................................................................................40
Distinguish between RF bandwidth and video (demodulated) bandwidth..........41
Power spectral density, average detection ....................................................41
Spectrum flatness determines total transmit power......................................42
Sweep time .....................................................................................................42
Use of average (rms) detector for power measurement ................................43
Peak power measurement using a swept spectrum analyzer........................44
Peak output power, CCDF ...............................................................................44
Baseband versus envelope (zoom) CCDF .......................................................45
Table of Contents
6. Transmitter Measurements continued...
Transmit output spectrum ...............................................................................46
Swept spectrum measurements of pulsed RF signals ...................................46
Effect of increasing the resolution bandwidth on display level ...............48
Peak and average detection of UWB signal ...................................................48
Comparing FFT-based and swept spectrum results .......................................52
Spectrograms and adjacent power measurements .......................................54
Two-channel (correlated) spectrum measurement ........................................54
Spectrum mask testing ...................................................................................55
Modulation tests...............................................................................................56
Baseband pulsed ............................................................................................56
Pulse modulated RF ........................................................................................56
OFDM ..........................................................................................................................58
Extending the capture period..........................................................................59
Frequency measurement
CW and long pulsed signals ...........................................................................60
ODFM modulated signals ...............................................................................61
Short pulsed signals ......................................................................................61
7. Transceiver Spurious Tests ...............................................................................62
8. Receiver Measurements ...................................................................................63
Test conditions and setup ...............................................................................63
Frequency hopping ...........................................................................................65
Receiver EVM measurements and BER........................................................66
Receiver sensitivity (RSSI) ..............................................................................67
Clear channel assessment test......................................................................68
9. Power Supply Measurements ..........................................................................69
Appendix A: Agilent Solutions for UWB ..............................................................70
Appendix B: Recommended Reading ...................................................................72
Appendix C: Glossary .............................................................................................73
Appendix D: Symbols and acronyms ....................................................................74
Appendix E: References..........................................................................................75
1. Basic Concepts
Behind UWB Radio
Definition of UWB
For the purpose of this application note, UWB is taken as a radio signal
with an instantaneous bandwidth of ≥ 500 MHz or a fractional occupied
bandwidth of ≥ 0.2, where
Fractional bandwidth = 2(fH - fL)/(fH + fL)
One of the key requirements for an UWB radio is the need for a broad, flat
power spectrum. A flat transmit spectrum within the chosen frequency
range will maximize the total transmitter power. Consumer UWB devices
will transmit at very low powers. As an example, an indoor device operating
from 3.1 to 4.8 GHz will need to transmit less than
–41.3*10.log(4800-3100) = –9 dbm
to satisfy the U.S.A.’s spectrum regulation requirements. The peak RF
voltages are < 1 volt.
Spectrum occupancy and channel capacity
Most readers will know that Shannon-Hartley derived a simple expression
to relate the basic data transfer capacity of a channel to the instantaneous
signal bandwidth as:
C=B.log2 (1+S/N)
C = Channel capacity
B = Occupied bandwidth
S/N = Signal-to-noise (linear power ratio, not dB)
Another variable available with digital radio is the option to transmit data at
a higher rate than the user needs to overcome practical problems. Assuming
the hardware is adequate, the performance of a real radio receiver is limited
by either interference or thermal noise at the input. The processing gain
from direct sequence code spreading can, for example, allow a receiver to
operate with input power spectral densities below the thermal noise floor of
the input circuitry.
The majority of existing radio applications have an occupied bandwidth
which is similar to the user data transfer rate. They rely on a good signal-tonoise ratio (SNR) and complex modulation formats for higher data rates.
Some radio applications like CDMA, GPS, and the original 802.11b use direct
sequence spreading to create a much wider signal bandwidth than
Shannon’s equation requires. This is variously used to deal with interference, multiple users, or in the case of GPS, extreme path propagation loss.
See Figure 1.
Bandwidth here depends on
the spectrum allocation
Radio link
Incoming data
formatted into
Bandwidth here depends
on the user's data rate &
error protection
User data
This is where the [correlated]
signal is recovered more strongly
than the [uncorrelated] noise
Figure 1. In spread spectrum radio, the SNR is improved by the correlation in the receiver.
An occupied bandwidth in the GHz range allows for some novel combinations of radio hardware and digital processing, while still addressing some
of the most demanding applications.
Spectrum regulations constrain what is allowable, because along with the
available spectrum, they define a maximum radiated power spectral density
and peak power. This determines the received SNR, because the environment will determine the path loss.
The regulations governing UWB in the United States envisage it as an
underlay technology, where the power spectral density is low enough to
avoid interference with existing systems.
Frequencies, power levels, and applications
UWB systems operate across licensed and license-exempt bands, within the
frequency bands shown in Table 1. The maximum transmit powers are also
shown. The limits shown are only applicable in the United States. Other
regions are exploring what limits they should set to suit local conditions.
Currently some regions like Japan and Europe are more cautious, while
Singapore is considering higher transmit powers. To allow CMOS implementations, and avoid 802.11a interference, the first UWB devices for consumer
electronics will operate below 5 GHz.
Historically, UWB has been widely applied to location-sensitive applications.
This is because the short pulses needed for the simplest ranging systems
inherently occupy a large bandwidth. The FCC 02-48 ruling is causing this to
Streaming video and wireless universal serial bus (USB) are key target
applications for UWB in communications. They will not be the only uses, but
act as useful, consumer-oriented ways to assess the capabilities of an UWB
Unlike the wireless local area network (WLAN) standards of IEEE 802.11,
IEEE 802.15 standards are for wireless personal area networks, WPANs. In a
WPAN, wireless devices form temporary piconets to enable data transfer.
The well-known Bluetooth standard is in this family (IEEE 802.15.1), and a
number of the principles governing system operation are shared.
The main impact of the distinction between WLAN and WPAN is felt in the
software between the radio and the appliance it serves. However, there is
also an effect on the RF because more than one piconet has to be able to
operate in the same area, at the same time. This is known as simultaneous
operating piconets, or SOPs. Unlike WLAN, there is no central access point
to coordinate network activity. The piconets must be able to operate independently and asynchronously, which immediately places burdens on the
system design. Each radio must be able to quickly identify RF packets meant
for it, and minimize the effect of unwanted signals on its data throughput.
Table 2. System frequency bands and applications for the U.S.A.
FCC Part 15 freq band2 Max power (1 MHz) Restrictions
1. Ground penetrating radars,
3.1 to 10.6 GHz
wall imaging, medical imaging
GPR < 960 MHz
2. Thru-wall imaging and
1.99 to 10.6 GHz
surveillance systems
Communication and measurement
3. Indoor
3.1 to 10.6 GHz
4. Outdoor handheld
3.1 to 10.6 GHz
24 to 29 GHz and
59 to 66 GHz1
Vehicular radar
5. Vehicular radar
24 to 29 GHz
Collision avoidance, improved
airbag activation, suspension
–41.3 dBm
Yes, usage
–51.3 dBm
–41.3 dBm
–41.3 dBm
No separate
or outdoor
–41.3 dBm
1. Unconfirmed.
2. Band edge is –10 dB relative to the maximum in-band signal.
IEEE 802.15.3a (alternate PHYsical layer) selection criteria
The development of a new standard has to satisfy many criteria, some of
which are difficult to accurately compare between different proposals.
Listed below are some of the more tangible factors that have to be
addressed, and provide some insight into why the process can be a lengthy
General solution
cost, signal robustness, technical feasibility,
scalability, location awareness
MAC supplements
MAC changes needed, power management, power
PHY layer
size and form factor, bit rate and throughput,
simultaneous piconet operation, signal
acquisition, range, sensitivity, multi-path,
antenna practicality
Signal generation and modulation
The 802.15.3 (high rate) RF physical layer is fairly conventional, but has not
been widely adopted. It is 802.15.3a (alternative high rate PHY) that is
specifically targeted at using extreme radio bandwidths, and the one this
application note addresses.
Most engineers are used to data being modulated onto a radio carrier before
transmission. There are many ways of generating and modulating that
carrier. Two of particular interest for UWB are discussed here, but the first
description is of a technique that does not use a carrier at all, and it is what
was initially thought might be the basis for UWB radios.
Baseband pulsed
Here, the RF energy is derived from the spectral components of a baseband
Pulse shape
The pulse shape determines the spectrum shape, or envelope. The most
desirable pulse shapes have a broad flat top in the spectrum, since this
maximizes the total transmit power allowed under the regulations. UWB is
not the first technology to adopt innovative pulse shapes and structures to
give specific spectral characteristics. DC (zero frequency) energy is difficult
to send accurately over any significant distance. As an example, Manchester
encoding was adopted many years ago to avoid this problem.
Figure 2 shows the time and spectrum waveforms of a simple bipolar pulse.
While the low frequency energy is lower than a uni-polar pulse, the second
lobe is only 10 dB below the main one. The frequency band of the second, or
higher, lobe may be most useful. Considerable additional filtering will be
Generating a UWB RF signal using a short, very fast pulse is conceptually
the simplest method. The relationship between the time domain and the
spectrum is derived from basic Fourier analysis. With the ultra fast
switching speeds of modern digital devices it is no longer necessary to use
specialist components like step recovery diodes or avalanche transistors. A
combination of a band-pass filter and time domain pulse shaping can be
used to remove unwanted spectral energy. Some innovative pulses shapes
even allow notches in the spectrum to be created, but these will require
more sophisticated implementations than described here. The right hand
plot in Figure 2 gives an indication of how these pulse shapes might appear.
The promise of a low cost implementation will be at the root of R&D work
for many years to come, but it seems unlikely a baseband pulsed approach
will be used for early mainstream devices because reliably shaping the
spectrum is difficult.
Complex pulse shaping
Long amplitude
Transmit band
Figure 2. Time and spectrum plots for a bipolar pulse, and an example of a complex pulse with a
more desirable spectrum shape.
Pulse spacing
The pulse spacing determines the frequency between adjacent signal
components seen on a spectrum analyzer. Since user data is applied to
change some characteristic of the pulse, the spacing also determines the rate
data may be sent.
Spectrum regulation measurements most often use a 1 MHz resolution bandwidth. For a repetitive signal, this means signal components of 1 MHz and
above can be seen (resolved) discretely on the analyzer's display, while low
frequency components cannot. However a low repetition rate does not suit
fast data transfer, and also will require a higher pulse voltage for the same
power spectrum density. Considering the signal in the time domain,
whenever the voltage is zero, there is no energy being transmitted.
Any repetitive element in the time domain will show as spikes (discrete
tones) in the frequency spectrum, so it essential to “whiten” the pulse
structure regardless of user data. What are required are tightly spaced
pulses with their amplitude, timing, or shape adjusted in a way that cancels
out any discrete spectral activity. Figure 18 on page 24 shows how this may
look in practice.
The block diagram in Figure 3 shows the main components within a baseband pulsed radio. Going from right to left in the lower half of the diagram,
the incoming user data is packaged into a formatted signal with a preamble,
header, and footer. The data stream is then passed for modulation. The
simplest schemes might use pulse position modulation, but amplitude and
even shape modulation may be employed.
Signal timing is derived from a crystal reference. An allowance is made
within the radio standards for static frequency errors, but timing jitter
caused by noise in the oscillator or the circuits it drives, will reduce the
radio link’s performance. Timing jitter and phase noise are different views
of the same thing – spectral noise.
Sample & hold
Direct pulse
generation &
Pulse position
/ pulse polarity
limit filter
data bus
Figure 3. Block diagram of basic baseband pulsed UWB radio. Some designs omit the sample and
hold circuit.
The pulse generator shown in this diagram is very simple. A pulse generator
like this will require sophisticated filtering to meet spectrum regulation
The UWB devices envisaged for IEEE 802.15.3a still use a TDMA packetbased transmit/receive technique. Being able to turn off as much circuitry as
possible when it is not in use will remain vital to meet the battery consumption expectations.
A single RF switch and antenna is shown. Spatial diversity transmission
and reception is not appropriate for UWB, because an UWB signal does not
suffer from the narrowband fades that antenna switching tackles.
Interference is the biggest problem for most radio designs, and an UWB
receiver is particularly sensitive to high-level signals simply because of the
wide input frequency range. For UWB consumer applications, an IEEE
802.11a transmitter or 1.9 GHz cell-phone is likely to be the hardest with
which to deal. A good demodulator can separate a wanted signal from
interference, as long as the distortion is linear. This means the amplifier
chain in the receiver must be well protected from high-level signals that
would cause them to distort the combined signal.
After band-pass filtering, the pulsed signal goes into a correlator, which
multiplies the signal by an ideal version of itself. The correlator can take
different forms, one of the simplest being a very high-speed sample and hold.
The baseband timing circuit needs to synchronize the timing of the sampling.
It does this by looking for readily identifiable parts of the radio signal in the
Multi-path reflections mean the pulse waveforms that arrive at the receiver
input will be far more complex than what was transmitted. Figure 4 gives an
indication of what might be seen for an isolated pulse. More sophisticated
correlators and multi-tap rake receivers can be used to capture more energy,
but the designer has to trade off performance with complexity and associated
power consumption.
Figure 4. Example of the complexity of an isolated pulse waveform (due to multi-path reflections)
as seen by a receiver.
Pulse modulated RF
Examination of the pulse waveforms needed to create a banded spectrum
shows they look like bursts of a few cycles of a carrier. The simplest extension from the bipolar pulse of Figure 2, the Gaussian mono-pulse shown in
Figure 5, looks like a single sine-wave cycle.
Log amplitude
Transmit band
Figure 5. Time and spectrum plots of Gaussian mono-pulse, showing approximation to a carrier
This points to the use of conventional frequency mixing as a way to generate
the UWB signal, which has become more popular. Figure 6 shows spectrum
analyzer plots for signals created in this way.
81132/81134 pulse generator
ESA, PSA spectrum analyzer
ESG signal generator
Wideband mixer: mini-circuits
ZEM 4300, Marki M2-0006MA
or similar
Pulse Rise time determined
the slope of this line
20 MHz span
1 MHz rectangular pulse
50% mark space ratio
Carrier Leakage
through mixer
Spectrum energy
is well spread
1 GHz span
1 MHz rectangular pulse
50% mark space ratio
1 GHz span
1 MHz rectangular pulse
20 ns on period
1 GHz span
1 MHz rectangular pulse
5 ns on period
Figure 6. Spectra of a pulsed 500 MHz carrier, with different turn-on periods. The carrier leakage
(shown as a discrete tone) circled in lower right plot is due to imperfect mixer balance.
Figure 6 shows how the energy gets distributed very broadly across the
spectrum as the on-period is reduced to the point where there are only a few
cycles of the carrier (2.5 in this case). This spectrum behavior is entirely
predictable, but few RF engineers will have had reason to experiment with
pulses this narrow. Pulse modulation usually involves hundreds of cycles of
the carrier.
On paper the implementation looks more complicated, but multiplying a
fixed carrier by a shaped pulse eases some of the significant problems
related to realizing low-cost, reproducible performance. For a given RF
bandwidth only half that bandwidth value is needed at IF. In the time
domain, the pulse rise time can be half the speed needed for the same
bandwidth generated using a baseband pulse system.
Band reject filter
Amplitude only or IQ
Low pass/
band pass
~4 GHz local
Pulse generation
& frequency
Amplitude only or
IQ modulation
Pulse position/
Emission limit filter
Pulse shaping
RF control
Data recovery
data bus
Baseband data
+1, 0, –1 for
DS-UWB scheme
Spectrum shaping
Low jitter [phase noise]
crystal reference
Figure 7. Block diagram of pulse modulated RF UWB radio.
Figure 7 shows how the RF front end now looks similar to other superheterodyne receivers. Differential signal paths are shown at various points,
to indicate how this has become an essential part of circuit design and is
progressively moving beyond the IC itself. The usual issues of carrier
generation apply.
Unlike the baseband pulsed system, the voltage of the transmitted RF waveform – as seen on an oscilloscope – no longer shows the shape of the pulse
the receiver recovers. Pulse shape measurements will require the signal to
be demodulated.
Figure 8. 89601A Vector Signal Analyzer software using Zoom (Demodulation) mode to show both
the time (voltage) waveform of pulsed RF signal, and the recovered pulse shape.
In Figure 8, a root raised cosine pulse is shown with ternary amplitude
modulation. The –1 amplitude multiplier means it is actually an extension of
binary phase shift keying (BPSK). The inter-pulse spacing may be very short.
The bottom trace shows how the pulses may partially overlap in a practical
UWB implementation.
Direct sequence UWB (DS-UWB) modulation
The options for signal modulation are very similar to the baseband pulsed
system. The direct sequence UWB (DS-UWB) proposal for IEEE 802.15.3a3
addresses the need for closely shaped pulses with long repeat intervals by
using code sequence modulation. Modulation involves choosing between
symbols made up of carefully chosen pulse sequences, alternating between
+1, –1, and possibly 0. These may be applied as BPSK or QPSK. Figure 8
shows a short part of one of these types of sequence. The pulse spacing in
the center plot of this figure was chosen to show how the pulse shape may
be recovered and does not represent the real signal.
A common crystal reference will be used for carrier and pulse timing, giving
a fixed relationship between the carrier frequency and pulse period. Some
designs may use a variety of local oscillator frequencies to provide the right
combinations of spectrum use and data throughput.
Post down-conversion filtering provides some additional interference
protection, but the IF bandwidth has to be so wide it is not as effective as a
normal narrow band radio. The recovered pulse can either be fed to a simple
correlator or, as shown in Figure 8, sent to an analog-to-digital converter
(ADC). Digital signal processing (DSP) on the ADC output is used to recover
the original signal. Over time, designs will change to use ever-faster ADCs.
This will allow more signal filtering and recovery to be done digitally, rather
than relying on analog circuit performance. For DS-UWB, it is the receiver’s
correlation of symbol pulse sequences that is more important than
individual pulses.
Orthogonal frequency division multiplexing (OFDM)
Based on the availability of high speed DSP, OFDM is becoming a very
popular format, used in technologies such as digital video broadcast (DVB)
and IEEE802.11a and g. The basic mechanism is to divide the payload data
between many (synchronous) sub-carriers, resulting in a reduced symbol
rate for each carrier, rather than using a much higher rate for a single carrier.
In the time domain, this extends the time period over which a data bit is
received, and makes it less affected by multi-path and narrow-band interference. The delay spread that has to be accommodated is considerably less
than for WLAN due to the shorter (10 m) expected operating range.
While the radio again looks complex, OFDM has a number of characteristics
that make it a realistic possibility for UWB. As well being potentially robust
to multi-path interference, it has a well-defined spectrum shape and is
scalable according to the data rates required. The minimum 500 MHz instantaneous bandwidth set by United States’ regulations determines the lowest
DSP processing rates that can be used.
Amplitude only or IQ demodulation
-: 8
:- 2
~4GHz Local
Freq switch control
RF control
Data recovery
Figure 9 shows a typical block diagram for an OFDM radio, with a choice of
single path or IQ mixing. Given the bandwidth of the signals being sent to
the ADC/DAC, it is possible for them to become part of the RF section
(e.g. in a silicon-gemanium IC), but the digital interface is then a challenge
to implement.
IF gain control
data bus
Baseband data
Amplitude only or IQ modulation
External amplifiers
and RF filtering
Figure 9. Block diagram of OFDM UWB radio.
Frequency switching
One of the unusual parts of this radio is the carrier generation. The
maximum DSP circuits currently realizable can only generate signals
with approximately 500 MHz RF bandwidth. To make use of more of the
spectrum, the frequency is hopped at the OFDM symbol rate. See Figure 10.
Unlike technologies like Bluetooth, the frequency is changed so rapidly it is
not possible to use a single phase-locked oscillator. All the frequencies needed are generated continuously, and a switch selects the one required. There
are too few frequencies to benefit from using a random frequency selection
pattern. The frequency is switched in on a small number of patterns, which
identifies a particular piconet.
4488 MHz
3960 MHz
1.875 µs
Tail & pad
Channel estimation
3432 MHz
Figure 10. Frequency switching for each symbol of UWB-OFDM burst.2
Lower order modulations like BPSK and QPSK will be used in an UWB OFDM
radio. This is because the wide bandwidths provide sufficient capacity and
the poor SNR does not support higher order modulation. The number of bits
in the ADC has to be limited due to the very high sampling rates. Four or
five bits should be sufficient.
As with other schemes using OFDM the preamble will use the most robust,
lowest order, modulation. The preamble is spread over all the frequencies to
be used, allowing the equalizer to form the best estimate of the channel.
For lower user data rates a further simplification is possible, where only the
real component of the DSP signal is needed. As shown in Figure 11, it is
noticeable that the spectrum becomes symmetric when this is happening.
The single baseband signal has half the bandwidth of the modulated RF
Figure 11. Time gated spectrum using 54855 oscilloscope and 89601A VSA software, with a
Gaussian RBW filter. It shows four points during an OFDM burst with real-only modulation data.
The spectrum is symmetrical around the center. This effect is seen with any scheme that allows
for real-component only modulation, including 2BOK DS-UWB.
TDMA and packet structures
The discussion so far has been largely about the RF signal. Many layers of
protocol are laid on top of this. The details of the frame structure depend on
the PHY format being employed and these are still being developed. This
application note will therefore not attempt to describe these in detail. The
medium access control protocol for 802.15.3a will draw heavily on that
developed for 802.15.3, which has a number of mechanisms to reduce
signalling overhead.
An UWB device for 802.15 data communication only transmits or receives at
any point in time. Transmissions occur as packets (frames), which vary in
length and spacing, usually for a few hundred microseconds. This means the
frame contains hundreds of thousands of DS-UWB pulses, or around a
thousand OFDM symbols.
Notes on MB-OFDM
The basic structure proposed2 to IEEE 802.15.3a for multi-band OFDM is
shown in Figure 12. It is very similar to existing WLAN frames. The preamble is used by the receiver to acquire and adapt to impairments on the input
signal. Depending on the modulation format, this can involve frequency and
phase error equalizing, and time alignment. Since the signal is spread over
multiple frequency bands, the path correction has to be calculated for all
these bands.
The header contains a lot of information including the destination address
and the format of the remainder of the burst. User data is transferred from
the original packets, which are fed to the MAC layer. Long packets may be
fragmented (broken up) if the radio determines this will improve the link
PLCP Reamble
30 OFDM symbols
55 Mbps
Frame payload
Variable length 0-4095 bytes
55, 80, 110, 160, 200, 320, 480 Mbps
11.5625 µs
Figure 12. Frame structure for UWB-OFDM format.
Notes on pulse modulated RF DS-UWB
The frame for a pulse stream system3 looks similar, see Figure 13, but there
are significant operational differences. For example, individual piconets are
identified by a small frequency offset (±3, ±9 MHz) in the local oscillator.
This is designed to be rapidly identifiable during the synchronization
process. In recovering the data, the pulse data coding is such that it is the
correlation of the code sequence that is most important.
Equalizer training
PHY header
FEC type
MAC header
Frame body [0-4096 bytes, includes FCS]
and frame check sequence
bits and
10, 15, or 30 µs
Figure 13. Frame structure for DS-UWB format.
In both schemes, each frame is recovered in isolation. The channel equalization is done on a small part of each frame and may need to use only a few
microseconds worth of data. This means special care is required to ensure
this part of the signal is stable relative to the remainder of the frame.
2. Simulation
Circuit and channel simulation are vital elements in the design of a new
radio system. It is important to be able to build up a complex system from
accurate component models that can themselves be verified against real-life
measurements. Integrated links to test equipment (including logic analyzers)
and the 89601A VSA software make this easier. It is the concept behind
Agilent’s Connected Solutions. The system shown in Figure 14 is an example
of a DesignGuide, which allows block-by-block construction of the simulated
system. It shows how a complete path, from bits-in to bits-out can be created.
Figure 14. ADS UWB DesignGuide schematic, showing the building blocks of a MB-OFDM signal
generator. The 89601 VSA software is built-in to allow analysis of the simulated signals.
DesignGuides are being developed for other UWB formats. Check for further
details on
3. RF (PHYsical)
Layer Test
The definition of what are suitable measurements also evolves during the
lifecycle of a new standard. At the time of writing, UWB is still at a fairly
early stage of development. Existing tools can be used for a wide range of
analysis based on the most fundamental properties of a design, such as
power, frequency, and existing modulation formats. This document describes
some of the latest techniques available.
A lot of work has been done to define the appropriate spectrum regulation
limits. In the United States this has resulted in some test limits being adopted, but work continues (such as in the ITU-R committee) to reach agreement
in other regions.
Test metrics will need to be agreed upon to ensure radios from different vendors meet a minimum level of performance for interoperability. This has not
yet happened, but there are a number of measurements that can be expected
to be useful, particularly around parameters that are undefined in the standard, such as what happens before the packet preamble.
UWB is an underlay technology, and therefore interference testing is a significant issue. The main issue is specifying the conditions that will be usefully
representative, without seeing an explosion in the number of test cases.
Many different types of UWB radios will be encountered. Existing equipment
can be used to approximate many of these. This provides a means to test for
areas of vulnerability, while retaining flexibility.
The radio standard determines how effectively the available spectrum is
used. Adjacent channel spectrum testing is only applicable if the RF is split
into frequency bands. With pulsed radio this is not the case, but multiple
piconets still have to work simultaneously. In CDMAone cellular a similar
issue is dealt with using a peak code domain error measurement, where the
effective leakage onto other codes is assessed. Different techniques, based on
the pulse timing and shape may be of use, but have yet to be defined.
Radio development
Spectrum regulation
Spectrum occupancy
PSG, Wideband Mod.
Signal. Gen.
Network Analyzer
81134 Dual Channel
Data /Pulse Gen.
89604 Distortion
Analysis software
89601 Vector Signal
Analysis software
DCA/Sampling Scope
Real Time Scope
Spectrum Analyzer
ADS Simulation
Design Guides
Table 3. Equipment suitable for different tasks. (Some restrictions apply.)
4. Interference Testing for
Non-UWB Devices
If UWB is to act as an underlay technology, it needs to be shown to have a
minimal impact on existing spectrum users. As is clear from the descriptions
earlier in this application note, there are many ways to generate an UWB
signal. When considering interference testing, allowing for some flexibility in
the test source is therefore likely to be beneficial, after making some basic
choices about which format to verify.
The first step to consider for an interference test is if it is an in-band or
out-of-band test as far as the victim receiver is concerned. Figure 15 identifies which is which. An in-band test is of probably the most interest, unless
the UWB device is to be co-located in the same appliance as the potential
Out-of-band UWB spurious
is in-band for victim
UWB signal is inband for victim
Mixed case
1st generation
3.1 GHz
IEEE 802.11a
Possible UWB
spectrum shape
Satellite receiver
GPS receiver
Possible UWB
spectrum shape
2nd generation
UWB band
8.2 GHz
Figure 15. In-band versus out-of-band Interference for different victims.
Out-of-band (for the victim)
It is possible to test the rejection of the DUT to out-of-band frequencies, but
for modern radios, like cell-phones, it should be very good. Ironically, some
high power cellular TDMA systems, like GSM, do cause interference, but with
non-RF circuitry. This is because the radiated field strengths are high
enough for unintentional reception in low frequency circuits that are not
well-screened. Non-linearities in the circuits provide unwanted amplitude
demodulation. The relatively low repetition rate of the RF bursts makes it
easy to literally hear the result.
WLAN signals are also transmitted as bursts of RF, but the signal levels are
lower, and the distribution of the bursts is more random. There have been
few obvious effects. Each frame of a WPAN UWB signal is still transmitted as
an RF burst, but the amplitude is even lower than WLAN. With MB-OFDM,
where the signal is transmitted in shorts bursts throughout each frame, the
repetition rate is very high. It seems unlikely that there will be a general
problem. Investigation continues for specific situations such as satellite
In-band (for the victim)
The interference is in-band if the victim’s input frequency is within the UWB
transmit frequency band, or if a practical transmit filter on the UWB device
leaves measurable unwanted sideband components. We cannot assume the
UWB device will be any better than the normal regulatory requirement and
we therefore can use this as the nominal test limit.
Device test configuration
The effect of interference on the victim needs to be assessed in as quantitative a method as possible. Four factors need to be understood:
• type of victim receiver (digital or analog). Protection provided by
modulation or coding formats – what is the most sensitive format to
• operating link margin for the victim system
• nature of the interference
• the power level of interference at victim receiver input
For a digital receiver, a practical test of the impact may take the form of a
bit error rate (BER) or packet error rate (PER) test in the device under test
(DUT). The drawback of a PER test can be that it does not show how much
margin there is between good and poor levels of operation. This can be
partly addressed by attenuating the wanted signal until it is on the verge of
failing. Alternatively, note that systems using error correction frequently
have a mechanism to show how much correction is being applied by the
digital signal processor (DSP) in the receiver. The signal quality indication
may even be made available to the normal user, as is the case for digital TV
in the United Kingdom. It provides an improved metric compared to PER
or watching the video signal. Special test software can also provide the
information needed, although it may not be widely available.
Even using this technique, the result depends on the specific implementation of the victim receiver. The most thorough understanding of how the
interference affects the victim is to measure the analog signal recovered by
the victim. Monitoring EVM results using a signal analyzer gives an insight
into the reasons why bit errors occur, especially if a time capture of the
combined signal is available to show the relative timing of bit error and EVM
RF signal coupling
In practice, the interference will usually (not always, we need to watch the
isolation of connecting leads) get into the victim via its antenna, so this is an
attractive test configuration. However, a cabled connection to the victim’s
receiver will provide far more repeatable measurements. Measurement
variations will be reduced if the victim system is left as close to complete as
A way to cross-calibrate the overall measurement path is to make use of the
absolute level accuracy of the RF signal generator, using the configuration
in Figure 16. With a suitable arbitrary (ARB) waveform, the generator can
produce a signal the DUT can recognize. The receive signal strength
indication (RSSI) result from the application software associated with the
DUT allows a reference power level to be defined at the DUT input. It may be
necessary to contact the DUT supplier for suitable test software. The ARB
waveform may also need to come from the DUT supplier, but refer to Use of
captured time records on page 39 for details on time-captured waveforms.
Recovering an
anlog signal
allows in-depth
analysis of RF
Ferrite Isolator
for cable
RF down conversion
user data
Calibration path
Signal Generator generates a "near
normal" signal for Victim DUT
Recovering data after
error correction has
been applied gives a
"User View", but is
unlikely to show partial
degradation - unless
the wanted signal is
already close to its
acceptable limit
Victim recovers data and reports
RSSI using PC and software link
or internal application
Figure 16. Calibrating the path loss using a signal generator.
The path loss needs to be separately measured using calibrated antennas.
There are a number of methods the path loss can be found, depending on
access to antenna feeds. Some are described in Antenna and channel
response measurements on page 26.
Generating the interference signal
While the radio standards are evolving, generating a test signal that has similar characteristics may be more useful than trying to exactly emulate a particular signal. For a radio module with good RF isolation from the power
supply and baseband, a signal that is wide relative to the input bandwidth of
the DUT will often be sufficient. Different set-ups are needed for pulsed or
MB-OFDM simulation.
Frequency switched OFDM
Within the bandwidth of the victim receiver, a MB-OFDM signal can be
approximated with an RF-modulated noise source that is wider than the
victim’s input bandwidth. If it is switched on for 312.5 ns and off for 625 ns
it simulates a worst-case effect of the out-of-band UWB emissions changing,
for a three-frequency system. The equipment needed for this is:
• ESG-C with wideband ARB, noise Option 403 and external pulse
modulation input
• 3323x function generator
Set the bandwidth of the white noise to 80 MHz, using the internal noise
function in the ESG. Select External Pulse Modulation and use the function
generator to switch the ESG RF on and off. The PRF should be 1.066 MHz,
and the pulse duration ~312.5 ns. Figure 17 shows what this will look like.
External loss needs to be accommodated by increasing the RF output power.
The ESG output power should not increased beyond ~+10 dBm to avoid
compressing the signal peaks. For testing out of band, the RF level should be
adjusted to be the maximum allowed by the spectrum regulations for the
particular frequency range.
Figure 17. Pulsed noise spectrum and time (linear magnitude) displays.
Pulsed and pulse modulated
Within the receiver bandwidth of the victim, the spectrum of the pulsed
signal, as measured with a spectrum analyzer will look quite flat, but the
statistics of the time domain signal may not be Gaussian.
The real system DS-UWB system uses shaped pulses of approximately 1 ns
duration to synchronously modulate a local oscillator. The position, polarity,
and possibly even the shape of the pulse are changed to modulate data onto
the RF signal. Accurately simulating this with general-purpose equipment is
very difficult, but several approximations should highlight any sensitivity in
the DUT to this or other pulsed interference.
UWB pulsed source
A wideband solution is to use a high-speed pulse generator. The dual
channel 81134 pulse generator can be configured to generate a noise-like
bipolar data stream, or its two channels can be used to create IQ signals
using a timing offset between them. The equipment needed is:
• 81134 pulse generator
• either ESG-C with a suitable external double-balanced mixer, or
PSG Option 015.
The configuration is basically that shown in Figure 6 on page 11. The I or Q
input may be used with PSG Option 015, or both to create QPSK, if the dual
channel 81134 is available. Sensitivity testing can start with simply generating narrow pulses and noting any effect on the DUT, but this kind of signal is
not related to DS-UWB.
To create an approximation to the DS-UWB signal, Ch1 and Ch2 on the
81134 are coupled using a power splitter. Using the Data Mode, the data
patterns are programmed to provide the +1, –1 and 0 states required. The
channel output voltage needs to be doubled to take account of the loss in the
splitter. Timing differences between the channels can be corrected using the
Delay adjustment. A small DC offset may be needed to minimize carrier
Channel 1
Channel 2
The RF output should be set to give the required power spectral density
(PSD) in a 1 MHz bandwidth using an average detector (for example,
–41.3 dBm/MHz for center frequencies between 3.1 and 10 GHz.)
Restricted bandwidth pulsed source
As with the OFDM interferer example, this technique is based on creating
what the DUT’s receiver will be exposed to within its RF input bandwidth.
Just part of the spectral content of the DS-UWB signal is created using the
ARB in the ESG. This signal could be from a simulation or even a captured
waveform using the technique discussed in Use of time captured records on
page 39. Reducing the capture span will increase the maximum length of
time recording.
Figure 18 shows how the signal amplitude with a 100 MHz span is reduced
from the full bandwidth signal, and how the modulation looks more noiselike as a time-domain waveform. The pulse shaping is no longer visible. The
CCDF (see Peak output power, CCDF on page 44) confirms the signal amplitude is statistically more evenly distributed in a narrower analysis span.
Some evidence of the (4104 MHz) carrier may be found if the capture is done
based on the center frequency. Tuning away from the center frequency
removes this effect, since the carrier is no longer contained in the
measurement. The VSA software is able to perform this calculation even on
a captured time record because of the re-sampling algorithms it uses.
Figure 18. Spectrum and time domain plots of a DS-UWB-like signal for 3 GHz (top traces) and
100 MHz (bottom traces) frequency spans. The lower traces of voltage (middle) and CCDF (righthand) show the increasingly noise-like nature of the signal as the analysis bandwidth is reduced.
The band power markers used in Figure 18 indicate how the PSD may be
measured. They have been spaced more widely than the normal 1 MHz
requirement to avoid the spectral effects due to the very short FFT time
lengths. A 10 log scaling should be applied to the selected bandwidth to
convert to a 1 MHz bandwidth.
5. Component and
Network Measurements
Component impedance and reflection measurements
Unintended RF signal loss degrades the system operation and has to be
avoided. Impedance mis-matches (such as between the antenna and DUT
input), are one of the ways signal loss occurs. Matching takes extra attention
when operating over a multi-GHz range.
Using a very fast voltage step, or pulse, is a well-known technique for
examining the performance of transmission lines. It is a cost-effective and
intuitive way of assessing UWB impedance matching.
Figure 19. TDR reflections along a printed circuit board filter.
The basic assumption is that the bandwidth of the DUT is very wide and
goes down to low frequencies. Otherwise, the picture becomes distorted. As
the voltage step of the test signal travels down the line, if the line impedance
changes, some of the signal is reflected. The pulse repetition rate has to be
slow enough to cope with the longest delay expected.
Using an 86100 as the test tool, Figure 19 shows a typical response. In this
simple example it is easy to translate what happens on the screen to problems with the circuit. More advanced techniques have been introduced to
improve the accuracy of the measurement. See references in Appendix B:
Recommended Reading on page 72 for more details.
There are some situations when a voltage pulse approach may cause
• there is an active device in the circuit that cannot cope with the test
• the circuit is band-limited, which distorts the reflected signal, making the
display difficult to interpret
• there is RF attenuation before the circuit that is to be tested, which
reduces the amplitude of the reflected component
An alternative technique uses a vector network analyzer. The source power
can be varied and a tuned receiver gives increased dynamic range. Using an
inverse Fast Fourier transform it is possible to switch from frequency to
time domain. This technique has been further developed to allow advanced
“windowing” of specific parts of the system being tested.
The choice of frequency domain parameters affects what will be seen in the
time-domain window. If the S11 plot is affected, so will the time domain. For
• the highest test frequency determines the time/distance resolution
• the number of frequency points determines the resolution between
adjacent items on the trace
The measurement works on the principle that the voltage at one frequency
point is the vector sum of all the system responses as the signal progresses
through it. The frequency of the test signal therefore has to be stable for long
enough for all significant response to have settled. This may be a problem
that is not often encountered, but filters with steep frequency responses
may cause it to happen.
Antenna and channel response measurements
The IEEE process requires a channel model to be agreed upon, to allow the
comparison of different radio proposals4. A new model was needed because
earlier narrow-band models assumed frequency-independent scattering,
which becomes invalid over the very wide frequency range of an UWB signal.
The IEEE model is a complex expression, with the main variables being the
excess delay, rms delay spread, and the number of significant multi-path
Some of the simplest channel measurements use a sampling oscilloscope,
with a high-level pulse being used as the stimulus. A network analyzer is
normally used for more detailed analysis, and was one of the methods used
to derive the IEEE channel model. As shown in Figure 20, it is possible to
incorporate a separate, triggered RF source to allow for longer range testing.
A reference antenna allows antenna pattern measurement to be included.
Antenna receive
Antenna transmit
RF source
Antenna reference
Test IF
(to Rcvr A in)
PSG series signal gen
PSG trig out
PSG series NA
Reference IF
(to Rcvr R1 in)
PSG trig in
LO source
(from PNA source out)
PSG trig in
PSG trig out
Figure 20. Extended range test configuration for antenna pattern and channel response
measurements, using a vector network analyzer and triggered external RF source.
A third option is to use a real-time oscilloscope in combination with postcapture analysis. Subject to dynamic range limitations of the oscilloscope
and external amplification, this approach allows the most complete set of
channel data to be acquired. Capturing signals using a two-channel oscilloscope and then correlating the data between the measurements allows the
relative timing of the signals to be seen. Vector averaging allows the effect
of noise to be reduced if the channel is stable. Table 4 summarizes the
Table 4. Comparison of different techniques for UWB path loss measurements.
Sampling scope
and pulse
Network analyzer
Real time scope
(54855)and analysis
software (89601)
Simple. real time. Good
for intuitive
understanding if path
not too complex/test
signal pulses spaced
well apart
Used for IEEE model
vector information. Swept
sine wave stimulus. Use
FFT to extract time
Allows use of DUT
transmit signal. Good for
understanding time delay
Separation of
Limited, amplitude only
contributing to
the measurement
Good. Use standard
calibration techniques to
improve accuracy of port
measurements like S21
Good. Use equalizer to
remove antenna
response. Compare
equalizer data with model
Dynamic range
Highest, especially with
external source. Ensure
cables do not suffer RF
Limited by scope ADC
resolution. Requires
external LNA. Ensure
displayed trace amplitude
is as high as possible
Limited by broadband
noise floor or
interference signals
including unwanted LO
Use of equalizer characteristics
Fixed equalization is useful for making two measurement channels appear
to have identical responses. This allows you to make stimulus-response
measurements in either the frequency domain or the time domain. It is a
more powerful technique than frequency-domain normalization, but must be
applied with care. An example of using fixed equalization for two-channel
measurements is illustrated in Figure 21.
54855 Oscilloscope
Low noise amplifier
e.g. Agilent 8449B
Test signal
Power splitter
e.g. Agilent
Path or device under test
Figure 21. Using digitized signal for time correlation/coherence measurements.
To obtain stimulus response measurements using the 89601 VSA software
with a 58455 oscilloscope:
1. Connect a wide-band signal via a splitter to Channel 1 and Channel 2.
The signal must have energy (on average) throughout the band of
frequencies for which you wish to equalize. The best signals are white
noise, chirps, or noise-like digitally modulated signals (OFDM). They need
not be periodic.
2. MeasSetup > Average > RMS (Video)
Choose sufficient averages to reduce the display variance noise to a
satisfactory level.
3. Trace > Data > ChX > Coherence
The coherence should be close to one across the span. Coherence usually
increases with the number of averages. If it is consistently low in some
portions of your span, you may need to change your signal so that is has
energy in these areas.
4. Trace > Data > ChX > Freq Response > Save > D1
The Data can be extracted for external use.
5. Utilities > Fixed Equalization > Equalization > Ch2 > D1, or
Utilities > Fixed Equalization > Equalization > Ch1 > D1 > Invert
6. If you now repeat the measurement, the frequency response should be flat
in amplitude and phase.
Multi-path reflection and wavelets
Wavelet analysis is a type of windowed spectrum analysis. Unlike short-time
Fourier analysis, with its uniform time-frequency regions, wavelet analysis
divides the time-frequency plane into non-uniform regions.
In the context of UWB, the term wavelet has been used in connection with
multi-path propagation and the composite signal levels a victim receiver
experiences. This is of interest because of the need to understand the effect
of a large number of UWB devices on existing radio systems.
The configuration in Figure 21 allows the time correlation of such signals to
be examined. The impact on a specific type of radio depends on its bandwidth and modulation format, as described in Device test configuration on
page 20. As shown in Figure 18, the UWB signal is noise-like for a narrow
band receiver, but the instantaneous vector sum will also depend on the
frequency response of the path. Due to the correlation time intervals
involved, the envelope amplitude detector of a spectrum analyzer will usually
respond to multi-path signals as if they were uncorrelated noise. This topic
is still under discussion in the spectrum regulatory agencies, but may affect
the transmission power allowed for certain devices.
Differential network analysis
Driven by low voltage power supplies and the need to reduce coupling
between digital and analog circuitry, differential signal paths are rapidly
becoming a standard practice in radio device design. A UWB radio places
special demands on the measurement of differential components for two
reasons. First, building a practical balun with the necessary phase matching
is difficult. This means the differential connection cannot be easily converted
to a single-ended connection. Second, the separation between RF and IF/
baseband signal frequencies is reduced, making it more important to fully
understand any limits in the isolation of different components through mode
There are two modes the device has to be tested in: differential mode and
common mode. Figure 22 shows the signal relationships for a fully balanced
Differential to common
mode conversion
Differential mode signal
S CD21
Port 1
Common mode signal
- EMI or noise
Port 2
Common mode to
differential conversion
S DC21
Figure 22. Terminology for signals that apply to a differential device. Signals referenced to each
other are differential mode; signals referenced to ground are common mode. Some devices may
convert to a single-ended, unbalanced, signal on the input or output.
Vector network analyzer techniques have evolved to address differential
analysis, by adding additional ports, new measurements, and error
correction techniques. Using a single port stimulus, mixed mode analysis is
available. It allows the display of the conversion of signals in one mode to
another. Instead of just S21, the stimulus and response mode is now
specified. The example shown in Figure 22 is for SDC21 and SCD21. As with
the original S parameters, the convention is for the response mode to be
written first in the subscript.
For linear analysis, the measurement accuracy is excellent, and many
important characteristics such as common mode rejection, can be thoroughly assessed.
For active components, both linear and non-linear analysis may be required.
The generation of a true differential (180 degree) test signal over a very wide
bandwidth is not trivial. Careful consideration of the practical error mechanisms is needed.
Different methods have advantages and issues as summarized in Table 5.
Table 5. Summary of issues for three methods of driving a differential device.
Linear analysis
Non-linear analysis
Mixed mode
Independently drive each
port. Mathematically derive
and correct the results
Excellent parameter
coverage and accuracy
Good. Common mode input
parameters can show errors
at high signal levels. These
may not be of practical concern
Build a custom physical
part with best possible
performance, since error
correction is limited
Cannot isolate common mode information
Balun imperfections give
Fair, if it is possible to get
errors in differential mode
correct phase relationship
over the bandwidth to be
Hybrid junction
Single port drive, but
feedback allows linear error
Good. Design of hybrid
junction limits performance
Figure 23 shows the configuration for a new technique, using a hybrid
junction. The technique offers more than a plain balun, because the errors
from the hybrid junction can be removed from all the linear measurements.
Figure 23. Basic configuration for device test using hybrid junction.
Research continues to make differential device measurements more straightforward. Updated information may be found at
including references to hybrid junction suppliers.
Delta (additive) EVM
Conventional network analysis uses swept frequency or amplitude sine wave
test signals. Moving beyond this to analyze a system with a variety of life-like
complex test signals reveals more about the effect of non-linearities, and
makes the transition from modeling and simulation to real hardware
analysis more straightforward.
54855 Oscilloscope
& 89604A software
Set sample rate
to maximum
Test signal
Power splitter
Device under test
NOTE: The input signal may also
simulated data from ADS, MATLAB®, etc,
Figure 24. Test configuration for using 89604 distortion suite.
By using a time record of the voltage of a signal at the input and output of a
device, see Figure 24, the 89604 Distortion Suite software is able to show the
nature of the non-linearity, and provide the coefficients of a best fitting
Figure 25. Sample plot from 89604 software using an 802.11 OFDM waveform as the test signal.
89604 results include delta EVM, which provides a generic figure of merit. It
is different to the EVM in a radio standard (see Table 6), which makes an
assumption about the type of signal being used and the amount of equalizing
that is done to create the EVM reference.
For UWB devices, the EVM result may be limited by sampling noise. To
minimize this effect, ensure the sampling rate in the oscilloscope is set to
the maximum available. This increases the amount of averaging that can be
done on the data.
Table 6. Comparison of traditional and delta EVM measurements.
Traditional EVM
Delta EVM
Requires demodulator to detect bits
Uses one input channel as a reference –
no demodulation required
Bits used to synthesize a perfect,
noise-free reference
Does not assume an ideal signal, or that the
signal is a modulated carrier
Assumes ideal signal at the input to the DUT
Noise on the reference channel measurement may degrade measurement's accuracy
Computed at symbol intervals
Computed over all time samples. Reduces the
numeric value. No measurement filter to limit
signal bandwidth
In Figure 25, the time domain input and output waveforms are shown at the
bottom. The example used an OFDM signal as the stimulus. Gain and phase
distortion is shown at the top, including a best-fitting curve for a fifth order
polynomial. Figure 26 shows the polynomial extraction. The amplitude
probability density function and the CCDF are the plots in the middle of
Figure 25. The analysis done in this software removes linear phase and timing differences, but does not perform the equivalent of adaptive equalization
in removing linear distortion from the signal.
Figure 26. Example of configuration options and curve fitting results for 89604.
6. Transmitter
The measurements described below are split into groups according to the
type of DUT (pulse oriented or OFDM) and the test objective. The details of
many areas of testing are still being developed, but it is reasonable to expect
they will fall into the following categories:
Test objective
Output power, power spectral density
Peak output power, CCDF
Spectrum occupancy, spectrum mask
Adjacent channel performance
Modulation analysis
Frequency accuracy and stability
Range, in-band interference
Interference, interoperability, range
Out-of-band interference
Range, interoperability
Range, interoperability
Range, interoperability
Test conditions and measurement setup
Parametric tests of the antenna/channel and the transmitter or receiver will
generally be done separately. Unless carefully controlled, using an antenna
or a live network introduces considerable uncertainty. While a number of
the tests described here can be performed live, it is expected that generally a
cabled RF connection will be used. This is essential for repeatable receiver
sensitivity measurements.
Microwave signals act very differently from the audio and digital signals
with which many people are used to dealing. The reader is advised seek
advice if they wish to perform measurements, but are new to RF testing.
There are two main configurations used for testing the transmitter path.
They are distinguished by the signal interfaces, and the way the device is
controlled. One is suitable for RF/analog only circuitry, the other for a
complete UWB device. Figure 27 shows the configuration for the RF/analog
only case. Control of the circuit will require proprietary hardware.
If the baseband signal is three-level digital, it can be generated using two
channels of an 81134 pattern generator. Details are provided in the UWB
pulsed source description on page 23. The 81134 can be programmed to
create specific patterns, or generate very long random sequences.
An external ARB or a proprietary device (from a real radio) can be used as
the modulation source for an OFDM design. For modulation accuracy, the
measurement has to be able to recognize the format. The delta EVM technique described in Delta (additive) EVM (beginning on page 31) can use any
signal, and will not degrade the result because the test signal is not perfect.
An alternative is to adapt the modulation format of the test signal to suit the
analysis capability available.
swept spectrum analyzer
Device under test
RF up/
12 Mb data pattern/
pulse generator
20 GSa/sec 6 GHz
real time scope
Power amp
& detector
Ultra high speed
formatted data
e.g. LVDS
For +1, 0, –1
signal levels
I, Q
Note: The 2 channels of the 81134 can
be used to create either 3 level pulses
or I and jQ signals. Setting the dattern
pattern to a long prbs (up to 2^31)
creates a wideband noise-like signal.
Dual arb wavefrom generator
[> 250 MHz bandwidth]
IQ path calibration
Waveform generation
Figure 27. Transmitter test configuration for RF/analog circuitry.
Equivalent isotropic radiated power (EIRP)
The antenna in a real-life system may be designed to focus the transmit
power in certain directions and will have a radiation efficiency that
depends on the implementation. This can make it difficult to compare the
performance of different hardware. Therefore, some measurements refer to
equivalent isotropic radiated power, EIRP. Physical measurements involve
the use of a remote antenna for testing, which can be impractical for anything except (pre-) certification testing. The designer needs to understand
the individual transmitter and antenna characteristics. EIRP measurements
may require offset factors depending on the propagation in the test chamber.
See also wavelet notes at the end of Antenna and channel response measurements section.
The FCC regulations do not allow externally mounted antenna, for indoor or
outdoor use. ETSI require certification tests to include the antenna. See
Antenna and channel response measurements on page 26 for antenna
Interoperability testing
There are many transmitter parameters, which, if not controlled, can reduce
the performance of the UWB system, or even prevent different devices working together. Tests will be devised to help stop this from happening. These
are not yet available.
The transmitter tests are described first in this application note because
there are several problems in a transceiver that can be found more readily
by analyzing the transmitted output.
Examination of the block diagrams in Signal generation and modulation
(beginning on page 8), show why this is the case. The local oscillator(s) for
frequency up and down conversion is shared. Many impairments on the LO,
which could affect the receiver, will be visible on the transmissions.
Hardware probing
Debugging problems with modules requires very high-speed probes, and
often the signal lines will be routed as differential signals. The latest differential probing systems provide very wide bandwidth and good common
mode rejection. Single-ended and differential probe heads are shown in
Figure 28. For differential measurements to 6 GHz with Agilent spectrum
analyzers and network analyzers, see the Agilent E2696A general-purpose 6
GHz probing solution.
Figure 28. Photograph of 1134A single ended and differential probe heads.
When examining the practical implementation of a differential probe it is
found that there is a bonus. The bandwidth of the differential probe is
considerably wider than its single-ended equivalent. This is because it avoids
the problem of creating a very low inductance ground connection. Figures
29a and 29b show plots of key performance parameters for the 1134A probe.
Figure 29a. Frequency response of single-ended and differential probes.
b. CMRR of single-ended and differential probes.
Measurement triggering
Triggering on a pulsed RF carrier, for time domain measurements
The most robust way of making measurements is to generate a trigger signal
from the baseband circuit that is driving the DUT. Beyond prototypes, this
kind of signal is not always available, so level-sensitive triggering on the RF
has been used.
Pulsed UWB presents some unique challenges for stable time domain RF
measurements. This is because of a varying relationship between the phase
of the RF carrier and the modulation signal.
The trace in Figure 30 shows the problem. A fixed voltage trigger will fire at
different points during the pulsed waveform, causing jitter on the recovered
Trigger voltage
THoldoff = n/PRF – 1/fRF
= 3/75 – 1/400 µs = 37.5 ns
Figure 30. A 4 ns, 400 MHz pulsed waveform 75 MHz pulse repetition frequency (PRF) rate.
Trigger hold-off may offer some improvements to the stability of the displayed waveform, if the pulse modulation and the RF carrier have a defined
and stable relationship. This will require them to share a reference frequency oscillator.
On the oscilloscope, the trigger hold off value should be set to
THoldoff = n/PRF – 1/fRF
where n is the lowest common multiplier of the RF and
pulse periods.
Once a stable trigger is established, envelope detection can be used for
amplitude-only measurements. Statistical distribution analysis may help
identify unexpected behavior in the pulse. Figure 31 shows an example of a
double pulse using the 81134 pulse generator.
Figure 31. Double pulse, 400 MHz carrier (75 MHz PRF).
As noted in Pulse modulated RF (page 11), post-capture analysis software
allows the envelope of the modulated signal to be displayed, suiting more
complex modulation formats. The trigger issues are eased, although some
extended trigger capability of the scope can be used to trigger on more complex events. Figure 32 shows the user interface available to do this. SCPI
commands for the oscilloscope are typed into the command line.
Figure 32. Entry fields for complex trigger commands in 89601.
Note: Allowable trigger settings depend on the hold-off type being used by the 89601
before the customer trigger command is entered. Select <Low Duty Cycle> and enable
Triggering in the main user menu before entering new trigger conditions. There are
differences in trigger hold-off operation when using 89601 and the oscilloscope
standalone. Check details in the 89601 on-line help.
Triggering on MAC data frame and MB-OFDM symbols for spectrum measurements
The spectrum of any modulated RF signal changes with time. In many burstbased radio systems there are specific events, such as the preamble, which
have quite different spectral characteristics to the data content. As shown in
Figure 33, there are a multitude of different timing intervals from which to
select. In WLAN, some of these can be measured using a conventional
spectrum analyzer and time-gating; with UWB they are likely to be too short
but some experiments may be done. Measuring such specific events requires
the use of a trigger signal. Many spectrum analyzers already generate this
internally using envelope detection. It may be wideband in the same way a
power sensor is wideband, but it is the video bandwidth of the trigger circuit
that determines how fast a pulse can be reliably triggering. Trigger options
vary with the model of spectrum analyzer. Refer to the spectrum analyzer
block diagram in Figure 44 to consider what trigger signals may be available.
Interframe spacing
[IFS] - variable length
MAC frame [variable length]
MB-OFDM symbol periods
Signal envelope
during each symbol
Inter symbol drop-out
Figure 33. Timing intervals and dropouts in signal envelope for MB-OFDM.
An RF frame is typically 200 to 1000 µs long, with a highly variable interframe spacing. In MB-OFDM, each symbol is 312.5 ns long, transmitted on a
different frequency at a rate of 457 kHz or 1.066 MHz. The envelope trigger
bandwidth of the ESA/PSA spectrum analyzers is fast enough for these
signals, but not individual pulses. An oscilloscope combined with 89601 software should be used for pulse spectrum diagnostics.
Frequency selective triggering has historically been achieved using the video
trigger. In UWB testing a problem can apparently arise because signal path
switching means the video trigger is available when the peak detector is
used, but may not be when using the average detector. In practice, the swept
analyzer’s peak detector is likely to be sufficient for this kind of diagnostic
Use of captured time records
By combining the 54855 digitizing oscilloscope with the 89601 software, it is
easy to capture signals of particular interest. Once captured, this technique
• analysis using multiple parameters, regardless of the settings during
• slowed down replaying of the signal spectrum, to identify specific events
Meas Setup > Time > Max Overlap (A larger number slows down the
trace update rate)
Control > Player displays a running pointer of what part of the time
record is being displayed
• change of analysis center frequency, span, and measurement bandwidth
after capture, as long as the desired span is within the initial capture
• troubleshooting using a remote expert, by e-mailing the captured file
• transfer to ADS simulation software and integration into device models
• creation of the signal using an ARB and PSG combination (within the
limits of modulation bandwidth), for many forms of device testing
including interference tests
Figure 34 shows the test configuration to capture data like this. See also
OFDM on page 58.
PSG signal
Dual ARB
Real signal
analysis and
Create RF
Figure 34. Method for capturing signals for troubleshooting or for use in receiver testing.
Test modes
Test modes are invariably used during prototyping stages of a design. They
are designed to allow verification of isolated system components without
requiring the whole radio to work, and may be needed for certification
testing. Modified versions of these tests may also be used to manufacture
sub-system components.
Some standards incorporate over-the-air test modes, such as signal loopback,
to ease type conformance testing, and receiver testing in particular. While
test modes are an additional development task, they significantly ease the
path of a radio from R&D through to integration in the host device and
At the time of writing, no standardized interfaces have been defined for the
UWB radios under discussion. Table 7 has been included to indicate what
functions have proved useful historically in design evaluation. A number of
transmitter test functions are usually mandatory to confirm the DUT meets
spectrum regulatory requirements.
For device testing, even simpler test signals may be used, for example
selecting specific groups of sub-carriers in OFDM-based systems.
Table 7. Basic test mode functions.
Test function
Device control
Output power
Transmit power control
Max power used for regulatory test.
Simple transmitter test using a power
Bursting on/off
meter. Generally tests are best carried
out with bursting on.
Hopping off (where applicable) Allows in-band spectrum mask testing
Spectrum characteristics PN9, 15 data sequences
Whitens signal. Use value of 0 as the
seed, for repeatable results
Defined bit patterns, 0, 1, 01,
Allows identification of specific issues
10, PN9, PN15
Scrambling/Encryption on/off Reduces reading to reading variations of
spectrum and EVM
Hopping off (where use)
Ack packets on/off
The DUT should be able to recover
arbitrary packets, or define required
Ack packets can give a simply way of
externally checking PER. Switch them
off to increase test speed
IEEE 802.11 WLAN receiver testing has not been standardized because no
test mode was defined. Even without a loopback mechanism, UWB testing
will be made more straightforward if the DUT is made to respond with an
ACK packet to a properly configured, but isolated, test packet. The payload
should be chosen to be easily generated, such as a repeating PN sequence
with a “0” seed.
All the power measurements described here are affected by the loss and
impedance mismatches of cables and other RF components used in the
measurement set-up. It is important to use parts suitable for the frequency
range. Even what looks like an SMA power divider may only be a Tee Piece.
The simplest, and most accurate, way to record the true average power of
any signal is to use a power meter with a thermal sensor. For IEEE 802.15.3a
radios, the result will be in the region of –10 to –3 dBm. The drawback is
that it tells you very little about the characteristics of the signal against time
and against frequency, which is what is important for an underlay technology.
Distinguish between RF bandwidth and video (demodulated) bandwidth
The RF measurement bandwidth is not the same as the video (demodulated)
bandwidth. Depending on the signal’s timing characteristics, a peak power
meter can be used to show the response against time, but again not against
frequency. A wideband digitizer (oscilloscope) can give both. There may be
some differences in the measurement results compared to a swept spectrum
analyzer due to the way the signal is detected. Table 8 summarizes the
options to measure different characteristics.
Table 8. Equipment choices for measuring UWB signal power characteristics.
Power meter
with thermal
Power meter
with peak
Swept spectrum No
detector needed)
Oscilloscope and
True peak
and CCDF
Basic power
Very good
bandwidth of
signal shown
Power spectral density, average detection
Power spectrum density (PSD), is the main regulatory performance test for
an UWB transmitter. It measures the power within a narrow portion of the
spectrum. PSD can be measured in an arbitrary bandwidth. It is often scaled
to dBm/Hz even though the measurement bandwidth is not 1 Hz. If the
measurement bandwidth is narrow however, measurements take longer. If
it is too wide, the measurement may not identify unwanted peaks in the
spectral response.
In the United States, FCC document 02-48, (CFR Part 15, August 2003,
Appendix F), requires measurement of the 1 ms time-averaged PSD of an
UWB transmitter in a 1 MHz bandwidth, and an assessment of the peak
power in a 3 MHz bandwidth. Both measurements anticipate the use of a
swept spectrum analyzer. By ensuring the PSD of the DUT does not exceed a
pre-defined maximum value across the permitted frequency band, an upper
limit is set on the signal-interferer ratio seen by another receiver operating
within the UWB frequency band. A different test safeguards those devices
operating at other frequencies. Graphically, a PSD measurement looks like a
spectrum plot.
Spectrum flatness determines total transmit power
Working with a limit value that applies to the whole usable frequency range,
the designer has to ensure the transmitter generates as flat a frequency
response as possible. This gives maximum total power, and therefore the
optimum transmission range for the user.
The example in Figure 35 is of a poor noise modulated signal suffering both
from LO feed-through, and amplitude unflatness. The output power in the
signal would have to be reduced to allow the DUT to pass the PSD test.
Carrier leakage spur
Unflatness in
(due to mixer)
Figure 35. An example of poor PSD flatness.
Sweep time
For regulatory tests, the sweep is not triggered by any part of the data
structure with a packet. A measurement interval of 1 ms determines the
sweep time. The signal should be within any given 1 MHz portion of the span
for 1 ms. This substantially removes the effect of modulation artifacts, and,
when present, rapid frequency switching. For a 1.5 GHz span, the sweep
should be 1.5 s.
Use of average (rms) detector for power measurement
Only more recently designed spectrum analyzers implement an average
detector. It is important to realize the measurement result truly is the average power calculated for each part of the span. It means if the DUT is not
transmitting continuously on one frequency, the trace position will shift.
Frequency switching or packet based transmissions cause non-continuous
Using an rms detector, it is also possible to measure the average power of
the RF signal within any user-chosen frequency range, and get the same
reading as a power meter. The plots in Figure 36 show the results of measurements on a broadband noise source. One is on a fixed carrier frequency.
The carrier in the other is rapidly hopping between two frequencies. The
levels recorded on the traces and the band powers are reported correctly. It
serves to indicate this kind of measurement is only indirectly sensitive to
the signal pulsing on and off.
Still 0 dBm average power
0 dBm average power
from band pwr markers
Figure 36a. A 5 MHz noise source on a 500 MHz carrier.
b. Same signal hopping between two frequencies, with 50:50 ratio.
In Figure 37, the time spent on the lower frequency has been increased to
67 percent. Whether the signal is pulsed or hopped, the trace level drops
from the static case according to the expression
10log(ton/(ton+ toff))
Note: There is a 0.25 to 0.5 dB difference
between the normal marker (using average detector) and the noise power marker,
due to slight differences between the
noise BW and the RBW filter’s 3 dB
bandwidth. See Application Note 1303
for details.
When measuring noise-like signals, the average detector behaves well and
gives the results one would expect, but older spectrum analyzers may not
have such a detector choice. Swept spectrum measurements of pulsed RF
signals on page 46 looks at some of the differences that will be seen, and
provides important notes about the video bandwidth setting. Having
configured the spectrum analyzer to use the rms detector, a marker can be
used with the peak search function to find the maximum PSD.
0 dBm total average power
67:33% ratio
= -4.8 dB
Figure 37. Frequency switched signal 67:33 ratio.
Peak power measurement using a swept spectrum analyzer
Unless the resolution bandwidth exceeds the occupied bandwidth of the
UWB signal, a swept spectrum analyzer cannot truly measure the signal’s
peak power. It can, however, approximate the response of another radio.
Some measurement methods refer to a 50 MHz RBW. The reason is to make
the RBW at least as wide as the widest victim receiver. In practice, great
care needs to be taken if the results obtained are to be predictable and
repeatable. The accuracy of the RBW filter generally degrades for wider
bandwidths and the video (impulse) response may not increase to match it.
Setting the analyzer to zero span should allow the amplitude step and
impulse response to be examined.
A peak measurement with a smaller bandwidth can give a useful indication
of DUT transmissions that might cause problems with common radio
systems. A 3 MHz resolution bandwidth is typically sufficient to perform
this type of measurement. The FCC specify that the result should be scaled
to a 50 MHz RBW, using a 20.log(RBW/50) scaling factor.
Details of how UWB signals interact with a swept spectrum analyzer are
discussed in Transmit output spectrum on page 46.
Peak output power, CCDF
The result obtained when measuring the peak of a signal depends on the
bandwidth of the detection system (see Figure 18 on page 24) and, if the
signal varies with time, how long you are prepared to wait.
Power meters and spectrum analyzers can identify bursts, and simulate the
effect on most other radios, but are unable to capture the true peak of a
UWB signal, because their resolution and/or video bandwidth is too small.
Compressed signal peaks will degrade the link.
Using a high-speed oscilloscope as a digitizer, it is possible to capture the
complete signal. Since the signals do vary with time, and linear devices will
find it harder to deal with peaks, the next requirement is to plot the power
on a scale that shows how often the signal reaches a particular level. This is
what is done with the CCDF. The measurement has to be gated to only show
what happens when the signal is present. Figure 38 shows some indicative
results for pulsed and OFDM test signals.
Average power
Figure 38. Peak power, average power, and CCDF of full bandwidth OFDM-like (top) and
DS-UWB-like signal envelopes (bottom).
Baseband versus envelope (zoom) CCDF
Traditionally, the CCDF curve plots the power distribution of the demodulated envelope of the signal. Using an oscilloscope as the input device, it is
possible to see the baseband CCDF too. It is not the same. The difference is
due to the number of degrees of freedom in the amplitude distribution of the
As an example, to help visualize why the plots are different, a 50 percent AM
signal is shown in Figure 39. The demodulated waveform is on the left side.
The lower plots show the amplitude probability density function, which is
closest to the waveform seen on an oscilloscope. In qualitative terms, the
broader shape of the modulated signal in the lower right corresponds to the
wider range of voltage excursions, and the wider peak-average power in
shown in the CCDF plot. For background information on CCDF plots, refer
CCDF of demodulated
sinewave with DC offset
Peaks associated with
sinewave signal
CCDF of the full am
modulated carrier
Wider skirts due to more complex
amplitude distribution of the carrier
and its modulating sinewave
Figure 39. Envelope and baseband CCDFs of a 50 percent AM waveform.
Transmit output spectrum
The power spectrum density and peak power indication measurements
described earlier are a subset of the spectrum measurements that may be
made. Often it will be necessary to look more deeply at the signal, and
compare results from simulations and real devices. The low power of UWB
devices for commercial communication can require attention to the signal to
noise ratio of measurements. Since there are many ways a UWB signal can
be created, and the spectrum measured, this section describes the main
techniques and why results can vary between them.
Swept spectrum measurements of pulsed RF signals
When operating normally, neither DS-UWB or MB-OFDM implementations
are simple pulsed UWB signals, but some aspects of their operation can
cause a swept spectrum measurement to respond as it would to a pulsed
system, particularly when using test modes.
Agilent Application Note 150 describes how the pulse repetition frequency,
pulse duration, and resolution bandwidth determine the display seen on a
swept spectrum analyzer. Figure 40 shows these interactions using a peak
detector. The video bandwidth setting is Auto, the sweep time manually set
to 500 ms.
Trace A is the simplest. The PRF is high compared to the RBW, producing
discrete spectral lines. The amplitude of an individual line depends on many
pulse-shape factors, and the RBW of the analyzer. If the RBW is reduced
from 1 MHz to 100 kHz, the signal display level drops by 20 dB. This is why
it is essential to define the RBW used for a measurement. For the same
change in RBW, the noise only drops by 10 dB. Keeping the RBW wide may
help to minimize measurement variations due to noise.
DS-UWB a special case
The very high-speed, very short duration RF pulses of DS-UWB put it into
the trace A category of signal. However, the spectral spreading from the
randomized BPSK modulation makes the signal appear more like noise. (See
Figure 18 on page 24.) The S/N of the peak-detected signal may not improve
as expected when the RBW is increased.
Trace D shown in Figure 40 is what would be seen for an un-modulated
MB-OFDM signal, pulsing on just one of the RF carriers. The amplitude rise
time determines the spectral width. In practice, this will represent the
spectrum at the start and end of each OFDM symbol, unless the modulation
is adapted for spectrum shaping.
Pulse repetition frequency
Pulse duration
ton << 1/RBW
Trace increases 20logRBW
Trace level unaffected by the RBW,
but varies as 20logPRF
ton ≥ 1/RBW
MB-OFDM carrier switching (no modulation)
Figure 40. Swept spectrum peak detector response to pulse signals with various PRF and pulse
duration. Note: These plots are for pulsed signals that do not have any modulation applied.
Pulse de-sensitization
This term dates back to the first RADAR spectrum measurements. Some
people find it misleading, because it has nothing to do with compression in
the spectrum analyzer. It refers to the effects shown in Figure 40, when the
display results do not directly reflect the actual signal power.
Effect of increasing the resolution bandwidth on display level
As noted, if the pulse duration of a repetitive signal is much less than
1/RBW, increasing the RBW increases the detected signal level. Figure 41
plots the transition between trace A and B in Figure 40.
Figure 41. Effect of changing RBW using peak detector with pulsed RF signal (ton<<1/RBW).
Peak and average detection of UWB signals
The transmission of a UWB device for communication should be noiselike. That is what allows it to be an underlay technology. In practice, real
transmitters will produce unwanted, discrete spurious signals. The average
detector, which is useful for noise-like signals, may not show the fixed
spurious components at the expected level.
The test configuration is that shown in Figure 6 on page 11. The PRF and
RBW have been chosen to highlight the differences between spectrum
displays using peak and average detection. It is an extreme example.
Peak detection
Peak measurements are very useful in indicating spectrum occupancy, but
generally not ideal for an absolute level indication when the signal is noiselike.
Figure 42 shows how the spectrum is made up from some form of UWB
signal (it would not be possible to tell from this picture alone) and discrete
spectral components, based on the mixer local oscillator and pulse clock
The amplitude of the noise and line spectra conform to the notes in Swept
spectrum measurements of pulsed RF signals on page 46.
Figure 42. Mixed noise and spurious peak detection.
Average detection
Figure 43 shows the average detector response measuring the same signal.
The spectrum shape is now correctly displayed, and the band power function
allows the total signal power to be easily measured. Some spurious signals
are evident, but their level is noticeably reduced from the trace using a peak
detector. The result is not wrong, but simply shows the true average of the
signal while the frequency is being swept. The average result depends both
on any time variation in the test signal, and the ratio of frequency span to
the number of display points (buckets). If the RBW is small compared to the
width of a bucket frequency span, a fixed frequency component will only be
detected for a fraction of the time corresponding to a specific display point.
Increasing the number of display points reduces this effect. If RBW > Freq
Span / 2*Display Points, the result will only depend on the time variation of
the signal.
Figure 43. Mixed noise and spurious average detection.
Average detector settings [PSA]
In the PSA, there are three choices for how the average is calculated. These
suit different types of signal. The rms setting should be used to measure
UWB signals.
Normal detector [PSA]
This is similar to the peak detector, but fills in more of the display to make
the result look like a purely analog spectrum analyzer. It is recommended
that the peak detector be used instead of the normal detector for UWB.
Mixed detector display
A simultaneous display addresses the need to see both signal responses
together. Figure 44 is from an enhanced display from the PSA spectrum
analyzer family.
Figure 44. Dual detector trace, showing a 660 MHz PRBS amplitude modulated signal with
unwanted spurious components.
The test configuration used for Figure 44 was that of Figure 6 on page 11.
The 800 ps rise time of the 81132A pulse generator used for the measurement determines the spectrum width seen in this display.
Pulse generator settings
Channel 1 of the pulse generator is used as the clock for the PRBS sequence
defined in Channel 2. The configuration for this test was:
Clear existing settings: Shift > Store > 0
Levels: Ch1 > High 450mV > Low –22mV > On > Ch2 > Off
Mode/Trg: Continuous > Pattern of > PRBS > 2E15-1 > Pulses Out 1 > NRZ
> Out 2 > RZ
Timing: Ch 1> Frequency > 660MHz > Lead Edge > 0.8ns
Pattern: Segment > Length > 4096
Spectrum analyzer settings
The UWB spectrum mask test is reached using:
Mode Setup > Radio Standard > UWB > UWB Indoor
The default settings can be adjusted to suit specific test needs by modifying
the range tables:
Measure > Spurious Emissions > Meas Setup > Range Table
The resolution bandwidth settings are the same for both traces in Figure 44.
It is possible to run sequenced tests with different settings using the ranges
in the spurious measurement. Preferred settings may then be saved using
File < Save < State.
Comparing FFT-based and swept spectrum results
Swept spectrum analyzers are very commonly used in practice, because the
dynamic range of the signals they can measure is far larger than that
obtained using a digitizing scope. The measurement frequency range can
easily exceed the UWB requirements, but because it is swept, it only views
part of the spectrum at any instant in time.
In simulations, and when using a time record from a digitizing oscilloscope,
the spectrum will be generated using an FFT. It can be the most informative
view of the spectrum, but often looks different to the spectrum seen on a
conventional swept spectrum analyzer.
The factors affecting what is displayed are
• the bandwidth and shape of the resolution bandwidth filter
• the amount of time data is collected for each frequency display point
• the point in time (during the frame) when the signal is sampled
• the way the signal is detected
Table 9. Spectrum measurement characteristics used in different tools.
Agilent Filters
example Resolution bandwidth/
Detector type
swept analyzer
Digital swept
Gaussian approximation
Selectable. Applies
display averaging
with average detector
Flat top
impulse (uniform)
Display averaging
(max hold)
Post capture
flat top
uniform: Not suitable
for spectrum analysis
Display averaging
(max hold)
Gaussian/Synchronously Selectable.
Gaussian approximation when using average
FFT (used for low RBW) to avoid log detector
Gaussian filter, average detector
Table 9 shows the variety of possibilities, not including simulation tools.
Selecting a Gaussian filter is the first step to getting the same results. Note
also from Table 9 that FFT-based solutions do not generally emulate the
peak detection function in a swept analyzer. This document will consider
average detection only.
With an FFT, the amount of time data used for a particular RBW is determined by
ENBW = normalized, equivalent noise bandwidth
(2.2 Hz-sec for Gaussian1)
RBW = the resolution bandwidth
T = the time-record length
An FFT spectrum is a time-gated view of the signal. For a 1 MHz RBW, only
2.2 µs of data is required. As seen earlier, in Figure 11 on page 14, the
spectrum changes shape dramatically over time, so the result depends on
when the FFT is triggered.
The physical components in a swept analyzer strongly influence the measurements that are available, and these vary from one design to another.
Figure 45 shows the main system components, including RF path switching,
low noise amplification, triggering options, and signal detection.
Figure 45. Block diagram of a swept spectrum analyzer. Like any tuned receiver, the local
oscillator is tuned to select specific frequencies. The output of the signal mixer is fed to either
analog or digital processing. In this diagram, the switching path for frequencies above 3 GHz is
also shown.
The rate the swept spectrum analyzer tunes over the chosen frequency span
(sweep speed), determines how much time data is used to represent the
spectrum at a specific frequency. The fastest sweep speed is set by the time
response of the RBW filter.
Sweep time = 2*span/RBW2
For a 1 MHz RBW, this gives a sweep rate of 500 GHz per second. This means
a 2 GHz span will take 4 ms. If the display is divided into 401 points, each
display point will show the combined effect of the spectrum at that point
over 10 µs.
Note: The PSA series use special techniques to increase the sweep rate by a
factor of ~2.
This is five times as long as the FFT would require, and implies more averaging is taking place. The regulatory requirements may require much longer
measurement periods, such as 1 ms per display point. The overall effect is
that a lot of somewhat hidden averaging takes place in the swept spectrum
Getting the FFT view to look like a swept analyzer
The main need is to ensure a similar amount of averaging takes place and
that any unintended gating effects are taken into account, especially if
spectrally-unusual events like the preamble of inter-symbol ramping are
included. The response of a peak detector can be approximated using a Max
Hold function based on multiple FFT results.
Practical effects, such as the exact processing used in real instruments mean
considerable care will be needed to achieve better than 2 dB matching
between results. A cross-reference using a defined modulation pattern can
be used to test the results.
Spectrograms and adjacent channel power measurements
The time gating inherent in an FFT-based spectrum measurement can give a
powerful insight into the dynamic characteristics of UWB signals. As an
example, Figure 46 shows a spectrogram of a frequency switch OFDM signal.
It comes from the 89601 software, running on the 54855 oscilloscope.
The spectral disturbances at the symbol transitions are clearly shown as
horizontal lines. The rising and falling edges of the bursts in a real device
would need to be controlled to reduce this effect.
Figure 46. Spectrogram of MB-OFDM.
Unless the frequency switching is turned off, the adjacent channel leakage of
a MB-OFDM signal needs to be made as a time-gated measurement. This
component is highlighted in Figure 46. The bandpower markers available in
the 89601A VSA software can be readily used to measure the relative power
levels of the wanted and unwanted signal components.
Two channel (correlated) spectrum measurement
Since the spectrum of an UWB signal is noise-like, and may have a PSD close
to the measurement noise floor, it is useful to consider techniques that can
distinguish between true random noise and the wanted signal.
The 89601A and 54855A combination provides this opportunity. Using the
same test configuration described in Antenna and channel response
measurements on page 26, Figure 47 shows an UWB signal used as the
Channel 1 reference waveform, while Channel 2 is a much lower level signal
(fed through an LNA). The Channel 2 signal is below the noise floor, but
using the frequency response function, its level relative to Channel 1 can be
Test signal (buried in noise)
0 dB
-40 dB
Correlated response
Figure 47. Correlated spectrum measurements.
Spectrum mask testing
Measurement of the RF spectrum generated by a transmitter addresses two
• Will the DUT interfere with other radio receivers?
• Will the DUT work effectively with another of the same type?
Spectrum masks specifications for interoperability have yet to be ratified,
and only make sense for MB-OFDM. Preliminary information for MB-OFDM
is –12 dBr with ±285 MHz offset from the carrier, and –20 dBr at ±330 MHz.
The reference level is the maximum PSD within the range ±260 MHz of the
center frequency. The detector type and sweep time have not been specified.
If an average detector is used, the mask measurement may require either a
time gated sweep, the DUT to be transmitting on a single frequency. The
absolute signal level will drop according to the mark-space ratio of the signal.
Out-of-band emission masks are currently only defined for the United States.
Those for indoor and outdoor devices are shown in Figure 48. The resolution
bandwidth may be changed depending on the frequency range being tested.
The notes in the differences between peak and average detectors should be
read prior to making these measurements.
UWB emission limit for outdoor hand-held systems
Figure 48a. Indoor FCC and proposed ETSI spectrum masks.
b. Outdoor FCC spectrum mask.
Below 1 GHz, a different detector is used, known as quasi-peak. For further
details on EMC measurements, refer to Application Note 1328.
Modulation tests
Baseband pulsed
Baseband signals may be measured using the time-domain tools available
within a digital oscilloscope. Features that may be useful include jitter
distribution and software clock recovery. Figure 49 shows examples of these
capabilities for the 54xxx family.
Figure 48a. Time domain jitter analysis.
b. Setup window for the software clock recovery application.
Pulse modulated RF
Figure 8 on page 12 shows how the pulse shape of a modulated RF carrier
may be displayed. It is possible to recover more information than just the
shape of the pulse. Which parameters give meaningful results depend on
the type of modulation. Figure 50 shows an example of a three-level BPSK
signal. In this case the 0 state is not recovered correctly, but useful qualitative information can be seen that will allow the differentiation between clean
and noisy signals. This plot shows the recovered baseband signal. The
data behind this trace can be extracted and used for post processing.
Use: File > Save > Trace.
Figure 50. Digital demodulation applied to three-state BPSK pulsed signal.
When amplitude and phase information is required, a technique like this is
essential. It can be applied to pulsed signals in general. Figure 51 shows the
double pulse of Figure 31 (see page 37) after is has been demodulated in
Figure 51. Demodulated double pulse
For sub-nanosecond pulses, the displayed time resolution may affect the
result. How significant the effect is depends on the actual bandwidth of the
test signal. The sampling rate for the demodulated result is one third as
large as that for the data capture. Part of this is due to the splitting into IQ
data pairs, the rest is related to data windowing.
The trace data can be extracted for post processing. To enhance the results
use averaging and increase the effective sampling rate.
Time alignment of capture waveforms
The reference point for phase in a captured waveform can be adjusted using
math functions on the trace data. An IQ plot will show the phase alignment
of the signal. In Figure 52, an example is shown that shifts the phase by
–90 degrees.
Figure 52. Use of math function to adjust captured results.
Ultimately, a demodulation measurement of the adopted standard will
provide the widest range of modulation performance indications. Prior to
that being available, there are a number of characteristics that can be
checked using existing tools. Some, such as delta EVM and CCDF have
already been described.
Time based characteristics of a frequency-switched signal are shown in
Figure 53. Figure 53a is a plot is of the un-modulated carrier. The plot in
Figure 53b has modulation applied. Selecting Group Delay as the
vertical parameter gives a frequency versus time trace.
Figure 53a. Frequency versus time results for frequency-switched RF carrier.
b. Spectrum versus time of modulated OFDM signal.
The MB-OFDM implementation is incompatible with existing format-specific
measurements and some valuable parametric results they show. Depending
on the component being tested, it may be possible to use a over-clocked
802.11a signal as an alternative test signal. This allows characteristics such
as settling, channel flatness, and pilot responses to be assessed. In Figure 54,
an 802.11a signal is being shown running 32 times its normal rate. This
means it occupies over 500 MHz of bandwidth.
Selecting MeasSetup > Demod Properties > Advanced allows the subcarrier spacing to be increased to 10 MHz, making the x32 signal compatible
with existing 89601 measurements.
Figure 54. Measuring demod characteristics, including channel flatness, using a x32 time scaled
802.11a test waveform.
Extending the capture period
Using the full sample rate of the oscilloscope to capture the entire RF signal
restricts the measurement period, and increases the amount of data that has
to be processed. Down-converting the RF signal to a lower frequency can be
used to either extend the capture period, or introduce some over-sampling.
Over-sampling tends to increase the dynamic range of the measurement
because it allows wideband noise to be averaged out.
[4-20 GSa/s] Digitizing scope
Wideband mixer:
Mini-circuits ZEM 4300,
Marki M2-0006MA
or similar
Signal generator generates an
LO of RF-500 MHz, +7 dBm
3 dB
DUT output
Figure 55. Down-conversion allows a lower sampling rate, and extended capture times.
In Figure 55, the ESG is set 500 MHz below the center frequency of the
signal from the DUT. This gives 1 GHz of measurement bandwidth. To avoid
aliasing, the DUT transmission needs to be filtered, or the measurement
gated, if frequency components are present more than 500 MHz from the
Depending on the level of the errors in the down-conversion path, it may be
possible to improve the measurement accuracy using normalization, or the
equalization described in Antenna and channel response measurements on
page 26.
This technique also eases triggering, by isolating individual symbol frequencies,
but it does not accommodate multiple symbol measurements on a frequency
hopping signal.
Frequency measurement
CW and long pulsed signals
Test modes are often used to switch the RF carrier on continuously, and
thereby allow the frequency to be measured without any special techniques.
Measurement of static errors in the crystal reference frequency are suited
this approach. The typical performance2 required is 20 ppm, which is
straightforward to achieve.
Figure 56. Using the FM demodulation in 89601.
Frequency measurements made on a pulsed signal will offer more insight
into how the DUT operates in practice. The length of the pulse will determine how much averaging can be done to reduce variations in the readings,
and hence the useful frequency resolution.
Many measurement methods are possible, with different requirements for
triggering and gating the frequency count interval. The FM demodulation
algorithm in the 89601 automatically recovers the center frequency. Shown
in Figure 56, this measurement is useful for checking the un-modulated
carrier of frequency switched OFDM. Phase stability may also be displayed.
In the example, a tuning frequency offset of 1 MHz has been deliberately
added. The markers show the corresponding –36 degree phase shift over a
100 ns interval. Note that the carrier frequency is still correctly reported.
The configuration of the FM demodulation is as follows:
MeasSetup > Center Freq: enter nominal value
MeasSetup > Freq > Span: 1.75 GHz (wide enough to avoid limiting main
time length)
MeasSetup > Time > Main Time: 150 ns (must be longer than the pulse
MeasSetup > Demod > Analog Demod: FM > Auto Carrier Frequency
Traced > Data > Ch1:Demod > Spectrum
Market > Function > Auto Carrier Frequency
OFDM modulated signals
The measurement of a modulated OFDM signal requires preamble recovery. If
the specific demod format is not available, an interim step is to scale the
clocking rate of the DUT transmission to approximate an 802.11a signal. In the
example shown in Figure 55 on page 58, it was multiplied by 32, to give a 10
MHz sub-carrier spacing. This will give a useful indicator of the DUT settling
characteristics, such as the frequency perturbation caused by the transition in
preamble sub-carriers, shown in the center trace. In Figure 57, the markers
are coupled to make it easy to see where in the time record the frequency
error occurs.
Figure 57. An 802.11a signal adapted to show preamble settling, by multiplying clock rate by 32.
Meas Setup > Demodulator > Wireless Networking: OFDM
Meas Setup > Demod Properties > Advanced > Sub-carrier spacing: 10 MHz
Trace > Data: Preamble Freq Error
Short pulsed signals
Measuring the frequency error of a very narrow pulsed signal can be difficult
because the gating period only encompasses a few RF cycles. It is made
straightforward using a different technique, still using the 89601 software
and the 54855 scope. The configuration adopts the same approach shown in
Delta (additive) EVM on page 31. We assume the signal is BPSK and set the
symbol rate to the pulse repetition frequency.
Figure 58 shows an example of a 200 MHz carrier that has been amplitude
modulated with a 10 ns pulse. A 3 kHz frequency error was deliberately added
to the signal and this is reported in the symbols/errors section of the display.
Figure 58. The 89601 can be used to show the frequency error of a narrow pulsed RF signal.
The basic configuration used is:
Meas Setup > Demodulator: Digital Demod
Meas Setup > Demod Properties: BPSK (set symbol rate, measurement
period, and filter types to suit the signal being tested)
7. Transceiver Spurious
The use of very high-speed digital circuitry means the overall system
emissions are often a combination of analog and digital effects. The tests,
described only in outline here, are often time-consuming and require close
attention to measurement configuration. Control lines that are nominally
digital can easily become unexpected antennas when RF signals couple onto
them. Unexpected variations in results often indicate RF signals being
present on cables.
Transceiver measurements consist of performing out-of-band spurious
emissions tests. These confirm the UWB radio is operating within regulatory
limits. Spurious emission testing can be performed using a spectrum analyzer.
Two types of emissions tests are carried out: conducted and radiated.
Conducted emissions are a measure of the unwanted signals generated by
the DUT from its output connector or any cabling the device normally uses.
Special signal coupling techniques are required for some measurements.
Radiated emissions are those emanating from the device and picked up by
an external antenna. Official testing often involves the use of an anechoic
chamber to remove background disturbances.
Separate standards are specified according to the region in which the
equipment is to be used. The United States follows the FCC standards,
where CFR47 part 15 Appendix D applies. Europe follows the ETSI. Task
Group 31a is working on document EN 302 065. It is at a draft stage. In
Japan, TELEC define operating limits. The ITU-R is also working on common
standards for UWB measurements.
Below 1 GHz, tests requiring compliance with the International Special
Committee on Radio Interference (CISPR) publication 16 may require
electromagnetic compatibility (EMC) spectrum analyzers with quasi-peak
detectors. These tests are not covered in this application note. Please
contact your local Agilent sales representative for more information on
Agilent EMC products.
8. Receiver Measurements
A receiver design is challenging since the designer has to allow for many
different input signal conditions, some of which are hard to predict. This
is especially true when operation includes unlicensed bands and multiple
chipset vendors. UWB receiver testing is particularly difficult because
there is little general-purpose equipment that has the modulation bandwidth or
multi-frequency switching required (for multiband OFDM), therefore
so-called golden radios will be used. This approach has a number of
drawbacks, but traditionally has been the only practicable solution for
reference design integration and manufacturing.
With the introduction of Agilent’s N7619A Signal Studio for multiband
OFDM UWB software, golden radios are no longer needed for receiver
testing. Signal Studio for multiband OFDM UWB generates accurate UWB
waveforms compliant with the MBOA proposal for 802.15.3a.
This application note describes those tests that can be run with test
equipment and a golden radio, and provides some suggestions on techniques
to minimize the drawbacks of using a golden radio. It also demonstrates
the setup for using the N7619A Signal Studio for multiband OFDM UWB
software in place of the golden radio.
Designers and users will want to know how the UWB DUT copes with
non-UWB transmission. For information on generating interference signals
refer to Generating the interference signal on page 22. Or, use the N7619A
Signal Studio for multiband OFDM UWB software to create the interference
signal. For more information, refer to the N7619A Technical Overview,
literature number 5989-2927EN.
Test conditions and setup
The basic receiver test configuration for the MB-OFDM proposal is described
below. DS-UWB and pulsed system receiver test can also be broken down into
the phases of basic timing and full system test. As described in Pulsed and
pulse modulated on page 23, the 81134 pulse generator can be used as the
modulation source. This allows deterministic impairments like jitter to be
added. Precise spectrum shaping may be relatively unimportant for this kind
of test.
Testing methods are typically not well defined in the IEEE 802 radio
standards. The use of asynchronous packet based transmission timing has
meant receiver test is generally done using a one-way signal path. Loopback
mechanisms are not defined. When it is available, loopback testing allows
external test equipment to demodulate the returned signal and do its own
BER measurement.
A one-way signal path has the potential for faster testing, because data does
not have to be returned, but places a greater burden on the device supplier
and system integrator. Care is required in the triggering and sequencing of
the measurement. For example, changes in level of the signal source need
time to settle before further measurements are begun.
Waveform generation
(using the N7619A Signal Studio
for multiband OFDM UWB)
IQ path calibration
Dual Arb waveform generator
Trigger signals
[> 250 MHz bandwidth]
RF up/
signal generator
Option 015
spectrum analyzer
data recovery
Calibration path
Figure 59. Arbitrary waveform-based receiver test configuration for the MB-OFDM format.
In Figure 59, the configuration for single-frequency OFDM testing is shown.
The test signal is created as an arbitrary waveform IQ file. The IQ files are
then downloaded into the hardware waveform generator, which is connected
to the I and Q inputs of the PSG signal generator. The I and Q signal bandwidth required is that of the modulated RF, or approximately 256 MHz for
Figure 60 shows the typical performance of the PSG wideband modulator
option. Depending on the EVM performance requirements, it may be necessary to calibrate the IQ path. The PSA spectrum analyzer can be used to do
this in conjunction with special calibration software. Contact your Agilent
representative for more details.
A fully operational receiver has to go through three steps to recover the
1. Symbol synchronization
2. Channel estimation
3. Packet recognition and data recovery
In design it will be important to confirm the performance for each stage,
especially using impaired signals. Testing with isolated sections of the
packet, starting with the synch symbols, will allow this. The timing of
individual symbols can be deliberately altered to test the recovery process.
For convenience the structure of the MB-OFDM packet, is reproduced here.
1.875 µs
Channel estimation
Figure 60. Sequence of symbols in the Mode 1 MB-OFDM packet.
Tail & pad
If the measurement is of BER or PER, the modulated RF signal is filtered
and down-converted in the DUT. The application software provided with the
DUT will determine what information is available for analysis. To operate
with the test setup of Figure 59, the radio will need to operate in a test
mode that only uses one frequency.
Figure 61. Specifications for PSG Option 015 wideband modulation.
Frequency hopping
Frequency hopping tests require careful synchronization of the test source,
and either a significant increase in the bandwidth of the ARB and modulation path, or a switched RF oscillator. These options may not be open to
many designers.
Therefore, if possible, it is recommended the source is left static, while the
DUT switches between the frequencies appropriate to its operating mode.
This allows any issues with LO switching in the receiver to be isolated. It
requires that the DUT is able to recover individual OFDM symbols and
depends on the appropriate DUT software being available.
Receiver EVM measurements and BER
A bit or packet error measurement shows the composite result of analog and
receiver demodulation. The correlation between modulation errors and bit
errors becomes more complex when using multiple carriers (OFDM). The
MB-OFDM proposal discussed here also use forward error correction to
reduce the probability of bit errors caused by poor signal to noise ratio
(energy per bit/noise or Eb/No). At lower data rates, it duplicates symbols
across two frequencies. DSP algorithms, such as Viterbi, improve the raw
data bit recovery performance by using a short amount of data history to
predict what was most likely to have been sent.
The effect of this combination of data protection measures is to hide analog
impairments and reduce the link margin. Processing gain that could be used
to increase the range of the device is used to cope with hardware design
Analog measurements of the output of the receiver down-conversion chain
can provide a lot more information than BER and PER about any impairment suffered by the recovered signal. Figure 62 shows how a two-channel
oscilloscope can be used for IQ signal recovery prior to the ADC.
Waveform generation
(using the N7619A Signal Studio
for multiband OFDM UWB)
logic analyzer
IQ path calibration
Complex trigger signals
Dual Arb waveform generator
CCA indicator
Trigger signals
[> 250 MHz bandwidth]
signal generator
Option 015
RF up/
data recovery
Figure 62. Enhanced receiver test configuration for EVM measurements.
Bit errors are created when the signal vector is not at the right place on the
IQ plane, when the receiver reaches a decision point. The same techniques
described in the transmitter modulation measurement Delta (additive) EVM
(page 29) and OFDM (page 58) may be used to isolate the causes.
Receiver sensitivity and RSSI verification
With the emphasis being on packet transmission, the IEEE UWB proposals
do not directly refer to BER measurements. Unlike cellular (voice) systems
there are no unprotected bits sent as part of a normal UWB transmission.
Figure 63 shows the preliminary minimum sensitivity requirements for the
MB-OFDM proposal, and the variation with data rate.
Figure 63. An eight percent PER sensitivity level results from draft MB-OFDM proposal.
Of course, packet errors are caused by bit errors, and the longer a packet is,
the less likely it will be successfully recovered. Therefore a full system test is
needed. These tests will often need to be run using a golden radio, as shown
in Figure 64. Received signal strength indication tests can also be run using
the setup. An alternative configuration, which reduces some of the problems
associated with the RF section of the golden radio, is to feed the baseband
outputs of the golden radio to the IQ inputs of the PSG signal generator
EPM power meter
golden radio
ESA spectrum analyzer
54855 oscilloscope
Isolated digital
RF screened box
Solution for multiband
Alternate to signal
generation with golden radio
RF up /
Baseband I, Q
Option 015
Alternative mixed golden radio /
signal generator configuration
Figure 64. Golden radio receiver test configuration.
In addition, the baseband signal of the golden radio can be replaced with the
waveform creation software N7619A Signal Studio for multiband OFDM UWB
coupled with a wideband arbitrary waveform generator.
Repeatable measurements can only be obtained if care is taken to ensure the
golden radio performance is controlled and the RF signal level used for the
test is calibrated to an absolute standard.
The modulation quality and dynamic frequency accuracy of the golden radio
can be verified using the techniques described in OFDM (see page 58) and
Short pulsed signals (see page 61). The output power can be tested using the
techniques described in Power on page 41. A power meter and thermal
detector will allow small variations in absolute RF level to be detected,
which is what is needed for this specific test.
A network analyzer can be used to calibrate the attenuator and check for
impedance mismatches in the system.
Clear channel assessment test
Despite their operation as underlay technologies, the IEEE proposals may
still use clear channel assessment (CCA) to control the transmission periods.
For both DS-UWB and MB-OFDM proposals, the most basic requirement is to
choose a piconet operating code or frequency switching sequence. However,
these are selected at the time the piconet is established rather than being
applied actively during data transfer.
In WLAN, CCA involves a combination of energy detection and network
based information. Being a personal area network, the options to get
information from other devices are more limited. The UWB radio will need
to perform some form of spectrum monitoring.
The CCA test is designed to prevent the DUT from transmitting at the same
time as another UWB device of the same type, although this may be modified
to allow different piconets to use alternative channel switching frequencies.
The test configurations of Figures 62 and 64 can be combined to run a CCA
test. The golden radio needs to be programmable to simulate different
piconets. For MB-OFDM, the DUT must respond to a valid signal at the eight
percent PER sensitivity level, within < 5 µs. If the preamble is not identified,
the test signal level is 20 dB higher.
The DUT must detect the signal with > 90 percent probability. This implies
an extended test will be needed to reduce measurement variations.
9. Power Supply
One of the criteria for PHY layer selection is power consumption. The more
portable the device, the more stringent the operational and quiescent current requirements.
All equipment designs need to be tested at extremes of supply voltage, even
if a particular specification does not make it explicit. Operating limits will
very according to the conditions imposed by the host device, whether it is a
personal computer or a combination cellular phone.
There are other power supply measurements that can be very informative.
These include the current consumption as a function of the operational state
of the device. Receiver power management is part of the specification,
because the current consumption when listening is similar to that used
during transmission. Careful timing is required for periods when the receiver
is active. The longer oscillators and digital circuitry can be turned off, the
longer the battery life.
Monitoring power supply current relative to the timing of radio transmission
or reception can help ensure firmware and hardware work together as
expected. It is also quite straightforward to do before and after comparisons
following firmware updates to ensure no unwanted changes have occurred.
Battery emulation allows repeatable testing of the DUT under realistic
Agilent offers a complete line of DC power supplies that are suitable for
these tests. The 63000 Series includes general-purpose supplies as well as
supplies specifically designed to meet the demands of mobile communication products. These DC voltage supplies also offer low-current measuring
capability, which is useful for evaluating current consumption during
standby operation.
The 11465 software works in conjunction with the 63000 Series power supplies. It is designed to make it easier to characterize the radio in different
modes of operation. A plot of current versus time is shown in Figure 65.
Figure 65. Sample plot of the 14565A software.
86100C DCA,
Power Meter
ESG/PSG Series
Signal Generators
RF Layer tests
● Some measurement limitations1
PSA, ESA Series
Spectrum Analyzers
◆ Full measurement capability
89601A VSA
Software with
Appendix A:
Agilent Solutions for UWB
Transmitter tests
Average Power, PSD
Peak Power, CCDF
Spectrum Mask
Correlated Spectrum
BaseBand Pulse Shape
Modulated Pulse Shape
Transmission Spurious
Frequency Error
Constellation Error
BB. Eye, Timing Jitter
Transceiver Tests
Output of band
spurious emission
Receiver Tests
Receiver EVM
Max Input Level
(Non) Adjacent Channel
Component Test
Common Mode rejection
Amplitude & Frequency
Linear analysis
Non-linear analysis
1. See the notes within the appropriate section of this document for a description of the capabilites and limitations
of specific items of equipment
Test equipment with UWB capability
• Simulation software, ADS with E5619A UWB DesignGuide
Software tool for the design and simulation of custom UWB systems.
Pre-defined UWB component models to speed the simulation process.
Can be linked with an external ARB, PSG Series signal generators, and
89601 VSA software
• Simulation software, N7619A Signal Studio for multiband OFDM UWB
Software provides flexible, fast waveform creation for your design and
verification of OFDM UWB transceivers and components. Signal Studio
for multiband OFDM UWB operates with the E8267C/D PSG vector
signal generators coupled with an external wideband arbitrary waveform
• 54855A 6-GHz real time oscilloscope
A 20 GSa/s, four-channel input, 6 GHz real time oscilloscope. The
89601 VSA software can be run internally, along with a range of other
digital signal analysis software
Recommended options:
• E2681A jitter analysis software
• 001: 1 Msa memory
• Vector Signal Analysis software, 89601A
Versatile and precise signal analysis.
Recommended options:
• 105: Dynamic links to EESof/ADS
• AYA: OFDM analysis
• Signal Generator, PSG E8267C/D
Modulation source for multiband OFDM signals for transmitter and
component test. Use in conjunction with an external arbitrary waveform
generator or IQ outputs from radio baseband. Recommended options:
• 015: 2 GHz wideband I/Q input
• UNR: enhanced phase noise performance
• 520: 20 GHz PSG vector signal generator
• Signal Generator, ESG 4438C
Create a wide variety of interference signals, including UWB, WLAN and
Bluetooth. Use in conjunction with 89601A VSA software to replay any
recorded test signal.
Recommended option:
• 403: wideband noise source
• Spectrum Analyzer, PSA (6.7 to 50 GHz) and ESA-A Series
Semi-automated, one-button test execution for swept spectrum transmitter
measurement. The frequency range for analysis can be extended
above 50 GHz using 11970 (un-preselected) and 11974 (preselected)
Recommended PSA options:
• E4440A 26.5 GHz
• H26: 50 GHz low noise preamplifier
• Pulse Generator, 81134A
Able to create a wide range of pulse signals, and known timing errors.
Combine Channel 1 and Channel 2 signals to create a three-level signal, or
use both channels to create an IQ signal
Recommended option:
• Dual channel
• EPM-P power meter and 8482A thermal sensor, E9327 peak power sensor
Make accurate average power measurements with the thermal sensor.
Measure and inspect framed signals with the peak power sensors.
Other Test Equipment
• TS 50 shielded RF enclosure
Allows repeatable RF measurements to be made, without interference
from external environment.
• DC sources, 66319, 66321 B/D with test software 11465
Fast programmable dynamic DC power sources with battery emulation.
• Logic Analyzers, 1680/1690 Series
Provides comprehensive system-level debugging for digital hardware
design and verification.
• Logic Analyzers, 16700 Series
Provides comprehensive system-level debugging for multiple
processor/bus designs. Use E5904B with emulation trace Macrocell port
for ARM processor triggering.
• Network Analyzers, PNA and ENA Series
Recommended Option 010 time domain analysis
Provides measurement of antenna VSWR, and performance of PA, LNA,
and RF switch.
• Function Generator, 33250A, 80 MHz function/arbitrary waveforms
MB-OFDM frequency switching signal.
• Oscilloscope Probe –113xA
Ultra high speed active probes. Differential and single-ended.
• E2696A general purpose 6-GHz probing solution
Single ended and differential probes with external power supply and DC
offset capability.
Appendix B:
Recommended Reading
Useful Web links
Agilent UWB application and product information:
Agilent application on differential device measurement:
Agilent information on PSG signal generators:
Agilent information on CCDF plots:
IEEE 802.15.3 Home page:
DesignGuides developments for other UWB formats:
Multi-band OFDM Alliance:
Demo software
• 89601A Software demo is software available on CD or downloadable (130 Mb)
• Download the N7619A Signal Studio for multiband OFDM UWB software
to your PC (3.59 Mb). The signal configuration and graphing capabilities
can be evaluated by navigating the user interface prior to purchase,
go to
Application notes
• Spectrum Analyzer Measurement and Noise, Application Note 1303,
literature number 5966-4008E
• Making Pre-Compliance Conducted & Radiated Emissions Measurements
with EMC Analyzers, Application Note 1328, literature number
• RF Testing Of Wireless LAN Products, Application Note 1380-1,
literature number 5988-3762EN
• Equalizer Techniques & OFDM Troubleshooting for Wireless LANs,
Application Note1455, literature number 5988-9440EN
• Improving TDR/TDT Measurements using Normalization, Application
Note 1304-5, literature number 5988-2490EN
• High Precision Time Domain Reflectometry, Application Note 1304-7,
literature number 5988-9826EN
• Spectrum Analysis, Application Note 150, literature number 5952-0292
• Easy Frequency Extension to 110 GHz Using Agilent 83550 Series
Millimeter Wave Source Modules, literature number 5988-1098
• Measuring Jitter in Digital Systems, Application Note 1448-1, literature
number 5988-9109EN
• Finding Sources of Jitter with Real-Time Jitter Analysis, Application
Note 1448-2, literature number 5988-9740EN
Product notes
• Agilent PSA series Swept and FFT analysis, literature number 5980-3081
• Agilent Infiniium Oscilloscopes Performance Guide Using 89601A VSA
software, literature number 5988-4096EN
• Agilent E2696A General Purpose 6 GHz Probing Solution, literature
number 5988-9889EN
• Agilent 89600 Series Wide Bandwidth Vector Signal Analyzers, literature
number 5980-0723E
• Using Vector Modulation Analysis in the Integration, Troubleshooting
and Design of Digital RF Communications Systems, product note
89400-8, literature number 5091-8687E
• Jitter Analysis Using and Agilent Infiniium Oscilloscope, literature
number 5988-6109EN
Appendix B:
Recommended Reading
– continued
RF background reading
Appendix C:
Acknowledgement – the short frame sent by a receiver when it is able to
correctly decode a packet
• Effects of Physical Layer Impairments on OFDM Systems RF Design,
May, 2002, p. 36,, Cutler, Robert
• Antenna measurements: Triggering the PNA series Network Analyzer, for
use with PSG as remote RF source, whitepaper, literature number
• 8 Hints for Making Better Spectrum Analyzer Measurements, Application
Note 1286-1, literature number 5965-7009E
• Cookbook for EMC Pre-compliance Measurements, Application
Note 1290-1, literature number 5964-2151E
• Testing and Troubleshooting Digital RF Communications Receiver
Designs, Application Note 1314, literature number 5968-3579E
• Testing and Troubleshooting Digital RF Communications Transmitter
Designs, Application Note 1313, literature number 5968-3578E
Bluetooth – a frequency hopping WPAN radio system, operating in the 2.4
GHz unlicensed band
CDMAone – spread spectrum cellular technology, using code domain
modulation, based on TIA/EIA IS-95
Convolution – a technique to determine the system output when the system
impulse response and input signal are known
Correlation – a technique to measure the similarity between two signals
Direct sequence code division multiple access – UWB transmission scheme
that amplitude or IQ modulates an RF carrier with a very high speed, digital
signal that has been fed through a spectrum-shaping filter. Payload symbols
are created from selected 24-bit data sequences
Medium access control – the function of the software that adapts wired
data transmissions, so they are suitable for sending over a RF link
Mixed mode – term used for a vector network measurement technique
giving singled ended and differential transmission and reflection results,
without needing a differential RF source
Multi-band OFDM – an UWB transmission scheme that transmits data packets
as a sequence of individual OFDM symbols on adjacent RF frequencies
Multi-path propagation – the dominant effect controlling RF transmission
inside buildings. A single transmit signal arrives at the receiver having been
reflected from many surfaces. The difference in delay between the many
reflections can lead to inter-symbol-interference
OFDM – a modulation scheme that is insensitive to multi-path radio wave
propagation. A high-speed data signal is divided amongst many sub-carriers,
thereby increasing the transmit duration for each data symbol
Pulse de-sensitization – a potentially mis-leading term, historically used to
describe why the amplitude shown on a spectrum analyzer screen is lower
than the average power of the transmitted signal. Caused by the resolution
bandwidth of the spectrum analyzer being much less than signal bandwidth
Appendix C:
Glossary – continued
Ultra wide band – informally, an RF signal with 20 percent ratio of occupied
bandwidth to center frequency, or an instantaneous bandwidth of at
least 500 MHz. Formally, the definition comes from spectrum regulations
documents, which may vary by region
Uncoordinated piconet – a pair of wireless devices involved in data
transmission, operating without the benefit of any centralized network
management functions, such as to determine transmission timeslots or
power level
Underlay technology – a system that can be added to an existing environment, and as a design feature must cause minimum impact. The definition
of minimum is determined by regulatory bodies
Wavelet – an RF signal that is of short duration relative to the transmission
propagation delay
Appendix D:
Symbols and acronyms
Analog-to-digital converter
Advanced design system
Access point
Arbitrary (waveform generator)
Bit error rate
Bits per second
Binary orthogonal keying
Binary phase shift keying
Clear channel assessment
Complementary cumulative distribution function
Code domain multiple access
International Special Committee on Radio Interference
Complementary metal-oxide semiconductor
Common mode rejection ratio
Cyclic redundancy check
Carrier sense multiple access with collision avoidance
Carrier wave
Digital-to-analog converter
Differential quadrature phase shift keying
Direct sequence code domain multiple access
Digital signal processor
Device under test
Digital video broadcast
Equivalent isotropic radiated power
Electro magnetic compatibility
Equivalent noise bandwidth
Electronic signal generator
European Technical Standards Institute
Error vector magnitude
Federal Communications Commissions
Fast Fourier transform
Ground penetrating RADAR
Global positioning system
Integrated circuit
Institute of Electrical and Electronics Engineers, Inc.
Intermediate frequency
Inter-frame spacing
Inter symbol interference
Industrial, Scientific and Medical
International Telecommunications Union - Radio
Joint Electron Device Engineering Council
Low noise amplifier
Appendix D:
Symbols and acronyms
– continued
PN (9,15)
Local oscillator
Medium access control
Multi-band orthogonal frequency division multiplexing
Non line of site (propagation path)
Orthogonal frequency division multiplexing
Packet error rate
Physical (layer)
Phase locked loop
Pseudo random number (2N-1); seed often 0
Pseudo random binary sequence
Pulse repetition frequency
Power spectral density
Quadrature amplitude modulation
Quaternary phase shift keying
Radio detection and ranging
Resolution bandwidth
Radio frequency
Radio frequency identification
Root mean square
Root raised cosine
Receive signal strength indication
Signal to noise
Standard commands for programmable instruments
Spectrum emission mask
Sub-miniature (RF connector) version A
Signal-to-noise ratio
Simultaneous operating piconet
Short range device
Time division duplex
Time division multiple access
Transmit power control
Unlicensed National Information Infrastructure
Universal serial bus
Ultra wide band
Video bandwidth
Voltage control interface
Vector spectrum analyzer
Voltage standing wave ratio
Wireless local area network
Wireless personal area network
Wireless MEDIA
Appendix E:
1. Supplement to IEEE Standard for Information Technology IEEE Std
802.15.3a–1999 (supplement to IEEE Std 802.15.3-1999)
2. IEEE P802.15-03/268 Multi-Band OFDM Physical Layer Proposal for
IEEE 802.15 Task Group 3a
3. IEEE P802.15-03/154 XtremeSpectrum CFP document
4. IEEE P802.15-02/490r1 document
5. Reference FCC 02-48 Section 15.521(d)
6. FCC CFR47 part 15, sub-part F, August 2003
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