Keysight Technologies 10 Hints for Making Successful

Keysight Technologies 10 Hints for Making Successful
Keysight Technologies
10 Hints for Making Successful
Noise Figure Measurements
Application Note
Optimize Your
Measurements and
Minimize the Uncertainties
Noise figure is often the key to characterizing a receiver and its ability to
detect weak incoming signals in the presence of self-generated noise. The
process of reducing noise figure begins with a solid understanding of the
uncertainties in your components, subsystems and test setups. Quantifying
those unknowns depends on flexible tools that provide accurate, reliable
Keysight Technologies, Inc. noise figure solution set—instruments, applications
and accessories—helps you optimize test setups and identify unwanted
sources of noise. We’ve been providing noise-figure test solutions for more
than 50 years, beginning with basic noise meters and evolving into solutions
based on spectrum, network, and noise-figure analyzers.
The 10 hints presented here will help minimize the uncertainties in your
noise figure measurements—whether you are designing for good, better or
best performance in your device. Key topics include minimizing extraneous
signals, mismatch uncertainties, nonlinearities, and path losses.
The checklist in
Appendix A is a
helpful tool to
verify that all hints
have been considered
for a particular
Table of Contents
Introduction................................................................................................. 2
HINT 1: Select the Appropriate Noise Source..................................... 4
HINT 2: Minimize Extraneous Signals.................................................. 5
HINT 3: Minimize Mismatch Uncertainties.......................................... 6
HINT 4: Use Averaging to Minimize Display Jitter............................. 7
HINT 5: Avoid Non-Linearities............................................................... 8
HINT 6: Account for Mixer Characteristics.......................................... 9
HINT 7: Use Proper Measurement Correction.................................. 12
HINT 8: Choose the Optimal Measurement Bandwidth.................. 13
HINT 9: Account for Path Losses........................................................ 14
HINT 10: Account for the Temperature of the Measurement
Components.............................................................................................. 15
Appendix A: Checklist............................................................................ 16
Appendix B: Total Uncertainty Calculations...................................... 17
Appendix C: Abbreviations.................................................................... 18
Appendix D: Glossary and Definitions................................................ 18
Additional Resources.............................................................................. 19
Find out more
HINT 1: Select the Appropriate Noise Source
Frequency range
The output of a noise source is
defined in terms of its frequency
range and excess noise ratio (ENR).
Nominal ENR values of 15 dB and
6 dB are commonly available. ENR
values are calibrated at specific spot
Commercial noise sources cover
frequencies up to 50 GHz with
choices of co-axial or waveguide
connectors. The frequency range of
the noise source must include the
input frequency range of the DUT,
of course. If the DUT is a mixer or
frequency translation device, the
output frequency range of the DUT
must also be addressed. If one source
does not include both frequency
ranges, a second source will be
required. A second noise source may
also be necessary when measuring a
non-frequency translating device with
low noise and high gain. Low ENR is
best for the measurement, however,
high ENR is necessary to calibrate the
full dynamic range of the instrument.
In either case, a full-featured noise
figure analyzer can account for the
different ENR tables required for
calibration and measurement.
If possible, use a noise source
with the lowest change in output
impedance between its ON and OFF
states. The noise source’s output
impedance changes between its ON
and OFF states, which varies the
match between the noise source
and the DUT. This variation changes
the gain and noise figure of the DUT,
especially for active devices like
GaAs FET amplifiers. To minimize
this effect, 6 dB ENR noise sources
are commercially available that limit
their changes in reflection coefficient
between ON and OFF states to better
than 0.01 at frequencies to 18 GHz.
Use a 15 dB ENR noise source for:
–– General-purpose applications to
measure noise figure up to 30 dB.
–– User-calibrating the fullest dynamic
range of an instrument (before
measuring high-gain devices)
Use a 6 dB ENR noise source when:
–– Measuring a device with gain that
is especially sensitive to changes
in the source impedance
–– The device under test (DUT) has a
very low noise figure
–– The device’s noise figure does not
exceed 15 dB
A low ENR noise source will minimize
error due to noise detector nonlinearity. This error will be smaller
if the measurement is made over a
smaller, and therefore more linear,
range of the instrument’s detector.
A 6 dB noise source uses a smaller
detector range than a 15 dB source.
Use a noise source with the correct connector for the DUT rather
than use an adapter, particularly for
devices with gain. The ENR values
for a noise source apply only at its
connector. An adapter adds losses to
these ENR values. The uncertainty
of these losses increases the overall
uncertainty of the measurement. If an
adapter must be used, account for the
adapter losses.
A low ENR noise source will require
the instrument to use the least internal attenuation to cover the dynamic
range of the measurement, unless
the gain of the DUT is very high.
Using less attenuators will lower
the noise figure of the measurement
instrument, which will lower the
uncertainty of the measurement.
Figure 1-1. Automatic download of ENR data to the instrument speeds up overall set-up time
HINT 2: Minimize Extraneous Signals
A noise figure analyzer measures the
noise power from the noise source
as affected by the DUT. It uses the
power ratio at two detected noise
levels to measure the noise figure of
whatever is between the noise source
and the instrument’s detector. Any
interference, airborne or otherwise,
is measured as noise power from the
DUT and can cause an error of any
Figure 2-1 demonstrates the types
of stray signals that can get coupled
into the signal path and affect the
measurement. Fluorescent lights,
adjacent instruments, computers,
local TV and Radio stations, pocket
pagers, mobile phones and base stations are notorious for their adverse
effects on noise measurements.
Random stray signals can cause
several tenths of a dB difference
between individual readings, and
result in unstable measurements
(i.e. jitter that will not average to a
stable mean).
Lights (esp. fluorescent)
RF comms basestation
– Double-shielded cables for IF (ordinary braid is too porous)
– Shielded GPIB cables
– Enclose all circuits
– Test connectors by shaking leads
Figure 2-1
Follow these guidelines:
a. Ensure that mating connectors are clean and not worn or damaged
(see reference 6 in the Additional Resources section for more information).
If the measurement becomes unstable when the cables and connectors are
shaken lightly by hand, try other cables or connectors.
b. Use threaded connectors in the signal path whenever possible,
as BNC connectors are very susceptible to stray signals.
c. Use double shielded cables,
as common flexible braided coaxial cables are too porous to RF.
d. Use shielded GPIB cables.
e. Move the measurement setup to a screened room.
If a transmitter that has any frequency content within the measurement
bandwidth is nearby and any covers are off of the DUT, move the measurement
setup to a screened room. Test for such signals with a spectrum analyzer with a
simple wire antenna on the input. Attenuate these stray signals by 70 to 80 dB.
f. Use shielding.
This is especially important for making measurements on an open PC
breadboard. (see reference 5 in the Additional Resources section for more
g. Use an analyzer with minimal electromagnetic emissions.
Devices may be susceptible to stray emissions from some measurement
HINT 3: Minimize Mismatch Uncertainties
Mismatch at connection planes will
create multiple reflections of the
noise signal in the measurement
and calibration paths (as shown in
Figure 3-1).
Isolators also add to path losses.
As a result compensation is required.
Keysight X-Series signal analyzers
and noise figure analyzers have a loss
compensation feature to account for
the insertion losses of any isolators.
One method to reduce the mismatch
uncertainty is to place an isolator
in the RF path between the noise
source and the DUT. This isolator
can prevent multiple re-reflections
from reaching the DUT and can suppress the build-up of error vectors.
Isolators, however, operate over
restricted frequency ranges. Several
may be needed for the frequency
range of interest.
Alternately, insert a well-matched
attenuator (pad) between the noise
source and the DUT to attenuate
multiple reflections. As an example,
with a 10 dB attenuator, the rereflections are attenuated by 20 dB.
The advantage of an attenuator vs.
an isolator is broadband response.
The disadvantage is that the noise
source’s ENR values will be reduced
by the attenuator’s insertion loss
(10 dB in this example).
ρ = reflection coefficient at a reference plane
Figure 3-1
HINT 4: Use Averaging to Minimize Display Jitter
Mismatch at connection planes
with noise measurement inherently
displays variability or jitter because of
the random nature of the noise being
measured. Averaging many readings
can minimize displayed jitter and
bring the measurement closer to the
true mean of the noise’s Gaussian
Decreasing the bandwidth of the
measurement increases proportionately the number of readings
necessary to obtain the same level of
jitter reduction. For example, half the
bandwidth requires twice as many
readings of noise figure averaged
together to obtain the same jitter
Selecting the number of readings
that will be averaged for each
measurement will reduce jitter in the
measurement by the square root of N,
where N is the number of measurements in the average. The table below
shows some examples of the effect of
averaging on jitter. For example, jitter
may be reduced by almost 70% by
averaging approximately 10 readings.
If time constraints limit the number
of averages during DUT measurement
(e.g. in manufacturing), use more
averages during calibration. This
will make the correction for all
subsequent DUT measurements
more accurate.
% jitter
Modern noise figure analyzers have
the ability to do point averaging or
trace averaging. Point averaging
makes all measurements for the
first frequency point, calculates and
displays their average, then moves to
the next frequency point. The process
is repeated until each frequency
has been measured and averaged
the number of times specified in the
measurement setup. Trace averaging measures only once at each
frequency point through one entire
sweep of the frequency range. It then
begins the second sweep, averaging
each individual measurement at each
frequency with the previous average
for that frequency, as it sweeps. It
repeats this process until the number
of sweeps or “traces” averaged
together equals the number of
averages specified in the
measurement setup.
Both types of averaging give the
same answer. Point averaging is
faster overall since the analyzer’s
tuner has to retune fewer times for
all measurements to complete. Trace
averaging displays a rough measurement over the entire frequency range
faster. This enables a user to see any
obvious problems with the measurement (e.g. an extraneous signal)
Use trace averaging first. Watch a
few sweeps across the display and
look for indications of RF interference
such as a spike in the response at a
single frequency, or even a small step
in the response.
Hint 5: Avoid Nonlinearities
Avoid all predictable sources of
–– Circuits with phase lock loops
(and any circuit that relies on signal
presence to set its operating
–– Circuits that oscillate (even if at
a far-removed frequency)
–– Amplifiers or mixers that are
operating near saturation
–– AGC circuits or limiters
(AGC circuits have been known to
contribute additional noise power
at the power levels near its
operational point, even when
A Y-factor noise figure analyzer
assumes a linear change in the
detected noise power as the noise
source is switched between Thot and
Tcold. Any variations from linearity
in either the DUT or in the detector
directly produce an error in the Y
value and hence in the noise figure
that is displayed. The instrumentation
uncertainty specification accounts for
the linearity of the analyzer’s detector.
The linearity of the DUT, however,
should be carefully considered when
making the measurement.
The measured noise figure is
determined by the power in the
instrument’s resolution bandwidth.
The instrument’s attenuation settings,
however, may be determined by the
power in the instrument’s overall frequency range that reaches the range
detector. The analyzer is therefore
susceptible to being overdriven by
noise outside the bandwidth of any
one individual measurement, and
therefore vulnerable to non-linearity
errors. In such cases, attenuate any
broadband power outside the analyzer’s resolution bandwidth. Use a
filter wider than the measurement
frequency range and before the DUT.
Consider measuring a familiar “
reference” or “gold standard”
device at the beginning of each day
to assure that the same result is
obtained as prior days for the same
device, to add assurance that the
measurement instrument is warmed
up sufficiently.
–– High-gain DUTs without in-line
attenuation (Attenuate the output
of the DUT if necessary. See Hint 7
for details)
–– Power supply drifts
–– DUTs or measurement systems
that have not warmed up
–– Logarithmic amplifiers
(The standard Y-factor
measurement is invalid for
amplifiers in a logarithmic mode.)
HINT 6: Account for Mixer Characteristics
If the device under test is a mixer:
–– Measure the same sideband(s) that
will be used in the application of
the mixer.
–– For single sideband measurements,
select a LO far from the RF band of
interest, if possible.
Noise power
–– For double sideband measurements,
select a LO frequency close to the
RF band of interest.
–– Choose the LO to suit the mixer.
–– Filter the RF signal (i.e. the noise
source) if necessary to remove
unwanted signals that would mix
with the LO’s harmonics or
spurious signals.
–– Always document a frequency
plan to identify which of the above
precautions are necessary.
Figure 6-1
a. Select a double sideband or a
single sideband measurement.
A mixer will translate input signals
and noise from the upper sideband
(USB) and lower sideband (LSB) as
Figure 6-1 shows. (Note that the
LSB and USB are separated by twice
the IF.) A double sideband (DSB)
measurement, shown in Figure 6-2,
measures the noise powers for both
the USB and LSB. Some receiver systems, like those in radio astronomy,
intentionally use both sidebands.
DSB noise figure measurement is
appropriate in these cases. In many
applications the desired signal will
be seen in only one sideband. A
single sideband (SSB) measurement
is appropriate in these cases. In
SSB measurement setups, the noise
power in the unwanted sideband is
suppressed by appropriate “image
rejection” filtering at the input of the
DSB measurements are easier to
perform since they don’t require the
additional burden of image rejection
filter design and matching. When
a mixer with a DSB specified noise
figure is going to be used in an
SSB application, careful correction
is needed. (see reference 5 in the
Additional Resources section for more
b. For double sideband measurements,
select a LO frequency as close as
possible to the RF band of interest.
The choice of LO frequency, and the
resulting IF, can make a dramatic
difference in the results of DSB measurements. Instruments typically
display the average of the LSB and
USB noise figures. They measure the
power in the IF band (which is the
sum of both sidebands after conversion), diminish that power by half (3
dB), and display the result. The closer
the USB and LSB are together, the
more likely they are to be equal (as
shown in Figure 6-1) and the more
likely the default 3 dB correction
will be accurate. Since noise power
versus frequency for a mixer is rarely
flat, if too wide an IF is used the error
and its correction will be unknown.
To minimize this error, choose the LO
frequency as close as possible to the
RF band of interest, within the limitation that the resulting IF cannot be
below the lower frequency limit of the
instrument being used (often 10 MHz).
Measurement system
Noise source
f RF
f LO
IF amplifier
f IF
Figure 6-2
To determine if sideband-averaging
error is a problem, set up the noise
figure measurement for a mixer with
swept LO and fixed IF (modern noise
figure analyzers allow this). Monitor
the noise figure reading. If the noise
figure values change dramatically as
the LO sweeps, then SSB measurement is recommended (see reference
5 in the Additional Resources section
for more information)
c. For single sideband
measurements, select a LO far from
the RF band of interest, if possible.
The farther the LO is from the RF,
the higher will be the IF, and the less
stringent will be the rolloff requirement for the filter that suppresses
the unwanted sideband as shown in
Figure 6-3.
d. Choose the LO to suit the mixer.
Choose the LO to avoid spurious
output where the mixer is sensitive
and to avoid high broadband noise
floor. Filter the LO if necessary
to diminish spurious signals and
broadband noise, since the IF pass
band will include noise power one IF
away from the LO frequency and one
IF away from any spurious signals or
harmonics of the LO.
Balanced and double-balanced mixers
have more than one diode to perform
the frequency conversion. This
improves the LO to IF isolation
(20 dB and better), but usually
requires more LO power or biasing.
This higher power may raise the LO’s
noise output, produce spurs away
from the set frequency, or produce
LO harmonics inside the mixer. Any
of these can mix additional, yet
unwanted, RF signal into the IF, or
leak through to the IF. This may raise
the measured noise and possibly
overdrive the analyzer’s first stage
to a non-linear state. Experiment
with the LO to see what improves
performance and produces the lowest
noise figure.
e. Filter the RF signal (i.e. the
noise source) if necessary.
It may be necessary to filter the RF
noise source to remove frequencies
that are outside the band of interest
as shown in Figure 6-4. This can
remove unwanted signals that would
mix with the LO or its harmonics
or spurious signals and raise the
measured noise figure.
g. Document a frequency plan
to evaluate which of the above
precautions are necessary.
A frequency plan will help identify
which of the precautions above are
the most important before the
measurement is made.
Noise power
f. Filter the RF signal (i.e. the noise
source) if necessary.
Keep the LO outside of the frequency
range of the instrument if possible.
LO power will almost certainly leak
through to the IF port of the mixer.
Assume that LO to IF isolation will be
insufficient. If this leakage is within
the band of the measurement, it will
add to the noise figure measured.
If it is outside the band of the
measurement but within (or close
to the overall frequency range of the
instrument, it could cause the instrument to autorange itself to use more
attenuation, which would increase
the uncertainty of the measurement.
Filter the output of the mixer to
remove this LO leakage and any
LO harmonics created in the mixer
without appreciably attenuating the
wanted mixer output signal.
Figure 6-3
Measurement system
IF amplifier
Noise source
f IF
Figure 6-4
HINT 7: Use Proper Measurement Correction
Take the following steps to ensure
the measurement system itself does
not add error to the measurement.
–– Remove the noise figure of the
measurement system with regular
user calibration.
–– Avoid exceeding the maximum
input power of the measurement
instrument. Modern instruments
can handle around 65 dB of device
gain for narrow band devices. For
wider band devices with high gains
it is likely that an attenuator will be
required after the device to keep
the overall power within the instrument’s range. Use the analyzer’s
compensation feature to account
for the losses of the attenuator.
Use appropriate filters/isolators/
circulator to suppress out-of-band
responses that would otherwise
contribute noise power at the
high gain level and overpower the
instrument’s input.
F1 and F2 are linear noise figure
values for the DUT and the measurement system, respectively, and G1 is
the gain of the DUT. User calibration
(termed “second stage correction”)
determines F2; measurement determines F12 and G1. The analyzer
calculates F1 from the cascade
Perform user-calibration prior to the
measurement to remove the second
stage contribution. Calibrate out the
second stage contribution at regular
intervals depending on how sensitive
the noise figure and gain are to
temperature drifts.
The cascade equation shows how
F12 is very sensitive to uncertainty
margins in the second stage term
[(F2 - 1) / G1]. (To see how F12
would vary with marginal changes
in F2 or G1, see reference 5 in the
Additional Resources section)
A Y-factor noise figure analyzer
measures the noise figure of the
measurement system and the DUT
combined (see Figure 7-1). Below is
F12 in the cascade equation:
F12 = F1 + [(F2 - 1 )/G1]
If the DUT has insertion loss (e.g.
a mixer, attenuator, etc.), use a
low noise pre-amplifier before the
instrument to reduce the uncertainty
margin. Choose a pre-amp with the
lowest noise figure and a gain of
more than 100 (20 dB +). (see reference 5 in the Additional Resources
section for more information on
selecting the pre-amp.) Make the
noise figure of the second stage as
low as practical and the uncertainty
of F12 (and hence F1), as low as
possible. Adding a pre-amp also gives
the measurement some resilience
against noise figure variations versus
frequency of the second stage.
In the case of a high-gain DUT, there
may not be a need for a pre-amp. In
order to make that decision, place the
DUT’s linear gain and the measurement system’s linear noise figure
(i.e. noise factor) into the Cascade
Equation. Notice the noise figure
of the cascade will converge to the
noise figure of the DUT.
Calibration path
(Eq. 7-1)
Gain, G
F2 = Noise figure of the measurement system
F1 = Noise figure of the device under test (DUT)
Figure 7-1
HINT 8: Choose the Optimal Measurement Bandwidth
Select a measurement bandwidth
no larger than the pass band of the
DUT. Modern noise figure analyzers
provide a selection of various
measurement bandwidths to enable
measurements that are more relevant
to current practical applications (e.g.
individual wireless GSM channels).
4 MHz bandwidths were common
in past generations of noise figure
instruments; modern analyzers
can measure down to at least 1 Hz
Lowest possible actual
measurement bandwidth
due to downward
frequency drift
If the measurement bandwidth is
similar to the DUT’s operational
pass band, then any instabilities
between the center frequency of
the analyzer’s final IF and the DUT’s
operational bandwidth will lead to
an error in the displayed gain. The
analyzer may recognize the noise
power outside of the DUT’s pass band
during calibration and inside during
measurement. The extent of this
effect depends on the shape of the
DUT’s pass band overlapping with the
shape of the analyzer’s final IF pass
band. This issue is less significant if
the analyzer’s pass band is narrower
than the DUT’s pass band.
Analyzer's frequency instability
due to drift during time required
for all measurements to be
made and averaged
Highest possible actual
measurement bandwidth
due to upward frequency
Noise power detected
during calibration but not
during DUT measurement
under-states DUT's noise
bandwidth setting
Figure 8-1
Past generations of noise figure
instruments required additional time
to obtain the same level of jitter
reduction in a narrower bandwidth.
For the same jitter reduction, half
the bandwidth required twice the
time, one fourth required four times
as long, etc. In modern full-featured
analyzers, this time penalty is
mostly eliminated due to digital
signal processing (DSP) techniques
that measure a group of adjacent
frequencies simultaneously.
HINT 9: Account for Path Losses
Adapters must be used if the connectors between the noise source,
DUT, and measurement system do not
mate, as in Figure 9-1. It is most
important to avoid adapters where
the signal is smallest in the measurement setup. For devices with gain,
avoid adapters before the DUT. For
devices with loss, avoid adapters
after the DUT.
Keep track of the insertion losses of
adapters and any additional components such as cables, baluns, filters,
pads, and isolators that are not part
of the DUT. These insertion losses
have to be taken into account by subtracting them from the measurement
result. Any adapters used between
the noise source and the DUT will
introduce some loss, effectively
reducing the ENR of the noise source.
Modern noise figure solutions can
subtract these losses from the
measurement automatically. The
analyzer will also need to know the
temperature of these components
and whether they are before or after
the DUT. The analyzer will correct
the displayed noise figure for all loss
elements and their temperatures that
are entered.
Losses in the measurement system
are corrected by user calibration.
Adapters and cables that are connected to the instrument during both
user calibration and measurement are
considered part of the measurement
system and are accounted for during
user calibration.
If a pre-amp is needed prior to the
measurement system then choose a
pre-amp with the correct connector
for the noise source, and include the
pre-amp as part of the calibration
setup that measures the second
stage noise figure.
Connection diagram
Cable C1
Coax/WG adapter
Coax/WG adapter
Figure 9-1
Cable C 2
HINT 10: Account for the Temperature of
the Measurement Components
Figure 10-1 shows the measurement
response curve where the noise powers N1 and N2 are due to input noise
temperatures Tcold and Thot, respectively. The solid line on this graph
represents the response when T0 is
considered as the reference ambient
temperature of 290 K. If the Tcold of
the noise source (for a solid state
noise source, its surface temperature)
is not equal to T0, then the dotted line
may become the response curve. The
noise power added by the DUT would
then be Na´ and not Na.
If the actual local ambient temperature is not entered into the analyzer,
it will make its calculations with the
wrong temperature assumption. An
extra error term in the over-all uncertainty margin will be present. This can
be a significant value if the true noise
figure of the DUT is low. Figure 10-2
shows typical differences between
the displayed and actual noise figures
for different ambient temperatures.
N a´
To Tc
Figure 10-1
NF (dB) = F mea - F actual
Y-factor noise figure analyzers
assume that the surface temperatures
of all components in the measurement (noise source, DUT, connectors,
cables, etc.) are the default value for
Tcold, 290 K (16.8 °C, 62.2 °F). If this is
not the case, enter the correct temperature of each component into the
analyzer and monitor them regularly.
Full featured noise figure solutions
allow the entry of the temperatures
of any components added before and
after the device.
T amb = 305K (= 89 F)
T amb = 296.5K (= 74 F)
NF (dB)
Figure 10-2
Appendix A: Checklist
Use this checklist to assist with
locating the hint relating to specific
issues or considerations. Reading
the 10 Hints Application Note
sequentially is not necessary.
1 Select the appropriate noise source. See Hint # 1.
Use a low ENR source whenever possible.
Avoid adapters between the noise source and DUT.
Double check manually entered ENR values.
2 Reduce EMI influence. See Hint # 2.
Use clean, undamaged connectors.
Use threaded connectors.
Use double shielded cables.
Use shielded GPIB cables.
Use a screened room.
Use shielding.
Use an analyzer with minimal electromagnetic emissions.
3 Minimize mismatch uncertainties. See Hint # 3.
Use an attenuator or isolator
if needed.
Use a pre-amp if needed.
4 Use averaging to minimize display jitter. See Hint # 4.
Select enough averages to stabilize the measurement.
Use “trace averaging” first to spot measurement setup problems soonest.
Look for spikes or even small steps in the display that indicate RF interference.
If time constrained, use more
averaging during calibration than during DUT measurement.
5 Avoid nonlinearities. See Hint # 5.
Avoid the following:
Circuits with phase lock loops.
Circuits that oscillate.
Amplifiers or mixers operating near saturation.
AGC circuits or limiters.
High-gain DUTs without in-line attenuation.
Power supply drifts.
DUTs or instruments that have not warmed up.
Logarithmic amplifiers.
6 Account for mixer characteristics. See Hint # 6.
Measure the same sideband(s) that will be used in the application.
For double-sideband measurements, select a LO
frequency close to the RF band of interest.
For single-sideband measurements, select a LO far from the RF band of interest,
if possible.
Choose the LO to suit the mixer.
Filter the LO if necessary to diminish spurious signals and broadband noise.
Keep the LO outside of the measurement bandwidth
if possible.
Filter the IF if necessary to remove LO harmonics created
inside the mixer.
Filter the RF to prevent unwanted mixing.
Test for DSB error by changing IF.
Experiment with different LOs to get the most accurate
(i.e. lowest) noise figure.
Document a frequency plan to
evaluate which of the above
precautions are necessary.
7 Enter proper measurement correction. See Hint # 7.
Calibrate regularly.
Keep overall gain below the instrument’s spec.
Filter out-of-band power.
If 2nd stage effect is large,
add a low-noise pre-amp (with proper connectors).
8 Choose the optimal bandwidth. See Hint # 8.
Select a measurement bandwidth no larger than the pass band of the DUT.
9 Account for path losses.
See Hint # 9.
Avoid adapters as much as possible.
If used, enter their losses into the instrument.
10 Account for the temperature of
measurement components.
See Hint # 10.
Enter physical temperatures
of the noise source and
the components of the measurement setup into
the instrument.
Appendix B: Total Uncertainty Calculations
The error model in the spreadsheet
shown in Figure B-1 is obtained from
the derivative of the Cascade equation
(F1 = F12 - [(F2-1)/G1]). It takes into
account the individual mismatch
uncertainty calculations at each
reference or incident plane of the
DUT, noise source and measurement
system. This example represents
the total noise figure measurement
uncertainty, RSS analysis, for a
microwave transistor with an S11 of
0.5, S22 of 0.8 and S21 of 5 (14 dB).
Noise figure uncertainty, here is calculated as ±0.48 dB. The uncertainty
dramatically improves to ±0.26 dB, in
this instance, if the gain is improved
to 20 dB. The calculation for this error
model is derived from reference 5 in
the Additional Resources section.
Figure B-2 shows the typical display
for the entry point to this interactive
model. Figures B-3, B-4 and B-5 show
simulated results for the uncertainty
calculator. There are a number of graphs
that can be plotted. These figures
show examples of RSS uncertainty
with respect to instrument match,
Figure B-1
Figure B-2
Figure B-3
Figure B-4
Figure B-5
DUT input match and instrument
noise figure, respectively. This interactive uncertainty calculator and the
spreadsheet version shown in Figure
B-1 can be accessed via the Internet
by using the URL on the back page of
this application note.
Appendix C:
Appendix D: Glossary and Definitions
AGC Automatic gain control
DUT Device under test
DSB Double sideband
ENR Excess noise ratio
F Noise factor
(linear expression of NF)
LSB Lower sideband
NF Noise figure (noise factor [F]
expressed in dB)
RSS Root sum of squares
SSB Single sideband
Tc Cold temperature
Tcold Cold temperature
Th Hot temperature
Thot Hot temperature
USB Upper sideband
URL Universal reference locator
1. Excess noise ratio (ENR):
ENR is the measure of how much
more noise power is output from a
noise source when “ON” (i.e. operating at virtual temperature Thot) than
is output when “OFF” (i.e. operating
at ambient temperature Tcold),
normalized by its output power at the
standard temperature 290 K.
ENR = 10 log (Th - Tc)/T0
For example, an ENR of 15 dB means
that the noise source output when
ON is greater than when OFF by the
antilog of 15/10 times the noise power
output at 290 K. This is equivalent
to a resistor at 9171 K, calculated as
antilog (15/10) x 290.
2. Second stage contribution:
During the measurement process
the noise source is connected to
the input of the DUT and the DUT’s
output is connected to the measurement system. In this cascade,
the DUT is the first stage and the
measurement system is the second
stage. The measurement system
will measure the noise figure of the
cascade. Correction requires that
the noise figure (F1) value of the
DUT has to be de-embedded” from
the cascade’s combined noise figure
(F12) by removing the contribution of
the second stage in the cascade or
Friis equation:
F12 = F1 + [(F2 - 1)/G1]
The expression in the brackets is the
second stage contribution. G1 is the
gain of the DUT.
3. Y-factor:
This is the linear ratio of the noise
power seen by the measurement
system when the noise source is
turned ON, to that when the noise
source is turned OFF. This is the basis
of the calculation of noise figure.
After the instrument recognizes the
ENR, the noise figure is derived from
the equation:
NF = ENR - 10 log (Y - 1)
Additional Resources
For more information on noise figure and Keysight Technologies’ noise figure
solutions, see below.
Noise figure solutions:
2. Hints for Making Better Noise Figure Measurements – video series:
3. Noise Figure Selection Guide, literature number 5989-8056EN
4. Fundamentals of Noise Figure Measurements (AN 57-1),
Application Note, literature number 5952-8255E
5. Noise Figure Measurement Accuracy (AN 57-2), Application Note,
literature number 5952-3706E
6. Principles of Microwave Connector Care, Application Note,
literature number 5954-1566
20 | Keysight | 10 Hints for Making Successful Noise Figure Measurements - Application Note
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© Keysight Technologies, 2011 - 2014
Published in USA, August 1, 2014
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