Contents i

Contents i
Contents
i
ii Contents
D P Kothari is presently Vice Chancellor of VIT University, Vellore. He obtained
a BE (Electrical) in 1967, ME (Power Systems) in 1969 and PhD in 1975 from the
Birla Institute of Technology and Science (BITS) Pilani, Rajasthan. From 1969
to 1977, he was involved in teaching and development of several courses at BITS
Pilani. Prior to his assuming charge as Vice Chancellor of VIT University, Dr Kothari
served as Director In-charge and Deputy Director (Administration) as well as Head
Centre for Energy Studies at Indian Institute of Technology, Delhi; and as Principal,
Visvesvaraya Regional Engineering College, Nagpur. He was Visiting Professor at the
Royal Melbourne Institute of Technology, Melbourne, Australia, during 1982–83 and
1989 for two years. He was NSF Fellow at Purdue University, US in 1992.
Dr Kothari, who is a recipient of the Most Active Researcher Award, has published and presented 625
research papers in various national as well as international journals, conferences, guided 30 PhD scholars and
63 MTech students, and authored 21 books in Power Systems and other allied areas. He has delivered several
keynote addresses and invited lectures at both national and international conferences on Electric Energy
Systems. He has also delivered 42 video lectures on science and technology on YouTube with a maximum
of 35,000 hits!
Dr Kothari is a Fellow of the Indian National Academy of Engineering (FNAE), Fellow of Indian National
Academy of Sciences [FNASc], Fellow of Institution of Engineers (FIE) and Senior Member, IEEE.
His many awards include the National Khosla award for Lifetime Achievements in Engineering for 2005
from IIT Roorkee. The University Grants Commission (UGC), Govt. of India, has bestowed the UGC National
Swami Pranavananda Saraswati award for 2005 on Education for his outstanding scholarly contributions.
He is also a recipient of the Lifetime Achievement Award (2009) by the World Management Congress, New
Delhi, for his contribution to the areas of educational planning and administration. His fields of specialization
are Optimal Hydro-thermal Scheduling, Unit Commitment, Maintenance Scheduling, Energy Conservation
(loss minimization and voltage control), Power Quality and Energy Systems Planning and Modelling.
I J Nagrath is Adjunct Professor, BITS Pilani, from where he retired in July 1999 as
Professor of Electrical Engineering and Deputy Director. He is now actively engaged
in writing books related to his long teaching and research experience.
He obtained his BE with Honours in Electrical Engineering from Birla Engineering
College in 1951 and MS from the University of Wisconsin, USA in 1956.
He has co-authored several successful books which include Electric Machines, 3/e,
Modern Power System Analysis, Power System Engineering, Signals and Systems,
Electrical Machines, Sigma Series and has authored Basic Electrical Engineering (all
published by TMH). He has also co-authored Control System Engineering and authored Electronics: Analog
and Digital. Besides he has these, published several research papers in prestigious national and international
journals and continues to be active in studies and writing.
Contents iii
D P Kothari
Vice Chancellor
Vellore Institute of Technology (VIT)
Vellore, Tamil Nadu
I J Nagrath
Adjunct Professor
Birla Institute of Technology and Science (BITS)
Pilani, Rajasthan
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iv Contents
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Electric Machines, 4/e
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Contents
Dedicated
to
Shobha
— D P Kothari
and
Pushpa
— I J Nagrath
v
vi Contents
Contents
Preface
vii
xiii
1. Introduction
1
1.1 Introduction 1
1.2 Basic Principle, Types and Constructional Features of Electric Machines
1.3 Recent Trends in Research and Developments in Electric Machines 7
2. Magnetic Circuits and Induction
2.1
2.2
2.3
2.4
2.5
2.6
2.7
2.8
12
Introduction 12
Magnetic Circuits 12
Magnetic Materials and Their Properties 25
Magnetically Induced EMF and Force 27
AC Operation of Magnetic Circuits 31
Hysteresis and Eddy-Current Losses 33
Permanent Magnets 35
Application of Permanent Magnet Materials 40
Summary 42
Practice Problems 43
Review Questions 47
3. Transformers
3.1
3.2
3.3
3.4
3.5
3.6
3.7
3.8
3.9
3.10
3.11
3.12
3
Introduction 48
Transformer Construction and Practical Considerations
Transformer on No-Load 54
Ideal Transformer 58
Real Transformer and Equivalent Circuit 62
Transformer Losses 71
Transformer Testing 72
The Per Unit System 80
Efficiency and Voltage Regulation 82
Excitation Phenomenon in Transformers 91
Autotransformers 94
Variable Frequency Transformer 97
48
49
viii Contents
3.13
3.14
3.15
3.16
3.17
3.18
3.19
3.20
3.21
3.22
Three-Phase Transformers 101
Parallel Operation of Transformers 116
Three-Winding Transformers 120
Phase Conversion 124
Tap Changing Transformers 127
Voltage and Current Transformers 131
Audio-Frequency Transformer 135
Grounding Transformer 136
Welding Transformer 136
Transformer as a Magnetically Coupled Circuit
Summary 146
Practice Problems 148
Review Questions 156
Multiple-Choice Questions 157
137
4. Principles of Electromechanical Energy Conversion
4.1
4.2
4.3
4.4
4.5
4.6
4.7
Introduction 158
Energy in Magnetic System 158
Field Energy and Mechanical Force 162
Multiply-Excited Magnetic Field Systems 176
Forces/Torques in Systems with Permanent Magnets
Energy Conversion via Electric Field 187
Dynamical Equations of Electromechanical Systems
Summary 193
Practice Problems 194
Review Questions 196
184
190
5. Basic Concepts in Rotating Machines
5.1
5.2
5.3
5.4
5.5
5.6
5.7
5.8
5.9
5.10
5.11
5.12
Introduction 197
Elementary Machines 198
Generated EMF 205
MMF of Distributed ac Windings 216
Rotating Magnetic Field 223
Torque in Round Rotor Machine 230
Operation of Basic Machine Types 234
Linear Machines 245
Magnetic Leakage in Rotating Machines 247
Losses and Efficiency 250
Rating and Loss Dissipation 255
Matching Characteristics of Electric Machine and Load
158
197
261
Contents ix
5.13 Resume 263
Summary 263
Practice Problems 266
Review Questions 270
Multiple-Choice Questions
272
6. AC Armature Windings
273
6.1 Introduction 273
6.2 AC Windings 275
Summary 283
Practice Problems 283
Review Questions 284
7. DC Machines
7.1
7.2
7.3
7.4
7.5
7.6
7.7
7.8
7.9
7.10
7.11
7.12
7.13
7.14
7.15
7.16
7.17
7.18
7.19
7.20
7.21
7.22
7.23
Introduction 285
Armature Winding and Commutator 287
Certain Observations 301
EMF and Torque 301
Circuit Model 305
Armature Reaction 310
Compensating Winding 316
Commutation 318
Methods of Excitation 322
Operating Characteristics of dc Generator 326
Self-Excitation 332
Characteristics of dc Generators 335
Shunt Generator–Predetermination of External Characteristic
Parallel Operation of dc Generators 357
Characteristics of dc Motors 361
Starting of dc Motors 382
Speed Control of dc Motors 390
Braking of dc Motors 408
Efficiency and Testing 410
Testing of dc Machines 412
DC Machine Dynamics 423
Permanent Magnet dc (PMDC) Motors 426
DC Machine Applications 430
Summary 431
Practice Problems 433
Review Questions 441
Multiple-Choice Questions 442
285
339
x Contents
8. Synchronous Machines
8.1
8.2
8.3
8.4
8.5
8.6
8.7
8.8
8.9
8.10
8.11
8.12
8.13
8.14
8.15
8.16
8.17
8.18
8.19
8.20
8.21
8.22
444
Introduction 444
Basic Synchronous Machine Model 445
Circuit Model of Synchronous Machine 451
Determination of the Synchronous Reactance 454
MMF Method 462
Determination of Armature Reaction Ampere-Turns and
Leakage Reactance of a Synchronous Machine—Potier Method 465
ASA (American Standards Association) Method (Latest) 473
Nature of Armature Reaction 475
Synchronizing to Infinite Bus-Bars 476
Operating Characteristics 478
Efficiency of Synchronous Machines 494
Power Flow (Transfer) Equations 497
Capability Curve of Synchronous Generator 518
Salient-Pole Synchronous Machine Two-Reaction Model 521
Staying in Synchronizm – The Synchronizing Power (Torque) 536
Determination of XD And XQ —Slip Test 543
Parallel Operation of Synchronous Generators 545
Hunting in Synchronous Machines 549
Starting of Synchronous Motors 554
Short-Circuit Transient in Synchronous Machine 555
Single-Phase Synchronous Generators 563
Brushless DC Motors 575
Summary 582
Practice Problems 585
Review Questions 590
Multiple-Choice Questions 591
9. Induction Machine
9.1
9.2
9.3
9.4
9.5
9.6
9.7
9.8
9.9
9.10
9.11
Introduction 593
Construction 593
Flux and MMF Waves in Induction Motor—Principle of Operation
Development of Circuit Model (Equivalent Circuit) 601
Power Across Air-Gap, Torque and Power Output 605
Tests to Determine Circuit-Model Parameters 614
The Circle Diagram (Approximate) 630
Starting 638
Cogging and Crawling 645
Speed Control 647
Deep-Bar/Double-Cage Rotor 663
593
596
Contents xi
9.12
9.13
9.14
9.15
9.16
9.17
Classes of Squirrel-Cage Motors 666
Induction Generator 667
Induction Machine Dynamics: Acceleration Time
Inverted Induction Machine 685
High Efficiency Induction Motors 687
Linear Induction Motor (LIM) 688
Summary 691
Practice Problems 694
Review Questions 699
Multiple-Choice Questions 701
670
10. Fractional Kilowatt Motors
10.1
10.2
10.3
10.4
10.5
10.6
10.7
Introduction 702
Single-Phase Induction Motors 702
Single-Phase Synchronous Motors 722
Circuit Model of Single-Phase Induction Motor 725
Balanced 2-Phase Motor Fed from Unbalanced Supply
Stepper Motors 740
Series Motor—Universal Motor 746
Summary 751
Practice Problems 752
Review Questions 752
702
734
11. Generalised Theory of Electrical Machines
11.1
11.2
11.3
11.4
11.5
11.6
11.7
11.8
11.9
Introduction 753
Convention 753
Basic Two-Pole Machine 753
Transformer with a Movable Secondary Winding 755
Kron’s Primitive Machine 757
Linear Transformations in Machine 758
Three-Phase to Two-Phase (a, b, c To a, b. 0) Transformation 761
Rotating Axis (a, b. 0) to Stationary Axis (d, q, 0) Transformation 762
Physical Concepts of Park’s Transformation 765
Review Questions 766
12. Motor Control by Static Power Converters
12.1
12.2
12.3
12.4
12.5
753
Introduction 767
Solid State Devices 769
Electrical Drives 782
Power Converters 783
Thyristor Motor Control 785
767
xii Contents
12.6
12.7
12.8
12.9
12.10
12.11
12.12
DC Motor Control Through Converters 786
DC Motor Control Through Choppers 800
Converter Topologies for dc Motor Drives 811
AC Motor Control 813
Inverters 819
Forced Commutation 828
Vector Control of an Induction Motor 831
Summary 837
Practice Problems 837
Review Questions 839
Multiple-Choice Questions 839
Appendix I: AC Steady-State Circuit Analysis 841
Appendix II: Three-Phase Systems 851
Appendix III: Special Topics in Transformers 863
Appendix IV: Cross-Field Machines 866
Appendix V: AC Commutator Machines 869
Appendix VI: Resistance 875
Appendix VII: Sample Examples Solved Using Matlab 877
Appendix VIII: Table of Constants and Unit Conversion 891
References
Answers to Problems
892
897
Index
907
Contents
xiii
The aim of this book is to give deep exposition of the theory of electromechanical devices, with specific
emphasis on the theory of rotating electric machines. The basic concepts have remained more or less the same
over the years since the first edition of this text appeared in 1985.
Since the appearance of the third edition in 2004, most of the advances in the application and control
of electric machines have taken place owing to the further breakthroughs in power electronics and
microprocessor/computer-based control systems. As a result, a much broader spectrum of electric machine
types are now available. Particularly, permanent-magnet and variable-reluctance machines are now used in
many applications and this is bound to increase further in future. AC drives are becoming more and more
attractive in many applications, such as those requiring variable speed and flexible control, while earlier dc
machines were the only choice. Realising this fact, these machine types find increased coverage in the fourth
edition.
This book presents simple, explicit, and yet rigorous and comprehensive treatment of transformers and
electric machines in a single volume. Considerable emphasis is laid on the fundamentals, physical concepts,
principles and on rigorous development of circuit model equivalents of both transformers and machines.
Each circuit model is closely related to the physical reality, the underlying assumptions are sharply focussed
and consequent limitations on the range of operation over which the model is valid are fully explained.
The clarity of the physical basis of models developed would be most satisfying to the reader and it would
enable him to make intelligent use of the models in the solution of machine problems and in the design of
systems using these devices. The prediction of device performance follows as an immediate sequel to its
model. Furthermore, as a next step (not covered in this book), the circuit parameters could be conveniently
related to the physical dimensions and properties of the materials used in the device. While the circuit theory
approach to electro-mechanical devices is introduced early in Chapter 2, the machine analysis in the bulk of
the book follows the field-theory approach which, as is well-known, is better understood and appreciated by
undergraduates and provides a deep insight into and a clear understanding of the electric machine.
This is the only book which clearly brings home to the reader the distinction in the sign convention
between the synchronous machine model and the transformer-type model, also employed for the induction
machine. Another distinguishing feature of the book is the clarity with which it brings out the difference
between a sinusoidally spaced distributed quantity (field) represented as a vector and sinusoidally timevarying quantities represented as phasors and how a rotating vector creates a time phasor. In order that the
teacher and student can both make convenient use of symbols on the blackboard or on paper, the phasors are
symbolically represented by capital letters with superbars and the vectors are represented by capital letters
with superarrows.
The book covers all the essential ingredients of machine knowledge expected of a modern-day
undergraduate in electrical engineering. With new and vital topics crowding the curriculum in electrical
engineering, machine courses have rightly been squeezed into two time slots of one-semester duration each.
xiv Contents
Preface
The book is designed to meet this need. The book is primarily designed to cater to a one-semester core course
common for all engineering disciplines and a one-semester topping off course for those majoring in electrical
engineering. The core course may comprise Chapters 1, 2, Secs 3.1 to 3.9 (except Sumpner’s test), 3.11, 3.12
(partly), Chapters 4,5 and Secs 7.1 to 7.4 for dc machine coverage along with a quick resume of armature
reaction, commutation, methods of excitation and characteristics of generators and motors. These topics are
covered in initial portions of the relevant sections of Chapter 7. The dc machine winding can be explained to
the class by merely projecting the two developed winding diagrams of Chapter 6. The remaining portions of
the book would then comprise the second course. The book is written in a flexible style and a high degree of
selectivity is inbuilt so that the teacher may leave out advanced articles of various topics in coping with the
time factor without any loss of continuity. It is even possible to select a single one-semester course out of the
book where time exigencies so demand.
The theory and applications of various machines as control-system actuators is treated at appropriate
places in the book. The methods of control-system analysis have not been included as these form a full
course in a modem curriculum. Linear approximations are employed for tackling non-linearities associated
with most machines. Wherever warranted, the effect of magnetic nonlinearity is accounted for in steady-state
analysis.
Although the models advanced are strictly applicable for steady-state analysis of device performance,
these are extended to the dynamic case at a few places by making strong assumptions. The transient analysis
of the synchronous machine is treated qualitatively and a graphical picture of the phenomenon is presented.
The reader is expected to have a prior grounding in electricity and magnetism, introductory circuit theory,
basic mechanics and elementary differential equations. However, appendices on ac steady-state circuit
analysis and three-phase systems have been included for ready reference.
New to this Edition
The chapters on dc machines and synchronous machines are re-written completely. The highlights of this
edition are large number of solved problems and practice problems that have been added in all the chapters.
The key features of this edition are
∑ New chapter on ‘Generalized Theory of Electric Machines’
∑ Detailed description of Transformers, dc Machines, dc Machines Excitation, Predetermination of
external characteristics of dc Generator, Parallel operation of dc Generators, Efficiency and Testing of
dc Machines, Speed control of Induction Motor, Linear Induction Motor
∑ Enhanced coverage of Permanent Magnet dc Motors, Permanent Magnet Materials and their
applications
∑ Discussion on Silicon Controlled Rectifier (SCR), Insulated Gate Bipolar Transistor (IGBT), MOS
Turn off Thyristor (MTO) and Emitter Turn off Thyristor (ETO) to cover new trends
∑ Synchronous generator (alternator), MMF Method, ASA Method, V curves and inverted V curves,
Rating of alternator, phasor diagrams, Reactive power flow from generator
∑ MATLAB examples to facilitate problem-solving skills
∑ Excellent pedagogy including
Contents
Preface
xv
Though no sophisticated knowledge of mathematics is required for the reader of this book, the mathematics
involved in this subject at times can get messy and tedious. This is particularly true in the analysis of ac
machines in which there is a significant amount of algebra involving complex numbers. One of the significant
positive developments in the recent years is the widespread availability of software such as MATLAB which
greatly facilitates the solution of such problems. MATLAB is freely used in many institutions and colleges
and is also available in a student version (http://www.mathworks.com). This edition, therefore, incorporates
MATLAB in some sample solved examples. It should be emphasized here that the use of MATLAB is not
a prerequisite for using this book. It is just an enhancement, an important one though! Further, it may be
noted that even in the cases where it is not specifically suggested, some of the problems in the book can be
attempted using MATLAB or an equivalent program. Some additional programs for solving problems using
MATLAB are included in this book.
The introductory chapter discusses electrical–electrical and electromechanical energy conversion
processes and devices from a general point of view with the explicit purpose of motivating the reader for
studying transformers and electric machines. This chapter, however, is not a prerequisite for the rest of the
book. Chapter 2 brushes up magnetic circuits and the principle of induction.
In Chapter 3, the transformer is treated exhaustively. The circuit-model approach is emphasized and for
obvious reasons the role of the phasor diagram is underplayed. This chapter lays the ground work for the
understanding of electromechanical energy conversion processes in machines and the circuit model of the
induction machine in particular.
Then follows Chapter 4 on the underlying principles of electromechanical energy conversion in the end
of which is answered the question, “Why is electric field not used as a coupling medium instead of the
magnetic field?” Cases of both linear and nonlinear magnetization (saturation) are treated.
Exposition of the basic concepts of rotating machines from a generalized point of view as well as engineering
aspects, such as cooling, rating and load mechanics is advanced in Chapter 5. General expressions for
emf and torque are derived. The torque production is explained here via interaction of two magnetic fields
assumed to be sinusoidally distributed. An alternative current-sheet approach is also given for the interested
reader. Elementary treatment of specific machine types—synchronous and induction—then follows and their
important characteristics are visualized on a field-interaction basis. Since interacting fields are assumed to be
sinusoidal, which is justified in these two classes of machines only, a most rudimentary treatment of the de
machine is given here because the fields in this class of machines are essentially nonsinusoidally distributed.
While Chapter 5 gives the essential treatment of ac windings, the details including important practical
features are dealt with in Chapter 6 devoted entirely to ac windings. Also given is a reasonably detailed
account of dc armature windings in Chapter 7. Where time is a limiting factor, ac winding details can be
skipped and dc winding directly introduced via the two developed diagrams with a brief explanation of
parallel paths, commutation and brush location.
Chapters 7–9 cover in depth the three basic machine types—the dc machine, synchronous machine and
induction machine. The approach adopted in all the three is one of rigorous modelling with due stress on
explanation of the underlying assumptions. The dc machine is the first to be dealt with as its steady-state
model is the simplest. The modelling in each machine results in a circuit model of the linear kind by virtue of
the assumptions made, which for all practical purposes are quite valid for steady-state performance analysis
as well as under certain transient situations. In Chapter 8, on the synchronous machine, a heuristic methods
are advanced to account for the effect of strong magnetic nonlinearity on the machine performance.
xvi Contents
Preface
Tests to determine circuit-model parameters are advanced at appropriate places. Assumptions involved
in machine modelling are once again stressed at this stage. Once the circuit model of the machine has been
arrived at, the discussion is then focussed on power flow and operating characteristics. The constructional
features and important practical details are included at suitable places and the circumstances under which a
particular machine would be employed as a motor are discussed.
With the availability of electronic calculators, circle-diagram methods have lost their significance. However,
the circle diagram for the induction machine is included as it gives the complete machine performance at a
glance and is quite useful in qualitative reasoning.
A simple approach to machine dynamics is given in all the three machine types. In the case of the
synchronous machine, dynamics is restricted to the phenomenon of “hunting”, while transient stability
receives elementary treatment.
Chapter 10 deals with the important topic of fractional-kW motors. A qualitative-cum-heuristic analysis
of a single-phase induction motor and its circuit model are followed by a rigorously developed circuit model
for a two-winding motor. This rigorous coverage may be skipped when time does not permit it. A variety of
single-phase induction, synchronous and series commutator types of motors are treated. Comprehensiveness
is imparted to this chapter by the inclusion of stepper motors, ac servomotor and ac tachometer; the latter two
follow simply as a corollary from the two-winding motor analysis.
Chapter 11 is an entirely new chapter and deals with the generalised theory of electrical machines.
Probably the most significant development in recent years in the allied area of motor control is the use of
power semiconductors—diodes, power transistors and thyristors. The growth in this area has already qualified
for a separate undergraduate level course. However, for the sake of completeness, a comprehensive chapter is
included in this book. This in our view is a better approach than to burden the previous chapters by spreading
out the relevant details. Chapter 12 on this topic has a wide coverage and includes all the three varieties
of SCR (silicon controlled rectifier) circuitry, namely converters, choppers and inverters. The contents and
effects of non-smooth dc and nonsinusoidal ac outputs of these control equipment on the circuit behaviour
and on machine performance are beyond the scope of this book.
With the phenomenal developments in SCR circuitry for power control, cross-field machines and ac
commutator machines have become almost obsolete. However, to fulfil the need of such universities which
still include these topics in their curriculum, fairly detailed appendices (IV and V) on these topics are added.
A number of cross-sectional views of built-up machines and their parts are included and the student is
exhorted to carefully study these to help him visualize the physical picture of the machine being modelled.
Laboratory exercises always associated with a machines course will further aid this process.
A large number and variety of illustrative examples are spread throughout the book. These would greatly
help in imprinting a clear physical picture of the devices and associated physical reasoning on the student’s
mind. An equally large number of unsolved problems are given as exercises at the end of each chapter.
Answers to all the unsolved problems are given. Some of these problems are devised to illustrate some points
beyond what is directly covered in the text.
International Standard (SI) units are used throughout the book. The list of symbols is necessarily large.
Apart from being illustrated at the point of occurrence, the symbols used are listed in the beginning of the
book.
Web Supplements
The web supplements can be accessed at http://www.mhne.com/electmach4e and contain the following
material:
Contents
Preface
xvii
For Instructors: Solution Manual and Power Point Lecture Slides
For Students: Interactive Quiz and Web links for Study Material.
Acknowledgements
While revising the text, we have had the benefit of valuable advice and suggestions from many teachers,
students and other readers who used the earlier editions of this book. All these individuals have influenced
this edition. We express our thanks and appreciation to them. We hope this support / response would continue
in future also.
We are grateful to the authorities of VIT University, Vellore, for providing all the facilities necessary for
writing the book.
One of us (D P Kothari) wishes to place on record the thanks he owes to his colleagues—Mr K Palanisamy,
Mr Dilip Debnath, Mr Umashankar, Mr N Murali and Mr N Sreedhar—for their help in preparing and typing
rough drafts of certain portions of the manuscript, writing MATLAB programs and solving problems using
Simulink (MATLAB) and for helping in preparing the solutions of examples and unsolved problems of
certain chapters. We also express our appreciation for all those reviewers who took out time to review the
book. Their names are given below.
R K Jarial
National Institute of Technology, Hamirpur, Himachal Pradesh
K N Vaishnav
National Institute of Technology, Jaipur, Rajasthan
P P Tarang
JSS College of Technical Education, Noida, Uttar Pradesh
Imtiaz Ashraf
Aligarh Muslim University, Aligarh, Uttar Pradesh
Sanjay Parida
Indian Institute of Technology, Patna, Bihar
S N Mahto
National Institute of Technology, Durgapur, West Bengal
Urmila Kar
Netaji Subhash Engineering College, Kolkata
K K Ghosh
Dream Institute of Technology, Kolkata
N Kumaresan
National Institute of Technology, Trichy, Tamil Nadu
Ashok S
National Institute of Technology, Calicut, Kerala
A Nirmal Kumar
Bannari Amman Institute of Technology, Tamil Nadu
B K Murthy
National Institute of Technology, Warangal, Andhra Pradesh
T B Reddy
GPR Engineering College, Kurnool, Andhra Pradesh
K S Pawar
BSD College of Engineering, Dhule, Maharashtra
We also thank TMH personnel and our families who supported us during this period and given all possible
help so that this book could see the light of the day.
Feedback
We welcome any constructive criticism of the book and will be grateful for any appraisal by the readers. The
suggestions can be sent on my email: [email protected]
D P KOTHARI
I J NAGRATH
xviii Contents
Preface
Publisher’s Note
Tata McGraw-Hill invites suggestions and comments from you, all of which can be sent to
[email protected] (kindly mention the title and author name in the subject line). Piracy-related
issues may also be reported.
Introduction 1
1
1.1
INTRODUCTION
Electricity does not occur naturally in usable form and it also cannot be stored*
in usefully large quantities. Therefore, it must be generated continuously to
meet the demand (of power) at all times. An efficient and convenient way to
generate electric power is by conversion of mechanical power into electrical form
in a rotating device** called a generator. In the process a small part of power is lost in the generator
(efficiencies in large generators are above 90%). The mechanical power is itself obtained from heat power
by thermodynamical means in a steam turbine (efficiency in the range of 40–50% as the present upper limit)
or by conversion of potential energy of water in a hydraulic turbine with very little loss. The basic source
of mechanical power—steam/hydraulic turbine is called the prime mover. Electricity can also be generated
directly from hot gases in plasma form, obviating the need of converging heat power to intermediate
mechanical power. This process† is still in an experimental stage. The electromechanical process of electric
power generation is shown schematically in Fig. 1.1. Under steady conversion conditions,
Pelectrical
Shaft
w
Heat power
Electric
generator
Prime
mover
Pmechanical
TPM
TG
Losses
Fig. 1.1
Electric generator
TPM (prime mover) = TG (generator) and the turbine and generator run at steady speed.
Other than lighting and heating††, the major use of electric energy is made by converting it back to the
mechanical form to run the wheels of industry as well as tiny household appliances. The intermediary, the
* Attempts are on to store a sizeable amount of electric energy in large superconducting coils. While these attempts
are not likely to succeed in the near future, this stored energy would only be sufficient to meet sharp load peaks.
** The device always has an outer stationary member (refer to Sec. 1.2).
† The process is known as magnetohydrodynamics (MHD) which uses the Hall effect to generate electric power.
The process is inefficient because the outlet gases are at high temperature. By utilizing the hot gases in a convenient gas turbine, the composite process could be made more efficient than the conventional steam turbine.
† † It is expensive to use electricity for heating except in special processes (e.g. electric arc furnaces) and where
highly accurate controlled heating is required (e.g. induction heating).
2
Electric Machines
electric power, permits the use of large efficient central generating stations, while it is easily transported
to the myriads of use points. The electromechanical energy conversion process is a reversible one and
simple adjustment of mechanical shaft and
Pelectrical
Shaft
electrical conditions reverses the flow of power as
TL
w
illustrated in Fig. 1.2. In this mode of operation,
the electromechanical device, in general called
Load
Electric
(mechanical)
motor
the electric machine, is known as the motor and
the machine is said to be in the motoring mode.
TM
Under steady speed operation, again TM (motor)
Losses
Pmechanical
= TL (load). Both in generating and motoring
modes, losses occur in the electric machine but Fig. 1.2 Motoring mode of operation of an electric machine
the overall conversion efficiencies are very high
(close to or above 90%).
Electric machines are employed in almost every industrial and manufacturing process. Pages can be
filled in listing the applications of electric machines right from giant-size generators (500 MW and above),
industrial motors ranging up to a few megawatts to fractional-kW domestic appliances and to sophisticated
aerospace applications requiring stringent reliability in operation.
This book deals with the important topic of electric machines, the indepth understanding of which is
necessary to tackle the problems of energy, pollution and poverty that presently confront the whole of
mankind.
Since Thomas Alva Edison developed an electric generator, more than hundred years ago, engineers
have continually strived and successfully reduced the size and revised upwards the efficiencies of electric
machines by the use of improved materials and optimal design strategies. We appear to have reached close
to the upper limit imposed by nature.
A transformer is a static device that transforms electric energy from one ac voltage level to another. It
is this device that has made the electric system almost universally ac. The electric power is generated at
relatively low voltages (up to a maximum of 33 kV) which then is raised to very high voltages (e.g. 756 kV)
by means of a transformer and then transmitted. High voltages are associated with low currents and reduced
transmission losses. Geographically close to the use points, the electric power is transformed back to safe
low utility voltages (400/231 V). A transformer consists basically of two coils (three sets of coil pairs for
a 3-phase system) tightly coupled by means of magnetic (steel) core. Figure 1.3(a) gives the symbolic
Electric power
Electric power
(a) Transformer
Transmission line
Generator
(ac)
Transformer
(step-up)
Load
Transformer
(step-down)
(b) Simple electric power system
Fig. 1.3
Introduction 3
representation of a transformer and Fig. 1.3(b) shows a simple electric power generation transmission and
reception system. A practical electric power system is an integrated one, far more complex than the simple
diagrammatic representation of Fig. 1.3(b), and is in the form of an interconnected network for reasons of
economy, operational efficiency and reliability.
Because the principle of rotating ac machines is akin to that of a transformer, these two are always
studied together in a book. Further, since the transformer analogy can be extended to both the ac machine
types, the transformer study usually precedes the machine study.
1.2
BASIC PRINCIPLE, TYPES AND CONSTRUCTIONAL FEATURES
OF ELECTRIC MACHINES
There are three basic rotating electric machine types, namely
1. the dc machine,
2. the polyphase synchronous machine (ac), and
3. the polyphase induction machine (ac).
Three materials are mainly used in machine manufacture; steel to conduct magnetic flux, copper (or
aluminium) to conduct electric current and insulation to insulate the voltage induced in conductors confining
currents to them.
All electric machines comprise of two parts: the cylindrical rotating member called the rotor and the
annular stationary member called the stator with the intervening air-gap as illustrated in Fig. 1.4. The rotor
has an axial shaft which is carried on bearings at each end located in end covers bolted to the stator. The shaft
extends out of the end cover usually at one end and is coupled to either the prime mover or the load.
The stator and rotor are both made of magnetic material (steel) which conducts the magnetic flux upon
which depends the process of energy conversion. In both dc and synchronous machines, the main field is
created by field poles excited with direct current.
Air-gap
The winding on the field poles is called the field
Rotor
winding. The relative motion of the field past
a second winding located in the other member
Shaft
induces emf in it. The winding interchanges
Stator
current with the external electric system
depending upon the circuit conditions. It is this
winding, called the armature winding, which
handles the load power of the machine, while
the field winding consumes a small percentage
(0.5% to 2%) of the rated load power. The load
dependent armature current is known as load
Fig. 1.4 An electric machine
current.
In a dc machine the field poles are on the stator while the rotor is the armature as shown in the crosssectional view of Fig. 1.5. The field poles are symmetrical and are even in number, alternately north and
south. As the armature rotates, alternating emf and current induced in the armature winding are rectified to dc
form by a rotating mechanical switch called the commutator, which is tapped by means of stationary carbon
brushes. The commutator is cylindrical in shape and comprises severel wedge-shaped copper segments
4
Electric Machines
bound together while they are insulated from each other. The armature is made of laminated steel with slots
cut out on the periphery to accommodate the insulated armature winding. The ends of each armature coil
are connected to the commutator segments to form a closed winding. The armature when carrying current
produces stationary poles (same as number of field poles) which interact with the field poles to produce the
electromagnetic torque.
Main pole
Field winding
Pole shoe
Yoke
N
Armature
winding
+
Va(dc)
la
S
–
S
lf
Vf (dc)
N
Armature
Commutator
Brushes
Fig. 1.5 Cross-sectional view of dc machine
In a synchronous machine the field poles could be either on the stator or rotor, but in all practical machines
the rotor carries the field poles as shown in the cross-sectional view of Fig. 1.6. The field poles are excited
Armature winding
Field winding
N
la
3-phase
ac
lb
lc
S
S
Stator
lf
Vf (dc)
N
Rotor
Main pole
Pole Shoe
Fig. 1.6 Cross-sectional view of synchronous machine
Introduction 5
by direct current. The stator forms the armature carrying a 3-phase winding wound for the same number of
poles as the rotor. All the three phases have identical windings with the same angular displacement between
any pair of phases. When the rotor rotates, it produces alternating emf in each phase forming a balanced set
with frequency given by
f=
nP
120
(1.1)
f = frequency in Hz
n = rotor speed in rpm
P = number of field poles
For a given number of poles, there is a fixed correspondence between the rotor speed and the stator
frequency; the rotor speed is therefore called the synchronous speed. When balanced 3-phase currents are
allowed to flow in the armature winding, these produce a synchronously rotating field, stationary with respect
to the rotor field as a result of which the machine produces torque of electromagnetic origin. The synchronous
motor is, however, nonselfstarting.
In both dc and synchronous machines the power handling capacity is determined by the voltage and
current of the armature winding, while the field is excited from low power dc. Thus these machine types are
doubly excited. Quite different from these, an induction machine is singly excited from 3-phase mains on the
stator side. The stator winding must therefore carry both load current and field-producing excitation current.
The stator winding is 3-phase, similar to the armature winding of a synchronous machine. When excited it
produces a synchronously rotating field. Two types of rotor constructions are employed which distinguish the
type of induction motor.
where
Here the rotor has copper (or aluminium) bars embedded in slots which are shortcircuited at each end as shown in Fig. 1.7(a). It is a rugged economical construction but develops low starting
torque.
1. Squirrel-cage rotor
2. Slip-ring (or wound-rotor) rotor The rotor has a proper 3-phase winding with three leads brought out
through slip-rings and brushes as shown in Fig. 1.7(b). These leads are normally short-circuited when the
motor is running. Resistances are introduced in the rotor circuit via the slip-rings at the time of starting to
improve the starting torque.
The rotating field created by the stator winding moves past the shorted rotor conductors inducing currents
in the latter. These induced currents produce their own field which rotates at the same speed (synchronous)
with respect to the stator as the stator-produced field. Torque is developed by the interaction of these two
relatively stationary fields. The rotor runs at a speed close to synchronous but always slightly lower than it.
At the synchronous speed no torque can be developed as zero relative speed between the stator field and the
rotor implies no induced rotor currents and therefore no torque.
Single-phase ac motors are employed for low-voltage, low-power applications—fractional-kW motors.
They operate on the same basic principles as the 3-phase motor, but the pulsating single-phase field produces
additional losses, reducing motor torque and the pulsating torque component increases the noise level of the
motor.
An induction machine connected to the mains when driven at supersynchronous speed behaves as a
generator feeding power into the electric system. It is used in small hydroelectric stations and wind and
aerospace applications.
The insulation of a machine (or transformer) is its most vulnerable part because it cannot be stressed beyond
a certain temperature. For a given frame size, the steady temperature rise is determined by the machine loading,
6
Electric Machines
End
rings
Conducting bars embedded in
slots and shorted at both ends
by end rings
Brushes
(a) A squirrel-cage rotor (schematic diagram)
Slip-rings
Windings (details not shown)
embedded in slots, leads brought
out to slip-rings
(b) A wound rotor (schematic diagram)
Fig. 1.7
the associated power loss (this appears in the form of heat) and the cooling provided. Thus the maximum
machine loading called its rating for a given frame size is limited by the permissible temperature rise which
is dependent upon the class of insulation used. In the case of high-speed dc machines poor commutation
(reversal of current in armature coils) may become a limiting factor even before the temperature limit is
reached. The speed itself may be a limiting factor in very high-speed machines on account of the centrifugal
forces developed. This limit is more stringent in dc machines with complicated armature construction than
in the rugged rotor induction motor. Because of their high thermal capacity, machines are quite capable of
withstanding a fair amount of overloads for short durations.
Motor Control
There is great diversity and variety in the components and systems used to control rotating machines. The
purpose of a motor control may be as simple as start/stop or the control of one or more of the motor output
parameters, i.e. shaft speed, angular position, acceleration, shaft torque and mechanical power output. With
the rapid development of solid-state power devices, integrated circuits and cheap computer modules, the
range, quality and accuracy of electronic motor control has become almost infinite. Machines and other
electromechanical systems having the highest possible precision and reliability have been developed for
nuclear power and space applications. Using solid-state power converters, schemes have been devised to start,
stop or reverse dc motors in the megawatt range in a matter of seconds. Finally, as the nonconventional and
Introduction 7
renewable sources of energy, such as solar, windmill, etc., would become economical, viable electromechanical
energy converters will be required with matching characteristics.
Economic and Other Considerations
As in other devices, economics is an important consideration in the choice of electric machines and the
associated control gear. The trade-off between the initial capital investment and the operating and maintenance
cost must be taken into account in this choice; the decision may be in favour of a high-efficiency high-cost
motor, particularly in an environment of rising energy costs. While the transformer produces magnetic noise,
the rotating machines, in addition, produce mechanical noise arising from bearings, windage, etc. In presentday noise-pollution levels, the noise figure in decibels can be an important factor in motor choice. These
considerations are not the subject matter of this book which emphasises electromechanical principles and the
theory and application of electric machines including transformers.
1.3
RECENT TRENDS IN RESEARCH AND DEVELOPMENTS IN ELECTRIC MACHINES
Design and operation of electrical machines become easier and cheaper with suitable electric drives. This
electric drive converts and feeds the input energy to the machine according to the desired operation. A
power electronic converter constitutes the heart of the drive system which uses the power semiconductor
devices. These converters help to convert the power from one form to another. Various advancements in
converter topologies, and control methods have been proposed to convert and control the energy efficiently.
Other intelligent techniques such as Neural Network, Artificial Intelligence, Expert system, Fuzzy logic and
Evolutionary Computing are used to make the control most accurate and fast.
Research in converter topologies has also improved the power quality at the supply end. Various multipulse
and multilevel power electronic converters have been developed for power-quality improvement along with
cost-effective ac-dc converters for power factor improvement in electric drive system.
Electrical energy offers the most flexible, economic and efficient mode of power generation, transmission,
distribution and utilization compared to other forms of energy systems. Most of the power required for
human activities round the globe continues to come from electrical machines from the very large generators
installed in power stations to the very small motors in automotive control systems. The rapid depletion and the
increased cost of conventional fuels have given a thrust in the research on isolated asynchronous generators
as alternative power sources, converting from wind energy, biogas, hydro units and biomass. Asynchronous
generators operated in isolated systems for supplying electricity to the remote areas, where grid supply is
not accessible, are best options because of having certain advantages such as low cost, less maintenance and
brushless constructions.
In a wind-energy conversion system, the voltage and frequency variation at the generator terminals is
due to varying consumer loads as well as change in wind speeds. Therefore, the controller should have the
capability to control the voltage and frequency of isolated generators under dynamic conditions. Various
types of voltage and frequency controllers are proposed for constant-speed, constant-power applications.
Friction, vibration and noise can be eliminated in industrial drive by using permanent magnet direct-drive
technology. Both ac and dc PM motors are more suitable for high performance and wide speed variation
applications. Motors with higher torque and low speed are highly appreciated. PM brushless motors are of
this category.
The electric machines market is rapidly growing because of various developments and emerging areas
such as wind energy, marine, traction and offshore.
8 Electric Machines
The materials used for making the electrical machines play a major role in their performance. Particularly,
maximum temperature rise of the material (permanent magnet and insulation materials) used will affect
the rated torque of an electrical machine. The operating temperature strongly affects the performances of
electrical machines. Finite Element Analysis (FEA) can be used to compute the temperature of the material
and machines. A thermal network of the electrical machines is considered and input to this network is
calculated from magnetic and electrical loading of the machines. An Object Oriented Program (OOP) is used
to develop thermal network and it allows a convenient organization of thermal analysis process.
Modern commercial electrical steels can be grouped under non-oriented, grain-oriented and rapidly
quenched alloy types, of which the first two dominate the applications. Because of the limitations on the
shape of the magnetic path, grain-oriented types are used predominantly in large power and distribution
transformers, while non-oriented ones are used in rotating machines and small apparatus. The silicon content
is critical to the performance of electrical steels because it increases resistivity but it, also reduces the
anisotropy and so reduces losses and magnetostriction and permeability and makes the material brittle. In
non-oriented type, this content varies between 2.9% to 3.2%.
The continuing development of high permeability silicon-steel, metallic glass, ferrites, aluminum
ceramics and high temperature insulating materials, permanent magnetic materials like Neomax (Nd-Fe-B)
and rare-earth cobalts has been influencing the construction and design of many large and small machines and
apparatus from large generators to small step motors. Not only the cost of material per kVA has come down
in many cases, more reliable and highly efficient machines have been successfully designed.
Areas which have revolutionized the growth of research in electrical machines could be grouped as
(a)
(b)
(c)
(d)
(e)
(f )
(g)
(h)
Design of power electronic converters for motor drive and with better efficiency and control
Increase in rating of power semiconductor solid-state devices
Development of cheap digital signal processing controllers for operation and control
The development of evolutionary computing techniques, artificial intelligent techniques for machine
design, operation and control
Power quality and power factor improvement using improved power quality converters for various
motor-drive applications
Design of motor drives of electric and hybrid electric vehicles
Condition monitoring of electrical machines using Artificial Intelligence techniques
Design of electric machines using CAD/CAM techniques and validation through FE analysis
Thyristor is still unbeaten in high voltage and current rating among power semiconductor devices with
voltage and current rating of 12 kV and 6 kV, respectively. New devices such as IGCT, GCT, power MOSFET,
power IGBT have opened up new vistas for the electronic switching and control of energy-converting devices;
ac/dc, dc/ac, ac/ac and dc/dc converters are at their peak in various applications and totally changed the
way of operating the machines from the conventional mode of operation. Development of self-commutated
devices overcome the drawbacks of line commutated converters and it is used in both Voltage Source
Converters (VSC) and Current Source Converters (CSC). High voltage dc transmission (HVDC), Flexible ac
Transmission System (FACTS), variable frequency operation of machines, voltage control of dc machines,
Switched Mode Power Supply (SMPS), offer a wide variety of applications for converters and for power
engineers to develop new and realiable system configurations. The most common application is the variable
speed drives using dc and ac machines. Power electronics appear to have shifted the emphasis of electrical
engineers from the design of special types of variable speed machines to the use of special electronic circuits
to make an existing machine to give the desired variable speed characteristics and performances.
Introduction 9
Electric vehicles and hybrid electric vehicles are one of the most recent and potential applications of
electrical machines. These vehicles use the motor drives in association with power electronic converters.
Application of electrical drives in system automobile began with 6 V in the first quarter of the 20th century.
Initially, it was used only for basic and necessary functions such as ignition, cranking and lighting loads.
Since then, there has been a constant increase in power demand and now the whole vehicle is driven by the
electrical system. Electrical system has replaced all mechanical, pneumatic and hydraulic systems, thereby
increasing the efficiency and performance of the automobile system.
Present voltage level being used is 12 V, and in the future it is expected to switch over to higher levels
such as 42 V and 300 V. There is a immense challenge in design of power electronic converters and suitable
high-efficiency machines for hybrid electric vehicles.
To make the squirrel-cage induction motor run like a separately excited dc motor has remained a dream of
generations of electrical engineers. Direct Torque Control (DTC) has fulfilled this dream and simplified the
control circuit of induction motor to a great extent. The availability of cheap microprocessor-based system
has further improved the control of electrical devices of all forms for obtaining the desired steady state,
transient as well as dynamic characteristics from the existing drive machine with the help of suitable designed
solid-state control circuits.
Electrical machines and drive systems are subjected to many different faults. They may include stator
faults, rotor electrical faults and rotor mechanical faults, failure of power electronics system and damage of
mechanical parts. Most of industrial processes demand continuous operation. This is mainly influenced by
the condition monitoring leading to fault diagnosis and prediction of performance of electrical machines and
drives. Fast and accurate diagnosis of machine faults results in prevention of failure and avoiding processes
interrupt and reduce the idle time of machines. It also helps in reducing financial loss, avoids harmful effects
and devastation of the system.
For many years, the manufactures and users had relied on protective relays such as over-current and overvoltage relays to trip fault machines. This scheme may lead to machine damage and other harmful effects.
Various intelligence techniques such as Artificial Neural Networks (ANN), Expert System (ES), Fuzzy
System (FS) are being used for machines analysis for monitoring and control to make the process continuous,
fast and accurate.
One of the latest developments in ac motor research has been in the direction of field-oriented control
or making ac motors perform like dc motors with highly accurate torque and power control. Another recent
development in ac motor drives is a system called Direct Torque Control and Direct Self-Control. There is
no modulator and no need for an encoder to feedback information about motor shaft speed and position.
The DTC sensorless-type control incorporates fast digital signal processing hardware, resulting in a torque
response which is ten times faster than any ac or dc drive.
One of the reasons why ac technology will continue to make inroads into dc dominance is reduced power
consumption. Brushless dc will survive eventually but it is not dc at all but a completely different technology.
It is called dc because of the concept of external commutation.
Now there are increased levels of customer support. With the emphasis on just-in-time production,
downtime is unacceptable. With today’s systems, if the machine goes down one can, via modem, have a
technician at a remote site use system diagnosis software to troubleshoot the entire system from anywhere
in the world.
Large Rating Machines
The innovative design features as direct water-cooled armature windings, gap-pickup rotor winding cooling,
Micapal II stator insulation, Class F rotor and stator insulation, advanced Tetraloc stator endwinding
10
Electric Machines
support systems, and the side ripple-spring armature bar slot support structure were developed for heavy
rating machines. The Finite Element Model (FEM) is exercised to interrogate the generator assembly for all
loads encountered during assembly and operation. Electromagnetic force is the most likely force to cause
structural issues. This force is cyclic and acts in the radial direction at the inside diameter of the stator core
with a magnitude of one million pounds. The resulting stator core vibration and the transmission to the
generator structure and foundation is a significant design consideration. Forced harmonic response analyses
are performed to ensure that the electromagnetic forces cannot excite the machine’s natural frequencies.
The structural design of the stationary components also must be considered when calculating the dynamic
behavior of the rotor, since the rotor is supported on bearings located in the end shields of the machine. In
this load configuration, the structural vibratory loads caused by the rotor, and the loading caused by stator
vibration that drives rotor behavior, are interrogated. Once again, a forced harmonic analysis is performed
to understand and optimize the interactions. Lastly, the structural design has a major impact on the overall
producibility and serviceability of the generator. The complexity of the fabrication determines the unit’s
machining cycles as well as its accessibility for thoroughly cleaning the inner cavities of the machine before
shipping.
Generator 2 pole machine running at 3000/3600 rpm and 4 pole machine running at 1500/1800 rpm have
become common for large outputs or 1000 MW of more. The main dimension of 1000 MVA machine is the
output coefficient C which will be 2 MVA s/m3 and D2L will be 10 m3. The higher value of output coefficient
is made possible by enhanced cooling techniques. The excitation current is 7 kA at 650 V for 1500 MVA
machine and 5.7 kA, 640 V for 1000 MVA machine respectively.
The stator core, made from grain-oriented silicon steel for low loss and high permeability, is mounted
rigidly on the inner frame. Isolation of the core vibration from the remainder of the structure is accomplished
through the use of flexible pads between the feet on the inner frame and the base structure. The end windings
are secured to a strong structure of insulating materials to be used. A solid cone of filament-wound resin
bonded fibre glass is used to generate strength and long-term rigidity. The coils are bedded to the structure
with comfortable packing material. The complete structure is bolted to the end of the core. Axial movement
may be allowed to accommodate expansion of coils relative to the core. Low-loss stator core, of grainoriented silicon steel, minimizes electrical losses within the core to increase machine efficiency.
The insulation between turns is usually provided by interleaves of resin-bonded glass fabric material. The
stator winding arrangements for 1000 MVA should be four parallel paths in 2-pole machines and in 4-pole
machines, it can be in parallel or series parallel combination. The efficiency of 1000–1500 MVA is normally
very high. Due to this megawatt loss, good cooling medium is required. Direct hydrogen cooling of rotors is
developed for 1000 MVA. The cooling of the rotor can be done by 1). Each coil is wound with continuous
length of copper strap bent on edge at the four corners, 2). Larger section conductors of silver bearing coppercontaining grooves and holes to provide the passage of gas. Cool, deionized water, supplied by a closed-loop
auxiliary system, flows through copper strands in the stator winding, and the warm water is discharged at the
turbine end of the generator. The hydrogen-cooled machines can be designed with a higher electrical loading
than the air-cooled machines due to the better cooling, and tend to have a larger subtransient reactance than
an air-cooled machine.
The generator’s performance is heightened by optimization of its bar strand configuration, including
hollow-to-solid-strand ratio. Spring bar stator core support system isolates vibration of the stator core, to
minimize vibration transmitted to the foundation. Stator winding support features top wedges and ripple
springs to secure stator bars in the slot and eliminate bar vibration. This maximizes insulation life and reduces
maintenance requirements. Core-end cooling is enhanced through proven design concepts to control core
Introduction 11
P = K ¢Di 2 ¢L ¢n
High TS 4Ni-Cr
shaft forging
Core end with
flux shunt
Unequal section
mix strand water
cooled stator
coil
Improved coil end
support
High TS 18 Mn-18Cr RR
Optimized L/D
rotor
Maximized field
conductor area
Center grooved
elliptical
journal bearing
Fig. 1.8
2 piece babbitt
sealing ring
Low Fn compact
frame
Hydrogen
cooled HVB
Internal view of 1000 MW generator (Courtesy, Toshiba Corporatin)
temperatures and minimize eddy current losses. Approaches include split tooth, stepped core, flux shields
and non-magnetic materials. Retaining rings of 18-Mangaanese/18-Chromium, non-magnetic stainless steel
resists stress-corrosion cracking.
Internal view of a 1000 MW generator is shown in Fig. 1.8.
Generator Parameters (600 MW)
Parameters
Unit
Rated capacity
Rated power factor
Synchronous reactance
Transient reactance
Sub-transient reactance
Nagative-seq reactance
Zero-seq reactance
Direct axis time constant
Open circuit time constant
Short circuit time constant
Open circuit sub-transinet
Short circuit sub-transient
Qudrature axis time constant
Open circuit time constant
Short circuit time constant
Open circuit sub-transinet
Short circuit sub-transient
MW
Xd%
Xd¢%
Xd≤%
X2%
X0%
600
0.85
240.313
28.281
21.582
21.295
10.131
Tdo¢ Sec
Td¢ Sec
Tdo≤ Sec
Td≤ Sec
8.724
1.027
0.046
0.035
Tdo¢ Sec
Td¢ Sec
Tdo≤ Sec
Td≤ Sec
0.969
0.169
0.068
0.035
12
Electric Machines
2
2.1
INTRODUCTION
The electromagnetic system is an essential element of all rotating electric
machinery and electromechanical devices as well as static devices like the
transformer. The role of the electromagnetic system is to establish and control
electromagnetic fields for carrying out conversion of energy, its processing and
transfer. Practically all electric motors and generators, ranging in size from fractional horsepower units
found in domestic appliances to the gigantic several thousand kW motors employed in heavy industry and
several hundred megawatt generators installed in modern generating stations, depend upon the magnetic
field as the coupling medium allowing interchange of energy in either direction between electrical and
mechanical systems. A transformer though not an electromechanical conversion device, provides a means
of transferring electrical energy between two electrical ports via the medium of a magnetic field. Further,
transformer analysis runs parallel to rotating machine analysis and greatly aids in understanding the latter.
It is, therefore, seen that all electric machines including transformers use the medium of magnetic field for
energy conversion and transfer. The study of these devices essentially involves electric and magnetic circuit
analysis and their interaction. Also, several other essential devices like relays, circuit breakers, etc. need the
presence of a confined magnetic field for their operation.
The purpose of this chapter is to review the physical laws governing magnetic fields, induction of emf
and production of mechanical force, and to develop methods of magnetic-circuit analysis. Simple magnetic
circuits and magnetic materials will be briefly discussed. In the chapters to follow, how the concepts of this
chapter are applied in the analysis of transformers and machines will be shown.
2.2
MAGNETIC CIRCUITS
The exact description of the magnetic field is given by the Maxwell’s equations* and the constitutive
relationship of the medium in which the field is established.
—◊B = 0
and
—◊D = r
∂D
–12
and
—¥H = J +
D = e0 E ; e0 = 8.85 ¥ 10
∂t
wherein J = conduction current density and D = displacement current density, negligible for slowly-varying
fields D = e0 E ; e0 = 8.85 ¥ 10–12 F/m).
Magnetic Circuits and Induction
13
Such description apart from being highly complex is otherwise not necessary for use in electric machines
wherein the fields (magnetic and electric) are slowly varying (fundamental frequency being 50 Hz) so that
the displacement current can be neglected. The magnetic field can then be described by Ampere’s law and is
solely governed by the conduction current. This law is in integral form and is easily derivable from the third
Maxwell’s equation (by ignoring displacement current) by means of well-known results in vector algebra.
The Ampere’s law is reproduced as follows:
J ◊ ds = H ◊ dl
(2.1)
s
wherein J = conduction current density
H = magnetic field intensity
s = the surface enclosed by the closed path of length l
d s = differential surface
d l = differential length
Ú
Ú
Consider the example of a simple electromagnetic system comprising an exciting coil and ferromagnetic
core as shown in Fig. 2.1. The coil has N turns and carries a constant (dc) current of i A. The magnetic field is
established in the space wherein most of the total magnetic flux set up is confined to the ferromagnetic core
for reasons which will soon become obvious. Consider the flux path through the core (shown dotted) which in
fact is the mean path of the core flux. The total current piercing the surface enclosed by this path is as follows:
J ◊ ds = Ni
Ú
s
Hc
i
Core mean
length lc
(Ferromagnetic)
+
e
N
Leakage fiux
–
ca
b
Exciting coil
d
Fig. 2.1 A simple magnetic system
Hence Eq. (2.1) acquires the form
Ni =
Ú H ◊ dl
(2.2)
l
Since N is the number of coil turns and i the exciting current in amperes, the product F = Ni has the units of
ampere-turns (AT) and is the cause of establishment of the magnetic field. It is known as the magnetomotive
force (mmf ) in analogy to the electromotive force (emf ) which establishes current in an electric circuit.
14 Electric Machines
The magnetic field intensity H causes a flux density B to be set up at every point along the flux path which
is given by
and
B = mH = m0 mrH
B = m0H
(for flux path in core)
(for flux path in air)
(2.3a)
(2.3b)
The units of flux density are weber (Wb )/m2 called tesla (T). The term m0 is the absolute permeability of
free space and has a value of
m0 = 4p ¥ 10–7
henry (H)/m
The permeability m = m0 mr of a material medium is different from m0 because of a certain phenomenon
occurring in the material. The term mr is referred to as relative permeability of a material and is in the range of
2000-6000 for ferromagnetic materials (see Sec. 2.3). It is, therefore, seen that for a given H, the flux density
B and, therefore, the flux over a given area
f=
Ú
B◊ds
s
will be far larger in the magnetic core in Fig. 2.1 than in the air paths. Hence, it is safe to assume that the
magnetic flux set up by mmf Ni is mainly confined to the ferromagnetic core and the flux set up in air paths
is of negligible value. The flux set up in air paths is known as the leakage flux as if it leaks through the core;
some of the leakage flux paths are shown chain-dotted in Fig. 2.1. There is no way to avoid magnetic leakage
as there are no magnetic insulators in contrast to electric insulators which confine the electric current to the
conductor for all practical purposes. The effect of the leakage flux is incorporated in machine models through
the concept of the leakage inductance.
The direction of field intensity is H and so the direction of flux f is determined from the Right Hand Rule
(RHR). It is stated as:
Imagine that you are holding a current carrying conductor in your right hand with the thumb pointing in
the direction of current. Then the direction in which the fingers curl gives the direction of flux. In case of a
coil you imagine that you are grasping the coil in right hand with the thumb in the direction of current; then
the fingers curl in the direction of flux.
The reader may apply RHR to the exciting coil in Fig. 2.1 to verify the direction of flux as shown in the
figure.
The magnetic field intensity H is tangential to a flux line all along its path, so that the closed vector
integration in Eq. (2.2) along a flux-line reduces to closed scalar integration, i.e.
Ni =
Ú H · dl
(2.4)
l
With the assumption of negligible leakage flux, the flux piercing the core cross-section at any point
remains constant. Further, from the consideration of symmetry it immediately follows that the flux density
over straight parts of the core is uniform at each cross-section and remains constant along the length; such
that H is constant along the straight parts of the core. Around the corners, flux lines have different path lengths
between magnetic equipotential planes (typical ones being ab and cd shown in Fig. 2.1) so that H varies from
a high value along inner paths to a low value along outer paths. It is reasonable to assume that H shown dotted
along the mean path will have the same value as in straight parts of the core (this mean path technique renders
simple the analysis of magnetic circuits of machines and transformers).
Magnetic Circuits and Induction
15
It has been seen previously that the magnetic field intensity along the mean flux path in the core can be
regarded constant at Hc. It then follows from Eq. (2.4) that
F = Ni = Hc lc
where F = mmf in AT
From Eq. (2.5)
and
(2.5)
lc = mean core length (m)
Ni
AT/m
(2.6)
lc
If one now imagines that the exciting current i varies with time, Eq. (2.6) would indicate that Hc will vary
in unison with it. Such fields are known as quasi-static fields in which the field pattern in space is fixed but
the field intensity at every point varies as a replica of the time variation of current. This simplified field picture
is a consequence of negligible displacement current in slowly-varying fields as mentioned earlier. In a quasistatic field, the field pattern and field strength at a particular value of time-varying exciting current will be the
same as with a direct current of that value. In other words, a field problem can be solved with dc excitation
and then any time variation can be imparted to it.
Now, the core flux density is given by,
Hc =
Bc = mc Hc
tesla (T)
and core flux (assumed to be total flux) is given by,
f = B ◊ d s = Bc Ac
Ú
Wb
s
where Ac = cross-sectional area of core and flux in the limbs is oriented normal to cross-sectional area. Then
from Eq. (2.6)
f = mc HcAc =
where
R=
Ni
Ê lc ˆ
ÁË m A ˜¯
c c
or
f=
F
= FP
R
F
l
= c = reluctance* of the magnetic circuit (AT/Wb)
f
mc Ac
(2.7)
(2.8)
and P = 1/R = permeance of the magnetic circuit. It is, therefore, seen that by certain simplifying assumptions
and field symmetries, it has been possible to lump the distributed magnetic system into a lumped magnetic
circuit described by Eq. (2.7) which is analogous to Ohm’s law in dc circuits. The electrical circuit analog
of the magnetic system (now reduced to a magnetic circuit) is shown in
f (∼ i)
Fig. 2.2 wherein F (mmf ) is analogous to E (emf ), R (reluctance) is
analogous to R (resistance) and f (flux) is analogous to i (current).
F (∼ E)
R (∼ R)
The analogy though useful is, however, not complete; there being two
points of difference: (i) magnetic reluctance is nondissipative of energy
unlike electric resistance, (ii) when F is time-varying, the magnetic Fig. 2.2 Electrical analog of the
circuit still remains resistive as in Fig. 2.2, while inductive effects are
simple magnetic circuit of
bound to appear in an electric circuit. This is because there is no timeFig. 2.1
lag between the exciting current and the establishment of magnetic flux
(quasi-static field).
* Unit of reluctance is AT/Wb and will not be specified every time in examples.
16
Electric Machines
The lumped magnetic circuit and its electrical analog are useful concepts provided the permeability (m) of
the core material and, therefore, the core reluctance is constant as is tacitly assumed above. This, however,
is not the case with ferromagnetic materials, but when air-gaps are involved, the assumption of constant
reluctance is generally valid and leads to considerable simplicity in magnetic circuit analysis.
In more complicated magnetic circuits—with multiple excitations and series-parallel core arrangement—
the general theorems of electric circuits apply, i.e. Kirchhoff’s voltage (mmf ) law and Kirchhoff’s current
(flux) law. This is illustrated in Example 2.3.
B-H Relationship (Magnetization Characteristic)
In free space (also nonmagnetic materials), the permeability m0 is constant so that B-H relationship is linear.
This, however, is not the case with ferromagnetic materials used in electric machines, wherein the B-H
relationship is strictly nonlinear in two respects—
(T )
saturation and hysteresis. Hysteresis non-linearity B
Saturation zone
is the double valued B-H relationship exhibited in
cyclic variation of H (i.e. exciting current). This
nonlinearity is usually ignored in magnetic circuit
calculations and is important only when current
wave shape and power loss are to be accounted
Linear zone
for. This is discussed in Sections 2.3 and 2.6. A
(constant m)
typical normal B-H relationship (magnetization
characteristic) for ferromagnetic materials is shown
lnitial nonlinear zone
in Fig. 2.3. It has an initial nonlinear zone, a middle
0
H(AT/m)
almost linear zone and a final saturation zone in
which B progressively increases less rapidly with H Fig. 2.3 Typical normal magnetization curve of
ferromagnetic material
compared to the linear zone. In the deep saturation
zone, the material behaves like free space.
Due to considerations dictated by economy, electric machines and transformers are designed such that
the magnetic material is slightly saturated (i.e. somewhat above the linear zone). In exact magnetic circuit
calculations the nonlinear magnetization curve has to be used necessitating graphical/numerical solutions.
Core with Air-gap
Transformers are wound on closed cores as in Fig. 2.1. Rotating machines have a moving element and must
therefore have air-gaps in the cores out of necessity. A typical magnetic circuit with an air-gap is shown in
Fig. 2.4. It is assumed that the air-gap is narrow and the flux coming out of the core passes straight down the
air-gap such that the flux density in the air-gap is the same as in the core. Actually as will soon be seen, that
the flux in the gap fringes out so that the gap flux density is somewhat less than that of the core. Further, let
the core permeability mc be regarded as constant (linear magnetization characteristic).
The mmf Ni is now consumed in the core plus the air-gap. From the circuit model of Fig. 2.4(b) or directly
from Fig. 2.4(a)
Ni = Hc lc + Hglg
or
Ni =
Bg
Bc
lc +
lg
mc
m0
(2.9a)
(2.9b)
Magnetic Circuits and Induction
17
Hc
Mean core length, lc
i
Cross-sectional
area A
+
Air-gap; Hg, lg
e
N
Rc
_
f
Rg
Ni
(b)
(a)
Fig. 2.4 A typical magnetic circuit with air-gap and its equivalent electric circuit
Assuming that all the core flux passes straight down the air-gap (it means no fringing (see Fig. 2.5))
\
Bg = Bc
f = Bc A = Bg A
(2.10)
Substituting Eq. (2.10) in Eq. (2.9b)
Ê lg ˆ
Ê l ˆ
Ni = f Á c ˜ + f Á
Ë mc A ¯
Ë m0 A ˜¯
(2.11)
Recognizing various quantities in Eq. (2.11)
F = f (Rc + Rg) = fReq
where
Rc =
Rg =
(2.12)
lc
= core reluctance
mc A
lg
m0 A
= air-gap reluctance
From Eq. (2.12)
f=
But
R
c
Rg
=
F /R
F
Rc + Rg
0 lc
clg
=
g
1 + Rc / Rg
(2.13)
1
because mc is 2000 to 6000 times m0 in ferromagnetic materials. The permeability effect predominates the
usual core and air-gap dimensions even though lc lg. It then follows from Eq. (2.13), that
f ª F/Rg
(2.14)
which means that in a magnetic circuit with air-gap(s), core reluctance may be neglected with no significant
loss of accuracy. This assumption will be generally made in modelling rotating machines. The effect of core
saturation (reduction of core permeability) will be introduced as a correction wherever greater accuracy is
desired.
18 Electric Machines
Magnetic Circuit Calculations
Normally magnetic circuit calculations involve two types of problems. In the first type of problem it is
required to determine the excitation (mmf ) needed to establish a desired flux or flux density at a given point
in a magnetic circuit. This is the normal case in designing electromechanical devices and is a straight forward
problem. In the second category, the flux (or flux density) is unknown and is required to be determined for a
given geometry of the magnetic circuit and specified mmf. This kind of problem arises in magnetic amplifiers
wherein this resultant flux is required to be determined owing to the given excitation on one or more control
windings. A little thought will reveal that there is no direct analytical solution to this problem because of the
non-linear B-H characteristic of the magnetic material. Graphical/numerical techniques have to be used in
obtaining the solution of this problem.
Leakage Flux
In all practical magnetic circuits, most of the flux is confined to the intended path by use of magnetic cores but
a small amount of flux always leaks through the surrounding air. This stray flux as already stated is called the
leakage flux, Leakage is characteristic of all magnetic circuits and can never be fully eliminated. Calculations
concerning the main magnetic circuit are usually carried out with the effect of leakage flux either ignored or
empirically accounted for. Special studies of leakage must be made for ac machines and transformers since
their performance is affected by it.
Fringing
At an air-gap in a magnetic core, the flux fringes out into neighbouring air paths as shown in Fig. 2.5;
these being of reluctance comparable to that of the gap. The result is nonuniform flux density in the airgap (decreasing outward), enlargement of the effective airFringing flux
gap area and a decrease in the average gap flux density. The
fringing effect also disturbs the core flux pattern to some
depth near the gap. The effect of fringing increases with the
Air
Core
Core
gap
air-gap length. Corrections for fringing in short gaps (as used
in machines) are empirically made by adding one gap length
to each of the two dimensions making up its area. For the
Fig. 2.5 Flux fringing at air-gap
example of the core with the air-gap previously presented, the
gap reluctance would now be given by
Rg =
lg
m0 Ag
which will be less than the previous value as Ag > A.
It can be shown theoretically that the magnetic flux leaves and enters the surface of an infinitely permeable
material normally. This will be nearly so in ferromagnetic materials which have high permeability. In electric
machines a small amount of the tangential flux component present at iron surfaces will be neglected.
Stacking Factor
Magnetic cores are made up of thin, lightly insulated (coating of varnish) laminations to reduce power loss
in cores due to the eddy-current phenomenon (explained in Sec. 2.6). As a result, the net cross-sectional area
Magnetic Circuits and Induction
19
of the core occupied by the magnetic material is less than its gross cross-section; their ratio (less than unity)
being known as the stacking factor. Depending upon the thickness of laminations, stacking factor may vary
from 0.5–0.95, approaching unity as the lamination thickness increases.
EXAMPLE 2.1 The magnetic circuit of Fig. 2.4(a) has dimensions: Ac = 4 ¥ 4 cm2, lg = 0.06 cm, lc =
40 cm; N = 600 turns. Assume the value of mr = 6000 for iron. Find the exciting current for Bc = 1.2 T and
the corresponding flux and flux linkages.
SOLUTION
From Eq. (2.9), the ampere-turns for the circuit are given by
Ni =
Bg
Bc
lc +
lg
m0 m r
m0
(i)
Neglecting fringing
Ac = Ag
Then
i=
therefore
Bc = Bg
ˆ
Bc Ê lc
+ lg ˜
Á
m0 N Ë m r
¯
1.2
ˆ
Ê 40
+ 0.06˜ ¥ 10–2
Á
¯
4 p ¥ 10- 7 ¥ 600 Ë 6000
= 1.06 A
=
(ii)
Ê 2/3ˆ
= 0.11 of the reluctance of the
The reader should note that the reluctance of the iron path of 40 cm is only Á
Ë 6 ˜¯
0.06 cm air-gap.
Flux linkages,
f = Bc Ac = 1.2 ¥ 16 ¥ 10–4 = 19.2 ¥ 10–4 Wb
l = Nf = 600 ¥ 19.2 ¥ 10–4 = 1.152 Wb-turns
If fringing is to be taken into account, one gap length is added to each dimension of the air-gap constituting the area.
Then
Ag = (4 + 0.06) (4 + 0.06) = 16.484 cm2
Effective Ag > Ac reduces the air-gap reluctance. Now
Bg =
19.2 ¥ 10- 4
16.484 ¥ 10- 4
= 1.165 T
From Eq. (i)
i=
=
ˆ
1 Ê Bclc
+ Bg l g ˜
m0 N ÁË m r
¯
(iii)
Ê 1.2 ¥ 40 ¥ 10- 2
ˆ
+ 1.165 ¥ 0.06 ¥ 10- 2 ˜
Á
-7
6000
4 p ¥ 10 ¥ 600 Ë
¯
1
= 1.0332 A
EXAMPLE 2.2 A wrought iron bar 30 cm long and 2 cm in diameter is bent into a circular shape as
shown in Fig. 2.6. It is then wound with 600 turns of wire. Calculate the current required to produce a flux
of 0.5 mWb in the magnetic circuit in the following cases:
(i) no air-gap;
(ii) with an air-gap of 1 mm; mr (iron) = 4000 (assumed constant); and
20
Electric Machines
(iii) with an air-gap of 1 mm; assume the following data for the magnetization of iron:
H in AT/m
B in T
2500
1.55
3000
1.59
3500
1.6
4000
1.615
SOLUTION
(i) No air-gap
Rc =
30 ¥ 10- 2
4000 ¥ 4 p ¥ 10- 7 ¥ p ¥ 10- 4
= 1.9 ¥ 105
Ni = fRc
or
30 cm, core length
i
1 mm
N = 600
0.5 ¥ 10- 3 ¥ 1.9 ¥ 105
i = fRc /N =
= 0.158 A
600
–4
Ac = p ¥ 10 m
(ii) Air-gap = 1 mm, mr (iron) = 4000
2
Rc = 1.9 ¥ 105 (as in part (i)
Rg =
1 ¥ 10- 3
= 25.33 ¥ 105
4 p ¥ 10- 7 ¥ p ¥ 10- 4
Fig. 2.6
R(total) = Rc + Rg = 27.1 ¥ 105
\
i=
0.5 ¥ 10- 3 ¥ 27.1 ¥ 105
= 2.258 A
600
(iii) Air-gap = 1 mm; B-H data as given
Bc = Bg =
Hg =
Bg
m0
0.5 ¥ 10-3
p ¥ 10- 4
=
ATg = Hg lg =
= 1.59 T (fringing neglected)
1.59
4 p ¥ 10- 7
1.59 ¥ 1 ¥ 10- 3
4 p ¥ 10- 7
= 1265
From the given magnetization data (at Bc = 1.59 T),
Hc = 3000 AT/m
ATc = Hc lc = 3000 ¥ 30 ¥ 10–2 = 900
AT (total) = ATc + ATg
= 900 + 1265 = 2165
2165
= 3.61 A
i=
600
EXAMPLE 2.3
The magnetic circuit of Fig. 2.7 has cast steel core with dimensions as shown:
Mean length from A to B through either outer limb = 0.5 m
Mean length from A to B through the central limb = 0.2 m
In the magnetic circuit shown it is required to establish a flux of 0.75 mWb in the air-gap of the central
limb. Determine the mmf of the exciting coil if for the core material (a) mr = (b) mr = 5000. Neglect
fringing.
Magnetic Circuits and Induction
21
Mean flux path
f1
f2
A
1 ¥ 1 cm
f
i
N
0.02 cm
2
2 cm
0.02 cm
3
0.025 cm
1
1 ¥ 1 cm
B
Fig. 2.7
SOLUTION
(a) mr = , i.e. there are no mmf drops in the magnetic core. It is easy to see from Fig. 2.7 that the two outer limbs
present a parallel magnetic circuit. The electrical analog of the magnetic circuit is drawn in Fig. 2.8(a). Various
gap reluctances are:
Rg1 =
Rg2 =
Rg3 =
0.025 ¥ 10- 2
= 1.99 ¥ 106
4 p ¥ 10- 7 ¥ 1 ¥ 10- 4
0.02 ¥ 10- 2
1.592 ¥ 106
4 p ¥ 10- 7 ¥ 1 ¥ 10- 4
0.02 ¥ 10- 2
= 0.796 ¥ 106
4 p ¥ 10- 7 ¥ 2 ¥ 10- 4
From Fig. 2.8(b),
Ni = 0.75 ¥ 10–3 (Rg3 + Rg1 || Rg2)
= 0.75 ¥ 10–3 (0.796 + 0.844) ¥ 106
= 1230 AT
f = 0.75 mWb
0.75 mWb
N
Rg
2
Rg
3
Ni
Rg
1
R g || R g
2
3
(a)
Fig. 2.8
1
Rg
(b)
Electrical analog of Fig. 2.7
(b) mr = 5000. This means that the reluctance of magnetic core must be taken into consideration. The analogous
electric circuit now becomes that of Fig. 2.9.
22
Electric Machines
Since gap lengths are negligible compared to core lengths, various core reluctances can be calculated as follows:
Rc1 =
0.5
4 p ¥ 10
-7
¥ 5000 ¥ 1 ¥ 10- 4
= 0.796 ¥ 106
Rc2 = Rc1 = 0.796 ¥ 106
Rc3 =
0.75 mWb
Ni
R c2
0.2
4 p ¥ 10- 7 ¥ 5000 ¥ 2 ¥ 10- 4
= 0.159 ¥ 10
R c3
6
R g2
R g3
The equivalent reluctance is
Req = (Rc1 + Rg1) || (Rc2 + Rg2) + Rc3 + Rg3)
R c1
R g1
Fig. 2.9
27.86 ¥ 23.86
¥ 106 + 0.955 ¥ 106 = 1.955 ¥ 106
=
51.72
Now Ni = fReq
= 0.75 ¥ 10–3 ¥ 1.955 ¥ 106
= 1466 AT
EXAMPLE 2.4 The magnetic circuit of Fig. 2.10 has cast steel core. The cross-sectional area of the
central limb is 800 mm2 and that of each outer limb is 600 mm2. Calculate the exciting current needed
to set up a flux of 0.8 mWb in the air gap. Neglect magnetic leakage and fringing. The magnetization
characteristic of cast steel is given in Fig. 2.16.
1 mm
400
mm
500 turns
160 mm
400
mm
Fig. 2.10
SOLUTION
Air gap
Central limb
From Fig. 2.16
1
0.8 10- 3
¥
= 1 T and Hg =
AT/m
800 10- 6
4 p ¥ 10- 7
1
Fg =
¥ 1 ¥ 10–3 = 796 AT
4 p ¥ 10- 7
Bg =
B c = Bg = 1 T
Hc = 1000 AT/m
Fc = 1000 ¥ 160 ¥ 10–3 = 160 AT
Magnetic Circuits and Induction
23
Because of symmetry, flux divides equally between the two outer limbs. So
f (outer limb) = 0.8/2 = 0.4 mWb
B (outer limb) =
0.4 ¥ 10- 3
600 ¥ 10- 6
= 0.667 AT
F (outer limb) = 375 ¥ 400 ¥ 10–3 = 150 AT
F (total) = 796 + 160 + 150 = 1106 AT
Exciting current = 1106/500 = 2.21 A
EXAMPLE 2.5
The magnetic circuit of Fig. 2.11 has a cast steel core whose dimensions are given below:
Length (ab + cd) = 50 cm
Cross-sectional area = 25 cm2
Length ad = 20 cm
Cross-sectional area = 12.5 cm2
Length dea = 50 cm
Cross-sectional area = 25 cm2
Determine the exciting coil mmf required to establish an air-gap flux of 0.75 m Wb. Use the B-H curve
of Fig. 2.16.
b
F
e
c
d
Fig. 2.11
SOLUTION
Assuming no fringing the flux density in the path abcd will be same, i.e.
B=
Fbc =
0.75 ¥ 10- 3
25 ¥ 10- 4
= 0.3 T
0.3 ¥ 0.25 ¥ 10- 3
B
= 60 AT
lbc =
m0
4 p ¥ 10- 7
Hab = Hcd (from Fig. 2.16 for cast steel for B = 0.3 T) = 200 AT/m
Fab+cd = 200 ¥ 50 ¥ 10–2 = 100 AT
\
Fad = 60 + 100 = 160 AT
160
Had =
= 800 AT/m
20 ¥ 10- 2
0.25 cm
a
24
Electric Machines
Bad (from Fig. 2.16) = 1.04 T
fad = 1.04 ¥ 12.5 ¥ 10–4 = 1.3 mWb
fdea = 0.75 + 1.3 = 2.05 mWb
2.05 ¥ 10- 3
Bdea =
25 ¥ 10- 4
= 0.82 T
Hdea (from Fig. 2.16) = 500 A T/m
Fdea = 500 ¥ 50 ¥ 10–2 = 250 AT
F = Fdea + Fad = 250 + 160 = 410 AT
EXAMPLE 2.6 A cast steel ring has a circular cross-section of 3 cm in diameter and a mean circumference
of 80 cm. A 1 mm air-gap is cut out in the ring which is wound with a coil of 600 turns.
(a) Estimate the current required to establish a flux of 0.75 mWb in the air-gap. Neglect fringing and
leakage.
(b) What is the flux produced in the air-gap if the exciting current is 2 A? Neglect fringing and leakage.
Magnetization data:
H (AT/m)
B (T)
200
0.10
400
0.32
600
0.60
800
0.90
1000
1.08
1200
1.18
1400
1.27
1600
1.32
1800
1.36
2020
1.40
SOLUTION
f = 0.75 ¥ 10–3 Wb
0.75 ¥ 10
Ê 0.03 ˆ
p ¥Á
Ë 2 ˜¯
2
=1.06 T
1.2
1.0
Bc = Bg (no fringing)
Reading from the B-H curve drawn in Fig. 2.12,
lc = 0.8 m (air-gap length can be neglected)
ATc = Hc lc = 900 ¥ 0.8 = 720
1.06
ATg =
¥ 10–3 = 843
4 p ¥ 10- 7
Ni = ATc + ATg = 720 + 843 = 1563
i=
0.8
0.6
Hc = 900 AT/m
Therefore
B(T)
Bg = f/A =
1.6
1.5
1.4
-3
Ni = HcIc +
Bc
Bc
I
m0 g
0.4
0.2
0
0
1563
= 2.6 A
600
400
Hc 800
1200
1600
2000
H in AT/m
Fig. 2.12
(b) The excitation is now given and the flux is to be determined from the B-H curve given. The problem must,
therefore, be solved numerically/graphically. It is solved here graphically. Now
B2
Ni =
l + Hc lc; (Bg = Bc)
(i)
m0 g
This is a linear equation in Bc and Hc; the second equation is the nonlinear B-H curve. The intersection of the
two for a given Ni will yield the solution. For this problem
Ni = 600 ¥ 2 = 1200 AT
Magnetic Circuits and Induction
25
Substituting various values in Eq. (i)
1200 =
Bc
4 p ¥ 10- 7
¥ 10–3 + 0.8 Hc
(ii)
This equation is plotted in Fig. 2.12, by locating the points
Hc = 0,
Bc = 0,
Bc = 1.5
Hc = 1500
The intersection gives the result
Bc = 0.78 T
f = BcA = 0.78 ¥
2.3
p
(0.03)2 = 0.55 mWb
4
MAGNETIC MATERIALS AND THEIR PROPERTIES
From the magnetic point of view* a material is classified according to the nature of its relative permeability
(mr). All nonmagnetic materials are classified as paramagnetic, mr slightly greater than 1, and diamagnetic,
mr slightly less than 1. For all practical purposes, mr of these materials can be regarded as unity, i.e. their
magnetic properties are very much similar to that of free space. Such materials are not of interest to us in this
treatise.
Materials which are of interest to us are those whose relative permeability is much higher than that of free
space. These can be classified as ferromagnetic and ferrimagnetic. Ferromagnetic materials can be further
subdivided as hard and soft. Hard ferromagnetic materials include permanent magnet materials, such as
alnicos, chromium steels, certain copper-nickel alloys and several other metal alloys. Soft ferromagnetic
materials are iron and its alloys with nickel, cobalt, tungsten and aluminium. Silicon steels and cast steels
are the most important ferromagnetic materials for use in transformers and electric machines. Ferrimagnetic
materials are the ferrites and are composed of iron oxides—MeO. Fe2O3, where Me represents a metallic
ion. Ferrites are also subgrouped as hard (permanent magnetic) and soft (nickel-zinc and manganese-zinc)
ferrites. Soft ferrites are quite useful in high frequency transformers, microwave devices, and other similar
high-frequency operations. There is a third category of magnetic materials, known as superparamagnetic,
made from powdered iron or other magnetic particles. These materials are used in transformers for electronics
and cores for inductors. Permalloy (molybdenum-nickel-iron powder) is the best known example of this
important category of magnetic materials.
Properties of Magnetic Materials
Magnetic materials are characterized by high permeability and the nonlinear B-H relationship which
exhibits both saturation and hysteresis. The physics of these properties is explained by the domain theory of
magnetization usually taught in junior level courses.
The B-H relationship for cyclic H is the hysteresis loop shown in Fig. 2.13 for two values of maximum
flux density. It is easily observed from this figure that B is a symmetrical two-valued function of H; at any
given H, B is higher if H is reducing compared to when H is increasing. This is the basic hysteresis property
in which B lags behind H. It can also be recognized as a memory-type non-linearity in which the material
* For the theory of magnetization based on atomic structure of materials a suitable book on material science may be
consulted.
26
Electric Machines
remembers its previous history. Further, it is observed that the hysteresis loop becomes wider for increasing
maximum flux densities. The dotted curve drawn through the positive and negative tips of the hysteresis
loops with increasing maximum flux densities is the normal magnetization curve and is obtainable in virgin
(unmagnetized) material by increasing the dc magnetization in either direction. The normal magnetization
curve exhibits the saturation phenomenon as discussed in Sec. 2.2.
B (T)
B2, max
b
a
B1, max
Normal magnetization
curve
Residual flux
density
– H1, max
– H2, max
H1, max H2, max
H (AT/m)
Hysteresis loops
B1, max
B2, max
Fig. 2.13 Typical B-H curve and hysteresis loops
It is easily seen that because of hysteresis and saturation, the magnetic characteristic of a given material
cannot be described by a few overall parameters but must be expressed in the form of a set of curves. It
will be shown in Sec. 2.6 that the area of the hysteresis loop is the energy loss (it appears in the form of
heat energy) per unit volume in one cycle of magnetization. This loss depends upon the quality of material
and the maximum flux density at which the material is operated. Hysteresis loop as such is of little use in
engineering applications except in illustrating the waveform of the exciting current. It is the normal (dc)
magnetization curve which is of direct application in magnetic circuit calculations and design. In short form
it will be referred as the magnetization curve. The hysteresis loss represented by the loop area is usually
lumped with the eddy-current loss (Sec. 2.6) and the two together are known as the core (or iron) loss which
is parameterized by material thickness and frequency and is expressed as loss per unit volume (specific loss).
Sheet Steels
Nearly all transformers and certain parts of electric machines use silicon steel in the form of sheets, thinly
insulated on each side. Silicon is added to steel to increase its resistivity thereby reducing the eddy-current
loss. Building the core out of thin sheets (laminations) greatly aids in reduction of the eddy-current loss (see
Sec. 2.6).
The crystalline structure of silicon steel is body centred cubic—an atom at each corner of the cube and an
atom at the cube centre. The cubic structure presents considerable ease of magnetization (high mr) along the
cube edge as compared to the diagonal on the cube side and the diagonal through the cube body which is most
difficult to magnetize. Therefore, if silicon steel crystals are aligned so that the cube edges lie parallel to the
Magnetic Circuits and Induction
27
direction of magnetization, the material has a much higher
relative permeability and can be operated at much higher
flux densities with moderate exciting currents and also
has superior core-loss qualities. This crystal arrangement
is illustrated in Fig. 2.14. The crystal arrangement is
practically achieved by cold-rolling steel sheets. The
material is then termed as cold-rolled grain-oriented steel
(crgos) and is invariably employed in electromechanical
devices. Annealing of silicon steel sheets further helps in
proper alignment of crystals. The saturation flux density
Fig. 2.14 Crystal arrangement in grain-oriented
and electric resistivity are of course independent of the
steel
grain orientation.
Magnetostriction
When ferromagnetic materials are subjected to magnetizing mmf, these undergo small changes in dimensions.
The lengthwise change is of the order of 10–5 m and is accompanied by transverse changes of the opposite sign.
These changes are caused by magnetostriction. Their nature is hysteric with consequent dissipation of energy
when magnetization is alternating. Further, there are associated mechanical stresses which produce noise
in audible bandwidth which can be a nuisance for high flux densities employed in transformers in modern
practice. Magnetostriction noise may, therefore, be the subject of limiting specifications in transformers.
Magnetization Curves
Typical magnetization curves of transformer steel are given in Fig. 2.15 and a comparative set of magnetization
curves are given in Fig. 2.16.
2.0
crgos
1.8
B (T)
1.6
1.4
1.2
1.0
0.05
Fig. 2.15
2.4
0.1
0.2
0.3 0.5
H (kAT/m)
1
2
3
5
Magnetization curves: transformer steel
MAGNETICALLY INDUCED EMF AND FORCE
Faraday’s law of induction, which is the integral form of the fourth Maxwell’s equation, is given as
∂B ◊ds
E ◊ dl = ∂t
Ú
Ú
(2.15)
28
Electric Machines
2.0
crgos
Flux density B (T)
1.6
ms
1.2
cs
0.8
ci
0.4
0
Fig. 2.16
0
1
2
3
4
5
6
Magnetic fields strength H (kAT/m)
7
8
Magnetization curves: comparative [Cold-rolled grain-oriented steel (crgos), mild steel, cast steel, cast iron]
The integrated form of Eq. (2.15) for a coil of N turns is
df
dl
= dt
dt
l = Nf = flux linkages of the coil (Wb-Turns)
e= -N
where
(2.16)
The positive direction of current in the coil is that direction which establishes positive flux and flux linkages.
The negative sign in Eq. (2.16) means that the induced emf owing to an increase in l is in opposite direction
to that of positive current. If this fact is separately remembered (it is known as Lenz’s law), Eq. (2.16) may
be written
e= N
dl
df
=
dt
dt
(2.17)
with the sign of the emf determined by Lenz’s law.
Change in flux linkages of a coil may occur in three ways:
(i) The coil remains stationary with respect to flux, but the flux through it changes with time. The emf
induced is known as statically induced emf.
(ii) Flux density distribution remains constant and stationary but the coil moves relative to it. The emf
induced is known as dynamically induced (or motional) emf.
(iii) Both changes (i) and (ii) may occur simultaneously, i.e. the coil moves through time-varying flux. Both
statically and dynamically induced emfs are then present in the coil.
The dynamically induced emf (case (ii) above) in a conductor of length l placed at 90° to a magnetic field
of flux density B and cutting across it at speed v is given by
e = v ¥ B l = Blv sin q
(2.18a)
where q is the angle between the direction of flux density and conductor velocity, and l the conductor along
which the flux density is assumed uniform. In electric machines q = 90°, so that
e = Blv
(2.18b)
Magnetic Circuits and Induction
29
This is known as the flux-cutting rule and the direction of emf is given by v ¥ B or by the well-known
Fleming’s right-hand rule*.
Inductance
Equation (2. 17) may be written as
df
d f di
di
= N
= L
(2.19a)
dt
di dt
dt
dl
df
H
(2.19b)
where
L= N
=
di
di
is the self-inductance of the circuit. For a magnetic circuit having a linear B-H relationship (constant
permeability of material) or with a dominating air-gap, the inductance L is a constant, independent of current
and depends only on the geometry of circuit elements and permeability of the medium. In this case Eq. (2.19)
can also be expressed as
e= N
l
i
The inductance can be written in terms of field quantities as
L=
L=
N2
N 2 BA
A
=
= N 2P
= N 2m
R
Hl
l
(2.20)
H
(2.21)
Thus self-inductance is proportional to N 2.
The inductance concept is easily extendable to the mutual inductance of two coils sharing a common
magnetic circuit. Thus,
l12
¸
HÔ
i2
Ô
˝
l 21
M 21 =
HÔ
Ô˛
i1
M12 =
where
(2.22)
l12 = flux linkages of coil 1 due to current in coil 2
l21 = flux linkages of coil 2 due to current in coil 1
For a bilateral* magnetic circuit,
M = M12 = M21
It can also be shown that for tight coupling, i.e. all the flux linking both the coils (no leakage)
M=
L1 L2
(2.23a)
In general,
where
From Eq. (2.20)
M = k L1 L2
k = coupling coefficient (which can be at most unity)
l = Li
(2.23b)
* Extend the thumb, first and second fingers of the right hand so that they are mutually perpendicular to each other.
If the thumb represents the direction of v (conductor with respect to B) and the first finger the direction of B, then
the second finger represents the direction of emf along l.
30
Electric Machines
In static magnetic configuration, L is fixed independent of time so that the induced emf is given by
Eq. (2.19a). In rotating devices both L and i vary with time giving the induced emf
di
e=L +
dt
Statically
induced
emf
dL
i
dt
(2.24)
Dynamically
induced
emf
Force
Force of electromagnetic origin is given by the Lorentz force equation
dF = Idl ¥ B
(2.25)
where I is the current flowing in the differential conductor of length dl. Integrating over the conductor length
along which B is assumed uniform, the total force is obtained as
F = Il ¥ B = BIl sin q aF N
(2.26a)
where q is the angle between the direction of conductor and the magnetic field and aF is unit vector in the
direction defined by the cross product. For q = 90° which is used in most machine configurations, Eq. (2.26a)
reduces to
F = BIl
N
(2.26b)
The direction of force being given by Il ¥ B or by the well-known Fleming’s lefthand rule*. It immediately
follows from Eq. (2.26b) that B can be imagined to have unit of N/Am.
EXAMPLE 2.7 For the magnetic circuit of Fig. 2.17 find the self and mutual inductances between the two
coils. Core permeability = 1600.
1 cm
6 cm
3 cm
R2
N1 = 500
turms
R0
4 cm
2 cm
N2 = 1000
turms
R1
Thickness
2 cm
Fig. 2.17
* Extend the thumb, first and second fingers of the left hand so that they are mutually perpendicular to each other.
If the first finger represents the direction of B and the second finger the direction of I, then the thumb points in the
direction of force on the conductor.
Magnetic Circuits and Induction
31
SOLUTION
l1 = (6 + 0.5 + 1) ¥ 2 + (4 + 2) = 21 cm
l2 = (3 + 0.5 + 1) ¥ 2 + (4 + 2) = 15 cm
l0 = 4 + 2 = 6 cm
R1 =
R2 =
R0 =
21 ¥ 10- 2
4 p ¥ 10- 7 ¥ 1600 ¥ 2 ¥ 2 ¥ 10 - 4
15 ¥ 10- 2
4 p ¥ 10- 7 ¥ 1600 ¥ 2 ¥ 2 ¥ 10 - 4
6 ¥ 10- 2
4 p ¥ 10
-7
¥ 1600 ¥ 1 ¥ 2 ¥ 10 - 4
= 0.261 ¥ 106
= 0.187 ¥ 106
= 0.149 ¥ 106
(i) Coil 1 excited with 1A
R = R1 + R0 || R2
= 0.261 + 0.1871 || 0.149 = 0.344 ¥ 106
f1 = (500 ¥ 1)/(0.344 ¥ 106) = 1.453 mWb
f21 = f2 = 1.453 ¥ 0.149/(0.149 + 0.187) = 0.64 mWb
L11 = N1fl = 500 ¥ 1.453 ¥ 10–3 = 0.7265 H
M21 = N2f21 = 1000 ¥ 0.649 ¥ 10–3 = 0.64 H
(ii) Coil 2 excited with 1 A
R = R2 + (R0R1)/R0 + R1)
= [0.187 + (0.149 ¥ 0.281)/(0.149 + 0.281)] ¥ 106
= 0.284 ¥ 106
f2 = (1000 ¥ 1)/(0.284 ¥ 106) = 3.52 mWb
L22 = N2f2 =1000 ¥ 3.52 ¥ 10–3 = 3.52 H
M12 = M21 (bilateral) = 0.65 H
2.5 AC OPERATION OF MAGNETIC CIRCUITS
The magnetic circuits of transformers, ac machines and several other electromagnetic devices are excited
from ac rather than dc sources. With ac operation, inductance is effective even in steady-state operation.
Often, the flux is determined by the impressed voltage and frequency, and the magnetization current has to
adjust itself in accordance with the flux so that B-H relationship is satisfied.
Except when linearity is desirable, economic utilization of material demands that working flux density
should lie in the nonlinear zone (but not in the region of deep saturation). Exact and accurate analysis,
therefore, cannot be predicted on the basis of constant inductance. Still circuit models (equivalent circuits)
with constant parameters are often used. It will be seen in later chapters that these not only provide simplified
approach but at the same time yield the desired accuracy for engineering applications.
Consider the N-turn iron-core coil of Fig. 2.1. Complete linearity of the magnetic circuit will be assumed.
Magnetic flux f is produced by the exciting current i. Let the current and so the flux vary sinusoidally with
time. Then
(2.27)
f = fmax sin w t
32
Electric Machines
fmax = maximum value of flux in core (Wb)
where
w = 2pf, where f is frequency in Hz
The induced emf in the coil as per Faraday’s law (Eq. (2.17)) is
e= N
df
= w Nfmax cos wt V
dt
(2.28)
and its rms value is
E=
2p
2
f Nfmax = 4.44 f Nfmax
(2.29)
E = 444 f NAc Bmax
(2.30)
where Bmax is the maximum value of the flux density and Ac is the core’s area of cross-section.
The polarity of the emf must, in accordance with Lenz’s law, oppose the flux change and, therefore, is
as shown in Fig. 2.1 when the flux is increasing. Since the current produces the flux instantaneously and in
proportion to it (quasi-static field), they are in phase. From Eqs. (2.27) and (2.28), it is found that the induced
emf leads the flux (hence the current) by 90°. The induced emf and coil resistance drop oppose the impressed
voltage. However, resistance drop in many ac electromagnetic devices is quite small and may be neglected
to a close approximation.
The electric power input into the magnetic circuit of Fig. 2.1 through the coil terminals is
dl
dt
The electric energy input which gets stored in the magnetic field* in the time interval tl to t2 is
p = ie = i
Wf =
Ú
t2
t1
p dt =
Ú
l2
l1
idl
(2.31)
(2.32)
where Wf = increase in field energy as the coil flux linkages change from l1 to l2. In field quantities
Wf =
Ú
H c lc ˆ
ÁË N ˜¯ (Ac N )dBc = Aclc
B2 Ê
B1
Ú
B2
B1
H c dBc
(2.33)
wherein Aclc is the volume of the core. Thus the energy density in the field is given by
wf (density) =
EXAMPLE 2.8
(a)
(b)
(c)
(d)
Ú
B2
B1
H c dBc
J/m3
(2.34)
For the magnetic circuit of Example 2.1 and Fig. 2.4(a), find the following:
Induced emf e for Bc = 1.2 sin 314t T,
reluctance Rc and Rg
coil inductance, L and
magnetic field energy at Bc = 1.2 T
* It will be seen in Chapter 4 that a part of this energy is converted to mechanical form if mechanical motion is
permitted between parts of the magnetic system.
Magnetic Circuits and Induction
33
SOLUTION
(a) In Example 2.1 the value of l was found as 1.152 Wb-T for Bc = 1.2 T. Therefore, for sinusoidai variation of Bc,
l = 1.152 sin 314t Wb-T
The emf is
dl
=361.7cos 314t V
dt
lc
0.4
=
Rc =
-7
m0 m r Ac
4 p ¥ 10 ¥ 6000 ¥ 16 ¥ 10- 4
e=
(b)
= 3.317 ¥ 104
Rg =
lg
m0 Ag
=
6 ¥ 10- 4
4 p ¥ 10- 7 ¥ 16 ¥ 10- 4
= 29.856 ¥ 104
(c) From Example 2.1
i = 1.06 A
l
1.152
=
=1.09 H
L=
i
1.06
\
It can also be found by using Eq. (2.21). Thus
L = N 2P =
N2
N2
(600) 2
=
=
R
Rc + Rg
(3.316 + 29.84) ¥ 104
= 1.08 H
(d) The energy stored in the magnetic field is from Eq. (2.32)
Wf =
=
Ú
l
0
id l =
Ú
1 (1.152)
¥
2
1.08
l
0
1 l2
l
dl =
2 L
L
2
= 0.6144 J
When a magnetic material undergoes cyclic magnetization, two kinds of power losses occur in it—hysteresis
and eddy-current losses—which together are known as core-loss. The core-loss is important in determining
heating, temperature rise, rating and efficiency of
B
transformers, machines and other ac run magnetic
Bm
g
b
devices.
Hysteresis Loss
c
Figure 2.18 shows a typical hysteresis loop of a
ferromagnetic material. As the mmf is increased from
zero to its maximum value, the energy stored in the field
per unit volume of material is
Ú
Bb = Bm
-Bf
HdB = area ofabgo
dw1= HdB
d
0
a
Hm
Bf
e
Fig. 2.18
Hysteresis loss
H
34
Electric Machines
As H is now reduced to zero, dB being negative, the energy is given out by the magnetic field (from the
exciting coil back to the voltage source) and has a value
Ú
Bc
Bb = Bm
HdB = area cbg
The net energy unrecovered in the process is area ofabco which is lost irretrievably in the form of heat and
is called the hysteresis loss. The total hysteresis loss in one cycle is easily seen to be the area of the complete
loop (abcdefa) and let it be indicated as wh (hysteresis loss/unit volume). Then hysteresis loss in volume V of
material when operated at f Hz is
Ph = whVf W
(2.35)
In order to avoid the need for computation of the loop area, Steinmetz gave an empirical formula for
computation of the hystersis loss based on experimental studies according to which
Ph = kh f B nm W/m3
(2.36)
where kh is a characteristic constant of the core material, Bm is the maximum flux density and n, called the
Steinmetz exponent, may vary from 1.5 to 2.5 depending upon the material and is often taken as 1.6.
Eddy-current Loss
When a magnetic core carries a time-varying flux, voltages are induced in all possible paths enclosing the flux.
The result is the production of circulating currents in the core (all magnetic materials are conductors). These
currents are known as eddy-currents and have power loss (i 2R) associated with them called eddy-current loss.
This loss, of course, depends upon the resistivity of the material and lengths of the paths of circulating currents
for a given cross-section. Higher resistivity and longer paths increase the effective resistance offered by the
material to induced voltages resulting in reduction of eddy-current loss. High resistivity is achieved by adding
silicon to steel and hence silicon steel is used for cores conducting alternating flux. Dividing up the material
into thin laminations along the flow of flux, with each lamination lightly insulated (varnish is generally used)
from the adjoining ones, increases the path length of the circulating currents with consequent reduction in
eddy-current loss. The loss in fact can be shown to depend upon the square of lamination thickness. The
lamination thickness usually varies from 0.3 to 5 mm for electromagnetic devices used in power systems and
from about 0.01 to 0.5 mm for devices used in electronic applications where low core-loss is desired.
The eddy-current loss can be expressed by the empirical formula
pe = ke f 2B2 W/m3
(2.37)
ke = K¢e d 2/r
(2.38)
wherein
d being the thickness of lamination and r the resistivity of material.
It is only an academic exercise to split the core-loss into its two components. The core loss in fact
arises from two types of flux variations: (i) flux that has a fixed axis and varies sinusoidally with time as
in transformers (this is the type visualized in the above discussion), (ii) flux density is constant but the flux
axis rotates. Actually in ac machines as well as in armature of dc machines the flux variation comprises both
these types occurring simultaneously. The core-loss is measured experimentally on material specimen and
presented graphically. Typical values of the specific core-loss (W/kg of material) are displayed in Figs 2.19
(a) and (b) for cold-rolled grain-oriented (crgos) steel. It is easy to see from these figures that for reasons
mentioned above specific core loss is much higher in machines than in transformers.
5
50
4
40
Specific loss (W/kg)
Specific loss (W/kg)
Magnetic Circuits and Induction
3
0.35 mm
2
crgos
1
0.8
1.0
1.2
1.4
B Flux density (T )
(a)
20
0.35 mm
10
1.8
1.6
0.5 mm 0.4
mm
30
0
0
35
0.8
2.0
1.2
1.6
Flux density (T )
(b)
2.4
Fig. 2.19 Core-loss at 50 Hz: (a) transformers, (b) machines
EXAMPLE 2.9 The total core loss of a specimen of silicon steel is found to be 1500 W at 50 Hz. Keeping
the flux density constant the loss becomes 3000 W when the frequency is raised to 75 Hz. Calculate separately
the hysteresis and eddy current loss at each of those frequencies.
SOLUTION
From Eqs. (2.36) and (2.37) for constant flux density, total core loss can be expressed as
P = Af + Bf 2
1500/50 = A + 50 B
3000/75 = A + 75 B
or
or
or
P/f = A + Bf
30 = A + 50 B
40 = A + 75 B
(i)
(ii)
Solving Eqs. (i) and (ii), we get A = 10, B = 2/5
Therefore
At 50 Hz
P = 10f + 2/5 f 2 = Ph + Pe
Ph = 10 ¥ 50 = 500 W
(iii)
Pe = 2/5 ¥ 2500 = 1000 W
At 75 Hz
Ph = l0 ¥ 75 = 750 W
Pe = 2/5 ¥ (75)2 = 2250 W
2.7
PERMANENT MAGNETS
The permanent magnet is an important excitation source (life long) commonly employed for imparting energy
to magnetic circuits used in rotating machines and other types of electromechanical devices. There are three
classes of permanent magnet materials (or hard magnetic materials) used for permanent magnet dc (PMDC)
motors: Alnicos, ceramics (ferrites) and rare-earth materials. Alnico magnets are used in motors up to
200 kW, while ceramic magnets are most economical in fractional kW motors. The rare-earth magnetic
materials are very costly, but are the most economic choice in very small motors. Latest addition is
neodymium-iron boron (Nd FeB). At room temperature, it has the highest energy product (to be explained
later in this section) of all commonly available magnets. The high permeance and coercivity allow marked
reductions in motor frame size for the same output compared to motors using ferrite (ceramic) magnets. For
very high temperature applications Alnico or rare-earth cobalt magnets must be used.
36
Electric Machines
Two important qualities of a permanent magnet (PM) are defined below with reference to the second
quadrant of its hysteresis loop.
Permanent Magnetization or Residual Flux Density (Br)
It is the flux density trapped in closed magnetic structure if the applied mmf (and therefore the magnetic field
intensity, H) were reduced to zero.
Coercivity
It is the measure of mmf (or H) which, when applied to the magnetic circuit, would reduce its flux density to
zero, i.e. it would demagnetize the material. Its value is negative and in units of kA/m.
The second quadrant of the hysteresis loops for Alnico 5 and M-5 steel are shown respectively in
Figs. 2.20(a) and (b). Their residual flux densities and coercivities are given below:
: Br ª 1.25 T, Hc ª –50 kA/m
: Br ª 1.4 T, Hc ª –6 kA/m
Alnico 5
M-5 steel
It is therefore observed that while Br of M-5 steel is higher than that of Alnico 5 but the latter (Alnico 5)
has a far greater coercivity. As we shall see below that materials with high coercivity qualify as PM materials.
An important measure of the capability of permanent magnet is known as its maximum energy product.
This corresponds to the largest BH product, (BH)max, which is a point on the second quadrant of the hysteresis
loop; see Fig. 2.20(a). It has the dimensions of energy density (J/m3) and it can be shown that operation of
a given PM material at this point will result in the minimum volume of material required to produce a given
flux density in the air gap.
kJ/m3
B (T )
B(T )
1.5
1.5
50
Point of
maximum
energy
product
40
Br
30
b
.
a
Br
1.0
1.0
0.5
0.5
.
.
.
Hc
H(kA/m)
–50
– 40
–30
(a) Alnico 5
–20
–10
Hc
0
–5
H(A/m) –10
(b) M-5 electrical steel
0
Fig. 2.20 Second quadrant of hysteresis loop for (a) Alnico 5 and (b) M-5 electrical steel
Magnetic Circuits and Induction
37
EXAMPLE 2.10 A magnetic circuit (Fig. 2.21a) consists of a core of very high permeability, an air-gap
length of lg = 0.4 cm and a section of permanent magnet (made of Alnico 5) of length lm = 2.4 cm. Assume
m of core = .
Calculate the flux density Bg in the air-gap. Given: Am = 4 mm2.
m
Area Am
Magnet
lm
Ag
lg
Fig. 2.21(a) A Magnetic circuit with a PM
SOLUTION
mcore =
fiHcore = 0
From Ampere’s circuital law
Hm lm + Hg lg = 0 = F
or
Êl ˆ
Hg = - Á m ˜ Hm
Ë lg ¯
(2.39)
(2.40)
where Hg and Hm are the magnetic field intensities in the air-gap and the PM respectively. Thus the existence of an air-gap
is equivalent to the application of a negative field to the PM material.
As the flux must be continuous around the path
f = Bm Am = Bg Ag
(2.41)
Also
Bg = m0 Hg
We obtain from Eqs. (2.40) and (2.41)
Bm = –m0
Ê Ag ˆ Ê lm ˆ
H
ÁË A ˜¯ Á l ˜ m
m Ë g¯
(2.42)
Substituting values we get
Bm = –6m0 Hm = –7.54 ¥ 10–6 Hm
(2.43)
This is a straight line (also called load line) shown in Fig. 2.20(a), where its intersection with the demagnetization
curve at point ‘a’ gives the solution for Bm.
Thus
Bg = Bm = 0.33 T
Note: If we repeat the above problem for M-5 electrical steel, it is easy to find the answer since the load line is the same
as given by Eq. (2.43). It can be shown that Bm = 4 ¥ 10–5 T. This is much less than the value of Bm for Alnico 5.
38 Electric Machines
From Eq. (2.40) we can get the expression for Bg as
Êl ˆ
Bg = m0 Hg = – m0 Á m ˜ Hm
Ë lg ¯
(2.44)
From Eqs. (2.41) and (2.44) we get
Êl A ˆ
B 2g = m0 Á m m ˜ (–Hm Bm)
Ë l g Ag ¯
Ê
ˆ
= m0 Á Volm ˜ (–Hm Bm)
Vol
Ë
g¯
Volm =
Bg Vol g
m0 (- H m Bm )
(2.45)
; a positive value as Hm is negative (Fig. 2.20)
(2.46)
Thus, to produce a flux density Bg in an air-gap of volume Volg, minimum volume of magnet material would be
required if the material is operated in the state represented by the maximum value of the product Bm Hm.
From Eq. (2.46) it may appear that one can get an arbitrarily large air-gap flux density just by reducing the air-gap
volume. However, in practice this cannot be achieved because the on increasing flux density in the magnetic circuit
beyond a given point, the magnetic core gets saturated and the assumption of infinite core permeability becomes invalid.
It may be noted that in Fig. 2.20(a) a set of constant BH product curves (hyperbolas) is also plotted.
EXAMPLE 2.11 For the magnetic circuit of Fig. 2.21(a) if Ag = 3.0 cm2 find the minimum magnet volume
required to produce an air-gap flux density of 0.7 T.
SOLUTION The smallest magnet volume will be obtained with the magnet operating at point as shown in Fig. 2.20(a)
which corresponds to Bm = 1.0 T and Hm = –40 kA/m.
From Eq. (2.41)
Am = Bg Ag/Bm = (0.7 ¥ 3)/1 = 2.1 cm2
From Eq. (2.39)
lm = –
= -
H g lg
Hm
=–
Bg l g
m0 H m
0.7 ¥ 0.4
4 p ¥ 10- 7 ¥ (- 40 ¥ 103 )
= 5.57 cm
Minimum magnet volume = 2.1 ¥ 5.57 = 11.7 cm3
Examples 2.10 and 2.11 depict the operation of hard (PM) magnetic materials.
However, the situation is more complex as discussed in the next section.
Now consider the case when an exciting coil is placed on the core of the
permanent magnet. Circuit of Fig. 2.21(a) with Ni ampere-turns. The circuit is
shown in Fig. 2.21(b).
From the magnetic circuit
F = Ni = Hmlm + Hg lg
Bm Am = Bg Ag = m0 Hg Ag
Ag
lg
lm
N
i
(2.47)
Exciting coil
As flux lines are continuous and no leakage is assumed.
So,
m
Am
(2.48)
Fig. 2.21(b)
Magnetic Circuits and Induction
39
Equation (2.47) can be written as follows:
Ni = Hm lm +
Bm Am
m0 Ag
(2.49)
This can be reorganized as follows:
Ê Ag ˆ 1
Ê A ˆÊl ˆ
Bm = –m0 Á g ˜ Á m ˜ Hm + m0 Á
(Ni)
Ë Am ˜¯ l g
Ë Am ¯ Ë l g ¯
(2.50)
This equation is the general form of the load line. The intersection of this line gives the operational point B, H. By
adjusting the current in the exciting coil, the permanent magnet can be brought to desired magnetization state. This is
accomplished as per steps below.
B-H (hysteresis) curve with desired value of Br (residual magnetization).
B-H curve, find Bmax and Hmax at the hysteresis loop tip.
imax. Usually the desired B-H curve may not be available. So from
Br we have to estimate Bmax and Hmax. This can be done by extrapolating the B-H curve from Br to about 4 times
Hc (into positive H-side).
While magnetizing a permanent magnet the exciting current is raised to imax and then reduced to zero gradually. The
exciting coil may then be removed.
EXAMPLE 2.12 Consider the magnetic circuit of Fig. 2.21(a). The permanent magnet material Alinco-5
is in demagnetized state. It is required to be magnetized to a reduced flux density Br = 1.25 T. Magnetic
circuit dimensions are: Am = Ag = 2.5 cm2, lm = 4 cm, lg = 0.2 cm. Excitation coil turns, N = 200.
SOLUTION
To find the first quadrant tip of the hysteresis loop, we assume
Hmax = 3.5 Hc
From Fig. 2.20(a)
So,
Hc = 50 (magnitude)
Hmax = 170 kA/m
Extrapolating B-H curve into first quadrant, we get
Bmax ª 2T
Substituting values in Eq. (2.50)
È
˘
Bmax = m0 Í- Ê 2 ˆ Ê 4 ˆ H max + Ê 2 ˆ 1 ¥ 200 i ˙
ÁË 2 ˜¯
ÁË 2 ˜¯ ÁË 0.2 ˜¯
-2
0.2 ¥ 10 ˙˚
ÍÎ
–7
m0 = 4p ¥ 10
\
Bmax = – 2.51 ¥ 10–5 Hmax + 12.57 ¥ 10–2 i
Substituting for Bmax and Hmax
2 = –2.51 ¥ 10–5 ¥ 170 ¥ 103 + 12.57 ¥ 102 i
(2.51)
Solving Eq. (2.51), we get
i = 49.86 A
Note: Equation (2.51) represents a straight line in B-H plane for a given i. Its intersection with B-H curve gives the state
of the permanent magnet at that value of exciting current.
40
Electric Machines
2.8
APPLICATION OF PERMANENT MAGNET MATERIALS
Typical values of the properties of four classes of PM materials are given in Table 2.1. These properties are:
residual flux density Br, coercive magnetizing force Hc, maximum energy stored (BH)max and resistivity.
Table 2.1
Material
Br
(T)
Hc
kA/m
BHmax
kJ/m3
r
mm
Barium or
Strontium ferrite
carbonate powders
Alcomax
Hycomax
Somarium-cobalt
0.39
200
30
High
1.25
0.8
0.75
60
100
600
45
35
130
500
500
60
Type
A:
Ceramic
B
C
D
Metallic
Metallic
Rare-earth
Type A material is cheap but heavy, and suitable for low-rated production-run motors. Types B and C
materials are hard and can be given simple shapes only. Type D material can be easily moulded and machined
and is used for most electric motors but is costly. Materials with higher coercivities are much less prone to
demagnetization. Figure 2.22 shows DC magnetization curves for a few commonly used materials.
B (T )
1.3
1.2
1.1
1.0
0.8
Aln
ico
0.9
lt
m
Co
ba
0.6
m
ar
iu
0.5
So
Ne
od
ym
ium
-ir
on
-b
or
on
0.7
0.4
0.3
Ce
ra
m
ic
0.2
H, kA/m –1000 –900 –800 –700 –600 –500 –400 –300 –200 –100
Fig. 2.22
0.1
0
–0
DC magnetization curves for some commonly used PM materials
Magnetic Circuits and Induction
The latest of the rare-earth magnetic materials is the
neodymium-iron-boron material. It has larger Br, Hc
and (BH)max than Somarium cobalt. It is cheaper and
has good mechanical properties and this is expected to
be used in a big way for PM applications.
Consider the magnetic circuit of Fig. 2.23 consisting
of a section of PM material in a core of highly permeable
soft magnetic material with an N-turn exciting winding.
Figure 2.24 shows that the PM is initially unmagnetised and current is applied to the exciting winding.
As core is of infinite permeability, the X-axis represents
both i and H.
Core m
PM
lm
41
i
N turns
Fig. 2.23
Practical magnetic circuit having a PM
B (T )
a
Bmax
Br
b
c
Recoil
line
B1
Minor loop
d
– H1
– i1
Fig. 2.24
0
i=
Hlm
N
Hmax H, kA/m
i,A
imax
Demagnetization curve with recoil line
As current i is raised to its maximum value (saturation), the BH trajectory rises from the origin to its
maximum value at point ‘a’. Now if the current is reduced to zero, the BH characteristic starts forming a
hysteresis loop meeting Y-axis at point b at which B = Br and H = 0. As the current is further decreased to
a negative value the BH curve continues to follow a hysteresis loop as shown in Fig. 2.24. For i = –i1, the
operating point is c. It may be noted that the same operating point c would be arrived (see Ex. 2.10) if the
material were to start at point b with zero excitation and air-gap length of lm (Ag/Am) (–m0 H1/B1) (Eq. (12.42))
were then inserted in the core.
If the current is further reduced, the trajectory would further trace the BH curve toward d as shown in
Fig. 2.24. However, if the current is reduced to zero, the trajectory does not normally retrace the loop toward
point b. Instead it starts to trace out a minor hysteresis loop, as shown in Fig. 2.24. As the current varies from
0 to i1. The minor hysteresis loop may usually be replaced with little error by a straight line called the recoil
line. This line has a slope called the recoil permeability mrec, which is approximately the same as that of the
original BH curve at H = 0, i.e., at B = Br. In fact the recoil line is essentially tangent to the BH curve for
a large portion of the useful operating region for many materials such as Somarium Cobalt, Ceramic 7 etc.
having large values of coercivity.
42 Electric Machines
As long as the negative value of applied magnetic field intensity does not exceed H1, the magnet may be
regarded as reasonably permanent. If, however, a negative field intensity greater than H1 is applied, the flux
density will be reduced to a value lower than B1 and a new and lower minor loop will be created with a new
recoil line and recoil permeability.
The demagnetization effects of negative excitation which have been discussed above are equivalent to
those of an air-gap in the magnetic circuit.
Thus, we see that these materials produce enough magnetic flux even in magnetic circuits with air-gaps.
With proper design they can be operated stable even when subjected to a wide range of destabilizing forces
and mmf’s, Permanent magnets are increasingly finding greater applications in many small devices such
as loud speakers, ac and dc motors, microphones, analog electric meters, driving, windshield wipers, radio
antennas, airconditioners, etc.
The role of the electromagnetic system is to establish and control electromagnetic fields for carrying
out conversion of energy, its processing and transfer.
The Ampere’s law,
J ◊ ds = H ◊ dl
Ú
where
Ú
S
=
conduction
current
density
J
H = magnetic field intensity
S = the surface enclosed by the closed path of length l
ds = differential surface
dl = differential length
In all practical circuits most of the flux is confined to the intended path by use of magnetic cores but
a small amount of flux always leaks through the surrounding air. The stray flux is called the leakage
flux.
The effect of the fringing field is to increase the effective cross-sectional area Ag of the air-gap.
Magnetic circuit law
F
f=
= FP = flux (Wb)
R
F = mmf (AT)
lc
(AT/Wb)
mc Ac
P = permeance = 1/R
Electrical analog of magnetic circut
F ~ E, R ~ R, f ~ i
R = reluctance =
Hysleresis loss, ph = kh f B nm W/m3
n = Stenmetz exponent, 1.5 to 2.5 upto taken as 1.6
Eddy current loss, pe = ke f 2B 2m W/m3
Magnetic Circuits and Induction
43
Magnetic cores are made up of thin, lightly insulated (coating of varnish) laminations to reduce power
loss in cores due to the eddy current phenomenon. As a result, the net cross-sectional area of the core
occupied by the magnetic material is less than its gross cross-section, their ratio (less than unity) being
known as stacking factor.
Super paramagnetic materials are made from powdered iron or other magnetic particles. These
materials are used in transformers for electronics and cores for inductors.
Magnetically induced EMF and FORCE is given by
e = BlV—Fleming’s right hand rule determines the direction of emf
F = BIl—Fleming’s left hand rule determines the direction of force
B = flux density (Wb/m2 or T)
V = conductor speed m/s relative to flux
l = conductor length (m)
I = conductor current (A)
Energy stored in magnetic field
Wf =
where
1 2 1 l2
LI =
2
2 L
L – self-inductance, (H)
I = current (A),
l = flux linkages (Wb-T)
Permanent Magnet – Br = residual flux
Coercerlity – H (negative) needed to reduce
B to zero
Note: Unless otherwise specified, neglect leakage and fringing.
2.1 A square loop of side 2d is placed with two of
its sides parallel to an infinitely long conductor
carrying current I. The centre line of the
square is at distance b from the conductor.
Determine the expression for the total flux
passing through the loop. What would be the
loop flux if the loop is placed such that the
conductor is normal to the plane of the loop.
Does the loop flux in this case depend upon
the relative location of the loop with respect
to the conductor?
2.2 For the magnetic circuit of Fig. P.2.2, find the
flux density and flux in each of the outer limbs
and the central limbs. Assume the relative
permeability of iron of the core to be (a) ,
(b) 4500.
Core thickness
5 cm
A
1000
turns
0.5 A
2 mm
40cm
1 mm
5cm
30 cm
10 cm
B
Fig. P 2.2
30 cm
5cm
44
Electric Machines
2.3 For the magnetic circuit shown in Fig. P.2.3,
calculate the exciting current required to
establish a flux of 2 mWb in the air-gap. Take
fringing into account empirically. Use the B-H
curve of Fig. 2.15.
in the core?
A
C
20 cm
B
Ac = 5 ¥ 4 cm2
0.1 cm
200 turns
15cm
Fig. P 2.5
20 cm
Fig. P 2.3
2.4 A steel ring has a mean diameter of 20 cm, a
cross-section of 25 cm2 and a radial air-gap
of 0.8 mm cut across it. When excited by a
current of 1 A through a coil of 1000 turns
wound on the ring core, it produces an airgap flux of 1 m Wb. Neglecting leakage and
fringing, calculate (a) relative permeability of
steel, and (b) total reluctance of the magnetic
circuit.
2.5 The core made of cold rolled silicon steel
(B-H curve of Fig. 2.15) is shown in Fig. P.2.5.
It has a uniform cross-section (net iron) of
5.9 cm2 and a mean length of 30 cm. Coils A,
B and C carry 0.4, 0.8 and 1 A respectively in
the directions shown. Coils A and B have 250
and 500 turns respectively. How many turns
must coil C have to establish a flux of 1 mWb
2.6 In the magnetic circuit shown in Fig. P.2.6, the
coil F1 is supplying 4000 AT in the direction
indicated. Find the AT of coil F2 and current
direction to produce air-gap flux of 4 mWb
from top to bottom. The relative permeability
of iron may be taken as 2500.
2.7 For the magnetic circuit shown in Fig. P.2.7,
the air-gap flux is 0.24 mWb and the number
of turns of the coil wound on the central limb
is 1000.
Calculate (a) the flux in the central limb, (b)
the current required. The magnetization curve
of the core is as follows:
H(AT/m)
B(T)
200
800
0.4
1.2
400
1060
0.8
1.3
F2
F1
2 mm
50
cm
Fig. P 2.6
50 cm
20 cm
Area of crosssection 40 cm2
throughout
500
1400
1.0
1.4
600
1.1
Magnetic Circuits and Induction
10 cm
45
2 cm
10 cm
15 cm
Air-gap, 1 mm
15 cm
2cm
2 cm
2cm
4 cm
Core thickness
3 cm uniform
2 cm
Fig. P 2.7
2.8 The magnetic circuit shown in Fig. P.2.8 has
a coil of 500 turns wound on the central limb
which has an air-gap of 1 mm. The magnetic
path from A to B via each outer limb is
100 cm and via the central limb 25 cm (airgap length excluded). The cross-sectional area
of the central limb is 5 cm ¥ 3 cm and that
each outer limb is 2.5 cm ¥ 3 cm. A current
of 0.5 A in the coil produces an air-gap flux
of 0.35 mWb. Find the relative permeability
of the medium.
H(AT/m) 0
200
1000 1200
B(T)
0
0.11
1.0 1.18
B
Fig. P 2.8
2.9 A cast steel ring has an external diameter of
32 cm and a square cross-section of 4 cm
side. Inside and across the ring a cast steel
bar 24 ¥ 4 ¥ 2 cm is fitted, the butt-joints
being equivalent to a total air-gap of 1 mm.
Calculate the ampere-turns required on half of
the ring to produce a flux density of 1 T in the
other half. Given:
600
1600
0.6
1.32
800
0.8
2.10 In Prob. 2.2 the B-H curve of the core material
is characterized by the data given below. Find
now the flux and flux densities in the three
limbs of the core.
H(AT/m)
B(T)
A
400
1400
0.32
1.27
50
250
0. 14
1.22
100
300
0.36
1.32
150
350
0.66
1.39
200
1.00
Hint: This problem can be solved by the
graphical-cum-iterative technique.
2.11 A ring of magnetic material has a rectangular
cross-section. The inner diameter of the
ring is 20 cm and the outer diameter is
25 cm, its thickness being 2 cm. An air-gap of
1 mm length is cut across the ring. The ring
is wound with 500 turns and when carrying a
current of 3 A produces a flux density of 1.2 T
in the air-gap. Find (a) magnetic field intensity
in the magnetic material and in the air-gap. (b)
relative permeability of the magnetic material,
and (c) total reluctance of the magnetic circuit
and component values.
2.12 For the magnetic ring of Prob. 2.11, the
exciting current is again 3 A. Find the
following:
46 Electric Machines
(a) Inductance of the coil,
(b) energy stored in the magnetic material
and in the air-gap, and
(c) rms emf induced in the coil when it
carries alternating current of 3 sin 314t.
density of 1.4 T in the core in the indicated
direction?
f
Thickness 5 cm
10 cm
2.13 Assume that the core of the magnetic circuit
of Fig. P.2.3 has mr = 2500.
(a) Calculate the energy stored in the core
and in the air-gap for an excitation
current of 5 A.
What will be these values if mr = ?
(b) What will be the excitation current to
produce a sinusoidally varying flux of
0.5 sin 314t mWb in the air-gap?
(c) Calculate the inductance of the coil.
What will be the inductance if mr = ?
2.14 The magnetic circuit of Fig. P.2.14 has a
magnetic core of relative permeability 1600
and is wound with a coil of 1500 turns
excited with sinusoidal ac voltage, as shown.
Calculate the maximum flux density of the
core and the peak value of the exciting current.
What is the peak value of the energy stored in
the magnetic system and what percentage of it
resides in the air-gap?
2.15 The material of the core of Fig. P.2.15, wound
with two coils as shown, is sheet steel (B-H
curve of Fig. 2.15). Coil 2 carries a current
2 A in the direction shown. What current (with
direction) should coil 1 carry to establish a flux
400 turns
1
10 cm
800 turns
20 cm
2
25 cm
10 cm
10 cm
Fig. P 2.15
2.16 The flux in a magnetic core is alternating
sinusoidally at a frequency of 600 Hz. The
maximum flux density is 2 T and the eddycurrent loss is 15 W. Find the eddy-current
loss in the core if the frequency is raised to
800 Hz and the maximum flux density is
reduced to 1.5 T.
2.17 The core-loss (hysteresis + eddy-current loss)
for a given specimen of magnetic material is
found to be 2000 W at 50 Hz. Keeping the flux
density constant, the frequency of the supply
is raised to 75 Hz resulting in core-loss of
3200 W. Compute separately hysteresis and
eddy-current losses at both the frequencies.
Hint: PL = Pc + Pk = ke f 2B 2mV + kh f B nmV; V =
fixed core volume
f
i
+
200 V
f = 50 Hz
0.15 mm
E
–
20 cm
Fig. P 2.14
Cross- sectional
area = 5 cm2
Magnetic Circuits and Induction
Since Bm remains constant
PL = k¢e f + k h¢ f
PL / f = k¢e f + k h¢
2
or
which gives a straight line from which k¢e and
k¢h can be determined.
2.18 A permanent magnet (PM) made of neodymium-iron boron alloy is placed in the magnetic
circuit of Fig. P.2.18. Given
Ag = 4 cm2, lg = 0.4 cm
It is desired to have air gap flux density Bg =
0.5 T. For optimum design (minimum volume
of PM) determine lm,
Note
47
Am = Ag.
2.19 In the PM circuit of Fig. P.2.18
Ag is reduced to 2 cm2,
Determine Bg and Bm.
2.20 The armature in the PM circuit Fig. 2.18 is
now taken out and its height reduced so that
when it is placed back in the circuit the air gap
length lg is now 0.5 cm. Determine Bg and Bm.
2.21 On the core of Fig. P.2.18 an exciting coil
is wound with 200 turns and is fed with an
exciting current of 1 A. Determine air-gap
flux density Bg. Note that direction of exciting
current is such that it aids magnetization.
Magnetic core
Am
m
Total air gap lg
lm
Armature
can be shifted
Ag values for
P 2.19
PM
Fig. P 2.18
1. State Ohm’s law for magnetic circuits.
2. Define magnetic reluctance.
3. Explain why a ferromagnetic material exhibits
its typical B-H behaviour.
4. Explain the practical use made of magnetic
saturation.
5. Explain the origin of magnetostriction noise
in ferromagnetic materials.
6. Advance a qualitative explanation for reduction of eddy current loss by using a core composed of silicon steel laminations.
7. What are the important qualities of a PM?
8. What is the phase angle between flux and induced emf in an ac excited coil wound on an
iron core?
9. Write the expression for inducted emf (rms) in
an ac excited coil wound in an iron core. Use
stantard symbols.
10. Write the expression for the self inductance of
a coil wound on an iron core.
48 Electric Machines
3
3.1
INTRODUCTION
A transformer is a static device comprising coils coupled through a magnetic
medium connecting two ports at different voltage levels (in general) in an electric
system allowing the interchange of electrical energy between the ports in either
direction via the magnetic field. The transformer is one of the most important
component of a variety of electrical circuits ranging from low-power, low-current
electronic and control circuits to ultra high-voltage power systems. Transformers
are built in an astonishing range of sizes from the tiny units used in communication systems to monsters
used in high-voltage transmission systems, weighing hundreds of tons. A circuit model and performance
analysis of transformers is necessary for understanding of many electronic and control systems and
almost all power systems. The transformer being an electromagnetic device, its analysis greatly aids in
understanding the operation of electromechanical energy conversion devices which also use magnetic field
but the interchange of energy is between electrical and mechanical ports.
The most important tasks performed by transformers are: (i) changing voltage and current levels in
electric power systems, (ii) matching source and load impedances for maximum power transfer in
electronic and control circuitry, and (iii) electrical isolation (isolating one circuit from another or isolating
dc while maintaining ac continuity between two circuits). Transformers are used extensively in ac power
systems because they make possible power generation at the most desirable and economical level (10–
20 kV), power transmission at an economical transmission voltage (as high as 400–1000 kV) and power
utilization at most convenient distribution voltages (230/400 V) for industrial, commercial and domestic
purposes but in industrial applications voltages may have to be as high as 3.3, 6.6 or 11 kV for large
motors. In communication and electronic systems where frequency ranges from audio to radio and video,
transformers are used for a wide variety of purposes. For instance input/output transformers (used to connect
the microphone to the first amplifying stage/to connect the last amplifying stage to the loudspeaker) and
interstage transformers are to be found in radio and television circuits. Indeed the transformer is a device
which plays an important and essential role in many facets of electrical engineering.
A transformer, in its simplest form, consists essentially of two insulated windings interlinked by a
common or mutual magnetic field established in a core of magnetic material. When one of the windings,
termed the primary, is connected to an alternating-voltage source, an alternating flux is produced in the
core with an amplitude depending on the primary voltage, frequency and number of turns. This mutual flux
links the other winding, called the secondary. A voltage is induced in this secondary of the same frequency
as the primary voltage but its magnitude depends on the number of secondary turns. When the number
of primary and secondary turns are properly proportioned, almost any desired voltage ratio, or ratio of
transformation can be achieved. The subscript “1” will be associated with the primary and “2” with the
Transformers
49
secondary. The reader should note that these are arbitrary terms and in no way affect the inherent properties
of a transformer.
If the secondary voltage is greater than the primary value, the transformer is called a step-up transformer;
if it is less, it is known as a step-down transformer; if primary and secondary voltages arc equal, the
transformer is said to have a one-to-one ratio. One-to-one transformers are used to electrically isolate
two parts of a circuit. Any transformer may be used as a step-up or step-down depending on the way it is
connected.
In order to ensure the largest and most effective magnetic linkage of the two windings, the core, which
supports them mechanically and conducts their mutual flux, is normally made of highly permeable iron
or steel alloy (cold-rolled, grain oriented sheet steel). Such a transformer is generally called an iron-core
transformer. Transformers operated from 25–400 Hz are invariably of iron-core construction. However,
in special cases, the magnetic circuit linking the windings may be made of nonmagnetic material, in
which case the transformer is referred to as an air-core transformer. The air-core transformer is of interest
mainly in radio devices and in certain types of measuring and testing instruments. An intermediate type,
exemplified by a type of induction coils and by small transformers used in speech circuits of telephone
systems, utilizes a straight core made of a bundle of iron wires on which the primary and secondary coils
are wound in layers.
3.2
TRANSFORMER CONSTRUCTION AND PRACTICAL CONSIDERATIONS
The type of construction adopted for transformers is intimately related to the purpose for which these are to
be used; winding voltage, current rating and operating
frequencies. The construction has to ensure efficient
removal of heat from the two seats of heat generation—
core and windings, so that the temperature rise is limited
to that allowed for the class of insulation employed.
Further, to prevent insulation deterioration, moisture
ingress to it must not be allowed. These two objectives
are simultaneously achieved in power transformers,
other than those in very small sizes, by immersing
the built-up transformer in a closed tank filled with
noninflammable insulating oil called transformer oil.
To facilitate natural oil circulation and to increase the
cooling surface exposed to the ambient, tubes or fins
are provided on the outside of tank walls. In large-size
transformers tubes may be forced-cooled by air. For still
larger installations the best cooling system appears to
be that in which the oil is circulated by pump from the
top of the transformer tank to a cooling plant, returning
when cold to the bottom of the tank. In small sizes
the transformers are directly placed in a protective
housing or are encased in hard rubber moulding and
are air-cooled. Figures 3.1. (a), (b) and (c) show the
constructional details of practical transformers.
Fig. 3.1 (a) Single-phase transformer core and
windings
50
Electric Machines
Power transformers are provided with a conservative through which the transformer breathes into the
atmosphere The conservative is a smaller-sized tank placed on top of the main tank. This arrangement ensures
that surface area of transformer oil exposed to atmosphere is limited so as to prevent fast oxidization and
consequent deterioration of insulating properties of the oil.
Fig. 3.1 (b) Three-phase transformer core and windings
The magnetic core of a transformer is made up of stacks of thin laminations (0.35 mm thickness) of coldrolled grain-oriented silicon steel sheets lightly insulated with varnish. This material allows the use of high
flux densities (1–1.5 T) and its low-loss properties together with laminated construction reduce the core-loss
to fairly low values. The laminations are punched out of sheets and the core is then built of these punching.
Before building the core, the punched laminations are annealed to relieve the mechanical stresses set in at
the edges by the punching process; stressed material has a higher core-loss. Pulse transformers and highfrequency electronic transformers often have cores made of soft ferrites.
The primary and secondary coils are wound on the core and are electrically insulated from each other
and from the core. Two types of cores are commonly employed in practice—core-type and shell-type. In
core-type construction shown in Fig. 3.2(a) the windings are wound around the two legs of a rectangular
magnetic core, while in shell-type construction of Fig. 3.2(b), the windings are wound on the central leg of
a three-legged core. Though most of the flux is confined to a high permeability core, some flux always leaks
through the core and embraces paths which partially lie in the air surrounding the core legs on which the coils
are wound. This flux which links one of the windings without linking the other, though small in magnitude,
Transformers
51
Conservator
Top core
clamp
L.V. winding
H.V. winding
Oil ducts
H.V. SIDE
L.V. SIDE
L.V. insulating
cylinder
H.V. insulating
cylinder
Tapping leads
to switches
Coil stack end
insulation
Bottom core clamps
Fig. 3.1 (c) Transformer showing constructional details
has a significant effect on the transformer behaviour. Leakage is reduced by bringing the two coils closer.
In a core-type transformer this is achieved by winding half low-voltage (LV) and half high-voltage (HV)
winding on each limb of the core as shown in Fig. 3.2(a). The LV winding is wound on the inside and HV on
outside to reduce the amount of insulation needed. Insulation between the core and the inner winding is then
stressed to low voltage. The two windings are arranged as concentric coils. In shell-type construction leakage
is reduced by subdividing each winding into subsections (wound as pancake coils) and interleaving LV and
HV windings as shown in Fig. 3.2(b).
52
Electric Machines
The core-type construction has a longer mean length of core and a shorter mean length of coil turn. This
type is better suited for EHV (extra high voltage) requirement since there is better scope for insulation. The
shell-type construction has better mechanical support and good provision for bracing the windings. The
shell-type transformer requires more specialized fabrication facilities than core-type, while the latter offers
the additional advantage of permitting visual inspection of coils in the case of a fault and ease of repair
at substation site. For these reasons, the present
Core yoke
1/2 LV
1/2 LV
practice is to use the core-type transformers in large
1/2 HV
1/2 HV
high-voltage installations.
Transformer windings are made of solid or
stranded copper or aluminium strip conductors. For
electronic transformers, “magnet wire” is normally
Windings
used as conductor. Magnet wire is classified by an
insulation class symbol, A, B, C, F and H, which is
indication of the safe operating temperature at which
the conductor can be used. Typical figures are the
Windings
Core
lowest 105 °C for class-A and highest 180 °C for
(a) Core-type transformer
class-H.
Core yoke
The windings of huge power transformers use
conductors with heavier insulation (cloth, paper,
etc.) and are assembled with greater mechanical
support and the winding layers are insulated from
each other—this is known as minor insulation for
f/2
Sandwiched LV
which pressed board or varnished cloth is used.
HV windings
f/2
Major insulation, insulating cylinders made of
specially selected pressed board or synthetic resin
Core
bounded cylinders, is used between LV and core and
LV and HV. Insulating barriers are inserted between
(b) Shell-type transformer
adjacent limbs when necessary and between coils
Fig. 3.2
and core yokes.
Transformer Cooling (Large Units)
Some idea of transformer losses, heating and cooling has been presented above. Details of transformer losses
will be presented in Section 3.6. Basically, there are two seats of losses in a transformer namely:
(1) Core, where eddy current and hysteresis losses occur (caused by alternating flux density).
(2) Windings (primary and secondary) where I 2 R or copper loss occurs because of the current flowing in
these.
Heat due to losses must be removed efficiently from these two main parts of the transformer so that steady
temperature rise is limited to an allowable figure imposed by the class of insulation used. The problem
of cooling in transformers (and in fact for all electric machinery) is rendered increasingly difficult with
increasing size of the transformer. This is argued as below:
The same specific loss (loss/unit volume) is maintained by keeping constant core flux density and
current density in the conductor as the transformer rating is increased. Imagine that the linear dimensions
Transformers
53
of transformer are increased k times. Its core flux and conductor current would then increase by k2 times
and so its rating becomes k4 times. The losses increase by a factor of k 3 (same as volume), while the surface
area (which helps dissipate heat) increases only by a factor of k2. So the loss per unit area to be dissipated is
increased k times. Larger units therefore become increasingly more difficult to cool compared to the smaller
ones. This can lead to formation of hot spots deep inside the conductors and core which can damage the
insulation and core properties. More effective means of heat removal must therefore be adopted with ducts
inside the core and windings to remove the heat right from the seats of its generation.
Natural Cooling
Smaller size transformers are immersed in a tank containing
transformer oil. The oil surrounding the core and windings
gets heated, expands and moves upwards. It then flows
downwards by the inside of tank walls which cause it to
cool and oil goes down to the bottom of the tank from
where it rises once again completing the circulation cycle.
The heat is removed from the walls of the tank by radiation
but mostly by air convection. Natural circulation is quite
effective as the transformer oil has large coefficient of
expansion. Still for large sizes, because of the arguments
presented earlier, the cooling area of the tank must be
increased by providing cooling fins or tubes (circular or
elliptical) as shown in Fig. 3.3. This arrangement is used
for all medium size transformers.
Fig. 3.3
Forced Cooling
Natural cooling in transformers
For transformer sizes beyond 5 MVA additional cooling would be needed which is achieved by supplementing
the tank surface by a separate radiator in which oil is circulated by means of a pump. For better cooling oil-toair heat exchanger unit is provided as shown in Fig. 3.4(a). For very large size transformers cooling is further
strengthened by means of oil-to-water heat exchanger as shown in Fig. 3.4(b).
Water inlet
Conservator
Oil/air heat
exchanger
Oil
pump
Oil/water
heatexchanger
Fan
(a)
(b)
Fig. 3.4
Forced cooling in transformers
54
Electric Machines
As already pointed out, ducts are provided in core and windings for effective heat removal by oil. Vertical
flow is more effective compared to horizontal flow but for pancake coils some of the ducts will have to be
horizontal.
The problem of cooling in transformers is more acute than in electric machines because the rotating
member in a machine causes forced air draft which can be suitably directed to flow over the machine part for
efficient heat removal. This will be discussed in Section 5.10.
Buchholz Relay
Buchholz relay is used in transformers for protection against all kinds of faults. It is a gas-actuated relay and
installed in oil-immersed transformers. It will give an alarm in case of incipient faults in the transformer.
This relay also disconnects the transformer in
Conservator
case of severe internal faults. A Buchholz relay
looks like a domed vessel and it is placed between
main tank of transformer and the conservator.
The upper part of the relay consists of a mercuryBuchholz relay
type switch attached to a float. The lower part
9.5°
contains mercury switch mounted in a hingedtype flat located in the direct path of the flow of
Transformer
oil from the transformer to the conservator. The
main
upper mercury-type switch closes an alarm circuit
tank
during incipient fault, whereas the lower mercury
switch is used to trip the circuit breaker in the
case of sever faults. The Buchholz relay is shown
Fig. 3.4 (c) Buchholz relay set up
in Fig. 3.4 (c).
Figure 3.5 shows the schematic diagram of a two-winding transformer on no-load, i.e. the secondary terminals
are open while the primary is connected to a source of constant sinusoidal voltage of frequency f Hz. The
simplifying assumption that the resistances of the windings are negligible, will be made.
f
i0
+
+
e1
N1
N2
–
+
+
–
–
Secondary
terminals
open
e 2 v2
–
Secondary
Primary
Core (magnetic material)
Fig. 3.5 Transformer on no-load
Transformers
55
The primary winding draws a small amount of alternating current of instantaneous value i0, called the
exciting current, from the voltage source with positive direction as indicated on the diagram. The exciting
current establishes flux f in the core (positive direction marked on diagram) all of which is assumed confined
to the core i.e., there is no leakage of flux. Consequently the primary winding has flux linkages,
l1 = N1f
which induces emf in it is given by
d l1
df
(3.1)
= N1
dt
dt
As per Lenz’s law, the positive direction of this emf opposes the positive current direction and is shown by
+ and – polarity marks on the diagram. According to Kirchhoff’s law,
e1 =
v1 = e1 (winding has zero resistance)
(3.2)
and thus e1 and therefore f must be sinusoidal of frequency f Hz, the same as that of the voltage source. Let
where
f = fmax sin w t
fmax = maximum value of core flux
w = 2pf
rad/s ( f = frequency of voltage source)
(3.3)
The emf induced in the primary winding is
df
= w N1fmax cos wt
e1 = N1
(3.4)
dt
From Eqs (3.3) and (3.4) it is found that the induced emf leads the flux by 90°*. This is indicated by the
phasor diagram of Fig. 3.6. The rms value of the induced emf is
2 pf N1fmax = 4.44 f N1fmax
E1 =
(3.5)
Since E1 = V1 as per Eq. (3.2),
fmax =
E1 (=V1 )
4.44 f N1
(3.6)
Even if the resistance of the primary winding is taken into account,
E1 ª V1
(3.7)
as the winding resistances in a transformer are of extremely small order. It is, therefore, seen from Eq. (3.6)
that maximum flux in a transformer is determined by V1 /f (voltage/frequency) ratio at which it is excited.
According to Eq. (3.6) the flux is fully determined by the applied voltage, its frequency and the number of
winding turns. This equation is true not only for a transformer but also for any other electromagnetic device
operated with sinusoidally varying ac and where the assumption of negligible winding resistance holds.
All the core flux f also links the secondary coil (no leakage flux) causing in it an induced emf of
df
(3.8)
e2 = N2
dt
The polarity of e2 is marked + and – on Fig. 3.5 according to Lenz’s law (e2 tends to cause a current flow
whose flux opposes the mutual flux f). Further, it is easily seen from Eqs. (3.1) and (3.8) that e1 and e2 are in
* cos w t leads sin w t by 90°
56
Electric Machines
phase. This is so indicated by phasor diagram of Fig. 3.6 where E1 , and E2 are the corresponding phasors in
terms of the rms values. As the secondary is open-circuited, its terminal voltage is given as
v2 = e2
From Eqs (3.1) and (3.8) we have the induced emf ratio of the transformer windings as
e1
N
= 1 =a
e2
N2
E1
N
= 1 = a ratio of transformation
E2
N2
(3.9)
This indeed is the transformation action of the transformer. Its current transformation which is in inverse
ratio of turns will be discussed in Section 3.4.
The value of exciting current i0 has to be such that the required mmf is established so as to create the flux
demanded by the applied voltage (Eq. (3.6)). If a linear B-H relationship is assumed (devoid of hysteresis and
saturation), the exciting current is only magnetizing in nature and is proportional to the sinusoidal flux and
in phase with it. This is represented by the phasor I m , in Fig. 3.6, lagging the induced emf by 90°. However,
the presence of hysteresis and the phenomenon of eddy-currents, though of a different physical nature, both
demand the flow of active power into the system and as a consequence the exciting current I 0 has another
component I i in phase with E1 . Thus, the exciting current lags the induced emf by an angle q0 slightly less
than 90° as shown in the phasor diagram of Fig. 3.6. Indeed it is the hysteresis which causes the current
component I i leading I m by 90° and eddy-currents add more of this component. The effect of saturation
nonlinearity is to create a family of odd-harmonic components in the exciting current, the predominant being
the third harmonic; this may constitute as large as 35–40% of the exciting current. While these effects will
be elaborated in Sec. 3.10, it will be assumed here that the current I o and its magnetizing component I m
and its core-loss component I i are sinusoidal on equivalent rms basis. In other words, I m is the magnetizing
current and is responsible for the production of flux, while I i is the core-loss current responsible for the
active power* being drawn from the source to provide the hysteresis and eddy-current loss.
To account for the harmonics, the exciting current I o is taken as the rms sine wave equivalent of the
actual non-sinusoidal current drawn by the transformer on no-load. Since the excitation current in a typical
transformer is only about 5% of the full-load current, the net current drawn by the transformer under loaded
condition is almost sinusoidal.
From the phasor diagram of Fig. 3.6, the core-loss is given by
Pi = E1 I0 cos q0
(3.10)
In a practical transformer, the magnetizing current (Im) is kept low and the core-loss is restrained to an
acceptable value by use of high permeability silicon-steel in laminated form.
From the no-load phasor diagram of Fig. 3.6, the parallel circuit model** of exciting current as shown
in Fig. 3.7 can be easily imagined wherein conductance Gi accounts for core-loss current I i and inductive
susceptance Bm for magnetizing current Im. Both these currents are drawn at induced emf E1 = V1 for
resistance-less, no-leakage primary coil; even otherwise E1 ª V1 .
* Suffix i is used as this current provides the core-loss which occurs in the iron core and is also referred as iron-loss.
** Series circuit is equally possible but not convenient for physical understanding.
Transformers
li
E2
q0
+
E1 = V1
V1
a0
Im
l0
+
l0
li
Gi
57
lm
Bm
E1
–
–
f
Fig. 3.6
Phasor relationship of induced emf,
Fig. 3.7
Circuit model of transformer on
EXAMPLE 3.1 A transformer on no-load has a core-loss of 50 W, draws a current of 2 A (rms) and has
an induced emf of 230 V (rms). Determine the no-load power factor, core-loss current and magnetizing
current. Also calculate the no-load circuit parameters of the transformer. Neglect winding resistance and
leakage flux.
SOLUTION
cos q0 =
Power factor,
Magnetizing current,
50
= 0.108 lagging;
2 ¥ 230
q0 = 83.76°
Im = I0 sin q0 = 2 sin (cos–1 0.108) = 1.988 A
Since q0 ª 90°, there is hardly any difference between the magnitudes of the exciting current and its magnetizing
component.
Core-loss current, Ii = I0 cos q 0
= 2 ¥ 0.108 = 0.216 A
In the no-load circuit model of Fig. 3.7 core loss is given by
Gi V 12 = Pi
Pi
or
Gi =
Also
Im = Bm V1
or
Bm =
=
V12
=
50
(230) 2
= 0.945 ¥ 10–3 Im
V1
1.988
= 8.64 ¥ 10–3 230
EXAMPLE 3.2 The BH curve data for the core of the transformer shown in Fig. 3.8 is given in Problem 2.10.
Calculate the no-load current with the primary excited at 200 V, 50 Hz. Assume the iron loss in the core to be
3 W/kg. What is the pf of the no-load current and the magnitude of the no-load power drawn from the mains?
Density of core material = 7.9 g/cc.
58
Electric Machines
5 cm thick
10 cm
150
turns
200 V
20 cm
75 turns
25 cm
Fig. 3.8
SOLUTION
Substituting values in Eq. (3.5)
200 = 4.44 ¥ 50 ¥ 150 ¥ fmax
fmax = 6.06 mWb
or
Bmax =
6.06 ¥ 10- 3
10 ¥ 5 ¥ 10- 4
= 1.212 T
From the data of BH curve of Problem 2.10, we get
Hmax = 250 AT/m
ATmax = 250 lC = 250 ¥ 2 (30 + 35) ¥ 10–2
= 325
325
Im(max) =
= 2.17 A
150
2.17
= 1.53 A
Im(rms) =
2
Core volume = 2(20 ¥ 10 ¥ 5) + 2(45 ¥ 10 ¥ 5)
= 6500 cm3
Weight of core = 6500 ¥ 7.9 ¥ 10–3 = 51.35 kg
Core loss = 51.35 ¥ 3 = 154.7 W
154.7
Ii =
= 0.77 A
200
Referring to the phasor diagram of Fig. 3.7
No-load
3.4
I 0 = 0.77 – j 2.17 = 2.3 ––70.5°
I0 = 2.3 A (no-load current)
pf = cos 71.5° = 0.334 lagging
IDEAL TRANSFORMER
In order to visualize the effect of flow of secondary current in a transformer, certain idealizing assumptions
will be made which are close approximations for a practical transformer. A transformer possessing these
Transformers
59
ideal properties is hypothetical (has no real existence) and is referred to as the ideal transformer. It possesses
certain essential features of a real transformer but some details of minor significance are ignored which will
be reintroduced at a refined stage of analysis. The idealizing assumptions made are listed below:
(i) The primary and secondary windings have zero resistance. It means that there is no ohmic power loss
and no resistive voltage drop in the ideal transformer. An actual transformer has finite but small winding
resistances. It will also be assumed that there is no stray capacitance, though the actual transformer has
inter-turn capacitance and capacitance between turns and ground but their effect is negligible at 50 Hz.
(ii) There is no leakage flux so that all the flux is confined to the core and links both the windings. An
actual transformer does have a small amount of leakage flux which can be accounted for in detailed
analysis by appropriate circuit modelling.
(iii) The core has infinite permeability so that zero magnetizing current is needed to establish the requisite
amount of flux (Eq. (3.6)) in the core.
(iv) The core-loss (hysteresis as well as eddy-current loss) is considered zero.
Figure 3.9 shows an ideal transformer having a primary of N1 turns and a secondary of N2 turns on a
common magnetic core. The voltage of the source
Primary
Secondary
to which the primary is connected is
v1 =
2 V1 cos wt
f
(3.11)
while the secondary is initially assumed to be
an open circuited. As a consequence, flux f is
established in the core (Eq. (3.6)) such that
i1
i2
+
F1
+
v1
–
df
(3.12)
e1 = v1 = N1
dt
but the exciting current drawn from the source is
zero by virtue of assumption (iii) above. The flux
f which is wholly mutual (assumption (ii) above)
causes an emf
+
F2
e2
e1
N1
–
Fig. 3.9
N2
v2
Load
z
–
Ideal transformer on load
df
(3.13)
dt
to be induced in the secondary of polarity* marked on the diagram for the winding direction indicated. The
dots marked at one end of each winding indicate the winding ends which simultaneously have the same
polarity due to emfs induced. From Eqs (3.12) and (3.13)
e2 = N2
e1
N1
=
=a
e2
N2
(3.14)
Since a, the transformation ratio, is a constant, e1 and e2 are in phase. The secondary terminal voltage is
v2 = e2
Hence
v1
e1
N1
=
=
=a
v2
e2
N2
* The reader may check these polarities by applying Lenz’s law while assuming that flux f is increasing.
(3. 15)
(3. 16)
60
Electric Machines
It is, therefore, seen that an ideal transformer changes (transforms) voltages in direct ratio of the number
of turns in the two windings. In terms of rms values Eq. (3. I 6) implies
V1
E1
N1
=
=
= a (same as Eq. (3.9))
(3.17)
V2
E2
N2
Now let the secondary be connected to a load of impedance Z2 so that the secondary feeds a sinusoidal
current of instantaneous value i2 to the load. Due to this flow of current, the secondary creates mmf F2 =
i2 N2 opposes the flux f. However, the mutual flux f cannot change as otherwise the (v1, e1) balance will be
disturbed (this balance must always hold as winding has zero leakage and resistance). The result is that the
primary draws a current i1 from the source so as to create mmf F1 = i1 N1 which at all time cancels out the load
caused mmf i2 N2 so that f is maintained constant independent of the load current flow, Thus
i1 N1 = i2 N2
(3.18)
i1
N2
1
(3.19)
=
=
i2
N1
a
Obviously i1 and i2 are in phase for positive current directions marked on the diagram (primary current
in at the dotted terminal and secondary current out of the dotted terminal). Since flux f is independent of
load, so is e2, and v2 must always equal e2 as the secondary is also resistanceless. Therefore, from Eqs (3.17)
and (3.19)
or
or
i1
N2
v2
=
=
i2
N1
v1
v1 i1 = v2 i2
(3.20)
which means that the instantaneous power into primary equals the instantaneous power out of secondary, a
direct consequence of the assumption (i) which means a loss-less transformer.
In terms of rms values Eq. (3.19) will be written as
I1
N2
1
(3.21)
=
=
I2
N1
a
which implies that currents in an ideal transformer transform in inverse ratio of winding turns.
Equation (3.20) in terms of rms values will read
V1 I1 = V2 I2
(3.22)
i.e. the VA output is balanced by the VA input.
Figure 3.10(a) shows the schematic of the ideal transformer of Fig. 3.9 with dot marks identifying similar
polarity ends. It was already seen above that V1 and V2 are in phase and so are I1 and I 2 . Now
and
N1
V1
=
N2
V2
(3.23a)
N2
I1
=
N1
I2
(3.23b)
Dividing Eq. (3.23a) by Eq. (3.23b)
N1 / N 2
V1 /V2
=
N 2 / N1
I1 / I 2
Transformers
I2
I1
a
+
61
Z2
V2
V1
–
b
N1 : N2
(a)
I1 (N1IN2)2 Z2
I2
I1
a
+
a
+
–
b
–
b
N1 : N 2
(b)
Fig. 3.10
(N1IN2)2 Z2 = Z¢2
V1
V1
(c)
Ideal transformer; referring impedance from secondary to primary
2
2
or
Ê N1 ˆ V2
Ê N1 ˆ
V1
= Á
˜¯ I = ÁË N ˜¯ Z 2
N
Ë
I1
2
2
2
or
Ê N1 ˆ
2
Z 2 = a Z 2 = Z 2¢
Z1 = Á
Ë N 2 ˜¯
(3.24)
2
(3.25)
It is concluded from Eq. (3.25) that the impedance on the secondary side when seen (referred to) on the
primary side is transformed in the direct ratio of square of turns. Equivalence of Eqs (3.24) and (3.25) to the
original circuit of Fig. 3.10(a) is illustrated through Figs 3.10(b) and (c). Similarly an impedance Z1 from the
primary side can be referred to the secondary as
2
Ê N2 ˆ
1
Z1 = 2 Z1
Z1¢ = Á
(3.26)
Ë N1 ˜¯
a
Transferring an impedance from one side of a transformer to the other is known as referring the impedance
to the other side. Voltages and currents on one side have their counterpart on the other side as per Eqs (3.23(a)
and (b)).
In conclusion it may be said that in an ideal transformer voltages are transformed in the direct ratio of
turns, currents in the inverse ratio and impedances in the direct ratio squared; while power and VA remain
unaltered.
Equation (3.25) illustrates the impedance-modifying property of the transformer which can be exploited
for matching a fixed impedance to the source for purposes of maximum power transfer by interposing a
transformer of a suitable turn-ratio between the two.
EXAMPLE 3.3 Assume the transformer of Fig. 3.8 to be the ideal transformer. The secondary is connected
to a load of 5 –30°. Calculate the primary and secondary side impedances, current and their pf, and the
real powers. What is the secondary terminal voltage?
62 Electric Machines
The circuit model of the ideal transformer is drawn in Fig. 3.11.
Z 2 = 5 –30° W
+
a = N1/N2 = 150/75 = 2
2
Z1 = Z 2¢ = (2) 5 –30° = 20 –30° W
V2 = 200/2 = 100 V;
(secondary terminal voltage)
or
or otherwise
l2
150 : 75
V2
200 V
–
I 2 = 100 –0°/5 –30° = 20 ––30° A
I2 = 20 A; pf = cos 30° = 0.866 lagging
or
3.5
l1
5–30° W
SOLUTION
Fig. 3.11
I1 = I 2¢ = 20 –30°/2 = 10 ––30° A
I1 = 10 A; pf = cos 30° = 0.866 lagging
P2 (secondary power output) = (20)2 ¥ Re 5 –30°
= 400 ¥ 4.33 = 1.732 kW
P1 (primary power input) = P2 (as the transformer is lossless)
= 1.732 kW
P1 = V1I1 cos q1 = 200 ¥ 10 ¥ 0.866
= 1.732 kW
REAL TRANSFORMER AND EQUIVALENT CIRCUIT
Figure 3.12 shows a real transformer on load. Both the primary and secondary have finite resistances R1 and
R2 which are uniformly spread throughout the winding; these give rise to associated copper (I 2R) losses.
While a major part of the total flux is confined to the core as mutual flux f linking both the primary and
secondary, a small amount of flux does leak through paths which lie mostly in air and link separately the
individual windings. In Fig. 3.12 with primary and secondary for simplicity assumed to be wound separately
on the two legs of the core, leakage flux fl1 caused by primary mmf I1N1 links primary winding itself and f l2
caused by I2 N2 links the secondary winding, thereby causing self-linkages of the two windings. It was seen
in Sec. 3.2 with reference to Fig. 3.2(a) that half the primary and half the secondary is wound on each core
leg. This reduces the leakage flux linking the individual windings. In fact it can be found by tracing flux paths
that leakage flux is now confined to the annular space between the halves of the two windings on each leg.
In shell-type construction, the leakage will be still further reduced as LV and HV pancakes are interleaved.
Theoretically, the leakage will be eliminated if
Mutual flux
the two windings could be placed in the same
f
physical space but this is not possible; the
l1
l2
practical solution is to bring the two as close
+
as possible with due consideration to insulation
+
N2
and constructional requirements. Shell-type V
V2
Z2
1
N1
construction though having low leakage is still
–
–
not commonly adopted for reasons explained in
Sec. 3.2. Actually some of the leakage flux will
link only a part of the winding turns. It is to be
Secondary
Primary
leakage fI2
understood here that fl1 and fl2 are equivalent
leakage fI1
leakage fluxes linking all N1 and N2 turns
Fig. 3.12 Real transformer
respectively.
Transformers
63
As the leakage flux paths lie in air for considerable part of their path lengths, winding mmf and its selflinkage caused by leakage flux are linearly related in each winding; therefore contributing constant leakage
inductances (or leakage reactances corresponding to the frequency at which the transformer is operated) of
both primary and secondary windings. These leakage reactances* are distributed throughout the winding
though not quite uniformly.
Both resistances and leakage reactances of the transformer windings are series effects and for low operating
frequencies at which the transformers are commonly employed (power frequency operation is at 50 Hz only),
these can be regarded as lumped parameters. The real transformer of Fig. 3.12 can now be represented as
a semi-ideal transformer having lumped resistances R1 and R2 and leakage reactances symbolized as X ll
and X l2 in series with the corresponding windings as shown in Fig. 3.13. The semi-ideal transformer draws
magnetizing current to set up the mutual flux f and to provide for power loss in the core; it, however, has no
winding resistances and is devoid of any leakage. The induced emfs of the semi-ideal transformer are E1 and
E2 which differ respectively from the primary and secondary terminal voltages V1 and V2 by small voltage
drops in winding resistances and leakage reactances (R1, X l1 for primary and R2, Xl2 for secondary). The ratio
of transformation is
a=
N1
E1
V1
=
ª
N2
E2
V2
(3.27)
–
f
l1
+
R1
X/2
X/1
+
V1
–
+
E1
N1
N2
–
E2
R2
l2
+
V2
–
–
Fig. 3.13 Circuit model of transformer employing semi-ideal transformer
because the resistances and leakage reactance of the primary and secondary are so small in a transformer that
E1 ª V1 and E2 ª V2.
Equivalent Circuit
In Fig. 3.13 the current I1 flowing in the primary of the semi-ideal transformer can be visualized to comprise
two components as below:
(i) Exciting current I 0 whose magnetizing component I m creates mutual flux f and whose core-loss
component I i provides the loss associated with alternation of flux.
(ii) A load component I 2¢ which counterbalances the secondary mmf I 2 N2 so that the mutual flux remains
constant independent of load, determined only by E1 . Thus
I1 = I 0 + I 2¢
(3.28)
* The transformer windings possess inter-turn and turns-to-ground capacitances. Their effect is insignificant in the
usual low-frequency operation. This effect must, however, be considered for high-frequency end of the spectrum
in electronic transformers. In low-frequency transformers also, capacitance plays an important role in surge phenomenon caused by switching and lightning.
64
Electric Machines
where
N2
I 2¢
=
N1
I2
(3.29)
The exciting current I 0 can be represented by the circuit model of Fig. 3.7 so that the semi-ideal
transformer of Fig. 3.13 is now reduced to the true ideal transformer. The corresponding circuit (equivalent
circuit) modelling the behaviour of a real transformer is drawn in Fig. 3.14( a) wherein for ease of drawing
the core is not shown for the ideal transformer.
The impedance (R2 + jX l2) on the secondary side of the ideal transformer can now be referred to its
primary side resulting in the equivalent circuit of Fig. 3.14(b) wherein
2
Ê N1 ˆ
X l2
¢ = Á
X
Ë N 2 ˜¯ l2
(3.30a)
2
Ê N1 ˆ
R¢2 = Á
R
Ë N 2 ˜¯ 2
(3.30b)
The load voltage and current referred to the primary side are
Ê N1 ˆ
V2
V2¢ = Á
Ë N 2 ˜¯
(3.31a)
Ê N2 ˆ
I2
I 2¢ = Á
Ë N1 ˜¯
(3.31b)
Therefore there is no need to show the ideal transformer reducing the transformer equivalent circuit to the
T-circuit of Fig. 3.14(c) as referred to side 1. The transformer equivalent circuit can similarly be referred to
side 2 by transforming all impedances (resistances and reactances), voltages and currents to side 2. It may be
noted here that admittances (conductances and susceptances) are transformed in the inverse ratio squared in
contrast to impedances (resistances and reactances) which as already shown in Sec. 3.4 transform in direct
ratio squared. The equivalent circuit of Fig. 3.14(c) referred to side 2 is given in Fig. 3.14(d) wherein
V1¢ =
N2
V1
N1
I1¢ =
N1
I1
N2
2
2
2
2
ÊN ˆ
ÊN ˆ
G¢i = Á 1 ˜ Gi; B¢m = Á 1 ˜ Bm
Ë N2 ¯
Ë N2 ¯
Ê N2 ˆ
ÊN ˆ
R¢2 = Á
R2; X¢l1 = Á 2 ˜ X l1
˜
Ë N1 ¯
Ë N1 ¯
With the understanding that all quantities have been referred to a particular side, a superscript dash can be
dropped with a corresponding equivalent circuit as drawn in Fig. 3.14(d).
In the equivalent circuit of Fig. 3.14(c) if Gi is taken as constant, the core-loss is assumed to vary as E 21 or
2
f max f 2 (Eq. (3.6)). It is a fairly accurate representation as core-loss comprises hysteresis and eddy-current
Transformers
l1
+
N1 : N 2
+
l0
R2 l2
+
+
lm
li
Gi
V1
X/2
l¢2
X/1
R1
E1
Bm
E2
–
–
V2
–
–
Ideal transformer
(a)
l1
R1
X¢/2
X/1
+
I0
E1
V1
li
Gi
R¢2 I¢
2 N1 : N 2
+
+
l2
+
lm
V¢2
Bm
–
–
–
–
V2
Ideal transformer
(b)
+
l1
R1
X l1
X¢/2
R¢2
I¢2
+
l0
V1
E1 Gi
V¢2
Bm
–
–
(c)
l¢1
R¢1
X¢/1
+
X/2
R2
I2
+
I0
V¢1
E2
G¢i
V2
B¢m
–
–
(d)
I1
R1
X/1
+
X¢l2
E1
Gi
Bm
–
I¢2
+
I0
V1
R¢2
V¢2
–
(e)
Fig. 3.14
Evolution of transformer equivalent circuit
65
66
Electric Machines
loss expressed as (Khf 1.6max f + Kef 2max f 2). The magnetizing current for linear B-H curve varies proportional
E
to fmax μ 1 . If inductance (Lm) corresponding to susceptance Bm is assumed constant.
f
Im =
E1
= Bm E1
2p f Lm
It therefore is a good model except for the fact that the saturation effect has been neglected in which case
Bm would be a nonlinear function of E1/f. It is an acceptable practice to find the shunt parameters Gi, Bm at
the rated voltage and frequency and assume these as constant for small variations in voltage and frequency.
The passive lumped T-circuit representation of a transformer discussed above is adequate for most
power and radio frequency transformers. In transformers operating at higher frequencies, the interwinding
capacitances are often significant and must be included in the equivalent circuit. The circuit modification to
include this parameter is discussed in Sec. 3.12.
The equivalent circuit given here is valid for a sinusoidal steady-state analysis. In carrying out transient
analysis all reactances must be converted to equivalent inductances.
The equivalent circuit developed above can also be arrived at by following the classical theory of
magnetically coupled circuits, section 3.22. The above treatment is, however, more instructive and gives a
clearer insight into the physical processes involved.
Phasor Diagram of Exact Equivalent Circuit of Transformer [Fig. 3.14(a)]
The KVL equations for the primary and secondary circuits are
V1 = E1 + I1 R1 + j I1 X1
V2 = E2 - I 2 R2 - j I 2 X1
The ideal transformer relationships are
1
E1
I¢
=a; 2 =
a
I2
E2
The nodal equation for the primary side currents is
(
I1 = I 2¢ + I 0¢ = I 2¢ + I i + I m
where
)
I i is in phase with E1
Im is 90° lagging E1
The exact phasor diagram from these equations is drawn in Fig. 3.15.
Alternative Phasor Diagram
dl
; direction of E1 and E 2 will reverse in Fig. 3.13 and these will lag the flux
dt
phasor by 90°. The direction of secondary current is now into the dotted terminal. So for mmf balance
Alternatively if we use e = -
I 2¢ = - I 2 . The KVL equation on the primary side is
V1 = (- E1 ) + I 2¢ R2 + j I 2¢ X 2
The corresponding phasor diagram is drawn in Fig. 3.16. Note that the polarity of V2 reverses in Fig. 3.13
but it is of no consequence.
Transformers
67
V1
I1 X1
E1
E2
O
Ii
q2
I1 R1
I2 X2
v2
q1
I2 R2
I0
Im
I¢2
I1
I2
I0
Fig. 3.15
–
f
I1
Im
E2
I¢2
Ii
q1
I2 X2
O
I1 R1
I1 X1
V2
q2
I2 R2
–E1
I2
V1
Fig. 3.16
EXAMPLE 3.4 Consider the transformer of Example 3.2 (Fig. 3.8) with load impedance as specified in
Example 3.3. Neglecting voltage drops (resistive and leakage reactive drops), calculate the primary current
and its pf. Compare with the current as calculated in Example 3.3.
SOLUTION
As per Eqs (3.28) and (3.29)
I1 = I 0 + I 2¢
and
As calculated in Example 3.3
I 2¢ = 10 ––30° A
N2
I 2¢
=
N1
I2
68
Electric Machines
Further as calculated in Example 3.2
I 0 = 1.62 ––71.5°
I1 = 1.62 ––71.5° + 10 ––30°
= (0.514 – j 1.54) + (8.66 – j 5)
= 9.17 – j 6.54 = 11.26 ––35° A
I1 = 11.26 A, pf = cos 35° = 0.814 lagging
Hence
Compared to the primary current computed in Example 3.3 (ignoring the exciting current) the magnitude of the
current increases slightly but its pf reduces slightly when the exciting current is taken into account.
In large size transformers the magnitude of the magnetizing current is 5% or less than the full-load current and so its
effect on primary current under loaded conditions may even be altogether ignored without any significant loss in accuracy.
This is a usual approximation made in power system computations involving transformers.
EXAMPLE 3.5 A 20-kVA, 50-Hz, 2000/200-V distribution transformer has a leakage impedance of 0.42
+ j 0.52 W in the high-voltage (HV) winding and 0.004 + j 0.05 W in the low-voltage (LV) winding. When
seen from the LV side, the shunt branch admittance Y0 is (0.002 – j 0.015) (at rated voltage and frequency).
Draw the equivalent circuit referred to (a) HV side and (b) LV side, indicating all impedances on the circuit.
SOLUTION
The HV side will be referred as 1 and LV side as 2.
N1
2000
Transformation ratio, a =
=
= 10 (ratio of rated
N2
200
voltages; see Eq. (3.27))
0.42 + j 0.52 W
(0.002 – j 0.015)
W
¥ 10 –2
2
Y0¢ =
1
1¢
2¢
(a)
(0.002 – j 0.015) (Notice that in
(10) 2
transforming admittance is divided by a2)
The equivalent circuit is drawn in Fig. 3.17(a).
(b) Equivalent circuit referred to LV side (side 2).
1
(0.42 + j 0.52) = 0.0042 + j 0.0052
Z1¢ =
(10) 2
The equivalent circuit is drawn in Fig. 3.17(b).
10:1
2
(a) Equivalent circuit referred to HV side (side 1)
Z 2¢ = (10) (0.004 + j 0.005) = 0.4 + j 0.5 W
0.4 + j 0.5 W
1
10:1
0.0042 + j 0.0052 W
0.004 + j 0.005 W
1
2
(0.002 – j 0.015)
1¢
W
2¢
(b)
Fig. 3.17
Approximate Equivalent Circuit
In constant frequency (50 Hz) power transformers, approximate forms of the exact T-circuit equivalent of
the transformer are commonly used. With reference to Fig. 3.14(c), it is immediately observed that since
winding resistances and leakage reactances are very small, V1 ª E1 even under conditions of load. Therefore,
the exciting current drawn by the magnetizing branch (Gi || Bm) would not be affected significantly by shifting
it to the input terminals, i.e. it is now excited by V1 instead of E1 as shown in Fig. 3.18(a). It may also be
observed that with this approximation, the current through R1, X l1 is now I ¢2 rather than I1 = I 0 + I 2¢ . Since I 0
is very small (less than 5% of full-load current), this approximation changes the voltage drop insignificantly.
Thus it is basically a good approximation. The winding resistances and reactances being in series can now
Transformers
69
be combined into equivalent resistance and reactance of the transformer as seen from the appropriate side (in
this case side 1). Remembering that all quantities in the equivalent circuit are referred either to the primary
or secondary dash in the referred quantities and suffixes l, 1 and 2 in equivalent resistance, reactance and
impedance can be dropped as in Fig. 3.18(b).
l1
Req = R1 + R¢2 Xeq = X/1 + X¢/2
Here
Req (equivalent resistance) = R1 + R2
Xeq (equivalent reactance) = X l1 + Xl2
Zeq (equivalent impedance) = Req + j Xeq
+
I0
V1
Gi
I¢2
Bm
+
V¢2
In computing voltages from the approximate equivalent
–
circuit, the parallel magnetizing branch has no role to play –
(a)
and can, therefore, be ignored as in Fig. 3.18(b).
–
–
Xeq
I1
Req
The approximate equivalent circuit offers excellent
l
+
computational ease without any significant loss in the +
accuracy of results. Further, the equivalent resistance and
V2
reactance as used in the approximate equivalent circuit
V1
offer an added advantage in that these can be readily
–
measured experimentally (Sec. 3.7), while separation of –
(b)
X l1 and X l2 experimentally is an intricate task and is rarely
attempted.
Xeq
–
–
l
l
The approximate equivalent circuit of Fig. 3.18(b) in +
+
which the transformer is represented as a series impedance
is found to be quite accurate for power system modelling
V2
V1
[7]. In fact in some system studies, a transformer may be
–
represented as a mere series reactance as in Fig. 3.18(c). –
(c)
This is a good approximation for large transformers which
always have a negligible equivalent resistance compared to Fig. 3.18
the equivalent reactance.
transformer
The suffix ‘eq’ need not be carried all the time so that R and X from now onwards will be understood to be
equivalent resistance and reactance of the transformer referred to one side of the transformer.
Phasor Diagram
For the approximate equivalent circuit of Fig. 3.18(b), (suffix ‘eq’ is being dropped now),
V2 = V1 - I Z
(3.32a)
V1 = V2 + I (R + jX )
(3.32b)
The phasor diagram corresponding to this equation is drawn in Fig. 3.19(a) for the lagging power factor
(phase angle* f between V2 and I ) and in Fig. 3.19(b) for the leading power factor (pf ). It is immediately
observed from these phasor diagrams that for the phase angle indicated
or
and
V2 < V1
V2 > V1
for lagging pf
for leading pf
It will be shown in Sec. 3.9 that V2 > V1 only when leading phase angle f is more than tan–1(R/X ) (see
Eq. (3.67).
* Phase angle f should not be confused with flux though the same symbol has been used.
70
Electric Machines
I
V1 A
lZ
V1
f
O
d
B
f
V2
I
IR
90°
f
IX
D
C
E
F
f
IX
d
O
IR
V2
(b) leading pf
(a) Lagging pf
Fig. 3.19
In the plasor diagrams of Figs. 3.19(a) and (b) (these are not drawn to scale), the angle d is such that V1
leads V2. This is an indicator of the fact that real power flows from side 1 to side 2 of the transformer (this
is proved in Section 8.9). This angle is quite small and is related to the value of the equivalent reactance:
resistance of the transformer being negligible.
Name Plate Rating
The voltage ratio is specified as V1 (rated)/V2 (rated). It means that when voltage V1 (rated) is applied to the
primary, the secondary voltage on fullload at specified pf is V2 (rated). The ratio V1 (rated)/V2 (rated) is not
exactly equal to N1/N2, because of voltage drops in the primary and secondary. These drops being small are
neglected and it is assumed that for all practical purposes
N1
V1 (rated )
=
N
V2 (rated )
2
(3.33)
The rating of the transformer is specified in units of VA/kVA/MVA depending upon its size.
V (rated ) ¥ I (full-load)
(3.34)
1000
where V and I are referred to one particular side. The effect of the excitation current is of course ignored.
The transformer name plate also specifics the equivalent impedance, but not in actual ohm. It is expressed
as the percentage voltage drop (see Sec. 3.8) expressed as
kVA(rated) =
I (full-load ) Z
¥ 100%
V (rated )
where all quantities must be referred to anyone side.
EXAMPLE 3.6 The distribution transformer described in Example 3.5 is employed to step down the
voltage at the load-end of a feeder having an impedance of 0.25 + j 1.4 W. The sending-end voltage of
the feeder is 2 kV. Find the voltage at the load-end of the transformer when the load is drawing rated
transformer current at 0.8 pf lagging. The voltage drops due to exciting current may be ignored.
Transformers
SOLUTION The approximate equivalent circuit referred
to the HV side, with the value of transformer impedance from
Fig. 3.17(a), is drawn in Fig. 3.20. The feeder being the HV
side of the transformer, its impedance is not modified.
Rated load current (HV side) =
20
= 10 A
2
0.25 + j 1.4
0.82 + j 1.02
Feeder
Transformer
+
V1 = 2000V
Z = (0.25 + j 1.4) + (0.82 + j 1.02)
I = 10 A 0.8 pf lag
Load
–
= 1.07 + j 2.42 = R + jX
71
+
V2
–
Fig. 3.20
One way is to compute V2 from the phasor Eq. (3.32). However, the voltage drops being small, a quick, approximate
but quite accurate solution can be obtained from the phasor diagram of Fig. 3.19(a) without the necessity of carrying out
complex number calculations.
From Fig. 3.19(a)
OE =
(OA) 2 - ( AE ) 2
From the geometry of the phasor diagram
AE = AF – FE
= IX cos f – IR sin f
= 10(2.42 ¥ 0.8 – 1.07 ¥ 0.6) = 12.94 V
OE =
Now
(2000) 2 - (12.94) 2 = 1999.96 V
It is therefore seen that
OE ª OA = V1 (to a high degree of accuracy; error is 2 in 105)
V2 can then be calculated as
V2 = OE – BE
ª V1 – BE
BE = BD + DE
Now
= I(R cos f + X sin f)
= 10(1.07 ¥ 0.8 + 2.42 ¥ 0.6) = 23.08 V
V2 = 2000 – 23.08 = 1976.92 V
Thus
Load voltage referred to LV side =
Remark
as
1976.92
= 197.692 V
10
It is noticed that to a high degree of accuracy, the voltage drop in transformer impedance can be approximated
V1 – V2 = I(R cos f + X sin f); lagging pf
(3.35a)
V1 – V2 = I(R cos f – X sin f); leading pf
(3.35b)
It will soon be shown that
3.6 TRANSFORMER LOSSES
The transformer has no moving parts so that its efficiency is much higher than that of rotating machines. The
various losses in a transformer are enumerated as follows:
72 Electric Machines
Core-loss
These are hysteresis and eddy-current losses resulting from alternations of magnetic flux in the core.
Their nature and the remedies to reduce these have already been discussed at length in Sec. 2.6. It may be
emphasized here that the core-loss is constant for a transformer operated at constant voltage and frequency
as are all power frequency equipment.
Copper-loss (I2R-loss)
This loss occurs in winding resistances when the transformer carries the load current; varies as the square of
the loading expressed as a ratio of the full-load.
Load (stray)-loss
It largely results from leakage fields inducing eddy-currents in the tank wall, and conductors.
Dielectric-loss
The seat of this loss is in the insulating materials, particularly in oil and solid insulations.
The major losses are by far the first two: Pi, the constant core (iron)-loss and Pc, the variable copper-loss.
Therefore, only these two losses will be considered in further discussions.
It will be seen in Sec. 3.7 that transformer losses and the parameters of its equivalent circuit can be easily
determined by two simple tests without actually loading it.
3.7 TRANSFORMER TESTING
Two chief difficulties which do not warrant the testing of large transformers by direct load test are: (i) large
amount of energy has to be wasted in such a test, (ii) it is a stupendous (impossible for large transformers)
task to arrange a load large enough for direct loading. Thus performance characteristics of a transformer
must be computed from a knowledge of its equivalent circuit parameters which, in turns, are determined by
conducting simple tests involving very little power consumption, called nonloading tests. In these tests the
power consumption is simply that which is needed to supply the losses incurred. The two nonloading tests
are the Open-circuit (OC) test and Short-circuit (SC) test.
In both these tests voltage, current and power are measured from which the resistance and reactance of
the input impedance can be found, as seen in each test. Thus only four parameters can be determined which
correspond to the approximate equivalent circuit of Fig. 3.16(a).
Before proceeding to describe OC and SC tests, a simple test will be advanced for determining similar
polarity ends on the two windings of a transformer.
Polarity Test
Similar polarity ends of the two windings of a transformer are those ends that acquire simultaneously positive
or negative polarity of emfs induced in them. These are indicated by the dot convention as illustrated in
Sec. 3.4. Usually the ends of the LV winding are labelled with a small letter of the alphabet and are suffixed
1 and 2, while the HV winding ends are labelled by the corresponding capital letter and are suffixed 1 and 2
as shown in Fig. 3.21. The ends suffixed 2 (a2, A2) have the same polarity and so have the ends labelled 1
(a1, A1).
Transformers
73
In determining the relative polarity of the two-windings of a transformer the two windings are connected
in series across a voltmeter, while one of the windings is excited from a suitable voltage source as shown in
Fig. 3.21. If the polarities of the windings are as marked on the diagram, the voltmeter should read V = V1 ~
V2. If it reads (V1 + V2), the polarity markings of one of the windings must be interchanged.
The above method of polarity testing may not be convenient in field testing of a transformer. Alternatively
the polarity testing can be easily carried out by a dc battery, switch and dc voltmeter (permanent magnet type
which can determine the polarity of a voltage) as shown in the simple setup of Fig. 3.21(b). As the switch
on the primary side is closed, the primary current increases and so do the flux linkages of both the windings,
inducing emfs in them. The positive polarity of this induced emf in the primary is at the end to which the
battery pasitive is connected (as per Lenz’s law). The end of secondary which (simultaneously) acquires
positive polarity (as determined by the dc voltmeter) is the similar polarity end. The reverse happens when the
switch is opened, i.e. the similar polarity end of the secondary is that end which acquires negative potential.
A2
+
+
V1
V1
V2
–
–
S
–
A1
a1
V
i
V
(a)
Fig. 3.21
DC voltmeter
a2
+
(b)
(a) Polarity test on two-winding transformer
(b)
Open-circuit (OC) or No-load Test
The purpose of this test is to determine the shunt branch parameters of the equivalent circuit of the transformer
(Fig. 3.14(c)). One of the windings is connected to supply at rated voltage, while the other winding is kept
open-circuited. From the point of view of convenience and availability of supply the test is usually performed
from the LV side, while the HV side is kept open circuited as shown in Fig. 3.22. If the transformer is to be
used at voltage other than rated, the test should be carried out at that voltage. Metering is arranged to read.
voltage = V1; current = I0 and power input = P0
W
I0
+
AC supply
LV
A
HV
V
–
Fig. 3.22 Connection diagram for open-circuit test
(3.36)
74
Electric Machines
Figure 3.23(a) shows the equivalent circuit as
seen on open-circuit and its approximate version
in Fig. 3.23(b). Indeed the no-load current I0 is so
small (it is usually 2-6% of the rated current) and
R1 and X1 are also small, that V1 can be regarded
as = E1 by neglecting the series impedance. This
means that for all practical purposes the power
input on no-load equals the core (iron) loss i.e.,
I0
X1
R1
I0
+
+
V1
Gi
(a)
Fig. 3.23
P0 = Pi (iron-loss)
Ii
BmE1 ª V1 = E1Gi
–
–
+
Im
Bm Y0
–
(b)
Equivalent circuit as seen on open-circuit
(3.37)
The shunt branch parameters can easily be determined from the three readings (Eq. (3.36)) by the following
circuit computations and with reference to the no-load phasor diagram of Fig. 3.6.
Y0 = Gi – jBm
I0
V1
2
V 1 Gi = P0
Y0 =
Now
Gi =
or
It then follows that
Bm =
P0
V12
Y02 - Gi2
(3.38)
(3.39)
(3.40)
(3.41)
These values are referred to the side (usually LV) from which the test is conducted and could easily be
referred to the other side if so desired by the inverse square of transformation ratio. The transformation ratio
if not known can be determined by connecting a voltmeter on the HV side as well in the no-load test.
It is, therefore, seen that the OC test yields the values of core-loss and parameters of the shunt branch of
the equivalent circuit.
Short-circuit (SC) Test This test serves the purpose of determining the series parameters of a transformer.
For convenience of supply arrangement* and voltage and current to be handled, the test is usually conducted
from the HV side of the transformer, while the LV is short-circuited as shown in Fig. 3.24. The equivalent
circuit as seen from the HV under short-circuit conditions is drawn in Fig. 3.25(a). Since the transformer
resistances and leakage reactances are very small, the voltage VSC needed to circulate the full-load current
under short-circuit is as low as 5-8% of the rated voltage. As a result the exciting current I0 (SC) under these
* Voltage needed for the SC test is typically 5% of the rated value. For a 200 kVA, 440/6600- V transformer, test on
the HV side would require
200 ¥ 1000
6600 ¥ 5
= 30 A supply
= 330 V
and
6600
100
while if conducted from the LV side it would need
440 ¥ 5
200 ¥ 1000
= 22 V
and
= 445 A supply
100
440
Low-voltage, high-current supply needed for conducting the SC test from the LV side is much more difficult to
arrange than the supply required for the same test from the HV side.
Transformers
75
W
Isc
LV
HV
A
Low-voltage supply
(variable)
V
Fig. 3.24 Short-circuit test on transformer
conditions is only about 0.1 to 0.5% of the full-load current (I0 at the rated voltage is 2-6% of the full-load
current). Thus the shunt branch of the equivalent circuit can be altogether neglected giving the equivalent
circuit of Fig. 3.25(b).
While conducting the SC test, the supply voltage is gradually raised from zero till the transformer draws
full-load current. The meter readings under these conditions are:
voltage = VSC; current = ISC; power input = PSC
Isc
R1
X1
X2
+
Vsc
R2
+
I0(sc)
Gl
R
lsc
Bm
R1
X
X1
R2
X2
ª Vsc
–
–
(b)
(a)
Fig. 3.25
Equivalent circuit under short-circuit conditions
Since the transformer is excited at very low voltage, the iron-loss is negligible (that is why shunt branch is
left out), the power input corresponds only to the copper-loss, i.e.
PSC = Pc (copper-loss)
(3.42)
From the equivalent circuit for Fig. 3.25(b), the circuit parameters are computed as below:
Z=
Equivalent resistance,
R=
Equivalent reactance,
X=
VSC
=
I SC
R2 + X 2
(3.43)
PSC
(3.44)
Z 2 - R2
(3.45)
( I SC ) 2
These values are referred to the side (HV) from which the test is conducted. If desired, the values could
be easily referred to the other side.
76
Electric Machines
It is to be observed that the SC test has given us the equivalent resistance and reactance of the transformer;
it has not yielded any information for separating* these into respective primary and secondary values.
It was observed that OC and SC tests together give the parameters of the approximate equivalent circuit of
Fig. 3.16(a) which as already pointed out is quite accurate for all important computations.
EXAMPLE 3.7 The following data were obtained on a 20 kVA, 50 Hz, 2000/200 V distribution transformer:
Draw the approximate equivalent circuit of the transformer referred to the HV and LV sides respectively.
Table 3.1
Voltage
Current
Power
(V)
(A)
(W)
200
60
4
10
120
300
OC test with HV open-circuited
SC test with LV short-circuited
SOLUTION
OC test (LV side)
Y0 =
120
4
= 0.3 ¥ 10–2 = 2 ¥ 10–2 ; Gi =
200
(200) 2
Bm =
Y02 - Gi2 = 1.98 ¥ 10–2 Z=
300
60
=3W
=6W;R=
10
(10) 2
SC test (HV side)
X=
Transformation ratio,
Z 2 - R 2 = 5.2 W
NH
2000
=
= 10
NL
200
Equivalent circuit referred to the HV side:
Gi (HV) = 0.3 ¥ 10–2 ¥
Bm (HV) = 1.98 ¥ 10–2 ¥
1
(10) 2
= 0.3 ¥ 10–4 1
(10) 2
= 1.98 ¥ 10–4 The equivalent circuit is drawn in Fig. 3.26(a).
Equivalent circuit referred to the LV side:
R(LV) = 3 ¥
1
(10) 2
= 0.03 W
* Resistances could be separated out by making dc measurements on the primary and secondary and duly correcting
these for ac values. The reactances cannot be separated as such. Where required, these could be equally apportioned to the primary and secondary, i.e.
X1 = X2 (referred to anyone side)
This is sufficiently accurate for a well-designed transformer.
Transformers
0.3 ¥ 10 –2
0.3 ¥ 10 –4
V¢H
V¢L
lL
VL
(b) Referred to LV
(a) Referred to HV
Fig. 3.26
1
X(LV) = 5.2 ¥
W
lOL
W
VH
I¢L
W
W
IOH
0.03 W 0.052 W
I¢H
1.98 ¥ 10 –2
5.2 W
3W
1.98 ¥ 10 –4
IH
77
(10) 2
Equivalent circuit
= 0.052 W
The equivalent circuit is drawn in Fig. 3.26(b).
EXAMPLE 3.8 The parameters of the equivalent circuit of a 150-kVA, 2400/240-V transformer are:
R1 = 0.2 W
X1 = 0.45 W
Ri = 10 kW
R2 = 2 ¥ 10–3 W
X2 = 4.5 ¥ 10–3 W
Xm = 1.6 kW (as seen from 2400-V side)
Calculate:
(a) Open-circuit current, power and pf when LV is excited at rated voltage
(b) The voltage at which the HV should be excited to conduct a short-circuit test (LV shorted) with fullload current flowing. What is the input power and its pf?
SOLUTION
Note:
Ri =
1
,
Gi
Xm =
1
Bm
Ratio of transformation, a =
2400
= 10
240
(a) Referring the shunt parameters to LV side
Ri (LV) =
Xm (LV) =
10 ¥ 1000
(10) 2
1.6 ¥ 1000
(10) 2
= 100 W
= 16 W
I 0 (LV) = 240–0∞ - j 240–0∞
100
or
16
= 2.4 – j 15 = 15.2–– 80.9° A
I0 = 15.2 A, pf = cos 80.9° = 0.158 lagging
(b) LV shorted, HV excited, full-load current flowing: Shunt parameters can be ignored under this condition.
Equivalent series parameters referred to HV side:
R = 0.2 + 2 ¥ 10–3 ¥ (10)2 = 0.4 W
X = 0.45 + 4.5 ¥ 10–3 ¥ (10)2 = 0.9 W
Z = 0.4 + j 0.9 = 0.985 –66° W
78
Electric Machines
I fl (HV) =
150 ¥ 1000
= 62.5 A
2400
VSC (HV) = 62.5 ¥ 0.958 = 59.9 V or 60 V (say)
PSC = (62.5)2 ¥ 0.4 = 1.56 kW
pfSC = cos 66° = 0.406 lagging
Sumpner’s (Back-to-Back) Test
While OC and SC tests on a transformer yield its equivalent circuit parameters, these cannot be used for the
‘heat run’ test wherein the purpose is to determine the steady temperature rise if the transformer was fully
loaded continuously; this is so because under each of these tests the power loss to which the transformer is
subjected is either the core-loss or copper-loss but not both. The way out of this impasse without conducting
an actual loading test is the Sumpner’s test which can only be conducted simultaneously on two identical
transformers*.
In conducting the Sumpner’s test the primaries of the two transformers are connected in parallel across
the rated voltage supply (V1), while the two secondaries are connected in phase opposition as shown in
Fig. 3.27. For the secondaries to be in phase opposition, the voltage across T2 T4 must be zero otherwise it will
be double the rated secondary voltage in which case the polarity of one of the secondaries must be reversed.
Current at low voltage (V2) is injected into the secondary circuit at T2T4. The supply (1) and supply (2) are
from the same mains.
W1
2l0
A1
+
T3
T1
AC supply (1)
V1
T2
–
T4
lfl
V2
A2
W2
Low
Voltage supply (2)
Fig. 3.27 Sumpner’s test on two identical single-phase transformers
* In very large sizes two identical transformers may not be available as these are custom-built.
Transformers
79
As per the superposition theorem, if V2 source is assumed shorted, the two transformers appear in opencircuit to source V1 as their secondaries are in phase opposition and therefore no current can flow in them.
The current drawn from source V1 is thus 2I0 (twice the no-load current of each transformer) and power is 2P0
(= 2Pi, twice the core-loss of each transformer). When the ac supply (1) terminals are shorted, the transformers
are series-connected across V2 supply (2) and are short-circuited on the side of primaries. Therefore, the
impedance seen at V2 is 2Z and when V2 is adjusted to circulate full-load current (I fl), the power fed in is 2Pc
(twice the full-load copper-loss of each transformer). Thus in the Sumpner’s test while the transformers are
not supplying any load, full iron-loss occurs in their cores and full copper-loss occurs in their windings; net
power input to the transformers being (2P0 + 2Pc). The heat run test could, therefore, be conducted on the two
transformers, while only losses are supplied.
In Fig. 3.27 the auxiliary voltage source is included in the circuit of secondaries; the test could also be
conducted by including the auxiliary source in the circuit of primaries.
The procedure to connect a 3-phase transformer for the back-to-back test will be explained in Sec. 3.12.
EXAMPLE 3.9 Two transformers of 20 kVA each with turn-ratios respectively of 250 : 1000 and 250 :
1025 are connected in back-to-back test; the two primaries being fed from a 250 V supply and secondaries
being connected in phase opposition. A booster transformer connected on primary side to the same 250 V
supply is used to inject voltage in the circuit of secondaries such as to circulate a current of 20 A. The
core losses of each transformer are 350 W and each transformer has a reactance 2.5 times its resistance.
Calculate the possible readings of the wattmeter connected to measure the input to the primaries.
SOLUTION Using the principle of superposition the currents on the primary side are first found, caused by the
circulating current in the secondaries with the primary voltage source shorted; the voltage injected on the secondary side
being intact. The primary currents necessary to balance the secondary circulating current are shown in Fig. 3.28; these
being in phase with each other. The difference of these currents is 2 A which flows in the lines connecting the primaries
to the main (refer to figure). This current has a power factor of
cos tan–1 2.5 = 0.371
Therefore, the power exchanged with the mains by this current is
250 ¥ 2 ¥ 0.371 = 185.5 W
This power will be drawn from or fed into the mains depending on the polarity of the injected voltage.
Mains
250 V
2A
250:1000
250: 1025
20 ¥ 1025
= 82 A
250
20 ¥ 1000
= 80 A
250
20 A
Fig. 3.28
80
Electric Machines
Consider now the currents owing to the voltage source connected to the primaries with the secondary injected voltage
source shorted. The primaries now draw the magnetizing currents from the mains with associated core-loss of both
the transformers equal to 2 ¥ 350 = 700 W. The currents in the secondaries because of the small voltage unbalance
(transformers have a slightly different turn ratio) would be small with very little associated loss.*
Hence power drawn from the mains is
700 ± 185.5 = 885.5 W or 514.5 W
3.8 THE PER UNIT SYSTEM
While carrying out the analysis of electrical machines (or electrical machine systems), it is usual to express
voltage, current, VA and impedance in per unit (or percentage**) of the base or reference values of these
quantities. The per unit (pu) value of any quantity is defined as the ratio of:
The actual value in any units
The base or reference value in the same units
While the base values can be selected arbitrarily, it is normal to choose the rated value of the device as its
base values.
There are two important advantages that accrue from the use of the pu system. First, the parameters of
transformers as well as rotating machines lie roughly in the same range of numerical values irrespective of
their ratings if expressed in per unit of their ratings; correctness of analysis becomes immediately obvious in
this system.
Second, the pu system is most convenient in power systems as it relieves the analyst of the need to refer
circuit quantities to one or other side of the transformers. It is a universal practice to use the pu system for
modelling real-life large integrated power systems and in computer simulation of machine systems for their
transient and dynamic analysis.
Base values of various quantities are related to each other by the usual electrical laws. For a single-phase
system,
Pbase, Qbase, (VA)base = Vbase /Ibase
(3.46)
Rbase, Xbase, Zbase = Vbase/Ibase
(3.47)
Gbase, Bbase, Ybase = Ibase/Vbase
(3.48)
Always, (VA)base and Vbase are first selected and their choice automatically fixes the other base values as
per Eqs (3.46)-(3.48). It immediately follows from these equations that
ZB =
VB2
(VA) B
* The unbalanced voltage in secondary circuit because of unequal turn ratio is 25 V, while the rated secondary voltage is 1000 V. Assuming a high voltage side transformer impedance of 40 W, the circulating current caused by the
unbalanced voltage would be
25
= 0.3125 A
2 ¥ 40
The ratio of losses because of this current to the losses caused by the 20A current is
Ê 0.3125 ˆ
ÁË 20 ˜¯
2
= 2.44 ¥ 10–4
This being of negligible order, there is no error of consequence in neglecting the effect of unequal turn ratio.
** Per cent values are not preferred as a factor of 100 has to be carried.
Transformers
Then
81
Z(pu) =
Z (W) ¥ (VA) B
Z(pu) =
Z (W) ¥ (kVA) B
(3.50a)
Z(pu) =
Z (W) ¥ (MVA) B
(3.50b)
(3.49)
VB2
In large devices and systems it is more practical to express the bases in kVA/ MVA and kV. Then Eq. (3.49)
is written as
or
1000 (kV) 2B
(kV) 2B
When (MVA)B and (kV)B are modified, the new pu impedance is given by
Z(pu)new = Z(pu)old
(MVA) B ,new
(MVA) B , old
¥
(kV) 2B ,old
(kV) 2B , new
(3.51)
It can be easily shown that in a transformer equivalent circuit using pu notations, the need for an ideal
transformer is eliminated because the pu impedance of a transformer is the same whether computed from
the primary or secondary side so long as the voltage bases on the two sides are selected in the ratio of
transformation (see Example 3.10). A procedure universally adopted is to translate all quantities to pu values
for carrying out analysis and to convert the results obtained back to actual units.
As has been mentioned earlier the pu parameters of transformers (and electric machines) lie within a
narrow range. For example, the magnetizing current normally lies between 0.02 and 0.05 pu, the equivalent
resistance between 0.005 pu (large transformers) and 0.02 pu (small transformers) and the equivalent
reactance usually varies from 0.05 (large) to 0.1 pu (small transformers). This information helps a great deal
in comparing units of a given size made by various manufacturers.
In the 3-phase system, the bases are chosen as
(MVA)B = 3-phase MVA
(kV)B = line-to-line kV
Assuming star connection (equivalent star can always be found),
ZB =
Then
Z(pu) =
((kV) B / 3 ) 2
(kV) 2B
=
1
(MVA) B
(MVA) B
3
Z (W) ¥ (MVA) B
(kV) 2B
(3.52)
which relationship is indeed the same as Eq. (3.50) for the single-phase system.
Consider three impedances, Z each, connected in delta. Then with 3-phase (MVA)B and line-to-line (kV)B,
ZB(D) =
Therefore
Z(pu) =
(kV) 2B
3 (kV) 2B
=
(MVA) B / 3
(MVA) B
( Z / 3) (MVA) B
(kV) 2B
Since Z/3 is the equivalent star impedance, the pu impedance for delta or its equivalent star is the same for
a given 3-phase MVA base and line-to-line kV base.
82 Electric Machines
EXAMPLE 3.10 The exciting current was found to be 3 A when measured on the LV side of a 20-kVA,
2000/200- V transformer. Its equivalent Impedance (referred to the HV side) is 8.2 + j 10.2 W. Choose the
transformer rating as the base.
(a) Find the exciting current in pu on the LV as well as HV side.
(b) Express the equivalent impedance in pu on the LV as well as HV side.
SOLUTION
VB (HV) = 2000 V
IB(HV) = 10 A
VB(LV) = 200 V
IB(LV) = 100 A
2000
= 200 W
10
200
2W
ZB(LV) =
100
3
= 0.03 pu
I0 (LV) =
100
ZB(HV) =
(a)
The exciting current referred to the HV side is 0.3 A
0.3
I0 (HV) =
= 0.03 pu
10
8.2 + j 10.2
= 0.041 + j 0.051
(b)
Z(HV)(pu) =
200
8.2 + j 10.2
Z(LV) =
= 0.082 + j 0.102
(10) 2
Z(LV)(pu) =
0.082 + j 0.102
= 0.041 + j 0.051
2
Remark The earlier remark is, therefore, confirmed that pu values referred to either side of the transformer
are the same so long as voltage bases on the two sides are in the ratio of transformation of the transformer.
3.9
EFFICIENCY AND VOLTAGE REGULATION
Power and distribution transformers are designed to operate under conditions of constant rms voltage and
frequency and so the efficiency and voltage regulation are of prime importance.
The rated capacity of a transformer is defined as the product of rated voltage and full load current on the
output side. The power output depends upon the power factor of the load.
The efficiency h of a transformer is defined as the ratio of the useful power output to the input power. Thus
h=
output
input
(3.53)
The efficiency of a transformer is in the range of 96–99%. It is of no use trying to determine it by measuring
the output and input under load conditions because the wattmeter readings are liable to have an error of 1–2%.
Transformers
83
The best and accurate method of determining efficiency would be to find the losses from the OC and SC tests.
With this data efficiency can then be calculated as
output
losses
= 1(3.54)
output + losses
output + losses
In Eq. (3.54) the effect of meter readings is confined to losses only so that the overall efficiency as obtained
from it is far more accurate than that obtained by direct loading.
Various losses in a transformer have already been enumerated in Sec. 3.6 and the two important losses
(iron-loss Pi and copper-loss Pc) are shown in Fig. 3.29 of a loaded transformer.
h=
l1
l2
+
Pinput (Pin)
V1
Load
+
V2
Poutput (P0)
–
–
Piron (Pi)
Pcopper (Pc)
Fig. 3.29(a) Transformer on load
The iron (core) losses depend upon the flux density and so on the induced emf. As E1 ª V1 at all loads, these
losses can be regarded as constant (independent of load) for constant primary voltage.
Copper losses in the two windings are
Pc = I 21 R1 + I 22 R2
= I 22 Req(2)
where Req(2) = equivalent resistance referred to the secondary side. Thus it is found that copper losses vary
as the square of the load current.
Transformer output, P0 = V2 I2 cos q2
cos q2 = load pf
From Eq. (3.54)
h=
V2 I 2 cos q 2
V2 I 2 cos q 2 + Pi + I 22 Req (2)
(3.55)
where V2 is the rated secondary voltage. It varies slightly with the load but the variation is so small (about
3-5%) that it can be neglected for computing efficiency.
Equation (3.55) shows that for a given power factor, efficiency varies with load current. It can be written as
h=
V2 cos q 2
ÊP
ˆ
V2 cos q 2 + Á i + I 2 Req (2)˜
Ë I2
¯
(3.56)
84 Electric Machines
For maximum value of h for given cos q2 (pf), the denominator of Eq. (3.56) must have the least value. The
condition for maximum h, obtained by differentiating the denominator and equating it to zero, is
I 22 Req (2) = Pi
or
(3.57)
Copper-loss (variable) = core-loss (constant)
It means that the efficiency is maximum at a load when the copper-loss (variable loss) equals the core-loss
(constant loss). Thus for maximum efficiency,
I 22 =
Dividing by I 22fl on both sides
Ê I2 ˆ
Á
˜
Ë I 2 fl ¯
2
=
Pi
Req (2)
Pi
2
I 2 f l Req (2)
Pi
Pc ( fl )
or
k=
where
k = I2/I2fl
and
= k2
(3.58)
(3.59a)
Pc( fl) = full-load copper-loss
Thus the efficiency is maximum at a fractional load current given by Eq. (3.59a). Multiplying the numerator
and denominator of Eq. (3.59a) by rated V2
k=
V2 I 2
V2 I 2 fl
(3.59b)
Thus the maximum efficiency is given at a load k(V2I2fl) or kS2, where S2, is VA (or kVA) rating of the
transformer. The expression for maximum efficiency is given by
kS2 cos q 2
(3.60)
kS2 cos q 2 + 2 Pi
It can be easily observed from Eq. (3.60) that hmax increases with increasing pf (cos q 2) and is the highest
at unity pf. Also hmax = 0 when cos q = 0, i.e. at zero pf (purely reactive load). Therefore, knowledge of
transformer losses is as important as its efficiency.
Efficiency
Power transformers used for bulk power
transmission are operated near about full load 100
at all times and are therefore designed to have
PF = 1.00
P.F = 0.80
maximum efficiency at full-load. On the other
P.F = 0.60
hand, the distribution transformers supply load
which varies over the day through a wide range.
Such transformers are, therefore, designed to
50
have maximum efficiency at about three-fourths
the full load.
Normally transformer efficiency is maximum
when the load power factor is unity. From
Load
Fig. 3.29(b), it is seen that the maximum efficiency
I2 (rated)
current
occurs at same load current independent of power
hmax =
Fig. 3.29(b)
Transformers
85
factor, because the total core loss Pc, and equivalent resistance Req(2) are not affected by load power factor. Any
way reduction of load power factor reduces the transformer output and the transformer efficiency also reduced.
The all-day efficiency of a transformer is the ratio of the total energy output (kWh) in a 24-h day to the
total energy input in the same time. Since the core losses are constant independent of the load, the all-day
efficiency of a transformer is dependent upon the load cycle; but no prediction can be made on the basis of
the load factor (average load/peak load). It is an important figure of merit for distribution transformers which
feed daily load cycle varying over a wide load range. Higher energy efficiencies are achieved by designing
distribution transformers to yield maximum (power) efficiency at less than full load (usually about 70% of
the full load). This is achieved by restricting the core flux density to lower values by using a relatively larger
core cross-section. (It means a larger iron/copper weight ratio.)
EXAMPLE 3.11 For the transformer of Example 3.7 calculate the efficiency if the LV side is loaded fully
at 0.8 power factor. What is the maximum efficiency of the transformer at this power factor and at what pu
load would it be achieved?
SOLUTION
Power output = V2 I2 cos q2
= 200 ¥ 100 ¥ 0.8 = 16000 W (independent of lag/lead)
Total loss PL = Pi + k 2Pc
= 120 + 1 ¥ 300 = 420 W
PL
h =1–
P0 + PL
420
=1–
= 97 44%
16000 + 420
\
For maximum efficiency
Pi
=
Pc
120
= 0.632
300
i.e. at 0.632 pu load (this is independent of power factor). Now
2 Pi
hmax (cos q2 = 0.8) = 1 –
P0 + 2 Pi
k=
2 ¥ 120
16000 ¥ 0.632 + 2 ¥ 120
= 97.68%
=1–
EXAMPLE 3.12 A 500 kVA transformer has an efficiency of 95% at full load and also at 60% of full
load; both at upf .
(a) Separate out the losses of the transformer.
(b) Determine the efficiency of the transformer at 3/4th full load.
SOLUTION
(a)
h=
500 ¥ 1
= 0.95
500 ¥ 1 + Pi + Pc
(i)
86
Electric Machines
500 ¥ 0.6
Also
500 ¥ 0.6 + Pi + (0.6) 2 Pc
= 0.95
(ii)
Solving Eqs (i) and (ii) we get
Pi = 9.87 kW
Pc = 16.45 kW
(b) At 3/4th full load upf
500 ¥ 0.75
h=
500 ¥ 0.75 + 9.87 + (0.75) 2 ¥ 16.45
= 95.14%
EXAMPLE 3.13 A transformer has its maximum efficiency of 0.98 at 15 kVA at upf. Compare its all-day
efficiencies for the following load cycles:
(a) Full load of 20 kVA 12 hours/day and no-load rest of the day.
(b) Full load 4 hours/day and 0.4 full-load rest of the day.
Assume the load to operate on upf all day.
SOLUTION
hmax =
P0
P0 + 2 Pi
or
Pi = 0.153 kW
Now
k2 =
P0
Time, h
Pc = 0.272 kW
W0
20
12
240
0
12
or
(a)
Pi
Pc
or
Ê 15 ˆ
ÁË 20 ˜¯
or
0
0.98 =
2
=
15
15 + 2 Pi
0.153
Pc
Pin = P0 + Pi + k 2Pc
Win
20 + 0.153 + 0.272 = 20.425
245.1
0 + 0.153 = 0.153
1.8
246.9 kWh
240 kWh
hallday =
(b)
SW0
240
=
= 97.2%
SWin
246.9
P0
Time, h
W0
Pin = P0 + Pi + k 2Pc
20
4
80
20 + 0.153 + 0.272 = 20.425
8
20
160
240 kWh
Ê 8ˆ
8 + 0.153 + Á ˜
Ë 20 ¯
hallday =
2
Win
¥ 0.272 = 8.196
81.7
163.9
245.6 kWh
240
= 97.7%
245.6
Remark
Even though the load factor is the same in each case [(240/24)/20 = 0.5] the all day efficiencies still differ
because of the difference in the nature of the two load cycles.
Transformers
87
Voltage Regulation
Constant voltage is the requirement of most domestic, commercial and industrial loads. It is, therefore,
necessary that the output voltage of a transformer must stay within narrow limits as the load and its power
factor vary. This requirement is more stringent in distribution transformers as these directly feed the load
centres. The voltage drop in a transformer on load is chiefly determined by its leakage reactance which must
be kept as low as design and manufacturing techniques would permit.
The figure of merit which determines the voltage drop characteristic of a transformer is the voltage
regulation. It is defined as the change in magnitude of the secondary (terminal) voltage, when full-load (rated
load) of specified power factor supplied at rated voltage is thrown off, i.e. reduced to no-load with primary
voltage (and frequency) held constant, as percentage of the rated load terminal voltage. In terms of symbols
% Voltage regulation =
where
V20 - V2. f l
V2. f l
¥ 100
(3.61)
V2, fl = rated secondary voltage while supplying full load at specified power factor
V20 = secondary voltage when load is thrown off.
Figure 3.30(a) shows the transformer equivalent circuit* referred to the secondary side and Fig. 3.30(b)
gives its phasor diagram. The voltage drops IR and IX are very small in a well-designed transformer (refer
Example 3.6). As a result the angle d between V1 and V2 is of negligible order, so that**
V1 ª OE
V1 – V2 ª BE = I(R cos f + X sin f); f lagging
= I(R cos f – X sin f); f leading
(3.62a)
(3.62b)
A
V1
+
V1
–
X
l
IX
+
V2
Load
R
O
–
(a) Equivalent circuit referred to secondary
f
f
B
d
V2
E
f IR
I
(b) Phasor diagram (not proportional)
Fig. 3.30
When the load is thrown off
\
V20 = V1
V20 – V2 = I(R cos f ± sin f)
(3.63)
* In approximate equivalent circuit, the magnetizing shunt branch plays no role in determining voltages and hence
is left out.
** V1 could be calculated from the phasor equation
V1 = V2 + I –f (R + jX)
but the approximate method is very much quicker and quite accurate (see Example 3.6).
88 Electric Machines
where I is the full-load secondary current and V2, the full-load secondary voltage (equal to the value of V2
(rated)). Thus
% Voltage regulation, Reg =
I ( R cos f ± X sin f )
¥ 100
V2
IR
IX
= R(pu) and
= X(pu)
V2
V2
=
(3.64)
Per unit voltage regulation = R(pu) cos f ± X(pu) sin f
(3.65)
Recognizing that
We have
V20 - V2
¥ 100
V2
It is seen from Eq. (3.64) that the voltage regulation varies with power factor and has a maximum value
when
or
or
d Reg
= 0 = –R sin f + X cos f
df
X
tan f =
R
R
cos f =
; lagging
2
R + X2
% Reg
5
4
(3.66)
3
Equation (3.66) implies that voltage regulation is
the maximum when the load power factor (lagging)
angle has the same value as the angle of the
equivalent impedance. From Eq. (3.64), the voltage
regulation is zero when
2
1
Leading pf
0
0
0.2
0.4
0.6
0.8
R cos f – X sin f = 0
1 0.8
–1
0.6
0.4
0.2
0
Lagging pf
–2
R
; leading
(3.67)
X
–3
For f (leading) larger than that given by
Eq. (3.67), the voltage regulation is negative (i.e.
–4
the secondary full-load voltage is more than the no–5
load voltage).
The complete variation of % regulation with Fig. 3.31 Percentage regulation versus power factor; R =
power factor is shown in Fig. 3.31.
or
tan f =
EXAMPLE 3.14
Consider the transformer with data given in Example 3.7.
(a) With full-load on the LV side at rated voltage, calculate the excitation voltage on the HV side. The
load power factor is (i) 0.8 lagging, (ii) 0.8 leading. What is the voltage regulation of the transformer
in each case?
(b) The transformer supplies full-load current at 0.8 lagging power factor with 2000 V on the HV side.
Find the voltage at the load terminals and the operating efficiency.
Transformers
SOLUTION
(a) The HV side equivalent circuit of Fig. 3.26(a) will be used.
VL = 200 V,
V L¢ = 2000 V,
Now
(i)
200 ¥ 1000
= 100 A
200
I L¢ = 10 A
IL =
VH = V L¢ + I¢L (RH cos f ± XH sin f);
RH =: 30 W
XH = 5.2 W
cos f = 0.8 lagging, sin f = 0.6
VH = 2000 + 10(3 ¥ 0.8 + 5.2 ¥ 0.6) = 2055.2 V
2055.2 - 2000
¥ 100 = 2.76%
2000
cos f = 0.8 leading, sin f = 0.6
Voltage regulation =
(ii)
VH = 2000 + 10 (3 ¥ 0.8 – 5.2 ¥ 0.6) = 1992.8 V
Voltage regulation =
1992.8 - 2000
¥ 100 = –0.36%
2000
(b) IL(full-load)= 100 A, 0.8 lagging pf
or
V¢L = V H¢ – I¢L (RH cos f + XH sin f)
= 2000 – 10(3 ¥ 0.8 + 5.2 ¥ 0.6) = 1944.8 V
VL = 194.48 V
Efficiency
\
Output, P0 = VL IL cos f
= 194.48 ¥ 100 ¥ 0.8 = 15558.4 W
PLOSS = Pi + Pc
Pi = 120 W (Ex. 3.7)
Pc = (10)2 ¥ 3 = 300 W
PLOSS = 420 W
420
h =1–
= 97.38%
15558.4 ¥ 420
Example 3.14 is solved by writing the following MATLAB code.
clc
clear
S=20*1000;
V1=200;
V2=2000;
I1=S/V1;
I2=S/V2;
RH=3;
XH=5.2;
Cosine-phi=0.8;
Sin-phi=0.6;
VH=V2+I2*(RH*cosine-phi+XH*sin-phi)
Vreg=(VH-V2)*100/V2
89
90 Electric Machines
%% case2
VH=V2+I2*(RH*cosine-phi-XH*sin-phi)
Vreg=(VH-V2)*100/V2
Il=100;
Vll=V2-I2*(RH*cosine-phi+XH*sin-phi);
Vl=Vll/10
Ploss=120+10*10*3;
Pop=Vl*Il*cosine-phi;
eff=(1-(Ploss/(Ploss+Pop)))*100
Answer:
VH = 2.0552e+003
Vreg = 2.7600
VH = 1.9928e+003
Vreg = –0.3600
Vl = 194.4800
eff = 97.3715
EXAMPLE 3.15 For the 150 kVA, 2400/240 V transformer whose circuit parameters are given in
Example 3.8, draw the circuit model as seen from the HV side. Determine therefrom the voltage regulation
and efficiency when the transformer is supplying full load at 0.8 lagging pf on the secondary side at rated
voltage. Under these conditions calculate also the HV side current and its pf.
SOLUTION
R(HV) = 0.2 + 2 ¥ 10–3 ¥ (10)2 = 0.4 W
X(HV) = 0.45 + 4.5 ¥ 10–3 ¥ (10)2 = 0.9 W
The circuit model is drawn in Fig. 3.32.
150 ¥ 1000
= 625 A, 0.8 pf lagging
240
V2 = 240 V
625
+
I2 =
= 62.5 A, 0.8 pf lagging
10
V 2¢ = 2400 V
I2( fl) =
= 53.75 V
53.75
¥ 100 = 2.24%
Voltage regulation =
2400
V1 = 2400 + 53.75 = 2453.75 = 2454 V
P(out) = 150 ¥ 0.8 = 120 kW
Pc(copper loss) = (62.5)2 ¥ 0.4 = 1.56 kW
Pi(core loss) =
(2454) 2
= 0.60 kW
10 ¥ 1000
PL = Pi + Pc = 0.60 + 1.56 = 2.16 kW
0.4 W
0.9 W
I¢2
+
I0
V1
10 kW
Voltage drop = 62.5(0.4 ¥ 0.8 + 0.9 ¥ 0.6)
I1
1.6 kW
–
V¢2
–
Fig. 3.32
Transformers
h=
91
120
= 98.2%
120 ¥ 2.16
2454 –0∞
2454 –0∞
- j
10 ¥ 1000
1.6 ¥ 1000
= 0.245 – j 1.53 A
I 2¢ = 62.5 (0.8 – j 0.6) = 50 – j 37.5 A
I1 = I 0 + I 2 = 50.25 – j 39.03
= 63.63 ––37.8° A
I1 = 63.63 A, pf = 0.79 lagging
I0 =
or
3.10
EXCITATION PHENOMENON IN TRANSFORMERS
It was stated in Sec. 3.3 that the no-load current in a transformer is nonsinusoidal. The basic cause for
this phenomenon, which lies in hysteresis and saturation non-linearities of the core material, will now be
investigated; this can only be accomplished graphically.
Assume that the voltage v1 applied to the transformer of Fig. 3.5 is sinusoidal. Since the ohmic drop (r1 i 0)
is assumed negligible compared to the magnitude of the applied voltage, the induced emf which balances
the applied voltage must also be sinusoidal and so must be the flux established in the core (see Eqs (3.3)
and (3.4)). Further, the flux must lag the induced emf by 90° as shown in the emf and flux waveforms drawn
in Fig. 3.33. The current necessary to set up sinusoidal flux can be obtained graphically by looking up the
hysteresis curve (f-i0 curve) also drawn in Fig. 3.33.
fmax
90°
e0
f
i02
– f1
t0
t1
t2
fmax
f2
f2
f1
i0
i03
i01
f
l0,max
f1
t3
t
i03 i
02
i0 max
i0
– f1 i01
a0
fmax
– fmax
Fig. 3.33
Assume that the steady-state operation has been reached so that hysteresis loop of Fig. 3.33 is being
repeated in successive cycles of the applied voltage. Consider the instant when the flux has a value – f1, the
corresponding exciting current being zero. When the flux becomes zero (at time instant t1), the current is a
small positive value i01. When the flux has a positive value f2 as shown in the figure, there are two possible
values of current, i02 when the flux is on the increasing part of the hysteresis loop and i03 when the flux is
on the decreasing part of the loop; i02 > i03. The flux maximum + fmax coincides with the current maximum
92
Electric Machines
+ i0 max. The current becomes zero once again for flux + f1. So far the positive half of exciting current has
been traced out; the negative half will be symmetrical (odd symmetry) to it because of the inherent symmetry
of the magnetic hysteresis loop. The complete cycle of the exciting current is sketched in Fig. 3.33.
From the exciting current wave shape of Fig. 3.33, it is observed that it is nonsinusoidal and peaky *.
While odd symmetry is preserved and the current and flux maximas occur simultaneously, the current zeros
are advanced** in time with respect to the flux wave shape. As a consequence the current has fundamental
and odd harmonics, the strongest being the third harmonic which can be as large as 40% of the fundamental.
Further, the fundamental of the exciting current leads the flux by a small angle a0 (also refer Fig. 3.6); so that
the current fundamental has a component in phase with flux (Im of Fig. 3.6) and a much smaller component
in quadrature to the flux (leading) or in phase with voltage (Ii of Fig. 3.6). While Im is responsible for creation
of core flux, Ii accounts for the power lost in the core due to hysteresis.
Current Ii must of course be modified to account for the eddy-current loss in the core. The corresponding
current component apart from being in phase with V1 is sinusoidal in nature as it balances the effect of
sinusoidal eddy-currents caused by the sinusoidal core flux. It is, therefore, seen that eddy-currents do not
introduce any harmonics in the exciting current.
When the transformer feeds current to a linear load, the load current is sinusoidal and being much larger
than the excitation current would ‘swamp out’ the nonsinusoidalness in the resultant primary current; as a
consequence the primary current on load is sinusoidal for all practical purposes.
In certain 3-phase transformer connections, third-harmonic current cannot flow (Sec. 3.12), as a result the
magnetizing current im is almost sinusoidal. To satisfy the B-H curve, the core flux must then be nonsinusoidal;
it is a flat-topped wave. This can be verified by assuming a sinusoidal im and then finding out the f wave shape
from the f-im relationship, the normal magnetizing curve†. Since the flux is flattopped, the emf which is its
derivative will now be peaky with a strong third-harmonic content. The various waveforms are illustrated in
Fig. 3.34.
im
Fundamental
f
Third-harmonic
e
Fundamental
Third-harmonic
Fig. 3.34 Case of sinusoidal magnetizing current
In the discussion above steady-state operation was assumed so that v1 and f are both sinusoidal, f lagging
v1 by 90° as shown once again in Fig. 3.35(a). The f-i0 relationship is shown in Fig. 3.35(b). The normal
* Peakiness in exciting current is due to saturation phenomenon and would be present even if hysteresis were absent.
** This shift in phase is contributed by the hysteric nature of f-i0 curve.
† Hysteresis contributes ii in phase with v1 which is not being considered here.
Transformers
93
exciting current under these conditions is about 0.05 pu if the transformer is designed with Bmax about 1.4 T.
When the voltage v1 is switched on to the transformer, the core flux and the corresponding exciting current
undergo a transient before reaching steady-state values. The severity of the switching transient is related to
the instant when the voltage wave is switched on; the worst conditions being when the applied voltage has
zero value at the instant of switching as shown in Fig. 3.35(c). It is assumed here that the initial flux in the
transformer at the instant of switching has zero value. It is seen from this figure that the steady-state value of
flux demanded at this instant is – fm, while the flux can only start with zero value (in the inductive circuit).
As a consequence, a transient flux component (off-set flux) ft = fm originates so that the resultant flux is
(ft + fss) which has zero value at the instant of switching. The transient component ft will decay according
to the circuit time constant (L/R) which is generally low in a transformer. If the circuit dissipation (core-loss)
is assumed negligible, the flux transient will go through a maximum value of 2fm, a phenomenon called
f
v1m
fm
fm
t
i0
i0, max
fss
v1
(a)
v
(b)
2fm
f
2fm
ft = fm
fm
t
i0,max i0
– fm
fss
(d) Low hysteresis loss
(c)
2fm + fr
fm + fr
f
2fm + fr
ft = fr + fm
fr
fr
t
i0,max i0
– fm
fss
(e)
(f) Very low hysteresis loss
Fig. 3.35 Transformer inrush current
94
Electric Machines
doubling effect. The corresponding exciting current will be very large as the core goes into deep saturation
region of magnetization (Bm = 2 ¥ 1.4 = 2.8 T); it may indeed be as large as 100 times the normal exciting
current, i.e, 5 pu (normal exciting current being 0,05 pu) producing electromagnetic forces 25 times the
normal. This is why the windings of large transformers must be strongly braced. In subsequent half-periods
ft gradually decays till it vanishes and the core flux acquires the steady-state value, Because of the low
time constant of the transformer circuit, distortion effects of the transient may last several seconds. The
transformer switching transient is referred to as the inrush current.
The initial core flux will not be zero as assumed above but will have some residual value f r because of
retentivity. As shown in Figs. 3.35(e) and (f ), the transient will now be even more severe; ft = fm + fr and the
core flux will now go through a maximum value of (2fm + fr).
It is observed from Figs. 3.35(c) and (e) that the offset flux is unidirectional so that the transient flux and
exciting current are unidirectional in the initial stage of the transient. A typical oscillogram of the inrush
current is shown in Fig. 3.36.
Normal
magnetizing
current
Time
Fig. 3.36
Inrush current wave shape
3.11 AUTOTRANSFORMERS
So far two-winding transformers have been discussed wherein the windings are electrically isolated. When
the primary and secondary windings are electrically connected so that a part of the winding is common to
the two as shown in Fig. 3.36 (core is not shown here), the transformer is known as an autotransformer.
Such a transformer is particularly economical where the
l1
A
voltage ratio is less than 2 in which case electrical isolation +
of the two windings is not essential. The major applications
R1
l1
are induction motor starters, interconnection of HV systems at
X1
B l2
voltage levels with ratio less than 2, and in obtaining variable
V1 N1
+
voltage power supplies (low voltage and current levels). The
R
2
N2
autotransformer has lower reactance, lower losses, smaller
(l2 – l1) V2
X2
exciting current and better voltage regulation compared to its
–
two-winding counterpart, All this is on account of the fact that –
C
in an autotransformer a part of the energy transfer is through
Fig.
3.37
Autotransformer
the conduction process.
Transformers
95
Figure 3.37 shows a single-phase autotransformer having N1 turns primary with N2 turns tapped for a
lower voltage secondary. The winding section BC of N2 turns is common to both primary and secondary
circuits. In fact it is nothing but a conventional two-winding transformer connected in a special way. The
winding section AB must be provided with extra insulation, being at higher voltage.
It will be assumed here that the magnetizing current is negligible; but it can easily be determined by a noload test and accounted for.
With reference to Fig. 3.37 the two-winding voltage and turn-ratio is
V1 - V2
N1 - N 2
=
; N1 > N2
V2
N2
As an autotransformer its voltage and turn-ratio is
a=
a¢ =
V1
N1
=
>1
V2
N2
(3.68)
(3.69)
It is easy to see that Eqs (3.68) and (3.69) are related as
a¢ = 1 + a
(3.70)
Visualizing that in Fig. 3.37 a two-winding transformer is connected as an autotransformer, let us compare
the VA ratings of the two. As a two-winding transformer
(VA)TW = (V1 – V2)I1 = (I2 – I1)V2
(3.71)
When used as an autotransformer
(VA)auto = V1 I1 = V2 I2
(3.72)
Equation (3.71) can be written as
Ê
V2 ˆ
(VA)TW = Á1 - ˜ (V1 I1) =
V1 ¯
Ë
or
Ê
N2 ˆ
ÁË1 - N ˜¯ (VA)auto
1
È
˘
1
(VA)auto = Í
˙ (VA)TW; a¢ = N1/N2 > 1
Î1 - (1 / a ¢ ) ˚
(3.73)
(VA)auto > (VA)TW
(3.74)
It immediately follows that
It is therefore seen that a two-winding transformer of a given VA rating when connected as an
autotransformer can handle higher VA. This is because in the autotransformer connection (Fig. 3.37) part of
the VA is transferred conductively. It is also noted from Eq. (3.74) as a¢ = N1/N2 (the autotransformation ratio)
approaches unity,
(VA)auto >> (VA)TW
(3.75)
It is for this reason that autotransformer is commonly used when turn-ratio needed is 2 or less, like
in interconnecting two high-voltage systems at different voltage levels. For low voltage, low VA rating
autotransformer is used to obtain a variable voltage supply for testing purposes. Here a¢ = N1/N2 is varied by
changing the N2-tap.
It will also be shown in the example that follows that an autotransformer compared to its two-winding
counterpart has a higher operating efficiency.
96 Electric Machines
Let us see the problem from the design point of view by comparing winding copper needed for a given
voltage ratio and VA rating for a two-winding transformer and an autotransformer. Assuming constant
conductor current density, we can write
I1 ( N1 - N 2 ) + ( I 2 - I1 ) N 2
Gauto
=
I1 N1 + I 2 N 2
GTW
2 I1 N 2
(∵ I1 N1 = I2 N2)
2 I1 N1
N
V
=1– 2 =1– 2
N1
V1
=1–
(3.76)
where G stands for weight of winding material. It then follows from Eq. (3.76) that
1
GTW
a¢
= saving of conductor material in using autotransformer
GTW – Gauto =
If a¢ = 10, saving is only 10% but for a¢ = 1.1, saving is as high as 90%. Hence the use of autotransformer
is more economical when the turn-ratio is close to unity.
The interconnection of EHV systems (e.g. 220 kV and 132 kV) by the autotransformers results in
considerable saving of bulk and cost as compared to the conventional two-winding transformers. Of course,
a 3-phase autotransformer will be required.
It can be easily shown with reference to Fig. 3.37 that
2
2
Ê N1 ˆ
ÊN
ˆ
- 1˜ R2; X 2¢ = Á 1 - 1˜ X2
R¢2 = Á
Ë N2 ¯
Ë N2 ¯
(3.77)
as seen on the primary side.
EXAMPLE 3.16 The 2000/200-V, 20-kVA transformer of Ex. 3.7 is connected as a step-up autotransformer
as in Fig. 3.38 in which AB is 200 V winding and BC is 2000-V winding. The 200-V winding has enough
insulation to withstand 2200-V to ground. Calculate (a) the LV and HV side voltage ratings of the
autotransformer; (b) its kVA rating; (c) kVA transferred inductively and conductively; (d) its efficiency at
full-load 0.8 pf.
(a)
(b)
I2
A
SOLUTION
+
V1 = 2000 V;
V2 = 2000 + 200 = 2200 V
20 ¥ 1000
I2 =
= 100 A
200
I1 – I2 = 10 A; I1 = 110 A
kVA rating =
2200 ¥ 100
= 220
1000
It is, therefore seen that a 20-kVA two-winding
transformer has a rating of 220 kVA as autotransformer, an
11 times increase.
I1
B
+
V2
(l1 – l2)
V1
–
–
C
Fig. 3.38
Transformers
97
V1 ( I1 - I 2 )
2000 ¥ 100
=
1000
1000
kVA transferred coductively = 220 – 20 = 200
(d) With data given in Ex. 3.7;
Core-loss (excitation voltage 2000 V) = 120 W
Full-load copper loss
300 W
(I2 = 100 A, I1 – I2 = 10 A)
420 W (Total loss)
(c) kVA transferred inductively =
Full-load output = 2200 ¥ 100 ¥ 0.8 = 176 kW
420
h =1–
= 99.76%
176000
It was shown in Ex. 3.11 that this transformer as a two-winding transformer has a full-load efficiency of
97.44%. The reason for such high efficiency (99.76%) for the autotransformer is its higher output for the same
excitation voltage and winding currents i.e., for the same losses.
EXAMPLE 3.17 A 240V/120V, 12 kVA transformer has full-load unity pf efficiency of 96.2%. It is
connected as an auto-transformer to feed a load at 360 V. What is its rating and full-load efficiency at
0.85 pf lagging?
SOLUTION
in Fig. 3.39
240 V/120 V, 12 kVA has rated currents of 50 A/100 A. It’s connection as an autotransformer as shown
Auto-transformer rating = 360 ¥ 100 ¥ 10–3 = 36 kVA
It is 3-times 2-winding connection.
As 2-winding connection,
Output, P0 = 12 ¥ 1 = 12 kW
1
P0
=
h=
= 0.962
P
P0 + PL
1+ L
P0
from which find full-load loss
Ê PL ˆ
1 = 0.962 + 0.962 Á ˜
Ë P0 ¯
or
100 A
+
+
120 V
–
150 A
+
+
360 V
240 V
240 V
50 A
PL
0.038
0.038
=
; PL = 12 ¥
0.474 kW
P0
0.962
0.962
–
–
In auto connection full-load loss remains the same. At 0.85 pf
–
Fig. 3.39
P0 = 36 ¥ 0.85 = 30.6 kW
1
h=
= 0.985 or 98.5%
0.474
1+
30.6
3.12 VARIABLE FREQUENCY TRANSFORMER
So far we have considered transformers which operate at fixed frequency (50 Hz). Their purpose is to transform
electric power from one voltage level to another; their performance measures being high efficiency and
low voltage regulation. Small transformers (usually iron-cored) are used for coupling purposes in electronic
98 Electric Machines
circuits for communication, measurement and control. These transformers process signals which contain a
wide band of frequencies (the width of band depends upon the signal measurement and control, audio, video,
etc). The two basic applications of these transformers are:
the impedance transforming property of the transformer. Under condition of impedance matching the
over-all efficiency of the system is as low as 50%. But in electronic circuit applications the performance
criterion is the maximum power unlike the maximum efficiency in power system applications. Such
transformers are known as output transformers while in audio applications these are known as audiotransformers.
An important requirement of these transformers is that the amplitude voltage gain (ratio of output/input
voltage amplitude) should remain almost constant over the range of frequencies (bandwidth) of the signal.
Further, it is desirable that the phase shift of output signal from the input signal over the signal bandwidth be
small. We shall now investigate the gain and phase frequency characteristics of the transformer. This would
of course include the effect of the output impedance (resistance) of the electronic circuit output stage. In these
characteristics as the frequency range is quite large the frequency scale used is logarithmic.
The circuit model of a transformer fed from a source of finite output resistance is drawn in Fig. 3.40(a)
where the transformer core loss is ignored and leakage and magnetizing effects are shown in their frequency
dependent form i.e., X = wL. It may be observed here that Lm (megnetizing inductance) = L11 (self inductance
of the primary coil).
Amplitude and phase response can be divided into three regions wherein the response calculations are
simplified by making suitable approximations as below.
Mid-band Region
In this region the series leakage inductances can be ignored (as these cause negligible voltage drops) and the
shunt inductance (magnetizing inductance) can be considered as open circuit. With these approximations the
equivalent circuit as seen on the primary side is drawn in Fig. 3.40(b). It immediately follows from the circuit
analysis that VL and VS are in phase, the circuit being resistive only. As for the amplitude gain, it is given as
È RL¢ ˘
V L¢ = VS Í
˙ ; R = RS + R1 + R¢2
Î R + RL¢ ˚
È RL¢ ˘
È N1 ˘
Í ˙ VL = VS Í R + R ¢ ˙
L˚
Î
Î N2 ˚
A0 =
È N 2 ˘ È RL¢ ˘
VL
= Í
˙Í
˙ ; – A0 = 0
VS
Î N1 ˚ Î R + RL¢ ˚
(3.78)
High-frequency Region
In this region the series inductances must be taken into account but the shunt inductance is an effective open
circuit yielding the approximate equivalent circuit of Fig. 3.40(c). Amplitude and phase angle as function of
frequency are derived below.
AH =
È N2 ˘
RL¢
VL
= Í
˙
N
R
R
(
+
VS
L¢ ) + jw L
Î 1˚
Transformers
Rs
wLl1
R1
wLl2
R2
+
+
wLm
Vs
VL
–
RL
–
N1 : N 2
(a) Circuit model of transformer
R
Rs
R¢2
R1
+
+
R¢L
V¢L
Vs
–
–
(b) Approximate circuit model in mid-frequency region
Rs
R
R1
wL
R2
wLl1
wL¢l2
+
+
V¢L
Vs
–
R¢L
–
(c) Approximate circuit model in high-freuency region
R
Rs
R1
R¢2
+
+
wLm
Vs
V¢L
R¢L
–
–
(d) Approximate circuit model low-frequkency region
Fig. 3.40
L = Ll1 + Ll2 = total leakage inductance as seen on primary side.
where
Further rearrangement leads to
È N 2 ˘ È RL¢ ˘
1
AH = Í ˙ Í
˙
N
R
+
R
1
+
j
w[
L
/( R + RL¢ )]
¢
L˚
Î 1˚Î
We can write
R + RL¢
= wH = corner frequency of high-frequency region
L
99
100
Electric Machines
Also recognizing
or
È N 2 ˘ È RL¢ ˘
Í
˙Í
˙ = A0, we get
Î N1 ˚ Î R + RL ˚
A0
AH =
1+ jw /w H
A0
AH =
; – AH = tan–1 w/wH
[1 + (w / w H ) 2 ]1/ 2
(3.79)
(3.80)
As per Eq. (3.80) the gain falls with frequency acquiring a valve of A0 / 2 at w/wH = 1 and a phase angle
of ––45°. This indeed is the half power frequency (wH).
In this region the series effect of leakage inductances is of no consequence but the low reactance (wLm)
shunting effect must be accounted for giving the approximate equivalent circuit of Fig 3.39(d). Amplitude
and phase angle of frequency response is derived below.
The corner frequency w L of this circuit is obtained by considering the voltage source as short circuit. This
circuit is Lm in parallel with R||R¢L. Thus
È R || RL¢ ˘
wL = Í
˙
Î Lm ˚
The complex gain can then be expressed as
AL =
AL =
or
A0
1+ j (w /w L )
A0
(3.81)
; – AL = tan–1 w/wL
2 1/ 2
[1 + (w /w L ) ]
(3.82)
Again the lower corner frequency is the half power frequency.
The complete amplitude and phase response of the transformer (with source) on log frequency scale
are plotted in Fig. 3.41. At high frequencies the interturn and other stray capacitances of the transformer
windings begin to play a role. In fact the capacitance-inductance combination causes parallel resonance effect
1.0 Low-frequency
region
Relative
voltage
ratio
High-frequency
region
80°
60°
0.8
A
40°
20°
Phase
angle
0°
0.4
20°
40°
0.2
60°
80°
0
0.2
0.5
1.0
2
w /wL
5
0.2
0.5
1.0
2
w /wH
5
Fig. 3.41 Normalized frequency characteristic of output transformers
Lead –A Lag
0.6
Transformers
101
on account of which an amplitude peak shows up in the high-frequency region of the frequency response. No
reasonably accurate modelling of these effects is possible and best results are obtained experimentally. The
frequency response of Fig. 3.41 gives a general guidance as to its nature.
In generation, transformation, transmission and utilization of electric energy it can be shown that it is
economical to use the three-phase system rather than the single-phase. For three-phase transformation,
three single-phase transformers are needed. Two arrangements are possible: a bank of three single-phase
transformers or a single three-phase transformer with the primary and secondary of each phase wound on
three legs of a common core. The three-phase transformer unit costs about 15% less than that of a bank
and furthermore, the single unit occupies less space. There is little difference in reliability, but it is cheaper
to carry spare stock of a single-phase rather than a three-phase transformer. In underground use (mines) a
bank of single-phase units may be preferred as it is easier to transport these units. The bank also offers the
advantage of a derated open-delta operation when one single-phase unit becomes inoperative. Reduced cost
being an overweighing consideration, it is common practice to use a three-phase transformer unit.
In a three-phase bank the phases are electrically connected but the three magnetic circuits are independent.
In the more common three-phase, 3-limb core-type transformer (Fig. 3.42(a)), the three magnetic circuits
are also linked. Where delinking of the magnetic circuits is desired in a three-phase unit, a 5-limb shell type
transformer could be used (Fig. 3.42(b)).
(a) Core type (commonly used)
(b) Shell type, 5 -limb core
Fig. 3.42 Three-phase transformer cores
Three-phase Transformer Connections
A variety of connections are possible on each side of a 3-phase transformer (single unit or bank). The three
phases could be connected in star, delta, open-delta or zigzag star. Each of the three phases could have two
windings or may have autoconnection. Further, certain types of connections require a third winding known
as tertiary (refer Sec. 3.14).
Labelling of Transformer Terminals
Terminals on the HV side of each phase will be labelled as
capital letters A, B, C and those on the LV side will be labelled
as small letters a, b, c. Terminal polarities are indicated by
suffixes 1 and 2 with 1’s indicating similar polarity ends and
so do 2’s Labelling of terminals is illustrated in Fig. 3.43 for
phase a. Assuming the transformer to be ideal, VA2A1 (voltage
of terminal A2 with respect to A1) is in phase with Va2a1 and
IA is in phase with Ia.
–
+
IA
A1
A2
a1
a2
–
+
Fig. 3.43
Ia
Va a
2 1
VA A
2 1
102
Electric Machines
Star/Star (Y/Y) Connection
A
Star connection is formed on each side by connecting together phase winding terminals suffixes 1 as in Fig. 3.44(a).
The phasor diagram is drawn in Fig. 3.44(b) from which it
is easily seen that the voltages of the corresponding phases
(and therefore of the corresponding lines) are in phase. This
is known as the 0°-connection. The letters within brackets on
the phasor diagram indicate the lines, to which the terminals
are connected. If the winding terminals on secondary side are
reversed, the 180°-connection is obtained.
It is also observed from Fig. 3.44 that if the phase
transformation ratio is x : 1, the line transformation (line-toline voltages, line currents) the ratio is also x : 1.
A2
A1 B2
x
V
3x
V/x
a
C
N
x
1
A2
xl
B
V
V
3
I
B1C2
1
a1 b2
1
b 1 c2
N
b
C1
x
c1
c
(a)
A2(A)
To line A
a2(a)
N
n
B2(B) c2(c)
(b)
C2(C)
Delta/Delta (D/D) Connection
To line a
b2(b)
Fig. 3.44 Star/star 0°-connection
Figure 3.45(a) shows the delta/delta connection* and the
corresponding phasor diagram is given in Fig. 3.45(b). The
sum of voltages around the secondary delta must be zero;
otherwise delta, being a closed circuit, means a short circuit.
With polarities indicated on the primary and secondary
sides, voltages Va2a1, Vb2b1 and Vc2c1 add to zero as per the
phasor diagram if the delta is formed by connecting a1b2,
b1c2 and c1a2. It is easily seen from the phasor diagram that
the primary and secondary line voltages are in phase so it is
the 0°-connection. However, if the secondary leads a, b, c are
taken out from the delta nodes a1b2, b1c2, c1c2, the secondary
voltages are in phase opposition to the primary voltages can
be visualized from the phasor diagram of Fig. 3.45(b). This is
the 180°-connection.
It is also seen from Fig. 3.45(a) that if the phase transformation ratio is x : 1, the transformation ratio for line quantities
is also x : 1.
A
l
B
C
V
V
l/ 3
A2
A1 B2
x
1
a2
a1
x
B1 C 2
1
b2
b1 c2
x
C1
1
c1
V/x
Ix/ 3
Vlx
a
b
Ix
c
(a)
a2(a)
A2(A)
C2(C)
N
B2(B)
c2(c)
n
b2(b)
(b)
Fig. 3.45
Delta/delta connection
* The star and delta connections in later parts of the book will generally be indicated as in Fig. 3.46. The style
temporarily adopted here is for the sake of clarity of identifying the primary and secondary of each phase. Furthermore, it also stresses the fact that physical disposition of the windings in the connection diagram has no relationship to the phasor diagram.
Star
Delta
Fig. 3.46
Transformers
103
In the delta/delta connection if one of the transformers is disconnected, the resulting connection is known
as open-delta. Supposing the b-phase transformer in Fig. 3.47(a) is removed, and the open-delta is excited
from balanced 3-phase supply, then it easily follows from the phasor diagram of Fig. 3.47(b) that the voltage
Vb2b1 = Vbc does not change as it equals – (Vca + Vab); thus the voltages on the secondary side still remain
balanced 3-phase. The open-delta connection supplying a balanced load is shown in Fig. 3.47(a). If the
maximum allowable secondary phase current is Iph, the transformer can handle VA of
Sopen-delta =
3 VIph ; Iph = Iline
which for normal delta/delta connection is
Sdelta = 3VIph
Thus the open-delta connection has a VA rating of 1/ 3 = 0.58 of the rating of the normal delta/delta
connection.
a
A
T2
T1
Iph
B
C
V
Iph
b
c
Fig. 3.47(a) Open-delta or V-connection
The phasor diagram of open-delta is drawn in Fig. 3.47(b). Vba and I ab pertain to transformer T1 and Vca
and I ca to T2.
Vca
Vc
Ic = Iac
(30° + f)
Va
f
(30° – f)
Ia
Iab
Vba
Fig. 3.47(b)
Vb
f
104
Electric Machines
P1 = VIph cos (30° – f)
P2 = VIph cos (30° + f)
P = P1 + P2 = VIph [cos (30° – f) + cos (30° + f)]
Power output of T1
and that of T2
Total power delivered
P=
Upon simplification, we find
3 VIph cos f
The two transformers supply equal power at upf, i.e., f = 0.
Star/Delta (Y/D) Connection
Star connection is formed on primary side by connecting together 1 suffixed terminals; 2 suffixed terminals
being connected to appropriate lines; the delta is formed by connecting c1a2, a1b2 and b1c2 with the lines
connected to these junctions being labelled as a, b and c respectively as shown in Fig. 3.48(a). The phasor
diagram is drawn in Fig. 3.48(b). It is seen from the phasor diagram on the delta side that the sum of voltages
around delta is zero. This is a must as otherwise closed delta would mean a short circuit. It is also observed
from the phasor diagram that phase a to neutral voltage (equivalent star basis) on the delta side lags by – 30°
to the phase-to-neutral voltage on the star side; this is also the phase relationship between the respective lineto-line voltages. This connection, therefore, is known as – 30°-connection.
The + 30°-connection follows from the phasor diagram of Fig. 3.49(a) with the corresponding connection
diagram as in Fig. 3.49(b).
A
B
A2(A)
C
N
V
l
a2(a)
V
3
A2
a2
xl
x 1
V
x 3
A1 B2
x 1
a1 b2
B1 C2
x 1
b 1 c2
+30°
n
N
C1
b2(b)
c2(c)
c1
C2(C)
B2(B)
(a)
3xl
a
V
x 3
c
b
–30°
a2(a)
C2(C)
c2(c)
B2(B)
Fig. 3.48
C
N
(a)
A2(A)
N
B
A
(b)
A2
A2 B2
B1 C2
C1
a2
a1 b2
b1 c2
c1
b
c
n
b2(b)
a
(b)
Fig. 3.49
Similarly ± 90°-connections are also possible in the star/delta connection by relabelling the delta side
lines. For example for + 90° connection relabel c Æ a, b Æ c and a Æ b. Reader may work out relabelling for
– 90° connection.
In Indian and British practices ± 30°-connections are employed. The American practice is to use
± 90°-connections.
Transformers
105
It follows from Fig. 3.48(a) that if the phase transformation ratio of the star/delta connection is x : 1, the
line transformation ratio in magnitude is 3 x: 1.
Delta/Star (D/Y) Connection
This connection is simply the interchange of primary and secondary roles in the star/delta connection.
One just interchanges capital and small letter suffixings in Figs 3.48 and 3.49. Of course what was the
– 30°-connection will now be the + 30°-connection and vice versa. If the phase transformation ratio is x : 1
(delta/star), the transformation ratio for line quantities will be (x/ 3 ) : 1.
Delta/Zig-zag Star Connection
The winding of each phase on the star side is divided into two equal halves with labelling as in Fig. 3.50.
Each leg of the star connection is formed by using halves from two different phases. The phasor diagram for
this connection is given in Fig. 3.51 from which the connection diagram easily follows. Obviously it is the
0°-connection. Reversal of connections on the star side gives us the 180°-connection.
Phase transformation = x : 1
3
Line transformation = x :
2
or
2
3
x:1
a2 a 3
a1
a4
Va a
2 1
Va a
4 3
–
–
+
+
Fig. 3.50
a4(a)
Line voltage = 3/2
1/2
c1
A2(A)
b1
x
1/2
n
N
C2(C)
B2(B)
Fig. 3.51
c4(c)
Delta/zig-zag star 0°-connection
Star/Zig-zag Star
The connection is indicated by the phasor diagram of Fig. 3.52.
Phase transformation = x : 1
Line transformation =
3x:
2
3
x:1
or
3
2
a1
b4 (b)
106
Electric Machines
a4(a) a4(a)
1/2
30°
b1
A2(A)
30°
c1
c1 b
1
1/2
n
c4(c)
x
n
b4(b)
N
a1
a1
B2(B)
C2(C)
–30° connection
b4(b)
c4(c)
+30° connection
Fig. 3.52 Star/zig-zag star
Phase Groups
Various transformer connections with the same phase shift are grouped together. Thus there are Group I (0°),
Group II (180°), Group III (30°) and Group IV (–30°).
In star connection with earthed neutral, the maximum voltage of the phase winding to ground is 1/ 3 or 58%
of the line voltage, while in delta connection this is equal to the line voltage in case of earthing of one of the
lines during a fault. Therefore, for very high voltage transformers the star connection on the HV side is about
10% cheaper than delta connection on account of insulation cost. A delta-connected primary is necessary for
a star-connected LV secondary feeding mixed 3-phase and 1-phase (line-to-neutral) loads. This is because the
lines on the primary side can only carry current which add to zero. In the case of unbalanced 1-phase loads on
secondary, delta-connected primary is needed to allow the flow of zero sequence current
I0 =
In
1
= ( I a + Ib + I c )
3
3
as shown in Fig. 3.53 so that
I A + I B + IC = 0
This means that only positive and negative sequence currents flow in the lines on the delta side.
IA
la
l0 = ln l3
IB
ln
lb
lo
lc
Fig. 3.53
Transformers
107
This could also be achieved by star-connected primary provided the primary and secondary star points are
grounded. But this is not recommended on account of flow of ground current for unbalanced secondary loads.
Choice of Transformer Connections
Star/star
This is economical for small HV transformers as it minimizes the turns/phase and winding insulation.
A neutral connection is possible. However, the Y /Y connection is rarely used* because of difficulties
associated with the exciting current.
Delta/delta
This suits large LV transformers as it needs more turns/phase of smaller section. A large load unbalance can
be tolerated. The absence of a star point may be a disadvantage. This connection can operate at 58% normal
rating as open-delta when one of the transformers of the bank is removed for repairs or maintenance. (This
has already been explained.)
Star/delta
This is the most commonly used connection for power systems. At transmission levels star connection is on
the HV side, i.e. D/Y for step-up and Y/D for step-down. The neutral thus available is used for grounding on
the HV side. At the distribution level the D/Y transformer is used with star on the LV side which allows mixed
3-phase and 1-phase loads, while delta allows the flow of circulating current to compensate for neutral current
on the star side (Fig. 3.53).
The Y/D connection has an associated phase shift of ± 30° which must be accounted for in power system
interconnections.
Harmonics
It was seen in Sec. 3.10 that when the third-harmonic current is permitted to flow, by circuit conditions, along
with the sinusoidal magnetizing current in a transformer, the core flux is sinusoidal and so is the induced
emf. On the other hand, when the circuit does not permit the flow of the third-harmonic current, i.e. the
magnetizing current is sinusoidal, the flux is flat-topped containing “depressing” third-harmonic and as a
consequence third-harmonic voltages are present in the induced emfs. This problem in 3-phase transformers
will now be examined.
It is to be observed here that the phase difference in third-harmonic currents and voltages on a 3-phase
system is 3 ¥ 120° = 360° or 0° which means that these are cophasal. Therefore, third-harmonic (in general
harmonics of order 3n called triplens) currents and voltages cannot be present on the lines of a 3-phase
system as these do not add up to zero.
Three-phase Bank of Single-phase Transformers
Delta/delta connection
The supply voltage provides only sinusoidal magnetizing current so that core flux is flat-topped; but the
third-harmonic emfs induced (cophasal) cause circulating currents in deltas restoring the flux to almost
sinusoidal. The third-harmonic voltages are very small as the transformer offers low impedance to thirdharmonic currents.
* Recently a favourable trend is developing for reasons of economy.
108 Electric Machines
Star/delta and delta/star connection
Because of one delta connection the same conditions are obtained as in D/D connection except that the
impedance offered to the flow of third-harmonic currents in delta is now larger and so are third-harmonic
voltages.
Star/star connection
In the case of isolated neutrals, third-harmonic voltages are present in each phase as explained earlier. Further,
since these voltages are cophasal, no third-harmonic voltages are present between lines. The voltage of phase
a to neutral can now be expressed as
eaN = ea sin wt + ea3 sin 3wt
While fundamental frequency voltages in the three phases have a relative phase difference of 120°, the
third-harmonic voltages in them are cophasal (with
a
respect to each other), but their phase with respect to the
EaN
Ea
fundamental frequency (voltage changes at the rate of
2 w, twice the fundamental frequency). This situation is
illustrated in the phasor diagram of Fig. 3.54 from which
Cophasal third-harmonic
it is immediately observed that the voltage of the neutral
voltage (E3)
2w
point oscillates at frequency 2w. The phenomenon is
N
known as oscillating neutral and is highly undesirable
because of which the star/star connection with isolated
neutrals is not used in practice.
c
b
If the neutrals are connected, it effectively separates
Fig. 3.54 Oscillating neutral
the three transformers. Third-harmonic currents can now
flow via the neutrals.
Three-phase Transformer
In core type transformer (Fig. 3.42(a)), the third-harmonic fluxes in all the three limbs are simultaneously
directed upwards or downwards so that this flux must return through air (high-reluctance path). The highreluctance path tends to suppress the third-harmonic flux. The phenomenon gets more complex now and at
core densities exceeding 1.5 T, the total harmonic content (particularly fifth) is very marked in the magnetizing
current (fifth harmonic currents can flow on lines as their relative phase difference is 5 ¥ 120° = 600° or 120°).
To reduce the strong fifth harmonic in the magnetizing current for the star/star connection with isolated
neutral, a path must be provided through iron for the third-harmonic flux. Hence, the use of a 5-limb core as
in Fig. 3.42(b).
Back-to-Back Test on Three-phase Transformers
Fig. 3.55(a) shows the connection arrangement for the back-to-back test on two identical 3-phase transformers.
The two secondaries must be connected in phase opposition and in proper phase sequence. The auxiliary
transformer for circulating full-load current is included in the circuit of the two secondaries; it could also be
included in the circuit of the primaries. Thus with only losses (core-loss and copper-loss) supplied from the
mains, a “heat run” test could be conducted on the transformers.
Transformers
109
3-phase
mains
a11
T1
a21
Auxiliary
transformer
a22
a12
T2
Fig. 3.55(a)
Delta/Delta Connected Transformers
The primaries are normally excited from the mains. Each secondary delta is opened at one junction and
a single-phase transformer can be employed to circulate full-load current in both the deltas as shown in
Fig. 3.55(b).
1-phase supply
Delta secondaries
Auxiliary transformer
Fig. 3.55(b)
D/D transformers
110 Electric Machines
EXAMPLE 3.18 A 3-phase transformer bank consisting of three 1-phase transformers is used to stepdown the voltage of a 3-phase, 6600 V transmission line. If the primary line current is 10 A, calculate the
secondary line voltage, line current and output kVA for the following connections: (a) Y/D and (b) D/Y. The
turns ratio is 12. Neglect losses.
lLY = 10 A = lPY
SOLUTION
lLD
12:1
(a) The Y/D connection is drawn in Fig. 3.56(a).
6600
VPY =
3
VLY = 6600
VLD = VPD
VPY
lPD
6600
= 317.55 V
3 ¥ 12
= 10 ¥ 12 = 120 A
VPD = VLD =
IPD
(a)
ILD = 120 3 = 207.84 A
6600
1
Output kVA = 3 ¥
¥ 120 3 ¥
1000
3 ¥ 12
lPY = lPY
lLD = 10 A
12:1
VLD = 6600 V = VPD
VPY
lPD
VLY
= 66 3 = 114.3
(b) The D/Y connection is drawn in Fig. 3.56(b).
IPD =
ILY =
VPY =
VLY =
10
A
3
(b)
Fig. 3.56
12 ¥ 10
= 69.28 A
3
6600
V
12
6600 3
12
925.6 V
6600 3
120
1
¥
¥
12
1000
3
= 114.3 (same as in part (a))
Output kVA =
3¥
EXAMPLE 3.19 A D/Y connected 3-phase transformer as shown in Fig. 3.57 has a voltage ratio of
22 kV (D)/345 kV(Y) (line-to-1ine). The transform is feeding 500 MW and 100 MVAR to the grid (345 kV).
Determine the MVA and voltage rating of each unit (single-phase). Compute all currents and voltages of
both magnitude and phase angle in all the windings (primaries and secondaries). Assume each single-phase
transformer to be ideal.
C
a
a
Load
1:
b
B
c
A
Fig. 3.57
Transformers
111
SOLUTION
Load MVA, S = 500 + j 100
S = 510
MVA rating of each (single phase) transformer = 510/3 = 170
345 / 3
Voltage rating of each transformer =
= 9.054
22
Let us choose voltage of star phase A as reference then
345
–0° = 199.2 –0° kV
Star side
VA = VAN =
3
VB = 199.2 ––120° kV, VC = 199.2 ––240° kV
Note: Phase sequence is ABC
VAB = VA - VB = 345 –30° kV
VBC = 345 ––90° kV
or
VCA = 345 ––210° kV
500 + j 100
= 0.837 + j 0.167 kA; as S = VI*
I A* =
3 ¥ 199.2
I A = 0.837 – j 0.167 = 0.853 ––11.3° kA
I B = 0.853 ––131.3° kA
I C = 0.853 –– 251.3° kA
Delta side
VA
199.2
=
–0° = 22–0° kV
a
9.054
= 22 ––120° kV
Vab =
Vbc
Vca = 22 ––240° kV
I ab = 9.054 ¥ 0.853 ––11.3° =7.723 ––11.3° kA
I bc = 7.723 ––131.3° kA
I ca = 7.723 ––251.3° kA
I a = I ab - I bc = 3 ¥ 7.723 –(–11.3° – 30°) = 13.376 ––41.3°
I b = 13.376 –(–120° – 11.3°) = 13.376 ––131.3° kA
I c = 13.376 –(–240° – 11.3°)=13.376 ––251.3° kA
Note
It is easily observed from above that line voltages and currents on the star side lead those on the delta side
by 30°.
EXAMPLE 3.20 Three 1-phase 20-kVA, 2000/200-V transformers identical with that of Ex. 3.3 are
connected in Y/D in a 3-phase, 60 kVA bank to step-down the voltage at the load end of a feeder having
impedance of 0.13 + j 0.95 W/phase. The line voltage at the sending-end of the feeder is 3464 V. The
transformers supply a balanced 3-phase load through a feeder whose impedance is 0.0004 + j 0.0015 W/
phase. Find the load voltage (line-to-line) when it draws rated current from transformers at 0.8 lagging
power factor.
112 Electric Machines
SOLUTION Figure 3.58 gives the circuit diagram of the system. The computations will be carried out on per phase-Y
basis by referring all quantities to the HV (Y-connected) side of the transformer bank.
LV feeder impedance referred to the HV side is
2
Ê 2000 3 ˆ
Á
˜ (0.0004 + j 0.0015) = 0.12 + j 0.45 W/phase
Ë 200 ¯
0.13 + j 0.95 W/phase
0.0004 + j 0.0015 W/phase
3464 V = 2000 3
Load
Fig. 3.58
The total series impedance of the HV and LV feeders referred to the HV side is
ZF = (0.13 + j 0.95) + (0.12 + j 0.45)
= 0.25 + j 1.4 W/phase
From Ex. 3.5, the equivalent impedance of the transformer bank is referred to the HV side
ZT = 0.82 + j 1.02 W/phase Y
3464
= 2000 V/phase Y
3
Load current on the HV side = rated current of transformer
= 10 A/phase Y
Sending-end feeder voltage =
It is now seen that the equivalent circuit for one phase referred to the Y-connected HV side is exactly the same as in
Ex. 3.6, Fig. 3.20. Thus the load voltage referred to the HV side is 197.692 V to neutral. The actual load voltage is
197.688 V, line-to-line (since the secondaries are D-connected).
PU method In such problems it is convenient to use the pu method. We shall choose the following base values:
(MVA)B =
3 ¥ 20
= 0.06
1000
Voltage base on HV side = 2 3 kV (line-to-line)
Voltage base on LV side = 0.2 kV (line-to-line)
Note
The voltage base values are in the ratio of line-to-line voltages (same as phase voltages on equivalent star basis).
Z1 (LV line) (pu) = (0.0004 + j 0.0015) ¥
0.06
(0.2) 2
= (0.06 + j 0.225) ¥ 10–2
Z 2 (HV line) (pu) = (0.13 + j 0.95) ¥
0.06
(2 3 )2
= (0.065 + j 0.475) ¥ 10–2
Transformers
ZT (star side) (pu) = (0.82 + j 1.02) ¥
113
0.06
(2 3 )2
= (0.41 + j 0.51) ¥ 10–2
Note: Suppose the transformer impedance was given on the delta side
Ê 200 ˆ
ZT (delta side) = ÁË
2000 ˜¯
2
¥ (0.82 + j 1.02)
= (0.82 + j 1.02) ¥ 10–2 W (delta phase)
1
Equivalent star impedance = (0.82 + j 1.02) ¥ 10–2 W
3
0.06
1
¥ (0.82 + j 1.02) ¥ 10–2
ZT (pu) =
(0.2) 2 3
= (0.41 + j 0.51) ¥ 10–3 (same as calculated above)
Z (total) (pu) = 0.06 + j 0.225
0.065 + j 0.475
0.41 + j 0.51
(0.535 + j 1.21) ¥ 10- 2
V1 (sending–end voltage) = 2 3 kV (line) or 1 pu
I1 (= rated current) = 1 pu; pf= 0.8 lag
V2 (load voltage) = 1 – 1 ¥ (0.535 ¥ 0.8 + 1.2 ¥ 0.6) ¥ 10–2
= 0.98846 pu
= 0.98846 ¥ 200 = 197.692 V (line)
EXAMPLE 3.21 Three transformers, each rated 20 kVA, 2 kV/200 V, are connected D/D and are fed
through a 2000 V (line-to-line) 3-phase feeder having a reactance of 0.7 W/phase. The reactance of each
transformer is 0.0051 pu. All resistances are to be ignored. At its sending-end the feeder receives power
through the secondary terminals of a 3-phase Y/D connected transformer whose 3-phase rating is 200 kVA,
20/2 kV (line-to-line). The reactance of the sending-end transformer is 0.06 pu on its own rating. The
voltage applied to the primary terminals is 20 kV (line-to-line).
A 3-phase fault takes place at the 200 V terminals of the receiving-end transformers. Calculate the fault
current in the 2 kV feeder lines, in the primary and secondary windings of the receiving-end transformers
and at the load terminals (200 V terminals).
SOLUTION Choose a common 3-phase base of 60 kVA. Line-to-line voltage bases are in ratio of transformation
20 kV : 2 kV : 200 V. It is observed that
XT (sending-end*) = 0.06 ¥
60
= 0.018 pu
200
For the 2 kV feeder
VB =
2000
= 1154.7 V (line-to-neutral)
3
* Transformer impedance in pu is independent of the connection.
114 Electric Machines
IB =
ZB =
60 ¥ 1000
= 17.32 A, phase Y
3 ¥ 2000
1154.7
= 66.6 W/phase Y
17.32
0.7
= 0.0105 pu
66.6
XT (receiving-end*) = 0.0051 pu
Xfeeder =
Total reactance from the sending-end to the fault point (on the secondary side of the receiving-end transformer) =
0.018 + 0.0105 + 0.0051 = 0.0336 pu
20
= 1.0 pu
20
1.0
Fault current =
= 29.76 pu
0.0336
The current in any part of the system can be easily computed as below:
Sending-end voltage =
Current in 2 kV feeder = 29.76 ¥ 17.32 = 515.4 A
Current in 2 kV winding of D/D transformer = 515.3 3 = 297.56 A
Current in 200 V winding of D/D transformer = 297.56 ¥ 10 = 2975.6 A
Current at load terminals = 2975.6 3 = 5154 A
EXAMPLE 3.22 A 3-phase bank of three single-phase transformer are fed from 3-phase 33 kV (line-toline). It supplies a load of 6000 kVA at 11 kV (line-to-line). Both supply and load are 3-wire. Calculate the
voltage and kVA rating of the single-phase transformer for all possible 3-phase transformer connection.
SOLUTION
1. Star-Star connection
Primary-side phase voltage, VP1 =
33
= 19.05 kV
3
11
= 6.35 kV
3
Transformer voltage rating = 19.05/6.35 kV
Secondary-side phase voltage, VP2 =
kVA rating =
6000
= 2000
3
2. Star-Delta connection
VP1 = 19.05 kV, VP2 = 11 kV
Transformer rating = 19.05/11 kV, 2000 kVA
3. Delta-Star connection
Transformer rating = 33/6.35 kV, 2000 kVA
4. Delta-Delta connection
Transformer rating = 33/11 kV, 2000 kVA
* Transformer impedance in pu is independent of the connection.
Transformers
115
EXAMPLE 3.23 A 6.6 kV/400 V, 75 kVA single-phase transformer has a series reactance of 12%
(0.12 pu).
(a) Calculate the reactance in ohms referred to LV and HV sides.
(b) Three such transformers are connected in Star-Star, calculate (i) the line voltage and kVA rating,
(ii) pu reactance of the bank, (iii) series reactance in ohms referred to HV and LV sides
(c) Repeat part (b) if the bank is connected star on HV side and delta on LV side.
SOLUTION
X(pu) =
(a)
HV side
0.12 =
X(W) =
LV side
0.12 =
X (W) MVA
(kV )2
X (W) ¥ 75 ¥ 10-3
(6.6) 2
0.12 ¥ (6.6) 2
75 ¥ 10-3
= 69.696 W
X (W) ¥ 75 ¥ 10-3
(0.4) 2
X(W) = 0.256 W
(b) Star-Star connection
(i) Line voltage
HV 6.6 3 = 11.43 kV
LV
400 3 = 692.8 V
Rating = 3 ¥ 75 = 225 kVA
(ii)
X(pu) =
X (W) MVA (3 - phase)
(kV (line))2
=
69.696 ¥ 225 ¥ 10-3
(6.6 3 ) 2
= 0.12
(iii) HV side
X(W) = 69.696 W/phase
LV side
X(W) = 0.256 W/phase
(c) Star-Delta
(i) Line voltages
Star side 6.6 3 = 11.43 kV
(ii)
(iii) Star side
Delta side
Delta side
Rating
X(pu)
X
X
= 400 V
= 3 ¥ 75 = 225 kVA
= 0.12
= 69.69 W/phase
= 0.256 W/phase
X(pu), calculated from delta side
X(pu) =
(0.256 / 3) ¥ 225 ¥ 10-3
(0.4) 2
= 0.12
116 Electric Machines
3.14
PARALLEL OPERATION OF TRANSFORMERS
When the load outgrows the capacity of an existing transformer, it may be economical to install another one
in parallel with it rather than replacing it with a single larger unit. Also, sometimes in a new installation, two
units in parallel, though more expensive, may be preferred over a single unit for reasons of reliability—half
the load can be supplied with one unit out. Further, the cost of maintaining a spare is less with two units in
parallel. However, when spare units are maintained at a central location to serve transformer installations in a
certain region, single-unit installations would be preferred. It is, therefore, seen that parallel operation of the
transformer is quite important and desirable under certain circumstances.
The satisfactory and successful operation of transformers connected in parallel on both sides requires that
they fulfil the following conditions:
(i) The transformers must be connected properly as far as their polarities are concerned so that the net
voltage around the local loop is zero. A wrong polarity connection results in a dead short circuit.
(ii) Three-phase transformers must have zero relative phase displacement on the secondary sides and
must be connected in a proper phase sequence. Only the transformers of the same phase group can be
paralleled. For example, Y/Y and Y/D transformers cannot be paralleled as their secondary voltages
will have a phase difference of 30°. Transformers with +30° and –30° phase shift can, however, be
paralleled by reversing the phase-sequence of one of them.
(iii) The transformers must have the same voltage-ratio to avoid no-load circulating current when
transformers are in parallel on both primary and secondary sides. Since the leakage impedance is low,
even a small voltage difference can give rise to considerable no-load circulating current and extra I 2R
loss.
(iv) There should exist only a limited disparity in the per-unit impedances (on their own bases) of the
transformers. The currents carried by two transformers (also their kVA loadings) are proportional
to their ratings if their ohmic impedances (or their pu impedances on a common base) are inversely
proportional to their ratings or their per unit impedances on their own ratings are equal. The ratio
of equivalent leakage reactance to equivalent resistance should be the same for all the transformers.
A difference in this ratio results in a divergence of the phase angle of the two currents, so that one
transformer will be operating with a higher, and the other with a lower power factor than that of the
total output; as a result, the given active load is not proportionally shared by them.
Parallel Transformers on No-load
V1
V2
2
E1
E2
Load
The parallel operation of transformers can be easily
Primary
conceived on a per phase basis. Figure 3.59 shows
two transformers paralleled on both sides with proper
1
polarities but on no-load. The primary voltages V1
Secondary
and V2 are obviously equal. If the voltage-ratio of
the two transformers are not identical, the secondary
ZL
induced emf’s, E1 and E2 though in phase will not
be equal in magnitude and the difference (E1 – E2)
will appear across the switch S. When secondaries
are paralleled by closing the switch, a circulating Fig. 3.59
current appears even though the secondaries are
not supplying any load. The circulating current will
S
Transformers
117
depend upon the total leakage impedance of the two transformers and the difference in their voltage ratios.
Only a small difference in the voltage-ratios can be tolerated.
Equal voltage-ratios
When the transformers have equal voltage ratio, E1 = E2
in Fig. 3.59, the equivalent circuit of the two transformers
would then be as shown in Fig. 3.60 on the assumption that
the exciting current can be neglected in comparison to the
load current. It immediately follows from the sinusoidal
steady-state circuit analysis that
Z2
IL
Z1 + Z 2
(3.83)
Z1
IL
I2 =
Z1 + Z 2
(3.84)
I1 =
and
Of course
Z1
I1
Z2
I2
IL
ZL
VL
I1 + I 2 = I L
(3.85)
Taking VL as the reference phasor and defining complex
power as V * I , the multiplication of VL* on both sides of
Eqs (3.83) and (3.84) gives
where
V1
Fig. 3.60
S1 =
Z2
SL
Z1 + Z 2
(3.86)
S2 =
Z1
SL
Z1 + Z 2
(3.87)
S1 = VL* I1
S2 = VL* I2
S L = VL* IL
These are phasor relationships giving loadings in the magnitude and phase angle. Equations (3.86) and
(3.87) also hold for pu loads and leakage impedances if all are expressed with reference to a common base.
It is seen from Eqs (3.83) and (3.84) that the individual currents are inversely proportional to the respective
leakage impedances. Thus, if the transformers are to divide the total load in proportion to their kVA ratings, it
is necessary that the leakage impedances be inversely proportional to the respective kVA ratings, i.e.
Z1
S2 (rated )
=
Z2
S1 (rated )
(3.88)
This condition is independent of the power factor of the total load. The condition of Eq. (3.88) can be
written as
Z1
VL I 2 (rated )
=
Z2
VL I1 (rated )
118 Electric Machines
or
or
Z1 I1 (rated )
Z 2 I 2 (rated )
=
VL
VL
Z1 (pu) = Z2 (pu); on own rating
(3.89)
It means that if individual transformer loadings are to be in the ratio of their respective kVA ratings, their
pu impedances (on their own ratings) should be equal. If
Z1 < Z2
S2 (rated )
S1 (rated )
(3.90a)
the transformer 1 will be the first to reach its rated loading as the total kVA load is raised. The maximum
permissible kVA loading of the two in parallel without overloading anyone is then given by
S1(rated) =
or
Z2
S (max)
| Z1 + Z 2 | L
SL(max) = S1 (rated)
| Z1 + Z 2 |
Z2
(3.90b)
Similarly if
Z2 < Z1
then
S1 (rated )
S2 (rated )
SL(max) = S2 (rated)
| Z1 + Z 2 |
Z1
(3.91a)
(3.91b)
In either case (Eq. (3.90a) or (3.91a))
SL(max) < S1(rated) + S2(rated)
(3.92)
Unequal Voltage Ratios
It has already been mentioned that a small difference in voltage ratios can be tolerated in the parallel operation
of transformers. Let E1 and E2 be the no-load secondary emfs of two transformers in parallel. With reference
to Fig. 3.58, if a load current I L is drawn at voltage VL , two mesh voltage balance equations can be written
as
E1 = I1Z1 + I L Z L = I1Z1 + ( I1 + I 2 ) Z L
and
\
E2 = I 2 Z 2 + I L Z L = I 2 Z 2 + ( I1 + I 2 ) Z L
E1 - E2 = I1Z1 - I 2 Z 2
(3.93)
(3.94)
(3.95)
On no-load I L = 0, so that the circulating current between the two transformers is given by
I1 = - I 2 =
E1 - E2
Z1 + Z 2
On short-circuit
I1 =
E1
E
, I2 = 2
Z1
Z2
On loading
I1 =
( E1 - E2 ) + I 2 Z 2
Z1
(3.96)
(3.97)
Transformers
119
Substituting for I1: in Eq. (3.94) we get
È ( E1 - E2 ) + I 2 Z 2
˘
+ I2 ˙ ZL
E2 = I 2 Z 2 + Í
Z1
Î
˚
E2 Z1 - ( E1 - E2 ) Z L
I2 =
Z1Z 2 + Z L ( Z1 + Z 2 )
\
I1 =
Similarly
E1Z 2 - ( E1 - E2 ) Z L
Z1Z 2 + Z L ( Z1 + Z 2 )
(3.98)
(3.99)
Normally E1 and E2 are in phase or their phase difference is insignificant. Severe results of paralleling
transformers not belonging to the same phase-groups (say Y/Y and Y/D transformers) are immediately
obvious from Eq. (3.96) for no-load circulating current. When many transformers are in parallel, their load
sharing can be found out using the Millman theorem [2].
EXAMPLE 3.24 A 600-kVA, single-phase transformer with 0.012 pu resistance and 0.06 pu reactance is
connected in parallel with a 300-kVA transformer with 0.014 pu resistance and 0.045 pu reactance to share
a load of 800 kVA at 0.8 pf lagging. Find how they share the load (a) when both the secondary voltages are
440 V and (b) when the open-circuit secondary voltages are respectively 445 V and 455 V.
SOLUTION
(a) The pu impedances expressed on a common base of 600 kVA are
Z1 = 0.012 + j 0.06 = 0.061 –79°
Z 2 = 2(0.014 + j 0.045) = 0.094 –73°
Z1 + Z 2 = 0.04 + j 0.15 = 0.155 –75°
The load is
S L = 800(0.8 – j 0.6) = 800 ––37° kVA
From Eqs (3.86) and (3.87)
S1 = 800 – –37° ¥
S2 = 800 ––37° ¥
0.094 –73∞
= 485 ––39° = 377 – j 305.2
0.155 –75∞
0.061–79∞
= 315 ––33° = 264 – j171.6
0.155 –75∞
It may be noted that the transformers are not loaded in proportion to their ratings. At a total load of 800 kVA,
the 300 kVA transformer operates with 5% overload because of its pu impedance (on common kVA base) being
less than twice that of the 600 kVA transformer.
The maximum kVA load the two transformers can feed in parallel without any one of them getting overloaded
can now be determined. From above it is observed that the 300 kVA transformer will be the first to reach its fullload as the total load is increased. In terms of magnitudes
\
0.061
SL(max) = S2(rated) = 300 kVA
0.155
300 ¥ 0.155
SL(max) =
= 762.3 kV A
0.061
while the sum of the ratings of the two transformers is 900 kVA. This is consequence of the fact that the transformer
impedances (on common base) are not in the inverse ratio of their ratings.
120 Electric Machines
(b) In this case it is more convenient to work with actual ohmic impedances. Calculating the impedances referred to
secondary
440
= 0.0039 + j 0.0194
600 ¥ 1000
440
= 0.0198 –79°
440
Z 2 (actual) = (0.028 + j 0.09) ¥
= 0.009 + j 0.029
600 ¥ 1000
440
= 0.0304 –73°
Z1 (actual) = (0.012 + j 0.06) ¥
Z1 + Z 2 = 0.0129 + j 0.0484 = 0.05 –75°
The load impedance Z L must also be estimated. Assuming an output voltage on load of 440 V,
–3
–3
2
VL* I L ¥ 10 = (VL / Z L ) ¥ 10 = 800 ––37°
ZL =
\
(440) 2
800 ¥ 103 – - 37∞
= 0.242 –37°
= 0.1936 + j 0.1452
From Eqs (3.98) and (3.99)
I1 =
445 ¥ 0.0304 –73∞ - 10 ¥ 0.242 –37∞
0.0198 –79∞ ¥ 0.0304 –73∞ + 0.242 –37∞ ¥ 0.05 –75∞
= 940 ––34° A
445 - 0.0198 –79∞ - 10 ¥ 0.242 –37∞
0.0198 –79∞ ¥ 0.0304 –73∞ + 0.242 –37∞ ¥ 0.05 –75∞
= 883 ––44° A
I2 =
The corresponding kV As are
S1 = 440 ¥ 940 ¥ 10–3 ––34° = 413.6 ––34°
–3
S2 = 440 ¥ 883 ¥ 10 ––4444° = 388 ––44°
The total output power will be
413.6 cos 34° + 388 cos 44° = 621.5 kW
This is about 3% less than 800 ¥ 0.8 = 640 kW required by the load because of the assumption of the value of the
output voltage in order to calculate the load impedance.
The secondary circulating current on no-load is
( E1 - E2 )
-10
=
= – 200 A
0.05
| Z1 + Z 2 |
which corresponds to about 88 kVA and a considerable waste as copper-loss.
Transformers may be built with a third winding, called the tertiary, in addition to the primary and secondary.
Various purposes which dictate the use of a tertiary winding are enumerated below:
(i) To supply the substation auxiliaries at a voltage different from those of the primary and secondary
windings.
Transformers
121
(ii) Static capacitors or synchronous condensers may be connected to the tertiary winding for reactive
power injection into the system for voltage control.
(iii) A delta-connected tertiary reduces the impedance offered to the zero sequence currents thereby
allowing a larger earth-fault current to flow for proper operation of protective equipment. Further, it
limits voltage imbalance when the load is unbalanced. It also permits the third harmonic current to
flow thereby reducing third-harmonic voltages.
(iv) Three windings may be used for interconnecting three transmission lines at different voltages.
(v) Tertiary can serve the purpose of measuring voltage of an HV testing transformer.
When used for purpose (iii) above the tertiary winding is called a stabilizing winding.
The star/star transformer comprising single-phase units or a single unit with a 5-limb core offers high
reactance to the flow of unbalanced load between the line and neutral. Any unbalanced load can be divided
into three 3-phase sets (positive, negative and zero sequence components). The zero-sequence component
(cophasal currents on three lines, I0 = In/3) caused by a line-to-neutral load on the secondary side cannot
be balanced by primary currents as the zero-sequence currents cannot flow in the isolated neutral starconnected primary. The zero-sequence currents on the secondary side therefore set up magnetic flux
in the core. Iron path is available for the zero sequence flux* in a bank of single-phase units and in
the 5-limb core and as a consequence the impedance offered to the zero-sequence currents is very high
(0.5 to 5 pu) inhibiting the flow of these currents. The provision of a delta-connected tertiary permits the
circulation of zero-sequence currents in it, thereby considerably reducing the zero-sequence impedance.
This is illustrated in Fig. 3.61.
I0
0
I0
I0
0
0
In = 3l0
Stabilizing
tertiary winding
I0
Fig. 3.61
Equivalent Circuit
The equivalent circuit of a 3-winding transformer can be represented by the single-phase equivalent circuit
of Fig. 3.62 wherein each winding is represented by its equivalent resistance and reactance. All the values
are reduced to a common rating base and respective voltage bases. The subscripts 1, 2 and 3 indicate the
primary, secondary and tertiary respectively. For simplicity, the effect of the exciting current is ignored in the
equivalent circuit. It may be noted that the load division between the secondary and tertiary is completely
* In a 3-limb core the zero-sequence flux (directed upwards or downwards in all the limbs) must return through the
air path, so that only a small amount of this flux can be established; hence a low zero-sequence reactance.
122 Electric Machines
arbitrary. Three external circuits are connected between terminals 1, 2 and 3 respectively and the common
terminal labelled 0. Since the exciting current is neglected, I1 + I 2 + I 3 = 0.
Z2
1
I1
Z1
l2
2
+
A
Z3
+
V1
l3
3
+
V2
V3
–
–
–
Common 0
Fig. 3.62
The impedance of Fig. 3.62 can be readily obtained from three simple short-circuit tests. If Z12 indicates
the SC impedance of windings 1 and 2 with winding 3 open, then from the equivalent circuit,
Similarly
where
Z12 = Z1 + Z 2
(3.100)
Z 23 = Z 2 + Z 3
Z13 = Z1 + Z 3
(3.101)
(3.102)
Z 23 = SC impedance of windings 2 and 3 with winding 1 open.
Z13 = SC impedance of windings 1 and 3 with winding 2 open.
All the impedances are referred to a common base.
Solving Eq. (3.100) to Eq. (3.102) yields
1
Z1 = ( Z12 + Z13 - Z 23 )
2
1
Z 2 = ( Z 23 + Z12 - Z13 )
2
1
Z 3 = ( Z13 + Z 23 - Z12 )
2
(3.103)
(3.104)
(3.105)
The open-circuit test can be performed on anyone of the three windings to determine the core-loss,
magnetizing impedance and turn-ratio.
EXAMPLE 3.25 The primary, secondary and tertiary windings of a 50-Hz, single-phase, 3-winding
transformer are rated as 6.35 kV, 5 MVA; 1.91 kV, 2.5 MVA; 400 V, 2.5 MVA respectively. Three SC tests on
this transformer yielded the following results:
(i) Secondary shorted, primary excited: 500 V, 393.7 A
(ii) Tertiary shorted, primary excited: 900 V, 393.7 A
(iii) Tertiary shorted, secondary excited: 231 V, 21 312.1 A
Resistances are to be ignored.
(a) Find the pu values of the equivalent circuit impedances of the transformer on a 5 MVA, rated voltage
base.
Transformers
123
(b) Three of these transformers are used in a 15 MVA, Y-Y-D, 3-phase bank to supply 3.3 kV and 400 V
auxiliary power circuits in a generating plant. Calculate the pu values of steady-state short-circuit
currents and of the voltage at the terminals of the secondary windings for a 3-phase balanced shortcircuit at the tertiary terminals. Use 15 MVA, 3-phase rated voltage base.
SOLUTION
(a) Let us first convert the SC data to pu on 5 MVA base/phase.
VB = 6.35 kV
For primary,
IB =
5000
= 787.4 A
6.35
VB = 1.91 kV
5000
IB =
= 2617.8A
1.91
Converting the given test data to pu yields:
For secondary,
Test No.
Windings involved
V
I
1
2
3
P and S
P and T
S and T
0.0787
0.1417
0.1212
0.5
0.5
0.5
From tests 1,2 and 3, respectively. 0.0787
X12 =
0.0787
= 0.1574 pu
0.5
X13 =
0.1417
= 0.2834
0.5
X23 =
0.1212
= 0.2424 pu
0.5
From (3.103) – (3.105)
X1 = 0.5(0.1574 + 0.2834 – 0.2424) = 0.0992 pu
X2 = 0.5(0.2424 + 0.1574 – 0.2834) = 0.05825 pu
X3 = 0.5(0.2834 + 0.2424 – 0.1574) = 0.1842 pu
(b) The base line-to-line voltage for the Y-connected primaries is 3 ¥ 6.35 = 11 kV, i.e. the bus voltage is 1 pu.
From Fig. 3.62, for a short-circuit at the terminals of the tertiary, V3 = 0. Then
V1
V1
1.00
=
ISC =
=
= 3.53 pu
X1 + X 3
X13
0.2834
SC current primary side = 3.53 ¥ 787.4 = 2779.5 A
SC current tertiary side = 3.53 ¥
5000 ¥ 1000
= 76424 A (line current)
400 / 3
Neglecting the voltage drops due to the secondary load current, the secondary terminal voltage is the voltage at
the junction point A (Fig. 3.63), i.e
VA = ISC X3 = 3.53 ¥ 0.1842 = 0.6502 pu
VA(actual) = 0.6502 ¥ 1.91 3 = 2.15 kV (line-to-line)
124 Electric Machines
0.05825
0.0992
A
0.1842
V2
V1
V3 = 0
lsc
Fig. 3.63
3.16
PHASE CONVERSION
Phase conversion from three to two phase is needed in special cases, such as in supplying 2-phase electric
arc furnaces.
The concept of 3/2-phase conversion follows from the voltage phasor diagram of balanced 3-phase
supply shown in Fig. 3.64(b). If the point M midway on VBC could be located, then VAM leads VBC by 90°. A
2-phase supply could thus be obtained by means of transformers; one connected across AM, called the teaser
transformer and the other connected across the lines B and C. Since VAM = ( 3 /2) VBC, the transformer
primaries must have 3 N1/2 (teaser) and N1 turns; this would mean equal voltage/turn in each transformer.
A balanced 2-phase supply could then be easily obtained by having both secondaries with equal number of
turns, N2. The point M is located midway on the primary of the transformer connected across the lines B and
C. The connection of two such transformers, known as the Scott connection, is shown in Fig. 3.64(a), while
the phasor diagram of the 2-phase supply on the secondary side is shown in Fig. 3.64(c).
The neutral point on the 3-phase side, if required, could be located at the point N which divides the primary
winding of the teaser in the ratio 1 : 2 (refer Fig. 3.64(b)).
A
lA
a2 la
3 N1/2
N
+
N2
lA
a1
B
C
lA /2
l l2
lB A
lC
N1l2
M
Va
–
A
IBC
N1l2
Va
b1
b2
Ib
+
N
N2
Vb
(a)
–
C
M
(b)
Fig. 3.64 Scott connection
Vb
B
(c)
Transformers
125
Load Analysis
If the secondary load currents are I a and I b , the currents can be easily found on the 3-phase side from
Fig. 3.64(a).
2 N2
IA =
I BC =
3 N1
Ia =
2
3
I a (for N1/N2 = 1)
N2
I b = I b (for N1/N2 = 1)
N1
I B = I BC - I A /2
I C = – I BC - I A /2
The corresponding phasor diagram for balanced secondary side load of unity power factor is drawn
in Fig. 3.65 from which it is obvious that the currents drawn from the 3-phase system are balanced and
cophasal with the star voltages. The phasor diagram for the case of an unbalanced 2-phase load is drawn
in Fig. 3.66.
A
IA = 2 3
–lBC
Va
lBC = 1
–lAl2
la
1
–IA/2 = 1/ 3
C
B
lC
1
IB = 1+1/3 = 2/ 3
lb
Fig. 3.65
–lBC
Va
A
la
lA
–lA/2
fa
lC
fb
C
Vb
B
IBC
IB
Fig. 3.66
lA/2
lb
Vb
126 Electric Machines
Three/One-phase Conversion
A single-phase power pulsates at twice the frequency, while the total power drawn by a balanced 3-phase
load is constant. Thus a 1-phase load can never be transferred to a 3-phase system as a balanced load without
employing some energy-storing device (capacitor, inductor or rotating machine). Suitable transformer
connections can be used in distributing a 1-phase load on all the three phases though not in a balanced
fashion. For large 1-phase loads, this is better than allowing it to load one of the phases of a 3-phase system.
A variety of transformer connections are possible. Figure 3.67(a) shows how Scott-connected transformers
could be used for this purpose and Fig. 3.67(b) shows the corresponding phasor diagram.
B
A
C
2
3–1
3
3
I
A
3+1
I
3
1
IB = 1 –
=
3
3
Va
I
IA =
l
3–1
I
V
2
l=1
3
lBC = 1
V
B
C
IC = 1 +
(a)
VB
–lBC = 1
1
=
3+1
3
(b)
3
Fig. 3.67
Three Phase/Six-phase Conversion
Each secondary phase is divided into two equal halves with polarity labelling as in Fig. 3.50. Six-phase
voltages (characteristic angle 360°/6 = 60°) are obtained by means of two stars in phase opposition, each
star being formed from three respective half-windings as shown in Fig. 3.68. This connection is employed in
rectifiers and thyristor circuits where a path for the dc current is needed.
b1
c4
C
C2
B2
B
A2
a4
a1
b4
c1
A
Fig. 3.68
EXAMPLE 3.26 Two single-phase furnaces A and B are supplied at 100 V by means of a Scott-connected
transformer combination from a 3-phase 6600 V system. The voltage of furnace A is leading. Calculate the
line currents on the 3-phase side, when the furnace A takes 400 kW at 0.707 pf lagging and B takes 800 kW
at unity pf.
Transformers
SOLUTION
127
With reference to Fig. 3.64(a)
N1
6600
=
= 66
N2
100
3
N1
2
= 57.16
N2
\
Furnace currents are
Ia =
400 ¥ 1000
= 5658 A;
100 ¥ 0.707
fa = 45° lagging
800 ¥ 1000
= 8000 A;
fb = 0°
100 ¥ 1
Furnace voltages and currents are drawn in the phasor diagram of Fig. 3.69(a).
Ib =
Va
IA = 99A
45°
la
lb
49.5A
Vb
IBC = 121.2
–IBC = 121.2
45°
IB
IC
45°
49.5 A
(a)
(b)
Fig. 3.69
On the 3-phase side
5658
= 99 A
57.16
8000
=
= 121.2 A
66
IA =
IBC
From the phasor diagram of Fig. 3.69(b)
I B = 121.2 – 49.5(0.707 + j 0.707)
= 86.2 – j 35
or
I B = 93 A
or
I C = 121.2 + 49.5 (0.707 – j 0.707)
= 156.2 – j 35
IC = 160 A
3.17 TAP CHANGING TRANSFORMERS
Voltage variation in power systems is a normal phenomenon owing to the rapid growth of industries and
distribution network. System voltage control is therefore essential for:
(i) Adjustment of consumers’ terminal voltage within prescribed limits.
128 Electric Machines
(ii) Control of real and reactive power flow in the network.
(iii) Periodical adjustment (1–10%) to check off-set load variations.
Adjustment is normally carried out by off-circuit tap changing, the common range being 5% in 2.5% steps.
Daily and short-time control or adjustment is carried out by means of on-load tap changing gear.
Besides the above, tapping are also provided for one of the following purposes:
(i)
(ii)
(iii)
(iv)
(v)
For varying the secondary voltage.
For maintaining the secondary voltage constant with a varying primary voltage.
For providing an auxiliary secondary voltage for a special purpose, such as lighting.
For providing a low voltage for starting rotating machines.
For providing a neutral point, e.g. for earthing.
The principal tapping is one to which the rating of the winding is related. A positive tapping means more,
and a negative tapping implies less turns than those of the principal tap. Tap changing may be achieved in one
of the three conditions, viz.
(i) voltage variation with constant flux and constant voltage turn,
(ii) with varying flux,
(iii) a mix of (i) and (ii). In (i) the percentage tapping range is same as the voltage variation.
Location
The taps may be placed on the primary or secondary side which partly depends on construction. If tappings
are near the line ends, fewer bushings insulators are required. If the tappings are placed near the neutral ends,
the phase-to-phase insulation conditions are eased.
For achieving large voltage variation, tappings should be placed near the centres of the phase windings
to reduce magnetic asymmetry. However, this arrangement cannot be put on LV windings placed next to the
core (as in core type transformer) because of accessibility and insulation considerations. The HV winding
placed outside the LV winding is easily accessible and can, thus, be tapped easily.
It is not possible to tap other than an integral number of turns and this may not be feasible with LV side
tappings. For example 250 V phase winding with 15 V/turn cannot be tapped closer than 5%. It is therefore
essential to tap the HV windings which is advantageous in a step-down transformer.
Some of the methods of locating tappings are depicted in Fig. 3.70(a) and (b).
(a) Taps at one end for small transformers
Fig. 3.70
(b) Large transformer taps centrally placed for
both delta and star transformer
Location of transformer tappings
Transformers
129
Axial mmf unbalance is minimized by thinning out the LV winding or by arranging parts of the winding
more symmetrically. For very large tapping ranges a special tapping coil may be employed.
Tap changing causes changes in leakage reactance, core loss, I 2R loss and perhaps some problems in
parallel operation of dissimilar transformers.
The cheapest method of changing the turn ratio of a transformer is the use of off-circuit tap changer. As the
name indicates, it is required to deenergize the transformer before changing the tap. A simple no-load tap
changer is shown in Fig. 3.71. It has eight studs marked one to eight. The winding is tapped at eight points.
The face plate carrying the suitable studs can be mounted at a convenient place on the transformer such as
upper yoke or located near the tapped positions on the windings. The movable contact arm A may be rotated
by handwheel mounted externally on the tank.
If the winding is tapped at 2% intervals, then as the rotatable arm A is moved over to studs 1, 2; 2, 3; …
…6, 7; 7, 8 the winding in circuit reduces progressively by it from 100% with arm at studs (1, 2) to 88% at
studs (7, 8).
The stop F which fixes the final position of the arm A prevents further anticlockwise rotation so that stud
1 and 8 cannot be bridged by the arm. Adjustment of tap setting is carried out with transformer deenergized.
For example, for 94% tap the arm is brought in position to bridge studs 4 and 5. The transformer can then be
switched on.
7
5
F
8
3
7
6
1
1
2
2
4
A
3
5
4
6
8
Fig. 3.71
No-load tap changer
To prevent unauthorized operation of an off-circuit tap changer, a mechanical lock is provided. Further,
to prevent inadvertent operation, an electromagnetic latching device or microswitch is provided to open the
circuit breaker so as to deenergize the transformer as soon as the tap changer handle is moved; well before
the contact of the arm with the stud (with which it was in contact) opens.
130
Electric Machines
On-load Tap Changing
On-load tap changers are used to change the turn ratio of transformer to regulate system voltage while the
transformer is delivering load. With the introduction of on-load tap changer, the operating efficiency of
electrical system gets considerably improved. Nowadays almost all the large power transformers are fitted
with on-load tap changer. During the operation of an on-load tap changer the main circuit should not be
opened to prevent (dangerous) sparking and no part of the tapped winding should get short-circuited. All
forms of on-load tap changing circuits are provided with an impedance, which is introduced to limit shortcircuit current during the tap changing operation. The impedance can either be a resistor or centre-tapped
reactor. The on-load tap changers can in general be classified as resistor or reactor type. In modern designs
the current limiting is almost invariably carried out by a pair of resistors.
On-load tap changing gear with resistor transition, in which one winding tap is changed over for each
operating position, is depicted in Fig. 3.72. The figure also shows the sequence of operations during the
transition from one tap to the next (adjoining) (in this case from tap 4 to tap 5). Back-up main contractors are
provided which short-circuit the resistor for normal operation.
6
5
5
5
4
4
3
4
l/2 – i
l
2
l/2 + i
1
r1
l
r2
l
Diverter
switch
(a) Tap 4
r1
i
r2
l
(b) Tap 4 and r1
5
l
(c) Taps 4 and 5
5
r2
r2
l
l
l
(d) Tap 5 and r2
l
(e) Tap 5
Fig. 3.72 Simple switching sequence for on-load tap changing
To ensure that the transition once started gets completed, an energy storage (usually a spring device) is
provided which acts even if the auxiliary power supply happens to fail. In resistor-aided tap changing the
current break is made easier by the fact that the short-circuit resistor causes the current to be opened to have
unity power factor.
On-load tap changer control gear can be from simple push-button initiation to complex automatic control
of several transformers operating in parallel. The aim is to maintain a given voltage level within a specified
tolerance or to raise it with load to compensate for the transmission line voltage drop. The main components
are an automatic voltage regulator, a time delay relay, and compounding elements. The time delay prevents
Transformers
131
unwanted initiation of a tap change by a small transient voltage fluctuation. It may be set for a delay upto
1 min.
At present tap changers are available for the highest insulation level of 1475 kV (peak) impulse and
630 kV power frequency voltage. Efforts are underway to develop tap changers suitable for still higher
insulation levels. More compact tap changers with high reliability and performance are being made by
employing vacuum switches in the diverter switch. Also, now thyristorized tap changers are available for
special applications where a large number of operations are desired.
3.18 VOLTAGE AND CURRENT TRANSFORMERS
These transformers are designed to meet the specific need of measurement and instrumentation systems,
which accept voltages in the range of 0–120 V and currents upto 5 A. Power system voltages can be as high
as 750 kV and currents upto several tens of kA. Their measurement requires accurate ratio voltage and current
transformations, which is accomplished by potential and current transformers.
Potential Transformer (PT)
It must transform the input voltage accurately to output voltage both in magnitude and phase. The impedance
presented by the instrument on measurement system to the transformer output terminals is called burden. It
is mainly resistive in nature and has a large value, e.g. the impedance (practically a resistance) of a voltmeter.
The circuit model of a PT is drawn in Fig. 3.73. It is the same as that of an ordinary transformer but ideally
should have
V1
N
= 1 – 0°
V2
N2
I1
V1
R1
R¢2
X1
X¢2
l2
N 1: N 2
Xm
V2
Zb
(burden)
Fig. 3.73 Circuit model of a PT
The current drawn by the burden causes a voltage drop in (R¢2 + j X¢2) and this current referred to primary
plus the magnetizing current (all phasors) causes a voltage drop in (R1 + j X1). Therefore V2 /V1 differs
from the desired value (N1/N2) in magnitude and phase resulting in magnitude and phase errors. The errors
are to be kept within the limit defined by the precision required. In order to achieve this a PT is designed
and constructed to have low leakage reactance, low loss and high magnetizing reactance (low magnetizing
current).
Low reactance is achieved by interlacing primary and secondary both on core limb. High magnetizing
reactance requires minimum iron path and high permeability steel. Low loss requires low-loss steel and very
thin laminations.
Most important thing for low PT errors is to make the burden (Zb) as high as feasible.
132
Electric Machines
Current Transformer (CT)
It is the current ratio transformer meant for measuring large currents and provide a step down current to
current measuring instruments like an ammeter. Such instruments present a short-circuit to the CT secondary.
It means that burden Zb ª 0. An ideal CT current ratio is
N1
I2
=
–0°
N2
I1
Causes of CT errors and their remedy are the same as for a PT discussed earlier in this section.
In power system applications CT has a single-turn primary which
Line
current
is the line itself as shown in Fig. 3.74. The secondary is rated 1–5 A.
The burden impedance (which in fact is practically resistive)
cannot be allowed to exceed beyond a limit. Most important
precaution in use of a CT is that in no case should it be open
circuited (even accidently). As the primary current is independent
A
of the secondary current, all of it acts as a magnetizing current when
Burden
the secondary is opened. This results in deep saturation of the core
Fig. 3.74 Current transformer for
which cannot be returned to the normal state and so the CT is no
power-line current
longer usable.
EXAMPLE 3.27 A 250 A/5 A, 50 Hz current transformer has the following parameters as seen on 250 A
side
X1 = 505 mW,
X¢2 = 551 mW,
Xm = 256 mW
R1 = 109 mW,
R¢2 = 102 mW
(a) The primary is fed a current of 250 A with secondary shorted. Calculate the magnitude and phase of
the secondary current.
(b) Repeat part (a) when the secondary is shorted through a resistance of 200 mW.
SOLUTION
(a) The equivalent circuit with secondary shorted is drawn in Fig. 3.75.
I1
R1
X1
R¢2
Xm
N2 250
= 50
=
N1
5
Fig. 3.75
By current division
Ê
ˆ
j Xm
I
I 2¢ = Á
Ë R2¢ + j X 2¢ + j X m ˜¯ 1
X¢2
I¢2
Transformers
133
I1 = 250 –0° A
I 2¢ =
j 256 ¥ 103
109 + j (551 + 256 ¥ 103 )
¥ 256
Ê
ˆ
j 256
I 2¢ = Á
¥ 250 A
Ë 256.51–89.975 ˜¯
I2 =
Ê N1 ˆ
j 256
1
I¢ =
¥ 5; I2 = Á
¥ 250° = 5 A
Ë N 2 ˜¯ 2 50
256.5 –89.975∞
I2 = 4.989 A phase = 0.025° (negligible)
5 - 4.989
Error =
¥ 100 = 0.22 %
5
2
(b)
Ê 1ˆ
R¢b = 200 mW in series with R¢2, X¢2, Rb = Á ˜ ¥ 200 = 0.08 mW
Ë 50 ¯
j 256 ¥ 103
¥ 250 A
I 2¢ =
(109 + 0.08) + j (551 + 256 ¥ 103 )
I 2¢ =
256 –90∞
¥ 5 = 4.989 –0.025°
256.551–89.975∞
No change as R¢b = 0.08 mW is negligible
EXAMPLE 3.28
HV side.
A 6000 V/100 V, 50 Hz potential transformer has the following parameters as seen from
R1 = 780 W
R¢2 = 907 W
X1 = 975 W
X 2¢ = 1075 W
Xm = 443 kW
(a) The primary is excited at 6500 V and the secondary is left open. Calculate the secondary voltage,
magnitude and phase.
(b) The secondary is loaded with 1 k W resistance, repeat part (a)
(c) The secondary is loaded with 1 k W reactance, repeat part (a)
SOLUTION
The potentiometer equivalent circuit as seen from HV side is drawn on Fig. 3.76.
Turn ratio,
(a) Secondary open;
6000
N1
=
= 60
1000
N2
Zb =
V1 = 6500 V
Ê j Xm ˆ
V 2¢ = Á
V
Ë R1 + j X1 ˜¯ 1
V¢2 =
j 443 ¥ 103
780 + j 443 ¥ 103
¥ 6500
V¢2 = 1 –0.1° ¥ 6500 = 6500 –0.1° V
Ê N2 ˆ
V2 = 6500 ¥ Á
= 108 V, –0.1°
Ë N1 ˜¯
134
Electric Machines
X1
R1
Vm
R¢2
X¢2
+
+
Zm
V1
Xm
Z¢b
V2¢
–
–
Fig.3.76
Zb = Rb = 1 kW, R¢b = (60)2 ¥ 1 = 3600 kW
(b)
As R¢b is far larger than R¢2 and X¢2, we can ignore R¢2, X¢2
Then
Z m = jXm || R¢b
Ê j 443 ¥ 3600 ˆ
Z m = ÁË 3600 + j 443 ˜¯ = 439.7 –83° = 53.6 + j 436.4 kW
(R1 + j X1) + Z m = (0.78 + j 0.975) + (53.6 + j 436.4)
= 54.38 + j 473.4 = 440.77 –82.9° kW
È
˘
Zm
˙ V1
Vm = Í
(
R
+
j
X
)
+
Z
1
1
m
Î
˚
È 439.7 –83∞ ˘
˙ ¥ 6500 = 6484 –0.1°
Vm = Í
Î 440.77 –82.9∞ ˚
V2¢ = Vm = 6484 –0.1° V
V2 =
Exact value should be
6484
= 108.07 V ; phase 0.1°
60
6500
= 108.33
60
108.33 - 108.07
= 0.26%
Error =
108.33
Z b = jXb ; Xb = 1 kW
(c)
Z b¢ = j 3600 kW
Ignoring R¢2, X¢2 in comparison
Z m = j 443 || j 3600 = j
443 ¥ 3600
= j 394.45 kW
443 + 3600
(R1 + jX1) + Z m = (0.78 + j 0.975) + j 394.45 = 0.78 + j 395.425
= 395.426 –89.89°
Transformers
V2¢ = Vm =
V2 =
135
394.45 –90∞
¥ 6500 = 6484 –0.01°
395.426 –89.89
6484
= 108.07, phase 0.01°
60
V2 is same as in resistive load (part (b) except for change in phase. In any case phase is almost zero.
It is used at the output stage of audio frequency electronic amplifier for matching the load to the output
impedance of the power amplifier stage. Here the load is fixed but the frequency is variable over a band
(audio, 20 Hz to 20 kHz), the response being the ratio V2/V1. A flat frequency response over the frequency
band of interest is most desirable. The corresponding phase angle (angle of V2 w.r.t. V1) is called phase
response. A small angle is acceptable.
The transformer is used in electronic circuits (control, communication, measurement etc.) for stepping up
the voltage or impedance matching. They are normally small in size and have iron cores. It is essential that
distortion should be as low as possible.
Figure 3.77 shows the exact circuit model of a
wL1
wL¢2
r1
r¢2
transformer with frequency variable over a wide range.
Here the magnetizing shunt branch is drawn between +
+
I0
primary and secondary impedances (resistance and
V2
Bm = wLm
Gi
Load
leakage reactance). Also represented is the shunting V1
Bs
effect of transformer windings stray capacitance Cs.
Bs = 1/wCs
–
In the intermediate frequency (IF) range the shunt –
IT
branch acts like an open circuit and series impedance
drop is also negligibly small such that V2/V1 remains
Fig. 3.77
fixed (flat response) as in Fig. 3.77.
V2lV1
LF
range
Phase
2
10
HF
range
IF
range
50
100
angle
103
0
104
105
log f
Fig. 3.78
/V
In the LF (low frequency) region the magnetizing susceptance is low and draws a large current with a
consequent large voltage drop in (r1 + jwL1). As a result V2/V1 drops sharply to zero as Bm = 0 (Fig. 3.78). In
the HF (high frequency) region Bs = 1/w Cs (stray capacitance susceptance) has a strong shunting effect and
V2/V1 drops off as in Fig. 3.78, which shows the complete frequency response of a transformer on logarithmic
frequency scale.
136
Electric Machines
3.20 GROUNDING TRANSFORMER
In case the neutral of the power transformer is not available for grounding (e.g. when a D-D transformer is
used), a special Y-D transformer is employed only for neutral grounding as shown in Fig. 3.79(a). Such a
transformer is called a grounding transformer and it is a step down transformer. The star connected primaries
are connected to the system and its neutral is grounded. The secondaries are in delta and generally do not
supply any load but provide a closed path for triplen harmonic currents to circulate in them. Under balanced
conditions the current in a grounding transformer is its own exciting current. Under fault conditions (such
as LG fault) large current may flow in it. Hence a grounding transformer should be of sufficient rating to
withstand the effects of LG (line to ground) faults. Transformers with ‘zigzag’ connection are sometimes
employed as grounding transformers as shown in Fig. 3.78(b).
R
R
Y
B
N
y
B
N
Fig. 3.79(a) Grounded transformer connections
Fig. 3.79(b) Zig-zag grounding connections
Welding transformer is basically a step-down transformer with high reactance both in primary and secondary.
Its primary and secondary winding are placed in separate limbs or in the same limbs but spaced distance
apart. This high reactance causes steeply drooping V-I characteristics. That is with increase in current, the
leakage flux increase and the induced emf will come down. This is why the increase in primary or secondary
current increases the reactance voltage drop across the respective windings, which is essential to limit the
welding current as the weld is practically a short circuit. The schematic of a welding transformer is shown
in Fig. 3.80.
Electrodes
Py
Sy
Fig. 3.80 Welding transformer
Transformers
137
3.22 TRANSFORMER AS A MAGNETICALLY COUPLED CIRCUIT
In Sections 3.3, 3.4 and 3.5 the equivalent circuit of a transformer was developed in terms of primary and
secondary resistances, leakage reactances, magnetizing shunt reactance, core loss shunt resistance and an
ideal transformer. This development was through magnetizing current needed to setup core flux and emf s
induced in the winding by sinusoidally varying core flux, the concept of the ideal transformer and representing
core loss by an equivalent shunt resistance. The following section treats the transformer as mutually coupled
circuit wherein voltages and currents are related in terms of resistances and inductances. Here the core
is assumed to have constant permeability so the magnetic saturation is neglected. This model gives more
physical meaning of equivalent circuit parameters, in terms of transformer magnetic field.
A two winding transformer is shown in Fig. 3.81, where R1 and R2 are primary and secondary winding
resistances.
fc1
fc2
R1
+
R2
+
i1
v1
i2
N1
fl
1
fl
N2
v2
2
–
–
Fig. 3.81 Two winding transformer
The primary current i1 into the dotted terminal produces
core flux (mutual flux) = fc1
leakage flux = fl1
total flux, f1 = fc1 + fl1
in the direction indicated. The secondary current i2 out of the dotted terminal produces
core flux (mutual flux) = fc2
leakage flux = fl2
total flux, f2 = fc2 + fl2
The core flux fc2 is in apposite directions to fc1.
Then net core flux due to i1 and i2 is f = fc1 – fc2
The primary and secondary windings have self inductances L1 and L2 and mutual inductance M. In a
bilateral circuit
M12 = M21 = M
refer Section 2.4
138
Electric Machines
It can be easily proved as the core offers the same permeance to (N1i1) and (N2i2).
The transformer as a coupled circuit is drawn in Fig. 3.82(a). From the basic laws the kVL equations of
primary and secondary are written below.
v1 = R1i1 + L1
Primary
di1
di
– M 2 ; minus sign results from the fact that
dt
dt
i2 flows out of dotted terminal
v2 = M
Secondary
di1
di
– L2 2 – R2i2
dt
dt
(3.106a)
(3.106b)
we will now refer the secondary side quantities to the primary side.
a =
Turn ratio,
N1
N2
v¢2 = a v2
then
i¢2 =
i2
a
Equation (3.106(a)) is then written as
v1 = R1i1 + L1
Adding and subtracting aM
di1
d Êi ˆ
- aM Á 2 ˜
dt
dt Ë a ¯
di1
, we get
dt
v1 = R1i1 + (L1 – aM)
di1
d Ê
i ˆ
+ aM Á i1 - 2 ˜
dt
dt Ë
a¯
(3.107)
It is easily recognized that
i1 –
i2
= im, the core magnetizing current
a
(3.108)
Converting secondary equation to primary side
av2 = aM
Adding and subtracting aM
di1
d Êi ˆ
Êi ˆ
- a 2 L2 Á 2 ˜ - a 2 R2 Á 2 ˜
Ë a¯
dt
dt Ë a ¯
(3.109)
d Ê i2 ˆ
and reorganizing we get
dt ÁË a ˜¯
v¢2 = av2 = aM
d Ê
i ˆ
d Êi ˆ
Êi ˆ
i1 - 2 ˜ - (a 2 L2 - aM ) Á 2 ˜ - a 2 R2 Á 2 ˜
Ë a¯
dt ÁË
a¯
dt Ë a ¯
From Eqs (3.107) and (3.110) the equivalent circuit is drawn in Fig. 3.82(b).
(3.110)
Transformers
R1
+
v1
i
i m = ÊË i1 - 2 ˆ¯
a
v2 v1
M
L1
a2R2
i2
a
i1
i2
i1
a2L2 – aM
L1 – aM
R1
R2
139
aM
v¢2 = av2
L2
–
(a)
(b)
Fig. 3.82
We now need to recognize leakage inductance. Using the basic definitions
L1 – aM =
N1 (fc1 + fl ) N1 N 2 fc1 N1 fl1
◊
=
= l1
i
N2
i1
i1
Mˆ
Ê
È N (f + f l 2 ) N 2 N1 fc 2 ˘
(a2L2 – aM) = a2 Á L2 - ˜ = a 2 Í 2 c 2
◊
Ë
a¯
i2
N1
i2 ˙˚
Î
ÊN f ˆ
= a 2 Á 2 l 2 ˜ = a2l2
Ë i2 ¯
aM =
N1 N 2 fc1 N1 fc1
◊
=
= Lm1
N2
i1
i1
Equations (3.107) and (3.110) can now be written as
di
di
v1 = R1i1 + li 1 + Lm1 m
dt
dt
av2 = Lm1
(3.111a)
dim
d Ê i2 ˆ
Ê i2 ˆ
2
- a 2 l2
ÁË ˜¯ - a R2 ÁË ˜¯
dt
dt a
a
(3.111b)
From these equations the equivalent circuit is drawn in Fig. 3.83.
a2 l2
l1
R1
a2 R2
+
+
i1
i2
a
v1
im
Lm1
–
av2 = v2¢
–
Fig. 3.83 Conductively coupled equivalent circuit of transformer
140
Electric Machines
Sinusoidal Applied Voltage When the sinusoidal applied voltage is considered, the equivalent leakage
reactance X1 and X2 are equal to wl1 and wl2 respectively and magnetizing reactance Xm is equal to wLm1. The
shunt resistance Rc is takes care of core losses and it is parallel with Xm. Now the equivalent circuit is shown
in Fig. 3.84 is similar to Figure wherein voltages and currents are phasors.
R1
X 2¢
X1
I1
I0
V1
R2¢
I2/a
Ii
Im
Ri
Xm
+
aV2 = V2¢
E1
–
Fig. 3.84
Equivalent circuit of a transformer with sinusoidal applied voltage
Induced emf in primary winding, E1 = Xm Im = wLm1 Im
It may be noted that Ri has been connected in parallel to Xm to account for core loss.
Ri =
E12
Pi
For the coupled-circuit of Fig. 3.82(a), Eqs 3.106(a) and (b) take the form
V1 = R1 I1 + jw L1 I1 - jw MI 2 Ô¸
˝
V2 = jw MI 2 - jw L2 I 2 - R2 I 2 ˛Ô
(3.112)
It is a measure of leakage fluxes in a magnetically coupled-circuit. We begin by
defining the coupling factor of each winding as
k=
mutual flux due to winding current
fc
=
total flux due to windingg current
f
The factor is less than unity as f = fc + fl.
For primary winding
k1 =
or
k1 =
fc1
( N 2fc1 )/ i1
=
f1
( N 2f1 )/ i1
Ê N1 ˆ M
M
= Á
Ë N 2 ˜¯ L1
Ê N 2 ˆ Ê N1f1 ˆ
ÁË N ˜¯ ÁË i ˜¯
1
1
(i)
(ii)
Transformers
141
Ê N2 ˆ M
◊
k2 = Á
Ë N1 ˜¯ L2
Similarly,
Taking the geometric mean yields the coupling coefficient as
k=
k1k2 =
M
L1 L2
M = k L1 L2 ; k < 1
or
(iii)
For a tight coupling k = 1 as fc1 = f1 and fc2 = f2, no leakage. It follows from Eqs (i) and (ii) that
Ê N2 ˆ M
Ê N1 ˆ M
ÁË N ˜¯ L = ÁË N ˜¯ L
1
2
2
1
N1
=
N2
or
L1
L2
To reduce primary and secondary voltage drop leakage flux of both
winding should be kept low which would result in high M (tight coupling). The methods for reducing leakage
flux have already been discussed in Section 3.2.
EXAMPLE 3.29 A transformer has turn ratio of a = 10. The results of two open-circuit tests conducted
on the transformer are given below:
(a) The primary on application of 200 V draws 4 A with secondary open circuited which is found to have
a voltage of 1950 V.
(b) The secondary on application of 2000 V draws 0.41 A with the primary open circuited.
Calculate L1 and L2 and coupling coefficient. What is the voltage of primary in part (b).
SOLUTION
(a)
Xm =
240
= 50 W, Xm = 2p f L1
4
L1 =
200
= 0.159 H
2p ¥ 50
1950 =
ymax =
2 p N2fmax =
2 pymax
1950
= 8.78 Wb–T
2p
142 Electric Machines
M=
E1 =
(b)
8.78
y max
=
= 1.55 H
2 ¥4
i1 (max)
2 p f N2fmax =
y max
= M,
i2 (max)
\
EXAMPLE 3.30
2 ¥ 0.42 ¥ 1.55
2 ¥ 0.41 ¥ 1.55 = 199.6 A
E1 =
2 p ¥ 50 ¥
L2 =
2000
1
◊
= 15.53 H
2 p ¥ 50 2 ¥ 0.41
k=
Coupling coefficient,
ymax =
2 p fymax
1.55
= 0.986
0.159 ¥ 15.53
A 150 kVA transformer 2400/240 V rating has the following parameters:
R1 = 0.2 W,
X1 = 0.45 W,
Ri = 10 kW
R2 = 2 ¥ 10–3
X2 = 4.5 ¥ 10–3
Xm = 1.6 kW (referred to HV)
Calculate the leakage inductances, magnetizing inductance, mutual inductance and self-inductances.
SOLUTION
a=
N1 2400
ª
= 10
N2
240
X1 = 2p f l1, l1 =
0.45
¥ 10–3 = 0.01433 mH
314
X2 = 2p f l2, l2 =
4.5 ¥ 10-3
= 0.01433 mH
314
Magnetizing inductance
2p f Lm1 = Xm = 1.6 ¥ 103
Lm1 = 5.096 H
Self inductances
l1 = L1 – Lm1
L1 = 5.096 + 0.01433 ¥ 10–3 = 5.096 H
Lm1 = aM, M =
Lm1
5.096
=
= 0.5096 H
a
10
Transformers
l2 = L2 –
M
a
L2 = l2 +
M
0.5096
= 0.01433 ¥ 10–3 +
a
10
143
= 0.05098 H
k=
Coupling factor
M
=
L1L2
0.5096
= 0.09998 ª 1
5.096 ¥ 0.05098
EXAMPLE 3.31 Solve Problem 3.8 using Matlab*. Also calculate % voltage regulation and h at full load
and 0.8 pf lagging.
SOLUTION Steps for computing circuit model parameters, voltage regulation and Efficiency at full load for a
Transformer using MATLAB.
Open-Circuit Test The equivalent circuit as seen on open-circuit test is given in Fig. 3.23(b).
Applied voltage = V1 (rated) Current drawn = 10
Power input = I0
Power input = P0 = Pi (core loss)
Y0 =
I0
P
, Gi = 02
V1
V1
Bm =
Y02 - Gi2
Short-Circuit Test The equivalent circuit as seen during short-circuit test is drawn in Fig. 3.23(b).
Applied voltage = Vsc (a fraction of rated value)
Current drawn = Isc (nearly full load value)
Power input = Psc = Pc (copper loss)
Z=
X=
Vsc
,
I sc
R=
Psc
( I sc ) 2
Z 2 - R2
Voltage Regulation
% voltage regulation =
voltage drop
¥ 100
rated secondary voltage at full load and speciffied pf
* For detailed write-up on MATLAB, the reader is encouraged to read Appendix G of the authors’ book “Modern
Power System Analysis”, 3rd ed. Tata McGraw-Hill, New Delhi, 2003.
144 Electric Machines
VR =
or,
I ( R cos f ± X sin f )
¥ 100; + for lagging pf;
V2
– for leading pf
where I = secondary current
R = equivalent resistance referred to secondary
X = equivalent reactance referred to secondary
f = power factor angle
Efficiency at full load
Efficiency at full load =
=
MATLAB PROGRAM
P=50000;
V1=2200;
V2=110;
V0=110;
I0=10;
P0=400;
Y0=I0./V0
Gi=P0./(V0^2)
Bm=sqrt (Y0^2-Gi^2)
Vsc=90;
Isc=20.5;
Psc=808;
Z=Vsc./Isc
R=Psc./Isc^2)
X=sqrt (Z^2-R^2)
TR=2200/110;
Gi_HV=Gi./(TR^2)
Bm_HV=Bm./(TR^2)
R_LV=R./(TR^2)
X _LV=X./(TR^2)
I2=P./V2
pf=0.8;
Full load output ¥ 100
Full load output + Core loss + Copper loss at full load
P
¥ 100
P + Pi + Pc
Transformers
Th=acos(pf)
dV=I2.*(R_LV.*cos(Th)+X_LV.*sin(Th))
VR=(dV./V2)*100
Pi=P0
Pc=Psc
EFF_Full_load=(P*100)./(P+Pi+Pc)
y0 =
0.0909
Gi =
0.0331
Bm =
0.0847
Z =
4.3902
R =
1.9227
X =
3.9468
Gi_HV =
8.2645e-005
Bm_HV =
2.1171e-004
R_LV =
0.0048
X_LV =
0.0099
I2 =
454.5455
Th =
0.6435
dV =
4.4389
VR =
4.0354
Pi =
400
Pc =
808
EFF_Full_load =
97.6410
Note: For manual solution, refer solved Problem 3.3 of the Authors’ book [76].
145
146
Electric Machines
A transformer is a static device comprising coupled coils (primary and secondary) wound on common
magnetic core. The arrangement transfers electric energy from one coil (primary) at a particular
voltage level to the other coil (secondary) at another voltage level via the magnetic flux carried by the
core.
In a transformer, all voltages and currents are sinusoidal. The device is bilateral i.e. electric energy can
be made to flow in either direction with reversal of roles of the two coils.
Ideal transformer
(a) The core is infinitely permeable and is lossless.
(b) Both windings have no resistance and there is no leakage flux; so no voltage drop in either winding.
Two types of transformer cores are commonly employed in practice - core type and shell type. In the
core type, the windings are wound around the two legs of a rectangular magnetic core, while in shell
type, the windings are wound on the central leg of a three legged core.
Transformer windings are made of solid or stranded copper or aluminium strip conductors. For
electronic transformers, ‘magnetic wire’ is normally used as conductor.
Transformer ratio a =
E1
N
I
= 1 = 2 ; (Ideal transformer)
E2 N 2 I1
Primary induced emf
E1 =
2 p f N1fmax = 4.44 f N1fmax
Correspondingly, E2 = E1/a
When a current I2 is drawn from secondary, the current I1 drawn from primary comprises three
compound
Im = magnetizing current to establish core flux. It lags by 90°
Ii = core (iron) loss current in phase with E1
Ê N2 ˆ
1
I2 = I2
I¢2 = current to counter secondary AT = Á
˜
a
Ë N1 ¯
Then
I1 = ( I m + I i ) + I 2¢ = I 0 + I 2¢
I 0 = I m + I i = no-load current, secondary open circuits.
Voltage drop in a transformer is due to primary and secondary resistance and leakage reactance (a
series effect).
Impedance is transferred from one side to the other in direct square of turn-ratio. susceptance to
transfer in inverse square of turn-ratio. Thus
Secondary impedance as seen on primary side
Z¢2 = a2Z2, a =
N1
N2
Transformers
Z ¢1 =
147
1
Z1
a2
The transformer equivalent circuit as seen from any side;
Shunt branch (Gi || Bm) draws Im and Ii
Series branch (R + jX) carries load current I¢2
Similarly,
R = R1 + a2R2, X = X1 + a2X2
The equivalent circuit parameters are determined by two non-loading tests:
Open-circuit test – rated voltage applied on one side the other left open
Determines: Gi , Bm and core loss, Pi
Short-circuit test – one side shorted, reduced voltage applied on other to carry full-load current
Determines: R, X and full-load copper loss, Pc
Transformer losses
Pi = core (iron) loss, constant at constant primary voltage
Pc = copper loss (I2R loss) proportional to square of load current
Transformer efficiency
h=
For h(max)
P0 ( = V2 I 2 cosq )
P0 + Pi + Pc
Pc = I 22 R2(eq) = Pi
or
Voltage registation,
I2(load) =
VR =
=
Pi
R2 (eq )
V20 - V2
¥ 100
V2
I ( R cos f ± X sin f )
¥ 100;
V2
+ for laging pf
– for leading pf
In an auto-transformer the primary and secondary windings are electrically connected so that a part
of the windings is common to the two. As a result, a part of the power is transferred conductively. It,
therefore has higher efficiency and kVA compared to the corresponding two-winding transformer.
In a 3-phase transformer, three single-phase transformers are connected in star/delta (various possible
connection). Depending on labeling and phase sequence, the line voltage undergoes a phase shift of
± 30 or ± 90°. The common practice is to use 30° phase shift.
148 Electric Machines
3.1 The emf per turn for a single-phase 2200/220V, 50-Hz transformer is approximately 12 V.
Calculate
(a) the number of primary and secondary
turns, and
(b) the net cross-sectional area of core of a
maximum flux density of 1.5 T.
3.2 A transformer has primary and secondary
turns of 1250 and 125 respectively. It has
core cross-section of 36 cm2 and its rms flux
density is to be limited to 1.4 T (to prevent
core saturation). What maximum 50 Hz
voltage can be applied on the primary side
and the corresponding open-circuit secondary
voltage?
The core has a mean length of 150 cm and
its relative permeability can be assumed to be
8000. What would be the rms exciting current
when the transformer’s primary winding is
excited at a voltage as calculated above? Also
calculate the magnetizing susceptance as seen
from primary and secondary sides.
If the transformer were to be excited at 60
Hz, what should be the maximum primary
voltage for the core flux density limit not to
be exceeded? What would be the magnetizing
susceptance as seen on each side in this case?
3.3 A single-phase transformer is rated 600/200 V,
25 kVA, 50 Hz. The transformer is supplying
full load on secondary side at 0.707 pf lagging.
What is the load impedance? Assuming the
transformer to be ideal what impedance is
seen on the primary side; also the primary
current and its pf.
3.4 A single-phase 50 Hz transformer has a
voltage ratio of 230/2300 V. A capacitor rated
30 kVAR is connected on the 2300 V side.
Calculate the value of this capacitor. What is
the kVAR of the capacitor and the value of its
capacitance as seen on 230-V side. Assume
the transformer to be ideal.
3.5 A transformer has 200 primary and 400
secondary turns. The primary draws a current
of (44.68 + j 2.37) A when the secondary
supplies a load current of (21.62 + j0) A. (a)
Find the exciting current. (b) If the core has a
permeance of 7.69 ¥ 10–5 H/T2, find the peak
value of core flux. (c) Find the primary and
secondary induced emfs if the frequency is
50 Hz. (d) Find the core loss.
Remark: This is a learning exercise. This is
not how the exciting current is measured in a
transformer because this needs differentiating
of two nearly equal quantities which introduces
large measurement error. As elaborated in this
chapter the exciting current is measured by a
no-load test.
3.6 A 23 kVA, 50 Hz, 2300/230 V transformer
has primary and secondary turns of 200/20.
When rated voltage is applied, calculate
the mutual core flux neglecting the winding
voltage drops. At full load the leakage flux
linking each winding is 1% of the mutual flux.
Calculate the primary and secondary leakage
reactances and the total reactance as referred
to either side.
Hint: fl1 is caused by I1N1 and by I1N2, while
the mutual flux is caused by (I1N1 – I2N2).
Refer Fig. 3.11.
3.7 A 100 kVA, 1100/230 V, 50-Hz transformer
has an HV winding resistance of 0.1 W and a
leakage reactance of 0.4 W. The LV winding
has a resistance of 0.006 W and a leakage
reactance of 0.01 W. Find the equivalent
winding resistance, reactance and impedance
referred to the HV and LV sides. Convert
these to pu values.
3.8 A 50 kVA, 2200/110 V transformer when
tested gave the following results:
OC test, measurements on the LV side: 400 W,
10 A, 110 V
SC test, measurements on the HV side; 808 W,
20.5 A, 90 V
Transformers
3.9
3.10
3.11
3.12
Compute all the parameters of the equivalent
circuit referred to the HV and LV sides of
the transformer. Also calculate % voltage
regulation and efficiency at full load and
0.8 pf lagging.
A 22/127 kV, 125 MVA transformer has
primary and secondary impedances of 0.015
+ j 0.06 pu each. Its magnetizing reactance is
j 120 pu. The pu values are expressed on the
base of the transformer rating. Calculate the
primary and secondary impedances in ohms
and also the magnetizing reactance in ohms
on the LV side.
A 20 kVA, 2000/200 V, 50 Hz transformer is
operated at no-load on rated voltage, the input
being 150 W at 0.12 power factor. When it is
operating at rated load, the voltage drops in the
total leakage reactance and the total resistance
are, respectively, 2 and 1 per cent of the rated
voltage. Determine the input power and power
factor when the transformer delivers 10 kW
at 200 V at 0.8 pf lagging to a load on the LV
side.
A single-phase load is fed through a 66-kV
feeder whose impedance is 120 + j 400 W and
a 66/6.6 kV transformer whose equivalent
impedance (referred to LV) is 0.4 + j 1.5 W.
The load is 250 kW at 0.8 leading power
factor and 6 kV.
(a) Compute the voltage at the sending-end of
the feeder.
(b) Compute the voltage at the primary
terminals of the transformer.
(c) Compute the complex power input at the
sending-end of the feeder.
An audio-frequency ideal transformer is
employed to couple a 60-W resistance load
to an electric source which is represented by
a constant voltage of 6 V in series with an
internal resistance of 2400 W.
(a) Determine the turn-ratio required to ensure
maximum power transfer by matching
149
the load and source impedances (i.e. by
making the 60 W secondary impedance to
2400 W when referred to the primary).
(b) Find the load current, voltage and power
under the conditions of maximum power
transfer.
3.13 Draw a clear phasor diagram of a transformer
operating at rated values. Refer to Fig. 3.14(a)
and assume N1/N2 = 1.5 and I1R1 = 0.15 E1,
I2R2 = 0.15 V2, I1X1 = 0.3 E1, I2 X2 = 0.25 V2,
Ii = 0.1 I 2¢ , Im = 0.25 I 2¢
Consider the load power factor to be (a)
0.8 lagging (b) 0.8 leading. Use V2 as the
reference phasor.
3.14 An ideal transformer has a primary winding
of 200 turns. On the secondary side the
number of turns between A and B is 600 and
between B and C is 400 turns, that between A
and C being 1000. The transformer supplies
a resistor connected between A and C which
draws 10 kW. Further, a load of 2000 –45° W
is connected between A and B. The primary
voltage is 2 kV. Find the primary current.
3.15 A 5-kVA, 400/80-V transformer has Req (HV)
= 0.25 W and Xeq (HV) = 5 W and a lagging
load is being supplied by it resulting in the
following meter readings (meters are placed
on the HV side).
I1 = 16 A, V1 = 400 V, P1 = 5 kW
For this condition calculate what a voltmeter
would read if connected across the load
terminals. Assume the exciting current to be
zero.
3.16 A 25-kVA, 230/115-V, 50-Hz transformer has
the following data
R1 = 0.12 W
X1 = 0.2 W
R2 = 0.04 W
X2 = 0.05 W
Find the transformer loading which will make
the primary induced emf equal in magnitude
to the primary terminal voltage when the
transformer is carrying the full load current.
Neglect the magnetizing current.
150
Electric Machines
3.17 The resistances and leakage reactances of
a 10 kVA, 50 Hz, 2200/220 V distribution
transformer are as follows:
R1 = 4 W
X1 = 5 W
R2 = 0.04 W
X2 = 0.05 W
Each quantity is referred to its own side of the
transformer. (Suffix ‘1’ stands for HV and ‘2’
for LV).
(a) Find the total leakage impedance referred
to (i) HV side (ii) LV side.
(b) Consider the transformer to give its rated
kVA at 0.8 pf lagging to a load at rated
voltage. Find the HV terminal voltage and
% voltage regulation.
(c) Repeat (b) for a pf of 0.8 leading.
(d) Consider the core-loss to be 80 W. Find
the efficiency under the conditions of part
(b). Will it be different for the conditions
under part (c)?
(e) If the load in part (b) gets short-circuited,
find the steady-state current in the HV
lines, assuming that the voltage applied to
the transformer remains unchanged.
3.18 For Problem 3.10, assume that the load power
factor is varied while the load current and
secondary terminal voltage are held fixed.
With the help of a phasor diagram, find the
load power factor for which the voltage
regulation is zero.
3.19 A 20 kVA, 2000/200 V, single-phase
transformer has the following parameters:
HV winding: R1 = 3 W
L V winding: R2 = 0.05 W
X1 = 5.3 W
X2 = 0.05 W
(a) Find the voltage regulation at (i) 0.8 pf
lagging (ii) upf (iii) 0.707 pf leading.
(b) Calculate the secondary terminal voltage
at (i) 0.8 pf lagging (ii) upf (iii) 0.707 pf
leading when delivering full-load current
with the primary voltage held fixed at
2 kV.
3.20 The approximate equivalent circuit of a
4 kVA, 200/400 V single-phase transformer,
referred to the LV side, is shown in Fig. P3.20.
(a) An open-circuit test is conducted by
applying 200 V to the LV side, keeping
the HV side open. Calculate the power
input, power factor and current drawn by
the transformer.
(b) A short-circuit test is conducted by passing
full-load current from the HV side keeping
the LV side shorted. Calculate the voltage
required to be applied to the transformer
and the power input and power factor.
0.15 W
0.4 W
+
+
V1
800 W
400 W
–
V¢2
–
Fig. P 3.20
3.21 A 20 kV A, 2000/200 V transformer has name
plate leakage impedance of 8%. What voltage
must be applied on the HV side to circulate
full-load current with the LV shorted?
3.22 Derive the condition for zero voltage
regulation. Also show that the magnitude of
maximum voltage regulation equals the pu
value of equivalent leakage impedance.
3.23 The following test results were obtained for
a 20 kVA, 50 Hz, 2400/240 V distribution
transformer:
Open-circuit test (LV): 240 V, 1.066 A, 126.6 W
Short-circuit test (HV): 57.5 V, 8.34 A, 284 W
(a) When the transformer is operated as a stepdown transformer with the output voltage
equal to 240 V, supplying a load at unity
power factor, determine the maximum
efficiency and the unity power factor load
at which it occurs.
Transformers
3.28 A 20 kVA, 200/500 V, 50 Hz, single-phase
transformer is connected as an auto-transformer as shown in Fig. P3.28. Determine its
voltage-ratio and the kVA rating. Mark on the
diagram, the magnitudes and relative directions of the currents in the winding as well as
in the input and output lines when delivering
the rated kVA to load.
(a) Calculate the efficiency on unity powerfactor at (i) full-load (ii) half-load.
(b) Determine the load for maximum
efficiency and the iron-and the copper-loss
in this case.
3.25 The efficiency of a 1000 kVA, 110/220 V,
50 Hz, single-phase transformer is 98.5% at
half full-load at 0.8 pf leading and 98.8% at
full-load upf.
Determine: (a) iron-loss, (b) full-load copperloss, and (c) maximum efficiency at upf.
3.26 Open and short-circuit tests performed on a
500 kVA, 6600/2300 V, 50 Hz transformer
yielded the following data:
No-load loss = 3 kW
Full-load short circuit loss = 4 kW
(a) Calculate the load (kVA) at which the
transformer efficiency would be maximum
for a given power factor. Calculate this
efficiency for a pf of 0.85.
(b) The transformer supplies the following
load cycle.
12 hours, full load 0.8 pf.
12 hours, half full load 0.9 pf.
Calculate the energy efficiency of the
transformer.
3.27 A transformer has its maximum efficiency of
0.98 at 20-kVA at unity power factor. During
the day it is loaded as follows:
12 hours; 2 kW at power factor 0.6
6 hours; 10 kW at power factor 0.8
6 hours; 20 kW at power factor 0.9
Find the ‘all-day’ efficiency of the transformer.
500 V
Load
Input
200V
(b) Determine the power-factor of the rated
load, supplied at 240 V, such that the
terminal voltage observed on reducing the
load to zero is still 240 V.
3.24 In a 25 kVA, 2000/200 V transformer, the
iron and copper losses are 300 and 400 W
respectively.
151
Fig. P3.28
3.29 A 400/100 V, 10 kVA, 2-winding transformer
is to be employed as an autotransformer to
supply a 400 V circuit from a 500 V source.
When tested as a 2-winding transformer at
rated load, 0.85 pf lagging, its efficiency is
0.97%.
(a) Determine its kVA rating as an autotransformer.
(b) Find its efficiency as an autotransformer.
3.30 A 20 kVA, 2000/200 V, two-winding transformer is to be used as an autotransformer,
with a constant source voltage of 2000 V.
At full-load of unity power factor, calculate
the power output, power transformed and
power conducted. If the efficiency of the twowinding transformer at 0.7 pf is 97%, find the
efficiency of the autotransformer.
3.31 A 200/400 V, 20 kVA, and 50 Hz transformer is
connected as an autotransformer to transform
600 V to 200 V.
(a) Determine the autotransformer ratio a¢.
(b) Determine the kVA rating of the autotransformer.
152
Electric Machines
(c) With a load of 20 kVA, 0.8 pf lagging
connected to 200 V terminals, determine
the currents in the load and the two
transformer windings.
3.32 An audio frequency output transformer
couples a variable frequency source of output
resistance 4.5 kW to a load of 10 W. The
transformer has a turn ratio of 25.4. On test
the following inductance data are measured
on the transformer.
(i) Inductance seen on the primary side with
secondary open = 18.7 H.
(ii) Inductance seen on the primary side with
secondary shorted = 0.215 H.
In terms of the frequency response calculate
(a) lower corner frequency (b) upper corner
frequency and (c) voltage gain and phase
angle at the geometric mean of the frequencies
in parts (a) and (b)
Hint: It is sufficiently accurate to assume that
test
(i) yields Lm, the magnetizing inductance and
test
(ii) yields the leakage inductance seen on
primary side. Transformer winding resistance is ignored.
3.33 A 20 kVA, 4400/220 V transformer with an
equivalent impedance of 0.01 W is to operate
in parallel with a 15 kVA, 4400/220 V
transformer with an equivalent impedance of
0.015 W. The two transformers are connected
in parallel and made to carry a load of 25 kVA.
Assume both the impedances to have the same
angle.
(a) Find the individual load currents.
(b) What per cent of the rated capacity is used
in each transformer?
3.34 Two single-phase transformers, rated
1000 kVA and 500 kVA respectively, are
connected in parallel on both HV and LV sides.
They have equal voltage ratings of 11 kV/400
V and their per unit impedances are (0.02 + j
0.07), and (0.025 + j 0.0875) W respectively.
What is the largest value of the unity power
factor load that can be delivered by the parallel
combination at the rated voltage?
3.35 Two
single-phase
transformers
rated
600 kVA and 500 kVA respectively, are
connected in parallel to supply a load of
1000 kVA at 0.8 lagging power factor.
The resistance and reactance of the first
transformer are 3% and 6.5% respectively,
and of the second transformer 1.5% and 8%
respectively. Calculate the kVA loading and
the power factor at which each transformer
operates.
3.36 An ideal 3-phase step-down transformer,
connected delta/star delivers power to a
balanced 3-phase load of 120 kVA at 0.8
power factor. The input line voltage is 11 kV
and the turn-ratio of the transformer, phase-tophase is 10. Determine the line voltages, line
currents, phase voltages and phase currents on
both the primary and the secondary sides.
3.37 A D/Y connected bank of three identical
60 kVA 2000/100 V, 50 Hz transformers
is fed with power through a feeder whose
impedance is 0.75 + j 0.25 W per phase. The
voltage at the sending-end of the feeder is held
fixed at 2 kV line-to-line. The shortcircuit test
when conducted on one of the transformers
with its LV terminals shortcircuited gave the
following results:
VHV = 40 V
IHV = 35 A
f = 50 Hz
P = 800 W
(a) Find the secondary line-to-line voltage
when the bank delivers rated current to a
balanced 3-phase upf load.
(b) Calculate the currents in the transformer
primary and secondary windings and in the
feeder wires on the occurrence of a solid
3-phase short-circuit at the secondary line
terminals.
3.38 Each phase of 3-phase transformer is rated
6.6 kV/230V, 200 kVA with a series reactance
of 8%.
Transformers
(a) Calculate the reactance in ohm referred to
HV/LV sides.
(b) The transformer is connected Y/Y. What is
its 3-phase rating (voltage and kVA) and
the per unit reactance.
(c) Calculate the pf of load (rated) at which
voltage regulation would be maximum. If
this load is fed at rated voltage on LV side,
what should be the HV side line voltage?
3.39 A 2400/220 V, 300 kVA, 3-phase transformer
has a core loss of 33 kW at rated voltage. Its
equivalent resistance is 1.6%. Calculate the
transformer efficiency at 1.8 pf at (i) full load
(ii) at half load.
What is the load at which the transformer
efficiency would be maximum? Calculate its
value at a pf of 0.8.
Hint: Use the pu method.
3.40 A 3-phase 50 kVA, 6.6/0.4 kV 50 Hz
transformer is D/Y connected. It yielded the
following test results:
OC Test
SC Test
P0 = 520 W
PSC = 610 W
I0 = 4.21 A
ISC = 4.35 A
V0 = 400 V
VSC = 340 V
Calculate the pu circuit parameters of the
transformer. Determine its efficiency and
voltage regulation at full load 0.8 pf lagging.
Calculate also the maximum efficiency and
the load (0.8 pf ) at which it will occur.
3.41 A 6.6/0.4 kV, 100 kVA distribution transformer
is connected D/Y. The transformer has 1.2%
resistance and 5% reactance. Find the voltage
regulation at full load, 0.8 pf leading. With
0.4 kV as secondary voltage (on load), what is
the primary voltage?
Hint: Use pu system.
3.42 A single-phase, 50 Hz, three-Winding transformer is rated at 2200 V on the HV side with
a total of 250 turns. Of the two secondary
windings, each can handle 200 kVA, one is
rated at 550 V and the other at 220 V. Compute
153
the primary current when the rated current
in the 220 V winding is at upf and the rated
current in the 550 V winding is 0.6 pf lagging.
Neglect all leakage impedance drops and
magnetizing current.
3.43 A small industrial unit draws an average load
of 100 A at 0.8 lagging pf from the secondaries
of its 2000/200 V, 60 kVA Y/D transformer
bank. Find:
(a) The power consumed by the unit in kW,
(b) the total kVA used,
(c) the rated line currents available from the
transformer bank,
(d) the rated transformer phase currents of the
D-secondaries,
(e) per cent of rated load on transformers,
(f ) primary line and phase currents, and
(g) the kVA rating of each individual
transformer.
3.44 The HV terminals of a 3-phase bank of three
single-phase transformers are connected to a
3-wire, 3-phase, 11 kV (line-to-line) system.
The LV terminals are connected to a 3-wire,
3-phase load rated of 1000 kVA and 2200 V
line-to-line. Specify the voltage, current and
kVA ratings of each transformer (both HV and
LV windings) for the following connections:
(a) HV – Y, LV – D
(b) HV – D, LV – Y
(c) HV – Y, LV – Y
(d) HV – D, LV – D.
3.45 A 3-phase bank consisting of three singlephase 3-winding transformers (Y/D/Y) is
employed to step-down the voltage of a
3-phase, 220 kV transmission line. The data
pertaining to one of the transformers are given
below:
Ratings
Primary 1: 20 MVA, 220 kV
Secondary 2: 10 MVA, 33 kV
Tertiary 3: 10 MVA, 11 kV
Short-circuit reactances on 10 MV A base
X12 = 0.15 pu
X23 = 0.1 pu
154 Electric Machines
X13 = 0.2 pu
3.46
3.47
3.48
3.49
Resistances are to be ignored. The D-connected
secondaries supply their rated current to a
balanced load at 0.85 power factor lagging,
whereas the tertiaries provide the rated current
to a balanced load at upf (constant resistance).
(a) Compute the primary line-to-line voltage
to maintain the rated voltage at the
secondary terminals.
(b) For the conditions of part (a) find the lineto-line voltage at the tertiary terminals.
(c) If the primary voltage is held fixed as in
part (a), to what value will the tertiary
voltage increase when the secondary load
is removed?
A 500-kVA, 11/0.43-kV, 3-phase delta/star
connected transformer has on rated load HV
copper-loss of 2.5 kW and LV loss of 2 kW.
The total leakage reactance is 0.06 pu. Find
the ohmic values of the equivalent resistance
and leakage reactance on the delta side.
Two transformers each rated 250-kVA,
11/2-kV and 50-Hz are connected in opendelta on both the primary and secondary.
(a) Find the load kVA that can be supplied
from this transformer connection.
(b) A delta connected three-phase load of
250 kVA, 0.8 pf, 2 kV is connected to
the low-voltage terminals of this openvoltage transformer. Determine the
transformer currents on the 11 kV side of
this connection.
Two 110-V, single-phase furnaces take loads
of 500 kW and 800 kW respectively at a
power factor of 0.71 lagging and are supplied
from 6600 V, 3-phase mains through a Scottconnected transformer combination. Calculate
the currents in the 3-phase lines, neglecting
transformer losses. Draw the phasor diagram.
Figure P3.49 shows a Scott-connected
transformer, supplied from 11 kV, 3-phase,
50 Hz mains. Secondaries, series connected as
shown, supply 1000 A at a voltage of 100 2
to a resistive load. The phase sequence of the
3-phase supply is ABC.
(a) Calculate the turn-ratio of the teaser
transformer.
(b) Calculate the line current IB and its phase
angle with respect to the voltage of phase
A to neutral on the 3-phase side.
Teaser
1000 A
A
11 kv, 3-phase
supply
Resistive load
100 2
Volts
B
C
M
Main
Fig. P3.49
3.50 A 15 kVA, 2200/220 V, 50 Hz transformer
gave the following test results:
OC (LV side) V = 220 V
I = 2.72 A
P = 185 W
SC (HV side) V = 112 V
I = 6.3 A
P = 197 W
Compute the following:
(a) Core loss
(b) Full-load copper loss
(c) Efficiency at full-load 0.85 lagging of
(d) Voltage regulation at full-load 0.8 lagging/
leading pf
3.51 A transformer of rating 20 kVA, 2000/200 V
has the following parameters:
Req(HV side) = 2.65 W
Zeq(HV side) = 4.23 W
Core loss at rated voltage = 95 W
(a) Calculate transformer efficiency when
delivering 20 kVA at 200 V at 0.8 pf
lagging.
Transformers
(b) What voltage must be applied on the HV
side for load as in part (a).
(c) Find the percentage voltage regulation.
3.52 A 100 kVA, 11 kV/231 V transformer has HV
and LV winding resistances of 8.51 W and
0.0038 W respectively. It gave the following
test results:
OC (LV side)
SC (HV side)
231 V
440 V
15.2 A
9A
1.25 kW
Not
measured
3.56
3.57
Calculate
(a) Equivalent leakage reactance of the
transformer
(b) Full load copper loss
(c) Efficiency at full-load and half full-load at
0.85 lagging power factor
3.53 A 100 kVA, 2200 V/220 V transformer has the
following circuit parameters.
R1 = 0.23 W
R2 = 0.0023 W
X1 = 1.83 W
X2 = 0.013 W
R1 (HV side) = 5.6 kW
Xm(HV side) = 1.12 kW
The transformer is subjected to the following
daily load cycle = 4 h on no load, 8 h on 1/4th
full-load at 0.8 pf, 8 h on 1/2 full-load at upf,
and 4 h on full-load at 0.9 pf.
Determine the all-day energy efficiency of the
transformer.
3.54 A 400/200 V, 50 Hz transformer has a primary
impedance of 1.2 + j 3.2 W and secondary
impedance of 0.4 + j 1.0 W. A short-circuit
occurs on the secondary side with 400 V
applied to the primary. Calculate the primary
current and its power factor.
3.55 A 50 Hz, 3-winding transformer can be
considered as an ideal transformer. The
primary is rated 2400 V and has 300 turns.
The secondary winding is rated 240 V,
400 kVA and supplies full-load at upf. The
3.58
3.59
155
tertiary is rated 600 V, 200 kVA and supplies
full-load at 0.6 pf lagging. Determine the
primary current.
An ideal transformer has 200 primary turns
and 360 secondary turns, the primary being
excited at 600 V. The full secondary has a
resistive load of 8 kW. The secondary is also
tapped at 240 turns which supplies a pure
inductive load of 10 kVA. Find the primary
current and its pf.
A 50 kVA, 2300 V/230 V transformer draws
power of 750 W at 0.5 A at no load when
2300 V is applied to the HV side. The HV
winding resistance and leakage reactance are
1.8 W and 4 W respectively. Calculate:
(a) the no load pf
(b) the primary induced emf
(c) the magnetizing current and
(d) the core loss component of current.
Two single-phase transformers operate in
parallel to supply a load of 44 + j 18.6 W. The
transformer A has a secondary emf of 600 V
on open circuit with an internal impedance
of 1.8 + j 5.6 W referred to the secondary.
The corresponding figures for transformer
B are 610 V and 1.8 + j 7.4 W. Calculate the
terminal voltage, current and power factor of
each transformer.
Each phase of a 3-phase transformer is rated
6.6 kV/230 V, 200 kVA with a series reactance
of 8%
(a) Calculate the reactance in ohm referred to
HV/LV sides.
(b) The transformer is connected Y/Y. What is
its 3-phase rating (voltage and kVA) and
the per unit reactance.
(c) Calculate the pf of load (rated) at which
voltage regulation would be maximum. If
this load is fed at rated voltage on LV side,
what should be the HV side line voltage?
156
Electric Machines
1. What is a transformer? Explain the functions
it fulfils as an element of a power system.
2. Differentiate between core and shell-type
transformers.
3. Explain briefly the ideal transformer as a
circuit element. Can voltage and current ratios
be adjusted independently?
4. Explain the operation and application of the
impedance transforming property of an ideal
transformer.
5. State how the LV and HV windings are
arranged in a core-type transformer. Advance
the reason why?
6. What is the phase relationship between the
core flux; the magnetizing current and the
induced emfs in the primary and secondary
winding of a transformer? Draw the phasor
diagram.
7. What determines the maximum value of
flux in a transformer core when it is excited
from the primary side? Does the value of flux
change substantially when the secondary is
loaded? Explain the reason why.
8. Why cannot the SC test separate out the
primary and secondary resistances and
leakage inductances?
9. Justify that under SC test that the core loss is
negligible.
10. Prove that in the system, if the voltage bases
are selected in the ratio of transformation, the
pu impedance of the transformer is same on
either side.
11. State and prove the condition from maximum
efficiency of a transformer.
12. Draw the phasor diagram of a transformer
13.
14.
15.
16.
17.
18.
19.
20.
21.
22.
23.
24.
25.
26.
as seen from any one side for zero voltage
regulation.
From the phasor diagram of Question 12,
derive the approximate condition for zero
voltage regulation.
Justify the statement that in the circuit model
of a transformer in a power system, the
magnetizing branch can be ignored.
Explain the meaning of all the items in the
nameplate of a transformer.
How can we refer the transformer winding
resistance and leakage reactance from one
side to the other?
From the percentage impedance given as the
nameplate, find the voltage to be applied for
full load current to flow in SC test.
Why are transformers needed in a power
system?
Why are transformers placed in oil-filled tanks?
Describe how the primary current adjusts
itself as the load on a transformer is increased.
Explain why the core flux in a transformer is
almost independent of load current.
Why is an OC test generally performed at
rated voltage on LV side of a transformer?
Why is the SC test performed at reduced
voltage on the HV side?
Where is an autotransformer employed in a
power system? Why?
In a transmission system the star side of a
star/delta transformer is HV side, while in a
distribution system the star side is the LV side.
Explain.
Explain the basic purpose of a tertiary
winding. To what additional use can it be put?
Transformers
3.1 The core in a large power transformer is built
of
(a) cast iron
(b) mild steel
(c) ferrite
(d) silicon steel
3.2 Cruciform shape is used in transformer core
(a) to reduce core loss
(b) to reduce winding copper
(c) to provide mechanical strength
(d) to reduce core reluctance
3.3 No load current in a transformer
(a) lags the applied voltage by 90°
(b) lags the applied voltage by somewhat less
than 90°
(c) leads the applied voltage by 90°
(d) leads the applied voltage by somewhat
less than 90°
3.4 A 200/100 V, 50 Hz transformer is to be
excited at 40 Hz from the 100 V side. For
the exciting current to remain the same, the
applied voltage should be
(a) 150 V
(b) 125 V
(c) 100 V
(d) 80 V
3.5 Power input to a transformer on no load at
rated voltage comprises predominantly
(a) copper loss
(b) hysteresis loss
(c) core loss
(d) eddy current loss
157
3.6 A 2/1 ratio, two-winding transformer is
connected as an auto transformer. Its kVA
rating as an auto transformer compared to a
two-winding transformer is
(a) same
(b) 1.5 times
(c) 2 times
(d) 3 times
3.7 The high frequency hum in the transformer is
mainly due to
(a) laminations being not sufficiently tight
(b) magnetostriction
(c) oil of the transformer
(d) tank walls
3.8 The efficiency of a transformer at full-load
0.85 pf lag is 95%. Its efficiency at full-load
0.85 pf lead will be
(a) less than 95%
(b) more than 95%
(c) 95%
(d) 100%
3.9 Under balanced load conditions, the main
transformer rating in the Scott connection is
greater than that of the teaser transformer by
(a) 5%
(b) 15%
(c) 57.7%
(d) 85%
3.10 Non-loading heat run test on transformer is
performed by means of
(a) SC test
(b) OC test
(c) half time on SC and half time on OC
(d) Sumpner’s test
158 Electric Machines
4
4.1
INTRODUCTION
The chief advantage of electric energy over other forms of energy is the relative
ease and high efficiency with which it can be transmitted over long distances. Its
main use is in the form of a transmitting link for transporting other forms of energy,
e.g. mechanical, sound, light, etc. from one physical location to another. Electric
energy is seldom available naturally and is rarely directly utilized. Obviously two
kinds of energy conversion devices are needed—to convert one form of energy
to the electric form and to convert it back to the original or any other desired form. Our interests in this
chapter are the devices for electromechanical energy conversion. These devices can be transducers for
low-energy conversion processing and transporting. These devices can be transducers for processing and
transporting low-energy signals. A second category of such devices is meant for production of force or
torque with limited mechanical motion like electromagnets, relays, actuators, etc. A third category is the
continuous energy conversion devices like motors or generators which are used for bulk energy conversion
and utilization.
Electromechanical energy conversion takes place via the medium of a magnetic or electric field—the
magnetic field being most suited for practical conversion devices. Because of the inertia associated with
mechanically moving members, the fields must necessarily be slowly varying, i.e. quasistatic in nature. The
conversion process is basically a reversible one though practical devices may be designed and constructed
to particularly suit one mode of conversion or the other.
This chapter is mainly devoted to the understanding of the principle of electromechanical energy
conversion. Simple examples will be used for illustrative purposes. In later chapters the analysis of
continuous energy conversion equipment will be carried out.
4.2
ENERGY IN MAGNETIC SYSTEM
Energy can be stored or retrieved from a magnetic system by means of an exciting coil connected to an
electric source. Consider, for example the magnetic system of an attracted armature relay of Fig. 4.1. The
resistance of the coil is shown by a series lumping outside the coil which then is regarded as an ideal loss-less
Principles of Electromechanical Energy Conversion
159
coil. The coil current causes magnetic flux to be established in the magnetic circuit. It is assumed that all the
flux f is confined* to the iron core and therefore links all the N turns creating the coil flux linkages of
l = Nf
(4.1)
Core
The flux linkage causes a reaction emf of
dl
(4.2)
dt
to appear at the coil terminals with polarity (as per
Lenz’s law) shown in the Fig. 4.1. The associated
circuit equation is
e=
R
+
Flux f
x
i
+
v
Armature
e
–
N
–
Hinge
v = iR + e
dl
Fig. 4.1 Attracted armature relay
(4.3)
dt
The electric energy input into the ideal coil due to the flow of current i in time dt is
= iR +
dWe = ei dt
(4.4)
Assuming for the time being that the armature is held fixed at position x, all the input energy is stored in
the magnetic field. Thus
dWe = ei dt = dWf
(4.5)
where dWf is the change in field energy in time dt. When the expression for e in Eq. (4.2) is substituted in
Eq. (4.5), we have
dWe = idl = F df = dWf
(4.6)
where F = Ni, the magnetomotive force (mmf ).
The relationship i-l or F-l is a functional one corresponding to the magnetic circuit which in general is
nonlinear (and is also history-dependent, i.e. it exhibits hysteresis). The energy absorbed by the field for finite
change in flux linkages for flux is obtained from Eq. (4.6) as
DWf =
Ú
l2
l1
i(l) dl =
Ú
f2
f1
F(f) df
(4.7)
As the flux in the magnetic circuit undergoes a cycle f1Æf2Æf1, an irrecoverable loss in energy takes
place due to hysteresis and eddy-currents in the iron, assuming here that these losses are separated out and
are supplied directly by the electric source. This assumption renders the ideal coil and the magnetic circuit
as a conservative system with energy interchange between themselves so that the net energy is conserved.
The energy absorbed by the magnetic system to establish flux f (or flux linkages l) from initial zero flux is
Wf =
Ú
l
0
i(l) dl =
Ú
f
0
F(f) df
(4.8)
This then is the energy of the magnetic field with given mechanical configuration when its state corresponds
to flux f (or flux linkages l).
* The leakage flux (which is of course small) does not take part in the energy conversion process. It can be accounted for by placing an imaginary coil in series with the ideal coil which produces exactly the flux linkages
corresponding to the leakage flux. As in the case of transformers, the inductance of such a coil is referred to as the
leakage inductance. Here the leakage inductance is assumed to be negligible.
160
Electric Machines
The i-l relationship is indeed the magnetization curve
which varies with the configuration variable x (Fig. 4.1: the
air-gap between the armature and core varies with position
x of the armature. The total reluctance of the magnetic path
decreases as x increases). The i-l relationship for various
values of x is indicated in Fig. 4.2. It immediately follows
that this relationship can be expressed as
l
x1
x2
x3
x1 > x 2 > x 3
i = i(l, x)
i
0
if l is the independent variable or as
Fig. 4.2
l = l(i, x)
i -l relationship with variable x
if i is the independent variable.
Therefore, the field energy (Eq. (4.8)) is in general a function of two variables,
Wf = Wf (l, x)
Wf = Wf (i, x)
i.e.
or
(4.9a)
(4.9b)
According to Eqs (4.9a) and (4.9b) field energy is determined by the instantaneous values of the system
states ((l, x) or (i, x) and is independent of the path followed by these states to reach the present values. This
means that the field energy at any instant is history independent.
A change in l with fixed x causes electric-magnetic energy interchange governed by the circuit Eq. (4.3)
and the energy Eq. (4.6). Similarly, if x is allowed to change with fixed l, energy will interchange between
the magnetic circuit and the mechanical system. The general case of such energy interchanges (electricmagnetic-mechanical) is the subject matter of Sec. 4.3.
As per Eq. (4.8) the field energy is the area between the l-axis and i-l. curve as shown in Fig. 4.3. A new
term, co-energy is now defined as
W f¢ (i, x) = il – Wf (l, x)
(4.10)
wherein by expressing l as l(i, x), the independent variables of W f¢ become i and x. The coenergy on Fig. 4.3
is shown to be the complementary area of the i-l curve. It easily follows from Fig. 4.3 that
W f¢ =
l-axis
i
Ú l di
(4.11)
0
Wf = field energy
l
i-l curve for fixed x
W'f = coenergy
0
i
Fig. 4.3
Field energy and coenergy
i-axis
Principles of Electromechanical Energy Conversion 161
Linear Case
Electromechanical energy conversion devices are built with air-gaps in the magnetic circuit which serve to
separate the stationary and moving members. As a result the i-l relationship of the magnetic circuit is almost
linear; also the losses of magnetic origin are separately accounted for by semi-empirical methods. With the
linearity assumption the analysis is greatly simplified. Losses and certain nonlinear effects may then be
incorporated at a later-stage.
Assuming linearity, it follows from Eq. (4.8) or Fig. 4.3 that
1
1
1
il = Ff = Rf 2
2
2
2
where, as it is known, R = F/f = reluctance of the magnetic circuit. Since the coil inductance
Wf =
(4.12)
L = l/i
the field energy can be expressed as
1 l2
(4.13)
2 L
In the linear case the inductance L is independent of i but is a function of configuration x. Thus the field
energy is a special function of two independent variables l and x, i.e.
Wf =
1 l2
(4.14)
2 L( x)
The field energy is distributed throughout the space occupied by the field. Assuming no losses and constant
permeability, the energy density* of the field is
Wf (l, x) =
wf =
Ú
B
0
HdB =
1 B2
1
HB =
2 m
2
J/m3
(4.15)
H = magnetic field intensity (AT/m)
B = magnetic flux density (T)
The energy density expression of Eq. (4.15) is important from the point of view of design wherein the
capability of the material is to be fully utilized in arriving at the gross dimensions of the device.
For the linear case it easily follows from Eq. (4.11) that coenergy is numerically equal to energy, i.e.
where
1
1
1
li = Ff = PF 2
2
2
2
where P = f/F = permeance of the magnetic circuit.
Also in terms of the coil inductance
W f¢ = Wf =
W f¢ =
or in general
W f¢ (i, x) =
Ú
i
0
(l = Li)di =
1 2
Li
2
1
L(x)i2
2
(4.17)
* If A(m2) and l(m) are the area and length dimensions of the field, then from Eq. (4.8)
wf =
Wf
Al
=
Ú
l
0
(4.16)
iN Ê l ˆ
=
d
l ÁË NA ˜¯
Ú
B
0
H dB
162
Electric Machines
The expression for coenergy density is
w¢f =
Ú
H
B dH
(4.18a)
1 B2
1
mH 2 =
2 m
2
(4. 18b)
0
which for the linear case becomes
w f¢ =
4.3
FIELD ENERGY AND MECHANICAL FORCE
Consider once again the attracted armature relay excited by an electric source as in Fig. 4.4. The field produces
a mechanical force Ff in the direction indicated which drives the mechanical system (which may be composed
of passive and active mechanical elements). The mechanical work done by the field when the armature moves
a distance dx in positive direction is
dWm = Ff dx
( 4.19)
x
Ff
R
Mechanical
system
+
+
Electric
source –
i
v
e
N
Armature
–
Fig. 4.4
Production of mechanical force
This energy is drawn from the field by virtue of change dx in field configuration. As per the principle of
energy conservation
Mechanical energy output = electrical energy input – increase in field energy
(4.20)
or in symbolic form
Ff dx = idl – dWf
(4.21)
It may be seen that Ff dx is the gross mechanical output, a part of which will be lost in mechanical friction.
From Eq. (4.10)
Then
Wf = il – W f¢ (i, x)
dWf = d(il) – dWf (i, x)
∂W f¢ ˆ
Ê ∂W f¢
di +
dx˜
= idl + ldi – Á
∂x
Ë ∂i
¯
Substituting for dWf from Eq. (4.22) in Eq. (4.21), we have
È
∂W f¢ ˆ ˘
Ê ∂W f¢
Ff dx = idl – Íid l + l di - Á
di +
dx˜ ˙
∂x
Ë ∂i
¯ ˙˚
ÍÎ
(4.22)
Principles of Electromechanical Energy Conversion
or
∂W f¢
È ∂W f¢
˘
- l ˙ di +
Ff dx = Í
dx
∂x
Î ∂i
˚
163
(4.23)
Because the incremental changes di and dx-are independent and di is not present in the left-hand side of
Eq. (4.23), its coefficient on the right-hand side must be zero i.e.
∂W f¢
∂i
–l=0
∂W f¢
l=–
(4.24)
∂x
It then follows from Eq. (4.23) that
Ff =
∂W f¢ (i, x)
(4.25)
∂x
This expression for mechanical force developed applies when i is an independent variable, i.e. it is a
current excited system
If (l, x) are taken as independent variables,
Wf = Wf (l, x)
dWf =
∂W f
∂l
dl +
∂W f
∂x
dx
(4.26)
Substituting Eq. (4.26) in Eq. (4.21)
Ff dx = idl –
Ff dx = –
or
∂W f
∂l
dl -
∂W f
∂x
dx
∂W f
Ê ∂W f ˆ
dx + Á i dl
∂x
∂l ˜¯
Ë
(4.27)
Since dl, the independent differential, is not present on the left hand side of this equation,
ior
Hence
∂W f
∂l
=0
i=
∂W f ( l , x )
Ff = -
∂l
∂W f ( l , x )
∂x
(4.28)
(4.29)
In this form of expression for the mechanical force of field origin, l is the independent variable, i.e. it is a
voltage-controlled system as voltage is the derivative of l.
In linear systems where inductances are specified it is more convenient to use coenergy for finding the
force developed (Eq. (4.25)). If the system is voltage-controlled, the current can be determined by writing the
necessary circuit equations (see Examples 4.11 and 4.12).
It is needless to say that the expressions of Eqs (4.25) and (4.29) for force in a translatory system will
apply for torque in a rotational system with x replaced by angular rotation q.
164
Electric Machines
Direction of Mechanical Force Developed
With reference to Eq. (4.29) it immediately follows that Ff is positive (i.e. it acts in the positive reference
direction of x) if ∂Wf (l, x)/∂x is negative which means that stored energy of the field is reduced with increase
of x while flux linkages l are held fixed. In the particular case of Fig. 4.4 as x increases (i.e., the armature
moves towards left), the field energy for fixed l is reduced because the air-gap is reduced. It means that Ff in
this case acts in the positive direction. It is therefore, concluded that the mechanical force produced by the
field acts in a direction to reduce field energy or in other words the system seeks a position of minimum field
energy. Similarly, it can be concluded from Eq. (4.25) that the system seeks a position of maximum coenergy.
Also in Fig. 4.4, the force acts in a direction to increase x thereby reducing the magnetic circuit reluctance
and increasing the coil inductance.
Determination of Mechanical Force
Nonlinear case It was seen above that the mechanical force is given by the partial derivatives of coenergy
or energy as per Eqs (4.25) and (4.29). In the general nonlinear case, the derivative must be determined
numerically or graphically by assuming a small increment Dx. Thus
Ff ª
DW f¢
Dx
Ff ª –
or
(4.30a)
i = cost
DW f
Dx
(4.30b)
l = cost
These two expressions will give slightly different numerical values of Ff because of finite Dx. Obviously
Ff is the same in each case as Dx Æ 0. Calculation of Ff by Eq. (4.30a) is illustrated in Ex. 4.2.
Linear case
From Eq. (4.17)
1
L(x)i 2
2
∂W f¢
1 ∂L ( x )
\
Ff =
= i2
(4.31)
∂x
2
∂x
From Eq. (4.31), it is obvious that the force acts in a direction to increase the inductance of the exciting
coil, a statement already made.
Alternatively from Eq. (4.14)
W f¢ (i, x) =
Wf (l, x) =
\
1 l2
2 L( x)
Ff = -
∂W f
∂x
2
=
1 Ê l ˆ ∂L ( x )
2 ÁË L( x) ˜¯ ∂x
(4.32)
It may be seen that Eqs (4.31) and (4.32) are equivalent as i = l/L.
Also from Eq. (4.12)
1
R(x)f 2
2
∂W f
1 ∂R( x)
Ff = = - f2
∂x
2
∂x
Wf (f, x) =
\
(4.33)
Principles of Electromechanical Energy Conversion 165
It must be remembered here that there is no difference between l as independent variable or f as
independent variable as these are related by a constant (l = Nf). It follows from Eq. (4.33) that the force acts
in a direction to reduce reluctance of the magnetic system, a statement that has been made already.
Another expression for Ff can be derived as below:
From Eq. (4.12)
Wf (l, x) =
1
li(x)
2
Ff = -
\
∂W f
= -
∂x
1 l∂i ( x)
2 ∂x
(4.34)
Mechanical Energy
When the armature in Fig. 4.4 is allowed to move from position xa to xb with the coil current remaining
constant at io, the mechanical energy output is
DWm =
Ú
DWm =
Ú
xb
xa
F f dx
(4.35)
Integrating Eq. (4.25)
xb
xa
F f dx = DWf¢ (i remaining constant)
= increase in coenergy
(4.36)
The graphical representation of Eq. (4.36) is given in Fig. 4.5(a) for the general nonlinear case while
Fig. 4.5(b) gives the linear case. In each case the electrical energy input is
DWe = i0(l2 – l1)
(4.37)
For the linear case, it follows from the geometry of Fig. 4.5(b) that
1
1
i0(l2 – l1) = DWe
(4.38)
2
2
which means that half the electrical energy input gets stored in the field and the other half is output as
mechanical energy. In this kind of operation the armature must move from position xa to xb infinitely slowly
for the excitation coil current to remain constant.
DWf = DW f¢ = DWm =
l
l
DWe
DWe
l2
b
l2
b
l1
a
l1
a
DWm = DW'f
0
i0
(a) Nonlinear case
i
DWm = DW'f
0
i0
(b) Linear case
Fig. 4.5
i
166
Electric Machines
Let now the armature in Fig. 4.4 be allowed to move from xa and xb with coil flux linkage l remaining
constant. Integrating Eq. (4.29),
DWm =
Ú
xb
xa
Ff dx = –DWf (l remaining constant)
= decrease in field energy
(4.39)
This is illustrated in Fig. 4.6(a) for the general nonlinear case and in Fig. 4.6(b) for the linear case. In each
case
DWe = 0
(4.40)
For the linear case
1
l 0 (i1 – i2)
(4.41)
2
For l to remain constant, the armature must move from xa to xb in zero time. Since there is no electrical
input, the mechanical energy output is drawn from the field energy which reduces by an equal amount.
DWm =
l
l
DWm = – DWf
b
l0
0
a
b
l0
i2
i1
(a) Nonlinear case
i
0
DWm = – DWf
a
i2
i1
i
(b) Linear case
Fig. 4.6
The actual armature movement lies between the two ideal cases illustrated above. The corresponding i-l
relationship is a general path from a to b as shown in Figs. 4.7(a) and (b). In this general case
DWe = area cabd
DWf = area obd – area oac
l
l2
l1
l
d
b
l2
DWm
c
a
e
i2
f
i1
l1
i
d
b
DWm
c
a
e
i2
(b) Linear case
(a) Nonlinear case
Fig. 4.7
f
i1
i
Principles of Electromechanical Energy Conversion 167
Now
or
DWm = DWe – DWf
= area cabd – area obd + area oac
= area oab
Mechanical energy output = shaded area in Fig. 4.7
Since ab is a general movement, this area which represents the mechanical energy output has to be
computed graphically or numerically.
Flow of Energy in Electromechanical Devices
Electromechanical energy conversion is a reversible process and Eqs (4.25) and (4.29) govern the production
of mechanical force. In Fig. 4.4 if the armature is allowed to move on positive x direction under the influence
of Ff, electrical energy is converted to mechanical form via the coupling field. If instead the armature is moved
in the negative x direction under the influence of external force, mechanical energy is converted to electrical
form via the coupling field. This conversion process is not restricted to translatory devices as illustrated but is
equally applicable to rotatory devices (see Ex. 4.4). Electrical and mechanical losses cause irreversible flow
of energy out of a practical conversion device. The flow of energy in electromechanical conversion in either
direction along with irrecoverable energy losses is shown in Figs. 4.8(a) and 4.8(b).
Electrical losses
(ohmic and iron-loss)
Mechanical losses
Net electrical input
ei¢dt
Gross mechanical output
Tw dt
Fvdt or
Electrical
source
Mechanical
sink
Coupling
field
Gross electrical
input
Total conversion
process
vidt
Net mechanical
output
F¢vdt or
(a)
Electrical losses
T ¢vidt
Mechanical losses
Gross electrical output eidt
Net mechanical input
F¢vdt or
Electrical
sink
Mechanical
source
Coupling
field
Net electrical
output vi¢dt
Fig. 4.8
T ¢wdt
Ideal conversion
process
(b)
Gross mechanical
input
Fvdt or
Tw dt
168 Electric Machines
EXAMPLE 4.1 Figure 4.9 shows the cross-sectional view of a cylindrical iron-clad solenoid magnet. The
plunger made of iron is restricted by stops to move through a limited range. The exciting coil has 1200 turns
and carries a steady current of 2.25 A. The magnetizing curve of the iron portion of the magnetic circuit is
given below:
Flux, Wb
MMF, AT
0.0010
60
0.00175
120
0.0023
210
0.0025
300
0.0026
390
0.00265
510
Calculate the magnetic field energy and coenergy for air-gap of g = 0.2 cm and g = 1 cm with exciting
current of 2.25 A in each case.
Solenoid
g
Plunger
0.02 cm 1.2 cm
5 cm
dia
Fig. 4.9
SOLUTION The magnetic circuit has two air-gaps. The reluctance of each of these are calculated as follows:
Case 1: g = 0.2 cm
Reluctance of the circular air-gap =
Reluctance of the annular air-gap =
0.2 ¥ 10- 2
= 810.5 ¥ 103
-7 p
2
4 p ¥ 10 ¥ ¥ (0.05)
4
0.02 ¥ 10- 2
4 p ¥ 10- 7 ¥ p ¥ 0.05 ¥ 0.012
= 84.4 ¥ 103
Total air-gap reluctance Rag = 895 ¥ 103
Fag = Rag f = 895 ¥ 103f AT
The combined magnetization curve of iron and air-gaps for g = 0.2 cm is calculated below:
l (WbT)
AT
i(A)
1.2
955
0.796
2.1
1686
1.405
2.76
2269
1.891
3.0
2538
2.115
The i-l curve is plotted in Fig. 4.10 from which the field coenergy found graphically is
\
Area oea = 3.73 J
Energy area oaf = 3.11 ¥ 2.25 – 3.73 = 3.27 J
3.12
2717
2.264
3.18
2882
2.40
Principles of Electromechanical Energy Conversion 169
Case 2: g = 1 cm
Reluctance of circular air-gap = 4052.5 ¥ 103
Reluctance of annular air-gap = 84.4 ¥ 103
Rag = 4136.7 ¥ 103
Because of such high reluctance of air-gap, ampere-turns absorbed by iron part of the magnetic circuit can be
neglected, e.g. for
f = 0.0025 Wb
AT (air-gap) = 10342
AT (iron) = 390
Thus it is seen that even in the near saturation region, AT (iron) is less than 5% of the total AT required for establishing
the flux.
i = 2.25 A
F = 1200 ¥ 2.25 = 2700 AT
2700
f=
= 0.653 ¥ 10–3 Wb
4136.7 ¥ 103
For
l = Nf = 1200 ¥ 0.653 ¥ 10–3 = 0.784 Wb-T
1
Field energy = coenergy = il
2
=
1
¥ 2.25 ¥ 0.784 = 0.882 J
2
EXAMPLE 4.2 In Ex. 4.1 calculate the force on the plunger for g = 0.2 cm with an exciting current of
2.25 A.
SOLUTION
Ff =
∂W f¢ (i, g )
∂g
where g is the air-gap. Given g = 0.2 ¥ 10 m.
Case a: Reluctance of the iron path accounted for.
This is the nonlinear case where the derivative desired can be found numerically.
Assume
–2
Dg = –0.05 ¥ 10–2 m (a decrease)
g + Dg = 0.15 ¥ 10–2 m
= 810.5 ¥ 103 ¥
Air-gap reluctance
0.15
+ 84.4 ¥ 103
0.2
= 692 ¥ 103
For this air-gap
l
AT
i
1.2
752
0.626
2.1
1331
1.110
2.76
1802
1.501
3.0
2030
1.692
3.12
2189
1.824
3.18
2344
1.953
170
Electric Machines
This is plotted in Fig. 4.10 from which increase in co energy for i = 2.25 A is the area oab, i.e.
DW f¢ = 0.718 J
Ff =
∂W f¢
∂g
=
0.718
- 0.05 ¥ 10
-2
Since a decrease in g causes an increase in Wf, the force on plunger
acts in negative direction (positive direction is in the increasing
direction of g). Hence the force is attractive (tending to reduce g).
It may be noted that better results will be obtained by choosing a
smaller value of Dg.
Case b: The magnetization curve of iron is assumed linear
(corresponding to the initial slope).
This is the linear case so that we can proceed analytically.
DW'f
g = 0.2 cm
1.5
1.0
0.5
0
0
= 144.4 (1 + 28.1 ¥ 10 g) ¥ 10
1200i
f=
Wb
144.4(1 + 28.1 ¥ 102 g ) ¥ 103
Ff =
g = 0.15 cm
2.0
3
l = Nf =
a
2.5
= 144.4 ¥ 103 + 4053g ¥ 105
2
d
f
3.0
60
g
Total reluctance R =
+ 84.4 ¥ 103 +
p
0.001
4 p ¥ 10- 7 ¥ (0.05) 2
4
W f¢ (i, g) =
b
= –1436 N
l(WbT)
\
l
3.5
0.5
1.0
1.5
i (A)
2.0 e
2.5
Fig. 4.10
(1200) 2 i
144.4(1 + 28.1 ¥ 102 g ) ¥ 103
WbT
(1200) 2 i 2
1
il =
2
2 ¥ 144.4(1 + 28.1 ¥ 102 g ) ¥ 103
∂W f¢
∂g
=–
(1200) 2 i 2 ¥ 28.1 ¥ 102
2 ¥ 144.4(1 + 28.1 ¥ 102 g ) 2 ¥ 103
For g = 0.2 ¥ 10–2 m
Ff = –
(1200) 2 ¥ (2.25) 2 ¥ 28.1 ¥ 102
2 ¥ 144.4(1 + 28.1 ¥ 0.2) 2 ¥ 103
= –1619 N
The linearity assumption causes the values of force to differ by about 13% from that obtained by the nonlinear method.
The actual difference will be still less as in the nonlinear case the derivative is obtained by approximation (Dg = 0.05 ¥
102 m). It may be noted here that the linearity assumption renders great computational saving and is hence commonly
employed.
EXAMPLE 4.3 In Ex. 4.1, assume the reluctance of the iron path to be negligible. The exciting current
is 2.25 A. The plunger is now allowed to move very slowly from g = 1 cm to g = 0.2 cm. Find the electrical
energy input to the exciting coil and the mechanical output.
SOLUTION
As already calculated in Ex. 4.2, the reluctance of the magnetic path as a function of the air-gap length is
R = (84.4 + 4053 ¥ 102 g) ¥ 10–3
Principles of Electromechanical Energy Conversion
171
Flux linkages for an exciting current of 2.25 A are
=
(1200) 2 ¥ 2.25
2
2
3
(84.4 + 4053 ¥ 10 g ) ¥ 10
=
38.4
(1 + 48 ¥ 102 g )
WbT
Since the plunger moves very slowly, the exciting current remains constant at 2.25 A. Hence
DWe = i0 (l2 – l1)
Ê
ˆ
1
1
= 2.25 ¥ 38.4 Á
Ë 1 + 48 ¥ 0.2 1 + 48 ¥ 1˜¯
= 2.25 ¥ 38.4 (0.094 – 0.020) = 6.39 J
1
DWm = DWe = 3.195 J
2
EXAMPLE 4.4 The magnetic flux density on the surface of an iron face is 1.6 T which is a typical
saturation level value for ferromagnetic material. Find the force density on the iron face.
SOLUTION Let the area of the iron face be A(m)2. Consider the field energy in the volume contained between the two
faces with a normal distance x. From Eq. (4.15)
Wf (B, x) =
1 B 2 Ax
2 m
From Eq. (4.29), the mechanical force due to the field is
∂W f ( B , x )
Ff = –
∂x
=–
1 B2 A
2 m
The negative sign indicates that the force acts in a direction to reduce x (i.e. it is an attractive force between the two
faces). The force per unit area is
|Ff | =
=
1 B2
2 m
1
(1.6) 2
¥
= 1.02 ¥ 106 N/m2
2 4 p ¥ 10- 7
EXAMPLE 4.5 In the electromagnetic relay of Fig. 4.11 excited from a voltage source, the current and
flux linkages are related as
i = l2 + 2l(1 – x)2 ; x < 1
Find the force on the armature as a function of l.
SOLUTION
Wf (l, x) =
Ú
l
0
idl
1 3
l + l2 (1 – x)2
3
∂W f
Ff = = 2l2(l – x)2
∂x
=
172
Electric Machines
EXAMPLE 4.6
The electromagnetic relay of Fig. 4.11 is excited from a voltage source
v = 2 V sin wt
Assuming the reluctance of the iron path of the magnetic circuit to be constant, find the expression for the
average force on the armature, when the armature is held fixed at distance x.
Armature
i
+
v
~
N
–
A
x
Fig. 4.11
SOLUTION
Reluctance of the iron path = a (say)
2x
Reluctance of the air path =
= bx
m0 A
Total reluctance of the magnetic path, R = a + bx
1
(Note that l = Nf)
Wf (f, x) = R(x)f2
2
∂W f (f , x)
1 2 ∂R
1
= - bf 2
Ff = = - f
2
∂
x
2
∂x
(4.42)
Notice that ∂R /∂x = b is positive, so that Ff is negative, i.e. it acts in a direction to reduce x (which means in a direction
to reduce reluctance R ).
Now i and v are related by the circuit equation
di
v =R+L
dt
whose steady-state solution is
V
I =
2
2 2
R +w L
– - tan -1
wL
R
(4.43)
Then
2V
wLˆ
Ê
sin Á w t - tan -1
Ë
r ˜¯
R +w L
i=
2
2 2
(4.44)
From Eq. (2.21) L = N2/R
Then
f=
Ni
=
R
Ê
wN2 ˆ
sin Á w t - tan -1
˜
RR ¯
Ë
( RR )2 + ( N 2w ) 2
2 NV
Substituting f in Eq. (4.42)
Ff = –
bN 2V 2
2 ˆ
Ê
2
-1 N w
t
sin
w
tan
Á
˜
RR ¯
( RR )2 + ( N 2w ) 2
Ë
(4.45)
Principles of Electromechanical Energy Conversion 173
Time-average force is then
Ff (av) =
1
T
Ú
T
0
Ff dt; T =
2p
w
bN 2V 2
1
= 2 ( RR )2 + ( N 2w ) 2
(4.46)
EXAMPLE 4.7 Figure 4.12 shows a rotational electromechanical device called the reluctance motor. It
is required to determine the torque acting on the rotor as a function of current input to the exciting coil and
the angle (q) of rotor and stator overlap.
Obviously the torque expression from coenergy, i.e. W¢f (i, f) must be developed. It is assumed that the
cast steel magnetic path has negligible reluctance so that the reluctance encountered in the magnetic path
is that due to the two annular air-gaps.
SOLUTION
1 ˆ
Ê
Air-gap area normal to flux, A = Á r + g ˜ q l
Ë
2 ¯
Total air-gap length along the flux path = 2g
Reluctance of the air-gaps =
2g
1 ˆ
Ê
m0 Á r + g ˜ q l
Ë
2 ¯
(4.47)
1 ˆ
Ê
Ni m0 Á r + g ˜ q l
Ë
2 ¯
Flux established, f =
2g
(4.48)
Flux linkage, l = fN
1 ˆ
Ê
N 2im0 Á r + g ˜ q l
Ë
2 ¯
=
2g
Stator
(4.49)
i
1
Coenergy, W¢f (i, q) = li
2
1 ˆ
Ê
N i m0 Á r + g ˜ q l
Ë
2 ¯
=
4g
∂W f¢ (i, q )
Torque developed, Tf =
∂q
q
r
2 2
1 ˆ
Ê
N 2i 2 m0 Á r + g ˜ l
Ë
2 ¯
=
4g
Axial length
(normal to paper) = l
T
Rotor
N
(4.50)
g
Cast steel
(4.51)
Fig. 4.12
Elementary reluctance machine
This torque acts in a direction to increase coenergy for a given coil current i. This happens (see Eq. (4.50)) when q
increases, i.e. the rotor tends to align itself with the stator. Also observe that the torque in this case is independent of
angle q.
Consider the problem from the design point of view. The design question is posed as to the maximum torque that can
be developed, when the magnetic material is stressed to its saturation value (1.6 T). The torque expression of Eq. (4.51)
174 Electric Machines
above must therefore be expressed in terms of the flux density rather than the exciting coil current.
B = f/A = F/R A
m0 Ni
=
2g
(4.52)
Using Eq. (4.52) in Eq. (4.51),
1 ˆ
Ê
B 2 gl Á r + g ˜
Ë
2 ¯
Tf =
m0
(4.53)
Since the maximum value of B is fixed from consideration of the magnetic material, any desired torque can be
achieved by a suitable combination of g, r and l. The considerations in relative adjustment of these three dimensions of
the device are beyond the scope of this book.
Let
Then
g = 0.0025 m
l = r = 0.025 m
B = 1.6 T
Tf =
1
Ê
ˆ
(1.6) 2 ¥ 0.0025 ¥ 0.025 Á 0.025 + ¥ 0.0025˜
Ë
¯
2
4 p ¥ 10- 7
= 3.34 Nm
In the nonoverlapping region of the rotor and stator, the field geometry is very complex and an analytical expression
for torque is not possible. It is interesting to examine this problem from the point of view of the coil inductance as a
function of the rotor’s angular position. From Eq. (4.49)
L(q) = l/i
1 ˆ
Ê
N 2 m 0l Á r + g ˜ q
Ë
2 ¯
=
2g
(4.54)
With constant air-gap length, in the overlapping region, the inductance increases linearly with q acquiring a maximum
value when rotor is in the vertical position in Fig. 4.12 and then decreases linearly. From Eq. (4.54), the inductance is
zero at q = 90°, i.e. no overlap between rotor and stator pole
faces. However, it is known that the coil inductance is not zero Main pole
for q = 90° which means that the inductance model of Eq. (4.54)
is not valid in the region of low q. In fact, the inductance has
a least value in the horizontal position of the rotor and rises to
a maximum value when the rotor goes to the vertical position
travelling from either direction. In a practical device the region
between the two pole faces of the stator is narrow (Fig. 4.13) and
Quadrature
axis
the rotor and stator pole faces are so shaped that the reluctance
w¢
of the magnetic circuit and therefore the coil inductance varies
almost sinusoidally. The coil inductance has a maximum value
when the rotor is aligned along the main pole axis (called the
q
direct axis) and a minimum value when the rotor is at 90° to
d
Position
of rotor at
Rotor axis
the main pole axis (called the quadrature axis) as shown in
t=0
Fig. 4.13. It is convenient to choose the direct axis as the
Direct axis
reference for angle q.
Fig. 4.13
Principles of Electromechanical Energy Conversion
175
It is readily seen from Fig. 4.13 that the coil inductance is a double frequency function of q (there are two cycles of
L-variation in one complete rotation of the rotor). Therefore, L(q) can be written as:
L(q) = L1 + L2 cos 2q
(4.55)
This variation of inductance is depicted in Fig. 4.14.
Lq (rotor aligned to quadrature axis)
L(q)
Ld (rotor aligned to direct axis)
L2 =
1
(L – Lq)
2 d
L1
0
p/2
p
3p/2
2p
q
Fig. 4.14 Variation of the coil inductance with rotor position in a reluctance machine
Assume the excitation current to be sinusoidal,
i = Im cos wt
(4.56)
Field coenergy is (Eq. (4.17))
W f¢ (i, q) =
1
L(q)i2
2
The mechanical torque due to field is then
Tf =
∂W f¢
∂q
=
1 2 ∂L(q )
i
∂q
2
= –I 2m L2 sin 2q cos2 wt
(4.57)
In terms of the angular speed of the rotor (w¢)
q = w¢t – d
(4.58)
where 0 is the position of rotor at t = 0 when current i is maximum. Then
Tf = – Im2 L2 sin 2 (w¢t – d) sin wt
1 2
Im L2 sin 2 (w¢t – d) (1 + cos 2 wt)
2
1
= – Im2 L2{sin 2 (w¢t – d) + sin 2 (w¢t – d) cos 2 wt}
2
1
1
Ï
¸
= - I m2 L2 Ìsin 2 (w ¢t - d ) + [sin 2 (w ¢t + w t - d ) + sin 2 (w ¢t - w t - d )]˝
2
2
Ó
˛
=–
(4.59)
176
Electric Machines
It is observed from Eq. (4.59) that the torque is time-varying with the average value zero if of w¢ π w. However, when
the rotor runs at speed w¢ = w, called the synchronous speed, the average torque is
Tf (av) =
1 2
I m L2 sin 2d
4
(4.60)
From Fig. 4.14
1
(Ld – Lq)
2
1
Tf (av) = I 2m (Ld – Lq) sin 2d
8
L2 =
\
(4.61)
Thus, for example, with Im = 5 A, Ld = 0.25 H, Lq = 0.15 H, the maximum value of the average torque is
1
¥ 25 ¥ (0.25 – 0.15) = 0.3125 Nm
8
d = 45°
Tf (av)| max =
when
Sinusoidal torque – d variation is typical of synchronous machines.
Singly-excited devices discussed earlier, are generally employed for motion through a limited distance or
rotation through a prescribed angle. Electro-mechanical transducers have the special requirement of producing
an electrical signal proportional to forces or velocities or producing force proportional to electrical signal
(current or voltage). Such transducers require two excitations—one excitation establishes a magnetic field
of specified strength while the other excitation produces the desired signal (electrical or mechanical). Also
continuous energy conversion devices—motors and generators—require multiple excitation. One continuous
energy conversion device has already been studied in Ex. 4.6 which is singly-excited (reluctance motor).
Figure 4.15 shows a magnetic field system with two electrical excitations—one on stator and the other
on rotor. The system can be described in either of the two sets of three independent variables; (l1, l2, q) or
(i1, i2, q). In terms of the first set
Tf = -
∂W f (l1, l2 , q )
(4.62)
∂q
Stator
q
i1
Tf
+
–
2
l1
v1
1
v2
l2
+
–
Rotor
Fig. 4.15
i2
Principles of Electromechanical Energy Conversion 177
where the field energy is given by
Wf (l1, l2, q) =
Ú
l1
0
i1d l1 +
Ú
l2
0
i2 d l2
(4.63)
Analogous to Eq. (4.28)
i1 =
i2 =
∂W f (l1 , l2 , q )
∂l1
∂W f (l1 , l2 , q )
∂l 2
l1 = L11i1 + L12i2
l2 = L21i1 + L22i2; (L12 = L21)
Assuming linearity
(4.64a)
(4.64b)
Solving for i1 and i2 in terms of l1, l2 and substituting in Eq. (4.63) gives upon integration*
1
1
b11l 21 + b12l1l2 + b22l 22
2
2
b11 = L22/(L11L22 – L 212)
Wf (l1, l2, q) =
where
(4.65)
2
b22 = L11/(L11L22 – L12
)
b12 = b21 = –L12/(L11L22 – L 212)
The self- and mutual-inductance of the two exciting coils are functions of angle q.
If currents are used to describe the system state
Tf =
∂W f¢ (i1 , i2 , q )
(4.66)
∂q
where the coenergy is given by
W f¢ (i1, i2, q) =
Ú
i1
l1 di1 +
0
Ú
i2
0
l2 di2
(4.67)
In the linear case
1
1
L11i 21 + L12i1i2 + L22 i 22
2
2
where inductances are functions of angle q.
W f¢ (i1, i2, q) =
*
i1 = b11l11 + b12l2
i2 = b21l1 + b22l2; b21 = b12
Wf (l1, l2, q) =
Ú
l1
0
= b11
= b11
=
(b11l1 + b12 l2 ) d l1 +
Ú
l1
Ú
l1
0
0
È
l1d l1 + b12 Í
ÍÎ
l1d l1 + b12
Ú
l1
0
Ú
0
(b12 l1 + b 22 l2 ) d l2
l2 d l1 +
l1 , l2
0
Ú
l2
Ú
l2
0
˘
l1d l2 ˙ + b 22
˙˚
d (l1l2 ) + b 22
1
1
b11l12 + b12 l1l2 + b 22 l22
2
2
Ú
l2
0
l2 d l2
Ú
l2
0
l2d l2
178 Electric Machines
EXAMPLE 4.8
For the system of Fig. 4.15, various inductances are:
L11 = (4 + cos 2q) ¥ 10–3 H
L12 = 0.15 cos q H
L22 = (20 + 5 cos 2q) H
Find the torque developed if i1 = 1 A, i2 = 0.02 A.
SOLUTION
1
1
(4 + cos 2q) ¥ 10–3 ¥ i12 + (0.15 cos q)i1i2 + (20 + 5 cos 2q)i 22
2
2
∂W f¢
= (sin 2q) ¥ 10–3 i21 – 0.15 (sin q)i1i2 – 5(sin 2q)i 22
Tf =
∂q
W¢f (i1, i2, q) =
= –10–3 sin 2q – 3 ¥ 10–3 sin q
The first term – 10–3 sin 2q is the reluctance torque which arises if the self-inductances are functions of space angle
q. If L1 and L2 are independent of q (the rotor and stator are round with uniform air-gap, known as the round rotor
construction) the reluctance torque becomes zero. The second term is the torque produced by the mutual component. It is
also seen that the reluctance torque is a double frequency of the space angle as compared to the second term. The negative
sign indicates that the torque is restoring in nature, i.e. it opposes the displacement q.
EXAMPLE 4.9
In the electromagnetic relay shown in Fig. 4.16
L11 = k1/x, L22 = k2/x, L12 = k3/x
Find the expression for the force on the armature, if
i1 = I1 sin w1t, i2 = I2 sin w 2t
write an expression for the average force. For what relationship between w1 and w2, the average force is (i)
maximum (ii) minimum.
SOLUTION
W f¢ (i1, i2, x) =
Ff =
1 k1 2 k2
1 k3 2
i1i2 +
i1 +
i2
2 x
x
2 x
∂W f¢
∂x
= -
1 k1 2 k2
1 k3 2
i –
i
ii –
2 x2 1 x2 1 2 2 x2 2
Substituting for i1, i2
Ff = -
1 k1 2 2
k
1 k3 2 2
I 1 sin w1 t – 22 I1I2 sin w1t sin w2t –
I 2 sin w 2 t
2 x2
2 x2
x
Ff = -
1 k12 2 1 k12
1 k2
I +
cos 2w1 t –
I1I2 cos(w1 – w2)t
4 x2 1 4 x2
2 x2
1 k1
1 k2 2 1 k3 2
I1I2 cos (w1 + w2)t –
I2 –
I 2 cos 2w2 t
2 x2
4 x2
4 x2
Since these are mixed frequency terms
+
Ff (av) = lim
T
1
T
Ú
T
0
l1
L11
i2
L12
L22
x
F f (t ) dt
Fig. 4.16
Principles of Electromechanical Energy Conversion
179
If w1 π w2,
Ff (av) = -
1 k12 2 1 k2 2
I1 –
I 2 (minimum force)
4 x2
4 x2
Ff (av) = -
1 k12 2 1 k2
1 k2 2
I1 –
I1I2 –
I 2 (maximum force)
4 x2
2 x2
4 x2
If w1 = w2,
EXAMPLE 4.10
Two coupled coils have self- and mutual-inductance of
1
;
2x
L11 = 2 +
L22 = 1 +
1
;
2x
L12 = L21 =
1
2x
over a certain range of linear displacement x. The first coil is excited by a constant current of 20 A and the
second by a constant current of –10 A. Find:
(a) Mechanical work done if x changes from 0.5 to 1 m.
(b) Energy supplied by each electrical source in part (a).
(c) Change in field energy in part (a).
Hence verify that the energy supplied by the sources is equal to the increase in the field energy plus the
mechanical work done.
SOLUTION
Since it is the case of current excitations, the expression of coenergy will be used
W f¢ (i1, i2, x) =
1
1
L11i 21 + L12i1i2 + L22i 22
2
2
1ˆ
Ê
1
= Á 2 + ˜ ¥ 200 +
¥ (–200) +
Ë
2x ¯
2x
= 450 +
Ff =
(a)
DWm =
∂W f¢
∂x
Ú
1
0.5
1ˆ
Ê
ÁË1 + 2 x ˜¯ ¥ 50
25
x
= -
25
x2
F f dx =
Ú
1
0.5
-
25
x2
dx = –25 J
l1 ( x =1)
DWe1 =
(b)
Ú
i1 dl1 = i1[l1 (x = 1) – l1(x = 0.5)]
l1 ( x = 0.5)
l1 = L11i1 + L12i2
1ˆ
1
5
Ê
= Á 2 + ˜ ¥ 20 +
¥ (–10) = 40 +
Ë
2x ¯
2x
x
l1(x = 0.5) = 50, l1(x = 1) = 45
\
DWe1 = 20(45 – 50) = –100 J
Similarly
DWe2 = i2[l2(x = 1) – l2(x = 0.5)]
l2 = L12i1 + L22i2
180 Electric Machines
l2 =
1
¥ 20 +
2x
1ˆ
5
Ê
ÁË1 + 2 x ˜¯ ¥ (–10) = –10 + x
l2 (x = 0.5) = 0, l2 (x = 1) = –5
DWe2 = –10(–5) = 50 J
Net electrical energy input, DWe = DWe1 + DWe2
= –100 + 50 = –50 J
(c) For calculating the change in the field energy, b’s have to be obtained.
L22
2
; D = L11L22 – L12
b11 =
D
=
Similarly,
4x + 1
4x + 3
1
= 4x + 3
2
3
= , b22 = ,
5
5
b22 =
b12
At x = 0.5;
2x + 1
4x + 3
b11
3
,
7
5
,
7
b12 = -
1
5
1
7
The values of l have already been calculated at x = 0.5, 1 m.
As per Eq. (4.65), the field energy is given by
At x = 1;
b11 =
b22 =
b12 = -
1
b11l 21 + b12 l1l2 + b22l 22
2
The field energy at x = 0.5 m and x = 1 m is then calculated as
Wf =
1 2
¥ ¥ (50)2 = 500 J
2 5
1 5
1 3
1
Wf (x = 1) = ¥ ¥ (45)2 –
¥ 45 ¥ (–5) + ¥ ¥ (–5)2
2 7
2 7
7
= 475 J
Wf (x = 0.5) =
Hence
DWf = Wf (x = 1) – Wf (x = 0.5) = 475 – 500 = –25 J
DWf + DWm = –25 – 25 = –50 = DWe (verified)
Note: In the linear case with constant current excitation
DWf = DW f¢
DWf can be easily calculated from part (a) without the need of calculating b’s. Thus
25
x
DW f¢ = W f¢ (x = 1) – W f¢ (x = 0.5)
W f¢ = 450 +
= 475 – 500 = – 25 J
EXAMPLE 4.11 Two coupled coils have self- and mutual-inductances as in Ex. 4.10. Find the expression
for the time-average force of field origin at x = 0.5 m if:
Principles of Electromechanical Energy Conversion
(a)
(b)
(c)
(d)
181
both coils are connected in parallel across a voltage source of 100 cos 314t V,
both coils are connected in series across a voltage source of 100 cos 314t V,
coil 2 is shorted and coil 1 is connected to a voltage source of 100 cos 314t V, and
both coils are connected in series and carry a current of 0.5 cos 314t A.
SOLUTION Though cases (a), (b) and (c) pertain to voltage excitation, the coenergy approach works out to be more
convenient and will be used here.
W f¢ (i1, i2, x) =
=
Ff =
1
1
L11i12 + L12i1i2 + L22i 22
2
2
1Ê
1ˆ
1ˆ
1Ê 1 ˆ
Ê
2 + ˜ i12 + Á1 + ˜ i1i2 + Á ˜ i22
Ë
2 ÁË
2x ¯
2x ¯
2 Ë 2x ¯
∂W f¢ (i1, i2 , x)
∂x
= -
1
4x
i2
2 1
-
1
2x
ii
2 12
-
1
4 x22
i22
For x = 0.5 m
Ff = –i 21 – 2i1i2 – i 22
The force acts in a direction to decrease x.
(a) Both coils connected in parallel across the voltage source:
L11 = 2 +
L22 = 1 +
1
2x
x = 0.5
1
2x
x = 0.5
L12 = L21 =
=3
=2
1
2x
=1
x = 0.5
From Eqs (4.64a) and (4.64b)
v = e1 =
d l1
di di
= 3 1 + 2 = 100 cos 314t
dt
dt
dt
v = e2 =
d l2
di
di
= 1 + 2 2 = 100 cos 314t
dt
dt
dt
Solving we get
di1
= 20 cos 314t
dt
di2
= 40 cos 314t
dt
Integrating
i1 =
20
sin 314t
314
i2 =
40
sin 314t
314
182
Electric Machines
Substituting for i1 and i2 in the expression for Ff,
1
[(20)2 + 2 ¥ 20 ¥ 40 + (40)2] sin2 314t
Ff = (314) 2
2
Ê 60 ˆ
= -Á
sin2 314t
Ë 314 ˜¯
1
T
But
\
Ú
T
0
sin 2 wt dt =
1
2
2
1 Ê 60 ˆ
= 0.0183 N
Ff (av) = - Á
2 Ë 314 ¯˜
(b) Both coils connected in series across the voltage source:
v=
d l1 d l2
+
dt
dt
di ˆ
Ê di1 di2 ˆ Ê di1
+Á
+2 2˜
+
= Á3
˜
Ë
¯
Ë dt
dt
dt
dt ¯
But i1 = i2 = i (series connection)
\
v =7
di
= 100 cos 314t
dt
Integrating we get
i=
100
sin 314t
7 ¥ 314
Substituting in the expression for Ff,
2
Ê 100 ˆ
Ff = –4 ¥ Á
sin2 wt
Ë 7 ¥ 314 ˜¯
or
Ê 100 ˆ
Ff (av) = – 2 Á
Ë 7 ¥ 314 ˜¯
2
(c) Coil 2 shorted, coil 1 connected to voltage source:
100 cos 314t = 3
0=
di1
di
+2 2
dt
dt
di1
di
+2 2
dt
dt
Solving we have
di1
= 40 cos 314t
dt
di2
= 20 cos 314t
dt
Upon integration we get
40
sin 314t
314
20
sin 314t
i2 = 314
i1 =
= –0.00144 N
Principles of Electromechanical Energy Conversion
183
Substituting for i1 and i2 in the expression for Ff ,
1
Ff = [(40)2 – 2 ¥ 40 ¥ 20 + (20)2] sin2 314t
3142
2
1 Ê 20 ˆ
Ff (av) = - Á
= –0.0203 N
2 Ë 314 ˜¯
(d) Both coils in series carrying current:
i = 0.5 cos 314t
Substituting in the expression for Ff,
Ff = –(l + 2 + 1) ¥ (0.5)2 cos2 314t
Ff (av) = –0.5 N
EXAMPLE 4.12
A doubly-excited magnetic field system has coil self-and mutual-inductances of
L11 = L22 = 2
L12 = L21 = cos q
where q is the angle between the axes of the coils.
(a) The coils are connected in parallel to a voltage source v = Vm sin wt. Derive an expression for the
instantaneous torque as a function of the angular position q. Find therefrom the time-average torque.
Evaluate for q = 30°, v = 100 sin 314t.
(b) If coil 2 is shorted while coil 1 carries a current of i1 = Im sin wt, derive expressions for the
instantaneous and time-average torques. Compute the value of the time-average torque when q = 45°
and i1 = 2 sin 314t.
(c) In part (b) if the rotor is allowed to move, at what value of angle will it come to rest?
SOLUTION
Tf =
=
∂W f¢ (i1, i2 ,q )
∂q
1 Ê ∂L11 ˆ 2 Ê ∂L12 ˆ
1 Ê ∂L22 ˆ 2
i1 + Á
ii +
i
Ë ∂q ˜¯ 1 2 2 ÁË ∂q ˜¯ 2
2 ÁË ∂q ˜¯
Substituting the values of inductances,
Tf = –(sin q) i1i2
From circuit equations
di1
di
+ (cos q) 2
dt
dt
di1
di2
Vm cos wt = (cos q)
+2
dt
dt
Vm cos wt = 2
Solving these we get
Vm sin w t
di1
di
= 2 =
(2 + cos q )
dt
dt
Integrating
i 1 = i2 =
Vm sin w t
w (2 + cos q )
184 Electric Machines
Substituting in Tf ,
Tf = Tf (av) = -
Vm2 sin q
(2 + cos q ) 2 w 2
sin2 wt
Vm2 sin q
2(2 + cos q ) 2 w 2
q = 30°, v = 100 sin 314t
Given:
\
Tf (av) = –
(100) 2 sin 30∞
2 (2 + cos 30∞) 2 ¥ (314) 2
= –0.069 Nm
(b) From circuit equations
di1
di
+2 2
dt
dt
1
di1
= - (cos q)
2
dt
1
= - (cos q)i1
2
= Im sin wt
1
= - Im(cos q) sin wt
2
0 = (cos q)
or
di2
dt
or
i2
Given:
i1
\
i2
Substituting in Tf ,
Tf = –(sin q) ¥
1 2
I m (cos q) sin2 wt
2
1 2
I (sin q)(cos q) sin2 wt
2 m
1
Tf (av) = - I2m(sin 2q)
8
= -
Given:
\
q = 45°, Im =
2
1
Tf (av) =
¥ 2 sin 90° = 0.25 Nm
8
(c) The average torque is zero and changes sign at q = 0°, 90°, 180°. The rotor can come to rest at any of these values
of q but the position of stable equilibrium will only be q = 90°, 270°, … (The reader should draw Tf (av) versus q
and reason out).
4.5
FORCES/TORQUES IN SYSTEMS WITH PERMANENT MAGNETS
Method of finding forces in systems with permanent magnets is best illustrated by an example. Figure 4.17(b)
shows a moving armature relay excited by a permanent magnet (PM).
The dc magnetizing curve of the permanent magnet is drawn in Fig. 4.17(a) which upon linear extrapolation
at the lower B-end can be expressed as
Bm = mR (Hm – H c¢ ) = mR Hm + Br
mR = recoil permeability of the PM material
= Br /Hc¢; on linearized basis
(4.68)
Principles of Electromechanical Energy Conversion 185
Soft iron portion of the magnetic circuit including the armature is assumed to have m = . For finding the
force on the armature, it will be convenient to use Eq. (4.25) for which we will need the expression of system
coenergy Wf¢ (i, x) which must be independent of i as there is no exciting current in the system. Thus coenergy
will be a function of the space variable x only i.e.,
Wf¢ (i, x) fi Wf¢ (x)
Coenergy is given by the expression of Eq. (4.11), i.e.
Wf¢ =
i
Ú l di
0
This expression needs to be carefully interpreted for the case of permanent magnetic excited systems. The
limits of integration in this expression mean from state of zero flux to a state of certain flux (but with zero
current). The state of zero flux is imagined by means of a fictitious exciting coil (of Nf turns) carrying current
if as shown in Fig. 4.17(c). The current is assumed to be adjusted to value If causing the core flux to reduce to
zero and the original state is then reached by imaging the current (if) to reduce to zero. Thus
x
Bm
PM
Br
d
mR = Br/H¢C
m=
–H¢c,–H c
x
Hm Crossm=
sectional area A (uniform)
(a) B-H curve of permanent magnet (PM)
Armature
(b)
x
if
Nf
PM
d
m=
m=
x
(c)
Fig. 4.17
186 Electric Machines
Wf¢ =
Ú
0
If
l f dif
(4.69)
At any value of if, we can write
Nf if = Hm d + 2Hg x
(4.70)
Continuity of flux allows us to write
Bm A = Bg A
Bm = mR Hm = Bg = m0 Hg
or
(4.71)
Substituting for Hg in Eq. (4.70)
Nf if = Hmd + 2(x/m0)Bm
Substituting for Hm from Eq. (4.68) and solving we get
Bm =
m R ( N f i f - H c¢ d )
(4.72)
d + 2( m R / m0 ) x
Flux linkages of the fictitious coil are given by
lf = ABm Nf =
m R N f A( N f i f - H c¢ d )
d + 2( m R / m0 ) x
(4.73)
Flux and flux linkages would be zero at
if = (H c¢ d/Nf )
(4.74)
Substituting Eq. (4.69), we get
Wf¢ (x) =
=
0
m R N f A( N f i f - H c¢ d )
if
d + 2( m R / m0 ) x
Ú
dif
m R AH c¢ d 2
2 [d + 2( m R / m0 ) x]
(4.75)
The force on the armature is then given by
Ff =
=
dW f¢ ( x)
dx
( m R H c¢ ) 2 d 2 A
m0 [d + 2( m R / m0 ) x]2
mRH c¢ = Br
But
Ff =
Therefore,
ABr2
(4.76)
m0 [1 + 2( m R / m0 ) ( x / d ) 2 ]
Let us now calculate the magnitude of the force for typical numerical values as below,
Br = 0.96 T, H ¢c = 720 kA/m,
(i) x = 0 cm
Now
d = 2 cm,
A = 6 cm2
(ii) x = 0.5 cm
mR
0.96
107
B /H ¢
¥
= r c =
= 1.06
m0
m0
720 ¥ 103 4p
Principles of Electromechanical Energy Conversion
187
Substituting values in force equation (Eq. (4.76))
(i) x = 0
Ff = -
6 ¥ 10- 4 ¥ (0.96) 2
4 p ¥ 10- 7
= –440 N
(ii) x = 0.5 cm
Ff = -
6 ¥ 10- 4 ¥ (0.96) 2
4 p ¥ 10- 7 [1 + 2 ¥ 1.06 ¥ (0.5 / 2)]2
= –188 N
Note: Negative sign in force is indicative of the fact that the force acts in a direction to reduce x (air-gap).
Similar treatment could be used for mixed situation where system has both permanent magnets and
exciting coils. It should be stressed here that an alternative procedure is to use the finite element method to
evaluate the coenergy from the vector form of Eq. (4.18a) integrated over the volume, i.e.,
B0 B ◊ dHdV
(4.77)
W f¢ =
ÚÚ
V
0
To calculate force –dW f¢ /dx is obtained by numerical differentiation (see Example 4.2). This method is of
general applicability wherever the magnetic circuit analysis cannot be carried out.
4.6
ENERGY CONVERSION VIA ELECTRIC FIELD
Electromechanical energy conversion via the electric field is analogous to the magnetic field case studied
earlier. Charge in the electric field is analogous to flux linkages and voltage to current in the magnetic field case.
Electric Field Energy
Figure 4.18 shows a parallel plate condenser with a fixed and a movable plate. The condenser is fed from
a current source. The leakage current of the condenser is represented outside by conductance so that the
condenser’s electric field is conservative. Let us assume that the movable plate of the condenser is held fixed
in position x. The electric energy input to the ideal condenser gets stored in the electric field so that
dWe = v dq = dWf
(4.78)
The total field energy is
Wf =
Ú
q
0
v dq
(4.79)
x0
Movable plate
x
Mechanical
system
Fixed plate
q,i
Ft
+
l
v
G
–
Fig. 4.18
Fm
188 Electric Machines
In a condenser v and q are linearly related as
C = q/v = capacitance of the device
\
1 q2
(4.80)
2 C
The capacitance C is a function of configuration (position x of the movable plate) and can be expressed as
Wf =
e0 A
( x0 - x)
where A = plate area and e0 = permittivity of free space. Thus Wf , the field energy is a function of two
independent variables q and x, i.e.
C=
1 q2
(4.81)
2 C ( x)
The expression of Eq. (4.81) for the field energy immediately reveals that the electric field energy can be
changed electrically by changing q or mechanically by changing x, i.e. moving the movable plate.
The field energy can also be written as
Wf (q, x) =
1
1
vq = C(x)v 2
2
2
The energy density in the electric field can be expressed as
Wf (v, x) =
1 D2
1
= e0 E 2
2 e0
2
0
E = electric field intensity or potential gradient
= D/e0
D = electric field flux density
wf =
where
Ú
D
EdD =
(4.82)
(4.83)
Energy Conversion
Let the movable plate of the device be now permitted to move under the action of the electric field force Ff.
As per the principle of energy conservation:
Mechanical energy output (work done by the field force)
= electric energy input – increase in the field energy
or
Ff dx = v dq – dWf
(4.84)
Let us choose (v, x) as independent variables. Then
q = q(v, x)
∂q
∂q
dv +
dx
∂v
∂x
Wf = Wf (v, x)
dq =
and
∂W f
dv +
∂W f
dx
∂v
∂x
Substituting Eqs (4.85) and (4.86) in Eq. (4.84) and reorganizing
dWf =
Ê ∂q ∂W f ˆ
Ê ∂q ∂W f ˆ
dx + Á v
dv
Ff dx = Á v
˜
∂x ¯
∂v ˜¯
Ë ∂x
Ë ∂v
(4.85)
(4.86)
(4.87)
Principles of Electromechanical Energy Conversion
189
Since v and x are independent variables, the coefficient of dv in Eq. (4.87) must be zero. Hence
v
Fdx =
so
∂W f
∂q
(v, x) –
(v, x) = 0
∂x
∂x
∂
(vq (v, x) – Wf (v, x))
∂x
(4.88)
Defining coeneregy as
W¢f (v, x) = vq(v, x) – Wf (v, x)
(4.89)
the electric field force Ff can be written as
Ff =
∂W f¢ (v, x)
(4.90)
∂x
If instead (q, x) are taken as independent variables
Wf = Wf (q, x)
dWf =
∂W f
∂q
dq +
∂W f
∂x
dx
(4.91)
Substituting in Eq. (4.84)
∂W f ˆ
∂W f
Ê
dq dx
Ff dx = Á v ˜
∂q ¯
∂x
Ë
(4.92)
Since v and q are independent variables, the coefficient of dq in Eq. (4.92) must be zero. Hence Eq. (4.92)
gives
∂W f ( q , x )
Ff = -
∂x
(4.93)
From Eqs (4.89) and (4.79), the coenergy* can be written as
v
W f¢ (v, x) =
Ú
q dv
(4.94)
W f¢ (v, x) =
1
C(x)v 2
2
(4.95)
0
For a linear system q = Cv
q
*
W¢f (v, x) = vq –
Ú v dq
0
Integrating the second term by part
W¢f (v, x) = vq – vq +
v
=
Ú q dv
0
Ú
v
0
q dv
190 Electric Machines
The coenergy density is given by
W f¢ =
1
e E2
2 0
(4.96)
EXAMPLE 4.13 Find an expression for the force per unit area between the plates of a parallel plate
condenser in terms of the electric field intensity. Use both the energy and coenergy methods. Find the value
of the force per unit area when E = 3 ¥ 106 V/m, the breakdown strength of air.
SOLUTION
With reference to Fig. 4.18, the energy in the electric field is
Wf (q, x) =
1 q2
1 q 2 ( x0 - x)
=
2 C
2
Ae 0
From Eq. (4.95)
Ff = -
∂W f ( q , x )
∂x
q = DA = e0EA
But
=
1 q2
2 Ae 0
1
1
e0E 2A or Ef /A = e0E2
2
2
1
=
¥ (3 ¥ 106)2 ¥ 8.85 ¥ 10–12 = 39.8 N/m2
2
Ff =
From Eq. (4.95), the field coenergy is
W¢f (v, x) =
1 2 1 2 Ae 0
Cv = v
2
2 ( x0 - x)
Now from Eq. (4.90)
Ef =
∂W f¢ (v, x)
∂x
=
1 2 Ae 0
v
2 ( x0 - x) 2
v = E(x0 – x)
1
1
\
Ff = e0E2A or Ff /A = e0E 2 (as before)
2
2
It may be observed here that while the force density on the bounding surface in a magnetic field near saturation
was found to be 1.02 ¥ 106 N/m2, it has a value* of only 39.8 N/m2 in an electric field with the electric intensity at its
breakdown value. This indeed is the reason why all practical energy conversion devices make use of the magnetic field as
the coupling medium rather than the electric field. Electric field devices are sometimes used as transducers.
But
4.7
DYNAMICAL EQUATIONS OF ELECTROMECHANICAL SYSTEMS
Figure 4.19 shows an electromagnetic relay whose armature is loaded with spring K, damper B, mass M and
a force generator F. Figure 4.20 shows the abstracted diagram of a general electromechanical system. It is
easily noticed that the electromechanical device has one electrical port and one mechanical port (one terminal
of the mechanical port being the ground) through which it is connected to the electrical source on one side
and mechanical load on the other side. In general there can be more than one electrical port (multiply excited
system).
* Such a low value results from e0 = 8.85 ¥ 10–12 compared to m0 = 4p ¥ 10–7
Principles of Electromechanical Energy Conversion 191
l0
x
K
i
B
+
N
v
M
–
A
f
Load
Fig. 4.19
R
i
Electromechanical x
conversion F
f
device
+
v
–
Mechanical
load
Fig. 4.20 Abstracted electromechanical system diagram
Let the electromechanical device has an inductance
L = L(x)
The governing electrical equation is
dl
d
= iR +
[L(x)i]
dt
dt
di
dL( x) dx
+i
◊
= iR + L( x)
dt
dx
dt
v = iR +
self-inductance
voltage
Now
\
W f¢ (i, x) =
Ff =
(4.97)
speed voltage
1
L(x)i 2
2
∂W f¢
∂x
=
1 2 dL( x)
i
2
dx
(4.98)
The mechanical power output is given by
Pm = Ff
=
1 2 dL( x) dx
dx
¥
= i ¥
2
dx
dt
dt
1 Ê dL( x) dx ˆ
i i
¥ ˜
2 ÁË dx
dt ¯
(4.99a)
192 Electric Machines
1
current ¥ speed voltage*
(4.99b)
2
Mechanical power output results (i.e. electrical power is converted to mechanical form) when the current
in the device flows in opposition to the speed voltage. When the current in the device is in the same direction
as the speed voltage, electrical power is output, i.e., mechanical power is converted to electrical form.
The governing differential equation of the mechanical system is
=
Ff = M
d2x
+B
2
dt
d2x
dx
+ Kx + f
dt
(4.100a)
dx
1 2 dL( x)
i
= M 2 +B
+ Kx + f
(4.100b)
dt
2
dx
dt
Now for the specific system of Fig. 4.19, when the armature is in position x, the self-inductance L is found
below:
or
P=
m0 A
2 ( I 0 - x)
m0 AN 2
2 ( I 0 - x)
L(x) = N 2P =
m0 AN 2
dL( x)
=
dx
2 (l0 - x) 2
Substituting in Eqs (4.97) and (4.98), the two differential equations defining the system’s dynamic
behaviour are obtained as:
v = iR +
m0 AN 2 i 2
4(l0 - x)
2
= M
m0 AN 2
2 (l0 - x)
d2x
dt
2
+B
2
¥
di m0 AN 2 i
dx
+
¥
dt 2(l0 - x) 2 dt
dx
+ Kx + f
dt
(4.101)
(4.102)
These are nonlinear differential equations which can be solved numerically on the digital computer.
However, for small movement around the equilibrium point the following procedure can be adopted.
Let the equilibrium point be (V0, I0, X0, f0). At equilibrium the system is stationary and all derivatives are
zero. Thus from Eqs (4.101) and (4.102), the following relationships between equilibrium values are obtained.
V0 = I0 R
m0 AN 2 I 02
4(l0 - x0 ) 2
= KX0 + f0
(4.103)
(4.104)
Let the departure (small) from the equilibrium values be
(v1, i1, x1, f1)
* Half the electrical input (ei) is stored in the magnetic field. This agrees with Eq. (4.38). In continuous energy conversion devices (electric motors and generators), however, the average energy stored in the magnetic field remains
constant over a cycle of operation, so that the electrical power input (EI for a dc device where E and I are dc speed
voltage and current input or EI cos f for an ac device where E is the rms speed voltage and I is the rms current
input while cos f is the power factor) is fully converted to mechanical form or vice versa.
Principles of Electromechanical Energy Conversion
Then
m0 AN 2
V0 + v1 = (I0 + i1)R +
m0 AN 2 ( I 0 + i1 ) 2
4(l0 - X 0 - x1 )
= M
2
d 2 x1
dt
2
2(l0 - X 0 - x1 ) 2
+B
◊
193
di1 m0 AN 2 ( I 0 + i1 ) dx1
+
dt 2(l0 - X 0 - x1 ) 2 dt
dx1
+ K(X0 + x1) + f0 + f1
dt
Neglecting products of small departures and small departures compared to equilibrium values, and also
cancelling out equilibrium terms as per Eqs (4.103) and (4.104),
v1 = i1R +
2 m0 AN 2 I 0 i1
4 (l0 - X 0 )
= M
2
m0 AN 2 di1
m AN 2 I 0 dx1
+ 0
2(l0 - X 0 ) dt 2(l0 - X 0 ) 2 dt
dx1
dt
2
+B
dx1
+ Kx1 + f1
dt
(4.105)
(4.106)
Equations (4.105) and (4.106) are linear differential equations governing the system behaviour for small
incremental values around the equilibrium values (called the operating point). These can be easily solved
analytically for dynamic or steady-state conditions.
Electromechanical energy conversion takes place via the medium of a magnetic or electric field–the
magnetic field being most suited for practical conversion devices.
Energy can be stored or retrieved from magnetic system by means of an exciting coil connected to an
electric source. The field energy can be given by
Wf = Wf (l, x)
l – flux linkage
x – air-gap between the armature and core
i – current
Mechanical force is given by
∂ W f (l , x)
Wf = Wf (i, x)
or
∂ W ¢ (l , x)
∂x
∂x
For doubly excited magnetic field system, field energy is given by
Ff = -
or Ff =
l1
Wf = (l1, l2, q) =
l2
Ú i dl + Ú i dl
1
0
1
2
2
0
Electromechanical energy conversion via the electric field is analogous to the magnetic field, charge in
the electric field is analogous to flux linkages and voltage to current in the magnetic field case
Wf =
q – charge in Coulombs
C – capacitance of the device
1 q2
2 C
194 Electric Machines
4.1 In the electromagnetic relay of Fig. 4.11,
the exciting coil has 1000 turns. The crosssectional area of the core is A = 5 cm ¥
5 cm. Reluctance of the magnetic circuit may
be assumed negligible. Also neglect fringing
effects.
(a) Find the coil inductance for an air-gap of
x = 1 cm. What is the field energy when
the coil carries a current of 2.0 A? What
is the force on the armature under these
conditions?
(b) Find the mechanical energy output when
the armature moves from xe = 1 cm to xb
= 0.5 cm assuming that the coil current is
maintained constant at 2.0 A.
(c) With constant coil current of 2.0 A, derive
an expression for force on armature as a
function of x. Find the work done by the
magnetic field when x changes from xe
= 1cm to xb = 0.5 cm from
Ú
xb
xe
Ff dx.
Verify the result of part (b).
(d) Find the mechanical energy output in part
(b) if the flux linkages are maintained
constant corresponding to a coil current
of 2.0 A.
4.2 In Fig. 4.7(b) if the i-l curve ab is assumed
to be a straight line, find the expression for
the mechanical energy output. If this figure
pertains to the electromagnetic relay of
Fig. 4.11, find the value of the mechanical
energy output, given that: i1 = 2.0 A, i2 = 1.5
A, xa = 1 cm and xb = 0.5 cm.
4.3 Consider the cylindrical iron-clad solenoid
magnet of Fig. 4.9. The data for magnetizing
curve of the iron part of the solenoid are given
in Ex. 4.1. For g = 0.2 cm, find the force on the
plunger if l is assumed constant corresponding
to an exciting current of 2.25 A. Why does this
value differ from that calculated in Ex. 4.2?
4.4 For the cylindrical iron-clad solenoid magnet
of Fig. 4.9, assume that the magnetic path
4.5
4.6
4.7
4.8
reluctance remains constant at a value
corresponding to the linear part of the
magnetization curve.
(a) Derive an expression for the force in
terms of g for constant coil current
2.25 A. Calculate the value of the force
for g = 1 and 0.2 cm.
(b) What is the electrical energy input to
the system when g changes from 1 to
0.2 cm, while the coil current is
maintained constant at 2.25 A.
(c) Calculate the work done on the plunger
during the movement specified in part
(b).
(d) With the coil current maintained constant
at 2.25 A. What is the direction and
magnitude of the electrical energy flow if
the plunger is made to move from g = 0.2
to 1 cm?
Repeat part (c) of Problem 4.4 with the
nonlinear magnetization curve for the iron
path. Compare the two results and comment.
For the electromagnetic relay of Fig. 4.11,
calculate the maximum force on armature if
saturation flux density in the iron part is 1.8 T.
Given: cross-sectional area of core = 5 cm ¥
5 cm, coil turns = 1000.
For the electromagnetic device shown in
Fig. P4.7, assume the reluctance of the iron
part of the magnetic circuit to be negligible.
Determine the time-average force on the
movable member at any fixed position of the
moving member, if
(a) i = I cos wt
(b) v = V cos wt
Two coils have self- and mutual-inductances
of
2
(1 + 2 x)
L12 = (1 – 2x)
L11 = L22 =
Principles of Electromechanical Energy Conversion
x
(c) the two coils are connected in series
across a voltage source of 100 cos 314t.
4.10 The doubly-excited magnetic field system of
Fig. 4.15 has coil self- and mutual-inductances
of
Movable member
N
Cross-sectional
area A
Air-gap
negligible
L11 = L22 = 2 + cos 2q
L12 = cos q
i
R
v
+
where q is the angle between the axes of the
coils. The coils are connected in series and
–
Fig. P4.7
carry a current of i = 2 I sin wt. Derive an
expression for the time-average torque as a
function of angle q.
4.11 In the rotary device of Fig. 4.15, when the rotor
is the region of q = 45°, the coil inductances
can be approximated as
The coil resistances may be neglected.
(a) If the current I1 is maintained constant at
5 A and I2 at –2 A, find the mechanical
work done when x increases from 0 to
0.5 m. What is the direction of the force
developed?
(b) During the movement in part (a), what is
the energy supplied by sources supplying
I1 and I2?
4.9 Two coils have self- and mutual-inductances
of
L11 = L22 =
L11 = L12 = 2 +
L12 = L21 =
2
(1 + 2 x)
pÊ
q ˆ
12 ÁË 90 ˜¯
i1 = 5 A, i2 = 0
i1 = 0, i2 = 5 A
i1 = 5 A, i2 = 5 A
i1 = 5 A, i2 = –5 A
Find the time-average torque if coil 1
carries a current of 5 sin 314t while
coil 2 is short circuited.
4.12 Figure P4.12 shows the cross-sectional view
of a cylindrical plunger magnet. The position
(a)
(b)
(c)
(d)
(e)
R
t
x
+
V
Plunger of mass M
Spring constant k
d
–
N
pÊ
q ˆ
12 ÁË 45 ˜¯
where q is in degrees
Calculate the torque of field origin if the rotor
is held in position q = 45° with
1
L12 =
1 + 2x
Calculate the time-average force and coil
currents at x = 0.5 m if:
(a) both the coils connected in parallel
across a voltage source of 100 cos 314t,
(b) coil 2 is shorted while coil 1 is connected
across a voltage source of 100 cos 314t,
and
i
195
D
h
Cylindrical core
Fig. P4.12
Coefficient of friction B
Nonmagnetic sleeve
196
Electric Machines
of the plunger when the coil is unexcited is
indicated by the linear dimension D. Write
the differential equations describing the
dynamics of the electromagnetic system.
Determine the equilibrium position of the
plunger and linearize the describing equation
for incremental changes about the equilibrium
point. Assume the iron to be infinitely
permeable.
4.13 For the electromagnet of Fig. P4.13 write
the dynamical equation. Assume the crosssectional area of each limb of the magnet to be
A and that the coupling between the two coils
to be tight. Iron is to be taken as infinitely
permeable.
4.14 For the electromechanical system shown
in Fig. P4.14, the air-gap flux density under
steady operating condition is
B(t) = Bm sin wt
Find
(a) the coil voltage,
(b) the force of field origin as a function of
time, and
(c) the motion of armature as a function of
time.
A
D
Rest position of
armature
x
i
Armature, mass M
+
D
i2
Rest position of
armature
x
A
+ i1
v1
–
–
K
N
R2
Armature, mass M
+
v2
–
v1
R1
N1
K
A
B
Coil resistance
negligible
Fig. P 4.14
N2
A
B
Fig. P4.13
1. Define field energy and co-energy.
2. Why do all practical energy conversion
devices make use of the magnetic field as a
coupling medium rather than an electric field?
3. What are the special applications where the
electric field is used as a coupling medium
for electromechanical energy conversion?
Also explain why electric field coupling is
preferred in such applications.
4. Elaborate the statement, “In a round rotor
machine (uniform air-gap) with exciting coil
placed in stator slots no reluctance torque is
developed”.
Basic Concepts in Rotating Machines
5
5.1
197
INTRODUCTION
It was seen in Ch. 4 that electromechanical energy conversion takes place
whenever a change in flux is associated with mechanical motion. Speed voltage is
generated in a coil when there is relative movement between the coil and magnetic
field. Alternating emf is generated if the change in flux linkage of the coil is
cyclic. The field windings which are the primary source of flux in a machine are,
therefore, arranged to produce cyclic north-south space distribution of poles.
A cylindrical structure is a natural choice for such a machine. Coils which are the seats of induced emf’s
are several in number in practical machines and are suitably connected in series/parallels and in star/delta
3-phase connection to give the desired voltage and to supply the rated current. This arrangement is called
the armature winding. When the armature coils carry currents they produce their own magnetic field which
interacting with the magnetic field of the field winding produces electro-magnetic torque tending to align
the two magnetic fields.
The field winding and armature winding are appropriately positioned on a common magnetic circuit
composed of two parts—the stator (stationary member) and the rotor (rotating member). The stator is
the annular portion of a cylinder in which rotates a cylindrical rotor; there being an appropriate clearance
(air-gap) between the two. The rotor axle is carried on two bearings which are housed in two end-covers
bolted on the two sides of the stator as shown in Fig. 5.1. The stator and rotor are made of high permeability
magnetic material—silicon steel. Further, the member in which the flux rotates is built up of thin insulated
laminations to reduce eddy-current loss.
Since electromechanical energy conversion requires relative motion between the field and armature
winding, either of these could be placed on the stator or rotor. Because of practical convenience, field
windings are normally placed on the rotor in the class of machines called the synchronous machines; the
cross-sectional view of one such machine is shown in Fig. 5.1. The armature winding is housed in suitably
shaped slots cut out in the stator. The field winding is supplied with dc from an external source, called the
exciter, through a pair of slip-rings as shown in Fig. 5.1. The exciter is generally coupled directly to the
rotor shaft of the synchronous machine.
In an induction machine the stator has a 3-phase winding which draws a component of current from the
mains to set up a cyclic flux pattern in the air-gap which rotates at a speed corresponding to supply frequency
(synchronous speed ) and the rotor is either properly wound and the winding is short-circuited or is merely
a set of copper (or aluminium) bars placed in rotor slots short-circuited at each end by means of end-rings.
198 Electric Machines
Armature winding
Stator (laminated)
End cover
Pole shoe (laminated)
End Cover
Brushes
Slip-rings and
brushes (for feeding
direct current to the
field winding)
Field windings
Field pole
Bearing
Rotor
Bearing
Insulation
Coupled to primemover (supplying
mechanical power)
Coupled to dc
generator to
supply the field
winding
Fig. 5.1 AC machine-synchronous type
In dc machines a most convenient and practical arrangement is to generate alternating voltages and to
convert these to dc form by means of a rotating mechanical rectifier. Therefore, it is necessary that the
armature winding in a dc machine is on the rotor and the field winding on the stator.
In this chapter, through suitable approximations, the physical machine will be reduced to a simple
mathematical model.
5.2
ELEMENTARY MACHINES
Synchronous Machine
Figure 5.2 shows the simplified version of an ac synchronous generator with a 2-pole field winding on the
rotor and a single coil aa¢ on the stator. The type of rotor poles are known as salient (projecting) poles; and
are excited by means of dc fed to the concentrated field winding. The current is fed to the rotor via two
slip-rings and carbon brushes as shown in Fig. 5.1. Figure 5.2 also shows two mean paths of magnetic flux
(shown dotted). The magnetic neutral regions are located in the interpolar gaps. Without any significant loss
of accuracy, the reluctance of the iron path will be neglected. Assuming that the air-gap over the pole-shoes
is uniform, the flux density in the gap over the pole-shoes is constant (as mmf acting along any flux path is
constant for the concentrated field winding); the flux density in air layer along the stator periphery gradually
falls off to zero in the interpolar region. The result is a flat-topped flux density wave as shown in Fig. 5.3.
Since the magnetic flux enters the stator normally, the relative movement between the stator conductors (coilsides of the stator coil) and the air-gap flux density is mutually perpendicular. This results in the emf being
induced along the stator conductors as per the Blv rule, with emf direction on being governed by v ¥ B or
the Fleming’s right-hand rule. It is to be further observed that the magnetic flux density is constant along the
axial length; the end effects (fringing) at either end of the cylindrical structure are neglected in this model.
Basic Concepts in Rotating Machines
199
Magnetic axis of
the field
a
+
a = w t, w = speed in rad elect/s
N
n
e
Stator coil
axis
S
–
a¢
Fig. 5.2
Elementary synchronous generator—salient-pole 2-pole rotor
B
0
p
q
2p
Fig. 5.3
In ac machines it is desirable for the induced emfs to
be sinusoidal in waveform, therefore, flux density wave in
the machine air-gap must be sinusoidal. This is achieved
in the salient pole construction (with concentrated field
coils) by providing nonuniform air-gap above poleshoes; minimum air-gap in the middle of the pole-shoe
progressively increasing towards outer edges. Here it
will be assumed the air-gap flux density wave is perfectly
sinusoidal as shown in Fig. 5.4.
Another method of obtaining a sinusoidal B-wave in
air-gap is the use of the nonsalient pole structure, i.e. a
cylindrical rotor with uniform air-gap but with a suitably
B
B - wave in space
w
a¢
a
0
p
Coil-span = 1/2
B-wave length
or p rad
Fig. 5.4
2p
q
200
Electric Machines
distributed field winding along the rotor periphery as shown in Fig. 5.5. In this structure as one moves away
from the pole-axis, the flux paths link progressively smaller number of field ampere-turns. The ampere-turns
can be so distributed as to give a nearly sinusoidal* B-wave in space as shown in Fig. 5.4.
The armature coil in the elementary 2-pole machine of Fig. 5.2 is placed in two diametrically opposite slots
notched out on the inside of the stator. The coil can have many
Magnetic axis
turns. Various shapes of coils are employed; Fig. 5.6 shows a
of the field
diamond-shaped coil. Each coil has two sides termed coil sides†.
The active length of coil-sides equals the magnetic length of
the stator over which the B-wave acts to induce emf. No emf is
produced in end connections which are suitably formed so as
N
to be neatly accommodated on the stator ends away from the
rotating parts.
The two coil-sides of our elementary machine are shown in
the cross-sectional developed diagram of Fig. 5.6. Since the
coil-sides are 1/2 B-wave length (p radians) apart, the voltages
induced‡ in the two coil-sides (Blv, where v is the peripheral
S
velocity of the pole-faces) are identical in value but opposite in
sign so that the total coil voltage is double the coil-side voltage
and has the same waveform as the B-wave and is shown in
Fig. 5.7. One cycle of the alternating emf is generated in one
Fig. 5.5 Nonsalient pole (cylindrical) rotor
revolution of the rotor.
End connections
(overhang)
Coil sides
Active conductor
length /
Stator
length
a
a¢
a
a¢
Coil-span (pitch) =
1/2 B-wave length
(b) Multi-turn coil
(a) Single-turn coil
Fig. 5.6
* Actual B-wave in this case will be a stepped wave, whose fundamental is being considered in our model. The high
frequency harmonics corresponding to the steps are ignored as they do not affect machine performance significantly.
† A coil-side in a single-turn coil is a conductor and in a multi-turn coil it is a bundle of conductors (equivalently it
can be considered as one conductor).
‡ The direction of induced emf can be established by Fleming’s right-hand rule in which the motion is that of a
conductor with respect to the field.
Basic Concepts in Rotating Machines
201
Consider now a machine with a 4-pole structure, as shown in Fig. 5.8, the poles being alternately north
and south. The flux-density space wave of this structure is drawn in Fig. 5.9. It has two complete cycles in the
total angle of 2p radians which will now be referred to as the mechanical angle. It is obvious from Figs 5.8
e
a1
Coil-span = 1/2
B-wave length or
p rad elect
N
n
0
p
wt
2p
S
S
a¢1
a¢2
N
a2
Fig. 5.7
Coil emf in the elementary machine of
Fig. 5.2 (w
Fig. 5.8
An elementary 4-pole synchronous generator
and 5.9 that now two coils can be placed symmetrically-one coil under each pair of poles. Each coil has a
span ( pitch) of 1/2 B-wave length and the axes of the two coils are spaced one full B-wave length apart. It is
immediately seen that magnetic and electrical conditions existing under one pair of poles are merely repeated
under every other pole-pair. The emf’s induced in coils a1 a¢l and a2 a¢2 are both alternating, equal in magnitude
and in time phase to each other. It is convenient to work in terms of one pole pair of a multi-polar machine
(number of poles must obviously be even). The total angular span of one pole pair is therefore taken as 2p
and is referred to as electrical angle as different from the actual mechanical angle. Let
q = electrical angle
q m = mechanical angle
B
Spacing between coil axis
(B - wave length or 2p rad elect)
B-wave
a1
a¢1
0
p
p
a2
2p
Coil-span = 1/2 B-wave length = p rad elect
Fig. 5.9
a¢2
2p
4p
qm (mech rad)
q (elect rad)
202
Electric Machines
These angles are both shown in Fig. 5.9 Taking the ratio of total electrical and mechanical angles of a
P-pole machine,
2 p ¥ ( P / 2) P
q
=
=
2p
2
qm
Ê Pˆ
q = Á ˜ qm
(5.1)
Ë 2¯
The span of the coil, called the coil-pitch or coil-span, which was indicated to be 1/2 B-wavelength, will be
p rad (180°) in electrical angle and is fixed irrespective of the number of machine poles. Such a coil is called
full-pitched. For the time being, it will be assumed that all coils are full-pitched. Short-pitched (chorded)
coils, i.e., coils with angular span less than 180∞ elect. are also employed in practice and will be discussed
in Sec. 5.3.
In a P-pole machine, one cycle of alternating emf is generated in each coil as one pole-pair of the rotor
poles glides past the stator. Thus for one complete revolution of the rotor, P/2 cycles of emf are generated in
the coil. Therefore, the frequency of the voltage wave (it is a time wave) is
\
P
n
nP
¥
=
Hz
2 60 120
n = speed of the rotor in revolution per minute (rpm)
f=
where
(5.2)
Differentiating each side of Eq. (5.1) with respect to time
Ê Pˆ
w = Á ˜ wm
Ë 2¯
where
(5.3)
w = rotor speed in electrical rad/s
wm = rotor speed in mechanical rad/s
Obviously
w = 2p f
rad (elect.)/s
The two coils of the 4-pole generator of Fig. 5.8 are seats
of identical emfs and can be connected in series or parallel
as shown in Figs 5.10(a) and (b). The series connection
gives double the voltage of one coil and can handle the same
maximum current as any one coil. The parallel connection
has the same voltage as that of each coil and has twice the
maximum current-carrying capacity of one of the coils. The
designer exploits the series-parallel arrangement of coil
groups to build a machine of desired voltage and current
rating.
(5.4)
a¢1
a1
a¢2
a2
(a)
a1
a¢1
a
a¢
a2
a¢2
(b)
Fig. 5.10
Three-phase Generator (Alternator)
Practical synchronous generators are always of the 3-phase kind because of the well-known advantages*
of a 3-phase system. If two coils were located at two different space locations in the stator of Fig. 5.2, their
emf’s will have a time phase difference corresponding to their electrical space displacement. If three coils
* All power generation and transmission systems are of 3-phase. Except for fractional-kW and for certain special
purposes all motors are of 3-phase.
Basic Concepts in Rotating Machines
203
are located in the stator of the 2-pole machine of Fig. 5.2 at relative electrical spacing of 120∞ (or 2p/3 rad),
an elementary 3-phase machine results as is shown in Fig. 5.11(a). The corresponding 4-pole arrangement is
depicted in Fig. 5.11(b) where each phase has two symmetrically placed coils corresponding to each pair of
poles. The coils of each phase are series/parallel connected and the three phases of a synchronous generator
are generally connected in star as shown in Figs 5.12(a) and (b).
60°elect
60°elect
a1
a
b¢2
N
b¢
120°elect
c¢2
N
c2
c¢
120°
a¢2
b1
S
S
a¢1 120°
elect
120°
b2
S
c
c1
N
b
c¢1
a¢
b¢1
a2
(a) Three-phase, 2-pole synchronous
generator
(b) Three-phase, 4-pole synchronous
generator
Fig. 5.11
A
A
a1
c1
c¢1
c2
c¢2
b¢2
C
a¢1
a1
a2
a2
a¢1
a¢2
c¢2
a¢2
b¢1
b2
120°
B
(a)
b¢1
b1
c2
b1
c¢1
C
c1
b¢2
b2
B
(b)
Fig. 5.12 Star connected elementary 3-phase generator.
The process of torque production in a machine will be explained in Sec. 5.5 after gaining some familiarity
with the rotating magnetic field produced by a current-carrying 3-phase winding.
The dc Machine
Figure 5.13 shows a 2-pole elementary dc machine with a single coil rotating armature. It may be seen that
the field winding is stationary with salient poles whose pole-shoes occupy a major part of the pole-pitch.
An alternating emf is induced in the coil due to rotation of the armature past the stator (field) poles. The
two ends of the armature coil are connected to two conducting (copper) segments which cover slightly less
than a semicircular arc. These segments are insulated from each other and from the shaft on which they are
Electric Machines
204
mounted. This arrangement is called the commutator. Current is collected from the commutator segments by
means of copper-carbon brushes. The connection of the coil to the outside circuit reverse each half cycle in
such a manner that the polarity of the one brush is always positive and the other negative. This indeed is the
rectification action of the commutator-brush arrangement. As already stated, it is because of this requirement
of the mechanical rectification action, to obtain direct voltage from the alternating induced emf, that the
armature in a dc machine must always be a rotor and the field winding must always be placed on the stator.
Mean flux paths
N
dc excitation
Brush
a
+
dc output
n
–
a¢
Commutator
segments
Armature
S
Pole-shoe
Pole
Fig. 5.13 A 2-pole elementary dc machine
The air-gap under the poles in a dc machine is almost uniform and further the pole-shoes are wider than in
a synchronous machine (pole-arc is about 70% of the pole-pitch). As a result, the flux density in air around
the armature periphery is a flat-topped wave as shown in Fig. 5.14(a). The voltage induced in the coil (fullpitched) has a similar wave shape. However, at the brushes both half cycles of voltage wave are positive,
(Fig. 5.14(b)) because of the commutator’s rectification action. It is easy to see that such a waveform has a
B
B-wave
Brush
voltage
Bpeak
2p/P
a¢
a
0
p
4p/P
2p
qm
q
0
p
1/2 B-wave length or
p rad elect
Fig. 5.14(a)
Fig. 5.14(b)
2p
wt
Basic Concepts in Rotating Machines
205
higher average value for a given value of Bpeak’ which therefore explains why a dc machine is designed to
have a flat-topped B-wave.
The actual dc machine armature has several coils placed in slots around the armature which are connected
in the from of a lap or wave winding. The addition of coil emfs (waveform as in Fig. 5.14(b)) with time phase
displacement results in almost constant dc voltage at brushes. The reader may add two such waveforms at 90∞
elect displacement and find that dc (average) voltage becomes double but per cent voltage variations reduce
and the frequency of variations doubles.
5.3 GENERATED EMF
The quantitative expressions will now be derived for the generated emf in synchronous and dc machine
armatures. Some idea of ac windings will be advanced here and certain sweeping statements made, but the
discussion on dc winding will be postponed to Ch. 7.
Generated Voltage of ac Winding
The B-wave of a synchronous machine (in general multi-polar) assumed sinusoidal is drawn in Fig. 5.15 and
a single full-pitched coil (coil-side space separation p rad (180∞) elect.) is shown in cross-sectional from. The
B-wave moves towards left with a speed of w elect.
B
rad/s or wm mech. rad/s. At the origin of time the
B-wave
coil-sides are located in the interpolar region where
Bp
the pole flux links the coil. At any time t the coil has
w,wm
relatively moved by
a = wt elect. rad
a
to the right of the B-wave. The B-wave can be
expressed as
B = Bp sin q
ÊP ˆ
= Bp sin Á q m ˜
Ë2 ¯
qm
a¢
a
p+a
p
2p
q
Fig. 5.15
coil at any time t
where Bp = peak flux density
Since the flux is physically spread over the mechanical angle, the flux f linking the coil can be computed
by integrating over the mechanical angle. Thus
2 (p + a ) / P
f=
Ú
2a / P
where
ÊP ˆ
B p sin Á q m ˜ lr dqm
Ë2 ¯
l = active coil-side length (axial stator length) and
r = mean radius of the stator at the air-gap.
Since
qm =
2
q
P
Eq. (5.5) modifies to
Ê 2ˆ
f= Á ˜
Ë P¯
p +a
Ú
a
Bp lr sin q dq
(5.5)
206
Electric Machines
Ê 2ˆ
= Á ˜ 2B p lr cos a
Ë P¯
Ê 2ˆ
= Á ˜ 2Bp lr cos wt = F cos wt
Ë P¯
(5.6)
It is, therefore, seen that the flux linking the coil varies sinusoidally and has a maximum value of
4
B lr (flux/pole)
P p
at a = wt = 0, which indeed is flux/pole. The flux linkages of the coil at any time t are
F=
(5.7)
l = Nf = NF cos wt
(5.8)
where N = number of turns of the coil.
Hence the coil induced emf is
dl
= wNF sin wt
(5.9)
dt
The negative sign in Eq. (5.9) (e = –dl/dt) accounts for the fact that the assumed positive direction of emf
and current in the coil aa¢ of Fig. 5.2 produces flux along the coil axis causing positive flux linkages. In case
of the transformer the positive direction of emf was assumed such as to cause a current which would produce
negative flux linkages and therefore the induced emf law used was e = +dl/dt.
It may be observed that the spatial flux density wave upon rotation causes time-varying flux linkages with
the coil and hence the production of emf, an effect which is produced by a fixed-axis time-varying flux in a
transformer. The time-variation factor is introduced by rotation causing the phenomenon of electromechanical
energy conversion which is not possible in a transformer with fixed-axis time-varying flux.
The rms value of emf induced in the coil from Eq. (5.9) is
e= -
E=
2 p fNF = 4.44 fNF volts
which is the same result as in a transformer except for the fact that F here is the
flux/pole.
It may be observed from Eqs (5.6) and (5.9) that the sinusoidally varying flux
linking the coil (represented by the phasor F ) leads the sinusoidally varying emf
(represented by the phasor E ) by 90∞. The phasor relationship is illustrated by the
phasor diagram of Fig. 5.16. This is in contrast of the transformer case wherein the
flux phasor lags the emf phasor by 90° (refer Fig. 3.4). This difference is caused
by the negative sign in the induced emf of Eq. (5.9) while a positive sign was used
for the transformer.
(5.10)
F
E
Fig. 5.16
Distributed Winding
It may be seen from Eq. (5.7) that the flux/pole is limited by the machine dimensions and the peak flux density
which cannot exceed a specified value dictated by saturation characteristic of iron. Therefore, for inducing
an emf of an appropriate value in a practical machine (it may be as high as 11/ 3 to 37/ 3 kV phase), a
large number of coil turns are needed and it is not possible to accommodate all these in a single slot-pair.
Furthermore, it may be also noticed that with one coil/pole pair/phase, i.e. one slot/pole/phase, the periphery
Basic Concepts in Rotating Machines
207
of the stator is far from being fully utilized. It is, therefore, natural to create more slots/pole/phase (SPP) on
the stator periphery. In a practical machine with S slot distributed uniformly round the stator periphery,
SPP = m =
S
qP
q = number of phases (generally q = 3)
(5.11)
Figure 5.17 illustrates a 2-pole, 3-phase machine with m = 3. The angle between adjacent slots is
pP
elect. rad
(5.12)
S
The winding of phase a in the machine has three coils (11¢, 22¢ and 33¢) which are placed in three slotpairs distributed in space with an angular separation of g elect. rad. The total angle s = mg occupied by the
phase winding along the armature periphery is called the phase spread. Such a winding is referred to as the
distributed winding. Since the machine is always wound with identical coils, the sinusoidal emfs induced in
coils 11¢, 22¢ and 33¢ have the same rms value (E ) but have a progressive time phase difference of g because
coil are uniformly distributed in space.
These coils are series connected to yield the phase voltage Ea which is the phasor sum of the coil emf’s
as shown in Fig. 5.18. It is observed from this figure that because of distribution, the rms phase voltage
is less than the algebraic sum of the rms coil voltages. This reduction ratio called the breadth factor (also
distribution factor) is to be determined now, for the general case of SPP = m.
g=
Phase 'a' belt
phase spread s
= mg = 60°
2
1
3
g
b¢
c¢
N
n
B
g E2
E1
b
Ea
A
g /2
3¢
2¢
g
E3
S
c
C
g /2
D
mg
1¢
O
Fig. 5.17
Fig. 5.18
Phasor diagram of coil emf’s to yield
It is easily seen that in Fig. 5.18 the coil emf phasors form sides of a regular polygon, the centre of whose
circumscribing circle is constructionally located in the diagram. The phase voltage Ea is given by the resultant
phasor (AD in Fig. 5.18). The breadth factor is then defined as
Kb =
AD
3 AB
or
AD
(in general)
mAB
(5.13)
208
Electric Machines
From the geometry of Fig. 5.18
AB = 2 OA sin g /2
AD = 2 OA sin g m/2
Kb =
Hence
sin mg / 2
< 1 for m > 1
m sin g / 2
(5.14)
It is seen from Eqs (5.11) and (5.12) that
p
p
or ; a fixed value independent of P and S
q
3
The induced phase emf for a distributed winding is obtained by multiplying Eq. (5.10) by Kb. Thus
mg =
E = 4.44 Kb f Nph F
Nph = total turns (in series) per phase
where
(5.15)
Harmonic Content in the Distributed Winding
The flux density wave of a synchronous machine is never exactly sine wave. Because of odd symmetry of
poles (alternately north-south), the space harmonic content of the B-wave comprises odd harmonics only,
which induce the corresponding harmonic emf’s in the winding. Figure 5.19 shows the fundamental B-wave,
and its third-harmonic; because of somewhat flat-topped nature of the B-wave, its third-harmonic is of the
“dipping’’ kind, producing a flux density dip of in the middle of the main pole. It follows from Fig. 5.19 that
n poles of the nth harmonic occur in the space occupied by one of pole the fundamental. Thus,
q (nth harmonic) = nq (fundamental)
Therefore from Eq. (5.14)
Kb (nth harmonic) =
sin mng / 2
m sin ng / 2
(5.16)
It can be easily shown (see Ex. 5.1) that Kb (nth harmonic) is less than Kb (fundamental) for important
harmonics resulting in reduction of harmonic content of the voltage of the distributed winding–an incidental
advantage.
B
Fundamental B-wave
Resultant B-wave
0
p
p
2p 3p
Third-harmonic
B-wave
Fig. 5.19
2p
6p
q (fundamental)
q (3rd-harmonic)
Basic Concepts in Rotating Machines
209
EXAMPLE 5.1 Calculate the fundamental, third and fifth harmonic breadth factors for a stator with
36 slots wound for 3-phase, 4-poles.
SOLUTION
m=
36
=3
3¥ 4
g =
and
180∞ ¥ 4
= 20° elect.
36
3 ¥ 20∞
2
= 0.96
20∞
3 sin
2
sin
(i)
Kb (fundamental) =
3 ¥ 3 ¥ 20∞
2
(ii)
Kb (third harmonic) =
= 0.667
3 ¥ 20∞
3 sin
2
3 ¥ 5 ¥ 20∞
sin
2
(iii)
Kb (fifth harmonic) =
= 0.218
5 ¥ 20∞
3 sin
2
It is noticed that Kb for the third-harmonic is sufficiently less than the fundamental and is far less for the fifth-harmonic.
sin
Short-pitched (Chorded) Coils
So far it was assumed that the stator coils are full-pitched (a span of p rad elect). Coils may have a span of
less than the full-pitch. This arrangement offers certain advantages. Consider that the coil-span is less than the
full-pitch by an elect. angle qsp (short-pitching angle) as shown in Fig. 5.20(a). With reference to Fig. 5.15,
Eq. (5.5) for the flux linking the coil now modifies as under:
2(p + a - q sp ) / P
f=
Ú
2a / P
Since
2
q
qm =
P
Ê 2ˆ
f= Á ˜
Ë P¯
ÊP ˆ
B p sin Á q m ˜ lr dqm
Ë2 ¯
(p + a - q sp )
Ú
Bp sin q lr dq
a
Ê 2ˆ
= Á ˜ 2Bp lr cos (a – qsp /2) cos qsp/2
Ë P¯
= F cos (qsp/2) cos (wt – qsp/2)
The flux linking the coil and therefore the coil emf reduces by multiplicative factor of
Kp = cos qsp/2 = pitch factor
(5.17)
(5.18)
The meaning of the pitch factor can be arrived at from another angle. In Fig. 5.20(a) with positive direction
of coil-side emfs marked in opposite direction, the coil emf is the phasor sum of coil-side emfs, i.e.
Ec = Ea + Ea ¢
In the case of a full-pitch coil E a¢ and Ea are in phase (they are p rad apart but their positive direction are
marked oppositely) so that
Ec = 2 Ea
210
Electric Machines
as shown in Fig. 5.20(b).
qsp
(p – qsp)
(short-pitch)
a
a¢
a¢
p
(full-pitch)
Ea¢
Ea
0
Ec
(b)
(a) Short-pitched coil
Ec
Ea¢
qsp
qspi2
0
Ea
(c)
Fig. 5.20
In a coil short-pitched by qsp, Ea¢ will lead (or lag) Ea by the angle qsp depending upon direction of
rotation as shown in Fig. 5.20(c). From the geometry of the phasor diagram
Ec = 2Ea cos qsp/2
Hence the reduction of the coil emf due to short-pitching is governed by the factor
Kp =
2 Ea cos q sp / 2
2 Ea
= cos qsp/2 (same as Eq. (5.18))
Because of short-pitching the expression of Eq. (5.15) for the phase voltage modifies to
E = 4.44 Kb Kp f N phF
Let
(5.19)
Kw = KbKp = winding factor; then
E = 4.44 Kw f NphF
(5.20)
For the nth harmonic the pitch factor would be
Kp = cos nqsp/2
(5.21)
Any particular harmonic can be eliminated by coil short-pitching by making
p
; k = 1, 3, … (odd)
2
For example, for elimination of the 13th harmonic
nqsp /2 = k
13q sp/2 = 90°
or
q sp ª 14°
Basic Concepts in Rotating Machines
211
Thus short-pitching helps in elimination of any particular harmonic or in reduction of the harmonic content
of the induced voltage in general. It is easily seen from Fig. 5.20(a) that short-pitched coils have shorter end
connections. Thus there is a saving in copper per coil but part of this saving is wiped off by the fact that more
series coils/ phase would now be needed to generate a specified phase voltage. A designer has to weigh these
factors in arriving at a decision on the angle of short-pitching.
Two-layer Winding
One important and commonly used way of neatly
arranging the end connection of coils in a winding
is to place two coil-sides per slot. Each coil then has
one coil-side in the bottom half of one slot and the
other coil-side in the top half of another slot (one pole
pitch away for full-pitched coils). Such a winding
is known as a two-layer winding. All the coils of a
two-layer winding are of similar shape so that these
can be wound separately and then placed in the slots.
The end connections at each end of such a coil are
given an involute kink so that the coil-sides can be
conveniently placed in the bottom of one slot and top
Fig. 5.21
of the other. A formed two-layer, multi-turn coil is
shown in Fig. 5.21.
The winding connections of a two-layer winding are best illustrated by means of a developed winding
diagram which imagines that the stator has been cut and laid out flat. One phase of a 4-pole, two-layer lap
winding is shown in Fig. 5.22. The upper layer is indicated by a continuous line and the lower layer by a
dotted line and both are drawn side-by-side in the diagram. This figure illustrates the simplest case of 4-poles
with 1 slot/pole/phase. The four coils could be series connected or various series-parallel connections could
be employed. In Fig. 5.22 the four coils of one phase are connected in parallel in two series groups of two
coils each. Obviously the phase current ia and the conductor current ic = are related as ic = ia/2.
Pole
pitch
ic
ia
a
a¢
Fig. 5.22
212
Electric Machines
EXAMPLE 5.2 A 3-phase, 16-pole synchronous generator has a star-connected winding with 144 slots
and 10 conductors per slot. The flux per pole is 0.04 Wb (sinusoidally distributed) and the speed is 375 rpm.
Find the frequency and phase and line induced emf’s. The total turns/phase may be assumed to be series
connected.
SOLUTION
f=
nP
375 ¥ 16
=
50 Hz
120
120
Total number of conductors = 144 ¥ 10 = 1440
Total number of turns =
Number of turns (series)/phase, Nph =
Slots angle, g =
Slots/pole/phase, m =
1440
= 720
2
720
= 240
3
180∞P
180∞ ¥ 16
=
= 20°
S
144
144
=3
16 ¥ 3
3 ¥ 20∞
2
= 0.96
20∞
3 sin
2
Phase emf, Ep = 4.44 Kb f Nph (series) F
sin m g / 2
Breadth factor, Kb =
=
m sin g / 2
sin
= 4.44 ¥ 0.96 ¥ 50 ¥ 240 ¥ 0.04 = 2046 V
Line voltage, EL =
3 Ep = 3543.6 V
EXAMPLE 5.3 A 3-phase, 50 Hz, star-connected alternator with 2-layer winding is running at 600 rpm.
It has 12 turns/coil, 4 slots/pole/phase and a coil-pitch of 10 slots. If the flux/pole is 0.035 Wb sinusoidally
distributed, find the phase and line emf’s induced. Assume that the total turns/phase are series connected.
SOLUTION
120 f
120 ¥ 50
=
= 10
n
600
Total slots, S = 4 ¥ 3 ¥ 10 = 120
P=
120 ¥ 12
= 480
3
SPP, m = 4
180∞ ¥ 10
Slot angle, g =
= 15°
120
N ph =
4 ¥ 15∞
sin
sin m g / 2
2
= 0.958
Kb =
=
15∞
m sin g / 2
4 sin
2
120
= 12 slots
Pole-pitch =
10
Coil-pitch = 10 slots
Basic Concepts in Rotating Machines
213
Short-pitching angle, qsp = (12 – 10) ¥ 15° = 30°
30∞
Kp = cos qsp/2 = cos
= 0.966
2
Phase emf induced, Ep = 4.44 Kb Kp f Nph (series)F
= 4.44 ¥ 0.958 ¥ 0.966 ¥ 50 ¥ 480 ¥ 0.035 = 3451 V
Also
3 ¥ 3451 = 5978 V
EL =
EXAMPLE 5.4 A 2-pole, 3-phase, 50 Hz, 2300 V synchronous machine has 42 slots. Each slot has two
conductors in a double layer winding. The coil pitch is 17 slots. Each phase winding has two parallel paths.
Calculate the flux/pole required to generate a phase voltage of 2300/ 3 V.
SOLUTION
SPP, m =
42
=7
3¥ 2
2 ¥ 180∞
= 8.57°
42
sin(7 ¥ 8.57∞ / 2
sin mg / 2
= 0.956
=
Kb =
7 ¥ sin 8.57∞ / 2
m sin g / 2
Slot angle, g =
Coil pitch = 17 slots
42
Pole pitch =
= 21 slots
2
Short pitching angle, qsp = (21 – 17) ¥ 8.57∞ = 34.28∞
Kp = cos 34.28°/2 = 0.956
42 ¥ 2
Nph (series) =
=7
2¥3¥2
Ea = 4.44 Kb Kp f Nph (series)F
2300
= 4.44 ¥ 0.952 ¥ 0.956 ¥ 50 ¥ 7 ¥ F
3
or
F = 0.94 Wb
The dc Machine
With reference to the single-coil elementary dc machine of Figs 5.13 and 5.14(a) which shows the B-wave of
the machine relative to the elementary full-pitched coil, let
F = flux/pole
Consider that the coil is lying in the interpolar region so that the full/pole (F) links is positively. Let it now
move through one pole-pitch (p rad (elect.)) in time Dt so that its flux linkages reverse in sign. The change in
flux linkages during this movement is
Dl = –2NcF
where Nc are the coil turns. For a P-pole machine
Dt =
2p
s
Pw m
(5.22)
214 Electric Machines
Hence the average coil-induced emf is
Fw m N c P
Dl
(5.23)
=
p
Dt
The dc machine armature is always wound with 2-layer winding forming a closed circuit. The coils are
suitably connected to the commutator segments made of copper, insulated from each other and from the shaft
and formed into a cylinder. The current from the armature winding is tapped through brushes placed on the
commutator periphery, 180° elect. apart. The alternate brushes are of opposite polarity and all the brushes of
one polarity are connected in parallel resulting in two armature terminals. This arrangement causes the closed
armature winding to form a series-parallel circuit. The dc voltage between the brushes of opposite polarity
is the sum of the average voltages of series turns between the brushes, each turn having the same average
voltage. The number of parallel paths depends upon the type of armature winding wave and lap type and are
Ea = -
A = 2; for wave winding
A = P; for lap winding
(5.24)
Let the machine be wound with Z conductors (a conductor is the active part of the side of a turn). Since
two conductors form a turn,
Z
2A
Using Eq. (5.23), the armature emf which equals the parallel path emf is given by
Series turns/parallel path, N =
Fw m Z Ê P ˆ
2 p ÁË A ˜¯
2p n
rad (mech.)/s
wm =
60
FnZ Ê P ˆ
Ea =
V; n = speed in rpm
60 ÁË A ˜¯
Ea =
But
Hence
(5.25)
(5.26)
It follows from Eq. (5.24) that
P
P
=
for wave winding
A
2
= 1 for lap winding
(5.27)
Since P has a minimum value of 2, the wave winding will have a larger armature emf than a lap winding
for the same values of F, n and Z.
The details of the dc machine winding, where the brushes are placed on the commutator and how the
parallel paths are formed, will be taken up in Ch. 7.
The circuit schematic (circuit model) of a de generator is drawn in Fig. 5.23. As the current in the field and
armature circuits are dc, the circuit inductances do not play any role. The armature circuit has a voltage Ea
induced in it by rotation of armature in the flux/pole F as per the relationship of Eq. (5.25). When the armature
feeds current to the external circuit (the load), there is a voltage drop Ia Ra in the armature circuit, where Ra
is the effective resistance of the armature including the voltage drop at brush contacts. The Kirchhoff voltage
law applied to the armature circuit yields
Ea = Vt + Ia Ra
where Vt is the voltage at generator terminals. This equation is usually written as
Vt = Ea – Ia Ra
(5.28)
Basic Concepts in Rotating Machines
215
EXAMPLE 5.5 A 1500 kW, 600 V, 16-pole, dc generator runs at 200 rpm. What must be the useful flux per
pole, if there are 2500 lap-connected conductors and full-load copper losses are 25 kW? Also calculate the
area of the pole-shoe if the average gap flux density is 0.85 T. The generator is feeding full load of 1500 kW
at the terminal voltage of 600 V.
SOLUTION
Figure 5.23 shows the circuit schematic of the dc generator
la
+
Armature
+
lf
Field
Load
Ea
wm
Ra
Vt
–
–
Fig. 5.23 Schematic circuit diagram of a dc generator
Full-load armature current, Ia =
1500 ¥ 1000
= 2500A
600
Copper-loss = I a2 Ra = 25 ¥ 1000
where Ra is the effective resistance of the armature circuit,
25 ¥ 1000
= 4 ¥ 10–3 = W
2500 ¥ 2500
Kirchhoff’s voltage law equation for the armature circuit is
Ra =
Ea = Vt + IaRa
= 600 + 2500 ¥ 4 ¥ 10–3 = 610 V
FnZ Ê P ˆ
But
Ea =
60 ÁË A ˜¯
where F is the useful flux/pole which links armature coil. Some of the pole flux will complete its circuit bypassing the
armature—called the leakage flux.
Substituting the values,
F ¥ 200 ¥ 2500 Ê 16 ˆ
610 =
ÁË 16 ˜¯
60
or,
F = 0.0732 Wb
Area of pole-shoe =
0.0732
= 861 cm2
0.85
EXAMPLE 5.6 A 4-pole, lap-wound dc machine has 728 armature conductors. Its field winding is
excited from a dc source to create an air-gap flux of 32 mWb/pole. The machine (generator) is run from a
primemover (diesel engine) at 1600 rpm. It supplies a current of 100 A to an electric load.
(a) Calculate the electromagnetic power developed.
(b) What is the mechanical power that is fed from the primemover to the generator?
(c) What is torque provided by the primemover?
216
Electric Machines
SOLUTION
Ea =
(a)
FnZ Ê P ˆ
¥Á ˜
Ë A¯
60
32 ¥ 10- 3 ¥ 1600 ¥ 728
¥ l = 621.2 V
60
Ia = 100 A
=
Electromagnetic power developed = Ea Ia
621 ¥ 100
= 62.12 kW
1000
(b) Mechanical power provided by prime mover, Pm
= electromagnetic power developed = 62.12 kW
But
Pm = Twm
=
Primemover torque, T =
5.4
62.12 ¥ 100
= 370.75 Nm
Ê 2p ¥ 1600 ˆ
ÁË
˜
¯
60
MMF OF DISTRIBUTED AC WINDINGS
It has been seen earlier that the armature of a practical machine has distributed winding wound for the
same number of poles as the field winding. As the armature carries current, the resultant field of its currentcarrying coils has the same number of poles as the field winding. Our approach will be to find the mmf space
distribution of the current-carrying armature by superimposing the mmf space waves of individual coils.
MMF Space Wave of a Single Coil
A cylindrical rotor machine with small air-gap is shown in Fig. 5.24(a). The stator is imagined to be wound
for two-poles with a single N-turn full-pitch coil carrying current i in the direction indicated. The figure
shows some flux lines of the magnetic field set up. A north and corresponding south pole are induced on
the stator periphery. The magnetic axis of the coil is from the stator north to the stator south. Each flux line
radially crosses the air-gap twice, normal to the stator and rotor iron surfaces and is associated with constant
mmf Ni. On the assumption that the reluctance of the iron path is negligible, half the mmf (Ni/2) is consumed
to create flux from the rotor to stator in the air-gap and the other half is used up to establish flux from the
stator to rotor in the air-gap. Mmf and flux radially outwards from the rotor to the stator (south pole on stator)
will be assumed to be positive and that from the stator to rotor as negative.
The physical picture is more easily visualized by the developed diagram of Fig. 5.24(b) where the stator
with the winding is laid down flat with rotor on the top of it. It is seen that the mmf is a rectangular space
wherein mmf of + Ni/2 is consumed in setting flux from the rotor to stator and mmf of – Ni/2 is consumed
in setting up flux from the stator to the rotor. It has been imagined here that the coil-sides occupy a narrow
space on the stator and the mmf changes abruptly from – Ni/2 to + Ni/2 at one slot and in reverse direction at
the other slot. The mmf change at any slot is
Ni = ampere-conductors/slot
and its sign depends upon the current direction.
Basic Concepts in Rotating Machines
217
Ni ampere-conductors
a
q
N
S
N
Coil magnetic axis
S
a¢
(a)
Fa1, Fundamental
MMF
Ni/2
Ni
q
0
–Ni/2
Rotor
Stator
a¢
a¢
a
S
South pole
N
North pole
(b) Developed diagram
Fig. 5.24
The mmf space wave of a single coil being rectangular, it can be split up into its fundamental and
harmonics. It easily follows from the Fourier series analysis that the fundamental of the mmf wave as shown
in Fig. 5.24(b) is
Fa1 =
4 Ê Ni ˆ
cos q = F1p cosq
p ÁË 2 ˜¯
(5.29)
218 Electric Machines
where q is the electrical angle measured from the magnetic axis of the coil which coincides with the positive
peak of the fundamental wave.
From Eq. (5.29)
Fal (peak) = F1p =
4 Ê Ni ˆ
p ÁË 2 ˜¯
(5.30)
From now onwards only the fundamental mmf wave of a current-carrying coil will be considered,
neglecting its associated harmonic space waves whose amplitudes are small*. Furthermore, as will soon be
seen, in a distributed winding, the rectangular mmf waves of individual phase group coils add up to produce
a mmf wave closer to a sine wave, i.e. the harmonics tend to cancel out.
It may be noted here that with the assumption of the narrow air-gap, the mmf distribution will be the same
if the coils were located in the rotor slots instead.
MMF Space Wave of One Phase of a Distributed Winding
Consider now a basic 2-pole structure with a round rotor, with 5 slots/pole/phase (SPP) and a 2-layer winding
as shown in Fig. 5.25. The corresponding developed diagram is shown in Fig. 5.26(a) along with the mmf
diagram which now is a stepped wave–obviously
60°
closer to a sine wave than the rectangular mmf wave of
a
a single coil (Fig. 5.24(b)). Here since SPP is odd (5),
half the ampere-conductors of the middle slot of the
phase group a and a¢ contribute towards establishment
of south pole and half towards north pole on the stator.
c¢
b¢
At each slot the mmf wave has a step jump of 2Nc ic
ampere-conductors where Nc = coil turns (equal to
conductors/layer) and ic = conductor current.
Now F1p, the peak of the fundamental of the mmf
wave, has to be determined. Rather than directly
finding the fundamental of the stepped wave, one can
c
b
proceed by adding the fundamentals of the mmf’s
of individual slot-pairs (with a span of one polepitch). These fundamentals are progressively out
of phase (space phase as different from time phase)
a¢
with each other by the angle g. This addition is easily
accomplished by defining the breadth factor Kb, which Fig. 5.25
will be the same as in the case of the generated emf of
a coil group.
Let
Nph (series) = series turns per parallel path of a phase
A = number of parallel paths of a phase
* For a rectangular wave the normalized peaks of the harmonic waves are:
4
4
4
, F5p =
, F7p =
,…
3p
5p
7p
It is needless to say that there cannot be any even harmonics.
F3p =
Basic Concepts in Rotating Machines
219
Fa1
2 Ncic
F1p
p/2
– p/2
0
N cic
(a)
2 ¥ 2 (2Ncic) + 1 ¥ (2Ncic)
2
= 5 Ncic
c
a¢
North
pole
b¢
South pole
a
North
pole
(b)
Fig. 5.26
Then
where
AT/parallel path = Nph (series) ic
AT/phase = A(Nph (series)ic) = Nph (series) ia
ia = phase current = A ic
It now follows that
Ê N ph (series) ˆ
AT/pole/phase = ÁË
˜¯ ia
P
provided the winding is a concentrated one. The fundamental peak of the concentrated winding is then
F1p (concentrated winding) =
4 Ê N ph (series) ˆ
Á
˜¯ ia
pË
P
Since the actual winding is a distributed one, the fundamental peak will be reduced by the factor Kb. Thus
F1p (distributed winding) =
Hence
Fa1 =
Ê N ph (series) ˆ
4
Kb ÁË
˜¯ ia
P
p
(5.31)
Ê N ph (series) ˆ
4
Kb ÁË
˜¯ ia cos q
P
p
(5.32)
where the pole axis is taken as the angle reference (Fig. 5.26(b)).
The effect on the mmf wave of short-pitched coils can be visualized by Fig. 5.27 in which the stator has
two short-pitched coils (ala¢1, a2a¢2) for phase a of a 2-pole structure*. The mmf of each coil establishes one
pole. From the developed diagram of Fig. 5.27(b) it is seen that the mmf wave is rectangular but of shorter
* The arrangement of coils for 3-phase 2-pole, two-layer winding, with S = 12, m = 2 and coil-pitch = 5 (chorded
by one slot) is shown in Fig. 5.28.
It is seen from Fig. 5.28 that the coil-sides in a given slot do not belong to the same phase. This is in contrast to
Fig. 5.25 depicting a full-pitch winding.
220
Electric Machines
space length than a pole-pitch. The amplitude of the fundamental peak gets reduced by a factor Kp, called the
pitch factor, compared to the full-pitch rectangular mmf wave. It can be shown** by Fourier analysis that
4
F cos qsp/2
p pole
Kp = cos qsp/2
F1p =
Hence
(5.33)
a1
a2
qsp/2
Axis of 'a' phase
a¢2
a¢1
(a) Stator with short-pitched coil at phase 'a' (single-layer winding)
Fa1
Fpole
F1p
– p/2
p/2
3 p/2
q
0
qsp/2
a¢2
a¢1
a¢2
a2
a1
a¢1
North pole
South pole
(b) Effect of short-pithced coil on mmf wave
Fig. 5.27
a
a
c¢
c¢
b
b
a¢
a¢
c
c
b¢
b¢
a
c¢
c¢
b
b
a¢
a¢
c
c
b¢
b¢
a
Fig. 5.28
** From the Fourier series
F1p
2
=
p
p /2 - q sp /2
Ú
Fpole cos q dq
- p /2 + q sp /2
4
=
Fpole cos qsp/2
p
Basic Concepts in Rotating Machines
221
It is not surprising that the same result is obtained for Kp for the space mmf wave as for the generated emf
in a short-pitched coil.
In general when the winding is both distributed and short-pitched, the fundamental space mmf of phase a
as in Eq. (5.32) generalizes to
Fa1 =
where
Ê N ph (series) ˆ
4
Kw ÁË
˜¯ ia cosq
p
P
(5.34)
Kw = KbKp = winding factor
Like in the case of the induced emf, distribution of stator winding and short-pitching both help reducing
harmonics in the space mmf wave. In the analysis from now onward, it will be assumed that the space mmf
wave created by each phase of the stator winding when carrying current is purely sinusoidal. Also Kb and Kp
are defined in the same manner as for the induced emf.
When the phase a carries sinusoidal current
ia = 1m cos wt; (Im = maximum value of the phase current)
(5.35)
the mmf wave is given by
Ê N ph (series) ˆ
4
Kw ÁË
˜¯ Im cos wt cosq
p
P
= Fm cos wt cos q
Fal =
Fm =
where
where
(5.36)
Ê N ph (series) ˆ
4 2
Ê N ph (series) ˆ
4
Kw ÁË
Kw Á
˜¯ I
˜¯ Im =
Ë
P
p
P
p
(5.37)
I = Im / 2 = rms value of phase current.
As per Eq. (5.36), the mmf of one phase is a standing wave (pulsating wave) in space whose peak always
coincides with the phase axis while the peak amplitude varies sinusoidally with time. This is illustrated in
Fig. 5.29, where a half-cycle of pulsation is indicated.
MMF
Axis of phase 'a'
–p/2
wt = 2p/3
wt = p
Fig. 5.29
wt = 0
wt = p/3
0
p/2
3 p/2
q
222
Electric Machines
Current-sheet Concept
It was seen above that a distributed winding gives rise to a stepped mmf wave having a strong fundamental
component which will be considered in machine modelling while all the harmonic components will be
neglected ( justification for the same has been advanced). Now the kind of space distribution of current on the
stator (or rotor) has to be found which will produce a purely sinusoidally distributed mmf wave. An intuitive
answer is that it would be a sinusoidally distributed current.
Figure 5.30 shows a sinusoidally distributed current-sheet with a peak linear density of A A/m length of
the air-gap periphery. The current flows from one end of the stator (or rotor) to the other end parallel to the
stator axis. With respect to the angle reference shown, the current distribution is given by
i = A cos q
(5.38a)
i = A cos q
– p/2
0
3p/2
p
p/2
q
Cross-sectional view of current-sheet-developed
F (q )
Current-sheet
A
a
MMF wave
p/2
p
q
da
q
Fig. 5.30
The mmf at angle q is contributed by the current contained in angular spreads of p/2 on either side of
it. The current in the differential angle da at angle a is found by converting it to mechanical angle and
multiplying it by mean radius. Thus the differential current is
(A cos a)
D Ê2 ˆ
AD
d
a =
cos a da
2 ÁË P ˜¯
P
where D = mean air-gap diameter
The mmf at angle q is then given by
p / 2 +q
q
È
˘
1 Í AD
AD
F(q) =
cos a d a cos a d a ˙
˙
2Í P
2
-p / 2 + q
q
Î
˚
AD
sin q
=
P
Ú
Ú
(5.38b)
Basic Concepts in Rotating Machines
223
whose peak value is
AD
P
As per Eqs (5.38a) and (5.38b), the mmf wave associated with the sinusoidally distributed current-sheet is
also sinusoidal but displaced 90°(elect.) from the current-sheet. Also the peak value of the mmf wave equals
half the current contained in one loop of the current-sheet, i.e.
Fpeak =
1Ê 2
pDˆ
AD
A¥
= Fpeak
=
2 ÁË p
P ˜¯
P
(5.39)
EXAMPLE 5.7 Trace out the variations in mmf due to a belt of current-carrying conductor representing
one phase of a 2-pole, 3-phase winding. The belt may be assumed as a current sheet of uniform current
density. What is the peak amplitude of the mmf wave and also the rms amplitude of the fundamental wave.
Given: Total current in the phase belt = A amperes; phase spread s = 60°.
SOLUTION The mmf wave is trapezoidal as sketched in Fig. 5.31(a). Its peak amplitude is A/2 ampere-turns.
As the current in the phase belt is spread uniformly the phasor diagram of fundamental phasors is arc of 60° as shown
in Fig. 5.31(b). It follows from this diagram that
Kb =
where
chord AB
sin s / 2
=
arc AB
s /2
s = phase spread in rads =
Kb =
sin 30∞
= 0.827
p /3
RMS amplitude of fundamental mmf wave of one phase = 0.827
p ¥ 60∞
= p /3
180∞
A
= 0.4135 A
2
A/2
a
c¢
b
a¢
c
(a)
d
A
B
s/2 s/2
0
(b)
Fig. 5.31
5.5
ROTATING MAGNETIC FIELD
It was seen in Sec. 5.4 that the sinusoidal current in any phase of an ac winding produces a pulsating mmf wave
in space whose amplitude varies sinusoidally with time. The expression for the fundamental component of
this mmf is given in Eq. (5.36). The harmonics of the mmf wave are rendered inconsequential in a distributed
224
Electric Machines
winding and are still further reduced in amplitude if short-pitched coils are used. Therefore, the fundamental
of the mmf wave will only be considered in the machine model given here.
Consider now that the three phases of an ac winding are carrying balanced alternating currents.
ia = Im cos wt
ib = Im cos (wt – 120°)
ic = Im cos (wt – 240°)
(5.40)
Three pulsating mmf waves are now set up in the air-gap which have a time phase difference of 120° from
each other. These mmf’s are oriented in space along the magnetic axes of phases a, b and c as illustrated by
the concentrated coil in Fig. 5.32. Since the magnetic
Axis of phase b
axes are located 120° apart in space from each other, the
three mmf’s can be expressed mathematically as
Fa = Fm cos wt cos q
Fb = Fm cos (wt – 120°) cos (q – 120°)
Fc = Fm cos (wt – 240°) cos (q – 240°)
a
(5.41)
b¢
q
120°
wherein it has been considered that three mmf waves
differ progressively in time phase by 120°, i.e. 2 p/3 rad
elect. and are separated in space in space phase by 120°,
i.e. 2p /3 rad elect. The resulting mmf wave which is the
sum of the three pulsating mmf waves is
or
c¢
F = Fa + Fb + Fc
F (q, t) = Fm [cos wt cos q
+ cos(wt – 120°) cos(q – 120°)
+ cos(wt – 240°) cos(q – 240)]
120°
Axis of phase a
120°
c
b
a¢
Axis of phase c
Fig. 5.32
three phases
It simplifies trigonometrically to
3
1
Fm cos(q – wt) + Fm [cos (q + wt) + cos(q + wt – 240°) + cos(q + wt – 480°)]
2
2
Recognizing that
F(q, t) =
(5.42)
cos (q + wt – 480°) = cos (q + wt – 120°)
it is found that the terms within the capital bracket of Eq. (5.42) represent three unit phasors with a progressive
phase difference of 120° and therefore add up to zero. Hence, the resultant mmf is
3
F cos (q – wt)
(5.43)
2 m
It is found from Eq. (5.43) that the resultant mmf is distributed in both space and time. It indeed is a
travelling wave with sinusoidal space distribution whose space phase angle changes linearly with time as w t.
It, therefore, rotates in the air-gap at a constant speed of w rad (elect.)/s.
The peak value of the resultant mmf is
F (q, t) =
Fpeak =
3
Fm
2
(5.44a)
Basic Concepts in Rotating Machines
225
Using the value of Fm from Eq. (5.37)
Fpeak = 3 ¥
2 2
Ê N ph (series) ˆ
Kw Á
˜¯ I
Ë
p
P
(5.44b)
At wt = 0, i.e. when the a phase current has maximum positive value, it follows from Eq. (5.43) that
3
Fm cosq
2
It means that the mmf wave has its peak value (at q = 0) lying along the axis of phase a when it carries
maximum positive current. At wt = 2 p/3, the phase b (assumed lagging) has its positive current maximum,
so that the peak of the travelling mmf lies along the axis of b phase. By the same argument the peak of the
mmf will travel and coincide with the axis of c phase at wt = 4 p/3. It is, therefore, seen that the resultant mmf
travels from the axis of the leading phase to that of the lagging phase, i.e., from phase a towards phase b and
then phase c when the phase sequence of currents is abc (a leads b leads c). The direction of rotation of the
resultant mmf can be reversed by simply changing the phase sequence of currents.
The speed of the rotating mmf is
F (q, 0) =
2
wm = ÊÁ ˆ˜ w rad (mech.)/s
Ë P¯
or
2
2p n
=
¥ 2p f
P
60
120 f
rpm
(5.45)
P
This indeed is the rotor speed to induce emf of frequency f in the ac winding. Hence this is called the
synchronous speed and now onwards would be denoted by the symbol ns.
Conclusion: Whenever a balanced 3-phase winding with phases distributed in space so that the relative
space angle between them is 2p/3 rad (elect.), is fed with balanced 3-phase currents with relative time phase
difference of 2p /3 rad (elect.), the resultant mmf rotates in the air-gap at a speed of ws = 2pf elect. rad/s
where f is the frequency of currents in Hz. The direction of rotation of the mmf is from the leading phase axis
towards the lagging phase axis.
The above conclusion is easily generalized to q phases (q > 2), where the relative space-phase angle
between the phase winding is 2 p /q rad (elect.) and the relative time phase angle between currents is also
2 p/q rad (elect.) For a 2-phase winding the time and space phase angles should be p/2 rad or 90° (elect.).
or
n=
Physical Picture
Sinusoidally distributed mmf along space angle q can be regarded as a two dimensional space vector oriented
along its positive peak (towards south pole on the stator) having an amplitude equal to its peak value. For
example, the mmf of a phase a in Fig. 5.33(a) is represented by the vector Fa along the positive direction of
the axis of coil aa¢. Sinusoidally distributed mmf’s can be added by the vector method.
The mmf of phase a varies in the time and space as (Eq. (5.41))
Fa = Fm cos wt cos q
(5.46a)
It is a pulsating wave with a fixed axis (Fig. 5.29), i.e. the amplitude of the vector Fa oscillates sinusoidally
with time (the exciting current is ia = Im cos wt with fixed axis (axis of phase a). Trigonometrically, we can
226
Electric Machines
a
Fa (pulsating)
F–
w
F+
q
q
w
Fa
w
F+
w
F–
a
a¢
a¢
(a) Rotating component fields of a pulsating field
Axis of phase b
a
+Im
w
w
F+
Axis of Phase a
w
F+
Fa (= Fm)
F–
b¢
w
120°
b
– 1/2/m
a¢
Field of phase a
F–
Fb (= 1/2Fm)
Field of phase b
a
+/m
c¢
F–
120°
w
Fc (= 1/2Fm)
b¢
c¢
w
w
F+
Fr (= 3 Fm)
2
–1/2/m
b
c
– 1 Im
2
Axis of phase c
Field of phase c
a¢
Resultant field
(b) Location of component rotating fields of all the
three phases and the resultant field at time t = 0
/a
w
+/m
– 1 Im
2
/c
(c)
Fig. 5.33
/b
c
– 1 Im
2
Basic Concepts in Rotating Machines
227
write Eq. (5.46a) as
1
1
Fm cos (q – wt) + Fm cos (q + wt)
2
2
+ Fa = F + F
Fa =
or
(5.46b)
(5.46c)
It may be seen from Eq. (5.48b) that a pulsating mmf can be considered as the sum of two (sinusoidally
distributed) mmf’s of same amplitude (1/2)Fm which rotates at constant speed of w elect. rad/sec. in opposite
directions; F + of amplitude l/2Fm rotates in the positive direction of q and F - also of amplitude 1/2Fm
rotates in the negative direction of q. The resolution of a pulsating sinusoidally distributed mmf into two
sinusoidally distributed rotating mmf’s is illustrated in Fig. 5.33(a). As the two component mmf’s rotate, their
resultant ( Fa ) oscillates sinusoidally in time with its axis remaining fixed.
The mmf’s along the axes of coils bb¢ and cc¢ can be similarly resolved into two oppositely rotating mmfs
of equal amplitude. Consider the instant when the phase a current is maximum positive, i.e. + Im(ia = Im
1
cos wt), the phase b current is at that time – Im and is becoming less negative (ib = Im cos (wt – 120°)) and
2
1
the phase c current is – Im but is becoming more negative (ic = Im cos (wt – 240°)). This is also obvious
2
from the phasor diagram of Fig. 5.33(c). Instantaneous locations of the component rotating of all three phases
are shown in Fig. 5.33(b). It is immediately observed from this figure that the three F + are coincident and
are rotating in the positive direction at speed w adding up to a positively rotating field of amplitude 3/2 Fm.
The three F - are location at relative angles of 120° and are rotating in the negative direction at speed w
and thereby cancel themselves out (zero resultant field). It may be, therefore, concluded that the resultant
of three pulsating mmf’s of maximum amplitude Fm with 120° time phase difference having axes at relative
space angles of 120° is a single rotating field of strength 3/2Fm rotating at speed w elect. rad/s in the positive
direction.
If the reluctance of the iron path of flux is neglected, the sinusoidal rotating mmf wave creates a coincident
sinusoidal rotating B-wave in the air-gap with a peak amplitude of
Bpeak =
3m0 Fm
2g
where g is the length of the machine air-gap.
EXAMPLE 5.8 Consider a 3-phase ac machine with star connected stator (neutral not connected). It is
excited by 3-phase balanced currents of rms magnitude I. If one phase is disconnected while the other two
carry the same magnitude current (by adjustment of voltage across the two lines), determine the relative
peak magnitude of rotating mmf waves.
SOLUTION Let for single-phase rms current I, peak value of oscillating mmf be Fm. With only two phases carrying
currens, the phase current will be equal and opposite.
Fa1 = Fm cosq cos wt
Fa2 = –Fm cos(q – 120°) cos wt
Fnet = Fa1 + Fa2 = Fm[cosq cos wt – cos(q – 120°) cos wt]
= Fm(cos q – cos(q – 120°))cos wt
228
Electric Machines
= 2Fm[sin(q – 60°)sin (–60°)] cos wt
Fnet =
Fm
Fnet
3 Fm cos(q + 30°) cos wt
=
3
2
F+
Fm cos(q + 30° – wt)
F–
3
cos(q + 30° + wt)
2
+
=
+
F
F≠
≠
Forward Backward
+
Fa
120°
3
Fm
2
Vector diagram The vector diagram is drawn in Fig. 5.34.
Relative strength of rotating mmf =
30°
Fm
F2
Fig. 5.34
EXAMPLE 5.9 A 3-phase, 400 kVA, 50 Hz, star-connected alternator (synchronous generator) running
at 300 rpm is designed to develop 3300 V between terminals. The armature consists of 180 slots, each slot
having one coilside with eight conductors. Determine the peak value of the fundamental mmf in AT/pole
when the machine is delivering full-load current.
SOLUTION
120 f
120 ¥ 50
=
= 20
ns
300
400 ¥ 1000
= 70 A
IL = I P =
3 ¥ 3300
P=
Maximum value of the phase current, Im = 70 2 A
Slot angle, g =
SPP, m =
Kb =
Turns/phase (series) =
Fm =
=
Fpeak =
180∞ ¥ 20
= 20°
180
180
=3
3 ¥ 20
sin mg / 2
= 0.96
m sin g / 2
180 ¥ 8
= 240
2¥3
Ê N ph (series) ˆ
4
Kb Á
˜¯ Im
Ë
p
P
4
Ê 240 ˆ
¥ 70 2 = 1452 AT /pole /phase
¥ 0.96 ¥ Á
Ë 20 ˜¯
p
3
F = 2178 AT /pole
2 m
EXAMPLE 5.10 A 3-phase, 4-pole, 50 Hz synchronous machine has its rotor winding with 364 distributed
conductors having a winding factor Kb = 0.957. The machine has an air-gap of 0.8 mm. It is desired to have
a peak air-gap flux density of 1.65 T. Rotor length = 1.02 m, rotor diameter = 41 cm.
Basic Concepts in Rotating Machines
229
(a) Calculate the field current required and flux/pole.
(b) The armature has 3 full-pitch 11-turn coils/pole pair per phase. Calculate the open-circuit phase
voltage and the line voltage of the phases are connected in star.
(c) It is desired to have an open-circuit rms 3/4th of that found in part (b). What should be the field
current?
SOLUTION As the field has a distributed winding we can use the result for armature winding except that the field
current is a fixed value i.e. 2 will not be there.
(a) For the field (on fundamental basis)
4 È Kb N field (series) I f ˘
Í
˙ ; Eq. (5.31)
P
pÎ
˚
= Fpeak /g
= m0 Hpeak
Fpeak =
Hpeak
Bpeak
or
È Kb N field (series) I f ˘
Í
˙
P
Î
˚
Bpeak =
4m0
pg
1.65 =
4 ¥ 4p ¥ 10- 7 È 0.957 ¥ (364 / 2) ¥ I f ˘
Í
˙
4
˚
p ¥ 0.8 ¥ 10- 2 Î
If =
(ii)
(iii)
(iv)
1.65 p ¥ 8 ¥ 10- 3 ¥ 4 ¥ 2
4 ¥ 4 p ¥ 10- 7 ¥ 0.957 ¥ 364
= 18.95 A
4
Flux/pole, F =
Bpeak lr
P
(b)
(i)
(Eq (5.7))
(v)
Substituting values
F = 1.65 ¥ 1.02 ¥ 0.41/2 = 0.345 Wb
P
= 66
Nph (series) = 3 ¥ 11 ¥
2
60∞
= 20°, m = 3
Slot angle g =
3
sin 3 ¥ 10∞
= 0.96
Breadth factor, Kb =
3 sin 10∞
Eph =
2 p Kb f Nph (series) F
Eph =
2 p ¥ 0.96 ¥ 50 ¥ 66 ¥ 0.345 = 4855 V
Substituting values
Eline = 4853 3 = 8409 V
(c) Open-circuit voltage required is 3/4th of Eline as above. On linear basis
If (new) = 0.75 ¥ 18.95 = 14.21 A
EXAMPLE 5.11 A 3-Phase 50 kW, 4-pole, 50 Hz induction motor has a winding (ac) designed for delta
connection. The winding has 24 conductors per slot arranged in 60 slots. The rms value of the line current
is 48 A. Find the fundamental of the mmf wave of phase-A when the current is passing through its maximum
value. What is the speed and peak value of the resultant mmf/pole?
230
Electric Machines
SOLUTION
180∞ ¥ 4
= 12°
60
60
=5
SPP =
4¥3
g=
5 ¥ 12∞
2
Kb =
= 0.957
12∞
5 sin
2
IL
48
IP =
A
=
3
3
sin
IP, max =
Turns/phase =
48 2
3
= 39.2 A
60 ¥ 24
= 240
2¥3
Ê N ph (series) ˆ
4
Kb Á
˜¯ IP, max
Ë
p
P
4
Ê 240 ˆ
¥ 39.2
=
¥ 0.957 ¥ Á
p
Ë 4 ˜¯
= 2866 AT/pole
3
Fpeak = Fm = 4299 AT/pole
2
120 f
120 ¥ 50
Speed of rotation of the resultant mmf =
=
P
4
= 1500 rpm
Fm (phase a) =
5.6 TORQUE IN ROUND ROTOR MACHINE
When the stator and rotor windings of a machine both carry
currents, they produce their own magnetic fields along their
respective axes which are sinusoidally distributed along the
air-gap. Torque results from the tendency of these two fields
to align themselves. The flux components set up by the stator
and rotor currents cross the air-gap twice and complete their
circuits via the stator and rotor iron. These components fields
cause the appearance of north and south poles on stator and
rotor surfaces; the field axes being along north-south and out
of the north pole. This is illustrated in Fig. 5.35(a) for a 2-pole
structure. The torque tending to align the two fields is produced
only if the two fields have the same number of poles and are
stationary with respect to each other. Two relatively rotating
fields will produce alternating torque as they cross each other
so that the average torque is zero. All varieties of electric
machines (synchronous, induction and dc) are therefore
devised to produce interacting fields with zero relative velocity.
Stator (1)
Rotor (2)
T
S
N
Axis of the stator field
(peak value F1)
S
a
N
Axis of the rotor field
(peak value F2)
(a) Stator and rotor fields
a
d
g
F1
a
a
g
d
Fr
F2
(b) Vector sum of stator and rotor fields
Fig. 5.35
Basic Concepts in Rotating Machines
231
Certain underlying assumptions are made at this stage:
1. Stator and rotor mmf’s are sinusoidal space waves; this is sufficiently ensured by distributed windings.
2. Rotor is cylindrical (nonsalient pole) so that the air-gap is uniform throughout.
3. The air-gap is narrow so that flux established in it is radial (negligible tangential component) and
further the flux density does not vary significantly* along a radial path in the gap. As a result, the field
intensity H along any radial path is constant in the air-gap. The mmf across the air-gap at any space
point is
Fair-gap = Hg
where g is the radial air-gap length.
4. Reluctance of the iron path of flux is assumed negligible. As a consequence of assumptions 1–3,
a sinusoidal space mmf wave produces a sinusoidal flux density wave in space in phase with it.
5. Most of the resultant flux is common to both stator and rotor windings, i.e. it is mutual flux. The
leakage flux** linking either winding produces the leakage inductance as in a transformer. These
affect only the net voltages applied to the ideal machine.
Let F1 F2 be the peak values of the spatial sinusoidal mmf waves of the stator and rotor respectively as
shown in Fig. 5.35(a) for a 2-pole machine; the angle between their respective positive peaks being denoted
by a. As stated earlier, these mmf’s can be represented as space vectors with magnitudes equal to their
peak values and angles corresponding to their positive peaks. The resultant space mmf (which will also be
sinusoidal being the sum of sinusoids) can be obtained by the vector summation as shown in Fig. 5.35(b).
Using trigonometric relations, it easily follows that the peak value of the resultant mmf is
F 2r = F 12 + F22 + 2F1F2 cos a
(5.47a)
Since the reluctance of the iron path is negligible, the peak value of the resultant field intensity is
Fr
g
From Eq. 4.18(b) it is known that the coenergy density is
Hr =
1
m0 H2
2
The average value of the coenergy density over the air-gap volume is
1
m0 (average value of H 2)
2
For a sinusoidal distribution,
1 2
Hr
2
1
Average coenergy density = m0 H 2r
4
Volume of air-gap = p Dlg
Average value of H 2 =
\
* This is so because the cylindrical area presented to gap flux does not vary appreciably with radius.
** Refer Sec. 5.7.
(5.47b)
232
Electric Machines
where D is the mean diameter at air-gap and l = rotor length = stator length. The total coenergy of the field
is then
2
ÊF ˆ
p
p
W f¢ = m0 H 2r Dlg = m0 Á r ˜ Dlg
4
4 Ë g¯
=
m0p Dl 2
Fr
4g
(5.48)
m0p Dl 2
(F 1 + F 22 + 2F1 F2 cos a)
4g
(5.49)
Substituting for Fr from Eq. (5.47a),
W f¢ =
The torque developed is given by
T= +
∂W f¢
∂a
= -
m0p Dl
F1F2 sin a
2g
(5.50)
For a P-pole machine
Ê P ˆ m0p Dl
F1F2 sin a
T = -Á ˜
Ë 2 ¯ 2g
(5.51)
From the torque expression (5.51) it is seen that the torque developed is proportional to the peak values of
the stator and rotor mmfs and is proportional to the sine of the angle between the axes of the two fields. The
negative sign in Eq. (5.52) indicates that the torque acts in a direction to reduce a, i.e. to align the two fields.
Obviously, equal and opposite torques will act on the stator and rotor.
With reference to the geometry of Fig. 5.34(b) it is found that
and
F1 sin a = Fr sin d
F2 sin a = Fr sin g
(5.52)
(5.53)
The torque expression of Eq. (5.51) can therefore be expressed in two more alternative forms,
Ê P ˆ m0p Dl
F1Fr sin g
T = -Á ˜
Ë 2 ¯ 2g
(5.54)
Ê P ˆ m0p Dl
F2 Fr sin d
T = -Á ˜
Ë 2 ¯ 2g
(5.55)
The torque in the expressions of Eqs (5.54) and (5.55) can be immediately imagined to be the result of
the interaction of the stator and the resultant mmf’s or the rotor and the resultant mmf’s and is proportional
to the sine of the angle between the interacting mmf’s. The expression of Eq. (5.55) is generally preferred in
simplified machine models.
Since m0 Fr /g = Br ; (peak value of resultant (or air-gap) flux density) the expression of Eq. (5.55) can be
written as
Ê P ˆ p Dl
T = -Á ˜
F2 Br sin d
Ë 2¯ 2
(5.56)
This expression reveals a design viewpoint; the torque of electromagnetic origin is limited by the saturation
value of flux density (Br) in the magnetic material and maximum permissible mmf (F2). The maximum value
of F2 is dictated by the maximum allowable winding current without exceeding the specified temperature rise.
Basic Concepts in Rotating Machines
233
An alternative and more useful form of the torque expression is in terms of the resultant flux/pole, Fr,
rather than the peak flux density Br Now
Fr = Bav (over a pole) ¥ (pole area)
2Br Dl
Ê 2 ˆ Ê p Dl ˆ
= Á Br ˜ Á
; same as Eq. (5.7)
=
˜
P
Ëp ¯Ë P ¯
D = 2r
(5.57)
Substituting for Br from Eq. (5.57) in Eq. (5.55),
2
p Ê Pˆ
F F sin d
2 ÁË 2 ˜¯ r 2
where Fr = resultant (i.e. air-gap) flux/pole due to superimposition of the stator and rotor mmf’s
Equation (5.57) can be written as
T= -
Ê 2 Dl m0 ˆ
¥
Fr = Á
F = P Fr
g ˜¯ r
Ë P
(5.58)
(5.59)
Ê 2 Dl m0 ˆ
¥
P= Á
= effective permeance per pole
g ˜¯
Ë P
where
The effective permeance/pole relates the peak value of sinusoidal mmf wave and the flux/pole created by
it. It is a constant quantity* so long as the permeability of iron is assumed infinite, otherwise it is a function
of Fr (i.e. P (Fr)) which decreases as Fr increases due to saturation of iron.
Alternative Derivation
Figure 5.36(a) shows the stator mmf wave and the corresponding air-gap flux density wave. The rotor mmf
wave makes an angle of a with the stator mmf wave. The rotor current wave which is the cause of rotor mmf
leads it by an angle of 90° as shown in Fig. 5.36(b). The force (and torque) is produced by the interaction
of B1-wave and A2-wave as per the Bli rule. It may be seen that positive B1 and positive A2 produce negative
force (opposite to the positive direction of q). An angular element dq of the machine located at an angle q
from the origin produces a torque
Substituting values
dt = –rBldi
dT = -
where
1
D(B1 cos q) l
2
D ˆ
Ê
ÁË - A2 sin (q - a ) P dq ˜¯
di
di = elemental current in differential periphery
P
D
dqm =
dq
2
P
1 D 2l
B A cos q sin (q – a) dq
=
2 P 1 2
Differential periphery =
* So long as the machine has uniform air-gap (round rotor), the permeance/pole offered by it is independent of the
spatial orientation of the axis of the mmf wave. In the salient-pole construction two different permeances—one
along the axis of the projecting poles and the other at 90° elect. to it—will be accounted for in Sec 8.11.
234 Electric Machines
F1
B1 cos q
B1
– p/2
p/2
0
p
3p/2
q
(a) Stator mmf and gap flux density
a
– A2 sin (q – a)
F2
A2
F2 cos ( q – a)
q
0
(b) Rotor mmf and current
Fig. 5.36
The torque for one pole-pair is obtained by integrating the above equation from q = 0 to q = 2p. It yields
1
p D2lB1 A2 sin a
2P
B1 = m0 F1/g
(see Eq. (5.39))
A2 = F2P/D
T= -
Now
It then follows
T= -
m 0p
Dl F1F2 sin a
2g
(5.60)
(5.61)
a result already established in Eq. (5.50).
5.7 OPERATION OF BASIC MACHINE TYPES
An elementary explanation of the torque-production process of basic machine types through the general
torque expression of Eq. (5.58) will be given in this section.
Synchronous Machine
Figure 5.37 shows a synchronous machine with a round rotor. The rotor is initially stationary with fixed northsouth poles created by dc excitation. Let the 3-phase winding of the stator be connected to a 3-phase supply
of fixed voltage V (line) and fixed frequency f (this is known as the infinite bus). As a result, 3-phase currents
flow in the stator winding creating a rotating magnetic field rotating at synchronous speed ns (=120f/P) in
the counter-clockwise direction (say). Since the rotor is stationary and cannot pick up speed instantaneously
Basic Concepts in Rotating Machines
235
(inertia effect), the two fields move relative to each other resulting in zero average torque. As such the motor
is non-self-starting.
Consider now that the rotor is run by auxiliary means to a
3 phase supply (V, f)
(infinite bus)
speed close to synchronous in the direction of rotation of the
stator field. The two fields now have the opportunity of locking
into each other or, in other words, the rotor pulls into step with
S
ns
T
N
the stator field and then on runs at exactly synchronous speed.
It is easily seen from Fig. 5.37 that the electromagnetic torque
ns
developed (T ) acts on the rotor in the direction of rotation of
TL
S
rotor and balances the load torque TL. The mechanical power
N
therefore flows to the load (motoring action) and, by the principle
a
F1 (Stator)
of conservation of energy, an equal amount of electrical power
d
F
(Rotor)
(plus losses in the device) are drawn from the electric supply.
2
ns
It is also seen from Fig. 5.37 that for a given TL, the rotor field
Fr (resultant)
lags behind the stator field by an angle a or behind the resultant
Fig. 5.37 Torque production in synchronous
field by an angle d. The torque developed by the synchronous
machine (motoring)
motor is given by the expression of Eq. (5.58), i.e.
2
T=
p Ê Pˆ
F F sin d
2 ÁË 2 ˜¯ r 2
(5.62)
It may be seen that the negative sign has been deleted from the torque expression with the understanding
that the torque acts in a direction to align the fields (it is indicated by an arrow sign in Fig. 5.37).
If the stator winding resistance and leakage reactance are assumed negligible (a fair assumption), the
induced emf of the stator winding balances the terminal voltage, i.e.
Vª
3 ¥ 4.44 Kw fFr Nph (series)
(5.63)
For a fixed terminal voltage, therefore, the resultant flux/pole is almost constant, independent of the shaft
load. F2, the peak of rotor mmf wave being dependent upon the rotor current (dc), is constant for fixed
excitation. Equation (5.62) under conditions of constant terminal voltage and constant rotor excitation can
therefore be written as
T = K sin d
(5.64)
where d is positive when the rotor field lags behind the resultant field and would be negative otherwise. The
angle d is known as the torque angle or power angle.
The plot of electromagnetic torque developed by the synchronous machine is shown in Fig. 5.38. The
machine operates at fixed d for a given mechanical load torque (say d1 for TL1) and runs at synchronous speed.
As the load torque is increased to TL2 > TL1, the rotor decelerates and the angle d increases to a new steady
value d 2 > d1 as shown in Fig. 5.38. Of course, the machine settles at the new operating angle in an oscillatory
manner and its steady speed is once again synchronous. The field coupling of the stator-rotor acts like a spring
coupling and combined with rotor inertia, the system is oscillatory in nature. However, these oscillations die
out after every disturbance because of the damping contributed by the mechanical and electrical dissipative
effects that are present in the machine.
It is also observed from Fig. 5.38 that the maximum torque developed by the motoring machine is at
d = 90° and is called the pull-out torque (or pull-out power); power being proportional to torque as the
236
Electric Machines
machine speed is synchronous, independent of load. If the
Torque (power)
motor is loaded with torque (power) more than Tpull-out’ the
developed torque reduces (Fig. 5.38) the rotor lag angle
Motoring
Tpull-out
increases monotonically till the rotor-stator field-bond
TL2
snaps, i.e. the rotor falls out of step. The machine will finally
TL1
come to a stop and must, as a precautionary measure, be
disconnected from the supply much before that. It is easily
–180°
–90°
seen from the expression of Eq. (5.62) that the pull-out
180° d
90°
d2
torque can be increased by increasing the stator terminal
d1
voltage, Fr increases with terminal voltage (Eq. (5.63)) and/
or rotor field excitation.
Generating
Tpull-out
With negative d, i.e. rotor field leading the resultant field
in the direction of rotation of the rotor, the electromagnetic Fig. 5.38 Torque-angle (T – d) characteristic of
synchronous machine
torque as seen from Fig. 5.39 is now developed in a
direction opposite to that of the rotor rotation and must be
V, f
balanced by an external mechanical torque TPM (provided
by a prime-mover) for the rotor to run at synchronous speed
F2 (Rotor)
maintaining the locking of the rotor and stator fields. The
ns
mechanical power now flows into the rotor and the electrical
d
T
power flows out of the stator to the infinite bus. The rotor
Fr (resultant)
readjusts its angle of lead such that the electrical output
TPM
equals the mechanical input minus losses. If the mechanical
input is more than the maximum electrical power developed
F1 (Stator)
(corresponding to generator pull-out torque, d = 90°), the
rotor accelerates and falls out of step, i.e. the synchronism Fig. 5.39 Torque production in synchronous
machine (generating)
between the rotor and stator fields is lost.
To summarize, a synchronous machine has a synchronous locking between the stator and rotor fields
with the rotor field lagging the resultant air-gap field in motoring operation and leading the resultant field in
generating operation. The electromagnetic torque developed is a sine function of angle d, between the rotor
field and the resultant field; as a result the machine falls out of step and loses synchronism if conditions are
created for d to increase beyond ±90°. Within this range the machine operates at synchronous speed under
varying load conditions. Further, the machine is non-self-starting as a motor.
EXAMPLE 5.12 In a certain electric machine F2 = 850 AT, F1 = 400 AT, a = 123.6° and P (permeance/
pole) = 1.408 ¥ 10 –4 Wb/ AT. Calculate the value of the resultant air-gap flux/pole.
SOLUTION
Fr = (F12 + F22 + 2F1F2 cos a)1/2
= [(400)2 + (850)2 + 2 ¥ 400 ¥ 850 cos 123.6°]1/2 = 711.5 AT
Fr = P Fr = 1.408 ¥ 10 –4 ¥ 711.5
= 0.1 Wb
EXAMPLE 5.13 A 50 Hz, 400 V, 4-pole cylindrical synchronous generator has 36 slots, two-layer winding
with full-pitch coils of 8 turns each. The mean air-gap diameter is 0.16 m, axial length 0.12 m and a uniform
air-gap of 2 mm. Calculate the value of the resultant AT/pole and the peak air-gap flux density. The machine
Basic Concepts in Rotating Machines
237
is developing an electromagnetic torque of 60 Nm as a generator at a torque angle 26°. What should be
the rotor AT/pole? What is the stator AT and the angle it makes with the resultant AT? Also find the stator
current.
SOLUTION
Nph (series) =
36 ¥ 8 ¥ 2
= 96
2¥3
4 ¥ 180∞
= 20°
36
36
=3
m=
4¥3
g=
Kb =
V
400
3
Fr
D
sin (3 ¥ 20∞ / 2)
= 0.96
3 ¥ sin (20∞ / 2)
ª E = 4.44 Kd f Fr Nph
= 4.44 ¥ 0.96 ¥ 50 ¥ Fr ¥ 96
= 0.0113 Wb/pole
= 0.16 m, l = 0.12 m
p ¥ 0.16 ¥ 0.12
= 0.0151m2
4
0.0113
Br (av) =
= 0.749 T
0.0151
p
¥ 0.749 = 1.18 T
Br (peak) =
2
g Br (peak )
2 ¥ 10- 3 ¥ 1.18
=
Fr =
m0
4 p ¥ 10- 7
Pole area =
= 1878 AT/pole (peak)
Torque developed, T =
60 =
p Ê Pˆ
Fr F2 sin d
2 ÁË 2 ˜¯
p Ê 4ˆ
¥
2 ÁË 2 ˜¯
2
¥ 0.0113 F2 sin 26°
F2 = 1928 AT/pole (peak)
Mmf phasor diagram is drawn in Fig. 5.40 with F2 leading Fr by d = 26°. This is the case in a generating machine.
From the phasor diagram we find
or
F 12 = (1928)2 + (1878)2 – 2 ¥ 1928 ¥ 1878 sin 26°
or
F1 = 858 AT/pole (peak)
2
(858) + (1878) - (1928)
2 ¥ 858 ¥ 1878
y = 80∞
cos y =
or
2
y
(1928)
2
F1 lags Fr by 80°
F1 =
F1
F2
3 4 2 Ê N ph (series) ˆ
¥
kw Á
˜¯ Ia
2
p
P
Ë
26°
Fig. 5.40
Fr
(1878)
238 Electric Machines
Kw = Kb = 0.96; Nph (series) = 96
3 4 2
96
¥
¥ Ia
¥ 0.96 ¥
2
p
4
Ia = 13.8 A
858 =
or
EXAMPLE 5.14
A 3f, 4-pole, 50 Hz synchronous machine has the data as given below.
Kw (rotor) = 0.976
Air-gap = 1.5 mm (cylindrical rotor)
Mean air-gap diameter = 29 cm
Axial length = 35 cm
Rotor winding turns = 746
Stator winding SPP = 4; conductor/slot = 20
Field current = 20 A
(a) The machine is being used in motoring mode with Br = 1.6 T. Determine
(i) F2, peak rotor AT
(ii) Maximum torque
(iii) Electrical input at maximum torque
(b) If the machine is used as generator with same field current, determine its open-circuit (no-load)
voltage.
SOLUTION
Motoring mode
Ê N pole (series) ˆ
4
Kw (rotor) Á
˜¯ If
Ë
P
p
(a)
F2 =
(b)
4
746
¥ 0.976 ¥
¥ 20
p
4
= 4635 AT
Br = 1.6 T (given)
=
Then
Ê P ˆ Ê p DL ˆ
F2 Br
Tmax = Á ˜ Á
Ë 2 ¯ Ë 2 ˜¯
4 p ¥ 0.29 ¥ 0.35
¥
¥ 4635 ¥ 1.6
2
2
= 2364
=
Synchronous speed (motor speed)
2w
4p f
4p ¥ 50
=
=
P
P
4
= 50 p rad (mech)/sec
Electrical power input, Pin = Tmax ¥ wm
= 2632 ¥ 50p W
= 371 kW
wm =
The machine is assumed loss-less, so
Pin = Pin, max = 371 kW
Basic Concepts in Rotating Machines
239
Generating Mode
Open Circuit Voltage The armature is not carrying any current, i.e. F1 = 0, F2 = 4630 AT as calculated above.
SPP =
g=
S
=4
3P
180∞P
180
=
= 15°
S
12
Kw = Kb =
sin 30∞
= 0.958
3 sin 15∞ / 2
Ê 2 Dl m0 ˆ 2
¥
Fr = Á
F
g ˜¯
Ë P
=
2 ¥ 0.29 ¥ 0.35 ¥ 4p ¥ 10- 7
4 ¥ 1.5 ¥ 10- 3
¥ 4635
= 0.197 Wb
20 ¥ 4 ¥ 4
= 160
2
= 4.44 Kb f Nph F
= 4.44 ¥ 0.958 ¥ 50 ¥ 160 ¥ 0.197
= 6704 V
= 11610 V or 11.61 kV
Nph =
Eph
Eline
Induction Machine
This machine has not been introduced so far. Consider a cylindrical rotor machine with both the stator and
rotor wound for three phases and identical number of poles as shown in Fig. 5.41. Assume initially the
rotor winding to be open-circuited and let the stator be
F1
connected to an infinite bus (V, f ). The stator currents
set up a rotating magnetic field in the air-gap which
runs at synchronous speed inducing emf in the stator
ns
winding which balances the terminal voltage under
a
ns
the assumption that the stator resistance and leakage
T
A
Fr
reactance are negligible. Also the rotating field induces
emf in the rotor winding but no rotor current flows
V, f
B
d n
because the rotor is open-circuited. The frequency of
TL
C
rotor emf is of course f. Since the rotor mmf F2 = 0,
b
c
no torque is developed and the rotor continues to be
stationary. The machine acts merely as a transformer
(ns – n) wrt rotor or ns wrt stator
where the stator (primary) and rotor (secondary) have
emfs of the same frequency induced in them by the
F2
rotating magnetic flux rather than by a stationary timeFig. 5.41 Illustrating the principle of induction machine
varying flux as in an ordinary transformer.
Let the rotor be now held stationary (blocked from rotation) and the rotor winding be short-circuited.
The rotor now carries 3-phase currents creating the mmf F2 rotating in the same direction and with the same
speed as the stator field. F2 causes reaction currents to flow into the stator from the busbar ( just as in an
240
Electric Machines
ordinary transformer) such that the flux/pole Fr of the resultant flux density wave (rotating in the air-gap at
synchronous speed) induces a stator emf to just balance the terminal voltage. Obviously Fr must be the same
as when the rotor was open-circuited. In fact, Fr will remain constant independent of the operating conditions
created by load on the motor. The interaction of Fr and F2, which are stationary with respect to each other,
creates the torque tending to move the rotor in the direction of Fr or the stator field F1. The induction motor
is therefore a self-starting device as different from the synchronous motor.
Let the short-circuited rotor be now permitted to rotate. It runs in the direction of the stator field and
acquires a steady speed of n. Obviously n < ns, because if n = ns, the relative speed between the stator field
and rotor winding will be zero and therefore the induced emfs and rotor currents will be zero and hence no
torque is developed. The rotor thus cannot reach the synchronous speed ns and hence cannot exceed ns. With
the rotor running at n, the relative speed of the stator field with respect to rotor conductors is (ns – n) in the
direction of ns. The frequency of induced emfs (and currents) in the rotor is therefore
Ê ns - n ˆ Ê ns P ˆ
(ns - n) P
= Á
Á
˜
Ë ns ˜¯ Ë 120 ¯
120
= sf
f2 =
(5.65)
ns - n
= slip of the rotor
(5.66)
ns
The slip s is the per unit speed (with respect to synchronous speed) at which the rotor slips behind the
stator field. The rotor frequency f2 = sf is called the slip frequency. From Eq. (5.66), the rotor speed is
where
s=
n = (l – s)ns
(5.67)
The slip frequency currents in the rotor winding produces a rotor field rotating with respect to rotor in the
same direction as the stator field at a speed of
120(ns - n) f
120sf
=
= (ns – n)
(5.68)
ns P
P
Since the rotor is running at a speed n and the rotor field at (ns – n ) with respect to the rotor in the same
direction, the net speed of the rotor field as seen from the stator (ground reference) is
n + (ns – n) = ns
i.e., the same as the stator field. Thus the reaction field F2 of the rotor is always stationary with respect to the
stator field F1 or the resultant field Fr (with flux Fr per pole). Since the rotor mmf F2 is proportional to the
rotor current I2 and the resultant flux/pole Fr is fixed by terminal voltage independent of operating conditions,
the induction motor torque is given by (see Eq. (5.58))
T = KI2 sin d
(5.69)
It is observed here that the torque is produced by the induction motor at any mechanical speed other than
synchronous; such a torque is called the asynchronous torque.
The angle d by which F2 lags behind Fr’ the resultant mmf needs to be known. Before proceeding to
determine d, it must be observed that shorting the rotor winding is equivalent to shorting all the winding
conductors individually. As a result the rotor does not necessarily have to be properly wound; it may be
constructed of conducting bars placed in the rotor slots slightly skewed and shorted by conducting end-rings
on each side of the rotor. Such a rotor is called the squirrel-cage rotor; the conducting cage is separately
Basic Concepts in Rotating Machines
241
illustrated in Fig. 5.42 The squirrel-cage rotor has a
cheap and rugged construction and is adopted in a
vast majority of induction motor applications. The
machine with a properly wound rotor is called the
End ring
wound-rotor induction motor and is provided with
End ring
three slip-rings which provide the facility of adding
external resistance in the rotor winding before
Rotor bars
shorting these. Such motors are used in on-load
(slightly skewed)
starting situations (see Ch. 9).
Normally the full-load slip of a squirrel-cage
Fig. 5.42 Squirrel-cage rotor
induction motor is small 3-10%. Consequently the
rotor impedance is mainly resistive, the rotor leakage reactance being proportional to f2 = sf is negligible.
Furthermore, the rotor induced emf is proportional to the rotor-slip as Fr is fixed and rotates at speed ns – n
= sns with respect to rotor. This results in the rotor current being very nearly in phase with the rotor emf and
proportional to the rotor slip. This conclusion would obviously apply to individual rotor conductors as well.
Figure 5.43 shows the resultant flux density wave Br gliding past the rotor conductors at speed (ns – n)
= sns in a developed diagram. The induced currents in the shorted rotor conductors are sinusoidally distributed–
the distribution moves at speed (ns – n) with respect to the rotor in synchronism with the Br-wave. Further,
because the rotor conductors are assumed resistive, i.e. currents in them are in phase with their respective
emf’s, the rotor current distribution is therefore in space phase with Br - wave. The sinusoidal rotor current
distribution produces a sinusoidal rotor mmf wave F2 which lags 90° behind rotor current distribution or 90°
behind Br - wave. It is, therefore, concluded that for small values of slip, the angle d in the induction motor
is 90°. Hence,
T = kI2 (for small slip)
(5.70)
Br (resultant flux density wave)
(ns – n)
F2 (rotor mmf wave)
(ns – n)
d = 90°
Fig. 5.43
Since rotor emf is linearly proportional to slip*, so is the rotor current for mainly a resistive rotor at small
values of slip. Hence, the torque developed in the induction motor is a linearly increasing function of slip for
small value of slip, being zero for s = 0, i.e. at synchronous speed.
* Since Fr, is practically constant, independent of operating conditions, the rotor emf is proportional to the relative
speed between the resultant field and the rotor. i.e.
(ns – n) = sns
242 Electric Machines
As slip increases further, the leakage reactance of the rotor can no longer be neglected. Its value at slip s
is sX2, where X2 is the rotor leakage reactance per phase at frequency f, i.e. when the rotor is at a stand-still.
The rotor current now lags the rotor induced emf by
q = tan–1
sX 2
R2
where R2 is the rotor resistance per phase.
Since the currents in the rotor conductors lag the induced emf’s by angle q, the rotor conductor current
distribution and therefore the rotor mmf F2 shifts to the left in Fig. 5.41 by an angle q, so that
d = 90° + q
(5.71)
Torque (in % full-load torque)
It means sin d < 1. Further, since the rotor impedance
is increasing with s, rotor current is less than proportional
Break-down torque
to slip. These two factors cause the motor torque to pass
250
through a maximum value and then begin to decrease
200
gradually as s is continuously increased.
The nature of the complete torque-slip characteristic of
150
the induction motor is exhibited in Fig. 5.44. The maximum
torque is known as the break-down torque. The motor would
100
come to rest if loaded for short time with torque load larger
than the breakedown value.
50
As already mentioned, the slip of an induction motor is
3-10% at full-load. Therefore, it is substantially a constant
0
speed drive unlike the synchronous motor which runs at
1.0
0.8
0.6
0.4
0.2
0
Slip
constant speed independent of load.
Fig. 5.44 Torque-slip characteristic of induction
Generating action results if an induction machine is
motor
run at negative slip or at speed n > ns, i.e. at a speed above
synchronous.
EXAMPLE 5.15 A 4-pole synchronous generator driven at 1500 rpm feeds a 6-pole induction motor
which is loaded to run at a slip of 5%. What is the motor speed?
SOLUTION
Frequency of the synchronous generator,
f=
4 ¥ 1500
= 50 Hz
120
Synchronous speed of the induction motor,
120 ¥ 50
= 1000 rpm
6
Motor slip, s = 0.05
Motor speed = (1 – s)ns
ns =
= 0.95 ¥ 1000 = 950 rpm
EXAMPLE 5.16 A 6-pole, 50-Hz wound-rotor induction motor when supplied at the rated voltage and
frequency with slip-rings open-circuited, developed a voltage of 100 V between any two rings. Under the
same conditions its rotor is now driven by external means at
Basic Concepts in Rotating Machines
243
(a) 1000 rpm opposite to the direction of rotation of stator field, and
(b) 1500 rpm in the direction of rotation of stator field.
Find the voltage available between slip-rings and its frequency in each of these cases.
SOLUTION
(a)
Synchronous speed,
120 ¥ 50
= 1000 rpm
6
n = –1000 rpm
ns =
ns - n
1000 - (-1000)
=
=2
ns
1000
Slip frequency, sf = 2 ¥ 50 = 100 Hz
s=
The given open-circuited voltage v2 = 100 V corresponds to a slip of s = 1, the motor being stationary with the
rotor open-circuited. Since the induced emf in ac winding is proportional to frequency, the rotor induced emf at
slip s (frequency sf ) is sv2. Therefore,
Slip-ring voltage = sv2 = 2 ¥ 100 = 200 V
n = 1500 rpm
(b)
1000 - 1500
= –0.5
1000
Slip frequency = 0.5 ¥ 50 = 25 Hz
s=
It may be seen that the negative sign has been dropped, which merely implies a reversal in the phase angle of
the voltage.
Slip-ring voltage = 0.5 ¥ 100 = 50 V
It is found that the induction machine can be regarded as a generalized transformer with rotor voltage and
frequency both being proportional to slip.
EXAMPLE 5.17 The stator of the induction motor of Ex. 5.16 is fed at the rated voltage and frequency
while its slip-rings are connected to a 25-Hz supply.
(a) Will there be a starting torque?
(b) At what speed will steady operation result?
(c) At what speed will steady operation result if the rotor is also fed with a 50-Hz supply?
SOLUTION
(a) Under stationary conditions of the rotor, no torque will be developed as stator and rotor fields will rotate relative
to each other, i.e. no starting torque.
(b) For steady operation, the stator and rotor fields must be stationary relative to each other.
Speed of the stator field (with respect to the stator surface)
120 ¥ 50
= 1000 rpm
6
Speed of the rotor field (with respect to the rotor surface)
=
120 ¥ 25
= 500 rpm
6
Steady synchronous operation will result when the rotor is run at 500 rpm in the same direction as the stator field.
=
244
Electric Machines
(c) Speed of the rotor field (with respect to the rotor surface)
= 1000 rpm
(i) If the rotor field rotates in the same direction as the stator, steady (synchronous) operation is only possible at
zero speed. At any other speed of the two fields will have relative motion and will produce zero torque.
(ii) If the rotor field rotates opposite to the stator field, the steady (synchronous) operation will result when the
rotor moves at 2000 rpm in the direction of the stator field; it is only at this speed of the rotor that the two
fields are relatively stationary.
Note: The induction motor fed as above from both stator and rotor sides operates in the synchronous mode as different
from the induction mode.
EXAMPLE 5.18
A 3-phase, 50 Hz induction motor runs at a speed of 576 rpm at full load.
(a) How many poles does the motor have?
(b) What is its slip and frequency of rotor currents at full load? Also find rotor speed with respect to the
rotating field.
(c) What is the motor speed at twice full-load slip?
(d) By what factor should the rotor resistance be increased for the motor to run at a speed of 528 rpm at
full-load torque?
SOLUTION
(a) Nearest synchronous speed,
ns = 600 rpm
120 ¥ 50
P=
=10
600
600 - 576
= 0. 04
600
f2 = 0.04 ¥ 50 = 2 Hz
s=
Rotor speed with respect to the rotating field = 0.04 ¥ 600 = 24 rpm
s = 2 ¥ 0.04 = 0.08
n = (1 – s)ns = (l – 0.08) ¥ 600 = 552 rpm
(c)
600 - 528
= 0.12
600
s (new )
0.12
=
=3
s (old )
0.04
s (new) =
(d)
For the same motor torques, the rotor current must remain constant. As the rotor slip becomes 3 times, the rotor
induced emf increases by the same factor. Therefore, for rotor current to remain the same, its resistance must be
increased 3 times.
The dc Machine
Figure 5.13 showed the essential constructional features of an elementary 2-pole dc machine. The stator has
a fixed pole structure with dc excitation which means that the stator-created flux density wave acting on the
rotor periphery remains fixed in space. For torque to be created, the armature (rotor) when carrying currents
must produce an mmf pattern that remains fixed in space while the armature moves. After the study of the
dc winding in Ch. 7 and how it is connected to commutator segments, it will be seen that the armature mmf
Basic Concepts in Rotating Machines
245
in a dc machine is indeed fixed in space and makes an angle of 90° with the main field. As the dc machine
structure is necessarily of the salient pole type, the main pole flux density wave is far from sinusoidal* (see
Fig. 5.14) and the armature mmf is stepped triangular (to be shown in Ch. 7). Equation (5.58) will not be used
in finding the torque expression for the dc machine as this result applies to sinusoidally distributed fields.
5.8
LINEAR MACHINES
So far we have dealt with rotary machines which find universal use. Each of these machines can have their
linear motion version. The developed diagrams that we have used extensively are linear versions of the
corresponding machine which are employed for specific purposes where linear motion is the requirement
like in transportation and reciprocation, machine tools, and also in limited range linear motion in robotics.
In these applications, linear induction motors are used because of constructional convenience and low cost.
In rail transportation, the ‘rotor’ of the normal induction motor is the conducting stationary rail, which
acts as short circuited conductor. The wound stator is on the moving vehicle. The details and performance
characteristics of the linear induction motor shall be taken up in Ch. 9. Here we shall present the basic
analysis of a linearly moving field. The mmf diagram for one phase of linear concentrated winding is the
same as the developed diagram of Fig. 5.24(b) and is redrawn in Fig. 5.45 where the linear dimension is z in
place of angle q. Of course, torque would now be force.
Fa1,fundamental
MMF
Nil2
b/2
Ni
b/2
b
0
3b
2
Z
–Nil2
b
wave length
Stationary
Movable
a¢
a¢
a
S
South pole
N
North pole
Fig. 5.45
* In fact it is desired to make the B-wave as flat-topped as possible to yield high value of flux/pole for given physical
dimensions.
246 Electric Machines
It is seen from Fig. 5.45 that the wavelength of the fundamental of the mmf wave is b, which corresponds
to 2 poles with electrical angle 2p. The fundamental mmf can be expressed as
Fa1 =
Ê 2p
4 Ê Ni ˆ
cos Á
p ÁË 2 ˜¯
Ë b
ˆ
z˜
¯
Fa1 =
Ê N ph (series) ˆ
Ê 2p
4
Kw Á
˜¯ cos Á
Ë
p
P
Ë b
(5.72)
For a distributed winding,
Fa1 =
or
4 2
p
ˆ
z˜
¯
Ê 2p
Ê N ph (series) ˆ
Kw Á
˜¯ I cos wt cos Á
Ë
Ë b
P
ˆ
z ˜ ; I = rms current
¯
(5.73)
For a 3-phase winding carrying 3-phase balanced currents at frequency w = 2pf, it can be shown (on the
lines of rotating magnetic field) that
F (z, t) =
Fm =
where
3
Ê 2p
ˆ
Fm cos Á
b - wt˜
Ë z
¯
2
4 2
p
Ê N ph (series) ˆ
Kw Á
˜¯ I
Ë
P
(5.74)
(5.75)
It is observed from Eq. (5.76) that the resultant field is a travelling wave, whose speed (linear) is found as
2p
z – wt = K (any constant value)
b
Ê 2p ˆ dz
ÁË b ˜¯ dt – w = 0
dz
wb
=
= fb
or
v=
dt
2p
The field thus travels at speed v.
EXAMPLE 5.19
The data of a 3-phase ac linear motor is as under:
Wave length, b = 0.5 m; gap = 1 cm
Distributed 3-phase winding spread over 2m length
Nph (series) = 48; Kw = 0.925
Supply frequency, 25 Hz, 3-phase balanced currents, I = 750/ 2 A (rms)
Calculate:
(a)
(b)
(c)
(d)
Amplitude of travelling mmf wave
Peak value of air-gap flux density
Velocity of the travelling mmf wave
Current, frequency if the desired velocity is 72 km/h
SOLUTION
(a)
Peak amplitude =
Ê N ph (series) ˆ
3 4 2
Kw Á
◊
˜¯ I
Ë
P
2 p
(5.76)
Basic Concepts in Rotating Machines
Winding length = 2 m or
247
2
= 4 wavelength
0.5
One wavelength = 2 pole
P =2¥4=8
Fpeak =
3 4 2
48 750
¥
¥ = 7.95 ¥ 103 A/m
◊
¥ 0.925 ¥
2 p
8
2
Observe the units.
(b)
Bpeak =
(c)
v=
g
=
4p ¥ 10- 7 ¥ 7.95 ¥ 103
1 ¥ 10- 2
= 0.999 or 1 T
wb
= 25 ¥ 0.5 = 12.5 m/s
2p
72 ¥ 103
= 20
3600
20 = f ¥ 0.5 or f = 40 Hz
v=
(d)
5.9
m0 Fpeak
MAGNETIC LEAKAGE IN ROTATING MACHINES
The leakage flux in rotating machines is that flux which links only the stator or only the rotor windings.
Because of the presence of air-gap in the magnetic circuit of machines, the leakage in these is quite significant
and cannot be neglected in analysis as could be done in the case of transformers. The leakage in machines
falls into the following two broad categories:
1. Leakage in main poles, and
2. Leakage in armature.
Leakage in Main Poles
The main poles of a dc machine and that of a synchronous machine are excited by means of dc to produce a
steady properly distributed flux density; the chief difference between the two being that while the poles of a
dc machine form the stator and that of a synchronous machine are located on rotor. The useful flux is that flux
which coming from the main poles crosses the air-gap and enters the armature. Some of the flux leaks through
via two typical paths indicated in Fig. 5.46(a) without entering the armature. This then constitutes the leakage
flux whose only effect is to increase the flux density in the roots of the poles without contributing to useful
flux and therefore must be accounted for in the magnetic circuit design of the machine. Similar leakage takes
place in the poles of a synchronous machine as shown in Fig. 5.46(b).
Leakage in Armature
The complexity of identifying the leakage paths in the wound armature arises from the fact that the winding is
distributed and the armature surface slotted. The total leakage flux of the armature can be divided into several
components identified in the following.
Slot leakage
This is the flux which follows the path from tooth to tooth across the slots as shown in Fig. 5.47
and in the process links stator/rotor windings only. It is observed that the path of the slot-leakage flux is
perpendicular to that of the main flux, which passes radially down the teeth, and a very small part of it straight
248
Electric Machines
N
S
S
Useful (mutual)
flux
N
Leakage flux
(a) dc machine
Useful (mutual) flux
Stator
Fm
Leakage
flux
Rotor
Leakage flux
(front and back of poles)
(b) Synchronous machine
Fig. 5.46
Leakage in main poles
Fig. 5.47
Slot leakage
down the slots. It is further to be noticed that a smaller amount of leakage flux links the bottom conductors in
slots than the top conductors. The slot leakage is very much dependent upon the shape of slots. It is larger in
semi-closed slots (Fig. 5.48(a)) used in induction machines, because of narrow (low reluctance) slot opening,
compared to open slots (Fig. 5.48(b)) used in synchronous and dc machines.
(b) Open slots
(a) Semiclosed slots
Fig. 5.48
Tooth-tip leakage This flux follows the path from the tip of one tooth to the adjoining one enclosing all
the conductors in the slot as shown in Fig. 5.49. This type of leakage flux is larger for larger stator to rotor
Basic Concepts in Rotating Machines
249
air-gap as more area for the leakage flux is available. Therefore, the tooth-tip leakage is smaller in induction
machines with a narrow air-gap than in synchronous machines which use much larger air-gaps.
Over-hang leakage This is the leakage flux which surrounds the end conductors of the winding (stator/
rotor) as shown in Fig. 5.50. Its path mainly lies through air but a part of it may be located in the core-iron
or the iron of end shields. The amount of this leakage depends upon the proximity of conductors and their
relative location with respect to both core and end-shields. This leakage is generally small because of the
large air paths involved. It is particularly insignificant in the squirrel-cage induction machine rotor which has
no over-hang.
Over-hang
Core
Fig. 5.49 Tooth-tip leakage
Fig. 5.50
Zig-zag leakage In the case of induction machines both the stator and rotor are slotted so that some of the
flux follows the path alternating between stator and rotor teeth as shown in Fig. 5.51. This flux therefore
alternately links conductors in stator and rotor slots and is known as zig-zag leakage. Because of its nature
it cannot be clearly assigned to either the stator or rotor windings. It is usually considered empirically that
half of this flux links the stator winding while the other half links the rotor winding. This type of leakage is
an exclusive feature of the induction machine and its value is a function of the percentage of the slot-pitch
occupied by tooth in the rotor and stator and upon the length of the air-gap.
Fig. 5.51 Zig-zag leakage
250
Electric Machines
Harmonic leakage This kind of leakage results when the winding distribution on the stator and rotor are
dissimilar. The main flux then has a harmonic component not corresponding to either winding and this excess
flux has the effect of leakage flux. The detailed treatment of this kind of leakage is beyond the scope of this
book.
Leakage reactance The leakage flux of various kinds as enumerated above, linking one of the windings,
causes that winding to possess leakage reactance which can be considered as a lumped parameter in series
in the circuit model of the machine whose effect is to cause a voltage drop in the machine. Since the field
windings of dc and synchronous machines carry direct current, the leakage flux linking them in no way
affects the machine steady-state performance.
5.10
LOSSES AND EFFICIENCY
The losses and efficiency of a transformer have been studied in Sec. 3.6. As in the case of transformers, it is
more accurate to determine the efficiency of a rotating machine by determination of its losses rather than by
the direct load test in which the input and output are required to be measured. Furthermore, in large and even
in medium-size machines, it is not practically possible to arrange for the actual loading of the machine. Once
the losses have been determined, the machine efficiency (h) can be computed from the relationships:
h=
Output
Losses
=1; (for generators)
Output + losses
Output + losses
h=
Input - losses
Losses
= 1; (for motors)
Input
Input
The efficiency thus determined is more accurate because the error involved is only in losses, whereas in
the direct method there is error in measurement of both the input and output.
The study of losses is essential for design purposes because (i) losses directly influence the economy of
operation of the machine; and (ii) the rating of a machine depends on the maximum temperature that the
insulation can withstand, which in turn is dictated by the heat developed in the core and conductors by the
losses. Of course, the rating of a machine for a given frame size and losses can be raised by proper design of
the ventilation system.
The process of energy conversion in rotating machines involves currents, fluxes and rotation which cause
losses in conductors and ferromagnetic materials, and mechanical losses of rotation. Various losses can be
classified conveniently by the tree-diagram shown in Fig. 5.52.
Constant Losses
A machine is normally designed to run at constant voltage mains and at a substantially constant speed
(variable speeds are also required for certain applications). As a result, some of the losses remain nearly
constant in the working range of the machine and are, therefore, named constant losses. The constant losses
can be further classified as no-load core-loss and mechanical-loss.
No-load Core (lron)-Loss
This loss consists of hysteresis and eddy-current loss caused by changing flux densities in the iron core of
the machine when only the main winding is excited. The core-loss is largely confined to the armature of a
Basic Concepts in Rotating Machines
251
Losses
Constant losses
No-load core
(iron) loss
Hysteresis
loss
Variable losses
Copper(l 2R)
loss
Mechanical
loss
Eddycurrnet
loss
Stator
copper loss
Windage loss
(including ventilation)
Rotor
copper loss
Friction loss
Brush friction
loss
Stray-load
loss
Brushcontact loss
Copper
stray-load
loss
Core
stray-load
loss
Bearing friction
loss
Fig. 5.52
dc machine, the armature of a synchronous machine and the stator of an induction machine. The frequency
of flux density variation in the rotor core of the induction machine is so low (sf ) under normal operating
conditions that it has negligible core-loss.
While in the case of transformers the core-loss arises because of time-variation of the flux density with
the axis of flux remaining fixed; in the case of rotating machines, this loss results from both time-variation of
the flux density and rotation of its axis. As a consequence the specific core-loss is larger in rotating machines
than that in transformers.
The time- and axis-variation of the flux density in a rotating
Main pole
machine is illustrated by means of the cross-sectional view of
N
a dc machine as shown in Fig. 5.53. It is easily seen from this
Pole-shoe
figure that as the machine armature rotates, the flux density in
the elemental volume of the core shown shaded varies cyclically
Armature
in magnitude as well as in direction.
Additional hysteresis and eddy-current loss called
pulsation loss also occurs in rotating machines on account
of high-frequency flux density variations caused by slotting
of the stator/rotor or both. In the case of dc and synchronous
machines, the relative movement between the slotted armature
and the poles causes high-frequency flux density variation
in the pole-shoes because of the difference in reluctance of
S
the flux paths corresponding to the teeth and slots. In case of
induction machines where both the stator and rotor are slotted,
the pulsation frequency is different in the two. In order to reduce Fig. 5.53
the pulsation loss, it is a common practice to use laminated
machine
252
Electric Machines
pole-shoes for dc and synchronous machines; also for small machines of this type, the main pole itself may be
built-up of laminations. Ofcourse, much thicker laminations are used in pole-shoe than in the machine core.
Hysteresis and eddy current losses in the core cause the flux density wave to somewhat lag behind the mmf
wave producing a torque which acts as a drag on the rotating member. In this regard the core-loss appears as
if it is mechanical loss as the hysteresis and eddy-current torque absorb mechanical power from the shaft. The
torque caused by these losses is relatively small. Practical use is made of this torque in small motors known
as hysteresis motors (Sec. 10.3).
Mechanical Loss
This comprises brush friction, bearing friction, windage and ventilation system losses, all of which are selfexplanatory. Mechanical loss may be relatively large in a machine of large diameter or high speed.
The no-load core-loss and mechanical loss together are represented in literature by the term no-load
rotational loss.
Variable Losses
These losses vary with the load supplied by the machine and are hence called “variable losses”. These can be
split into copper loss (I 2R) and stray-load loss.
Copper-loss (I 2R)
All windings have some resistance (though small) and hence there are copper-losses
associated with current flow in them. The copper-loss can again be subdivided into the stator copper-loss,
rotor copper-loss and brush-contact loss. The stator and rotor copper-losses are proportional to the current
squared and are computed with the dc resistance of windings at 75°C.
The conduction of current between the brushes (made of carbon) and the commutator of a dc machine is
via short arcs in the air-gaps which are bound to exist in such a contact. As a consequence, the voltage drop
at the brush contact remains practically constant with load; its value for positive and negative brushes put
together is of the order of 1 to 2 V. The brush-contact loss in a dc machine is therefore directly proportional
to current. The contact losses between the brushes (made of copper-carbon) and slip-rings of a synchronous
machine are negligible for all practical purposes.
Copper-losses are also present in field windings of synchronous and dc machines and in regulating the
rheostat. However, only losses in the field winding are charged against the machine, the other being charged
against the system.
Stray-load loss Apart from the variable losses mentioned above, there are some additional losses that vary
with load but cannot be related to current in a simple manner. These losses are known as “stray-load loss” and
occur both in the windings and the core.
(i) Copper stray-load loss Additional copper-loss occurs in the conductors due to nonuniform
distribution of alternating currents which increase the effective resistance of conductors and is known
as skin-effect. Further, when the conductors carry load current, the teeth of the core get saturated and as
a consequence more flux passes down the slots through the copper conductors setting up eddy-current
losses in them. Eddy-current losses are also present in the winding overhang.
(ii) Core stray-load loss Due to the flow of load current in a machine, the flux pattern in teeth and core
gets distorted. The flux density decreases at one end of the flux density wave and increases at the
other. Since the core-loss is almost proportional to the square of the flux density, its reduction due to
a reduction in the flux density is less than the increase due to an increase in the flux density and as a
Basic Concepts in Rotating Machines
253
consequence there is a net increase in the core-loss, predominantly in the teeth, which is known as the
stray-load loss in the core.
Under loaded conditions, the teeth are highly saturated and as a result more flux leaks through the stator
frame and end-shields causing eddy-current loss in them which, indeed, is another component of the core
stray-load loss.
The stray-load loss is difficult to calculate accurately and therefore it is taken as 1 % of the output for a dc
machine and 0.5% of the output for both synchronous and induction machines.
Because of the presence of fixed and variable losses in a machine, the machine efficiency continuously
increases with the load acquiring a maximum value at a particular load related to the design of the machine.
Further, the full-load efficiency varies with the rating of a machine and is considerably higher for large-size
machines; for example, the efficiency is close to 75% for 1 kW machine, 90% for 35 kW, 93% for 350 kW
size and as high as 97% for 3500 kW. Efficiency of low-speed machines is usually lower than that of highspeed machines, the spread being 3 to 4%.
For a machine operating at a substantially constant voltage and speed, the various losses as enumerated
earlier are:
(1) Constant losses*,
Pk = Pi0 + Pwf
where
(5.77)
Pi0 = no-load core (iron)-loss (constant)
Pwf = windage and friction loss (constant)
(2) Variable losses,
Pv = Pc + Pst + Pb
where
(5.78)
2
Pc = 3I R, the copper-loss (factor 3 will not be present in a dc machine);
R is the resistance parameter of the machine.
Pst = stray-load loss (copper + iron) = a I 2
(Here the stray-load loss is assumed proportional the square of the load current)
Hence
Pb = Vb I = brush-contact loss (in dc machines); Vb being the brush-contact voltage drop
Pv = 3I2 R + a I 2 + Vb I = (3R + a) I2 + Vb I = Kv I2 + VI
(5.79)
Thus total machine losses can be expressed as a function of the current as
PL = Pk + Kv I 2 + VbI
Generating machine
Power output, Pout = CVI
C = constant ( 3 ¥ pf for 3-phase ac machine)
V = machine voltage (line)
Efficiency, h =
Pout
Pout + PL
* Field copper-loss for a dc and synchronous machine is constant and can be lumped with constant losses.
(5.80)
254
Electric Machines
=
=
CVI
CVI + Pk + K v I 2 + Vb I
CV
ÊP
ˆ
(CV + Vb ) + Á k + K v I ˜
Ë I
¯
(5.81a)
(5.81b)
The maximum efficiency is obtained at (minimum denominator in Eq. (5.81))
or
Pk
= Kv I
I
Pk = Kv I2
(5.82)
Thus the maximum efficiency is reached at a load when the losses proportional to the square of current
equal the constant losses. This is the same conclusion as arrived at for a transformer (Eq. (3.59(a))).
Motoring machine
Power input, Pin = CVI
Efficiency, h =
=
Pin - PL
Pin
CVI - Pk - K v I 2 - Vb I
CVI
ÊP
ˆ
(CV - Vb ) - Á k + K v I ˜
Ë I
¯
=
CV
This also reaches the maximum value when
i.e.
Pk = Kv I 2
Constant losses = losses proportional to the square of current
(5.83a)
(5.83b)
(5.84)
As per the condition of Eq. (5.82) or (5.84) for maximum efficiency, the constant losses and variable
losses (proportionality constant Kv) are so proportioned by the choice of machine dimensions as to yield
maximum efficiency near about the full-load. The constant losses are mainly determined by the choice of
flux density and the volume of iron used and the variable losses are governed by the choice of current density
and the volume of copper used. Further, the flux density used is limited to slightly saturated values and
the current density is limited by the allowable temperature rise (depending upon the class of insulation).
Therefore, adjusting the machine efficiency to yield the maximum value at a particular load is an exercise in
proportioning iron and copper to be used in the machine.
Maximum Output
Consider for example the motoring machine. The power output is expressed as
Pout = CVI – Pk – Kv I2–VbI
It is a fairly good assumption to neglect Vb; in fact this term is not present in ac machines. Then
Pout = CVI – Pk – Kv I2
Basic Concepts in Rotating Machines
255
For maximum power output
dPout
= CV – 2KvI = 0
dI
CV
or
I=
2Kv
The maximum power output is then given by
Ê CV ˆ
Ê CV ˆ
Pout (max) = CV Á
– PK – Kv Á
˜
2
K
Ë v¯
Ë 2 K v ˜¯
=
The power input is
2
(CV ) 2
– Pk
4Kv
(CV ) 2
2Kv
The efficiency at maximum power output is given by
Pin = CVI =
h=
(CV ) 2 / 4 K v - Pk
(CV ) 2 / 2 K v
Obviously this will be slightly less than 50%. This is too low a value to be acceptable for a powerdelivering device. Further, under maximum output operation, the losses being almost half the input, it would
be impossible to limit the temperature rise to the allowable value. Thus the electromechanical power devices
are never operated to deliver maximum output. In fact these are operated at a load (nearly full-load) at
which the efficiency is maximum. This is in contrast to electronic devices (low power) which are usually
operated to deliver maximum power output as the total power being very small, the efficiency is of secondary
consideration. Further, the problem of heat (caused by losses) dissipation is not so intense as in large power
rotating machines.
5.11
RATING AND LOSS DISSIPATION
Rating
The rating of a synchronous generator is its operating voltage, frequency, speed and kVA/MVA output at a
specified power factor*. In case of motors the output rating is given in kW (older practice was to specify the
rated output in horse-power (1 hp = 746 W)). The dc machines are rated in terms of voltage and power output
in kW. The frequency specification of ac machines is normally the standard supply frequency, i.e. 50 Hz—it
is 60 Hz on the American continent. The rated voltage is specified as a standard value, viz 230 V/ 400 V/
3.3 kV/6.6 kV/11 kV. Most manufacturers build small and medium sized motors in standard kW sizes (consult
ISI: 325–1970).
Insulation which is used in intricate forms is the most vulnerable part of a machine being highly susceptible
to temperature which is the major factor** in determining its lifetime and therefore that of the machine. As
* kVA/MVA rating and specified pf determine the mechanical power rating of the prime mover to which it is
coupled. It can of course be operated at other power factors.
** Other factors affecting the life of insulation are oxidation and ingress of dirt and moisture.
256
Electric Machines
a rule of thumb the lifetime of a machine is reduced to one-half for every 8–10 °C rise in temperature.
Acceptable life expectancy of electric power equipment being 10–30 years, the highest temperature of
insulation anywhere in the machine has to be limited to a value depending upon the class of insulation
employed. The maximum output that a machine can supply without exceeding a specified temperature rise
above the ambient (40°C as per ISI), is known as the continuous rating of the machine. The continuous
rating, as determined by temperature rise, limits the machine losses generally assuring an acceptable value of
machine efficiency. Any other specific performance figure(s) (say the breakdown torque of induction motor)
must be independently met by the machine designer.
Both iron and copper being the seats of losses in a machine, the temperature distribution in the machine
is quite complex. This problem is greatly simplified by assigning a single temperature to define the thermal
state of the machine. The single temperature is determined by the rise in resistance of windings as measured
from the machine terminals. In super-size synchronous generators, thermocouples are embedded during
manufacture to track temperature of hot-spots predetermined by the designer through heat transfer studies.
ISI specifications classify insulation for industrial machines as Class B, Class F and Class H. Class B
insulation includes mica, glass fibre, asbestos, and other similar materials with suitable binding substances.
The highest temperature rise (40°C above ambient) allowed for this class of insulation is 130°C. Class F
insulation also includes mica, glass fibre, and synthetic substances, but it must be able to withstand a higher
temperature of 155°C. Class H can withstand a still higher temperature of 180°C and may consist of materials
like silicone elastomer and combinations in various forms of mica, glass fibre, asbestos, etc. with silicone
resins for bonding.
A continuous-rated motor must operate successfully ±10% variation of the rated voltage and ±5% variation
of the rated frequency. The combined variation of voltage and frequency in a direction to adversely affect
losses (V increases, f reduces) must not exceed 10%. It is further expected that the continuous-rated motors
are built with ample safety margin so as be capable of withstanding short-time overloads of 25% with 10%
reduction in voltage without excessive temperature rise.
In industrial applications certain situations require the motor to be loaded for short time periods followed
by long cooling intervals. Such motors are short-time rated for standard periods of 5, 15, 20, 30 and
60 min. These motors are specially designed with higher flux densities in iron and higher current densities in
copper. As a result they have better torque producing capability but lower thermal capacity when compared
to continuous-rated motors.
Short-time rating of a continuous-rated motor is much more than its continuous rating because of the
heat storage in thermal capacity of the machine during the heat transient under short-time loading. This is
illustrated in Fig. 5.54. The thermal transient* has an exponential growth (a single time constant) with a
steady temperature rise of (PLRT) where PL represents the motor loss in heat units at a particular load and RT
is the overall thermal resistance of the cooling system. It is evident from Fig. 5.54 that P stL (short-time loading
loss) allowed for a specified loading period is more than PLc (continued loading loss) without the machine
* Figure 5.55 gives the simplified, lumped, analogous circuit model of the machine for heat transfer. The temperature (single temperature representing the overall thermal state of the machine) is measured with respect to ground
reference (ambient). Here
PL = motor loss in heat units
RT = overall thermal resistance of the cooling system
CT = thermal capacity of the machine
(Contd. on next page)
Basic Concepts in Rotating Machines
257
Temp. rise
PLst RT
Short-time rating
T max = PLc RT
Continuous rating
Time
Loading period
Fig. 5.54 Continuous and short-time rating
exceeding the allowable temperature rise. Hence as the loading period is reduced, the motor short-time rating
increases.
In industrial applications of motors a typical problem is the determination of the size of a continuous-rated
motor for a given duty cycle—the duty cycle of a “planer” may be visualized as a simple example; during the
forward stroke the motor is on full-load while it is practically unloaded during the return stroke. A crude yet
reliable method of motor selection is to assume that the motor losses are proportional to the square of loading
(this overemphasises I2R loss as compared to the constant coreloss). The average loss during a duty cycle is
proportional to
È S (kW ) 2 ¥ time ˘
Í
˙
S time
ÍÎ
˙˚
(Contd. from previous page)
The differential governing the thermal transient is
dT
T
+
= PL
dt RT
whose solution for initial temperature rise of 0 °C (machine starting from cold conditions) is
CT
where
T(t) = T, (1 – e–t/t)
t = RT CT, the thermal time-constant
Ts = PLRT the steady-state temperature rise
T
PL
CT
RT
g
Fig. 5.55
258
Electric Machines
where kW = motor loading in a period of duty cycle. The continuous-rated motor which has the same loss is
given by
È S (kW ) 2 ¥ time ˘
(kW)2continuous-rating = Í
˙
S time
ÍÎ
˙˚
or
(kW)continuous-rating =
S (kW ) 2 ¥ time
= (kW)rms (of the duty cycle)
S time
If stand-still period(s) are involved as part of the duty cycle (as in a crane) the above relationship must be
modified as under
(kW)rms =
S(kW ) 2 ¥ time
running time + (stand-still time/k )
(5.85)
where the constant (k > 1) accounts for poor ventilation (cooling) during the stand-still period(s) where there
is no forced cooling. For open-type motors, k = 4. It is tacitly assumed in Eq. (5.85) that the duty cycle period
is sufficiently less than the time for the motor to reach almost its steady temperature rise when continuously
loaded to its continuous-rating.
The errors involved in the (kW)rms-method are swamped out when the nearest higher standard rating, e.g.
90 kW motor is selected for (kW)rms = 85.
A duty cycle requiring high torque peaks cannot be satisfied by (kW)rms choice which has a thermal basis.
Short-time rated motors as already mentioned are better suited for such applications because of their better
torque-producing capabilities.
Loss Dissipation (Cooling)
To prolong insulation life-time to an acceptable value, the heat generated owing to loss in a machine must be
dissipated fast enough so that the temperature rise does not exceed the allowable limit for a specified ambient
temperature. In fact it is the improvement in heat transfer technology that has helped in a major reduction in
machine sizes for given ratings, in particular for large-size machines.
Combined conduction and forced convective cooling are the practical means of removing heat of losses
from all electric machinery. Because of limited allowable temperature rise, radiation does not make any
significant contribution to loss dissipation.
Radial ventilation Radial ventilation is commonly employed wherein the natural centrifugal action of the
rotor may be supplemented by the rotor fan. Figure 5.56 shows the radial ventilation scheme suitable for
machines up to 20 kW.
Axial ventilation Axial ventilation scheme of Fig. 5.57 is suitable for machines of moderate outputs and
high speeds.
Combined radial and axial ventilation
This is employed for large machines as shown in Fig. 5.58 for an
induction motor.
Totally-enclosed Totally enclosed machine presents a special ventilation problem as the inside of the
machine has no air-connection with outside. In such machines heat is transferred to the enclosure (called
Basic Concepts in Rotating Machines
259
Shaft
Shaft
Fig. 5.56
Fig. 5.57
Fig. 5.58
carcass) by an internal fan and from where it is removed to the ambient by an external fan mounted on the
shaft. The cooling in a totally enclosed machine cannot be as efficient as in an open-type machine.
Losses being roughly proportional to the volume of material increase as the cube of the linear dimensions,
while the cooling surface increases as the square of the same. Therefore, the loss dissipation problem becomes
more intense in large turbo-generators.
For large machines, which may require several tonnes of cooling air/hour, forced ventilation is used
wherein air is passed through a cleaning filter before being forced into the machine for cooling purposes.
A more compact scheme of securing clean cooling air is the closed-circuit system as employed for turbogenerators of small rating. In this system hot air is cooled by a water-cooled heat exchanger.
Hydrogen Cooling
For large turbo-generators, hydrogen is commonly used as a cooling medium in a closed circuit. The following
properties of hydrogen make it most suited for this purpose.
1. Hydrogen has a density of 1/14 of that of air at the same temperature and pressure, reducing thereby
windage losses and noise.
2. On an equal weight basis, the specific heat of hydrogen is 14 times that of air. Therefore, for the same
temperature and pressure, the heat-storing capacity/ unit volume of hydrogen is the same as that of air.
3. The heat-transfer capability of hydrogen by forced convection over a hot surface is 1.5 times that of air.
4. The thermal conductivity of hydrogen is seven times that of air.
260
Electric Machines
5. By use of hydrogen environment, the life of insulation is prolonged and the maintenance cost goes
down because of the absence of dirt, moisture and oxygen.
6. The hydrogen-air mixture does not explode so long as air content is less than 30%.
To avoid air leaking into the hydrogen circuit, hydrogen pressure is maintained above 1 atm. Hydrogen
cooling at 1, 2 and 3 atm can raise the rating of a machine by 15%, 30% and 40% respectively. Hydrogen
cooling reduces the temperature and resistance of windings and hence the losses to be dissipated. This fact
marginally raises the full-load efficiency of the machine (by about 0.5%).
The machine and its water-cooled heat exchanger for cooling hydrogen are enclosed in a gas-tight
envelope; the most intricate problem being that of sealing the bearings. Oil-filled gas-seals are used for this
purpose. Further, the envelope must be explosion-proof.
Direct Gas cooling
For a machine of 100 MW or more, the temperature gradient over the conductor insulation is high enough
to call for direct contact between the coolant and conductor. For this purpose, the rotor conductor comprises
hollow tubes as shown in Fig. 5.59 through which hydrogen is circulated by means of flexible connections.
Direct Water Cooling
Turbo-generators of the highest rating have a hydrogen-cooled stator core and a direct water-cooled stator
and rotor windings. The speed of circulating water must be limited to 2.5 m/s to avoid erosion and cavitation.
Figure 5.60 shows the arrangement for direct water-cooling of the rotor winding which is most desirable
Fig. 5.59
Direct gas cooling
Fig. 5.60
Basic Concepts in Rotating Machines
261
because of high electric loading of rotor and is mechanically most difficult. Direct water-cooling of the
stator winding requires flexible water-tube connections with insulation against high voltages and low water
conductivity.
5.12
MATCHING CHARACTERISTICS OF ELECTRIC MACHINE AND LOAD
The machine and the load are the two components of an electro-mechanical energy-conversion system, and
the machine characteristics, generally, play a predominant part in the operating behaviour of the complete
system.
Speed
In choosing an electric motor its speed-torque characteristic
is needed to be known to a fair degree of accuracy and further
Load
P
it has to be properly matched to the speed-torque characteristic
ns
of the mechanical load. Figure 5.61 shows the speed-torque
characteristic* of an induction motor with a fan-type load
Motor
(load torque roughly proportional to square of speed). The
steady operating point is the intersection point P of the two
characteristics. As can be seen from Fig. 5.61, it is a stable
operating point and the machine-load system returns to it when
0
Torque
subjected to a short-duration disturbance.
The characteristics of mechanical loads can be classified as Fig. 5.61 Steady operating point of a motorload system
below:
1. Constant-speed loads These can be of two kinds. Certain loads require approximately constant speed as
the load torque varies, e.g. machine tools, hydraulic pumps, fans, etc. Certain special loads like paper mill
drives require exactly a constant speed independent of the load torque.
2. Variable-speed (or constant kW) loads Certain loads, such as cranes, hoists and other traction-type
drives, demand high torque at low speeds and low torque at high speed so that the kW demanded from the
mains remains substantially constant. This nature is imparted to the load wherever heavy inertias are to be
accelerated.
3. Adjustable-speed loads These are of a constant adjustable speed kind as in certain machine tool
applications or of a variable adjustable speed kind as in cranes. The range of speed adjustment in certain
drives can be highly demanding.
The motor characteristics can be classified as:
1. Constant-speed type The speed remains exactly constant independent of torque as in Fig. 5.62(a). This
characteristic is possessed by the synchronous motor.
2. Shunt-type Here the motor speed drops by a few per cent from no-load to full-load as in Fig. 5.62(b). The
ac induction motor (over the operating region) and dc shunt motor both possess this characteristic.
3. Series-type Here the speed rises sharply as the load torque reduces as in Fig. 5.62(c). This type of
characteristic is possessed by a dc series motor ideally suitable for traction-type loads.
Adjustable speed drives require the adjustment (raising/lowering) of the three motor characteristic types.
It will be seen in later Chapters (7, 8 and 9) that this is much more easily accomplished in dc motors than in
ac motors. Solid-state power control (Ch. 12) has contributed a lot to adjustable speed drives.
* This is the same characteristic as in Fig. 5.43 except with the ordinates reversed.
262
Electric Machines
n
n
n0
ns
0
0
T
(a) Synchronous-type characteristic
T
(b) Shunt characteristic (n0 = ns for
induction motor)
n
0
(c) Series characteristic
T
Fig. 5.62 Types of motor characteristic
The accelerating (starting) and decelerating (braking) characteristics of motor-load systems are also of
equal importance in their industrial applications. The system should be capable of coming to full speed from
rest and be able to be stopped in an acceptable time period. These requirements are stringent in starting onload and in fast braking and reversal in certain special applications (rolling mill drives). A motor has three
regions of operation—generating, motoring and braking. In generating region it returns the decelerating
intertial energy back to electric main preventing the system from acquiring dangerously high speeds—as in
lowering a hoist or down-the-gradient traction. In braking region the machine absorbs mechanical energy (as
well as some electric energy) in form of losses in it appearing as heat. A dc motor offers excellent starting and
braking characteristics, much superior to those of an ac induction motor.
Similar to the case of motors, the operating point of a generator-load system is determined by the
characteristics of the two as shown in Fig. 5.63 for a dc
shunt generator (Sec. 7.11). Similar is the case with ac Voltage
(synchronous) generators. In modern systems, generators
operating in parallel feed loads spread over geographically
wide areas through transmission lines. The system must
meet the requirement of a substantially constant voltage as
load varies over a wide range. A captive generator feeding
Load
P
a single motor is used in certain speed-control schemes in
which the terminal voltage may be required to vary in a
Generator
peculiar fashion.
It is therefore seen that among the features of great
importance are the torque-speed characteristic of a motor
0
Current
and the V-I characteristic of a generator. Equally important
can be the limits through which these characteristics can Fig. 5.63 Steady operating point of a generatorload system
be varied. Other relevant important economic features of
Basic Concepts in Rotating Machines
263
an electric machine are efficiency, power factor, comparative cost and effect of losses on heating and rating
(already discussed in Sec. 5.10). Apart from machine modelling, it is these performances that form the subject
matter of a major part of the chapters that follow.
Other than the steady-state operation, which will be discussed in depth in this book, the electrical transient
response and dynamic response of the machine-load system are of outstanding importance in power systems
and in automatic control systems. The in-depth study of these topics forms a separate study and will only be
touched upon here and there in this book.
5.13
RESUME
In this chapter the following common features of rotating machines have been studied—ac windings
(elementary treatment), emf and mmf of ac winding, concepts of rotating magnetic field and process of torque
production by two interacting fields. Differences have also been discovered, particularly in a dc machine
which has a commutator, the study of whose action has been postponed to Ch. 7. Through expressions of
emf and torque, it was seen that the machine capability for a given frame size is limited by (i) the saturation
flux density in iron parts of the machine, and (ii) the current-carrying capability of windings which is limited
by temperature rise. Improvement in machine performance and the cost per unit power have over the years
resulted mainly from improvement in quality and characteristics of magnetic, conducting and insulating
materials. Another important thrust forward has been in the direction of heat removal from the seats of
heat generation (because of inherent power loss) so as to limit the temperature rise to that permitted in the
insulating material which in fact is the most vulnerable part of a machine.
A theoretical level has now been reached at which simple mathematical models of various types of rotating
machines can be built and their performance characteristics studied through these models.
Constructional Feature – Electric Machines
Rotating electric machines have two flux carrying parts which are made of laminated silicon steel.
These two are the following:
Stator: It is a stationary annular cylinder
Rotor: It rotates within the stator supported by a shaft, ball bearings and end rings bolted to the stator.
There is a narrow air-gap between the stator and rotor.
Windings: There are two windings made of copper. These are placed in stator and rotor slots or in one
of these wound on projecting poles. In synchronous and dc machines the main field is created by the
field poles (even in number) and dc excited. The other winding which interchanges electric power with
the external circuit and so carries the load current is called the armature winding and in the seat of
induced emf.
In a synchronous machine, the field poles are on the rotor and armature winding on the stator. It
is the preferred construction and universally adopted. Excitation current is passed to the field poles
through slip-ring brush arrangement. In a dc machine it is a must that the field poles are on the
stator and the armature on the rotor. The rotor also carries a commutator whose segments are suitably
connected to the armature windings and act to convert the alternating armature current to dc for the
external connection.
264
Electric Machines
In an induction machine both stator and rotor are slotted and carry armature windings; rotor may
carry just slot conductors shorted by end rings.
Thus the electric machines are of two types:
- ac Machines: Synchronous and induction
- dc Machines
Field-pole types
Salient projecting poles, non-salient cylindrical poles
Synchronous machine can have both types, dc machine has only salient poles
Induced dc emf in rotating machines. It is the speed rmf. The relative motion between B-wave and
coils, which causes change in flux linkage and emf induction
Mechanical and electrical angles
qe
2
; P = number of poles
=
qm
P
Speed-Poles-Frequency
120 f
, synchronous speed in rpm
P
np
Or
f=
Hz
120
Armature Coils Could be single-turn or multi-turn with two end connections.
Coil-side – each active side of a coil
Coil span ( pitch) – full-pitched, angle between coil sides is p rad or 180° electrical
– short-pitched; angle between coil sides is less than p in terms of number of slots
Two-layer windings–two coil sides per slot
Induced emf (ac) of a single N turn full-pitch coil
n=
E(rms) = 2 p f NF = 4.44 f NF
F = flux/pole
Induced emf phasor lags the flux phasor by 90°
Distributed winding More than one coil/phase
Slots/pole/phase, SPP = m =
S
; S = slot, q = number of phases, generally three
qP
pP
rad (elect.)
S
Phase spread, s = mg
Slot angle, g =
Breadth factor, kb Because of distributed winding the phase emf is less than the algebraic sum of
series turns/phase by the breadth factor
kb =
sin mg / 2
<1
m sin g / 2
Generally kb (harmonics) < kb (fundamental)
Therefore, distributed winding incidentally reduces the harmonic content of emf induced.
Basic Concepts in Rotating Machines
265
Short-pitched (corded) coils The emf of a short-pitched coils is less than that for a full-pitched coil
by the pitch factor
q sp
< 1; qsp = short-pitching angle in rad elect.
2
Chording of coil saves in overhang copper by proper choice of qsp any particular harmonic can be
eliminated
Winding factor
Kp cos
Kw = Kb Kp < 1
General formula for induced emf phase
Ep =
2 p Kw f Nph (series) F V
It is applicable for synchronous machine (stator) and induction machine stator and rotor. For 3-phase
synchronous and induction machine the windings of the three phase are laid 120° elect. apart from
each other.
MMF of AC Winding A single coil produces rectangular mmf wave (with north and south poles,
strength (Ni/2)
Fundamental emf wave
Fa1 =
For sinusoidal current (ia =
Fa1 =
4
(Ni/2) cos q,
p
q = space angle elect.
2 I cos wt)
2 K¢ I cos wt cos q
= Fm cos wt cos q
It is a standing pulsating wave
For a distributed winding
Fa1 =
4 2
Ê N ph (series) ˆ
Kw Á
˜¯ I cos w t cos q
Ë
p
P
= Fm cos w t cos q
Rotating Magnetic Field A 3-phase winding with their axis located at 120° elect space phase
difference from each other and fed with 3-phase balanced currents with a time phase difference of 120°
elect. The mmf-wave rotates at synchronous speed ws = 2p f rad (elect.)/s (or ns = 120 f /P rpm). The
direction of rotation of mmf wave is from the leading phase to the lagging phase axis. The number of
poles of the mmf wave is same as for which the winding is wound.
Torque in Round Rotor Machine Necessary conditions for production of steady torque by two
interacting magnetic fields.
1. The two fields must be relatively stationary
2. The two fields must have the same number of poles
Torque expression F1, F2 are peak values of sinusoidally distributed fields rotating at synchronous
speed and Fr is the resultant field.
266
Electric Machines
Torque, T = k F1 F2 sin a,
= k Fr F2 sin d ;
a = angle between F1 and F2
d = angle between Fr and F2
It is Fr that produces the air-gap flux, Fr /pole
Synchronous Machine
Generating F2 leads Fr by angle d. Electromagnetic torque T = TPM in opposite direction to
synchronous
Motoring F2 lags Fr by angle d. Electromagnetic torque T = TL (load torque) in opposite direction to
synchronous speed. Non-self starting
Terminal voltage (equal to induced emf )
3 ¥ 4.44 Kw f Fr Nph (series)
V(line) =
For fixed terminal voltage Fr is constant; same as in a transformer. Therefore
T = KT sin d
Pullout torque, Tmax = KT, d = 90°
Higher torque load causes loss of synchronism or the machine pullout.
Induction Machine Two types: 1. wound rotor, terminals shorted externally 2. squirrel cage rotor,
copper or aluminum bars in slots shorted by conducting end rings.
Operation When the stator is excited from 3-phase mains (V, f ), the rotating field induces currents
in shorted windings and their interaction produces torque. The machine is therefore self-starting. The
steady rotor speed n must be less than ns, the synchronous speed for induction currents in rotor and
torque production. The induction motor is therefore asynchronous motor.
Slip, s =
ns - n
ns
Rotor frequency, f2 = sf
The air-gap flux Fr is determined by the applied voltage. The torque-slip (T – s) characteristic is
non-linear and slip at full-load is 2-5%. So speed less than synchronous is nearly constant (shunt
characteristic).
Maximum torque is the break-down above which the motor stalls.
5.1 Determine the breadth and pitch factors for
a 3-phase winding with 2 slots per pole per
phase. The coil span is 5 slot-pitches.
If the flux density wave in the air-gap consists
of the fundamental and a 24% third-harmonic,
calculate the percentage increase in the rms
value of the phase voltage due to the harmonic.
5.2 A 50-Hz, 6-pole synchronous generator
has 36 slots. It has two-layer winding with
full-pitch coils of 8 turns each. The flux per
pole is 0.015 Wb (sinusoidally distributed).
Determine the induced emf (line-to-line) if
the coils are connected to form (a) 2-phase
winding (b) star-connected 3-phase winding.
5.3 The air-gap flux density distribution of a
6-pole, 50-Hz synchronous generator is
B(q) = B1 (sin q + 0.3 sin 2q + 0.15 sin 5q)
The total flux/pole is 0.015 Wb. Find the fundamental, third-harmonic and fifth harmonic
flux/pole.
Basic Concepts in Rotating Machines
5.4 Show that the limiting value of the breadth
factor for the fundamental is
1
sin s
2
Kb =
1
s
2
where s = mg = phase spread
and m, the slots pole per phase, tend to be
large.
5.5 A 50-Hz synchronous salient pole generator
is driven by a hydro-electric turbine at a speed
of 125 rpm. There are 576 stator slots with
two conductors per slot. The air-gap diameter
is 6.1 m and the stator length is 1.2 m. The
sinusoidally distributed flux density has a peak
value of 1.1 T. (a) Calculate the maximum rms
single-phase voltage that can be produced by
suitably connecting all the conductors. (b)
Find the per phase emf if the conductors are
connected in a balanced 3-phase winding.
5.6 Find the number of series turns required
for each phase of a 3-phses, 50-Hz, 10-pole
alternator with 90 slots. The winding is to be
star-connected to give a line voltage of 11 kV.
The flux/pole is 0.2 Wb.
5.7 A dc armature is built up of laminations having
an external diameter of 80 cm and the internal
diameter of 42 cm. The length of the armature
is 32 cm. The flux density in the armature core
is 0.85 T. The armature is wave connected with
72 slots and 3 conductors/slot. If the number
of poles is 6, find the emf induced when the
armature is rotated at a speed of 600 rpm.
(Hint: Flux passing through armature core is
half the flux/pole. See Fig. 5.13.)
5.8 A 6-pole, wave-connected dc armature has
300 conductors and runs at 1200 rpm.
The emf generated is 600 V. Find the useful
flux/pole.
5.9 A 4-pole, dc machine has a lap-connected
armature having 60 slots and eight conductors
per slot. The flux per pole is 30 mWb. If the
armature is rotated at 1000 rpm, find the emf
available across the armature terminals. Also
5.10
5.11
5.12
5.13
5.14
267
calculate the frequency of emf in the armature
coils.
Trace out the variations in mmf due to a belt
of current-carrying conductors representing
one phase of a 2-pole, 3-phase winding. The
belt may be assumed to be a current-sheet
with uniform current density. What is the peak
amplitude of the mmf wave if the total current
in the belt is A amperes?
(Hint: The mmf wave is trapezoidal.)
Each phase belt of a 2-pole, 3-phase winding
carrying balanced 3-phase currents can be
assumed to be a current-sheet with uniform
density (as in Prob. 5.10). Sketch the resultant
mmf wave at wt1 = 0, wt2 = p /3 and wt3, =
2p/3.
Phase a of a 3-phase stator at the instant of
carrying maximum current has 60 amperesconductors in the phase belt. Sketch the mmf
wave of the phase when the slots/pole/phase
are 1, 2, 3, 4 and 5 respectively. Comment
upon the change in shape of mmf wave with
the number of slots/pole/phase.
A 2-pole, 3-phase ac winding is housed in
18 slots, each slot having 12 conductors.
Consider the time instant at which the current
in phase a has its maximum value of 10 A.
(a) Sketch all the 18 slots on a horizontal
axis. Mark the direction of currents in the
conductors occupying the slots relevant to
phase a. Make a proportional sketch of the
mmf wave of phase a only.
(b) Mark the maximum value of the mmf
wave on the sketch.
(c) Calculate the peak value of the fundamental of the mmf wave.
A 4-pole, 50-Hz induction motor has 24 stator
slots with 2-layer winding. It has 16-turn
coils chorded (short-pitched) by one slot. The
machine is delta-connected and connected to a
440 V, 3-phase supply. If the stator resistance
and leakage reactance are assumed negligible,
find the flux/pole of the rotating flux density
wave.
268
Electric Machines
5.15 The induction machine of Prob. 5.14 has a
stator length of 28 cm and a mean air-gap
diameter of 18 cm. The machine air-gap is
1 mm. What line current will it draw when
running at no-load? Assume the iron to be
infinitely permeable.
(Hint: At no-load the machine draws only the
magnetizing current to establish the flux/pole
as calculated in Prob. 5.14.)
5.16 In Ex. 5.6 what will be the peak value of
resultant mmf/pole, if the winding is 3-phase
and is chorded by one slot.
5.17 A 3-phase induction motor runs at a speed of
1485 rpm at no-load and at 1350 rpm at fullload when supplied from a 50-Hz, 3-phase
line.
(a) How many poles does the motor have?
(b) What is the % slip at no-load and at fullload?
(c) What is the frequency of rotor voltages at
no-load and at full-load?
(d) What is the speed at both no-load and fullload of: (i) the rotor field with respect to
rotor conductors, (ii) the rotor field with
respect to the stator, and (iii) the rotor field
with respect to the stator field.
5.18 A 4-pole, 3-phase synchronous motor fed
from 50-Hz mains is mechanically coupled
to a 24-pole, 3-phase synchronous generator.
At what speed will the set rotate? What is the
frequency of the emf induced in the generator?
5.19 A 20-pole synchronous generator running at
300 rpm feeds a 6-pole induction motor which
is loaded to run at a slip of 5%. Find the speed
at which the induction motor runs and the
frequency of the currents induced in its rotor.
5.20 A slip-ring induction motor runs at 285 rpm
on full-load when connected to 50-Hz supply.
Calculate: (a) the number of poles; (b) the
slip; and (c) the slip for full-load torque if
the total resistance of the rotor circuit is
doubled. Assume the rotor leakage reactance
to be negligible in the range of slips being
considered.
5.21 Consider the synchronous machine with
dimensional and other data. The machine is
excited with same field current If = 20 A.
Calculate the air gap (resultant) flux density
needed to generate a terminal voltage of 3 kV
(line). What is the corresponding maximum
torque it can develop as a motor? What would
be the mechanical load for a torque angle of
25°?
5.22 Consider a synchronous machine which
is used in motor mode. Its mechanical
dimensions and winding particulars are:
Mechanical dimension
Air-gap length = 1.3 mm
Mean air-gap diameter = 22 cm
Axial length = 41 cm
Rotor winding
Total field series turns = 880
Kw (rotor) = 0.965
Stator winding
SSP = 3
Conductors/slot = 12
Operating conditions
Field current, If = 4 A
Peak air-gap flux density, Br = 1.35 T
(a) Find peak rotor ampere-turns F2
(b) Find open-circuit voltage.
5.23. A three-phase ac linear motor has armature
winding wavelength of 30 cm. The supply
currents have a frequency of 75 Hz.
(a) Calculate the linear velocity of mmf wave.
(b) The vehicle velocity of the motor is
synchronous type.
(c) The vehicle velocity of the motor is
induction type with a slip of 0.05.
Ans. (a) 358.1 m/s (b) 358.1 m/s
(c) 340.2 m/s
Solution
wb
75 ¥ 23
=
= 358.1 m/s
2p
2p
(b) Synchronous velocity = vs = 358.1 m/s
(a) v =
Basic Concepts in Rotating Machines
(c) Induction motor
s=
vs - v
v
=1–
vs
vs
5.27
v
=1–s
vs
v = (1 – 0.05)vs = 340.2 m/s
5.24 The armature of three-phase linear motor has
a winding wavelength of 25 cm and winding
length of six wavelength. The 3-phase winding
has 240 turns/phase with a winding factor of
0.92. For an air-gap 0.95 cm it is desired to
have peak fundamental air-gap flux density of
1.25 T. Calculate the rms value of armature
currents needed.
Ans. 190 A
Bpeak =
4p ¥ 10- 7
-2
=
5.28
5.29
3 4 2
Ê 240 ˆ
◊
¥ 0.92 Á
Ë 12 ˜¯
2 p
0.95 ¥ 10
I = 1.25
5.25 The dimensions, stator and rotor windings of
a synchronous motor are:
Phases = 3, Frequency = f, No. of poles = P
Rotor length (m) = l1
Axial length (m) = l
Air-gap length (mm) = lg
Rotor winding turns (series) = N1
Rotor winding factor = Kw1
Field current (A) = If
Peak air-gap flux density = Br
Peak rotor mmf = F2
Stator winding turns (series) = N2
Stator short pitching angle = qsp
Peak stator mmf = F1
Torque angle = d
Write METLAB script to calculate
(i) Open-circuit voltage, Voc
(ii) Developed torque Tdev
(iii) Developed power Pdev
5.26 For the synchronous motor of Problem 5.22,
determine
(a) developed torque Tdev as a function of d
5.30
5.31
5.32
5.33
5.34
269
(b) power developed Pdev as a function of d
(c) maximum Tdev and Pdev.
The outside diameter of the rotor of an
alternation is 0.74 and the axial length is
1.52 m. The machine has four poles and the
flux density at the rotor surface is given by
1.12 cos qe where qe = elect. angle.
(a) Find the flux/pole (b) If the peak value
of Fr, is 18000 AT, calculate the permeance/
pole.
A synchronous generator of 50 Hz with
6 poles has a flux/pole of 0.15 Wb. Each stator
coil has two turns and a coil pitch of 150°
elect. Calculate the coil voltage (rms).
Calculate the short-pitching angle to
eliminate the fifth harmonic in the induced
emf of a synchronous generator. What is the
corresponding reduction in the fundamental
and the thirteenth harmonic?
A 50 Hz, 8-pole, pole 3-phase synchronous
generator has 48 slots. Calculate the %
reduction in the fundamental, third and fifth
harmonic strengths on account of distributed
windings.
A synchronous generator has 12 poles and
3-phase winding placed in 144 slots; the coil
span is 10 slots. Determine the distribution
factor, pitch factor and winding factor.
The phase voltage of a 50 Hz synchronous
generator is 3.3 kV at a field current of 10 A.
Determine the open-circuit voltage at 50 Hz
with a field current of 8 A. Neglect saturation.
A 50 Hz, 3-phase hydroelectric generator has
a rated speed of 100 rpm. There are 540 stator
slots with two conductors per slot. The air-gap
dimensions are: D = 6.25 m, L = 1.16 m. The
maximum flux density Bm = 1.2 T. Calculate
the generated voltage/phase.
Calculate the voltage induced in the armature
of a 4-pole lap-wound dc machine having
728 conductors and running at 1600 rpm. The
flux/pole is 32 mWb.
Electric Machines
270
5.35
5.36
5.37
5.38
If this armature carries a current of 100 A,
what is the electromagnetic power and torque
developed?
A 240 V dc motor takes 25 A when running at
945 rpm. The armature resistance is 0.24 W.
Determine the no-load speed assuming
negligible losses. Flux/pole is constant.
A 4-pole dc motor has a lap-connected
armature with 60 slots and 8 conductors/ slot.
The armature has an applied voltage of 240
V. It draws a current of 50 A when running at
960 rpm. The resistance of the armature is 0.1
W. Find the flux/pole that would be necessary
for this operation.
In a given machine F2 (rotor mmf ) 850 AT
and F1 (stator mmf ) 400 AT, a (included
angle) = 123.6° and P ( permeance/pole) 1.408
¥ 10–4 Wb/AT. Find the value of the resultant
air-gap flux/pole.
A P-pole machine has a sinusoidal field distribution as shown in Fig. P5.38. The armature carries a uniform current sheet of value
J A/m causing a triangular mmf distribution as
shown in the figure.
Bp sin q
J
F
2p
0
p
q
d
Fig. P 38
1. What measures are adopted to make the
B-wave in a synchronous machine nearly
sinusoidal? Why should the B-wave be
sinusoidal?
2. A synchronous machine has P poles and
generate voltage of frequency f Hz. Write the
The machine has an axial length of l and a
mean air-gap diameter of D.
(a) Find the peak value of the armature mmf.
(b) Derive an expression for the electromagnetic torque developed.
5.39 A 3-phase, 50 Hz, 4-pole, 400 V wound
rotor induction motor has a stator winding
D-connected and a rotor winding Y-connected.
Assume effective turn ratio stator/rotor = 2/1
( phase basis). For a rotor speed of 1440 rpm,
calculate:
(a) the slip
(b) the standstill rotor induced emf/phase
(c) the rotor induced emf/phase at this speed
(d) the rotor frequency in (b) and (c)
5.40 A 50 Hz induction motor runs at 576 rpm at
full load. Determine:
(a) the synchronous speed and the number of
poles
(b) the frequency of rotor currents
(c) the rotor speed relative to the revolving
field
5.41 A 3-phase induction motor runs at a speed
of 940 rpm at full-load when supplied with
power at 50 Hz, 3-phase.
(a) How many poles does the motor have?
(b) What is its slip at full-load?
(c) What is the corresponding speed of:
(i) the rotor field wrt the rotor surface
(ii) the rotor field wrt the stator
(iii) what is the rotor speed at twice fullload slip?
expressions for its speed in rad (elect.)/s, rad
(mech)/s and rpm
3. Explain the terms, coil span, coil pitch, shortpitching and cording of coils.
4. What is SPP? Write the expressions for a
stator having S slots and P poles.
Basic Concepts in Rotating Machines
5. In a distributed winding why is the phase emf
less than the algebraic sum of phase current in
series?
6. Derive the expression for the breadth factor by
means of a phasor diagram.
7. Repeat Question 6 for the pitch factor.
8. What is the purpose of using short-pitched
coils in ac windings?
9. Write the expression for phasor emf in a
synchronous machine. Use standard symbols
and explain what each symbol stands for.
10. Taking the B-wave to be sinusoidal, derive the
expression for flux/pole.
11. Write the expression for flux linkages of an
N-turn coil if the B-wave is sinusoidal and
rotating at synchronous speed w rad/s.
12. Draw the phasor diagram relating emf phasor
to flux phasor.
13. Sketch the mmf wave of an N-turn coil
carrying current i. Write the expression for its
fundamental if i is sinusoidal. What kind of
wave is this?
14. Write the expression for standing pulsating
p 2p
space wave and sketch it at wt = 0, ,
3 3
and p rad.
15. Rotate the peak value of a rotating magnetic
field and the maximum value of the
fundamental of the mmf space wave of one
phase.
16. State the conditions or a 3-phase winding of
a stator to create a rotating magnetic field and
its speed and direction of rotation.
17. State the conditions for two interacting
rotating fields to create steady torque.
18. Two interacting fields F1 and F2 have a
resultant field Fr . Write the expression for
torque developed in terms of F2 and Fr .
Explain the significance of the angle between
them.
271
19. Write the expression for stator line voltage
of a synchronous machine and show that
it determines the air-gap flux/pole of the
machine.
20. What is the pull-out torque of a synchronous
machine and the meaning of loss of
synchronism?
21. Advance the reason why a synchronous motor
is not self-starting.
22. Explain the process of how an induction motor
develops torque when ac supply is connected
to its stator. Why it cannot develop torque at
synchronous speed?
23. Define slip of an induction motor. At full-load
what is the range of the value of slip.
24. What is the frequency of the rotor currents of
an induction motor?
25. Why an induction motor is called
asynchronous motor?
26. List the type of losses in electric machine.
What is the nature of each loss?
27. What is the relative speed between stator and
rotor rotating fields in an induction motor?
28. Sketch the torque-slip characteristic of an
induction motor. Explain the nature of the low
slip part of the characteristic. Locate on the
characteristic the full-load torque operating
point.
29. Explain how an induction motor can self-start
but cannot run at synchronous speed.
30. Explain why rotor induced emf is proportional
to slip.
31. Distinguish between time phase difference
and space phase difference.
32. State the condition of maximum efficiency of
an electric machine. Compare it with that of a
transformer.
33. When a conducting coil is moving past
a sinusoidal B-wave, what is the relative
position of the coil axis when the induced emf
in it is (1) maximum, and (ii) zero?
272
Electric Machines
5.1 A full-pitched coil in a 6-pole machine has a
mechanical angle span of
(a) 30°
(b) 60°
(c) 90°
(d) 180°
5.2 To eliminate the fifth harmonic a short pitched
coil should have a short-pitching angle of
(a) 36°
(b) 18°
(c) 15°
(d) 12°
5.3 Armature winding of a synchronous generator
can be connected (i) single-phase and (ii)
3-phase. Compare the kVA rating of the
generator in the two cases:
(a) both will have the same kVA
(b) kVA (single-phase) > kVA (3-phase)
(c) kVA (3-phase) > kVA (single-phase)
(d) armature winding cannot be connected in
both the ways stated
5.4 Phase relationship between mmf phasor F
and emf phasor E in a synchronous machine
is
(a) F leads E by 90°
(b) F lags E by 90°
(c) F and E are in phase
(d) this angle depends upon the pf of the load
5.5 A full-pitched coil of Ni ampere-turns placed
in stator slots causes a fundamental mmf wave
of peak amplitude:
(a)
4
(Ni)
p
(c)
4 Ê Ni ˆ
p ÁË 2 ˜¯
p
(Ni)
4
p Ê Ni ˆ
(d)
4 ÁË 2 ˜¯
(b)
5.6 In a dc machine the angle between the stator
and rotor fields is
(a) dependent upon the load
(b) 45°
(c) 90°
(d) 180°
5.7 A 4-pole 50 Hz induction motor runs at a
speed of 950 rpm. The frequency of rotor
currents is
(a) 47.5 Hz
(b) 50 Hz
(c) 5 Hz
(d) 2.5 Hz
5.8 If the rotor of an induction motor is assumed
to be purely resistive, the angle between the
resultant flux density wave and rotor mmf
wave is
(a) dependent upon the load
(b) 180°
(c) 90°
(d) 45°
5.9 In a non-salient pole synchronous machine
the distribution of field mmf around the airgap is a
(a) sinusoidal wave
(b) rectangular wave
(c) stepped triangular wave
(d) flat topped stepped wave
5.10 For a cyclic load variation of a motor the
rating is determined by
(a) average load
(b) the peak load
(c) the rms load
(d) 3/4th of the peak load
AC Armature Windings
6
6.1
273
INTRODUCTION
Chapters 4 and 5 emphasized that field and armature windings are the essential
features of electric machines. The field windings are simple arrangements with
concentrated coils (i.e coils in which all the turns have the same magnetic axis).
Armature windings on the other hand comprise a set of coils (single or multiturn)
embedded in the slots, uniformly spaced round the armature periphery. The emfs
are induced in armature coils due to relative motion between them and the B-wave in the air-gap established
by the field windings. In an ac machine (3-phase) the armature coils are symmetrically connected to form
a set of three balanced phases (equal emf magnitudes with a relative phase displacement of 2p/3 rad). In
a dc machine the armature coils are connected via commutator segments which are tapped by stationary
brushes so as to give a constant (dc) voltage between brushes. It was also seen in Chapter 5 that when the
armature winding carries current, it establishes the same number of alternating (north-south) poles for
which it is wound.
A coil may be of single-turn having two conductors with end connections or multiturn with two coilsides each composed of several conductors. The active coilside (or conductor) length in which the emf is
induced equals the armature length (over which the flux density is established). The pitch of a coil is the
space angle (electrical) between its two sides and must equal an integral number of slots. The coil pitch
may be full (equal to one pole pitch or 180° elect.) or short-pitch (chorded ) coils may be used. The pitch of
a coil could be expressed in terms of its angular span or in terms of slots. The slots/pole must be an integral
number for a full-pitch coil.
Practically there are two types of windings, viz. single-layer and two-layer (or double-layer). In a singlelayer winding each coil—side of a coil occupies the whole slot as shown in Fig. 6.1(a). In a double-layer
winding one coil-side of a coil occupies the upper position in one slot and the second coil-side occupies
the lower position in a slot displaced from the first coil-side by the coil-span as shown in Fig. 6.1(b). In a
double-layer winding each slot is occupied by two coil-sides, one placed on top of the other, referred to as
top and bottom coil-sides. It easily follows from Fig. 6.1 that
where
U
C
C
C
S
= 2 coil-sides/slot
= S/2 (single-layer winding)
= S (double-layer winding)
= number of armature coils
= number of armature slots
(6.1a)
(6.1b)
(6.1c)
The primary difference in single-and double-layer windings is in the arrangement of the overhang. In
single-layer the coils are arranged in groups and the overhang of one group of coils is made to cross the
274 Electric Machines
other appropriately by adjusting the size (corresponding to axial length of armature) and shape of individual
coil groups. This means a variety of coils differing both in size and shape resulting in inconvenience and
higher cost in production. Single-layer windings are, therefore, rarely used in modern machine practice
except in small sizes. Machines are of course still found in use with single-layer windings.
Overhang
Overhang
Bottom coilside
Top coil-side
Coil-span
6 slots (say)
1
2
(a) Single-layer coil
Coil-span
6 slots (say)
7
8
1
2
7
8
(b) Double-layer coil (bottom coilside is
shown dotted)
Fig. 6.1
In a double-layer winding all the coils are identical in shape and size (diamond shape as shown in
Fig. 5.21 is employed) with two coil-sides lying in two different planes. Each slot has one coil-side entering
its bottom half from one side and the other coil-side leaving its top half on the other side. A special kink
at each end of the diamond shape allows neat symmetrical packing of coil overhangs and the problem
of overhang crossing as in a single-layer winding is avoided. Because of identical coils, production is
facilitated and results in a reduction of cost.
DC machine windings are invariably double-layered.
Nomenclature
S = number of slots (must be divisible by 3 in a 3-phase ac machine)
C = number of coils
2C = number of coil-sides
Nc = number of turns/coil
S/P = slots/pole, i.e. pole pitch in terms of slots (is noninteger for fractional slot winding)
180∞P
slot angle (electrical) or slot-pitch (angular displacement between midpoints of adjacent
g=
S
slots)
q = number of phases (generally q = 3)
b = 2p/q, time phase displacement between emfs of successive phases (generally b = 2p/3)
m = number of slots/pole/phase (SPP)
A = number of parallel paths
In ac machines it is possible to have only one path or more paths in parallel-phase (each phase is an
open-circuit); but dc machine windings are always of closed-circuit type with two or more (even) number
of parallel paths.
AC Armature Windings
275
In large dc machines windings may be arranged with more than one coil-side in top and bottom halves
of the slots. Nomenclature specific to dc machine winding is:
U = 2C/S, number of coil-sides/slot (even)
Zs = UNc , number of conductors/slot
Z = 2CNc , total number of armature conductors
Remark
The arrangement of coils round the armature periphery and their interconnections is best illustrated in
form of a winding diagram. For the purpose of drawing a winding diagram, it is convenient to imagine
the armature to be laid out flat in a developed form with slots parallel to each other. Slots under the
influence of each pole can then be marked out; all coil-sides under one pole will have emfs induced in the
same direction with a progressive time phase difference corresponding to the slot angle, Cross-sectional
developed view is also handy in illustrating the underlying ideas of ac windings. In a developed form of dc
winding, the field poles are also indicated.
6.2 AC WINDINGS
Ac windings are generally of a 3-phase kind because of the inherent advantages of a 3-phase machine. The
armature coils must be connected to yield balanced (equal in magnitude and successive phase difference of
2p /3 rad) 3-phase emfs. To begin with the slots around the armature periphery must be divided into phasebands.
Phase Grouping
Initially a simple case will be assumed where SPP is an integral number; such winding is referred to as
integral-slot winding. For illustrative purposes, let m = 2 which means 12 slots per pole pair for a 3-phase
armature. Slot angle is 360°/12 = 30°. Further let the coil-pitch be full six slots. Figure 6.2(a) shows the
12 slots numbered from left to right; six slots are under the influence of one pole with a particular direction
of emfs in coil-sides and the remaining six slots are under opposite pole with opposite direction of emfs as
indicated.
In Fig. 6.2(a) the 12 armature slots are divided into six phase-bands of two (= m) slots each having an
angular spread of s = 60°; in fact each pole is divided into three band (as many as the number of phases).
If coil-sides in slots (1, 2) belong to phase band A, those in slots (5, 6) which are 120° (or four slots) away
belong to phase band B and those in slots (3, 4) are 60° (or two slots) away from A which when reverseconnected would belong to phase C. (See phasor diagram of Fig. 6.2(b).) Therefore coil-sides in slots-(3,
4) are said to belong to phase band C¢. As a result of this arrangement the phase-band sequence is AC¢ BA¢
CB¢ which will repeat for each pair of poles. This arrangement of phase-band is called 60° phase grouping.
The four coil-sides of each pair of coils of a phase can be connected additively in any order. For example,
the order of the coil-side connection for phase A could be (1–8–2–7) as used in a single-layer winding with
concentrated coils or it could be (1–7–2–8) in a two-layer lap winding (these are explained soon after). The
phasor diagrams of all the three phases for the former kind of connection is given in Fig. 6.2(c).
The 60° phase grouping discussed above can be used for single-layer or double-layer windings. It is also
possible to use a 120° phase grouping where the slots under a pole pair are divided into three phase-bands
276
Electric Machines
as in Fig. 6.2(d). For the example in hand there are four slots per phase-band. It is obvious that slots for
return coil-side for this phase-grouping will not be available in single-layer winding. It can only be used for
a double-layer winding. A phase-grouping of 120° is rarely adopted and will not be discussed any further.
1
3
2
s = 60°
A
Pole
4
C¢
5
6
7
8
Pole
9
10
11
12
+
+
+
+
+
+
B
A¢
C
(a) Phase-bands 60° phase grouping
B¢
A
–7
A
2
120°
–8
C¢
60°
1
120°
–4
9
B
C = –C¢
10
5
6
C
–11
(b)
3
2
A
B
60° phase grouping
C¢ reverse connected
1
–12
–3
(c)
4
5
6
7
8
9
10
11
12
+
+
+
+
+
+
B
C
120° phase grouping
Fig. 6.2
Single-layer Windings
Single-layer windings are not commonly used in practice except for machines of a few kW because of the
disadvantages mentioned earlier. Single-layer winding may be concentric, lap or wave type. Here only the
concentric type winding will be illustrated while the lap type will be explained in two-layer winding. Wave
winding because of certain problems in end connections is not used in ac machines.
Concentric Windings
Concentric windings may be classified into two main categories, viz. unbifurcated (or half-coiled) and
bifurcated (whole-coiled). In the former type the coils comprising a phase group in adjacent pole pitches are
concentric as indicated in Fig. 6.3(a). The individual coils may have a span greater or less than a pole pitch
but the average coil-span equals one pole-pitch. This kind of arrangement is provided to avoid crossing of two
coils under one phase-group. In bifurcated winding, each coil group is split into two sets of concentric coils
and the return coil-sides are shared with those of another group as shown in Fig. 6.3(b). It is clearly evident
from the figure that this kind of arrangement is only possible when SPP is even.
AC Armature Windings
277
It is easily seen from Figs 6.3(a) and (b) that for accommodating the windings for all the three phases, the
overhang must be arranged in two or three planes. Figure 6.4(a) which corresponds to unbifurcated winding
(Fig. 6.3(a)), the overhang is arranged in a continuous chain with sequence (if seen from a fixed reference)
A ≠ B Ø C ≠ A Ø B …, where upward and downward arrows indicate the upper and lower planes. The three
plane overhang arrangement of a bifurcated winding is depicted in Fig. 6.4(b).
A
C
B
Concentric coils
Core
Pole
pitch
(a) Unbifurcated winding with two-plane overhang (continuous chain)
(a) Unbifurcated winding for one phase; S/P = 6, m = 2
Pole
pitch
A
B
C
Pole
pitch
Core
(b) Bifurcated winding with three-plane overhang
(b) Bifurcated winding for one phase; S/P = 6, m = 2
Fig. 6.3
Single-layer winding with concentric coils
Fig. 6.4
Arrangement of overhang in single-layer
concentric winding
Single-layer coils can be arranged in semi-closed slots (the coil is opened and pushed in slots from one
side, the coil then being reformed and reconnected by buff-welding).
Chording and the use of fractional SPP is not possible in a single-layer winding. As will be seen in the next
section, it is a serious drawback.
Double-layer Windings
Double-layer windings are the most widely used class of windings. Though both lap and wave types are
possible, because of inherent problems of a wave winding*, it is now an accepted practice to use the lap type
* Figure 6.5 shows a double-layer wave winding with single-turn coils.
In this type of winding, after traversing the armature once the winding closes on to the start of the first coil (i.e. after connecting P/2
coils in series). To overcome this difficulty, the connection is made to
the second coil-side of the first phase group and a similar procedure
is continued until all the coils are exhausted. However, in the case
of fractional slot windings this problem is even more complicated
because, after all the turns around the armature are completed, some
coils remain unconnected.
2 pole
pitches
Front
pitch
Coil
span
Lower
Upper
Fig. 6.5
278 Electric Machines
for double-layer ac winding. Double-layer windings fall into two main classes depending upon the value
of SPP-integral slot winding when SPP is an integer and fractional slot winding when SPP is a fractional
number. To meet the requirement of symmetry among phases, the number of slots/phase (S/3) must be a
whole number.
Integral Slot Winding
Here SPP is an integer. This type of winding has already been illustrated in Fig. 5.22. The winding arrangement
is further illustrated through an example. Let
m = 2 slots
and
S/P = 6 slots
Coil-pitch = 6 slots (full-pitch coils)
Phase spread, s = 60° elect.
The winding diagram for one phase is shown in Fig. 6.6. The first set of phase-group coils (coil-group 1)
lying under one pole-pair (NS) are connected in series (finish end of the first coil is connected to the start of
the next coil lying to the right of the first). The second coil-group of the phase lies under SN poles and must
therefore be connected in reverse to the first coil-group for additive emf. It may be noticed that alternate coilgroups are reverse connected. It is observed that the winding appears like a bifurcated one. It is also observed
that coil-sides lying in any given slots pertain to the same phase. All the coil-groups of the phase could be
connected in series or in series-parallel.
1
2
3
A2
A1
Fig. 6.6
4
Double-layer lap winding; 4 poles, m = 2, s = 60° coil-pitch = 6 slots ( full-pitch)
In practice, however, it is common to use chorded or short-pitched coil. As already mentioned in
Ch. 5, this type of arrangement offers certain inherent advantages such as reduction in copper needed for
end connections. Further, certain harmonics present in the emf wave are greatly suppressed. However, coil
chording lead to a reduction in the emf generated (refer Eqs (5.18) and (5.20). This type of winding is best
illustrated by means of a cross-sectional developed diagram. Here letters (a, b, c) refer to the coil-sides of the
corresponding phase. For the example considered,
S/P = 9 slots
and
m=3
Coil-span = 8 slots (chorded by one slot)
= 60° electrical
AC Armature Windings
279
The cross-sectional view of winding for one pole-pair is drawn in Fig. 6.7. Certain observations can be
made from this figure. The top bottom coil-side phase grouping is merely displaced by one slot (equal to
chording). Further, in each-group of three slots, the coil-side of two different phases are placed in one of the
slots.
Pole pitch
Pole pitch
a a a c¢ c¢ c¢ b b b a¢ a¢ a¢ c c c b¢ b¢ b¢
Top
Bottom
a¢ a¢ c¢ c¢ c¢ b b b a¢ a¢ a¢ c
c
c b¢ b¢ b¢ a
Coil-sides of different phases in same slot
Fig. 6.7 Cross-sectional view of two-layer winding: m = 3; s = 60°, coil-pitch = 8 slots (chorded by one slot)
Fractional Slot Windings
So far these types of windings have been studied in which SPP is an integer. Windings, wherein SPP is a
fractional number, are known as fractional slot windings. Fractional slot winding is easily adopted with a
double-layer arrangement.
While m = S/3P is a noninteger in a fractional slot winding, S, the number of slots, must be divisible by
3, i.e. slots per phase must be integral in order to obtain a symmetrical 3-phase winding. The pole pitch, S/P,
is also fractional, so that the coil-span cannot be of full-pitch. For example if S/P = 10.5, then the coil-span
can be either 11 or 10 slots. The coil-span chosen is of course less than the full-pitch because of the inherent
advantages of chording elaborated earlier.
In the previous discussion it has been learnt that winding constituting a basic unit under a pole-pair
(N and S ) is repeated for any number of pole pairs when m is integral. In order to obtain the “basic unit” for
fractional slot winding, S/3P is reduced to the irreducible fraction by cancelling out the highest common
factor in S and P. Thus
S¢
S
=
(6.2)
3P ¢
3P
Coil-sides under P¢ poles constitute the basic unit whose connections will be repeated P/P¢ times in the
complete winding. It will soon be evident that P cannot be a multiple of three in a 3-phase winding.
In a double-layer winding, the phase-grouping of coil-sides for the top layer is repeated in the bottom layer
with corresponding coil-sides being located one coil-span away. Therefore, all that is needed is to establish
the phase-grouping of top layer of coil-sides.
Fractional slot winding is easily understood by means of an example Let
m=
S = 108,
then
m=
P = 10
3
S
108
=
= 3 slots
5
3P
3 ¥ 10
108
54
S¢
S
=
=
=
3 ¥ 10
3¥5
3P ¢
3P
Thus the basic unit in the winding has 5 poles (note that this need not be even) covering 54 slots.
A coil-group table (Table 6.1) is now perpared on the following lines.
Now
280
Electric Machines
Table 6.1
Slot No.
1
Angle
0
Phase
[a]
2
2
16
3
a
3
1
33
3
a
4
5
50
a
2
66
3
c¢
6
1
83
3
c¢
7
100
c¢
8
9
10
2
116
3
c¢
1
133
3
b
11
150
19
20
166
2
3
b
b
21
1
153
3
b¢
22
2
136
3
b¢
170
31
32
33
Pole-ptich 1
Slot No.
Angle
Phase
12
1
3
3
a¢
13
20
a¢
14
1
36
3
a¢
15
1
53
3
a¢
16
70
c
17
2
86
3
c
18
1
103
3
c
120
[b¢]
b¢
Pole-Pitch 2
Slot No.
Angle
Phase
23
6
2
3
a
24
1
23
3
a
25
40
a
26
56
2
3
a
27
1
73
3
c¢
28
29
90
106
c¢
c¢
30
1
123
3
b
40
41
2
3
140
156
2
3
173
b
b
b
43
44
2
3
Pole-pitch 3
Slot No.
34
Angle
10
Phase
a¢
35
2
3
a¢
26
36
1
43
3
a¢
37
38
60
76
[c]
c
2
3
39
1
93
3
c
c
b¢
42
1
143
3
b¢
51
1
113
3
c¢
52
53
110
126
2
3
160
b¢
176
2
3
b¢
Pole-ptcih 4
Slot No.
Angle
Phase
45
1
13
3
a
46
30
a
47
2
46
3
a
48
1
63
3
c¢
49
80
c¢
50
2
96
3
c¢
130
b
2
146
3
b
54
1
163
3
b
Pole-pitch 5
1. Calculate the slot angle,
10 ¥ 180∞
50
2∞
=
= 16
(Keep in fractional form)
108
3
3
2. Beginning with 0°, calculate the angle for serially arranged slots in a table. Every time the angle
exceeds 180°, subtract 180°, the angle of one pole-pitch. This takes care of the fact that the positive
direction of emfs under adjacent poles are in opposite direction.
3. Phase-group A is located corresponding to:
g=
0 £ angle < 60°
Phase-group C:
60° £ angle < 120°
Phase-group B:
120° £ angle < 180°
Phase-group ordering is AC¢ BA¢ CB¢ º
AC Armature Windings
281
4. The starting points of phases are located at 0°, 60° and 120°. While 0° is the start, the other two angles
are located at S¢/3 = 18 and 2S¢/3 = 36 slots away.
g S¢
180 P¢ S ¢
=
¥
= 60 P¢
3
S¢
3
180 P¢ 2 S ¢
2g S¢
¥
Angle 2S¢/3 slots away =
=
= 120 P¢
S¢
3
3
Angle S¢/3 slots away =
These angles are 60°, 120° or their multiples for any P¢ except when P¢ is multiple of 3 (= number of
phase). Thus if P¢ (or P) is multiple of 3, balanced winding is not possible.
It is immediately seen from the last rows of Table 6.1 that there are three cycles of (4, 4, 3, 4, 3) slot
distribution starting at slot no. 1(0°, phase A), slot no. (l + S¢/3) = 19 (120°, phase B) and slot no. (1 + 2S¢/3)
= 37 (60°, phase C ). So this pattern results in a balanced winding.
Figure 6.8 shows the layout of the winding for the basic unit of five poles indicating the number of coils
and their phase-groups. The dashed phase-groups are reverse connected. The start of all the three phases
is made at the start of an undashed coil-group. This winding pattern would repeat for every 5-pole unit. It,
therefore, represents the winding diagram for 10, 20, º pole armature.
4(a) 4(c¢) 3(b) 4(a¢) 3(c) 4(b¢) 4(a) 3(c¢) 4(b) 3(a¢) 4(c) 4(b¢) 3(a) 3(c¢) 3(b)
A
B
C
Fig. 6.8
Though the fractional slot winding may appear to be somewhat complicated, it has certain technical
advantages and can be easily manufactured. The number of armature slots chosen need not necessarily be an
integral multiple of the number of poles. Consequently one may choose a particular number of slots for which
the notching gear is available. This results in saving in machine tools. This flexibility can be effectively used
where the number of poles (machine speed) varies over a wide range as in the case of synchronous machines.
Furthermore, the high-frequency harmonics caused by slotting are considerably reduced by the use of
fractional slot winding. As the poles move past a slotted armature, the flux/pole fluctuates (does not move) as
the slot-teeth pattern facing a pole repeats. This causes induction (static) of emf harmonics of high frequency.
If S/P is integral, the disposition of armature slots relative to pole simultaneously repeats (in time phase) at
every pole so that the harmonics in phase-groups of a given phase are in phase. Nonintegral S/P causes the
harmonics in phase-group to become out-of-phase thereby reducing their strength in the phase voltage.
The generation of harmonics of slot origin is explained in detail below for the interested reader.
Tooth Ripple
Because of lower reluctance of the magnetic path corresponding to the teeth as compared to the slots, the flux
density wave in the air-gap is rippled as shown in Fig. 6.9. Unlike the space harmonics of the B-wave which
move at the same peripheral speed as the fundamental, the tufts of flux embracing the teeth move back and
282
Electric Machines
forth as the teeth move relatively past the pole-shoes. Since there is no net movement of B-wave ripples, no
space harmonic is produced by these. However, due to cyclic variation in reluctance of the magnetic path
offered to the field poles by the toothed armature, the total flux/pole contains a space stationary but timevarying component. This effect is most pronounced when the pole-arc covers an integral plus one-half slotpitches as illustrated in Figs 6.10(a) and (b). The relative position of the armature and poles in Fig. 6.10(a)
corresponds to low reluctance and maximum flux/pole and Fig. 6.10(b) corresponds to high reluctance and
minimum flux. It is thus seen that the flux has a stationary time-varying component. A complete cycle of
flux variation occurs when the pole moves through one slot-pitch, while one cycle of the fundamental wave
is generated when the poles move through two pole-pitches. Therefore, the tooth ripple frequency in flux is
Ê 2S ˆ
fs = f ¥ slots/pole-pair = Á ˜ f
Ë P¯
where
f = fundamental frequency
fs = frequency of flux variation due to slotting
3 21 slot-pitches
Tufting of flux
Pole
(a) Four teeth opposite field pole (low reluctance, maximum flux)
Armature
Ripple
(b) Three teeth opposite field pole (high reluctance, minimum flux)
Fig. 6.9 Tooth ripple in B-wave
Fig. 6.10
Flux linking an armature coil can be expressed as
f = (F + fs sin 2pfs t) cos 2p ft
and the emf due to this flux is
df
= 2p fNF sin 2p ft
dt
1
– Nfs [2p ( fs – f ) cos 2p ( fs – f )
2
+ 2p ( fs + f ) cos 2p ( fs + f )]
e = –N
(6.3)
Thus, apart from the fundamental, the two frequencies present in the emf wave owing to armature slotting
are
Ê 2S ˆ
± 1˜
( fs ± f ) = f Á
(6.4)
Ë P
¯
AC Armature Windings
283
If S/P = 9, then
( fs ± f ) = 50(18 ± 1) = 950, 850 Hz
These frequencies are high enough to cause interference in telecommunication in the neighbourhood of
power lines.
If S/P is fractional, the space relation between the slotted armature and a given field pole is not the same
as in succeeding poles as a result of which the emfs of ripple frequencies ( fs ± 1) in various armature coils
become out-of-phase and tend to cancel out in an interconnected set of phase coils.
The pitch of a coil is the space angle (electrical) between its two sides and must equal an integral
number of slots.
Two types of winding
- Single layer: each coil side of a coil occupies the whole slot
- Double layer: each slot is occupied by two coil sides
Winding diagram is used to illustrate the arrangement of coils round the armature periphery and their
interconnections.
Fractional slot winding is used to reduce the high frequency harmonics caused by slotting.
In lap winding the finish of one coil is connected to the start of the adjoining coil.
Wave winding the finish end of one coil under one pole pair is connected to the start of a coil under the
next pole pair.
6.1 Draw a single-layer unbifuracated winding for
a 3-phase, 4-pole machine having 24 armature
slots. Assume one coil-side. Cleary show the
end connection of a continuous chain arrangement is used.
6.2 For the same number of slot and poles as
in Problem 6.1, draw a bifurcated type of
winding. If the number of slots are changed
from 24 to 36, is it possible to have bifurcated
winding? If not, why?
6.3 Give a developed view of double-layer armature winding for a 3-phase machine with
6 poles and 36 slot, If full-pitched coils are
used. Indicate all end connection and the start
and finish of each phase.
6.4 A 3-phase machine has 4 poles and 48 armature
slots. If the coils are chorded by one slot, draw
the double-layer winding diagram for all three
phases. Why is it that chording more than
1/3 pole-pitch is not used in practice?
6.5 The armature of a 3-phase machine with
16 poles and 180 slots is wound with fractional
slot windings. Construct the winding table
for one basic unit of poles. Indicate the start
of each phase. For the basic unit determine
the distribution of coil groups and the phase
sequence.
6.6 A 3-phase, 10-pole machine has 72 slots.
Consturct the winding table for fractional slot
winding. Draw the winding diagram with a
coil-span of seven slots.
6.7 A 3-phase, 50-Hz, 10-pole machine has
120 armature slots. What harmonic
frequencies will be present in the generated
emf on account of slotting ? How do these
affect the operation of the machine?
284
Electric Machines
1. What is the significance of a winding diagram?
2. When do you use concentric winding?
3. What are the advantages of fractional slot
winding over integral slot winding?
4. Compare lap and wave winding. Where each
type is used and why?
5. Why double layer winding is preferred?
6. Explain how fractional winding reduce the
emfs of ripple frequencies.
DC Machines
7
7.1
285
INTRODUCTION
A dc machine is constructed in many forms and for a variety of purposes, from
the 3 mm stepper drawing a few mA at 1.5 V in a quartz crystal watch to the
giant 75000 kW or more rolling mill motor. It is a highly versatile and flexible
machine. It can satisfy the demands of load requiring high starting, accelerating
and retarding torques. A dc machine is also easily adaptable for drives with a wide
range of speed control and fast reversals.
DC motors are used in rolling mills, in traction and in overhead cranes. They are also employed in many
control applications as actuators and as speed or position sensors. With ac being universally adopted for
generation, transmission and distribution, there are almost no practical uses now of the dc machine as a
power generator. Its use as a motor-generator (ac motor-dc generator) for feeding dc drives has also been
replaced in modern practice by rectifier units. In certain applications dc motors act as generators for brief
time periods in the “regenerative” or “dynamic braking” mode, especially in electric traction systems.
The basic principles underlying the operation and constructional features of a dc machine were discussed
in Sec. 5.2 (refer to Fig. 5.13) while the emf equation was given in Eq. (5.26). It was stated there that the
field winding (concentrated type) is mounted on salient-poles on the stator and the armature winding
(distributed type) is wound in slots on a cylindrical rotor. Constructional features of a practical machine
are brought out by half cross-sectional views of Figs 7.1 and 7.2 wherein all important machine parts are
named. Both small and large industrial machines have generally the conventional heteropolar cylindrical
rotor structure, although some unconventional homopolar machines have also been devised.
The magnetic circuit of a dc machine consists of the armature magnetic material (core), the air-gap, field
poles and yoke as shown in Figs 5.13 and 7.2. The yoke of a dc machine is an annular ring on the inside
of which are bolted field poles and the interpoles. The interpoles or commutation poles are narrow poles
fixed to the yoke, midway between the main field poles. Interpoles and compensating windings, which
will be described later in this chapter in connection with commutation problems, are required to be excited
suitably.
The use of an electric field winding, which supplies electric energy to establish a magnetic field in the
magnetic circuit, results in the great diversity and a variety of performance characteristics. The armature
winding is connected to the external power source through a commutator-brush system (see Fig. 7.1
item 6), which is a mechanical rectifying (switching) device for converting the alternating currents and
induced emfs of the armature to dc form. Figure 7.4(a) shows a single commutator segment and Fig. 7.4(b)
is the cross-sectional view of a built-up commutator. The longitudinal and perpendicular to the machine
axis cross-sectional view of a dc machine, indicating the location and nomenclature of machine parts are
presented in Figs 7.1 and 7.2.
286
Electric Machines
DC Machines
5
9
2
3
4
7
8
6
1
Fig. 7.1
1. Armature Core
2. Main Field Pole
3. Interpole
Sectional view of a dc machine
4. Main Pole Winding
5. Interpole Winding
6. Commutator
7. Brush and Brush Holder
8. Armature Winding Overhang
9. Fan
Interpole
Yoke
Interpole coil
Main field
coil
Main pole
Pole shoe
Inter pole
Armature
slots
Commutator
Armature
Fig. 7.2 Cross section of a dc machine
DC Machines
287
The cylindrical-rotor or armature of a dc machine is mounted on a shaft which is supported on the
bearings. One or both ends of the shaft act as input/output terminal of the machine and would be coupled
mechanically to a load (motoring machine) or to a prime-mover (generating machine). Usually parallelsided axial slots (evenly spaced normally) are used on the rotor surface. In these slots armature coils
are laid as per winding rules. The magnetic material between slots are the teeth. The teeth cross-section
influences significantly the performance characteristics of the machine and parameters such as armature coil
inductance, magnetic saturation in teeth, eddy-current loss in the stator poles and the cost and complexity
of laying armature winding.
The design of electrical machines has become a very interesting and challenging topic and is continuously
changing with new and improved magnetic, electrical and insulating materials, the use of improved heattransfer techniques, development of new manufacturing processes and the use of computers. There are
full-fledged excellent texts [9, 46] dealing with the design aspects. The objective of this chapter is to
analyse the behaviour of the dc machine in detail and present the physical concepts regarding its steadystate performance.
7.2 ARMATURE WINDING AND COMMUTATOR
A dc machine is a heteropolar structure with stationary
poles and the rotating armature (Fig. 5.13). An alternating
emf of the same wave shape as that of B-wave is induced in
every coil. As the armature rotates, the emfs induced in the
belt of coil-sides under a given pole is unidirectional and
the pattern alternates from pole to pole as shown in Fig. 7.3
for a 4-pole machine.
The coil side current pattern is the same as the emf
pattern. The only difference is that while the coil-side emf
reduces towards the outer side of poles, the current remains
the same in all the coil-sides except for alternations from
pole to pole, while the coil-side current reverses, the current
exchanged with external circuit must be unidirectional and
voltage must be constant and of same polarity (d.c.). This is
the rectification process which is carried out by mechanical
rectifier comprising commutator-brush assembly.
EMF
S
N
Total current
of both layers
N
S
Fig. 7.3
4-pole dc machine
Commutator-Brush Assembly
The commutator is a cylindrical assembly of wedge-shaped copper segments (Fig. 7.4(a)) insulated from one
another and the shaft by thin mica or micanite sheets. In high-speed machines the segments are so shaped
that they can be clamped by two cast-iron V-shaped rings as shown in Fig. 7.4(b). Each commutator segment
forms the junction between two armature coils (“finish” of one coil and “start” of the other). In large machines
flat copper strips known as risers are used forming clip connections to armature bar conductors (Fig. 7.4(b)).
A double-layer winding is universally adopted in dc machines. The coils are continuously connected
“finish” to “start” to form a closed (re-entrant) winding. Depending upon the type of connection (lap or
wave), pairs (one or more) of parallel paths exist.
288 Electric Machines
Riser
Commutator segment
V-ring
V-ring
Locking
ring
Mica
Riser
(b) Commutator segment
(a) Commutator assembly
Fig. 7.4
Stationary carbon brushes are placed in contact with the commutator under spring pressure; see item 7 of
Fig. 7.1. The brushes are electrically placed in the magnetic neutral regions where the armature coils have
almost zero induced emf. Because of the diamond shape of coils, the brushes are physically placed opposite
midpoles. With this placement of brushes, the commutator segment contacted is either fed current from both
coil-sides connected to it or it feeds current to both the coil-sides. Thus, at one brush the current constantly
flows out and at the next brush the current flows in. This occurs at all brush pairs. The adjoining brushes are
at constant dc voltage and the coils in series between the two constitute one parallel path. As the armature
rotates, the number of coils in series tapped by the brush pairs remains constant and also their disposition
relative to the poles is the same. As a result constant (dc) voltage appears across brush pairs. As a coil crosses
the brush position, the current in it must reverse which is the commutation process.
Armature Winding Commutator Connections*
For elementary explanation, cross-sectional view of armature winding and commutator may be employed.
As only conductor cross-sections appear in the diagram, clarity does not emerge with out lengthy detailed
account and general winding rules get left out.
We shall adopt the alternate powerful method of drawing the developed diagrams of windings by the use
the general winding rules.
* These who do not want to study winding rules may skip these and follow the study as under
Lap winding diagram Fig. 7.8; trace out parallel paths
Ring diagram Fig. 7.9; commutation
Wave winding diagram Fig. 7.12. Go to Fig. 7.13 the coils in the two parallel paths at an instant. Trace out the
parallel paths
Then go to section 7.3
DC Machines
289
Developed diagram Imagine that the machine (dc) is cut out axially and laid out on a plane. The poles will
appear underneath the armature winding with coil ends suitably connected to the commutator segments. This
is not a scaled version of the machine but a schematic representation of the poles, armature winding and
commutator segments; see Fig. 7.8.
Coil-side Numbering Scheme
Coil-sides are numbered continuously—top, bottom, top, …. The first top coil-side is numbered 1 so that
all top ones are odd and all bottom ones even. Figure 7.5 shows the coil-side numbering scheme for U =
4 coil-sides/slot.
1
1 3
2
5 7
2
6
4
8
3
9 11
4
13 15
5
17 19
10 12
14 16
18 20
Ycs = 4 slots
Fig. 7.5
Coil-pitch/Back-pitch
In Fig. 7.5 the coil-span Ycs is assumed 4 slots (or the
coil spans 4 teeth). Therefore, coil-side 1 will form a
coil with coil-side 18 and 3 with 20, and so on. This
is depicted in Fig. 7.6. The coil-span in terms of coilsides is ycs = 18 – 1 = 17 or 20 – 3 = 17. This indeed
is the distance measured in coil-sides between two
coil-sides connected at the back end of armature (end
away from commutator) to form a coil. It is known
as back-pitch denoted as yb. Obviously coil-span
Back end
3
2
ycs = yb
19
17
18
4
20
Front end
(commutator end)
'Start'
Ycs = yb = 17 (odd)
'Finish'
Fig. 7.6
which is odd in this case and must always be so. It indeed equals
or in general
where Ycs = coil-span in slots
yb = 4 ¥ 4 + 1 = 17
yb = UYcs + 1 (odd because U is even)
(7.1)
Commutator-pitch
The junction of two coils (“finish” - “start”) is connected to one commutator segment. Therefore,
Number of commutator segments = C (number of armature coils)
(7.2)
The number of commutator segments spanned by the two ends of a coil is called commutator-pitch, yc. In
Fig. 7.7(a), yc = 2 – 1 = +1.
290
Electric Machines
Coil-span
The coil-span in terms of slots is always nearly full-pitch. This ensures that coil-side voltages around the coil
are additive most of the time (except when coil-sides lie near the magnetic neutral region). Thus
S
(nearest lower integer)
P
which means that for nonintegral S/P, the coils are short-pitched.
Ycs =
(7.3)
Lap Winding
In a lap winding (as in case of ac) the “finish” of one coil (coming from the bottom coil-side) is connected
to (lapped on) the “start” of the adjoining coil as illustrated for single-turn coils in Fig. 7.7. The coil-side
displacement of the front end connection is called the front-pitch, yf . In a lap winding the resultant-pitch
yr = yb ~ yf = 2
(7.4)
The direction in which the winding progresses depends upon which is more, yb or yf . Thus
yb > yf
yb < yf
(Progressive winding, Fig. 7.7(a)
(Retrogressive winding, Fig. 7.7(b)
(7.5(a))
(7.5(b))
There is not much to choose between progressive or retrogressive winding; either could be adopted.
Winding progresses
(1 + Yb)
Back end
1
2
Yr
Yb
3
4
Yf
Front end
(1 + Yr)
Yc = +1
C
1
2
3
Commutator
(a)
Winding retrogresses
–1
Yr
0
(1 + Yb)
Yb
1
2
Yr
(1 – Yr)
Yc = –1
C
1
(b)
Fig. 7.7
2
DC Machines
291
As shown in Figs 7.7(a) and (b), the two ends of a coil are connected across adjacent commutator segments.
Thus commutator-pitch,
yc = ±1 (+1 for progressive, –1 for retrogressive)
(7.6)
Coils in lap winding are continuously connected as per the above rule and in the end it closes onto itself
(as it must). In the process all coils have been connected.
To learn certain further aspects of lap winding—location of brushes, etc., an example is worked out.
EXAMPLE 7.1 Draw the lap-winding diagram in the developed form for a 4-pole, 12-slot armature with
two coil-sides/slot. Assume single-turn coils.
Indicate the number and position of brushes on the commutator. What is the number of parallel paths?
SOLUTION
C = S = number of commutator segments = 12
12
= 3 slots
4
Ycs =
Yb = 2Ycs + 1 = 7
yf = 7 – 2 = 5 (we choose progressive winding)
This information is sufficient to draw the developed winding diagram of Fig. 7.8. The ends of coil formed by coilsides (1-8) are connected to commutator segments 1, 2, and so on. The four poles are shown on the developed diagram.
All coil-sides under one pole have emf induced in the same direction and the pattern alternates. The arrows on the coil
sides indicate the direction of current flow. If the armature moves left to right, the emfs in coil sides are induced in the
same direction as currents, the machine is generating, supplying power to the external circuit. On the other hand, if the
armature moves right to left, the induced emfs are in opposite direction to currents, the machine is motoring; receiving
power from the supply.
Equalizer
Parallel path
From 23
To 4
From 21
To 6
From 20
To 2
1
2
3
4
5
6
7
8
9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24
1
1
2
2
To 23
To 21
Ic =
From 2 From 4
Ia
4
11
12
1
3
2
4
–
B1
Ia/2
Ia
A1
Fig. 7.8
5
B2
6
7
+
8
9
10
–
B3
11
12
B4
Ia/2
Ia/2
Ia/2
A2
Lap winding for 4-poles, 12-slot armature (single-turn coils, 2 coil-sides/slot)
292
Electric Machines
Parallel Paths/Brushes
It is easily found from the winding diagram that three coils (C/P = 12/4) are located under one pole-pair
(N1S1) and are series connected so that their emfs add up. This constitutes one parallel path. The complete
winding can be divided into four such parallel paths lying under four different pole-pairs (N1S1, S1N2, N2S2,
S2N1). It is, therefore, concluded that the number of parallel paths (A) in a lap-wound machine in general
equal the number of poles (P), i.e.
IaI2
A=P
(7.7)
The four parallel paths in the winding
B1
Ia
+
of Fig. 7.8 electrically form a close ring
E1(N1S1)
E4(S2N1)
A1(+)
as shown in Fig. 7.9 in which the parallel
path emfs and currents around the loop
–
B4
Ic = (Ia/4)
alternate.
–
B2
The ends of parallel paths meet at
commutator segments 1, 4, 7 and 10 at
E2(S1N2)
E3(N2S2)
the instant shown. These are the locations
A2(–)
B3
+
of the brushes (equal to the number of
IaI2
poles) and are alternately positive and
IaI2
negative. It is also found that the brushes
are physically located opposite the pole
Fig. 7.9 Equivalent ring diagram of 4-pole, lap-wound armature
centres and the electrical angle between
them is, therefore, 180°. The spacing between adjacent brushes in terms of the commutator segment is
C
12
=
=3
(7.8)
P
4
It may also be noted that C/P need not necessarily be an integer. It is further noticed that because of the
diamond shape of coils, the brushes which are physically opposite the pole centres are electrically connected
to coil-sides lying close to the interpolar region. Thus electrically the brushes are displaced 90° elect. from
the axes of the main poles.
The two positive and two negative brushes are respectively connected in parallel for feeding the external
circuit.
From the ring diagram of Fig. 7.9, which corresponds to the winding diagram of Fig. 7.8, it immediately
follows that the current in armature conductors is
Ic =
Commutation
Ia
A
(7.9)
Consider any parallel path, say the one tapped at commutator segments 1 and 4. As the armature rotates, one
coil moves out of this parallel path at one brush and another coil enters the parallel path at the other brush. The
brush pair now taps commutator segments 2 and 5. This process happens simultaneously at all the brushes
and can be more easily imagined from the ring diagram of Fig. 7.9 wherein the coils can be considered to
rotate in a circular fashion. In this way a brush pair always taps C/P coils (in series) and as a consequence the
voltage available at each brush pair is maintained constant (dc). This indeed is the commutation action. As
a matter of fact the voltage tapped varies slightly for a brief period when the changeover of coils in parallel
paths takes place. However, this voltage variation is negligible in practical machines which have a large
number of coils.
DC Machines
293
It easily follows from the winding diagram of Fig. 7.8 and from the equivalent ring diagram of Fig. 7.9 that
as a coil moves out of one parallel path into another, the current in it must reverse. In Fig. 7.8 as the armature
moves by one commutator segment, currents in four coils—(l, 8), (7, 14), (13, 20) and (19, 2)—must reverse.
These coils are said to undergo current commutation. It is also observed that during the brief period in which
a coil undergoes commutation, its coil-sides are passing through the interpolar region so that negligible emf
is induced in the commutating coil. At the same time, in this period the coil remains short-circuited by the
brush, bridging the adjoining commutator segments to which the coil ends are connected. Ideal commutation
in which the conductor current change from +Ic to –Ic is sketched in Fig. 7.10(a). If the current in a coil
does not reverse fully at the end of commutation period, there will be sparking at the brush contact. This
phenomenon and its remedy will be discussed in a Section 7.8
Commutation
+Ic
Time to
travel one
pole
Time
–Ic
Commutation
time
Fig. 7.10(a) Ideal Commutation
Symmetry Requirement
To avoid no-load circulating currents and certain consequential commutation problems, all the parallel paths
must be identical so as to have the same number of coil-sides. Symmetry therefore requires that
Remark
2C
US
=
= integer
P
P
To practically wind a dc armature the above winding rules are
not needed except that the coils of the double-layer winding are
to be continuously connected from “finish” to “start” till the
winding closes onto itself. Further the “finish-start” junction is
connected to the commutator segment physically opposite the
midpoint of the coil; the ends of each coil being connected to the
adjacent commutator segment. This is illustrated in Fig. 7.10(b).
Of course the winding rules given above help in providing an
insight to the reader into the winding and commutator action to
produce dc at the brushes.
(7.10)
Fig. 7.10(b)
Equalizer Rings
The poles in a dc machine cannot be made identical so as to have the same value of flux/pole. Any dissymmetry
among poles leads to inequality in parallel path emfs e.g. in Fig. 7.9, E1(N1S1), E2(S1N2), E3(N2S2) and
294
Electric Machines
E4(S2N1) will not be identical. As a result the potential of the positive and negative brush sets are no longer
equal so that circulating current will flow in the armature via the brushes to equalize the brush voltage even
when the armature is not feeding current to the external circuit. Apart from causing unbalanced loading
around the armature periphery when the armature feeds current, the circulating currents also interfere with
“commutation” resulting in serious sparking at the brushes.
The circulating currents even out pole dissymmetry by strengthening the weak poles and by weakening
the strong poles. The remedy is therefore to allow the circulating currents to flow at the back end of the
armature via low resistance paths inhibiting thereby the flow of these currents through carbon brushes which
in comparison have considerably higher resistance. This remedy is applied by connecting several sets of
points which would be “equipotential” but for the imbalance of field poles via equalizer rings. Currents
flowing in the rings would of course be ac as only ac voltage exists between points on coil back ends.
Equipotential points are 360° (elect.) apart and would be found only if
S
= integer
( P / 2)
i.e. winding exactly repeats for each pair of poles. The distance between equipotential points in terms of
number of commutator segments is C/(P/2), i.e. 12/2 = 6 in the example. It is too expensive to use equalizer
rings equal to the number of equipotential point-pairs; a much smaller number is employed in actual practice.
Two equalizer rings are shown properly connected in Fig. 7.8 for the example.
While equalizer rings inhibit the flow of circulating currents via brushes, they are not a prevention for the
circulating currents which do cause additional copper losses.
It will soon be seen that wave winding scheme does not have the need of equalizer rings and would
naturally be preferred except in large heavy current machines.
EXAMPLE 7.2 Give the relevant details for drawing lap winding for a dc machine with 4 poles, 22 slots
and 6 coil-sides/slot. What should be the spacing between brushes?
SOLUTION
22
ª 5 slots
4
Back pitch = 6 ¥ 5 + 1 = 31
Front pitch = 31 – 2 = 29 (progressive winding)
Coil span, Ycs =
These data are sufficient to draw the winding diagram.
1
Number of commutator segments = US = 3 ¥ 22 = 66
2
P =A=4
Number of brushes = 4
66
1
Spacing between adjacent brushes =
= 16 segments
4
2
Wave Winding
In wave winding the “finish” end of one coil under one pole-pair is connected to the start of a coil under the
next pole-pair as shown in Fig. 7.11. The process is continued till all the armature coils are connected and
the winding closes onto itself. Certain conditions must be fulfilled for this to happen. The winding has the
appearance of a wave and hence the name. The ends of each coil spread outwards and span yc (commutator
pitch) segments. As the number of coil-sides is double the number of segments, the top coil-side of the second
coil will be numbered (1 + 2yc). The numbering of other coil-sides is clear from the figure. It follows that
DC Machines
1 + 2yc – yf = 1 + yb
or
yf + yb = 2yc = yr
(7.11)
Starting at segment 1 and after going through P/2
coils or yc (P/2) segments, the winding should end up in
segment 2 for progressive winding or segment (C ) for
retrogressive winding. This means that for the winding
to continue and cover all the coils before closing onto
itself, i.e.
1+ 2Yc – Yf
1+ 2Yc
295
1+ 4Yc – Yf
Yb
Yr
Yf
Yc
Yc
C 1 2
Ê Pˆ
yc Á ˜ = (C ± 1)
1+ Yc
1+ 2Yc
(C + 2)
Ë 2¯
2(C ± 1)
Fig. 7.11
or
yc =
(must be integer)
(7.12)
P
In Eq. (7.12) the winding is progressive if + sign is used and is retrogressive otherwise.
Once yb is known from the coil-span and yc is determined from Eq. (7.12), the back-pitch yf is calculated
from Eq. (7.11). The winding diagram can now be drawn.
In wave winding the coils are divided into two groups—all coils with clockwise current are series connected
and so are all coils with counter-clockwise current–and these two groups are in parallel because the winding
is closed. Thus a wave winding has always two parallel paths irrespective of the number of poles; also only
two brushes are required, i.e.
A=2
(7.13)
These statements are corroborated by an illustrative example below:
EXAMPLE 7.3 For a 6-pole dc armature with 16 slots having two coil-sides per slot and single-turn coils,
calculate the relevant pitches for a wave winding and draw the developed winding diagram.
SOLUTION
16
ª 2 slots (nearest lower integral value)
6
=2¥2+1=5
= 16
2(16 ± 1)
=
= 5 segments
6
= 2yc – yb = 5
Ycs =
yb
C
yc
yf
As per the above values of various pitches, the developed diagram of the winding is drawn in Fig. 7.12. It is
observed that the armature winding has two parallel paths; in Fig. 7.29)—current is going in at segment 6 and
is coming out at segment 14 (coil-side 27 has negligible emf and the direction of current in it is determined
by the next coil-side in series with it, i.e. 32). Only two brushes are, therefore required—one brush is opposite
a north pole and the other opposite the diametrically opposite south pole. The spacing between brushes is
C
C
=
= 8 segments
(7.14)
A
2
The coils are continuously numbered at top ends in Fig. 7.12. Between the two brushes there are two
parallel paths each comprising 8 coils any time as shown in Fig. 7.13 where coil of each parallel path are
numbered at the instant corresponding to Fig. 7.12.
30
28
1
32
29
2
1
31
2
2
3
4
5
6
7
8
9
10
11
12
13
14
15
3
5
–
6
A2
7
8
9
10
11
12
13
+
14
A1
15
16
Fig. 7.12 Wave winding for 6-pole, 16-slot armature (single-turn coil-sides/slot)
4
1
3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32
1
2
1
16
3
3
5
296
Electric Machines
DC Machines
297
We thus conclude that number parallel path in a wave winding is A = 2, irrespective of number of poles.
Therefore conductor current in a wave would machine is
Ic =
Spacing between the two brushes is
Ia
2
(7.15)
C
16
=
= 8 segments
A
2
Electrical spacing = 180°
6
Ia
11
5
16
10
15
4
9
–
A2
+
1
12
7
13
2
Fig. 7.13
8
3
14
Ia
A1
Parallel paths of wave winding
In practical wave-wound machines as many brushes as number of poles are used with spacing between
adjacent brushes being C/P commutator segments and the brushes are alternatively positive and negative.
All positive and all negative brushes are respectively connected in parallel to feed the external circuit. This
reduces the current to be carried by each brush to a value.
IBrush =
Ia
( P / 2)
(7.16)
For a given commutator segment width this reduces the segment length required for maximum allowable
brush current density. In small machines, however, economy—cost of brush gear relative to commutator—
dictates in favour of two brushes only, which are placed opposite two adjoining poles.
Equalizer rings not needed The armature coils forming each of the two parallel paths are under the influence
of all pole-pairs so that the effect of the magnetic circuit asymmetry is equally present in both the parallel
paths resulting in equal parallel-path voltages. Thus equalizer rings are not needed in a wave winding.
In a lap winding yc = ±1 irrespective of the number of armature coils so that coils can always
1
be chosen to completely fill all the slots (C = US).
2
In a wave winding from Eq. (7.12), the number of coils must fulfil the condition
Dummy coils
C=
P
y ±1
2 c
(7.17)
1
US
2
(7.18)
while at the same time C must also be governed by
C=
For a certain-design value of P and the choice of S restricted by manufacturing considerations (availability
of a certain notching gear for armature stamping), the values of C as obtained from Eqs (7.17) and (7.18) may
298 Electric Machines
not be the same. In such a situation the number of coils C ¢ is dictated by
C¢ =
1
US
2
in such a manner that
C¢ > C
and yc is so selected that (C¢– C ) is the least possible. Electrically only C coils (Eq. (7.17)) are needed, but
C¢ coils are accommodated in the armature slots to ensure dynamic (mechanical) balancing of the armature.
The difference (C¢ – C ) are called dummy coils and are placed in appropriate slots with their ends electrically
insulated.
As an example, if P = 4n (multiple of four), C can only be odd (Eq. (7.17)), while C¢ may be even if an
even number of slots are used. In this case then at least one dummy coil would be needed.
EXAMPLE 7.4 For a 4-pole dc armature with 28 slots and 8 coil-sides per slot, find the winding pitches
and the commutator pitch for a wave winding. What is the distance between brushes in terms of commutator
segments?
SOLUTION
The number of coils that can be accommodated in slots,
28 ¥ 8
= 112
2
2
(C ± 1)
yc =
P
2
1
1
= (112 ± 1) = 55 or 56
4
2
2
Since no integral value of yc is possible for C = 112, dummy coil would be present.
Now
C¢ = 112
P
C=
yc ± 1
2
Choosing
yc = 55
C = 111 (number of coils needed electrically
= number of commutator segments)
Therefore
(112 – 111) = 1 dummy coil will be placed on the armature
C=
28
=7
4
yb = 8 ¥ 7 + 1 = 57
yf = 2yc – yb = 2 ¥ 55 – 57 = 53
Ycs =
Distance between brushes if four brushes are used = 111/4
= 27
3
segments
4
Remark
As per Fig. 7.11, to wind a wave armature, the coils are prepared as per yb (= UYcs + 1). Once the coils are
ready only yc is needed to connect the coils. Practical winders may prefer coil numbering instead of coil-side
numbering–a coil beginning with a top coil-side numbered (l + 2yc) in Fig. 7.11 is indeed a coil numbered
(1 + yc).
DC Machines
299
Comparative Summary of Lap and Wave Winding
Table 7.1
Lap Winding
Wave Winding
S
(lower integer)
P
2. Back-pitch, yb = UYcs + 1
S
(lower integer)
P
yb = UYcs + 1
Ycs =
1. Coil-span, Ycs =
2(C ± 1)
(must be integral)
P
(+ for progressive, – for retrogressive)
yf = 2yc – yb
3. Commutator pitch, yc = ±1
yc =
(+ for progressive, – for retrogressive)
4. Front-pitch, yf = yb ± 2
(+ for progressive, – for retrogressive)
5. Parallel paths, A = P
Conductor current, Ic = la/A
6. Number of brushes A = P
7. No dummy coil needed
8. Equalizer rings needed
A=2
Ic = Ia/2
Number of brushes = 2 (Large machine use P brushes)
Dummy coil may be needed
Equalizer rings not needed
Choice between Lap and Wave Windings
Wave winding’s greatest attraction is that it does not require equalizer rings* which means a less expensive
machine compared to lap winding. Lap winding has the advantage of a larger number of parallel paths and
lower conductor current (Ic = Ia/A) and is therefore adopted for low-voltage high-current machines. The
use of wave winding is prohibited for armature currents exceeding 300 A because of certain commutation
difficulties.
Steps for Designing Armature Winding Using MATLAB
Designing steps of a lap connected dc winding are given below:
S
P
Back-pitch, yb = UYcs + 1
Commutator pitch, yc = ±1
Front-pitch, Yf = Yb ± 2
Parallel paths, A = P
1. Coil span Ycs =
2.
3.
4.
5.
(+ for progressive – for retrogressive)
(+ for progressive – for retrogressive)
Ia
A
7. A loop program to construct winding table.
Given in a DC machine:
No of slots (S ) = 18
Poles (P ) = 6
Coil sides/slot (U ) = 2
Armature current (Ia ) = 25 A
6. Conductor current, Ic =
* A duplex wave winding not discussed here would require equalizer rings.
300
Electric Machines
MATLAB PROGRAM
S = 18;
P = 6;
Ycs =S./P
U = 2;
Yb = U.*Ycs + 1
Yc_progressive = 1
Yc_retrogressive=–1
Yf_progressive = Yb + 2
Yf_retrogressive = Yb–2
A = P
Ia = 25;
Ic = Ia./A
for i = 1: 2 : 2 (2*S –Yb)
[i i + Yb]
end
Ycs =
3
Yb =
7
Yc_progressive =
1
Yc_retrogressive =
- 1
Yf_progressive =
9
Yf_retrogressive =
5
A =
6
Ic =
4.1667
ans =
1
8
ans =
3
10
ans =
5
12
ans =
7
14
ans =
9
16
ans =
11
18
ans =
13
20
% Winding Parameters
% Winding Table
DC Machines
ans =
15
ans =
17
ans =
19
ans =
21
ans =
23
ans =
25
ans =
27
ans =
29
301
22
24
26
28
30
32
34
36
7.3 CERTAIN OBSERVATIONS
From this discussion in sections 7.1 and 7.2, certain observations about the dc machine are summarized as
follows. These will help in visualizing the machine behaviour in this chapter.
1. Brushes in a dc machine are normally placed electrically in the interpolar regions and therefore make
an angle of 90° elect. with the axes of the adjoining field poles.
2. A lap winding has A = P parallel paths such that the armature current Ia divides out into A paths giving
a conductor current of Ic = Ia/A. In the case of wave winding, A = 2, independent of P.
3. The brushes are alternately positive and negative (elect. angle between adjacent pair being 180° elect).
Only two brushes are needed in wave winding though P brushes are commonly provided for heavycurrent armatures.
4. The armature periphery is divided into “belts” (P in number) each under influence of a pole. Emfs
and currents in all the conductors of a belt are unidirectional–conductors that go out of a belt due to
rotation are simultaneously replaced by an equal number coming into the belt. Magnitude of conductor
emfs in a belt follows the pattern (wave) of flux density in the airgap while the current in all these
conductors (Ic) is the same in all the belts except that the current pattern in the belts alternate in space
but remain fixed in time. This basically results from the action of the commutator.
5. If the conductor current flows in the same direction as the conductor emf, the machine outputs electrical
power (and absorbs mechanical power), i.e. the machine is operating in generating mode. On the other
hand, when the conductor current and emf oppose each other, the machine absorbs electrical power
and outputs mechanical power, i.e. it operates in the motoring mode.
6. Barring irrecoverable losses (of both electric and magnetic origin), there is a balance between
electrical and mechanical powers of the machine; the average energy stored in the magnetic field
remains constant independent of the armature rotation.
7.4
EMF AND TORQUE
It was shown in Sec. 5.2 (Fig. 5.14(a) that in a dc machine the magnetic structure is such that the flux density
wave in the air-gap is flat-topped with quarter-wave symmetry so long as the armature is not carrying any
302
Electric Machines
current. It will be seen in Sec. 7.7 that the flux density wave gets distorted when the armature carries current
(armature reaction effect destroys quarter-wave symmetry). However, this fact does not affect the constancy
of emf (between brushes) and torque developed by the machine with magnitudes of each of these being
determined by the flux/pole independent of the shape of the B-wave.
EMF Equation
As per Eq. (5.23), the average coil emf is
E ac =
where
F = flux/pole
wm = armature speed in rad/s
Nc = number of coil turns
P = number of poles
Fw m N c P
p
(7.19)
Let
Cp = coils/parallel path
This number is fixed independent of the armature rotation; as one coil moves out of the parallel path
another comes in and takes its place. Thus, the parallel path emf which equals the armature emf is given by
Ea =
Here
where
Fw m (C p N c ) P
p
=
Fw m N P P
p
Z
2A
Z = total armature conductors
A = number of parallel paths
Np = turns/parallel path =
Hence
Ea =
Fw m Z Ê P ˆ
= KaFwm
2p ÁË A ˜¯
where
Ka =
ZP
2p A
or
Ea =
where
(7.20)
FnZ Ê P ˆ
2p n
; wm =
60 ÁË A ˜¯
60
n = armature speed in rpm
(7.21)
(7.22)
Ea remains constant (dc) by virtue of fixed coils per parallel path independent of rotation of armature
(commutation action). It is also observed that Ea depends upon the flux/pole and not upon the shape of the
flux density wave.
Torque Equation
Figure 7.14 shows the flux density wave in the air-gap and the conductor current distribution in the developed
armature for one pole-pair. It is immediately seen that the force on conductors is unidirectional. Each
conductor as it moves around with the armature experiences a force whose time variation is a replica of the
B-wave.
Therefore, the average conductor force
fc,av = Bav lIc
(7.23)
DC Machines
303
Force on conductors
Flux density B
tp
Conductor current (Ic)
Fig. 7.14 Torque production in dc machine
Bav = average flux density over pole
l = active conductor length
Total force
F = Zfc, av = BavIclZ
where
This force (and therefore torque) is constant (independent of time) because both the flux density wave and
current distribution are fixed in space at all times. Now the torque developed
where
T = Bav IclZr
r = mean air-gap radius
(7.24)
The flux/pole* can be expressed as
F = Bav tp l
where
tp = pole-pitch (Fig. 7.14) =
\
Ê 2p r ˆ
l
F = Bav ÁË
P ˜¯
Bav =
or
FP 1
¥
2p
rl
2p r
P
(7.25)
Substituting for Bav in Eq. 7.24,
1
F IcZP
2p
Ê Pˆ
1
F IaZ Á ˜ Nm
=
Ë A¯
2p
= KaF Ia Nm
T=
(7.26)
(7.27)
It is, therefore, seen that the machine torque is uniform for given flux/pole and armature current. Further,
it is independent of the shape of the B-wave, which in fact gets distorted by the armature mmf when it carries
current.
It is convenient to use force on each conductor in deriving the expression for armature torque. However,
the mechanism of torque production is different in an actual machine in which conductors are placed in
* It is to be noted that the flux/pole F is not dependent on the actual shape of flux density distribution in the airgap.
The flux density distribution in fact gets distorted by the armature current.
304
Electric Machines
armature slots. The force is produced by the interaction of the main flux and the flux produced by current
carrying conductors placed in armature slots. Due to the large reluctance of the air-path of slots, the main flux
passing through the conductors is negligible and so is the force acting on the conductor. Force is produced
mainly by the distortion of the flux lines passing through the teeth, and this force acts on the teeth of the
armature as shown in Fig. 7.15. It is rather fortunate that there is very little force acting on conductors. If all
the force were to act on conductors, it would crush the insulation between conductors and slots.
N
(a) Main flux (solid) and flux (dotted) produced
by armature current-carring conductors
Tangential component of force
(b) Resultant flux
Fig. 7.15
Power Balance
Mechanical power
Twm = Ka FwmIa
= EaIa W
(7.28)
This is nothing but a statement of energy conservation, i.e. electrical and mechanical powers must balance
in a machine.
Ea Ia is referred to as electromagnetic power. From Eq. (7.28) we get the electromagnetic torque as
T = (EaIa)/wm
Sum-up
Armature emf, Ea = Ka F wm
Electromagnetic torque, T = Ka F Ia
where
Power balance
ZP
; machine constant
2p A
2p n
wm =
rad/s
60
n = speed in rpm
Twm = Ea Ia = Electromagnetic power
(7.29)
(7.30)
Ka =
In frequent use we may drop the suffix m in wm, i.e., write w in place of wm.
Linear Magnetization
If the magnetic circuit of the machine is assumed linear*
F = Kf If
* Presence of air-gap justifies this approximation so long as iron is lightly in saturated state.
(7.31)
DC Machines
Then
and
If = field current
Kf = field constant
Ea = Ka Kf Iwm = Ke If n V
T = Ka Kf If Ia = Kt If Ia Nm
where
Ke =
where
305
(7.32)
(7.33)
2p
(K K )
60 a f
Kt = KaKf
(7.34)
(7.35)
The derivation of torque developed (Eq. 7.26) using magnetic field interaction is carried out in Sec. 7.6
after armature reaction ampere turns are determined.
EXAMPLE 7.5 A 4-pole dc motor is lap-wound with 400 conductors. The pole shoe is 20 cm long and
average flux density over one-pole-pitch is 0.4 T, the armature diameter being 30 cm. Find the torque and
gross mechanical power developed when the motor is drawing 25 A and running at 1500 rpm.
SOLUTION
Flux/pole =
Induced emf =
p ¥ 30 ¥ 10- 2
¥ 20 ¥ 10–2 ¥ 0.4 = 0.0188 Wb
4
FnZ Ê P ˆ
60 ÁË A ˜¯
0.0188 ¥ 1500 ¥ 400 Ê 4 ˆ
¥ Á ˜ = 188 V
Ë 4¯
60
Gross mechanical power developed = Ea Ia
=
=
Torque developed =
188 ¥ 25
= 4.7 kW
1000
4.7 ¥ 1000
= 29.9 Nm
Ê 2p ¥ 1500 ˆ
ËÁ
¯˜
60
7.5 CIRCUIT MODEL
The parallel paths of dc machine armature are
symmetrical and each has an induced emf Ea and a
resistance Rp. as shown in Fig. 7.16 below for A = 4. Its
Thevenin equivalent is drawn by the side in which
Voc = Ea, RTH = Rp/A = Ra
(7.36)
The armature can therefore be represented by the
symbol as shown in Fig. 7.16 with Ea within circle and
the series resistance Ra written by its side. We may later
on skip writing Ra with the understanding that it present
within.
The armature resistance is quite small so as to limit
the copper-loss to an acceptable value. Figure 7.16 also
shows the field circuit of the machine and the field coil
Rp
Rp
Rp
+
+
Ea
Ea
Rp
Rp
+
+
Ea
A
= Ra
∫
+
Ea
Ea
Ia
+
Rf
F
+
Ea
–
If
Ra
Vf
Fig. 7.16 Circuit model of dc machine
Va
–
Electric Machines
306
axis is placed at 90° to the brush axis as per the actual arrangement in the machine*. From circuit point of
view it is not necessary to rigidly follow this scheme. Since most of the time steady-state dc behaviour of the
machine will be considered, the inductances of field and of armature (this is negligible any way) circuits are
of no consequence and are not shown in the circuit model. The armature induced emf and machine torque are
governed by the relationships of Eqs (7.29) and (7.30).
The voltage drop at brush-commutator contact is fixed (1–2 V), independent of armature current as the
conduction process is mainly through numerous short arcs. However, this voltage being small is modelled
as linear resistance and lumped with Ra. From now onwards it will be assumed that Ra includes the effect of
brush voltage drop.
Generating Mode
The machine operates in generating mode (puts out electrical power) when Ia is in the direction of induced
emf Ea as in Fig. 7.17(a). For the armature circuit
Vt (armature terminal voltage) = Ea – IaRa; Ea > Vt
(7.37)
Thus a dc machine is generating if its armature induced emf (Ea) is more than its terminal voltage (Vt)
The electromagnetic power converted from mechanical to electrical from is
Ea Ia = Pmech (in)| net = Pelect (out)| gross
(7.38)
The net electrical power output is
Also
and
P0 = Vt Ia
Ea Ia – Vt Ia = I2a Ra = armature copper-loss (inclusive of brush loss)
Pmech(in)|gross = shaft power = Pmech(in)|net + rotational loss
(7.39)
(7.40)
(7.41)
In this mode torque (T) of electromagnetic origin is opposite to the direction of rotation of armature, i.e.,
mechanical power is absorbed and a prime-mover is needed to run the machine.
The conductor emf and current are also in the same direction for generating mode as shown in the crosssectional view of Fig. 7.17(c).
Motoring Mode
In this mode, Ia flows in opposition to induced emf Ea as in Fig. 7.17(b). Ea is now known as the back emf to
stress the fact that it opposes the armature emf. For the armature circuit
Vt (armature terminal voltage) = Ea + Ia Ra; Vt > Ea
(7.42)
Thus a d.c. machine is motoring if armature terminal voltage (Va) is more than its induced emf (Ea).
The electromagnetic power converted from mechanical to electrical from is
EaIa = Pelect (in)|net = Pmech (out)|gross
(7.43)
Pi = Vt Ia
Vt Ia – Ea Ia = I2aRa = armature copper-loss (inclusive of brush loss)
Pmech (out)|net = shaft power = Pmech (out)|gross – rotational loss
(7.44)
(7.45)
(7.46)
The electrical power input is
Also
and
* In actual machine this angle is 90° elect.
DC Machines
Ia
Ia
+
+
+
F
Ea
F
Va
Ra
–
If
+
Vf
–
T
307
+
Ea
–
Va
–
If
Vf
+
n
Ra
–
–
n
T
Pmech
Pmech
(a) Generating mode
(b) Motoring mode
Current gen
emf
+
Current mot
N
n
S
If
(c) Generating/motoring modes
Fig. 7.17
In this mode torque (T) of electromagnetic origin is in the direction of armature rotation, i.e., mechanical
power is put out and is and absorbed by load (mechanical).
Conductor emf and current are also in opposite directions for motoring mode as shown in Fig. 7.17(c).
EXAMPLE 7.6 A 220 V dc generator supplies 4 kW at a terminal voltage of 220 V, the armature resistance
being 0.4 W. If the machine is now operated as a motor at the same terminal voltage with the same armature
current, calculate the ratio of generator speed to motor speed. Assume that the flux/pole is made to increase
by 10% as the operation is changed over from generator to motor.
SOLUTION
From Eq. (7.22)
Ea
F
4 ¥ 1000
Ia =
= 18.18 A
220
Eag = 220 + 0.4 ¥ 18.18 = 227.3 V
Eam = 220 – 0.4 ¥ 18.18 = 212.7 V
Fm = 1.1 Fg
nμ
As a generator
As a motor
Also
Substituting in Eq. (i)
ng
nm
227.3 Fm
227.3
¥
=
¥ 1.1
212.7 Fg
212.7
= 1.176
=
(i)
(ii)
(iii)
(iv)
308
Electric Machines
EXAMPLE 7.7 A dc shunt generator driven by a belt from an engine runs at 750 rpm while feeding
100 kW of electric power into 230 V mains. When the belt breaks it continues to run as a motor drawing 9
kW from the mains. At what speed would it run?
Given armature resistance 0.08 W and field resistance 115 W.
Note: In a shunt machine the field is connected across the armature and is also connected directly to the
230 V mains. The field excitation therefore remains constant as the machine operation changes as described
above.
SOLUTION
The operation of a dc shunt generator/motor is indicated in the circuit models of Figs 7.18(a) and (b).
IL
If
115 W
IL
If
Ia
+
Ea(g)
230 V
115 W
Ia
+
Ea(m)
230 V
0.08 W
0.08 W
–
–
PM
n(?)
n = 750 rpm
(a) Generating
(b) Motoring (prime-mover disconnected)
Fig. 7.18
230
= 2 A; remains constant in operation change-over.
Field current If =
115
Running as generator (feeding power to mains)
100 ¥ 1000
= 434.8 A
230
=2A
= IL + If = 434.8 + 436.8 A
= 230 + 0.08 ¥ 436.8 = 264.9 V
= 750 rpm
IL (line current) =
If
Ia
Ea(g)
n(g)
Running as motor (drawing power from mains)
9 ¥ 1000
= 39.13 A
230
If = 2 A
Ia = IL – If = 39.13 – 2 = 37.13 A
Ea(m) = 230 – 0.08 ¥ 37.13 = 227 V
IL =
As field current (and so flux/pole) do not change during the two kinds of operation, the induced imf (Ea) is proportional
to armature speed. Hence
n (motor )
227
n (motor )
=
=
750
n (generator )
264.9
n (motor) = 642.7 rpm.
DC Machines
309
Lap Versus Wave Winding
Consider a P pole machine having flux/pole F and rotating at wm rad/s. It has a total of Z conductors and
maximum permissible conductor current is Ic. Let us derive the expression for power converted and torque
developed.
Ê ZP ˆ
Ea = Á
Fwm
Ë 2p A ˜¯
Ia (permitted) = A Ic
Ê ZP ˆ
Power converted = Ea Ia = ÁË
Fwm Ic
2p ˜¯
Torque developed,
Ê ZP ˆ
T= Á
FI
Ë 2p A ˜¯ a
or
Ê ZP ˆ
T= Á
FI
Ë 2p ˜¯ c
(i)
(ii)
We find that the power converted and torque developed are independent if the number of parallel paths. It
means that these values are that same whether the conductors are lap connected or wave. These in fact depend
on number of conductors and permissible conductor current.
EXAMPLE 7.8 Consider a dc machine whose circuit model is drawn below. It is a separately excited
machine as its field coil (winding) is excited from a voltage source independent of the armature circuit.
A 25 kW, 250 V dc machine is separately excited. The field current is held constant at a speed of 3000 rpm.
The open circuit voltage is 250 V. Calculate the terminal power, electromagnetic power and torque at
terminal voltage of (a) 255 V, and (b) 248 V. The armature resistance is 0.05 W. Speed is held constant at
3000 rpm.
SOLUTION
Open circuit (Ia = 0), Then
Vt = Ea = 250 V at 3000 rpm
Ia (g)
+
Ea remains constant
(a)
Ia (m)
Vt = 255 V
Ea
As Vt > Ea, the machine is acting as a motor
Vt - Ea
255 - 250
Ia =
=
= 100 A
Ra
0.05
n
If
+
Vt
Vf
–
–
The current flowing into the positive terminal in opposition to
Ea. Therefore
Fig. 7.8(P)
Electromagnetic power (in) = Ea Ia = 250 ¥ 100 = 25 kW = mechanical power output
Speed = 3000 rpm or
Electromagnetic torque,
T=
3000 ¥ 2p
= 314.16 rad/s
60
Ea I a
25 ¥ 103
=
= 79.58 Nm
wm
314.16
The torque is in the direction of rotation driving the mechanical load which absorbs the mechanical power
produced by the motor.
310
Electric Machines
(b) Vt = 248 V, Ea >Vt, the machine is acting as generator
250 - 248
= 40 A
0.05
Ia flows out of positive terminal and is in same direction as Ea.
Ia =
Electromagnetic power (out) = Ea Ia = 248 ¥ 40 = 9.92 kW = mechanical power in
9.92 ¥ 103
= 31.58 Nm
314.16
The torque is opposite direction to direction of rotation. The mechanical is absorbed by the machine. It is supplied
by the prime mover.
Electromagnetic Torque,
T=
EXAMPLE 7.9 In the machine of Example 7.8 the field is held constant with a terminal voltage of 250 V,
the armature speed is found to be 2900 rpm. Is the machine motoring or generating? Calculate terminal
current, terminal power and electromagnetic power.
SOLUTION
Ea = Ka Fw
As
With F constant, at 2950 rpm
Ea = 250 ¥
2950
= 245.8 V
3000
Vt = 250 V
As Vt > Ea, the machine is motoring
Terminal quantities
250 - 245.8
= 84 A (in)
0.05
Power, Pin = 250 ¥ 84 = 21 kW
Electromagnetic power = Ea Ia = 245.8 ¥ 84 = 20.65 kW
Ia =
Note: The speed reduces as the motor shaft carries the load.
7.6 ARMATURE REACTION
When the armature of a dc machine carries current, the distributed armature winding produces its own
mmf (distributed) known as armature reaction. The machine airgap is now acted upon by the resultant mmf
distribution caused by simultaneous action of the field ampere-turns (ATf) and armature ampere-turns (ATa).
As a result the air-gap flux density gets distorted as compared to the flat-topped (trapezoidal) wave with
quarter-wave symmetry when the armature did not carry any current.
Figure 7.19 shows the cross-sectional view of a 2-pole machine with single equivalent conductor in each
slot (current/conductor = UNcIc where U = coil-sides/slot, Nc = conductors/coil-side (turns/coil) and Ic =
conductor current). Two axes can be recognized—the axes of main poles called the direct axis (d-axis) and
the axis at 90° to it called the quadrature axis (q-axis). Obviously the q-axis is the geometric neutral axis
(GNA) of the machine. The brushes in a dc machine are normally located along the q-axis.
Because of commutator action, armature current distribution is as shown in Fig. 7.19 for a 2-pole machine
(or Fig. 7.3 for a 4-pole machine). All the conductors on the armature periphery between adjacent brushes
carry currents (of constant value, UNc Ic) in one direction and the current distribution alternates along the
periphery. This current pattern remains fixed in space independent of armature rotation. Since the brushes are
DC Machines
311
placed along GNA, the stationary armature current pattern is congruent with the main poles. Also, the current
pattern can be shifted by moving all the brushes simultaneously to either side; this is not a normal operation
in a dc machine.
It is easy to see from Fig. 7.19 that the axis of ATa lies along the q-axis at 90° elect. to the of main poles
which lies along the d-axis (ATa lags behind ATf with respect to the direction of armature rotation for the
motoring mode and vice versa for the generating mode). It may be noticed that ATa and ATf as shown by
arrows are not vectors as their space-angle distribution is non-sinusoidal (though periodic). Armature reaction
with axis at 90° to the main field axis is known as cross-magnetizing mmf. Figure 7.19 also shows the flux
pattern, in dotted lines, caused by armature reaction acting alone. It is immediately observed that the armature
reaction flux strengthens each main pole at one end and weakens it at the other end (crossmagnetizing effect). If
the iron in the magnetic circuit is assumed unsaturated (therefore linear), the net flux/pole remains unaffected
by armature reaction though the air-gap flux density distribution gets distorted. If the main pole excitation is
Electric machines
Current = UNcIc A
N
3
A
4
b
2
1
ATa
Gen
ATf
1¢
Mot
B
2¢
3¢
4¢
S
Ia
d-axix
Fig. 7.19
GNA
q-axis
a
312 Electric Machines
such that iron is in the saturated region of magnetization (this is the case in a practical machine), the increase
in flux density at one end of the poles caused by armature reaction is less than the decrease at the other end,
so that there is a net reduction in the flux/pole, a demagnetizing effect; the decrement being dependent upon
the state of magnetization of iron and the amount of ATa (i.e. the armature current).
It may be summarized here that the nature of armature reaction in a dc machine is cross-magnetizing
with its axis (stationary) along the q-axis (at 90° elect. to the main pole axis). It causes no change in flux/
pole if the iron is unsaturated but causes reduction in flux/pole (demagnetizing effect) in presence of iron
saturation.
Graphical Picture of Flux Density Distribution
For a better understanding of the interaction between the field and the armature magnetic field, consider the
developed diagram of Fig. 7.20(a) for one pole-pair with brushes placed in geometrical neutral axis (GNA),
which is also magnetic neutral axis (MNA) when armature is not carrying current. Using the principles
evolved in Sec. 5.4, the armature mmf distribution is drawn in Fig. 7.20(b) which is a stepped wave with axis
shift of 90° elect. from the main pole axis (d-axis), i.e. it is cross-mangnetizing. Each step of the wave has a
height of UNcIc, where U = coil-sides/slot, Nc = conductors/coil-side (turns/coil) and Ic = conductor current.
The stepped-wave of mmf can be well approximated as a triangular wave as shown in Fig. 7.20(b). The peak
value of armature ampere-turns is obtained as follows:
Zlc
Zl
= a
P
AP
Zla
= ATa (peak)
Ampere-turns/pole =
(7.47)
2 AP
ATa ( total)
Also
ATa (peak) =
P
The exact way to find the flux density owing to the simultaneous action of field and armature ampere-turns
is to find the resultant ampere-turn distribution
Ampere-conductors/pole =
ATresultant (q) = ATf (q) + ATa(q)
where q is electrical space angle.
A simpler procedure, however, will be adopted by assuming linearity of the magnetic circuit making
possible the superposition of individual flux density waves to obtain the resultant flux density as illustrated
in Fig. 7.20(c).
The flux density of ATa(q) is shown in Fig. 7.20(b) which, because of large air-gap in the interpolar region,
has a strong dip along the q-axis even though ATa (peak) is oriented along it. The flux density of the main field
alone (trapezoidal wave) and the resultant flux density are both drawn in Fig. 7.20(c). It is found from this
figure that the armature reaction mmf causes the flux density wave to get distored so as to be depressed in one
half of the pole and causes it to be strengthened equally (linearity effect) in the other half (cross-magnetizing
effect) because of the odd symmetry with respect to the d-axis of the flux density wave of the armature mmf.
It is, therefore, seen that while the resultant flux density wave is distorted, the flux/pole remains unchanged
at its value in the absence of the armature current. Figure 7.20(c) also reveals that apart from distortion
of the resultant flux density wave, its MNA also gets shifted from its GNA by a small angle a so that the
DC Machines
d-axis
313
q-axis
GNA
N
(a)
S
Current UNcIc
Brush
Motoring
Generating
Armature mmf
distribution
UNcIc
Flux density distribution
(armature mmf only)
ATa(peak)
(b)
90°
elect
Saturation effect
Flux density distribution
(main field only)
Resultant flux density
distribution
MNA
(c)
B¢
A
B
A¢
a
GNA
Shift in magnetic
neutral axis, a
Fig. 7.20
a
brushes placed in GNA are no longer in MNA as is the case in the absence of armature current. This effect is
countered by interpoles placed in GNA (Sec. 7.8).
The effect of iron saturation can now be brought into picture. At angle q on either side of the d-axis,
ampere-turn acting on elemental magnetic path are (ATf ± ATa). It is found from the magnetization curve
314 Electric Machines
of Fig. 7.21 that the increase in flux density on one side of the d-axis caused by additive ATa is less than the
decrease in flux density on the other side by subtractive ATa. As a consequence in presence of saturation,
armature reaction apart from being crossB
magnetizing also causes a net reduction in flux/
pole, a demagnetizing effect. However, no simple
B(+ATa)
quantitative relationship can be established
between demagnetization and ATf and ATa. The B(AT = 0 )
a
reduction in the resultant flux density caused by
saturation of iron is shown by the cross-hatched
B(–ATa)
areas in Fig. 7.20(c).
ATa
To summarize, the armature reaction in a dc
machine is cross-magnetizing causing distortion
ATa
in the flux density wave shape and a slight
shift in MNA. It also causes demagnetization
ATf
AT
(ATf – ATa)
(ATf + ATa)
because a machine is normally designed with
iron slightly saturated.
Fig. 7.21 Magnetisation curve
(i) Increase in iron loss Increase if flux density under one half of the pole and decrease on the other causes
iron loss in armature teeth to increase as the loss is proportional to square of flux density.
(ii) Commutation Shift of MNA causes induced emf in coils undergoing commutation to oppose the current
reversal. It is seen from Fig. 7.20 that for generating case the coil-side to the right of the brush will have ≈
emf while it should be – emf. The same holds for the motoring case. The details will be discussed in Sec. 7.8
on commutation.
(iii) Possibility of commutator sparking Under heavy load (large armature current) and so deep distortion
both coil sides of the coil passing the maximum flux density region will have much larger induced emf than
the average coil emf. If the emf value exceeds 30-40 V, there may be sparkover of the commutator segment
connected to the coil to the adjacent segment. The result may be flash-over of the complete commutator.
Remedies
The cross-magnetizing effect of the armature reaction can be reduced by making the main field ampere-turns
larger compared to the armature ampere-turns such that the main field mmf exerts predominant control over
the air-gap flux. This is achieved by:
(i) Introducing saturation in the teeth and pole-shoe.
(ii) By chamfering the pole-shoes which increases the air-gap at the pole tips. This method increases the
reluctance to the path of main flux but its influence on the cross-flux is much greater. This is because
the cross flux has to cross the air-gap twice: see Fig. 7.19.
(iii) The best yet the most expensive method is to compensate the armature reaction mmf by a compensating
winding located in the pole-shoes and carrying a suitable current. This method is discussed in detail in
Sec. 7.7.
Brush Shift
To counter the effect of shift in MNA due to armature reaction , the brushes could be shifted. A small brush
shift in appropriate direction, in the direction of rotation for generator and in opposite direction for motor, also
DC Machines
315
helps in commutation; Sec. 7.8. The effect of brush shift by angle b is illustrated in Fig. 7.22. The armature
conductor current pattern changes accordingly. It is seen from the figure that the current belt of angle 2b has
direct demagnetizing action. The remaining current belt of angle (180° – 2b ) is cross-magnetizing. This is
illustrated by the following example.
2b
b
b
Brush axis (shifted)
S
N
Fig. 7.22
EXAMPLE 7.10 A 250 kW, 400 V, 6-pole dc generator has 720 lap wound conductors. It is given a brush
lead of 2.5 angular degrees (mech). from the geometric neutral. Calculate the cross and demagnetizing
turns per pole. Neglect the shunt field current.
SOLUTION
25 ¥ 103
= 625 A
400
Number of parallel paths = 6
625
Conductor current, Ic =
= 104.2 A
6
Armature current, Ia =
Total armature ampere-turns, ATa =
1 Ê 720 ¥ 104.2 ˆ
˜¯
Á
2Ë
6
= 6252 AT/pole
With reference to Fig. 7.22, it is easily observed that in 180° elect., conductors in that belt of 2b elect. degrees are
demagnetizing where b is the brush shift in electrical degrees.
b = 2.5 (6/2) = 7.5° elect.
Hence,
2 ¥ 7.5 ˆ
Ê
cross-magnetizing ampere-turns = 6250 Á1 Ë
180 ˜¯
= 5731 AT/pole
2 ¥ 7.5
Demagnetizing ampere-turns = 6250 ¥
= 521 AT/pole
180
Torque Equation (Based on magnetic field interaction)
The magnetic field interaction torque given by Eq. (7.48) is reproduced below
T = (p/2) (P/2)2 Fr F2 sin d
(7.48)
316
Electric Machines
It has been shown above that the resultant flux/pole F in a dc machine is always oriented at 90° to the armature
reaction AT (i.e.F2). Thus
d = 90°(fixed)
This indeed is best value of d for torque production
As per Eq. 7.46
F2 (triangular peak) = ATa (peak)
ZI a
per pole
2 AP
It can be shown that for a triangular periodic wave, the fundamental is 8/p 2 of the peak value. Thus
8 ZI a
F2 (fundamental peak) = 2
p 2 AP
Substituting in torque equation
=
2
or
T=
8 ZI a
p Ê Pˆ
F 2
2 ÁË 2 ˜¯
p 2 AP
T=
1 Ê Pˆ
Z Ia; same as Eq. (7.26)
F
2p ÁË A ˜¯
7.7 COMPENSATING WINDING
It was seen in Sec. 7.6, Fig. 7.20(c) that armature reaction causes the flux density wave to be so badly
distorted that when a coil is passing through the region of peak flux densities, the emf induced in it far
exceeds the average coil voltage. If this emf is higher than the breakdown voltage across adjacent segments,
a sparkover could result which can easily spread over and envelop the whole commutator as the environment
near the commutator is always somewhat ionized and conditions are favourable for flashover. The result
is complete short circuit of armature. The maximum allowable voltage between adjacent segments is 30–
40 V, limiting the average voltage between them to much less than this figure. The choice of the average coil
voltage determines the minimum number of commutator segments for its design.
In spite of the above safe design of the commutator there is another factor which can cause severe
overvoltages to appear between commutator segments. This is the time variation of the armature reaction and
its associated flux owing to sudden changes in machine load. Consider coil aa¢ of Fig. 7.23 located midway
Fa(flux linking coil aa¢)
S
N
Gen
Dynamical emf gen
Ba
a
a¢
Brush
Current
Mot
Dynamical emf motor
Statically induced emf
motor (load added)
Statically induced emf
generator (load dropped)
Fig. 7.23
DC Machines
317
between the main poles so that the full armature flux/pole, Fa (shaded area), links the coil. If the load on the
machine undergoes a fast change, Ia and Fa change accordingly resulting in statically induced emf in the
coil proportional to dFa/dt. The voltage is over and above the dynamically induced emf in the coil. Worst
conditions occur when these two emfs are additive. This happens when load is dropped from a generator or
added to a motor. (The reader should verify by application of the right-hand rule and Lenz’s law to coil aa¢
of Fig. 7.23.) The only way to remedy this situation is to neutralize the armature reaction ampere-turns by a
compensating winding placed in slots cut out in pole faces such that the axis of this winding coincides with the
brush axis (along which lies the axis of ATa). For automatic neutralization of ATa at any current, it is necessary
that the compensating winding be series excited with armature current in such a direction as to oppose ATa.
The compensating winding appropriately connected is shown schematically in Figs 7.24(a) and (b).
If
q-axis
d-axis
la
d-axis
Compensating
winding
la
q-axis
(b)
(a)
Fig. 7.24
Compensating Winding
It is found from Fig. 7.24(a) that complete neutralization of the armature mmf is not possible with this
arrangement, since the distributions of armature and compensating mmfs are not identical. It is customary
to compensate part of the armature mmf directly under the pole shoes. The number of ampere-turns required
for this purpose is
Ê pole arc ˆ
ATcw /pole = ATa (peak) ¥ Á
Ë pole pitch ˜¯
=
Ia Z
pole arc
¥
2 AP pole pitch
(7.49)
The compensating winding neutralizes the armature mmf directly under the pole while in the interpolar
region, there is incomplete neutralization. Further, the effect of the resultant armature mmf in interpolar region
is rendered insignificant because of large interpolar gap. The compensating winding, therefore, practically
eliminates the air-gap flux density distortion. The small flux density remaining unneutralized in GNA will be
appropriately modified by the interpole windings discussed in Sec. 7.8.
Compensating windings, though expensive, must be provided in machines where heavy overloads are
expected or the load fluctuates rapidly, e.g. motors driving steel-mills are subjected to severe duty cycles with
rapid changes.
318 Electric Machines
EXAMPLE 7.11 Calculate the number of conductors on each pole piece required in a compensating
winding for a 6-pole lap-wound dc armature containing 286 conductors. The compensating winding carries
full armature current. Assume ratio of pole arc/ pole pitch = 0.7.
SOLUTION
As per Eq. (7.48)
ATcw /pole =
\
Ncw /pole =
I a Z Ê pole arc ˆ
2 AP ÁË pole pitch ˜¯
Z Ê pole arc ˆ
286
¥ 0.7 = 2.78
=
2 AP ÁË pole pitch ˜¯
2¥6¥6
Compensating conductors/pole = 2 ¥ 2.78 = 6 (nearest integer).
7.8 COMMUTATION
One coil each under an adjoining pole-pair is connected between adjacent commutator segments in a lapwound dc armature, while in a wave-wound armature the only difference is that P/2 coils under the influence
of P/2 pole-pairs are connected between adjacent segments. Coil(s) current is constant and unidirectional
so long as the coil is under the influence of given pole-pair(s), while it reverses (commutates) when the coil
passes onto the next pole-pair as the armature rotates. The process of current reversal called commutation
takes place when the coil is passing through the interpolar region (q-axis) and during this period the coil is
shorted via the commutator segments by the brush located (electrically) in the interpolar region. Commutation
takes place simultaneously for P coils in a lap-wound machine (it has P brushes) and two coil sets of P/2 coils
each in a wave-wound machine (electrically it has two brushes independent of P).
Figure 7.25 shows the schematic diagram of commutator segments in developed form connected to
armature coils. Attention will now be focussed on coil Cc as it undergoes commutation. Various symbols used
in the figure are:
Ic = coil current
Ib = 2Ic = brush current
wc = width of one commutator segment
wm = width of mica insulation between segments
wb = brush width
= (wc + wm) for the case illustrated; in practice, however, wb = 1.5 wc
vc = peripheral speed of commutator
At the instant the commutation of the coil Cc begins, the leading tip of the brush is making full contact with
the segment x and is just going to make contact with segment y as shown in Fig. 7.25(a). At this instant all
coils to the right of segment x carry current Ic flowing from left to right and those on the left current Ic in the
opposite direction. During the period of commutation as the coil passes from right to the left of the brush, the
coil current must reverse. During this period the brush short-circuits the coil via segments x and y as shown
in Fig. 7.25(b). The contact width, xc, between brush and segment x reduces linearly while the contact width,
yc, between brush and segment y increases. The coil current ic(t) during this period is changing. If at the end
of the commutation period, when the trailing tip of the brush is going to break contact with segment x as
shown in Fig. 7.25(c), the coil current has not reversed and acquired full value Ic but as Ic¢ < Ic, the breaking
DC Machines
319
of current (Ic – Ic¢) at the trailing brush tip takes place causing sparking. This is known as under-commutation
(or delayed-commutation). It is easy to see from Fig. 7.25, that the period of commutation is given by
wb - wm
(7.49)
vc
even when wb > (wc + wm) in which case more than one coil undergoes commutation simultaneously.
tc =
Cc
lc
lc
lc
vc
lc
y
x
wc
wm
Trailing tip
lb = 2lc
wb
Leading tip
(a) Beginning of commutation period
ic(t)
lc
Cc
lc
vc
yc
xc
2lc
(b)
l¢c
lc
vc
x
lc
lc
y
(lc + I¢c)
(lc – I¢c)
Sparking
2lc
(c) End of commutation period
Fig. 7.25
Commutation process
Before the causes underlying under-commutation and consequent sparking are explained, the deleterious
effects of sparking and why it cannot be tolerated to any large degree may now be studied. Sparking leads
to destructive blackening, pitting and wear and eventual burning of commutator copper and brush carbon.
It must, therefore, be limited to a tolerable intensity to prolong life of commutator-brush assembly to an
320
Electric Machines
acceptable value. As completely sparkless commutation is not possible practically (for reasons advanced
below), the carbon brushes must be replaced after some time and less frequently commutator “turned” to a
slightly smaller diameter to prepare a fresh clean surface.
Ideal Commutation (also called straight-line commutation) is that in which the current of the commutating
coils changes linearly from + Ic to – Ic in the commutation period as shown in Fig. 7.26. The figure also shows
delayed commutation and the current (Ic – I ¢c) in the spark. In a machine without commutation aids (described
later in this section) the commutation is delayed for the following reasons:
1. The leakage inductance Lc of the coil (see Sec. 5.7) undergoing commutation has induced in it
reactance voltage Lc (dic/dt) which opposes the change in current thereby delaying commutation.
Also, usually more than one coil undergo commutation simultaneously, the induced voltage due to
mutual inductance among them also tends to prevent current reversal.
2. The effect of armature reaction causes
Delayed (under)-commutation
+lc
shift in MNA as shown in Fig. 7.20(c)
from A, B to A¢, B¢. Since the brushes are
located at A, B(GNA’s), a small voltage
is induced in the commutating coil. It
tc
opposes current commutation (both for
0
t
generating/motoring machine) as the
–l¢c
commutating coil is cutting the flux which
has the same sign as that of the pole being
Current in spark
left behind. It could be partially remedied
by shifting the brushes towards MNA but
Ldeal commutation
–lc
that causes direct demagnetization and is
Fig. 7.26 Straight-line commutation
therefore not employed in practice.
There are two ways of achieving good commutation–close to straight-line commutation. These are
resistance commutation and voltage commutation. The former is always used to give marginal support to the
latter.
Resistance Commutation
High contact resistance between commutator segments and brushes, achieved by using carbon brushes, adds
resistance to the circuit of the commutating coil thereby reducing the time-constant (L/R) of the current
transient (ic(t)), helping it to change faster in the desired direction. Carbon brushes are invariably used in dc
machines. They also help reduce commutator wear and are themselves easily replaceable.
Voltage Commutation
To speed up the commutation process, the reactance voltage must be neutralized by injecting a suitable polarity
dynamical (speed) voltage into the commutating coil. In order that this injection is restricted to commutating
coils, narrow interpoles (also called commutating poles or compoles) are provided in the interpolar region.
These apply a local correction to the air-gap flux density wave such that a pip of appropriate flux density
exists over the commutating coil to induce in it a voltage of the same sign as that of coil current after
commutation. For neutralization of reactance voltage at all loads, the interpoles must be excited by armature
current by connecting them in series with armature. Arrangement of interpoles, their polarity relative to the
DC Machines
321
main poles, flux pattern of both sets of poles and one commutating coil are shown in Fig. 7.27. It is easy to
observe from this figure that polarity of an interpole is that of the main pole ahead in the direction of armature
rotation for the generating mode and that of the main pole left behind with respect to the direction of rotation
for motoring mode. The interpolar air-gap is kept larger than that of the main pole so that their magnetic
circuit is linear resulting in cancellation of the reactance voltage (a linear derivative term) at all loads. Large
air-gap results in greater amount of leakage flux which is accommodated by tapering the interpoles with a
wider base as shown in Fig. 7.27.
Main flux
Interpole flux
N
Gen
b
n
s
a
Commutating
emf (gen)
Mot
S
Current
Leakage flux
Fig. 7.27
For cancellation of reactance voltage on an average basis
2[Bi (av)liva] Nc = Lc
where
Ê 2lc ˆ
dic
= Lc ÁË t ˜¯
dt
c
(7.50)
Bi (av) = average flux density in interpolar air-gap
li = iron length of interpoles (it is less than that of main poles)
va = armature peripheral speed
Nc = number of turns of commutating coil.
With Bi determined from Eq. (7.50), the ampere-turns needed to cancel the armature reaction ampere-turns
and then to create the necessary flux density are given by
ATi = ATa (peak) +
where
Bi
l
m0 gi
(7.51)
lgi = air-gap of interpoles.
As relationships of Eqs (7.50) and (7.51) are based on sweeping approximation and also accurate estimate
of the coil leakage inductance Lc cannot be obtained, the attainment of good commutation is more an
322
Electric Machines
empirical art than an analytical science. Furthermore, only good commutation can be achieved but not perfect
commutation. One can always observe some sparking at the brushes of a dc machine in operation.
It is not necessary to have interpoles equal to the number of main poles and to reduce cost, especially in
low-power dc machines, interpoles of the same polarity are often fitted in alternate interpolar spaces only.
EXAMPLE 7.12 A 440 V, 4-pole, 25 kW, dc generator has a wave-connected armature winding with
846 conductors. The mean flux density in the air-gap under the interpoles is 0.5 Wb/m2 on full load and the
radial gap length is 0.3 cm. Calculate the number of turns required on each interpole.
SOLUTION
As per Eq. (7.51)
ATi = ATa(peak) +
Assuming Ia = Iline
\
\
Bi
I Z
B
lgi = a + i lgi
2 AP m0
m0
25 ¥ 103
= 56.82 A
440
56.82 ¥ 846
0.5
+
¥ 0.3 ¥ 10–2 = 4198
ATi =
2¥2¥4
4p ¥ 10- 7
Ia =
Ni =
ATi
4198
=
= 73.88
Ia
56.82
= 74
7.9
METHODS OF EXCITATION
The performance characteristics of a dc machine are greatly influenced by the way in which the field winding
is excited with direct current. There are two basic ways of exciting a dc machine.
Here the field winding is provided with a large number (hundreds or even thousands)
of turns of thin wire and is excited from a voltage source. The field winding, therefore, has a high
resistance and carries a small current. It is usually excited in parallel with armature circuit and hence
the name shunt field winding. Since the armature voltage of a dc machine remains substantially
constant, the shunt field could be regulated by placing an external series resistance in its circuit.
Here the field winding has a few turns of thick wire and is excited from armature current
by placing it in series with armature, and therefore it is known as series field winding. For a given field
current, control of this field is achieved by means of a diverter, a low resistance connected in parallel
to series winding. A more practical way of a series field control is changing the number of turns of the
winding by suitable tappings which are brought out for control purpose.
Figure 7.28 shows the physical arrangement of shunt
and series field windings on one pole of a machine.
Shunt field winding
Excellent and versatile ways of controlling the shunt and
series excitations are now possible by use of solid-state devices
and associated control circuitry.
The dc machine excitation is classified in two ways—separate
excitation and self-excitation explained below.
Note: In drawing the excitation diagrams of a dc machine, the
field winding will be drawn at 90° to the armature circuit. As
Series field winding
Fig. 7.28
DC Machines
323
pointed out earlier the actual spatial orientation of the magnetic fields produced by the field and armature
circuits is 90° elect.
Where the excitation diagrams are to be drawn repeatedly, we may not necessarily use this convention.
Separate excitation The field is excited from a source independent of the armature circuit as shown in
Fig. 7.29(a). Permanent magnet excitation fall into this category.
Shunt excitation The shunt field is excited from the armature voltage as shown in Fig. 7.29(b).
The series field is excited from the armature current as in Fig. 7.29(c).
Series excitation
A1
A1
Field
F1
F2
Armature
F1
F2
A2
A2
(a) Separate excitation
(b) Shunt self excitation
A1
A1
S1
S2
F1
A2
(c) Series self excitation
Shunt
F2
S1
S2
Series
A2
(d) Compound excitation (cumulative compound)
A1
F1
F2 S2
A1
F1
S1
Rf
F2 S1
Rse
S2
Ea
A2
Ra
A2
(e) Compound excitation (differential compound)
(f) Compound excitation long shunt
+
A1
F1
Rf
F2
S1
Vt
Va
Rse S2
A2
(g) Compound excitation; short shunt
Fig. 7.29
+
Methods of excitation of dc machine
–
–
324
Electric Machines
Remark
As in a dc generator there is no initial voltage or current the shunt field resistance or total circuit resistance
in series excitation and the generator speed must meet certain condition for the generation to excite and build
up voltage; to be discussed in Section 7.10.
Compound Excitation
In compound excitation both shunt and series field are excited. If the two field aid each other (their ampereturn are additive), the excited is called cumulative compound as shown in Fig. 7.29(d). The shunt field is
much stronger than the series field. The air gap flux increases with armature current.
If the two fields oppose each other, the excitation is called differential compound as in Fig. 7.29(e). The air
gap flux/pole decreases with armature current.
The series field is so designed that the increase or decrease in flux/pole is to a limited extent.
There are two type of compounding connections. In long shunt compound of Fig. 7.29(f ) the shunt field
is connected across terminals. In short shunt compound, the shunt field is connected directly across the
armature as shown in Fig. 7.29(g). There is no significant difference in machine performance for the two
types of connections. The choice between them depends upon mechanical consideration or the reversing
switches.
Important Note
If a dc compound machine connected as a generator is run as a motor, the series field connections must
be reversed as the armature current reverses. The motoring action as cumulative/differential would then be
preserved (same as in the generator). This equally applies vice versa – motor to generator.
Self–excitation
It means that its shunt field winding is excited by its own voltage and series field winding.
Steady-state Circuit Equations In steady-state operation of a dc machine the field winding inductances do
not play any role. The schematic diagrams of long and short compound machine are shown in Figs 7.30(a)
and (b). Applying Kirchhoff’s voltage and current laws, the circuit equation.
If
If
IL
IL
Va
+
+
Rse
Ia
Ia
Vt
Rf
Ra
Rse
Ea
Rf
Ra
Ea
Va
Vt
Va
–
–
(a) Long-shunt compound motoring
(b) Shot-shunt compound generating
Fig. 7.30
Long-shunt compound (Fig. 7.30(a)) motoring
Vt = Ea + Ia (Ra + Rse)
Va = Ea + Ia Ra
(7.52a)
(7.52b)
DC Machines
If =Vt /R f
IL = Ia +If
325
(7.52c)
(7.52d)
Generating
Directions of Ia and IL reverse while If direction does not changes. Correspondingly, in Eqs. (7.52)(a) and (b)
+ sign changes to – sign.
Short-shunt compound (Fig. 7.30(b)) Though the figure is drawn for generating the equations for both
operations can be directly written down
Vt = Ea ∓ Ia Ra ∓ ILRse
Va = Ea ∓ Ia R a
If = Va /Rf
IL = Ia ∓ If
(7.53a)
(7.53b)
(7.53c)
(7.53d)
where – sign is for generating and + sign for motoring.
General Connection Diagram of a Compound DC Machine
The connection diagram of a compound d.c. machine with commuting winding and compensating windings
is drawn in Fig. 7.31. Total resistance in series with armature is (Ra + Ri + Rc); Ri = interpole (compole)
winding resistance, Rc = compensating winding resistance.
Commutating
winding
Series
field
Shunt field
Armature
Diverter
Compensating
winding
Regulating
registance
Tapped series
field
(alternative)
Fig. 7.31 Compound d.c machine, long shunt
Control of Excitation
Shunt field: by a series regulating resistance
Series field: For small armature by a diverter resistance connected in parallel with series field. For large
armature by tapped field winding so the winding turns can be changed.
Nomenclature
The dc machines are named according to the method of excitation. Thus, we have
dc shunt generator/motor
dc series generator/motor
dc compound generator/motor
326
Electric Machines
As we have seen in section 7.2 a dc machine can operate as a generator or motor; but the machine designed
specifically as generator/motor has certain distinguishing features.
Net Excitation
For a machine on load, the next AT excitation is
AT (net) = AT (shunt) ± AT (series) – ATd
= Nf If ± Nse Ise – ATd
where
Nf = shunt field turns
Nse = series field current can be reduced by tapping
Ise = Series winding Current Ia or IL; can be reduced by a diverter (a resistance in parallel to the
series winding)
ATd = demagnetizing ampere-turn caused by armature reaction
Usually the net excitation is stated in terms of equivalent shunt field current
If, eq = If (net) = AT (net)/Nf
If, eq determines the flux/pole F whose measure is the induced emf Ea at specified speed (Ea = Ka Fwm). It
will be discussed in section 7.10.
7.10 OPERATING CHARACTERISTICS OF DC GENERATOR
In the operation of a dc generator, the four basic variable of concern are terminal voltage Vt, armature current
Ia, field current If and the speed n. To investigate their interrelationship the generator is run (by a prime
mover) at rated speed (n constant) of the remaining three variables one is held constant at a certain value
and of the last two one is varied to study its relationship with the other. The relationship has to be presented
graphically because of the magnetic saturation effect. Four characteristics of importance are the following:
No-load Characteristic
With Ia = 0 (no load) at constant n, it is the presentation of Vt (=Ea) vs If . This is the most important
characteristic as it reveals the nature of the magnetization of the machine. It is easy to determine as the
generator is on no load and so only low rated prime mover will serve the purpose. It is commonly called the
open–circuit/magnetization characteristic.
Load Characteristic
If the plot of Vt vs If with Ia with held is constant at rated value and constant speed, it is indeed the magnetization
characteristic on load.
External Characteristic
With If and n constant at present, the variation of Vt vs Ia is indeed the characteristic when the generator feeds
a load (load is normally variable)
Armature Characteristic
It is the presentation of Ia vs If with Vt held constant (at rated value) and generator run at constant n and load
varied. It reveals the armature reaction affect on the flux/pole. It is also called regulation characteristic.
The characteristics 2, 3 and 4 are to be determined by on load tests which cannot be easily conducted on
a large size machine as both prime mover and load are to be fully rated.
DC Machines
327
Experimental Set-up for Determining Characteristics
The experimental set-up for determining the dc generator characteristics is shown in Fig. 7.32. The generator
is run by a prime mover at rated speed. The field current can be varied form a low value to full value by a
potentiometer arrangement. The load can be switched ‘ON’ or ‘OFF’ and when ‘ON’ can be varied.
If only no load test is to be conducted, a small motor (usually an induction motor) is used as a prime motor.
S
Ia
A
+
Ea
If
A
V
Load
–
DC Source
Coupled
to
Prime
mover
Fig. 7.32
No Load Test–Open Circuit Test
The switch S in Fig. 7.32 is kept open. The open circuit voltage
VOC = Ea μ F at n = constant
Thus, VOC is a measure of F. Therefore, the plot, of VOC vs If is the magnetization characteristic of the
machine. It is rightly called the Open Circuit Characteristic (OCC).
In conducting the OCC test, If must be raised gradually only in the forward direction otherwise the curve
would exhibit local hysteresis loops. Further, as the machine would have been previously subjected to
magnetization, a small residual voltage would be present with field unexcited. As will be seen presently, this
is necessary for generator self-excitation.
Air-gap line
Voc = Ea
A typical magnetization characteristic is
n = nrated
shown in Fig. 7.33. It is known as open-circuit
n = n1 < nrated
characteristic (OCC) because of the method by
which it is determined. It exhibits all the important
characteristic of the magnetization curve of
iron, modified by the presence of air-gap in the
magnetic circuit.
The extension of the liner portion of the
magnetization curve, shown dotted in Fig. 7.33, is
known as the air-gap line as it represents mainly
Residual
the magnetic behaviour of the machine’s air-gap, voltage
the iron being unsaturated in this region consumes
If
negligible ampere-turns; in any case the effect of
Fig. 7.33 Open-Circuit Characteristic (OCC)
iron is also linear here.
328 Electric Machines
The open-circuit characteristic at a speed other than the one at which the test is conducted is a mere
proportional translation of the characteristic as shown in Fig. 7.33. This is because of the direct proportionality
between Ea and n.
Under load conditions Ea cannot be determined from the OCC for If in the saturation region because of
the demagnetizing effect of armature reaction. We must therefore determine experimentally the equivalent
demagnetizing ampere-turns ATd due to armature reaction under actual load conditions. The induced emf Ea
can then be found from
ATnet = ATf + ATse – ATd
(7.54)
Indeed Ea is Ea (If, Ia), non-linear function of I f and Ia. To determine ATd there is no choice but to conduct
a load test.
Load Characteristic
In Fig. 7.32, the switch S is closed. At every value of If the load current Ia is adjusted to the rated value and
the corresponding terminal voltage Vt is read. The load characteristic V vs If |Ia constant and the OCC V vs
If | Ia = 0 are both plotted in Fig. 7.34.
B
Ia = 0 (OCC)
Ea , V
C
IaRa
Ia(rated)
D
Load
characteristic
Ea
O
F
A
If
Ifd
Fig. 7.34 OCC and load characteristic
To the load characteristic we add I aRa drop to get Ea induced emf with load. At If = OA, Ea = AD + DC,
DC = IaRa. Therefore, the voltage drop caused by armature reaction is BC. In the low If region the magnetic
circuit is unsaturated and armature reaction drop is almost zero and so OCC and Ea vs If merge.
At If = OA, the voltage induced with load is AC. With no load the same voltage is induced at If = OF.
Therefore AF = Ifd, the demagnetizing field current equivalent of armature reaction. Thus
ATd = Ifd Nf
DC Machines
329
We need only one point (D) on the load characterization. Further, it is sufficiently accurate to assume that
ATd μ Ia
(7.55)
Another type of load characterization is described below:
Armature Characteristic
With switch S open in Fig. 7.32, If is adjusted to give VOC = V (rated). The switch is then closed the load
current Ia is increased and also If is increased so as to keep the terminal voltage constant at rated value. The
plot of If vs Ia sketched in Fig. 7.35 is the armature characteristic.
If
F
D
A
B
O
Ia (rated)
Ia
Fig. 7.35
It is seen from the figure that at low values of Ia, the increase in If is very small to provide for increasing
Ia Ra drop. At large values of Ia there is a sharp increase in If to compensate for voltage drop caused by
armature reaction. Thus at Ia (rated) DIf = AF provides for Ia R a drop plus armature reaction drop.
Let us examine this data on the OCC plotted in Fig. 7.36.
The voltage drop Ia Ra + Vd is indicated on
the figure. By subtracting Ia Ra, we can then
OCC
(IaRa + Vd)
V(rated)
calculate the constant
Vd
K=
I a (rated )
(full load)
(7.56)
The proportionality constant can be used as
an approximation for other values of Ia. From
the DIf we can compute the Nse needed for a
cumulative compound generator as
Ê DI f ˆ
Nse = Nf Á
Ë I a ˜¯
F
A
O
(7.57)
If
Fig. 7.36
The external characteristics of various types of dc generators will be taken up in Section 7.12.
EXAMPLE 7.13 A 240-V compound (cumulative) dc motor has the following open-circuit magnetization
characteristic at normal full-load speed of 850 rev/min:
330 Electric Machines
Excitation, AT/pole
Generated emf, V;
1200
76
2400
135
3600
180
4800
215
6000
240
The resistance voltage drop in the armature circuit at full-load is 25 V. At full-load the shunt and the
series winding provide equal ampere-turn excitation.
Calculate the mmf per pole on no load. Estimate the value to which the speed will rise when full-load
is removed, the resistance voltage drop in the armature circuit under that condition being 3 V. Ignore
armature-reaction and brush-contact effects. Assume long-shunt cumulative compounding.
SOLUTION
At full load, from Fig. 7.37,
Ea (full-load) = V – Ia (Ra + Rse) = 240 – 25 = 215 V
Corresponding ATnet from magnetizing of Fig. 7.38,
240
215
At 850 rpm
IL
Ea(v)
200
If
160
148
120
+
80
la
Vt
Rsa
Shunt field
Ea
Series field
40
0
Ra
0
1000
2000
3000
4000
5000
2688
4800
Excitation AT/pole
–
Fig. 7.37
Then
Now
Now
Hence,
Long-shunt compound dc motor
Fig. 7.38
ATnet (full-load) = 4800
ATsh = ATse (full-load) = 2400
3
Ia (no-load) =
I (full-load); Ia (no-load) = If (shunt field)
25 a
3
¥ 2400 = 288
ATse (no-load) =
25
ATsh
ATnet (no-load)
Ea (from the magnetizing curve)
Ea (no-load)
Ea
= 2400 (no change)
= 2400 + 288 = 2688
= 148 V at 850 rpm, at ATnet (no-load)
= 240 – 3 = 237 V
μ n at given ATnet
n(no-load) = 850 ¥
237
= 1361 rpm
148
6000
DC Machines
331
It will be seen that in a cumulatively compound dc motor, full-load speed is much less than no-load speed.
EXAMPLE 7.14
If (A)
Voc(V)
The following OC test data was recorded for a separately dc generator:
0.0
10
0.2
52
0.4
124
0.6
184
0.8
220
1.0
244
1.2
264
1.4
276
Its load test data is as under
Ia ( fl) = 50 A, V = 240 V and If = 1.4 A
The armature resistance inclusive of the brush voltage drop is Ra = 0.3 W
Estimate at full load
(a) The internal induced emf
(b) The voltage drop caused by armature reaction
(c) The field current equivalent of armature reaction demagnetization
SOLUTION
The OCC is plotted in Fig. 7.39. The load test point (V = 240 V, If = 1.4 A) for Ia = 50 A is located at A.
Ia Ra = 50 ¥ 0.3 = 15 V
Armature voltage drop,
V
320
C
OCC
D
B
A
240
160
0.36 A
80
0
0.4
0.8
1.2
1.4
1.6
2.0
If (A)
Fig. 7.39
(a) Adding 15 V to 240 V located the point B which is the internal induced emf,
Ea = 240 + 15 = 255 V
(b) At If = 1.4 A VOC = 276 at point C. Therefore armature reaction voltage drop,
Vd = BC = 276 – 255 = 21 V
332 Electric Machines
(c) To induce Ea = 255 V (point B) the field current corresponds to point D on OCC. The field current equivalent of
the armature reaction demagnetization is
Assuming linear relationship
If = BD = 0.36 A
I fd
0.36
=
Kar =
= 7.2 ¥ 10–3
I a ( fl)
50
Rather than arranging a separate dc source for excitation purposes, practical generators are always excited
from their own armature terminals, this method of excitation being known as self-excitation. A self-excited
generator with connection as shown in Fig. 7.40 is known as a shunt generator by virtue of the parallel
connection of the field winding with the armature.
lL = 0
lf
Assume that the field is introduced into the
+
circuit after the armature has been brought to
la
Rf
rated speed. At the instant of switching on the
field, the armature voltage corresponds to a small
Ea
V0 (no-load)
residual value which causes a small field current
to flow. If the field is connected such that this
–
current increases the field mmf and therefore the
induced emf, the machine will continuously buildup. This indeed is a positive feedback connection
nrated(constant)
and the machine builds up to a final steady value
only because of the saturation characteristic of
Fig. 7.40
the machine’s magnetic circuit. Analytical study
of voltage build-up is presented in Section 7.21.
Since the generator is assumed to be on no-load during the build-up process, the following circuit
relationships apply (Fig. 7.40).
Ia = If
(7.56(a))
V = Ea – If R a
(7.56(b))
The field current in a shunt generator being very small, the voltage drop If R a can be neglected so that
V ª Ea(If) (magnetization characteristic)
(7.57)
For the field circuit
(7.58)
V = If Rf
which is a straight line relationship, called the Rf -line, in V-If plot of Fig. 7.41.
The no-load terminal voltage is the solution of Eqs (7.58b) and (7.59). Thus the intersection point P of the
Rf -line with the magnetization characteristic as shown in Fig. 7.41 gives the no-load terminal voltage (V0)
and the corresponding field current. Further, it is easy to visualize from Fig. 7.41 that the no-load voltage can
be adjusted to a desired value by changing the field resistance. It can also be seen with reference to Fig. 7.42
that as the field resistance is increased the no-load voltage decreases. The no-load voltage is undefined for a
field resistance (Rf3 = Rfc) whose line coincides with the linear portion of the magnetization curve. With field
resistance even slightly more than this value, the machine does not excite to any appreciable value and would
give no-load voltage close to the residual value. The machine with this much resistance in the field fails to
excite and the corresponding resistance is known as the critical resistance (Rfc).
DC Machines
333
V
V
Air-gap line
Rf 3 = Rfc (critical resistance)
Rf – line
P
V0
Ea ªV (magnetization curve)
Rf 2
Rf1
V01
V02
Rf 3 > R f 2 > R f 1
lf
0
Field current
Fig. 7.41
lf
0
Fig. 7.42
resistance
Consider now the operation with fixed Rf and variable armature speed. It is observed from Fig. 7.43 that
as the speed is reduced the OCC proportionally slides downwards so that the no-load voltage reduces. At a
particular speed, called the critical speed, the OCC is tangential to the Rf -line and as a result the generator
would fail to excite.
V
Rf –line
n1
V01
n2
V02
n3 = nc = critical speed
n3 < n 2 < n 1
lf
0
Fig. 7.43
To summarize, a dc shunt generator may fail to self-excite for any of the three reasons mentioned below:
1. Residual magnetism is absent.
2. The field connection to the armature is such that the induced emf due to the residual magnetism tends
to destroy the residual magnetism (i.e. the feedback is negative).
3. The field circuit resistance is more than the critical value.
334 Electric Machines
Condition (2) can be remedied simply by reversing the field connection to the armature or reversing the
direction of rotation. In large dc generators with permanent connections and a fixed direction of rotation, the
problem is overcome by temporarily exciting the field from a battery source. (This is known as flashing.)
EXAMPLE 7.15
300 rpm:
If (A)
Voc (V)
The following figures give the open-circuit characteristics of a dc shunt generator at
0
7.5
0.2
93
0.3
135
0.4
165
0.5
186
0.6
202
0.7
215
The field resistance of the machine is adjusted to 354.5 W and the speed is 300 rpm.
(i) Determine graphically the no-load voltage.
(ii) Determine the critical field resistance.
(iii) Determine the critical speed for the given field resistance.
(iv) What additional resistance must be inserted in the field circuit to reduce the no-load voltage to 175 V.
SOLUTION
(i)
From the magnetization characteristic drawn in Fig. 7.44
V (no-load)|Rf
= 195 V
90
Rf (critical) =
= 450 W
0.2
71
= 236.7 rpm
n (critical) = 300 ¥
90
(ii)
(iii)
=354.5W
Rf = 354.5 W
225
195 V
Rf 2
200
175
150
VOC (V) h
125
100
90
75
71
50
25
0.44
0
0
0.1
0.2
0.3
0.4
lf (A)
Fig. 7.44
0.5
0.6
DC Machines
(iv)
335
V (no-load) = 175 V
175
Rf 2 =
397.7 W
0.44
Additional resistance to be inserted in field circuit
= 397.7 – 354.5 = 43.2 W
7.12 CHARACTERISTICS OF DC GENERATORS
With the advent of silicon-controlled rectifiers, the importance of the dc machine as a generator has
considerably reduced as SCRs can be employed to draw ac power from standard ac supply and convert it to
dc; also the dc voltage can also be varied with ease. For the sake of completeness the characteristics of dc
generators will be briefly discussed here, which are still found in older installations in industry (as a motorgenerator set for speed control of dc motors; see Sec. 7.14).
The load characteristic of a dc generator at a particular speed is the relationship between its terminal voltage
and load current (line current) and is also termed as the external characteristic. The internal characteristic is
the plot between the generated emf and load current.
IL = Ia
Separately-excited dc Generator
Figure 7.45 is that of a separately-excited dc generator. The
operation considered here assumes that the armature is driven at
constant speed (by means of prime mover) and the field excitation
(If) is adjusted to give rated voltage at no-load and is then held
constant at this value throughout the operation considered. The
armature circuit is governed by the equation
V = Ea – Ia R a ; Ia = IL
(7.59)
Rf
Ea
Ra
+
V
lf
Vf
–
n(constant)
Fig. 7.45
In spite of fixed excitation, Ea drops off with load owing to
the demagnetizing effect of the armature reaction (see Sec. 7.5). As the voltage drop is caused by magnetic
saturation effect, it increases with load nonlinearity. The internal characteristic (Ea – IL) is shown dotted in
Fig. 7.46. The external characteristic differs from the internal by the armature voltage drop la R a which is also
shown in Fig. 7.46.
V
V0
Armature reaction drop
Ea, Internal characteristic
laRa drop
V,external characteristic
laRa
laRa drop
0
ln
Fig. 7.46
IL = Ia
336 Electric Machines
Voltage Regulation
The voltage regulation of a generator (independent of the kind of excitation employed) is defined as
% regulation =
where
V0 - (V fl = Vrated )
(7.60)
Vrated
Vf l = full-load voltage = Vrated
V0 = no-load voltage corresponding to rated voltage at full-load excitation remaining unchanged*
Shunt Generator
A dc shunt generator is a self-excited generator.
The phenomenon of voltage buildup on no-load
and the conditions necessary for the same have
already been discussed in Sec. 7.11. Figure 7.47
shows a shunt-connected generator. With field
resistance adjusted to a certain value by means
of the regulating resistance, the desired no-load
voltage can be obtained (refer to Fig. 7.41). The
external characteristic of the generator can then be
obtained by a load test with total field resistance
remaining fixed in the process. The terminal
voltage drops off much more rapidly with load
in a shunt generator than in a separately-excited
generator because of fall in field current with
terminal voltage. The external characteristic is
a double-valued curve with a certain IL (max)
as shown in solid line in Fig. 7.48. As indicated
in Fig. 7.48, the useful parts of the external
characteristic is much before the turning point.
Internal characteristic is obtained from the
external characteristic by adding Ia Ra. At any
point P (V1, IL1) we find
+
la
+
Ea
Ra
V
–
–
Shunt field regulator
n = constant
Fig. 7.47
DC shunt generator on load
V
IaRa
V0
Internal characteristic
V1
External characteristic
Ia1 = IL1 + (V1/R f)
This locates the corresponding point on the
internal characteristic shown by dotted curve in
Fig. 7.48.
lL
lf
Rf = Total field resistance
Short
circuit
Fig. 7.48
IL1
IL2 (max)
IL
External Characteristic of dc shunt generator
Compound Generator
The causes of voltage drop in the terminal voltage from no-load to full-load in a shunt generator can be
partially/fully/over-compensated by use of an aiding series field (cumulative compound), which can be
connected in a long- or short-shunt (long-shunt shown in Fig. 7.49). The aiding ampere-turns of the series
* In case a shunt field winding is provided, it would mean that the total resistance in a filed circuit remains
unchanged in the operation.
DC Machines
337
field automatically increase with the load, compensating the armature voltage drop. In level-compound
generator full-load voltage equals no-load voltage. Steady-state volt-ampere (V-I) characteristics of a
compound generator are shown in Fig. 7.50. Differential compounding is not used in practice as the terminal
voltage falls off steeply with load. Long or short-shunt connection of series winding makes only a marginal
difference in the V-I characteristic of a compound generator. Compounding level can be adjusted by a diverter
in parallel to the series field; see Section 7.9, Fig. 7.31.
If
IL
+
Ia
Rf
ATse
Ea
ATsh
Rse
Ra
V
–
n(constant)
Fig. 7.49 Compound generator (long-shunt)
V
Over-compound
V0
Level-compound
Under-compound
Differential-compound
IL(rated)
0
IL
Fig. 7.50
Series Generator
A series generator is employed in series with a dc line to compensate for the line voltage drop which is
proportional to line current.
338 Electric Machines
Connection Diagram
A series self-excited dc generator connection has been given in Fig. 7.29(c). It is drawn in Fig. 7.51 along
with a load which can be switched on. The terminal voltage is given by
V = Ea – Ia (Ra + Rse)
(7.61)
IL = Ia
+
Rse
Series field
Ea
Load
R
V
Ra
–
n
Fig. 7.51
As the load current incurrent increase the excitation Nsc Ia increase and Ea rises sharply but levels off with
saturation and begins to drop due to armature reaction.
Characteristics
V
No-load (1)
No load test is conducted by exciting the series
field separately from a low voltage source. It is the
saturation curve VOC vs If = Ia drawn in Fig. 7.52
curve 1. Internal characteristic.
It is the induced emf Ea which is less than VOC by
the armature reaction voltage drop. The voltage drop
sharply increases with saturation and Ea beings to
reduce; curve 2. External characteristic lies below lies
below curve 2 by Ia (Ra + Rse) voltage drop, curve 3.
Condition for Self-excitation
Armature reaction drop
Internal (2)
External (3)
Ia(Ra + Rse)
IL = Ia
Fig. 7.52
V
Separately excited
The total resistance (Ra + Rse + RL) should be less
than the critical resistance RC as determined from the
saturation characteristic; curve 1.
It is observed from the external characteristic,
curve 3, that the maximum line boost is limited.
Series
Vrated
Shunt
Compound (cum)
Comparison of Volt Ampere (V-I ) Characteristics of
dc Generators
The VI characteristic (steady state) of various types of
dc generators is shown in Fig. 7.53. These and drawn
with rated terminal voltage at full-load.
IL
Fig. 7.53 V-I characteristics of dc generators
DC Machines
339
The external characteristic of a shunt generator can be predetermined from a knowledge of the OCC and
armature and field resistances (which can be determined by simple dc test), without actually performing
the load test which in fact is not feasible for a large generator. Assume for the time being that the armature
reaction voltage drop is negligible. With reference to Fig. 7.47.
Ia = IL + If
ª IL (shunt field current can be neglected in comparison to load current)
Ea = V + Ia Ra
V = If R f , (Rf – line)
Ea = Ea (If), the OCC
(7.62)
(7.63)
(7.64)
(7.65)
Because of the nonlinearity (saturation in the OCC), the solution of Eqs (7.62)-(7.65) to determine the
terminal voltage for given Rf and IaRa can be carried out graphically as shown in Fig. 7.54.
(V + IaRa) (Constant)
Rf-line
V
P
R
Ea(If), the OCC
No-load point
R¢
V1
Q
IaRa (Constant)
(IaRa)max
S
V2
Q¢
S¢
Vr
O
D
C
If
Predetermination of external characteristic of dc shunt generator
For given Ia Ra, Eq. (7.64) is represented by a straight line parallel to the Rf line and above it by vertical
intercept of IaRa. The intersection of this line with OCC gives two solution points R and S for the induced emf.
For these solution points the two terminal voltage values V1 and V2 can be read from the Rf line corresponding
to points R¢ and S¢. Maximum possible armature current corresponds to point Q at which a line parallel to
the Rf line is tangent to the OCC. Corresponding to each solution point, values of V and If can be read and
IL = Ia – If , can be calculated and subsequently the external characteristic plotted as shown in Fig. 7.48. The
short-circuit point on the external characteristic corresponds to IaRa = Vr , the residual voltage.
340 Electric Machines
It has been shown in Section 7.10 that the demagnetization caused by armature reaction can be quantified by
equivalent field current Ifd which can be taken as proportional to the armature current Ia. There are two ways
in which we can account for Ifd in predetermining the external characteristic of a shunt generator starting from
its OCC. With reference to Fig. 7.55
Shift the origin by Ifd to left along the If –axis. This is equivalent to shifting the OCC to right parallel
to the If –axis by Ifd from the new origin.
We shall use the first approach and illustrate the second through an example. A typical OCC is sketched in
Fig. 7.55 and the Rf line is drawn from the origin. Their intersection point P determines the no load voltage V0.
Constructional Steps Method
fi Located D to the left of origin such that OD = Ifd
fi Locate A vertically above D such that AD = IaRa for an assumed value of Ia (about the rated value)
V
Rf -line
P
V0
V1
OCC
No-load
R
R¢¢
Q
VQ
D¢
Q¢
S
V2
S¢
A
D
IaRa
O
If
Ifd
Fig. 7.55
DC Machines
341
fi From A draw a line parallel to the Rf line intersecting OCC at R and S
fi Shift DODA at R and S as shown in Fig. 7.55. This step locates R¢ and S¢ on the Rf line which yield the
two terminal voltages V1 and V2 for the assumed value of Ia.
fi For V1, V2 calculate If 1 = V1/Rf and If 2 = V2 /Rf
fi Calculate the corresponding line currents IL1 = Ia – If 1 and IL2 = Ia – If 2
fi Plot V1 vs IL1 and V2 vs IL2 on the V–IL coordinate plane
fi Repeat the above steps for different values of Ia
fi The complete plot is the external characteristic of double-valued shape as sketched in Fig. 7.48
fi To find Ia (max), draw a tangent to OCC parallel to the Rf line giving the point Q (corresponding
to Ia (max)). Draw now QQ¢ parallel to OA and the DQ¢D¢Q similar to DODA. QD¢ then gives max
Ia Ra. This construction is made possible by the fact the IaRa and Ifd are both proportional to Ia. It then
follows that
Ia (max) =
QD¢
Ra
The corresponding terminal voltage is VQ. The line current is
ILQ ¢ = Ia (max) – (VQ /R f)
This may be not be the exact IL (max)
For plotting the internal characteristic plot Ea1 = (V1 + Ia Ra) vs IL1 and Ea2 = (V2 + IaRa) vs IL1; similarly
for other assumed values of Ia. The internal characteristic is also plotted in Fig. 7.48 in dotted line.
EXAMPLE 7.16 A 100 kW, 200 V, long-shunt commutative compound generator has equivalent armature
resistance of 0.03 W and a series field resistance of 0.004 W. There are 1200 shunt turns per pole and
5 series turn per pole. The data of its magnetization characteristic at 1000 rpm is given below:
If (A)
Ea (V)
0
11
1
33
2.2
105
3.3
150
4.2
175
5.3
200
7.1
225
(a) Calculate the terminal voltage and rated output current for shunt field current of 5 A and a speed of
950 rpm. Neglect armature reaction effect.
(b) Calculate the number of series turns per pole for the generator to level compound with same
shunt field current, series field resistance and generation speed to be the same as in part (a). The
demagnetization due to armature reaction in equivalent shunt field current is 0.001875 Ia where Ia is
the armature current.
SOLUTION
The schematic diagram of long-shunt compound generator is drawn in Fig. 7.56(a)
Line current, IL =
100 ¥ 103
= 500 A
200
Shunt field current, If = 5 A (given)
From Fig. 7.56(a)
Ia = 500 + 5 = 505 A
N se
If, eq = If +
I
N f se
=5+
5
¥ 505 = 7.1 A
1200
342 Electric Machines
If
IL
+
Rse
Ia
Vt
Rf
Ea
Ra
–
Fig. 7.56(a)
From the magnetization characteristic of Fig. 7.56(b) (as per data given)
(a)
Ea = 225 V at 1000 rpm
950
Ea (950 rpm) = 225 ¥
= 213. 75 V
1000
From Fig. 7.56(a)
Ea(V)
Vt = Ea – Ia (Ra + Rse)
= 213.75 –505 (0.03 + 0.004)
250
The generator is under – compounded (armature
reaction ignored)
200
Ifd = 0.001875 ¥ 505
150
(b)
= 0.95 A (demag)
For level compound
Vt = 200 V
Ea = 200 + 505 (0.03 + 0.04)
= 217 .17 (950 rpm)
1000
Ea (1000 rpm) = 217.17 ¥
= 228.6 V
950
From the magnetization characteristic
100
50
1
2
3
4
5
6
7
8
9 If (A)
Fig. 7.56(b)
If (net) = 7.5 A
Excitation balance equation
If +
N se
I – I = I (net)
N f a fd f
Ê N ˆ
5 + 505 Á se ˜ – 0.95 = 7.5
Ë 1000 ¯
Nse = 6.87 or 7 turns
which gives
EXAMPLE 7.17 The data for magnetization characteristic of a dc shunt generator is as under:
If (A)
VOC(V)
0
7.5
0.1
62.5
0.2
120
0.3
152.5
0.4
175
0.5
192.5
0.6
205
0.7
215
DC Machines
343
The shunt field has a resistance of 354.5 W and the armature resistance is 0.5 W.
Determine
(a) the no-load voltage
(b) the terminal voltage at an armature current of 40 A
(c) the maximum possible armature current and the corresponding terminal voltage
(d) the short circuit armature current
SOLUTION
The magnetization characteristic (OCC) is drawn in Fig. 7.57.
Rf = 354.5 W
225
OCC
Q
200
VOC(V)
175
Q¢
150
P
125
100
75
50
25
P¢
R
R¢
0.1
0.2
0.3
0.4 0.5
If (A)
0.6
0.7
Fig. 7.57
(a) Rf = 354.5 W line is drawn from the origin. Its intersection with OCC gives no-load voltage V0 = 200 V
(b) Ia = 40 A, Ia Ra = 40 ¥ 0.5 = 20 V
We have drawn a line parallel to the Rf line and 20 V (vertically) above it. It intersects the OCC at Q and R. The
points Q¢ and R¢ vertically below on the Rf line give the voltages at Ia = 40 A, 170 V, 25 V
The generation will be operated at 170 V.
(c) Draw a line tangent to the OCC. It locates the point P from which we locate P¢ vertically below it on the Rf line.
It is found that
IaRa (max) = PP¢ = 50 V
or
Ia (max) = 50/0.5 = 100 A
(d) Short circuit current
7.5
Vr
Ia(sc) =
=
= 15 A
Ra
0.5
344 Electric Machines
EXAMPLE 7.18 A 100 kW, 200 V, 1000-rpm dc shunt generator has Ra = 0.03 W. The generator is driven
at rated speed and excitation is such that it gives a rated voltage of 200 V at no-load. The data for the
magnetization characteristic of the generator are as follows:
I *f (A)
EMF(V)
0
10
1
32.5
2.2
100
3.3
167.5
4.5
200
7.1
225
Determine the voltage appearing across the generator terminals when Ia = 500 A
A series field of 5 turns per pole having a total resistance of 0.005 W is to be added in a long-shunt
compound. There are 1200 turns per pole in the shunt field. The generator is to be level-compounded so that
the full-load voltage is 200 V when the resistance in the shut field circuit is adjusted to give a no load voltage
of 200 V. Find the value of the series field diverter resistance to obtain the desired performance.
Assume that the armature reaction AT has been compensated for.
Rf -line
R
225
17.5 V
P
200
VOC(V)
175
Q
15 V
Q¢
150
125
If, se 98 = 1.7 A
100
75
P¢
50
4.5 A
25
1
2
3
4
5
If (A)
6
7
Fig. 7.58
SOLUTION
The magnetization characteristic is drawn in Fig. 7.58. Drawing the Rf -line corresponding to 200 V.
Rf =
200
= 44.4 W
4.5
Ia = 500 A
IaRa = 500 ¥ 0.03 = 15 V
Drawing a line parallel to the Rf -line at 15 V (= QQ¢)
V(terminal) = 165 V (corresponding to Q¢)
DC Machines
345
With series winding added in the long-shunt compound (cumulative):
Armature circuit resistance = 0.03 + 0.005 = 0.035 W
Armature circuit voltage drop = 500 ¥ 0.035 = 17.5 V
To compensate for this voltage drop, the operating point on OCC must lie at R, i.e. (200 + 17.5) = 217.5 V. Both fullload and no-load voltage will now be 200 V. From Fig. 7.58 the series field current measured in terms of shunt of field
current is
If, se = 1.75 A
ATse = If, se ¥ 1200 = 1.75 ¥ 1200 = 2100
2100
= 420 A
5
= 500 – 420 = 80 A
420
= 0.005 ¥
= 0.0265 W
80
Ise =
Idiverter
Rdiverter
EXAMPLE 7.19
If (A)
VOC (V)
1
22
The open-circuit characteristic of a shunt generator at speed is
2.5
231
5
400
7
479
9
539
12
605
15
642
18
671
The field and armature resistances are 46 W and 0.12 W respectively. Estimate the terminal voltage when
the armature current is 360 A in two cases (a) Armature reaction ignored (b) 1 A, field current is needed to
counteract the effect of armature reaction.
SOLUTION
OCC is plotted in Fig. 7.59 Rf -line is also drawn thereon.
700
VOC(V)
OCC
C
A
600
OCC (shifted by 1A)
B
556 V
520
D
500
1A
Rf = 46 W
400
43.2 V
300
200
100
If = 10.8 A If = 12 A
4
8
12
16
20
If (A)
Fig. 7.59
346 Electric Machines
(a) Ia = 360 A
Armature voltage drop = 360 ¥ 0.12 = 43.2 V
Drawing line parallel to Rf -line 43.2 V above it along the V axis, the terminal voltage and corresponding field
current are at point B on the Rf -line. These values are
Vt = 566 V, If = 12 A
(b) Field current equivalent of armature reaction caused demagnetization = 1 A. To account for this the OCC shifted
to right by 1 A. The shifted OCC intersects Rf -line at C. The corresponding point D on the Rf -line yields the
terminal voltage and field currents as
Vt = 520 V, If = 10.8 A
The empirical procedure to shift OCC to right by a certain amount of field current applies in the
region around full load current which as is seen from Fig. 7.59 is the region of interest.
Remark
EXAMPLE 7.20 The OCC and Rf = 167 W line are drawn to scale in Fig. 7.60 for a dc shunt generator
with armature resistance of 0.5 W. The generator is driven at constant speed. It is estimated that at armature
V
300
Rf = 167W
Parallel to OCC
P
R
250
R¢
D
200
A
D¢
150
100
50
1.5 A
0.4
0.8
1.2
1.4
1.6
If
Fig. 7.60
occ
DC Machines
347
current of 40 A, the armature reaction demagnetization has field current equivalent of 0.14 A.
Estimate thee acceptable terminal voltage when the armature is delivering 50 A. Also find the line current.
SOLUTION
Using proportionality
50
= 0.175 A
40
Ia Ra = 50 ¥ 0.5 = 25 V
Ifd|at Ia = 50 A = 0.14 ¥
Choose an arbitrary point D¢ on the Rf line. Draw a right angle triangle D¢AD with D¢A = 0.175 A (horizontal) –D¢AD
= 90°, AD = 25 (vertical). Through D draw a line parallel to the Rf line interacting the OCC at R (higher voltage point)
Now draw RR¢ parallel to DD¢ locating R¢ on the Rf line. Reading from R¢
V (terminal) = 232.5 V
If = 1.5 A
IL = Ia – If = 50 – 1.5 A = 48.5 A
Then
EXAMPLE 7.21 A 20 kW, 240 V shunt generator is driven by a prime mover whose speed drops uniformly
from 1190 rpm at no-load to 1150 rpm at full-load. By adding a series winding it is converted to a short
shunt compound such that its voltage rises from 230 V at no-load to 240 V at Ia = 83.3 A. The resistance
of the series winding is estimated to be 0.045 W. The armature resistance including brush voltage drop is
found to be 0.12 W. The shunt field turns are 550/pole.
The generator is separately excited and the following tests are conducted.
No-load Test (1190 rpm)
VOC (V)
If (A)
220
1.0
230
1.15
240
1.35
250
1.50
Load Test (1150 rpm)
Armature terminal voltage = 240 V
Armature current = 83.3 A
Field current = 2.1 A
Determine
(a) the demagnetizing AT/pole at armature current of 83.3 A
(b) the number of series turns required
SOLUTION
The no-load test data is plotted in Fig. 7.61(b)
(a) Load Test – 1150 rpm
Ea = 240 + 83.3 ¥ 0.12 = 250 V
Looking up mag. curve we find Ea at 1190 rpm (OCC prime mover speed)
1190
= 258.7 V
1150
If = 2.1 A (given)
Ea (1190 rpm) = 250 ¥
The field current needed to induce 258.7 V on no-load
If (net) = 1.65 A
Demagnetization in terms of field current
Then
Ifd = 2.1 – 1.65 = 0.45 A
ATd = 550 ¥ 0.45 = 247.5
260
1.69
270
2.02
348 Electric Machines
(b) Series turns added – compound generator – short shunt. The connection diagram is drawn in Fig. 7.61(a). The
performance required:
No-load (1190 rpm)
We can assume
+
Vt = 230 V
Ea ª 230 V
Ia
Consulting mag. curve, we find
Rf =
Ea
Rf
If = 1.43 ; ATf = 1.43 ¥ 550 = 786.5
Then
Rse
Va
If
230
= 161 W remains constant
1.43
Fig. 7.61(a)
Vt = 240 V, Ia = 83.3 A
V
IL = Ia – If = 83.3 – a ; Va = armature voltage
Rf
V ˆ
Ê
Va = 240 + 0.045 I L = 240 + Á 83.3 - a ˜ ¥ 0.045
Ë
161¯
Ê 0.045 ˆ
Va Á1 +
= 240 + 83.3 ¥ 0.045 = 243.7
Ë
161 ˜¯
1190 rpm
280
260
Voc
240
230
220
210
200
1.5
2.0
If (A)
Fig. 7.61(b)
Vt
–
Load (1150 rpm)
1.0
IL
DC Machines
or
Va = 243.7 V
Then
Ea = 243.7 + 83.3 ¥ 0.12 = 253.3 V
349
To consult mag. curve
Ea (1190 rpm) = 253.3 ¥
1190
= 262.1 A
1150
If (needed) = 1.675 A
AT (needed) = 550 ¥ 1.675 = 921.25
If =
Va
243.7
= 1.51 A
=
Rf
161
ATf = 550 ¥ 1.51 = 830
AT balance equation
ATf + ATse – ATd = AT (needed)
830 + ATse – ATd = 925.25
AT se = 343
or
IL = 83.3 – 1.51 ª 81.0 A
Nse =
343
ª 4 turns /pole
81.8
EXAMPLE 7.22 A 50 kW, 250 V, 200 A long shunt cumulative compound dc generator has the following
data:
Ra (inclusive of brush voltage drop) = 0.05 W
Series field resistance Rse = 0.01 W
The magnetization characteristic at 1200 rpm is drawn in Fig. 7.62(b)
Shunt field turns, Nf = 1000
Series field turns, Nse = 3
The generator is run at 1150 rpm and its shunt field current is adjusted to 5.6 A.
compute the terminal voltage at rated voltage current.
SOLUTION
The connection diagram of long shunt compound generator is drawn in Fig. 7.62(a).
Load current, IL = 200 A
IL
Ia = 200 + 5.6 = 205.6 A
Excitation ampere-turns 1000 ¥ 5.6 + 205.6 ¥ 3 = 6217
Equivalent shunt field current, If, eq =
6217
ª 6.22 A
1000
Rf
From the mag. characteristic
Ea = 282 V at 1200 rpm
At
Terminal voltage
If
1150 rpm, Ea = 282 ¥
Rse
Ia
Vt
Ea
1150
= 270 V
1200
Vt = 270 – 205.6 ¥ (0.05 + 0.01)
= 257.7 V
+
–
Fig. 7.62(a)
350
Electric Machines
320
Voc
n = 1200
300
Nf = 1000
280
260
240
220
200
180
160
140
120
100
80
60
40
20
0
1
2
3
4
5
6
7
8
9
10
If
Fig. 7.62(b)
EXAMPLE 7.23 Consider the dc generator of Example 7.22. What should be the number of series for a
terminal voltage or 250 A at IL = 200 A. The demagnetizing ampere-turns of armature reaction at Ia = 200 A
are 400. Shunt field current is adjusted to 5.6 A.
DC Machines
351
ATf = 5.6 ¥ 1000 = 5600
Ia = 205.6 A
205.6
ATd = 400 ¥
= 411.2
200
Ea = 250 + 205.6 ¥ 0.06 = 262.3 V at 1150 rpm
1200
1200 rpm, Ea = 262.3 ¥
= 273.7 V
1150
SOLUTION
At
From the mag. characteristic
If (net) = 6.2 A
AT (net) = 1000 ¥ 6.2 = 6200
AT balance equation
AT (net) = ATf + ATse – ATd
6200 = 5600 + ATse – 411.2
ATse = 1011.2
1011.2
Nse =
= 4.9 or 5 turns
205.6
We conclude that 2 additional series turns are needed are to counter armature reaction effect.
or
EXAMPLE 7.24 A 20 kW, 250 V separately excited dc generator is run at constant speed of 1000 rpm has
an armature resistance of 0.16 W. The magnetization characteristic is presented in Fig. 7.63.
(a) At rated armature current find the generator output at constant field current of (i) 1 A, (ii) 2 A, and
(iii) 2.5 A.
(b) Repeat part (a) if the generator speed is decreased to 800 rpm.
Note: Neglect armature reaction.
20 ¥ 103
= 80 A
250
(a) (i) If = 1 A From the mag. characteristic Ea = 150 V
Ia = 80 A
Ra = 0.16
Internal voltage drop = 80 ¥ 0.16 = 12.81
Vt = 150 – 12.8 = 137.2 V
Generator output, P0 = 137.2 ¥ 80 = 10.976 kW
Ea = 257.5 V
(ii)
If = 2 A
Vt = 257.5 – 12.8 = 238.2
P0 = 238.2 ¥ 80 = 19 kW
SOLUTION
(iii)
Rated armature current =
If = 2.5 A
Ea = 297.5 V
Vt = 297.5 – 12.8 = 284.7 V
P0 = 284.7 ¥ 80 = 22.8 kW
(b) Generator speed increases to 1200 rpm
The magnetization characteristic at each point shifts downwards by a factor of 800/1000 = 0.8
(i)
Ea = 150 ¥ 0.8 = 120 V
Vt = 120 – 12.8 = 107.2 V
P0 = 107.2 ¥ 80 = 8.576 kW
(ii)
Ea = 257.5 ¥ 0.8 = 206 V
P0 = 206 ¥ 80 = 16.5 kW
352 Electric Machines
300
occ
Ea (1)
250
Vo (V)
Ia = 80 A
200
Vt
Ia Ra = 12.8 V
150
100
50
0
0.5
1.0
1.5
If (A)
2.0
2.5
Fig. 7.63
(iii)
Ea = 295 ¥ 0.8 = 236 V
Vt = 236 – 12.8 = 223.2 V
P0 = 223.2 ¥ 80 = 17.856 kW
EXAMPLE 7.25 The separately excited generator of Example 7.24 feeds a load resistance of 3 W. Find
the power fed to the load at field current 1 A, 2 A and 25 A. Neglect armature reaction.
SOLUTION
Load resistance = 3 W
If = 1 A
Ea = 150 V
150
Ia =
= 47.67 A
0.16 + 3
Power fed load, P0 = (47.67)2 ¥ 3 = 6.808 kW
If = 2 A
Ea = 240 V
DC Machines
Ia =
353
257.5
= 81.5 A
0.16 + 3
P0 = (81.5)2 ¥ 3 = 19.93 kW
If = 2.5 A
Ea = 297.5 V
297.5
Ia =
= 94.15 A
0.16 + 3
P0 = (94.15)2 ¥ 3 = 26.56 kW
EXAMPLE 7.26 The separately excited dc generator of Example 7.24 is run at 1000 rpm and the load
test is conducted. At each field current. the load current is adjusted to 80 A and the readings of the terminal
voltage taken. The data recorded is as under:
If (A)
Vt (V)
2.5
240
2.0
200
1.0
120
Find the armature reaction demagnetization in terms of field current (Ifd ) at each data point.
SOLUTION On Fig. 7.63 the magnetization characteristic Vt – If is now plotted on it. To this plot, we add IaRa = 80 ¥
0.16 = 12.8 V (vertically). This gives us the internal characteristic Ea(i) – If ; shown dotted.
Ifd is the field current difference between Ea(i) and Ea(open circuit) at given field current. Ifd at the three values of If
are shown by horizontal lines between magnetization characteristic (Ea(open circuit)) and internal characteristic (Ea(i)).
The data is recorded below:
If (A)
Ifd (A)
1
0.2
2
0.75
2.5
1
It is observed that Ifd increases as the magnetic circuit gets into saturation region.
EXAMPLE 7.27 The OCC of a dc shunt generator is drawn to scale in Fig. 7.64. The generator has an
armature resistance of 0.5 W. The generator is run at constant speed.
(a) Find the field resistance for a no-load voltage of 230 V and the field current.
(b) At an armature current of 80 A find the terminal voltage, field current and load power.
(c) It is required that the armature current be 80 A. What would be the terminal voltage accounting for
5% flux reduction ? Also calculate load power.
SOLUTION
Locate 230 V, point P on the OCC. Draw the Rf line from the origin to P. We find
\ Rf = 230/1.3 = 176. 9 W
(a)
If = 1.3 A
(b)
Ia = 80 A, Ra = 0.5,
Ia Ra = 40 V
Draw a line parallel to Rf line 40 V above it, It intersects OCC at P.
The point S below it on Rf line in the terminal voltage
Vt = 202 V
Corresponding field current is
If = 202/176.9 = 1.14 A
354
Electric Machines
Ia = 80 A,
Ra = 0.5 W
Ia Ra = 80 ¥ 0.5 = 40 V
(b) Given
Between OCC and Rf line locate (by scale) vertical line RS = 40 V. The point given the terminal voltage
Vt
If
Line current, IL
Load power
Corresponding
= 200 V
= 1.07 A
= 80 – 1.07 = 78.93 A
= 200 ¥ 78.93 = 15.786 kW
Rf line
Ea(V)
R
260
OCC
P
220
R¢
200
S
S¢
180
140
100
80
60
40
0
0.2
0.4
0.6
0.8
1.0
If(A)
Fig. 7.64
1.2
1.4
DC Machines
355
(c) Induced emf curre with 5% reduction (Corresponding to 5% reduction in flux/pole) is drawn in dotted line in
Fig. 7.64. Its intersection with line parallel to Rf line 20 V above it is R¢. The corresponding point S on Rf line gives
Vt = 182 V
If = 182/176.9 = 103 A
IL = 40 – 1.03 = 38.97 A
Load power = 182 ¥ 38.397 = 7.09 kW
EXAMPLE 7.28
A 75 kW, 250 V compound dc generator has the following data:
Ra = 0.04 W, Rse = 0.004 W
Brush contact drop, Vb = 2 V (1 volt each brush)
Rf = 100 W
Compare the generator induced emf when fully loaded in (i) long shunt compound, and (ii) short shunt
compound
SOLUTION
Long shunt and short shunt connection are drawn in Figs 7.65(a) and (b).
If
If
IL
IL
+
Rse
Ia
Rse
Ia
1V
1V
Rf
Ea
Vt
Ra
Rf
Ra
Ea
Vb
Vt
1V
1V
–
(a) Long shunt
(b) Short shunt
Fig. 7.65
At full load
Vt = 250 V, IL =
75 ¥ 103
= 300 A
250
Long Shunt (LS)
If = 250/100 = 2.5 A, Ia = 300 + 2.5 = 302.5 A
Ea = 250 + (0.04 + 0.004) ¥ 302.5 + 2 = 265.31 A
Short Shunt (SS)
Vb
If
Ia
Ea
= 250 + 0.004 ¥ 300 = 251.2 V
= 251.2 /100 = 2.512 A
= 300 + 2.512 = 302. 512 A
= 250 + 0.04 ¥ 302.512 + 2 = 264.1 V
356 Electric Machines
Ea (LS) – Ea (SS) = 265.31 – 264.1 = 1.21 V
1.21 ¥ 100
Difference as percentage of Vt =
= 0.484%
250
Conclusion: There is no significant difference in LS and SS.
EXAMPLE 7.29 A 25 kW, 220 V, 1600 rpm dc shunt generator with Ra = 0.1 W has magnetization
characteristic data given below:
If (A)
Ea (V)
0.0
10
0.25
90
0.50
150
0.75
190
0.1
220
1.25
243
1.5
250
(a) what would be the current and field resistance at terminal voltage of 220 V?
(b) At rated current and rated terminal voltage, find the value of field current and field resistance. Ignore
the effect of armature reaction.
(c) Find the value of electromagnetic power and torque in part (b).
(d) Under load conditions in part (b), the field current is If = 1.25 A. Find the field current needed to
counter the effect of armature reaction.
SOLUTION
The magnetization characteristic drawn in Fig. 7.66.
260
240
220
200
160
120
80
40
0
0.25
0.5
0.75
1.0
Fig. 7.66
1.25
1.5
1.75
2.0
DC Machines
(a) No-load voltage,
Corresponding,
(b)
357
V0
If
Rf
Vt
= 250 V
= 1.5 A
= 250/1.5 = 167 W
= 220 V
25 ¥ 103
IL =
= 113.6 A
220
As If is small, we can assume
Ia ª IL = 113.6 A
Ea = 220 + 113.6 ¥ 0.1 = 231.4 V
Ignoring armature reaction, we find from magnetization characteristic
If = 1.1 A (about 1% of IL)
R f = 220/1.11 = 198.2 W
(c) Electromagnetic power = Ea Ia = 231.4 ¥ (113.6 – 1.1)
= 231.4 ¥ 112.5 = 26.033 kW
26.033 ¥ 103
= 155.37 Nm
2p
1600 ¥
60
(d) Under load condition as in part (b)
Electromagnetic torque =
If = 1.1 A, armature reaction ignored
Actual If = 1.25 A
If needed to counter effect of armature reaction = 1.25 – 1.1 = 0.15 A
7.14
PARALLEL OPERATION OF DC GENERATORS
For supplying a large dc load it is desirable to use more than one generator in parallel. This arrangement
provides the security that if one generator gives way, the other(s) can feed part load.
Desirable Conditions for dc Generators in Parallel
fi Same voltage rating
fi Same percentage voltage regulation
fi Same percentage speed regulation of the prime movers
As the generator voltage is easily adjustable (in a range) so the above conditions are not a must.
Paralleling a dc Generator to Busbars (Mains)
We will consider the case of dc shunt generators.
A dc shunt generator (G1) is connected to the busbars (switches S, S1 closed) and feeding the load as shown in
Fig. 7.67(a). I1 = IL. Another shunt generator G2 is to be connected to the busbars to share part load relieving
the load on G1. The stepwise procedure for this operation is described below:
1. The generator G2 is driven by its prime mover and brought to rated speed
2. The switch S of G2 is closed. The centre-zero voltmeter across the switch S2 reads the voltage difference
between busbars and voltage of G2
358
Electric Machines
3. The voltage of G2 is adjusted by its shunt field regulator till the voltage across S2 is nearby zero. The
switch S2 is then closed. G2 now floats on the busbars without exchanging any current.
4. The regulator resistance of G2 is now reduced (increasing its field current) so that it feeds current I2
to busbars. The regulating resistance of G1 is now increased (reducing its field current) so that it feeds
smaller current to the busbar. The adjustment process finally results in proper current sharing between
the two generators such that I1 + I2 = IL
5. In the above adjustment the bus bar voltage is maintained constant
Determination of Load Sharing
The load sharing between the generators is determined by their external characteristics. Shown in Fig. 7.67(b)
are the external characteristics of the two generators with their field currents adjusted for the same no-load
voltage. The combined characteristic of the system is also drawn. At busbar voltage Vbus the total load current
is supplied by the two operators such that
I1 + I2 = IL
The adjustments of the field currents modifies the external characteristic changing the load sharing.
+
I1
S1
I2
V
IL
S2
Load
G1
G2
S
S
–
Fig. 7.67(a)
Parallel Operation of Compound Generators
Two compound generators in parallel feeding a load are sketched in Fig. 7.68. With switches S1 and S2
closed they are sharing the load. This system is found to be unstable because of the positive voltage feedback
through series windings. We will present the qualitative arguments.
If because of any reason the emf of G1 increases, it causes its load current I1 to increase and correspondingly
I2 to decrease. The series excitation of G1 increases and that of G2 decreases and so the internal voltage of G1
increases further and that of G2 decreases. Consequently current I1 fed by G1 to load sharply increases and
that of G2 sharply decreases. Finally all the load shifts to G1 from G2. In fact G2 may begin to act as a motor.
All this leads to heavy overloading of G1, an unacceptable operation.
DC Machines
V0
359
Combined
V(bus)
2
1
I1
I2
IL
Load current
Load sharing: IL = I1 + I2
Fig. 7.67(b)
Remedy
A low-resistance equalizer connection is made directly between the two armatures before the series fields as
shown in Fig. 7.68. Any emf variations of the armatures causes equalizing circulating current which do not
affect the current through the series windings. Thereby the parallel operation is stabilized.
+
I1
I2
S2
S1
IL
Equalizer
Load
G1
G2
–
Fig. 7.68 Compound generators in parallel S1 and S2 closed
360
Electric Machines
EXAMPLE 7.30 Two dc shunt generators are rated 230 kW and 150 kW, 400 V. Their full load voltage
drops are 3% and 6% respectively. They are excited to no load voltages of 410 V and 420 V respectively.
How will they share a load of 1000 A and the corresponding bus voltage?
SOLUTION
G1
If1 =
250 ¥ 103
3
= 625 A; Full load voltage drop = 400 ¥
= 12 V
400
100
G2
If2 =
150 ¥ 103
6
= 375 A; Full load voltage drop = 400 ¥
= 24 V
400
100
Voltage of G1 at load current I1
Ê 12 ˆ
I1
V1 = 410 – Á
Ë 625 ˜¯
(i)
Voltage of G2 at load current I2
Ê 24 ˆ
I2
V2 = 420 – Á
Ë 375 ˜¯
Load on generation in parallel, IL = I1 + I2 = 1000 A, Bus voltage = V (?)
V1 = V2 = V
In parallel operation
(ii)
(iii)
From Eqs (i) and (ii)
Ê 12 ˆ
Ê 24 ˆ
I1 = 420 – Á
I2
410 – Á
Ë 625 ˜¯
Ë 375 ˜¯
0.064 I2 – 0.0192 I1 = 10
I1 + I2 = 1000
(iv)
(v)
Solving Eqs (iv) and (v), we get
I1 = 649 A, I2 = 351 A
Ê 12 ˆ
¥ 649 = 397.5 V
Bus voltage, V = 410 – Á
Ë 625 ˜¯
Load = 103 ¥ 397.5 = 397.5 kW
EXAMPLE 7.31 In Example 7.30, the two generators are excited to equal no-load voltages. What should
be the percentage voltage drop of 150 kW generator in order that the share load is in the ratio of their
ratings? What is the no load voltage for a bus voltage of 400 V and load current of 1000 A?
SOLUTION
Let the full load voltage drop of G2 be x volts. Then
Ê 12 ˆ
I1
V1 = V 0 – Á
Ë 625 ˜¯
(i)
Ê x ˆ
I2
V2 = V0 – Á
Ë 375 ˜¯
(ii)
For parallel operation
V1 = V2 = V
So from Eqs (i) and (ii)
Ê x ˆ
Ê 12 ˆ
ÁË 375 ˜¯ I2 = ÁË 625 ˜¯ I1
(iii)
DC Machines
We want
I1
250
5
=
=
I2
150
3
361
(iv)
Solving (iii) and (iv)
I1
625 x
5
=
=
I2
12 ¥ 375
3
or
x = 12 V
12
Percentage voltage drop of G2 =
¥ 100 = 3%
400
Hence the percentage voltage drop of the two generators must be equal
I1 + I2 = 1000
I1
5
=
I2
3
which give
or
I1 = 625 A, I2 = 375 A
Ê 12 ˆ
¥ 625 = 400 V
VBus = V0 – Á
Ë 625 ˜¯
V0 = 412 V
7.15 CHARACTERISTICS OF DC MOTORS
The power of the dc motor lies in its versatility and ease with which a variety of speed-torque characteristics
can be obtained, and the wide range of speed control which is possible without the need of elaborate control
schemes while a high level of operating efficiency is maintained. In a dc generator the speed is fixed by the
primemover and remains nearly constant throughout the operating part of the characteristics,
While the field excitation is adjusted to yield the desired terminal voltage at a given load. In a motor, on the
other hand, the need is to match the speed-torque characteristic of the load and to run the load at a specified
speed or speed by adjustment of the field and the armature voltage in case the speed control over a wide range
is required.
The fundamental emf and torque relationships of Eqs. (7.29) and (7.30) are reproduced below:
Induced emf, Ea =
Electromagnetic torque, T =
where
and
Ka =
FnZ Ê P ˆ
= Ka Fw V
60 ÁË A ˜¯
(7.66)
1
Ê Pˆ
FIa Z Á ˜ = KaFIa Nm
Ë A¯
2p
(7.67)
ZP
2p A
Ê 2p ˆ
w = Á ˜ n = speed in rad (mech)/s and n = speed in rpm
Ë 60 ¯
(7.68)
In place of electromagnetic torque may also use the term developed torque with symbol T. For repeated
use we may use the term torque only.
In motoring operation it is convenient to express armature speed in rpm. So we write the above relationships
as
362
Electric Machines
n=
1
K a¢
Ê 2p ˆ
Ê Ea ˆ
ÁË F ˜¯ ; K¢a = ÁË 60 ˜¯ Ka
(7.69)
T = Ka F Ia
(7.70)
The induced emf Ea in a motor is known as back emf as it opposes the applied terminal voltage Vt . It is
related to the terminal voltage by the armature circuit equation. For the short-compound motor (given in
Fig. 7.30(b); Eq. 7.53(a))
Ea = Vt – Ia (Ra + Rse)
(7.71)
The flux/pole F is found from the magnetization characteristic against the equivalent excitation
ÊN ˆ
If,eq = I f ± Á se ˜ Ia
Ë Nf ¯
(7.72)
corrected for the demagnetizing effect of armature reaction.
As in the case of dc generators depending on the type of excitation there are three types of dc motors –
shunt motor, series motor and compound motor. Self-excitation has no meaning for a dc motor.
Three important operating characteristic of dc motor are
In this section, we shall examine the nature of the three characteristics mentioned above on quantitativequalitative basic because of magnetic saturation and armature reaction caused demagnetization.
Shunt Motor
The connection diagram of dc shunt motor is shown in Fig. 7.69. Its operation with fixed terminal voltage and
constant field current (fixed field resistance) will now be considered.
IL
If
Shunt field
winding
+
+
Ia
Ea
Ra
Vt
Rf
–
–
Fig. 7.69 Shunt motor
Speed-current characteristic
The armature circuit equation is
Ea = Vt – Ia Ra = K¢a F n
(7.73)
which gives
n=
Vt - I a Ra
K a¢ F
(7.74)
DC Machines
363
which indeed is the speed-current characteristic. At no load the armature current Iao is quite small (2 – 5% of
Ia ( f l )), we will, therefore, assume Iao = 0 for sketching the characteristic. The no load speed is then
Ê Vt ˆ
n0 = Á
Ë K a¢ F ˜¯
(7.75)
n, T
n
n0
n0
Speed AR
neglected
Speed
Speed, armature
reaction neglected
Torque
Torque, armature reaction neglected
Ia
0
T
0
(a) n versus Ia and T versus Ia
(b) n versus T
Fig. 7.70 Shunt motor characteristics (Vt , If constant)
Armature reaction effect ignored.
F remains constant. It is seen from Eq. (7.74) that speed drops linearly due to IaRa drop. The characteristic
is drawn in dotted line in Fig. 7.70(a).
Effect of armature reaction caused demagnetization.
F reduces with increasing Ia. It then follows from Eq. (7.74) that the speed at any Ia is higher than in the
armature reaction neglected case. However, the effect of IaRa drop predominates and the speed drop with Ia.
Of course, the no load speed is the same as given by Eq. (7.77a). The characteristic is drawn in solid line in
Fig. 7.70(a) which lies above the characteristic with armature reaction effect ignored.
Torque-current characteristic
As per Eq. (7.70), motor torque is
T = Ka FIa
(7.76)
The characteristic is linear if armature reaction effect is ignored as in shown Fig. 7.70(a) by dotted line.
Otherwise reduction in F causes the characteristic to bend downwards as shown by solid line in Fig. 7.70(a).
Speed-torque characteristic
Eliminating Ia from Eqs (7.74) and (7.76), the speed-torque characteristic is
obtained as
Ê Vt ˆ
È
Ra
˘
-Í
˙T
n= Á
Ë K a¢ F ˜¯ ÍÎ K a¢ K aF 2 ˙˚
(7.77)
If the armature reaction is ignored F remains constant. Therefore the speed drops linearly as per Eq. (7.77).
The no load speed is
Ê V ˆ
n0 = Á t ˜
Ë K a¢ F ¯
The characteristic is drawn in dotted line in Fig. 7.70(b).
(7.77a)
364
Electric Machines
Armature reaction effect causes F to reduce with increasing torque (and so increasing current). The speed
drops more sharply with torque than in the linear case because of F2 in the denominator of the second them
in Eq. (7.77). The characterstic is skeeched in solid line in Fig. 7.70(b).
Sum-up The speed drops from no load to full load by a few per cent; in fact the speed remains substantially
constant. Such a characteristic is known as “shunt characteristic”.
EXAMPLE 7.32 An 8 kW, 230 V, 1200 rpm dc shunt motor has Ra = 0.7 W. The field current is adjusted
until, on no load with a supply of 250 V, the motor runs at 1250 rpm and draws armature current of 1.6 A.
A load torque is then applied to the motor shaft, which causes the Ia to rise to 40 A and the speed falls to
1150 rpm. Determine the reduction in the flux per pole due to the armature reaction.
SOLUTION
Equation (7.73) gives
Ê V - I a Ra ˆ
ÊE ˆ
n = K¢a Á a ˜ = K¢a Á
˜¯
Ë
ËF¯
F
Ê V - I a Ra ˆ
F = K¢a Á
˜¯
Ë
n
or
Ê 250 - 1.6 ¥ 0.7 ˆ
F(no-load) = K¢a Á
˜¯ = 0.2 K¢a
Ë
1250
Ê 250 - 40 ¥ 0.7 ˆ
F(load) = K¢a Á
˜¯ = 0.193 K¢a
Ë
1150
Reduction in F due to the armature reaction
=
0.2 - 0.193
¥ 100 = 3.5%
0.2
EXAMPLE 7.33 A 20 kW, 250 V dc shunt motor has a full-load armature current of 85 A at 1100 rpm.
The armature resistance is 0.18 W.
Determine:
(a) the internal electromagnetic torque developed;
(b) the internal torque if the field current is suddenly reduced to 80% of its original value;
(c) The steady motor speed in part (b) assuming the load torque to have remained constant.
Assume: magnetic circuit to be linear.
SOLUTION
Ea = 250 – 0.18 ¥ 85 = 234.7 V
Ea Ia = Tw
(a)
2p ¥ 1100
60
T = 173.2 Nm
234.7 ¥ 85 = T ¥
or
(b) Magnetic circuit linearity is assumed i.e. F μ If . Field suddenly reduced to 0.8 of original value. Ia is assumed to
remain constant for that instant. Then
T1 = 0.8 ¥ 173.2 = 138.6 Nm
DC Machines
365
(c) Under steady condition for the motor internal torque to build to the original value, new values of armature current
and speed are established. These are obtained below:
T = Ka¢ Kf If ¥ 85 = Ka¢ Kf ¥ 0.8 If ¥ Ia1 = 173.2 Nm
Ia1 = 106.25 A
or
Ea1 = 250 – 0.18 ¥ 106.25 = 230.9 V
234.7 = Ka¢ Kf If ¥ 1100
and
230.9 = Ka¢ Kf¢ 0.8 If n1
230.9
0.8 n1
=
234.7
1100
n1 = 1353 rpm.
Dividing we get
or
Observe that both motor speed and armature current increase by reducing field current with constant load torque.
EXAMPLE 7.34 A dc shunt motor runs at 1200 rpm on no-load drawing 5 A from 220 V mains. Its
armature and field resistances are 0.25 W and 110 W respectively. When loaded (at motor shaft), the motor
draws 62 A from the mains. What would be its speed? Assume that the armature reaction demagnetizes the
field to the extent of 5%.
Also calculate the internal torque developed at no-load and on load. What is the motor shaft torque at
load (this torque drives the mechanical load).
SOLUTION
In a dc shunt motor
IL = Ia + If
Shunt field current,
If =
220
= 2 A (constant)
110
At no load
Ia0 = 5 – 2 = 3 A
Ea0 = 220 – 0.25 ¥ 3 = 219.25 V
n0 = 1200 rpm
(i)
Ea0 = K¢a F n0
or
219.25 = K ¢a F ¥ 1200
T0w0 = Ea0 Ia0
Ê 2p ¥ 1200 ˆ
˜¯ T0 = 219.25 ¥ 3
ÁË
60
or
T0 = 5.23 Nm
This torque is absorbed in iron loss, and windage and friction losses of the motor (the shaft being at no load).
On load
Ia1 = 62 – 2 = 60 A
Ea1 = 220 – 0.25 ¥ 60 = 205 V
(ii)
flux/pole F1 = 0.95 F
Ea1 = K¢a F1 n1
205 = K¢a ¥ 0.95 F n1
Dividing Eq. (ii) by Eq. (i)
205
0.95 n1
=
219.25
1200
(iii)
366
Electric Machines
n1 = 1181 rpm
or
Drop in motor speed is only 1.6% on being loaded. This because the reduction in flux/pole due to armature reaction
counters the drop in speed caused by armature resistance drop.
T1 w1 = Ea1 Ia1
Ê 2p ¥ 1181ˆ
˜¯ T1 = 205 ¥ 60
ÁË
60
T1 = 99.45 Nm
T1 (shaft) = 99.45 – 5.23
= 94.22 Nm
or
+
IL = Ia
Series Motor
The connection diagram of a series motor is drawn in
Fig. 7.71. From the armature circuit equation
Ea = Vt – Ia (Ra + Rse) = K¢a F n
Speed-current characteristic
From Eq. (7.78), we can
express motor speed as
Vt
+
Ea
Series field
winding
Ra
n
–
Ê Vt ˆ Ê Ra + Rse ˆ
n= Á
I
Ë K ¢ F ˜¯ ÁË K ¢ F ˜¯ a
a
Rse
(7.78)
a
–
(7.79)
Fig. 7.71 Series Motor
This is the exact speed-current equation.
In a series motor F – Ia is the magnetization characteristic, which as we know has an initial linear region
beyond which saturation sets in. In the linear region
F = Kf Ia; Kf is constant
(7.80)
Substituting in Eq. (7.79), the speed can be expressed as
1 È Vt
˘
( ) - ( Ra + Rse ) ˙
(7.81)
Í
K a¢ K f Î I a
˚
This is a shifted rectangular hyperbola sketched in Fig. 7.72(a) in solid line. As the armature current
increases rate of increase of F reduces. It means
F < Kf Ia
n=
As a result, the actual speed found from Eq. (7.79) is higher than based on linear assumption. The actual
speed-current characteristic lies above the characteristic based on linear magnetization; the characteristic is
sketched in dotted line in Fig. 7.72(a).
Some observations
1. As the armature current increases with load the speed comes down sharply
2. At no load
Ia Æ 0, F Æ 0, n
as shown from Eq. (7.79); the second term is very small because of (Ro + Rse)
This is a dangerous situation and the centrifugal forces will destroy the armature and may harm the
personnel. Hence, a series motor must never be allowed to run at no load even accidently.
DC Machines
n
367
T
Linear
magnetic
Actual
Actual
Linear magnetic
Ia » IL
O
Ia » IL
(a)
(b)
Fig. 7.72 Series Motor Characteristics
Torque-current characteristic
T = Ka FIa
(7.82)
T = KaKf I 2a : a parabola
(7.83)
For linear magnetization
The characteristic is sketched in Fig. 7.72(b) in solid line.
Because of saturation and demagnetization caused by armature reaction, the torque tends to level off in the
actual characteristic shown in dotted line in Fig. 7.72(b).
Speed-torque characteristic
1
Ia =
Ka
From Eq. (7.82)
n
ÊTˆ
ÁË F ˜¯
Substituting in Eq. (7.79) and organizing
Ê Vt ˆ Ê Ra + Rse ˆ
-Á
n= Á
˜T
Ë K a¢ F ˜¯ Ë K a¢ K aF 2 ¯
(7.84)
For linear magnetization
Eliminating Ia between Eqs (7.81) and (7.83) we get
n=
˘
1 È Vt K a K f
Í
- ( Ra + Rse ) ˙
K a¢ K f Í
˙
T
Î
˚
Actual
Linear magnetic
T
(7.85)
Fig. 7.73 Speed-Torque characteristics of series motor
The speed-torque characteristic as per Eq. (7.85) based on linear magnetization is sketched in Fig. 7.73
in solid line. Due to saturation and demagnetize the rate of increase of F reduces with increasing torque. At
large torque, F would be less than that based on linear assumption and so the speed would be higher. This is
because in Eq. (7.84) the contribution of the negative term is small as there is F2 in the its denominator and
so the first term predominates. The actual speed is almost constant at large torque.
Remark It is observed from the series motor speed-torque characteristic that as the load torque increases
the speed drops heavily relieving thereby the few load (n ¥ T) on the motor. This type of characteristic is
known as “series” characteristic and is ideally suited for traction, cranes etc—applications where in large
368
Electric Machines
accelerating is demanded by the load at starting, while when running a small torque is needed to maintain
steady speed. In the two applications cited the load is always there so there is no danger of under-loading or
no-loading.
EXAMPLE 7.35 The magnetization characteristic of a 4-pole de series motor may be taken as proportional
to current over a part of the working range; on this basis the flux per pole is 4.5 mWb/A. The load requires a
gross torque proportional to the square of the speed equal to 30 Nm at 1000 rev/min. The armature is wavewound and has 492 active conductors. Determine the speed at which the motor will run and the current it
will draw when connected to a 220 V supply, the total armature resistance of the motor being 2.0 W.
SOLUTION
Referring to Eq. (7.22)
Ea =
FnZ Ê P ˆ
(4.5 ¥ 10- 3 ¥ I a )n ¥ 492 Ê 4 ˆ
=
˜
Á
ÁË 2 ˜¯
60 Ë A ¯
60
= 0.0738 nIa
(i)
Recalling Eq. (7.26), the torque developed
T=
Further
Ê 4ˆ
1
1
Ê Pˆ
FIa Z Á ˜ =
(4.5 ¥ 10–3 Ia) Ia ¥ 492 Á ˜
Ë 2¯
Ë A¯
2p
2p
= 0.705 I 2a
Ea = Vt – Ia (Ra + Rse) = 220 – 2Ia
(ii)
(iii)
Substituting Eq. (i) in (iii),
0.0738 nIa = 220 – 2Ia
Ia =
or
220
2 + 0.0738n
(iv)
Substituting this expression for Ia in Eq. (ii),
Given:
220
È
˘
T = 0.705 Í
Î 2 + 0.0738n ˙˚
Load Torque TL = KLn 2
2
From the given data KL can be evaluated as
KL =
30
(1000) 2
= 3 ¥ 10–5 Nm /rpm
Under steady operation conditions, TL = T (developed)
or
Solving,
220
ˆ
Ê
3 ¥ 10–5 n2 = 0.705 Á
Ë 2 + 0.0738n ˜¯
2
n = 662.6 rpm
Substituting for n in Eq. (iv)
220
2 + 0.0738 ¥ 663.2
= 4.32 A
Ia =
EXAMPLE 7.36 A 250-V dc series motor with compensating winding has a total armature circuit
resistance of (Ra + Rse) = 0.08 W. It is run at 1200 rpm by means of a primemover with armature circuit
open and series field separately excited. This test yielded the following magnetisation data:
DC Machines
Ise(A)
VOC (V)
40
62
80
130
120
180
160
222
200
250
240
270
280
280
320
288
360
290
369
400
292
Sketch the speed-torque characteristic of the series motor connected to 250 V main by calculating the
speed and torque values at armature currents of 75, 100, 200, 300, 400 A.
SOLUTION Rather than drawing the magnetisation curve, we shall usee the above data by linear interpolation between
the data point given.
Sample Calculation
Ia = Ise = 75 A
Ea = 250 – 0.08 ¥ 75 = 244 V
Using linear interpolation, at Ise = 75 A
Ea (1200 rpm) = 130 –
130 - 62
¥ 5 = 121.5 V
40
1200
¥ 244 = 2410 rpm
121.5
Tw = Ea Ia
n=
Ê
ˆ
60
T= Á
˜¯ ¥ 244 ¥ 75 = 72.5 Nm
2
p
¥
2410
Ë
or
Computations can be carried in tabular from below:
Ia = Ise (A)
Ea (V)
Ea (V) at 1200 rpm
n (rpm)
T (Nm)
100
242
155
1874
123
75
244
121.5
2410
72.5
300
226
283
958
676
200
234
250
1123
398
The speed-torque (n – T ) characteristic is drawn in Fig. 7.74.
2500
2000
1500
n(rpm)
1000
500
0
200
400
Fig. 7.74
600
T(Nm)
800
1000
400
218
292
902
923
370 Electric Machines
EXAMPLE 7.37 A 220 V, 7.5 kW series motor is mechanically coupled to a fan. When running at
400 rpm the motor draws 30 A from the mains (220 V). The torque required by the fan is proportional to the
square of speed. Ra = 0.6 W, Rse. = 0.4 W. Neglect armature reaction and rotational loss. Also assume the
magnetisation characteristic of the motor to be linear.
(a) Determine the power delivered to the fan and torque developed by the motor.
(b) Calculate the external resistance to be added in series to the armature circuit to reduce the fan speed
to 200 rpm. Calculate also the power delivered to the fan at this speed.
SOLUTION
Pfan
P
Tdev = Ka¢Kf IseIa
Tfan
Tdev
Ia
But
\
= Pdev = P; no rotational loss
= Ea Ia
= Ka¢Kf I a2; Ise = Ia; linear magnetization
= KF n 2
= Tfan = T
μn
(i)
(ii)
(a) Operation at 400 rpm (Rext = 0)
Ea
Ia
P
Tw
= 220 – (0.6 + 0.4) ¥ 30 = 190 V
= 30 A
= 190 ¥ 30 = 5.7 kW
= Ea Ia
T=
or
5700
= 136 Nm
2p ¥ 400
60
(b) Operation at 200 rpm (Rext = ?)
Ê 200 ˆ
T = 136 ¥ Á
Ë 400 ˜¯
2
= 34 Nm
Ê 200 ˆ
Ia = 30 ¥ Á
= 15 A
Ë 400 ˜¯
Tw = EaIa
Ê 2p ¥ 200 ˆ
= [220 – (0.6 + 0.4 + Rext) ¥ 15] ¥15
34 ¥ Á
Ë
60 ˜¯
Solving we get
Rext = 10.5 W
Ê 2p ¥ 200 ˆ
P = Tw = 34 ¥ Á
Ë
60 ˜¯
= 0.721 kW
EXAMPLE 7.38 A 180 kW, 600 V dc series motor runs at 600 rpm at full load current of 300 A. The total
resistance of its armature circuit 0.105 W. The magnetization curve data at 500 rpm is as below. The series
field excitation is provided separately.
VOC = Ea (V)
Series field
current (A)
440
250
470
277
500
312
530
356
560
406
590
466
DC Machines
371
Determine the starting torque developed when the starting current is limited to 500 A. Assume that the
armature reaction mmf is proportional to the square of armature current.
SOLUTION
The magnetization curve at 500 rpm is drawn in Fig. 7.75.
Full load current, Ia =
180 ¥ 103
= 300 A, Speed, n = 600 rpm
600
Back emf
Ea = 600 – 300 ¥ 0.105 = 568.5 V
To look up magnetization curve, we should find the emf at 500 rpm
Ea (500 rpm) = 568.5 ¥
500
= 473 V
600
From the magnetization curve to induce 473 V, the series field current needed is
Ise = Ia = 282 A; we will write Ia
Therefore the series field equivalent demagnetizing current is
Iad = 300 – 282 = 18 A
As Iad is proportional to the square of armature current,
Iad /ampere of armature current =
18
(300) 2
= 0.2 ¥ 10–3A/A
Ia = 500 A
Effective Ia = 500 – Iad = 500 – (500)2 ¥ 0.2 ¥ 10–3
At start
= 450 A
From the magnetization curve, the corresponding induced emf is
Ea(V)
600
500
400
200
220
300
400
Fig. 7.75
500
If (A)
372
Electric Machines
Ea = 590 V at n = 500 rpm
Power balance equation
Tw = EaIa
Therefore
Tstart =
Ea I a
w
Tstart =
590 ¥ 500
= 5634 Nm
2p
500 ¥
60
(i)
(ii)
Substituting values
Compound Motor
It has been said earlier that there is no significant difference in the performance of long and short shunt
connection. We shall proceed on the basis of long shunt whose circuit diagram is given in Fig. 7.30(a).
Further as differential compound motor is not used in practice, for which we will advance the reason later, we
will consider only cumulative compound motor.
Speed equation
n=
Vt - I a ( Ra + Rse )
K a¢ F
(7.84)
Torque equation
T = KaF Ia
(7.85)
In a compound motor F is determined from the magnetization characteristic for the combined (additive)
net excitation
Ê N se ˆ
If (net) = If + Á
˜ Ia, If = constant
Ë Nf ¯
It will help matters if we could separate out the flux created by shunt and series fields but the superposition
cannot be applied. To overcome this problem we consider the magnetization characteristic (F vs If) sketched
in Fig. 7.76 it is found from this figure that F = Fsh + Fse
where
Fsh = flux of shunt excitation
Fse = flux of series excitation
Further Fse can be taken to be proportional to Ia corresponding to slope of the magnetization characteristic
in the region beyond Fsh.
It may be noted that the above method accounts for saturation but not demagnetizing effect of armature
reaction.
Speed-current characteristic
n=
1 È Vt - I a ( Ra + Rse ) ˘
Í
˙
K a¢ Î
Fsh + Fse
˚
(7.86)
DC Machines
373
F
Fse
Fsh
Nse
I
Nf a
If
It (net)
If
Fig. 7.76
As Ia increases, numerator decreases and Fse in the denominator increases and as a result the motor speed
falls much more sharply than in shunt motor. At large values of current the characteristic is similar to that of a
series motor. The characteristic lies in between those of shunt and series motors, closer to one whose field is
stronger near full load. The compound motor has a great advantage over series motor because it has a definite
no load speed given by
n0 =
1 Ê Vt ˆ
; shunt motor no load speed
K a¢ ÁË Fsh ˜¯
(7.87)
Yet its characteristic can made closer to the relieving characteristic of series motor.
A differentially compound motor has flux/pole F = (Fsh – Fse). It is seen from Eq. (7.84) that on over-load
F reduces sharply and so the motor acts like a series motor on no load. This is why the differential compound
motor is not used in practice.
Comparison of compound motor speed-current characteristic with shunt at the same full load speed and
also the series motor is depicted in Figs. 7.77(a) and (b) in which the conclusions made above are borne out.
Certain observations are made below:
1. Compound and shunt motors (Fig. 7.77(a)) Because of Fse increasing with Ia, the speed of the
compound motor falls much more sharply than the shunt motor. Therefore, the n – Ia characteristic of
the compound motor lies above that of the shunt motor for Ia < Ia ( f l) and lies below for Ia >Ia ( f l)
2. Compound and series motors (Fig. 7.77(b)) Because the constant shunt flux Fsh, n – Ia characteristic
of compound motor starts at definite speed n0 and drop gradually, while that series motor drop steeply.
Therefore, the compound motor characteristic lies below that of the series motor for Ia < Ia ( f l) and
lies above that of the series motor for Ia >Ia ( f l)
Torque –current characteristic
From Eq. (7.85)
T = Ka (Fsh + Fse) Ia
(7.88)
374 Electric Machines
n
n0
Commutative
compound
Differential
compound
n(fl)
n
Commutative
compound
Series
n0
Differential
compound
n(fl)
Shunt
n0
Ia(fl)
(a) Compound and shunt motor
Ia(fl)
(b) Compound and series motor
Ia
Ia
a characteristics
Fig. 7.77
= KaFsh Ia + KaFse Ia
Fse ª Kse Ia
Thus
T = KaFsh Ia + KaKse I 2a
(7.89)
shunt type + series type
Due to saturation and demagnetization both torque components level off. The torque-current characteristic
of the compound motor as sum of shunt and series components is sketched in Fig. 7.78. This approach
of dividing of F into Fsh and Fse is heuristic and so reveals the nature of characteristic; can not to used
quantitatively. Quantitative approach (graphical) will be illustrated in examples that follow.
T
Commutative
compound
T(fl)
Series
Shunt
Ia(fl)
Ia
Fig. 7.78 Torque-current characteristic of compound motor.
Incidentally the figure provides the comparison of the nature of the (T – Ia) characteristics of the three
kinds of dc motors.
DC Machines
Speed-torque characteristic
375
Eliminating Ia between Eqs 7.86 and (7.88) we obtain
n=
1
1
Vt
Ra + Rse
T
K a¢ Fsh + Fse K a¢ K a (Fsh + Fse ) 2
(7.90)
1 Ê Vt ˆ
K a¢ ÁË Fsh ˜¯
(7.91)
where as before no load speed is
n0 =
It may be noted that the first term in Eq. (7.90) is not a constant quantity.
As torque increases, Ia and so Fse increases, the first term reduces (sharply if the series field is stronger
than shunt field); the second negative term reduces but it does not have significant effect on speed as (Ra +
Rse) is very small. The speed reduces with torque and at large torque becomes almost constant because of
saluration and demagnetigation. Comparison of speed-torque characteristic of compound motor with shunt
and series motor at same full load speed is shown in Fig. 7.79.
Speed
Series motor
Compound motor
(strong series field)
Compound motor
(medium series field)
n(fl)
Shunt motor
T(fl)
Torque
Fig. 7.79 Comparison of speed-torque characteristic of cumulative compound motor with shunt and series motor
EXAMPLE 7.39 For the series motor of Example 7.38 determine the speed and mechanical power and
torque developed, when the motor draws 250 A from the mains.
SOLUTION It has been shown in Example 7.38 that the demagnetizing effect of armature reaction in terms of series
field current is
At
Therefore
lad
la
Iad
Ia (net)
= 0.2 ¥ 10–3 I2a
= 250 A
= 0.2 ¥ 10–3 ¥ (254)2 = 12.5 V
= 250 – 12.5 = 237.5 A
376 Electric Machines
From the magnetization curve
Ea (at 500 rpm) = 428 V
Ea (actual) = 600 – 250 ¥ 0.105 = 573.75 V
Therefore, motor speed
n = 500 ¥
573.75
= 670 rpm
428
Mechanical power developed
Pmech = EaIa = 573.75 ¥ 250
= 143.44 kW
Torque developed =
143.44 ¥ 103
= 2044 Nm
2p
670 ¥
60
EXAMPLE 7.40 A 240 V compound (cumulative) dc motor has the following open circuit magnetization
characteristic at normal full load speed of 850 rev/min:
Excitation, AT/pole
Generated emf, V
1200
76
2400
135
3600
180
4800
215
6000
240
The resistance voltage drop in the armature circuit at full load is 25 V. At full load the shunt and the
series windings provide equal ampere-turn excitation.
Calculate the mmf per pole on no load. Estimate the value to which the speed will rise when full load is
removed, the resistance voltage drop in the armature circuit under that condition being 3 V. Ignore armaturereaction and brush-contact effects. Assume long-shunt cumulative compounding.
SOLUTION
At full load, from Fig. 7.81,
Ea (full load) = V – Ia (Ra + Rse) = 240 – 25 = 215 V
+
IL
If
Ia
Shunt
field
Rse
Vt
Series field
Ea
Ra
–
Fig. 7.80
Long-shunt compound dc motor
Corresponding ATnet from magnetizing curve of Fig. 7.81.
Then
Now
ATnet (full load) = 4800
ATsh = ATse (full load) = 2400
3
Ia (no load) =
I (full load); Ia (no load) = If (shunt field)
25 a
DC Machines
377
240
At 850 rpm
215
200
160
Ea(V)
148
120
80
40
0
1000
2000
3000
4000
5000
6000
Excitation AT/Pole
Fig. 7.81
3
¥ 2400 = 288
25
= 2400 (no change)
= 2400 + 288 = 2688
= 148 V at 850 rpm, at ATnet (no load)
= 240 – 3 = 237 V
μ n at given ATnet
ATse (no load) =
Now
ATsh
ATnet (no load)
Ea(from the magnetizing curve)
Ea (no load)
Ea
237
= 1361 rpm
148
It may be seen that in a cumulatively compound dc motor, full load speed is much less than no load speed (see
Fig. 7.71).
n(no load) = 850 ¥
Hence,
EXAMPLE 7.41
A 240 V, 10 kW dc shunt motor has
Ra = 0.18 W,
R(inter poles) = 0.025 W and
Shunt field turns = 2000
Rf = 320 W (field only),
No load test conducted at 240 V by varying the field current yielded the following data:
Speed, n
Field current, If (A)
1160
0.664
1180
0.632
1220
0.584
1250
0.558
1300
0.522
At no load armature voltage drop is negligible.
(a) At full load the motor field current is adjusted to 0.6 A for a speed of 1218 rpm. Calculate the
demagnetizing ampere-turns.
378
Electric Machines
(b) In part (a) calculate the electromagnetic torque.
(c) The field current is adjusted to the maximum value. The starting armature current is limited to 75 A.
Calculate the starting torque. Assume demagnetizing ampere-turns to be 165.
(d) The shunt field current is adjusted to give a no load speed of 1250 rpm. A series field is provided to
give a speed of 1200 rpm at full load current. Calculate the number of turns of the series field. Assume
the series field resistance to be 0.04 W.
SOLUTION
Full load current,
At no load
or
10 ¥ 103
= 41.7 A
240
Ea ª Vt = 240 V
Ea = K¢a Fn
1 Ea
, F = F (If )
n=
K a¢ F
Ia =
n=
1
Ea
; F = F (If )
◊
K a¢ F( I f )
Ê Ea ˆ
, pulled in Fig. 7.82
Thus n – If characteristic is the inverse of F – If characteristic scaled by Á
Ë K a¢ ˜¯
(a) At full load current,
Ia
Ea
Ea
n
= 41.7 A, If = 0.6 A
= Vt – Ia (Ra + Ri) = 240 – (0.18 + 0.025) ¥ 41.7
= 231.45 V
= 1218 rpm
240
n (at Ea = 240 V) = 1218 ¥
231.45
= 1262 rpm
The corresponding field current required is found from the n – If characteristic at
Actual field current required is,
If = 0.548 A
If = 0.6 A
Therefore, equivalent demagnetizing field current is
Demagnetizing ampere-turns,
Ifd = 0.6 – 0.548 = 0.052 A
ATd = 0.052 A ¥ 2000 = 104
(b) Torque developed
Tw = Ea Ia
or
(c) Field current (max),
231.45 ¥ 41.7
= 75.7 Nm
2p
1218 ¥
60
240
=
= 0.75 A
320
= 165 (given)
165
=
= 0.0825 A
2000
= 0.75 – 0.0825 = 0.6675 A
T=
If
ATd
Ifd
If (net)
DC Machines
379
n
(rpm)
1300
1200
1100
1000
0.4
0.5
0.6
0.7
If (A)
Fig. 7.82
At this field current at Ea = 240 V, the motor speed is from the n – If plot as
We know that
and
n = 1150 rpm or w = 121.27 rad/s
Ea = Ka Fw
T = KaFIa
(i)
(ii)
380
Electric Machines
At If = 0.75, ATd = 165, F has the same value in Eqs (i) and (ii).
Ea
240
=
From Eq. (i)
Ka F =
= 1.98
w
121.27
Using this value in Eq. (ii) with Ia (start) = 75 A, we get
T (start) = 1.98 ¥ 75 = 148.5 Nm
n0 = 1250 rpm
Ea = 240 V
(d) No load speed,
Corresponding field current, If = 0.56 A ; from n – If characteristic of Fig. 7.82.
n
Rse
Total resistance in armature circuit
Ea
At full load speed specified,
= 1200 rpm
= 0.04 W
= 0.18 + 0.25 + 0.04 = 0.245 W
= 240 – 0.245 ¥ 41.7 = 229.8 V
240
At,
Ea = 240 corresponding speed = 1200 ¥
= 1150 rpm
229.8
From n – If characteristic
If (net) = 0.684 A
Equivalent If to be produced by the series field = 0.684 – 0.56 = 0.124 A
Nse =
We then field
0.124 ¥ 2000
= 5.95 or 6 turns
41.7
EXAMPLE 7.42 A shunt motor draws a full load armature current of 56.5 A from 230 V supply and runs
at 1350 rpm. It has an armature-circuit resistance of 0.15 W and shunt field turns of 1250. Its OCC data at
1350 rpm is given below:
VOC (V)
If (A)
180
1.18
200
1.4
220
1.8
240
2.4
250
2.84
(a) Determine the shunt field current of the motor at no load speed of 1350 rpm.
(b) Determine the demagnetizing effect of armature reaction in ampere-turns/pole at full load current.
(c) By adding series-field turns and connecting the motor in long shunt compound, it is required to have a
speed of 1230 rpm when drawing an armature current of 56.5 A from 230 V supply. Assume the series
field resistance to be 0.033 W. Determine the series field turns/pole.
(d) In part (c) the series field turns provided are 25/pole with series field resistance of 0.025 W. Determine
the speed at which the motor would run when connected to 230 V Supply drawing armature current
of 56.5 A.
SOLUTION The shunt motor connections are drawn in Fig. 7.84 No regulating resistance in the shunt, so the shunt
field current remains constant. The magnetization characteristic from the OCC data is plotted in Fig. 7.85
(a) At no load
n0 = 1350 rpm
The no load armature voltage drop Iao Ra can be neglected. So
We find from the OCC at
Ea = Vt = 230 V
Ea = 230 V
If = 1.08 A
The shunt field current remains constant throughout as the field has fixed resistance.
DC Machines
381
VOC(V)
300
If
IL
Ia
200
+
100
Ea
230 V
1
–
Fig. 7.84
Speed n
Ia
Ea
Ea
If (net)
If
Actual
3
If (A)
Fig. 7.83
(b) Full load
Armature current,
Therefore
At
2
= 1350 rpm
= 56.5 A
= 230 – 56.5 ¥ 0.15 = 221.5 V
= 221.5 V (speed 1350) it is found from the OCC that
= 1.8 A
= 2.08 A
The difference is the demagnetizing effect of armature reaction in terms of field current. So
Demagnetizing ampere-turns,
Ifd = 2.08 – 1.8 = 0.28 A
ATd = 0.28 ¥ 1200 = 336
(c) Long shunt compound connections are drawn in Fig. 7.85. Shunt field current is same as calculated in part (a)
At full load
Speed required,
If = 2.08 A
Ea = 230 – 56.5 (0.15 + 0.033)
= 219.7 V
n = 1230 rpm
IL
230 V
To look up OCC, we find Ea at 1350 rpm
1350
Ea(1350 rpm) = 219.7 ¥
= 241.1 V
1230
To induce this emf
Ea
If (net) = 2.41 A or 2.41 ¥ 1200 = 2982 AT
Fig. 7.85
or
1296 AT
AT balance equation
or
Series field turns,
(d)
n
–
Actual shunt field current
If = 1.08 A
+
Ia
If
ATnet = ATsh – ATd + ATse
2892 = 1296 – 336 + ATse
ATse = 1932
1932
Nse =
= 34/pole
65.5
ATse = 25 ¥ 56.5 = 1412.5
ATnet = 1296 – 334 + 1412.5 = 2374.5
382
Electric Machines
2374.5
= 1.98 A
1200
Ea (at 1350 rpm) = 226 V
If (net) =
From the OCC
Actual induced emf
Ea (actual) = 230 – 56.5 (0.15 + 0.025) = 221.1 V
We then find
Speed, n = 1350 ¥
221.1
= 1320 rpm
226
7.16 STARTING OF DC MOTORS
At the time of starting (n = 0), the induced emf of a motor is zero such that the current drawn from rated
voltage supply would be
I as =
V
Ra
(7.92)
for a shunt motor. The series field resistance would be included in the denominator for a series motor. For
large motors the armature resistance may be 0.01 pu or less; even for lower kW motors it ranges from
0.0625 to 0.125. Thus in full-voltage starting of a dc motor, the armature current can be several times (about
100 times for large motors) the rated value. For several reasons mentioned below such a large current cannot
be allowed to flow in a motor even for the short starting period.
(i) It would cause intolerably heavy sparking at the brushes which may destroy the commutator and
brush-gear.
(ii) Sudden development of large torque causes mechanical shock to the shaft, reducing its life.
(iii) Such heavy current cannot be generally permitted to be drawn from the source of supply.
For the above-mentioned reasons all motors, except for small and fractional-kW motors, must be started
with external resistance included in armature circuit to limit the starting current safe values. Where variablevoltage dc supply is available, e.g. in the Ward-Leanard control scheme, this can be used for motor starting
and no starting resistance would be needed.
One point in favour of direct starting must be mentioned here. Since the motor torque with direct start is
much higher, the motor starts much more quickly. As a consequence the Joule input per start is much less
than that with resistance start. This would be helpful in repeated starting–saving energy and causing less
temperature rise.
Maximum allowable starting current is not more than 1.5 to 2.0 times the rated value. These values are
safe and yet at the same time permit a high starting torque for quick acceleration of the motor. Obviously the
dc motor can start on load. Where a variable-voltage supply is available for speed control, starting poses no
problem at all.
Shunt and Compound Motor Starters
In shunt and compound motors starting the shunt field should be switched on with full starting resistance in
armature circuit. A short time delay in this position allows the field current to build up to the steady value of
the inductive field transients. Also all the regulating resistance in the field circuit must also be cut out before
starting for the field current to be maximum as Tstart μ If .
DC Machines
383
We shall illustrate shunt motor starters, but these are applicable for compound motors as well. There are
two types of shunt motor starters:
fi Three-point starter – employed where motor field current can be varied in a narrow range and so does
the motor speed
fi Four-point starter – motor field current can vary over a wide range and so does the motor speed
Three-point starter The connection diagram of a three-point shunt motor starter is shown in Fig. 7.86. The
starter terminals to be connected to the motor are A (armature); F (field) and L (line). The starting resistance
is arranged in steps between conducting raised studs. As the starting handle is rotated about its fulcrum, it
moves from one stud to the next, one resistance step is cut out, and it gets added to the field circuit. There
is a short time wait at each stud for the motor to build up speed. This arrangement ensures a high average
starting torque.
Electric machines
3-point starter
A
ON
A1
M
Handle
A2
F1
F
Start
NVC
F2
OL
release
L
Spring
OFF
+
Fig. 7.86
–
DC mains
DC shunt motor there point starter (manual)
At start the handle is brought to stud one. The line voltage gets applied to the armature with full starting
resistance in series with armature and to the field with NVC in series. Thus the motor starts with maximum
torque. As it pick up speed the handle is moved from stud to stud to the ‘ON’ position shown in Fig. 7.86.
The starting resistance has been fully cut out and is now included in the field circuit; being small it makes
little difference in the field current. The resistance of NVC is small and forms part of the field resistance. The
voltage across the armature is the line voltage. The handle is held in this position by the electromagnet excited
by the field current flowing through NVC.
Two protections are incorporated in the starter.
1. NVC (no volt coil): In case of failure of field current (due to accidental or otherwise open circuiting),
384 Electric Machines
this coil releases the handle (held electromagnetically), which goes back to the OFF position under the
spring action.
2. OL (over-load) release: The contact of this relay at armature current above a certain value (over load/
short circuit) closes the NVC ends, again bringing the handle to OFF position.
In the three-point starter if the field regulator is used to reduce the field current to low values for high
motor speed NVC may release the handle causing the motor to shut down where such variation of field
current is desired a four-point starter is used.
Four-point starter To overcome the problem caused when the field current is low, NVC is connected across
the two lines, one line connected to F terminal through the starter and other directly to the second line from
another L terminal of the starter. To limit the NVC current a protective resistance R is connected in series with
it. The starter diagram is drawn in Fig. 7.87. It now has four terminals = A F LL. The rest operation remains
the same.
3-point starter
A
ON
A1
A2
M
Handle
F1
F
Start
NVC
R L
OL
release
OFF
F2
L
Spring
+
Fig. 7.87
–
DC mains
DC shunt motor four point starter (manual)
The modern practice is to use a push-button type automatic starter in industries. Automatic starters carry
out essentially the same functions as the manual ones with electromagnetic relays that short out sections of
the robust metallic starting resistors either in a predetermined time sequence or when the armature current
has dropped to a preset value. Such an arrangement is shown in Fig. 7.88. Most automatic starters embody
extra control and safety features.
Starter Step Calculation for dc Shunt Motor
From Fig. 7.88 it is seen that the instant the starter is moved to stud 1 or conductor* CM is closed, the current
in the circuit reaches a value I1, designated as the upper current limit, given by
* Resistance arranged in a circular arc with studs on which a conductor arm moves are employed in a manual starter.
Contactors as shown in Fig. 7.88 are used in automatic starters.
DC Machines
V
(7.93)
R1
Thereafter the current value decreases as the
motor speeds up and its back emf rises as shown in
Fig. 7.89. The current is allowed to reduce to 12, the
lower current limit, given by.
I1 =
C1
1
2
r1
C3
3
r2
CM
Ea
4
Ra
r2
R4 = R a
V - Ea (n1 )
I2 =
(7.94)
R1
where Ea(n1) is the back emf at speed n1 reached by
the motor. At this instant the starter is moved to stud
2 or contactor C1 is closed. The current increases
instantaneously to I1 as shown in Fig. 7.89 and
satisfies the relationship
I1 =
C2
385
R3
R2
R1
Vt
Fig. 7.88 Automatic Shunt motor starting
V - Ea (n1 )
R2
(7.95)
From Eqs (7.96) and (7.97)
I1
R1
=
I2
R2
(7.96a)
Ia
I1
I2
0
ILoad
0
t1
t2
t3
t4
t5
t
0
t1
t2
t3
t4
t5
t
Speed
n
n2
n1
Fig. 7.89 Variation of armature current and speed versus time in shunt motor starting
386
Electric Machines
By induction, for a k-stud ((k – 1) sections of external resistance) starter,
Rk -1
I1
R1
R2
=
=
=º=
I2
R2
R3
Rk
(7.96b)
It immediately follows from Eq. (7.98b) that
ÊI ˆ
R1 R2 Rk -1
◊ º
= Á 1˜
R2 R3
Rk
Ë I2 ¯
g=
where
k -1
= g k–1
I1
upper current limit
=
I2
lower current limit
R1
= g k–1
Rk
Hence
(7.96c)
(7.96d)
From Fig. 7.89 it is obvious that
Rk = rk = Ra (armature resistance)
R1
= g k–1
Ra
\
(7.96e)
Figure 7.89 shows the plot of the armature current and speed versus time as the starter resistance is cut out
in steps. During waiting time at each step, the current falls and the speed rises exponentially according to a
single dominant time-constant (Sec. 7.21). This is the reason why the waiting time at each step progressively
reduces as is easily evident from Fig. 7.89.
Once the designer has selected the upper and lower limits of armature currents during starting, starter step
calculations can proceed on the following lines:
(i) From Eq. (7.95) calculate R1.
(ii) From Eq. (7.96e) calculate the number of steps k choosing the nearest integral value.
(iii) Calculate resistances R1, R2 … from Eq. (7.98(b)). From these the resistance values of various sections
of the starter can be found out.
OR
It is convenient to calculate the step resistance from the recursive relationship derived below
From Eqs (7.96(b)) and (7.96(d))
R2 = R1/g
Then
Ê 1ˆ
r1 = R1 – R2 = Á1 - ˜ R1
Ë g¯
Next
Ê 1ˆ
r2 = R2 – R3 = Á1 - ˜ R2
Ë g¯
1 Ê 1ˆ
1R1 = r1/g
g ÁË g ˜¯
rn = r n–1/g
(7.97)
=
By induction
(7.98)
Sometimes the number of sections are specified in addition to the upper current limit. This problem can be
handled by a slightly different manipulation of Eqs (7.95) to (7.96(e)).
DC Machines
387
In the series motor the flux/pole changes with the armature current making the starter step resistance
calculation somewhat more involved. These are not dealt with in this book.
EXAMPLE 7.43 A starter is required for a 220-V shunt motor. The maximum allowable current is 55 A
and the minimum current is about 35 A. Find the number of sections of starter resistance required and the
resistance of each section. The armature resistance of the motor is 0.4 W.
SOLUTION
I1 = 55 A;
I2 = 35 A
I1
g=
= 1.57
I2
From Eq. (7.95)
R1 =
V1
200
=
4W
I1
55
From Eq. (7.98(e))
R1
4
=
=10
Ra
0.4
n = 6.1
g n–1 =
or
For an integral choice of n = 6,
g = 1.585,
1/g = 0.631
Using Eq. (7.98(b)) the values of the resistances are obtained as
R1 = 4 W
r1 = (1 – 0.631) ¥ 4 = 1.476 W
r2 = 1.476 ¥ 0.631 = 0.931 W
Proceeding on the above lines
r3 = 0.587 W,
r4 = 0.370 W,
r5 = 0.235 W
EXAMPLE 7.44 A 25 kW, 230 V has an armature resistance of 0.12 W and a field resistance of 120 W. Its
speed at full load is 2000 rpm. The field current may be neglected.
(a) For a four step (resistance steps) calculate the resistances of the steps if the allowable maximum
armature current is 1.5 times the full load current. What is the lower current limit?
(b) Calculate the motor speed at each stud when the current reaches the lower limit and starting handle
is to be moved to the next stud
25 ¥ 103
= 108.7 A
230
(a) Four resistance steps require five studs (k = 5) upper current limit,
SOLUTION
Ia ( fl) =
I1 = 108.7 ¥ 1.5 = 163 A
230
230
=
R1 =
= 1.41 W
I1
163
Ra = 0.12 W (given)
R1
= g k–1
Ra
388 Electric Machines
1/ 4
1.41
Ê 1.41 ˆ
= g 4 or g = Á
Ë 0.12 ¯˜
0.12
g = 1.851
I1
= g = 1.851, 1/g = 0.54
I2
1.63
= 88.1 A
or
I2 (lower limit) =
1.85
Resistance steps
r1 = (1 – 0.54) ¥ 1.41 = 0.6486 W = 0.649 W
r2 = 0.6486 ¥ 0.54 = 0.3502 W = 0.350 W
r3 = 0.1891 W = 0.189 W
r4 = 0.1021 W = 0.102 W
(b) At full load
Ia( fl) = 103.7 A
Ea = 230 – 103.7 ¥ 0.12 = 217.6 V
Speed n = 2000 rpm
Ea = K¢a Fn = K ¢¢a n ; filed current is constant, armature reaction effect ignored.
or
217.6 = K ¢¢a ¥ 2000 or K ¢¢a = 0.1088
At stud 1 when armature current reduces to I2 = 88.1 A
Ea (1) = Vt – I2 ¥ R1 = 230 – 88.1 ¥ 1.41 = 105.8 V
\
n1 =
105.8
= 971 rpm
0.1088
At stud 2
Ea (2) = 230 – 88.1 ¥ (1.41 – 0.649) = 230 – 88.1 ¥ 0.761
= 162.96 V
n2 =
162.96
= 1498 rpm
0.1088
At stud 3
Ea(3) = 230 – 88.1 ¥ (0.761 – 0.35) = 230 – 88.1 ¥ 0.411
= 193.8 V
193.8
n3 =
= 1778 rpm
0.1088
At stud 4
Ea (4) = 230 – 88.1 ¥ (0.411 – 0.189) = 230 – 88.1 ¥ 0.222
= 210.4 V
210.4
n4 =
= 1934 rpm
0.1088
At stud 5 (ON)
The armature current instantaneously rises to I1 = 163 A. The speed increases from 1934 rpm to 2000 rpm and
armature current falls to 108.7 A to match the full load on the motor of the load is any different the armature
current and speed adjust accordingly.
Note: To determine the times at which speeds n1, º, n4 are reaches needs electromagnetic study; Section 7.21.
DC Machines
The Example 7.42 is also solved, by MATLAB.
clc
clear
Pop=25*l000;
Vt=230;
Ra=0.12;
rf=120;
k=5;
I1=Iamax;
R1=Vt/ I1;
r=(R1/Ra)^(l/(k–l));
I2 = I1/r ;
for i=l:(k–l)
if i==l
rest(i)=(l–(l/r) )*Rl
else
rest(i)=rest(i–l)*(l /r)
end
end
Answer:
rest =
0.6488
rest =
0.6488
rest =
0.6488
rest =
0.6488
0.3504
0.3504
0.1892
0.3504
0.1892
0.1022
MATLAB code for Example 7.44 is given below:
clc
clear
Pop=25*l000;
Vt=230;
Ra=0.12;
rf=120;
%
%
%
%
Power Output
Terminal Voltage
Armature resistance
Field Resistance
Iamax=1.5*Iaf1;
% Max armature current
k=5;
% No of studs
I1=Iamax;
R1=Vt/I1;
r=(R1/Ra)^(l/(k–1));
I2=I1/r;
389
390
Electric Machines
for i=1 : (k–1)
if i==1
rest (i)=(1–(l/r))*R1; % rest=resistance of the studs
else
rest (i)=rest (i–l)*(l /r);
end
end
% part (b)
Iaf1 = 103.7;
Ea=Vt–Iaf1*Ra; % back EMF
Ka=Ea/Nf1;
for i=1: (k–1)
if i==1
Eastud (1)=Vt–I2*R1;
Rnew=R1
else
Eastud (i)=Vt– I2*(Rnew-rest(i–1));
Rnew=Rnew–rest(i–1)
end
end
N=Eastud/Ka;
display (Renw)
Answer:
Rnew=1.4107
Rnew=0.7618
Rnew=0.4114
Rnew=0.2222
Rnew=0.2222
7.17 SPEED CONTROL OF DC MOTORS
The dc motors are in general much more adaptable speed drives than ac motors which are associated with
a constant-speed rotating field. Indeed one of the primary reasons for the strong competitive position of dc
motors in modem industrial drives is the wide range of speeds afforded.
From Eq. (7.69)
1 Ea
1 Ê Vt - I a Ra ˆ
=
˜¯
K a¢ F
F
K a¢ ÁË
Since the armature drop is small, it can be neglected.
n=
\
nª
1 Vt
K a¢ F
(7.99)*
(7.100)
This equation gives us two methods of effecting speed changes, i.e. the variation of field excitation, If and
that of terminal voltage, Vt. The first method causes a change in the flux per pole, F and is known as the field
control and the second method is known as the armature control.
* In case a series field is also provided, the armature drop would be Ia(Ra + Rse).
DC Machines
391
Base Speed It is the speed at which the motor runs at rated terminal voltage and rated field current. It is
indeed the name plate speed of the motor.
Speed Regulation
no - n f l
¥ 100
nfl
no = no load speed
nf l = full load speed i.e. rated speed.
% speed regulation =
Field Control
For fixed terminal voltage, from Eq. (7.102)
n2
F1
=
n1
F2
(7.101)
I f1
n2
F1
=
=
If2
n1
F2
(7.102)
which for linear magnetization implies
Certain limitations of the field control method are:
1. Speeds lower than the rated speed cannot be obtained because the field cannot be made any stronger;
it can only be weakened.
2. Since the speed is inversely proportional to the flux/pole while the torque is directly proportional to it
for a given armature current, it can cope with constant kW drives only where the load torque falls with
speed.
3. For motors requiring a wide range of speed control, the field ampere-turns are much smaller than the
armature ampere-turns at high speeds causing extreme distortion of the flux density in the air-gap. This
leads to unstable operating conditions or poor commutation. Compensating winding can be used to
increase the speed range which can be 2 to 1 for large motors, 4 to 1 for medium sized ones and 8 to 1
for small motors. Even then the field control is restricted to small motors.
4. This control method is not suited to applications needing speed reversal; since the only way to reverse
speed is to disconnect the motor from the source and reverse the field/armature polarity. The field
circuit being highly inductive, it is normally the armature which is reversed.
Shunt Motor
Figure 7.90(a) illustrates the field control for shunt motors; the control being achieved by means of a rheostat
(regulator) in the field circuit. Reproducing Eq. (7.77) here for convenience:
Ê
ˆ
Vt
Ra
-Á
T
(7.103)
2
K a¢F Ë K a¢ K aF ˜¯
The speed-torque characteristic which has a small linear drop due to the second term (Ra effect) and
translates upwards as the field is weakened due to the armature reaction is shown in Fig. 7.90(b). The
demagnetizing effect of the armature reaction causes the characteristics to somewhat bend upwards with
increasing torque (increasing load current). The working range of the speed-torque characteristic reduces
with increasing speed in order for the armature current not to exceed the full-load value with a weakening
field.
n=
392
Electric Machines
n3
If
IL
+
Limit imposed by
full-load current
n2
If 3
Ia
Rf
If 2
n1
Ea
Vt
AR effect
considered
If 3 < If 2 < If 1
If1
Rreg
–
0
T
(b)
(a) Connection diagram
Fig. 7.90
Field control for shunt motor
EXAMPLE 7.45 A 400-V dc shunt motor takes a current of 5.6 A on no-load and 68.3 A on full-load.
Armature reaction weakens the field by 3%. Calculate the ratio of full-load speed to no-load speed. Given
Ra = 0.18 W, brush voltage drop = 2 V, Rf = 200 W.
SOLUTION
If =
400
=2A
200
No-load
Ia0 = 5.6 – 2 = 3.6 A
Ea0 = 400 – 0.18 ¥ 3.6 – 2 = 397.4 V
Full-load
Ia( f l) = 68.3 – 2 = 66.3 A
Ea( f l) = 400 – 0.18 ¥ 66.3 – 2 = 386.1 V
n( fl)
386.1
1
=
¥
=1
n(nl )
397.4 0.97
Observation Because of field weakening caused by armature reaction, full-load speed is equal to no-load speed.
EXAMPLE 7.46 A 750 kW, 250 V, 1200 rpm dc shunt motor has armature and field resistances of 0.03 W
and 41.67 W respectively. The motor is provided with compensating winding to cancel out the armature
reaction. Iron and rotational losses can be ignored. The motor is carrying constant torque load drawing
126 A and running at 1105 rpm.
(a) Assuming magnetic linearity, what would be the line current and motor speed if the field current is
reduced to 5 A. Also calculate the load torque.
(b) The motor has magnetization data as below. Calculate motor speed and line current as in part (a).
Compare and comment upon the results of parts (a) and (b)
Speed: 1200 rpm
If (A)
VOC (V)
4
217
5
250
6
267
7
280
DC Machines
393
SOLUTION
(a) Linear magnetization characteristic
250
=6A
41.67
Ia1 = 126 – 6 = 120 A
Ea1 = 250 – 0.03 ¥ 120 = 246.4 V
n1 = 1105 rpm, w1 = 115.7 rad /s
If 1 =
Now we know
Ea = KaFw = K ¢¢a If w
T = Ka FIa = K¢¢a If Ia
From Eq. (i)
or
Ea1 = 246.4 = K¢¢a ¥ 6 ¥ 115.7
K a¢¢ = 0.355
T = 0.355 ¥ 6 ¥ 120 = 255.6 Nm (constant)
Field current reduced
If 2 = 5 A
T (constant) = K¢¢a If1Ia1 = K¢¢a If 2Ia2
or
6
= 144 A
5
= 144 + 2 = 146 A
Ia2 = 120 ¥
IL2
From Eq. (i)
or
Ea2 = 250 – 0.03 ¥ 144 = 245.68 V
245.68 = 0.355 ¥ 5 ¥ w2
w2 = 138.4 rad/s or 1322 rpm
(b) Rather than drawing the magnetisation curve, we shall linearly interpolate between data points.
or
or
If 1
267
KaF1
If 2
250
= 6 A fi VOC = 267 V at 1200 rpm or 125.7 rad/s
= Ka F1 ¥ 125.7
= 2.124
= 5 fi VOC = 250 V at 1200 rpm or 125.7 rad/s
= KaF2 ¥ 125.7
250
= 1.989
125.7
T(constant) = KaF2Ia2 = KaF1 Ia1
KaF2 =
or
Ê K a F1 ˆ
Ia2 = Ia1 Á
Ë K a F 2 ˜¯
2.124
= 128.1 A
1.989
IL2 = 128.1 + 2 = 130.1 A
Ea2 = 250 – 0.03 ¥ 128.1 = 246.16 V
246.16 = KaF2 w2
= 120 ¥
or
w2 =
246.16
= 123.76 rad /s or 1182 rpm
1.989
(i)
(ii)
394 Electric Machines
1322 - 1182
¥ 100 = 11.8% higher than obtained
1182
by consideration of the actual magnetisation characteristic. So the linearization is not very good for speed estimation.
As armature voltage drop is very small the armature induced emf is nearly the same in both cases (245.68 V and
246.16 V). Therefore, the speed is mainly governed by the flux/pole. On linear basis the flux reduces by a factor of 5/6
= 0.833 while the actual reduction (obtained from the magnetization characteristic) is 250/267 = 0.936. This is why the
actual speed is lower than that calculated on linear basis.
Comparison of results
Speed as calculated by linear assumption is
EXAMPLE 7.47 A 250-V dc shunt motor has Rf = 150 W and Ra = 0.6 W. The motor operates on noload with a full field flux at its base speed of 1000 rpm with Ia = 5 A. If the machine drives a load requiring
a torque of 100 Nm, calculate armature current and speed of motor. If the motor is required to develop
10 kW at 1200 rpm, what is the required value of the external series resistance in the field circuit? Neglect
saturation and armature reaction.
SOLUTION
Assuming linear magnetization characteristic
Ea = KaFwm = Kwm
T = KaFIa = KIa
(i)
(ii)
At no load
250 – 5 ¥ 0.6 = K ¥
2p ¥ 1000
60
K = 2.36
or
When driving a load of 100 N m,
T
100
=
= 42.4 A
K
2.36
250 - 42.4 ¥ 0.6
E
= 95.15 rad/s
wm = a =
2.36
K
60
wn = 909 rpm
n=
2p
Output = 10 kW at 1200 rpm
Ia =
Now
\
Given:
Assuming linear magnetization, Eqs (i) and (ii) can be written as
Ea = K¢If wm
T = K¢ lf Ia
(iii)
(iv)
From the data of no-load operation
250
= 1.67 A
150
K
2.36
=
K¢ =
= 1.413
1.67
If
If =
\
Now
Solving
Substituting values in Eq. (iii)
(250 – 0.6 I¢a) I¢a = 10 ¥ 1000
I¢a = 44.8 A (the higher value is rejected)
250 – 0.6 ¥ 44.8 = 1.413 ¥ I¢f ¥
or
I¢f = 1.257 A
2p ¥ 1200
60
DC Machines
395
250
= 199 W
1.257
Rf (external) = 199 – 150 = 49 W
\
Rf (total) =
Hence
Series Motor
Speed control is achieved here by adjusting the field ampere-turns. There are three ways of changing them:
A diverter resistor is
connected across the field winding as shown in
Fig. 7.91. By varying Rd the field current and
hence the field ampere-turns can be reduced.
From the circuit, it is obvious that
Ê Rd ˆ
Ise = Ia Á
= Kd Ia
Ë Rse + Rd ˜¯
where
Kd =
(7.104)
Rd
Id
Ise
Ia
Rse
Ea
Vt
Rd
1
=
Rse + Rd
Rse /Rd + 1
For linear magnetization
Fig. 7.91 Diverter resistor control circuit
F = Kd K f Ia
(7.105)
Hence Eq. (7.79) can be rewritten as
n=
1
K a¢ K f K d
È Vt K a K f K d
˘
Í
- {Ra + ( Rse || Rd )}˙
Kd T
ÍÎ
˙˚
From Eq. (7.106), speed-torque characteristics for
decreasing values of Kd (decreasing Rd) are plotted in
Fig. 7.92. These can be corrected for saturation and
armature reaction effects.
One precaution to be taken in this method in order to
avoid oscillations in speed initiated by load changes is to
use an inductively wound diverter resistor.
Here the field ampere-turns are
adjusted in steps by varying the number of turns included
in the circuit. The circuit is shown in Fig. 7.93, from which
the following relations are obtained:
N se
¢
I = Kse Ia
Ise (effective) =
N se a
where
Now
(7.106)
n
Decreasing Kd
0
T
Fig. 7.92
torque characteristics
N¢se = tapped field turn with resistance R¢se = Kse R se.
F = Kf Kse Ia, T = Ka Kf Kse J2a, linear case
substituting F in Eq. (7.79 and eliminating Ia with Rse replaced by Kse Rse)
n=
1
K a¢ K f K se
È Vt K a K f K se
˘
Í
- ( Ra + K se Rse ) ˙
T
ÍÎ
˙˚
(7.107)
396
Electric Machines
Nse
Ia
Ea
+
Vt
N¢se
–
Fig. 7.93
The controlled speed-torque characteristics are similar to those of Fig. 7.92 except for the marginal effect
of the resistance term.
3. Series-parallel control Here the field windings are divided into two equal halves and then connected
in series or parallel to control the field ampere-turns. The circuits shown in Figs 7.95(a) and (b). For any
armature current, parallel connection of half windings gives
I ˆ
ÊN
ATparallel = 2 Á se ¥ a ˜
Ë 2
2¯
=
1
1
NseIa = ATseries
2
2
1
Ê Rse Rse ˆ
||
R¢se = Á
= Rse
˜
Ë 2
¯
2
4
and
Nse /2
Nse /2
Nse/2
Ia
(7.108)
Ia/2
Ia
+
Ea
Ea
V
Ia/2
+
Nse/2
V
–
–
(b) Parallel
(a) Series
Fig. 7.94
For the linear case
1
K I
2 f a
In speed control Eq. (7.107), it follows from above
F=
1
2
i.e. only two speeds are possible; parallel field connection gives the higher speed.
Kse = 1 or
(7.109)
DC Machines
397
EXAMPLE 7.48 The following open-circuit characteristic was obtained by separately exciting the field
of a series motor and driving the armature at a speed of 900 rev/min.
Generated emf (V)
Field current (A)
0
0
78
50
150
100
192
150
220
200
The armature and series-field resistances are 0.035 and 0.015 W respectively. Determine the speed and
torque of the motor for a current of 200 A when fed from a 220 V supply and operated with (a) the full fieldwinding, and (b) the field turns reduced by half. (c) diverter of resistance 0.03 W.
Total armature resistance = 0.035 + 0.015 = 0.05 W
Ea = 220 – 200 ¥ 0.05 = 210 V
(a) Full field winding:
At 200 A current, 900 rpm, induced emf = 220 V (from magnetization characteristics)
SOLUTION
Motor speed = 900 ¥
T=
Torque developed,
Ea I a
w
210
= 859.1 rpm
220
210 ¥ 100
= 159.24 Nm.
=
2p ¥ 859.1
60
(b) Field winding terms reduced to half;
0.015
= 0.0075 W
2
= 0.035 + 0.0075 = 0. 0425 W
= 220 – 200 ¥ 0.0425 = 211.5 V
200
=
= 100 A
2
= 150 V (at 900 rpm); from magnetization characteristic
211.5
= 900 ¥
= 1269 rpm
150
211.5 ¥ 200
= 159.24 Nm
=
2p ¥ 1269
60
Rse =
R (total)
Ea
Equivalent full field turns current
Ea
Motor speed
Torque developed,
T
(c) Diverter across series field
Ra = 0.03, Rse = 0.15
1
1
2
=
Kd =
=
Rse /Rd + 1 1 / 2 + 1
3
2
¥ 200 = 133.3 A
Ise = Kd Ia =
3
From the open-circuit characteristic (by interpolation) at Ise = 133.3 A
192 - 150
¥ (133.3 – 100)
150 - 100
Ea = 178 V at 900 rpm
2
Rse || Rd =
¥ 0.015 = 0.1 W
3
Ea = 220 – 200 (0.035 + 0.01) = 211 V
Ea = 150 +
or
398 Electric Machines
Motor speed = 900 ¥
Torque developed =
211
= 1067 rpm
178
211 ¥ 200
= 377.87 Nm.
2p
¥ 1067
60
EXAMPLE 7.49 A 4-pole series-wound fan motor draws an armature current of 50 A, when running
at 2000 rpm on a 230 V dc supply with four field coil connected in series. The four field coils are now
connected in two parallel groups of two coils in series. Assuming the flux/pole to be proportional to the
exciting current and load torque proportional to the square of speed, find the new speed and armature
current. Neglect losses. Given: armature resistance = 0.2 W, resistance of each field coil = 0.05 W.
SOLUTION
Ea = KE Ise n
TL = KLn2 = TM = KM Ise Ia
(i)
(ii)
Field coils in series
and
Ise
Rse
RA
Ia1
Ea1
210
KL ¥ (2000)2
= Ia
= 4 ¥ 0.05 = 0.2 W, Ra = 0.2 W
= 0.2 + 0.2 = 0.4 W,
= 50 A
= 230 – 0.4 ¥ 50 = 210 V
= KE ¥ 50 ¥ 2000
= KM ¥ (50)2
(iii)
(iv)
Field coils in two series groups in parallel
and
From Eqs (iv) and (vi)
Ise (effective) = Ia/2
2 ¥ 0.05
Rse =
= 0.05 W
2
RA = 0.2 + 0.05 = 0.25 W
Ea2 = 230 – 0.25 Ia2 = KE ¥ (Ia2 /2) n2
KLn22 = KM ¥ (Ia2/2) ¥ Ia2
n22
(2000)
or
2
=
(v)
(vi)
2
I a2
2 ¥ (50) 2
n2 = 28.3 Ia2
(vii)
From Eqs (iii) and (v)
230 - 0.25 I a 2
I a2 n2
=
2 ¥ 50 ¥ 2000
210
Substituting for n2 from Eq. (vii) and simplifying
or
I2a2 + 8.4 Ia2 – 7740 = 0
Ia2 = 83.9 A; negative value is rejected
n2 = 28.3 ¥ 83.9 = 2374 rpm
Armature Control
The main requirement of this control scheme is a variable voltage supply to the armature whose current rating
must be somewhat larger than that of the motor. It is superior to the field control scheme in three respects,
outlined below:
DC Machines
399
(i) It provides a constant-torque drive. In the shunt motor case by keeping the field current at maximum
value full motor torque can be obtained at full-load armature current at all speeds.
(ii) Since the main field ampere-turns are maintained at a large value, flux density distortion caused by
armature reaction is limited.
(iii) Unlike field control scheme, speed reversal can be easily implemented here.
There are three main types of armature control schemes. These are discussed below:
Rheostatic Control
Series armature-resistance control Here the applied armature voltage is varied by placing an adjustable
resistance Re in series with the armature as shown in Fig. 7.95 along with the speed-torque characteristics.
IL
If
+
Re
Rf
n
Ia
n0
Vt
Increasing Re
Ea
–
0
(a)
T
(b)
Fig. 7.95 Series armature resistance control and speed-torque characteristics
Some of the limitations of the rheostatic control method are enumerated below:
(i) Only speeds below the rated value can be obtained. This can be shown by using Eq. (7.100) (armature
resistance is negligible) that
n1 =
1 Vt
;
K a¢ F
n2 =
1 Vt - I a Re
K a¢
F
(7.110)
I a Re
n1 - n2
=
(7.111)
Vt
n1
(ii) Range of speeds available is limited because efficiency is reduced drastically for large speed reductions.
By definition of armature efficiency
which gives
n2
I R
(Vt - I a Re ) I a
(7.112)
=1– a e =
n1
Vt I a
Vt
(iii) The speed regulation of the method is poor as for a fixed value of series armature resistance, the speed
varies directly with load, being dependent upon the resistance voltage drop.
hª
In general, rheostatic control is economically feasible only for very small motors (fractional kW) or for
short-time, intermittent show-downs for medium-sized motors.
It is a variation of the rheostatic control. The principle of voltage division is used
to reduce the voltage across the armature as shown for a shunt motor in Fig. 7.96(a). The Thevenin equivalent
circuit as seen from the armature terminals is drawn in Fig. 7.96(b). The no-load armature speed is governed
Shunted armature control
400
Electric Machines
by VTH, which can be independently adjusted by the ratio R2 /R1. The series resistance (Thevenin resistance)
is Re = R1|| R2 = bR1 is very small so the control circuit gives better speed regulation compared to the circuit
arrangement of Fig. 7.96(a). VTH = b Vt
The potential divider circuit for series motor speed control is shown in Fig. 7.96(c) and its Thevenin
equivalent is drawn in Fig. 7.96(d).
On approximate basis neglecting voltage drop in (bR1 + Ra),
Ea ª bVt = K¢a F n
n=
or
1 bVt
; F(Ise) = F(b Ia)
K a¢ F
At light load through F is quite small the reduced value of bVt result in finite motor speed unlike a series
motor operating at rated Vt .
+
R1
R1
Vt
Vt
R2
Ea
R2
Ea
Rse
–
(a) Shunt motor
(c) Series motor
Re = R1||R2 = b R1
+
VTH
+
Ea
Vt
–
–
(b) VTH = bVt ; b =
+
1
1 + R1/R2
R1
Rse
Ia
VTH = bVt
R2
Ea
–
(d)
Fig. 7.96
Shunted armature speed control
DC Machines
401
EXAMPLE 7.50 A 230 V dc shunt motor having armature resistance of 2 W draws an armature current of
5 A to drive a constant torque load at 1250 rpm. At no load it draws a current of 1 A.
(a) A resistance of 15 W is added in series to the armature. Find the motor speed with load torque as
above. Also determine the speed regulation.
(b) A resistance of 15 W is shunted across the armature and 10 W in series with the supply line (as in
Fig. 7.96(a). Calculate the load speed and speed regulation.
(c) Compare the power wasted in external resistance (s) in parts (a) and (b).
Rotational loss torque is negligible. The armature reaction effect is to be ignored.
SOLUTION
or
Ea
n
Ea
220
KaF
(a)
= 230 – 2 ¥ 5 = 220 V
= 1250 rpm, w = 130.9 rad/s
= Ka Fw ; F is constant; constant shunt field current, no armature reaction effect
= KaF ¥ 130.9
= 1.68
Re = 15 W in series
No load speed
Iao = 1 A
Ea = 230 – (15 + 2) ¥ 1 = 213 V
213
w0 =
= 126.8 rad/s
1.68
Load torque constant
As F is constant
Ia = 5 A
Ea = 230 – (15 + 2) ¥ 5 = 145 V
145
w =
= 86.3 rad/s
1.68
126.8 - 86.3
¥ 100 = 46.9%
Speed regulation =
86.3
(b) From the Thevenin equivalent of Fig. 7.96(c)
R1 = 10 W , R2 = 15
R2
15
=
b =
= 0.6
R1 + R2
10 + 15
VTH = 230 ¥ 0.6 = 230 V
RTH = bR1 = 0.6 ¥ 10 = 6 W
No load speed
Iao = 1 A
Ea = VTH – (RTH + Ra) Iao
= 230 – (6 + 2) ¥ 1 = 130 V
130
w0 =
=77.38 rad/s
1.68
On load
Ia = 5 A
Ea = 138 – (6 + 2) ¥ 5 = 98 V
98
w =
= 58.33 rad/s
1.68
402
Electric Machines
77.38 - 58.33
¥ 100 = 32.6%
58.33
Observation Speed regulation is much better (less) in shunted armature control than in rheostatic control.
(c) Power loss
(i) Rheostatic control
Pe = (5)2 ¥ 15 = 375 W
(ii) Shunted armature control
Speed regulation =
Va(across armature) = 98 + 2 ¥ 5 = 108 V
(108) 2
= 777.6 W
15
108
=
= 7.24 A
15
= 7.2 + 5 = 12.2 A
= (12.2)2 ¥ 10 = 1488.4 W
= 777.6 + 1488.4 = 2266 W
P (15 W) =
I (15 W)
Then
I (10 W)
P (10 W)
Pe
Observation External power loss is far larger in shunted armature control than in rheostatic control. In fact it is
much larger than power of the motor (230 ¥ 5 = 1150 W) being controlled.
Remark
Shunted armature control is employed for very small motors where speed regulation requirement is stringent.
EXAMPLE 7.51 A dc shunt motor is connected to a constant voltage source and is driving a constant
torque load. Show that if Ea > 0.5 Vt increasing the resultant flux reduces the speed and if Ea < 0.5 Vt,
increasing the resultant flux increases the speed. The back emf Ea is changed by a series resistance in the
armature circuit.
SOLUTION
Let
Rt = Ra + Re
Ea = Vt – Ia Rt > 1/2 Vt
Ea
Vt
I R
- a 1
=
w=
K aF
Ka F Ka F
T
, Eq. (i) is converted to the form with F as the only variable (T is constant).
Ka F
Vt
T R1
Thus
w=
K a F K a2 F 2
dw
should be negative. It then follows from Eq. (ii)
For w to decrease with F,
dF
dw
2T R
Vt
+ 2 t3 < 0
= 2
dF
Ka F
Ka F
(i)
(ii)
As Ia =
T = KaF Ia, we get
Substituting
or
or
But
Vt
Ka F 2
+
2 K a F I a Rt
K a2 F 3
–Vt + 2 IaRt < 0
Vt – 2 I a R t > 0
Ia Rt = Vt – Ea, so
Vt – 2Vt + 2 Ea > 0
<0
(iii)
(iv)
DC Machines
403
Ea > 0.5 Vt
or
For speed to increase by increasing flux
dw
>0
dF
By increasing the inequality sign in Eq. (iv) it follows:
Ea < 0.5 Vt
Series-parallel control Here two identical motors are coupled together mechanically to a common load. Two
speeds at constant torque are possible in this method—one by connecting the motors armatures in series and
the other by connecting them in parallel as shown in Fig. 7.97. When connected in series, the terminal voltage
V
across each motor is t whereas when they are connected in parallel it is Vt. Thus armature control of speed
2
is achieved; speed (series): speed (parallel) ::1:2.
Ia
Ia
+
If
Vt
2
21a
+
Ia
Ea
If
Ea
Vt
Vt
2
Vt
Ea
Ea
–
–
(a) Armature in series (low speed)
(b) Armature in parallel (high speed)
Fig. 7.97 Series-parallel speed control (shunt-motors); case of constant load torque is illustrated; speed ratio 1:2
Figure 7.98(a) and (b) gives the connections for series-parallel speed control of two identical series motors.
Ia
2Ia
+
+
Ia
Vt
Vt
Ia
Vt
Vt
–
(a) Series connection (low speed)
–
(b) Parallel connection (high speed)
Fig. 7.98 Series-parallel speed control of series motors; case of constant load torque is illustrated; speed ratio 1:2
404
Electric Machines
This method is superior to the rheostatic control insofar as efficiency is concerned. It is, however, limited
to two speed steps. The method is commonly employed for speed control of series traction motors.
EXAMPLE 7.52 A dc shunt motor has speed control range of 1600 rpm to 400 rpm by rheostatic control.
All losses and armature reaction effect may be neglected.
(a) The motor drives a constant power load. It has a speed of 1600 rpm drawing 120 A armature current.
What would be the armature current at 400 rpm?
(b) Repeat part (a) if the load is constant torque.
(c) Repeat parts (a) and (b) if speed is controlled by armature voltage.
SOLUTION
All losses neglected means Ra = 0
(a) Constant power
\
(b) Constant torque
P
Ea
P
n
n
1600 ¥ 120
Ia
T
= Ea Ia
= Kn as shunt field current is constant
= KIan
= 1600
Ia = 120
= 400
Ia = ?
= 400 ¥ Ia
= 4 ¥ 120 = 480A
= Ka Ia, F constant
Therefore Ia is constant independent of speed. Thus at n = 400 rpm
Ia = 120 A
(c) Speed adjustment requires control of Va ª Ea . It does not how it is achieved–rheostatic or armature voltage.
Therefore the armature current is same as found in parts (a) and (b).
Ward Leonard Speed Control
It is combined armature and field control and is therefore, operationally the most efficient method of speed
control with a wide range. The dc motor armature is fed from a variable voltage and adjustable polarity
supply whose current rating must be somewhat higher than that of the motor. The field (shunt) of the motor
is separately excited from an independent dc source (low current rating). The variable voltage dc supply in
older installations is obtained from a dc generator driven by a 3-phase squirrel-cage motor. The field circuit
of the generator is separately excited from a small rectifier unit or by an excitor coupled to an extension of the
motor shaft. The complete arrangement is shown in the connection diagram of Fig. 7.99(a). The connection
of the potentiometer (Pot 1) makes it possible to easily reverse the generator excitation thereby reversing the
voltage polarity for reversal of the direction of rotation of the motor. This type of speed control is known as
Ward-Leonard speed control. Modem installations use SCR circuitry for variable-voltage dc supply drawing
power from ac mains through a transformer. Though expensive, this arrangement is neat and relatively free
from maintenance problems. It is also easily adopted to feedback schemes for automatic control of speed.
At the base speed nb the motor armature is fed at rated voltage and its field current is adjusted to the
maximum value, i.e. the field is excited at rated voltage. Reducing the armature voltage provides a constanttorque speed control where the speed can be reduced below the base value, while the motor has full torque
capability (as 1f = max and Ia can have rated value). For obtaining speeds above nb, the field is gradually
weakned maintaining armature voltage at rated value. The motor torque therefore reduces as its speed increases
DC Machines
405
which corresponds is to constant-kW (or hp) drive. The kind of control over torque-speed characteristic
achieved is illustrated in Fig. 7.99(b) where the nature of power-speed characteristic is also revealed.
AC suplly
DC
Gen
AC
Mot
+
DC
Mot
Exciter
M
Pot 2
Pot 1
+
(a) Ward-Leonard speed control system
P
T
T
P
nbase
Va control
If control
(b) Torque-speed and power speed characteristic
Fig. 7.99
Some of the attractive features of the Ward-Leonard system are listed below in addition to the advantages
mentioned for armature control in general:
(i) The absence of an external resistance considerably improves the efficiency at all speeds. Another
feature which enhances the efficiency is that when the generator emf becomes less than the back emf
of the motor, electrical power flows back from motor to generator, is converted to mechanical form and
is returned to the mains via the driving ac motor. The latter aspect makes it an ideal choice if frequent
starting, stopping and reversals are required.
(ii) No special starting gear is required. As the generator induced voltage is gradually raised from zero, the
motor starts up smoothly.
(iii) Speed reversal is smoothly carried out.
Explanation through Fundamental Relationships
Fundamental relationships are reproduced below:
Electromagnetic power
P = Ea Ia = Tw
(i)
T = Ka F Ia
(ii)
Electromagnetic torque
406
Electric Machines
Back emf
Ea = Ka F w
(iii)
Ea = Vt – Ia Ra
(iv)
Armature circuit equation
Constant torque operation
F = F(max) ; If max, all regulating resistance cut out
T = constant (max)
From Eq. (ii)
Ia = Ia (rated) = constant
As Vt is increased, Ea increases. It follows from Eq. (iii) w increases, P increases almost linearly (IaRa drop
ignored). At Vt = Vt (rated), n = nbase where maximum (rated) power is reached.
Constant power operation
Vt = Vt (rated) held constant
Ia = Ia (rated)
From Eq. (iv), Ea is constant and so P is constant. As If is reduced, F reduces. So from Eq. (iii) speed w
increases and from Eq. (ii) T reduces but P remains constant.
EXAMPLE 7.53 A 200-V shunt motor with a constant main field drives a load, the torque of which varies
at the square of the speed. When running at 600 rpm, it takes 30 A. Find the speed at which it will run and
the current it will draw, if a 20-W resistor is connected in series with armature. Neglect motor losses.
SOLUTION
current, i.e.,
Armature resistance is assumed negligible. Further field current is ignored in comparison to armature
IL = Ia
200 = Ke ¥ 600
T = Kt ¥ 30 = KL ¥ (600)2
As per the data given
(i)
(ii)
With a 20-W resistor added in the armature circuit
(200 – 20 Ia) = Ke ¥ n
Kt Ia = KLn2
(iv)
Dividing Eq. (iii) by (i) and (iv) by (ii)
200 - 20 I a
n
=
200
600
Solving
Ia
n2
=
30
(600) 2
n = 260.5 rpm
Ia = 5.66 A
(v)
(vi)
EXAMPLE 7.54 A 400 V series motor has a total armature resistance of 0.25 W. When running at
1200 rpm it draws a current of 25 A. When a regulating resistance of 2.75 W is included in the armature
circuit, it draws current of 15 A. Find the speed and ratio of the two mechanical outputs. Assume that the
flux with 15 A is 70% of that with 25 A.
DC Machines
Ea = K¢a F n
400 – 0.25 ¥ 25 = K¢a F1 ¥ 1200
400 – (2.75 + 0.25) ¥ 15 = K¢a F2 ¥ n2
SOLUTION
407
(i)
(ii)
Dividing Eq. (ii) by (i)
n2
n2
F
355
¥ 2 =
¥ 0.7
=
1200
F
1200
393.75
1
n2 = 1545.6 rpm
which gives
355 ¥ 15
P02
=
= 0.541
393.75 ¥ 25
P01
Ratio of mechanical outputs,
EXAMPLE 7.55 A dc shunt motor is driving a centrifugal pump whose load torque varies as square of
speed. The pump speed is controlled by varying the armature voltage of the motor with the field current
remaining constant. At full load with an armature voltage of 500 V, the armature current is 128 A. Calculate
the armature voltage required to reduce the speed to 1/ 2 of its original value. Ra = 0.28 W. Ignore the
effect of armature reaction and loss torque (reduction in torque output on account of rotation losses).
SOLUTION
Ia1 = 128 A
Ea1 = 500 – 0.28 ¥ 128 = 464.2 V
As the field current remains constant
T = KT n21 μ 128; field current constant
(i)
Speed is to be reduced to n2 = n1/ 2 . Then
KT (n1/ 2 )2 μ Ia2
(ii)
From Eqs (i) and (ii)
I a2
1
=
or
128
2
New applied armature voltage = Vt2
n
(Vt2 – 0.28 Ia2) μ 1
2
464.2 μ n1
Ia2 = 64 A
(iii)
(iv)
From Eqs (iii) and (iv) we get
Vt 2 - 0.28 ¥ 128
=
464.12
1
2
Vt 2 = 346.1 V
or
EXAMPLE 7.56 The Ward Leonard speed control system of Fig. 7.99 uses two identical machines of
rating 230 V, 4.5 kW, 1500 rpm. The generator is driven at a constant speed of 1500 rpm, Ra = 0.5 W each
machine.
The magnetization characteristic data obtained at 1500 rpm is as under
If (A)
Voc(V)
0.0
45
0.2
110
0.3
148
Neglect the effect of armature reaction.
0.4
175
0.5
195
0.6
212
0.7
223
0.8
230
1.0
241
1.2
251
408
Electric Machines
(a) The motor field current is held constant at 0.8 A. To obtain a motor speed at range of 300-1500 rpm
with power (mechanical) output of 4.5 kW, determine the range of the generator field current.
(b) The generator field current is kept constant at 1 A, while the motor field current is reduced to 0.2 A.
Determine the motor current and speed for a power output of 4.5 kW.
SOLUTION
OCC will not be drawn. Instead interpolation will be used in the regions between the given data points.
Ifm = 0.8 A fi Eam = 230 V at 1500 rpm
nm = 300 – 1500 rpm (range)
nm = 300 rpm
(a)
(i)
230 ¥ 300
= 46 V
1500
Pmot = 4500 = 46 Ia
Ia = 97.8 A
Eag = 46 + 2 ¥ 0.5 ¥ 97.8 = 143.8 V
Eam =
or
From the magnetisation characteristic we get
0.1
¥ (148 – 143.8) = 0.29 A
(148 - 110)
nm = 1500 rpm and Eam = 230 V
Pmot = 230 ¥ Ia = 4500
4500
Ia =
= 19.6 A
230
Eag = 230 + 2 ¥ 0.5 ¥ 19.6 = 249.6 V
If = 0.3 –
(ii)
or
From the magnetisation characteristic
If = 1.2 –
0.2
¥ (251 – 249.6) = 1.18 A
(241 - 230)
Hence range of If is 0.29 – 1.18 A
If g = 1 A fi Eag = 241 A, 1500 rpm (constant)
(b)
Ê 241 - Eam ˆ
ÁË 2 ¥ 0.5 ˜¯ ¥ Eam = 4500
or
or
E2am – 241 Eam + 4500 = 0
Eam = 220.5 V; lower value is rejected
Ifm = 0.2 A fi Eam = 110 V at 1500 rpm
Ê 220.5 ˆ
= 3007 rpm
nm = 1500 ¥ Á
Ë 110 ˜¯
Io =
7.18
241 - 220.5
= 20.5 D
2 ¥ 0.5
BRAKING OF DC MOTORS
Controlled slowing or stopping of a motor and its driven load is as important as starting in many applications
(e.g. cranes, traction on a slope to avoid excessive speed, etc.). Braking methods based on friction,
electromechanical action, eddy-currents, etc. are independent of the motor but sometimes electric braking is
better justified owing to its greater economy and absence of brake wear. The dc motor is still being widely
DC Machines
409
used for traction purposes. One of the main reasons for this is its excellent braking characteristics and ability
of smooth transition from the motor to the generator mode and vice versa. During the braking period, the
motor is operated as a generator and the kinetic or gravitational potential energy (cranes or hoists) is dissipated
in resistors (plugging) or returned to the supply (regenerative braking).
There are three methods of electrical braking:
(i) plugging or counter-current, (ii) dynamic or rheostatic, and (iii) regenerative. These will be discussed
here briefly. The dynamics of the braking problem is discussed in Ref. [9].
Plugging
This involves the sudden reversal of the connections of either the field or armature* winding during motor
operation. A strong braking torque is achieved by maintaining the supply voltage to the armature with
connections reversed (Fig. 7.100). The effective armature voltage
+
–
Vt
(Ea + Vt ) is initially ª 2Vt so that a limiting braking resistor (may
be a starting resistor) must be brought into the circuit. The kinetic
It
energy of the moving system is dissipated in the armature and
braking resistances.
Electrical braking of any variety becomes less effective as speed
decreases with a consequent decrease in the braking torque. This is
because the braking torque
Braking
Pb (breaking power)
n
= [E2a /Rb]/n
resistor, Rb
Tb =
(nK a ) 2 /Rb
n
2
= n(K a/Rb)
Ea
=
Ia
Fig. 7.100 Plugging connections for a
shunt motor
The supply must be switched off close to zero speed (unless the intention is to run the motor in the reverse
direction), using a current or speed directional relay and applying back-up mechanical or hydraulic brakes
to bring the motor to a halt. The large initial current and the resultant high mechanical stress restrict the
application of plugging to small motors only.
Dynamic Braking
The armature is disconnected from the supply
and then a braking resistor Rb is immediately
connected across it (Fig. 7.101). The motor acts
as a generator, driven by the stored kinetic energy
dissipating power in Rb. This is a simple method of
bringing a motor nearly to a standstill. The braking
time is a function of the system inertia, load torque
and motor rating. The field circuit is left connected
to the supply. The only danger is that if the supply
Ia
+
If
Vf
Rb
Ea
–
Fig. 7.101
Dynamic braking, shunt motor
* Because of the problem of interrupting highly inductive field current and the time needed for the field current to
build up in opposite direction, it is a common practice to reverse armature connections.
410
Electric Machines
fails, braking also fails. If the field is left connected across the armature, then initially the braking torque is the
same but starts falling sharply with speed, and the problem arises once the speed falls below the critical value
for self-excitation. For a series motor, it is necessary for braking to reverse either the field or the armature
winding connections for build-up of the armature emf. The value of Rb should be such that (Rb + Ra + Rse) is
less than the critical resistance for the speed at which the braking is commenced.
Regenerative Braking
In this method most of the braking energy is returned to the supply and is used specially where the duty cycle
requires the braking or slowing of the machine more frequently and is most useful in holding a descending
load of high potential energy at a constant speed. The condition for regeneration is that the rotational emf
is more than the applied voltage so that the current is reversed and the mode of operation changes from
motoring to generating.
Regeneration is possible with a shunt and separately excited motors and with compound motors with
weak series compounding. Series motors need a reversal of either the field or the armature connections.
Regeneration is achieved by increasing the field current or armature speed or reducing the supply voltage.
It has been shown in literature [9] that about 35% of the energy put into an automotive vehicle during
typical urban traction is theoretically recoverable by regenerative braking. However, the exact value of the
recoverable energy is a function of the type of driving, the terrain, the efficiency of the drive train, gear ratios
in the drive/train, etc. The method needs a supply capable of accepting the generated power without undue
rise of the terminal voltage.
7.19
EFFICIENCY AND TESTING
Machine efficiency, in general, has been discussed in Sec. 5.10. The approach here will be to apply the
general principles for the specific case of dc machines.
The power flow diagrams for the generating and motoring modes of a dc machine are shown in
Figs. 7.102(a) and (b).
Pe = Pm = EaIa
Pm
Pin
Pwf
Pi
Pout = VIL
Pe
Psh Pc
Pb
(a) Generating mode
Pm = Pe = EaIa
Pm
Pout
Pwf
Pin = VIL
Pe
Pi
Pb
Pc
Psh
(a) Motoring mode
Fig. 7.102
i and stray copper-loss in Pc)
DC Machines
411
Various losses indicated in these figures are:
Pwf = winding and friction in loss
Pi = total core loss
= Pio + (stray load iron loss)
(This break up is possible only for shunt machine)
where
Pio = no load core loss
(7.113)
We define rotational loss as
Prot = Pio + Pwf = rotational loss
Psh = shunt field loss in shunt and compound machine
Pc = armature copper losses including loss in series winding and stray load copper
Pb = brush contact losses
(7.114)
We will combine these losses as
Pk = (Pio + Pwf) + Psh
Pv = (Ra + Rse) I 2a + P stl
P stl = total stray load loss (iron plus copper)
ª proportional to square of armature current
Constant loss,
Variable loss,
where
(7.115)
(7.116)
We can than write
Pv = Kv I 2a
The brush contact loss Pb will be treated separately as it proportional to Ia.
The expressions of dc machine efficiencies are derived below:
Machine efficiency, h =
Output
Input
For generating machine
hG =
Output
Losses
=1–
Output + losses
Output + losses
In form of symbols in Fig. 7.102(a)
hG = 1 –
For motoring machine
hM =
Pk + K v I a2 + Vb I a
VI L + Pk + K v I a2 + Vb I a
(7.117)
Input - losses
Losses
=1–
Input
Input
In terms of symbols of Fig. 7.102(b)
Pk + K v I a2 + Vb I a
VI L
It is known from Eqs. (5.82) and (5.84) that the maximum efficiency occurs when
hM = 1 –
or
Variable loss = constant loss
K vI 2a = Pk
(7.118)
(7.119)
412 Electric Machines
Ia =
or
Pk
Kv
The ratio Ia /la (f l) can be adjusted in machine design by apportioning iron and copper content of the
machine.
7.20 TESTING OF DC MACHINES
Swinburne’s Test
There is a wide variety of non-loading tests that could be performed on dc machines. Swinburne’s test and
Hopkinson’s test are the most important and actually conducted in practice on shunt motors. For obvious
reasons the non-loading test cannot be conducted on a series motor.
This is a no-load test and hence cannot be performed on a series motor. Figure 7.103 gives the connections
for the test. The motor is run at no-load at rated speed by adjusting the field current to a rated value for
accurate determination of no-load loss (Pi0 + Pwf). The machine would run at higher than rated speed with a
rated armature voltage. Therefore a series in the armature circuit is employed to reduce voltage applied to the
motor armature such that it runs at rated speed.
A1
Ia0
A2
nrated
Supply, Vs
If
Va
Fig. 7.103 Swinburne’s test
Constant loss
New
Hence
In Fig. 7.103
Motor input, Va Ia0 = Pi0 + Pwf + I 2a0 Ra
Rotational loss = Pi0 + Pwf = Va Ia0 – I 2a0 Ra
Psh = shunt field loss
= I 2f R f = Vf If
Pk = constant loss
= (Va Ia0 – I 2a0 Ra) + I2f Rf
(7.120)
(7.121)
(7.122)
Variable loss The armature resistance (inclusive of brush contact drop assumed approximately linear) is
measured by a dc test by passing a rated armature current from a battery supply. Then
Pv = I 2a Ra
The stray load-loss can be neglected or estimated as 1% of rated output at full load.
Total loss
PL = Pk + Pv
2
= (Va Ia0 – I a0
Ra) + I f2Rf + I a2 Ra
(7.123)
DC Machines
413
Efficiency can now be calculated at any load current on the following lines:
Generator
Ia = IL + If
hG = 1 –
Then
PL
Vt I L + PL
(7.124)
Motor
Ia = IL – If
Ê
PL ˆ
hM = Á1˜
V
Ë
t IL ¯
Then
(7.125)
Note: Since the resistances (Ra and Rf ) are measured cold, temperature correction must be applied to these
before using in efficiency calculations.
Disadvantages
(i) The stray-load loss cannot be determined by this test and hence efficiency is over-estimated. Correction
can be applied by assuming the stray-load loss to be half the no-load loss.
(ii) Steady temperature rise of the machine cannot be determined.
(iii) The test does not indicate whether commutation would be satisfactory when the machine is loaded .
Separating out Windage and Friction Loss–Retardation Test
If both the armature and field are simultaneous by disconnected in Fig. 7.103 when the motor was running at
steady speed, the armature is governed by the homogeneous first order differential equation.
dw
J
+ fw = 0, motor torque is given
(i)
dt
where J = moment of inertia of the armature in Kg-m2, f = windage and friction coefficient in Nm/rad/s
The natural solution of this equation is
w(t) = Aest
(ii)
Substituting in Eq. (i), we get
or
or
Substituting in Eq. (ii)
Let
then
JAs est + f Aest = 0
(Js + f ) = 0
s = – f /J
w(t) = Ae–(f/J)t
J/f = t, time constant
w(t) = Ae–t/T
(iii)
(iv)
At t = 0, switch-off time w (0) = w0, then A = w0
Therefore at any time t,
w (t) = w0 e–t/t
where t is known as the time constant. At t = t, the speed will be
w(t = t ) = w0 e–1 = 0.368 w0
Thus at t = t, the speed reduces to 36.8% of the initial value. This result can be used to determine t = J/f.
or initial slope of w (t)
(v)
414 Electric Machines
dw
- t/t
= – (wot)
t = 0 = - (w 0 /t ) e
t =0
dt
Tangential line to w(t) at t = 0, intersects the t-axis at t.
This is the alternative way to determine t
(vi)
The Retardation Test
The motor is run to rated speed (or any high speed) and the supply is switched-off. As the motor decelerates
(retards), several speed-time readings are taken, by a speedometer and watch with seconds hand. Initial
readings are taken at small time intervals and the time interval
w
is increased as the motor slows down. The readings (w vs t) are
w0
plotted as shown in adjoining figure. From the graph we find the
time T at which the speed reduces to 36.8% of the initial value. Now
T = J/ f seconds
The moment of inertia is estimated from the measured
dimensions and estimated density of the armature, commutator,
fan and axle. Therefore, we find
f = J/T, Nm/rad/s
Slope (–w0/t)
0.368 w0
0
The windage and friction loss at any speed is then
Pwf = f w W
t
2t
3t
4t
w vs t plot
5t
t
(vi)
EXAMPLE 7.57 A 10 kW, 250 V, dc shunt motor with an armature resistance of 0.8 W and a field resistance
of 275 W takes 3.91 A, when running light at rated voltage and rated speed.
(a) What conclusions can you draw from the above data regarding machine losses?
(b) Calculate the machine efficiency as a generator when delivering an output of 10 kW at rated voltage
and speed and as a motor drawing an input of 10 kW. What assumption if any do you have to make in
this computation?
(c) Determine the maximum efficiencies of the machine when generating and when motoring.
SOLUTION
(a) Shunt field loss,
(b) Generator
(250) 2
= 227.3 W
275
Rotational loss = Prot = 250 ¥ 3.91 – (3.91)2 ¥ 0.8 = 965 W
Psh =
250
10 ¥ 103
= 0.91 A
= 40 A; If =
275
250
Ia = 40 + 0.91 = 40.91 A
PL = 965 + 227.3 + (40.91)2 ¥ 0.8 = 2.53 kW
2.53
hG = 1 –
= 79.8%
10 + 2.53
IL =
Motor
Ia = IL – If = 40 – 0.91 = 39.1 A
PL = 965 + 227.3 + (39)2 ¥ 0.8 = 415 kW
2.415
hM = 1 –
= 75.85%
10
DC Machines
Assumption: Stray-load loss has been neglected.
(c) The condition for maximum efficiency is
or
Total loss
Generator
I2a Ra
0.8Ia
Ia
PL
= Prot + Psh
= 965 + 227.3 = 1192.3 W
= 38.6 A
= 2 ¥ 1192.3 = 2384.6 W
IL = Ia – If = 38.6 – 0.91 = 37.69 A
Pout = 250 ¥ 37.69 = 9422.5 W
hG (max) = I –
2384.6
= 79.8%
9422.5 + 2384.6
Motor
IL = Ia + If = 38.6 + 0.91 = 39.51 A
Pin = 250 ¥ 39.51 = 9877.5 W
hM (max) = 1 –
2384.6
= 75.85%
9877.5
The MATLAB program for the Example 7.57 is shown below.
clc
clear
Pop=l0*l000; Vt=250; Ra=0.8; Rf=275; Ia=3.91;
%% part (a)
Psh=Vt^2/Rf;
Prot=Vt*Ia–Ia^2*Ra;
%% part (b)
%% generator
I1=Pop /Vt;
If=Vt/Rf;
Ia=I1 + If;
Ploss=Prot+Psh+Ia^2*Ra;
Eff_gen=(l–Ploss/(Ploss+Pop))*100
%% motor
Ia=I1–If;
Ploss=Prot+Psh+Ia^2*Ra;
Eff_rnotor=(1–Ploss/(Pop))*100
%% part (c)
Ia=sqrt((Prot+Psh)/Ra);
Ploss_tot=2*(Prot+Psh)
%% generator
I1=Ia–If;
Pout=Vt*I1;
Eff_gen_rnax=(1–Ploss_tot/(Ploss_tot+Pout))*100
% motor
I1=Ia+If;
Pin = Vt*I1;
Eff_motor_max=(1–Ploss_tot /Pin)*100
415
416
Electric Machines
Answer:
Eff_gen = 79.7996
Eff_ motor = 75.8498
EfC_gen_max = 79.8048
Eff_motor_max = 75.8585
EXAMPLE 7.58 A 50-kW, 250 V, 1200 rpm dc shunt motor when tested on no-load at 250 V draw an
armature current of 13.2 A, while its speed is 1215 rpm. Upon conducting other tests it is found that Ra =
0.06 W and Rf = 50 W while Vb (brush voltage drop) = 2 V.
Calculate the motor efficiency at a shaft load of 50 kW at rated voltage with a speed of 1195 rpm. Assume
that the stray load loss is 1% of the output.
What would be the load for the motor to have maximum efficiency and what would be its value?
SOLUTION
No-load test
Armature input = 250 ¥ 13.2 = 3300 W
Prot = Pi0 + Pfw = 3300 – 0.06 ¥ (13.2)2 – 2 ¥ 13.2
= 3263 W
As speed varies very little from no-load to full-load Prot almost remains constant.
On-load
Pout = 50 kW (at shaft)
Let the armature current be Ia. We write the power balance equation.
250 Ia – 0.06 I 2a – 2 Ia – 3263 – 0.01 ¥ 50 ¥ 103 = 50 ¥ 103
Pst
or
0.06
I 2a
– 248 Ia + 53763 = 0
Solving we get
Ia = 229.6 A
Pin = 250 ¥ 229.6 +
h=
(250) 2
= 58650 W
50
50000
¥ 100 = 85.25%
58650
We shall assume that Pst remains mainly constant in the range of load, we are investigating. Then
h=
For maximum efficiency
250 I a - 0.06 I a2 - 2 I a - 3763( Pout + Pst )
250 I a + 250 ¥ 5
dh
=0
dI a
Ia = 284 A
Solving we get
Substituting this value of Ia in h expression, we get
hmax = 86.36%
EXAMPLE 7.59 A 600 V dc shunt motor drives a 60 kW load at 900 rpm. The shunt field resistance is
100 W and the armature resistance is 0.16 W. If the motor efficiency at the load is 85%, determine
(a) the rotational loss
DC Machines
417
(b) the no load armature current and speed. Also find speed regulation
(c) the armature current for electromagnetic torque of 600 Nm
SOLUTION
Pout = 60 kW
Ê1 ˆ
ˆ
Ê 1
PL = Á - 1˜ Pout = Á
- 1 ¥ 60
Ëh ¯
Ë 0.85 ˜¯
= 10.59 W
Pin = 60 + 10.59 = 70.59 kW
70.59 ¥ 103
= 117.65 A
600
600
=
=6A
100
= 117.65 – 6 = 111.65 A
= 600 – 111.65 ¥ 0.16 = 582.14 V
= 900 rpm (given)
= I 2a Ra + Prot + Psh
= (111.65)2 ¥ 0.16 + Prot + 600 ¥ 6
= 4995 W or 4.995 kW
IL =
If
Ia
Ea
n
PL
10.59 ¥ 103
Prot
(a)
or
(b) No load
Armature resistance loss can be ignored
Rotational loss = Prot = 4995 W (loss in nearly independent of speed)
Input power, P0 ª Prot = 4995 W
4995
= 8.325 A
600
At no load Iao Ra drop can be neglected. Therefore
Iao =
Eao ª 600 V
n0 = 900 ¥
600
= 927.6 rpm
582.14
927.6 - 900
¥ 100 = 3.07%
900
Ea = Ka F w
Speed regulation =
(c)
Substituting full load values
2p
60
Ka F = 6.177 (constant as F is constant)
T = Ka F Ia
600 = 6.177 Ia
582.14 = Ka F ¥ 900 ¥
or
Electromagnetic torque
or
Ia = 97.13 A
EXAMPLE 7.60 A dc shunt motor rated 10 kW connected to 250 V supply is loaded to draws 35 A
armature current running at a speed of 1250 rpm. Given Ra = 0.5 W
(a) Determine the load torque if the rotational loss is 500 W.
(b) Determine the motor efficiency if the shunt field resistance is 250 W.
418 Electric Machines
(c) Determine the armature current for the motor efficiency to be maximum and its value. What is the
corresponding load torque and speed?
SOLUTION
(a)
Electromagnetic power,
Ea
Pe
Pout (gross)
Prot
Pout (net)
Speed
w
Load torque,
TL
If
(b)
IL
Pin
Efficiency,
(c) Constant loss,
or
For maximum efficiency
or
Pout (net )
7637.5
= 84 86%
=
9000
Pin
Pc = Prot + Psh ; Psh is neglected
Pk = 500 + 250 ¥ 1 = 750 W
I 2a Ra = Pk
h=
750
= 38.73 A
0.5
= 2 ¥ 750 = 1500 W
= 38.73 + 1 = 39.73 A
= 250 ¥ 39.73 A = 9932.5 W
1500
=1–
= 84.9%
9932.5
= 250 – 0.5 ¥ 38.73 = 230.64 V
230.64
= 1250 ¥
= 1240 rpm (marginally different)
232.5
2p
¥ 1240 = 129.85 rad/s
=
60
= 230.64 ¥ 38.73 = 8932.7 W
= 8932.7 – 500 = 8432.7 W
Ia =
Total loss
IL
Pin
hmax
Speed
= 250 – 0.5 ¥ 35 = 232.5 V
= Ea Ia = Pout (gross)
= 232.5 ¥ 35 = 8137.5 W
= 500 W
= 8137.5 – 500 = 7637.5 W
2p
=
¥ 1250 = 130.9 rad/s
60
7637.5
=
= 58.35 Nm
130.9
250
=
=A
250
= 35 + 1 = 36 A
= 250 ¥ 36 = 9000 W
Ea
n
w
Pout (gross)
Pout (net)
TL =
8432.7
= 84.94 Nm
129.85
EXAMPLE 7.61 A 250 V, 25 kW shunt motor has maximum efficiency of 89% at shaft load of 20 kW and
speed of 850 rpm. The field resistance is 125 W. Calculate the rotational loss and armature resistance. What
will be the efficiency, line current and speed at an armature current of 100 A?
DC Machines
SOLUTION
Total loss,
Power input,
Line current,
PL =
419
P(shaft )
Ê1 ˆ
– P(shaft) = Á - 1˜ P(shaft)
h
Ëh ¯
ˆ
Ê 1
PL = Á
- 1 ¥ 20 = 2.472 kW or 2472 W
Ë 0.89 ˜¯
Pin = 20 + 2.472 = 22.472 kW
22.472
¥ 103 = 89.89 A
250
250
If =
=2A
125
Ia = 89.89 – 2 = 87.89 A
IL =
At maximum efficiency
I2a Ra = Prot + Psh = 2472/2 = 1236 W
1236
= 0.153 W
(89.89) 2
Psh = 250 ¥ 2 = 500 W
Prot = 1236 – 500 = 736 W
Ea = 250 – 89.89 ¥ 0.153 = 236.25 V
n = 850 rpm
Ra =
\
At speed,
Now shaft load is raised till armature current becomes Ia = 100 A
Armature copper loss,
Pc = (100)2 ¥ 0.153 = 1530 W
PL = Pc + Prot + Psh
= 1530 + 1236 = 2766 W
Pin = 250 ¥ IL = 250 ¥ (100 + 2) = 25.5 kW
2.766 ˆ
Ê
¥ 100 = 89.15
h = Á1 Ë
25.5 ˜¯
Ea = 250 – 100 ¥ 0.153 = 234.7 V
n μ Ea, constant If
\
n = 850 ¥
234.7
= 844.4 rpm
236.25
Hopkinson’s Test
This is a regenerative test in which two identical dc shunt machines are coupled mechanically and tested
simultaneously. One of the machines is made to act as a motor driving the other as a generator which supplies
electric power to motor. The set therefore draws only loss-power from the mains while the individual machines
can be fully loaded. (Compare this test with Sumpner’s test on two identical transformers).
Figure 7.104 shows the connection diagram for Hopkinson’s test. One of the machines of the set is started
as a motor (starter connections are not shown in the figure) and brought to speed. The two machines are made
parallel by means of switch S after checking that similar polarities of the machine are connected across the
switch. If this is the case, the voltage across the switch can be almost reduced to zero by adjustment of the
field currents of the machines. Otherwise the polarities of either one of the armatures or one of the fields must
be reversed and the set restarted. The switch S is closed after checking that the voltage across it is negligible
so that heavy circulating current will not flow in the local loop of armatures on closing the switch.
420
Electric Machines
Ia
S
Ifm
Iam
Afm
Aag
Aam
+
DC Supply
Iag
V2
Gen
n
–
Fig. 7.104
Afg
+
Mot
V1
Ifg
–
Hopkinson’s test
The speed of the set and electric loading of the machines can be adjusted by means of rheostats placed in
the two field circuits. The cause-effect relationship to load variation is given below:
Ifg ≠ Æ Eag ≠ Æ Eag > Eam Æ Iag ≠, Iam ≠
Ifm Ø Æ n≠ Æ Eag > Eam Æ Iag ≠, Iam ≠.
Computation of losses and efficiencies
Current drawn from supply.
Ia = Iam – Iag; motor draws larger current as it runs the generator
Total input to armature circuit = Vt Ia
= total armature losses since set output is zero
Hence
Total stray loss = Vt Ia – I 2am Ram – I2ag Rag
= [(Windage and friction loss) + (no-load iron-loss)
+ (stray-load loss)] of both machines
(7.126)
(7.127)
(7.128)
Since the generating machine excitation is more than that of the motoring machine, their no-load iron
(Pi0) and stray-load loss (Pstl ) are not quite equal. As there is no way of separating these in this test and the
difference in any case is small, these are regarded as equal in the two machines. Hence
1
[Vt Ia – I 2am R am – I2ag Rag]
2
Motor field copper-loss = Vt Ifm
Generator field copper-loss = Vt Ifg
PLm = (Pstray + Vt Ifm) + I 2am Ram
PLg = (Pstray + Vt Ifg) + I 2ag Rag
Pstray (each machine) =
Therefore,
and
Now
hM = 1 –
and
hG = 1 –
PLm
Pin, m
PLg
Pout, g + PLg
(7.129)
(7.130)
(7.131)
(7.132)
(7.133)
DC Machines
421
Advantages of Hopkinson’s Test
(i) The two machines are tested under loaded conditions so that stray-load losses are accounted for.
(ii) Since it is a regenerative test, the power drawn from the mains is only that needed to supply losses. The
test is, therefore, economical for long duration test like a “heat run”.
(iii) There is no need to arrange for actual load (loading resistors) which apart from the cost of energy
consumed, would be prohibitive in size for large-size machines.
(iv) By merely adjusting the field currents of the two machines, the load can be easily changed and a load
test conducted over the complete load range in a short time.
Drawbacks of Hopkinson’s Test
(i) Both machines are not loaded equally and this is crucial in smaller machines.
(ii) Since a large variation of field currents is required for small machines, the full-load set speed is usually
higher than the rated speed and the speed varies with load. The full load in small machines is not
obtained by cutting out all the external resistance of the generator field. Sufficient reduction in the motor field current is necessary to achieve full-load conditions resulting in speeds greater than the rated
value.
(iii) There is no way of separating the iron-losses of the two machines, which are different because of
different excitations.
Thus the test is better suited for large machines.
EXAMPLE 7.62 The following test results were obtained while Hopkinson’s test was performed on two
similar dc shunt machines:
Supply voltage = 250 V
Field current of motor = 2 A
Field current of generator = 2.5 A
Armature current of generator = 60 A
Current taken by the two armatures from supply = 15 A
Resistance of each armature circuit = 0.2 W
Calculate the efficiency of the motor and generator under these conditions of load.
SOLUTION
Using Eq. (7:128)
Iam = Iag + Ia = 60 + 15 = 75 A
1
[Vt Ia – I2am Ram – I 2ag Rag]
2
1
= [250 ¥ 15 – 752 ¥ 0.2 – 602 ¥ 0.2] = 952.5 W
2
= Vt Iam + Vt Ifm = 250 ¥ 75 + 250 ¥ 2 = 19250 W
Pstray =
Pin,m
Recalling Eq. (7.129)
2
PLm = (Pstray + Vt Ifm) + I am
Ram
= (952.5 + 250 ¥ 2 ) + 752 ¥ 0.2 = 2577.5 W
The motor efficiency is given by
hM = 1 –
PLm
2577.5
= 86.66%
=1–
19250
Pin, m
422
Electric Machines
Using Eq. 7.130
PLg = (Pstray + Vt Ifg) + I2ag Rag
= (952.5 + 250 ¥ 2.5) + 602 ¥ 0.2 = 2297.5 W
Pout, g = Vt Iag = 250 ¥ 60 = 15000 W
hG = 1 –
PLg
Pout, g + PLg
=1–
2297.5
= 86.7%
15000 + 2297.5
Field’s Test: Two Identical Series Motors
Regenerative test on two identical series motors is not feasible because of instability of such an operation and
the possibility of run-away speed. Therefore, there is no alternative but to conduct a loading test. In Field’s
test the two motors are mechanically coupled with the motoring machine driving the generator which feeds
the electrical load. The connection diagram of the Field’s test is shown in Fig. 7.105. It is observed that
excited and its excitation is identical to that of motor at all loads. This ensures that the iron-loss of both
the machines are always equal.
load cannot be switch off accidentally.
Vm = Vrated
Ig
Im
Vg
Mot
Gen
+
Local (P0)
n
Vm
V supply
–
Im
Fig. 7.105
Under load conditions:
Input to the set,
Pi = Vm Im
Output of the set,
P0 = Vg Ig
Total loss,
PL = Vm Im – VgIg
2
Total copper loss,
Pc = (Ram + 2Rse) I m
Total rotational loss,
Prot (total) = Pi – P0 – Pc
Rotational loss of each machine
Prot = Prot (total)/2
Efficiencies
hM = 1 –
Prot + ( Ram + Rse ) I m2
Vm I m
(i)
(ii)
(iii)
(iv)
(v)
DC Machines
hG = 1 –
Prot + ( Rag I g2 + Rse I m2 )
423
(vi)
Vg I g + [ Prot + ( Rag I g2 + Rse I m2 ]
Note: Series traction motors are normally available in pairs because of the speed control needs. Being a load
test even though the load is electrical it can be conducted on small motors only.
7.21
DC MACHINE DYNAMICS
The DC machines are quite versatile and are capable of giving a variety of V-A and speed-torque characteristics
by suitable combinations of various field windings. With solid-state controls their speeds and outputs can be
controlled easily over a wide range for both dynamic and steady-state operation. By addition of the feedback
circuit, the machine characteristics can be further modified. The aim of this section is to study dc machines
with reference to their dynamic characteristics.
Ra
La
For illustration, let us consider the separately excited
Ia
dc machine shown schematically in Fig. 7.106. For +
ease of analysis, the following assumptions are made:
+
Lf
If
(i) The axis of armature mmf is fixed in space,
ea
Vt
along the q-axis.
TL
Rf
(ii) The demagnetizing effect of armature reaction is
–
T
wm
neglected.
–
Vf
+
–
(iii) Magnetic circuit is assumed linear (no hysteresis
Fig.
7.106
Schematic
representation
of
a
separately
and saturation). As a result all inductances
(which came into play in dynamic analysis) are
regarded as constant.
The two inductance parameters appearing in Fig. 7.106 are defined below:
La = armature self-inductance caused by armature flux; this is quite small* and may be neglected without
causing serious error in dynamic analysis
Lf = self-inductance of field winding; it is quite large for shunt field and must be accounted for
Mutual inductance (between field and armature) = 0; because the two are in space quadrature.
Further for dynamic analysis it is convenient to use speed in rad/s rather than rpm.
Applying Kirchhoff’s law to the armature circuit,
d
ia (t)
dt
ea(t) = K eif (t)w m; Ke = constant (f (t) μ if (t))
Vt = ea(t) + Raia (t) + La
where
(7.134)
(7.135)
Similarly for the field circuit,
d
if (t)
dt
For motoring operation, the dynamic equation for the mechanical system is
vf (t) = R f i f (t) + Lf
T(t) = Kt if (t)ia(t) = J
d
w m(t) + Dw m(t) + TL(t)
dt
* The armature mmf is directed along the low permeance q-axis.
(7.136)
(7.137)
Electric Machines
424
where
J = moment of inertia of motor and load in Nms2
D = viscous damping coefficient representing rotational torque loss, Nm rad/s
Energy storage is associated with the magnetic fields produced by if and ia and with the kinetic energy
of the rotating parts. The above equations are a set of nonlinear* (because of products if (t)wm and if (t)ia(t))
state equations with state variables if, ia and wm. The solution has to be obtained numerically.
Transfer Functions and Block Diagrams
In the simple linear case of motor response to changes in armature voltage, it is assumed that the field voltage
is constant and steady-state is existing on the field circuit, i.e. If = constant. Equations (7.134), (7.136) and
(7.137) now become linear as given below
v(t) = K¢e wm(t) + Raia (t) + La
T(t) = K¢t ia (t) = J
d
ia (t)
dt
(7.138)
d
wm(t) + Dwm(t) + TL(t)
dt
(7.139)
Laplace transforming Eqs (7.138) and (7.139)
V(s) = K¢ewm(s) + (Ra + sLa) Ia(s)
T(s) = K¢t Ia(s) = (sJ + D)w m(s) +TL(s)
(7.140)
(7.141)
These equations can be reorganized as
Ia(s) =
V ( s ) - K e¢w m ( s )
( Ra + sLa )
1/Ra
(1 + st a )
ta = La/Ra = armature circuit time-constant
= [V(s) – K¢ewm (s)] ¥
where
(7.142)
1/D
(1 + st m )
tm = J/D = mechanical time-constant
T(s) = K¢t Ia (s)
wm (s) = [T(s) – TL (s)] ¥
Also
where
(7.143)
(7.144)
From Eqs (7.142) – (7.144), the block diagram of the motor can be drawn as in Fig. 7.107. It is a secondorder feedback system with an oscillatory response in general. It is reduced to simple first-order system, if La
and therefore ta is neglected
TL(s)
V(s)
1/Ra
+
–
1+sta
Ia(s)
K¢t
T(s)
–
+
1/D
1+stm
wm(s)
K¢e
Fig. 7.107
* This is inspite of the fact that the magnetic circuit has been regarded as linear.
wm(s)
DC Machines
425
Shunt Generator Voltage Build-up
The qualitative explanation for the voltage build-up
process in a shunt generator has already been advanced
in Sec. 7.11. Here the mathematical treatment of this
problem will be given, which in fact boils down to the
solution of a nonlinear differential equation.
Referring to Fig. 7.108 it is seen that for any field
current the intercept ab, between the OCC and the
Rf -line gives the voltage drop caused by the rate of
change of Ff and the intercept bc gives the drop in
the field resistance. The two together balance out the
generated emf ea (neglecting if Ra, the armature drop).
Thus
Rf -line
OCC
ea0
a
b
Nf
Rf I f
dFf
dt
er
0
dF f
= ea – R f if
(7.145)
dt
Ff = field flux/pole
Nf = number of turns of field winding
Nf
where
ea
c
Fig. 7.108
If
Magnetization curve and Rf-line
The field flux Ff is greater than the direct axis air-gap flux Fd because of leakage.
Taking this into account
Ff = s Fd
(7.146)
Here s is known as the coefficient of dispersion.
Recalling Eq. (7.3),
Fd =
ea
Ka wm
(7.147)
Substituting Eqs (7.146) and (7.147) in Eq. (7.148),
Nfs
K aw m
◊
dea
= ea – if Rf
dt
(7.148)
Multiplying numerator and denominator by Nf Pag
where Pag is the permeance of the air-gap/pole
Nfs
K aw m
=
N 2f s Pag
K aw m Pag N f
It is easily recognized that the numerator is the unsaturated value of field inductance, Lf , and the
denominator is the slope of the air-gap line. Both are constants. Hence,
L f dea
= ea – R f if
K g dt
Rewriting Eq. (7.145)
dt =
L f Ê dea ˆ
Á
˜
K g Ë ea - R f i f ¯
(7.149)
426 Electric Machines
t=
or
Lf
Kg
ea
dea
a - Rf if
Úe
er
(7.150)
where the limits of integration,
er = residual voltage
ea = instantaneous generated voltage
This integral can be evaluated graphically by summing up the areas on a plot of 1/(ea – R f if ) against ea.
This approach is employed to plot ea against time. The
theoretical time needed for the generated emf to attain the
no-load value, ea0 would be infinite; hence in practice the
time needed to reach 0.95 ea0 is taken as the time needed
to reach ea0. The variation of ea with time is plotted in
Fig. 7.109.
The response is rather sluggish since only small
voltage differences (= ea – R f i f) contribute to the flux
build-up (Ff)·
ea
er
0
t
Fig. 7.109 Voltage build-up of a shunt generator
As has been discussed in Chapter 2, a number of new permanent magnet materials—ceramics, and rare earth
magnetic materials-have become available commercially. These materials have high residual flux as well as
high coercivity. Smaller fractional and sub-tractional hp dc motors are now constructed with PM poles. As
no field windings are needed, so no field current and continuous field loss. As a result PMDC motors are
smaller in size than the corresponding rated field wound type motors, this fact partially off-sets the high cost
of permanent magnets. Obviously, these motors offer shunt type characteristic and can only be armature
controlled. The risk of permanent magnetism getting destroyed by armature reaction (at starting/reversing or
heavy over-loads) has been greatly reduced by the new PM materials.
Constructional Features
The stator is an annular cylindrical shell of magnetic material on the inside of which are bonded fractional
cylindrical permanent magnets (usually two poles) as shown in the cross-sectional view of Fig. 7.110. The
magnetics are radially magnetized as shown by arrows. The rotor is laminated magnetic material with slotted
structure in which the winding is placed whose coil ends suitably connected to the commutator (usual
construction).
Magnetic Circuit
It is seen from Fig. 7.110 that the flux crosses the airgap length (lg) twice and the thickness of the permanent
magnet tm twice. The iron path in the shell and rotor teeth and core being highly permeable (m
) can be
assumed to consume any mmf. The mmf balance equation is then
or
2lg Hg + 2tm Hm = 0 ; no external mmf is applied
lg Hg + tm Hm = 0
DC Machines
427
Outer shell
Radially magnetized
permanent magnets
(arrows indicate direction
of magnetization)
Rotor
Fig. 7.110 Cross section view of permanent-magnet motor
or
Air-gap flux density,
Also
Êt ˆ
Hg = – Á m ˜ Hm
Ë lg ¯
Bg = m0Hg
Bg = Bm
(7.151)
(7.152)
it then follows from Eq. (7.151) that
Êt ˆ
Bg = – m0 Á m ˜ Hm = Bm
Ë lg ¯
(7.153)
which is the load line of the magnetic circuit.
The dc magnetization characteristics of various PM materials is presented in Fig. 2.20. Obvious choice of
PMDC motor is neodymium-iron-boron which has high coercivity and high retentivity. Its characteristic is
almost a straight line which can be expressed as
Ê
ˆ
1.25
Bm = – Á
m H + 1.25
3
-7 ˜ 0 m
Ë 940 ¥ 10 ¥ 4p ¥ 10 ¯
or
Bm = –1.06 m0Hm + 1.25
(7.154)
The solution of Eqs. (7.153) and (7.154) yields Bm = Bg from which we can find the flux/poles as
F = Bg Ag
Unlike a normal dc motor F is a constant quantity
Emf and torque equations
As F is constant
where Km = motor torque constant
Electromagnetic power,
Ea = Ka F w
T = KaF Ia
Ea = Km w
T = Km Ia
Pe = Ea I a
(7.155)
(7.156)
428
Electric Machines
Circuit model As there is no electrically excited field and the
permanent magnetic creates a constant flux/pole, the circuit of a
permanent magnet motor is as drawn in Fig. 7.111. where the armature
resistance Ra is shown in series with the armature which has induced
back emf. The armature circuit equation is Ea = Vt – Ia R a.
Ra
Ia
+
Ea
Vt
n
Rated output
(both motors)
Speed
Speed
fi In PMDC, even for wider range of armature voltage the torque
speed characteristics are linear, Fig. 7.112(b)
–
fi PMDC motor exhibits better speed regulation and efficiency
Fig. 7.111 Circuit model
than dc shunt motor.
fi The main problem of dc shunt motor is goes to run away when the field terminals are opened. But in
PMDC there is no run away problem, so it gives practical benefit to the industry applications.
fi PMDC produces high torque even at low speeds which is shown in Fig. 7.112(a) and also it produces
high starting torque compared to dc shunt motors.
V4 > V3 > V2 > V1
V1
Starting torque of
shunt motor
Starting torque of
PMDC motor
V4
Torque
Torque
(a)
Fig. 7.112
V3
V2
(b)
Speed torque characteristics of PMDC motors
EXAMPLE 7.63 A PMDC motor has an armature resistance of 4.2 W. When 6 V supply is connected to
the motor it runs at a speed 12,125 rpm drawing a current of 14.5 mA on no-load
(a) Calculate its torque constant
(b) What is the value of rotational loss?
With an applied voltage of 6 V,
(c) calculate the stalled torque and stalled current of the motor (motor shaft held stationary)
(d) at a gross output of 1.6 W, calculate the armature current and efficiency. Assume that the rotational
loss varies as square of speed.
(e) calculate the motor output at a speed of 10,250 rpm and the efficiency.
SOLUTION
No load
(a)
Vt = 6 V, Iao = 14.5 mA, n = 12125 rpm or w = 1269.7 rad/s
Ea = 6 – 14.5 ¥ 103 ¥ 4.2 = 5.939 V
5.939 = K m w = Km ¥ 1269.7
DC Machines
Km = 4.677 ¥ 10–3
Prot = Ea Ia ; there is no load
= 5.939 ¥ 14.5 ¥ 10–3 = 0.0861 W
or
(b) Rotational loss,
(c) Stalled current
w = 0 so Ea = 0
6
Ia (stall) =
= 1.4285 A
4.2
Torque (stall) = K mI a(stall) = 4.677 ¥ 10–3 ¥ 1.428
= 6.67 m Nm
Pout (gross) = 1.6 W = EaI a
(6 – 4.2 Ia) Ia = 1.6
4.2 I2a – 6 Ia + 1.6 = 0
(d)
Solving we find
Ia = 0.354 A, 1.074 A
Thus
Ia = 0.354 A; higher value rejected
Ea = 6 – 0.854 ¥ 4.2 = 4.513 V = Km w
4.513 ¥ 103
= 965 rad/s
4.677
Rotational loss (proportional to square of speed)
w=
2
Ê 965 ˆ
= 0.05 W
Prot = 0.0861 ¥ Á
Ë 1269.7 ˜¯
Power input,
Pout (net) = Pout (gross) – Prot
= 1.6 – 0.05 = 1.55 W
Pi = V t I a = 6 ¥ 0.354 = 2121 W
1.55
¥ 100 = 73%
2.124
n = 10250 rpm or w = 1073.4 rad/s
Ea = K mw = 4.513 ¥ 10–3 ¥ 1073.4 = 4.844 V
h=
(e) Motor speed,
6 - 4.844
= 0.275 A
4.2
Pout (gross) = Pe = Ea I a
= 4.844 ¥ 0.275 = 1.332 W
Ia =
2
Ê 1073.4 ˆ
= 0.0615 W
Prot = 0.0861 ¥ Á
Ë 1269.7 ˜¯
Pout (net) = 1.332 – 0.0615 = 1.27 W
Pin = 6 ¥ 0.275 = 1.65 W
h=
1.27
¥ 100 = 77%
1.65
429
430 Electric Machines
7.23
DC MACHINE APPLICATIONS
Whenever the application of any machine is considered, its operating characteristics along with its economic
and technical viability as compared to its competitors are the essential criteria. For a dc machine, of course,
the main attraction lies in its flexibility, versatility and ease of control. This explains why in spite of its rather
heavy initial investment it still retains its charm in strong competitive industrial applications. In the world,
today, around 25% of the motors manufactured are dc motors.
With the advent of various power electronic devices, there is no doubt that the importance of a dc generator
has gone down. Now, for ac to dc transformation, the dc generator as part of an ac-to-dc motor-generator
set has to compete with SCR rectifiers and various other types of controlled power electronic devices
which usually are less costly, compact, relatively noise-free in operation and need minimum maintenance,
but suffer from the disadvantages of having poor power factor, harmonic generation, poor braking, etc.
However, some of the important applications of a dc generator include—dynamometers, welding, crossfield generators for closed-loop control systems, tachogenerators, etc. Separately-excited generators are
still in use for a wide output-voltage control such as in the Ward-Leonard system of speed control.
In dc series motor the starting torque is very high, up to five times the full-load torque. It may be interesting
to note that the maximum torque in a dc motor is limited by commutation and not, as with other motors, by
heating. Speed regulation of a dc series motor can be varied widely. For drives requiring a very high starting
torque, such as hoists, cranes, bridges, battery-powered vehicles and traction-type loads, the series dc motor
is the obvious choice. Speed control is by armature resistance control. Its closest rival is the wound-rotor
induction motor with a rotor resistance control. But ultimately the availability and economics of a dc power
is the deciding factor rather than the motor characteristics.
Compound motor characteristics depend naturally upon the degree of compounding. Shunt field of course
restricts the no-load speed to a safe value. Its main competitor is the squirrel-cage high-slip induction motor.
A compound motor has a considerably higher starting torque compared to a shunt motor and possesses,
a drooping speed-load characteristic. Compound dc motors are used for pulsating loads needing flywheel
action, plunger pumps, shears, conveyors, crushers, bending rolls, punch presses, hoists, rolling mill, planing
and milling machines, etc.
A dc shunt motor has a medium starting torque. Speed regulation is about 5–15%. It is used essentially
for constant speed applications requiring medium starting torques, such as centrifugal pumps, fans, blowers,
conveyors, machine tools, printing presses, etc. Owing to the relative simplicity, cheapness and ruggedness
of the squirrel cage induction motor, the shunt motor is less preferred for constant-speed drives except at lowspeeds. At low speeds, dc shunt motors are comparable with synchronous motors. The outstanding feature
of a dc shunt motor however is its superb wide range flexible speed control above and below the base speed
using solid-state controlled rectifiers (discussed in Ch. 12).
In general, whenever a decision is to be made for a choice of a suitable motor for a given application, it
is necessary to make specific, analytic, economic, and technical comparison of all practical choices. Finally,
it should be mentioned to the credit of a dc machine that it still remains most versatile, flexible, easily
controllable energy conversion device whose demand and need would continue to be felt in industries in
future for various applications discussed above.
DC Machines
431
The main advantage of dc machines lies in their flexibility, versatility and high degree of control. The
disadvantages are complexity associated with armature winding and commutator/brush system, more
maintenance and less reliability.
Lap winding – number of parallel paths A = P
number of brushes = P, equalizer rings needed = P
Wave winding – number of parallel paths = 2, independent of number of poles
number of brushes, 2 needed but P used in practice
no equalizer rings needed
2p
Ea = K aFwm = ÊÁ ˆ˜ KaFn V
Ë 60 ¯
where
Ê ZP ˆ
Ka = Á
, machine constant
Ë 2p A ˜¯
T = KaFIa ; Ka same as in emf equation
T = electromagnetic torque developed
Twm = EaIa
Vt = Ea – Ia Ra ; Vt < Ea
Ra = armature resistance, very small 0.01 pu
There is constant voltage drop at brush about 1 to 2 V. Ia flow out of positive terminal.
Vt = Ea + Ia R a; Vt > Ea
Ia flows into positive terminal
d-axis – along the axis of main poles
q-axis – along the magnetic neutral axis at 90° electrical to d-axis
ATa along magnetic neutral axis (q-axis) at 90° to d-axis
Nature – cross magnetizing, weakens one side of poles and strengthens the other side of poles. F
remains constant in linear region of magnetizing. In saturation region of main poles F decreases,
a demagnetizing effect.
Armature reaction creates distortion in the flux density, shift in MNA, increased iron loss, commutation
problem and commutator sparking.
the harmful effects of armature reaction
432 Electric Machines
out the influence of one pole pair to the next pole pair,
di (coil) ˘
the current in it must reverse. The reactance emf ÈÍ
˙ opposes the change.
Î dt ˚
Interpoles – to aid current reversal in commutating coil narrow poles are placed in magnetic neutral
region with polarity such that the speed emf induce in the coil opposes reactance emf.
Excitation of machine poles Separate excitation from independent source, shunt excitation (from
armature voltage), series excitation (from armature current), compound excitation (combined); it
could be cumulative compound (series excitation aids shunt excitation) or differential compound
(series excitation opposes shunt excitation, not used in practice)
Machine types as per method of excitation
Generator: shunt, series, compound
Motor : shunt, series, compound
Compound connection – long shunt and short shunt
VOC (= Ea) vs I f. Machine is run as separately excited generator
at constant speed. This indeed is the magnetization characteristic.
Separately excited dc generators have the advantage of permitting a wide range of output voltages.
But self-excited machines may produce unstable voltages at lower output voltages, where the field
resistance line becomes essentially tangent to the magnetization curve.
Cumulative compounded generators may produce a substantially flat voltage characteristics or one
which rises with load, where as shunt or separately excited generators may produce a drooping voltage
characteristics.
Shunt Motor
1 Vt - I a Ra
Ê 2p ˆ
, T = Ka F Ia ; K¢a = Á ˜ Ka
Ë 60 ¯
K o¢
F
The speed n decreases slightly as load torque increases and so Ia R a drop increases. This is typical
slightly drooping shunt characteristic. However, armature reaction reduces F due to saturation
effect. This counters drop in speed.
Speed Control
n=
1 Vt
, IaRa ignored
K o¢ F
Field Control As shunt field current is reduced, F reduces and n increases and torque developed
reduces. Constant-kW control.
Armature Control As Vt is increased with Fmax’ n increases. The machine develops constant
torque. Constant – T control
Series Motor As the series field is excited by armature current
n=
1 ˆ Vt
n= Ê
, T = Ka F Ia = (KaKf )I a2, on linear basis
ÁË K ¢ K ˜¯ I a
o f
The speed reduces sharply with load torque (Ia increases). This is typical series characteristics. To
be noted that at no-load Ia becomes zero and speed tends to infinite. The series motor should never
be light loaded. Ideal for traction-type loads.
DC Machines
433
Speed Control
Field Control F is controlled by tapping series field turns or by a diverter resistance across the
series field.
Armature Control Vt is controlled in two steps by series-parallel connection of two mechanically
coupled identical series motors.
Compound Motor (cumulative) Speed-torque characteristic lies in between shunt and series
characteristic depending upon compounding. The motor has a definite no-load speed determined
by the shunt field.
Motor starting At start Ea = 0, direct starting current is unacceptably high. A series resistance is
included in motor armature circuit and is cutout in steps as the motor speeds up.
Motor efficiency Constant loss, Pk = core loss, windage and friction loss, shunt field loss
Variable loss, armature copper loss I a2 Ra
At h max
Variable loss = Constant loss
I 2a Ra = Pk or Ia =
Pk
Ra
Relationships to remember
nμ
Ea
, T μ FIa
F
If F μ I f (linear magnetization)
n μ Ea , T μ If Ia, T μ Ia2 (for series motor)
If
Approximation
On no-load, Ea ª Vt
This approximation may be used even on load where less accuracy is acceptable.
7.1 A compensated dc machine has 20000 AT/
pole. The ratio of the pole arc to pole pitch
is 0.8. The interpolar air-gap length and flux
density are respectively 1.2 cm and 0.3 T. For
rated Ia = 1000 A, calculate the compensating
winding AT per pole and the number of turns
on each interpole.
7.2 The no-load saturation curve for a generator
operating at 1800 rpm is given by the
following data
Eg (V)
8
213
40
234
74
248
113
266
152
278
If (A)
(a)
(b)
(c)
(d)
0
0.5
1.0
1.5
2.5
3.0 3.5
4.0
5.0
6.0
Plot the no-load saturation curve for
1500 rpm.
Calculate the generated voltage when the
generator is operating on no-load with a
field current of 4.6 A and at a speed of
1000 rpm.
What is the field current required to generate 120 V on no-load when the generator
is operating at 900 rpm?
This machine is operated as a shunt
generator at 1800 rpm with a field current
434 Electric Machines
of 4.6 A. What is the no-load voltage when
the generator is operating at 1500 rpm?
7.3 The accompanying data are given for
the saturation curve of an 80-kW, 220-V,
1200-rpm shunt generator, the data being for
1200 rpm:
If (A)
0
2.5
Eg (V) 10
188
0.4
3.2
38
222
0.8
4.0
66
248
1.2
4.5
96
259
1.6
5.0
128
267
2.0
5.5
157
275
(a) The shunt field resistance is adjusted to
50 W and the terminal voltage is found to
be 250 V, at a certain load at 1200 rpm.
Find the load supplied by the generator
and the induced emf. Assume that the flux
is reduced by 4% due to armature reaction.
Armature resistance is 0.1 W.
(b) For the same field resistance and an
armature current of 250 A obtain the
values of Eg, Vt and If .
7.4 Find the resistance of the load which takes a
power of 5 kW from a shunt generator whose
external characteristic is given by the equation
Vt = (250 – 0.5IL)
7.5 A dc shunt generator rated 175 kW, 400 V,
1800 rpm is provided with compensating
winding. Its magnetization data at 1800 rpm
is given below:
If (A)
VOC(V)
1
6
100
440
2
7
200
765
3
8
300
475
4
9
370
480
5
415
Other data of the generator are
Ra = 0.05 W, Rf = 20 W, Rext = 0 to 300
when Rext = regulating resistance in the shunt
field.
The generator connected as shunt is run by a
prime-mover at 1600 rpm.
(a) Find the value of Rext for the no-load
voltage to be 400 V.
(b) A load resistance of 1 W is connected
across the generator terminals. What
should be the value of Rext for the load
voltage to be 400 V.
(c) What would be the generator terminal
voltage when the load resistance is
disconnected?
7.6 In a 110 V compound generator, the resistance
of the armature, shunt and series windings are
0.06, 25 and 0.05 W respectively, The load
consists of 200 lamps each rated at 55 W,
100 V. Find the emf and armature current,
when the machine is connected for (a) long
shunt (b) short shunt (c) How will the ampereturns of the series windings be changed, if in
(a) a diverter of resistance 0.1 W is connected
across the series field? Ignore armature
reaction and brush voltage drop.
7.7 A dc shunt generator has the following opencircuit characteristic when separately excited:
Field current, A 0.2
1.0
Emf, V
80
210
0.4
1.4
135
228
0.6
2.0
178
246
0.8
198
The shunt winding has 1000 turns per pole
and a total resistance of 240 W. Find the
turns per pole of a series winding that will be
needed to make the terminal voltage the same
at 50 A output as on no-load. The resistance
of the armature winding, including the series
compounding winding, can be assumed to be
0.36 W and constant. Ignore armature reaction.
7.8 A dc compound generator has a shunt field
winding of 3600 turns per pole and a series
field winding of 20 turns per pole. Its opencircuit magnetization characteristic when it is
separately excited by its shunt field winding
and driven at its no-load rated speed is given
below:
AT/pole 3120
Emf (V) 289
4680
361
6240 7800
410 446
9360
475
The full-load armature current is 100 A and
the ohmic drop in the armature circuit for this
current is 20 V including brush drop. At no
load the ohmic drop may be ignored and the
DC Machines
terminal voltage is 415 V. The fall in speed
from no-load to full-load is 8%; the shunt
field circuit is connected across the output
terminals of the machine and its resistance is
kept constant.
Determine the terminal voltage and power
output for the full-load armature current
of 100 A. Neglect the effects of armature
reaction.
7.9 A 250 kW, 6-pole, dc compound generator is
required to give 500 V on no-load and 550 V
on full-load. The armature is lap-connected
and has 1080 conductors; the total resistance
of the armature circuit is 0.037 W. The opencircuit characteristic for the machine at rated
speed is given by:
Armature
voltage, V
Field ampereturns/pole
500
535
560
580
6000
7000
8000
9000
The field ampere-turns per pole to
compensate for armature reaction are 10%
of the armature ampere-turns per pole. The
shunt field winding is connected across the
output terminals and has a resistance of 85 W.
Determine the required number of series turns
per pole.
7.10 A 10 kW, 250 V shunt motor has an armature
resistance of 0.5 W and a field resistance of
200 W. At no load and rated voltage, the speed
is 1200 rpm and the armature current is 3 A.
At full load and rated voltage, the line current
is 47 A and because of armature reaction, the
flux is 4% less than its no-load value.
(a) What is the full-load speed?
(b) What is the developed torque at full load?
7.11 A 75 kW, 250 V, dc shunt motor has the
magnetisation data at 1200 rpm as below:
If (A)
Ea (V)
1
5
70
250
2
6
130
274
3
7
183
292
4
8
207
307
435
No-load test on the motor conducted at
1100 rpm yielded rotational loss (no-load iron
loss + windage and frictional loss) as 2200 W.
Stray load loss can be taken as 1% of output.
Given: Ra = 0.025 W
(a) Determine the motor speed and mechanical (net) output at an armature current
of 300 A. Assume that armature reaction
causes 5% voltage reduction in induced
emf.
(b) Shunt field turns/pole = 1200. Two series
turns are added in cumulative compound.
Assuming Rse to be negligible, calculate
speed and power output.
7.12 A 200 V shunt motor has Ra = 0, 1 W, Rf =
240 W and rotational loss 236 W. On fullload, the line current is 9.8 A with the motor
running at 1450 rpm.
Determine:
(a) the mechanical power developed
(b) the power output
(c) the load torque
(d) the full-load efficiency.
7.13 A 220 V unsaturated shunt motor has an
armature resistance (including brushes and
interpoles) of 0.04 W and the field resistance
of 100 W. Find (a) the value of resistance to be
added to the field circuit to increase the speed
from 1200 to 1600 rpm, when the supply
current is 200 A, (b) With the field resistance
as in (a), find the speed when the supply
current is 120 A. If the machine is run as a
generator to give 200 A at 220 V, find (c) the
field current at 1300 rpm, and (d) the speed
when the field current is 2 A.
7.14 Derive the standard torque equation of a dc
motor, from first principles. A 4-pole series
motor has 944 wave-connected armature
conductors. At a certain load the flux per pole
is 34.6 mWb and the total mechanical power
developed is 4 kW. Calculate the line current
taken by the motor and the speed at which it
will run with an applied voltage of 500 V. The
total motor resistance is 3 W.
436 Electric Machines
7.15 The following data pertain to 250 V dc series
motor:
P
=1
A
Flux/pole = 3.75 m Wb/field amp
Total armature circuit resistance = 1 W
The motor is coupled to centrifugal pump
whose load torque is
Z = 180,
TL = 10–4 n2 Nm
where n = speed in rpm.
Calculate the current drawn by the motor and
the speed at which it will run.
7.16 A dc shunt motor is being operated from 300 V
mains. Its no-load speed is 1200 rpm. When
fully loaded, it delivers a torque of 400 Nm
and its speed drops to 1100 rpm. Find its speed
and power output when delivering the same
torque; if operated with an armature voltage of
600 V. Excitation is assumed unchanged, i.e.
the motor field is still excited at 300 V. State
any assumption you are required to make.
7.17 A 50 kW, 230 V dc shunt motor has an armature resistance of 0.1 W and a field resistance of 200 W. It runs on no-load at a speed
of 1400 rpm, drawing a current of 10 A from
the mains.
When delivering a certain load, the motor
draws a current of 200 A from the mains.
Find the speed at which it will run at this load
and the torque developed. Assume that the
armature reaction causes a reduction in the
flux/pole of 4% of its no-load value.
7.18 A 250 V dc series motor has the following
OCC at 1200 rpm:
If (A)
5
10
VOC(V) 100 175
15
220
20 25
240 260
30
275
Ra = 0.3 W and series field resistance is 0.3 W.
Find the speed of the machine when (a) Ia =
25 A and (b) the developed torque is 40 Nm.
7.19 A 15 kW, 250 V, 1200 rpm shunt motor
has 4 poles, 4 parallel armature paths, and
900 armature conductors; Ra = 0.2 W. At rated
speed and rated output the armature current is
75 A and If = 1.5 A. Calculate (a) flux/pole, (b)
the torque developed, (c) rotational losses, (d)
h, (e) the shaft load, and (f ) if the shaft load
remains fixed, but the field flux is reduced to
70% of its value by field control, determine
the new operating speed.
7.20 A 115 kW, 600 V dc series wound railway
track motor has a combined, field and armature
resistance (including brushes) of 0.155 W. The
full-load current at rated voltage and speed is
216 A. The magnetization curve at 500 rpm is
as follows.
EMF (V) 375
If (A)
188
400
216
425
250
450
290
475
333
(a) Neglecting armature reaction, calculate
the speed in rpm at the rated current and
voltage.
(b) Calculate the full-load internal (developed)
torque.
(c) If the starting current is to be restricted to
290 A, calculate the external resistance
to be added in the motor circuit and the
starting torque.
7.21 A 100 kW, 600 V, 600 rpm dc series wound
railway motor has a combined field and
armature resistance (including brushes) of
0.155 W. The full-load current at the rated
voltage and speed is 206 A. The magnetization
curve at 400 rpm is as follows:
If (A)
188 206 216
EMF (V) 375 390 400
250
425
290 333
450 475
(a) Determine the armature reaction in
equivalent demagnetizing field current at
206 A.
(b) Calculate the internal (developed) torque
at the full-load current.
(c) Assuming demagnetizing armature reaction mmf proportional to (Ia2). determine
the internal starting torque at the starting
current of 350 A.
7.22 A 3 kW series motor runs normally at 800 rpm
on a 240 V supply, taking 16 A; the field coils
DC Machines
7.23
7.24
7.25
7.26
are all connected in series. Estimate the speed
and current taken by the motor if the coils are
reconnected in two parallel groups of two in
series. The load torque increases as the square
of the speed. Assume that the flux is directly
proportional to the current and ignore losses.
A 20 kW, 500 V shunt motor has an efficiency
of 90% at full load. The armature copperloss is 40% of the full-load loss. The field
resistance is 250 W. Calculate the resistance
values of a 4-section starter suitable for this
motor in the following two cases:
Case 1: Starting current £ 2If l
Case 2: Starting current (min) = 120% If l·
A starter is to be designed for a 10 kW,
250 V shunt motor. The armature resistance
is 0.15 W. This motor is to be started with
a resistance in the armature circuit so that
during the starting period the armature current
does not exceed 200% of the rated value or
fall below the rated value.
That is, the machine is to start with 200% of
armature current and as soon as the current
falls to the rated value, sufficient series
resistance is to be cut out to restore current to
200% (or less in the last step).
(a) Calculate the total resistance of the starter.
(b) Also calculate the resistance to be cut out
in each step in the starting operation.
A dc motor drives a 100 kW generator having
an efficiency of 87%.
(a) What should be the kW rating of the
motor?
(b) If the overall efficiency of the motor
generator set is 74%, what is the efficiency
of the motor?
(c) Also calculate the losses in each machine.
A 600 V dc motor drives a 60 kW load at
900 rpm. The shunt field resistance is 100 W
and the armature resistance is 0.16 W. If the
motor efficiency is 85%, determine:
(a) the speed at no-load and the speed
regulation.
(b) the rotational losses.
437
7.27 Enumerate the principal losses that occur in a
dc generator, and where appropriate, state the
general form of the physical law upon which
each loss depends. Calculate the efficiency
of a self-excited dc shunt generator from
the following data; Rating: 10 kW, 250 V,
1000 rpm.
Armature resistance = 0.35 W
Voltage drop at brushes = 2 V
Windage and friction losses = 150 W
Iron-loss at 250 V = 180 W
Open circuit characteristic:
EMF (V): 11
Field
current (A) 0
140
227 285
300 312
1.0
1.5
2.2
2.0
2.4
7.28 A 60 kW, 252 V shunt motor takes 16 A when
running light at 1440 rpm. The resistance of
armature and field is 0.2 and 125 W respectively
when hot. (a) Estimate the efficiency of the
motor when taking 152 A. (b) Also estimate
the efficiency if working as a generator and
delivering a load current of 152 A at 250 V.
7.29 A 200 V shunt motor takes 10 A when running
on no-load. At higher loads the brush drop is
2 V and at light loads it is negligible. The strayload loss at a line current of 100 A is 50% of
the no-load loss. Calculate the efficiency at
a line current of 100 A if armature and field
resistances are 0.2 and 100 W respectively.
7.30 Hopkinson’s test on two machines gave the
following results for full load; line voltage
230 V, line current excluding field current
50 A; motor armature current 380 A; field
currents 5 and 4.2 A. Calculate the efficiency
of each machine. The armature resistance of
each machine = 0.02 W. State the assumptions
made.
7.31 Calculate the efficiency of a 500 V shunt
motor when taking 700 A from the following
data taken when the motor was hot: motor
stationary, voltage drop in the armature
winding 15 V, when the armature current was
510 A; field current 9 A at normal voltage.
438 Electric Machines
Motor running at normal speed unloaded; the
armature current was 22.5 A, when the applied
voltage was 550 V; allow 2 V for brush contact
drop and 1 % of the rated output of 400 kW
for stray-load losses.
7.32 A 480 V, 20 kW shunt motor took 2.5 A, when
running light. Taking the armature resistance
to be 0.6 W, field resistance to be 800 W and
brush drop 2 V, find the full-load efficiency.
7.33 The open-circuit characteristic for a dc
generator at 1200 rpm is:
Field current (A) 0
Armature voltage (V)
13
2
4
6
8
140 230 285 320
Estimate the critical field resistance at this
speed. Keeping the field resistance fixed at
this speed, the generator is run at 1600 rpm.
What would be the no-load voltage?
The generator is now loaded to an armature
current of 500 A. Estimate the terminal
voltage and line current (at 1600 rpm). Neglect
armature reaction effects.
400 V, the resistance of the armature circuit
0.2 W and the speed 600 rpm. What will be
the speed and line current if the total torque
on the motor is reduced to 60% of its full-load
value and a resistance of 2 W is included in the
armature circuit, the field strength remaining
unaltered.
7.38 230 V dc is shunt motor has an armature
resistance of 0.25 W. What resistance must
be added in series with the armature circuit to
limit the starting current 90 A?
With this starting resistance in circuit, what
would be the back emf when the armature
current decreases to 30 A?
7.39 For the hoist drive system shown in Fig. P.7.39.
(a) Calculate the minimum size of the motor.
(b) If the line voltage drops to 180 V, what
would be the hoisting speed?
Rt = 350 W
230 V
Ra = 0.2 W
Given: Armature resistance = 0.05 W,
Brush voltage drop = 2 V
7.34 The open-circuit voltage of a dc shunt
generator is 130 V. When loaded the voltage
drops to 124 V. Determine the load current.
Given: Ra = 0.03 W, Rf = 20 W
7.35 A 220 V dc shunt motor runs at 1400 rpm on
no-load drawing an armature current of 2.4 A
from the supply. Calculate the motor speed for
an armature current of 60 A. Assume that the
flux/pole reduces by 4% armature reaction. It
is given that Ra = 0.24 W.
7.36 A dc shunt machine has an open-circuit voltage of 220 V at 1200 rpm. The armature resistance is 0.2 W and field resistance is 110 W.
Calculate its speed as a motor when drawing a line current of 60 A from 230 V mains.
Assume that armature reaction demagnetizes
the field to the extent of 5%.
7.37 The full-load current in the armature of a
shunt motor is 100 A, the line voltage being
Radius = 16 cm
2.5 m/s
750
kg
Fig. P.7.39
7.40 A 400 V dc shunt motor takes a current of
5.6 A on no-load and 68.3 A on full-load.
The load current weakens the field by 3%.
Calculate the ratio of full-load speed to noload speed. Given: Ra = 0.18 W, brush voltage
drop = 2 V, Rf = 200 W.
7.41 What resistance must be added in series with
the armature of the shunt motor of Prob. 7.40
to reduce the speed to 50% of its no-load value
with gross torque remaining constant. The
shunt field resistance also remains unchanged.
DC Machines
7.42 A 200 V dc series motor yielded the following
operational data:
Speed (rpm)
Current (A)
640
20
475
30
400
40
Find the speed at which the motor will
run when connected to 200 V mains with a
series resistance of 2 W while drawing 35 A.
Armature circuit resistance = 1.2 W.
7.43 A series motor takes 50 A at 400 V while
hoisting a load at 8 m/s. The total series
resistance of the armature circuit is 0.46 W.
What resistance must be added in series with
the armature circuit to slow the hoisting speed
down to 6 m/s? Assume magnetic linearity.
What other assumption you need to make.
7.44 The magnetization curve of a 4-pole dc series
motor with a 2-circuit winding was obtained
by separately exciting the field with currents
of the values given below, and loading the
armature to take the same current. the speed
being maintained constant at 800 rpm:
Current (A)
10
50
Voltage between 160
brushes (V)
460
20
60
295
485
30
70
375
505
40
80
425
525
Determine the speed and torque for this motor
for currents of 10 A, 50 A and 80 A for a
line voltage of 600 V, having been given that
the total number of conductors is 660, the
resistance of the armature circuit is 0.30 W
and that of the field coils is 0.25 W.
7.45 A dc series motor has the following
magnetization characteristic at 800 rpm:
Field current (A)
OC voltage (V)
0
120
40
910
40
160
390
1080
80
200
680
1220
Armature resistance = 0.2 W
Field resistance = 0.2 W
Brush voltage drop = 2 V (total)
The machine is connected to 750 V mains. If a
diverter of resistance 0.2 W is connected across
439
the field windings, calculate (a) the speeds,
and (b) the torques developed at armature
currents of 80 A and 120 A respectively.
7.46 The following test results were obtained while
Hopkinson’s test was performed on two similar
dc Shunt Machines: supply voltage is 250 V,
field current of generator and motor was 6 A
and 5 A respectively and the line current and
motor current were 50 A and 400 A. Calculate
the efficiency of motor and generator of each
having 0.0150 W armature resistance.
7.47 The Hopkinson’s test was performed at full
load on two similar shunt machines. The test
results are:
Line voltage = 110 V
Line current = 48 A,
Armative current = 230 A (motor)
Field currents = 3 A and 5 A
Armature resistance = 0.035 W (each)
Brush contact drop = 1 V per brush
Calculate the efficiency of both the machines.
7.48 At no load, a dc shunt motor takes 5 A from
400 V supply. Armature and field resistance
of that shunt motor is 0.5 W, 200 W. Find the
efficiency of the motor when it takes 60 A line
current.
7.49 The following test results are obtained while
Swinburne’s test is performed on dc shunt
motor with input voltage is 200 V. At no
load the input power is 1.1 kW, the current is
5.5 A, and the speed is 1150 rpm. Calculate
the efficiency at 50 A load. Given: armature
resistance is 0.6 W and shunt field resistance
is 110 W.
7.50 A 250 V dc shunt motor takes 5 A at no load.
The armature resistance including brush
contact resistance is 0.2 W, field resistance
is 250 W. Calculate the output and efficiency
when input current is 20 A.
7.51 Determine the minimum speed at which the
motor can hold the load by means of regenerative braking. The following parameters are:
440 Electric Machines
7.52
7.53
7.54
7.55
input voltage is 250 V, armature resistance is
0.1 W with full load condition. It has an emf
of 240 V at a speed of 1200 rpm. The motor
is driving an overhauling load with a torque of
180 Nm.
A series motor has an armature resistance of
0.7 W and field resistance of 0.3 W. It takes a
current of 15 A from a 200 V supply and runs
at 800 rpm. Find the speed at which it will run
when a 5 W resistance is added in series to
the motor and it takes the same current at the
supply voltage.
Find the efficiency of a long shunt compound
generator rated at 250 kW, 230 V when
supplying 76% of rated load at rated voltage.
The resistance of armature and series field are
0.009 W and 0.003 W. The shunt field current
is 13 A. When the machine is run as a motor
at no load the armature current is 25 A at rated
voltage?
A PMDC mator connected to 48 V supply
runs on no-load at 2000 rmp drawing 1.2 A. It
is found to have an armature resistance of 1.02
W.
(a) Calculate torque constant and no-load
relational loss
(b) It is now connected to 50 V supply and
runs at 2200 rpm.
Calculate its shaft power output and
electromagnetic torque
A reparality excited dc generator has OCC
data at 1200 rmp as below.
7.56
7.57
7.58
7.59
VOC 10 54 160 196 248 284 300 360
If (A) 0 1 2
3
4
5 6
7
At If = 6 A when loaded at Ia = 50 A, it has
a terminal voltage of 230 V at speed of 1200
rpm. Given: Ra = 0.5 W.
(a) Determine the armature voltage drop due
to the demagnetizing effect of armature
reaction
7.60
(b) Assuming that this voltage drop in
proportional to the square of armature
current, determine the field current for
a terminal voltage of 230 V at armature
current of Ia = 40 A
(c) In part (a) determine the field current
equivalent of demagnetization
A 260 V dc shunt mator has armature resistance
of Ra = 0.4 W and field resistance of Rf = 50 W.
It is drawing full load armature current of Ia =
50 A. Which has a demagnetizing field current
equivalent of 0.8 A. Calculate the motor speed.
The mator’s OCC data at 1200 rpm is as given
in Problem 7.55.
Design a suitable double-layer lap winding
for a 6-pole armature with 18 slots and two
coil-sides per slot. Give the values of frontpitch, back-pitch and commutator pitch. Draw
the developed diagram and show the positions
of brushes.
Design a suitable double-layer wave winding
for a 4-pole, 13-slot dc armature with two
coil-sides per slot. Give values of pitches,
Draw the developed diagram and show the
positions of brushes.
A 6-pole, dc armature with 36 slot and 2 coilsides per slot is to be wave-wound (doublelayer). Calculated the back-pitch, front-pitch
and commutator-pitch. How many dummy
coils, if any, are required? Draw the developed
diagram of the winding and show the locations
of brushes and the distance between them in
terms of commutator segments. If
(a) 2 brushes are used, and
(b) 6 brushes are used.
A 25 kW dc generator has a 6-pole lapconnected armature of 312 conductors. How
would you change the connections to the
commutator to form two armature circuits
and what effect would this change have on the
voltage, current and output of the machine?
DC Machines
1. What is the effect of magnetic saturation
on the external characteristics of a dc shunt
generator?
2. Discuss that emf and torque of a dc machine
depend on the flux/pole but are independent of
the flux density distribution under the pole.
3. Write the expressions for the induced emf
and torque of a dc machine using standard
symbol. What is the machine constant? What
is the value of the constant relating wm and n?
4. Explain the meaning and significance of the
critical field resistance of a shunt generator.
5. Each commutator segment is connected to
how many coil ends?
6. What are interpoles, their purpose, location
and excitation? Explain why of each item.
7. Compare the number of parallel paths in the
lap and wave windings.
8. State the condition which determines if a dc
machine is generating or motoring.
9. Write the expression relating the electrical
power converted to the mechanical form in a
dc motor. How are the electrical power input
and mechanical power output different from
these powers?
10. What is OCC and what information does it
reveal about a dc machine? At what speed is it
determined? What is the air-gap line?
11. Write the basic proportionality relationships
of a dc machine. What form these take for
linear magnetization?
12. State the types of dc motors. What is the basis
of the classification?
13. Based on emf and torque equation compare
and contrast the two methods of speed control
of dc motor.
14. How can one choose between a short shunt
and long shunt cumulatively compound dc
motor?
15. List the factors involved in voltage build-up in
a shunt generator.
441
16. Explain how the back emf of a motor causes
the development of mechanical power.
17. Sketch the speed-torque characteristic of a
shunt motor at fixed field current. Explain the
nature of the characteristic through relevant
fundamental relationships of the dc machine.
18. Sketch the speed-torque characteristic of a
dc series motor and advance the underlying
reasoning for the nature of the characteristic
based on fundamental relationships of the dc
machine.
19. Explain through sketch and derivations the
speed-torque characteristic of a differentially
compound dc motor.
20. Advance the methods of varying the shunt
field and the series field excitation of a dc
machine.
21. Discuss the method of speed control of a dc
series motor.
22. How is a shunt motor started? Why it should
not be started direct on line?
23. Why do we need a compensating winding to
nearly over come the armature reaction and
how is this winding excited and why?
24. Enumerate and classify the losses in a dc
shunt motor.
25. How to determine the load current of a dc
shunt motor at which the motor efficiency is
maximum?
26. Explain the different methods of braking of dc
motors.
27. What are the advantages of Hopkinson’s test
over Swinburne’s test and what are its limits?
28. By what test on dc reparalely excited generator
would you determine the armature reaction
equivalent field current at rated armature
current.
29. Assuming magnetic linearity derive the
expression for speed-torque characteristic of
a dc series motor using suitable symbols.
30. Compare the speed troque characteristices
of a series and cummulative compound
442 Electric Machines
motor. Why does the compound motor have a
defenite no-load speed?
31. Sketch the external characteristic of a shunt
generator and explain the reason for its special
nature: part of it is two-valued.
32. What is te significance of a winding diagram?
33. When do you use concentric winding?
34. What are the advantages of fractional slot
winding over integral slot winding?
35. Compare lap and wave winding. Where each
type is used and why?
36. Why double layer winding is preferred?
37. Explain how fractional slot winding reduces
the emfs of ripple frequencies.
7.1 Why is the armature of a dc machine made of
silicon steel stampings?
(a) To reduce hysteresis loss
(b) To reduce eddy current loss
(c) For the ease with which the slots can be
created
(d) To achieve high permeability.
7.2 Slot wedges in a dc machine are made of
(a) mild steel
(b) silicon steel
7.3 The armature reaction AT in a dc machine
(a) are in the same direction as the main poles
(b) are in direct opposition to the main poles
(c) make an angle of 90° with the main pole
axis
(d) make an angle with the main pole axis
which is load dependent.
7.4 A dc series motor has linear magnetization
and negligible armature resistance. The motor
speed is
7.6 What losses occur in the teeth of a dc machine
armature?
(a) Hysteresis loss only
(b) Eddy current loss only
(c) Both hysteresis and Eddy current loss
(d) No losses
7.7 The process of current commutation in a dc
machine is opposed by the
(a) emf induced in the commutating coil
because of the inter-pole flux
(b) reactance emf
(c) coil resistance
(d) brush resistance
7.8 In a level compound generator the terminal
voltage at half full-load is
(a) the same as the full load voltage
(b) the same as no load voltage
(c) is more than the no-load voltage
(d) is less than the no-load voltage
7.9 Field control of a dc shunt motor gives
(a) constant torque drive
(b) constant kW drive
(c) constant speed drive
(d) variable load speed drive
7.10 In a series-parallel field control of a dc series
motor with fixed armature current
(a) such connections are not used in practice
(b) both series and parallel connections give
the same speed
(c) series connection gives higher speed
(d) parallel connection gives higher speed
(a) directly proportional to
T;
T = load torque
(b) inversely proportion a to T
(c) directly proportional to T
(d) inversely proportional to T
7.5 The armature reaction mmf in a dc machine is
(a) sinusoidal in shape
(b) trapezoidal in shape
(c) rectangular in shape
(d) triangular in shape.
DC Machines
7.11 If the magnetic circuit of dc machine is in
saturation region, the armature reaction
(a) does not affect the flux/pole
(b) increases flux/pole.
(c) decreases flux/pole
(d) affects the flux/pole only when armature
current in very small.
7.12 In Hopkinson’s test
(a) iron loss in both machines are equal
(b) iron loss in motoring machine is more
than in generating machine
443
(c) iron loss in generating machine is more
than in motoring machine
(d) only stray load iron loss is equal in both
machine.
7.13 In a dc motor electromagnetic torque is
(a) wm/EaIa, in the direction of wm
(b) wm/EaIa, opposite to the direction of wm
(c) EaIa/wm, opposite to the direction of wm
(d) EaIa/wm, in the direction of wm.
444 Electric Machines
8
8.1
INTRODUCTION
A synchronous machine is one of the important types of electric machines; in fact
all generating machines at power stations are of synchronous kind and are known
as synchronous generators or alternators.
The structure and certain operational features of the synchronous machine have
already been explained in Ch. 5, while the 3-phase ac windings used in the stator of the synchronous
machine have been further elaborated in Ch. 6. In this chapter the basic model used for performance
analysis of the synchronous machine and some of the operational features peculiar to this type of machine
will be discussed. The methods of obtaining model parameters and active and reactive power transfer
characteristics will also be discussed at length.
It has been seen in Ch. 5 that essentially two types of constructions are employed in synchronous
machines–one of these is known as the cylindrical-rotor type and the other is called the salient-pole type.
The cylindrical-rotor machine has its rotor in cylindrical form with dc field windings (distributed type)
embedded in the rotor slots. This type of construction provides greater mechanical strength and permits
more accurate dynamic balancing. It is particularly adopted for use in high-speed turbo-generators wherein
a relatively long but small diameter rotor is used to limit the centrifugal forces developed. Two or at most
4-pole machines use this type of construction.
The second type of synchronous machine, known as the salient-pole type, has its rotor poles projecting
out from the rotor core. This type of construction is used for low-speed hydroelectric generators and the
large number of poles necessary are accommodated in projecting form on a rotor of large diameter but
small length. This construction is almost universally adopted for synchronous motors.
Because of the distinguishing constructional features explained above, the cylindrical-rotor machine has
uniform air-gap, so that the permeance offered to the mmf acting on the magnetic circuit is independent
of the angle between the axis of the mmf and that of the rotor poles. On the other hand, in the salient-pole
construction, the permeance offered to the mmf varies considerably with the angle between the mmf axis
and that of the rotor poles. It is on this account that the cylindrical-rotor machine is simpler to model
and analyse, compared to the salient-pole type. The modelling and analysis of both these types will be
presented in this chapter.
It has been seen in Ch. 5 that the essential feature that distinguishes the synchronous machine from the
other types of electric machines is the synchronous link between the rotor and stator rotating fields. As a
Synchronous Machines
445
result there is a fixed relationship between the rotor speed and the frequency of emfs and currents on the
stator, which is reproduced below:
Frequency f =
where
Pns
Hz
120
(8.1)
P = number of poles
ns = speed of rotor in rpm (called synchronous speed)
ws = 2pf = synchronous speed in rad/s.
A synchronous generator when supplying isolated load acts as a voltage source whose frequency is
determined by its primemover speed as per Eq. (8.1). Synchronous generators are usually run in parallel
connected through long distance transmission lines. The system (called power system) is so designed as
to maintain synchronism in spite of electrical or mechanical disturbances. Such an interconnected system
offers the advantages of continuity of supply and economies in capital and operating costs. Synchronous
motors find use in industry wherever constant-speed operation is desired. Another advantage of the
synchronous motor is that its power factor can be controlled simply by variation of its field current. This is
the reason why in most large industrial installations a part of the load is usually handled by synchronous
motors which are operated at a leading power factor so as to yield an overall high power factor for the
complete installation.
8.2
BASIC SYNCHRONOUS MACHINE MODEL
As stated above the cylindrical-rotor synchronous machine offers constant permeance to mmf waves
irrespective of the mechanical position of the rotor and is, therefore, simpler to model. Figure 8.1 shows the
cross-sectional view of a 2-pole* cylindrical-rotor synchronous machine.
Positive direction of emf
(conductor ‘a’ under
influence of N-pole)
ws
a
Axis of field (at the
instant considered)
Æ
Ff
N
a = wst
ws
Axis of coil aa¢
(phase ‘a’)
S
Positive direction of emf
(conductor ‘a’ under
influence of S-pole)
Stator
a¢
Rotor
Fig. 8.1 Cylindrical-rotor synchronous machine
The rotor has distributed windings which produce an approximately sinusoidally distributed mmf wave
in space rotating at synchronous speed ws rad (elect.)/s (ns rpm) along with the rotor. This mmf wave is
represented by the space vector F f in the diagram and which at the instant shown makes an angle a = w st
* Multipolar structure is merely a cyclic repetition of the 2-pole structure in terms of the electrical angle.
446 Electric Machines
with the axis of coil aa¢ on the stator (coil aa¢ represents the phase a). The peak value of the vector F f
is Ff. As a consequence of the uniform air-gap, the mmf wave Ff produces sinusoidally distributed flux
density wave Bf , in space phase with it. Figure 8.2 shows the developed diagram depicting the space phase
relationship between Bf and Ff waves. As the rotor rotates (at synchronous speed ws), the Bf wave causes
sinusoidally varying flux f to link with the coil aa¢. The maximum value of this flux is Ff , the flux per pole.
Considering the time reference when F f lies along the axis of coil aa¢, this is a cosinusoidal variation, i.e.
f = F f cos ws t; ws = 2p f
(8.2)
Axis of field
Axis of coil aa¢(phase ‘a’)
MMF wave, Ff
a = w st
Flux density
wave, Bf
ws
ws
a
a¢
N
a
Stator
Rotor
S
¢ of stator
Fig. 8.2
It is, therefore, seen that the flux linking the coil aa¢ is a sinusoidal time variation and can be represented
as the time phasor F f . It will be referred to as flux phasor.
Consider now the space vector F f as seen from the axis of coil aa¢ on the stator. As F f rotates at
synchronous speed, it appears to be sinusoidally time-varying at ws = 2pf elect. rad/s as is evident from
the developed diagram of Fig. 8.2. Furthermore, when the maximum positive value of Ff space wave is
directed along the axis of coil aa¢, the flux linkage of
Axis of field
the coil has maximum positive value. It may, therefore,
w
s
be considered that the rotating space vector F f as
Ff
seen from the stator is a time phasor F f which is in
phase with the flux phasor F f as shown in Fig. 8.3.
Ff
The magnitude relationship between F f and F f will
be governed by the magnetization curve; this will be
linear if the iron is assumed to be infinitely permeable
in which case
F f = P Ff
where P = permeance per pole (see Eq. (5.59)).
The emf induced in the coil aa¢ of N turns is given
by the Faraday’s law,
eaf = – N
dl
, l = flux linkage of one coil
dt
90°
Axis of phase ‘a’
Ef
Fig. 8.3
Synchronous Machines
d
(Ff cos w s t)
dt
= Nws Ff sin wst
447
= –N
(8.3)
The positive direction of the emf is indicated on the coil aa¢. This is also verified by the flux cutting rule
when conductor a lies under the influence of N-pole of the rotor and conductor a¢ simultaneously lies under
the influence of the S-pole. It immediately follows from Eqs (8.2) and (8.3) that the emf eaf represented as
time phasor E f lags behind the mmf phasor F f and the flux phasor F f by 90° as shown in Fig. 8.3. This
figure also shows the relative location of the field axis and the axis of phase a wherein
the axis of phase a is
90° behind the rotor field axis. This is indicative of the fact that the field vector F f lies 90° ahead of the axis
of coil aa¢ in Fig. 8.1 when the emf in coil aa¢ has maximum positive value (projection of the phasor E f on
the axis of phase a). The rotor field axis is known as the direct-axis and the axis at 90° elect. from it is known
as the quadrature-axis.
It immediately follows from Eq. (8.3) that the rms value of the emf induced in coil aa¢ is
Ef =
wherein
2 pf NF f
Ff = Ff (Ff)
or
(8.4)
Ff (If)
flux/pole
(8.5)
If being the direct current in the rotor field. Equation (8.5) between the flux/pole and field current is indeed
the magnetization characteristic. Equation (8.4) suitably modified for a distributed (and also possibly shortpitched) stator winding is
Ef =
2 p Kw f Nph F f
(see Eq. (5.20))
(8.6)
It easily follows that the emfs produced in the other phases of the stator would progressively differ in time
phase by 120°.
The conclusions drawn from the above discussion are indeed general and are reproduced below.
Whenever the magnetic structure of a cylindrical rotor synchronous machine is subjected to rotating mmf
vector, it is seen as an mmf phasor from the stator with its flux phasor in phase with it, while the phasor
representing the phase emf induced lags behind both these phasors by 90° (see Fig. 8.3).
It was shown in Sec. 5.5 that when a 3-phase stator supplies a balanced load, it sets up its own mmf vector
Far (sinusoidally distributed in space), called the armature reaction, rotating at synchronous speed in the
same direction as the rotor. The magnetic circuit is now subjected to two rotating mmf vectors F f and Far ,
both rotating at synchronous speed with a certain angle between them. The objective here is to establish what
determines this angle.
To begin with, it is observed that in the generating operation of the machine, the emf and current have the
same positive direction. Consider the case when the current Ia supplied by the coil aa¢ is in phase with the
coil emf Ef . It means that at the time instant when the emf is maximum positive in the coil aa¢, its current
also
has maximum positive value. The emf in coil aa¢ will be maximum positive when the field mmf vector
F f is 90° ahead of the coil axis (in the direction of rotation) as shown in Fig. 8.4(a). Simultaneously, Far is
directed along* the axis of coil aa¢ which has maximum positive current at this instant. It is therefore, seen
* It was shown in Sec. 5.5 that the mmf vector of stator carrying balanced 3-phase currents is directed along the axis
of coil aa¢ when phase a carries maximum positive current.
448 Electric Machines
from Fig. 8.4(a) that F f is 90° ahead of Far when E f and I a are in phase. The resultant mmf vector Fr is
the vector sum
Fr = F f + Far
(8.7)
as shown in Fig. 8.4(a). It is observed that Fr lags F f by angle d. The corresponding phasor diagram,
drawn as per the general conclusions stated earlier, is shown in Fig. 8.4(b) wherein the phasor equation
corresponding to the vector Eq. (8.7) is
Fr = F f + Far ( Fr lags F f by angle d)
Field axis
Field axis
Tem torque
Æ
Far
ws
Æ
Æ
Ff
Fr
a
(8.8)
Ia (max positive)
d
ws
T
Ff
–
Fr
Far
Ff
Fr
Ia (max positive)
d
Axis of coil aa¢
(phase ‘a’)
Far
d
Ia
Ef
Axis of
phase ‘a’
a¢
Er
(a) MMF vector diagram
(b) Phasor diagram
Fig. 8.4 Synchronous machine on load (generating action), Ia in phase with E f
It is observed from the phasor diagram that I a and Far are in phase; this is logical* because Far is
produced by I a and is proportional to it. The resultant mmf vector Fr produces the resultant air-gap flux
phasor Fr which in turn induces the emf Er in phase a lagging 90∞ behind the phasor Fr . The phase emf
Er induced in the machine, called air-gap
emf, lags by angle d behind E f (emf induced by F f acting alone)
which is the same angle by which Fr lags behind F f in the vector diagram of Fig. 8.4(a). E f is known as
the excitation emf.
In case I a lags E f by an angle y, the
positive current maximum in the coil aa¢ will occur angle y later,
so that F f now lies (90° + y) ahead of Far as shown in the vector diagram of Fig. 8.5(a). The corresponding
phasor diagram is shown in Fig. 8.5(b) wherein F f leads Far by (90° + y). The phase angle between Er
and I a indicated by q is the power factor angle provided it is assumed that the armature has zero resistance
and leakage reactance so that the machine terminal voltage
Vt = Er
It is seen from Figs 8.5(a) and (b) that the field poles lie an angle d ahead of the resultant mmf (or
resultant flux) wave. The electromagnetic torque developed in the machine tries to align the field poles with
the resultant field and is, therefore, in a direction as shown in Fig. 8.5(a) as well as in Fig. 8.5(b). It is
immediately seen that the torque on the field poles is in opposite direction to that of rotation which means
* An observer on stator from the axis of phase a observes simultaneous occurrence of maximum positive value of
Far and Ia.
Synchronous Machines
449
that mechanical power is absorbed by the machine. This is consistent with the assumed condition of the
generating action (positive current in the direction of positive emf ). It may, therefore, be concluded that in
generating action, the field poles are driven ahead of the resultant flux wave by an angle d as a consequence
of the forward acting torque of the primemover. Also, the field poles are dragged behind by the resultant flux
from which results the conversion of mechanical energy into electrical form.
Field axis
Field axis
T
Æ
Far
ws
ws
Æ
Ff
Ff
T
Ff
Far
Ff
a
Ia (max
positive)
d
d
Axis of coil aa¢
(phase ‘a’)
y
yd
Far
a¢
Fr
Fr
Ef
q
Ia
(a) Vector diagram
Axis of
phase ‘a’
Er
(b) Phasor diagram
Fig. 8.5 Synchronous machine on load (generating action), Ia lags E f by angle y
As already shown in Sec. 5.6 Eq. (5.58), if the magnetic circuit is assumed linear, the magnitude of the
torque is given by
2
T=
p Ê Pˆ
Fr Ff sin d
2 ÁË 2 ˜¯
(8.9)
where Fr is the resultant flux/pole, Ff the peak value of field ampere-turns and d is the angle by which F f
leads Fr . It is also observed from Fig. 8.5(b) that E f also leads Er by the same angle d.
When Ff and Fr are held constant in magnitude, the machine meets the changing requirements of load
torque by adjustment of the angle d which is known as the torque (power*) angle.
The torque expression of Eq. (8.9) can also be written in terms of voltages as
T = KEr Ef sin d
(8.10)
where Er = emf induced in the machine under loaded condition; called air-gap emf.
Ef = emf induced by the field mmf Ff acting alone, i.e. the machine is on no-load with same Ff (or
rotor field current) as on-load; called excitation emf .
In motoring action of the synchronous machine, the positive current flows opposite to the induced emf.
Since the phasor diagrams above have been drawn with the convention of generating current (i.e. current in
the direction of emf ), the armature reaction phasor Far will now be located by phase reversing the motoring
current for consistency of convention. Accordingly the phasor diagram for motoring action is drawn in
Fig. 8.6. It is immediately observed from this figure that F f and E f now lag Fr and Er respectively by
angle d. The torque of electromagnetic origin therefore acts on the field poles in the direction of rotation so
that the mechanical power is output meaning thereby motoring action.
* Torque and power are proportional to each other as the machine runs at synchronous speed (ws) under steady
operating conditions.
450 Electric Machines
If the terminal voltage Vt = Er and its frequency
Field axis
T
is held constant by an external 3-phase source, called
Fr
ws
infinite bus, the machine operates as a generator
(Fig. 8.5) or as a motor (Fig. 8.6) depending upon the
Far F
Fr
f
mechanical conditions at a shaft.
Er
F
Figure 8.3 is representative of no-load conditions
f
d
when the machine is said to be floating on busbars with –I–
a
zero stator current and the rotor being run at synchronous
Axis of
d q
Far
y
phase ‘a’
speed by external means (prime mover). If mechanical
Ef
power from the primemover is now increased, the field
Ia (motoring)
poles (rotor) move ahead causing current to be fed into
Fig. 8.6 Synchronous machine on load (motoring
the bus-bars. Under steady condition, the field poles lie
action)
ahead of the resultant flux wave by angle d (Figs 8.4 and
8.5) creating electromagnetic torque in opposition to the direction of rotor rotation; the value of d corresponds
to the balance of torques (or power, P = Tws). The electrical power output is
3Vt (= Er)Ia cos q
(for 3 phases)
which balances the mechanical power input from the primemover because there are no losses in the stator
(resistance is assumed to be zero). Here Ia is the generating current taken to be positive in the direction of the
machine’s induced emf Er.
If instead, the shaft is mechanically loaded from the no-load condition (Fig. 8.3), the field poles lag behind
the resultant flux wave as in Fig. 8.6 creating electromagnetic torque in the direction of rotation thereby
outputting mechanical power; the electrical input power being 3 Vt (= Er)Ia cos q, where Ia is the motoring
current taken as positive in opposition to the positive direction of Er.
The torque (power)-angle (T – d ) relationship
T, P
of Eq. (8.10) for fixed Ef and Er (= Vt) is drawn
in Fig. 8.7 with d taken as positive for generating
action and negative for motoring action. The
Tpull-out
Generating
operating points as the generator and motor are
indicated by g and m on this curve corresponding
to the specific condition of loading. The
characteristic exhibits, both in generating and
– 180°
– 90° dm
motoring operation, a maximum torque at d =
d
90°, called the pull-out torque, beyond which
dg 90°
180°
the synchronous link between field poles and the
resultant flux wave is severed and the machine
m
falls out-of-step (or loses synchronism). The
Tpull-out
average developed machine torque now becomes
Motoring
zero and so the average electric power fed by the
generating machine to the bus-bars, (infinite) or
Fig. 8.7 T-d characteristic of synchronous machine
average electric power drawn by the motoring
machine from the bus bars reduces to zero. The generating machine thus accelerates and so overspeeds
(primemover power is assumed to remain constant) and the motoring machine decelerates and comes
to stop.
Synchronous Machines
451
Realistic Machine
A realistic synchronous machine will have resistance Ra and leakage reactance Xl per armature phase which
can be assumed to be lumped in series between the terminal voltage Vt and the air-gap emf Er for each
machine phase. The circuit diagram of the machine on a per phase basis is drawn in Fig. 8.8(a) for the
generating mode (armature current in same direction as Er). From Fig. 8.8(a)
Vt = Er - I a (Ra + jXl) (generating mode)
(8.11)
Vt = Er + I a (Ra + jXl) (motoring mode)
(8.12)
and from Fig. 8.8(b)
The resistance of a synchronous machine armature is usually very small and can generally be neglected in
performance analysis (see Sec. 8.3).
Ra
If
XI
Ra
If
Ia
XI
Ia
+
+
+
+
Vt
Er
Er
–
Vt
–
–
–
Field
Field
(a) Generating mode
(b) Motoring mode
Fig. 8.8
Voltage Regulation
The voltage regulation of a synchronous generator is defined on lines similar to that of a transformer. Consider
the generator supplying full-load current at a specified power factor and rated terminal voltage, Vt (rated). As
the load is thrown off keeping the field current constant, the terminal voltage rises to
Vt (no-load)
I f kept constant
as at full - load
specified power factor
= Ef, the excitation emf
The percentage voltage regulation is then defined as
E f - Vt (rated )
Vt (rated )
¥ 100%
(8.13)
At specified power factor
8.3 CIRCUIT MODEL OF SYNCHRONOUS MACHINE
By assuming linearity of the magnetic circuit, it is possible to obtain simple circuit model of the synchronous
machine. The validity of this assumption stems from the fact that air-gap is the predominant component of the
magnetic circuit of the machine. Approximate nonlinear analysis is the subject matter of Sec. 8.4.
As per Eq. (8.7), the resultant mmf phasor is given by
Fr = F f + Far
(8.14)
452 Electric Machines
The resultant flux Fr and the air-gap emf Er must in general be obtained from Fr . However, the assumption
of linear magnetic circuit (F = P F; P is constant permeance*), allows one to find the resultant flux by the
principle of superposition as
Fr = F f + Far
(8.15)
where F f = flux component produced by F f acting alone
Far = flux component produced by Far acting alone.
Since emf induced is proportional to the flux/pole (Eq. (8.6)) and lags behind it by 90°, from the flux
phasor Eq. (8.15), the emf phasor equation can be written as
Er = E f + Ear
where
(8.16)
E f = emf induced by F f acting alone (excitation emf )
Ear = emf induced by Far acting alone.
The emf phasors in Eq. (8.16) are proportional to the
corresponding flux phasors of Eq. (8.15) with the emf
phasors lagging the respective flux phasors by 90°. The
phasor Eqs (8.15) and (8.16) are represented by the phasor
diagram of Fig. 8.9. In this figure the flux phasor triangle
and emf phasor triangle are similar to each other with the
emf phasor triangle being rotated anticlockwise from the
flux phasor triangle by 90°.
As Far is in phase with I a (generating machine) and is
proportional to it, the emf Ear is proportional** to I a and
lags behind Far by 90°, i.e.
Ear
= – j I a Xar
Ff
Far
Ef
Fr
d
Ear = – jIaXar
d
Er
Ia
Fig. 8.9
(8.17)
components in synchronous machine
where Xar’ the constant of proportionality, is indeed an inductive reactance. Thus Eq. (8.16) can be written as
Er = E f – j I a Xar
(8.18)
Corresponding to Eq. (8.18), Fig. 8.10(a) gives the per phase circuit model of the synchronous machine.
Comparing Eqs (8.15), (8.16) and (8.18), it is concluded that the reactance Xar equivalently replaces the
effect of the armature reaction flux. If Xar is known for a machine, one can work in terms of voltages and
currents and need not represent fluxes on the phasor diagram.
* In cylindrical-rotor machine being considered here, P is independent of the direction along which F is directed
because of a uniform air-gap.
Ear = – jKa Far; Ka = emf constant of armature winding
**
= – jKa P Far ; P = permeance/pole
= – jKa P Kar I a ; Kar = constant of armature winding (see Eq. (5.44b))
= – jXar I a
where
Xar = KaP Kar
Xar will be more for a machine with higher permeance, i.e., smaller air-gap,
Synchronous Machines
453
The effect of armature resistance Ra and leakage reactance Xl are embodied in the phasor equation as (see
Eq. (8.11))
Vt = Er - I a (Ra + jXl)
(8.19)
From Eq. (8.19) and Fig. 8.10(a) follows the complete circuit model of Fig. 8.10(b) which can be reduced
to the simpler form of Fig. 8.10(c) by combining series reactances and by allowing the identity of Er to be
lost. In the circuit model of Fig. 8.10(c), the total reactance
Xs = Xar + Xl (per phase)
(8.20)
is known as the synchronous reactance of the machine, while
Zs =
Ra2 + X s2
(8.21)
is the synchronous impedance of the machine.
Xar
Ia
+
+
Ef
Er
–
–
(a)
Xar
Ra
Xl
Xs
Ia
Ra
Ia
+
+
+
Ef
Vt
Er
–
+
Ef
–
Vt
–
(b)
–
(c)
Fig. 8.10 Circuit model of synchronous machine
The synchronous reactance takes into account the flux produced by the flow of balanced 3-phase currents
in the stator as well as the leakage flux. The excitation emf, Ef , accounts for the flux produced by the rotor
field (dc excited). The magnitude of the excitation emf can be controlled by the dc field current (If) called
the excitation current. If the load on the machine is thrown off, Ef appears at the machine terminals which
then is the open-circuit voltage of the machine. With reference to Fig. 8.10(c) Ef is also called voltage behind
synchronous impedance or reactance (as Ra can be neglected).
It must be remembered here that the synchronous impedance model of the synchronous machine is based
on the linearity assumption and will hold for the unsaturated region of machine operation and is valid only
for the cylindrical-rotor machine.
In the above method, we have converted mmfs to emfs based on magnetic linearity assumption this method
is also known as the emf method.
454
Electric Machines
Range of Synchronous Impedance
Expressed in the pu system, the synchronous reactance of synchronous machines lies in a narrow range of
values. From practical data, it is observed that the armature resistance (Ra) is usually of the order of 0.01 pu,
i.e. the voltage drop in the armature resistance at the rated armature current is about 1% of the rated voltage.
The leakage reactance value ranges from 0.1 to 0.2 pu and the synchronous reactance (Xs = Xar + Xl) is of
the order of 1.0 to 2 pu. It is, therefore, seen that the armature resistance of a synchronous machine is so
low that it can be neglected for all practical purposes except in the computation of losses, temperature rise
and efficiency. It may be noted here that Ra must be small in order to minimize the I 2R loss and limit the
temperature rise of the machine, and Xs should be large to limit the maximum current that may flow under
fault (short-circuit) conditions. However, the modern practice is to design* synchronous machines with a
medium range of values for synchronous reactance as quick-acting** circuit breakers are now available to
disconnect a machine from the faulted line.
8.4
DETERMINATION OF THE SYNCHRONOUS REACTANCE
With the assumption of a linear magnetic circuit, the circuit model (per phase) of a synchronous machine is
as given in Fig. 8.10(c). If Ra is neglected, it then follows that
E f = Vt + j I a Xs
(8.22)
It is immediately seen from Eq. (8.22) that for a given field current under short-circuit condition (Ia = ISC,
Vt = 0),
Xs =
Ef
(8.23)
I SC
But Ef = VOC (open-circuit voltage, i.e. Ia = 0 with the same field current). Then with the linearity assumption
Xs =
VOC
I SC
(8.24)
I f = const
where VOC = open circuit voltage and ISC = short-circuit current on a per phase basis with the same field
current.
Since the magnetization characteristic of the machine is nonlinear, it is necessary to determine the
complete open-circuit characteristic (OCC) of the machine (VOC – If relationship). However, it will soon
be shown that it is sufficient to determine one point on the short-circuit characteristic (SCC) of the machine
(ISC – If relationship). as it is linear in the range of interest (for ISC up to 150% of the rated current).
Open-circuit Characteristic (OCC)
In this test the machine is run by a primemover at synchronous speed ns to generate voltage at the rated
frequency, while the armature terminals are open-circuited as in Fig. 8.11 with switch S open. The readings
of the open-circuit line-to-line armature voltage, VOC = 3 Ef , are taken for various values of If , the rotor
field current. It may be noted that If is representative of the net mmf/pole acting on the magnetic circuit of
* The designer can achieve the desired value of synchronous reactance by adjusting air-gap of the machine (see
footnote p. 450).
** The typical circuit breaker opening time is 1-1.5 cycles.
Synchronous Machines
455
Switch S
If
A
A
V
N
A
A
ns
Fig. 8.11 Schematic diagram for open-circuit and short-circuit tests
the machine. These data are plotted as OCC in Fig. 8.12 which indeed is the magnetization characteristic, i.e.
the relation between the space fundamental component of the air-gap flux and the net mmf/pole acting on the
magnetic circuit (space harmonics are assumed negligible).
ISC
VOC (line)
Air-gap line
OCC
SCC
Vt (rated)
Xs (unsaturated)
Ia (rated)
Xs (adjusted)
Isc (If = Of ¢)
VOC / 3
ISC
O
f¢ If
f¢¢
O¢
Fig. 8.12 Open-circuit and short-circuit characteristics
The OCC exhibits the saturation phenomenon of the iron in machine. At low values of If when iron is in
the unsaturated state, the OCC is almost linear and the mmf applied is mainly consumed in establishing flux
in the air-gap, the reluctance of the iron path being almost negligible. The straight-line part of the OCC, if
extended as shown dotted in Fig. 8.12, is called the air-gap line and would indeed be the OCC if iron did not
get saturated.
Open Circuit Loss
The loss in the open-circuit method conducted as above comprises no load (OC) core loss and windage and
friction loss. The power corresponding to these losses is drawn from the prime-mover running the machine,
which can be measured by a dynmometer or torque meter. The plot of POC versus field current is shown in
456 Electric Machines
POC = OC loss
Fig. 8.13. The winding and friction loss being constant gets separated out by reducing the field current to
zero. The OC core loss arises from hysteresis and eddy-current losses which very as 1.6th power of open
circuit voltage which is shown in the plot of Fig. 8.13(b).
OC (core) loss
OC core loss
Windage and friction loss
Open-circuit voltage
Field current
(b)
(a)
Fig. 8.13
Short-circuit Characteristic (SCC)
The short-circuit characteristic of the machine is obtained by means of the short-circuit test conducted as
per the schematic circuit diagram of Fig. 8.11 with switch S closed. While the rotor is run at synchronous
speed ns, the rotor field is kept unexcited to begin with. The field excitation is then gradually increased till
the armature current equals about 150% of its rated value. While the current in all the three ammeters should
be identical, practically a small unbalance will always be found on account of winding and field current
dissymmetries which cannot be completely avoided in a machine. Therefore ISC the short-circuit current per
phase is taken as the average value of the three ammeter readings.
It is to be noted that the machine must not be short-circuited under excited conditions with a near about
rated voltage. This can give rise to intolerably large transient currents in the machine.
The circuit model of the machine under short-circuit conditions is given in Fig. 8.14(a) and the corresponding
phasor diagram in Fig. 8.14(b) wherein Ra = 0. Since the armature circuit is assumed purely inductive, the
Ff
Ff
Xs
Xar
XI
Ia(SC)
Ia (SC) XI
Fr
+
Ef
–
Fr
Vt= 0
Er
Er
Ia (SC) Xs
Far
Ia (SC)
(a) Circuit model
Fig. 8.14 Short-circuit test
(b) Phasor diagram
Synchronous Machines
457
short-circuit current lags the air-gap voltage Er by 90° so that the armature reaction mmf phasor Far is in
direct opposition to F f , i.e, the armature reaction is fully demagnetizing in effect.
The air-gap voltage needed to circulate the short-circuit current in the armature is given by
Er = Ia (SC) Xl
(8.25)
As Xl is about 0.1 to 0.2 pu (while Xs may be as high as 1.0 pu), Er is very small even when Ef has a value
close to rated voltage of the machine. This implies that under the short-circuit condition with the armature
current as high as 150% of the rated value, and resultant air-gap flux is small and so the machine is operating
under the unsaturated magnetization condition, so that the SCC (ISC versus If) is linear and therefore only one
short-circuit reading is necessary for the complete determination of the SCC as shown in Fig. 8.12.
Since under the short-circuit condition the machine is highly underexcited, the losses as drawn in from
the mechanical shaft drive comprise mechanical loss and copper-loss in the resistance of the armature, the
iron-loss being negligible.
The unsaturated synchronous reactance can be obtained from the OCC and SCC of Fig. 8.12 as
Xs (unsaturated) =
VOC / 3
I SC
(8.26)
I f const
where If corresponds to the unsaturated magnetic region or VOC value corresponding to the air-gap line could
be used.
Since a synchronous machine under operating conditions works in a somewhat saturated region of the
magnetization characteristic, the performance of the machine as calculated from Xs, defined above, will differ
considerably from the actual value effective during normal operation. To account for the fact that the machine
actually operates in the saturated region, it is a must to resort to the nonlinear analysis (Sec. 8.4) or use a
heuristic technique of adjusting Xs to a suitable value.
If Xs, as defined in Eq. (8.26), is plotted for various values of the field current, the chain-dotted curve of
Fig. 8.12 will be obtained. Initially in the unsaturated region the value of the synchronous reactance remains
constant at Xs (unsaturated) and then drops off sharply because of saturation of the OCC.
The value of Xs corresponding to the field current, which gives rated voltage on open-circuit, is defined as
Xs (adjusted) =
Vt (rated )/ 3
I SC
I f corresponding to
Vt (rated) on OCC
; If = Of ¢
(8.27)
The value of Xs (adjusted) is less than Xs (unsaturated) as shown in Fig. 8.12. Obviously Xs (adjusted)
would yield the machine performance figures closer to those obtained under actual operation.
Short Circuit Ratio (SCR)
The short-circuit ratio (SCR) is defined as the ratio of the field current required to produce rated voltage
on open-circuit to the field current required to produce rated armature current with the armature terminals
shorted while the machine is mechanically run at synchronous speed. From the OCC and SCC as shown in
Fig. 8.15,
SCR =
of ¢
of ¢¢
(8.28)
458
Electric Machines
ISC
VOC (line)
Slope k1
OCC
Slope k2
SCC
Vt (rated)
Ia (rated)
c
f¢
o
f¢¢
o¢
Fig. 8.15 Short-circuit ratio (SCR)
as per the definition. It is further noted from this figure that
Xs (adjusted) =
Vt (rated )/ 3
o ¢c
(8.29)
and slopes of OCC and SCC are
k1 =
Vt (rated )
OC voltage
=
of ¢
field current
(8.30)
k2 =
SC current
I a (rated )
o ¢c
=
=
field current
of ¢¢
of ¢
(8.31)
Substituting Eqs (8.30), and (8.31) in Eq. (8.28),
SCR =
of ¢
of ¢¢
=
Vt (rated )
k2
◊
k1
I a (rated )
=
o ¢c
V (rated )
◊ t
Vt (rated ) I a (rated )
=
=
SCR =
o ¢c
Vt (rated )/ 3
◊
Vt (rated )/ 3
I a (rated )
1
◊ XBase
X s (adjusted )
1
X s (adjusted ) (pu )
(8.32)
Synchronous Machines
459
Equation (8.32) means that SCR is the reciprocal of Xs (adjusted) in pu. Therefore, a low value of SCR
implies a large value of Xs (adjusted) (pu) and vice versa.
Short Circuit Loss
During the short circuit test the loss can be obtained by measuring the mechanical power required to drive the
machine. The loss PSC comprises the following items:
1.
2.
3.
4.
I 2R loss in armature winding due to the flow of short circuit current (ac).
Local core loss caused by armature leakage flux.
Core loss due to resultant air-gap flux. As the flux is very small, this loss can be ignored.
Windage and friction loss.
Loss
The windage and friction loss can be separated out as explained in open circuit loss. The remaining loss
(items 1 and 2) is called short circuit load loss whose
SC load loss
plot with armature current is shown in Fig. 8.16. By
measuring armature dc resistance correcting it for ac and
the armature temperature during SC test, dc I 2R loss can
Stray load loss
then be subtracted leaving behind the stray load loss –
sum of load core loss and loss due to additional conductor
resistance offered to alternating current. The stray load
loss is also plotted in Fig. 8.16.
Armature current
The armature resistance as calculated from armature
Fig. 8.16
current and short circuit load loss is called effective
armature resistance. It can be calculated at rated current and then assumed to remain constant. Thus
Ra (eff ) =
short circuit load loss (per phase)
(8.33)
(short circuit armature current)2
EXAMPLE 8.1 Draw the open-circuit and short-circuit characteristics using the data given below for a
150 MW, 13 kV, 0.85 pf, 50 Hz synchronous generator.
Open-circuit characteristic
If (A)
VOC (line) (kV)
200
4
450
8.7
600
10.8
850
13.3
1200
15.4
Short-circuit characteristic
If = 750 A,
(a)
(b)
(c)
(d)
(e)
ISC = 8000 A
Determine the unsaturated synchronous reactance of the machine.
Determine the saturated synchronous reactance of the machine.
Convert the value of reactances in part (a) and (b) in their pu value.
What is the short circuit ratio of this machine?
The machine supplies full-load at a pf of 0.85 lagging and a terminal voltage of 13 kV.
Case I – Find the excitation emf and voltage regulation using the synchronous reactance (linear
circuit model).
Case II – Find again the excitation emf using saturated synchronous reactance and then find the field
460
Electric Machines
current needed to supply the specified load. With field current remaining constant, find the OC voltage
and therefrom calculate the voltage regulation of the machine.
( f ) Compare the results of part (e) cases I and II.
(g) For the load as in part (e) draw the phasor diagram showing voltages and current. Calculate the
angle between the terminal voltage and excitation emf.
SOLUTION
The OCC and SCC as per the data are drawn in Fig. 8.17.
18
Modified air-gap line
16
OCC
Air-gap line
14
Vt (rated)
SCC
10
10000
8
8000
6
6000
4
4000
2
2000
f¢¢
o
200
400
600
ISC (A)
VOC (Line)(kV)
12
f¢
800
If (A)
1000
1200
1400
1600
Fig. 8.17
Note: Values have been read from Fig. 8.17 drawn on enlarged scale
(a) Corresponding to VOC = 13 kV on the air-gap line, ISC = 7000 A for the same field current.
\
Xs (unsaturated) =
13 ¥ 1000
3 ¥ 7000
= 1.072 W
(b) Corresponding to VOC = 13 kV on the OCC, ISC = 8600 A for the same field current.
\
Xs (adjusted) =
13 ¥ 1000
3 ¥ 8600
= 0.873 W
Ia (rated) =
150 ¥ 106
3 ¥ 0.85 ¥ 13 ¥ 103
= 7837 A
Synchronous Machines
Base ohms =
Xs (unsaturated) (pu) =
(c)
Xs (adjusted) (pu) =
461
13 ¥ 1000
= 0.958 W
3 ¥ 7837
1.072
= 1.12
0.958
0.873
0.958
= 0.911 pu
If corresponding to VOC = Vt (rated) is of ¢ = 810 A
If corresponding to ISC = Ia (rated) is of ≤ = 750 A
(d)
\
SCR =
Vt =
(e)
810
1
= 1.08 =
750
X s (adjusted )( pu )
13 ¥ 1000
= 7505 V (phase value)
3
Vt = 7505 –0° V
cos f = 0.85, f = 31.8°
Ia (fl ) = 7837 A
I a = 7837 – – 31.8°
= 7837 (0.85 – j 0.527)
For generating operation
Case I:
\
E f = Vt + j I a Xs
Xs (unsaturated) = 1.072 W
Ef = 7505 + j 7837 (0.85 – j0.527) ¥ 1.072
= 11747 + j 6841
Ef = 13594 V or 23.54 kV (line) = Vt (OC) (linear model)
Regulation =
Vt (OC ) - Vt
¥ 100
Vt
13594 - 7505
¥ 100
7505
= 81%
Xs (adjusted) = 0.873 W
=
Case II:
\
E f = 7505 + j 7837 (0.8 – j 0.527) ¥ 0.873
= 11110 + j 5815 = 12540 –27.6° V
or
Ef = 12540 V
or
21.72 kV (line)
The field current needed for this Ef is found from the modified air-gap line. This cannot be found from the
OCC where Ef will correspond to highly saturated region. In fact the machine under lagging load operates in
lightly saturated region as the load current is demagnetizing. So on empirical basis field current is found from the
modified air-gap line (line joining origin to Vt (rated) in OCC). From this line use find.
If =
810
¥ 21.72 = 1353 A
13
462
Electric Machines
For this value of field current when the machine is open circuited
Vt (OC) = 16.3 kV
Voltage regulation =
16.3 - 13
¥ 100 = 25.4%
13
Comment: The voltage regulation as obtained by Xs
(adjusted) as found from the modified air-gap line
is far more realistic than that obtained by use of Xs
(unsaturated).
(f ) The phasor diagram is drawn in Fig. 8.18. It is
immediately seen that Ef leads Vt by d = 27.2°.
Observation The approximation used in transformer
(Eq. (3.60)) in calculating voltage drop cannot be
used in a synchronous machine as Xs is much larger
than in a transformer where Xeq is 0.05-0.08 pu. So
phasor calculation must be carried out.
8.5
Ef = 12540 V
IaXs = 7837 ¥ 0.873
= 6842 V
27.2°
31.8°
Vt = 7505
Ia = 7837 A
Fig. 8.18
MMF METHOD
In the synchronous reactance method of finding voltage regulation, we linearly converted all mmfs into emfs
and then heuristically adjusted the synchronous reactance to account for saturation. In the mmf we will follow
the reverse procedure.
Reproducing Eqs (8.14) and (8.19), Vt and Ia are rated valued throughout,
Fr = F f + Far
(8.34)
Vt = Er – j I a Xl – I a Ra
(8.35)
or
Vt = E r¢ – I a Ra
(8.36)
where
Er¢ = Er – j I a Xl ; Er¢ is not air-gap emf
(8.37)
In the synchronous machine, induced emf is found from associated mmf through OCC and lags it 90°, i.e.,
OCC
E ¨ææÆ – j F
If we assume magnetic linearity (that is operation on the air-gap line), we can write
Ê V ˆ
E = – jK F ; K = constant, units Á
Ë mmf ˜¯
(8.38)
(8.39)
This result helps to convert the leakage reactance voltage to equivalent mmf. Thus
– j I a Xl = – jK Fal
(8.40)
where Fal is equivalent mmf.
Ê mmf ˆ
Far = Kar I a , Kar = armature constant, units Á
Ë A ˜¯
Substituting I a from Eq. (8.41) in Eq. (8.40), we get
Far
X = K Fal
K ar l
(8.41)
Synchronous Machines
Xl ˆ
Fal = ÊÁ
Far
Ë K K ar ˜¯
W
Xl
= constant dimensionless
=
V
mmf
K K ar
◊
mmf
A
or
Units of
463
(8.42)
We can now write the mmf equivalent of Eq. (8.37) using Eq. (8.39)
Fr¢ = Fr + Fal ; – jK cancels out
Substituting Fr from Eq. (8.34), we have
(
Fr¢ = F f + Far + Fal
(8.43)
)
(8.44)
in which Far and Fal are in phase with I a
From Fr¢ we can find Er¢ from the OCC
For clarity of understanding let us draw the phasor diagram corresponding to the phasor Eqs (8.36) and
(8.44) where q = pf angle. If Ia Ra is ignored, b = q
Ff
Far + Fal
b
Fr¢
90°
90°
Vt
IaRa
q
b
E¢r
Far + Fal
Ia
Fig. 8.19 Phase diagram of mmf method
Important Note
mmfs are measured in units of field current
mmf
= If , Nf = number of effective field turns which need not be known
Nf
To determine (Far + Fal)
464
Electric Machines
Under short circuit at rated current, the field excitation is consumed in balancing armature reaction mmf
and the balance induces emf to balance ISC(rated) Xl voltage drop.
Thus
If | at rated SC current = If ,ar + If ,al
(
)
With Fr¢ and Far + Fal known as above F f can be found by drawing the mmf phasor diagram to scale.
Computationally it is convenient to use trigonometric relationship derived below.
The mmf phasor diagram is redrawn in Fig. 8.20 in convenient orientation. It easily follows that
Ff =
( AB + BC sin b ) 2 + ( BC cos b ) 2
(8.45)
Observation
In this (mmf ) method linear assumption has only been made in finding mmf equivalent of leakage reactance
voltage drop. Otherwise, saturation has been accounted for by finding emfs from mmfs and vice versa.
C
Steps to Compute Voltage Regulation
1. For Vt (rated) and Ia(rated) at specified pf
find E¢r
2. From OCC find F r¢
3. From SCC at rated current, find (Far + Fal)
4. From mmf phasor diagram or Eq. (8.44)
determine Ff
5. Consult OCC to find Ef = Vt (no load)
6. Compute voltage regulation
EXAMPLE 8.2
mmf method.
b
Ff
Far + Fal
b
A
Fr¢
B
D
Fig. 8.20
For the synchronous generator of Example 8.1, determine its voltage regulation by the
SOLUTION
PF = 0.85 lagging
Ia (rated) =
From Fig. 8.17 corresponding to
150 ¥ 106
3 ¥ 0.85 ¥ 13 ¥ 103
= 7837 A
ISC = Ia(rated) = 7837 A
If = 750 A = Far + Fal
At rated current, 0.85 pf lag
E¢r = Vt = 13 kV (line-to-line) ; Ia R a = 0.
From the OCC corresponding mmf is
If = 810 = F¢r ; in units (A) of field current
b = q = cos–1 0.85 = 31.8°
From the phasor diagram of Fig. 8.20 and Eq. 8.44
Ff =
(810 + 750 sin 31.8∞) 2 + (750 cos 31.8∞) 2
Synchronous Machines
465
= 1505 A = If
At If = 1500 A from the OCC, we find
Ef = 16.3 kV (line)
Vt (no load) = Ef = 16.3 kV (line)
Voltage regulation =
16.3 - 13
¥ l 00 = 25.4%
13
The simple circuit model of the synchronous machine was obtained in Sec. 8.3 by making the assumption that
the magnetic circuit of the machine is linear. However, it is known that, under normal operating conditions, the
machine operates in a somewhat saturated region. In order to take account of magnetic saturation a procedure
for heuristic adjustment of synchronous reactance was suggested so that the simple circuit model could still be
used with the reactance parameter Xs (adjusted). While this does give more accurate results for many practical
purposes compared to the use of Xs (unsaturated), it still does not fully account for magnetic saturation.
The mmf method presented in Section 8.5 gives more accurate results but it also uses linearity in converting
leakage reactance drop to equivalent mmf.
For taking saturation into account, superposition cannot be applied and, therefore, mmf phasor equation,
Fr = F f + Far
(see also Fig. 8.5 (b))
must be used.
The induced emfs Ef and Er corresponding to Ff and Fr can be found from the OCC (which takes magnetic
saturation into account). The problem, however, is to determine Far for a given armature current Ia, i.e. the
armature reaction constant
Far
(8.46)
Ia
This proportionality constant* must be found experimentally for a given machine.
Knowing the magnetization characteristic of the machine and the proportionality constant Kar, the phasor
diagram of Fig. 8.5(b) can be constructed for given operating conditions (say, for generating mode) wherein
no approximation of neglecting the nonlinearity need be made. The next step then is to find the terminal
voltage of the machine which immediately follows from the phasor equation
Kar =
Vt = Er – I a (Ra + jXl) (generating mode)
(8.47)
The armature resistance, Ra, can be easily measured under dc conditions and duly corrected to its ac value
and operating temperature; it can even be altogether neglected without any significant loss of accuracy of
analysis since its value is only about 0.01 pu as stated already. However, Xl, the leakage reactance of the
machine must be determined. This can be calculated from the design parameters of the machine (provided
these are known) but only to a low degree of accuracy. Therefore, it is necessary to obtain its actual value by
experimental methods.
* This proportionality constant is given by Eq. (5.44b) provided the design constants of the machine, Kw and Nph
(series) are known. Even then a designer would be interested to determine the experimental value of this proportionality constant to verify his design calculations.
466
Electric Machines
Accurate analysis of the synchronous machine performance can be carried out under any operating
conditions provided Kar (Eq. (8.34)) and Xl, the leakage reactance are known. Because of the nonlinearity of
iron, there are a variety of experimental methods of determining these quantities to high but varying degrees
of accuracy. One of the well-known methods called the Potier method is described here.
Potier Method
In the Potier method, tests are conducted to determine the following two characteristics with the machines
running at synchronous speed.
1. Open-circuit characteristic (OCC) as described in Sec. 8.3, with reference to Fig. 8.12. This is redrawn in
Fig. 8.21.
V
Air-gap line
OCC
S
Er
Ia (rated)Xl
Vt (rated)
Q
R
P
ZPFC
(Iarated)
S≤
R≤
Q≤
ar
Far /Nf = If
P≤
S¢
O
A
Q¢
B
P¢
Far /Nf = Ifar
If
Fr /Nf = Ifr
Ff /Nf = If
Fig. 8.21 Graphical features of the Potier method
2. Zero power factor (lagging) characteristic (ZPFC)
This is conducted by loading the machine as a
generator with pure inductive load* (balanced 3-phase) which is adjusted to draw rated current from
the machine while the field current is adjusted to give various values of terminal voltage. Figure 8.21
* The zero power factor test could be performed on a machine by employing loading inductors; these constitute the
load of nearly but not exactly zero of and also are impractical for large machines. The ZPF test at rated voltage
only could also be conducted by synchronizing it to the mains and regulating its excitation to yield zero pf operation (see Sec. 8.9).
Synchronous Machines
467
also shows the ZPFC. However, there is no need for conducting this test fully to determine the ZPFC.
All one needs is two points on this characteristic—P corresponding to a field current which gives the
rated terminal voltage while the ZPF load is adjusted to draw rated current, and the point P¢which
corresponds to the short-circuit conditions on the machine (Vt = 0) with the field current adjusted
to give rated armature current. Since the armature resistance is of negligible order, the short-circuit
current lags behind the resultant induced emf Er by almost 90°, Vt being equal to zero. Therefore, P¢
constitutes a point on the ZPFC. It will soon be shown that the complete ZPFC, if required, can be
constructed from the knowledge of the points P and P¢.
Figure 8.22 gives the phasor diagram under conditions of zero
power factor (lagging) load with the armature resistance neglected. It
is seen from this figure that the mmf and voltage phasor equations
Vt = Er – j I a Xl
and
Fr = F f + Far
These reduce to simple algebraic equations
Vt = Er – Ia Xl
and
Fr = Ff – Far
(8.48)
Ff
Far
Fr
(8.49)
(8.50)
(8.51)
IaXI
O
Vt
Er
The algebraic Eqs (8.50) and (8.51) can now be translated onto the
Far
OCC and ZPFC of Fig. 8.21. Further, in the test data the horizontal
Ia
axis of Fig. 8.21 being the field current If, Eq. (8.51) must be converted
into its equivalent field current form by dividing throughout by Nf, the Fig. 8.22 Phasor diagram for zero
power factor (lagging) load
effective number of turns/pole on the rotor field. It then modifies to
or
Fr /Nf = Ff /Nf – Far /Nf
I fr = If – Iar
f
(8.52a)
(8.52b)
Point P on the ZPFC corresponds to terminal voltage Vt (rated) and a field current of OB = Ff /Nf = If.
Corresponding to point P on the ZPFC, there will be a point S on the OCC which pertains to the emf Er and
the resultant excitation OA = Fr /Nf (= Ff /Nf – Far /Nf ).
From Fig. 8.21 and Eqs (8.50) and (8.51) it easily follows that
(i) SQ, the vertical distance between points P and S is nothing but the leakage reactance drop Ia (rated) Xl.
(ii) QP, the horizontal distance between points P and S, is in fact Far /Nf.
It is observed here that while point P is known from the ZPF test, the corresponding point S on the OCC is
not yet known because so far the numerical values of Ia (rated) Xl and Far /Nf are not known.
By constructing triangles parallel to SQP, points can be found on the ZPFC corresponding to the points
on the OCC, e.g. P≤ corresponds to S≤, and thereby construct the complete ZPFC, if desired. In the reverse
process, corresponding to P¢, the short-circuit point, S¢ can be located on the OCC by drawing S¢ P¢ parallel
to SP.
Obviously
S¢Q¢ = Ia (rated) Xl and Q¢P¢ = Far /Nf
Since the initial part of the OCC is almost linear, OS¢ is part of air-gap line. Therefore going back to point
P and by taking horizontal distance RP = OP¢ and drawing RS parallel to OS¢, the desired point S can be
located on the OCC corresponding to the known point P on the ZPFC.
Once point S on the OCC has been located, the following can be measured to scale:
(8.53)
SQ = Ia (rated) Xl and QP = Far /Nf = I arf
468
Electric Machines
from which Xl and Far can be calculated. Since I arf corresponds to Ia (rated), the armature mmf proportionality
constant can be found as
K¢ar = I arf /Ia (rated)
With the knowledge of Xl and K¢ar for the stator windings, the complete phasor diagram of the machine can
be constructed corresponding to any operating conditions, generating or motoring.
It must be observed here that the Potier method though elegant is not exact because of the following
explicit and implicit assumptions made therein:
1. In arriving at the algebraic Eqs (8.50) and (8.51)/(8.52), the armature resistance has been neglected.
This being a very valid assumption, introduces no error of any significance.
2. If inductors are used for conducting the ZPF test, the power factor is somewhat different from zero.
3. It has been assumed that in Fig. 8.21, S¢Q¢ = SQ = Ia (rated) Xl which means that in the ZPF test
corresponding to point P and the short-circuit test corresponding to point P¢, the leakage reactance
of the machine is assumed to remain unchanged. This is not altogether correct because the machine
excitation under short-circuit conditions is OP¢ while it is OA for point P (this point on the ZPFC
corresponds to rated terminal voltage and rated armature current). Since OA >> OP¢, point P
corresponds to saturated conditions on the machine with a larger leakage flux and hence a larger value
of leakage reactance contrary to the assumption made.
Better methods [48] are available in literature which attempt to overcome assumption 3 above but these
are beyond the scope of this book.
Construction Procedure—Potier Method (Fig. 8.23)
From the SC test ISC = Ia (rated) which locates P¢ on If – axis such that OP¢ = I SC
f
Air-gap line
V
OCC
Er
S
Ia (rated)Xl
Vt (rated)
R
P (ZPF at rated voltage)
Q
ar
If
O
P¢
A
B
Short circuit point,
Ia(rated)
Fig. 8.23 Construction procedure Potier method
If
Synchronous Machines
469
Vt (rated) and Ia (rated) locate the point P such that OB = I zpf
f
P draw a line parallel to If – axis and locate R such that RP = OP¢
R draw line parallel to the air-gap line which locates point S on OCC.
S draw SQ perpendicular to RP
PQ = I ar
f ; Xl =
SQ ( to scale)
3 I a (rated )
; OC voltage is line voltage
EXAMPLE 8.3 The following data were obtained for the OCC of a 10 MVA, 13 kV, 3-phase, 50 Hz, starconnected synchronous generator:
If (A)
VOC (line) (kV)
50
6.2
75
8.7
100
10.5
125
11.6
150
12.8
162.5
13.7
200
14.2
250
15.2
300
15.9
An excitation of 100 A causes the full-load current to flow during the short-circuit test. The excitation
required to give the rated current at zero pf and rated voltage is 290 A.
(a) Calculate the adjusted synchronous reactance of the machine.
(b) Calculate the leakage reactance of the machine assuming the resistance to be negligible.
(c) Determine the excitation required when the machine supplies full-load at 0.8 pf lagging by using
the leakage reactance and drawing the mmf phasor diagram. What is the voltage regulation of the
machine?
Also calculate the voltage regulation for this loading using the adjusted synchronous reactance. Compare
and comment upon the two results.
SOLUTION
Ia (rated) =
10 ¥ 106
3 ¥ 13 ¥ 103
= 444 A
(a) The OCC and SCC are plotted in Fig. 8.24 from which the short-circuit armature current corresponding to Vt
(rated) = 13 kV (line) on the OCC is
ISC = 688 A
\
Xs (adjusted) =
=
Vt (rated)
3 ¥ I SC (corresponding to Vt (rated))
13 ¥ 1000
= 10.9 W
3 ¥ 688
(b) To find the leakage reactance, the Potier triangle must be constructed. Figure 8.25 shows the plot of the OCC and
the location of the point P corresponding to ZPF at rated current and voltage and the point P¢ corresponding to
short-circuit at rated current.
At P draw parallel to the horizontal axis PR = OP¢. At R draw RS parallel to the initial slope of the OCC thereby
locating the point S on the OCC. Draw SQ perpendicular to RP. Then
SQ =
\
Xl =
3 Ia (rated) Xl = 1200 V
1200
= 1.56 W
3 ¥ 444
(c) From Fig. 8.22, the armature reaction in the equivalent field current is
I ar
f = QP = 90 A
470 Electric Machines
Modified air-gap line
16
Air-gap line
OCC
14
13 kV
12
SCC
8
1000
6
750
688
4
500
2
250
0
50
100
150
200
250
ISC(A)
VOC (line) (kV)
10
300
Field current (A)
Fig. 8.24
18
OCC
S
16
VOC (line)(kV)
14
R Q
Ifar
P
12
10
8
6
4
2 S¢
O
O Q¢
P¢
50
100
150
200
250
Field current (A)
300
Fig. 8.25
The phasor diagram is drawn in Fig. 8.26. Proceeding computationally
–3
Er = 13–0º + j 3 ¥ 444 ¥ 1.56 ¥ 10 = 13.75 –4º kV (line)
÷3IaXl
Synchronous Machines
ar
If
= 90° A
b
r
If = 178.5 A
If = 253.3 A
4°
36.9°
Er = 13.75 KV
j÷3IaX2
Vt = 13
KV
If = (rated)
b = 36.9° + 4° = 40.9°
Fig. 8.26
From the OCC corresponding to
Er = 13.75 kV
I fr = 185 A leads Er by 90º
From the phasor diagram
If = [(185 + 90 sin 40.9º)2 + (90 cos 40.9º)2]1/2
= 253 A
corresponding to which the value of Ef from the OCC is
\
Ef = 15.2 kV (line)
E f - Vt
Voltage regulation =
¥ 100
Vt
=
15.2 - 13
¥ 100 = 16.9%
13
Voltage regulation using Xs (adjusted):
E f = Vt + j 3 I a (rated) Xs (adjusted)
or
\
= 13000 +j 3 ¥ 444 (0.8 – j 0.6) ¥ 10.9
= 18029 + j 6706
Ef = 19.24 kV (line)
150
If (from modified air-gap line) =
¥ 19.24 = 222 A
13
Vt (OC)| If = 222 A = 14.8 kV
14.8 - 13
Voltage regulation =
¥ 100
13
= 13.85%
471
472 Electric Machines
Comment
The voltage regulation as calculated above by Potier’s method is quite accurate. In fact this value is somewhat
lower than the actual as it does not account for increase in leakage reactance under condition of load. This fact
has already been mentioned earlier. A lower value given by Xs (adjusted) method compared to Xs (unadjusted)
is explained by the fact that the method is empirical and the error caused by it depends upon the saturation
level of the machine at rated voltage.
EXAMPLE 8.4
If (A)
VOC (line) (V)
The OCC of a 3-phase, 50 Hz synchronous machine is given by the following data:
15
600
30
1200
50
2000
75
2900
90
3300
120
3700
160
4000
Under short-circuit conditions a field current of 40-A gives the full-load stator current. The armature
resistance and leakage reactance per phase are known to be 0.01 and 0.12 pu. When the machine is
operating as a motor drawing full-load current at the rated terminal voltage of 3.3 kV and 0.8 pf leading,
calculate the field current required.
SOLUTION The OCC as per the data given is drawn in Fig. 8.27. The field current required to circulate the full-load
short-circuit current is indicated by OP¢ = 40 A. Now
3 Ia Xl = 0.12 ¥ 3300 = 396 V (at rated current)
Corresponding to 396 V on OCC, a point S¢ is marked and a vertical line S¢Q¢ drawn. Q¢P¢ now corresponds to Far /
Nf = Iar
f = 30 A.
4000
Er (line)
OCC
VOC (line)(kV)
3000
2000
1000
0
Ifr
S¢
400
Q¢
25 P¢ 50
75
f
I ar
100
If (A)
Fig. 8.27
125
150
175
Synchronous Machines
473
Given that Ra = 0.01 pu, the voltage drop in resistance is so small that it can be neglected.
Vt =
Ia X l =
3.3
¥ 1000 = 1905 (phase)
3
396
= 229 V
3
The machine is operating as a motor at 0.8 pf leading. Therefore as per Eq. (8.12) and Fig. 8.8(b)
Er = Vt – jIa Xl ; Vt (phase) =
3300
= 1905 V
3
Er = 1905 –0º – j 229 (0.8 + j 0.6)
= 2042 – j 183.2
ar = 40
A
If
= 2050 –– 5.1º V or 3551 V (line)
42°
From the OCC for Er = 3551 V, we get
I fr = 108 A
Ifr
The phasor diagram is drawn in Fig. 8.28.
leads Er by 90º and I ar
f is drawn anti-parallel to Ia
(motoring operation) Angle b = 36.9º + 5.1º = 42º.
The construction lines are shown dotted. From the mf
phasor diagram geometry
or
or
If = 125 A
Ifr = 108 A
2
ar
2 1/2
If = [(If + I ar
f sin b ) + (I f cos b ) ]
If = [(108 + 30 sin 42º)2 + (30 cos 42º)2]1/2
If = 125 A
Ia
°
36.9
From the OCC at
If = 125 A
Ef = 3760 V (line)
V t = 1905 V
IaXL = 229 V
5.1°
Er = 2050 V
Fig. 8.28
ASA method of finding voltage regulation is heuristic modification of the mmf method which requires the
value of Xl. Therefore, OCC and ZPFC tests are conducted and Xl is determined from the Potier triangle by
the method of Section 8.6.
The OCC is sketched in Fig. 8.29 and the short circuit point is located on the If – axis.
The basic kVL equations of the mmf method are:
or
Er¢ = Vt + I a Ra
Er¢ = Vt ; I a Ra ignored; b = q
Eq. (8.36)
Er = Vt + j I a X l
Eq. (8.35)
where we know Xl.
In stead of finding Fr¢ from the OCC against E¢r = Vt (rated). F¢r is read on the air-gap line as shown in
Fig. 8.29(a). If means saturation has been ignored.
474
Electric Machines
Now draw the mmf phasar diagram of Fig. 8.29(b). with b = q, the pf angle. This determines
(
)
F f¢ = Fr¢ + Far + Fal ; saturation ignored
Saturation Correction
Against Er, the intersept LM = Fsat between air-gap line and OCC in found out. Heuristically this is the
saturation correction. We extend F f¢ on the phasar diagram by Fsat to yield
Ff = F¢f + Fsat; magnitude-wise
Having determined Ef we read Ef from the OCC and therefrom calculate the volatage regulation.
Air-gap line
V
Er
Vt (rated) = E¢r
K
Fsat
L
OCC
M
G
H
F¢r
O
P
If
(S.C.point)
Far + Fal
(a)
t
F sa
Ff
F¢f
b
Far + Fal
F¢r
(b)
Fig. 8.29
EXAMPLE 8.5
SOLUTION
Determination of voltage regulation by ASA (latest) method
For Example 8.4 calculate the voltage regulation by the ASA method.
For the data of Example 8.4 the OCC and air-gap are drawn in Fig. 8.30. Ignoring Ia R a drop
and
From the air-gap against
E¢r
Er
Vt
F¢r
= Vt (rated) = 13 kV (line)
= VL + j Ia (rated) Xl = 13.75 kV (line); calculated in Example 8.4
= 13 kV, we find
= 100 A, b = 36.9º + 4º = 40.9º
Synchronous Machines
475
V(kV)
16
Air-gap line
Er = 13.75 kV
Fsat = 80 A
14
E¢r = 13 kV
F¢r = 100 A
12
10
8
6
4
2
P
O
100
150
200
250
300
If (A)
Far + FaL = 100 A
Fig. 8.30
Against Er = 13.75 kV, the line segment between air-gap line and OCC, we find
Fsat = 80 A
OP¢ = Far + Fal = 100 A
Also
From the values of ASA, the mmf phasor diagram is sketched in Fig. 8.31 from which we find
F¢f = [(100 + 100 sin 40.9º)2
+ (100 cos 40.9°)2]1/2
= 182 A
Then
Ff = 182 + 80 = 262 A
From the OCC, we find
Ef = 15 kV (line)
Voltage regulation =
8.8
15 - 13
¥ 100 = 15.4%
13
t
Ff¢ F sa
A
= 80
Ff
0A
=
40.9°
+
10
F al
F ar
F¢r = 100 A
Fig. 8.31
NATURE OF ARMATURE REACTION
The nature of the armature reaction is dependent on the power factor at which the machine is operating modegenerating/motoring. For simplicity of explanation, it will be assumed here that the armature resistance and
leakage reactance are negligible so that
Vt = Er
Figure 8.32 shows the phasor diagrams, with component fluxes (i.e. the magnetic circuit is assumed linear)
indicated therein for a generating machine for (a) zero power factor lagging (Fig. 8.32(a)), (b) zero power
factor leading (Fig. 8.32(b)) and (c) for unity power factor (Fig. 8.32(c)). The following observations are
immediately made from these phasor diagrams.
476 Electric Machines
1. Armature reaction is demagnetizing (Far opposes Ff) when a generating machine supplies zero power
factor lagging current.
2. Armature reaction is magnetizing (Far aids Fr) when a generating machine supplies zero power factor
leading current.
3. Armature reaction is mostly cross-magnetizing (i.e. at 90° to Fr) though it has a small demagnetizing
component (see dotted curve in Fig. 8.32(c), when a generating machine supplies unity power factor
current.
Ff
Far
Fr
Fr
la
Vt = E r
Far
Far
Ff
Ff
Fr
Vt = E r
la
Vt = E r
la
(a) Zero power factor lagging
(b) Zero power factor leading
(c) Unity power factor
Fig. 8.32 Nature of armature reaction in generating machine
From the above discussion the following more general conclusions regarding a synchronous machine in
generating mode can be drawn:
(i) When the machine supplies a lagging power factor current, the armature reaction has both demagnetizing
and cross-magnetizing components.
(ii) When the machine supplies leading power factor current, the armature reaction has both magnetizing
and cross-magnetizing components.
Since in a motoring machine the armature reaction mmf and flux are in phase opposition to the armature
current, the nature of the armature reaction is just the reverse of what is stated above for the generating
machine. The corresponding conclusions for the motoring machine are stated below:
(i) When the machine draws a lagging power factor current, the armature reaction has both magnetizing
and cross-magnetizing components.
(ii) When the machine draws leading power factor current, the armature reaction has both demagnetizing
and cross-magnetizing components.
The effect of the above conclusions on machine operation will be seen in further detail in Sec. 8.10.
A definite procedure has to be followed in connecting a synchronous machine to bus-bars which for the
present purpose will be assumed to be infinite. Infinite busbars means a 3-phase supply of constant voltage and
frequency independent of the load exchanged (fed into the bus-bars or drawn from the bus-bars). Figure 8.33
Synchronous Machines
477
shows a synchronous machine with terminals a, b, c which is required to be connected to bus-bars with
terminals A, B, C by means of a switch S.
a S
synchronous
machine
A
3-phase mains
(infinite bus-bars)
Vbus
V machine
b
B
c
C
ns
Fig. 8.33
The machine is run as a generator with its terminals so arranged that its phase sequence is the same as that
of the bus-bars. The machine speed and field current are adjusted so as to satisfy the following conditions:
(i) The machine terminal voltage must be nearly equal to the bus-bars voltage.
(ii) The machine frequency is nearly equal to the bus-bars frequency, i.e. the machine speed is close to
synchronous speed.
After the above conditions are satisfied the instant of switching on (synchronising) must be determined
such that the two voltages are almost co-phasal (the acceptable phase difference is of the order of 5°). This
instant is determined with the help of the method described below.
Figure 8.34 shows the phasor diagram for phase voltages
A
a
(line-to-neutral) for the machine and bus-bars. As the two
frequencies are not exactly equal, the machine phasors are
rotating slowly with respect to the bus-bar phasors at 2 pDf
2 p D f rad/s
rad/s, where Df is the difference in the two frequencies. At the
instant when the two sets of phasors are coincident (cophasal),
the voltage
VaA = 0
VbC = VcB
c
(8.54)
The condition can be easily determined by connecting three
C
B
lamps—one across aA, the other across bC and the third across
cB (in order to use standard-voltage lamps it may be necessary
b
to employ potential transformers). The rms values of voltages Fig. 8.34 Determination of the synchronizing
VaA, VbC and VcB oscillate at the difference frequency D f so
instant
that each lamp is alternately dark and bright. At the instant
of synchronization, as per the condition (8.54) stated above, the lamp across aA is dark while the other two
lamps are equally bright. It is at this instant that switch S is closed. Instead of using lamps, generating stations
use an instrument called synchroscope.
Once switch S is closed the stator and rotor fields of the machine lock into each other (synchronize) and
the machine then onwards runs at synchronous speed. The real power exchange with the mains will now be
478
Electric Machines
governed by the loading conditions on the shaft while the reactive power exchange will be determined by the
field excitation.
As a generator is coupled to a primemover it is easy to follow the above procedure to connect it to the
bus-bars. The same procedure has to be followed for a synchronous motor which must be run initially by
an auxiliary device (may be a small dc/induction motor) and then synchronized to the bus-bars. It may
be pointed out here that the synchronous motor is non-self-starting. If, for example, switch S is closed in
Fig. 8.35 with the rotor stationary, the stator and rotor fields will be moving relative to each other at
synchronous speed so as to develop alternating torque with zero average value and as a result the motor
would not start. Synchronous motors are made self-starting by providing short-circuited bars on the rotor
which produce induction torque for starting (see Sec. 5.6).
The operating characteristics of a synchronous machine are examined here under conditions of variable
load and variable excitation. One of these quantities will be assumed to be held constant at a time while the
other will be allowed to vary over a wide range. Further, here too the armature resistance will be assumed
negligible. This does not significantly change the operating characteristic of the machine but leads to easier
understanding of the machine operation. The more general case of the machine with armature resistance
accounted for will be discussed in Sec. 8.11. By virtue of negligible resistance assumption, the electrical
power at the machine terminals and the mechanical power at its shaft are simply related as follows:
Pe (out) = Pm (in) (net)
where Pe (out) = electrical power output of the machine (electrical power developed)
Pm (in) = net mechanical power input to the machine after deducting iron-loss and windage and
friction loss
Motoring Machine
where
Pe (in) = Pm (out) (gross mechanical power developed)
Pe (in) = electrical power input to the machine
Pm (out) (gross) = gross mechanical power output of the machine; the net mechanical power output
will be obtained by deducting iron-loss and windage and friction loss
Power-angle Characteristic (Constant Excitation Variable Load)
Figure 8.35 shows the circuit diagrams and phasor diagrams of a synchronous machine in generating mode
(Figs 8.35(a) and (c) and motoring mode (Figs 8.35 (b) and (d)). The machine is assumed to be connected
to infinite bus-bars of voltage Vt. It is easily observed from the phasor diagrams that in generating mode, the
excitation emf Ef leads Vt by angle d, while it lags Vt in the motoring mode. It follows from the phasor triangle
OMP (Figs 8.35(c) and (d) that
Ef
sin (90 ± f )
=
Ia X s
; (90∞+ f ), generating
sin d
(90∞- f ), motoring
Synchronous Machines
Xs
Ia
479
Xs
Ia
+
+
+
+
Vt
Vt
Ef
Ef
–
–
–
–
(b) Motoring mode Ef = Vt – jIaXs
(a) Generating mode Ef = Vt + jIaXs
P
Ef
jIaXs
O
d
d
f
Vt
Vt
O
M
f
Ia
Ia
(c) Generating mode
M
Ef
P
jIaXs
(d) Motoring mode
Fig. 8.35 Synchronous machine operation (generating/motoring mode)
or
Ia cos f =
where f is the power factor angle.
Multiplying both sides of Eq. (8.55) by Vt
Vt Ia cos f =
Ef
Xs
sin d
Vt E f
Xs
(8.55)
sin d
Vt E f
sin d
(8.56)
Xs
where
Pe = Vt Ia cos f = electrical power (per phase) exchanged with the bus-bars
d = Angle between Ef and Vt and is called the power angle* of the machine (d has opposite sign
for generating/motoring modes).
The relationship of Eq. (8.56) is known as the power-angle characteristic of the machine and is plotted in
Fig. 8.36 for given Vt and Ef . The maximum power
or
Pe =
Pe, max =
Vt E f
(8.57)
Xs
occurs at d = 90° beyond which the machine falls out of step (loses synchronism). The machine can be taken
up to Pe, max only by gradually increasing the load. This is known as the steady-state stability limit of the
* The angle d in Eq. (8.56) is between Vt and Ef while in Fig. 8.9 it is the angle between Er and Ef . The difference
between these two angles is due to the fact that now the leakage reactance of the machine is being accounted for.
480
Electric Machines
machine. The machine is normally operated at d much less** than 90°. The phasor diagram of a generating
machine under condition of Pe,max is drawn in Fig. 8.37. Obviously Ia will be several times larger than the
rated machine current in this condition.
P
Generating
Pe, max
Ef
–180°
jlaXs
–90°
90°
180°
Ia
d
d = 90°
Pe, max
f
Vt
Motoring
Fig. 8.36
Power-angle characteristic
Fig. 8.37
Phasor diagram of generating machine at
steady-state stability limit
Operation at Constant Load with Variable Excitation
At constant load, from Eq. (8.56)
Ef sin d =
Also
or
Pe X s
= const
Vt
(8.58)
Vt Ia cos f = Pe = const
Ia cos f =
Pe
= const
Vt
(8.59)
It is therefore, observed that at constant load, as the excitation emf Ef is varied (by varying field current If),
the power angle d varies such that Ef sin d remains constant. The machine bahaviour is depicted by the
phasor diagrams of Figs 8.38(a) and (b)). As Ef varies, the tip of phasor E f moves on a line parallel to Vt
and at distance Ef sin d = Pe X s /Vt from it. Since Ia cos f = constant, the projection of the current phasor on Vt
must remain constant, i.e. the tip of the current phasor traces a line perpendicular to Vt at distance Ia cos f =
Pe /Vt from the origin. The current phasor I a is always located at 90° to phasor I a Xs (phasor joining tips to
E f and Vt in the direction of E f ). The effect of varying excitation (Ef) on machine operating characteristics
is brought out by Figs 8.38(a) and (b).
Normal excitation: At this excitation the machine operation meets the condition Ef cos d = Vt at which the
machine power factor is unity.
Over excitation:
Ef cos d >Vt
Under excitation:
Ef cos < Vt
** This is to prevent the machine from going into an unstable region (where it will fall out of step) during transient
power swings. This topic concerns transient stability of the machine and is discussed in books on power systems [7].
Synchronous Machines
481
The following conclusions* are drawn from the phasor diagrams of Figs 8.38(a) and (b).
Normal-excitation
Under-excitation
Unstable
region
Over-excitation
Ef1
1X
a
Ef2
jI
a
I
a3
Ef (min)
Ef3
o
PeXs
Vt
d1
f1
Ia2
Vt
Ia1
Pe
Vt
(a) Generating machine; Pe(out) = Pm (in) (net) = constant
Pe
Vt
Ia3
o
Ia2
f1
Ef (min)
Unstable
region
Vt
jIaXs
d1
PeXs
Vt
Ia1
Ef1
Under-excitation
Ef2
Ef3
Over-excitation
Normal-excitation
(b) Motoring machine; Pe (in) = Pm (out) (grass) = constant
Fig. 8.38
1. The machine supplies a lagging power factor current when over-excited.
2. The machine supplies a leading power factor current when under-excited.
* These conclusions are corroborated by the nature of the armature reaction discussed in Sec. 8.7. For example,
when a generator is overexcited it supplies lagging current which has a demagnetizing effect so that the air-gap
emf Er matches the applied voltage. Similarly an overexcited motor draws a leading current which has demagnetizing effect.
482
Electric Machines
Motoring Machine
1. The machine draws a leading power factor current when over-excited.
2. The machine draws a lagging power factor current when under-excited.
Minimum Excitation
From Figs 8.38(a) and (b) it is seen that as excitation is reduced, the angle d continuously increases. The
minimum permissible excitation, Ef (min), corresponds to the stability limit, i.e. d = 90°. Obviously
Ef (min) =
Pe X s
Vt
(8.60)
The reader is advised to draw a phasor diagram at Ef (min) for motoring machine corresponding to
Fig. 8.37.
V– Curves
Let us consider the phasor diagram of Fig. 8.38 (b) for the motoring machine. At low excitation, Ia is large and
pf is low lagging at a given constant Pe = Pm (load) say 1 pu. As the excitation is increased Ia reduces and pf
increases till at normal excitation Ia is minimum and pf is unity. As the excitation is increased further Ia begins
to increases and pf becomes leading and begins to reduce. The plot of Ia vs at Pm = 1 pu exhibits a V-curve
nature as shown in Fig. 8.39(a). At lower values of Pm the plot has same V-curve shape except that Ia (min)
is smaller and occurs at lower values of If as shown by the dotted curve passing that Ia (min). The minimum
excitation stability limit given by Eq. (8.60) is also indicated in the figure.
Armature
current, Ia
Stability limit (min. excitation)
1.0 pf
0.8 pf
0.8 pf 0.75
0.5
0.25
0
1pu load (real power)
If
Over-excitation
Lagging pf (generator)
Leading pf (motoring)
Under-excitation
Leading pf (generator)
Lagging pf (motoring) Normal-excitation
Fig. 8.39(a)
Synchronous Machines
483
It is easily seen that pf vs If curves for fixed Pm are inverse of Ia vs If plot. Thus these are inverted V-Curves
each having maximum value of unity (pf) also shown in Fig. 8.39(b).
The generator V-curves and inverted V-curves can be found to have the same form as for the motor except
for reversal of lagging and leading pf regions as shown in Figs 8.39(a) and (b).
pf
1
Stability
limit
1 Pu load
0 Pu load
If
Leading pf (generating)
Loagging pf (motoring)
Lagging pf (generating)
Leading pf (motoring)
Fig. 8.39(b)
Observation
In a synchronous machine the real electrical power exchanged with the bus-bars is controlled by the
mechanical shaft power irrespective of excitation. The excitation, on the other hand, governs only the power
factor of the machine without affecting the real power flow. For example, in a generator if it is desired to
feed more real power into the bus-bars the throttle must be opened admitting more steam into the turbine
(coupled to generator) thereby feeding more mechanical power into shaft. As a consequence the power angle
d increases and so does the electrical power output (Eq. 8.56)). However, if it is desired to adjust the machine
power factor, its excitation should be varied (well within the limit imposed by Eq. (8.60)).
The dotted curves of Fig. 8.39(a) pertain to constant
terminal voltage, constant power factor operation of a
synchronous machine. For a generating machine operation
these curves are called compounding curves. These are
presented once again in Fig. 8.40 as the field current
needed for a given armature current or kVA loading at
a particular power factor for constant terminal voltage.
These are useful guide for generator operation in a power
house.
If to maintain Vt constant
Compounding Curves
0.8 pf lag
1.0 pf
0.8 pf lead
Armature current (la) or kVA
Fig. 8.40
Compounding curves of a synchronous
generator
484 Electric Machines
While the shape of the compounding curves can be visualize from the dotted curves of Fig. 8.39(a), these
are better understood by means of the phasor diagrams of Fig. 8.41.
Locus of
Ef
Locus of Ef
Locus of Ef
Ia
Ef > 1
Et > 1
IaXs
IaXs
f
Ef
Vt = 1
Vt = 1
Ia
(b) pf = unity
(a) pf = 0.8 lag
IaXs
f
Vt = 1
(c) pf = 0.8 lead
Ia
Fig. 8.41
These phasor diagrams are drawn to determine excitation emf Ef and so the field current If to maintain
constant terminal voltage Vt = 1.0(say) for specified power factor with increasing armature current, Ia. The
three phasor diagrams pertain to lagging (0.8), unity and leading (0.8) power factors.
The following conclusions are drawn
(a) 0.8 pf lagging
Ef > 1 ; Ef and so If required increases with increasing Ia as shown by the locus of Ef tip.
(b) unity pf
Ef > 1 but less than Ef (lagging pf ), also increases at a slower rate. Therefore If needed is less than
lagging pf case and has to be increased.
(c) 0.8 pf lagging
Ef < Vt; it is seen from the locus the Ef reduces goes through a minimum and then continues to increase.
Therefore, If should be adjusted accordingly.
These conclusions corroborate the compounding curves of Fig. 8.41.
Zero power factor case
or
Ef = Vt – j I a Xs ,
I a = Ia – ∓ 90º = ∓ j Ia
Ef = Vt ± Ia X s, a scalar equation
Hence Ef increases linearly from Vt = 1 for increasing Ia with zero pf lagging but decreases linearly for
increasing Ia with zero pf leading.
It is instructive for the reader to draw the corresponding phasor diagram.
The corresponding compounding curves has no practical significance for a motor and are not drawn in
Fig. 8.40. It needs to be mentioned here that for zero pf leading load Ef cannot be reduced below Ef (min) as
per Eq. (8.60).
Synchronous Machines
485
Rating based on temperature rise is volt-amperes (in practical units of kVA, MVA). As the temperature rise
is related to losses, the iron loss determines the voltage rating and Ia2 Ra loss determines the current rating.
However, MVA rating unlike transformers is incomplete for alternators it gives no information on the real
power which is needed to determine the size of the prime mover (turbine) to drive the alternator. Therefore,
the alternator is rated in terms MW capacity and not in terms of MVA. The other alternator rating is the power
factor at which it supplies power. The pf rating is normally in the range of 0.8 to 0.9 lagging. It limits the
exciter output and the field current and so the heating of field winding. Of course, the terminal voltage must
remain within narrow limits (± 5%) of the rated value. As the alternator is normally connected to the bus-bars,
its terminal voltage is the bus-bar voltage.
The reactive power output of an alternator is Q = P tan (cos–1 pf ). The reactive power flow increases if
the alternator is operated at lower factor at rated real power. The reactive power flow is limited by armature
heating. At still larger reactive power flow (lower pf ), much larger field current is needed which would not
be permissible.
To Sum Up
Alternator ratings are Vt (kV line), MW and pf = 8.0 – 0.9 lagging (if unspecified it should be taken as lagging
because at lagging pf alternator requires larger field current)
For a unit system – boiler-turbine, alternator and step-up transformer form one generator unit.
Turbine rating = MW rating of alternator plus over-load margin
Transformer rating, MVA =
MW rating of alternator
pf rating
Synchronous Condenser
It has been seen above that a synchronous motor under over-excited condition operates at a leading power
factor. Synchronous motors are therefore employed in large power installation for overall high power factor
of the installation.
At no-load with losses assumed negligible, a synchronous motor operates at
d=0
(see Eq. (8.56))
which means that Ef and Vt are in phase. It is seen from the phasor diagram of Figs 8.42(a) and (b), that the
machine (motor) draws zero power factor leading current
Ia =
E f - Vt
Xs
(Ef >Vt, over-excited)
and draws zero power factor lagging current
Ia =
Vt - E f
(Ef < Vt, under-excited)
Xs
Thus a synchronous motor at no-load behaves as a variable condenser or inductor by simply varying
its excitation. The machine operated under such a condition (motor on no-load or light load) is known as
a synchronous condenser and finds application in large integrated power systems for improving the power
factor under heavy-load conditions and for deproving the power factor under light-load conditions, thereby
controlling the voltage profile of the power system within reasonable limits.
486
Electric Machines
la
jlaXs
jlaXs
Vt
Ef
Ef
Vt
la
(a) As capacitor (variable)
(over-excited)
(b) As inductor (variable)
(under-excited)
Fig. 8.42 Synchronous condenser
Dual-purpose Synchronous Motor
Synchronous motor is used in an industry/factory for serving two purposes. It drives a constant speed
mechanical load such as a large pump, a dc generator, etc. and at the same time it also corrects an otherwise
low lagging pf of the electrical load such as induction motors and fluorescent tubes. Such a synchronous
motor serving dual-purpose is called dual-purpose synchronous motor.
EXAMPLE 8.6 A, 3300 V, delta-connected motor has a synchronous reactance per phase (delta) of 18 W.
It operates at a leading power factor of 0.707 when drawing 800 kW from the mains. Calculate its excitation
emf.
SOLUTION
On equivalent-star basis,
Vt = 3300/ 3 = 1905 V
Xs = 18/3 = 6 W
Ia =
800 ¥ 1000
= 198 A
3 ¥ 3300 ¥ 0.707
Ia X a = 198 ¥ 6 = 1188 V
cos f = 0.707 or f = 45° (leading)
The phasor diagram is drawn in Fig. 8.43 from which Ef
can be measured if the diagram is drawn to scale, or directly by
calculating from the geometry of the phasor diagram.
Ia
O
f = 45°
M
d
Vt
–MPQ = 45°
MQ = PQ = 1188/ 2 = 840
Ef = OP =
or
(1905 + 840) 2 + (840) 2 = 2871
Et
j Ia Xs
Q
f
P
Fig. 8.43
4972 V (line)
It may be seen that the motor is operating over-excited.
EXAMPLE 8.7 A 1000 kW, 3-phase, star-connected, 3.3 kV, 24-pole, 50 Hz synchronous motor has a
synchronous reactance of 3.24 W per phase; the resistance being negligible.
Synchronous Machines
487
(a) The motor is fed from infinite bus-bars at 3.3 kV. Its field excitation is adjusted to result in unity pf
operation at rated load. Compute the maximum power and torque that the motor can deliver with its
excitation remaining constant at this value.
(b) The motor is now fed from a 1200 kVA, 3-phase star-connected, 3.3 kV, 2-pole, 50 Hz synchronous
generator with a synchronous reactance of 4.55 W per phase, the resistance being negligible. Compute
the field excitations of motor and generator when the set is operating at rated terminal voltage at unity
pf and the motor is delivering full-load power. The field excitations of both the machines remaining
constant, the motor load is gradually raised. Compute the maximum power and torque that the motor
can deliver. Also compute the terminal voltage when the motor is delivering maximum power.
SOLUTION
(a) The operation of motor at infinite bus-bars is shown in Fig. 8.44.
3.24 W
Vt = 3300/ 3 = 1905 V
Ia =
+
1000 ¥ 1000
= 175 A
3 ¥ 3300 ¥ 1
cos f = 1,
Ia
+
Vt
Efm
–
f = 0º
–
Taking the terminal voltage as reference,
Fig. 8.44
Vt = 1905 –0º V
I a = 175 –0º A
Then the excitation emf is computed as
which gives
Efm = 1905 –0° – j 175 –0º ¥ 3.24
= 1905 – j 567
Efm = 1987 V
Excitation remaining fixed, the maximum power delivered by the motor is
pe,max = Pm,max (gross)
=3¥
Vt E f
X sm
=3¥
1905 ¥ 1987
3.24 ¥ 1000
= 3505 kW (3-phase)
\
wsm =
120 ¥ 50 ¥ 2p
= 26.18 rad/s
24 ¥ 60
Tmax =
3505 ¥ 1000
= 133.9 ¥ 103 N m
26.18
(b) Figure 8.45(a) shows the generator feeding the motor. At rated
terminal voltage, unity pf, full-load operation
4.55 W
Vt = 1905 –0°
I a = 175 –0°
As calculated before
Now
Efm = 1905 V
E fg = 1905 –0° + j 175 –0° ¥ 4.55
= 1905 + j 796
3.24 W
Ia
+
+
+
Vt
Efg
–
Efm
–
–
Fig. 8.45(a)
488 Electric Machines
or
Efg = 2065 V
The total series reactance X = Xsm + Xsg
= 3.24 + 4.55 = 7.79 W
The maximum power output delivered by the motor is
Pe,max = Pm,max (gross)
=3¥
E fg ¥ E fm
X
2065 ¥ 1987
=3¥
= 1580 kW (3-phase)
7.79 ¥ 1000
1580 ¥ 1000
= 60.35 ¥ 103 Nm
26.18
The phasor diagram under condition of maximum power output is drawn in Fig. 8.45(b). For convenience,
choosing the motor excitation as reference,
At maximum power
j Ia Xsg
Output d = 90º
Ia
E fm = 1987 –0º V
Tmax =
E fg = 2065 –90º V
Then
Ia =
=
j I a Xsm =
E fg - E fm
jX
2065 –90∞ - 1987 –0∞
j 7.79
-1987 + j 2065
¥ j 3.24
j 7.79
Efg
Vt
j Ia Xsm
Efm
Fig. 8.45(b)
= – 826.4 + j 858.9
Now
Vt = E fm + j I a Xsm
or
= 1987 – 826.4 + j 858.9
= 1160.6 + j 858.9
Vt = 1443.85 or 2500 V (line)
Remark The reduction in Pe,max in case (b) compared to case (a) is explained by the fact that Vt in this case is
only 2500 V (line) compared to 3300 V in case (a).
Consideration of Armature Resistance
Figures 8.46(a) and (b) show the circuit model of synchronous machine for generating motoring modes of
operation with due consideration of armature resistance. Operational analysis can be carried out by means of
the phasor equations for Figs 8.46(a) and (b) respectively or by the phasor diagrams of Figs 8.46(c) and (d).
For the sake of clarity Ia Ra voltage drop is shown larger, i.e. out-of-proportion. (The armature resistance has
to be accounted for in efficiency calculations).
EXAMPLE 8.8 A synchronous generator feeds power to a power system. The generator and power
system data are:
Generator:
100 MVA, 11 kV
Synchronous Machines
Ia
Ra
Xa
Ra
Ia
+
489
Xa
+
+
Vt
+
Vt
Ef
Ef
–
–
–
–
(a) Generating mode Ef = Vt + Ia Ra + jIa Xs
(a) Motoring mode Ef = Vt – Ia Ra – jIa Xs
Ia
Ef
O
f
d
jIa Xs
M
Vt
Ia
Q
f
O
Vt
Q
Ia Ra
d
Ef
(c)
M
Ia Ra
j Ia Xs
P
(d)
Fig. 8.46 Synchronous machine operation; armature resistance considered (generating/motoring mode)
Unsaturated synchronous reactance
Xs = 1.3 pu
Power System: Thevenin’s equivalent as seen from the generator terminals is
VTH = 1 pu, XTH = 0.24 pu (on generator base)
Generator open circuit voltage 11 kV at a field current of If = 256 A
(a) Generator internal emf Ef is adjusted to 1 pu. What is the maximum power that the generator supplies
to the power system?
(b) The generator feeds power Pe = 1 pu to the power system at generator terminal voltage Vt = 1 pu.
Calculate the power angle d of the generator and the corresponding field current If.
Plot Vt as the load is varied from a to 0.8 pu. Ef held constant at 1 pu. Use MATLAB.
(c) The generator is fitted with automatic voltage regulator, which is set for Vt = 1 pu. Load is now varied.
Plot If versus Pe.Use MATLAB.
SOLUTION
The equivalent circuit model of the generator feeding the power system is drawn in Fig. 8.47(a).
Bases
(MVA)B = 100
(kV)B = 11
Power (MW)B = 100
(a) Ef =1 pu, Pe = 1 pu, specified VTH = 1 pu
Max. Power Supplied,
E f Vth
Pe,max =
X s + X TH
Xs
Ia
XTH
+
+
Vt
Ef
–
VTH
–
Fig. 8.47(a)
490 Electric Machines
or
1
= 0.649 pu or 64.9 MW
1.3 + 0.24
Pe = 1 pu, Vt = 1 pu; specified VTH = 1 pu
Pe,max =
(b)
For power transferred from generator terminals to load
Pe =
Vt VTH
sin d1
X TH
1=
1
sin d1
0.24
d1 = sin–1 (0.24) = 13.9º
Reference phasor VTH –0º
Vt = e j13.9°
Vt - VTH
Ia =
j X TH
=
(e j13.9∞ - 1)
= 1 + j 0.122 = 1.007 –7º pu
j 0.24
To find Ef and If
Slope of air-gap line,
For
(b)
Let
E f = VTH + j (Xs + XTH) I a
= (1 + j0) + j (1.54) (1 + j 0.122) = 1.74 e j62.2º
| E f | = 1.741 pu
= 19.15 kV
d = 62.2∞
Voc
11
=
V/A
If
256
Ff = 19.15 kV
256
If =
¥ 19.15
11
= 445.7 A
Ef = 1 pu (held constant), Pe varied
E f = Ef e jd
Pe =
Pe =
Ia =
E f VTH
X s + X TH
1
e jd
1.54
E f e jd - VTH –0∞
j ( X s + X TH )
(i)
=
e jd - 1
j 1.54
(ii)
We then get
Vt = VTH + j XTH I a
Vt = 1 +
0.24 jd
(e – 1)
1.54
Choose a value of Pe and find d from Eq. (i). Substitute in Eq. (ii) and find the magnitude Vt from Eq. (iii).
Vt versus Pe is plotted as shown in the Fig. 8.47(b) by Using MATLAB program given below.
(iii)
Synchronous Machines
Pe = 0 : 0.01 : 0.8
Vt = 1 + (0.24 /1.54). * (1.54 * Pe – 1);
Plot (Pe, Vt)
1.05
1
0.95
Vt
0.9
0.85
0.8
0
0.1
0.2
0.3
0.4
0.5
0.6
Load
Fig. 8.47(b)
V1 = 1 pu (controlled at this value)
(c)
Pe varied
Pe =
Vt VTH
1
sin d 1 =
sin d1
X TH
0.24
Vt = e jd1
Ia =
e jd1 - 1
j 0.24
E f = VTH + j 1.54 I a
= 1 + ( j 1.54)
(e jd1 - 1)
j 0.24
1.54 jd 1
– 1)
(e
0.24
256
=
Ef
11
=1+
If
=
256 È 1.54 jd1
˘
(3 - 1)˙
1+
11 ÍÎ 0.24
˚
0.7
0.8
491
492 Electric Machines
For If vs Pe plot MATLAB program is given as follows and its resultant plot is shown in the Fig. 8.47(c).
MATLAB Program to Plot If vs Pe
Pe=0: 0.01:0.8
d1=asin(0.24*Pe)
Ef=1+(1.54/0.24).*(exp(i*d1)-1)
If=(256/11).*Ef
plot(Pe,abs(If))
36
34
Excitation Current
32
30
28
26
24
22
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
Load
Fig. 8.47(c)
EXAMPLE 8.9 A 4 pole, 50 Hz, 24 kV, 600 MVA synchronous generator with a synchronous reactance of
1.8 pu is synchronized to a power system which can be represented by a Thevenin voltage of 24 kV in series
with Thevenin reactance of 0.24 pu on generator base. The generator voltage regulator adjusts the field
current to maintain its terminal voltage and 24 kV independent of load.
(a) The generator prime mover power is adjusted so that it feeds 400 MV.
(i) Draw the phasor diagram under this operating condition
(ii) Calculate the generator current and its power factor
(b) Repeat part (a) when the generator feeds 600 MW
(MVA)B = 600, (kV)B = 24, (MW)B = 600
The system diagram is drawn in Fig. 8.48(a).
Synchronous Machines
493
SOLUTION
(a) (i) As the generator is feeding power to the systems
1.8 pu
Vt leads VTH by angle d
VV
Pe = t TH sin d
0.24
400
Pe =
= 2/3 pu ,
600
2
1
=
sin d 1
3
0.24
Ia
0.24 pu
+
+
+
Vt = 1 pu
Ef
VTH = 1 pu
Vt = 1 pu,
–
–
–
or
d 1 = 9.2º
Fig. 8.48(a)
The phasor diagram is drawn in Fig. 4.48(b); d1 is not to scale
(i) From the phasor-diagram geometry
or
Ef
0.24 Ia = 2 ¥ 1 sin d1/2
2
Ia =
sin 9.2º/2 = 0.668 pu
0.24
(Ia)B =
Phase angle,
Check
(600 / 3) ¥ 106
= 14433 A
24
¥ 103
3
Ia = 0.668 ¥ 14433 = 9641 A
d
f=
= 4.6º lag
2
Power factor = cos 4.6º = 0.9968
Pe =
=
Observation
Pe =
3 kV Ia ¥ 10–3 cos f
Ia Xs
= 1.8 Ia
Vt
1 pu
d
f= 1
2
d1
Ia XTH
= 0.24Ia
Ia
1 pu
VTH
Fig. 8.48(b)
3 ¥ 24 ¥ 9.641 ¥ 0.9968 = 399.5 ª 400 MW
Vt VTH
sin d1 = Vt Ia cos f (in pu)
X TH
E f = Vt + j I a Xs
= 1 –0º + j 0.668 –– 4.6º ¥ 1.8 = 1 + 1.2 –85.4º = 1.622 –47.4º
or
Ef = 1.622
or
38.9 kV, d2 = 47.4º
Pe = 600 MW or Pe = 1 pu
1
1=
sin d 1, d 1 = 13.89º
0.24
(b)
2
sin d 1/2 = 1 pu
0.24
Ia = 14433 ¥ 1 = 14433 A
Ia =
or
Phase angle,
f = 13.89º/2 = 6.945º lag
Power factor = 0.993 lag
E f = 1–0º + j ¥ 1–– 6.95º ¥ 1.8 = 1 + 1.8 –83 .05º = 2.157 –55.6º
Ef = 2.157 pu
or
VTH
= 1 pu
Ef = 2.157 ¥ 24 = 31.77 kV, d2 = 55.6°
494 Electric Machines
Let us find power factor from Ef to VTH
Pe =
Pe =
or
E f VTH
X s + X TH
sin (d1 + d2)
2.157 ¥ 1
sin (13.89º + 55.6º) = 0.99 pu
1.8 + 0.24
It should be 1 pu. The difference is due to rounding off errors.
8.11
EFFICIENCY OF SYNCHRONOUS MACHINES
The losses in a synchronous machine have been dealt with elaborately in Section 8.4 while presenting OC and
SC test. These are summarized below:
OC Test
As it is no load test, the open circuit loss POC comprises two losses.
(i) Core loss, P(core, OC) (ii) windage and friction loss, Pwf . P(core, OC) is proportional to square of VOC
and Pwf is constant (machine speed is synchronous)
Pwf gets separately out during OC test by reducing the field current to zero, thereby making P(core) = 0.
It is to be noted that there is no armature ohmic loss in OC test.
SC Test
As the test is conduced at very much reduced field current and so P(core) is negligible. The components of
the PSC loss are
1. Armature copper loss I 2a Ra (dc, hot)
2. Stray load loss, Pst comprising stray core and armature teeth loss caused by leakage flux and stray
copper loss
3. Windage and friction loss
As Pwf is known from the OC test it can be subtracted from the total SC losses. The remaining loss is
Psc(load) loss. The stray loss is found as
Pst = PSC (load) – I a2 Ra (dc, hot)
Ra (dc) can be measured by a battery test and corrected for a temperature of 75°C.
Synchronous machine losses from OC and SC tests as separated out above are
P (core, OC)
Pwf constant
I a2 Ra (dc, hot), computed
Pst
Loss Measurement
OC and SC loss measurement can be carried out by measuring the mechanical input to the synchronous
generator during the tests. For this purpose, the generator is run at synchronous speed by a dynamometer
dc motor wherein the stator is free to rotate but is prevented by spring balances from the readings of which
Synchronous Machines
495
mechanical power input can be computed. The other convenient and accurate method is the torque meter
which measures the prime mover shaft torque calibrated in torque units.
Where such facilities are not available, a dc shunt motor of rating somewhat more than estimated machine
losses. Before coupling the dc motor to the synchronous machine the Swinburne’s test is performed on it at
synchronous speed to determine the rotational loss of the motor and also its armature resistances is measured.
The dc motor is now-coupled to the synchronous generator and the set driven at synchronous speed. The
motor terminal voltage and armature current is measured during the test.
The OC/SC loss is found as
Vt Ia – I a2 Ra – Prot = POC /PSC
The efficiency calculation are illustrated by a comprehensive example.
EXAMPLE 8.10 A 60 kVA, 400 V, 50 Hz synchronous generator is tested for by means OC and SC tests
whose data are given below:
Field current
OC voltage
(line)
SC current
(line)
SC loss
(3 phase)
2.85 A
1.21 A
400 V
––
108 A
3.95 kW
OC
SC (at 75°C)
The OC loss data are plotted in Fig. 8.49.
3.2
2.8
Losses, kW
2.4
2.0
1.6
1.2
OC loss
0.8
0.4
0
0
80
160
240
320
VOC (line) (V)
Fig. 8.49
400
480
560
496
Electric Machines
The armature (star connected) has dc resistance/phase at 25ºC of 0.075 W. The machine is operating at
full-load 0.8 pf lagging at rated terminal voltage with a field current of 3.1 A. The field resistance is 110 W
at 75ºC.
Calculate:
(a) Effective armature resistance and synchronous reactance (saturated).
(b) Full-load stray load loss.
(c) Ratio of effective armature resistance/dc resistance.
(d) Various category of losses at full-load (75ºC)
(e) Full-load efficiency at 75ºC
SOLUTION
(a) From SC test data
or
3 ¥ (108)2 ¥ Ra (eff ) = 3.95 ¥ 100
Ra (eff ) = 0.113 W
As SC current is linear function of field current
lSC (If = 2.85 A) = 108 ¥ 2.85/1.21 = 254 A
Zs = (400/ 3 )/254 = 0.91
Xs = [(0.91)2 – (0.113)2] 1/2 = 0.903 W (saturated)
Ê 75 + 273 ˆ
= 0.088 W
Ra (dc) (75ºC) = 0.075 ¥ Á
Ë 25 + 273 ˜¯
Ia (rated) = (60 ¥ 1000)/( 3 ¥ 400) = 86.6 A
(b)
SC loss at 86.6 A
3 I a2 Ra (dc)
Stray load loss
Ra (eff )/Ra (dc)
(c)
= 3.95 ¥ (86.6/108)2 = 2.54 kW
= 3 ¥ (86.6)2 ¥ 0.088 = 1.98 kW
= 2.54 – 1.98 = 0.56 kW
= 0.113/0.088 = 1.284
(d) From the OC loss curve of Fig. 8.49
Windage and friction loss (at zero excitation) = 0.9 kW
We will now find excitation emf on full-load
or
E f = (400/ 3 ) + 86.6 – – 36.9° ¥ j 0.903; Ignoring Ia R a voltage drop
Ef = 285 V or 494 V (line)
From the OC loss curve at
VOC
Core loss + windage and friction loss
Then core loss
Field copper loss (75°C)
= Ef = 494 V
= 2.44 kW
= 2.44 – 0.9 = 1.54 kW
= (3.l)2 ¥ 110 = 1.06 kW
Various losses on full load are summarised below:
Windage and friction loss = 0.9 kW
Core loss = 1.54 kW
Armature loss (in dc resistance) = 1.98 kW
Stray load loss = 0.56 kW
Field copper loss = 1.06 kW
Total loss = 6.04 kW
Synchronous Machines
497
Note: Losses in exciter* and field rheostat are not accounted against the machine.
Output = 60 ¥ 0.8 = 48 kW
Input = Output + losses = 48 + 6.04 = 54.04 kW
(e)
h = 1-
Efficiency,
6.04
= 88.8%
48 + 6.04
The flow of active and reactive power in a synchronous link will now be studied. The approach will be
analytical and armature resistance will be considered for generality of results. All quantities are per phase,
star connection
Figure 8.50(a) shows the schematic diagram of a synchronous generator wherein E f leads Vt by angle d.
The synchronous impedance** is
Z s = Ra + jXs = Zs –q
(8.61)
as shown by the impedance triangle of Fig. 8.50(b) wherein
Xs
q = tan–1
Ra
and
(8.62a)
a = 90º – q = tan–1
Ra
Xs
(8.62b)
The armature current in Fig. 8.50(a) can be expressed as
Ia =
E f –d - Vt –0∞
(8.63)
Z s –q
The complex power output is
Se = Pe + jQe = Vt –0º I a*
(8.64)
where
Pe = active power, W
Qe = reactive power, vars, positive for lagging pf and negative for leading pf
Q
Power factor = cos tan–1
P
Substituting for Ia from Eq. (8.63) in Eq. (8.64),
*
Ê E f –d - Vt –0 ˆ
Pe + jQe = Vt –0 Á
˜¯
Z –q
Ë
s
=
Vt E f
Zs
–(q – d ) –
Vt 2
–q
Zs
(8.65)
* In a practical arrangement field will be supplied from a dc exciter (coupled to machine shaft) through a rheostat
for adjusting the field current This field current (3.1 A) with field resistance of 110 W requires voltage at field
terminal of 341 V. So exciter voltage must be 400 V. The difference is dropped in the field rheostat.
** If a line is present as a part of the synchronous link, the line resistance and reactance will be included in Ra and
Xs respectively.
498 Electric Machines
S¢e = P¢e + jQ¢e
Xs
R
Se = Pe + jQe
a
Ia
a
+
+
Zs
Zs –q
Ef –d
Xs
Vt –0∞
–
q
–
Ra
Fig. 8.50(a)
Fig. 8.50(b)
Impedance triangle
Equating the real and imaginary parts of Eq. (8.65), the following expressions for real and reactive power
output are obtained as
Vt E f
Vt 2
cos q +
cos (q – d )
Zs
Zs
Vt E f
V2
Qe (out) = – t sin q +
sin (q – d )
Zs
Zs
The net mechanical power input to the machine is given by
Pe (out) = –
(8.66a)
(8.66b)
*
È
Ê E f –d - Vt –0 ˆ ˘
˙
Pm (in) = P¢e = Re Í Se¢ = E f –d Á
˜¯ ˙ ; Re = real of
Z s –q
Ë
Í
Î
˚
E 2f
Zs
cos q –
Vt E f
cos (d + q)
(8.67)
Zs
It is the mechanical power converted to electric power. It is convenient to express the above results in terms
of angle a defined in the impedance triangle of Fig. 8.50(b) (Eq. 8.62b). Equations (8.66a), (8.66b) and (8.67)
then modify as below
=
Qe(out) =
Pm (in) =
Pm(in) – Pe (out) =
Vt 2
Vt E f
R +
sin (d + a)
2 a
Zs
Zs
Vt E f
V2
– t 2 Xs +
cos (d + a)
Zs
Zs
E 2f
Vt E f
Ra +
sin (d – a)
2
Zs
Zs
I a2 Ra
Pe(out) = –
(8.68a)
(8.68b)
(8.69)
as the only loss is in resistance. This result can be proved by substituting Pm (in) and Pe (out) from
Eq. (8.67) and (8.68a) followed by several steps of manipulation and reference to the phasor diagram of
Fig. 8.46.
The real electrical power output, Pe, as per Eq. (8.68a) is plotted in Fig. 8.51 from which it is observed
that its maximum value is
Pe (out)| max = –
Vt 2 Ra
Z s2
+
Vt E f
Zs
, at d + a = 90°
(8.70)
Synchronous Machines
499
occurring at d = q, which defines the limit of steady-state stability. The machine will fall out of step for angle
d > q. Of course, q will be 90° if resistance is negligible in which case the stability limit will be at d = 90° as
already explained in Sec. 8.9.
Pe
Vt Ef
Zs
Pe (max)
Generator (Electrical
power output)
Vt2Ra
d
Zs2
d=q
Vt E f
Zs
a
Motor
(Electrical power input)
q + 2a
Fig. 8.51
At maximum electrical power output, the corresponding reactive power output is found from Eq. (8.68b)
by substituting d = q
Qe (out )| at Pe (out ), max = –
Vt 2
Z s2
Xs +
Vt E f
Zs
cos (d + a)
Vt 2
Xs ; as d + a = 90º
Z s2
As the Qe (out) is negative, it means the generator var output is leading i.e. the generator operates at
leading pf.
We can find Qe (out)max from Eq. (8.68 b). It occurs at d + a = 180º or d = 180º – a. Its value is
Qe (out) = –
or
Qe (out)max = –
Vt 2
Z s2
Xs -
Vt E f
Zs
Corresponding
Pe (out) = –
Vt 2
Ra ; insignificant value as Ra is very small
Z s2
As Qe (out)max is negative, the generator operates at very low, leading pf (close to 90º).
These result have no significance for generator operation as that not how a generator is operated.
It follows from Eq. (8.69) that
Pm (in)| max =
E 2f Ra
Z s2
+
Vt E f
Zs
; at d = 90º + a = q + 2a
(8.71)
500
Electric Machines
Since the angle d in Eq. (8.71) is more than q, the maximum mechanical power input (net) operation for a
generator lies in the unstable region.
Equations (8.68a), (8.68b), (8.69) and (8.70) simplify as below when armature resistance is neglected
Pe (out) =
Vt E f
Qe (out) = –
Xs
sin d
(8.72a)
Vt 2 Vt E f
+
cos d
Xs
Xs
Pm (in) = Pe (out) =
Vt E f
Xs
Pe (out)| max = Pm (in)| max =
(8.72b)
sin d
Vt E f
Xs
;
(8.72c)
at d = 90º
(8.72d)
Vt 2 Vt E f
+
cos d = 0
Xs
Xs
Ef cos d = Vt ; normal excitation (generator)
Qe (out) > 0
Ef cos d > Vt ; over-excited (generator)
Qe (out) < 0
Ef cos d < Vt ; under-excited (generator)
Qe (out) = –
For unity pf
or
For lagging pf
or
For leading pf
or
These results can be elloborated with the phasor diagram of Fig. 8.38(a)
Motor Operation
Figure 8.52 shows the operation of the synchronous machine as a motor*. Here the angle d by which Ef lags
Vt is defined as positive. Also the direction of power flow is now into the machine while the mechanical power
flows out at the machine.
It then follows from Eqs (8.68a), (8.68b) and (8.69) by changing the sign of d that for motoring operation
Pe (in) = – Pe (out)
Vt 2
Pe¢ = Pm
Vt E f
Ra +
sin (d – a)
Zs
Z s2
Qe (in) = – Qe (out)
=
=
Vt 2
Xs -
Vt E f
Z s2
Pm (out, gross) = –Pm (in)
=–
E 2f
Z s2
Ra +
Zs
Vt E f
Zs
Xs
Ra
(8.73a)
Ia
+
+
Zs –q
Et ––d
cos (d – a)
Pe, Qe
(8.73b)
Vt –0
–
–
sin (d + a)
(8.74)
Fig. 8.52
* The motoring operation can be analyzed by the generator power flow equation except that d will have a negative
sign and active and reactive powers will be negative of those obtained from Eqs (8.68a); (8.68b) and (8.69).
Synchronous Machines
501
The maximum mechanical power output from Eq. (8.74) is given by
Pm (out)| max = –
E 2f
Z s2
Ra +
Vt E f
Zs
(8.75)
It occurs at d = q which defines the limit of steady-state stability. It is easily seen from Eq. (8.73a) that the
maximum electrical power input occurs at d = q + 2a which lies outside the stability limit. The reader should
compare these results with that of the generator.
Qe (in)max =
Vt 2
Z s2
Xs +
(d – a) = 180º
Occurs at
Pe (in) =
Vt E f
Zs
d = 180º – a
or
Vt 2
Ra : very small
Z s2
As Qe is positive and Pe is very small, the motor operate a low lagging pf (close to 90º). The motor acts
as an inductor.
For the case of negligible resistance
Corresponding
Pe (in) =
Qe (in) =
Vt E f
sin d
(8.76a)
Vt 2 Vt E f
cos d
Xs
Xs
(8.76b)
Xs
Pm (out) = Pe (in) =
Vt E f sin d
Xs
(8.76c)
Unity pf
Qe = 0 fi Ef cos d = Vt ; normal excitation (motor)
Lagging pf
Qe > 0 fi Ef cos d < Vt ; under-excited (motor)
Leading pf
Qe < 0 fi Ef cos d > Vt ; over-excited (motor)
These results can be verified from the phasor diagram of Fig. 8.38 (b).
Conditions for Power Factor, Ra Accounted
These could be arrived at by the sign Qe from Eq. (8.68 b) for generator and from Eq. (8.73 b) for motor.
However, simple from of these conditions are found from the phasor diagrams of Fig. 8.46. These phasor
diagrams are redrawn in Fig. 8.53(a) for generator with lagging power factor and in Fig. 8.53(b) for motor
with leading power factor with some projections shown in dotted line.
Generator
AF = Ef cos d
AE = Vt + Ia R a cos f
From the phasor diagram geometry
AF > AE
502 Electric Machines
Ef cos d > (Vt + Ia R a cos f)
Ef cos d – Ia R a cos f > Vt : over-excited, lagging pf
or
or
(i)
Other two conditions are
Ef cos d – Ia R a cos f = Vt ; normal excitation, unity pf
Ef cos d – Ia R a cos f < Vt : under-excitation, leading pf
Motor
(ii)
(iii)
The pf condition that follow similarly from Fig. 8.52(b) are
Ef cos d + Ia Ra cos f > Vt ; over-excited, leading pf
Ef cos d + Ia Ra cos f = Vt ; normal excitation, unity pf
Ef cos d + IaRa cos f < Vt ; under-excitation, lagging pf
(i)
(ii)
(iii)
Ef D
d
A
B
f
Vt
IaRa
E
F
C
Ia
(a) Generator, lagging pf
Ia
Vt
f
A
E
d
C
B
F
IaRa
IaXa
Ef
(b) Motor, leading pf
Fig. 8.53
F
Determination of power factor conditions
EXAMPLE 8.11 A 400 V, 3-phase, delta-connected synchronous motor has an excitation emf of 600 V and
synchronous impedance per phase of 0.3 + j6 W. Calculate the net power output, efficiency, line current and
power factor when the machine is developing maximum mechanical power (gross). Windage, friction and
core losses may be assumed to be 2.4 kW.
SOLUTION
Zs (eq. star) =
1
(0.3 + j 6) = 0.1 + j2 = 2 –87.14°
3
Synchronous Machines
503
For maximum mechanical power output
d = q = 87.14° (Ef lags Vt)
Vt = 400/ 3 = 230.9 V
Ef = 600/ 3 = 346.4 V
From the phasor diagram of Fig. 8.54
IaZs =
IaZs =
\
Vt2 + E 2f - 2Vt E f cosq
2
2
(230.9) + (346.4) - 2 ¥ 230.9 ¥ 346.4 ¥ cos 87.14∞
= 406.6 V
Ia = 203.3 A
d=q
Ef
2
2
(230.9) + (406.6) - (346.4)
2 ¥ 230.9 ¥ 406.6
or
b = 58.31°
From the geometry of the phasor diagram
Further cos b =
Vt
f
90° – 87.14° = 2.86° = a
b
Ia
Ia Zs
Ia Xs
2
q
Ia Ra
Fig. 8.54
f
pf
Pe (in)
Ia2 R a = (203.3)2 ¥ 0.1
= 90° – 58.31° – 2.86° = +28.8°
= cos f = 0.876 lag
= 230.9 ¥ 203.3 ¥ 0.876 = 41.12 kW
= 4.13 kW, (stray load loss is included as Ra is effective value)
2.4
Pwf + Pcore =
= 0.8 kW
3
Pm (out)|net = 41.12 – 4.13 – 0.8 = 36.19 kW or 108.57 kW (3-phase)
36.59
h=
= 89%
41.12
Note: Field copper loss is not accounted for here.
EXAMPLE 8.12 A 3300 V, star-connected synchronous motor is operating at constant terminal voltage
and constant excitation. Its synchronous impedance is 0.8 + j5 W. It operates at a power factor of
0.8 leading when drawing 800 kW from the mains. Find its power factor when the input is increased to
1200 kW, excitation remaining constant.
SOLUTION
Figure 8.55 gives the circuit model of the motor
\
Vt
Pe (in)
cos f
Qe (in)
= 3300/ 3 = 1905 V
= 800/3 = 266.7 kW
= 0.8 leading
P¢e = Pm
a = 9º
Zs = 0.8 + j 5 = 5.06–81º ;
0.8 W
+
(per phase)
Ef ––d
Vt –0
–
–
Note that Qe (in) is negative because the power factor is leading.
From Eqs (8.73a) and (8.73b)
Qe (in) =
Pe , Qe
Ia
+
= – Pe (in) tan f = –200 kVAR
Pe (in) =
5W
Vt2
Z s2
Vt2
Z s2
Ra +
Xs -
Vt E f
Zs
E f Vt
Zs
Fig. 8.55
sin (d – a)
(i)
cos (d – a)
(ii)
504
Electric Machines
Substituting the values
266.7 ¥ 1000 =
– 200 ¥ 1000 =
(1905) 2 ¥ 0.8
(5.06) 2
(1905) 2 ¥ 5
(5.06)
2
-
+
(1905) E f
5.06
(1905) E f
5.06
sin (d – a)
cos (d – a)
These equations simplify as
Ef sin (d – a) = 407.2
Ef cos (d – a) = 2413.6
from which Ef is obtained as
Ef = 2447.7 V
Under new operating conditions
Pe (in) = 1200/3 = 400 kW
Substituting in (i),
400 ¥ 1000 =
(1905) 2 ¥ 0.8
(5.06) 2
+
(1905) ¥ (2447.7)
sin (d – a)
5.06
or
sin (d – a) = 0.31 or d – a = 18º
Now Eq. (ii) is used to find Qe for (d – a) = 18°
Qe =
(1905) 2 ¥ 5
(5.06) 2
-
1905 ¥ 2447.7
cos 18º
5.06
= – 167.7 kVAR
Power factor = cos (tan–1Qe /Pe)
= cos (tan–1167.7/400) = 0.92 leading
EXAMPLE 8.13 A three phase 10 kVA, 400 V, 4-pole, 50 Hz star connected synchronous machine has
synchronous reactance of 16 W and negligible resistance. The machine is operating as generator on 400 V
bus-bars (assumed infinite).
(a) Determine the excitation emf (phase) and torque angle when the machine is delivering rated kVA at
0.8 pf lagging.
(b) While supplying the same real power as in part (a), the machine excitation is raised by 20%. Find the
stator current, power factor and torque angle.
(c) With the field current held constant as in part (a), the power (real) load is increased till the steadystate power limit is reached. Calculate the maximum power and kVAR delivered and also the stator
current and power factor. Draw the phasor diagram under these conditions.
SOLUTION
Xs
The circuit equivalent of the machine is drawn in Fig. 8.56.
3
Ia =
(a)
pf angle,
10 ¥ 10
=14.43 A
3 ¥ 400
f = cos–1 0.8 = 36.9° lag
I a = 14.43 – – 36.9°
400
Vt =
= 231 V
3
+
+ jI X –
a s
Ia
+
Vt –0∞
Ef –d
–
–
Fig. 8.56
Synchronous Machines
505
From the circuit equivalent E f = 231–0º + j 14.43 – – 36.9° ¥ 16
= 231 + 231–53.1° = 369.7 + j 184.7
or
E f = 413.3–26.5°
Torque angle,
d = 26.5°, Ef leads Vt (generating action)
(b) Power supplied (source),
Pe = 10 ¥ 0.8 = 8 kW (3 phase)
Ef (20% more) = 413.3 ¥ 1.2 = 496 V
E f Vt
Pe =
Torque angle,
From the circuit equivalent
Xs
sin d
8 ¥ 103
496 ¥ 231
=
sin d
3
16
d = 21.9°
Ia =
=
E f –d - Vt –0∞
j Xs
=
496 – 21.9∞ - 231
j16
229 + j185
= 11.6 – j 14.3 = 18.4– – 50.9º
j16
Ia = 18.4 A,
pf = cos 50.9° = 0.63 lagging
(c) Ef = 413 V ; field current same as in part (a)
Pe (max) =
=
Ia =
E f Vt
Xs
; d = 90°
413 ¥ 231
¥ 10–3 = 5.96 kW/phase or 17.38 kW, 3-phase
16
413–90∞ - 231
= 25.8 + j 14.43
j16
= 29.56 –29.2º A
Ia = 29.56 A, pf = cos 29.2° = 0.873 leading
Ia Xs
Ia = 29.56 A
Ef = 413 V
The phasor diagram is drawn in Fig. 8.57
kVAR delivered (negative)
or
EXAMPLE 8.14
Qe
= tan – 29.2
Pe
Qe = 8 ¥ 0.559 = – 4.47 kVAR
Vt = 231 V
Fig. 8.57
The synchronous machine of Example 8.13 is acting as a motor.
(a) The motor carries a shaft load of 8 kW and its rotational loss is 0.5 kW. Its excitation emf is adjusted
to 750 V (line). Calculate its armature current, power factor and power angle. Also calculate the
developed and shaft torques.
(b) The motor is running at no load and its losses can be neglected. Calculate its armature and power
factor at excitation emf (line) of (i) 600 V, and (ii) 300 V. Calculate also the kVAR drawn in each case.
(c) The motor is on no load (losses to be ignored). What should be its excitation for it to draw a leading
kVAR of 6? Draw the phasor diagram.
506
Electric Machines
SOLUTION
Pm(gross) = 8 + 0.5 = 8.5 kW
Pe (in) = Pm(gross), as Ra = 0
(a)
Ef =
Pe (in) =
750
= 433 V,
3
E f Vt
Xs
Vt = 231 V (data Example 8.13)
sin d
433 ¥ 231
8.5 ¥ 103
sin d
=
16
3
d = 27º, Ef lags Vt
E f = 433 – – 27º V, Vt = 231–0º V
In a motor current Ia will flow in Fig. 8.56.
Ia =
231–0∞ - 433– - 27∞
= 15.6–38º A
j16
Ia = 15.6 A,
Synchronous speed,
ns =
Torque developed =
Shaft torque =
pf = cos 38.1º = 0.787 leading
120 ¥ 50
= 1500 rpm or 157.1 rad/s
4
8.5 ¥ 103
= 54.1 Nm
157.1
8 ¥ 103
= 50.9 Nm
157.1
(b) Motor running no load, no loss
d =0
(i)
Ef =
600
= 346.4 V,
3
Ia =
231 - 346.4
= j 7.213 A leading Vt by 90º
j16
Vt = 231 V
3 Vt I a* = 3 ¥ 231 ¥ (– j 7.213) ¥ 10–3
= – j 5 ; minus sign is for leading kVAR
The motor acts as a capacitor
7.213
1
¥
C=
= 99.4 mF
231
2p ¥ 50
kVAR drawn
Ef = 300/ 3 = 173.2 V, Vt = 231 V
(ii)
Ia =
kVA drawn
231 - 173.2
= j 3.61 A lagging Vt by 90°
j16
3 Vt I a* = 3 ¥ 231 ¥ ( j 3.61) ¥ 10–3
= j 2.5; plus sign means for lagging kVA
The motor acts as inductor
L=
231
1
◊
= 294.1 mH
2.5 2p ¥ 50
Synchronous Machines
507
3 Vt I a* ¥ 10–3 = – j 6, Vt = 231
(c)
2000
= – j 8.66
231
= j 8.66 A (leading Vt by 90º)
I a* = – j
Ia
E f = Vt – j I a Xs = 231 – j ( j 8.66) ¥ 16 = (231 + 138.6) –0°
E f = 369.6 –0ºV, Ef = 640 V (line)
The phasor diagram is drawn in Fig. 8.58.
Ia = j8.66 A
–
j IaXs = – j138.6 V
Vt –0∞
0
Fig. 8.58
Ef –0∞
Phasor diagram
EXAMPLE 8.15 A 6-pole, 3-phase, 4 MVA, 50 Hz, star connected synchronous motor is supplied from
6.6 kV bus-bars. It has a synchronous reactance of 4.8 W:
(a) The motor is operating at a power angle of 20º at rated current. Find the excitation emf if the power
factor is lagging leading.
(b) In part (a) find the mechanical power developed and power factor in each case.
SOLUTION
Vt =
6.6
= 3.81 kV
3
Ia (rated) =
4 ¥ 103
= 350 A
3 ¥ 6.6
Ia
Ia (rated) Xs = 350 ¥ 4.8
= 1680 V or 1.68 kV
(a) In the motor
Ef lags Vt by d = 20°
q2 = ?
A
E f = Vt –0º – j I a Xs
The phasor diagram is drawn in Fig. 8.59.
As Ia Xs = 1.68 kV constant, there are two possible solutions.
From the geometry of triangle ABD
q1
Vt = 3.81 kV
Ef
1
IaXs = 1.68 kV
B
V t2 + E 2f – 2 Vt Ef cos d = (Ia Xs)2
Substituting values
Ef
2
(3.81) +
or
Its solutions are
D
d = 20°
E 2f –
E 2f –
Ef 1
2 ¥ 3.81 Ef cos 20º = (1.68)
7.16 Ef + 11.69 = 0
= 2.52 kV, Ef 2 = 4.64 kV
2
2
Ia
Fig. 8.59
Phasor diagram
508
Electric Machines
or
Ef 1 = 4.36 kV (line),
Ef 2 = 8.04 kV (line)
(b) Ef 1 = 2.52 kV ; motor under-excited case, pf lagging
Ê 3.81 ¥ 2.52 ˆ
Pm = 3 ¥ Á
˜¯ sin 20° = 2.052 MW
Ë
4.8
3 ¥ 6.6 ¥ 350 cos q1 = 2.052 ¥ 103
pf 1 = cos q1 = 0.513 lagging
Ef 2 = 4.64 kV ; motor over-excited pf leading
Ê 3.81 ¥ 4.64 ˆ
Pm = 3 ¥ Á
˜¯ sin 20º = 3.779 MW
Ë
4.8
3 ¥ 6.6 ¥ 350 cos q2 = 3.779 ¥ 103
pf 2 = cos q2 = 0.944 leading; q2 = 19.2°
EXAMPLE 8.16 A synchronous motor is drawing 50 A from 400 V, three-phase supply at unity pf with a
field current of 0.9 A. The synchronous reactance of the motor is 1.3 W.
(a) Find the power angle.
(b) With the mechanical load remaining constant, find the value of the field current which would result in
0.8 leading power factor. Assume linear magnetization.
SOLUTION
We proceed on per basis, star connection
400
= 231–0 A
3
pf = unity, pf angle, q = 0°, I a = 50–0°
Vt =
(a)
E f = Vt – j I a Xs = 231 – j 50 ¥ 1.3 = 231 – j 65 = 240–– 15.7° V
d = 15.7°, Ef lags Vt (motor)
Power angle,
(b) Mechanical load in part (a)
Pm = Vt Ia cos q = 231 ¥ 50 ¥ 1 = 11550 W
It remains constant. As there is no ohmic loss
Pe = Pm = Vt Ia cos q
11550 = 231 ¥ Ia ¥ 0.8
or
Ia
Ia = 62.5 A
Ia Xs = 62.5 ¥ 1.3 = 81.25 V
q = cos–1 0.8 = 36.9° leading
The phasor diagram is drawn in Fig. 8.60.
E f2 = (Vt cos q)2 + (Vt sin q + Ia Xs)2
= (184.8) 2 + (219.85)2
or
Ef = 287.2 V
On linear magnetization basis
If = 0.9 ¥
287.2
= 1.077 A
240
q
Vt
d
IaXa
Ef
Fig. 8.60
Synchronous Machines
509
EXAMPLE 8.17 A 40 kVA, 600 V star-connected synchronous motor has armature effective resistance of
0.8 W and synchronous reactance of 8 W It has stray loss of 2 kW.
The motor is operating at 600 V bus-bar while supplying a shaft load of 30 kW, it is drawing rated
current at leading pf.
(a) Calculate the motor efficiency.
(b) What is its excitation emf and power angle?
(c) With this excitation calculate the maximum power output (gross) and corresponding net output and
the po