國立交通大學機構典藏

國立交通大學機構典藏
國
立
交
通
大
學
電信工程學系
碩士論文
超寬頻射頻傳收機模組
Ultra Wide Band RF Transceiver Module
研究生:江俊賢
指導教授:張志揚
中華民國
九十四
年
博士
六
月
超寬頻射頻傳收機模組
Ultra Wide Band RF Transceiver Module
研 究 生:江俊賢
指導教授:張志揚 博士
Student:Chin-Hsien Chiang
Advisor:Dr. Chi-Yang Chang
國立交通大學
電信工程學系
碩士論文
A Thesis Submitted to Institute of
Communication Engineering
College of Electrical Engineering and Computer Science
National Chiao Tung University
In Partial Fulfillment of the Requirements
for the Degree of Master of Science
In
Communication Engineering
June 2005
Hsinchu, Taiwan, Republic of China
中華民國 九十四 年 六 月
超寬頻射頻傳收機模組
研究生:江俊賢
指導教授:張志揚 博士
國立交通大學電信工程學系
摘要
軍用無線通信系統有別於商用無線通信系統,在使用情景上需
具靈活性、機動性與強韌性。本論文中將介紹超寬頻的射頻傳收機
模組,以完成 UWB 之通訊機。其中包括下列各個微波關鍵性電路,
包含:UWB 射頻濾波器、UWB 功率放大器、UWB 低雜訊放大器、頻率
合成器、寬頻升降頻混頻器、寬頻微波開關、及微波功率偵測器等。
首先,我們將介紹一般的無線通訊系統,並對傳送路徑與接收路徑
分開討論。最後,再實作所有主要的電路元件實作並整合成電路模
組。
I
Ultra Wide Band RF Transceiver Module
Student: Chin-Hsien Chiang
Advisor: Dr. Chi-Yang Chang
Department of Communication Engineering
National Chiao-Tung University
Abstract
Military wireless communication systems are fundamentally
different from commercial ones; they must offer flexibility, mobility,
and robustess in the applied scenarios. In this thesis, a new architecture
of UWB RF T/R module will be proposed to realize the target UWB
communication unit. There are several RF key components which
include UWB RF filter, UWB power amplifier, UWB low noise
amplifier, frequency synthesizer, broadband up/down mixer, broadband
microwave switch, broadband microwave detector, etc. We will
introduce general wireless systems and separate into transmit path and
receive path to analysis. Finally, implement the whole RF components
and combine them.
II
Acknowledgment
誌謝
首先要感謝指導教授張志揚博士在兩年的時間中辛勤地指導與鼓
勵,老師在微波電路豐富的知識與經驗使學生在學業上獲益良多。同時要
感謝口試委員邱煥凱教授、楊正任教授及林育德教授的不吝指導得以使得
此篇論文更為完善。
感謝陳慧諄學長對我的指導,雖然我的程度不好,但還是肯一步一步
的教我,對我的問題耐心的回答,還教了我很多在實作電路上的技巧。謝
謝金雄學長在我心情不好的時候陪我聊天,平時沒事時陪我說些五四三,
當然也有教我關於濾波器的學問。謝謝實驗室的同學子閔,不管如何,真
的很謝謝你,從碩一修課的作業到陪我打球打屁,還有其他課業以外的幫
忙。謝謝志偉同學,雖然平常沒幫上你什麼,但還是謝謝你。謝謝秀琴同
學,平常一些瑣事都麻煩你。謝謝思嫻~~~~。
最後特別感謝我那籃球挑輸我,但一直都很罩我的弟弟,還有他的小
學同學,論文的完成真的麻煩你們了,想不到我也要畢業了。
III
Contents
Abstract (Chinese) ------------------------------------------------------------------Ⅰ
Abstract -------------------------------------------------------------------------------Ⅱ
Acknowledgment --------------------------------------------------------------------Ⅲ
Contents ------------------------------------------------------------------------------Ⅳ
List of Figures ------------------------------------------------------------------------Ⅵ
Chapter 1 Introduction -------------------------------------------------------------1
1.1 Motivations
1
1.2 Objectives and Approaches
1
1.2.1 Project Background
1.2.2 UWB RF Module
1
2
1.3 Chapter Outline
3
Chapter 2 RF System Theory and Analysis --------------------------------------5
2.1 Introduction
5
2.2 Receiver System Considerations
5
2.2.1 Receiver Noise
7
2.2.2 Dynamic Range
8
2.2.3 Third-order Intermodulation
2.3 Transmitter System Considerations
2.3.1 Transmitter Noise
2.3.2 Adjacent Channel Power Ratio
2.4 Analysis of the System
9
11
13
14
14
2.4.1 Frequency Conversion
14
2.4.2 Stages Tracking
16
2.4.3 Practical Design
18
Chapter 3 Implementation and Measurement --------------------------------21
IV
3.1 Introduction
21
3.2 RF Bandpass Filters
21
3.2.1 3.3~3.6GHz Bandpass Filter
21
3.2.2 0.9~1.2GHz Bandpass Filter
23
3.3 Broadband Voltage Controlled Oscillator
25
3.4 Broadband Mixer
28
3.5 Broadband Low Noise Amplifier
31
3.6 Broadband Power Amplifier
32
3.7 Frequency Synthesizer
34
3.8 Measurement and Discussion
3.8.1 Transmit Path
42
43
3.8.2 Receive Path
45
3.8.3 Measurements
47
Chapter 4 Conclusion --------------------------------------------------------------49
References ---------------------------------------------------------------------------50
V
List of Figures
Figure 1.1 UWB RF transceiver system block diagram. --------------------------------3
Figure 2.1 (a) Transmitter and (b) receiver front ends of a wireless transceiver. ----5
Figure 2.2 Block diagram of a basic radio receiver. -------------------------------------6
Figure 2.3 Determining the noise figure of a noisy network. ---------------------------8
Figure 2.4 Cascaded noisy circuit with n networks. -------------------------------------8
Figure 2.5 Illustrating the dynamic range of realistic mixers, amplifiers, or receivers.
-------------------------------------------------------------------------------------9
Figure 2.6 Intermodulation in a nonlinear system. -------------------------------------10
Figure 2.7 Growth of output components in an intermodulation test. ---------------10
Figure 2.8 Cascaded n nonlinear stages. -------------------------------------------------10
Figure 2.9 Block diagram of a basic radio transmitter. --------------------------------11
Figure 2.10 Ideal signal and noisy signal. ----------------------------------------------13
Figure 2.11 Oscillator output power spectrum. ----------------------------------------14
Figure 2.12 Adjacent channel power. ----------------------------------------------------14
Figure 2.13 Problem of image. -----------------------------------------------------------15
Figure 2.14 The cascaded gain, noise figure, and IIP3 are plotted in a receive path.
--------------------------------------------------------------------------------17
Figure 2.15 The cascaded gain, P1dB, and IIP3 are plotted in a transmit path. ---17
Figure 2.16 Block diagram of the RF transceiver. -------------------------------------18
Figure 2.17 Design procedure of RF transceiver module. ----------------------------20
Figure 3.1 Block diagram of the RF transceiver. ---------------------------------------21
Figure 3.2 3.3~3.6 GHz bandpass filter simulation.
(a) Circuit configuration. ----------------------------------------------------22
(b) Simulated return loss and insertion loss. ------------------------------22
Figure 3.3 3.3~3.6 GHz bandpass filter physical implementation.
VI
(a) Circuit layout. -------------------------------------------------------------23
(b) Measured return loss and insertion loss. ------------------------------23
Figure 3.4 0.9~1.2 GHz bandpass filter simulation.
(a) Circuit configuration. ----------------------------------------------------24
(b) Simulated return loss and insertion loss. ------------------------------24
Figure 3.5 0.9~1.2 GHz bandpass filter physical implementation.
(a) Circuit layout. -------------------------------------------------------------24
(b) Measured return loss and insertion loss. ------------------------------25
Figure 3.6 Photograph of broadband voltage controlled oscillator. ------------------26
Figure 3.7 Controlled voltage tuning range. --------------------------------------------26
Figure 3.8 Measured phase noise.
(a) @ 2 MHz offset. ----------------------------------------------------------27
(b) @ 1 MHz offset. ----------------------------------------------------------27
(c) @ 500 kHz offset. --------------------------------------------------------28
Figure 3.9 Measured power of 2-order harmonic frequency. -------------------------28
Figure 3.10 Photograph of broadband mixer. ------------------------------------------29
Figure 3.11 Conversion loss V.S. LO power @ IF= 2.4 GHz, -10 dBm. -----------29
Figure 3.12 Conversion loss V.S. LO freq. @ IF= 2.4 GHz, -10 dBm. -------------30
Figure 3.13 Conversion loss V.S. LO freq. @ IF=2.5 GHz, -10 dBm. --------------30
Figure 3.14 Different freq. of IF to RF leakage @ IF= -10 dBm. -------------------31
Figure 3.15 Different power of IF to RF leakage @ IF= 2.4 GHz. ------------------31
Figure 3.16 Photograph of broadband low noise amplifier. --------------------------32
Figure 3.17 Measured S-parameters of broadband low noise amplifier. -----------32
Figure 3.18 Measured noise figure of broadband low noise amplifier. -------------32
Figure 3.19 Photograph of broadband power amplifier. ------------------------------33
Figure 3.20 Measured S-parameters of broadband power amplifier. ----------------34
VII
Figure 3.21 Measured P1dB of broadband power amplifier. -------------------------34
Figure 3.22 Photograph of frequency synthesizer. -------------------------------------35
Figure 3.23 Codeloader interface. -------------------------------------------------------35
Figure 3.24 Controlled voltage tuning range. ------------------------------------------36
Figure 3.25 Measured phase noise at 500 kHz offset.
(a) @ 3.3 GHz. --------------------------------------------------------------37
(b) @ 3.4 GHz. --------------------------------------------------------------37
(c) @ 3.6 GHz. --------------------------------------------------------------38
Figure 3.26 Measured power spectrum at 3.3 GHz.
(a) Phase noise with 100 kHz offset. -------------------------------------39
(b) 2-order harmonic power spectrum. -----------------------------------39
(c) 3-order harmonic power spectrum. -----------------------------------39
Figure 3.27 Measured power spectrum at 3.4 GHz.
(a) Phase noise with 100 kHz offset. -------------------------------------40
(b) 2-order harmonic power spectrum. -----------------------------------40
(c) 3-order harmonic power spectrum. -----------------------------------41
Figure 3.28 Measured power spectrum at 3.6 GHz.
(a) Phase noise with 100 kHz offset. -------------------------------------41
(b) 2-order harmonic power spectrum. -----------------------------------42
(c) 3-order harmonic power spectrum. -----------------------------------42
Figure 3.29 Photograph of transceiver module. ----------------------------------------42
Figure 3.30 Photograph of transmit path. -----------------------------------------------43
Figure 3.31 Measured S-parameters of transmit path. --------------------------------43
Figure 3.32 Transmit testing setup diagram. -------------------------------------------44
Figure 3.33 Transmit testing on spectrum analyzer.
(a) Power spectrum of transmitted carrier. ------------------------------45
VIII
(b) Power spectrum of transmitted packets. -----------------------------45
Figure 3.34 Photograph of receive path. ------------------------------------------------45
Figure 3.35 Measured S-parameters of receive path. ---------------------------------46
Figure 3.36 Receive testing setup diagram. --------------------------------------------46
Figure 3.37 The diagnostic program (MFP) interface. --------------------------------47
Figure 3.38 Photograph of transceiver module. ----------------------------------------47
IX
Chapter1 Introduction
1.1 Motivations
In the last twenty years, we have witnessed wireless communications with
dramatic down-scaling and price decreasing. This evolution has been enabled by
significant advances in radio and integrated circuit techniques. For example,
time-division or code-division multiple access enabled by modern digital signal
processing, together with vary large scale integrated circuit (VLSI) increased
significantly radio capacity and brought the radio costs down to the consumer level.
To accommodate different bandwidth signals, a multi-standard receiver/transmitter
must have a bandwidth equal to the largest one. In another words, the
receiver/transmitter has to be wideband.
1.2 Objectives and Approaches
1.2.1 Project Background
Military wireless communication systems are fundamentally different from
commercial ones; they must offer flexibility, mobility, and robustess in the applied
scenarios. Since fixed wireless stations cannot function effectively in an unpredictable
battlefield, military communications must reply on wireless ad-hoc technologies to
reply information. On the other hand, it is equally important to implement a system
that can collect, transfer, and distribute information effectively and quickly in order to
meet the demand of future “digital battlefield”, in which a great deal of message and
actions are to be shared among users.
According to above view, our proposal is designed to achieve the following
goals:
1. To meet the communication requirements of mobile military operation by means
of ad-hoc networking;
1
2. Design communication equipments for digital warfighters based on the UWB
technique to provide high-speed, secure, and robust data links;
3. Employee an SDR platform to provide flexible communication ability to
accommodate the demand of varying battlefield scenarios. More effective
utilization of system resources should be achieved via adaptive parameter
selection.
In this year, five modules which complete the system will be planned, designed
or implemented. They are given as following:
1. UWB antenna module
2. UWB RF module
3. UWB baseband module
4. Ad-Hoc wireless communication networks module
5. SDR module, each module will be planned.
1.2.2 UWB RF Module
UWB (Ultra Wide Band) was originally developed to serve for the military
purpose, and was applied in military radar the earliest time. However, the huge
commercial benefits that followed stimulate FCC (Federal Communication
Commission) to allow their people to use it. They even set 3.1~10.6GHz as the
frequency band. This decision brought about an extremely agitation in applying UWB
in the commercial wireless communication field. Many electric factories and stores
and computer companies come in great numbers to proceed to develop different kinds
of products.
After saying the popular and importance of UWB, you may ask what UWB
means? UWB is different from traditional narrow band and broadband, and it has
wider frequency band. Narrowband means the relative bandwidth is under 1%. On the
other hand, if the relative bandwidth is between 1% to 25%, we call it broadband. If
2
the relative bandwidth is over 25% and the central frequency is over 500MHz, it
would be Ultra Wide Band.
Fig. 1.1 is the first structure draft of UWB RF transceiver system block diagram.
As we can see in Fig. 1.1, it uses a broadband frequency synthesizer between 3.3 to
3.6GHz to produce local oscillated signal, and then sends it into the broadband
upconvert/downconvert mixer. When it sends to the receiver, it will raise 900 to
1200MHz of receiving signal to 2.4GHz, and then sends it into the 2.4GHz
receive/transmit chip. As for the transmitter, it will downconvert the frequency, which
is from the 2.4GHz to 900~1200MHz, and then sends it into the power amplifier to
enlarge the amplitude and transmits from the antenna.
Figure 1.1 UWB RF transceiver system block diagram.
1.3 Chapter Outline
There are four chapters in this thesis. Chapter one is an introduction. We will talk
about the research background first, and introduce the structure and standards of
UWB RF transceiver system. Chapter two will include the basic theory of wireless
systems and the parameter discussion that needed to be take into consider when
receiver and transmitter are independent in the wireless systems. After this, we will
3
introduce UWB RF T/R module designing procedures and circuit analysis. In chapter
three, we will show the fabrication of real circuits and measuring outcome, and have a
further discussion according to the outcome. Chapter four will be a conclusion of this
thesis.
4
Chapter2 RF System Theory and Analysis
2.1 Introduction
Any wireless system consists of a transmitter and a receiver. The transmitter
delivers the carrier signal modulated by information through an antenna. The receiver
recovers the information from the received signal from the antenna. Besides, as
depicted in Fig. 2.1 [1], the transmitter must employ narrowband modulation,
amplification, and filtering to avoid leakage to adjacent channels, and the receiver
must be able to process the desired channel while sufficiently rejecting strong
neighboring interferers. And, more detail parameters used in transmitter and receiver
systems will discuss as follow.
Power
amplifier
Transmitted
Channel
Bandpass
filter
w
Adjacent
Channels
(a)
I nterferers
BandpassLow noise
filter
amplifier
w
Desired
Channel
(b)
Figure 2.1 (a) Transmitter and (b) receiver front ends of a wireless transceiver.
2.2 Receiver System Considerations
The input to a wireless transmitter may be voice, video, data, or other
information to be transmitted to one or more distant receivers. So, the basic function
of the receiver is to demodulate, or decode, the transmitted baseband data. The
performance of the receiver depends on the system design, circuit design, and
5
working environment. To facilitate the discussion, a basic radio receiver as shown in
Fig. 2.2 [2] is used.
Antenna
Bandpass Low noise
filter
amplifier Mixer
I F filter
I F Demodulator
amplifier
Data
out
Local
oscillator
Figure 2.2 Block diagram of a basic radio receiver.
The antenna receives electromagnetic waves radiated from many sources over a
relatively broad frequency range. A preselector bandpass filter can minimize the
intermodulation and spurious responses by filtering out received signals at undesired
frequencies. The bandpass filter is followed by a low noise amplifier, which has a low
noise figure, high gain, and high intercept point, can amplify the possibly very weak
received signal and minimize the noise power that is added to the received signal.
Next, a mixer is used to downconvert the received RF signal to a low frequency signal
called intermediate frequency (IF). When IF signal is selected by a IF filter, a high
gain IF amplifier raises the power level of the signal so that the baseband information
can be recovered easily. This process is called demodulation. A local oscillator
provides a LO source which should have low phase noise and sufficient power to
pump the mixer. The receiver system considerations are listed below [3]:
1. Sensitivity: Receiver sensitivity quantifies the ability to respond to a weak signal.
The requirement is the specified signal-to-noise ratio (SNR) for an analog
receiver and bit error rate (BER) for a digital receiver.
2. Selectivity: Receiver selectivity is the ability to reject unwanted signals on
adjacent channel frequencies. This specification, ranging from 70 to 90 dB, is
difficult to achieve. Most systems do not allow for simultaneously active
adjacent channels in the same cable system or the same geographical area.
6
3. Spurious response rejection: The ability to reject undesirable channel response is
important in reducing interference. This can be accomplished by properly
choosing the IF and using various filters. Rejection of 70-100 dB is possible.
4. Intermodulation rejection: The receiver has the tendency to generate its own
on-channel interference from one or more RF signals. These interference signals
are called intermodulation (IM) products. Greater than 70 dB rejection is
normally desirable.
5. Frequency stability: The stability of the LO source is important for low
frequency modulated (FM) and phase noise. Stabilized sources using dielectric
resonators, phase-locked techniques, or synthesizers are commonly used.
6. Radiation emission: The LO signal could leak through the mixer to the antenna
and radiate into free space. This radiation causes interference and needs to be
less than a certain level specified by the FCC.
2.2.1 Receiver Noise
In many analog circuits, noise figure is a measure of the degradation in the
signal-to-noise ratio between the input and output of the component. The noise figure
of a system depends on losses in the circuit, kind of the solid-state device, bias
applied, and amplification. The noise figure, F, is defined as
Si
Ni
Signal − to − noise power ratio at input
F=
=
Signal − to − noise power ratio at output S o
No
where Si, Ni are the input signal and noise powers, and So, No are the output signal
and noise power. By definition, the input noise power is assumed to be the noise
power resulting from a matched resistor at T0=290K; that is, Ni=kT0B.
Consider Fig. 2.3 [4], which shows noise power Ni and signal power Si being
into a noisy two-port network. The network is characterized by a gain G, a bandwidth
7
B, and an equivalent noise temperature Te. If we define Nadded as the noise power
added by the network, then the output noise power can be expressed as
N 0 = G ( N i + N added ) .
The noise figure can be written as
Si
F=
GS i
Ni
= 1+
G ( N i + N added )
N added
Ni
T0= 2 9 0 K
R
Si
Noisy
network
G, B, Te
Pi= Si+ N i
N i= kT 0 B
R
P0 = S0 + N 0
Figure 2.3 Determining the noise figure of a noisy network.
For a cascaded circuit with n networks as shown in Fig. 2.4, the overall noise figure
can be expressed as
Fcas = F1 +
Fn −1
F2 − 1 F3 − 1
+
+…+
.
G1
G1G2
G1G2 … Gn −1
F1
G1
F3
G3
F2
G2
…
Fn
Gn
Figure 2.4 Cascaded noisy circuit with n networks.
2.2.2 Dynamic Range
Dynamic range (DR) is generally defined as the ratio of the maximum input level
that the circuit can tolerate to the minimum input level at which the circuit provides a
reasonable signal quality. The typical definition of DR is shown in Fig. 2.5 [4]. The
dynamic range (DR) is defined as the range between the 1dB compression point and
the minimum detectable signal (MDS). If the input power is above this range, the
output starts to saturate. If the input power is below this range, the noise dominates.
8
Pout( dB)
1 dB
Dynamic range
PD
N oise M DS
floor
Pin( dB)
1 dB
compression
point
Figure 2.5 Illustrating the dynamic range of realistic mixers, amplifiers, or receivers.
From the 1dB compression point, gain, bandwidth, and noise figure, the dynamic
range (DR) of a receiver can be calculated. Expressing the DR in dBm, we can write
as
DR = PD − MDS .
But, the definition is quantified in different applications differently. Another
definition is called the “spurious-free dynamic range” (SFDR). The upper end of the
dynamic range is defined as the maximum input level in a two-tone test for which the
third-order IM products do not exceed the noise floor. The SFDR is given by
SFDR =
2
(POIP3 − G − MDS ) .
3
where G is the gain of a receiver, POIP3 is the output power at the third-order, two-tone
intercept point in dBm.
2.2.3 Third-order Intermodulation
When two signals with different frequencies are applied to a nonlinear system,
the output in general exhibits some components that are not harmonics of the input
frequencies. Called intermodulation (IM), this phenomenon arises from mixing of the
two signals when their sum is raised to a power greater than unity. We are particularly
interesting in the third-order IM products at 2w1-w2 and 2w2-w1, illustrated in Fig. 2.6
[1]. These intermodulation products are a troublesome effect in RF systems because
9
they are difficult to filter from desired channel and may corrupt the desired signal.
w1 w2
w
w1 w2
w
2 w1 -w2 2 w2 -w 1
Figure 2.6 Intermodulation in a nonlinear system.
The third intercept point (IP3) is a figure of merit for intermodulation product
suppression. A high intercept point indicates a high suppression of undesired
intermodulation products. Also, it is an important measure of the system linearity. As
shown in Fig. 2.7, the magnitude of the IM products grows at three times the rate at
which the main signal increases. The third-order intercept point is defined to be at the
intersection of the two lines. The horizontal coordinate of this point is called the input
IP3 (IIP3), and the vertical coordinate is called the output IP3 (OIP3).
Pout( dB)
O I P3
M ain signal
power
IM
power
Pin( dB)
I I P3
Figure 2.7 Growth of output components in an intermodulation test.
For a cascaded circuit, as shown in Fig. 2.8, the following procedure can be used
to calculate the overall system intercept point:
x( t)
I I P3 ,1
I I P3 ,2
y1 ( t)
y2 ( t)
I I P3 ,n
yn( t)
…
Figure 2.8 Cascaded n nonlinear stages.
1. Transfer all intercept points to system input, subtracting gains and adding losses
10
dB for dB.
2. Convert intercept point to power (dBm to mW).
3. Assuming all intercept points are independent and uncorrelated, add powers in
parallel:
IP3 INPUT =
1
1
1
1
+
+…+
IP3(1) IP3(2)
IP3(n )
(mW ) .
4. Convert IP3INPUT from power (mW) to dBm.
2.3 Transmitter System Considerations
The basic function of the transmitter is to modulate, or encode, the baseband
information onto a high frequency sine wave carrier signal that can be radiated by the
transmit antenna. The reason for this is that signals at higher frequency can be
radiated more effectively, and use the RF spectrum more efficiently, than the direct
radiation of the baseband signal. A simple transmitter could have only an oscillator,
and a complicated one would include a phase-locked oscillator or synthesizer.
Generally, a transmitter consists of a modulator, an oscillator, an upconverter, filters,
and power amplifiers as shown in Fig. 2.9 [2].
Modulator I F filter
Mixer
Antenna
Bandpass Power
filter amplifier
Data
in
Local
oscillator
Figure 2.9 Block diagram of a basic radio transmitter.
First, the baseband data modulates an intermediate sine wave signal, then the IF
signal upconverts to the desired RF transmit frequency using a mixer. A bandpass
filter allows the RF signal to pass, while rejecting the spurious interferers. A power
amplifier is used to increase the output power of the transmitter. Finally, the antenna
radiates a propagating electromagnetic wave converted from the modulated carrier
11
signal. The transmitter system considerations are listed below [3]:
1. Power output and operating frequency: The output RF power level generated by
a transmitter at a certain frequency or frequency range.
2. Efficiency: The dc-to-RF conversion efficiency of the transmitter. Efficiency is
defined as η = ( PRF
Pdc
) × 100% . For a power amplifier, the power added
efficiency (PAE) is defined as η =
Pout − Pin
× 100% where Pout and Pin are the
Pdc
output and °input RF power, respectively.
3. Power output variation: The output power level variation over the frequency
range of operation.
4. Frequency tuning range: The frequency tuning range due to mechanical or
electronic tuning.
5. Stability: The ability of an oscillator/transmitter to return to the original operating
point after experiencing a slight electrical or mechanical disturbance.
6. Circuit quality (Q) factor: The loaded and unloaded Q-factor of the oscillator’s
resonant circuit.
7. Noise: The AM, FM, and phase noise. AM noise is the unwanted amplitude
variation of the output signal, FM noise is the unwanted frequency variations,
and phase noise is the unwanted phase variations.
8. Frequency variations: Frequency jumping, pulling, and pushing. Frequency
jumping is a discontinuous change in oscillator frequency due to nonlinearities in
the device impedance. Frequency pulling is the change in oscillator frequency
versus a specified load mismatch over 360° of phase variation. Frequency
pushing is the change in oscillator frequency versus dc bias point variation.
9. Post-tuning drift: Frequency and power drift of a steady-state oscillator due to
heating of a solid-state device.
12
10. Spurious signals: Output signals at frequencies other than the desired oscillation
carrier.
11. Adjacent channel power ratio (ACPR): A figure-of-merit to evaluate the
intermodulation distortion performance of power amplifiers designed for digital
wireless communication systems.
2.3.1 Transmitter Noise
In RF systems, local oscillators provide the carrier signal for both the receiver
and transmit paths. If the LO output contains phase noise, both downconverted and
upconverted signals are corrupted. So, it is important to concern about the noise of the
local oscillators. Consider the output power shown in Fig. 2.10 [3], for an ideal
sinusoidal oscillator operating at f0, the spectrum assumes the shape of an impulse.
But, for an actual oscillator, the spectrum exhibits “skirts” around the carrier
O utput power
O utput power
frequency.
I deal signal
f0
f
N oise signal
f0
f
Figure 2.10 Ideal signal and noisy signal.
As shown in Fig. 2.11, L(fm) is the difference of power between the carrier at f0
and the noise at fm. It is a ratio of single-sideband noise power normalized in 1Hz
bandwidth to the carrier power and is defined as
L( f m ) =
Noise power in 1Hz bandwidth at f m offset from carrier N
= .
Carrier signal power
C
The power is plotted in dB scale. The unit of L(fm) is dBc/Hz.
13
O utput power
( dB)
L( fm)
C
N
f0
f
f0 + f m
Figure 2.11 Oscillator output power spectrum.
2.3.2 Adjacent Channel Power Ratio
Adjacent Channel Power Ratio (ACPR) is normally used a figure of merit for
power amplifiers to characterize their linearity. ACPR is a measure of spectral
regrowth, appears in the signal sidebands, and is analogous to IM3/IM5 for an analog
RF amplifier. Fig. 2.12 [1] shows the power measurement frequency spectrum. And,
the ACPR is defined as
ACPR =
Power spectral density in the main channel
Power spectral density in the offset channel
and is expressed in dBc.
Signal
Channel
Adjacent
Channel
w
Figure 2.12 Adjacent channel power.
2.4 Analysis of the System
2.4.1 Frequency Conversion
First, before we design this RF T/R module, we need to decide the working
frequency of whole system. We already know that transmit/receive signals are
between 900MHz and 1200MHz, and use 2.4GHz of 802.11b as baseband module
14
signal; so the main factor of following is to consider how to set up LO frequency.
As shown in Fig. 2.13, the problem of image is a serious one which we will take
into account. The radio frequency (RF) input signals at frequency of (wLO + wIF) and
(wLO - wIF) will be downconverted to the same intermediate frequency (IF) wIF by
mixing them with a local oscillator at frequency of wLO. The image interferer must be
rejected to prevent aliasing with the desired signal. In addition, to ensure that the
image frequency is outside the RF bandwidth of the receiver, it is necessary to have
f IF >
BRF
,
2
where BRF is the RF bandwidth of the receiver. The separation between the RF and
image frequencies must be greater than the bandwidth of the system in order to filter
the image without affecting the RF response.
Desired
Band
Image
Mixer
wRF
wI F
wim
w
wI F
wI F
w
coswLO t
wLO
w
Figure 2.13 Problem of image.
Finally, because of considering the cost and system structure simplified, we
choose a voltage controlled oscillator between 3.3GHz and 3.6GHz as our LO
frequency.
f IF = 2.4GHz >
1.2GHz − 0.9GHz B RF
=
2
2
From this formula, we could know that if we choose LO frequency this way, image
frequency would not act within the wanted bandwidth. Because of above reason, a
band select filter between receiver and antenna is essential to pass the desired signal
and reject the interferences.
15
2.4.2 Stages Tracking
The receiver encounters two types of the noise: the noise picked up by the
antenna and the noise generated by the receiver. And, the noise that occurs in a
receiver masks weak signals and limits the ultimate sensitivity of the receiver. So,
consider the gain, noise figure, and third-order intercept point listed for each
component in Fig. 2.14, which is the typical value of the chip data sheet. Using the
cascade formulas for noise figure and third-order intercept, the gain, noise figure, and
third-order intercept are plotted in Fig. 2.14 at the output of each stage, versus
position through the receiver.
The specifications for a transmitter depend on the applications. For
communication systems, low noise and good stability are required. As discussed in
previous sections, the power level that exceed the 1 dB compression point P1 of an
amplifier will cause harmonic distortion, and power levels in excess of the third-order
intercept point P3 will cause intermodulation distortion. Thus it is important to track
P1 and P3 through the stages of the transmit path as shown in Fig. 2.15.
16
G(dB), NF(dB), IIP3(dBm)
Antenna
Bandpass Low noise Low noise
amplifier amplifier Mixer
filter
I F Demodulator
amplifier
I F filter
Data
out
CL= 9
G= 1 5
G= 1 5
L= 3
F= 1 .2
F= 6
F= 1 .2
F= 3
P3 = 1 0 0 P3 = 1 0 .1 P3 = 1 0 .1 P3 = 1 8
110
100
Gain
90
80
Noise
70
Figure
60
IIP3
50
40
30
20
10
0
-10
1
2
3
4
Gai n
-3
12
27
18
Noi s e Fi gur e
3
4. 1
4. 23
4. 24
100
13. 1
- 2. 033
- 9. 792
IIP3
Figure 2.14 The cascaded gain, noise figure, and IIP3 are plotted in a receive path.
Modulator
I F filter
Bandpass
Power
Gain Block
filter
amplifier
Mixer
Antenna
Data in
L= 3
P1 = 1 0 0
P3 = 1 0 0
G(dB ), P1dB (dB m), IIP3(dB m)
CL= 9
P1 = 8
P3 = 1 8
G= 1 6
P1 = 1 5
P3 = 3 0
G= 2 0
P1 = -1 .5
P3 = 1 3 .5
25
15
5
-5
-15
Gain
P1dB
IIP3
1
2
3
4
Gai n
-9
- 12
8
24
P 1dB
8
8
8
7
18
18
17. 289
16. 025
IIP3
Figure 2.15 The cascaded gain, P1dB, and IIP3 are plotted in a transmit path.
17
2.4.3 Practical Design
RF
( 2 . 4 GHz)
Gain
M ix er
Block
IF
( HM C2 1 3 M S8 )
PA
TX
( A H2 1 5 )
( RF20 4 4 )
( 0 .9 ~ 1 .2GHz)
LO
Gain
Power
Divider
( 3 .3 ~ 3 .6GHz)
VCO
( VCO 6 90 -40 0 0 T )
Block Synt hesizer
( RF2 0 4 4)
T/ R
Swit ch
A nt enna
Diversit y
( U PG2 0 09 )
( U PG2 0 09 )
( LM X2 4 3 3 )
LO
IF
M ix er
LN A
LN A
( SGL-01 6 3 ) ( SGL-0 1 6 3 )
( HM C2 1 3 M S8 )
( 0 .9 ~ 1 .2GHz)
RX
RF
( 2 .4 GHz)
Figure 2.16 Block diagram of the RF transceiver.
The block diagram of Fig. 2.16 shows the proposed RF transceiver module. It is
a half-duplex system, where the transmitter and receiver are not operating
simultaneously, and duplexing can be achieved with a T/R switch. A single-pole
double-throw switch can connect the antenna to either the transmitter or the receiver.
In transmitting path, we need to connect a bandpass filter after mixer to restrain the
intermodulation products, which is producing at mixer, and then connect it with a gain
block and a power amplifier to increase transmit power to reach requiring 30dBm. In
receiving path, signal from antenna need to be filtered by a band select bandpass filter,
which is chosen in-band signal and weaken out-of-band signal. In the LO path, in
order to increase stability of VCO, we need to add a synthesizer circuit to make it
oscillate stably. Next, in order to increase transmitting power for driving mixer, one
more gain block is required. Besides, a bandpass filter to restrain the harmonic signal
of VCO is also needed.
Finally, for a complete T/R module, except to have proper control of the entirety
functions, the jams and matching problems between circuits need also be overcome.
Whether circuit layout is suitable or not is another key point. We need to avoid
coupling between signals, and we also hope to reduce the size of circuit; each one
18
connects to others. So, we could say that this planning process is very complicated. As
a result, we bring up an effective design flow chart as shown in Fig. 2.17 as a
guideline for designing of our RF T/R module. When there are some functions of
whole module do not meet the specs, we need to re-select some of those key
components.
19
Start
Collect system process data and
relative standards
Establish circuit block diagram according
to system characteristics and standards
Search spec. of circuit components and
choose proper elements
Program link budget with parameters of
circuit elements
Does the whole function fit
the standards?
Rechoose the problem
components
No
Find the problem
components without
fitting
Yes
Circuits implementation
and measurement
Do they fit the
spec.?
No
Debug the
external circuits
Yes
The whole module layout
( with test port)
Implementation
with CW testing
Do RF parameters fit
expectation?
No
Yes
Examine RF parameters:
1. Examining mismatch with
failure Gain.
2. Examining P1dB and gain
with failure P1dB.
3. Examining IP3 and gain
with failure IP3 .
M odulate and test with interferers
M odify the module
layout
Does it fit the
standards?
No
Rechoose the critical
components in order to
increase the estimative margin
Yes
End
Figure 2.17 Design procedure of RF transceiver module.
20
Chapter3 Implementation and
Measurement
3.1 Introduction
RF
( 2.4GHz)
3
M ix er
Gain
Block
IF
( HM C21 3M S8 )
Power
Divider
2
1
( 3 .3 ~ 3 .6GHz)
VCO
Gain ( VCO 690-4000T )
Block Synt hesizer
( RF20 44)
( LM X2433 )
LO
IF
M ix er
TX
( AH215 )
( RF20 44)
( 0 .9 ~ 1 .2GHz)
LO
5
PA
LN A
6
T/ R
Swit ch
Ant enna
Diversit y
( UPG2 009 )
( UPG2 009 )
LN A
( SGL-0163 ) ( SGL-0163 )
( HM C21 3M S8 )
RF
( 2.4GHz)
4
( 0 .9 ~ 1 .2GHz)
RX
Figure 3.1 Block diagram of the RF transceiver.
Through the system budget calculated in previous chapter, the whole system with
appropriate ICs is shown in Fig. 3.1. The main purpose of this chapter is to describe
each individual circuit such as filter, VCO, mixer, LNA, PA, and synthesizer and its
measured results. The last thing is to estimate whole transceiver module, and discuss
the possible reasons of some of these results that are not as we expected.
3.2 RF Bandpass Filters
Filters are key components in all wireless transmitters and receivers. They are
used to reject interfering signals outside the operating band of receivers and
transmitters. Important filter parameters include the cutoff frequency, insertion loss,
and the out-of-band attenuation rate, measured in dB per decade of frequency. Filters
with sharper cutoff responses provide more rejection of out-of-band signals.
3.2.1 3.3~3.6 GHz Bandpass Filter
According to the system structure, we know that a 3.3~3.6GHz is required to
pass the desired LO signal and reject the harmonic signals. Following shows the filter
21
layout, which is designed by a 3-D EM simulator Sonet. The filter is designed with a
center frequency of 3.45GHz, bandwidth of 300MHz. The filter size is 244Χ470mil.
The passband insertion loss is expected to be within -1dB and the return loss to be
better than -10dB.
(a) Circuit configuration.
(b) Simulated return loss and insertion loss.
Figure 3.2 3.3~3.6 GHz bandpass filter simulation.
Fig. 3.3 shows the photograph of the filter and its measured results. The filter is
an interdigital filter with three quarter-wave resonators. The measured passband
insertion loss is within -3.2dB and the return loss is better than -10dB.
22
(a) Circuit layout.
(b) Measured return loss and insertion loss.
Figure 3.3 3.3~3.6 GHz bandpass filter physical implementation.
3.2.2 0.9~1.2 GHz Bandpass Filter
In the transmitting and receiving paths, we need a 0.9~1.2GHz bandpass filter.
The filter we proposed is a three-resonator combline filter. Figure 3.4 depicts the
schematic circuit and the simulated results where the circuit simulator is AWR’s
Microwave Office. The filter is designed with center frequency of 1.05GHz, and
bandwidth of 300MHz. The size of circuit is 330Χ515mil. The passband insertion loss
is expected to be within -1dB and the return loss to be better than -10dB.
23
(a) Circuit configuration
(b) Simulated return loss and insertion loss.
Figure 3.4 0.9~1.2 GHz bandpass filter simulation.
Figure 3.5 shows the circuit photo and measured results of the proposed filter. It
can be seen in Fig. 3.5(a) that there are two coupling capacitors to enhance the
coupling between resonators. The capacitance value is 3.3pF for both capacitors. The
measured passband insertion loss is within –2.7dB and the return loss is better than
-20dB..
24
(a) Circuit layout.
(b) Measured return loss and insertion loss.
Figure 3.5 0.9~1.2 GHz bandpass filter physical implementation.
3.3 Broadband Voltage Controlled Oscillator (VCO)
Oscillators are required in wireless receivers and transmitters to provide
frequency conversion, and to provide sinusoidal sources for modulation. Usually,
these sources need to be tunable over a frequency range, and must provide very
accurate output frequencies. Frequency can be tuned by adjusting the value of the LC
network, perhaps electronically with a varactor diode.
We choose Sirenza VCO690-4000T to be our broadband VCO, and change
capacitance value to make it oscillate in the wanted frequency range. A dc voltage of 5
volts, and controlling voltage of 0 volts to 2.5 volts is desired. Fig. 3.6 is the real
circuit photograph of the VCO. Figure 3.7 depicts the tuning voltage Vt versus
oscillating frequency, which is measured by maximum hold mode of spectrum
analyzer. It is shown in this figure that when control voltage increases slowly, the
oscillated frequency will also increase from 3.28GHz to 3.67GHz.
25
Figure 3.6 Photograph of broadband voltage controlled oscillator.
Figure 3.7 Controlled voltage tuning range.
Next, it is to measure the phase noise of the VCO. The phase noise is measured
at frequency 2MHz.1MHz, and 500KHz offset from center frequency respectively. It
can be seen in Fig. 3.8 that they are all over 100dBc/Hz.
26
Phase noise = -79.73 dBm-10logRBW= -129.73 dBc/Hz
(a) @ 2 MHz offset.
Phase noise = -66.94 dBm-10logRBW= -116.94 dBc/Hz
(b) @ 1 MHz offset.
27
Phase noise = -61.52 dBm-10logRBW= -111.52 dBc/Hz
(c) @ 500 kHz offset.
Figure 3.8 Measured phase noise.
Finally, the second harmonic signal is measured to be –16.96dBc from its
fundamental signal. This can be largely improved by the LO filter described
previously.
Figure 3.9 Measured power of 2-order harmonic frequency.
3.4 Broadband Mixer
The primary function of a mixer in a communication system is to translate signal
from one frequency (RF frequency) to another frequency (IF frequency). A passive
mixer always produces an output signal (IF) of less power than the input signal (IF).
28
This loss is characterized by the mixer conversion loss. Mixers that use active
components generally have lower conversion loss, and may even have conversion
gain. As in the case of amplifiers, harmonic distortion and noise are also important
considerations in mixer performance.
Fig. 3.10 shows the photograph of the broadband mixer. We choose
HMC213MS8 of Hittite as our mixer. It operates with LO frequency of 3.3 to 3.6GHz,
IF frequency of 2.4 to 2.5 GHz, and RF frequency of 900 to 1200MHz.
Figure 3.10 Photograph of broadband mixer.
Fig. 3.11 shows measured mixer conversion loss versus LO input power where
the IF frequency is 2.4GHz and with –10dBm of power. From this graph we could see
obviously that the conversion loss is better than 10dB as LO power over 10dBm. Fig.
3.12 and 3.13 shows the measured mixer conversion loss versus LO frequency where
the IF frequencies are 2.4GHz and 2.5GHz respectively. The IF input power is
from –5 to –13dBm as depicted with different curves in Fig. 3.12 and 3.13.
Figure 3.11 Conversion loss V.S. LO power @ IF= 2.4 GHz, -10 dBm.
29
Figure 3.12 Conversion loss V.S. LO freq. @ IF= 2.4 GHz, -10 dBm.
Figure 3.13 Conversion loss V.S. LO freq. @ IF=2.5 GHz, -10 dBm.
Fig. 3.14 depicts the measured IF to RF isolation of the mixer with IF power
of –10dBm. The isolation is better than –29dBm. Fig. 3.15 shows the leakage of IF
power to the RF port versus input IF power where the IF frequency is fixed at 2.4GHz.
According to this measured results, we need to further decrease the leakage IF signal
to an appropriate degree.
30
Figure 3.14 IF to RF isolation @ IF= -10 dBm.
Figure 3.15 Different power of IF to RF leakage @ IF= 2.4 GHz.
3.5 Broadband Low Noise Amplifier
A low noise amplifier is used in the input stage of a receiver. This is most critical
in the front end of a receiver, where the input signal level is very small, and it is
desired to minimize the noise added by the receiver circuitry. In addition, the noise
power in a receiver is affected more by the first few components than by later
components. Thus it is important that the first amplifier in a receiver have as low a
noise figure as possible.
We choose Sirenza SGL-0169 as the low noise amplifier. According to the data
sheet, we adjust the input and output capacitances to make it work in the desired
0.9GHz to 1.2GHz frequency band. Fig. 3.16 depicts the photograph of the LNA..
31
Figure 3.16 Photograph of broadband low noise amplifier.
Fig. 3.17 shows the measured gain and return loss of this LNA. The frequency
between dotted lines is the desired frequency band, 0.9GHz~1.2GHz. Within this area,
the insertion gain is beyond 17.54dB, and return loss is better than –13.58dB. The
measured noise figure and gain is shown in Fig. 3.18. it can be found in this figure
that the measured noise figure is 1.77 to 1.65 dB with associated gain of 19.66dB to
17.54dB.
Figure 3.17 Measured S-parameters of broadband low noise amplifier.
Figure 3.18 Measured noise figure of broadband low noise amplifier.
3.6 Broadband Power Amplifier
32
Power amplifiers are used in the final stages of wireless transmitters to increase
the radiated power level. So, high linearity is an important parameter for power
amplifier. Because transistors are nonlinear devices, transistor amplifier exhibit two
nonlinear effects: saturation and harmonic distortion. On the other hand, important
considerations for power amplifiers are efficiency, gain, and intermodulation effects.
We choose WJ AH215 as our power amplifier. The DC biase voltage is 5 volts.
Appropriated changing of the matching capacitances at input and output are required
to fit this PA to work in the desired frequency band. After adjusting, we could have
maximal output gain. Following is the circuit photograph.
Figure 3.19 Photograph of broadband power amplifier.
After adjusting a little bit of power amplifier circuit, we measure its small signal
S-parameters. As it is shown in Fig. 3.20 that the measured gain is higher than 13dB
and the measured return loss is better than –8dB. In order to insure the output power
reaches 30dBm, we execute PldB measurement. As we can see in Fig. 3.21, when input
power is 0dB, output power is around 30dBm. (PldB~31dBm)
33
Figure 3.20 Measured S-parameters of broadband power amplifier.
Figure 3.21 Measured P1dB of broadband power amplifier.
3.7 Frequency Synthesizer
Frequency synthesizers consist of phase-locked loops and other circuits that
provide accurate, stable, and tunable frequency outputs. Important parameters that
characterize frequency synthesizers are tuning range, frequency switching time,
frequency resolution, cost, and power consumption. Another very important parameter
is the noise associated with the output spectrum of the synthesizer, in particular the
34
phase noise.
We choose NS LMX2433 as our frequency synthesizer IC. Fig. 3.22 is the
picture of circuit layout, where the Vcc is worked at 5 volts, and input DSP signal pins
are located at place marked with “1” to control this frequency synthesizer. Use PLL to
lock the VCO with a reference crystal oscillator and the output frequency should
cover 3.3~3.6GHz. A DSP control program “Codeloader” is used here, which is free
from download at National Semiconductor website. Fig. 3.23 is interface of
Codeloader. Before starting to use it, we need to set up frequency synthesizer type
LMX2433, and choose RF PLL manual. Inside the RF PLL manual, input wanted
value at Counter field in PLL to decide the distance of each frequency jumping. Or
input the frequency that we want to lock it at VCO field directly. In the following, we
will introduce each special character of input waveform at spectrum analyzer.
Figure 3.22 Photograph of frequency synthesizer.
35
Figure 3.23 Codeloader interface.
Location marked by “2” in Fig. 3.22 is the output port of the frequency
synthesizer. Fig. 3.24 shows the measured tuning range corresponding to VCO’s
control voltage. If we want to generate LO frequency include 3.3GHz~3.6GHz, the
control voltage Vt needs to be within 0.5 to 2.5 volts. The next step is to observe the
phase noise of synthesized signal at frequency 3.3GHz, 3.4GHz, and 3.6GHz
respectively. The measured results are shown in Fig. 3.25. We measure the phase
noise at 500KHz offset from carrier frequency. The measured phase noises are all
better than –109dBc/Hz @500kHz offset.
Figure 3.24 Controlled voltage tuning range.
36
Phase noise = -60.93 dBm-10logRBW= -110.93 dBc/Hz
(a) @ 3.3 GHz.
Phase noise = -59.97 dBm-10logRBW= -109.97 dBc/Hz
(b) @ 3.4 GHz.
37
Phase noise = -60.33 dBm-10logRBW= -110.33 dBc/Hz
(c) @ 3.6 GHz.
Figure 3.25 Measured phase noise at 500 kHz offset.
Location marked with “3” in Fig. 3.22 is the output where the signal is amplified
and filtered to have the LO signal with proper power and clean from harmonic and
spur signals. The signal is measured again with different frequencies of 3.3GHz,
3.4GHz, and 3.6GHz. Fig. 3.26(a) shows the measured spectrum of 3.3GHz
signalwhere the frequency span is 1MHz. The phase noise at 100KHz offset is
calculated to be –104.87dBc/Hz, and transmit power is 9.39dBm.
Phase noise = -64.87 dBm-10logRBW= -104.87dBc/Hz @ 100kHz offset.
Fig. 3.26(b) and (c) shows the measured spur and harmonic signals. The results shows
that the second harmonic signal is –65.29dBc where the third harmonic signal
is –70.15dBc, and no apparent spur signals can be found.
38
Phase noise = -64.87 dBm-10logRBW= -104.87 dBc/Hz
(a) Phase noise with 100 kHz offset.
(b) 2-order harmonic power spectrum.
39
(c) 3-order harmonic power spectrum.
Figure 3.26 Measured power spectrum at 3.3 GHz.
Fig. 3.27(a) is the measured results at 3.4GHz. The measured phase noise
is –106.14dBc/Hz at 100KHz offset, and output power is 8.86dBm.
Phase noise = -66.14dBm-10logRBW= -106.14dBc/Hz @100kHz offset.
Fig. 3.27(b) and (c) shows the measured spur and harmonic signals. The results shows
that the second harmonic signal is –65.29dBc where the third harmonic signal
is –70.15dBc, and no apparent spur signals can be found.
Phase noise = -66.14 dBm-10logRBW= -106.14 dBc/Hz
(a) Phase noise with 100 kHz offset.
(b) 2-order harmonic power spectrum.
40
(c) 3-order harmonic power spectrum.
Figure 3.27 Measured power spectrum at 3.4 GHz.
Fig. 3.28(a) is the measured results at 3.4GHz. The measured phase noise
is –102.82dBc/Hz at 100KHz offset, and output power is 6.39dBm.
Phase noise = -62.82dBm-10logRBW= -102.82dBc/Hz @ 100kHz offset.
Fig. 3.28(b) and (c) shows the measured spur and harmonic signals. The results
shows that the second harmonic signal is –50.86dBc where the third harmonic signal
is –53.05dBc, and no apparent spur signals can be found.
Phase noise = -62.82 dBm-10logRBW= -102.82 dBc/Hz
(a) Phase noise with 100 kHz offset.
41
(b) 2-order harmonic power spectrum.
(c) 3-order harmonic power spectrum.
Figure 3.28 Measured power spectrum at 3.6 GHz.
3.8 Measurement and Discussion
Figure 3.29 Photograph of transceiver module.
42
Fig. 3.29 is the whole RF transceiver module after integrating all of above
described circuits. The whole project is planed to finish within three years. In Fig.
3.29, the two of right-hand side connectors need to be connected with antenna module.
Two connectors located at center of the board are 2.4GHz IF signal input and output
ports respectively. The module has many dc and logic pins needs to be connected with
baseband module. There are no special requirements and limitation about RF T/R
module, except the operating frequency and bandwidth and transmitted signal power
is controllable and can reach at least 30dBm. And, because there are some difficulties
in obtaining appropriate measuring equipments, we could only measure some simple
parameters, such as small signal gain, output power, and frequency spectrum to fit
them into practical situation. We need to find possible components that may have bad
influence on over all module performances. The following is the discussion of
transmit path and receive path.
3.8.1 Transmit Path
Figure 3.30 Photograph of transmit path.
Figure 3.31 Measured S-parameters of transmit path.
43
Fig. 3.30 is a circuit layout of transmit path. Thr RF signal is sent from input port
and passes through a bandpass filter, Gain Block, power amplifier, and then goes to
the output port. The DC bias voltage is five volts, and an extra DC power switch is
added to switch off the power amplifier during receiving mode for energy saving. The
baseband module will control this DC power switch in the future. Fig. 3.31 is the
measured small signal S-parameters of the whole transmitter module. The
approximate filter response curve can be found in the figure. The in-band small signal
gain is over 30dB. It is enough to amplify the modulated RF signal which we want to
transmit.
PC
Spectrum
Analyzer
802.11b
USB
Test
Cable
+ 5V
Test
Cable
T/ R
M odule
Figure 3.32 Transmit testing setup diagram.
We already know that IF frequency is 2.4GHz. So, we use a 802.11b WLAN card
to generate 2.4GHz CDMA IF signal and after down converting to RF frequency the
signal is send to the transmitter. The transmitter outputs are directly sent to a spectrum
analyzer. Fig. 3.32 depicts the transmitter test setup. Shown in Fig. 3.33(a) and (b) are
the output spectrums. Fig. 3.33(a) shows transmitted carrier only and the output
power is 28dBm. Fig. 3.33(b) shows spectrum of a transmitted packet. There are some
adjacent channel power is generated as shown in Fig. 3.33(b). These adjacent channel
power rejection decreases as the output power increases.
44
(a) Power spectrum of transmitted carrier.
(b) Power spectrum of transmitted packets.
Figure 3.33 Transmit testing on spectrum analyzer.
3.8.2 Receive Path
Figure 3.34 Photograph of receive path.
45
Figure 3.35 Measured S-parameters of receive path.
Fig. 3.34 is a receive circuit layout. The receiver gets signal from antenna. The
received signal goes through a bandpass filter to select appropriate frequency and the
signal is then amplified by a two-stage LNA. The bias voltage is also five volts. Fig.
3.35 shows the measured S-parameters of the receiver. The in-band small signal gain
is over 30dB, and it is enough to amplify the received signal to proper level..
PC
PC
802.11b
USB
802.11b
Test
Cable
Test
Cable
+ 5V
USB
Test
Cable
-50 dB
T/ R
M odule
+ 5V
T/ R
M odule
Figure 3.36 Receive testing setup diagram.
Figure 3.36 depicts the setup for the receiver testing. A 802.11b card together
with the transmitter described above is used to transmit the CDMA signal. The
connection between receiver and transmitter is a high loss test cable to simulate the
wireless environment. As shown in Fig. 3.37, we use a test software, which is
downloaded from website, to test the receiver. When we set transmitted signal to be
2Mbps, we get some BER of the received signal. It is possibly due to phase noise, and
spurious caused by nonlinearities of circuit.
46
Figure 3.37 The diagnostic program (MFP) interface.
3.8.3 Measurements
DSP control
PA O N / O FF
&
T/ R switch
5 volts Baseband
4 5 0 mA
46 . 91 mm
I F( 2 .4 GHz)
TX
LO ( 3 .3 ~ 3 .6 GHz)
RF( 0 .9 ~ 1 .2 GHz)
28 dBm
RX
1 3 6 .3 3 mm
I F( 2 .4 GHz)
LN A O N / O FF
Baseband
Figure 3.38 Photograph of transceiver module.
Fig. 3.38 shows the complete RF T/R module. The specifications are described
as following. The LO frequency is between 3.3GHz and 3.6GHz, IF frequency is
2.4GHz, and RF frequency is between 0.9GHz and 1.2GHz. The measured output
power is 28dBm, the size is 46.91mmΧ136.33mm, and relative bandwidth is 28.57%.
BPF
Return Loss(dB)
Insertion Loss(dB)
3.3~3.6GHz
< -15
-2.94 ~ -3.257
0.9~1.2GHz
-22.7 ~ -25
-2.623 ~ -1.408
47
VCO (Sirenza
VCO690-4000T)
Tuning Range
3.28~3.67GHz
(0~2.5V)
Mixer (Hittite
HMC213MS8)
Phase Noise
(2MHz
offset)
-129.73
dBc/Hz
Conversion Loss
(LO=10dBm)
10
LNA (Sirenza
SGL-0169)
Phase Noise
(1MHz
offset)
Phase Noise
(500kHz
offset)
-116.94
dBc/Hz
-111.52
dBc/Hz
IF to RF leakage
@ IF=-10dBm
< -29 dB
Return Loss (dB)
Insertion Loss
(dB)
Noise Figure
900MHz
-13.58
19.96
1.66
1200MHz
-16.82
17.84
1.72
PA (WJ AH215)
Return Loss (dB)
Insertion Loss
(dB)
900MHz
-7.99
13.334
1200MHz
-8.525
13.323
Synthesizer (NS
LMX2433)
P1dB (dBm)
> 31
Tuning Range
3.3 ~ 3.6GHz
(0.5~2.5V)
Phase Noise
(500kHz offset)
2nd-order
harmonic
suppression
3rd-order
harmonic
suppression
3.3GHz
-104.87 dBc/Hz
-65.29 dB
-70.15 dB
3.4GHz
-106.14 dBc/Hz
-57.41 dB
-52.13 dB
3.6GHz
-102.82 dBc/Hz
-50.86 dB
-53.09 dB
48
Ch4 Conclusion
In the future, this module will combine with baseband module and antenna
module. And, in order to make the performance better, the RF T/R module still have
room to be improved. Some suggestions are listed as following.
1. Load-pull Measurements of the PA: We can measure the source-pull and
load-pull data of the PA. Then, these data helps us to design the matching
networks for power amplifiers to achieve maximum power, better PAE, and
good IM3 and ACPR.
2. Insertion Loss of the Filter: On the receiving path, a preselect filter is placed in
fromt of LNA. So, in order to have lower receiver noise figure, the loss of the
bandpass filter should be as low as possible. Furthermore, on the transmit path,
the filter may also reduce the spurious and harmonic content of the output
signal.
3. LO power: Because the conversion loss of mixer decreases as LO power
decreases. Proper LO power level that saturates the loss of the mixer is essential.
We can increase the LO power by changing of gain block chip.
4. Size of the Whole Module: To make the whole module more compact, we can
change all passive components in LTCC. Reducing of test and ground pad can
also reduce the module size. Rough estimation shows that a 50% of size
reduction would be possible.
In conclusion, the RF T/R module is important and commonly used unit in general
wireless systems. Wider bandwidth and smaller size are always in demand. The
results of this thesis can be applied to other wireless systems.
49
References
[1] Behzad Razavi, RF Microelectronics, Prentice Hall PTR.
[2] David M. Pozar, Microwave and RF Wireless Systems, John Wiley & Sons, N. Y.
1998.
[3] Kai Chang, Inder Bahl, and Vijay Nair, RF and Microwave Circuit Component
Design for Wireless Systems, John Wiley & Sons, INC.
[4] David M. Pozar, Microwave Engineering, 2nd Edition, John Wiley & Sons, N. Y.
1998.
[5] Kai Chang, RF and Microwave Wireless Systems, John Wiley & Sons, INC.
50
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